# Frequency reconfigurable low-profile UWB magneto-electric dipole in VHF band

Abdul-Sattar Kaddour, Serge Bories, Anthony Bellion, Christophe Delaveaud

► To cite this version:

Abdul-Sattar Kaddour, Serge Bories, Anthony Bellion, Christophe Delaveaud. Frequency reconfigurable low-profile UWB magneto-electric dipole in VHF band. IEEE Access, 2021, 9, pp.61269-61282. 10.1109/ACCESS.2021.3073094. cea-04906978

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Submitted on 23 Jan 2025

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<figure><img src="image_1.png" /></figure>

Distributed under a Creative Commons Attribution 4.0 International License
Received March 23, 2021, accepted April 8, 2021, date of publication April 14, 2021, date of current version April 28, 2021.

Digital Object Identifier 10.1109/ACCESS.2021.3073094

# Frequency Reconfigurable Low-Profile UWB Magneto-Electric Dipole in VHF Band

ABDUL-SATTAR KADDOUR<sup>1</sup>, (Member, IEEE), SERGE BORIES<sup>1</sup>, ANTHONY BELLION<sup>1,2</sup>, AND CHRISTOPHE DELAVEAUD<sup>1</sup>, (Member, IEEE)

Corresponding author: Abdul-Sattar Kaddour (akaddour@fiu.edu)

This work was supported in part by the Centre national d'études spatiales (CNES), French Space Agency.

**ABSTRACT** Reducing the size of directive and wideband antennas operating in the Very High-Frequency band is a major challenge for small satellite integration. At 100 MHz, the wavelength is 3 m, and the size reduction will make the integration easier and reduces fabrication costs. The original technique presented in this paper enables a significant height reduction (up to 50%) of a crossed Magneto-Electric Dipole antenna while maintaining an ultra-wideband operation bandwidth from 93 to 360 MHz (118%). The proposed technique is based on adding frequency reconfigurable capability to a crossed Magneto-Electric Dipole antenna structure. The antenna has been prototyped and experimentally characterized in a large-sized anechoic chamber. The measured results are in excellent agreement with full-wave electromagnetic simulated results.

**INDEX TERMS** Compact and directive antenna, frequency reconfigurable, magneto-electric dipole, very high frequency, ultra-wideband.

## I. INTRODUCTION

In recent years, the space industry undergoes rapid growth with new players and new constellations that reduced the cost of telecom and space observation applications via smaller and larger-scale satellites (SpaceX Starlink constellation [1]). Therefore, a new generation of “smaller, faster available and cheaper” satellites [2], capable of carrying out many space missions in addition to conventional large satellite systems, has begun to emerge. One critical issue among the numerous challenges deals with the size and weight reduction of satellite antennas, particularly at VHF-UHF frequency bands, where the wavelength involves large-sized radiating elements (antennas of several meters). According to the fundamental theory of small antennas [3], [4], antenna miniaturization is a major scientific challenge that deteriorates the antenna gain, reduces efficiency and operation bandwidth. In this work, we aim to combine miniaturization techniques known as narrow-band techniques with frequency reconfigurable techniques. Thus, frequency agility techniques make it possible to envisage miniaturization of the radiating structure at the cost

of bandwidth reduction. Also, this technique ensures better stability of the radiation properties with frequency.

Crossed “Magneto-Electric Dipoles” (MED) [5], [6] Dual Polarized (DP) antennas are well-known antennas with excellent radiation characteristics. Their operation is based on the Huygens source’s concept and was demonstrated in [7], using spherical wave expansion analysis. The antenna presents the same behavior as a generalized Huygens source with an identical and stable contribution of Transverse Electric (TE) and Transverse Magnetic (TM) modes over a wide bandwidth. Traditionally, antennas with unidirectional radiation proprieties were achieved using director/reflector [8], periodic EBG structures [9]–[11] or metallic cavities [12]. However, as a result, the whole antenna system has a relatively large electrical size. Hence, Huygens source type due to the TE and TM contribution can achieve a unidirectional radiation pattern by its nature without any additional elements. Therefore, MEDs antennas offer outstanding features in terms of compact size, directional radiation and wideband operation. These antennas have been extensively studied in the past [13]–[18]. They can achieve a broad impedance bandwidth, broad half-power beamwidth, and stable gain. Thus, this antenna can be considered an excellent solution for space application. However, these antennas suffer from a significant height (λ₀/4 between

The associate editor coordinating the review of this manuscript and approving it for publication was Chinmoy Saha.

<footnote n="1">¹CEA, LETI, Minatec Campus, Université Grenoble Alpes, 38054 Grenoble, France</footnote>

<footnote n="2">²Centre National d'Etudes Spatiales (CNES), 31401 Toulouse, France</footnote>
the dipole and the reflector plane) and reducing the antenna's electrical size deteriorates the antenna impedance bandwidth. Numerous techniques have been developed to miniaturize the crossed MED antenna, such as the use of dielectric loading [19], the introduction of meta-materials [20] or a capacitive loading [21] between two vertical shorting walls, or by geometric folding of the structure [22]–[24]. Nevertheless, these methods systematically reduce the antenna's operation bandwidth.

Frequency band-reconfigurable antennas have been intensively studied in recent years [25]–[36]. Their attractive feature in terms of frequency band tuning can reduce the size, limiting the complexity and cost while improving a radiofrequency system's total performance. Using tunable materials (ferromagnetic [25], ferroelectric [26] and Liquid Crystal [27]), or adding tunable elements (varactor diodes [28]–[31], RF p-i-n diodes [32], RF MEMS (Micro Electro Mechanical System) [33], [34] and DTC (Digital Tunable Capacitor) [35] on the antenna structure currently constitute the most common techniques for frequency reconfigurability. In [36], a band-reconfigurable linearly polarized antenna is proposed with one wideband mode (99%) and four narrow-band modes using an RF switch (p-i-n diodes). The developed wideband mode is based on MED technology, and a length-reconfigurable thin dipole realizes the narrow-band modes. This article proposes a different agility solution for MED structure since we aim to obtain two instantaneous fractional bandwidths of 66% minimum according to a specific wide-band radio frequency transceiver. This limitation encourages a wideband switching strategy rather than continuous narrow-band tuning. The proposed frequency reconfigurable low-profile crossed MED antenna in the VHF band is designed to achieve the following: (a) a low-profile thickness, (b) a reconfigurable frequency capability with two instantaneous wideband operation modes and (c) a lightweight prototype. Notably, this work advances the design of [21] by (a) rescaling the low-profile antenna to the VHF band, (b) adding a novel wideband frequency reconfigurability associated with its DC biasing network and (c) discuss detailed prototype fabrication and measurements results. To the best of our knowledge, this is the first work that proposes a wide-band frequency reconfigurable low-profile crossed MED antenna. This concept is validated by the fabrication of a prototype and its complete radio measurements in the VHF band.

This paper is structured as follows. Firstly, the antenna miniaturization principle is presented in Section II. The antenna design is described in section III, focusing on band reconfiguration implementation and its ElectroMagnetic (EM) modelization. The simulation results of the antenna were carried from the time-domain CST® MWS solver. A practical validation is achieved by comparing simulation and measurement results in Section IV. Finally, concluding remarks are drawn in Section V.

<figure><img src="image_2.png" /><figcaption>FIGURE 1. 3-D view of (a) Proposed antenna, (b) metallic vertical wall, (c) feeding strips.</figcaption></figure>

## II. ANTENNA MINIATURIZATION

A 3-D view of the proposed compact crossed MED antenna operating in the VHF band is presented in Fig. 1(a). The two crossed electric dipoles are made from four horizontal square plates. Each horizontal plate is connected to the ground plane via two vertical plates, as shown in Fig. 1(b). The magnetic dipoles (slots) are made by each adjacent vertically oriented plate and the ground. This arrangement forms two crossed MEDs producing a DP at $\varphi = 0^\circ$ and $\varphi = 90^\circ$. Two “Γ” shaped feeding strips made from copper are orthogonally placed in the cavities between the vertical plates (see Fig. 1(c)) to properly excite the antennas. The shorter strip is the Port 1 and the higher strip is the Port 2. A 50 $\Omega$ SMA connector is connected to each feeding strips below the ground plane. The shorter (Port 1) and the higher (Port 2) strips excite the 0$^\circ$ and 90$^\circ$ field polarization, respectively. Further details about the two orthogonal “Γ” shaped feeding strips are available in [5].

Despite its complexity, orthogonal coupling strips as a feeding technique can achieve a broadband bandwidth (up to 90%) compared to other simple feeding techniques such as a microstrip line to a parallel strip line [37], where only 17% of bandwidth could be achieved. In order to improve the radiation characteristics, the MEDs are mounted inside a square metallic cavity. Therefore, the main objective of this section is to reduce the height of the crossed MED antenna without degrading its performance. The proposed miniaturization
technique (Fig. 1(a)) consists of reducing the height of the vertical walls by placing a pair of capacitive loads ($C_L$) between each vertical wall (magnetic dipoles) [21]. This height reduction is equally necessary to obtain a stable wide-band radiation property, notably at high frequencies, where the spacing of electric dipoles from the reflector became electrically large.

## A. OPERATION PRINCIPLE

The operating principle of a crossed MED antenna may vary from one antenna to another, depending on the excitation type. In our case, when Port 1 is excited, a planar electric dipole (two pairs of horizontal metal plates D₁-D₂ & D₃-D₄) and a magnetic dipole (vertical metal plates short-circuited with the reflector plane) are excited simultaneously. An electric current $\vec{J}$ is set in the direction $\varphi = 0^{\circ}$ on the electric dipoles and a magnetic current $\vec{M}$ in the direction $\varphi = 90^{\circ}$ in the slot between the two vertical walls [5]. Similarly, when Port 2 is excited, the current $\vec{J}$ is established on the electric dipole (two pairs of horizontal metal plates D₁-D₃ & D₂-D₄) is in the direction $\varphi = 90^{\circ}$ and the magnetic current $\vec{M}$ is in the direction $\varphi = 0^{\circ}$. As a result, this antenna generates two modes of the same amplitude excited on the electric and magnetic dipoles simultaneously. Electrical currents and magnetic currents are 90° out of phase and orthogonal to each other. These two modes resonate at two adjacent frequencies [5], revealing a remarkably enhanced operation bandwidth. Hence, the MED antenna achieves a wide impedance bandwidth and low back radiation due to the Huygens source configuration [7].

## B. EQUIVALENT ELECTRICAL MODEL

MED antennas exhibit wideband operation frequency; therefore, their input impedance can be modeled based on the first canonical form [38]. Hence, the matching bandwidth of wideband antennas such as MEDs results from several adjacent R-L-C parallel circuit resonances in series with an L-C resonance circuit. Each RLC parallel resonance circuit is equivalent to an anti-resonance where the real part of the input impedance is maximum. In contrast, the imaginary part is minimum (close to zero or at its inflection point). Since MED antennas have three resonance modes, only three RLC circuits were added. The first resonance is related to the feeding strip; the second and third resonances are related to electric and magnetic dipole electrical length, respectively [39]. An additional L-C series resonance stage is used to model an impedance transformation. Fig. 2 shows a simple equivalent circuit model of the input impedance of a classical crossed MED antenna at one of its ports.

The first resonance is modeled by a resistance $R_1$, inductance $L_1$ and a capacitor $C_1$. The electric dipole's equivalent circuit is the second parallel RLC resonator (resistor $R_2$, inductance $L_2$, and a capacitance $C_2$). The magnetic dipole is modeled by an RLC parallel resonator (resistance $R_3$, inductance $L_3$, and capacitance $C_3$). The input impedance $Z_{in}$

<figure><img src="image_3.png" /><figcaption>FIGURE 2. Equivalent circuit model of the MED antenna without capacitive loading $C_L$.</figcaption></figure>

<table><caption>TABLE 1. Element values of the equivalent circuit model at Port 1.</caption><thead><tr><th></th><th>RLC circuit</th><th>1</th><th>2</th><th>3</th></tr></thead><tbody><tr><td>Series resonator</td><td>$Q_n$</td><td>24.9</td><td>7.34</td><td>3</td></tr><tr><td>$C_f$ (pF)</td><td>$R_n$ (Ω)</td><td>143.7</td><td>192</td><td>28.4</td></tr><tr><td>150</td><td>$L_n$ (nH)</td><td>9.1</td><td>33</td><td>7.2</td></tr><tr><td>$L_f$ (nH)</td><td>$C_n$ (pF)</td><td>273.5</td><td>48.2</td><td>81.1</td></tr><tr><td>20</td><td>f (MHz)</td><td>100.8</td><td>126.1</td><td>207.5</td></tr></tbody></table>

of the proposed circuit model can be expressed by,

$$Z_{in} = j \left( L_f \omega - \frac{1}{c_f \omega} \right) + \sum_{n=1}^{3} \left( \frac{j R_n L_n \omega}{R_n + j \omega (L_n + R_n C_n \omega)} \right) \quad (1)$$

$$Z_{in} = j \left( L_f \omega - \frac{1}{c_f \omega} \right) + \sum_{n=1}^{3} \left( \frac{R_n}{1 + j Q_n \left( \frac{\omega}{\omega_n} + \frac{\omega_n}{\omega} \right)} \right) \quad (2)$$

where $\omega_n$ and $Q_n$ represent the radian frequency and the quality factor at each resonance. $\omega_n$ and $Q_n$ are defined in (3) and (4), respectively

$$\omega_n = \frac{1}{\sqrt{L_n C_n}} \quad (3)$$

$$Q_n = R_n C_n \omega_n \quad (4)$$

In this work, $Q_n$ is calculated using the approximation of Yaghjian and Best [40] given by,

$$Q_n \approx \frac{\omega_n}{2 * R_n} |Z'_{in}(\omega_n)| \quad (5)$$

where $Z'_{in}(\omega_n)$ is the frequency derivative of the input impedance. At each resonance, $Q_n$ is evaluated numerically from the input impedance extracted from CST MWS full-wave simulations. Additionally, $R_n$ is obtained using the real part of the input impedance. Hence, $L_n$ and $C_n$ values can be found by using $R_n$ and $Q_n$ values in the Eqs. (3) and (4). The circuit element values of the equivalent model of the MED at Port 1 are listed in TABLE 1.

Fig. 3 compares the simulated input impedance on Port 1 using CST MWS to the calculated by the circuit equivalent model. Despite the slight discrepancy at the frequencies below 100 MHz, the proposed equivalent model closely matches the CST MWS simulation in the intended operation bandwidth between 100 and 200 MHz. Therefore, it validates
<figure><img src="image_4.png" /><figcaption>FIGURE 3. Comparison of circuit model input impedance response at Port 1 with the CST MWS simulation.</figcaption></figure>

<figure><img src="image_5.png" /><figcaption>FIGURE 4. Equivalent circuit model of the MED antenna with capacitive loading $C_L$.</figcaption></figure>

the proposed equivalent circuit model. Moreover, the difference between CST MWS simulation and the analytical model is due to the simplification of the model and $Q_n$'s approximated expression usage. In order to increase the accuracy of the equivalent model, more parallel R-L-C parallel circuits could be added and optimized using data fitting tools [38], [41], [42], which lead to a long and complex calculation. However, a detailed model is not addressed in this paper since our proposed model showed acceptable EM simulation agreement. The behavior of input impedance will be discussed in section II.C.

### C. CAPACITIVE LOADING ANALYSIS

The capacitors $C_L$ were placed on the magnetic dipoles (slots); therefore, the previous circuit model is modified to include the four $C_L$ on the magnetic dipole RLC circuit as shown in Fig. 4. Introducing four capacitors $C_L$ on the top of the magnetic dipoles implies a change in the value of the magnetic dipole resonator's total capacitance ($C_3 + 4C_L$). According to the electrical equivalent model, the resultant resonance frequency of the magnetic dipole is equal to $1/(2\pi\sqrt{L_3(C_3 + 4C_L)})$. Hence, the magnetic resonance frequency is shifted towards lower frequencies by virtually extending the electric current path in the magnetic dipoles when $C_L$ is increased. This modification then causes a change in the behavior of the antenna, which modifies its impedance profile. Thus, it is possible to achieve broadband impedance matching with reduced thickness of the antenna by choosing the appropriate value of $C_L$.

In order to analyze the influence of capacitive loads value $C_L$ on the one-to-one correspondence between the antenna structure and circuit model, $C_L$ values are varied to optimize the operation bandwidth. Fig. 5 illustrates the simulated and

<figure><img src="image_6.png" /></figure>

<figure><img src="image_7.png" /></figure>

<figure><img src="image_8.png" /><figcaption>FIGURE 5. Effect of the value of the capacitive load $C_L$ on the input impedance at Port 1: (a) real part, (b) imaginary part and (c) SWR.</figcaption></figure>

modeled input impedance and the SWR on Port 1 for different values of $C_L$. It can be seen that the response of the proposed equivalent circuit model of the MED with capacitive loading matches the one from EM simulations. However, the magnitude of the real part and imaginary part of both electrical and magnetic resonance are affected when $C_L$ varies thereby, modifying their corresponding quality factors ($Q_2$ and $Q_3$). Besides, our proposed circuit model relies on the quality factor of each resonance to calculate its related RLC circuit components. Therefore, for each value of $C_L$ the resistive and inductive of the magnetic RLC circuit and the electrical RLC circuit components are recalculated to reproduce the magnitude of the resonances.

Nevertheless, this model validates the frequency modification of magnetic resonance. Thus, confirming our proposed equivalent circuit model. When the antenna's height is reduced from $0.25\lambda_0$ (i.e., a standard thickness of MED antennas) to $0.1\lambda_0$ (i.e., proposed design without capacitive loading $C_L$), the three resonance peaks of the antenna are shifted to the high frequencies. The feeding strip's resonance
<figure><img src="image_9.png" /><figcaption>FIGURE 6. Geometry of the proposed reconfigurable crossed MED antenna: (a) 3D view, (b) side view, (c) electrical dipoles, (d) vertical wall. Switch configuration on (e) electrical dipole and (f) magnetic dipole.</figcaption></figure>

shifts to 100 MHz, whereas the electrical resonance is ampli-
fied and moved to 125 MHz since the dipoles become very
close to the reflector plane. The magnetic resonance is shifted
to 208 MHz resulting in an impedance mismatch.

Furthermore, by adding the capacitive loads' $C_L$ on
each magnetic dipole (between vertical walls), the magnetic
resonance frequency can be controlled. As $C_L$ increases,
the magnetic resonance shifts to the lower frequencies and
improves the antenna impedance matching. For example,
using a $C_L$ = 3.5 pF, an impedance bandwidth of 66% from
100 MHz to 200 MHz is obtained.

### III. DESIGN OF FREQUENCY RECONFIGURABLE LOW-PROFILE ANTENNA

In this section, the proposed frequency reconfigurability solu-
tion is described. The aim is to cover an ultra-wide frequency
band from 100 to 370 MHz divided into two instantaneous
fractional sub-bands using the same compact antenna struc-
ture. To cover two wide bands, we opt for the strategy of
decomposition of the antenna structure with two different
electrical dimensions and propose to combine them by elec-
trical disconnections. Consequently, the design approach first
looks at the most complicated structure in terms of minia-
turization and wide bandwidth, i.e., the antenna covering
the lower sub-band from 100 to 200 MHz (State 1). Then,
the antenna's design over the upper sub-band from 200 to
370 MHz (State 2) is studied, and its integration into the low
band antenna is considered. The band switching is achieved
using RF switches (p-i-n diodes) to isolate each antenna

structure and create a different RF current path within the
antenna structure according to the desired frequency band.

#### A. ANTENNA GEOMETRY AND OPERATION PRINCIPLE

A 3-D view of the compact frequency reconfigurable crossed MED antenna and its associated feeding strips are presented in Fig. 6 (for a visual simplification, all the RF switches are hidden) and Fig. 7, respectively. To achieve a wideband frequency switching, the effective electrical length of the antenna should be changed. Therefore, the three different elements of the antenna must be reconfigured to address both sub-bands. Several 3.3 mm-wide gaps are cut on the antenna structure, and RF switches are placed to bridge these gaps. The antenna dimensions are given in TABLE 2.

##### 1) ELECTRIC DIPOLE ARMS RECONFIGURATION

The configuration of one arm of the four electrical dipoles is shown in Fig. 6(e). Three RF switches, “SW1”, are added on each electrical dipole arm to significantly reduce the current density in each RF switch, therefore preventing a significant heating and a non-linear operation mode of the p-i-n diodes. By turning the RF switches “SW1” ON and OFF, the dipoles’ length can be varied, which shifts the operating frequency band. When the RF switches “SW1” are ON, the electrical current flows in the longer dipoles allowing the antenna to operate in the lower sub-band and on the contrary, when the RF switches “SW1” are OFF, the electrical current flows in the smaller dipole allowing the antenna to operate in the upper sub-band. Therefore, the electrical current path on the
<table><caption>TABLE 2. Dimensions of the proposed reconfigurable antenna.</caption><thead><tr><th>Parameter</th><th>Value (cm)</th><th>Parameter</th><th>Value (cm)</th></tr></thead><tbody><tr><td>G</td><td>180</td><td>H<sub>g</sub></td><td>59</td></tr><tr><td>H<sub>p</sub></td><td>29</td><td>S</td><td>5.2</td></tr><tr><td>L</td><td>43.2</td><td>L<sub>1</sub></td><td>12</td></tr><tr><td>L<sub>2</sub></td><td>7.7</td><td>L<sub>3</sub></td><td>23</td></tr><tr><td>L<sub>4</sub></td><td>0.28</td><td>L<sub>5</sub></td><td>5.9</td></tr><tr><td>L<sub>6</sub></td><td>2.9</td><td>L<sub>d</sub></td><td>0.41</td></tr><tr><td>D<sub>11</sub></td><td>19</td><td>D<sub>12</sub></td><td>27</td></tr><tr><td>D<sub>13</sub></td><td>11.2</td><td>D<sub>14</sub></td><td>9.5</td></tr><tr><td>D<sub>15</sub></td><td>11</td><td>D<sub>16</sub></td><td>22.1</td></tr><tr><td>D<sub>17</sub></td><td>30.3</td><td>D<sub>21</sub></td><td>22</td></tr><tr><td>D<sub>27</sub></td><td>27</td><td>D<sub>22</sub></td><td>11</td></tr><tr><td>D<sub>23</sub></td><td>9.5</td><td>D<sub>24</sub></td><td>8</td></tr><tr><td>D<sub>25</sub></td><td>8</td><td>D<sub>26</sub></td><td>25.4</td></tr><tr><td>D<sub>27</sub></td><td>30.3</td><td>E<sub>d</sub></td><td>0.33</td></tr><tr><td>E<sub>s</sub></td><td>1</td><td>W<sub>c</sub></td><td>0.23</td></tr><tr><td>W<sub>d</sub></td><td>1.5</td><td>W<sub>p</sub></td><td>22.4</td></tr><tr><td>W<sub>m</sub></td><td>22.4</td><td>W<sub>s</sub></td><td>7.2</td></tr></tbody></table>

electrical dipoles can be controlled by the location of the RF switches “SW1”. Thus, RF switches “SW1” were placed to create electrical dipoles that resonate around 200 MHz (≈0.41λ). In addition, a gap of dimension L₂ between the two portions of each electric dipole’s arms is optimized to perform enough isolation between the two operating modes. We note that for a small gap (L₂ = 4 cm), a significant portion of the field is coupled with the external arm despite if the switches “SW1” are OFF, whereas sufficient decoupling is ensured from L₂ equal to 7.7 cm.

## 2) MAGNETIC DIPOLE SLOT RECONFIGURATION

Two RF switches, “SW2”, are introduced between both sides of the capacitive load C_L to control the magnetic dipoles’ electrical length, as shown in Fig. 6(f). The two antenna states are determined by either connecting or not the capacitive load C_L. When the RF switches “SW2” are ON, the electrical current flows in the capacitor C_L. Thus, the magnetic dipoles become capacitively-loaded, allowing the antenna to operate in the lower sub-band. When the RF switches “SW2” are OFF, the capacitor C_L becomes invisible, and as a consequence, the antenna operates in the upper sub-band. Thus, the slots’ capacitive loading is used both to miniaturize and make the antenna frequency reconfigurable. The location of RF switches “SW2” doesn’t affect the antenna performance, and it was chosen to be further possible from the capacitor C_L to simplify the antenna mounting.

## 3) FEEDING STRIPS RECONFIGURATION

One RF switch “SW3” is placed on each feeding strip to modify its electrical length, as seen in Fig. 7. The longer strip configuration excites the lower sub-band (State 1), and the shorter strip configuration excites the upper sub-band (State 2). The location of RF switch “SW3” on the feeding strips was chosen to switch between shorter and longer feeding strips already optimized to ensure impedance matching each operating state. Different states of switches are presented in TABLE 3.

<figure><img src="image_10.png" /><figcaption>FIGURE 7. Geometry and switch configuration of the feeding strips.</figcaption></figure>

<table><caption>TABLE 3. State of the RF switches for operation band.</caption><thead><tr><th rowspan="2">Band</th><th colspan="3">Switch</th></tr><tr><th>SW1</th><th>SW2</th><th>SW3</th></tr></thead><tbody><tr><td>State 1 (lower sub-band)</td><td>ON</td><td>ON</td><td>ON</td></tr><tr><td>State 2 (higher sub-band)</td><td>OFF</td><td>OFF</td><td>OFF</td></tr></tbody></table>

<table><caption>TABLE 4. Element values of the p-i-n diode equivalent circuit.</caption><thead><tr><th>Element</th><th>L<sub>d</sub></th><th>R<sub>s</sub></th><th>R<sub>p</sub></th><th>C<sub>t</sub></th></tr></thead><tbody><tr><td>Value</td><td>0.32 nH</td><td>1.2 Ω</td><td>1.5 kΩ</td><td>0.3 pF</td></tr></tbody></table>

## 4) RF SWITCHES

In this design, for higher isolation, each SW1 switch is realized by two p-i-n diodes (Model Infineon Bar63-02) [43] mounted in series, whereas SW2 and SW3 switches are made using one p-i-n diode. Each diode can be forward biased to the ON state with a DC voltage source that provides a 50 mA biasing current, whereas it will be OFF if left unbiased.

The equivalent circuit of the p-i-n diode [44] is described in Fig. 8, while the values of the elements at the chosen biasing condition (biasing current = 50 mA) are given in TABLE 4. In the CST® MWS simulation, the setting for the ON-OFF operation of the p-i-n diodes is modeled by selecting its equivalent circuit. All the DC lines and the DC decoupling components are not included here and will be discussed in section “DC lines Geometry”.

## B. ANTENNA PERFORMANCE

Although an SWR under 2 is sometimes necessary for devices that emit high power to prevent the reflected signal from damaging the radio front end or blinding the reception chain depending on the system. For example, radar antennas require a matching condition of an SWR < 1.3 [45]. However, in our case, the proposed antenna is expected to operate in receiving mode, where a matching condition of SWR below 3 is considered acceptable for the intended applications’ (e.g smartphone [46] or Internet of Things [47] applications). Fig. 9 shows the simulated SWR and realized gain for both ports and both operation states described in TABLE 3. The two wideband modes (close to one octave each) can be
<figure><img src="image_11.png" /><figcaption>FIGURE 8. Equivalent circuits of the p-i-n diode: (a) ON state (for State 1 or lower sub-band) and (b) OFF state (for State 2 or upper sub-band).</figcaption></figure>

<figure><img src="image_12.png" /><figcaption>FIGURE 9. Simulated SWRs and realized gain at $\theta = 0^\circ$ of the frequency reconfigurable antenna.</figcaption></figure>

achieved from 100 to 180 MHz and 180 to 370 MHz. The maximum gain achieved is approximately 9.5 dBi when the antenna operates in State 1 and 11.8 dBi in State 2.

### C. SURFACE CURRENT DISTRIBUTION

For an adequate understanding of the proposed reconfigurable MED antenna operation, the surface current distribution with input from *Port 1* and *Port 2* for both State 1 at 150 MHz and State 2 at 300MHz is shown in Fig. 10(a) and Fig. 10(b), respectively. The color indicates the amplitude of the current densities and the vectors show the orientation and direction of the flow. In both States, the electric current on the electrical dipoles is set in the direction $\varphi = 0^\circ$ and $\varphi = 90^\circ$ when *Port 1* and *Port 2* is excited, respectively. In State 1, when all the RF diodes are ON, the electrical current passes through “SW1” from the internal to the external electrical dipole’s arms. Therefore, the electrical current attains its maximum on the larger electric dipoles. Hence, a half wavelength current distribution mode is established on both internal and external dipole’s arms (see the arrows). Moreover, the current passes through “SW2” and flows into the capacitive load $C_L$ placed along with the current path, therefore loading the magnetic dipoles capacitively. As a result, the electrical and magnetic dipole resonates at the lower frequency band. The electrical current on the feeding strips passes through the switch “SW3” and its maximum dominates the longer feeding strips. In State 2, when all the RF diodes are OFF, the electrical currents on the electrical dipoles are blocked by the “SW1” and “SW2” switches, therefore mainly trapped in the internal dipole’s arms. Thus, the electrical and magnetic dipole resonance shifts to higher frequencies. It should be noted that the presence of a symmetrical smaller level current mode on the external dipole’s

arms is due to the field coupling from the inner dipole’s arms and the reflection of the ground plane. Their contribution to the radiated far-field is canceled out in the broadside direction given their phase opposition. The switch “SW3” blocks the electrical current on the feeding strips and its maximum dominates the shorter feeding strips.

### D. DC LINES GEOMETRY

DC lines have been the topic of a specific design aiming to limit their possible impact on antenna performances. Several ferrite beads (Model Murata BLM18G) [48] are used along the DC lines to cut off the radiated RF signals while allowing the DC signals to pass. The equivalent circuit of the ferrite bead is described in Fig. 11. This type of ferrite bead has a very low resistance to DC signals (approximately 1.3 Ω); nevertheless, it has a very high resistance to RF signals (over 470 Ω starting from 100 MHz).

The switches’ polarization structure placed on the electrical dipoles’ arms is presented in Fig. 12(a). In order to simplify the design, a vertical DC line structure is connected to the second portion of each dipole arm. A DC signal of 150 mA is needed to bias a group of three parallel switches, “SW1” placed on each electrical dipole arm, and the cathode is connected to the ground through the vertical walls. The DC line structure to bias the diodes on both sides of the capacitive load and feeding strips are presented in Fig. 11(b) and Fig. 11(c), respectively. Each DC line is divided into two lines: vertical DC line printed on the vertical portions of the feeding strips and horizontal DC line printed on the electric dipoles’ substrate. A DC signal of 50 mA is needed to bias “SW2” and “SW3” switches. The DC line position and the number of ferrites are optimized to minimize their impact on antenna performance.

Fig. 13 illustrates the simulated antenna efficiency with and without DC lines and switch resistance $R_s$. The efficiency degradation caused by DC lines is negligible for the State 2 mode; DC lines produce obvious power losses for the State 1 mode. These power losses caused by DC lines are attributed to the RF power absorption by the ferrite beads on DC lines. Also, the forward resistance $R_s$ produces power losses in State 1 mode at the electric and magnetic frequency resonance (about 10%). Accordingly, with high-performance switches presenting low forward resistance, high-performance RF chokes with a wide operating band and no power losses are desired in the lowest frequency band (where current densities are higher) to obtain high radiation efficiency. Additionally, the slight difference in the radiation efficiency between *Port 1* and *Port 2* is explained by the difference in both feeding strips’ electrical lengths.

### IV. SIMULATED ANTENNA PERFORMANCE AND EXPERIMENTAL RESULTS

A prototype is built to validate the proposed reconfigurable antenna, which is depicted in Fig. 14(a). In order to obtain a lightweight and robust structure, the reflector plane and cavity are decomposed into two parts; the first part is a central
<figure><img src="image_13.png" /><figcaption>FIGURE 10. Surface current distribution for both *Port 1* and *Port 2* in (a) state 1 and (b) state 2.</figcaption></figure>

<figure><img src="image_14.png" /><figcaption>FIGURE 11. Equivalent circuits of the ferrite bead: (a) RF mode and (b) DC mode.</figcaption></figure>

reflector plane of dimensions 110 cm × 110 cm made using four double-sided FR-4 PCB boards of 1.6 mm thickness. The second part is an extended reflector plane made of polystyrene panels (thickness 20 mm) covered with a copper tissue layer [49]. The thickness of the copper tissue is 0.4 mm and weighs around 0.8 kg/m². The prototype was built on several steps; first, the central reflector plane was mounted on a metallic structure followed by the feeding strips, as seen in Fig.14(b). The vertical walls, dipoles arms and vertical DC lines are soldered to each other and mounted on the central reflector plane (Fig. 14(c) shows one side of dipoles arm and vertical walls). Finally, the extended reflector plane is added to the structure. The total weight of the prototype was about 16 kg.

The radioelectric antenna properties are measured using a vector network analyzer R&S® ZVB4 inside the CEA Leti

VHF anechoic chamber of dimensions 12 × 12 × 20 m³, as depicted in Fig. 15. Three different DC power supplies placed outside the anechoic chamber were used for biasing the p-i-n diodes separately on the electrical dipoles, magnetic dipoles, and feeding strips. As mentioned above, a current intensity of 50 mA is required to bias each RF switch. This current intensity is equivalent to a 0.95 V forward voltage bias and a total dissipated power of 47.5 mW on each p-i-n diode. A photograph of the DC bias circuit implementation is shown in Fig.14(d). This configuration ensures the biasing network's proper operation to maintain a linear operation mode of the p-i-n diodes.

A. DUAL POLARIZATION MODE: SWRs AND ISOLATION

The SWR and isolation from *Port 1* and *Port 2* for both simulation and measurements are shown in Fig. 16(a) and Fig. 16(b), respectively. The antenna measured SWRs are in good agreement with simulated values. In State 1, when all the diodes are ON, the antenna operates from 97 to 193 MHz with 66.2% fractional bandwidth (SWR<3) and from 93.5 to 199 MHz with a bandwidth of 72.1% (SWR<3) for *Port 1* and *Port 2*, respectively. The common bandwidth of the two ports
<figure><img src="image_15.png" /></figure>

<figure><img src="image_16.png" /></figure>

<figure><img src="image_17.png" /><figcaption>**FIGURE 12.** DC lines configurations for (a) electric dipole, (b) magnetic dipole, and (c) feeding strips. (dimensions in cm).</figcaption></figure>

is 66.2% (SWR<3). In State 2, when all the diodes are OFF, the antenna operates from 174 to 367 MHz with 71.3% fractional bandwidth (SWR<3) and from 171 to 368 MHz with a bandwidth of 72% (SWR<3) for Port 1 and Port 2, respectively. The common bandwidth of the two ports is 71.3% (SWR<3). Additionally, the two operating states overlap from 174 MHz to 193 MHz.

A better impedance matching is obtained with the measured results. It is presumably due to slightly higher losses in the active components and ferrites beads. The isolation between Port 1 and Port 2 is always below – 20 dB in both states' operation frequency bands. Some differences between simulation and measurement are only observed for low isolation level (<-25 dB).

<figure><img src="image_18.png" /><figcaption>**FIGURE 13.** Antenna efficiency versus frequency for State 1 and State 2 with and without DC lines.</figcaption></figure>

<figure><img src="image_19.png" /><figcaption>**FIGURE 14.** Photographs of the (a) fabricated antenna, (b) central reflector and feeding strips, (c) dipole arm, vertical walls and vertical DC line and (d) DC lines circuit implementation.</figcaption></figure>

## B. CIRCULAR POLARIZATION (C.P.) MODE

CP radiation mode was obtained using an ultra-wideband (100-520 MHz) 90° hybrid coupler INOV IPP 7057 [50] connected to the antenna ports. The measured properties of the fabricated feeding system have been introduced in the EM simulator.

### 1) REALIZED GAIN

The simulated and measured realized gain in circular polarization in the broadside direction ($\theta = 0°$) are shown in Fig. 17. The measured maximum gain achieved is
<figure><img src="image_20.png" /><figcaption>FIGURE 15. Photographs of the fabricated antenna inside the CEA Leti VHF anechoic chamber.</figcaption></figure>

<figure><img src="image_21.png" /><figcaption>FIGURE 16. Measured and simulated values of (a) SWR and (b) isolation as functions of the frequency for the proposed antenna in both states.</figcaption></figure>

<figure><img src="image_22.png" /><figcaption>FIGURE 17. Measured and simulated circularly polarized realized gain at $\theta = 0^\circ$ as a function of the frequency for State 1 and State 2.</figcaption></figure>

approximately 9.7 dBic and 10.6 dBic when the antenna is operating in State 1 and State 2, respectively. The measurement is globally 1 dB under the simulation, and it's associated with additional losses in the prototype and/or a calibration

error uncertainty. In fact, the difference between simulated and measured is more significant (1.5 dBi) towards the low frequencies when the antenna is more compact and where the resistive parasitics have a possible higher impact.

## 2) RADIATION PATTERNS

The simulated and measured radiation patterns in the State 1 at 100, 150, and 200 MHz and in State 2 at 200, 275, and 350 MHz are shown in Fig. 18(a) and Fig. 18(b), respectively. Over the operating frequencies, the main beam is always pointing the broadside direction. Furthermore, the measured front-to-back ratios are approximately above 20 dB, and the cross-polarization levels are consistently below -15 dB over the entire frequency band. The radiation patterns are symmetric and nearly identical in both cut planes $\varphi = 0^\circ$ and $\varphi = 90^\circ$. In State 2, the reconfiguration in frequency contributes to improving the antenna radiation; the -3dB beamwidth has increased compared to State 1 at 200 MHz. At lower frequencies, the -3dB beamwidth is around $60^\circ$ and becomes narrower at higher frequencies. Besides, the radiation pattern is stable as a function of the frequency in the operating band of the antenna up to 300 MHz. Above this frequency, the secondary lobes begin to appear in both planes $\varphi = 0^\circ$ and $\varphi = 90^\circ$ (distance between the electric dipole and the reflector plane ($0.29\lambda$) becomes higher than $0.25\lambda$). Overall, a remarkable agreement is observed between simulated and measured results.

## 3) ANTENNA EFFICIENCY

The proposed antenna simulated and measured efficiency versus frequency at both operating states is illustrated in Fig. 19. The antenna efficiency is defined as the ratio of the total power radiated by the antenna to the power delivered to the antenna. Measured antenna efficiency is obtained using measured total realized gain patterns integration. The antenna has a measured antenna efficiency above 50% in both the operation state's bandwidth. Below 120 MHz, the measured antenna efficiency is lower than 50%, mainly due to the domination of the RF diodes and ferrite beads' ohmic losses. The acceptable difference between simulation and measurements is probably due to additional losses from fabrication (RF diode, ferrite) and some measurement inaccuracies.
<figure><img src="image_23.png" /><figcaption>FIGURE 18. Measured and simulated radiation patterns at $\phi = 0^\circ$ and $\phi = 90^\circ$ (a) State 1 and (b) State 2.</figcaption></figure>

<figure><img src="image_24.png" /><figcaption>FIGURE 19. Measured and simulated antenna efficiency as a function of the frequency for state 1 and state 2.</figcaption></figure>

### 4) AXIAL RATIO (AR)

A comparison between simulation and measurements of the AR in the broadside direction ($\theta = 0^\circ$) versus frequency and the AR patterns in both states are shown in Fig. 20(a) and Fig. 20(b), respectively. As we can see, the AR is less than 2 dB from 80 to 245 MHz and 175 to 360 MHz in State 1 and State 2, respectively. This excellent quality and wide bandwidth of the AR is due to the four-order symmetry of the radiating structure (i.e., electrical and magnetic dipoles), the stable radiation over the operating frequency band, and the use of the wideband hybrid coupler. The 3-dB AR beamwidth at 150 MHz and 250 MHz is 115° and 65°, respectively. These results show a very good agreement between simulations and measurements.

### C. KEY RESULTS

A compact and frequency reconfigurable UWB crossed MED antenna operating in the VHF band suitable for space

<figure><img src="image_25.png" /><figcaption>FIGURE 20. Comparison of simulated and measured (a) AR versus frequency, (b) AR cut plane at 150 MHz and 250 MHz in both planes $\phi = 0^\circ$ and $\phi = 90^\circ$.</figcaption></figure>

applications has been introduced and studied in this work. Our findings demonstrate that placing RF switches (p-i-n diodes) on the structure of the compact crossed MEDs provides two wideband operating states. In TABLE 5, the performance of our proposed antenna is quantitatively compared to previously published designs. This comparison includes
<table><caption>TABLE 5. Measured performance comparison of the proposed and the reference antennas.</caption><thead><tr><th rowspan="2">Antenna</th><th rowspan="2">Polarization</th><th colspan="2">Operation frequency band</th><th rowspan="2">Electrical size at the lowest operation frequency</th><th rowspan="2">Gain (dBi)</th></tr><tr><th>SWR&lt;3</th><th>AR &lt;3 dB</th></tr></thead><tbody><tr><td>This work</td><td>Circular</td><td>Band 1: 97 - 193 MHz (66.2%)<br>Band 2: 74 - 367 MHz (72%)</td><td>Band 1: 85-245 MHz (97%)<br>Band 2: 175-365 MHz (70%)</td><td>Ant.: 0.3λ₀×0.3λ₀×0.09λ₀<br>Cavity: 0.58λ₀×0.58λ₀×0.19λ₀</td><td>Band 1: 5-10<br>Band 2: 7.5-11</td></tr><tr><td>[5]</td><td>Circular</td><td>1600-3050 MHz (63%)</td><td>1.3 3.05 GHz (80%)</td><td>Ant.: 0.32λ₀×0.32λ₀×0.18λ₀<br>Cavity: 0.58λ₀×0.58λ₀×0.19λ₀</td><td>7.6-9.4</td></tr><tr><td>[6]</td><td>Circular</td><td>3000-4600 MHz (42%)</td><td>-</td><td>Ant.: 0.5λ₀×0.5λ₀×0.25λ₀<br>Reflector: 1.5λ₀×1.5λ₀</td><td>7-9.3</td></tr><tr><td>[12]</td><td>Dual</td><td>5600-7200 MHz (25%)</td><td>5.6-6.9 GHz (21%)</td><td>Ant.: 0.67λ₀×1.15λ₀×0.005λ₀<br>Cavity: 0.48λ₀×0.97λ₀×0.22λ₀</td><td>7.5-10</td></tr><tr><td>[16]</td><td>Circular</td><td>1380-3600 MHz (89%)</td><td>-</td><td>Ant.: 0.33λ₀×0.30λ₀×0.13λ₀<br>Reflector: 0.83λ₀×0.83λ₀</td><td>6-8.4</td></tr><tr><td>[17]</td><td>Linear</td><td>3000-8000 MHz (90%)</td><td>-</td><td>Ant.: 0.2λ₀×0.24λ₀×0.15λ₀<br>Reflector: 0.4λ₀×0.4λ₀</td><td>3.7-5.3</td></tr><tr><td>[18]</td><td>Linear</td><td>2450-5400 MHz (75%)</td><td>-</td><td>Ant.: 0.16λ₀×0.24λ₀×0.11λ₀<br>Reflector: 0.42λ₀×0.42λ₀</td><td>5-7</td></tr><tr><td>[19]</td><td>Linear</td><td>1600-2500 MHz (43.9%)</td><td>-</td><td>Ant.: 0.25λ₀×0.28λ₀×0.1λ₀<br>Reflector: 0.8λ₀×0.8λ₀</td><td>6-9</td></tr><tr><td>[20]</td><td>Dual</td><td>1500-2800 MHz (60.46%)</td><td>-</td><td>Ant.: 0.25λ₀×0.35λ₀×0.13λ₀<br>Reflector: 0.75λ₀×0.75λ₀</td><td>5-8.5</td></tr><tr><td>[21]</td><td>Linear</td><td>1600-3100 MHz (64%)</td><td>-</td><td>Ant.: 0.32λ₀×0.32λ₀×0.1λ₀<br>Cavity: 0.6λ₀×0.6λ₀×0.25λ₀</td><td>6-10</td></tr><tr><td>[22]</td><td>Dual</td><td>2000-3700 MHz (59.6%)</td><td>-</td><td>Ant.: 0.55λ₀×0.5λ₀×0.09λ₀<br>Reflector: 0.96λ₀×0.64λ₀</td><td>5-10</td></tr><tr><td>[23]</td><td>Linear</td><td>1610-2950 MHz (59%)</td><td>1.2-3 GHz (85%)</td><td>Ant.: 0.23λ₀×0.23λ₀×0.18λ₀<br>Cavity: 0.59λ₀×0.59λ₀×0.2λ₀</td><td>5-9.3</td></tr><tr><td>[24]</td><td>Circular</td><td>1700-3300 MHz (64%)</td><td>-</td><td>Ant.: 0.34λ₀×0.34λ₀×0.11λ₀<br>Cavity: 0.63λ₀×0.63λ₀×0.26λ₀</td><td>6-9</td></tr><tr><td>[36]</td><td>Dual</td><td>Narrow bands: 950, 1350, 1700,<br>and 2000 MHz<br>Wideband band: 800 - 2200 MHz<br>(93%)</td><td>-</td><td>Ant.: 0.44λ₀×0.26λ₀×0.14λ₀<br>Cavity: 0.64λ₀×0.64λ₀×0.13λ₀</td><td>7.7-9.7</td></tr></tbody></table>

the operating frequency band for an SWR<3, as well as an AR<3dB, the antenna (radiating element, cavity/reflector) electrical size, expressed as a function of λ₀ where λ₀ is the wavelength at the lowest operation frequency, polarization, and the realized gain variation over the operating frequency band. As can be inferred from this table, the proposed antenna offers the best compromise in terms of size and bandwidth. The electrical size of the radiator of our antenna is only 0.3λ₀ × 0.3λ₀ × 0.09λ₀ while the size of the cavity is about 0.58λ₀ × 0.58λ₀ × 0.19λ₀, where λ₀ represents the wavelength at 97 MHz (lowest operation frequency).

Concurrently, it's difficult to compare the proposed design to the one listed in TABLE 5 since our design deals with compact, wideband frequency reconfigurability and circular polarization features. As tabulated in TABLE 5, it can be seen that the contribution in [16]–[18] and [36] exhibits much wider fractional bandwidth than the ones listed in the table, where these designs operate in linear polarization and their overall thickness is more significant than the proposed antenna. Hence, compared with the low-profile designs, the proposed antenna is much compact than the dielectric resonator [18], dielectric-loaded [19], meta-material [20],

geometry folded [21] MED designs. The proposed frequency reconfigurable compact antenna yields to a better bandwidth of 120% with two fractional bandwidths of 66.2% and 77.3% in DP mode, a low-profile thickness of 0.09λ₀, which corresponds to a 50% reduction factor, in addition to a wide AR bandwidth (AR < 3 dB) of 85%. The proposed antenna achieved an effective bandwidth of 120% from 93 to 370 MHz while considering a common bandwidth limited by an SWR < 3 and an AR < 3 dB.

Moreover, none of the listed designs provide any wideband frequency reconfigurability. The key advantages of our proposed antenna compared to previous designs are summarized here. Specifically, our proposed design is the most compact (0.09λ₀), exhibits two wide bandwidths operating states reconfigurable and provides circular polarization. Hence, our design establishes a new type of frequency reconfigurable antennas well suited for small-satellite applications. To our knowledge, this has not been achieved before, specifically the validation of this concept by the manufacture of a prototype and its complete radio measurements in the VHF band, which requires unusual measurement means (large shielded anechoic chamber).
V. CONCLUSION

A novel frequency reconfigurable low-profile ultra-wideband crossed MED antenna in VHF Band is proposed. The crossed MED antenna height is reduced to 0.09λ₀ by adding a capacitive loading on the antenna structure. This height reduction is used to develop a wideband MED solution based on frequency band commutation using RF switches added to the antenna structure to modify the antenna length. Consequently, a combination of two operating MED states leads to a wide-band frequency extending from 93 to 370 MHz with an SWR <3 and directional radiation presenting broadside realized gain varying from 4 to 11 dBi. A prototype has been designed and fabricated in the VHF band. The switching system is the subject of specific developments aimed at reducing its impact on antenna performances. The experimental measurements carried out over an unusual VHF band validate the excellent antenna performances expected from simulations. The antenna finds potential applications in future wireless communications and space applications.

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**ABDUL-SATTAR KADDOUR** (Member, IEEE) received the B.S. degree in electronics from the Faculty of Science, Lebanese University, Lebanon, in 2011, the B.S. and M.S. degrees in electronics and embedded systems engineering from the Grenoble Institute of Technology, Grenoble, France, in 2012 and 2014, respectively, and the Ph.D. degree in optics and radiofrequency from the Grenoble Alpes University, Grenoble, in 2018. His Ph.D. has been realized with CEA-LETI, Grenoble, where he developed electrically small and frequency reconfigurable antennas for space applications. In 2018, he joined the Xlim Research Institute, Limoges, France. Since 2019, he has been a Research Fellow with the Transforming Antennas Center (TAC), Florida International University, Miami, Florida, USA. He has four issued patents in the US Patent Office. His main research interests include electrically small antennas, reconfigurable antennas, antenna arrays, reflectarrays, and wireless power transfer systems. He serves as a reviewer for the numerous IEEE journals in the field of microwave, antennas and propagation.

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**SERGE BORIES** received the Engineer Diploma degree in electronics from the Ecole Nationale de l'Aviation Civile, Toulouse, France, in 2002, with a focus on microwaves, and the Ph.D. degree in UWB antenna design for multimedia communications, in 2006. Since 2006, he has been working with the Antenna and Propagation Laboratory, CEA, LETI, Grenoble, France. His research interests include UWB antennas, miniature antennas design and characterization, low energy/low EM exposure radio technologies, and direction finding antennas.

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**ANTHONY BELLION** received the Engineer Diploma degree in electronics from the Ecole Nationale Superieure d'Ingénieurs de Limoges (ENSIL), Limoges, France, in 2003, and the Ph.D. degree in optical communication and microwave techniques from the University of Limoges, Limoges, France, in 2008. He was working on multi polarizations and UWB antenna design for direction finding systems. After industrial experience with Thales, France, he joined the Antenna Department of the French Space Agency (CNES). Since 2016, he has been headed the CNES Antenna Department. His research interests include UWB antennas and miniature antennas design and characterization.

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**CHRISTOPHE DELAVEAUD** (Member, IEEE) received the Ph.D. degree in optical communication and microwave techniques from the University of Limoges, Limoges, France, in 1996, and the Accreditation degree in direct physics research from the University Joseph Fourier, Grenoble, France, in 2010.

After industrial experience with Radiall Voreppe, France, he joined the Electronic and Information Technology Laboratory, French Alternative Energies and Atomic Energy Commission (CEA-LETI), Grenoble, in 2002. Since 2010, he has been headed the Antennas Propagation and Inductive Coupling Laboratory (LAPCI), CEA-LETI. He was appointed as the Research Director of the CEA, in 2019. Until now, he has contributed to more than 250 papers and communications in major revues and conferences in the field of antennas and radio frequency circuits. He is the author or coauthor of 19 patents of invention. His research interests include the range from electromagnetism applied to antennas (miniature antennas, multiantenna systems, and reconfigurable antennas) to propagation and microwave circuits for radio frequency transmitters.

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