# Development of a Modular High Current DC Power Supply for Plasma Torch

Noman Khan, Tanveer Abbas, Muhammad Mubassir Hassan, Nouman Saeed, Naseer Ahmed, Christophe Pasquier, Khalil El Khamlichi Drissi, Kambiz Tehrani

<figure><img src="image_1.png" /></figure>

## To cite this version:

Noman Khan, Tanveer Abbas, Muhammad Mubassir Hassan, Nouman Saeed, Naseer Ahmed, et al. Development of a Modular High Current DC Power Supply for Plasma Torch. The 26th European Conference on Power Electronics and Applications, GDR SEEDS France & EPE Association, Mar 2025, Paris, France. 10.34746/epe2025-0276. hal-05084846v2

HAL Id: hal-05084846

https://utc.hal.science/hal-05084846v2

Submitted on 26 May 2025

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Distributed under a Creative Commons Attribution 4.0 International License
# Development of a Modular High Current DC Power Supply for Plasma Torch

Noman Khan, Tanveer Abbas, Muhammad Mubassir Hassan, Nouman Saeed, Naseer Ahmed, Christophe Pasquier, Khalil El Khamlichi Drissi, Kambiz Tehrani

► To cite this version:

Noman Khan, Tanveer Abbas, Muhammad Mubassir Hassan, Nouman Saeed, Naseer Ahmed, et al. Development of a Modular High Current DC Power Supply for Plasma Torch. The 26th European Conference on Power Electronics and Applications, GDR SEEDS France & EPE Association, Mar 2025, Paris, France. 10.34746/epe2025-0276. hal-05084846

HAL Id: hal-05084846

https://utc.hal.science/hal-05084846v1

Submitted on 26 May 2025

HAL is a multi-disciplinary open access
archive for the deposit and dissemination of sci-
entific research documents, whether they are pub-
lished or not. The documents may come from
teaching and research institutions in France or
abroad, or from public or private research centers.

L'archive ouverte pluridisciplinaire HAL, est
destinée au dépôt et à la diffusion de documents
scientifiques de niveau recherche, publiés ou non,
émanant des établissements d'enseignement et de
recherche français ou étrangers, des laboratoires
publics ou privés.

<figure><img src="image_3.png" /></figure>

Distributed under a Creative Commons Attribution 4.0 International License
# Development of a Modular High Current DC Power Supply for Plasma Torch

Noman Khan¹, Tanveer Abbas¹, Muhammad Mubassir Hassan¹, Nouman Saeed¹,
Naseer Ahmed², Christophe Pasquier³, Khalil El Khamlichi Drissi³, Kambiz Tehrani³

¹Pakistan Institute of Engineering and Applied Sciences (PIEAS), Nilore, Islamabad 45650, Pakistan

²Pakistan Tokamak Plasma Research Institute (PTPRI), Islamabad 3329, Pakistan

³University of Clermont Auvergne, Clermont Auvergne INP, CNRS, Institut Pascal, F-63000 Clermont-Ferrand, France

Corresponding author e-mail: kambiz.tehrani@uca.fr or kambiz.tehrani@ieee.org

**Abstract**—This paper presents a modular and indigeneous design of a 400 A / 120 V power supply for a plasma torch. The design employs three LLC resonant converters operating in parallel to power the plasma torch. The proposed design results in reduced electromagnetic component size, and offers convenience in fabrication, testing, and maintenance. Detailed calculations, simulation validation, and fabrication of a single 135 A / 120 V LLC converter module are evaluated and presented. The main features include the surge repression circuit, soft-switching for high efficiency, and dynamic response of the supply. The proposed system demonstrates practicality for high-current application in resource-constraint environments.

**Index Terms**—Plasma torch, high current power supply, LLC resonant converter, high-current rectifiers.

## I. INTRODUCTION

Plasma, also known as the fourth state of matter, is a hot mixture of ionized gases and charged particles. Plasma is a good conductor of electricity and may be heated, and sustained by passing electrical current through it [1]. Plasma may be heated to a very high temperature for a high energy density ($10^6-10^7$ Jm$^{-3}$) and a high heat flux density ($10^7-10^9$ Wm$^{-2}$) [2]–[4]. With recent technological developments, plasma has been employed in many industrial applications, such as welding, cutting, waste treatment, energy conversion, aerospace and so on [5]. A plasma torch is one of the devices used to generate, heat, and sustain high-temperature plasma [5].

A plasma torch acts as a highly non-linear electrical load with a power rating in the range of 10s of kW to MW. A few commercial plasma torches are listed in Table I with their typical applications. Pakistan Tokamak and Plasma Research Institute (PTPRI) has indigenously designed and fabricated a plasma torch with specifications

<table><caption>TABLE I<br>TYPES AND APPLICATIONS OF PLASMA TORCHES [1], [6]–[8]</caption><thead><tr><th>Torch make</th><th>Power MW</th><th>Voltage (V)</th><th>Current (A)</th><th>Applications</th></tr></thead><tbody><tr><td>Huls 8</td><td>0.5–8.4</td><td>2500-7000</td><td>200-1200</td><td>Iron smelting</td></tr><tr><td>Linde-Retech</td><td>0.1–0.75</td><td>200-250</td><td>500-3000</td><td>Plasma melting</td></tr><tr><td>Westinghouse</td><td>0.15–2</td><td>500-1000</td><td>300-2000</td><td>Iron smelting</td></tr><tr><td>Trioxide</td><td>0.5–5</td><td>1700</td><td>350</td><td>TiO<sub>2</sub> Production</td></tr><tr><td>PEC</td><td>0.25–4.5</td><td>600-900</td><td>420-5000</td><td>Iron smelting</td></tr><tr><td>Aerospatiale</td><td>0.3–5</td><td>750-2630</td><td>400-1900</td><td>FeMn production</td></tr><tr><td>SKF</td><td>6–7</td><td>2500-3500</td><td>1800-2200</td><td>FeCr processing</td></tr><tr><td>Ours</td><td>0.048</td><td>120-150</td><td>400</td><td>Waste treatment</td></tr></tbody></table>

Conventional power supplies based on simple diode rectifiers and line frequency choppers offer trivial and simple solutions for several applications such as electrolyzers [9], [10]. However, such power supplies offer poor regulation, high input current THD, low power factor, and sluggish dynamic response. To address these issues, different approaches are used, such as multi-phase diode/thyristor rectifiers with on-load tap changing transformers (OLTC) and saturable reactors [11], [12]. However, OLTCs are used when the load variations are not sudden. Additionally, the maintenance cost is quite high due to the mechanical movements involved. A 12-pulse diode rectifier followed by DC link and series-connected IGCT-based DC-DC converters for the output voltage regulation is proposed in [12]. The input power factor is improved in this case, however, the current ratings of the IGCTs are very high and the switching takes place at a low frequency of 300 Hz resulting in the requirement of bulky inductors for filtering. So, the conventional DC power supplies are not suitable for the plasma torch because of their sluggish dynamic response.

An 1800 A / 160 V buck converter for a 288 kW
plasma torch is presented in [13]. The realization of a solid-state switch for 1800 A is a challenge. Multiple solid-state devices are used in parallel to realize the switch in simulation. A 2.4 kW (40A, 60V) hardware prototype is presented. The full-scale implementation requires dozens of solid-state devices and poses other issues such as synchronized switching and unequal current sharing. This solution is not a good choice for the plasma torch. A power supply design based on a resonant converter is presented in [14] to harvest the advantages of soft switching in the form of reduced current stresses on the switches. However, the proposed design is bulky for a phase-shifting transformer required to realize 18-pulse rectification. In [11], [15], [16] modular multi-level converters (MMCs) are suggested with transformers operating at medium/high frequency. Hardware complexity of MMCs makes them unattractive for applications such as the plasma torches. Active/hybrid rectification [17], rectifiers followed by choppers [18], [19], and third harmonic injection method [20] are also among the related works. These ideas are validated in simulations and/or low-power prototypes, and suffer from many challenges if realized in hardware for high-current applications such as the one under consideration [15].

Resonant converters are recognized for their high power density and high power efficiency. Zero current switching (ZCS) and zero voltage switching (ZVS) in resonant converters reduce switching loss, improving power efficiency, and reliability [21], [22]. This work considers an LLC resonant converter-based design for the required 400 A / 120 V power supply. Single high-current module-based solutions face challenges such as increased component size, complex fabrication, and limited scalability [23], [24]. The initial design calculations resulted in a large size of the magnetic components as presented in [23]. We foresaw the challenges of fabrication and testing for such large components and decided to opt for modular design. The desired power supply is then designed in three parallel connected modules to drive the common load of the plasma torch. The modular design approach achieves significant improvements in size, efficiency, and reliability. Unlike a single full-scale system, the modular architecture provides flexibility, scalability, and ease of maintenance. Additionally, the indigenous development of the power supply using locally available components and lab-fabricated magnetic components underscores the system's practicality for resource-constrained environments. The main contributions of this paper are listed as follows:

1) Hardware fabrication and evaluation of a 135 A / 120 V LLC resonant converter using locally sourced components.

2) Integration of a surge current suppression strategy to improve reliability and component longevity.

3) Demonstration of soft-switching in the resonant converter to reduce voltage/current stresses and improve power efficiency.

4) Experimental analysis of input current THD and load dynamics highlighting the practical viability of the proposed design.

The remainder of the paper is organized as follows. Section II describes the overall system design and proposed modular architecture. Section III presents the hardware implementation and experimental results for a 135 A/120 V module. Section IV presents the full-power hardware testing results. Section V concludes the paper with main findings and future directions.

## II. DESCRIPTION OF THE MODULAR APPROACH

The proposed modular LLC resonant inverter-based power supply for the plasma torch is shown in Fig. 1. The proposed converter consists of a three-phase hex-bridge rectifier, LC filter, H-bridge inverter, resonant tank, high-frequency step-down transformer, high-frequency bridge rectifier, output filter, and the equivalent plasma torch load. In addition to this, the converter consists of several auxiliary systems such as a gate-drive circuit, and associated measurement and control systems.

The three-phase AC input is rectified by a hex-bridge rectifier and filtered through an LC filter ($L_f, C_f$). $C_f$ acts as a DC-link capacitor with a constant voltage of 515 V. The rectifier output ripple frequency is 300 Hz, therefore, $L_f$ is wound on a silicon steel core. The DC-Link ($C_f$) maintains 515 V and acts as a source for the single-phase H-bridge inverter. The output of the H-bridge inverter excites the resonant tank with a high-frequency quasi-square wave voltage. A high-frequency step-down pulse transformer is employed to reduce the voltages to the desired level necessary for the operation of the plasma torch. The high-frequency inverter helps reduce the size of the transformer. As this is a high-frequency transformer, a ferrite core will be used to wound this transformer. The resonant tank current is then rectified through a high-frequency rectifier and filtered through the output filter ($C_{out}, L_{out}$) to provide a constant DC to the plasma torch.

The authors have performed a detailed analytical and simulation analysis in [23], and it is concluded that it is beneficial to use a modular approach rather than a single converter. The DC link energizes three identical modules each capable of supplying 135 A operating in parallel to meet the current requirements of the plasma torch. The design summary of System-I (input rectifier) is provided in Table II and the design
<figure><img src="image_4.png" /><figcaption>System-II: Resonant Inverter</figcaption></figure>

<caption>System-III: Output Rectifier</caption>

<caption>Fig. 1. Overall scheme of proposed converter for 48 kW (400 A / 120V) plasma torch</caption>

<table><caption>TABLE II
DESIGN SUMMARY OF SYSTEM-I (INPUT RECTIFIER)</caption><thead><tr><th colspan="2">Design Parameters</th><th>Values</th></tr></thead><tbody><tr><td rowspan="4">Input Rectifier Ratings</td><td>Peak inverse voltage</td><td>540 V</td></tr><tr><td>Peak repetitive current (I<sub>Dp</sub>)</td><td>122.5 A</td></tr><tr><td>RMS current (I<sub>Drms</sub>)</td><td>57 A</td></tr><tr><td>Average current (I<sub>Davg</sub>)</td><td>33 A</td></tr><tr><td rowspan="2">Capacitor (C<sub>f</sub>)</td><td>Capacitance</td><td>6000 μF</td></tr><tr><td>Voltage rating</td><td>515 V</td></tr><tr><td rowspan="6">Inductor (L<sub>f</sub>)</td><td>Inductance</td><td>722 μH</td></tr><tr><td>Peak current (I<sub>Lp</sub>)</td><td>122.5 A</td></tr><tr><td>RMS current (I<sub>Lrms</sub>)</td><td>99 A</td></tr><tr><td>Average current (I<sub>Lavg</sub>)</td><td>98 A</td></tr><tr><td>Area Product (A<sub>c</sub>A<sub>w</sub>)<sub>Lf</sub> (mm<sup>4</sup>)</td><td>5.78×10<sup>6</sup></td></tr><tr><td>Silicon-Steel core model</td><td>EI-150</td></tr></tbody></table>

summary of System-II (resonant inverter) and System-III (output rectifier) is provided in Table III and Table IV respectively (the detailed procedure is provided in [23]). The next section details the hardware implementation and testing of each system component to validate the modular design presented in [23].

### III. IMPLEMENTATION OF A 135 A/ 120 V MODULE

In this section, we present the implementation of a 135 A / 120 V power supply for the plasma torch. The steps involved in the hardware implementation of each component in the power supply are discussed in detail. Moreover, the challenges faced in hardware realization

<table><caption>TABLE III
DESIGN SUMMARY OF SYSTEM-II (RESONANT INVERTER)</caption><thead><tr><th colspan="2">Design Parameters</th><th>Values</th></tr></thead><tbody><tr><td rowspan="4">H-Bridge Ratings</td><td>Peak current (I<sub>Sp</sub>)</td><td>52 A</td></tr><tr><td>Average current (I<sub>Savg</sub>)</td><td>17 A</td></tr><tr><td>Forward blocking voltage</td><td>515 V</td></tr><tr><td>Inductance</td><td>131.16 μH</td></tr><tr><td rowspan="5">Resonant Inductor (L<sub>r</sub>)</td><td>Peak current (I<sub>Lp</sub>)</td><td>52 A</td></tr><tr><td>RMS current (I<sub>Lrms</sub>)</td><td>37 A</td></tr><tr><td>Area Product (A<sub>c</sub>A<sub>w</sub>)<sub>Lr</sub> (mm<sup>4</sup>)</td><td>0.9626 ×10<sup>6</sup></td></tr><tr><td>Ferrite core model</td><td>EE-100</td></tr><tr><td>Core volume (mm<sup>3</sup>)</td><td>222726</td></tr><tr><td rowspan="4">Resonant Capacitor (C<sub>r</sub>)</td><td>Core weight</td><td>1176 g</td></tr><tr><td>Capacitance</td><td>0.4824 μF</td></tr><tr><td>Peak voltage rating</td><td>806 V</td></tr><tr><td>Turn Ratio (N<sub>p</sub>:N<sub>s</sub>)</td><td>4.28:1</td></tr><tr><td rowspan="4">High Frequency Transformer</td><td>Area Product (A<sub>c</sub>A<sub>w</sub>)<sub>T,F</sub> (mm<sup>4</sup>)</td><td>1.83 ×10<sup>6</sup></td></tr><tr><td>Ferrite core model</td><td>EE-110</td></tr><tr><td>Core volume (mm<sup>3</sup>)</td><td>311810</td></tr><tr><td>Weight</td><td>1564 g</td></tr></tbody></table>

and their solutions are also discussed. A systematic procedure for hardware testing and evaluation is provided.

#### A. Three Phase Rectifier

The circuit is energized from a three-phase variac transformer. Since only one module is being considered for prototype development, the input rectifier's current ratings are reduced by 1/3. A hex-bridge rectifier module (RM20TPM-H) with an average current rating of 40 A / 800 V is used for rectification. The output of the rectifier
<table><caption>TABLE IV
DESIGN SUMMARY OF SYSTEM-III (OUTPUT RECTIFIER)</caption><thead><tr><th colspan="2">Design Parameters</th><th>Modular Unit</th></tr></thead><tbody><tr><td rowspan="3">Output Rectifier Ratings</td><td>Peak inverse voltage</td><td>150 V</td></tr><tr><td>Peak current (I<sub>Dp</sub>)</td><td>210 A</td></tr><tr><td>Average current (I<sub>Davg</sub>)</td><td>66 A</td></tr><tr><td rowspan="2">Output Capacitor (C<sub>out</sub>)</td><td>Capacitance</td><td>196 μF</td></tr><tr><td>Voltage rating</td><td>150 V</td></tr><tr><td rowspan="6">Output Inductor (L<sub>out</sub>)</td><td>Inductance</td><td>29.84 μH</td></tr><tr><td>Peak current (I<sub>p</sub>)</td><td>135 A</td></tr><tr><td>RMS current (I<sub>Lrms</sub>)</td><td>132 A</td></tr><tr><td>Area product (mm<sup>4</sup>)</td><td>2.0463 × 10<sup>6</sup></td></tr><tr><td>Ferrite core model</td><td>EE-130</td></tr><tr><td>Core volume (mm<sup>3</sup>)</td><td>489600</td></tr><tr><td colspan="2">Core weight</td><td>2340 g</td></tr></tbody></table>

<figure><img src="image_5.png" /><figcaption>Fig. 2. Strategy employed to reduce the three-phase rectifier stage surge current</figcaption></figure>

<figure><img src="image_6.png" /><figcaption>Fig. 3. Experimental waveforms of surge current (bottom) and DC-link voltages (top) without surge repression circuit</figcaption></figure>

<figure><img src="image_7.png" /><figcaption>Fig. 4. Experimental waveforms of charging current (bottom) and DC-link voltages (top) with surge repression circuit</figcaption></figure>

is filtered with an LC filter to achieve a clean DC for the inverter.

When the circuit is powered ON from a cold-start state, a huge surge current flows from the rectifier, and the circuit breaker with the AC mains trips. The strategy employed to overcome this issue is shown in Fig. 2. A three-phase AC circuit breaker (CB) is used at the input of the rectifier to protect the circuit in case of an overcurrent fault. Due to the high-power system instead of using a CB to connect and disconnect the circuit from AC mains, a magnetic contactor (contactor 1) is used. To suppress the surge current at the start a power resistor bank is added to the circuit, which is then bypassed with another contactor (contactor 2) once the DC-link capacitor is charged to its nominal value.

The results obtained with a three-phase supply (515 V<sub>DC</sub>, and no-load state) before and after using the surge repression circuit are provided in Fig. 3 and Fig. 4 respectively. In Fig. 3 the DC-Link capacitor charging time is relatively fast (10 ms), however, a surge current of 200 A flows from the circuit. Hence, it is desired that the surge repression circuit is included. After using the surge repression circuit, the response is recorded and shown in

Fig. 4. The rise time of the voltage is increased but the surge is suppressed. This increases the system reliability and the life span of the semiconductor devices. Moreover, here the first-time charging time of the capacitors is not important. Therefore, the experimental results for the input three-phase rectifier stage are acceptable and safe as desired.

## B. Resonant Inverter

The filter $C_f$ acts as a DC link for the H-bridge inverter. The DC link maintains a constant voltage of 515 V for the inverter. The typical behavior of a MOSFET/IGBT switch is shown in Fig. 5. When the switch is ON, i.e., conducting a nominal current $I_D$ flows through the switch and has a negligible voltage drop across it, or in an ideal case voltage drop is zero, i.e., $V_{DS} = 0$. Hence, power loss during the ON state = $P_{ON} = V_{DS} \times I_D = 0$. Similarly, during the OFF state, the current from the switch is zero which results in $P_{OFF} = 0$. However, during the transition state, i.e., when the switch changes its state from ON to OFF or from OFF to ON there is a significant overlap between
<figure><img src="image_8.png" /><figcaption>Fig. 5. Time-domain representation of switching and conduction losses in a semiconductor switch</figcaption></figure>

voltage and current. This power loss is determined as $P_{sw} = I_{sw} \times V_{sw}$. Where, $I_{sw}$ is the current from the switch, and $V_{sw}$ is the voltage across the switch during this transition. These losses can be more dominant at higher frequencies due to hard-switching in the conventional SMPS [25]. The term hard switching is associated with the concept that when a switch is forced to change its state either by,

* applying gate pulse to turn on the switch when there is a considerable amount of voltage across the switch (in OFF state) or,

* withdrawing the gate pulse to turn off the switch when a considerable amount of current flows through the switch.

Therefore, to reduce these switching losses resonant converter is used as it can support soft-switching. Soft switching refers to the technique that reduces the switching losses and the electromagnetic interference (EMI) in the power electronic circuits [21]–[23], [26]. In soft-switching the voltage or the current is shaped such that the transition of the switch from ON to OFF, or vice versa happens at the zero crossing. If this transition occurs when zero current is flowing through the switch, it is referred to as zero current switching (ZCS) [26]. Similarly, if the transition happens when the voltage across the switch is zero it is referred to as zero voltage switching (ZVS) [26].

The schematic circuit diagram shown in Fig. 1 is simulated in PSIM and the corresponding voltage and current waveforms in the resonant tank are recorded as shown in Fig. 6 [23]. The LLC resonant tank acts as a band-pass filter for this square pulse and allows only the sinusoidal component of current to pass through the resonant tank [21]–[24], [26]. The same current passes through the switching network of the H-bridge. It can be seen that the switching happens at the zero crossing, hence reducing the switching losses and improving the efficiency of the inverter. Soft-switching in resonant

<figure><img src="image_9.png" /><figcaption>Fig. 6. Simulation waveforms of resonant tank voltage and current</figcaption></figure>

inverters helps achieve higher powers at high frequencies which is difficult through other inverters.

For hardware implementation, the gate-drive circuit for the H-bridge inverter is shown in Fig. 7, and the PCB is developed in Altium Designer. TLP 251 optocouplers are used as gate drivers, and the control signal is generated by a Texas Instrument microcontroller (LaunchXL-F280025C). Two IGBT half-bridge modules (CM200DU-24H) make up the single-phase H-bridge inverter.

The inverter when tested at low voltages showed huge voltage spikes at the switching instances. It was due to the stray inductance introduced due to the length of the wires from the DC-link to the inverter module (between collector and emitter). A decoupling capacitor of 100 μF was used across each half-bridge module to resolve this issue. Moreover, the DC wires from the DC link to the switch modules were twisted and tied together so the inductance of each wire could be minimized, reducing the spikes across the inverter. Additionally, RC snubbers and varistors are employed across each IGBT for protection. Moreover, the length of the connecting cables from the gate drive circuit to the gate terminal of the inverter also introduces stray inductance which results in spikes across the base. This can permanently damage the switch if the spikes exceed the rated base-to-emitter voltage. To resolve this issue a small damping resistor (10-50 Ω) and a small capacitor ($C_{BE}$) were added near the gate terminals on the inverter module.

The inverter output is fed to the resonant tank, which causes the current to become sinusoidal. The inductive nature of the LLC-resonant network above resonance can be exploited to achieve ZVS. Fig. 8 shows the inductive behavior of the resonant current ($I_r$), which lags behind the voltage. When the voltage is switched, the current is non-zero. This non-zero current passes through the antiparallel diodes of the next pair of IGBT switches, which turn on. The current flow through the anti-parallel diodes ensures that the voltage across the IGBT switch is zero when the gate pulse is applied to turn it ON. The current is pure sinusoid and hence the phenomena of soft-switching in the resonant converter occurs.
<figure><img src="image_10.png" /><figcaption>Fig. 7. Bootstrapping based-gate drive circuit for H-bridge inverter</figcaption></figure>

<figure><img src="image_11.png" /><figcaption>Fig. 8. Experimental waveforms during resonant inverter testing. (Red) voltage ($V_{CE}$ across the switch (20 V/div). (Blue) resonant tank current waveforms (100 mA/div)</figcaption></figure>

<figure><img src="image_12.png" /><figcaption>Fig. 9. Zero voltage switching in resonant inverter. (Blue) gating signal ($V_{GE}$=5 V/div). (Red) voltage across the IGBT ($V_{CE}$=250 V/div)</figcaption></figure>

The voltage waveform across the IGBT and the gate signal is shown in Fig. 9. It is observed that when the voltage across the collector to the emitter drops to zero then the gate signal is applied to turn on the switch. The experimental results of the resonant tank voltage and current at the resonance frequency are shown in Fig. 10. In this result, the oscilloscope channel A (current) is inverted (180° phase shifted) for better visibility. It is observed that the switching in the resonant tank always occurs at zero-crossing at resonance. Hence, the switching losses are minimized in this manner.

<figure><img src="image_13.png" /><figcaption>Fig. 10. (Red) experimental waveform of resonant tank voltage (200 V/div) and (blue) resonant tank current (10 A/div)</figcaption></figure>

### C. High-frequency Step-down Transformer

As calculated in Table III, EE-130 is required for the fabrication of this transformer. However, due to the limitation of ferrite core availability a smaller core is

used, and the finalized configuration is shown in Fig. 11. In this configuration, three smaller transformers having a voltage ratio of 172:120 are fabricated. The primaries
<figure><img src="image_14.png" /><figcaption>Fig. 11. Circuit diagram for the realization of the high-frequency step-down transformer</figcaption></figure>

<figure><img src="image_15.png" /><figcaption>Fig. 12. Simulation waveforms at the primary and secondary winding of the transformer</figcaption></figure>

of these transformers are connected in series (to relieve the voltage stresses on the primary) and the secondaries are connected in parallel. Since this is a step-down transformer and the current on the secondary side is large, i.e., 135 A, therefore, the adopted configuration provided an additional benefit of a reduced wire gauge for winding the secondary of the transformer. Due to high frequency and high current, the skin effect in the winding will also increase, therefore, to avoid it Litz wire is used.

The configuration is first verified in circuit simulations by connecting a square voltage source of ±515 V / 20 kHz, and at the secondary winding, a resistive load of 0.9 Ω is connected. Fig. 12 shows the voltage and current waveforms at the primary and secondary winding of the transformer. It is observed that the voltage and shape remain the same at the primary and secondary winding of the transformer, moreover, the voltage and current ratios are also satisfied. Hence, these simulation results validate the idea of realizing a large rating transformer using multiple small transformers. This configuration helps in reducing the complexity (by lowering voltage insulation levels, current ratings, and thermal power dissipation in the transformer) and the summary of one such transformer is provided in Table V.

The transformers are fabricated in the lab and connected according to Fig. 11. The transformer working

<table><caption>TABLE V
DESIGN SUMMARY FOR STEP-DOWN TRANSFORMER</caption><thead><tr><td>Sr. No.</td><td>Parameters</td><td>Values</td></tr></thead><tbody><tr><td>1.</td><td>Area Product (A<sub>c</sub>A<sub>w</sub>)<sub>TF</sub></td><td>0.5778 × 10<sup>6</sup> mm<sup>4</sup></td></tr><tr><td>2.</td><td>Ferrite Core</td><td>EC-120</td></tr><tr><td>3.</td><td>Primary Turns (N<sub>p</sub>)</td><td>16 turns</td></tr><tr><td>4.</td><td>Primary Current (I<sub>p</sub>)</td><td>35 A</td></tr><tr><td>5.</td><td>Wire Gauge (SWG) / Strands</td><td>28 / 170</td></tr><tr><td>6.</td><td>Secondary Turns (N<sub>s</sub>)</td><td>12 turns</td></tr><tr><td>7.</td><td>Secondary Current (I<sub>s</sub>)</td><td>45 A</td></tr><tr><td>8.</td><td>Wire Gauge (SWG) / Strands</td><td>28 / 223</td></tr></tbody></table>

<figure><img src="image_16.png" /><figcaption>Fig. 13. Experimental validation of the high-frequency step-down transformer. (Top) transformer primary winding voltages (500 V/div). (Bottom) transformer secondary winding voltages (100 V/div)</figcaption></figure>

and turn ratio is then verified experimentally. The experimentally achieved results are shown in Fig. 13. An input of ±515 V / 20 kHz from the resonant inverter energizes the primary of the transformer and an output of ±120 V is recorded at the secondary winding of the transformer (which corresponds to a turn ratio of ≈4). Therefore, the experimental results also validate the adopted configuration.

### D. Output Single Phase Rectifier

The output of the high-frequency step-down transformer is rectified with a single-phase full bridge rectifier. Due to rectification, the frequency of the ripple at the output is doubled, i.e., 40 kHz. Therefore, high-frequency diodes (VS-75EPU12L-N3) having an average current rating of 75 A are used. Since the current at the output is high, therefore, to avoid thermal stresses the rectifier connections are made using copper strips.

The output of the rectifier is then filtered through a capacitive filter of 196 μF / 150 V. At the output a bank of power resistors having an equivalent resistance of 1 Ω is connected. The complete testing and input/output response of the power supply are discussed in the next section.
<figure><img src="image_17.png" /><figcaption>Fig. 14. Prototype developed for the single module of LLC resonant converter-based power supply (120 V / 135 A)</figcaption></figure>

<figure><img src="image_18.png" /><figcaption>Fig. 15. Experimental waveform of the phase current (20 A/div)</figcaption></figure>

<figure><img src="image_19.png" /><figcaption>Fig. 17. Power supply testing at full power. (Top) output voltage (100 V/div) and (bottom) output current (100 A/div)</figcaption></figure>

<figure><img src="image_20.png" /><figcaption>Fig. 16. FFT spectrum of the input current (vertical 5 A/div, center frequency 1.75 kHz, horizontal 250 Hz/div)</figcaption></figure>

## IV. FULL-POWER HARDWARE TESTING

In the previous section experimental verification of individual stages of the power supply is accomplished. This section presents the full-power testing of the power supply along with its dynamic response. Fig. 14 shows

the developed full-scale converter stage of 120 V / 135 A.

Fig. 15 shows the input current drawn by the power supply. Its FFT spectrum is recorded and shown in Fig. 16. The RMS value of the fundamental component is 18 A, and the overall THD of the system is ≈ 36%. The most dominant lower-order harmonic exists at 250 Hz (5^th^), and 350 Hz (7^th^) having a magnitude of 6 A and 2.5 A (RMS). The rest of the higher-order harmonics are relatively smaller in magnitude, therefore, only 5^th^ and 7^th^ are contributing the most in determining the overall THD of the power supply. Therefore, appropriate filtering needs to be considered in the future to reduce the THD and improve the power quality of the system.

The power supply's output voltage and current waveform at full load are recorded and shown in Fig. 17. The impact of connecting the load to the power supply is
<figure><img src="image_21.png" /><figcaption>Fig. 18. DC-link profile in dynamic loading condition</figcaption></figure>

<figure><img src="image_22.png" /><figcaption>Fig. 19. Output of the supply in dynamic loading condition. (Top) Output voltage (100 V/div) and (bottom) current (100 A/div)</figcaption></figure>

also studied. The DC-link is charged to 515 V and then the load is connected to the supply at 100 ms. As soon as the load is connected a dip in DC-link voltages is observed and the DC-link voltages reduce below 500 V (as shown in Fig. 18). Its impact can be seen in the output (Fig. 19) due to which the output voltage reduces from 120 V to 93 V. Since the load is very small (1 Ω) the stray resistance of the designed components plays a major role and hence the voltages on DC-link reduces at full load. In the developed setup, it can be compensated by reducing the step-down ratio of the designed high-frequency transfer. Therefore, the design of the 135 A/ 120 V resonant converter-based power supply for the plasma torch has been validated experimentally.

## V. CONCLUSIONS

This paper presented the design and development of a 400 A / 120 V DC power supply for a plasma torch. Such a power supply requires large magnetic components that result in inconvenience in fabrication, testing, and maintenance. Therefore, a modular design of three parallel-connected LLC converter-based power supplies was proposed. The proposed design helped reduce the switch ratings, magnetic component sizes, and

cooling requirements. Hence, it makes the indigenous development convenient in the lab. Although the component count increased, the aforementioned advantages outweighed the increased component count. A 135 A / 120 V module is developed based on the components available in the local/lab environment. The issues, design choice, and fabrication procedures are discussed. The developed hardware is thoroughly tested and evaluated. The developed hardware demonstrated a good dynamic response. Robustness against sudden changes in the load is also demonstrated. No significant overshoots or undershoots in the voltage are observed at critical nodes in response to large disturbances in the load. These results confirm the practicality of the proposed system for high-current applications in resource-constrained environments.

Future work will focus on the development of closed-loop control for output regulation and addressing power-sharing challenges between modules (that may arise due to mismatch between modules) to further improve system performance.

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