# Gigahertz, Low-Cost Current Shunt with Retrofitting Capability for Characterization of Fast-Switching GaN Devices

Sebastian Klötzer, Jim Honea, Nils Duchow, Georgii Iashin

► To cite this version:

Sebastian Klötzer, Jim Honea, Nils Duchow, Georgii Iashin. Gigahertz, Low-Cost Current Shunt with Retrofitting Capability for Characterization of Fast-Switching GaN Devices. The 26th European Conference on Power Electronics and Applications, GDR SEEDS France & EPE Association, Mar 2025, Paris, France. 10.34746/epe2025-0216. hal-05095881

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https://utc.hal.science/hal-05095881v1

Submitted on 3 Jun 2025

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# Gigahertz, Low-Cost Current Shunt with Retrofitting Capability for Characterization of Fast-Switching GaN Devices

Sebastian Klötzer, Jim Honea, Nils Duchow, Georgii Iashin

NEXPERIA GERMANY GMBH

Stresemannallee 101, D-22529

Hamburg, Germany

E-Mail: sebastian.kloetzer@nexperia.com

URL: http://www.nexperia.com

*Index Terms*—Current Sensor, Device characterization, Wide bandgap, Gallium Nitride (GaN), Switching losses

*Abstract*—A low-cost current shunt based on a flexible polyimide printed circuit board (PCB) design is presented. With low insertion inductance in the range of 300 pH and bandwidth in excess of 1 GHz, the design is suitable to be used for characterization of very fast GaN devices. Due to its very low thickness, it is possible to retroactively integrate the sensor into many existing designs that were not specifically prepared for current measurement. Thus, it offers an alternative to designing specific PCBs only for switching characterization.

## I. INTRODUCTION

One of the main challenges when evaluating fast-switching power semiconductors is the accurate measurement of the switching currents. Modern gallium-nitride (GaN) field effect transistors (FETs) enable very fast switching transitions of 1 ns or even below [1]. In addition to these high current slopes, state-of-the-art surface-mount package technology enables minimization of power loop inductances to only a few nanohenries. In combination with the reduction of the output capacitance of modern wide bandgap FETs, power loop oscillation frequencies can reach well beyond 100 MHz. To capture the details of these switching currents on the oscilloscope, current sensors with sensor bandwidth of several 100 MHz are necessary.

Various sophisticated solutions for this challenge can be found in literature and they have in common that a dedicated printed circuit board (PCB) with current sensing capability has to be designed with high focus on keeping the increase in power loop inductance as low as possible to not compromise the switching performance. This leads to additional design work and complicates

benchmarking of parts in different packages but also limits the usability of these test platforms to mainly evaluation purposes. Being able to retroactively add current sense capability to an existing platform would not only facilitate the characterization of individual devices but could also be used for other scenarios like paralleling or for troubleshooting. However, available retrofitting solutions are limited in bandwidth, might come with a high price tag and their integration usually requires modifications of the PCBs, potentially increasing loop inductance significantly. Especially when working with very low-inductance SMD packages, the introduced inductance can change the switching behaviour drastically and might not be acceptable. [5], [8]

As a solution to this challenge, the design of a low-cost current sensor with an easy to evaluate output is presented in this paper. The sensor features a ±3dB bandwidth beyond 1 GHz, is optimized for minimal insertion inductance and can be adapted to fit into many existing designs with only small rework.

## II. CURRENT SENSING STATE OF THE ART

The importance of having high-bandwidth current sensing capability with low insertion inductance for characterization of fast-switching devices is well known and there are various different commercial and experimental solutions to this challenge. Table I provides a non-exhaustive overview of current sensors, many of which are targeted specifically for the evaluation of fast GaN and SiC devices. The basic principle of the measurements can be split into two categories. On the one side are inductive sensors that measure the change of the magnetic flux through a wire or PCB trace by using coils formed of a single turn, up to a high number of turns. Common solutions either contain a ferromagnetic core and work as a current transformer [2], [3] or use
<table><thead><tr><th>Current sensor design</th><th>±3 dB frequency</th><th>insertion inductance</th><th>measurement principle</th><th>galvanic isolation</th><th>DC capability</th><th>cost</th></tr></thead><tbody><tr><td>Tektronix split-core probe TCP0030A [2]</td><td>120 MHz</td><td>unspec.</td><td>inductive</td><td>yes</td><td>yes</td><td>high</td></tr><tr><td>Pearson current monitor 6585 [3]</td><td>250 MHz</td><td>unspec.</td><td>inductive</td><td>yes</td><td>no</td><td>high</td></tr><tr><td>PEM UK Rogowski CWTMini50HF [4]</td><td>50 MHz</td><td>unspec.</td><td>inductive</td><td>yes</td><td>no</td><td>high</td></tr><tr><td>PCB integrated pick-up coil [1]</td><td>unspec.</td><td>&lt;0.3 nH</td><td>inductive</td><td>yes</td><td>no</td><td>low</td></tr><tr><td>Univ Bristol Infinity sensor V1 [5]</td><td>225 MHz</td><td>0.2 nH</td><td>inductive</td><td>yes</td><td>no</td><td>low</td></tr><tr><td>Univ Bristol Infinity sensor V2 [6]</td><td>&gt;400 MHz</td><td>&lt;0.2 nH</td><td>inductive</td><td>yes</td><td>no</td><td>low</td></tr><tr><td>T&amp;M Research coaxial shunt [7]</td><td>≤2 GHz*</td><td>2-7 nH [5], [8]</td><td>resistive</td><td>no</td><td>yes</td><td>medium</td></tr><tr><td>Univ. Cambridge UFCS shunt - E [8]</td><td>1.55 GHz</td><td>0.02 nH</td><td>resistive</td><td>no</td><td>yes</td><td>low</td></tr><tr><td>Univ. Cambridge UFCS shunt - I [8]</td><td>394 MHz</td><td>&lt;0.3 nH</td><td>resistive</td><td>no</td><td>yes</td><td>low</td></tr><tr><td>Univ. Tennessee coaxial PCB shunt [9]</td><td>2.23 GHz</td><td>&lt;0.6 nH</td><td>resistive</td><td>no</td><td>yes</td><td>low</td></tr><tr><td>TU Dresden SMD shunt [10]</td><td>unspec.</td><td>&lt;0.5 nH</td><td>resistive</td><td>no</td><td>yes</td><td>low</td></tr><tr><td>TU Bremen M-shunt [11]</td><td>&gt;200 MHz</td><td>ca. 1 nH</td><td>resistive</td><td>no</td><td>yes</td><td>low</td></tr></tbody></table>

<caption>TABLE I
COMPARISON OF STATE-OF-THE-ART CURRENT SENSORS</caption>

the induced voltage for current detection, similar to the Rogowski coil approach. [1], [4]–[6]. With the exception of the split-core probe that utilizes a hall-effect sensor for DC current sensing, all the inductive sensors have a minimal cut-off frequency and cannot be used for DC to low-frequency current measurement. This is typically not a problem when using exclusively for pulsed current switching characterization but it can limit the range of applications.

As the measured variable is already field based, galvanic isolation is provided by all the mentioned solutions, increasing safety and offering the capability and added flexibility to measure currents without reference to ground, for example at the drain side of a high-side device. However, especially the core-less sensors can be susceptible to stray fields near inductive components or capacitive coupling to areas of high $dv/dt$, such as the switch-node of a half-bridge circuit and careful placement is very important. Thus, all of the low-cost experimental variants need to be carefully designed into or onto the PCB at a fixed position during the design stage, which complicates retrofitting onto existing platforms. Also, as the performance depends significantly on the PCB design and post-processing has to be done manually, using these sensors requires a good level of expertise. In theory, the higher cost commercial probes [2]–[4] are more flexible in placement and easier to utilize. But due to their lower bandwidth and as they rely on wrapping around a conductor or device pin, which is typically not available in low-inductance SMD designs without compromising the power loop inductance, they are not well suited for use in an SMD commutation loop or for retroactively fitting onto a wide bandgap evaluation

board.

To achieve higher bandwidth, using the resistive method based on routing the current through a well-known distributed resistance [11] or discrete resistors is an very common alternative. The popular commercial coaxial shunt presented in [7] is often used for high speed switching evaluation and as a reference for evaluating custom sensor designs. The coaxial arrangement leads to a very good level of flux cancellation, which is a prerequisite for achieving low insertion inductance. Similar approaches are used in all of the mentioned sensor designs, either in a coaxial PCB arrangement [9] or by using adjacent PCB layers to create an in-plane cancellation effect. However, due to the size and termination of the commercial shunt, relatively high insertion inductance values of 2-7 nH are reported and significant impact on the switching waveforms has to be expected. The custom designs all demonstrate inductances of less than 1 nH, which makes them better suited for commutation loops with low-inductance packages that can be just as low. In common for all shunt resistors is their capability for DC measurement and a lack of galvanic isolation. Careful placement at the ground reference potential of the oscilloscope is therefore required, making them mainly suited for measuring the source current of low-side devices unless an external isolator is used. In addition, in case of failure of the switching devices, there is high risk of damaging the sensors and potentially other connected circuitry or equipment. Also, the power handling capability of the sensors is quite limited as typically a set of relatively small resistors are used to minimize inductance. An exception to this is design [11], which is able to handle higher power levels

<footnote>*Manufacturer claimed bandpass value. Bandwidth of 50-150 MHz has been reported in [12]</footnote>
thanks to the distributed sensing resistance.

The main advantage of a properly designed current shunt is its resistive behaviour up to very high ±3 dB frequencies which facilitates extraction of the current by simply dividing the measured voltage by the resistor value, but also leaves the door open for post-process optimization at higher frequencies if the transfer characteristic is known. On top of that, as mostly low-cost passive components are required, these designs can be very cost-effective. However, very good design knowledge and significant effort is required when aiming for good performance well beyond 100 MHz. This for example includes the estimation and compensation of shunt resistor parasitics [8], [10]. Thus, the current sensors are typically tied to very specific designs and while retrofitting onto a busbar can be achieved [8], [11], it is nearly impossible to integrate into an existing low-inductance SMD layout. The potential combination of very high bandwidth, low insertion inductance, low-cost and the ease of use for the evaluation makes a solution based on the current shunt principle very attractive for the scope of this paper. However, to achieve these goals in combination with retrofitting capability, instead of using standard PCB material such as FR-4, the sensor will instead be designed onto a very thin flexible PCB that can be placed beneath SMD components with only minimal impact on the behaviour of the circuit.

### III. DESIGN OF THE PROTOTYPES

For the design of the retrofit current shunt, similar design rules apply as for a normal high-frequency shunt on PCB but to keep the layer stack thin, only two layers are used. The low-medium frequency equivalent circuit of the current sensor shown in fig. 1 is comprised of the desired shunt resistance $R_s$ and three main parasitic circuit elements: the external inductance $L_{ext}$ of the power loop that does not reach into the measurement circuit, the common shunt inductance $L_s$ between power loop and measurement loop due to shared current paths or magnetic coupling and the parasitic layer capacitance $C_p$. The impedance $Z_{21}$ formed by the series connection of $L_s$ and $R_s$ with $C_p$ in parallel determines the transfer characteristic $v_{meas}^*/i_{meas}$ for the uncompensated shunt. At low frequencies, the shunt will behave like a purely resistive element but will become more inductive and its gain will increase towards higher frequency due the presence of $L_s$. Measured current slopes will appear steeper on the oscilloscope if this is not addressed. A simple first order low-pass filter formed by $R_c$ and $C_c$ will be used on the sensor, as proposed in [10]. When

<figure><img src="image_2.png" /><figcaption>Fig. 1. Simplified equivalent circuit of the retrofit current sensor</figcaption></figure>

assuming sufficiently small $C_p$, the +3dB gain frequency can be estimated as

$$f_{+3dB} \approx \frac{R_s}{2\pi L_s} \qquad (1)$$

and it is clear that choosing high resistor value and minimizing inductance are both required for good high frequency performance. To measure current peaks up to 100 A when connecting the sensor to the 50 Ω input of an oscilloscope, $R_s$ of 47 mΩ is selected to not exceed the typical 5 V input rating. For low $L_s$, a parallel connection of several resistors is required and a trade-off between inductance and space requirement must be made. Similar to the solutions shown in [13], [14], ten resistors in parallel are chosen for the first prototype. 0603 sized current sense resistors with sensing element on the bottom-side are selected to ensure that both power and measurement loop are not unnecessarily increased beyond the surface of the circuit board.

Especially the choice of layer stack highly influences all three parasitics. As priority is given to reducing inductance, a very thin $d_{sub} = 25\mu$ polyimide (PI) substrate with moderate relative permittivity $\epsilon_r$ of approximately 3.4 is selected. The power path uses wide tracks and due to strong coupling between forward and return path, both $L_s$ and $L_{ext}$ benefit from very good flux cancellation. Due to the way printed circuit boards are typically manufactured, the low layer-to-layer distance of a flexible setup on PI is nearly impossible to achieve with conventional PCBs. However, this leads to higher layer capacitance per area of

$$C_p = \frac{\epsilon_0 \epsilon_r A_{Cu}}{d_{layer}} \approx 1.2 A_{Cu} \frac{pF}{mm^2}. \qquad (2)$$

The power path trace width is adapted to fit the size of the pad for the attachment onto a PCB on the one side and to the width of the shunt array on the opposite side of the sensor. For the first prototype, trace width
$w_{trace}$ = 13 mm is selected. The distance between power pad and the shunt array should be as low as feasible to reduce both $L_{ext}$ and $C_p$ but it should be sufficiently long to ensure good handling of the sensor and to not require an excessive bending radius when fitting onto a tightly populated PCB. Also, keeping a minimum length is useful to give the conductive coaxial connectors some clearance from the potentially high voltage area around the point of attachment if no additional insulating film is used. A distance between the edge of the power pad and the sensing resistors of 10 mm is chosen for the prototype and a copper thickness of 70 µm per layer is selected. This results in a total copper area of ca. 260 mm² and an estimated layer capacitance slightly above 300 pF, according to equation 2. The copper thickness is mainly chosen to improve mechanical stability of the sensor and does not impact the power handling capability as this is limited by the low continuous power rating of the shunt resistors. A depiction of the arrangement of the power and sensing loop of the first prototypes are shown in fig. 2 and a side view of the sensor prototype is given in fig. 3. In addition to this variant with SMA coaxial connector, alternative versions that use MMCX connection targeted for very tight space or high-side operation in combination with an optical isolated probe have been designed.

The geometry of the sensor layout can be used to esti-

<figure><img src="image_3.png" /><figcaption>Fig. 2. Power loop and measurement loop in side view</figcaption></figure>

<figure><img src="image_4.png" /><figcaption>Fig. 3. Side view of a sensor variant with SMA connector. RC-compensation network is located in the center.</figcaption></figure>

mate its high-frequency power loop insertion inductance $L_{ins} = L_{ext} + L_s$. Assuming minimal skin-depth and using the length $l \approx 25$ mm of the $i_{meas}$ current loop

from top pad through the via array to bottom pad yields

$$L_{ins} \geq \frac{\mu_0 l d_{layer}}{w_{trace}} \approx \frac{4\pi 10^{-7} \cdot 25 \text{ mm} \cdot 25 \text{ µm}}{13 \text{ mm}} = 60 \text{ pH.} \quad (3)$$

Impedance analyzer measurement of the prototypes using an SMD fixture has been used to confirm the range of this estimation. However, at the highest useful measurement frequency of ca. 10 MHz, much lower than the relevant frequency for operation, skin-depth is in a similar range as the PI layer thickness of 25 µm, leading to increased effective $d_{layer}$ when using equation 3 [15]. Therefore, measured $L_{ins}$ from pad to pad of 121 pH, as shown in figure 4, is meant to be seen only as an indication, but it confirms that $L_{ins}$ will be very low.

<figure><img src="image_5.png" /><figcaption>Fig. 4. Insertion inductance estimation of the flexible board using an impedance analyzer with SMD fixture</figcaption></figure>

## IV. DESIGN OF THE COMPENSATION NETWORK

The transfer characteristic of the designed shunt array is captured well by the simple representation given in fig. 1 up to a frequency of approximately 100 MHz. At higher frequencies, significant deviations must be expected, particularly due to the presence of the skin effect [9], which leads to an increase in $R_s$ and a decrease in $L_s$. To get a good understanding of the parasitics of the designed prototype, the transfer characteristic of the uncompensated sensor of fig. 3 has been measured using a vector network analyzer (VNA). The measurement has identified a dominating resonance frequency of ca. 1.83 GHz, measurement loop inductance $L_s \approx 21$ pH and parallel capacitance $C_p \approx 360$ pF, 20% higher than
anticipated in equation 2. Without compensation, the sensor will exceed the +3dB = √2 threshold slightly beyond 300 MHz, as shown in fig. 6

Despite the non-linearity at high frequency, assuming constant parameters can be a valid approach for selection of the RC compensation. Around the +3dB threshold, the influence of C<sub>p</sub> is still sufficiently low, so it will be omitted in the first step. A decent starting value for optimization can be computed by matching the cutoff frequency of the RL part of the shunt array and the RC compensation network. An arbitrary, dimensionless scaling factor c<sub>c</sub> is added, as expressed by equation 4.

$$ \frac{L_s}{R_s} = c_c \cdot R_c C_c \quad (4) $$

A trade-off between both values in the compensation network has to be made. On the one hand, it is desirable to choose a low value for R<sub>c</sub> to not strongly attenuate the signal amplitude through the complex voltage divider formed by the compensation network and the 50 Ω input of oscilloscope. On the other hand, low R<sub>c</sub> requires higher capacitance C<sub>c</sub> to keep the corner frequency constant, which limits the usable frequency of the compensation network due to self-resonance [10]. The equivalent series inductance (ESL) of small discrete ceramic capacitors is typically in the range of 500-600 pH and C<sub>c</sub> should not be selected larger than 47 pF to avoid shifting the self-resonance frequency below 1 GHz. For the design of this prototype, 0402 sized filter components are used and arranged in similar locations on opposite sides of the boards to keep the inductance in the measurement loop as small as possible. R<sub>c</sub> = 10 Ω has been found to be a good minimal value for the present compensation network. To determine C<sub>c</sub>, LTspice simulations of the transfer characteristic of the sensor including 50 Ω termination have been performed for c<sub>c</sub> values between 1 and 1.8, as shown in fig. 5. By using c<sub>c</sub> = 1, as proposed in [10], the phase-shift of the sensor is kept very small and gain will remain close to unity up to well beyond 100 MHz. Instead of crossing the +3dB limit, the compensated sensor will reach the -3dB limit first due to its self-resonance at f<sub>-3dB</sub> ≈ 700 MHz. Selection of larger c<sub>c</sub> can further increase the bandwidth for a fixed selection of R<sub>c</sub> by shifting the self-resonance frequency upwards. However, this introduces additional phase-shift and the gain will deviate from unity earlier. In the present case, choosing c<sub>c</sub> up to 1.4 seems still acceptable and will help to increase the bandwidth up to 1 GHz although at slightly reduced accuracy in the range of 100-400 MHz. Whether these configurations will be more precise than c<sub>c</sub> = 1 or

not depends on the individual use case and should be evaluated based on the final use scenario of the sensor. Further discussion as to why looking exclusively at the bandwidth as performance indicator might not lead to the best results can be found in [16].

Using the extracted parasitics and R<sub>c</sub> = 10 Ω, effective capacitance C<sub>c</sub> = 45 pF/c<sub>c</sub> is calculated. C<sub>c</sub> is comprised of the discrete capacitor value plus a small adder of few picofarads due to parasitic capacitance of the compensation network and coaxial connector. Three different capacitor values are used for VNA measurements and in good agreement with the LTSpice simulation, highest ±3dB bandwidth >1 GHz as well as best gain stability among these is achieved for C<sub>c</sub> = 33 pF (c<sub>c</sub> ≈ 1.35), as shown in fig. 6. It should be noted that the VNA measurements have only been performed using the SMA variant of the prototype and proper calibration of the measurement setup has been performed, but no de-embedding of the sensor circuit itself. However, the results are expected to be fairly accurate due to the small size of the structure and should also match well for the MMCX variants.

<figure><img src="image_6.png" /><figcaption>Fig. 5. Simulated transfer characteristic of the current sensor for different scaling factors c<sub>c</sub>, R<sub>c</sub> = 10 Ω and 50 Ω termination</figcaption></figure>

## V. SWITCHING EVALUATION OF THE PROTOTYPE

For evaluation of the performance and usability of the proposed flexible PCB shunt, a commercial half-bridge double-pulse test platform with low-inductance 650 V GaN Cascode FETs is used [14]. This platform
<figure><img src="image_7.png" /><figcaption>Fig. 6. Vector network analyzer forward transmission measurement results for different compensation networks</figcaption></figure>

uses state-of-the-art top-side cooled copper-clip packages (CCPAK) with a total commutation loop inductance of less than 5 nH and features an uncompensated on-board current shunt for comparison. To demonstrate the flexibility of the approach as well as the performance, the board is reworked and current sensors are fitted beneath the drain pins of the high-side (HS) FET and the source pins of the low-side (LS) FET. The low-side position is especially sensitive as the platform does not utilize a Kelvin Source connection. In this spot, the shunt impedance is introduced as a common impedance for both the gate and source loop and it will measure the sum of both currents $i_{GS,ext}$. It is generally not advisable to introduce additional common source inductance as this will increase the coupling between power and gate loop, leading to reduced performance so this is mainly intended for demonstrative purpose. Although there are no dedicated areas on the test platform for retroactively adding a current sensor, the flexible shunts fit well onto the board with only slight tilting of the GaN FETs, as demonstrated in fig. 7. Use of standard lead-free solder is possible but if no high-temperature testing capability is required, the attachment can be simplified and thermal stress on the components reduced by choosing lower temperature solder alloys. In the presented case, indium based solder (In97/Ag3) with a melting point of 148°C has been used.

The double-pulse test setup uses a 1 GHz Tektronix MSO58 oscilloscope with passive probes TPP0850 for $v_{DS}$ and TPP1000 for $v_{GS}$ measurement, both on the low-side. The on-board current sensor and the LS shunt are

<figure><img src="image_8.png" /><figcaption>Fig. 7. Picture and schematic of the double-pulse test platform after addition of two sensor prototypes into the commutation loop</figcaption></figure>

connected using double shielded coaxial cables. For HS measurement, a PMK Firefly optical isolated probe with 10:1 MMCX tip is utilized. The GAN039-650NTB cascade GaN FETs on the double-pulse board are operated using gate resistance of 10 Ω in series with a 100 Ω (at 100 MHz) ferrite bead. Switching waveforms at 400 V and 50 A are shown in fig. 8 for comparison between the onboard current shunt and the external shunts, for both the uncompensated current sensor and the optimized variant using $R_c = 10$ Ω and $C_c = 33$ pF. The on-board shunt is designed in a very similar array as the external shunt and consists of the same resistor array formed by ten 0603 sense resistors in parallel. However, its parasitics are not compensated as it was initially designed to match well with results from the commercial coaxial current shunt in [7]. Due to the larger thickness of the pre-pregs on standard PCB compared to a flexible PCB, higher measurement loop inductance $L_s$ is to be expected and in case $i_{meas}$ is extracted only by division of the ohmic resistance, phase and gain error $e$ according to equation 5 occurs.

$$e = 1 + j\omega \frac{L_s}{R_s} \qquad (5)$$

In the current waveform, this error will show as increased amplitude at high frequencies and more importantly, a leading phase when measuring high current slopes. Such a dynamic phase shift can not be compensated by the standard de-skew process and leads to an increase of calculated turn-on switching energy $E_{on}$ and decrease of turn-off energy $E_{off}$. The effect can well be observed in fig. 8 for the on-board shunt current calculation and to a lesser extent also for the uncompensated flex shunt. While $di/dt$ is nearly identical at 6 A/ns for turn on, $E_{on}$ of 316 µJ is reported due to the phase shift compared
to 307 µJ for the uncompensated shunt and 299 µJ with compensation; an increase of 5.7%. As $E_{off}$ is very small compared to $E_{on}$ for fast GaN FETs, slight errors during turn-off energy calculation have lesser impact. Nevertheless, the effect of $L_s$ in the measurement loop can be observed well in the peak current slope of -24 A/ns of the on-board sensor compared to -17 A/ns for the compensated shunt. The comparison shows that the external shunt is not only able to match the on-board solution but it in fact can outperform it due to lower $L_s$ and adapted RC compensation. In addition, it can enable extra benefits like capturing both the active switch current and the freewheeling current in the same setup, enabling switching loss and reverse-recovery charge $Q_{rr}$ evaluation in a single measurement. As demonstrated in fig. 8, there is very good agreement between the waveforms taken on the source and on the drain-side using the retroactively fitted sensors. It is worth mentioning that source-side measurements are generally to be preferred as a measurement on the drain side will not be accurate without use of expensive high bandwidth isolated probes.

<figure><img src="image_9.png" /><figcaption>Fig. 8. Switching waveform comparison of the uncompensated on-board shunt with both external current sensors at $V_{dc} = 400$ V and $i_L = 50$ A</figcaption></figure>

To extract the insertion inductance of the proposed prototypes on the CCPAK evaluation board, waveforms of a measurement with both external current sensors in place are compared to the same board after removal of the sensors, as depicted in fig. 9. Intricate differences between both variants can be observed and it is clear that especially the external low-side sensor significantly impacts the switching behavior due to the common source impedance. However, upon closer inspection, the

difference in switching behavior can not be attributed to the insertion inductance of the current sensor but is mainly due to the dominating ohmic voltage drop at the sense resistors $R_s = 47$ mΩ, which reduces the effective gate drive voltage at high switching currents. Typical

<figure><img src="image_10.png" /><figcaption>Fig. 9. Switching waveforms of the test platform and comparison after addition of the two external sensors</figcaption></figure>

ways to extract the power loop inductance are to analyze the oscillation frequency after turn-off or by estimation of the dip on $v_{DS}$ during the $di/dt$ phase at turn-on. However, the ringing shown in fig. 9 is very small and not sufficient for analysis. To provoke higher ringing, the turn-off current is increased beyond 60 A and the waveform of the on-board shunt current measurement is used to estimate the total loop inductance $\Sigma L_{loop}$ on the board. Figure 10 shows a zoomed plot of the ringing for both configurations and estimated damped oscillations of 174 MHz in the unmodified setup and 163 MHz, when two sensors are used. The total capacitance of the board is formed by the output capacitance of the GaN HEMT of $C_{oss}$(400V) $\approx 150$ pF plus an additional 40 pF for stray capacitance of PCB and test inductor. Calculating the total loop inductance via the LC circuit resonance relationship $\omega^2 = 1/LC$ results in $\Sigma L_{loop,orig} \approx 4.4$ nH in the unmodified case and $\Sigma L_{loop,2s} \approx 5.0$ nH with two sensors. The difference of ca. 600 pH can be attributed to both sensors plus the additional inductance introduced by the extra solder layers and tilting of the transistors, as can be seen in fig. 7. To confirm the estimation, the 50 A turn-on waveforms of fig. 9 are used to match the reduction in $v_{DS}$ with the induced voltage $v_{dip} = L di_s/dt$, using the sensor current as input. Figure 11
<figure><img src="image_11.png" /><figcaption>Fig. 10. Estimation of the high-current turn-off oscillation frequency for the test board, without and with two external sensors</figcaption></figure>

shows the results of a least mean squares fit of the induced voltage, indicating a difference of ca. 300 pH between both setups. This is in very good agreement with the oscillation method since $v_{DS}$ is measured on the drain and source pads of the PCB, not at directly at the package pins and so it includes the external current sensor on the low-side. Therefore, both methods confirm that the insertion inductance for a single external sensor is in the range of 300 pH, an excellent value considering that the board was not designed for such kind of modification.

<figure><img src="image_12.png" /><figcaption>Fig. 11. Least mean squares fit of induced voltage $v_{ind}$ with $v_{DS}$ measurement for 50 A turn-on</figcaption></figure>

## VI. CONCLUSION

With careful layout and compensation of the parasitic measurement loop inductance, very high-bandwidth current sensors can be designed on standard low-cost flexible PCBs. The thin layer-stack and bendability of the proposed sensor enables retrofitting into existing designs without the need for a dedicated footprint. Prototypes developed in this paper demonstrate bandwidth of more than 1 GHz combined with very low insertion of only 300 pH. The ability to add and remove the sensors at different locations can significantly enhance investigations, e.g. when paralleling devices or during troubleshooting. The potential of adding a current sensor into low inductance commutation loops even of fast GaN FET designs comprised exclusively of SMD devices can be an alternative to designing specific PCBs only for switching characterization and can help to save time and resources during benchmarking.

## REFERENCES

[1] K. Wang, X. Yang, H. Li, L. Wang and P. Jain, "A High-Bandwidth Integrated Current Measurement for Detecting Switching Current of Fast GaN Devices," in IEEE Transactions on Power Electronics, vol. 33, no. 7, pp. 6199-6210, July 2018.

[2] Tektronix, TCP0030A Datasheet, https://download.tek.com/datasheet/TCP0030A-Datasheet_51W1904211.pdf, accessed on 10th September 2024.

[3] Pearson, Current Monitor Model 6585 Datasheet, https://www.pearsonelectronics.com/pdf/6585.pdf, accessed on 10th September 2024.

[4] PEM UK Ltd, CWTMini50HF Datasheet, https://www.pemuk.com/userfiles/cwtmini50hf/cwtmini50hfDs_feb_2020.pdf, accessed on 10th September 2024.

[5] J. Wang et al., "Infinity Sensor: Temperature Sensing in GaN Power Devices using Peak di/dt," 2018 IEEE Energy Conversion Congress and Exposition (ECCE), Portland, OR, USA, 2018, pp. 884-890.

[6] H. C. P. Dymond, Y. Wang, S. Jahdi and B. H. Stark, "Probing Techniques for GaN Power Electronics: How to Obtain 400+ MHz Voltage and Current Measurement Bandwidths without Compromising PCB Layout," PCIM Europe 2022; International Exhibition and Conference for Power Electronics, Intelligent Motion, Renewable Energy and Energy Management, Nuremberg, Germany, 2022, pp. 1-10.

[7] T&M research, Current Viewing Resistors, SERIES SDN-414, https://www.tandmresearch.com/, accessed on 10th September 2024.
[8] L. Shillaber, Y. Jiang, L. Ran and T. Long, "Ultrafast Current Shunt (UFCS): A Gigahertz Bandwidth Ultra-Low-Inductance Current Sensor," in IEEE Transactions on Power Electronics, vol. 37, no. 12, pp. 15493-15504, Dec. 2022

[9] W. Zhang, Z. Zhang, F. Wang, E. V. Brush and N. Forcier, "High-Bandwidth Low-Inductance Current Shunt for Wide-Bandgap Devices Dynamic Characterization," in IEEE Transactions on Power Electronics, vol. 36, no. 4, pp. 4522-4531, April 2021

[10] M. Meissner, J. Schmitz, F. Weiss and S. Bernet, "Current measurement of GaN power devices using a frequency compensated SMD shunt," PCIM Europe 2019; International Exhibition and Conference for Power Electronics, Intelligent Motion, Renewable Energy and Energy Management, Nuremberg, Germany, 2019, pp. 1-8.

[11] H. Lutzen, J. Müller, V. Polezhaev, T. Huesgen and N. Kaminski, "Reducing the Impact of Skin Effect Induced Measurement Errors in M-Shunts by Deliberate Field Coupling," 2022 24th European Conference on Power Electronics and Applications (EPE'22 ECCE Europe), Hanover, Germany, 2022, pp. P.1-P.11.

[12] W. Zhang, Z. Zhang and F. Wang, "Review and Bandwidth Measurement of Coaxial Shunt Resistors for Wide-Bandgap Devices Dynamic Characterization," 2019 IEEE Energy Conversion Congress and Exposition (ECCE), Baltimore, MD, USA, 2019, pp. 3259-3264

[13] J. Böcker, S. Schoos and S. Dieckerhoff, "Experimental Comparison and 3D FEM Based Optimization of Current Measurement Methods for GaN Switching Characterization," 2018 20th European Conference on Power Electronics and Applications (EPE'18 ECCE Europe), Riga, Latvia, 2018, pp. P.1-P.9.

[14] Nexperia, NX-DP-GAN039-TSC double pulse evaluation board with top-side cooled CCPAK GaN FETs, user manual, https://assets.nexperia.com/documents/user-manual/UM90028.pdf, Rev 1.0.

[15] P. Clayton, "Inductance Loop and Partial", John-Wiley & Sons, Inc., ISBN: 9780470461884

[16] H. Lutzen, P. Ziegler, J. Roth-Stielow and N. Kaminski, "Why the 3 dB bandwidth of a current sensor does not provide an exact statement about its measurement capability," CIPS 2024; 13th International Conference on Integrated Power Electronics Systems, Düsseldorf, Germany, 2024, pp. 182-189.