Magnetic field measurement for contactless current sensing in power electronic converters

Tien Anh Nguyen, Pierre-Yves Joubert, Stéphane Lefebvre, Laurent Dupont, Denis Labrousse, Gérard Chaplier, Mickael Petit, François Costa

► To cite this version:

Tien Anh Nguyen, Pierre-Yves Joubert, Stéphane Lefebvre, Laurent Dupont, Denis Labrousse, et al. Magnetic field measurement for contactless current sensing in power electronic converters. The 26th European Conference on Power Electronics and Applications, GDR SEEDS France & EPE Association, Mar 2025, Paris, France. 10.34746/epe2025-0210. hal-05100704

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# Magnetic field measurement for contactless current sensing in power electronic converters

Tien Anh NGUYEN¹,*, Pierre-Yves JOUBERT², Stéphane LEFEBVRE¹, Laurent DUPONT³
Denis LABROUSSE¹, Gérard CHAPLIER⁴, Mickael PETIT¹, François COSTA⁵

¹SATIE, CNRS UMR 8029, CNAM, Paris, France.

²Centre de Nanosciences et de Nanotechnologies, CNRS, Université Paris-Sud, Palaiseau, France.

³SATIE, CNRS UMR 8029, Université Gustave Eiffel, Versailles, France.

⁴SATIE, CNRS UMR 8029, ENS Paris-Saclay, Gif-sur-Yvette, France.

⁵SATIE, CNRS UMR 8029, Université Paris Est Créteil, France.

*Corresponding author e-mail: tien-anh.nguyen@lecnam.net

## Keywords

«Magnetic field measurement», «short-circuit», «contactless current sensor».

## Abstract

Current sensing plays an important role in power electronics for both protection and control of power electronic converters. This paper proposes an approach for detecting short-circuit (SC) events through the measurement of the magnetic field radiated by power converters. Two magnetic field sensor technologies (AMR and induction coils) were evaluated. The magnetic field mapping over the PCB is carried out to estimate favorable positions for detecting SC events. The obtained results show that the induction coil is a promising candidate, not only for detecting a SC event but also for performing contactless current measurement in power converters. This work plans to develop the integration of contactless current sensors for power converters, or for intelligent drivers integrating protection against SC event (or over-currents), especially dedicated to wide band gap devices such as GaN and SiC materials which could have lower SC robustness compared to silicon.

## Introduction

Improving the efficiency and reliability of power converters is one of the important goals in power electronics industry. Integrating current sensing is considered to assist in the protection and current measurement of high-density integrated power converters [1-2]. Current estimation is essential for control, diagnosis and protection of converters [3-4]. The selected current sensing techniques must be compact to be easily integrated, present a

high bandwidth to be able to protect wide band gap transistors (SiC and GaN) switching quickly [5-6] and less invasive.

Many current sensing technologies are currently available for power electronics applications [7]. Measurement by coaxial shunt [8] has a high bandwidth (DC up to a few hundred MHz) but presents disadvantages of not ensuring galvanic isolation. The method is also invasive and responsible for additional losses at high current and adds parasitic inductance [9].

Current transformers are widely used for AC current measurement with their bandwidth up to several hundred MHz. They provide galvanic isolation and consume low energy. However, hysteresis and saturation of the magnetic core can degrade measurement accuracy [10]. The magnetic core also limits the operating temperature of the sensor [11].

An alternative solution can be considered by using Rogowski sensors. Without a magnetic core, the Rogowski sensor is minimally invasive. The PCB circuit must however be designed to accommodate it, it exhibits low losses, and excellent linearity. To be integrated, the sensor must be specifically designed [4] [12]. As shown in [12], authors present an application up to 10 MHz for which a Rogowski sensor was buried in a four-layers PCB with a particular track configuration for the generation of the penetrating flux into the Rogowski coil.

Current measurement techniques based on direct magnetic field sensing therefore seem to give more freedom for the design of integrated and non-invasive sensors. Different techniques have also been applied for a long time in power electronics such as: Hall effect, magnetoresistance and induction coil.
Hall effect sensor has a moderate bandwidth (DC to less than 1 MHz) and is sensitive to temperature. The sensitivity of a Hall effect sensor to magnetic field is low and may require a magnetic circuit to concentrate the flux to be measured [13-14]. Hall effect sensors are popular, they are not however very suitable for integration. Different principles of magnetoresistance have also been studied in the context of integration: anisotropic magnetoresistance (AMR) and giant magnetoresistance (GMR). These two technologies present good sensitivity. The sensitivity (µV/V/T) of GMR is much higher than that of AMR, the maximum variation of AMR resistance is around 3 to 4%, while that of GMR reaches 50%. The GMR sensor reacts to the magnetic field intensity and not to its direction. The response of a GMR sensor is therefore unipolar [15]. In contrast, the response of AMR sensor is bipolar [15]. Finally, AMR sensors exhibit more linear response with less hysteresis than GMR sensors [15].

In [16], the authors present an application of GMR sensors to measure different types of currents (switch, DC link capacitor and AC node currents) in a GaN SYSTEMS evaluation board. The simulation of magnetic field above the several basic structures of PCB is to determine the favorable positions on PCB for measuring the currents "in situ", without added current track. The experimental results show that the bandwidth of the sensor system ranges up to 600 kHz. An active temperature compensation of the GMR sensor is necessary in high temperature applications [2].

The “hybrid” solutions for integrating current sensing in power converters have also been investigated. The current sensor combines different sensing principles [17-20]. For example, the combination of AMR and Rogoski coil allows the system to achieve an extended bandwidth (from DC up to 10 MHz) [17].

There are also several examples of implementation of induction coil sensor for contactless current sensing. Printed circuit board (PCB) air coils embedded in power PCB are presented in [21-22]. In [21], it requires a large placement of embedded air coil inside the busbars of power converter to sufficiently pickup the magnetic field generated by the busbar current, which can degrade the stray inductance of power loop. Without electronic conditioning associated with induction coil, the authors rebuilt the measured current offline by using a postprocessing method based on the mutual inductance between the busbar structure and the air coil [21]. In [22], a smaller air coil has been embedded in a PCB near a single interconnect track of high current density designed between

the power device and the DC busbar. PCB coil senses the rate of rising of the current through the power device for overcurrent protection function and implementation of peak current mode control in a buck converter [22].

The performances of different current measurement techniques are still controversial in the context of integration and need to be precisely evaluated and compared. The measurement solution selected may need to be based upon compatibility with the available technology of integration.

In this paper, we focus on current sensing through the measurement of magnetic field using two sensor technologies: AMR sensor and induction coil. The research aim is to detect short circuit (SC) events in power converters and to perform contactless measurement of current in a highly integrated converter. The AMR current probe was designed based on a commercial AMR sensor (Honeywell). We will detail the signal conditioning of sensor system and evaluate the responses of these two measurement techniques. The simplified PCBs are designed to test the performance of both types of sensors. The magnetic field mapping is carried out above PCB, which allows us not only to estimate the favorable positions for current measurement or detection of SC, but also to know the effect of PCB design on the distribution of the magnetic field.

## Experimental protocol

The sensor probes (AMR and induction coil sensors) are moved on a three-axis robot and connected to an oscilloscope for data acquisition. A PC is used to control the overall system and store the data. The experimentation was set up to map 3D magnetic field distribution (Bx, By and Bz components of the magnetic field induced above the PCB).

## AMR sensor system

Figure 1 shows the AMR sensor system. We used three axes magnetic sensor HMC 1053 manufactured by Honeywell. The sensitivity of AMR sensor HMC 1053 is 10 µV/V/µT. When the power supply is set to 5 V, we obtain a sensitivity of 50 µV/µT. The datasheet bandwidth of the AMR sensor is 5 MHz. To acquire the signal, several amplifiers are used: a wide-band differential instrumentation amplifier (AD8421), and a wide-band low-noise operational amplifier (ADA4817). The total gain of the system is set to 100. The electronic conditioning of the AMR sensor requires a Set/Reset system which allows the orientation of the magnetic moments in the magnetoresistance to be reset under the effect of
the strong magnetic field measured or the variation of ambient temperature.

<figure><img src="image_2.png" /><figcaption>Fig. 1: Signal conditioning of the AMR sensor.</figcaption></figure>

## Induction coil sensor system

Figure 2 shows the induction coil sensor system. Two types of induction coil are made for this study (coil with ferromagnetic core and air core).

<figure><img src="image_3.png" /><figcaption>Fig. 2: Induction coil system.</figcaption></figure>

The impedances of the induction coils are characterized by HP 4194A impedance analyzer. Figure 3 illustrates an electrical circuit model equivalent to the induction coil [23].

<figure><img src="image_4.png" /><figcaption>Fig. 3: Equivalent circuit of the induction coil.</figcaption></figure>

The impedance of the induction coil is therefore represented by the formula:

$$Z = \frac{R + j\frac{\omega}{\omega_0}\left(\frac{L}{C}\right)}{\left(1 + \frac{R}{R_0}\right) + j\frac{\omega}{\omega_0}\left(\frac{L+RR_0C}{R_0}\right)\left(\frac{1}{\sqrt{LC}}\right) - \frac{\omega}{\omega_0}} \quad (1)$$

Where R, L and C are the internal resistor, inductance and self capacitance of induction coil, respectively. The resistance R₀ in parallel with self capacitance represents the non-linear behavior of the internal resistance as well as the iron losses for the coil with ferrite core at high

frequencies. The resistance R₀ favors adjusting the quality factor of induction coil which is consistent with the experimental measurement.

In which $\omega_0 = \frac{1}{\sqrt{LC}}$, we note ($f_R$) the resonance frequency of the coil.

<figure><img src="image_5.png" /><figcaption>Fig. 4: Frequency characteristics of induction coils.</figcaption></figure>

Using this model, the parameters (R, L, C, R₀) of the two types of induction coil were estimated. For the coil with ferrite core, we obtained R = 0.3 Ω, L = 93.3 µH, C = 8.2 pF, corresponding to a resonance frequency, f_R = 5.8 MHz. And for the air coil, the parameters of induction coil are identified with R = 1.5 Ω, L = 29.5 µH, C = 3.75 pF, these correspond to the resonance frequency f_R = 15.2 MHz. The resistance R₀ of 100 kΩ and 50 kΩ are estimated for the air coil and the coil with ferrite core, respectively.

Figure 4 presents the diagrams of the impedances experimentally measured and modeled by using the equivalent model of the induction coil, characterized from 100 Hz to 40 MHz. To obtain a high cut-off frequency of the measurement system, we chose the air coil which has the highest resonance frequency in order to develop the signal conditioning system dedicated to magnetic field measurement. The electronic conditioning consists of two stages: an integrator circuit and an amplifier as shown in figure 2.

## Frequency characteristics of the two types of magnetic field sensors

Both types of sensors are characterized by their transfer function. A high frequency antenna is used to create the excitation magnetic field to be measured. The antenna is supplied by the internal oscillator of HP 4194A analyzer, the current passing through the antenna is measured by a current transformer Tektronix CT2 probe with bandwidth: 1.2 kHz-200 MHz. By putting the antenna closer to the investigated sensor along the preferred axis of the measured magnetic field component, the Bode diagrams are
experimentally characterized. The transfer functions are then corrected by taking into account the transfer function of the CT2 probe. Obtained results are illustrated in figure 5.

<figure><img src="image_6.png" /><figcaption>Fig. 5: Frequency characteristics of the studied sensor systems.</figcaption></figure>

The measured frequency responses show a 3dB bandwidth of 1 MHz for the AMR sensor system, and a higher bandwidth of 20 MHz for the induction coil system (Fig. 5).

## PCB design for detecting SC events

Two PCB prototypes were designed to analyze and study the possibility of detecting SC events. A single-sided PCB with a large current loop was used to validate the principle of the method (Fig. 6.a). The two gate resistors $R_{ext\_ON}$ and $R_{ext\_OFF}$ allow adjusting the switching speeds of the transistor (Fig. 7.a). Another more optimized double-sided PCB prototype was produced to reduce parasitic inductances (Fig. 6.b). The structure of the double-sided PCB is like that of half bridge inverter with two SiC MOSFETs (High side and Low side) (Fig. 7.b).

<figure><img src="image_7.png" /><figcaption>Fig. 6: Single-sided (a) and double-sided (b) PCB prototypes.</figcaption></figure>

<figure><img src="image_8.png" /><figcaption>Fig. 7: Schematic of single-sided PCB (a) and double-sided PCB (b).</figcaption></figure>

<figure><img src="image_9.png" /><figcaption>Fig. 8: Short circuit waveforms tested on single-sided PCB (a) and double-sided PCB (b).</figcaption></figure>

SC cycle is performed by turning on only SiC MOSFET of single sided PCB or simultaneously turning on two SiC MOSFET of double sided PCB, for on-state duration of 2 µs. SC tests were performed for 1.2 kV SiC MOSFET under low DC bus voltage of 30 V to ensure the repeatability of SC stress and to avoid important damage on the transistors. The SC current is measured in reference using a commercial Rogowski coil (PEM / CWT MiniHF06 of 30 MHz bandwidth). Figure 8 shows typical SC waveforms on the two PCB prototypes.
By decreasing the value of R<sub>ext_ON</sub> or R<sub>ext_OFF</sub>, we can clearly observe the effect of the stray inductance of the power circuit on SC waveform for both tested PCBs. In the single-sided PCB with R<sub>ext_ON</sub> = 22 Ω, a voltage drop V<sub>SC</sub> of more than 10 V is observed in turn-on state. In the double-sided PCB, even with low R<sub>ext_ON</sub> = 10 Ω, we observe a smaller voltage drop V<sub>SC</sub> of about 2 V at turn on. This signifies that the stray inductance of power loop in the double-sided PCB is significantly reduced.

## AMR sensor

In this section, we analyze the transient responses of the AMR as well as the magnetic field mapping above the PCBs during SC event.

## Analysis of pulse responses of the AMR sensor

Firstly, by placing the AMR sensor at position 1 on the single-sided PCB at a height of 4 mm (Fig. 6.a), we measure the pulse responses of the AMR sensor during SC cycle with different values of [R<sub>ext_ON</sub>, R<sub>ext_OFF</sub>]. For example, figure 9.a presents a typical pulse response of the AMR sensor (component B<sub>X</sub> of the magnetic field) with [R<sub>ext_ON</sub>, R<sub>ext_OFF</sub>] = [47 Ω, 220 Ω].

<figure><img src="image_10.png" /></figure>

<figure><img src="image_11.png" /><figcaption>Fig. 9: Pulse responses of the AMR sensor during SC stress at position 1 on the single-sided PCB at a height h = 4mm, with [R<sub>ext_ON</sub>, R<sub>ext_OFF</sub>] = [47Ω, 220Ω] (a), and with [R<sub>ext_ON</sub>, R<sub>ext_OFF</sub>] = [22Ω, 220Ω], [47Ω, 470Ω] (b).</figcaption></figure>

To find the origin of the measured magnetic field peaks appearing at the moments of SC current

switching, we propose a model based on the amplitude of the SC current and its derivative.

$$B_X = \alpha_X \cdot \left( \frac{dI_{CC}}{dt} \right) + \beta_X \cdot I_{CC} \quad (2)$$

The coefficient β<sub>X</sub> can be determined over the interval of [1.7 µs – 2 µs] when the SC current is approximately constant (the plateau of SC current). We can then deduce the value of coefficient α<sub>X</sub> by:

$$\alpha_X = \frac{(B_X - \beta_X \cdot I_{CC})}{\left( \frac{dI_{CC}}{dt} \right)} \quad (3)$$

The numerical calculation shows [α<sub>X</sub>, β<sub>X</sub>] = [1.85×10<sup>-8</sup>, 0.0465] which represents the magnetic coupling along the X axis of the magnetic field between the AMR sensor and the PCB, at position 1 of height h = 4 mm on the single-sided PCB. To validate this model, different values of [R<sub>ext_ON</sub>, R<sub>ext_OFF</sub>] are also applied in order to obtain different dynamic slopes of SC current. By applying the model above in the case of [R<sub>ext_ON</sub>, R<sub>ext_OFF</sub>] = [22 Ω, 220 Ω] and [47 Ω, 470 Ω], we note that the estimation of model is consistent with the experimental measurements (Fig. 9.b).

Moreover, to verify the generalization of the model in this case study, we measure the pulse responses (component B<sub>X</sub> of the magnetic field) during SC stress on a straight line between the position 1 and position 3 on the single-sided PCB, with a scanning step of 2 mm (Fig. 6.a). Figure 10.a illustrates such distribution of pulse responses. By transforming the above model into the matrix form, it therefore becomes:

$$[B_X] = [\alpha_X] \cdot \frac{dI_{SC}}{dt} + [\beta_X] \cdot I_{SC} \quad (4)$$

We obtain a distribution of the coefficients α<sub>X</sub> and β<sub>X</sub> on this scanned line (Fig. 10.b). Each measurement point is represented by its own values [α<sub>X</sub>, β<sub>X</sub>] which presents the magnetic coupling between the AMR sensor and the PCB at the determined position. From this model, with the coefficients [α<sub>X</sub>, β<sub>X</sub>] identified, we also estimate the responses of the AMR sensors in the cases of the different values [R<sub>ext_ON</sub>, R<sub>ext_OFF</sub>] = [22 Ω, 220 Ω] and [47 Ω, 470 Ω]. The estimation results using the model and experimental measurements are presented in figure 11.a and 11.b. A good agreement between the model and the experimental measurement is thus observed on the scanned line.

Then, an estimation using this model is also performed on the double-sided PCB (Fig. 6.b). Unfortunately, the results estimated by model are less consistent with the experimental measurements. It therefore seems that the model
(2) is only validated in a few favorable configurations of PCB. It depends on the PCB design and the relative position between the sensor and the PCB, so a particular model will need to be required and studied for each prototype. The development of such a reliable model dedicated to power converters will be mandatory, once it is desired to use the magnetic field sensor such as AMR sensor technique for contactless current sensing.

<figure><img src="image_12.png" /></figure>

<figure><img src="image_13.png" /><figcaption>Fig. 10: Distribution of pulse responses on the straight line between positions 1 and 3, on single-sided PCB with $[R_{\text{ext\_ON}}, R_{\text{ext\_OFF}}] = [47 \, \Omega, 220 \, \Omega]$ (a), coupling coefficient distribution (b).</figcaption></figure>

<figure><img src="image_14.png" /></figure>

<figure><img src="image_15.png" /><figcaption>Fig. 11: Distribution of pulse responses on the straight line between positions 1 and 3, on single-sided PCB, with $[R_{\text{ext\_ON}}, R_{\text{ext\_OFF}}] = [22 \, \Omega, 220 \, \Omega]$ (a), $[47 \, \Omega, 470 \, \Omega]$ (b).</figcaption></figure>

## Magnetic field mapping measured by the AMR sensor

3D magnetic field mapping using AMR sensor are performed during SC event. The mapped areas are shown in figure 6. Each measurement requires one SC cycle of excitation to measure the magnetic field induced above this point.

<figure><img src="image_16.png" /><figcaption>Fig. 12: Magnetic field mapping on the single-sided PCB, h = 4 mm.</figcaption></figure>
<figure><img src="image_17.png" /><figcaption>Fig. 13: Magnetic field mapping on the double-sided PCB, h = 4 mm.</figcaption></figure>

The average magnetic field calculated during the time interval [1.7 µs to 2 µs] of each SC cycle will then be illustrated. Figure 12 and 13 present the magnetic field mapping (components Bx, By, and Bz) at a height of h = 4 mm above the single-sided and double-sided PCB, respectively. This allows us to determine suitable positions for the detection of SC stresses by magnetic field measurements. The magnetic field mapping performed above the double-sided PCB shows that the magnetic field induced around several positions such as through-hole of transistor pin-out or vias, cannot be fully eliminated by PCB design (Fig. 13).

## Induction coil sensor system

In this section, we present the development of the signal conditioning dedicated to increase the bandwidth of measurement for the induction coil sensor (air coil) as well as the experimental results obtained.

## Analysis of signal conditioning of induction coil sensor

The signal conditioning of the induction coil sensor consists of two main stages: the integrator stage and the amplifier circuit. The bandwidth of the system also depends on the optimization of the bandwidth of the integrator stage.

<figure><img src="image_18.png" /><figcaption>Fig. 14: Schematic of signal conditioning for the induction coil sensor system.</figcaption></figure>

To optimize the integrator stage, it is made up of three elementary circuits as shown in figure 14 [24]. The first elementary stage of the integrator is called a “self-integrator”, charging the induction coil with a resistance $R_C = 56 \ \Omega$. The purpose of the self-integrator is to ensure the integration function in the frequency range of more than 300 kHz [25]. The transfer function

model (F1) of this integrator stage is expressed by:

$$F_1 = \left( \frac{\frac{R_C R_0}{j\omega R_0 R_C C + R_C + R_0}}{\frac{R_C R_0}{j\omega R_0 R_C C + R_C + R_0} + j\omega L + R} \right) (j\omega NS) \quad (5)$$

Where N is the number of turns of the coil and S the active area of the coil.
The second elementary stage is a passive integrator (R&C); it is designed to allow the signal to be integrated in the frequency range from 100 kHz to 300 kHz. Its transfer function ($F_2$) is given by:

$$F_2 = \frac{R_4 + \frac{1}{j\omega C_2}}{R_3 + R_4 + \frac{1}{j\omega C_2}} \quad (6)$$

The third elementary stage is an active integrator. It is designed for the frequency range from 700 Hz to 100 kHz. The expression of $F_3$ is:

$$F_3 = \frac{A_d}{1 + j \frac{\omega}{\omega_0} + \frac{A_d}{k_1}} \quad (7)$$

Where $A_d$ and $\omega_0$ are respectively the open-loop gain and the cutoff frequency of the OP Amp used in the active integrator, with the expression:

$$k_1 = \frac{R_1 + R_2 + j\omega R_1 R_2 C_1}{R_2 + j\omega R_1 R_2 C_1}$$

The model of the total transfer function is thus formulated as follows:

$$F_{total} = K \cdot F_1 \cdot F_2 \cdot F_3 \quad (8)$$

The modeled transfer function presents a fair agreement with experimental estimation over the frequency range up to 20 MHz (Fig. 15).

<figure><img src="image_19.png" /><figcaption>Fig. 15: Transfer function of induction coil sensor system.</figcaption></figure>

## Responses of the induction coil sensor

Firstly, the induction coil sensor has been employed to detect SC events on the single-sided PCB. Three dynamic behaviors of SC current were performed with three different values of the gate turn on resistance $R_{ext\_ON} = 10 \, \Omega$, 22 $\, \Omega$, and 47 $\, \Omega$, while the gate turn off resistance $R_{ext\_OFF}$ is always maintained at 220 $\, \Omega$. Figure 16.a illustrates three SC current waveforms to be detected.

<figure><img src="image_20.png" /><figcaption>Fig. 16: SC waveforms in the single-sided PCB (a), pulse responses measured by the induction coil at position 1 (b), position 2 (c) and position 3 (d) on the single-sided PCB during SC event.</figcaption></figure>
<figure><img src="image_21.png" /><figcaption>Fig. 17: Mapping of component B<sub>Z</sub> measured by the induction coil on the single-sided PCB (a) and on the double-sided PCB (b) in SC event.</figcaption></figure>

The induction coil sensor is finally moved to positions P1, P2 and P3 at a height of h = 4 mm above the single-sided PCB (Fig. 6.a) in order to measure the SC current. Figures 16.b, 16.c, and 16.d show the pulse responses of the sensor in position 1, position 2, and position 3, respectively. It is important to note that the induction coil sensor allows us to obtain a more directly usable signal than the AMR sensor (Fig. 9). A lower noise level is observed on the results measured by the induction coil system compared to those measured by the AMR sensor.

Then we map the component B<sub>Z</sub> of magnetic field on the PCBs. The B<sub>Z</sub> mappings measured by the induction coil sensor (Fig. 17) show similar distribution as observed with the AMR sensor (Fig. 12 and Fig. 13).

We note that the induction coil sensor presents more advantages compared to the AMR sensor. The induction coil system has higher bandwidth up to 20 MHz, with simple electronic conditioning, which is less sensitive to noise and has smaller magnetic coupling to the power PCB under test.

## Contactless current sensing for half bridge power converter

In this section, we investigate the approach of contactless current measurement using inductor sensor for EPC EPC9002C evaluation board with EPC 2001C GaN transistors. The multilayer PCB is not very easy to integrate the current sensor inside the PCB. Indeed, the distribution of the magnetic field above the PCB is very complex with different current sources such as currents passing through the transistors, DC bus currents, and decoupling capacitor currents (Fig. 18.b).

<figure><img src="image_22.png" /><figcaption>Fig. 18: EPC evaluation board (a) and schematic of power converter (b).</figcaption></figure>

<figure><img src="image_23.png" /><figcaption>Fig. 19: Waveforms of the Buck converter for V<sub>DC</sub> = 10 V, f = 100 kHz, δ = 20%, Load : R = 1.5 Ω, L = 100 µH.</figcaption></figure>

In the context of this study, the EPC evaluation board operates as a Buck converter (Fig. 19). A mini coil whose dimensions are: D<sub>int</sub> x D<sub>ext</sub> x E<sub>p</sub> x N<sub>tum</sub> = 0.52mm x 1.1mm x 1.1mm x 200 was implemented to improve the resolution of the induction coil sensor (Fig. 20).

<figure><img src="image_24.png" /><figcaption>Fig. 20: Mini air coil for current sensing above the EPC evaluation board.</figcaption></figure>

<figure><img src="image_25.png" /><figcaption>Fig. 21: Positions investigated on EPC evaluation board for contactless current sensing.</figcaption></figure>

At the positions A and B above the high side GaN transistor at a height of h = 1 mm (Fig. 21), the effects due to harmonics of I<sub>DC</sub> current is significant on the sensor response (Fig. 22).
However, at the position C of height h = 1 mm above the decoupling capacitor (Fig. 21), the sensor response presents more clearly the shape of the output current ripple corresponding to each complementary switching (δ) and (1-δ) of transistors high side and low side (Fig. 23).

<figure><img src="image_26.png" /><figcaption>Fig. 22: Responses of the induction coil for positions A (a) and B (b) (high-side transistor).</figcaption></figure>

<figure><img src="image_27.png" /><figcaption>Fig. 23: Responses of induction coil for the position C (over decoupling capacitor) with different duty cycle, δ = 20% (a), 50% (b).</figcaption></figure>

## Conclusions and perspectives

This article investigates contactless current measurement techniques using two types of magnetic field sensors (AMR and induction coil) to detect SC stress and indirect current measurements on PCB tracks. The first results obtained show that induction coils present good candidates, not only for detecting SC stresses, but also for contactless current sensing in an integrated power converter. Induction coil sensor allows to achieve the required frequency bandwidth of 20 MHz with simple signal conditioning. Magnetic field mapping performed above PCB has allowed to determine the favorable positions for detecting SC event and/or current measurement in power converter.

This study opens the way to the development of intelligent drivers integrating protection against SC (or overcurrents) specially dedicated to GaN and SiC devices, as well as integration of contactless current sensing into high-density integrated power converters.

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