# Wideband 1-bit Reconfigurable Circularly Polarized Transmitarrays in Ka-Band

Alessandro De Oliveira Cabral Junior, André Barka, Hamza Kaouach, Francesco Positano

► To cite this version:

Alessandro De Oliveira Cabral Junior, André Barka, Hamza Kaouach, Francesco Positano. Wideband 1-bit Reconfigurable Circularly Polarized Transmitarrays in Ka-Band. IEEE Access, 2025, 13, pp.170670-170680. 10.1109/ACCESS.2025.3615780. hal-05308660

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Submitted on 10 Oct 2025

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<figure><img src="image_1.png" /></figure>

Distributed under a Creative Commons Attribution 4.0 International License
Received 9 September 2025, accepted 21 September 2025, date of publication 29 September 2025, date of current version 3 October 2025.

Digital Object Identifier 10.1109/ACCESS.2025.3615780

RESEARCH ARTICLE

# Wideband 1-bit Reconfigurable Circularly Polarized Transmitarrays in Ka-Band

ALESSANDRO DE OLIVEIRA CABRAL JUNIOR<sup>1, ∗</sup>, ANDRÉ BARKA<sup>1, †</sup>, HAMZA KAOUACH<sup>1, ‡</sup>, AND FRANCESCO POSITANO<sup>1, §</sup>

Corresponding author: Alessandro de Oliveira Cabral Junior (alessandro.de_oliveira_cabral_junior@onera.fr)

This work was supported in part by the Grand équipement national de calcul intensif (GENCI) for the full wave simulation using high performance computing (HPC) resources under Grant A0161007558 and Grant AD011013741R1.

**ABSTRACT** This study details the design of an active Transmitarray (TA) operating in the Ka-band made of a reconfigurable linear-to-circular-polarization (LP-CP) unit-cell. The unit-cells incorporate a true-time-delay (TTD) polarization conversion mechanism and a stacked patch configuration to achieve wide transmission and axial ratio bandwidths. Each cell integrates a pair of PIN diodes for electronically controllable 180° phase shifts. This is the first demonstration of a wide-band reconfigurable and circularly polarized unit-cell in millimeter wave bands, without the use of an external polarizer. A 20 × 20 cells Ka-band circularly polarized TA was designed and fabricated. A random symmetric distribution of rotations applied to the inner patch antenna enhances CP purity and increases the scan range. Measurement results demonstrate a broadside gain of 20.55 dB, with both –3 dB gain bandwidth and 3 dB axial ratio bandwidth exceeding 22%, which surpasses all previous millimeter-wave reconfigurable design. In the case of the 20 × 20 cell array manufactured the scan range extends to ±60° in both E and H planes, with a maximum scan loss of 4.2 dB and axial ratio values under 2 dB for elevation angles up to ±50°. Additional full wave simulations of a larger 60 × 60 cell array demonstrate a better maximum scan loss of 2.4 dB for elevation angles reaching ±50°, 3.8 dB for an elevation angle of ±60° and 6 dB for an elevation angle of ±70° with axial ratio values consistently under 3 dB.

**INDEX TERMS** Transmitarray, phased array, unit-cell, stacked patch, polarization conversion, circular polarization, axial ratio, wideband, Ka-band.

I. INTRODUCTION

The continuous evolution of telecommunications demands advanced antenna technologies capable of supporting high-speed applications in Ka and higher frequency bands. These bands are critical enablers for next-generation services such as 8K video, autonomous mobility, and smart infrastructure. Low Earth Orbit (LEO) satellite constellations, in particular, promise to deliver global, high-capacity connectivity. However, this advancement brings challenges like increased propagation losses, requiring high-gain, beam-steerable antennas [1].

The associate editor coordinating the review of this manuscript and approving it for publication was Woorham Bae<sup>¶</sup>.

Phased arrays have traditionally addressed these challenges through dynamic beamforming, yet their complexity and cost, especially in millimeter-wave (mmWave) bands, remain major barriers. The requirement for individual RF chains and beamforming networks (BFNs) per element significantly complicates their implementation at high frequencies.

An attractive alternative is found in space-fed solutions like reflectarrays and transmitarrays (TAs), which retain the beam-steering benefits of phased arrays without the associated RF front-end complexity. Among these, TAs have garnered increasing interest because of their compactness and efficiency. By adjusting the phase of incident waves through their unit-cells, TAs allow dynamic beam shaping, which is essential, for instance, in LEO satellite tracking [2],

<footnote n="1"><sup>1</sup>DEMR, ONERA, Université de Toulouse, 3100 Toulouse, France</footnote>

<footnote n="2"><sup>2</sup>LAPLACE Laboratory, INP-ENSEEIHT, Université de Toulouse, 31000 Toulouse, France</footnote>
facilitating seamless handovers between satellites and ensuring a smooth user experience. However, many current designs are either passive [3], limited in scanning capability [4], or tailored for linear polarization [5], [6], restricting their suitability for satellite communications where circular polarization (CP) is critical to avoid polarization mismatch losses.

Although reconfigurable TA designs have emerged utilizing PIN diodes [7], [8], [9], [10], [11], varactors [12], [13], [14], [15], MEMS [16], or liquid crystals [17], most of these references target lower frequency bands (e.g., C, X, Ku) and struggle to scale into the Ka band due to insertion losses, integration complexity, or limited phase-shift capabilities. Although varactors enable continuous phase shifting, their use above 20 GHz is challenging due to the scarcity of commercial components with high capacitance ratios and low parasitics. In [15], a reconfigurable TA operating at 28 GHz is proposed. To meet the capacitance requirements, two varactors in parallel and a pair of PIN diodes are employed to achieve a full 360° phase range. The prototype attains 26.7% aperture efficiency, higher than typical 2-bit designs, but at the cost of increased complexity, requiring multiple bias lines per unit-cell, a complex control circuit, and precise voltage tuning for each varactor. The radiation patterns are poorly shaped, with side-lobe levels around -8 dB.

PIN diodes are widely used in Ka-band designs due to their simplicity and easy biasing. However, most implementations target linear polarization (LP), as in [10], which employs a 2-bit magnetoelectric dipole design for wideband operation and high aperture efficiency. Four phase states are achieved using five PIN diodes and dedicated delay lines, resulting in a complex unit-cell stack-up with multiple metal layers, two bias lines per element, and multiple decoupling layers. The design achieves 37% aperture efficiency at 15 GHz and a 32% 3 dB gain bandwidth. Its main drawbacks are the fixed LP, requiring external polarizers for CP, and the use of five diodes, which significantly increase cost and power consumption in large arrays.

Finally, very few approaches address both reconfigurability and CP simultaneously, particularly in the Ka-band and above. Existing methods often require external polarizers [18] or sequential rotation schemes [19], [20]. These limitations degrade TA performance due to polarizer losses and the need for precise assembly, which is difficult at high frequencies. Additionally, using sequentially rotated LP unit-cells introduces 3 dB losses [19], further reducing aperture efficiency.

In this work, we propose the first Ka-band reconfigurable TA unit-cell that enables both beamforming and circular polarization through a single aperture, without requiring external polarization conversion or complex stacking. The proposed design integrates within the unit-cell a linear-to-circular polarization (LP-CP) mechanism directly into the unit-cell and a pair of PIN diodes, achieving 1-bit electronic phase control and CP emission with minimal insertion loss.

This eliminates the need for additional layers for polarization conversion, reduces fabrication complexity, and supports wide-angle beam steering, up to ±60° while maintaining standard scan losses.

The proposed unit-cell targets the 5G FR2 bands (24.5-29.5 GHz). It demonstrates insertion losses inferior to 1 dB, low axial ratio (less than 2 dB) across a wide bandwidth. A 20 × 20 element TA prototype shows a measured gain of 20.55 dB at 27.5 GHz, a peak aperture efficiency of 18%, and an overlapped 3 dB gain and AR bandwidth exceeding 22%, with a scan range of ±60° in both E and H planes. These results are comparable to more complex 2-bit designs while offering a significantly simpler and lower-cost solution.

To the best of our knowledge, this is the first demonstration of a Ka-band reconfigurable LP-CP TA that combines beamforming and CP generation in a single radiating aperture, without relying on sequential rotation or external polarizers.

The rest of this paper is structured as follows. Section II details the unit-cell architecture and performance. Section III presents the full TA prototype and its comparative evaluation. Finally, conclusions are drawn in Section IV.

## II. CIRCULARLY POLARIZED RECONFIGURABLE UNIT-CELL DESIGN

### A. PHASE SHIFTING AND LP-CP CONVERSION PRINCIPLES

The unit-cell is based on a receiver-transmitter configuration. It consists on a 5 metallic layer structure, with 3 substrate layers and 2 prepeg used to bond the whole stack-up, which can be seen in Fig. 1.

<figure><img src="image_2.png" /><figcaption>FIGURE 1. LP-CP Unit-Cell multilayer stack-up structure, see Table 1 for detailed geometric parameters.</figcaption></figure>

The exploded view of the proposed unit-cell is shown in figure 2. The receiver (Rx) antenna is a linearly polarized patch (M5 layer, Fig 2a), which includes two PIN diodes. The Rx patch is connected by a central via to the transmitter (Tx) circularly polarized square patch (M2 layer, Fig. 2c). Both antennas are separated by a ground plane (M3 layer), which allows polarization isolation from
each side. Fig. 3 shows detailed geometrical information of the UC metal layers.

The Tx patch situated in layer M2 is also responsible for the polarization conversion. This is achieved by the dual inner excitation of the Tx square patch antenna. As exemplified in [5] and [21], the feeding structures of the patch antennas are modeled as transmission lines (TLs) terminated by the patch antenna input impedance. A true-time delay (TTD) phase shift is introduced between each excited LP by physically extending one of the inner TL. By designing these TLs to match the load impedance and to impose a ±90° phase difference, high transmission coefficients are achieved with circular polarization formation. By placing the polarization conversion mechanism inside the unit-cell, the need for an external polarizer and the losses associated with sequential rotation schemes are suppressed.

The parasitic annular ring patch (M1 Layer) is included on the design to enable high transmission and axial ratio bandwidths, by improving the current distribution on the driven element and balancing the excitation of each LP field, see [21] for detailed information.

<figure><img src="image_3.png" /><figcaption>FIGURE 2. 3D exploded view of the proposed reconfigurable LP-CP unit-cell.</figcaption></figure>

To address the imperative requirements of low cost, high reliability and efficiency in mmWave antenna applications, a 1-bit solution is proposed in this study. This solution exploits a single biasing line, offering comparable beam-forming capabilities compared to other solutions which implement higher quantization levels and require additional active elements and biasing lines, which in turn increase insertion losses and design complexity.

<figure><img src="image_4.png" /><figcaption>FIGURE 3. (a) Rx linearly polarized patch antenna (M5 Layer), (b) Biasing layer with decoupling RF/DC capacitors, (c) Tx Circularly polarized patch antenna (M2 layer) and (d) proximity coupled stacked patch antenna for wideband performance (M1 Layer).</figcaption></figure>

In this work, a pair of PIN diodes is integrated into the LP Rx patch antenna, arranged in an anti-parallel configuration, as shown in Fig. 3a. By controlling a singular DC tension applied to each patch, a current reversion mechanism is achieved [7], [8]. This configuration ensures that one diode is forward biased or on the “ON” state (low impedance state), while the other is reversely biased or on the “OFF” state (high impedance state), enabling an electronically controlled 180° phase shift, as illustrated by Fig. 4. Moreover, positioning the PIN diodes on the Rx layer ensures effective isolation from the Tx circularly polarized patch by means of the ground plane, thereby rendering their influence on the unit-cell axial ratio negligible.

<figure><img src="image_5.png" /><figcaption>FIGURE 4. E-Field (absolute value) distribution over the Rx patch antenna for both quantization states demonstrating the 180° phase shift, 27.5 GHz.</figcaption></figure>

The diode biasing system is integrated on the unit-cell simulation model to avoid undesired coupling between the biasing lines and the radiating parts. Therefore, as shown in Fig. 3b, the layer “M4” corresponds to the biasing layer, in which the bias lines are routed to each unit-cell.
The bias lines are routed orthogonally to the incoming LP field, reducing the direct coupling between the feed source field and the DC layer. Furthermore, a distributed capacitor is introduced to increase the RF/DC decoupling levels. The Rx patch antennas are connected to those capacitors by buried vias, which are placed on the electric wall plane of this patch antenna, avoiding undesired coupling. A microcontroller based control circuit inspired by [9] and [22] is used to independently set the biasing voltages of each unit-cell. The circuit is based on 8-bit shift registers (SN74HC164) and single-pole double-throw (SPDT) switches (ADG1434) that allow the selection of a biasing voltage of ±5V. The circuit includes a series resistance of 220Ω to set the forward biasing currents at 15 mA.

The biasing is completed by referencing the central pad of the Rx patch to DC ground through the central via and the Tx patch. The latter is connected to the ground plane through a quarter wavelength line and a buried via. The presence of the biasing components has negligible effects on the RF performances, as proven in the next subsection.

The novelty of our design lies in the development of a wideband, reconfigurable, circularly polarized unit-cell operating in the Ka band. The current state-of-the-art reconfigurable circularly polarized unit-cells in the Ka band are typically limited to either sequentially rotated linearly polarized patch antennas [19] or complex 2-bit designs incorporating external polarizers [18]. These approaches introduce additional losses, require higher quantization to compensate, and restrict both bandwidth and scan range. Our proposed unit-cell enables wide-angle scanning with a simplified 1-bit quantization, significantly reducing cost and biasing complexity of the active components.

## B. Ka-BAND ELEMENT PERFORMANCE

The proposed Ka-band unit-cell is designed aiming to cover the FR2 band from 24.5 to 29.5 GHz (K/Ka bands), the central frequency is set at 27.5 GHz. The periodicity ($L_{cell}$) is fixed at 5 mm, which corresponds to 0.5λ at 30 GHz. This value is chosen so that grating lobes are not created during scanning over the whole desired frequency band.

The substrate layers are shown in Fig. 1: there are two layers of Rogers RO3003 ($ε_r$ = 3, $tgδ$ = 0.001, $h_{sub}$ = 0.75 mm), one layer of Rogers RT5880 ($ε_r$ = 2.2, $tgδ$ = 0.0009, $h_{sup}$ = 0.76mm) and two layers of Rogers prepeg Speedwave 300P ($ε_r$ = 3, $tgδ$ = 0.002, $h_{preg}$ = 89μm).

The challenge when designing reconfigurable elements is to optimize the key geometrical parameters of the unit-cell in the presence of the active elements. Aiming to an easily scalable design towards higher mmWave bands, the selected diode is MADP-000907-14020 from MACOM, since it can operate up to 70 GHz according to its datasheet. The S-parameter file of the diode characterization provided by the manufacturer is used to extract the equivalent lumped model [23]. The values of the diode equivalent model around 27.5 GHz are $R_s$ = 6.1Ω, $L_s$ = 0.05nH, $R_p$ = 300kΩ

and $C_p$ = 30pF, for a 15 mA forward DC current and 5 V reverse biasing voltage. Under these biasing conditions, the “ON” diode presents about 0.5 dB insertion losses, while the “OFF” state provides about -8.15 dB of isolation.

The unit-cell is simulated in a periodic array environment with Floquet modes excitation using CST Microwave Studio [24]. The localized components are taken into account through a co-simulation (3D electromagnetic simulation and post-processing circuit simulation) scheme available on this software. The PIN diodes are modeled as lumped RLC circuits depending on their biasing state: in case of a forward bias (“ON” state) they behave as a series resistor ($R_s$) and inductor ($L_s$) circuit, whereas the reverse bias (“OFF” state) is considered as a parallel resistor ($R_p$) and capacitor circuit ($C_p$). The value of these parameters depends on the selected diode, the frequency band of operation and the DC biasing voltage value.

To allow a 1-bit quantization with good circular polarization purity without the need for a sequential rotation scheme, the unit-cell is optimized to achieve wide transmission and axial ratio bandwidths. The unit-cell is excited by a x-polarized field on the Rx side and it radiates a LHCP field on the Tx side. Furthermore, the unit-cell is optimized taking into account its performance under oblique incidence of 50° (E-Plane) in order to broaden its scanning range. The geometrical parameters of the optimized Ka-band unit-cell are available on Table 1.

<table><caption>TABLE 1. Ka-Band unit-cell key geometrical parameters.</caption><thead><tr><td>Parameter</td><td>Value</td><td>Description</td></tr></thead><tbody><tr><td>L<sub>Cell</sub></td><td>5 mm</td><td>Unit-cellCell periodicity</td></tr><tr><td>L<sub>Patch-el</sub></td><td>2.5 mm</td><td>Length of Tx patch</td></tr><tr><td>L<sub>Patch-sp</sub></td><td>2.8 mm</td><td>Length of Rx patch</td></tr><tr><td>W<sub>Patch-sp</sub></td><td>2.2 mm</td><td>Width of Rx patch</td></tr><tr><td>R<sub>hole</sub></td><td>1 mm</td><td>Radius of circular slot</td></tr><tr><td>L<sub>hor</sub></td><td>0.4 mm</td><td>Length of inclined line</td></tr><tr><td>d<sub>yvia</sub></td><td>0.6 mm</td><td>Length of Rx bias line</td></tr><tr><td>R<sub>ext</sub></td><td>1.35 mm</td><td>Outer radius of stacked patch</td></tr><tr><td>R<sub>int</sub></td><td>0.6 mm</td><td>Inner radius of stacked patch</td></tr><tr><td>W<sub>cap</sub></td><td>0.5 mm</td><td>Width of distributed capacitor</td></tr><tr><td>L<sub>cap</sub></td><td>2.25 mm</td><td>Length of distributed capacitor</td></tr><tr><td>h<sub>sub</sub>, h<sub>sup</sub></td><td>0.75 mm</td><td>Substrate thickness</td></tr></tbody></table>

The transmission parameters and phase response of each unit-cell state are shown on Fig. 5 for a normal incidence. As confirmed by the simulation results, both states achieve a constant 180° phase shift over the whole frequency band with a linear phase slope. Furthermore, these simulations show remarkable results, with a minimum insertion loss of 0.9 dB at the central frequency and a transmission magnitude superior to -3 dB over 22% BW. Finally, the decoupling between CPs is higher than 25 dB for the whole simulated band, meaning that an axial ratio smaller than 2 dB is achieved over a 22% BW. This showcases the polarization conversion
efficiency of the structure, with minimal losses due to the active components.

<figure><img src="image_6.png" /><figcaption>FIGURE 5. Ka-band unit-Cell transmission parameters for a normal incidence of floquet modes for both phase states.</figcaption></figure>

<figure><img src="image_7.png" /><figcaption>FIGURE 6. Reconfigurable Ka-band LP-CP Unit-Cell S-parameters under oblique incidence of Floquet modes. Solid lines represent the $S_{LHCP,x}$ values, while the dashed lines represent the $S_{RHCP,x}$ parameters.</figcaption></figure>

When considering oblique incidence angles, the results of Fig. 6 show stable transmission parameters up to 60° scanning range. At 50° the –3 dB transmission magnitude bandwidth is still 16%, with decoupling between co and cross-polarization transmission higher than 16 dB inside this band. At the extreme scanning angle of 60° the upper bound of the frequency range is greatly affected, but transmission above –3dB is still achieved up to 28 GHz, with a corresponding 14% transmission BW and isolation between CPs still at least equals to 16 dB.

Fig. 7 shows the unit-cell radiation pattern in the principal planes for both E and H planes at the central frequency. It shows that the optimization of the unit-cell under oblique incidence successfully improved the control of cross-polarization levels in the E-plane, achieving decoupling levels of at least 15 dB between ±50°. However, we observe that its radiation pattern is quite asymmetric, with better performance towards positive angles. Furthermore, in the

<figure><img src="image_8.png" /><figcaption>FIGURE 7. (a) Unit-cell radiation pattern in both principal planes and (b) its axial ratio over all pointing directions at 27.5 GHz.</figcaption></figure>

H-plane, the unit-cell performance at large angles is less effective, with cross-polarization discrimination exceeding 15 dB limited to ±40°. Fig. 7b shows the simulated axial ratio for each pointing direction, which confirms a narrower axial ratio below 3 dB in the H-plane compared to the E-plane. Nonetheless, these results confirm its significant scanning range in both planes with good CP fields.

The unit-cell presents superior performances when compared to previous realizations, since very good CP fields can be achieved with wideband transmission characteristics, even at highly oblique angles. This will allow the design of wide-angle and wideband TA at mmWave bands, with the possibility of scanning at least to ±50°.

### III. TRANSMIT ARRAY DESIGN, SIMULATION AND MEASUREMENT RESULTS

In this section the design of reconfigurable TAs using the developed LP-CP unit-cell is proposed. It is first simulated using our semi-analytical hybrid code [5] and then by full-wave solvers such as CST MWS and ONERA's HPC solver FACTOPO [25]. The use of multiple solvers aims to better predict the performance of the final antenna given that the unit-cell frequency response is closely related to the lumped component modeling of the PIN diodes.

In order to increase the CP purity over the whole scanning range, a modulated distribution of rotations of the
unit-cell over the TA aperture is then applied, allowing smaller axial ratio values and increasing the scanning range to more than 120° field-of-view in both principal planes. This final configuration was manufactured in a 20 × 20 cells array for experimental validation of the proposed design.

Finally, to estimate the performance of this unit-cell with respect to realistic gain values in LEO communication applications, we simulated a large TA panel containing 60 × 60 cells. This antenna aimed to achieve gain levels higher than 30 dB at broadside and allow scanning angles larger than 70° with good beam-formation and low axial ratio values. The simulations are carried out using HPC clusters and the FACTOPO solver from ONERA [25].

## A. 1-BIT WITH INTERNAL SEQUENTIAL ROTATION

It is well known in the literature that applying a sequential rotation scheme can improve the axial ratio values of CP antennas [19]. In this case, 4 rotations of the inner layer CP patch (layer M2) are distributed inside the aperture in order to mitigate the unbalance level between each LP component, and thus improve the axial ratio.

The proposed unit-cell directly radiates CP fields, unlike [19] where two sets of orthogonally polarized unit-cells are sequentially rotated to synthesize CP radiation. As a result, in our approach, all four rotational states contribute simultaneously to both beam formation and enhanced CP purity, therefore avoiding the 3 dB polarization conversion losses observed in [19].

For the proposed design, the distribution of rotational states is generated randomly, with symmetry constraints enforced along both the x- and y-axes. This imposes a similar scanning performances in terms of axial ratio in both principal planes. The generation of the distribution of rotations could be done with more degrees of freedom using optimization techniques such as a particle swarm optimization (PSO) [26] or a genetic algorithms, possibly leading to better scanning performances. The optimization itself is out of the scope of this paper.

Therefore, in this case, multiple random symmetric distributions were tested with the hybrid code and the best scanning performance obtained was used for the fabrication. Fig. 8a shows the distribution of rotations of the M2 layer, while Fig. 8b shows the applied 1-bit phase correction for a broadside scan. It is important to note that the phase corrections are calculated taking into account the fixed distribution of rotations on the inner layer. Therefore compensation for both spatial phase delay and phase rotation of each cell is required, Fig. 9 shows a schematic representation of the proposed reconfigurable TA. Equation 1 details the calculation of the active phase correction of the m-th cell ($\Psi_{Rx}^m$) as a function of the spatial phase delay ($k_0 R_m$), the inner rotation of the Tx patch ($\Psi_{Tx}^m$) and the beam-steering direction ($\vec{k}_{bf}$).

$$ \Psi_{Rx}^m = k_0 R_m - \Psi_{Tx}^m - \vec{r}_m \cdot \vec{k}_{bf} \quad (1) $$

<figure><img src="image_9.png" /><figcaption>FIGURE 8. (a) Random symmetric distribution of rotations on the inner CP patch layer (M2) and (b) 1-bit phase correction applied for a broadside scan considering the phase rotation of the inner layer.</figcaption></figure>

<figure><img src="image_10.png" /><figcaption>FIGURE 9. Proposed reconfigurable TA with double phase correction scheme, see equation 1.</figcaption></figure>

This 20 × 20 TA antenna is then fabricated and measured. The TA measurement setup is shown in Fig. 10, where it is possible to see the fabricated array and its control cables placed on the Rx side. The diode polarization control is done by a microcontroller based card, which independently bias each unit-cell. The power consumed by each bias
line is determined by the output voltage (5 V) and the current (15 mA), resulting in a total antenna consumption of approximately 30 W, including the losses introduced by the 220Ω limiting resistors in the control circuit. In this work, these operating conditions were deliberately adopted to validate the proof-of-concept unit cell, while acknowledging that more energy-efficient or optimized biasing strategies could be implemented in future designs.

<figure><img src="image_11.png" /><figcaption>**FIGURE 10.** Fabricated 1-bit reconfigurable sequentially rotated TA with its control cables attached.</figcaption></figure>

The TA control system is set to impose a broadside pointing ($\theta$ = 0°) at 27.5 GHz, with a focal ratio (F/D) of 1. The measured result are presented in Fig. 11a. It presents a comparison between the measured and simulated radiation patterns of the array. We observe a gain difference of 1.4 dB between the simulations and measurements, with a measured value of 20.55 dB on-axis, while the simulations expected 22.25 dB. However, the shape of the obtained pattern is very consistent with the simulation.

By normalizing each curve with respect to its maximum, Fig. 11b shows a significant agreement among all the curves. The main beam obtained is almost identical to the numerical results, and the measured relative level of cross-polarization is also consistent with the simulations. The SLL level is around 15 dB, and the decoupling between circular polarizations is about 23 dB, corresponding to an axial ratio of about 1.2 dB. We observe a slight beam misalignment, which is related to centering issues in the measurement chamber. This positioning error (less than 1%) causes the main beam shift and the asymmetry of the measured pattern.

Fig. 12 presents the measured and simulated gain and axial ratio bandwidths. We observe a very low level of cross-polarization over the entire measured band, consistent with full-wave simulations. Despite the trend of the curve differs, the measurement shows an axial ratio level consistently below 2 dB across the measured band. Fig. 12 shows that the relative variation in measured gain is very similar to the FACTOPO simulation, with a maximum loss of 3 dB at 30 GHz. The measured and simulated –3 dB gain bandwidth and axial ratio below 3 dB exceed 22%,

<figure><img src="image_12.png" /><figcaption>**FIGURE 11.** (a) Simulated and measured broadside radiation patterns in E-Plane at 27.5 GHz. (b) Normalized patterns with respect to their maximum gain value. Dashed lines correspond to the cross-polarized RHCP gain.</figcaption></figure>

<figure><img src="image_13.png" /><figcaption>**FIGURE 12.** Simulated and measured gain and axial ratio bandwidth of the 20 x 20 cells TA.</figcaption></figure>

demonstrating good agreement relative to normalized values. The peak measured gain is 22 dB at 25.5 GHz.

Following the measurements, retro-simulations were performed using our hybrid analytical code, incorporating the presence of the DC biasing lines in the unit-cell's radiation pattern. Although the biasing lines have a minor effect on unit-cell performance, the simulations show a gain reduction of 0.85 dB due to decreased transmission, as clearly illustrated in Fig. 13. Therefore, the measured gain deviation of 1.3 dB can be largely attributed to the presence of the biasing circuitry, while the remaining 0.45 dB
is likely due to fabrication tolerances and measurement uncertainties.

<figure><img src="image_14.png" /><figcaption>FIGURE 13. Comparison of the proposed 20 × 20 cells reconfigurable TA with and without the effects of the DC biasing lines.</figcaption></figure>

<figure><img src="image_15.png" /></figure>

<figure><img src="image_16.png" /><figcaption>FIGURE 14. Measured (solid lines) and simulated (dashed lines) LHCP (a) and RHCP (b) 20 × 20 cells Ka-band TA normalized gain patterns at 27.5 GHz in the E-plane.</figcaption></figure>

Figures 14 and 15 compare the measured and simulated results of the beam-scanning over the range of ±60° in both principal planes, the radiation patterns are normalized with respect to the axial gain. Good beam formation is achieved over all the desired angles, with the beams starting to get

deformed around 60° of elevation due to peaks in cross-polarization and the reduction of the unit-cell gain. The measured results are in great agreement with the simulation with respect to beam shape and direction.

<figure><img src="image_17.png" /></figure>

<figure><img src="image_18.png" /><figcaption>FIGURE 15. Measured (solid lines) and simulated (dashed lines) LHCP (a) and RHCP (b) 20 × 20 cells Ka-band TA normalized gain patterns at 27.5 GHz in the H-plane.</figcaption></figure>

The maximum scan loss in the E-plane is 4.5 dB, while in the H-plane a 5.5 dB loss is measured due to the unit-cell narrower beam in this plane. Fig. 16 and 17 show that the SLLs are measured around 15 dB for the broadside gain, while for the more oblique beams its value drops to 10 dB. The axial ratio is always smaller than 2 dB for scanning angles between ±50° in both planes. The simulation shows that scanning up to 60° is possible, but due to problems with the measurement setup, such as edge diffraction on the support structure, there are peaks in the cross-polarization levels that increase the measured axial ratio in the H-plane.

Excellent scanning performance is obtained, with a controlled axial ratio value, wide-angle scan with losses comparable to previous realizations and a peak aperture efficiency of roughly 19% at 25.5 GHz. Table 2 summarizes the key performance indicators of the proposed active TA compared with other available literature. As mentioned earlier, there are very few works on reconfigurable CP TAs
<figure><img src="image_19.png" /><figcaption>**FIGURE 16.** Simulated and measured Axial Ratio (AR) and SLL of the proposed 20 × 20 TA in E-plane for each steering configuration at 27.5 GHz.</figcaption></figure>

<figure><img src="image_20.png" /><figcaption>**FIGURE 17.** Simulated and measured Axial Ratio (AR) and SLL of the proposed 20 × 20 TA in H-plane for each steering configuration at 27.5 GHz.</figcaption></figure>

in the mmWave bands. Table 2 helps to show how our work presents similar scanning losses, with a peak aperture efficiency close to 2-bit designs [19], [20], a wider bandwidth and a smaller complexity due to a single pair of active elements per cell.

Therefore, this work presents an important contribution to the advancement of CP TA antennas for future mmWave applications with over the state of the art performances.

## B. FULL WAVE SIMULATIONS OF A 60 × 60 CELLS ARRAY

This section discusses the expected performance of a larger TA aperture size, with a focus on refining the gain and axial ratio levels for wide-angle scanning. Specifically, we investigate a 1-bit 60 × 60 cells TA using the proposed unit-cell design, maintaining the same focal ratio and primary source as in the previous section. The simulation was executed on an HPC cluster using 247 Genoa EPYC 9654 scalar nodes. The complete simulation, implemented using the FETI-2LM domain decomposition method (FACTOPO software [25]), incorporates the primary source (horn), the TA unit-cells, and the free space between the horn and the TA.

To ensure accurate 1-bit phase quantization combined with sequential rotation, we employ 10 distinct subdomain types, including 8 different unit-cell meshes (representing 2 phase states × 4 different inner rotations), as well as air-filled subdomains for the focal region and the primary source domain. In total, 12,289 subdomains are used in the complete simulation. Fig. 18 presents the beam steering simulation results in the E-plane for angles ranging from −70° to 70°. The broadside gain reaches 32 dB, with a well-formed beam observed across all steering directions, showing losses of 2.4 dB at 50°, 3.8 dB at −60°, and 6 dB at −70°.

<figure><img src="image_21.png" /><figcaption>**FIGURE 18.** Simulated radiation pattern of the proposed 60 × 60 cells 1-bit reconfigurable and sequentially rotated TA at 27.5 GHz.</figcaption></figure>

Moreover, the simulations indicate that even at −70°, the decoupling between circular polarizations remains greater than 15 dB, reinforcing the robust beam-steering capability demonstrated by the previously manufactured 20 × 20 cells TA. The gain diagrams over the half-sphere further confirm well-controlled beam formation across all cutting planes for both co-polarization and cross-polarization. Fig. 19 and 20 show LHCP and RCHP fields, respectively, over the upper half-sphere for a beam-steer at 50° in the E-plane. The values in these figures are normalized by the peak gain at the steering angle. It is possible to see that the SLL are better than 20 dB in all azimuth planes and that the cross-polarization is smaller than −18 dB.

<figure><img src="image_22.png" /><figcaption>**FIGURE 19.** LHCP gain radiation pattern of the 60 × 60 cells TA at 27.5 GHz, for a 50° scanning angle in the E-plane.</figcaption></figure>
<table><caption>TABLE 2. Comparison of reconfigurable TA performances.</caption><thead><tr><th>Ref.</th><th>f<sub>0</sub><br>(GHz)</th><th>N<sub>bits</sub></th><th>Pol.</th><th>Number of<br>Elements</th><th>Peak Gain<br>(dB)</th><th>Aperture<br>Efficiency (%)</th><th>3 dB Gain<br>BW (%)</th><th>3 dB AR<br>BW (%)</th><th>Overlapped<br>BW (%)</th><th>Scan<br>Range</th><th>Scan<br>Loss (dB)</th></tr></thead><tbody><tr><td>[7]</td><td>12.5</td><td>1-Bit<br>PIN</td><td>LP</td><td>12×12</td><td>19</td><td>20.5</td><td>32</td><td>N.A</td><td>N.A</td><td>±40°<br>(E/H)</td><td>2.7</td></tr><tr><td>[8]</td><td>17</td><td>1-Bit<br>PIN</td><td>LP</td><td>16×16</td><td>19.6</td><td>13.8</td><td>17.5</td><td>N.A</td><td>N.A</td><td>±60°</td><td>4.15</td></tr><tr><td>[15]</td><td>27.5</td><td>Continuous<br>Varactor</td><td>LP</td><td>12×12</td><td>19</td><td>26.37</td><td>9.45</td><td>N.A</td><td>N.A</td><td>±50°</td><td>3.9</td></tr><tr><td>[20]</td><td>26</td><td>2-Bit<br>PIN</td><td>CP</td><td>24×24</td><td>22.7</td><td>18.1</td><td>13.8</td><td>27.1</td><td>13.8</td><td>±60°<br>(E/H)</td><td>4.6</td></tr><tr><td>[19]</td><td>30</td><td>2-Bit<br>PIN</td><td>Dual<br>CP</td><td>24×24</td><td>25.5</td><td>19.2</td><td>10.4</td><td>13.33</td><td>10.4</td><td>±60°<br>(E/H)</td><td>4.6</td></tr><tr><td>[18]</td><td>30</td><td>2-Bit<br>PIN</td><td>Dual<br>CP</td><td>24×24</td><td>26.9</td><td>24.8</td><td>15</td><td>15</td><td>15</td><td>±40°<br>(E/H)</td><td>1.6</td></tr><tr><td>Our<br>Work</td><td>27.5</td><td>1-Bit<br>PIN</td><td>CP</td><td>20×20</td><td>22.0</td><td>19</td><td>22</td><td>22</td><td>22</td><td>±60°<br>(E/H)</td><td>5</td></tr></tbody></table>

<figure><img src="image_23.png" /><figcaption>FIGURE 20. RHCP gain radiation pattern of the 60 × 60 cells TA at 27.5 GHz, for a 50° scanning angle in the E-plane.</figcaption></figure>

## IV. CONCLUSION

This work presents a novel reconfigurable LP-CP unit-cell for Ka-band frequencies, integrating TTD conversion and a stacked patch design to achieve wide transmission and axial ratio bandwidths. The inclusion of PIN diodes enables electronically controlled 180° phase shifts. To the best of our knowledge, this is the first demonstration of a reconfigurable unit-cell achieving simultaneous wideband performance in both criteria at millimeter-wave frequencies, without requiring the use of sequential rotations schemes or external polarizers.

The concept is validated through simulations and measurements of a 20 × 20 reconfigurable TA enhanced with a random symmetric rotation of the inner CP patch. This rotation improves axial ratio and scan range without compromising gain. Measurements confirm a broadside gain of 20.55 dB, over 22% bandwidth in both gain and axial ratio, and a scan range of ±60° with minimal scan loss and axial ratio below 2 dB up to ±50° elevation. The difference between simulations and measurements is inferior to 0.45 dB, validating the numerical modeling of the antenna.

Further simulations on a larger 60 × 60 array demonstrate scalability and effective beam-steering up to 70° elevation,

maintaining axial ratio below 3 dB and sidelobe levels under 16 dB. These results establish the proposed design as a promising solution for next-generation circularly polarized mmWave TA antennas, outperforming current state-of-the-art alternatives.

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<figure><img src="image_24.png" /></figure>

ALESSANDRO DE OLIVEIRA CABRAL JUNIOR received the bachelor's degree in electrical engineering from the Federal University of Pernambuco (UFPE), the combined bachelor's and master's degree in electronics and microwave engineering from the École Nationale Supérieure d'Électrotechnique, d'Électronique, D'informatique, D'hydraulique et des Télécommunications (ENSEEIHT), Toulouse-INP, in 2021, and the Ph.D. degree from INP, University of Toulouse, on research into active transmit-array antennas in the Ka-band. In 2021 he joined ONERA, where his current research interests include reflectarrays, transmitarrays, periodic structures, SatCom, and radar antennas for small platforms. He received the CAPES/BRAFITEC Grant for a double degree in France.

<figure><img src="image_25.png" /></figure>

ANDRÉ BARKA received the M.Sc. and Ph.D. degrees in applied mathematics from the University of Bordeaux, France, in 1986 and 1990, respectively, and the H.D.R. (Habilitation à Diriger des Recherches) degree from the University of Toulouse, in 2008. In 1989, he joined ONERA, where he is currently in charge of developing advanced multi-domain and multi-method techniques combining integral equations, finite element methods, and asymptotic techniques. From September 2016 to September 2017, he was a Guest Researcher with the Telecommunications Institute, University of Lisbon, Portugal, through a research convention with ONERA, French DoD, and the University of Lisbon. He is also a Ph.D. Advisor of the GEET Doctoral School, Toulouse. He is a Senior Scientist with more than 20 years of research experience in computational electromagnetics, especially in domain decomposition methods in various application fields, such as electromagnetic compatibility, radar cross section, and antennas. He was the Work-Package Manager for advanced modeling activities in the FP6 IPAS European Project and the H2020 EPICEA European/Canadian Project. His current research activities include the design and realization of new antenna concepts based on metamaterials and advanced domain decomposition methods, including finite element tearing and interconnecting (FETI) and integral equation domain decomposition methods (IE-DDM). He has been a Board Member of Computer Physics Communications journal, since 2006; and a Specialist Editor of Computational Electromagnetic Journal.

<figure><img src="image_26.png" /></figure>

HAMZA KAOUACH received the M.S. degree in high frequency communications systems from the University of Paris-Est Marne-la-Vallée, France, in 2006, and the Ph.D. degree in signal processing and telecommunications from the University of Rennes 1, France, in 2009. His Ph.D. studies were partially supported by French Atomic Energy and Alternative Energy Commission CEA Grenoble, and French Government Space Agency CNES Toulouse, France. From 2006 to 2009, he was a Research Engineer with CEA-Léti, Grenoble, France. From 2010 to 2012, he was an Antenna and RF Engineer in industries at PSA Peugeot Citroën Group, Paris, France. From July 2012 to August 2017, he was an Assistant Professor with the Electronic and Communications Engineering Department, College of Engineering at Al-Lith (CEL), Umm Al-Qura University, Makkah, Saudi Arabia. From August 2015 to August 2017, he was an Associate Researcher with LATIS, National Engineering School of Sousse, Tunisia. Since September 2017, he has been an Associate Professor (MCF, HDR 2025) with the Institut National Polytechnique de Toulouse (INP/N7) and LAPLACE Laboratory, University of Toulouse. His current research interests include quasi-optic reconfigurable antennas, antenna arrays, optical control of reconfigurable antennas, transmit-arrays at millimeter-wave frequencies, frequency selective surfaces, measurement antennas, electromagnetic reverberation chamber (RVC and MSC), and metamaterial and numerical modeling (FDTD).

<figure><img src="image_27.png" /></figure>

FRANCESCO POSITANO received the M.Sc. degree in telecommunications engineering from the Politecnico di Bari, Bari, Italy, in 2021, and the Ph.D. degree from the Université Côte d'Azur, Sophia Antipolis, France, in 2024. He conducted his research with the Laboratory of Electronics, Antennas and Telecommunications (LEAT), Université Côte d'Azur. During the Ph.D. studies, he was a Visiting Researcher with Georgia Institute of Technology, Atlanta, USA, and the Université di Trento, Trento, Italy. Since 2025, he has been a Postdoctoral Researcher with ONERA-The French Aerospace Laboratory, Toulouse, France. His current research interests include miniature, low power and reconfigurable antennas, and antenna arrays at millimeter-wave.