# Design of an Integrated Magnetic Structure for Hybrid IPT Charger to Achieve Misalignment Tolerance and Constant Output Current

Armin Gheysari, Ali Yazdian Varjani, Amir Babaki, Thomas Ebel

► To cite this version:

Armin Gheysari, Ali Yazdian Varjani, Amir Babaki, Thomas Ebel. Design of an Integrated Magnetic Structure for Hybrid IPT Charger to Achieve Misalignment Tolerance and Constant Output Current. The 26th European Conference on Power Electronics and Applications, GDR SEEDS France & EPE Association, Mar 2025, Paris, France. 10.34746/epe2025-0032 . hal-05075762

HAL Id: hal-05075762

https://utc.hal.science/hal-05075762v1

Submitted on 20 May 2025

HAL is a multi-disciplinary open access
archive for the deposit and dissemination of sci-
entific research documents, whether they are pub-
lished or not. The documents may come from
teaching and research institutions in France or
abroad, or from public or private research centers.

L'archive ouverte pluridisciplinaire HAL, est
destinée au dépôt et à la diffusion de documents
scientifiques de niveau recherche, publiés ou non,
émanant des établissements d'enseignement et de
recherche français ou étrangers, des laboratoires
publics ou privés.

<figure><img src="image_1.png" /></figure>

Distributed under a Creative Commons Attribution 4.0 International License
# Design of an Integrated Magnetic Structure for Hybrid IPT Charger to Achieve Misalignment Tolerance and Constant Output Current

Armin Gheysari
ELECTRICAL AND COMPUTER
ENGINEERING DEPARTMENT, TARBIAT
MODARES UNIVERSITY Tehran, Iran
+98913-0173159.
a.gheysari@modares.ac.ir

Amir Babaki
INSTITUTE OF MECHANICAL AND
ELECTRICAL ENGINEERING,
UNIVERSITY OF SOUTHERN DENMARK
Denmark
+4565504857
amirbabaki@sdu.dk

## Keywords

«Wireless Power Transmission», «Hybrid», «Magnetic coupling», «Electric Vehicle (EV)», «Battery charger».

## Abstract

To address output current fluctuations caused by magnetic coupler mismatches in Inductive Power Transfer (IPT) chargers, a double-D quadrature (Double DDQ) coupler with LCC-S hybrid topology is proposed. This topology combines an LCC compensator (with an inductor and two capacitors) and series capacitors, along with added coupling between primary and secondary circuits. Integrating compensator inductance within the magnetic coupler enhances mismatch tolerance, ensures constant output current, and eliminates extra coils on the PCB, reducing size. This cost-effective approach is ideal for wireless charging of Electric Road Vehicles (ERVs), tolerating coupler mismatches from free movement. The magnetic coupler's performance in ensuring smooth output current under X- and Y-axis misalignments is analysed. A 1-kW prototype validates the design and demonstrates its effectiveness.

## Introduction

Inductive Power Transfer (IPT) technology has emerged as a promising battery charging approach for commercial applications ranging from a few watts to several kilowatts [1]. IPT systems can wirelessly transmit power through an air gap between the primary and secondary coils

Ali Yazdian Varjani
ELECTRICAL AND COMPUTER
ENGINEERING DEPARTMENT, TARBIAT
MODARES UNIVERSITY Tehran, Iran
+98912-1594654.
yazdian@modares.ac.ir

Thomas Ebel
INSTITUTE OF MECHANICAL AND
ELECTRICAL ENGINEERING,
UNIVERSITY OF SOUTHERN DENMARK
Denmark
+4565501288
ebel@sdu.dk

[2], making it a suitable and safe method for fully isolated charging or power supply [3]. This technology has gained popularity in various fields such as electronics, biomedical, unmanned aerial vehicles, and electric vehicles [4], [5]. However, achieving a balance between the primary and secondary coils in wireless charging of electric vehicles is challenging. The mismatch of the coils causes a significant reduction in mutual inductance, leading to deviations from the rated system performance and reducing system efficiency [6]. Improving the tolerance of IPT systems to coil mismatch is being investigated from three aspects: control methods, topology design, and magnetic coupling design.

From a control method perspective, references [7] and [8] obtain a constant output voltage by adding a DC/DC converter at the final stage of the IPT converter. However, this method may increase the complexity of the IPT system and reduce circuit efficiency. Self-oscillating frequency control is proposed in [9] to achieve output power adjustment and zero reactive power. However, frequency control may become unstable within specific parameter ranges, leading to power loss. Additionally, these control methods require voltage and current measurements at the output load and data exchange through wireless communication.

In terms of topology design, references [10] and [11] have well-analysed IPT converters using various topologies to demonstrate relatively small changes in inductances in different misalignment directions. Since the coupling channel is not
essentially changed, the efficient performance under misalignment phenomena is limited, and the output current varies with the load.

Regarding magnetic coupling design, references [12] and [13] propose bi-polar double-D (DD) pad, dual-square (DDQ) pad, and bipolar (BP) pad configurations to achieve uniform magnetic distribution. An LCC-S hybrid network, along with a BP coupler, is proposed in reference [13] to provide a constant output current with 20.6% magnetic coil mismatch in the X-axis direction. In similar research, reference [14] introduces an LCC-S topology with a Double-D quadrature (DDQ) coupler, in which a constant output voltage with 50% coil misalignment in the X-axis direction has been achieved. An LCC-S hybrid network is also presented in [15] with a DDQ coupler capable of providing a constant output current under 50% coil mismatch.

This paper introduces a hybrid LCC-S structure with a double magnetic coupling DDQ, shown in Fig. 1, to achieve constant output current by providing magnetic coupling between primary and secondary compensation coils. In fact, two main primary windings ($L_1$ for the LCC resonance loop and $L_5$ for the series resonance loop) form the DD part of the DDQ pad with a specific configuration to achieve zero mutual inductance. The Q pad at each side contains the compensator inductances ($L_1$ in primary and $L_4$ in secondary), which are integrated into the coupler pads. In this way, the output current remains smooth despite deviations in the coupling coefficient caused by changes in the air gap or misalignment. As a result, the voltage/current stress across the converter switches is reduced, as is the system's control complexity.

In this paper, first, the hybrid Battery Charger with LCC-S topology model with primary-secondary coupled compensation coils is presented, and the relationship between output

<figure><img src="image_2.png" /><figcaption>Fig. 1: LCC-S hybrid topology circuits.</figcaption></figure>

current and load inductance is analysed. The double DDQ coil structure is introduced to eliminate unwanted mutual inductances. Finally, a 1-kW prototype is built to achieve constant output current up to 53.3% mismatch between magnetic couplers, with a size of 60 mm in the Y-axis direction and up to 20% mismatch along the X-axis with pad size of 60 mm in this direction. It was also concluded that, around the optimum load, the overall system efficiency was kept within the acceptable range (higher than 90%) in all mismatch scenarios.

## Theoretical analysis

As shown in Fig. 2, there is mutual inductance between each pair of coils ($L_1$ to $L_6$), where $M_{14}$, $M_{23}$, $M_{56}$, $M_{13}$, $M_{16}$, $M_{24}$, and $M_{45}$ are the primary mutual inductances. The magnetic pads are designed in such a way that the mutual inductances $M_{12}$, $M_{25}$, $M_{15}$, $M_{35}$, $M_{46}$, $M_{36}$, $M_{35}$, and $M_{26}$ are all zero at all distances.

For each coil (L) and each capacitor (C) in Fig. 2, the resonance conditions are met as shown in (1).

$$ 
\begin{align} 
j\omega L_2 + 1/(j\omega C_2) &= j\omega L_1 = -1/j\omega C_1, \nonumber \\ 
j\omega L_5 &= -1/j\omega C_5, \nonumber \\ 
j\omega L_3 + 1/(j\omega C_3) &= j\omega L_4 = -1/j\omega C_4, \tag{1} \\ 
j\omega L_6 &= -1/j\omega C_6. \nonumber 
\end{align} 
$$

where, $\omega$ is the resonant frequency of the system, the coupling coefficient K between two coils follows (2), where $L_i$ and $L_j$ are the self-inductances of the first and second coils, respectively. $M_{ij}$ is also the mutual inductance between these two coils.

$$ K_{ij} = \frac{M_{ij}}{\sqrt{L_i L_j}}, \qquad (2) $$

To simplify the circuit, the equivalent resistance seen before the diode bridge is defined as $R_{eq}$, which can be obtained from (3).

<figure><img src="image_3.png" /><figcaption>Fig. 2: Equivalent fundamental wave circuit.</figcaption></figure>
$$R_{eq} = \frac{8}{\pi^2} R_L, \qquad (3)$$

where, $R_L$ is the output load resistance. To obtain the current in each mesh of the circuit, Equation (4) is used, which follows Kirchhoff's current law. To simplify the equations, it is assumed that $M_{13}$, $M_{16}$, $M_{24}$, and $M_{45}$ are zero.

$$ \left\{ \begin{aligned} \dot{I}_{p1} Z_{11} + \dot{I}_{p2} Z_{12} + \dot{I}_{s2} Z_{14} &= \dot{U}_1, \\ \dot{I}_{p1} Z_{21} + \dot{I}_{p2} Z_{22} + \dot{I}_{s1} Z_{23} &= 0, \\ \dot{I}_{p2} Z_{32} + \dot{I}_{s1} Z_{33} + \dot{I}_{s2} Z_{34} &= 0, \\ \dot{I}_{p1} Z_{41} + \dot{I}_{s1} Z_{43} + \dot{I}_{s2} Z_{44} + \dot{I}_{s3} Z_{46} &= 0, \\ \dot{I}_{p3} Z_{55} + \dot{I}_{s3} Z_{56} &= \dot{U}_2, \\ \dot{I}_{s2} Z_{64} + \dot{I}_{p3} Z_{65} + \dot{I}_{s3} Z_{66} &= 0. \end{aligned} \right. \qquad (4) $$

that $Z_{11}$ to $Z_{66}$ are Equivalent model impedances related to the capacitive and inductive components. by solving (4), the current in each circuit's loop can be expressed as (5).

$$ \left\{ \begin{aligned} I_{s2} &= \frac{U_1 M_{23}}{j\omega(L_1 L_4 + M_{14} M_{23})}, \\ I_{s3} &= \frac{U_2}{j\omega M_{56}}. \end{aligned} \right. \qquad (5) $$

The output current of the circuit is expressed as (6).

$$I_o = I_{s2} + I_{s3}. \qquad (6)$$

## Design of magnetic pads

In Fig. 3, magnetic pads designed using Maxwell software are depicted. The primary pad includes coils $L_2$ and $L_5$, designed in DD form, and coil $L_1$, designed in Q form. The secondary pad includes coils $L_3$ and $L_6$, designed in DD form, and coil $L_4$, designed in Q form. Ferrite cores with a square lattice shape are placed behind each magnetic pad to reduce leakage flux and improve the conduction of flux. The overall dimensions of the pads are 30 cm × 30 cm. A constant distance of 10 cm along the Z-axis between the magnetic pads is maintained, and the mismatch between the primary and secondary magnetic couplers in the Y and X directions is investigated.

In order to show the performance of the magnetic couplers in normal situations and misalignment, one winding from the primary side and its corresponding winding on the secondary side are

<figure><img src="image_4.png" /><figcaption>Fig. 3: DDQ coils structure; (a) both pads from front view(pad size is same for both sides), (b) Primary and secondary pad from top view, and (c) Assembled pads (all simulated in 3-D Maxwell).</figcaption></figure>

investigated using the finite element method, Ansys-Maxwell 3D. Other windings are assumed to have the same situation and are neglected in the initial simulation due to the long required running time.

In Fig. 4, the primary coil $L_2$ and its corresponding coil on the secondary side, $L_3$, are simulated with their actual sizes. Figs. 4(a)-(c) illustrate the cases of perfect alignment, misalignment of 160 mm in the Y direction, and 160 mm in the X direction, respectively. In Fig. 4(a), it can be observed that each of the coils is composed of two loops, 1 and 2. When two coils are exactly opposite to each other, the flux produced by coil $L_2$ in both loops 1 is upwards, and in both loops 2 is downwards. However, when coil $L_3$ moves 160 mm in the Y direction (Fig. 4(b)), loop 1 in coil $L_2$ and loop 2 in coil $L_3$ do not directly oppose each other. As a result, the coupling between these two loops decreases significantly. Due to the opposing flux between loop 1 of coil $L_3$ and loop 2 of coil $L_2$, negative coupling occurs at this distance. When coil $L_3$ deviates by 160 mm in the X direction (Fig. 4(c)), the flux between the two coils is still in the same direction but reduced in intensity. Therefore, for coils $L_2$ and $L_3$, a mismatch of 160 mm in the X direction only reduces the coupling between the two coils without switching the sign of the coupling coefficient. The same applies to the coil
<figure><img src="image_5.png" /><figcaption>Fig. 4: coil $L_2$ and $L_3$ in misalignment; (a) 0mm, (b) 160mm in Y-axis, and (c) 160mm in X-axis.</figcaption></figure>

pair of $L_5$ and $L_6$. However, due to the 90-degree difference in angle with respect to coils $L_2$ and $L_3$, a mismatch of 160 mm in the X direction for coils $L_5$ and $L_6$ leads to negative coupling between these two coils, while a mismatch of 160 mm in the Y direction only reduces the coupling between them without changing the coupling polarity. Meanwhile, due to the 90-degree rotation in angle between $L_2$ and $L_5$ on the primary side as well as $L_3$ and $L_6$ on the secondary side, the mutual inductance between each pair of coils at each side is almost zero at all times, as the upward and downward flux will fully cancel each other in all working scenarios, including alignment and a wide range of misalignments. The coupling between coils $L_1$ and $L_4$ only decreases with increased mismatch in both the X and Y directions, due to their symmetry in this direction because of their square coil shape.

In Fig. 5, the magnetic pads and the flux produced by coil $L_2$ are shown for misalignments of 0 mm and 160 mm in the Y direction and 60 mm in the X direction. In Fig. 5(a), it is observed that the primary and secondary pads are exactly opposite to each other; therefore, only the mutual inductances $M_{14}$, $M_{23}$, and $M_{56}$ have non-zero values, while all other mutual inductances are zero in this case. In Fig. 5(b), a mismatch of 60 mm in the X direction between the primary and secondary pads leads to non-zero values for mutual inductances $M_{13}$ and $M_{24}$, in addition to

<figure><img src="image_6.png" /><figcaption>Fig. 5: Double DDQ coils structure in misalignment; (a) 0mm, (b) 160mm in Y-axis, and (c) 60mm in X-axis.</figcaption></figure>

$M_{14}$, $M_{23}$, and $M_{56}$. In Fig. 5(c), a mismatch of 160 mm in the Y direction between the secondary and primary pads results in non-zero values for mutual inductances $M_{14}$, $M_{23}$, $M_{56}$, $M_{16}$, and $M_{45}$, while all other mutual inductances are zero.

The reason for investigating a mismatch of 60 mm in the X direction and a mismatch of 160 mm in the Y direction is that within these ranges, the output current remains almost constant.

The variation in mutual inductances between all coils $L_1$ to $L_6$ with respect to misalignment in the X direction is shown in Fig. 6. At a distance of 0 mm, mutual inductances $M_{14}$, $M_{23}$, and $M_{56}$ have non-zero values, while all other mutual inductances are almost zero. Accordingly, $K_{14} = 0.229$, $K_{23} = 0.244$, and $K_{56} = 0.227$, while other possible couplings are zero. At the maximum eligible misalignment in the X direction, equal to 60 mm, at which a constant output is obtained, mutual inductances $M_{14}$, $M_{23}$, $M_{56}$, $M_{24}$, and $M_{13}$ are non-zero, where $K_{14} = 0.174$, $K_{23} = 0.172$, $K_{56} = 0.206$, $K_{13} = -0.1$, and $K_{24} = 0.1$. The mutual inductance $M_{23}$ has significantly decreased due to misalignment; however, the increases in the mutual inductances $M_{13}$ and $M_{24}$ compensate for this reduction, resulting in a nearly constant output current. In other words, the increase in the mutual inductances $M_{24}$ and $M_{13}$ along the X-axis misalignment is advantageous and is contingent upon the design of the magnetic couplers.
<figure><img src="image_7.png" /><figcaption>Fig. 6: The changes of mutual inductances between coils $L_1$ to $L_6$ in term of the misalignment in X-axis.</figcaption></figure>

<figure><img src="image_8.png" /><figcaption>Fig. 7: The changes of mutual inductances between coils $L_1$ to $L_6$ in term of the misalignment in Y-axis.</figcaption></figure>

Changes in mutual inductances between all coils $L_1$ to $L_6$ with respect to misalignment in the Y direction are illustrated in Fig. 7. In the aligned mode, mutual inductances $M_{14}$, $M_{23}$, and $M_{56}$ are non-zero, while all other mutual inductances are zero, i.e., $K_{14} = 0.229$, $K_{23} = 0.113$, $K_{56} = 0.227$, and the rest are zero. As the mismatch reaches its maximum at 160 mm, mutual inductances $M_{16}$ and $M_{45}$ also acquire non-zero values in addition to $M_{14}$, $M_{23}$, and $M_{56}$, where $K_{14} = 0.2$, $K_{23} = 0.135$, $K_{56} = -0.067$, $K_{16} = 0.105$, and $K_{45} = -0.105$. Similarly, the increases in the mutual inductances $M_{16}$ and $M_{45}$ in the misalignment mode compensate for the reduction in the mutual inductances $M_{56}$, thereby enhancing the misalignment tolerance and ensuring that the output current remains approximately constant.

## Experimental evaluation

In order to investigate the accuracy and feasibility of the above method, a laboratory prototype (Figs. 8 and 9) has been constructed. As shown in Fig. 8, based on the magnetic design presented in Fig. 3, two magnetic couplers with dimensions of 30 mm × 30 mm have been built. For each coil, a plastic frame has been designed and constructed using a 3D printer to increase the precision in the construction of the magnetic coils.

<figure><img src="image_9.png" /><figcaption>Fig. 8: Experimental Magnetic couplers; (a) Q coil ($L_1$ and $L_4$), (b) DD coil($L_2$, $L_3$, $L_5$ and $L_6$), and (c) Double DDQ magnetic coupler in both sides.</figcaption></figure>

<figure><img src="image_10.png" /><figcaption>Fig. 9: The entire experimental platform.</figcaption></figure>

In Fig. 9, the overall laboratory platform is demonstrated, which uses an STM32F103 processor for generating gate signals and TIP460N MOSFETs as the power electronic switches. MKP capacitors are used as compensation capacitors. A variable autotransformer, Varia, is utilized for the system power supply to control the supply voltage, along with a diode bridge and a capacitor filter for rectifying the AC voltage. On the secondary side, a heater is employed as a DC load, and there is a 10 cm distance between the secondary and primary sides.
<table><caption>Table I. System Parameters.</caption><thead><tr><th>Parameter</th><th>Value</th><th>rating</th></tr></thead><tbody><tr><td>Operating frequency (fs)</td><td>85 kHz</td><td rowspan="2">8 A</td></tr><tr><td>Primary-side transmitting coil (L<sub>1</sub>)</td><td>17.5 uH</td></tr><tr><td>Primary-side transmitting coil (L<sub>2</sub>)</td><td>150 uH</td><td>8 A</td></tr><tr><td>Secondary-side transmitting coil (L<sub>3</sub>)</td><td>17.5 uH</td><td>8 A</td></tr><tr><td>Secondary-side transmitting coil (L<sub>4</sub>)</td><td>150 uH</td><td>8 A</td></tr><tr><td>Primary-side transmitting coil (L<sub>5</sub>)</td><td>156 uH</td><td>8 A</td></tr><tr><td>Secondary-side transmitting coil (L<sub>6</sub>)</td><td>156 uH</td><td>8 A</td></tr><tr><td>Primary-side compensation capacitor (C<sub>1</sub>)</td><td>206.5 nF</td><td>1.6 KV</td></tr><tr><td>Primary-side compensation capacitor (C<sub>2</sub>)</td><td>25 nF</td><td>1.6 KV</td></tr><tr><td>Secondary-side compensation capacitor (C<sub>3</sub>)</td><td>25 nF</td><td>1.6 KV</td></tr><tr><td>Secondary-side compensation capacitor (C<sub>4</sub>)</td><td>206.5 nF</td><td>1.6 KV</td></tr><tr><td>Primary-side compensation capacitor (C<sub>5</sub>)</td><td>22.5 nF</td><td>1.6 KV</td></tr><tr><td>Secondary-side compensation capacitor (C<sub>6</sub>)</td><td>22.5 nF</td><td>1.6 KV</td></tr><tr><td>Output filter capacitor (C<sub>L</sub>)</td><td>60 uF</td><td rowspan="2">200 V</td></tr><tr><td>Input DC voltage (V<sub>in</sub>)</td><td>100 V</td></tr></tbody></table>

The main parameters of the laboratory prototype are listed in Table I, where the values of the customized inductances and the used capacitors are specified, along with their current/voltage ratings. The output current graph with respect to mismatch in the X-axis direction for three different output loads is provided in Fig. 10. The 10 Ω load is the rated operating point, around which the system has been examined within a range of 8 Ω to 12 Ω. As the distance in the X-axis increases, the output current decreases. The highest output current corresponds to a distance

of 0 mm, where the current reaches up to 8.6 A, while the lowest output current corresponds to a distance of 60 mm, where the output current decreases to 6.85 A, with an average load current of 7.72 A. As a result, the output current tolerates up to 12% for a mismatch of 20% compared to the length of the pad in the X-axis direction.

The changes in the output current based on the misalignment in the Y-axis for the three above-mentioned loads of 8 Ω, 10 Ω, and 12 Ω are provided in Fig. 11. From a distance of 0 mm to 60 mm, i.e., a 53% mismatch compared to the size of the magnetic pads in the Y-axis direction, the output current remains almost constant at 8.47 A. The highest value is 8.75 A, and the lowest value is 8.2 A, which shows a maximum tolerance in output current of around 3.5% in the extreme case (53% misalignment between magnetic couplers).

The efficiency under misalignment conditions is also investigated to ensure the efficient performance of the proposed hybrid system with coupled compensator inductances. Fig. 12 shows the changes in circuit efficiency with respect to mismatch in the X-axis direction for three different output loads. The highest circuit efficiency corresponds to a 40 mm mismatch in the X-axis direction with η = 90.6%. It can be

<figure><img src="image_11.png" /><figcaption>Fig. 10: Variation in the output current relative to the X-axis misalignment.</figcaption></figure>

<figure><img src="image_12.png" /><figcaption>Fig. 11: Changes in the output current versus misalignment in Y-axis.</figcaption></figure>

<figure><img src="image_13.png" /><figcaption>Fig. 12: Variation in efficiency in term of the misalignment in X-axis.</figcaption></figure>
<figure><img src="image_14.png" /><figcaption>Fig. 13: Variation in efficiency in term of the misalignment in Y-axis.</figcaption></figure>

concluded from Fig. 12 that, unlike the conventional IPT system with a single resonance circuit, the maximum efficiency in this hybrid system occurs at a non-zero misalignment. This is because, as the misalignment in the X-axis direction increases, the mutual inductances $M_{23}$, $M_{14}$, and $M_{56}$ decrease, while the mutual inductances $M_{13}$ and $M_{24}$ increase. As a result, the maximum efficiency of the circuit does not occur at a misalignment of 0 mm.

In Fig. 13, changes in circuit efficiency with respect to mismatch in Y-axis direction for three different output loads are plotted. Like other type of IPT systems, the highest efficiency is related to 0mm mismatch between primary and secondary sides, which is equal to 90%, and gradually decreases as the mismatch increases in Y-axis direction.

To visually reflect the advantages of the proposed system, the structure proposed in this paper is compared with the recently most relevant reference, [15], with the same figure of merits. This includes the number of coils, coil structure, output power, number of compensator elements to be misalignment tolerant, the number of passive components, misalignment tolerance range, and output characteristic. The comparison results are presented in Table II from different aspects. It can be concluded that the proposed system has advantages in keeping efficient performance also in Y-axis misalignment and lower number of passive/active electronic components, in exchange for a bit weakness in efficient performing in wide range of X-axis misalignment and number of magnetic coils. Comparison with the most relevant work(reference[15]).

## Conclusion

This paper presents a Double DDQ magnetic coupler with an LCC-S hybrid topology, integrating primary and secondary compensation coils to enhance system performance tolerance to

<table><caption>Table II. Comparison of the proposed structure with the most relevant state of the arts.</caption><thead><tr><th></th><th>Ref [15]</th><th>Proposed structure</th></tr></thead><tbody><tr><td>Number of coils</td><td>4</td><td>6</td></tr><tr><td>Coil structure</td><td>DDQ</td><td>Duble DDQ</td></tr><tr><td>Output power</td><td>600-W</td><td>1-kW</td></tr><tr><td>Number of Compensator elements to be misalignment tolerant</td><td>8</td><td>6</td></tr><tr><td>Number of passive components</td><td>4</td><td>2</td></tr><tr><td>Misalignment tolerance range</td><td>X-axis misalignment: ±140 mm (50%)<br>Y-axis misalignment: N/A</td><td>X-axis misalignment: ±60 mm (20%)<br>Y-axis misalignment: ±160 mm (53.3%)</td></tr><tr><td>Output load characteristic</td><td>CC</td><td>CC</td></tr></tbody></table>

mismatches in the X and Y axes. This design eliminates the need for compensation coils on the PCB, reducing the on-board converter's size. A mathematical model was developed to analyse the system's output characteristics, and the effects of misalignment in the X and Y directions on mutual inductances were studied. A 1kW platform was constructed to validate the theoretical analysis. The results show that the output current remains nearly constant at around 8A, with a variation of 12% at a 60mm mismatch in the X-axis and 3.5% at a 160mm mismatch in the Y-axis. The system efficiency remains above 80%, reaching up to 90.6% for X/Y misalignment up to 60mm. Future research will focus on analysing circuit performance with varying air gap distances between the magnetic couplers under different loads, as well as implementing a closed-loop control system and investigating Zero Voltage Switching (ZVS) conditions.

## References

[1] W. Chen, A. Babaki, F. Lin and G. A. Covic, "Impact of Secondary Sizing on EV Dynamic Charging System Power Capability," IECON 2023-
49th Annual Conference of the IEEE Industrial Electronics Society, Singapore, Singapore, 2023.

[2] S. Bajelvand, A. Y. Varjani, S. Vaez-Zadeh and A. Babaki, "Design of a High-Efficient IPT System for Battery Charging Under CP-CV Charging Scenarios," in IEEE Transactions on Energy Conversion, vol. 38, no. 3, pp. 1650-1658, Sept. 2023.

[3] J. Wang and Z. Huang, "A Single-Switch WPT Circuit With Inherent CCO-CVO for Battery Charging," in IEEE Transactions on Transportation Electrification, vol. 10, no. 1, pp. 1957-1968, March 2024.

[4] M. Vinod, D. Kishan and B. D. Reddy, "Three-Leg DC-DC Converter for Efficient Inductive Power Transfer of Electric Vehicles for Wide-Range Battery Applications," in IEEE Transactions on Power Electronics, vol. 38, no. 8, pp. 9317-9321, Aug. 2023.

[5] R. Mai, Y. Chen, Y. Zhang, N. Yang, G. Cao and Z. He, "Optimization of the Passive Components for an S-LCC Topology-Based WPT System for Charging Massive Electric Bicycles," in IEEE Transactions on Industrial Electronics, vol. 65, no. 7, pp. 5497-5508, July 2018.

[6] B. Song, B. Du, S. Cui, Y. Li and C. Zhu, "Mechanism Analysis of Output Fluctuation in a Three-Phase Dynamic Wireless Charging System," in IEEE Transactions on Industrial Electronics, vol. 69, no. 3, pp. 2252-2264, March 2022.

[7] Z. Li, C. Zhu, J. Jiang, K. Song and G. Wei, "A 3-kW Transfer System for Sightseeing Car Supercapacitor Charge," in IEEE Transactions on Power Electronics, vol. 32, no. 5, pp. 3301-3316, May 2017.

[8] H. H. Wu, A. Gilchrist, K. D. Sealy and D. Bronson, "A High Efficiency 5 kW Inductive Charger for EVs Using Dual Side Control," in IEEE Transactions on Industrial Informatics, vol. 8, no. 3, pp. 585-595, Aug. 2012.

[9] L. Xu, Q. Chen, X. Ren, S. C. Wong and C. K. Tse, "Self-Oscillating Resonant Converter With Contactless Power Transfer and Integrated Current Sensing Transformer," in IEEE Transactions on Power Electronics, vol. 32, no. 6, pp. 4839-4851, June 2017.

[10] L. Zhao, D. J. Thrimawithana and U. K. Madawala, "Hybrid Bidirectional Wireless EV Charging System Tolerant to Pad Misalignment," in IEEE Transactions on Industrial Electronics, vol. 64, no. 9, pp. 7079-7086, Sept. 2017.

[11] Y. Chen, B. Yang, Z. Kou, Z. He, G. Cao and R. Mai, "Hybrid and Reconfigurable IPT Systems With High-Misalignment Tolerance for Constant-Current and Constant-Voltage Battery Charging," in IEEE Transactions on Power Electronics, vol. 33, no. 10, pp. 8259-8269, Oct. 2018.

[12] A. Zaheer, H. Hao, G. A. Covic and D. Kacprzak, "Investigation of Multiple Decoupled Coil Primary Pad Topologies in Lumped IPT Systems for Interoperable Electric Vehicle Charging," in IEEE Transactions on Power Electronics, vol. 30, no. 4, pp. 1937-1955, April 2015.

[13] S. Y. Choi, J. Huh, W. Y. Lee and C. T. Rim, "Asymmetric Coil Sets for Wireless Stationary EV Chargers With Large Lateral Tolerance by Dominant Field Analysis," in IEEE Transactions on Power Electronics, vol. 29, no. 12, pp. 6406-6420, Dec. 2014.

[14] X. Qu, Y. Yao, D. Wang, S. C. Wong and C. K. Tse, "A Family of Hybrid IPT Topologies with Near Load-Independent Output and High Tolerance to Pad Misalignment," in IEEE Transactions on Power Electronics, vol. 35, no. 7, pp. 6867-6877, July 2020.

[15] Z. Gong, Q. Shi, J. Li and X. Tong, "Parallel Hybrid Wireless Charging with Misalignment Tolerance and Constant Current Output," 2022 International Conference on Mechanical and Electronics Engineering (ICMEE), Xi'an, China, 2022.