# Compact Cassegrain Antenna based-on a Low-Profile Multi-faceted Reflectarray

Borja Imaz-Lueje, D. Gonzalez-Ovejero, Marcos R. Pino, Manuel Arrebola, Ronan Sauleau

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Borja Imaz-Lueje, D. Gonzalez-Ovejero, Marcos R. Pino, Manuel Arrebola, Ronan Sauleau. Compact Cassegrain Antenna based-on a Low-Profile Multi-faceted Reflectarray. IEEE Transactions on Antennas and Propagation, 2025, pp.1-1. 10.1109/tap.2025.3585279. hal-05233850

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# Compact Cassegrain Antenna Based on a Low-Profile Multifaceted Reflectarray

Borja Imaz-Lueje<sup>D</sup>, Member, IEEE, David González-Ovejero<sup>D</sup>, Senior Member, IEEE, Marcos R. Pino<sup>D</sup>, Member, IEEE, Manuel Arrebola<sup>D</sup>, Senior Member, IEEE, and Ronan Sauleau<sup>D</sup>, Fellow, IEEE

**Abstract**—This article presents the design of a compact Cassegrain antenna that uses a multifaceted reflectarray (MFRA) as a main reflector. The multifaceted surface comprises several panels with different angular orientations, to conform a cylindrical parabolic profile along one plane. A prototype of this antenna concept has been manufactured and tested to provide a high gain in dual-linear polarization at the Ka-band. When compared with an equivalent single-facet aperture, the multifaceted approach effectively benefits from the parabolic profile optics to reduce the differential spatial phase delay. Consequently, the required phase excursion is smaller compared with the single-facet flat topology, resulting in enhanced beamwidth stability in the sectorization plane. The proposed low-profile multifaceted antenna demonstrates better in-band performance than other similar works in the literature, while maintaining similar compactness.

**Index Terms**—Cassegrain system, compact antennas, Ka-band, multifacet reflector, reflectarray antennas.

## I. INTRODUCTION

ADVANCED wireless applications demand high-gain and broadband antennas to meet the challenging specifications of their systems, such as high throughput [1], [2]. Printed reflectarrays [3] have emerged as a competitive and flexible solution for providing high gain and efficiency compared with other antenna solutions, such as arrays or parabolic reflectors. On one hand, reflectarrays are spatially fed antennas, so they do not require complex and lossy corporate feeding networks as conventional arrays. On the other hand, they can achieve aperture efficiencies comparable to parabolic reflectors, while using low-profile and low-cost PCB technology with high

stowage capability [4], [5]. Furthermore, reflectarrays can be properly designed to generate different radiation patterns (single-beam, multibeam, or shaped beam) [5], [6], [7], [8], steer the beam [9], operate in different bands simultaneously [10], or provide polarization conversion [11], among other capabilities. All these different operations can be performed through proper design of the radiating elements in the reflectarray, so the manufacturing costs are not affected, in contrast to parabolic reflectors [12], [13]. Due to these attractive features, printed reflectarrays have been proposed in many applications, including remote sensing [12], cellular networks [15], [16], and satellite communications [6], [11], [17], [18].

Reflectarray antennas typically follow a center-fed or single-offset configuration [5], [6], [19], in which the primary feed is located at a distance from the reflectarray panel that is comparable to the aperture dimensions [3]. Nevertheless, such configurations exhibit limited stowage, making them unsuitable for applications where the compactness is critical [6], [12], [21]. An alternative is the use of a dual-reflector arrangement based on a Cassegrain configuration [22], [23], [24], [25], [26], [27], [28], [29], [30] or folded reflectarrays [32], [33], [34]. The optics of such structures significantly reduce the antenna profile and allow for better accommodation of the primary feed. However, folded reflectarrays are typically limited to operating in single polarization [3]. Regarding the Cassegrain configuration, the reflectarray aperture can perform the roles of the subreflector [22], the main reflector [23], [24], [25], [26], [27], [28], [29], or both [30]. For instance, several works propose the use of reflectarrays as the main reflector in a classical Cassegrain configuration [24], [25], [26], [27], [28], [29], where the feed and subreflector are positioned in the center of the main reflector. Here, the antenna typically features lower focal length to aperture ratio ($f/D$), providing a compact solution. This is particularly advantageous for some applications such as monopulse radar [24], [25] or satellite communication [29], [30].

The drawback of typical Cassegrain reflectarrays compared with their reflector counterparts is their inherently narrower bandwidth, which arises from the in-band behavior of the main printed reflectarray panel [3]. This bandwidth limitation is due to the performance of the unit-cell elements and the differential spatial phase delay effect [32]. The latter effect is particularly pronounced in electrically large apertures or reflectarrays with small $f/D$ (as occurs in conventional Cassegrain configurations). Several broadband strategies have

Received 3 September 2024; revised 2 June 2025; accepted 24 June 2025. Date of publication 10 July 2025; date of current version 14 October 2025. This work was supported in part by the MCIN/AEI/10.13039/501100011033 under Grant PID2020-114172RB-C21, Grant PID2023-146246OB-C32, and Grant TED2021-130650B-C22; and in part by European Union (UE) through NextGenerationEU/PRTR. (Corresponding author: Manuel Arrebola.)

Borja Imaz-Lueje is with the Department of Signal Theory and Communications, Universidad Rey Juan Carlos, 28032 Madrid, Spain (e-mail: borja.imaz@urjc.es).

David González-Ovejero and Ronan Sauleau are with the Univ Rennes, CNRS, Institut d'Électronique et des Technologies du Numérique (IETR), UMR 6164, 35000 Rennes, France (e-mail: david.gonzalez-ovejero@univ-rennes.fr; ronan.sauleau@univ-rennes.fr).

Marcos R. Pino is with the Department of Electrical Engineering, Group of Signal Theory and Communications, Universidad de Oviedo, 33206 Gijón, Spain (e-mail: mpino@uniovi.es).

Manuel Arrebola is with the Information Processing and Telecommunications Center (IPTC), Applied Electromagnetics Group (GEA), Universidad Politécnica de Madrid (UPM), 28040 Madrid, Spain (e-mail: manuel.arrebola@upm.es).

Digital Object Identifier 10.1109/TAP.2025.3585279
<figure><img src="image_2.png" /><figcaption>Fig. 1. Sketch of a possible application of the proposed antenna concept: deployable antenna onboard small satellite (3U CubeSat).</figcaption></figure>

proposed to tackle this issue [36], [37], [38], [39], [40], [41], [42]. Specifically, the use of true-time delay (TTD) unit-cells [36] or the optimization of radiating elements with multiple resonances [8] mitigates the differential spatial phase delay effect. However, both the methods are typically based on multilayer topologies, which increase the insertion losses and weight of the reflectarray. Another solution to reduce the differential spatial phase delay is to manipulate the antenna optics, using parabolic [37] or multifaceted apertures [38], [39], [40], [41], [42]. These latter antenna concepts partially correct the differential spatial phase delay in single-offset reflectarrays, using low-profile and low-loss planar apertures, as discussed in the literature [42]. Multifaceted reflectarrays (MFRAs) maintain most of the advantages above-mentioned for conventional reflectarrays, so they exhibit similar antenna efficiency to parabolic reflectors but with better flexibility in the field control and cost-effectiveness in its manufacturing.

This article explores the use of MFRAs as the primary reflector, building upon the very preliminary numerical study reported in [43]. The main contribution of this work is the full characterization of an MFRA in a dual-reflector Cassegrain configuration, including the design, fabrication, and testing of a demonstrator with a realistic feeder. Unlike other composite antenna solutions [39], [40], [41], [42], the multifaceted surface is integrated in a compact antenna structure, with a reduced *f/D* ratio and using a Cassegrain feed network (primary feed and subreflector), so these concerns have been taken into account during the design process. The performance of this demonstrator has been compared with that of a dual-reflector Cassegrain with single-facet reflectarray (SFRA) as the main reflector, showing superior capabilities, and with other similar works in the literature. Specifically, the proposed antenna demonstrates a reduction in phase excursion in the plane where the facets are bent to conform to a parabolic profile. This feature partially corrects the defocusing of the pencil beam in-band along the sectorization plane and, therefore, enhances the stability of the beamwidth in-band. Compared with other existing works, the multifaceted antenna

also exhibits an enhancement in the in-band performance, while maintaining a low-profile and compact structure. These features demonstrate the feasibility of the proposed multifaceted approach in different applications such as monopulse radars [24], [25] or as an antenna solution onboard small satellites. In the latter case, it could follow the deployment mechanism illustrated in Fig. 1, similar to those used in other antenna structures based on reflectarray technology [5], [6], [29], [30].

The rest of this article is organized as follows. Section II describes the proposed antenna structure, including its main parameters. Section III presents the antenna performance, including both the predicted and measured results. Section IV focuses on comparing the proposed demonstrator with an equivalent SFRA and with other similar works in the literature. Finally, Section V draws the conclusions of this work.

## II. DESIGN OF THE MULTIFACETED CASSEGRAIN REFLECTARRAY

The proposed Cassegrain system is schematically presented in Fig. 2, and its optical parameters are listed in Table I. The antenna is designed at 30 GHz to provide a high-gain pencil beam in the broadside direction and to operate in dual-linear polarization (X- and Y-polarizations). Sections II-A–II-C detail the features of the proposed Cassegrain design.

### A. Antenna Optics

As illustrated in Fig. 2, the MFRA comprises three elements: the main reflector, the primary feed, and the subreflector. Table I lists the optical parameters of each of these elements.

The main reflector consists of a multifaceted aperture (yellow consecutive blocks in the side view in Fig. 1) composed by three reflectarray panels of identical size. Each panel consists of 900 square unit-cells, with a side length of 4 mm, and arranged according to a Cartesian lattice of 18 × 50 cells. The reflectarray panels follow a parabolic profile along the XZ plane (see dashed black line in Fig. 1) with its focal point

<table><caption>TABLE I
OPTICAL PARAMETERS OF THE CASSEGRAIN MULTIFACETED STRUCTURE</caption><thead><tr><th colspan="2">Multi-faceted reflectarray main reflector</th></tr></thead><tbody><tr><td>Total size (D)</td><td>203 mm × 200 mm</td></tr><tr><td>Number of panels</td><td>3</td></tr><tr><td>Panel size</td><td>72 mm × 200 mm</td></tr></tbody><thead><tr><th colspan="2">Primary feed: pyramidal horn antenna</th></tr><tr><td colspan="2">(data in the reflectarray coordinate system shown in Fig. 1)</td></tr></thead><tbody><tr><td>Aperture location</td><td>(0, 0, 52) mm</td></tr><tr><td>Subtended angle (θ<sub>r</sub>)</td><td>39.3°</td></tr></tbody><thead><tr><th colspan="2">Sub-reflector: metallic plate with hyperbolic profile</th></tr><tr><td colspan="2">(data in the reflectarray coordinate system shown in Fig. 1)</td></tr></thead><tbody><tr><td>Distance with the main reflector (h<sub>s</sub>)</td><td>70 mm</td></tr><tr><td>Focal point (P<sub>f,1</sub>)</td><td>(0, 0, 52) mm</td></tr><tr><td>Focal point (P<sub>f,2</sub>)</td><td>(0, 0, 80) mm</td></tr><tr><td>Diameter of the plate (d<sub>s</sub>)</td><td>31.4 mm</td></tr><tr><td>Subtended angle (θ<sub>p</sub>)</td><td>60.6°</td></tr></tbody></table>
<figure><img src="image_3.png" /><figcaption>Fig. 2. Sketch of the multifaceted Cassegrain reflectarray and cross section of the structure (XZ plane).</figcaption></figure>

at $P_f = (0, 0, 80)$ mm. Thus, the lateral panels exhibit mirror symmetry in the XZ plane and are tilted 24.3°.

The Cassegrain feed system (i.e., the feed horn and the subreflector) has been designed properly to minimize volume and blockage of the field radiated by the main reflector. The primary feed is a pyramidal horn antenna (ATM-28-441) with a nominal gain of 15 dBi. The aperture of the horn is placed at 52 mm from the center of the central reflectarray panel. Since the available source is single linearly polarized, the feed is properly rotated around the z-axis of Fig. 2 to provide the dual-polarization feature of the antenna system. However, a dual-polarized source can be used to avoid this rotation. The subreflector is a metallic plate with a hyperbolic profile; one of its focal points ($P_{f,1}$) coincidences with the location of the horn aperture and the other one with the focal point of the parabolic profile ($P_{f,2} = P_f$). The diameter of the subreflector ($d_s$) corresponds to 15% of the total side length of the main reflector (D). The subtended angle between the primary feed and the subreflector is 39.3°, which translates into an edge taper illumination of 17 dB for the subreflector.

According to this antenna optics, the Cassegrain system exhibits an $f/D$ ratio of 0.4 and a compactness (ratio $h_s/D$) of 0.35. This configuration presents a significant reduction of the antenna volume compared with other single-facet single-offset reflectarray configurations [6], [17], [18], [19], [42], with $h_s/D$ ratios of about 1.0.

## B. Unit-Cell Characterization

Each reflectarray panel comprises unit-cells whose geometry is presented in Fig. 3(a). They consist of a variable-size rectangular patch, printed on a single layer of substrate DiClad880 ($\epsilon_r = 2.26; \tan \delta = 0.0025$) with thickness $h = 0.762$ mm. As already mentioned, the distance between unit-cells in both the dimensions ($d_x = d_y$) is 4.0 mm, which corresponds to $0.4\lambda_0$ at the design frequency. This unit-cell topology has been deliberately selected to demonstrate the added value of the MFRA optical structure, working in dual-linear polarization. More complex unit-cell topologies could be used instead to provide additional features to the MFRA design.

The performance of the unit-cell is computed by a home-made analysis tool based on the method of moments in spectral domain (MoM-SD) reported and validated in [44].

Fig. 3(b) and (c) depicts the direct reflection coefficient of the cell when the dimensions of the patch (a, b) vary, considering the impinging wave polarized in the X-direction. The low-profile unit-cell exhibits low losses at different frequencies and under various angles of incidence. The magnitude of the reflection coefficient is better than -0.4 dB (i.e., more than 90% of the incident energy is effectively reflected). Besides, the unit-cell topology provides an S-shaped phase response typical of these unit-cell topologies [28], [45], with good angular stability but a phase range restricted to 280°, less than a full phase cycle. The phase response is stable around frequencies close to the design frequency, but a significant phase excursion appears at off-center frequencies [see Fig. 3(b)]. A similar behavior has been observed in the Y-polarization.

## C. Reflectarray Design Procedure

To provide the desired radiation pattern, the reflectarray panels have been designed according to the following procedure.

First, the horn antenna is analyzed with CST software [46] to obtain the fields at the horn aperture. Subsequently, the field over the reflectarray surface is computed using the GRASP tool from TICRA [47], incorporating a model of the entire Cassegrain feed (horn and subreflector). Here, the horn is modeled based on the fields obtained in the previous step. Fig. 4 shows the incident field in each reflectarray panel expressed in terms of magnitude and phase. The difference in the illumination between polarizations is due to the projections of each field component onto the multifaceted aperture. However, the edge taper illumination of the multifaceted aperture is about 17 dB for both the polarizations, which ensures lower diffraction at the edges of the MFRA and low spillover.

Then, the reflectarray surface is designed to provide a high-gain pencil beam pattern in each direction $\hat{r}_b(\theta_b, \varphi_b)$ from the incident field depicted in Fig. 4. The phase distribution provided by the reflectarray aperture $\phi(\vec{r}_n)$ is computed analytically, according to the expression [3]

$$ \phi(\vec{r}_n) = \phi_{\text{inc}}(\vec{r}_n) - k_0 (\vec{r}_n \cdot \hat{r}_b) \quad (1) $$

with $\vec{r}_n$ being the position vector of the nth radiating element, $k_0$ the propagation constant in vacuum at the design frequency, and $\phi_{\text{inc}}(\vec{r}_n)$ the phase of the impinging wave on each element [see Fig. 4(c)]. Considering the coordinate system shown in Fig. 1, the phase distribution implemented by the reflectarray is set to generate a broadside pencil beam ($\theta_b, \varphi_b) = (0.0°, 0.0°)$ for both the polarizations. Fig. 5 provides the phase distribution of the aperture for the X-polarization.

After obtaining the phase distribution, the dimensions of the rectangular patches of each unit-cell are properly selected, following an iterative process element by element. A zero-finding routine is used to adjust the dimensions of the rectangular patch (a, b) to achieve the required phase response for each polarization. The response of the unit-cell is evaluated at the design frequency, using the MoM-SD [44] and considering the real incidence angle on each radiating element. As it can
<figure><img src="image_4.png" /><figcaption>Fig. 3. Phase-shifter used in the multifaceted structure. (a) Sketch of the unit-cell. (b) Response of the reflection component under normal incidence and different frequencies. (c) Response of the reflection component at 30.0 GHz with different angles of incidence.</figcaption></figure>

<figure><img src="image_5.png" /><figcaption>Fig. 5. Planar representation of the phase distribution in degrees (°) required in the MFRA. Design frequency: 30 GHz.</figcaption></figure>

<figure><img src="image_6.png" /></figure>

<figure><img src="image_7.png" /><figcaption>Fig. 4. Simulated incident field in the multifaceted aperture from the Cassegrain feed at 30 GHz. Normalized amplitude (dB) of the field for (a) X- and (b) Y-polarizations; (c) phase in degrees (°) of the incident field for X-polarization.</figcaption></figure>

<figure><img src="image_8.png" /><figcaption>Fig. 6. CST simulation model of the Cassegrain system.</figcaption></figure>

be seen in Section II-B, the maximum phase provided by the unit-cell is 280°, so this feature has been considered during the design process. For elements that require a reflection

phase larger than 280°, the algorithm adjusts the patch sizes to provide this limiting phase value, thereby minimizing the phase errors resulting from this restriction.

The layouts resulting from this process are shown in Figs. 2 and 6.

### III. ANTENNA PERFORMANCE

#### A. Numerical Analysis

The performance of the proposed antenna was evaluated through a full-wave simulation using CST software [46]. Fig. 6 depicts the model of the simulated antenna, which includes the pyramidal horn, the subreflector, and the main MFRA. The metallization of the reflectarray surface was modeled with cooper ($\sigma = 3.7 \cdot 10^7$ S/m) while the feed and subreflector were modeled with aluminum ($\sigma = 3.56 \cdot 10^7$ S/m).
<table><caption>TABLE II
PREDICTED RF PERFORMANCE OF THE CASSEGRAIN SYSTEM</caption><thead><tr><th rowspan="2">f [GHz]</th><th rowspan="2">Gain [dBi]</th><th colspan="3">SLL [dB]</th><th colspan="3">HPBW [°]</th><th rowspan="2">ε<sub>ap</sub> [%]</th></tr><tr><th>E-Plane</th><th>H-Plane</th><th>D-Plane</th><th>E-Plane</th><th>H-Plane</th><th>D-Plane</th></tr></thead><tbody><tr><td>27.0</td><td>28.7 / 28.0</td><td>-12.3 / -15.9</td><td>-18.4 / -18.2</td><td>-23.9 / -18.2</td><td>3.0 / 3.8</td><td>7.2 / 5.3</td><td>3.5 / 4.2</td><td>59.3/56.8</td></tr><tr><td>28.5</td><td>32.3 / 31.4</td><td>-12.4 / -17.3</td><td>-12.4 / -8.1</td><td>-18.3 / -15.7</td><td>2.9 / 3.2</td><td>3.0 / 3.2</td><td>2.9 / 3.2</td><td>59.9/57.4</td></tr><tr><td>30.0</td><td>31.1 / 31.7</td><td>-13.3 / -14.9</td><td>-16.0 / -16.0</td><td>-16.0 / -16.0</td><td>3.2 / 3.1</td><td>2.9 / 3.0</td><td>3.1 / 3.1</td><td>60.5/58.0</td></tr><tr><td>31.5</td><td>28.8 / 28.5</td><td>-13.0 / -18.5</td><td>-19.5 / -20.1</td><td>-17.2 / -20.1</td><td>3.0 / 3.3</td><td>3.8 / 3.3</td><td>3.3 / 3.3</td><td>59.1/56.2</td></tr><tr><td>33.0</td><td>25.7 / 26.7</td><td>-17.3 / -14.7</td><td>-15.4 / -16.7</td><td>-13.4 / -17.6</td><td>3.2 / 2.7</td><td>4.9 / 4.7</td><td>3.8 / 3.4</td><td>57.6/54.3</td></tr></tbody></table>

Blue data corresponds to Pol. X and red one to Pol. Y.

<figure><img src="image_9.png" /></figure>

<figure><img src="image_10.png" /></figure>

<figure><img src="image_11.png" /><figcaption>Fig. 7. Simulated radiation pattern in gain (dBi) evaluated at different frequencies. (a) f = 27 GHz. (b) f = 30 GHz. (c) f = 33 GHz. Polarization X.</figcaption></figure>

<figure><img src="image_12.png" /><figcaption>Fig. 8. Analysis of the illumination (εᵢ), spillover (εₛ), and aperture (εₐₚ) efficiencies as a function of f/D ratio.</figcaption></figure>

The S₁₁ of the multifaceted structure was computed using this model and its value at 30 GHz is roughly -9 dB. It is

relatively higher due to the center-fed configuration and the simple hyperbolic profile of the subreflector. This Cassegrain feed could be enhanced using other subreflector approaches such as ring-focus subreflectors [26].

Table II and Fig. 7 provide the simulated copolarized gain pattern of the antenna in the frequency band of interest, ranging from 27 to 33 GHz. At the central frequency, the Cassegrain approach generates a directive pencil beam pattern with a gain of about 31 dBi, a half-power beamwidth (HPBW) of 3°, and a sidelobe level (SLL) lower than -13 dB. At off-center frequencies, the MFRA maintains the directivity of the beam along the E-plane (φ = 0°) and in the D-plane (φ = 45°) as depicted in Fig. 7. Conversely, the radiation pattern in the H-plane (φ = 90°) suffers from evident defocusing of the beam (see Fig. 7), which mostly increases the HPBW up to 9.5° at 27 GHz. This results in a gain reduction of 4.2 and 4.5 dB for the X- and Y-polarizations, respectively, in the band under study.

Table II also provides the total aperture efficiency of the antenna εₐₚ, calculated considering the spillover εₛ and the illumination effects εᵢ in the reflectarray surface [48]. The aperture efficiency is roughly 60% in the bandwidth under study. To further analyze this parameter, Fig. 8 shows the behavior of the aperture, illumination, and spillover efficiencies varying the f/D ratio of the structure but maintaining the relative position between feed and subreflector. These results shown that the aperture efficiencies simulated are the maximum achieved using the feeding structure detailed in this
<table><caption>TABLE III
MEASURED RF PERFORMANCE OF THE CASSEGRAIN SYSTEM</caption><thead><tr><th rowspan="2">f [GHz]</th><th rowspan="2">Gain [dBi]</th><th colspan="3">SLL [dB]</th><th colspan="3">HPBW [°]</th><th rowspan="2">CP/XP [dB]</th></tr><tr><th>E-Plane</th><th>H-Plane</th><th>D-Plane</th><th>E-Plane</th><th>H-Plane</th><th>D-Plane</th></tr></thead><tbody><tr><td>27.0</td><td>27.0 / 27.0</td><td>-11.7 / -14.9</td><td>-19.4 / -19.0</td><td>-21.7 / -17.4</td><td>3.0 / 3.6</td><td>4.9 / 3.6</td><td>2.9 / 2.6</td><td>28.4 / 24.8</td></tr><tr><td>28.5</td><td>31.8 / 30.7</td><td>-15.2 / -15.9</td><td>-20.4 / -24.2</td><td>-20.3 / -22.6</td><td>3.2 / 3.8</td><td>3.6 / 3.0</td><td>2.9 / 2.7</td><td>26.3 / 24.5</td></tr><tr><td>30.0</td><td>30.9 / 30.5</td><td>-11.9 / -15.5</td><td>-18.0 / -13.3</td><td>-17.9 / -18.7</td><td>3.0 / 3.2</td><td>3.3 / 2.0</td><td>2.8 / 2.0</td><td>25.6 / 26.3</td></tr><tr><td>31.5</td><td>29.2 / 27.7</td><td>-11.0 / -15.3</td><td>-15.1 / -13.3</td><td>-16.5 / -14.3</td><td>3.2 / 3.1</td><td>3.5 / 2.0</td><td>3.0 / 1.9</td><td>18.5 / 19.9</td></tr><tr><td>33.0</td><td>25.9 / 26.6</td><td>-13.8 / -10.5</td><td>-10.0 / -14.4</td><td>-14.9 / -10.9</td><td>3.0 / 2.7</td><td>4.4 / 2.8</td><td>3.1 / 2.1</td><td>22.3 / 25.1</td></tr></tbody></table>

Blue data corresponds to Pol. X and red one with Pol. Y.

<figure><img src="image_13.png" /><figcaption>Fig. 9. Measurement setup used in the experimental validation. Detail of the manufactured Cassegrain system and the feeding structure (feed and subreflector).</figcaption></figure>

antenna design. The discrepancies between polarizations in $\epsilon_{ap}$ are due to different illumination of the multifaceted surface for each polarization, as illustrated in Fig. 4.

## B. Experimental Validation

The proposed MFRA design has been manufactured and tested at the Institut d'Électronique et Technologies du numéRique (IETR) in collaboration with the University of Oviedo. The antenna prototype (shown in Fig. 8) consists of a metallic base that supports different elements of the antenna system (reflectarray panels, primary feed, and subreflector) according to the optics shown in Fig. 1. The Cassegrain feed is placed at the center of the main reflector above a pedestal-shaped piece, shown in Fig. 9, which allows for the adjustment of the height of the horn antenna and its excitation from the back of the structure. The metallic subreflector is embedded in a foam piece (white block in Fig. 9), which is inserted into the primary feed to ensure it is positioned at the correct distance above the feeder. This foam piece has a negligible effect in the electrical performance of the antenna, since its permittivity is close to 1 (roughly 1.05). Both the primary feed and the foam piece have been rotated along the z-axis to provide both the linear polarizations of the antenna during the measurements' campaign.

The Cassegrain demonstrator is measured in the spherical acquisition range in the anechoic chamber at the University of Oviedo. The measured copolar and cross-polar far fields of the demonstrator at 30 GHz for each polarization are presented in Figs. 10 and 11. They are compared with the simulation results discussed in Section III-A. In addition, Table III lists the main parameters of the pattern within the bandwidth under study. The measured results exhibit overall good agreement with the simulations. At the central frequency, the MFRA achieves a pencil beam pattern with an HPBW of roughly 3°.

As predicted by the simulations, this parameter is more stable in the E- and D-planes compared with the H-plane. This effect is also evident in the defocusing of the beam at off-center frequencies (see Figs. 10 and 11) and is further explained in Section IV.

Table III also provides the SLL of the antenna at different frequencies. The MFRA exhibits a maximum SLL of -10.5 dB, which is in accordance with the predicted results. The higher side radiation may be the result of the effect of the Cassegrain feed and the field contributions generated by each panel at directions of space different from broadside.

The bottom rows in Figs. 10 and 11 depict the cross-polar field in the three main planes (E-, H-, and D-planes), and Table III provides the co/cross-polar ratio in the direction of maximum radiation. In both the polarizations, the MFRA exhibits good polarization purity, as the cross-polar values remain roughly 20 dB lower than the copolar field.

The measured peak gain at 30 GHz was 30.9 and 30.5 dBi for the X- and Y-polarizations, respectively. These gain values correspond to a radiation efficiency $\epsilon_{rad}$ of 24% and 22%, respectively. This latter parameter has been computed by the ratio between the maximum directivity ($D_{max}$) considering a uniform illumination of the multifaceted aperture [48] and the gain measured at the design frequency $G_0$

$$ \epsilon_{rad} = \frac{G_0}{D_{max}}. \qquad (2) $$

Table IV lists the gain budget analysis of the MFRA. The main sources of loss are the illumination and the blockage loss generated by the Cassegrain feed over the field radiated by the main reflector. The former is the result of the nonunifomity of the field that impinges on the reflectarray surface (see Fig. 4). The latter is due to the center-fed configuration and broadside radiation of the structure [48]. Several strategies can help mitigate these sources of loss, including the use of a lower gain primary feed, optical adjustment of the feeding network,
<figure><img src="image_14.png" /><figcaption>Fig. 10. Normalized radiation patterns of the Cassegrain system measured (solid lines) and simulated (dashed lines) for X-polarization. Copolar (top) and cross-polar (bottom) field values at (a) E-plane ($\varphi = 0^{\circ}$), (b) H-plane ($\varphi = 90^{\circ}$), and (c) D-plane ($\varphi = 45^{\circ}$).</figcaption></figure>

implementation of unit-cells that cover the full phase cycle [18], or the use of ring focus subreflectors [26], among others.

<table><caption>TABLE IV
GAIN BUDGET ANALYSIS OF THE MFRA AT 30 GHz</caption><thead><tr><th></th><th>X-polarization</th><th>Y-polarization</th></tr></thead><tbody><tr><td>Directivity [dB]</td><td>37.10</td><td>37.10</td></tr><tr><td>Illumination loss [dB]</td><td>1.75</td><td>1.93</td></tr><tr><td>Spillover loss [dB]</td><td>0.54</td><td>0.54</td></tr><tr><td>Phase losses [dB]</td><td>0.76</td><td>0.76</td></tr><tr><td>Material loss [dB]</td><td>0.10</td><td>0.10</td></tr><tr><td>Cross-polar loss [dB]</td><td>0.02</td><td>0.02</td></tr><tr><td>Feed mismatch [dB]</td><td>0.46</td><td>0.76</td></tr><tr><td>Blockage loss [dB]</td><td>2.24</td><td>1.48</td></tr><tr><td>Total loss [dB]</td><td>5.87</td><td>5.59</td></tr><tr><td>Gain predicted [dB]</td><td><strong>31.23</strong></td><td><strong>31.91</strong></td></tr><tr><td>Gain simulated (CST) [dB]</td><td>31.10</td><td>31.66</td></tr><tr><td>Gain measured [dB]</td><td>30.90</td><td>30.50</td></tr></tbody></table>

The gain obtained in this budget is roughly 31 dBi, which is slightly higher than the one measured experimentally. This could be attributed to sources of loss not considered in this analysis such as ohmic losses, feed insertion losses, and mechanical misalignments in the demonstrator.

Fig. 12 shows the peak gain values measured and predicted by the full-wave simulation detailed in Section III-A. Both the results exhibit similar values at central frequency and an evident loss of gain at off-center frequencies. Discrepancies between the predicted and measured gains can be attributed to uncertainties in the measurement setup and mechanical tolerances of the demonstrator not considered in the simulation. Regarding the latter point, the model presented in Fig. 6 has been resimulated for the X-polarization, incorporating a 0.4° rotation of the Cassegrain feed relative to the X-axis in the coordinate system of Fig. 2. The results, shown as green lines in Fig. 12, exhibit better agreement with the measurements (blue line) compared with the simulated gain with perfect alignment (red line). A similar analysis was conducted for the Y-polarization, with the measurements (dashed blue line), the simulation with perfect alignment (red dashed line), and the simulation including the rotation (dashed green line) also represented in Fig. 12. This analysis demonstrates that the gain response of the antenna is affected by mechanical tolerances in the position and orientation of the Cassegrain feed.

## IV. COMPARISON TO AN EQUIVALENT SINGLE-FACET APERTURE AND WITH OTHER WORKS

To further investigate the features of the proposed MFRA, an equivalent SFRA has also been designed, manufactured, and tested. It follows the design process described in Section II, and the experimental results have been obtained using an equivalent measurement setup to the one presented in Section III.
<figure><img src="image_15.png" /><figcaption>Fig. 11. Normalized radiation patterns of the Cassegrain system measured (solid lines) and simulated (dashed lines) for Y-polarization. In rows, copolar (top) and cross-polar (bottom) field values at (a) E-plane ($\varphi = 0^{\circ}$), (b) H-plane ($\varphi = 90^{\circ}$), and (c) D-plane ($\varphi = 45^{\circ}$).</figcaption></figure>

<table><caption>TABLE V
COMPARISON OF THE PROPOSED CASSEGAIN SYSTEM TO OTHER HIGH-GAIN REFLECTARRAY ANTENNAS</caption><thead><tr><th></th><th>[20]</th><th>[5]</th><th>[33]</th><th>[34]</th><th>[35]</th><th>[27]</th><th>[28]</th><th>[29]**</th><th>Proposed antenna</th></tr></thead><tbody><tr><td>Freq. (f<sub>0</sub>) [GHz]</td><td>32</td><td>8.4</td><td>61</td><td>44</td><td>400</td><td>94.5</td><td>12</td><td>77</td><td>30</td></tr><tr><td>Configuration</td><td>Centered fed</td><td>Single offset</td><td>Folded</td><td>Folded</td><td>Folded</td><td>Cassegrain</td><td>Cassegrain + TA</td><td>Cassegrain</td><td>Cassegrain</td></tr><tr><td>Operation</td><td>Sing. CP</td><td>Sing. CP</td><td>Sing. LP</td><td>Sing. LP</td><td>Sing. LP</td><td>Dual LP</td><td>Sing. LP</td><td>Sing. LP</td><td>Dual LP</td></tr><tr><td>Aperture (D) [λ<sub>0</sub>]</td><td>53 (circ.)</td><td>20 x 20 (rect)</td><td>20 (circ.)</td><td>22 (circ.)</td><td>26.5 x 26.5 (rect.)</td><td>22 x 22 (rect.)</td><td>6 x 6 (rect.)</td><td>39 x 39 (rect.)</td><td>20 x 20 (rect.)</td></tr><tr><td>Unit cell</td><td>1-layer patch rotated</td><td>1-layer dipoles</td><td>1-layer patch</td><td>2-layer patch</td><td>Cross &amp; coupled patches</td><td>2-layer patch</td><td>1-layer patch</td><td>1-layer patch</td><td>1-layer patch</td></tr><tr><td>Sub reflector</td><td>-</td><td>-</td><td>Strip-grid polarizer</td><td>Strip-grid polarizer</td><td>Strip-grid polarizer</td><td>Ring-focus</td><td>Ring-focus/TA</td><td>Hyp.</td><td>Hyp.</td></tr><tr><td>Compactness (h<sub>s</sub>/D)</td><td>0.75</td><td>0.93*</td><td>0.25</td><td>0.25</td><td>0.25</td><td>0.24*</td><td>0.30*</td><td>0.30*</td><td>0.35</td></tr><tr><td>Blockage (d<sub>s</sub>/D)</td><td>-</td><td>-</td><td>-</td><td>-</td><td>-</td><td>0.25*</td><td>0.24*</td><td>0.11*</td><td>0.15</td></tr><tr><td>SLL (f<sub>0</sub>) [dB]</td><td>&lt;-25*</td><td>&lt;-19.9</td><td>&lt;-24</td><td>&lt;-17.5</td><td>&lt;-17.8</td><td>&lt;-13</td><td>&lt;-12</td><td>&lt;-19</td><td>&lt;-12</td></tr><tr><td>Cross-polarization discrimination (f<sub>0</sub>) [dB]</td><td>&gt;35*</td><td>&gt;20</td><td>-</td><td>&gt;25</td><td>&gt;28</td><td>&gt;28</td><td>&gt;19</td><td>-</td><td>&gt;26</td></tr><tr><td>G (f<sub>0</sub>) [dBi]</td><td>41.5*</td><td>29.2</td><td>34.0</td><td>31.0*</td><td>32.8</td><td>31.9 (X)<br>30.9 (Y)</td><td>22.0*</td><td>36.9</td><td>30.9 (X)<br>30.5 (Y)</td></tr><tr><td>Rad. efficiency (f<sub>0</sub>) (%)</td><td>35.6*</td><td>41.6</td><td>62.0*</td><td>28.0*</td><td>29.3*</td><td>31.4 (X)<br>25.1 (Y)</td><td>35.3*</td><td>25.7*</td><td>24.1 (X)<br>22.0 (Y)</td></tr><tr><td>1 dB BW [%f<sub>0</sub>]</td><td>4.8*</td><td>-</td><td>7.0*</td><td>5.0*</td><td>4.6*</td><td>2.3 (X)<br>2.3 (Y)</td><td>6.7*</td><td>2.5</td><td>10.0 (X)<br>7.3 (Y)</td></tr><tr><td>3 dB BW [%f<sub>0</sub>]</td><td>5.8*</td><td>-</td><td>11.5*</td><td>8.0*</td><td>16.0</td><td>-</td><td>11.5*</td><td>-</td><td>19.0 (X)<br>17.0 (Y)</td></tr></tbody></table>

LP: linear polarization; CP: circular polarization; f₀: designed frequency; TA: transmitarray antenna; *Estimated from the data provided in the paper;

**The data listed from [29] corresponds to the reflectarray example IV.D: “6”-Square Cassegrain Reflectarray at 77 GHz”

Fig. 13 shows the layout of the SFRA. It exhibits several phase wraps (i.e., abrupt variations in the size of contiguous patches) in both the E- and H-planes. These phase wraps are due to two main factors: on one hand, the use of a unit-cell with a limited phase range; on the other hand, the compactness

of the structure (i.e., small *f*/*D* ratio), which demands several phase jumps in the reflectarray surface, in contrast to other reflectarray configurations, such as [42].

These issues introduce a phase error along the reflectarray surface both at the central frequency and across the band
<figure><img src="image_16.png" /><figcaption>Fig. 12. Predicted gain over frequency of the multifaceted antenna, with and without considering the mechanical tolerances. Comparison to the measured peak gain.</figcaption></figure>

<figure><img src="image_17.png" /><figcaption>Fig. 13. Layout of the equivalent single-facet aperture designed.</figcaption></figure>

of interest. Fig. 14 depicts the phase excursion of both the reflectarray designs at different frequencies. At the central one, both the antennas exhibit phase excursion due to the limited phase range of the unit-cell. However, the MFRA exhibits lower phase excursion than the SFRA along the E-plane (XZ plane in Fig. 13), and this effect also appears at off-center frequencies. This feature is due to the optics of the antenna system which mitigates the phase jumps required along this plane (see Fig. 5) and, therefore, the phase wraps (see Fig. 6). In the H-plane, both MFRA and SFRA exhibit a notable phase excursion.

The SFRA achieves similar RF performance than the proposed MFRA at the central frequency. It exhibits a beam of roughly 3° × 3°, SLLs lower than -12 dB, and a peak gain of 30 dB. Nevertheless, MFRA and SFRA exhibit some differences in their in-band performance. Fig. 15 shows the radiation pattern of both the antennas measured at off-center frequencies. In the E-plane, in which the MFRA exhibits less

<figure><img src="image_18.png" /><figcaption>(a)</figcaption></figure>

<figure><img src="image_19.png" /><figcaption>(b)</figcaption></figure>

<figure><img src="image_20.png" /><figcaption>(c)</figcaption></figure>

<caption>Fig. 14. Phase difference in degrees (°) between the required phase and the one provided by the layout for the SFRA (left) and the MFRA (right) at (a) 27 GHz, (b) 30 GHz, and (c) 33 GHz. Phase distribution to operate at the X-polarization.</caption>

phase errors, the MFRA achieves a more stable beamwidth and lower side radiation compared with the SFRA. In the H-plane, both the antennas show evident beam defocusing and high side radiation levels. According to the former parameter, the HPBW of the reflectarray designs has been analyzed in a wide frequency range (Fig. 16). In the bandwidth under study, the MFRA exhibits a variation in the HBPW in the E-plane (see the blue dotted lines of Fig. 16) of roughly 0.5° and 1.7° for THE X- and Y-polarizations respectively. Under the same conditions, the HPBW variation in the SFRA is 3.0° for the X-polarization and 7.2° for the Y-polarization. Therefore, the proposed MFRA exhibits at least four times lower in-band beamwidth variation than its equivalent SFRA.

Regarding the antenna gain response in-band, Fig. 17 depicts the peak gain measured and predicted in the band of interest (26–34 GHz) for the SFRA and the MFRA. Both the antenna systems feature similar gain within the designed frequency range (between 28 and 32 GHz) but the MFRA features higher gain than its single-faceted approach at the
<figure><img src="image_21.png" /><figcaption>Fig. 15. Normalized radiation pattern of the Cassegrain system measured using the MFRA and the equivalent single-facet (SFRA) reflectarray for X-polarization. (a) H-plane ($\varphi = 0^\circ$). (b) H-plane ($\varphi = 90^\circ$).</figcaption></figure>

<figure><img src="image_22.png" /><figcaption>Fig. 16. Measured HPBW over frequency, for the proposed Cassegrain MFRA and its equivalent SFRA. (a) X-polarization. (b) Y-polarization.</figcaption></figure>

<figure><img src="image_23.png" /><figcaption>Fig. 17. Measured peak gain over frequency of the multifaceted and its equivalent single facet. Solid and dotted lines correspond to multifaceted and single-facet results, respectively.</figcaption></figure>

extreme frequencies (26 and 34 GHz). The SFRA features almost the same performance in both the polarizations due

to the symmetry between the x- and y-axes (see Fig. 13). Conversely, the gain of the MFRA shows differences between polarizations due to the lack of symmetry between axes (note that the panels of the MFRA have a tilt along the XZ plane).

The 1-and 3-dB gain bandwidths, relative to the gain at the central frequency, have been computed for both the designs. For the MFRA, the 1-dB bandwidths were 10% and 7%, and the 3-dB bandwidths were 19% and 17% for the X- and Y-polarizations, respectively. The SFRA achieved same bandwidths in both the polarizations, with 10% and 17% for the 1-and 3-dB gain bandwidths, respectively. This similarity in gain bandwidth is primarily due to the defocusing of the beam in the H-plane, which appears in both the antenna designs and can be observed in the variation of the HPBW (red solid and dotted lines) in Fig. 16.

Table V shows comparison of the proposed multifaceted antenna with other high-gain reflectarray designs in the literature. The MFRA exhibits higher compactness than conventional reflectarray configurations, such as center-fed [19], or single-offset [5]. Moreover, it demonstrates a dual-polarization operability, while other compact solutions such as folded reflectarrays [32], [33], [34] can only provide single-polarization. In any case, the multifaceted demonstrator achieves higher 1-and 3-dB relative bandwidth than the works listed in Table V and maintains similar compactness, profile, blockage, and side radiation as other reflectarray designs based on Cassegrain optics [26], [27], [28].

V. CONCLUSION

This article has presented the analysis and design of a compact antenna configuration, based on a multifaceted aperture with parabolic profile. The antenna is based on a Cassegrain system, where the main reflector consists of multiple reflectarray panels. Each panel is a low-profile, single substrate with different angular orientation, so that the structure conforms to a cylindrical paraboloid reflector. A prototype has been successfully manufactured and experimentally tested.
The measured results agreed well with simulations, showing that the multifaceted prototype achieves satisfactory performance at the design frequency. It exhibits high-gain pencil beam, with low side radiation, and high cross-polar isolation. Furthermore, within the band of interest, these features are stable along the E-plane, where the panels are tilted.

The features of the proposed multifaceted structure have been compared with those of an equivalent Cassegrain system based on a single-faceted reflectarray. The results of this study indicate that the planar arrangement and the use of a simple unit-cell element introduce phase errors, especially at off-center frequencies. This produces an in-band degradation of the pattern in terms of beam defocusing and high side radiation. The multifacet optics partially mitigates the phase range required on the reflectarray surface, reducing the in-band phase excursion along the sectorization plane. This results in an enhancement in the far-field of the antenna along this plane, specifically in the stability of the pencil beam pattern. Indeed, the multifaceted approach demonstrates a clear improvement in the in-band stability of the beam pattern compared with its equivalent single-facet design. However, this enhancement is limited in terms of in-band gain, as the multifaceted approach compensates for the phase excursion along only one plane of the antenna structure.

The presented results demonstrate the feasibility of using multifaceted apertures together with dual-reflector configurations. Such configurations exhibit good in-band performance while maintaining low-profile and high compactness. These features make them suitable for storage and deployment in small platforms, such as those used in various space missions. Future work will focus on applications of advanced multifaceted structures, which compensate for phase excursion in several planes simultaneously.

## ACKNOWLEDGMENT

The authors are grateful to Laurent Cronier for his support in the fabrication and assembly of the antenna prototype.

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[37] M. Zhou, S. B. Sørensen, O. Borries, and E. Jørgensen, "Analysis and optimization of a curved transmit-receive contoured beam reflectarray," in *Proc. 9th Eur. Conf. Antennas Propag. (EuCAP)*, Lisbon, Portugal, Apr. 2015, pp. 1-5.

[38] A. Roederer, *Reflector antenna comprising a plurality of panels*, U.S. Patent 6 411 255 B2, Mar. 6, 2001.

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<figure><img src="image_24.png" /></figure>

**Borja Imaz-Lueje** (Member, IEEE) was born in La Hueria, Spain, in 1995. He received the B.Sc., M.Sc., and Ph.D. degrees in telecommunication engineering from the Universidad de Oviedo (UO), Gijón, Spain, in 2017, 2019, and 2023, respectively.

Since 2016, he has been a Research Assistant with the Electrical Engineering Department, UO. From 2018 to 2019, he spent several months in Rohde & Schwarz GmbH & Co. KG, Munich, Germany, where he was involved in the deployment of CATR systems and OTA measurements. In 2022, he was

in a research mission with the Institut d'Electronique et des Technologies du numérique (IETR), Rennes, France, where he was involved in the analysis and design of compact antennas. In 2023, he joined the Department of Signal Theory and Communications and Telematic Systems and Computing, Universidad Rey Juan Carlos (URJC), Madrid, Spain, where he is currently an Assistant Professor. His current research interests include development of efficient analysis and design techniques of reflectarrays in complex configurations, working in environments of near- and far-fields.

Dr. Imaz-Lueje was a recipient of a Pre-Doctoral Fellowship supported by the Spanish Government in 2019. He has also received the 2023 National Award for the Best Ph.D. Thesis on Telecommunication Engineering for Satellite Communications and the 2024 Outstanding Ph.D. Thesis Award from UO. He was also a recipient of IEEE Antennas and Propagation Society Fellowship in 2024.

<figure><img src="image_25.png" /></figure>

**David González-Ovejero** (Senior Member, IEEE) received the M.Sc. degree in telecommunication engineering from the Universidad Politécnica de València, Valencia, Spain, in 2005, the Ph.D. degree in electrical engineering from the Université catholique de Louvain, Louvain-la-Neuve, Belgium, in 2012, and the Habilitation à Diriger des Recherches degree from the Université de Rennes, Rennes, France, in 2024.

From 2012 to 2014, he was a Research Associate at the University of Siena, Siena, Italy. In 2014,

he joined the Jet Propulsion Laboratory, California Institute of Technology, Pasadena, CA, USA, where he was a Marie Curie Post-Doctoral Fellow. Since 2016, he has been a Tenured Researcher at the French National Center for Scientific Research (CNRS), Institut d'Electronique et des Technologies du numérique (IETR), Rennes. His current research interests include computational electromagnetics, large phased arrays, periodic structures, metasurfaces, and submillimeter-wave antennas.

Dr. González-Ovejero was awarded a Marie Curie International Outgoing Fellowship from the European Commission in 2013, the Sergei A. Schelkunoff Transactions Prize Paper Award from IEEE Antennas and Propagation Society in 2016, the Best Paper Award in Antenna Design and Applications at the 11th European Conference on Antennas and Propagation in 2017, the Best Paper Award in Electromagnetics at the 15th European Conference on Antennas and Propagation in 2021, and the Best Paper Award from the International Workshop on Antenna Technology (iWAT) in 2023. Since 2019, he has served as an Associate Editor for IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION and IEEE TRANSACTIONS ON TERAHERTZ SCIENCE AND TECHNOLOGY.

<figure><img src="image_26.png" /></figure>

**Marcos R. Pino** (Member, IEEE) received the M.Sc. and Ph.D. degrees in telecommunication engineering from the University of Vigo, Vigo, Spain, in 1997 and 2000, respectively.

In 1998, he was a Visiting Scholar with the ElectroScience Laboratory, The Ohio State University, Columbus, OH, USA. From 2000 to 2001, he was an Assistant Professor with the University of Vigo. Since 2001, he has been with the Electrical Engineering Department, University of Oviedo, Gijón, Spain, where he is currently a Full Professor,

teaching courses on communication systems and antenna design. From 2017 to 2023, he has spent several months as a Visiting Fellow with the Department of Information Engineering, University of Pisa, Pisa, Italy, collaborating in near-field UHF-RFID applications.

His current research interests include antenna design optimized for both near- and far-field applications, antenna measurement techniques, and efficient computational techniques applied to EM problems.
<figure><img src="image_27.png" /></figure>

**Manuel Arrebola** (Senior Member, IEEE) was born in Lucena, Spain. He received the M.Sc. degree in telecommunication engineering from the University of Málaga, Málaga, Spain, in 2002, and the Ph.D. degree from the Technical University of Madrid (UPM), Madrid, Spain, in 2008.

From 2003 to 2007, he was a Research Assistant with the Department of Electromagnetism and Circuit Theory, UPM. From 2007 to 2024, he was with the Department of Electrical Engineering, University of Oviedo, Gijón, Spain, where he became an Associate Professor. In 2024, he joined the Department of Signals, Systems and Radiocommunications, UPM, as an Associate Professor with the Applied Electromagnetics Group. He was a Visiting Researcher with the Department of Microwave Techniques, Universität Ulm, Ulm, Germany, in 2005; and with the European Space Research and Technology Center, European Space Agency, Noordwijk, The Netherlands, in 2009. He has been a Visiting Professor with several institutions, such as the Edward S. Rogers Sr. Department of Electrical and Computer Engineering, University of Toronto, Toronto, ON, Canada, in 2018; the Institute of Sensors, Signals and Systems, Heriot-Watt University, Edinburgh, U.K., in 2019; and the Antennas and Propagation Group, Instituto de Telecomunicações, Lisbon, Portugal, in 2023. He has authored and co-authored more than 250 peer-reviewed journals and conference papers. His current research interests include application of innovative manufacturing techniques in mm and sub-mm-wave antenna design and development of efficient analysis, design, and optimization techniques of spatial fed arrays, including reflectarrays and transmitarrays, and periodic structures for near- and far-field applications.

Dr. Arrebola was a co-recipient of the 2007 Sergei A. Schelkunoff Transactions Prize Paper Award from the IEEE Antennas and Propagation Society. He has also received the 2008 Outstanding Ph.D. Thesis Award from UPM and the 2009 National Award for the Best Ph.D. Thesis on Telecommunication Networks. Since 2022, he has served as an Associate Editor of IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION.

<figure><img src="image_28.png" /></figure>

**Ronan Sauleau** (Fellow, IEEE) received the M.Sc. degree in electrical engineering and radio communications from the Institut national des sciences appliquées de Rennes, Rennes, France, in 1995, the Agrégation degree from the École normale supérieure Paris-Saclay, Cachan, France, in 1996, and the Ph.D. degree in signal processing and telecommunications and the Habilitation à Diriger des Recherches degree from the University of Rennes 1, Rennes, in 1999 and 2005, respectively.

He was an Assistant Professor and Associate Professor at the University of Rennes 1, from September 2000 to November 2005, and from December 2005 to October 2009, respectively. He has been appointed as a Full Professor with the University of Rennes 1, since November 2009. He has been involved in more than 80 research projects at the national and European levels and has co-supervised 32 post-doctoral fellows, 65 Ph.D. students, and 50 master's students. He has received 25 patents and is the author or co-author of 310 journal articles and more than 600 publications in international conferences and workshops. He was the Co-Director of the research Department “Antenna and Microwave Devices” at IETR and the Deputy Director of IETR between 2012 and 2016. He is now the Director of IETR. His current research fields are millimeter-wave beam-steering antennas, substrate-integrated waveguide antennas, lens-based focusing devices, and periodic and nonperiodic structures (FSS, metasurfaces, polarizers, reflectarrays, and transmitarrays).

Prof. Sauleau received the 2004 ISAP Conference Young Researcher Scientist Fellowship in Japan and the first Young Researcher Prize in Brittany, France, in 2001, for his research work on gain-enhanced Fabry-Perot antennas. In September 2007, he was elevated to the Junior Member of the “Institut Universitaire de France.” He was awarded the Bronze Medal by CNRS in 2008 and the Silver Medal in 2020. He received the 2021 Antenna EurAAP Award. He was a co-recipient of several international conference awards with some of his students (International School of BioEM in 2005, BEMS in 2006, MRRS in 2008, E-MRS and BEMS in 2011, IMS and Antem in 2012, BioEM in 2015, EuCAP in 2019, EuCAP in 2021, and EuMW in 2022). He served as the Guest Editor for IEEE ANTENNAS AND PROPAGATION Special Issue on “Antennas and Propagation at mm and sub mm waves.” He served as a National Delegate for several EU COST actions. He has served as a National Delegate for EurAAP and as a member of the Board of Director of EurAAP from 2013 to 2018.