# Single-phase single-stage bidirectional PFC converter utilizing monolithic GaN bidirectional switches

Jan Sitnik, Matthias Kasper

► To cite this version:

Jan Sitnik, Matthias Kasper. Single-phase single-stage bidirectional PFC converter utilizing monolithic GaN bidirectional switches. The 26th European Conference on Power Electronics and Applications, GDR SEEDS France & EPE Association, Mar 2025, Paris, France. 10.34746/epe2025-0267. hal-05085166

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Submitted on 26 May 2025

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<figure><img src="image_1.png" /></figure>

Distributed under a Creative Commons Attribution 4.0 International License
Single-phase single-stage bidirectional PFC converter utilizing monolithic
GaN bidirectional switches

Jan Sitnik

Institute of Control and Industrial Electronics
Warsaw University of Technology
Warsaw, Poland

Matthias Kasper

Power and Sensor Systems
Infineon Technologies Austria AG
Villach, Austria

Acknowledgements

The authors would like to express their sincere
gratitude to Martin Wattenberg and Kennith Kin
Leong, employees of Infineon Technologies
Austria AG, for their invaluable support during
the work on the topics presented in this article.

Keywords

Bidirectional converters, On-board charger, V2G,
LLC resonant converter, Gallium Nitride (GaN).

Abstract

This article presents a comprehensive study on a bidirectional, single-phase, single-stage converter for Electric Vehicle (EV) charging applications. A converter topology providing galvanic insulation between the grid and the battery with Power Factor Correction (PFC) capabilities has been proposed. By utilizing new GaN transistors with bidirectional voltage blocking capabilities, the proposed converter topology allows for a reduction in the number of transistors, leading to a significant simplification of the converter in comparison to other state-of-the-art solutions. The system is successfully validated in a simulation study.

Introduction

The dynamic advancements in electromobility is
shaping the direction of future power electronics,
particularly in the area of On-Board Chargers
(OBC), which are also often required to operate
in Vehicle-to-Grid (V2G) mode [2]. The widely

sought-for features of modern OBCs include,
among others, high power density, limited
weight, and increased efficiency [1]. To achieve
these goals, in this paper, a new converter
topology has been developed, in which
employing transistors with bidirectional voltage
blocking capabilities allows for a reduction in the
number of transistors by over 30% compared to
the most common solutions.

Nowadays, the vast majority of OBCs are
implemented using a two-stage topology. This
concept involves dividing the entire system into
two independent sections, which are connected by
a DC-link circuit [3]. The first part is the PFC
stage, responsible for interfacing with the grid,
while the second part is the DC-DC converter
used for battery charging. The schematic diagram
of such a solution is presented in Figure 1.

A concrete example of this approach, as
illustrated in Figure 2, is the application of a full
bridge in conjunction with an LLC resonant
converter [5]. Although this solution is effective,
it has a notable disadvantage due to the relatively
high number of transistors used which leads to
increased volume and limited efficiency.

<figure><img src="image_2.png" /><figcaption>Fig. 1: Two-stage converter concept diagram [4].</figcaption></figure>
<figure><img src="image_3.png" /><figcaption>Fig. 2: Conventional two-stage, bidirectional converter realization with a full-bridge topology at AC-stage and LLC resonant at DC-stage.</figcaption></figure>

Consequently it has led to a search for innovative solutions that utilize fewer transistors. Therefore, various single-stage solutions have been presented in the literature. Commonly seen are matrix converters in various configurations. These can function as single-stage by configuring the primary-side transistors in either a common drain or common source arrangement [6], [7]. While those topologies are indeed single-stage, the number of utilized transistors remains high, at a comparable level to the two-stage topologies [8], [9], [10], as there is a need for blocking the voltage in both directions at the converter's AC-side. An exemplary schematic of a bidirectional matrix converter is depicted in Figure 3.

A straightforward solution for this requirement is to employ the recently introduced monolithic bidirectional switch (BDS) based on Infineon CoolGaN™ Transistor technology that allows for bidirectional current flow and blocking voltage in both directions [11], [12]. Moreover, the use of a common drain configuration enables independent gate control, which is essential for the proposed topology. Furthermore, this approach significantly reduces the parasitic inductance and minimizes the required PCB area, leading to better switching performance.

The proposed topology offers numerous advantages compared to the previously discussed solutions. Primarily, it facilitates control of the battery voltage and current during bidirectional energy transfer, while reducing the number of components required in the design. Additionally, it operates in PFC mode and ensures galvanic insulation between the battery and the grid, a critical design requirements for OBCs. Moreover, the integration of advanced GaN technology allows for higher switching frequencies, leading to a reduction in the size of passive components.

The paper is constructed as follows. After the first section briefly introducing the concept, the second section provides an in-depth analysis of

<figure><img src="image_4.png" /><figcaption>Fig. 3: Bidirectional matrix converter schematic</figcaption></figure>

<figure><img src="image_5.png" /><figcaption>Fig. 4: The proposed single-stage, bidirectional converter topology utilizing BDS.</figcaption></figure>

the proposed converter topology, including the control system. Furthermore, in the next section, a comprehensive simulation study is presented, validating the proposed concept. Finally, the paper is concluded in the last section.

## Topology overview

The proposed single-phase single-stage bidirectional PFC converter topology utilizing monolithic GaN bidirectional switches facilitates direct, single-stage energy conversion between an AC voltage source and a battery. The proposed topology, depicted in Fig. 4, integrates a full bridge on the primary side, employing monolithic bidirectional switches as the power electronic devices, along with an LLC resonant circuit. Additionally, it incorporates a full bridge on the secondary side utilizing standard discrete GaN transistors.

The converter control ensuring proper operation of the converter in all modes can be split into two distinct sections. The first section focuses on the control of the primary side. This involves the sequential switching of the switches at a fixed duty cycle of 50% and a constant frequency, while accounting for the polarity of the grid voltage. Figure 5 presents the control schematic for the primary side.

Through this switching approach, the voltage applied to the LLC resonant circuit approximates a rectangular waveform from the switching frequency perspective, while its amplitude
<figure><img src="image_6.png" /><figcaption>Fig. 5: Control scheme of the primary-side.</figcaption></figure>

<figure><img src="image_7.png" /><figcaption>Fig. 6: Voltage generated during the switching of the secondary side (a), Close-up view of the voltage generated during the switching of the secondary side (b)</figcaption></figure>

directly correlates to the instantaneous grid voltage value as illustrated in Figures 6a and 6b. The second section focuses on the control of the secondary side, which regulates all parameters of the converter through a variable duty cycle. This control strategy is structured around a regulation loop and a modulator (Figure 7). The control loop consists of two PI controllers, implementing a standard Power Factor Correction (PFC) control that tracks the waveform of the grid voltage, along with appropriate scaling to achieve the desired duty cycle. The secondary side modulator dictates the operation of the switches at a frequency that is twice that of the primary side. The exemplary PWM signals of the primary side and secondary side at 60% duty cycle are presented in Figure 8.

<figure><img src="image_8.png" /><figcaption>Fig. 8: Primary and secondary side PWM signals.</figcaption></figure>

The proposed converter consists of an LLC resonant circuit to provide soft-switching capabilities and increased efficiency. The resonant tank is designed to operate at a frequency of 400 kHz with a maximum power of 500 W. It should be noted that when operating at a constant frequency, which corresponds to the resonant frequency of the LLC circuit, the gain will always be 1. Figure 9 shows the schematic of the resonant circuit, which consists of a capacitor, two inductors, and an ideal transformer.

The selection of component values in the resonant circuit is not focused on fine-tuning the gain-frequency response with respect to the system's quality factor, but rather on reducing the size of the inductors while ensuring a reasonable voltage rating for the capacitor. Value of the inductor was selected based on simulation results to ensure that the current ripple does not exceed 20 A peak-to-

<figure><img src="image_9.png" /><figcaption>Fig. 7: Secondary side control loop schematic.</figcaption></figure>
<figure><img src="image_10.png" /><figcaption>Fig. 9: LLC resonant circuit schematic</figcaption></figure>

<table><caption>Table I: Parameters of the system</caption><thead><tr><th></th><th>Min value</th><th>Max value</th></tr></thead><tbody><tr><td>Grid voltage</td><td>184 V<sub>RMS</sub></td><td>276 V<sub>RMS</sub></td></tr><tr><td>Battery voltage</td><td>30 V</td><td>60 V</td></tr><tr><td>Power</td><td>50 W</td><td>500 W</td></tr></tbody></table>

peak. The capacitance value is therefore determined to achieve a resonance frequency of 400 kHz. Through this balanced approach, the following component values were determined:

* C<sub>S</sub>: 6 nF

* L<sub>S</sub>: 27 µH

* L<sub>M</sub>: 27 µH

## Simulation study

To validate the assumptions of this approach, simulation study was conducted using PLECS software. The simulation analysis assessed not only the parameters and functionality of the converter but also the operating conditions of the components used. The design assumptions of the converter are detailed in Table I.

The analysis was divided into three sub-sections: the first focuses on the output parameters of the converter during energy transfer from the grid to the load. The second sub-section addresses the converter's interaction with the grid, covering topics such as PFC and change in the direction of the energy flow. The third sub-section details the operating conditions of the LLC resonant circuit.

## Battery-side operation

Figure 10 presents the waveforms of the output voltage and output current of the converter under variable load and output voltage levels. The load transitions from 500 W to 50 W at a time of 0.1 s, while the output voltage remains at 60 V. At 0.2 s, the output voltage is reduced to 30 V. Subsequently, at 0.3 s, the load increases from

<figure><img src="image_11.png" /><figcaption>Fig. 10: Output voltage and output current of the converter.</figcaption></figure>

50 W to 250 W. The specified reference voltage is attained across the full range of load conditions. However, it is essential to consider the voltage ripples associated with the 50 Hz grid. The presence of those ripples is inevitable, and their amplitude is influenced by the output capacitance of the converter and the current drawn by the load. In the converter under discussion, the output capacitance is 4800 µF, and the voltage ripple at maximum load does not exceed 5 V (17% at the worst-case conditions).

## Grid-oriented parameters

A critical parameter for converters is the capability to operate at a unity power factor. In the considered converter, this functionality has been incorporated into the control loop. The waveforms depicting the current drawn from the grid and the grid voltage are presented in Figure 11, where the voltage and power levels drawn from the converter correspond to those shown in Figure 10. Throughout the entire power range, a sinusoidal current is obtained that remains in phase with the voltage.

<figure><img src="image_12.png" /><figcaption>Fig. 11: Grid current and voltage during full range of load and output voltage</figcaption></figure>

A close-up view of the grid current and voltage during operation at a power level of 500 W is presented in Figure 12a. It is evident that the current exhibits a smooth waveform with minimal
<figure><img src="image_13.png" /><figcaption>Fig. 12: Grid current and voltage during operation at: (a) maximum power of 500 W, (b) low power of 50 W.</figcaption></figure>

<figure><img src="image_14.png" /><figcaption>Fig. 13: Grid current and voltage during energy flow direction change at power of 500 W.</figcaption></figure>

ripples, even in the vicinity of the zero crossing. Figure 12b illustrates the same parameters for operation at a power level of 50 W. In this instance, significant distortion of the current waveform is observed near the zero crossing, accompanied by pronounced ripples; nonetheless, the current maintains a sinusoidal shape throughout.

The change in the direction of energy transfer does not disrupt the operation of the converter and occurs smoothly. Figure 13 illustrates this moment during operation at maximum load of 500 W.

## Resonant tank operating conditions

The integration of a resonant circuit within the converter enables soft switching or semi-soft

<figure><img src="image_15.png" /><figcaption>Fig. 14: LLC components operating conditions.</figcaption></figure>

switching, thereby substantially reducing switching losses. Figure 14 depicts the operating conditions of the LLC circuit components.

The peak amplitude of the capacitor voltage $C_S$ as well as the current through the inductors $L_S$ and $L_M$ is directly correlated with the grid voltage and exhibits a sinusoidal envelope. Figures 15a and 15b provide a detailed close-up of the operating conditions of the LLC resonant circuit components. The peak current through the inductors remains below 10 A under both maximum and minimum power conditions. However, the RMS value of the inductor current varies with the load. At maximum power, the RMS current reaches 3.46 A, while at minimum power, it is 2.94 A. The voltage across the capacitor $C_S$ is dependent on both the output voltage and the current drawn, but under worst-case conditions, it does not exceed 420 V.

## Conclusion

This article introduces a novel topology facilitating single-stage, bidirectional energy conversion between a single-phase AC voltage source and a battery, utilizing monolithic bidirectional switches. Comprehensive simulation studies were performed, validating the proposed concept. The system successfully achieved the regulation of battery current and voltage while maintaining a unity power factor, as well as seamless operation during changes in the energy transfer direction. Thus, it can be effectively and competitively employed in modern OBCs.
<figure><img src="image_16.png" /><figcaption>Fig. 15: LLC components operating conditions at: (a) maximum power of 500 W, (b) low power of 50 W.</figcaption></figure>

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