Interference-free broadband television tuner

An interference free local oscillator circuit is disclosed. A first local oscillator signal is generated in a first phase locked loop. A second local oscillator signal is generated in a second phase locked loop. Third and fourth phase locked loops provide inputs to the second phase locked loop to control the second local oscillator frequency. The operating frequencies of the first and second local oscillator signals are selected so that spurious signals generated in the phase locked loops do not interfere with a received RF signal in a conversion circuit.

RELATED APPLICATIONS 
This application is related to co-pending application entitled DUAL MODE 
TUNER FOR CO-EXISTING DIGITAL AND ANALOG TELEVISION SIGNALS, assigned Ser. 
No. 08/904,693, co-pending application entitled BROADBAND INTEGRATED 
TELEVISION TUNER, assigned Ser. No. 08/904,908, and co-pending application 
entitled BROADBAND FREQUENCY SYNTHESIZER, assigned Ser. No. 08/904,907, 
all of which are filed concurrently herewith and assigned to a common 
assignee, which applications are hereby incorporated by reference herein. 
TECHNICAL FIELD OF THE INVENTION 
This invention relates to local oscillators and more particularly to local 
oscillators that drive conversion circuits in such a way as to reduce 
interference from spurious signals. 
BACKGROUND OF THE INVENTION 
It is well known in the art to use local oscillators (LOs) to drive dual 
conversion circuits. In such circuits, a first LO signal is mixed with an 
RF signal in a first mixer to generate a first intermediate frequency (IF) 
signal. Then, the first IF signal is mixed with a second LO signal in a 
second mixer to generate a second IF signal. The frequencies of the first 
and second LO signals are usually selected so that the first and second IF 
signals occur either at a specific frequency or within a specified 
frequency range. 
The LOs may generate spurious signals at harmonic and subharmonic 
frequencies of the desired signal. One problem in the prior art occurs 
when these spurious signals feed back into the RF input or couple to some 
other part of the conversion circuit. Signal coupling is likely to occur 
when the local oscillators and the conversion circuit are constructed on 
an integrated circuit substrate. One method of eliminating the effects of 
spurious signals is to use band pass filters which attenuate the spurious 
frequencies in the IF signals. A problem with this method arises when the 
spurious signals are close in frequency to the incoming RF signal or to 
the selected IF signals. 
When a dual mixer conversion circuit is used in a television tuner, 
additional problems arise because there are more than one frequency 
associated with each television channel. For example, in the United 
States, the television system is based upon a signal that comprises a 
picture carrier at the signal frequency, a chroma carrier that is 3.6 MHz 
above the picture carrier frequency and a audio carrier that is 4.5 MHz 
above the picture carrier. As a result, spurious signals may interfere 
with any of these carrier frequencies. 
A television tuner that is constructed on a integrated circuit substrate is 
disclosed in the pending patent application entitled MONOLITHIC TELEVISION 
TUNER, filed Apr. 21, 1995 and assigned Ser. No. 08/426,080. However, that 
application does not disclose the present system and method for an 
interference free broadband tuner circuit. 
SUMMARY OF THE INVENTION 
For an RF signal, such as television signal, which has a predetermined 
shape, a conversion circuit can be controlled so that the shape of the 
output signal is proportionate to the shape of the input signal. Also, the 
output signal shape may have the same orientation as the RF signal shape 
or it may be a mirror-image of the RF signal shape. A conversion circuit 
is controlled by the local oscillator (LO) frequencies that are applied to 
the circuit's mixers. If the LO frequencies are selected to be lower than 
the desired intermediate frequency (IF) signal, then the resulting IF 
signal will have the same shape as the RF signal. On the other hand, if 
the LO frequencies are selected to be higher than the desired IF signal, 
then the mixer output will be an IF signal with a shape that is a 
mirror-image of the RF signal. When two or more mixers are used in a 
conversion circuit, various combinations of the LO frequencies can be used 
to generate IF signals having a desired shape. 
Spurious signals having harmonic and subharmonic frequencies are generated 
in local oscillator circuits and in conversion circuits. If the frequency 
of these signals are near the frequency of the desired RF signal, then 
interference will result. The present invention provides for monitoring 
local oscillator signals and the associated spurious signal frequencies in 
a conversion circuit. If the spurious signals frequencies are near the RF 
signal frequency, then the present invention provides for adjusting the 
local oscillator frequencies in such a way as to maintain a desired IF 
signal shape while minimizing interference signals at the same time. 
It is a technical advantage of the present invention to provide a system 
and method for adjusting the local oscillator frequencies of a conversion 
circuit in such a manner as to maintain a desired IF signal shape. 
It is a further technical advantage of the present invention to provide a 
system and method for calculating the frequencies of spurious signals in a 
conversion circuit. 
It is another technical advantage of the present invention to provide a 
system and method for adjusting the local oscillator frequencies of a 
conversion circuit to minimize interference from spurious local oscillator 
frequencies. 
The foregoing has outlined rather broadly the features and technical 
advantages of the present invention in order that the detailed description 
of the invention that follows may be better understood. Additional 
features and advantages of the invention will be described hereinafter. It 
should be appreciated by those skilled in the art that the conception and 
the specific embodiment disclosed may be readily utilized as a basis for 
modifying or designing other structures for carrying out the same purposes 
of the present invention. It should also be realized by those skilled in 
the art that such equivalent constructions do not depart from the spirit 
and scope of the invention as set forth in the appended claims.

DESCRIPTION OF THE PREFERRED EMBODIMENTS 
These and other objects, features and technical advantages are achieved by 
a system and method in which multiple phase locked loop (PLL) circuits are 
used to drive voltage controlled oscillators (VCOs) in order to generate 
the LO signals for a dual mixer conversion circuit. 
FIG. 1 shows conversion circuit 10 having dual mixers 102 and 104 which 
receive LO signals LO1 and LO2 on lines A and B from local oscillator 
circuit 20. In the preferred embodiment, conversion circuit 10 is used in 
a television tuner circuit as described in the above mentioned co-pending 
application entitled DUAL MODE TUNER FOR CO-EXISTING DIGITAL AND ANALOG 
TELEVISION SIGNALS. 
In a television system, signals representing individual channels are 
assigned to specific frequencies in a defined frequency band. For example, 
in the United States, television signals are generally transmitted in a 
band from 55 MHz to 806 MHz. The received RF signals pass through a 
front-end filter 100. In the prior art, filter 100 usually was a bandpass 
tracking filter that allowed only a narrow range of frequencies to pass. 
In the preferred embodiment, filter 100 is a low pass filter that is 
designed to remove all frequencies above an input cutoff frequency. The 
input cutoff frequency is chosen to be higher than the frequencies of the 
channels in the television band. The output of filter 100 then passes 
through amplifier 101 to adjust the signal level that is provided to mixer 
102. When conversion circuit 10 is used in a receiver circuit, amplifier 
101 may be an automatic gain control (AGC) amplifier that is adjusted to 
maintain an overall receiver gain. Following amplifier 101, the RF signal 
is provided to mixer 102 where it is mixed with a local oscillator signal 
LO1 from local oscillator circuit 20. The output of mixer 102 is first 
intermediate frequency signal IF1. Typically, the frequency of LO1 is 
variable and will be selected based upon the channel in the RF signal that 
is being tuned. LO1 is selected so that the mixing of LO1 and RF in mixer 
102 generates an IF1 signal either at a specified frequency or within a 
narrow range of frequencies. 
Following mixer 102, IF filter 103 is a band pass filter that is used to 
remove unwanted frequencies and spurious signals from the IF1 signal. The 
band of frequencies that are passed by filter 103 is a matter of design 
choice depending upon the IF1 frequency selected in each particular 
conversion circuit. In the preferred embodiment, IF filter 103 is centered 
at 1090 MHz and has a 14 MHz pass band. This allows the selected IF1 
frequency to vary within 1083-1097 MHz. Mixer 104 receives both the 
filtered IF1 signal from filter 103 and a second local oscillator signal 
(LO2) from oscillator circuit 20. These signals are mixed to generate a 
second intermediate frequency (IF2) at the output of mixer 104. In the 
preferred embodiment, mixer 104 is an image rejection mixer that rejects 
image frequencies from the IF2 signal. LO2 may be a variable or fixed 
frequency depending upon whether IF1 is at a fixed frequency or if it 
varies over a range of frequencies. In either case, the frequency of LO2 
is selected to generate a fixed frequency IF2 signal. The IF2 signal is 
provided through amplifier/ buffer 105 to additional processing circuitry 
(not shown) to generate either digital or analog television signals. In 
the preferred embodiment, the frequency of IF2 is selected to be 45.75 
MHz. 
An additional consideration when using a dual mixer conversion circuit in a 
television receiver is the relationship of the picture, chroma and audio 
carriers in an analog television signal. FIG. 3A illustrates the 
relationship of these carrier signals for a 73.25 MHz signal. FIGS. 3B and 
3C show the carrier signals after the 73.25 MHz RF signal of FIG. 3A has 
been converted to a first IF signal using different LO signals. FIGS. 3D 
and 3E show the carrier signals after the first IF signal of FIG. 3B is 
converted to a second IF signal at 45.75 MHz using different LO signals. 
FIGS. 3F and 3G show the carrier signals after the first IF signal of FIG. 
3C is converted to a second IF signal at 45.75 MHz using different LO 
signals. RF picture carrier 301a is at 73.25 MHz, chroma carrier 301b is 
3.6 MHz higher at 76.85 MHz and audio carrier 301c is 4.5 MHz above the 
picture carrier at 77.75 MHz. Signals 302a-c and 303a-c illustrate the 
result of mixing RF signal 30la-c with LO1 to obtain an IF1 frequency 
within the 1090 MHz.+-.7 MHz pass band of filter 103. 
IF1 301a-c results from mixing RF 301a-c with LO1a at 1160.25 MHz. As a 
result of this mixing, the relationship between the picture, chroma and 
audio carriers has been inverted. Picture carrier 301a has been translated 
to an IF1 picture carrier 302a at 1087 MHz. However, the audio carrier 
302a and chroma carrier 302c have been shifted to lower frequencies. IF1 
302a-c can be further mixed with LO2 to generate an IF2 signal with the 
picture carrier at the desired frequency of 45.75 MHz. Using LO2a at 
1041.25 MHz, the carriers in IF2 304a-c will maintain an inverse 
relationship to the RF signal 301a-c. Picture carrier 304a is at 45.75 MHz 
as desired and audio carrier 304c and chroma carrier 304b are at lower 
frequencies. On the other hand, if LO2b at 1132.75 MHz is used, then IF2 
305a-c will have the same relationship as the carriers in RF 301a-c. 
Chroma carrier 305b is 3.6 MHz above picture carrier 305a and audio 
carrier 305b is 4.5 MHz above picture carrier 305a. 
Another group of local oscillator frequencies can be used to obtain the 
same results. IF1 303a-c is generated by mixing RF 301a-c with LO1b at 
1018 MHz. IF1 303a-c maintains the same carrier relationship as RF 301a-c. 
An IF2 signal with the picture carrier at 45.75 can be generated from IF1 
303 by mixing it with LO2c at 1137.5 MHz or LO2d at 1046 MHz, as shown by 
IF2 306a-c and 307a-c. The carriers in IF2 306a-c have an inverted 
relationship to the carriers in RF301a-c and the carriers in IF2 307a-c 
have the same relationship as RF 301a-c. 
For analog television signals, it is desirable to choose a combination of 
LO1 and LO2 so that the relationship between the picture, chroma and audio 
carriers is always the same in the IF2 signal. When the IF2 signal is 
further processed after amplifier 105, it may be a consideration that the 
analog processing circuits are able to find the chroma and audio carriers 
in the same place, either above or below the picture carrier, for every 
channel. In the preferred embodiment, LO1 and LO2 are selected so that the 
IF2 shape is the inverse of the RF shape. That is, the picture carrier is 
converted from an RF signal of 55-806 MHz to an IF2 signal at 45.75 MHz 
with the audio carrier 4.5 MHz below the picture carrier and the chroma 
carrier 3.6 MHz below the picture carrier. 
As shown in FIGS. 3D and 3E, the audio and chroma carriers in IF2 signals 
304a-c and 306a-c are below the picture carrier frequency. This is 
accomplished by using the lower LO2 frequency (1041 MHz) with the higher 
LO1 frequency (1160.25 MHz) or using the higher LO2 frequency (1137.5 MHz) 
with the lower LO1 frequency (1018.5 MHz). 
LO1 is generated in local oscillator circuit 20 by PLL1 21 and LO2 is 
generated by PLL2 22. PLL3 23 and PLL4 24 provide reference inputs to PLL2 
22. I.sup.2 C 120 controls local oscillator circuit 20 and causes PLL1-4 
21-24 to select the correct LO1 and LO2 frequencies. Local oscillator 
circuit 20 receives reference signals from oscillator 122 and reference 
frequency generator 123. Oscillator 122 provides a 5.25 MHz output based 
on crystal 121. Frequency generator 123 divides the 5.25 MHz signal from 
oscillator 122 to generate additional reference signals at other 
frequencies. 
Local oscillator circuit 20 and PLL1-4 21-24 are shown in greater detail in 
FIG. 2. PLL1 21 provides the first local oscillator signal (LO1) to mixer 
102. PLL2 22, PLL3 23 and PLL4 24 cooperate to provide the second local 
oscillator signal (LO2) to mixer 104. PLL1 21 receives a 5.25 MHz 
reference signal at phase comparator 205. The output of phase comparator 
205 feeds loop amplifier 202 which, in turn, provides the input for VCO1 
201. There are two outputs from VCO1 201. One output provides the LO1 
signal to mixer 102 over line A. The other output goes into a divider 
network comprised of .div.8/.div.9 circuit 203 and .div.N circuit 204. 
Divider circuits 203 and 204 divide the output of VCO1 201 down to a 
signal having a frequency of 5.25 MHz. This divided-down signal is 
compared with the 5.25 MHz reference signal in phase comparator 205 to 
complete the phase locked loop. 
The output of VCO1 21 is variable between 1145-1900 MHz on the high side 
and 572-1033 MHz on the low side. Frequencies below 572 MHz are not used 
in LO1 to minimize the introduction of interference frequencies into the 
conversion circuit. LO1 is chosen from within these ranges so that IF1 
signal is within the 1090 MHz.+-.7 MHz pass band of filter 103. The 5.25 
MHz reference signal creates an output stepsize of 5.25 MHz in LO1 which 
is utilized for course tuning in conversion circuit 10. In the preferred 
embodiment, PLL1 21 has a bandwidth on the order of 500 KHz. A wide 
bandwidth is preferable to get good close-in phase noise characteristics. 
Fine tuning is accomplished by LO2 which is produced by the operation of 3 
phase lock loops PLL2 22, PLL3 23 and PLL4 24. PLL4 24 has the same basic 
configuration as PLL1 21. It has reference signal of 2.625 MHz which is 
input to phase comparator 235. The output of phase comparator 235 drives 
loop amplifier 232 which in turn drives VCO4 231. The output of VCO4 231 
has frequency range of 220-440 MHz with a 2.625 MHz stepsize and is 
provided to two divider circuits. One output of VCO4 231 goes to a divider 
network comprised of .div.6/.div.7 circuit 233 and .div.N circuit 234. The 
effect of divider network 233 and 234 is to divide the output signal of 
VCO4 230 back down to 2.625 MHz. This signal is then compared with the 
2.625 MHz reference signal in phase comparator 235 to complete the phase 
locked loop. The other output of VCO4 231 is provided to .div.42 circuit 
230. The output of divider 230 is a signal with a frequency range of 
5.25-10.5 MHz and having a 62.5 KHz stepsize. The output of divider 230 
serves as a reference signal for PLL2 22. 
In PLL3 23, a 5.25 MHz reference signal is input to phase detector 222. 
Phase detector 222 drives loop amplifier 221 which in turn drives VCO3 
220. The output of VCO3 23 is divided back down to 5.25 MHz by .div.N 
circuit 223 and then fed back into phase detector 222 to complete the 
loop. The output of VCO3 23 is selectable between 1128.75 MHz and 1034.25 
MHz. The selection between these two frequencies determines whether LO2 is 
on the high side or the low side. 
In PLL2 22, the signal from PLL4 24 is received by phase comparator 214 
which in turn drives loop amplifier 213. The output of loop amplifier 213 
controls VCO2 210. VCO2 210 provides the LO2 signal for mixer 104 over 
line B. The LO2 signal varies between 1134.75-1140.125 MHz on the high 
side and 1039.875-1045.125 MHz on the low side. Another output from VCO2 
210 passes through buffer amplifier 211 and then drives image reject mixer 
212. Mixer 212 receives its other input from PLL3 23. Since the signal 
from PPL3 23 is near the frequency of the LO2 signal in VCO2 210, it is 
important that the reverse isolation between mixer 212 and VCO2 210 is 
good to prevent the PLL3 23 signal from passing into the LO2 output of 
VCO2 210. The output of mixer 212 is provided to phase comparator 214 to 
complete the loop in PLL2 22. 
In the preferred embodiment, the loop bandwidths of PLL2 22, PLL3 23 and 
PLL4 24 are all wide to provide good overall close-in phase noise. PLL2 22 
and PLL3 23 have bandwidths of approximately 300-500 KHz. The bandwidth of 
PLL4 24 is approximately 200-300 KHz. These bandwidths give phase noise at 
100 KHz that is satisfactory for digital television. 
FIGS. 4 and 5 are tables showing the harmonic and spurious signal 
frequencies that occur in conversion circuit 10 for certain LO1 and LO2 
signals. In FIG. 4, column 401 represents potential picture carrier 
frequencies in the RF signal. Column 402 shows the associated LO1 
frequency that is required to generate an IF1 signal within the pass band 
of filter 103. The resulting IF1 frequencies are shown in column 403. 
Column 404 shows the frequencies of the various LO2 signals that are 
required to convert each IF1 to a 45.75 MHz IF2 signal. 
Columns 405 represent spurious signals that can be generated in PLL1 21 for 
each LO1 frequency. As indicated by the column headings, the spurious 
signals can occur at frequencies that are one-half, one-quarter, etc. of 
the LO1 frequency. Columns 406 represent similar spurious signals 
generated in PLL2-4 22-24 for each LO2 frequency. Some of these spurious 
signals are caused by the divider circuits in the phase locked loops. 
Interference problems can arise in television tuner circuits if harmonic 
or spurious signals are within -1 MHz to +5 MHz of the picture carrier 
frequency 401. 
The frequencies in FIG. 4 represent the situation where LO1 is on the high 
side and LO2 is on the low side as indicated by column 407. In row 41, the 
picture carrier frequency of 73.25 is shown along with the associated LO, 
IF and spurious signal frequencies. It can be seen from the table that a 
spurious signal at 72.5 MHz is generated in PLL1 21 as a subharmonic of 
the VCO1 210 frequency. Also, a 73.9 MHz spurious signal is generated in 
PLL3 23 from VCO3 220 and a 74 MHz spurious signal is generated in PLL4 24 
from VCO4 231. 
In FIG. 5, the same picture carrier frequencies are shown, however, for 
certain frequencies the LO1/LO2 combination has been switched so that LO1 
is on the low side and LO2 is on the high side as indicated in column 501. 
The benefits of switching the LO1 and LO2 frequencies are shown in line 51 
for the 73.25 MHz picture carrier. Now the spurious signals that fell 
within -1 MHz to +5 MHz of the picture carrier are gone and the closest 
signal is at 80.625 MHz which is more than 5 MHz higher than the picture 
carrier frequency. 
Although a television tuner circuit has been used to describe the 
advantages of the present invention, the same principles can be applied to 
prevent interference in other conversion circuits that operate in other 
frequency bands. I.sup.2 C 120 can be controlled by a processor, such as a 
computer, so that it is capable of monitoring the RF and local oscillator 
frequencies, determining potential spurious signal frequencies and 
adjusting the local oscillator frequencies as appropriate to reduce 
interference and to maintain the desired output signal shape. 
It will be understood that the present invention can be embodied to operate 
with either digital or analog signals in any frequency range. Generally, 
the orientation of the second IF signal will only be a consideration for 
analog television signals. It will be further noted the I.sup.2 C protocol 
need not be used but this circuit will work with any digital interface. 
FIG. 3A illustrates a typical analog television signal having three 
distinct carrier frequencies 301a-c. Analog RF television signal 301a-c is 
converted to a second IF signal, such as signal 304a-c of FIG. 3D. The 
second IF signal is then further processed by the video and audio sections 
of a receiver. Typically, the second IF signal will pass through a filter 
with an asymmetrical pass band that is designed specifically for analog 
television signals. Such a filter is disclosed in the above mentioned 
pending patent application entitled DUAL MODE TUNER FOR CO-EXISTING 
DIGITAL AND ANALOG TELEVISION SIGNALS. 
When a second IF signal comprising analog television carriers passes 
through an asymmetric filter, it is critical that the carriers are 
positioned properly in the filter pass band. In a typical analog 
television receiver, a second IF signal filter will be designed to provide 
different levels of attenuation for the picture carrier and the audio 
carrier. If the carriers are reversed when the second IF signal passes 
through the filter, then the signal will not be filtered correctly. This 
has the effect of limiting the choice of LO frequencies to those that 
produce the correct second IF signal shape. The first and second LO 
signals usually will have to be correlated so that one is in a high band 
and one is in a low band. 
On the other hand, a digital television signal is likely to have a flat, 
symmetrical shape. In this case, a second IF signal filter will also have 
a flat and symmetrical pass band. Therefore, the orientation of the second 
IF signal will not be a critical factor when it passes through a second IF 
signal filter. Accordingly, as long as the second IF signal is at the 
proper frequency, its shape will not be dependent upon a specific filter 
characteristic. This will allow a broader range of first and second LO 
frequencies to be used. In the case of a digital television signal, the 
first and second LO signals will not have to be correlated between 
low/high and high/low frequency bands. 
In a more broad application, beyond strict television signal receivers, 
anytime a receiver symmetrically filters the second IF signal, then all 
combinations of the first and second LO frequencies may be used. A control 
circuit that monitors the LO frequencies, such as I.sup.2 C control 120, 
can also provide a signal to the audio and video processing circuits of 
the receiver to indicate the actual shape or orientation of the second IF 
signal so that the signal is processed correctly. Additionally, this would 
apply to receivers having a single mixer conversion circuit and 
symmetrical filtering. If there is a potential for interference from one 
LO frequency choice, then the system could switch to the another LO 
frequency and still generate the same IF signal frequency. 
The present invention will be understood to operate with conversion 
circuits having any number of mixers and LO signals. Software interfacing 
through I.sup.2 C 120 directs the local oscillators to switch operating 
frequencies if there is a potential for interference. Unless there is a 
requirement to maintain a specific IF signal orientation or shape, the 
combination of the various local oscillator frequencies will be limited 
only by the IF signal frequencies to be generated by the conversion 
circuit. 
Although the present invention and its advantages have been described in 
detail, it should be understood that various changes, substitutions and 
alterations can be made herein without departing from the spirit and scope 
of the invention as defined by the appended claims.