Voltage regulation circuit for a solar cell charging system

A voltage regulation circuit includes an active integrator circuit which generates an output signal representing the time integral of the error between the voltage across a load and a reference voltage. The integrator circuit output is connected to a series of voltage comparators (each having hysteresis) each of which controls one of a series of separate switch elements each controlling a proportion of the total current flow from a source to the load.

This invention relates to a voltage regulation circuit which is primarily 
intended for use in a storage battery charging installation. 
If has been proposed to use solar energy collectors for charging storage 
batteries, particularly in inaccessible locations where there is a 
continuous current drain from the battery and no mains electric supply is 
available. Naturally, the amount of charging current which can be supplied 
by a solar energy collector depends on the time of day and usually on the 
season of the year and on weather conditions and some form of voltage 
regulation which will vary the current provided by the solar energy 
collectors in accordance with the battery state of charge is therefore 
required. 
Both conventional continuous series and continuous shunt type regulators 
have the disadvantage of high power dissipation and possible instability 
with certain regulator loads. Both conventional switching series and 
switching shunt type regulators have the disadvantage that the total 
available source current is switched by a single switch thereby causing 
large voltage disturbances across the load. 
It has also been proposed (see for example Belgium Pat. No. 853124 
published July 18, 1977) to utilize an error amplifier comparing the 
battery voltage with a reference and driving a plurality of comparators 
controlling switches each connected to shunt a proportion of the source 
output current. With such an arrangement, however, it is difficult to 
achieve very low frequency switch operation unless a capacitor of very 
large capacitance is connected across the source 
It is accordingly an object of the invention to provide a voltage regulator 
capable of use in an application such as that mentioned above which avoids 
or reduces the disadvantages mentioned. 
A voltage regulation circuit in accordance with the invention comprises a 
battery voltage sensing means including an electronic active integrator 
circuit producing an output representing the time integral of difference 
between the voltage of a battery under charge and a reference voltage, a 
plurality of comparator circuits comparing the output of said integrator 
circuit with a plurality of different reference levels, and a plurality of 
switch devices controlled by respective ones of the comparator circuits 
whereby, in use, each of said switch devices acts selectively to permit or 
prevent the flow of a portion of the output current from charging current 
source means to the battery the switch devices being sequentially operated 
by the comparator circuits as the fraction of the current output 
capability of said source means required to maintain the battery voltage 
and the reference voltage varies, and the integrator circuit operating so 
as to control the duty ratio of at least one of the switch elements. 
The voltage regulation circuit defined above may be used to control a 
battery charging installation in which the current source is constituted 
by a plurality of independent current source sections. 
Where the voltage regulation circuit is used with a plurality of 
independent current source sections, one or more switch devices may be 
arranged to shunt each current source section. 
Alternatively, where the voltage regulation circuit is used with a 
plurality of independent current source sections, each current source 
section may have an associated switch device connected in series 
therewith. 
The voltage regulation circuit defined above may also be used in 
conjunction with a single current source, the switch device, which is 
first to permit its associated current portion to charge the battery as 
the fraction of the available current required to attain the reference 
voltage increases, being connected directly across the current source, and 
each of the remaining switch devices being connected in series with an 
associated resistor, the switch element and resistor being connected 
across the current source. 
The comparator circuits are preferably connected to operate with hysteresis 
to provide said duty ratio control.

Referring firstly to FIG. 1 there is shown therein a battery charging 
installation including a plurality of independent solar energy 
conversation units 10, 11, 12, 13 and 14. These each have one output 
terminal connected to a negative bus 9 and the other output terminal 
connected by an associated one of a plurality of diodes 15, 16, 17, 18 and 
19 to a positive bus 20. Associated with each unit 10 to 14 is an 
associated one of a plurality of switching elements in the form of npn 
power transistors 21, 22, 23, 24 and 25. Each such transistor has its 
emitter connected to the negative bus 9 and its collector connected to the 
said other output terminal of the associated unit 10 to 14. Thus when each 
transistor 21 to 25 is on it shunts the current produced by the associated 
unit 10 to 14. 
The transistor 21 to 25 are controlled by a voltage sensing means as will 
now be described. 
The transistors 21 to 25 have their respective bases connected to the 
emitters of five npn drive transistors 26 to 30. Each transistor 26 to 30 
has its collector connected to the positive bus 20 by an associated one of 
a plurality of resistors 31 to 35 and an associated one of a plurality of 
light emitting diodes 36 to 40 respectively. The base of each transistor 
26 to 30 is connected by a respective resistor 41 to 45 to the output of 
an associated one of a plurality of voltage comparators 46 to 50. The 
non-inverting inputs of the comparators 46 to 50 are connected to points 
on a resistor chain 51 to 56 connected between the cathode of a zener 
diode 57 and the negative bus 9. The anode of the zener diode 57 is 
connected to the negative bus 9 and its cathode is connected by a resistor 
58 to the positive bus 20. 
To provide each comparator with hysteresis an associated one of a plurality 
of positive feedback resistors 59 to 63 connects its output to its 
non-inverting input. 
The inverting inputs of all the comparators 46 to 50 are connected together 
and are connected by a resistor 64 to the output of an operational 
amplifier 65 connected as an integrator. The inverting input of the 
amplifier 65 is connected by a resistor 66 to a point on a resistor chain 
consisting of a variable resistor 67 and two resistors 68, 69 connected in 
series between the positive and negative buses 20 and 9. A capacitor 70 is 
connected between the output of the operational amplifier 65 and the 
above-mentioned point to provide the required integrating action. The 
non-inverting input of the amplifier 65 is connected by a resistor 71 to 
the cathode of a voltage reference device 72 having its anode connected to 
the bus 9 and its cathode connected via a resistor 73 to the bus 20. 
A protective zener diode 74 having a break down voltage less than the 
voltage which can cause damage to the transistors 21 to 30 is connected 
between the buses 9 and 20. 
A capacitor 75 ensures normal operation in the event of battery 
disconnection during commissioning or testing. 
The output voltage of the amplifier 65 is proportional to the integral of 
the error between the battery voltage and a reference voltage set by the 
reference device 72. Thus when the battery voltage is less than the 
reference voltage the integrator output increases at a rate proportional 
to the magnitude of the error. Similarly when the battery voltage is 
higher than the reference voltage the output of the integrator dereceases 
at a rate proportional to the magnitude of the error. When the output of 
operational amplifier is very high (indicating that the battery has been 
low for some time for example overnight), the outputs of all the 
comparators 46 to 50 will be low so that all the transistors 21 to 25 are 
off and any current generated by all the units 10 to 14 can flow to the 
battery. As battery charging proceeds in the morning its voltage rises so 
that eventually it rises above the reference level. The output of 
integrator amplifier 65 then starts to fall until the output of comparator 
46 goes high. The current being produced by unit 14 is now shunted and 
this will cause the battery voltage to fall below the reference level so 
that the integrator output rises again and switches off transistor 21 
thereby restoring full current to the battery. This switching on and off 
continues with the duty ratio of current into the battery gradually 
decreasing (as a result of increasing sunlight falling on the units 10 to 
14) to maintain the average value of the battery voltage constant 
Eventually the point will be reached where switching on transistor 21 does 
not cause the battery voltage to fall below the reference level. The 
output of the amplifier 65 then continues to fall until the output of 
comparator 47 goes high and transistor 22 switches on so that the units 14 
and 13 are both shunted. Transistor 21 now stays on continuously (assuming 
steadily increasing sunlight) and the duty ratio of transistor 22 is 
controlled to maintain the average battery voltage at the reference level. 
It will be appreciated that as the sunlight intensity increases the 
transistors 21 to 25 are turned on successively, the last of the 
transistors turned on having its duty ratio controlled automatically. 
The reverse sequence of events occurs when the sunlight intensity 
decreases, or when the current required to maintain the battery voltage 
increases (e.g. when a load is connected across the battery). 
Thus as the fraction of the total current available from the units 10 to 14 
required to maintain the battery voltage reference level decreases each of 
the transistors 21 to 24 is switched on in turn and as the fraction 
increases each of the transistors 21 to 24 is switched off in turn. 
The use of an active integrator circuit in the circuit of FIG. 1 confers 
many benefits. Firstly, by choosing suitably large values for the 
resistors 66, 67, 68 and 69 and the capacitor 70, the switching frequency 
of the circuit can be kept low i.e. in the region of 0.1 to 1 Hz. To 
obtain a similar switching frequency utilizing merely the capacitor 75 for 
smoothing of the voltage would require capacitor 75 to have an excessively 
large value, so that it would need to be a bulky and expensive 
electrolytic device with all the known drawbacks of such devices. 
Furthermore, the use of a linear amplifier instead of an active integrator 
would give rise to problems of ensuring that the d.c. gain of the 
amplifier could be accurately predetermined. With an integrator d.c. gain 
is relatively unimportant because the error is cancelled out completely 
when an equilibrium condition exists. The compromise between sensitivity 
and stability, which arises when a linear amplifier is used, does not 
arise when an integrator is employed, and accurate voltage regulation is 
obtained. 
In the arrangement shown in FIG. 2 the voltage sensing means itself is 
identical to that shown in FIG. 1, but only one high power source 110 is 
used. To obtain switched stepwise current control, the transistors 21 to 
24 have their collectors connected by respective resistors 111 to 114 to 
the output terminal of the source 110 and the collector of transistor 25 
is connected directly to this output terminal. A single diode 115 connects 
the output terminal to the positive bus 20. The values of resistors 111 to 
114 are chosen so that when any transistor 21 to 24 is on its shunts one 
fifth of the maximum output current of the source 110. If with transistors 
21 to 24 conducting too much current still reaches the battery, then 
transistor 25 turns on intermittently. However, transistor 25 shunts all 
the output current of source 110 as it is connected in parallel with 
resistors 111 to 114. 
Turning now to the modification shown in FIG. 3 a resistor 210 connects the 
non-inverting input of the amplifier 65 to the collector of an npn 
transistor 211, the base of which is connected by a resistor 212 to the 
negative bus 9. The base of transistor 211 is also connected by two 
resistors 213, 214 in series to the cathode of a diode 215 the anode of 
which is connected to the output of comparator 46. A capacitor 216 
connects the junction of resistors 213, 214 to the bus 9. The emitter of 
transistor 211 is connected to bus 9. 
With this modification at the start of a days charging all the comparators 
46 to 50 have their outputs low so that capacitor 216 is in a discharged 
state. Thus, initially, the reference voltage has a value U.sub.1. When 
the battery voltage rises above U.sub.1, and eventually comparator 46 
output goes high, the capacitor 216 charges up and switches on the 
transistor 211. This causes the reference voltage to be reduced slightly 
to a lower value U.sub.2 which serves as the reference voltage for the 
remainder of the day light hours. Whilst the comparator 46 is switching on 
and off the capacitor 216 keeps transistor 211 on continuously. 
Turning now to FIG. 4, in this example the arrangement of the voltage 
sensing means and power transistors is generally similar to that shown in 
FIG. 1 except that the comparator 50, resistor 56, transistor 30 and power 
transistor 25 and the elements associated therewith have been emitted. In 
this example there are two solar energy conversion units 210, 211, and 
these have their one output terminal connected to the negative bus 9 and 
their other output terminals connected respectively through lines 212, 213 
and diodes 214, 215 to the positive bus 20. Also, the collector of the 
power transistor 21 is connected through a resistor 216 to line 212, the 
collector of power transistor 22 is connected directly to line 212, the 
collector of transistor 23 is connected through a resistor 217 to line 
213, and the collector of transistor 24 is connected directly to line 213. 
The values of resistor 216 and 217 are chosen so that when transistor 21 is 
conductive one quarter of the total available current is shunted, when 
transistors 21 and 22 are conductive one half of the available current is 
shunted, when transistors 21, 22 and 23 are conductive three quarters of 
the available current is shunted, and when all transistors are conductive 
the entire current is shunted. 
The examples shown in FIGS. 1 to 4 are suitable for use with low voltage 
installations e.g. 6, 12 or 24 volts. In these examples, as the power 
transistors shunt the solar energy conversion units, there is no voltage 
drop across the switching elements during charging. 
However, the general arrangement of the voltage sensing means shown in 
these examples is also suitable for use with switching elements connected 
in series with solar energy conversion units where the installation 
voltage is higher, for example 60 volts, and consequently the voltage drop 
across the switching elements is acceptable. Such an arrangement is shown 
in the example of FIG. 5 which will now be described. 
In FIG. 5, the voltage sensing means is generally similar to that shown in 
FIG. 1 except there are only two comparators 46 and 47 and two drive 
transistors 26 and 27, the comparators 48, 49 and 50, and the drive 
transistors 28, 29 and 30 being omitted together with associated elements. 
Also, in the voltage sensing means of this example the polarity of the 
inputs of the comparators 46, 47 is reversed, and the emitters of 
transistors 26 and 27 are connected directly to the negative bus 9. 
In this example, there are two solar energy conversion units 310, 311, the 
one output terminal of which are connected to the negative bus 9. The 
other output terminal of unit 310 is connected to the emitter of a pnp 
Darlington transistor 312, the collector of which is connected through a 
diode 313 to the positive bus 20, and the base of which is connected 
through a resistor 314 to the collector of drive transistor 27. The other 
output terminal of unit 311 is connected to the emitter of a pnp 
Darlington transistor 315, the collector of which is connected through a 
diode 316 to the positive bus 20, and the base of which is connected 
through a resistor 317 to the collector of drive transistor 26. 
In this example, when the current available from units 310, 311 is not 
sufficient for the battery to maintain its reference level, both 
transistors 312 and 315 will be conductive. Then, as the available current 
increases, and the battery voltage rises above the reference level, the 
output of amplifier 65 will decrease thereby causing the output of 
comparator 46 to go low, and transistors 26 and 315 to switch off, and 
thereby blocking current from unit 311 from charging the battery. As the 
available current increases further, the transistor 315 will switch on and 
off with a decreasing duty ratio until eventually the transistor 315 is 
switched permanently off and then transistor 312 will switch on and off. 
As may be appreciated, the circuit of FIGS. 4 and 5 may be modified in a 
manner similar to the modification shown in FIG. 3 with reference to FIGS. 
1 and 2. 
It will be appreciated that in all the examples described above each of the 
power transistors is only controlling a fraction of the current output of 
the current source and only one is being switched at any given instant. 
Consequently, the voltage disturbance problems which can arise in 
conventional switching series and switching shunt type regulators are less 
likely to arise. The change in battery voltage caused by one of the power 
transistors being switched on or off is relatively small so that switching 
can be effected at a relatively low frequency, no passive smoothing 
components being needed, to reduce ripple. 
Also, in all these examples, as no continuous power elements are involved, 
there are no stability or power dissipation problems. 
Further, there is no necessity for the circuit of any of these examples to 
be factory matched with the solar energy conversion units they are 
intended to control. Even if different units produce different amounts of 
current at the same intensity of illumination, the 
integrator/multicomparator voltage sensing means arrangement described 
above will cope adequately (provided the maximum current rating of an 
individual power transistor is not exceeded). 
Turning now to FIG. 6, elements which are common to the example shown in 
FIG. 1 have the same reference numerals and will not be redescribed. 
Each of the transistors 26, 27, 28 now drives a relay coil instead of an 
output transistor, there being three such coils 401, 402 and 403. The 
transistor 26 has its emitter connected to rail 9 and its collector 
connected via a switch contact 404a to one end of the relay winding 401 
the othe end of which is connected to a rail 405 which is connected by a 
diode 406 to the rail 20. A resistor 407 and a diode 408 (in this case a 
light-emitting diode) are connected in series between the rail 405 and the 
collector of the transistor 26. A resistor 409 connects the base of the 
transistor 26 to the rail 9. Similarly transistor 27 is associated with 
the relay winding 402, a resistor 411 and diode 412, and a resistor 413 
and transistor 28 is associated with the relay winding 403, a resistor 414 
and diode 415 and a resistor 416, but no switch corresponding to switch 
404a is associated with transistors 27 and 28. The relays 401, 402 and 403 
have normally closed contacts 401a, 402a, 403a connecting the three 
sources 10, 11, 12 via the diodes 15, 16, 17 to the rail 20. 
To prolong the life of the relay contacts the switching rate is reduced by 
connecting a further resistor 420 in series with the inverting input to 
the operational amplifier 65. In addition a switch contact 404b ganged 
with the switch contact 404a is connected between the output terminal of 
the amplifier 65 and the capacitor 70 and two resistors 421, 422 are 
connected in series between the junction of this contact 404b with 
capacitor 70 and a rail 423 connected by a diode 424 to the rail 20. A pnp 
transistor 425 has its emitter connected to the rail 423 and its base 
connected to the junction of resistors 421 and 422. The collector of the 
transistor 425 is connected by a resistor 426 to the junction of the 
resistors 68 and 69. A diode 427 has its anode connected to the junction 
of resistors 66 and 420 and its cathode connected by a capacitor 428 to 
the rail 9 and by a resistor 429 to the rail 423. 
A third ganged contact 404c of the switch 404a, 404b has its common pole 
connected to rail 20. One pole of contact 404c is connected to the end of 
resistor 67, an additional variable resistor 430 also being connected 
between rail 20 and the resistor 67 and being shorted out by contact 404c 
in one position of the latter. The other pole of contact 404c is connected 
by a resistor 431 and a light-emitting diode 432 to the earth rail 9. 
Two further switches 433 and 434 are incorporated of which the first 433, 
when closed connects the inverting inputs of comparators 46 and 47 to the 
rail 423 and the other of which 434, connects the inverting input of 
comparator 48 to the rail 9, an additional resistor 435 being connected 
between the resistor 64, are the inverting input of comparator 48. 
The resistor 429, capacitor 428 and diode 427 act to ensure correct 
operation when the circuit is first switched on or connected to the 
battery and current sources. Thus, initially, whilst the voltage on 
capacitor 428 is low, the output of the amplifier 65 will slew positively 
at a rapid rate (as compared with the very slow rate of slew which can be 
caused by resistor 66 and capacitor 70). Once capacitor 428 is charged to 
a voltage greater than that of zener diode 72, it ceases to have any 
effect on the operation of the circuit. 
The transistor 425 with the associated resistors 421, 422 and 426 provides 
the same function as the added components in FIG. 3, i.e. it ensures that 
at day break the output of amplifier 65 is fully positive and holds 
transistor 425 non-conducting. As before an initial period of charging at 
a higher voltage is provided before the integrator output goes low enough 
to switch on transistor 425. 
The switches 404, 433 and 434 and the associated resistors 430, 435 and 431 
and diode 432 are included for use during calibration and fault finding 
routines and normally occupy the positions shown. 
With the arrangement shown in FIG. 6 the resistor 66 and capacitor 70 are 
chosen to provide a switching frequency of about 0.001 to 0.01 Hz. 
Turning now to FIG. 7 this shows how the circuit of FIG. 6 can be modified 
to control a single source as in FIG. 2. The contacts 401a, 402a and 403a 
are used to switch resistors 501 and 502 into and out of the charge 
current circuit. With the three relays all de-energised, all the contacts 
401a, 402a and 403a are closed so that there is a direct connection from 
the source via the single diode 503 to the battery. When relay 401 is 
energised reduced charging current flows through resistors 501 and 502 in 
parallel. When relays 401 and 402 are both energised, the resistor 501 
alone is left in the charge current path and this path is interrupted when 
relay 403 is energised. 
The arrangement shown in FIG. 8 is used when it is required to charge a 
high voltage battery from a plurality of sources 600, 601 . . . 605 in 
series. With such an arrangement it is found possible to obtain the 
desired level of charging control by shorting out selected ones of the 
current sources. As shown a pnp output transistor 621 (corresponding to 
transistor 21 of FIG. 1) has its emitter-collector connected across two of 
the sources (604, 605) and another output transistor 622 has its 
emitter-collector connected across another two of the sources (602, 603). 
Each transistor 621, 622 has its base connected to the collector of a 
respective one of the transistors 26, 27 each of the latter having its 
emitter grounded and its base connected by a respective resistor 41, 42 to 
the output of an associated one of the comparators 46, 47 of which only 
two are necessary. When the output of the integrating amplifier 65 is high 
both transistors 621 and 622 are off so that the whole series array of 
sources is connected across the battery. As the output of the integrator 
falls during charging, the comparator 46 output goes high first, turning 
on transistor 26 and hence transistor 621 and thereby shorting out the two 
sources 604 and 605. If the integrator output falls still lower sources 
602 and 603 and shorted out. 
FIG. 9 shows how the relay arrangement of FIG. 6 can be applied to a series 
array of sources 601 . . . 605. In this case the negative end of source 
605 is connected by the normally closed relay contact 401a to ground and 
the junction of sources 603 and 604 is connected by a diode 701 and the 
normally closed contact 402a to ground. A further diode 702 connects the 
positive end of source 602 to earth. At high demand both contacts 401a and 
402a are closed and the diodes 701 and 702 are both reversed biased so 
that the whole current source array is connected across the battery. When 
relay 401 is energised contact 401a opens and the "return" path for 
charging current is then through the contact 402a and diode 701. When 
relay 402 is energised the return path is through diode 702. 
In both of FIGS. 8 and 9 removing four of the series sources from the array 
will result in the total source voltage being less than the battery 
voltage.