Static convertor apparatus

A static convertor with forced commutation is connected between an alternating voltage network and a direct voltage network. A filter is connected in parallel with a smoothing capacitor on the DC side of the convertor. The filter consists of the series-connection of an inductor, a filter capacitor and two controllable semiconductor valves that are connected in parallel in opposed relation. The natural frequency of the filter is higher than the product of the frequency of the alternating voltage network and the pulse number of the converter. The valves are made to carry current in time with the fundamental tone of the AC component present in the direct current of the convertor. In this way, the variations of the direct voltage of the convertor, caused by the AC direct current component, may be practically eliminated.

BACKGROUND OF THE INVENTION 
The present invention relates to static convertor equipment for 
transmitting energy between an AC network and a DC network, and, more 
particularly, to such equipment including a convertor with forced 
commutation that is connected to an AC network through an inductance 
element, and to a DC network having a smoothing capacitor and a filter 
connected in parallel with the capacitor, the filter having an inductor 
that is connected in series with a filter capacitor. 
Static convertor apparatus for transmitting energy between an AC and a DC 
network is known to the art, for example such apparatus is disclosed in 
the published German patent application No. 2,217,023. For such apparatus, 
the filter circuits connected on the DC side of the convertor may be 
assumed to maintain the direct voltage substantially constant. The 
alternating voltage generated by the convertor may then be controlled, for 
example by pulse width modulation of the convertor, so that the voltage 
becomes approximately sinusoidal and has the same frequency as the 
frequency of the AC network. By controlling the amplitude and the phase 
position of the alternating voltage of the convertor, the amplitude and 
the phase position of the alternating current flowing between the AC 
network and the convertor may be controlled and, thus, the magnitude and 
direction of the active and reactive power flowing between the AC network 
and the convertor equipment may be individually and arbitrarily 
controlled. In this way, complete control is obtained of the power that is 
transmitted between the AC and the DC networks, regardless of the 
direction of the power flux. At the same time the equipment may be 
controlled so that its reactive power consumption is maintained at a 
desired value, for example zero. 
The inductance element that is connected between the convertor and the AC 
network and that takes up instantaneous voltage differences between the DC 
and AC voltages of the convertor may consist of a separate inductor or of 
the inductance of a transformer that is used for connecting the convertor 
to the AC network. 
Equipment of the kind described above may, for example, be used to supply 
an AC motor with a variable frequency from an AC network with a constant 
frequency. The DC network then consists of an inverter connected to the DC 
side of the convertor. 
A smoothing capacitor connected on the DC side of the convertor limits 
variations in the direct voltage. The direct current of the convertor will 
contain an AC component with a fundamental tone that has a frequency equal 
to the frequency of the AC network, multiplied by the pulse number of the 
convertor. If the convertor is a single-phase convertor (low pulse number) 
and the frequency of the AC network is low, the fundamental tone of the AC 
direct current component will have an amplitude that is of the same order 
of magnitude as the mean value of the direct current and it will have a 
low frequency. 
In order to keep the variations in the direct voltage within reasonable 
limits an unrealistically high capacitance for the smoothing capacitor is 
often required or, at least, the required high capacitance seriously 
limits the maximum convertor power that may be installed, for reasons of 
space and weight. 
From the above-mentioned German application No. 2,217,023 it is known to 
tune an LC filter to twice the power frequency and to connect the filter 
in parallel with a smoothing capacitor. In this way, the required 
capacitance of the smoothing capacitor is reduced, since the filter can be 
operated to completely eliminate the voltage variations that are due to 
the fundamental tone component of the AC direct current component of the 
convertor. The smoothing capacitor then only needs to damp the current 
harmonics of a higher order. However, for such prior art circuits, the 
necessary smoothing components (the smoothing capacitor and the filter) 
become undesirably large and heavy, particularly in situations where the 
equipment must be built into a vehicle. Thus, the rated power of the 
filter components only, at the frequency of the fundamental tone, will be 
of the same order of magnitude as the maximum active power for which the 
convertor is dimensioned. 
Furthermore, for prior art filters there is the added risk that variations 
in operating temperature may change the tuning of the filter so that the 
filtering becomes less efficient. Accordingly, it is a primary object of 
the invention to provide a static convertor apparatus that transmits 
energy between an AC and a DC network and that employs filter components 
of reduced size. 
A further object of the invention is to provide such an apparatus that will 
not be affected in operation by variations in the impedance values of 
filter components due to such factors as temperature or aging. 
These and other objects of this invention will become apparent from a 
review of the detailed specification which follows and a consideration of 
the accompanying drawings. 
BRIEF SUMMARY OF THE INVENTION 
In order to achieve the objects of the invention and to overcome the 
problems of the prior art, the improved convertor apparatus, in accordance 
with the invention, includes a static convertor with forced commutation 
that has a characteristic pulse number. The convertor is connected on its 
AC side to a power-supplying alternating voltage network through an 
inductance element and on its DC side to a smoothing capacitor that is 
connected in parallel with a filter. 
The filter includes a series-connected capacitor and inductor and the 
filter has a natural frequency that is higher than the product of the 
frequency of the alternating voltage network and the pulse number of the 
convertor. The filter is connected in series with two thyristors that are 
connected together in parallel in opposed relation. 
A control circuit is connected to the thyristors and the circuit operates 
to alternately turn on the thyristors synchronously with the fundamental 
tone of the AC component of the direct current of the convertor.

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS 
The remaining portion of this specification will describe preferred 
embodiments of the invention when read in conjunction with the attached 
drawings, in which like reference characters identify identical apparatus. 
FIG. 1 shows a single-phase convertor SR that comprises four thyristors 
T11-T14 that may be turned on and off in a manner known to the art. Each 
thyristor is connected in parallel in an opposing relation to an 
associated diode D11-D14. The convertor SR is connected to an AC network 
that includes a transformer TR that is connected to a contact line KL 
through a current collector SA. The contact line KL carries AC voltage 
having a frequency that is assumed to be 162/3 Hz for purposes of 
explaining the operation of the invention. 
The convertor is connected to the AC network through an inductance element 
L.sub.i that may be either a separate inductor or the leakage inductance 
of the transformer TR, for example a leakage flux transformer. The DC 
terminals of the convertor are connected to a DC network that, in this 
case, is an intermediate link between the convertor SR and an inverter 
VXR. However, in general, the DC network may be any network or load 
object. 
The inverter VXR is a self-commutated, three-phase inverter having a 
controllable frequency for controlling the speed of an asynchronous motor 
M. The inverter includes thyristors T21-T26 which can be turned off in a 
manner known to the art and associated diodes D21-D26 that are connected 
in parallel and in opposed relation to their respective thyristors. The 
inverter is known to the art and may be controlled in a known manner to 
provide a desired value of frequency and/or amplitude of alternating 
voltage to the motor M. 
The convertor may be controlled by pulse width modulation, for example in 
the manner described in the above-mentioned German application No. 
2,217,023, so that it generates an alternating voltage with a fundamental 
tone having the same frequency as the main frequency and having a moderate 
harmonic content. By controlling the amplitude and phase position of the 
alternating voltage output of the convertor relative to the main voltage, 
it is possible to individually control the magnitude and direction of the 
active and the reactive components of the alternating current that flows 
between the network and the convertor. 
The active current component is controlled by influencing the phase 
position of the convertor voltage so that the mean value of the DC link 
voltage u.sub.d is maintained constant at a predetermined value that is 
higher than the peak value of the alternating voltage from the transformer 
TR. The reactive current component is controlled by influencing the 
amplitude of the convertor voltage so that the reactive current component 
of the current taken from the contact line becomes zero. 
Since the current flowing between the network and the convertor is 
approximately sinusoidal, the direct current i.sub.d of the convertor 
contains a strong AC component. The AC component has a fundamental tone 
with a frequency that is the power frequency multiplied by the pulse 
number of the convertor. A single-phase convertor of the type described 
has a pulse number 2. Therefore, at the power frequency 162/3 Hz the 
frequency of the fundamental tone is 331/3 Hz. 
In order to maintain the variations in the DC link voltage u.sub.d within 
reasonable limits despite the large AC component in the direct current of 
the convertor, a smoothing capacitor C.sub.p and a filter are connected on 
the direct voltage side of the convertor. The filter icludes an inductor 
L.sub.s that is connected in series with a capacitor C.sub.s and with two 
thyristors T1 and T2 that are connected in parallel in opposing relation. 
A control pulse device SPD is connected to control inputs of the thyristors 
T1 and T2. In operation, a current measurement device IM delivers a 
measurement signal i'.sub.d that is proportional to the instantaneous 
value of the direct current i.sub.d of the convertor SR. The measurement 
signal is differentiated in a differentiating circuit DV and the output 
signal di.sub.d /dt from the differentiating circuit is supplied to the 
summation devices S1 (with inverted sign) and S2. A constant comparison 
quantity i.sub.o is supplied to the summation devices, the quantity being 
obtained from a potentiometer P. The output signals from the summation 
devices are supplied to level flip-flops NV1 and NV2. Each of the 
flip-flops emits a logical one voltage signal when the input signal to the 
flip-flop is positive. 
The output signals of the flip-flops are supplied to the set inputs S 
(dynamic inputs) of two bistable circuits BV1 and BV2. When the output 
signal of a level flip-flop is changed from "zero" to "one", the 
corresponding bistable circuit is set to one, that is its output signal Q 
becomes a logic "one". The output signals Q and dynamic reset inputs R of 
the circuits BV1 and BV2 are cross-connected. Thus, when one of the output 
signals Q of the bistable circuits switches from "zero" to "one", the 
output Q of the opposite bistable circuit is set to "zero". The output 
signals Q of the bistable circuits BV1 and BV2 are amplified by drivers F1 
and F2 and are applied to the thyristors T1 and T2 in the form of control 
signals SP.sub.T1 and SP.sub.T2. 
The operation of the equipment shown in FIG. 1 will now be described with 
reference to FIG. 2. As shown in FIG. 2, the alternating current i.sub.N 
that is applied to the convertor SR from the AC network is assumed to be 
sinusoidal, which (apart from high frequency harmonics) is a good 
approximation of what applies in practice. Also, the direct current 
i.sub.d at the output of the convertor SR contains an Ac component, the 
fundamental tone of which has a frequency that is twice that of the power 
frequency, since in this case the convertor SR has the pulse number 2. If 
the convertor is controlled so that i.sub.N is in phase with the voltage 
on the network side of the convertor, i.sub.d assumes the shape shown by 
the continuous line, and the amplitude of the fundamental tone of the AC 
component becomes equal to the mean value I.sub.D of i.sub.d (dashed 
horizontal line). If there is a phase difference between the alternating 
voltage and current of the convertor, and if the amplitude of the AC 
component of i.sub.d is unchanged, then the mean value of I.sub.D is 
lowered; that is, the curve i.sub.d is moved downwards for example, to the 
dash-lined sine curve, when the phase difference increases. 
FIG. 2 shows the direct voltage u.sub.d on the direct voltage side of the 
convertor. The curve for the voltage u.sub.d is shown for the hypothetical 
case wherein the filter L.sub.s -C.sub.s is inactive. Thus, the AC 
component of the voltage is limited only by a capacitor C.sub.p having a 
moderate capacitance. It should be understood that u.sub.d has a direct 
voltage component U.sub.D and an alternating voltage component with a 
frequency that is twice the power frequency and with a relatively high 
amplitude. 
The third curve from the bottom of FIG. 2 shows the signal di.sub.d /dt 
that is obtained from the differentiating circuit DV. The di.sub.d /dt 
signal is a sine curve that leads the AC component of i.sub.d in phase by 
90.degree. and that is compared in the summation circuits S1 and S2 with 
the signals +i.sub.o and -i.sub.o. 
At t.sub.2 the output signal from NV1 becomes "one" and the flip-flop BV1 
is set to one, thereby applying a control signal SP.sub.T1 to the 
thyristor T1 that causes T1 to ignite. Of course, at the same time the 
flop-flop BV2 is set to zero and the control signal SP.sub.T2 to the 
thyristor T2 terminates. The filter circuit L.sub.s -C.sub.s then performs 
half a cycle of an oscillation and thereafter the thyristor T1 is 
extinguished. The filter current i.sub.s and the capacitor and inductor 
voltages u.sub.Cs and U.sub.Ls are shown in FIG. 2 for this oscillation, 
which takes place during the time interval t.sub.2 -t.sub.4. In the shown 
case, where the power frequency is 162/3 Hz and the frequency of the AC 
component (the fundamental tone) of the direct current i.sub.d is 331/3 
Hz, the filter C.sub.s -L.sub.s may be tuned, for example, to 50 Hz. The 
magnitude of the signal i.sub.o is chosen so that the thyristor T1 is 
ignited 1/4 cycle (of the natural frequency of the filter) before the time 
t.sub.3 at which the current i.sub.d has its maximum value. Therefore, the 
positive half cycle of the filter current i.sub.s is symmetrical around 
the time t.sub.3. 
At the time t.sub.6 the output signal from S2 becomes positive, NV2 becomes 
one, the flip-flop BV2 is set to one and the flip-flop BV1 is set to zero. 
A control signal SP.sub.T2 is then passed to the thyristor T2, which is 
ignited. The filter then oscillates for half a cycle of its natural 
frequency, and, thereafter, returns to its original position. The negative 
half cycle of the filter current i.sub.s (between t.sub.6 and t.sub.8) is 
symmetrical around the time t.sub.7 at which i.sub.d has its minimum 
value. 
At the time t.sub.10, T1 is again ignited, thereby starting a new 
operational cycle by providing a new positive half cycle of the filter 
current i.sub.s. Of course, the above-described operations are repeated in 
a cyclic fashion. Thus, it should be understood that the fundamental tone 
of the filter current i.sub.s will lie in phase with the current i.sub.d. 
If only the fundamental tones (with the frequency 331/3 Hz) of the AC 
components of i.sub.d, i.sub.s and u.sub.d are considered, the amplitude 
of u.sub.d is proportional to the difference between i.sub.d and i.sub.s. 
When the filter circuit L.sub.s -C.sub.s starts working, i.sub.s grows 
from zero and the amplitude of u.sub.d decreases from the value shown in 
FIG. 2 as i.sub.s grows. On the other hand, it is the AC component of 
u.sub.d that maintains the oscillation of the filter circuit. Thus, 
i.sub.s will automatically increase to a value close to zero. The 
resulting amplitude of the fundamental tone (331/3 Hz) in the AC component 
assumes a value in a stationary condition that is just capable of 
maintaining the oscillation in the filter circuit. If the losses of the 
filter circuit are low (such losses may be achieved in a practice in a 
simple manner), the amplitude of the fundamental tone is very low and the 
u.sub.d component with the frequency of 331/3 Hz is practically completed 
eliminated. 
In practice, it may be suitable to delay the ignition of the thyristors T1 
and T2 somewhat in relation to the times t.sub.2, t.sub.6, t.sub.10, etc., 
shown in FIG. 2 in order to obtain a safe supply of energy to the 
oscillating circuit, for example when the equipment is started. 
The filter for apparatus according to the invention should have a natural 
frequency that is greater than the pulse number times the power frequency. 
In the above-described example, the natural frequency of the filter was 50 
Hz and the pulse number (2) times the power frequency (162/3 Hz) was 331/3 
Hz. In prior art apparatus, for example as described in the German 
application No. 2,217,023, a filter is tuned to twice the power frequency, 
or 331/3 Hz for the present example. 
Because of the higher natural frequency for apparatus according to the 
invention, the inductance of the inductor L.sub.s, where the size of the 
capacitor C.sub.s is unchanged, is only 
##EQU1## 
of the inductance of the inductor for prior art apparatus. Thus, for the 
invention, the size of the inductor L.sub.s is at least halved, thereby 
resulting in a considerable reduction in the total weight and space 
required for filter components, which constitute a considerable portion of 
the total apparatus. It should be appreciated that the weight and space 
saving for apparatus operating in accordance with the invention is 
particularly significant where the apparatus is used on a vehicle. 
According to a preferred embodiment of the invention, thyristors T1 and T2 
are ignited at intervals (e.g. t.sub.2 -t.sub.6, t.sub.6 -t.sub.10), the 
lengths of which are half the time of a cycle (1/331/3 seconds) of the 
fundamental tone of the AC component of the direct current of the 
convertor. Thus, the second tone is completely eliminated from the filter 
current. Therefore, the filter current i.sub.s only has tones of the third 
and higher orders, which may be filtered off in a simple manner. 
The capacitor C.sub.p in FIG. 1 is dimensioned so as to reduce the 
variations in the direct voltage u.sub.d to the desired level. The 
variations are produced by resultant harmonic currents from the convertor 
SR, the load VXR, and the filter L.sub.s -C.sub.s. 
FIG. 1 shows how the load current of the convertor SR for controlling the 
thyristors T1 and T2 is sensed on the DC side of the convertor. 
Alternatively, this sensing may be performed on the AC side of the 
convertor. 
In the foregoing, the invention was described with respect to a single 
phase convertor having a pulse number of two and connected to a 
single-phase network. However, the invention may alternatively be applied 
with apparatus in which the convertor is a three-phase convertor, for 
example with a pulse number 6 and is connected to a three-phase network. 
Although the invention has been described above in connection with 
apparatus intended for vehicle operation, in which the DC network is an 
inverter apparatus according to the invention may also be employed in 
other fields as well. For such other applications, the DC network may 
consist of an arbitrary network or an object either supplied with or 
generating direct current. 
In the above-described apparatus, the power flows substantially from the AC 
network to the DC network, although the power may temporarily flow in the 
opposite direction, for example in connection with regenerative braking. 
However, the invention may be applied with apparatus in which the power 
flows either wholly or substantially from the DC side to the AC side of 
the convertor. 
In the apparatus described above, a single-phase AC network with the 
frequency 162/3 Hz has been selected as an example. However, the invention 
may also be applied with apparatus which is operated with either lower or 
higher frequencies. 
The invention may be embodied in other specific forms without departing 
from its spirit or essential characteristics. The present embodiments are, 
therefore, to be considered in all respects as illustrative and not 
restrictive, the scope of the invention being indicated by the claims 
rather than by the foregoing description, and all changes which come 
within the meaning and range of the equivalents of the claims are 
therefore intended to be embraced therein. 
The convertors SR and VXR of FIG. 1 may be formed by phase groups of the 
kind disclosed in the General Electric SCR Manual, 5th Edition. 1972, page 
384, FIG. 13.25. The convertor SR may thus be according to FIG. 13.28 of 
this reference, where the terminals designated "Load" are connected to the 
AC network (transformer TR) and the DC terminals are connected to the DC 
intermediate link. 
The inverter VXR may be built up from three phase groups, each according to 
the above-mentioned FIG. 13.25. 
The current sensing means IM of FIG. 1 may be, for instance, a simple 
current measuring shunt, that is, a small resistor traversed by the 
current to be measured. The voltage across the resistor will be 
proportional to the instantaneous value of the current.