Dual level error detection and correction employing data subsets from previously corrected data

A memory system that provides extra data bits without utilizing storage capacity. A first data word is fetched from memory and corrected to remove any single-bit errors. A second data word (which is a subset of the first data word as corrected) is then fetched, and new data correction bits (parity or ECC check bits) is generated for the second data word. Both the second data word and the newly-generated data correction bits are output. This structure amortizes the expense of in-system data correction over a greater data output, and over a smaller storage capacity relative to the data output.

CROSS-REFERENCE TO RELATED APPLICATIONS 
Reference is made to co-pending U.S. patent application Ser. No. 479,145, 
"On-Chip ECC With Optimized Bit And word Redundancy," by J. Barth et al 
and assigned to the assignee of the present invention, which relates to a 
memory chip with on-chip error correction code (ECC) circuitry, the 
teachings of which are incorporated herein by reference. 
Reference is also made to co-pending U.S. patent application Ser. No. 
479,145, "Optimized On-Chip ECC System," by J. Fifield and assigned to the 
assignee of the present invention, which relates to an on-chip ECC system 
that has been designed to minimize data delays without compromising data 
integrity, the teachings of which are incorporated herein by reference. 
BACKGROUND OF THE INVENTION 
1. Technical Field 
The present invention relates generally to fault--tolerant memory 
architectures, and more particularly to a memory system that utilizes 
on-chip fault recovery to enhance data rates. 
2. Background Art 
In memory systems, it has been a standard practice to provide some sort of 
fault recovery technique in order to enhance yield/performance. One of 
these techniques is redundancy, wherein faulty bit/word lines of cells are 
replaced with spares. Another of these techniques is ECC, wherein a data 
word is fetched from memory and corrected using generated syndrome bits 
that identify faulty bits within the data word. See for example the 
aforementioned U.S. patent application Ser. No. 479,145 and references 
cited therein. 
Another technique is parity, in which a parity bit is generated for a 
stored data word and is compared against a stored parity bit corresponding 
to that word. An example of such a system is shown in U.S. Pat. No. 
4,528,666, issued to Cline et al and assigned to Texas Instruments. 
Yet other memory systems have sought to combine ECC and parity techniques. 
Examples of such systems as shown in U.S. Pat. No. 3,568,153 to Kurtz et 
al and U.S. Pat. No. 3,573,728 to Kolankowsky et al, both of which are 
assigned to the assignee of the present invention. In both patents, new 
ECC check bits and parity bits are generated for the data as fetched from 
the memory. Comparing the stored check bits to those newly generated 
produces a string of syndrome bits that indicate the location of faulty 
bits within the data word. The syndromes are also used to correct the 
generated parity as a function of these faulty data bits, and the 
corrected parity is provided along with the corrected data word. 
In providing optimal fault coverage by combining different error coding 
techniques (such as ECC and parity), the designer must do everything 
possible to reduce the area and performance impact as much as possible. In 
other words, these coding techniques add cost by increasing chip size (the 
requisite logic circuitry consumes critical chip real estate that could be 
devoted to arrays) while decreasing performance (the data must pass 
through this logic before being output to the processor, adding delay 
directly to the data path). In order to reduce the cost of these error 
recovery techniques, the designer must minimize their impact by minimizing 
circuit complexity and reducing data delays. One way of reducing data 
delays is to provide more data in a given time. Neither of the above 
patents meet these goals; for example, both add appreciably to data delays 
because parity is generated and then data corrected as a function of the 
generated syndromes. 
SUMMARY OF THE INVENTION 
It is thus an object of the present invention to provide a fault-tolerant 
data processing system at minimum cost, without sacrificing fault 
coverage. 
It is another object of the invention to reduce the cost of fault tolerance 
by enhancing data rates. 
The above and other objects of the invention are achieved by taking 
advantage of the fault recovery system to provide extra data. 
Specifically, a first data word is read out along with error correction 
bits, and is corrected by a fault recovery system. Then data from the 
corrected data word is used to provide a second data word that has fewer 
bits than the first data word. Then error correction bits are generated 
for each of the plurality of second data words, and the newly-generated 
error correction bits are output along with the second data word. 
In this manner, the cost of on-chip fault recovery is reduced, because we 
rely on the correction of the first data word to provide new error 
correction bits that will not be faulty. Because the new error correction 
bits are generated as opposed to stored, the memory system provides more 
data than it is capable of storing. This enhances data rates, because more 
data is available in a unit of time from a given accessed portion of 
memory. 
In the invention, the second data word comprises eight of the bits of a 
corrected 128-bit ECC word. In a first embodiment of the invention, a 
single parity bit is generated for the second data word, and is provided 
as a ninth output bit. In a second embodiment of the invention, four of 
the second data words are combined, and new ECC check bits are generated 
for the combination. As a result, the 32 data bits are supplemented with 
seven check bits at the output.

DESCRIPTION OF THE BEST MODE FOR CARRYING OUT THE INVENTION 
1. RAM With On-Chip Error Correction 
The invention relates to a memory system (preferably a DRAM or SRAM) that 
has the capability of correcting data words fetched from storage, and then 
breaking down the corrected data word into a plurality of smaller data 
words for which parity or other correction bits can be generated. Thus, 
one element of the invention is to provide a means for fetching and 
correcting data words from memory. This section provides a description of 
the preferred single-chip embodiment of this element of the invention. 
With reference to FIG. 1, a general block diagram of a DRAM with on-chip 
ECC is shown. The DRAM array 10 is coupled to the ECC block 30 by a 
plurality of pre-data lines (PDLs) 15. Corrected data from the ECC 30 is 
sent to an SRAM 40, from which it is accessed through I/O 50. While the 
memory array of the invention could be of any configuration/density, it is 
preferred that array 10 comprise a 4 Mb quadrant of a 16 million bit (16 
Mb) DRAM chip. Thus, such a chip would have four separate ECC systems 
on-chip, one per quadrant. The memory cells are of the conventional "one 
device" DRAM type (i.e. an FET having its gate coupled to a word line, its 
drain coupled to a bit line, and its source coupled to the storage 
capacitor, wherein a sense amplifier coupled to the bit line compares the 
voltage from the capacitor to a reference voltage from a reference cell to 
determine the stored logic state), as generally described in U.S. Pat. No. 
3,387,286 issued June 1968 to Dennard and assigned to IBM. Although the 
cells can be construed using any one of a number of known techniques, it 
is preferred that substrate-plate cells be used (wherein the storage plate 
of the storage capacitor is formed by doped poly disposed in a trench that 
extends through epitaxial layers to the underlaying substrate that forms 
the charge plate--see U.S. Pat. No. 4,801,988, issued January 1989 to 
Kenney and assigned to IBM, the teachings of which are incorporated herein 
by reference). 
The DRAM array consists of 4096 word lines and 1096 bit line pairs. That 
is, in the memory array of the invention it is preferred to use the folded 
bit line configuration of U.S. Patent RE 32,708. The DRAM array receives 
control signals ROW ADDRESS STROBE (RAS) and COLUMN ADDRESS STROBE (CAS) 
from the memory controller. When RAS falls, memory operations commence, 
and address signals are buffered and decoded to couple two of the 4096 
word lines to the array sense amplifiers coupled to the bit lines 
(preferably the sense amplifiers comprise NMOS and PMOS cross-coupled 
devices). Subsequently, when CAS falls the input address signals are 
decoded to determine which bit lines are to be accessed. However, per the 
teachings of co-pending application Ser. No. 479,145, only a 1/8 decode is 
done at the array. That is, of the 1096 bit line pairs in the array 10, 
137 will be coupled to the pre-data lines 15. Thus, the array provides an 
error correction word ECW of 137 bits, of which 9 are check bits and 128 
are data bits. The remaining address bits at CAS falling are used to 
access one or more of the bits at the SRAM 40. 
The 137 pre-data lines 15 are driven by the cell data to provide inputs to 
the ECC 30. By "driven," we mean that the PDLs are precharged high, such 
that when the bit switches turn on to couple the PDLs to the selected bit 
lines, some of the PDLs are driven from the high voltage (e.g. 3.3 volts) 
to a low voltage (ground). While as a practical matter any data bussing 
configuration could be used, in practice it is preferred that the PDLs be 
disposed above (and criss-cross over) the bit lines of the DRAM array 10, 
so as to equalize capacitive coupling there between, co-pending 
application Ser. No. 479,145. 
The ECC block 30 utilizes an odd-weight Hamming code, which provides a 
double error detect, single error correct (DED/SEC) capability. While 
other codes (e.g., horizontal-vertical parity) could be used, odd-weight 
Hamming code is preferred because it provides the best error coverage at 
the lowest cost (for a more detailed comparison between odd-weight ECC 
codes and other codes, see the article by N. Jarwala et al entitled "Cost 
Analysis of On Chip Error Control Coding for Fault Tolerant Dynamic RAMs," 
Proceedings of the Seventeenth International Symposium on Fault-Tolerant 
Computing, Pittsb. Pa., Jul. 6-8 1987, pp. 278-283). 
While the operation of the ECC block will be discussed in detail with 
reference in FIG. 2, the general operation of the major functional blocks 
for the ECC will now be described with reference to FIG. 1. The ECC block 
30 comprises four main parts: syndrome generators 30S1-30S9, syndrome bus 
32, NOR gates 36, and XOR gates 38. 
As shown within syndrome generator 30S1, each generator (or "syndrome 
tree") is made up of three stage exclusive-OR (XOR) logic trees. The first 
stage 1S of the logic tree is made up of a first set of four-input XOR 
gates; the second stage 2S is made up of approximately four four-input XOR 
gates; and the final stage 3S is a single four-input XOR gate. Note that 
the syndrome generators 30S1-30S9 have different numbers of inputs 
(specifically 51, 59, 59, 59, 55, 59, 60, 47, and 50 respectively) to 
optimize the interconnect wiring layout. The three stages of XOR of one 
syndrome generator provide the parity of a subset of the one hundred and 
twenty eight data bits. This generated parity bit is then compared to a 
corresponding one of the stored check bits for that error correction word. 
The comparison operation, which is the XOR of a specific subset of PDL 
lines 15 and their corresponding stored check bits is executed by 1S, 2S 
and 3S. For the purposes of illustration, assume the arrowhead going into 
bus 32 is the result of this XOR operation. This XOR result is referred to 
as a syndrome bit, which is coupled to a respective line of a syndrome bus 
32. The syndrome bus 32 is 18 bits wide (it carries the true and 
complement of each of the 9 syndrome bits). The inputs to the first stage 
S1 of each syndrome generator 30S1-30S9 are subsets of the 128 data bits. 
Each syndrome generator receives a unique set of data bits, in accordance 
with the error correction code requirements. In other words, these XOR 
inputs are wired to calculate the parity of selected subsets of the 128 
bit data word according to a parity check matrix defining the error 
correction code used. 
The bits of the syndrome bus are provided to the inputs of 128 NOR gates 
36, one for each of the 128 data bits for the error correction word. The 
NOR gates work the same way as convention address decoders; the syndrome 
bits in combination indicate which of the 128 PDLs are carrying a bad bit. 
The outputs of the NOR gates 36 are sent to one input of XOR gates 38, 
each of which also receive a corresponding data bit. Should a given NOR 
gate indicate that its corresponding PDL is carrying bad data, the 
corresponding XOR 38 will simply invert the data on that PDL. 
The data bits as corrected by the ECC are then passed by data lines 35 at 
the output of the XORS 38 to SRAM 40. The SRAM (or data register) 40 
consists of a plurality of conventional four-device cross-coupled cells. 
From the SRAM, data is passed to the I/O pin 50 under the control of clock 
drivers (not shown) activiated during the CAS cycle to select and drive 
data from one or more of the SRAM cells. 
Referring now to FIG. 2, the interlocked ECC system utilized in the 
invention will be described in detail. In the description to follow, 
reference will be made to a "fetch" operation (wherein data is transferred 
from the DRAM array 10 through the ECC 30 to the SRAM 40) and to a 
"write-back" operation (wherein data is transferred from the SRAM 40 
through the ECC 30 to the DRAM array 10). The ECC circuitry utilized in 
the invention receives 128 data bits and 9 check bits for each ECC word. 
For ease of illustration, these bits are shown schematically as a single 
data bit DB and a single check bit CB. 
First, the fetch operation of the invention will be described with 
reference to both FIG. 2 and the waveform diagram of FIG. 3. Prior to 
initiation of the fetch cycle, both RAS and CAS are high, and the various 
clock drivers are in their restore state. The start of the fetch cycle is 
indicated by the falling edge of RAS. RAS going low causes the signal 
ARRAY RESTORE PHASE (ARN) to rise. ARN is used to take the ECC circuitry 
directed to fetching out of restore. Specifically, ARN rising drives the 
PC and PCNX generators 23, 25 high, which readies the syndrome generators 
30S1-30S9 as well as the NOR/XOR gates 36, 38 to receive inputs. At the 
same time, ARN rising enables the T/C receivers 20 to begin operation. At 
the falling edge of RAS, BUSRST 28 turns on to clamp the ECC busses 
21A-21D to ground via NMOS devices 28A-28D, respectively. 
As shown in FIG. 2, one of the PDLs from the DRAM array 10 is a dummy PDL 
(or DDL). The DDL provides the same general performance characteristics as 
the FDLs coupled to the memory cells. In other words, the DDL is coupled 
to a ground line via a device TA that has the same performance 
characteristics as the bit switch devices TB, TC that couple the normal 
PDLs to the bit lines b1A, b1B coupled to the selected word line w1 via 
memory cells MCA, MCB, respectively. Note that transfer devices TA-TC are 
enabled by the same signal T in practice, the signal enabling device TA 
would be derived from (e.g. a NOR of) all the transfer signals; signal T 
couples selected bit lines to the PDLs. The DDL conductor itself is formed 
the same time as the PDLs' thus, since it is driven by a device having 
approximately the same size as the PDL drivers TB, TC, it will have the 
same rise/fall times as the PDLs. 
The DDL is precharged high, as are the PDLs. When the bit switches TB, TC 
of the memory array are turned on by the column decoders providing signal 
T once CAS falls, the coupling device of the DDL TA turns on to discharge 
the DDL to ground. As a practical matter, the loading on the DDL is 
slightly greater than that of the normal PDLs such that the DDL simulates 
the worst-case delay associated with the normal PDLs being set to their 
respective logic states. The DDL is coupled to a large buffer 27 (actually 
a series of two conventional CMOS inverters) by means of a 2-input OR 
device having a second input coupled to logic restore phases PCNX, which 
is high during the early portion of the fetch cycle. The output PCR of 
buffer 27 is sent to the ECC T/C receivers 20. 
The ECC T/C receivers 20 are shown in detail in FIG. 4. The ARN signal 
rising turns off PMOS devices T4, T5, allowing the differential lines T,C 
to float. The S and SN signals from SGEN 26 (particularly, with S being 
high and SN being low) provide enable inputs to the CMOS transmission 
gates TG1-TG4. As shown in FIG. 5, SGEN 26 is enabled by PCNX, and 
generates buffered S, SN outputs when ODDL (the unbuffered version of PCR, 
taken from the output of OR gate 11) falls. Referring back to FIG. 4, the 
receiver 20 is not completely enabled until it receives the signal PCR 
from the dummy PDL that turns on NMOS T3. With T3 on, the CMOS inverter 
T1, T2 is activated, such that if the data from the PDL input is highs 
line T will be set to a low state and line C will be set to a high state, 
which will be passed by the respective CMOS transmission gates TG1-TG4 to 
the inverting outputs, such that ECCT will be high and ECCC will be low. 
Thus, by virtue of the interlocking function provided by the DDL, the ECC 
T/C receivers 20 will not set the ECC busses 21A, 21B until the PDL inputs 
thereto are valid. By "valid," we mean that the PDL has been pulled 
sufficiently low (at least to the "maximum positive down level (MDDL)," 
which is the highest voltage indicating a binary logic state of 0) such 
that the data thereon can be reliably read. In the case of CMOS, the MPDL 
is on the order of 0.7 volts, and the "minimum positive up level (MPUL)" 
(the lowest voltage indicating a binary logic state of 1) is on the order 
of 1.4 volts. This interlock prevents the input to ECC of erroneous data 
due to setting the ECC busses prior to adequate signal development on the 
PDLs. At the same time, because the remaining circuitry (the internal T/C 
lines, the CMOS transmission gates) of the receivers 20 are enabled just 
prior to the PCR signal by ODDL, once the PCR signal rises the receiver 
can operate without further delay. 
Note from FIG. 2 that the ECCT, ECCC outputs of ECC T/C receivers 20 
(corresponding to ECC busses 21A, 21B are held at ground by BUSRST during 
the early part of the fetch cycle. When PCR rises, the BUSRST generator 28 
lowers the BUSRST signal, such that the ECC bus lines 21A, 21B can be 
driven by the ECCT, ECCC outputs of the receivers 20. 
As previously described, the data from the ECC busses is passed to the DCVS 
syndrome generators 30S1-30S9, which in turn provide the syndromes onto 
the 9-bit syndrome bus 32. While as a practical matter the XOR gates 
within syndrome generators 30S1-30S9 could be provided using any 
conventional logic, it is preferred that differential cascode voltage 
switch (DCVS) logic be utilized. DCVS is described in detail in U.S. Pat. 
No. 4,570,084, issued February 1986 to Griffin et al, the teachings of 
which are incorporated herein by reference). FIG. 6 is a circuit diagram 
of a DCVS 4-input XOR. Transistors T7 through T2O form the N-type 
combinational logic of a 4-input XOR function with differential inputs AT, 
AC to DT, DC. Since phase PC is driven high at the start of the fetch 
cycle, the differential output of the XOR of differential inputs A,B,C and 
D from T/C receivers 20 is driven to nodes Qt and Qc by inverters formed 
by T21, T22, T25 and T26. Leakage protection is provided by soft latching 
action of T23 and T27. Note that the syndrome generators are self-timed; 
that is, there are no enable/trigger clock signals that activate the 
syndrome generators as there are for the T/C receivers 20. The syndrome 
generators are effectively synchronized by clocking of the the T/C 
receivers. That is, because the operation of the T/C receivers insures 
that the T/C inputs to the syndrome generators are valid, there is no need 
for independent clocking for the syndrome generators. 
The NOR/XOR logic 36, 38 are shown in more detail in FIG. 7. Note that the 
output of the NOR node within block 36 defined by NMOS transistors T1X-T9X 
is enabled by NMOS T31, which receives an interlock pulse SYNREDY from 
SYNREDY generator 24. The ERRC output will rise at the falling edge of 
NORNODE. On the other hand, ERRT will rise if NORNODE remains high, and 
only when SYNREDY enables clocked inverter formed by T29-T30. As shown in 
FIG. 8, the SYNREDY generator 24 produces an output when one of the 
syndrome bits SC, ST from syndrome bus 32 rise to indicate that syndrome 
bus 32 is active. Note that the relative sizes of devices T32-T34 are set 
such that the SYNREDY pulse is not generated until the SYNDROME inputs to 
T1X-T9X of the NOR node of NOR gate 36 are valid and NORNODE is at its 
valid level. Specifically, these devices are significantly longer/wider 
than the devices that make up the NOR node, to introduce a discrete delay. 
Briefly, when either SC or ST rise, the corresponding transistor T32, T33 
will turn on, coupling the gate of PMOS T35 low, such that the SYNREDY 
output rises through inverter T35, T36. Thus, the generation of the ERRT, 
ERRC pulses is interlocked with the generation of syndrome data. More 
specifically, the output of the NOR node within block 36 is not enabled 
until the SYNREDY pulse rises to indicate that sufficient time has passed 
since the syndrome bits were valid to assure proper operation of the NOR 
decode. Again, this prevents premature outputs from the NOR decoder from 
erroneously indicating an error condition. 
The remaining operations of the system (i.e., firing of the XORs within 
block 38 to correct the bad bit, and passage of the data bits as corrected 
to the SRAM registers) are self-timed, relying on the above self timed 
nature of DCVS logic gates. Specifically, the XOR 38 receives ERRT, ERRC 
from NOR 36 and ECCT, ECCC from the T/C receivers 20, and carries out an 
XOR operation to provide the outputs SRT, SRC that are sent to the data 
registers 40. 
After the fetch operation as described above, the ECC circuitry must be 
restored so that it can be driven quickly during a subsequent write-back 
cycle. This restore is triggered by the SRV generator 27. The SRV 
generator as shown in FIG. 9 receives its inputs SC, ST from the bus 32B. 
The SRV and SRVF outputs of generator 27 will rise when the syndrome bits 
on bus 32B are valid by turning on one of devices T37, T38. These SRV and 
SRVF signals are used for different purposes. First, SRVF rises to enable 
the clock driver (not shown) that controls the transfer of corrected data 
latched by the SRAM cells to the I/O pads of the chip. Because these clock 
drivers add considerable inverter delays, SRVF is generated before the 
SRAM nodes are actually valid. That is, the delay associated with the 
clock drivers is factored into the SRVF timing, such that by the time the 
clock drivers enable data transfer from the SRAM, the data therefrom will 
be valid. SRV rises approximately 3/10 nanoseconds after SRVF, to restore 
the ECC circuitry. Again, although SRV is generated prior to the SRAM 
nodes being actually valid, the inverter delays associated with restoring 
the ECC circuitry are such that by the time the circuit outputs are driven 
to their restore states the SRAM nodes will be valid. Thus, ECC restore is 
timed to occur at the end of the RAS cycle, and is interlocked to the 
provision of valid data to the SRAM register blocks. This prevents the ECC 
from being restored until it has had an opportunity to process the DRAM 
data. Moreover, the driver devices of SRV are sized such that the SRV 
signal will rise after the SRT and SRC outputs of the ECC error indicator 
and corrector blocks have updated the SRAM cells with the correct data. 
As shown in more detail in FIG. 10, SRV rising will turn turn off the PCNX 
output of generator 23, to disable the ECC error detection and correction 
circuitry 36, 38. Specifically, when SRV rises NMOS T40 will turn on, 
coupling node PCOFF to ground. PCOFF forms the input to four inverter 
stages I1-I4, that buffer the PCNX signal to drive the large load 
presented by the NOR/XOR block 36, 38. Thus, the PCNX output is driven 
low. Referring back to FIG. 7, note that the PCNX input disables the ERRC, 
ERRT outputs by turning off devices T41, T42; disables the NOR node by 
turning on device T43; and disables the XOR drivers by turning on PMOS 
devices T43-T44. The falling edge of PCNX also causes the S, SN outputs of 
SGEN 26 to change state, which turns off the CMOS transmission gates 
TG1-TG4 of the ECC T/C receivers 20. The falling edge of PCNX also turns 
off the OR gate 11, such that PCR falls to both disable the PDL inputs of 
the ECC T/C receivers 20 and restore the ECC busses 21A, 21B to ground by 
raising the output of the BUSRST generator 
Finally, SRV also drives the output of PC generator 25 low, disabling the 
DCVS logic of the syndrome generators (See FIG. 6). 
The write-back cycle will now be described. With reference to FIGS. 2 and 
3, the start of the write-back cycle is indicated by RAS rising. RAS going 
high restores the SRV generator 27, causing both SRV and SRVF outputs to 
fall to ground shortly after RAS rises. The rising edge of RAS also serves 
to reset the BUSRST generator 28, such that the ECC busses 21A, 21B are 
disconnected from ground by turning off the NMOS devices 28A-28D. The fall 
of SRV serves to pull the PC generator 25 out of restore, to again enable 
the syndrome generators 30S1-30S9. 
As shown in FIG. 11, the rising edge of RAS also serves to turn on device 
T45 of the write generator WGEN 29. Since ARN is still high at this time, 
the node WG is pulled low, pulling the W output high and the WN output 
low. These signals are fed to SRAM buffers 29A-29D. During a write-back 
cycle, the SRAM buffers 29A, 29B receive data bits SRT, SRC from each of 
the SRAM cells. This reception is enabled by the W, WN signals, which 
cause the SRAM buffers 29A, 29B to pass the SRT, SRC bits to respective 
lines within the ECC bus 21A. However, in the case of the SRAM buffers 
29C, 29D, note that their inputs are wired to ground and Vdd, respectively 
(that is, when enabled by the W, WN signals, the SRAM buffers 29C, 29D 
pass ground and Vdd, respectively, to the ECC bus lines 21B). In 
combination, these signals provide a logical input of "0" to the ECC bus 
21B, such that all the check bit inputs to the syndrome generators 
30S1--30S9 are zero. 
This is done because the syndrome generators are used to generate new check 
bits for the data during the write-back cycle. As previously described, 
the input data bits are applied to the ECC bus 21A. As described with 
reference to the fetch cycle, the syndrome generators 30S1-30S9 will 
generate check bits based on this data. However, because all of the input 
check bits are "0," the generated check bits will be directly passed to 
form new check bits (that is, unlike the fetch operation, a comparison 
between the newly-generated check bits and the old check bits is not 
carried out). 
Once the new check bits are generated by syndrome generators 30S1-30S9, 
they are passed onto the syndrome bus 32 as described for the fetch cycle. 
As the check bits are passed to the bus, one of them is sent to the check 
bit read (CBR) generator 60. The CBR generator is configured the same way 
as the syndrome ready SYNREDY generator 24, and it performs the same 
function of providing an output (in this case, by switching its output 
states such that CKBRED is high and CKBREDN is low) when the bits on the 
syndrome generator 24 have risen sufficiently such that their logic states 
can be reliably processed. These signals enable the write-back inverter 62 
to pass the newly-generated check bits from the syndrome bus 32 to the PDL 
corresponding to the ECC bus 21B. 
Write-back inverter 62 is shown in detail in FIG. 12. When CBRED is high, 
NMOS T47 is on. If the input check bit CB is high, NMOS T48 will turn on, 
pulling the input to T49 and T50 low. Thus, T47 will be isolated from 
ground while T50 turns on to clamp node CBH high, so as to supply a high 
signal to the PDL output. If the check bit CB is low, device T47 is 
connected to ground via NMOS T50, such that node CBH (which was set high 
during standby by signal PC) will be pulled to ground so as to provide a 
high signal to the PDL. Note that when CKBREDN falls (and CKBRED rises) at 
the end of the write-back cycle, NMOS T51 rises and PMOS T52 falls, to 
disable the inverter output. Thus, the new check bits are written to the 
corresponding PDL. Similarly, write-back inverter 64 couples the 
"complement" signal on ECC bus 21A (i.e., the input to the ECCC line of 
the T/C receiver 20) to the corresponding PDL. 
An interlock is used to make sure that the DRAM arrays receive accurate 
data. In this case, the same dummy data line DDL is driven high, 
indicating to signal that the PDLs as driven by the write-back drivers 62, 
64 have valid logic states. A dummy write-back driver 66 is coupled to the 
DDL, and is configured the same as the write-back driver of FIG. 12 except 
its CB input is permanently tied to GND. When the CBRED, CBREDN signals 
are generated, the dummy write-back driver 66 will drive the dummy data 
line up to Vdd with the same general timing as the other write-back 
generators. The DDL output is sent to clock drivers (not shown) that will 
provide enable signals to the sense amplifiers of the bit lines within the 
DRAM. Thus, similarly to the PDL--ECC T/C receiver data transfer, the 
transfer of data from the ECC circuits to the PDLs is interlocked by a 
dummy PDL, insuring that the data will not be read before it is valid. 
The system is reset after write-back as follows. The rise of the DDL also 
enables the clock generator (not shown) that produces ARN to switch state. 
When this happens, the PC generator 25 is restored low, to bring signal PC 
low to turn off the syndrome generators 30S1-30S9. The fall of ARN also 
restores the CBR generator 60, disabling the write-back inverters 62 and 
64 as well as the dummy write-back inverter 66. The fall or ARN also 
resets the write generator WGEN 29, to disable the SRAM buffers 29A-29D. 
Thus, all the circuitry is reset at the end of the cycle, facilitating 
enhanced performance. 
2. Remaining Elements of the Invention 
Section 1 above describes the preferred system and process by which data is 
fetched from memory and corrected to provide "clean" data. This Section 
discusses alternate embodiments of the remaining elements of the 
invention, by which subwords of the "clean" data are fetched and new 
correction bits are generated therefor. 
FIG. 13 is a block diagram of a first embodiment of the invention. The RAM 
Storage and Error Correction (EC) block 100 is the same as the circuitry 
described in Section 1 above the data lines 35 feeding the SRAM 40 as 
shown in FIG. 1. Similarly, the data lines 35 and SRAM 40 are the same as 
the like-numbered elements of FIG. 1. 
In FIG. 13, each of the 128 SRAM cells within SRAM 40 has its own output. 
As such, 128 output lines 50A originate from SRAM 40; in FIG. 1, the 128 
SRAM cells shared a single output line 50. The 128 output lines pass 
through a 1:16 decoder 60, such that only 8 of the 128 SRAM outputs are 
sent to the parity generator 70. The 1:16 decoder receives column address 
signals A7-A9 to select which eight of the 128 data signals are to be 
passed to the parity generator. The decoder 60 comprises 16 conventional 
NOR decode nodes, each of which receive different combinations of the 
true, complement versions of the address signals. Each NOR node has an 
output enable line connected to the gate electrodes of PFET transmission 
gates coupled between a group of eight of the output lines 50A and the 
eight output lines 65. In any given access cycle, only one of the 10 NOR 
decoders will be active, such that only one group of 8 of the 128 SRAM 
outputs will be passed along lines 65 to the parity generator 70. 
FIG. 14 is a detailed circuit diagram of the parity generator 70. The 
parity for the group of eight output lines 65 (inputs) A-H is generated by 
the exclusive OR (XOR) gates XOR0-XOR6. The four 2-input XOR gates XOR0-3 
have their outputs XOR-ed by XOR4-6 to produce the output parity bit 80P. 
Note that XOR gates with greater than 2 inputs can be used to optimize the 
design for speed. 
Thus, the nine outputs 80 and 80P from parity generator 70 to the data 
processing system utilizing the memory system of the Invention include a 
bit 80P that is generated without taking up extra memory space. At the 
same time, this extra bit is provided without compromising the fault 
recovery capabilities of the memory system. That is, the 128 data plus 9 
check bit ECC system optimizes fault recovery while minimizing cost; by 
generating extra bits from a sub-word of the 128 corrected data bits, 
these efficiencies are retained. Since the memory storage 100 in FIG. 13 
is actually a 4 Mb quadrant of a 16 Mb device, the nine outputs here would 
repeat four times (1/quadrant), such that the entire chip has 36 outputs. 
Obviously, the size of the RAM storage in 100 can be any size. In this 
example it was 4 Mb but could be 16 Mb, 64 Mb, etc. 
Note the UE signal from the storage/error correction block 100 to the 
parity generator 70. The circuitry described in Section 1 above is capable 
of correcting single errors in the fetched data word. At the same time, it 
is capable of detecting when there are two errors in a data word, even 
though it is not capable of correcting both errors. Should a double error 
be detected, the logic produces an uncorrectable error (UE) signal, 
indicating that the fetched data word contains two errors and hence cannot 
be corrected. The UE signal is fed to the parity generator 70 to 
complement it. That is, we do not want to generate the correct parity bit 
for bad data. The UE signal is used to generate invalid parity, indicating 
the presence of faulty data. 
FIG. 15 is a block diagram of a second embodiment of the invention. 
Elements in FIG. 15 that have the same reference numerals as those of FIG. 
13 have the same structure/function as those elements. In FIG. 15, the 128 
SRAM outputs 50 are divided into 16 8-bit sections, each of which forms 
the input to a separate parity generator 70A-70P. That is, instead of a 
single parity generator servicing all of the SRAM outputs as in FIG. 13, 
here each group of eight SRAM outputs has its own dedicated parity 
generator. The 9-bit outputs 75 of the sixteen parity generators 70A-70P 
are then provided to a 1:16 decoder 60, to select a single set of outputs 
at the output 80, 80P. A particular aspect of this embodiment of the 
invention is that because the parity bit is generated for each group of 
eight bits from the SRAM, more than one group can be output at the same 
time without a performance penalty. That is, instead of a 1:16 decoder, we 
could use a 1:8 or 1:4 decoder, which would output 15 and 32 bits at a 
time from each quadrant of the chip, without increasing the cycle time due 
to multiplexing a single parity generator amongst the selected bit groups. 
In FIG. 15, note that the UE signal is now sent to all of the parity 
generators 70A-70P. 
FIG. 10 is a block diagram of a third embodiment of the present invention. 
The RAM and ECC blocks have been omitted for clarity. Here, instead of 
showing a single quadrant of the chip, the output system for the entire 
chip is shown. Similarly to the first embodiment of the invention shown in 
FIG. 13, the 128 outputs of the SRAMs 40A-40D are sent to respective 1:16 
decoders 60A-60D to select a single group of eight outputs from each 
quadrant. However, instead of generating a parity bit for each group as in 
FIG. 13, here the four groups of eight outputs are combined to form a 
32-bit data word, which is processed through a Hamming ECC check bit 
generator 110 (having the same circuitry and operation as the check bit 
generator discussed in Section 1 above) to yield seven check bits. That 
is, the check bit generator 110 works the same general way as that 
described in conjunction with FIGS. 1-12, except that it works off of a 
different Hamming code, such that a 32-bit data word yields seven check 
bits. The 32 data bits are passed through lines 120D to the output; the 
seven check bits are passed through lines 120C to the output through seven 
XOR gates 132. The XORs receive a control signal UEC (or "UE Composite") 
from an OR gate 130, which receives UE signals from each of the four 
memory array quadrants. Is one of the quadrants sees an uncorrectable 
error, the corresponding UE signal will rise, causing the UEC signal to 
rise. UEC high causes one or more of the XORs to invert the value of the 
check bits to indicate faulty data. As a result, the chip has a 39-bit 
output, of which 32 bits originated from storage and 7 were generated. Put 
another way, in this embodiment slightly less than 20% of the output bits 
were generated as opposed to stored, such that the chip is 20% more area 
efficient. 
FIG. 17 shows a fourth embodiment of the invention. Similarly to FIG. 16, 
an ECC tree that generates seven check bits based on 32 data bits is used. 
However, instead of having one of these trees for the chip, each quadrant 
has its own, wherein the outputs are multiplexed to the chip outputs by a 
decoder. Within a quadrant A, a 1:4 decoder 65A selects 32 of the 128 
outputs from SRAM 40A. These 32 bits are the 32 data bits fed to the ECC 
tree 110A, which outputs data bits 120D and check bits 120C as discussed 
above. Note that the other quadrants B-D have similar circuitry. As a 
result, each quadrant produces its own group of 39 bits 140A--140D. A 
second 1:4 decoder then selects which of these four groups of 39 bits are 
to be passed to the outputs 80, 80C. 
Accordingly, in the present invention the area/performance penalty 
presented by using error correction techniques is reduced by amortizing 
such penalties over more output bits. By increasing the number of output 
bits, access times can be decreased because less cell address cycles are 
needed. At the same time, this effect is enhanced by generating output 
bits instead of storing them. This makes the memory more area efficient, 
because less storage is required to output a given number of bits. 
Various modifications can be made to the structures and teachings rendered 
above without departing from the spirit and scope of the present 
invention. For example, while Hamming codes have been used to correct the 
data originally fetched from memory, other techniques such as 
horizontal--vertical parity could be used. This also applies to the check 
bits generated for the second data word selected from the "clean" data. 
While the invention has been described as embodied on a single chip, it 
could also be practiced on a memory card or other substrate populated with 
conventional memory chips, wherein their respective outputs are 
multiplexed into a central error correction system that provides the 
"clean" data to the circuitry of the invention.