Mixer circuit, receiver circuit, and frequency comparison circuit

A mixer circuit contains a first terminal and a second terminal to which a first differential input signal is applied, and an active element switching a short-circuit between the first terminal and the second terminal. By driving the active element by a second differential input signal having a predetermined frequency, the first terminal and the second terminal are intermittently short-circuited at twice the frequency of a predetermined frequency.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention generally relates to mixer circuits and communication apparatuses employing a mixer circuit, and more particularly to a direct demodulation circuit and a communication apparatus employing a direct demodulation circuit.

2. Description of the Related Art

Regarding a communication system having a low-frequency component in a baseband of a broadband such as WCDMA (Wideband Code Division Multiplex Access), it has recently become desirable to use a direct demodulation circuit without any external filter part so as to realize miniaturization, low cost and low electric power consumption with respect to a receiver. A direct demodulation circuit has the problem that a weak local current mixed into an RF signal is detected together with the RF signal and there arises a DC offset.

It is possible to reasonably construct a small subharmonic pumping mixer employing an APDP (anti-parallel diode pair) in which diodes are connected in anti-parallel. Also, when the frequency of the local current is set as half of a carrier frequency, it is possible to perform direct demodulation. At the same time, even if the local current with half of the carrier frequency leaks an RF signal, it is impossible for the local current to return and become a DC component. As mentioned above, the subharmonic pumping mixer has the advantage that the DC offset is not generated in principal. Thus, the subharmonic mixer can be used as a direct demodulation mixer.

FIG. 1shows a structure of a subharmonic pumping mixer.

An RF is supplied to an APDP10formed with diodes D1and D2via a filter formed with an L1and a C1. In a down-conversion mixer, a local Lo is additionally supplied to the APDP via a filter formed with an L2and a C2. When the APDP demodulates the RF, the demodulated signal is supplied to a terminal of a resistor R via a low-pass filter formed with an L3and a C3.

FIG. 2shows a characteristic of the subharmonic pumping mixer inFIG. 1with respect to a conversion gain. As is shown inFIG. 2, the conversion gain has a maximum of no more than about −20 DB within a range in which local power is increased.

FIG. 3shows a characteristic of the subharmonic pumping mixer inFIG. 1with respect to a noise index. As is shown inFIG. 3, the noise index has a minimum of no less than about 20 DB.

As mentioned above, while a conventional subharmonic pumping mixer has the advantage that it is possible to implement the subharmonic pumping mixer with low cost and no DC offset arising due to a leak of a local signal, the subharmonic pumping mixer has some problems with respect to the gain and the noise characteristics.

Therefore, it is necessary to develop a small size, low cost and a low electric power consuming mixer circuit capable of suppressing any DC offset caused by a local leak signal and achieving better gain and noise characteristics.

SUMMARY OF THE INVENTION

It is a general object of the present invention to provide a mixer circuit and others in which at least one problem caused by the above limits and disadvantages of the related art are virtually eliminated.

A mixer circuit according to the present invention comprises: a first terminal and a second terminal to which a first differential input signal is applied; and an active element switching a short-circuit between the first terminal and the second terminal, wherein by driving the active element by a second differential input signal having a predetermined frequency, the first terminal and the second terminal are intermittently short-circuited at a predetermined multiple frequency of the predetermined frequency (for example, twice the predetermined frequency).

For example, the active element is a MOS transistor. Also, the active element is formed with two active elements connected in parallel between the first terminal and the second terminal, which are driven with a reversed phase signal of the second differential input signal.

The above mixer circuit can be realized at a small size, a low cost and with low electric power consumption. Additionally, since a signal having half of a frequency to be detected (carrier frequency) is used as a local signal, it is possible to prevent a DC offset caused by a local leak signal and achieve good gain and noise characteristics because the signal path is always maintained at less than a predetermined impedance.

Additionally, a receiver circuit according to the present invention comprises: a phase comparison circuit containing an active element switching a short-circuit between two terminals to which a first differential input signal is applied, the phase comparison circuit, by driving the active element by a second differential input signal having a predetermined frequency, intermittently switching on a short-circuit between the terminals at a predetermined multiple frequency of the predetermined frequency; a feedback circuit generating the first differential input signal based upon a signal arising between the terminals; and a latch circuit latching the second differential input signal in sync with the first differential input signal.

According to the above-mentioned receiver circuit, since the mixer circuit according to the present invention is used as a phase comparison circuit, it is possible to implement a phase comparison circuit achieving small size, low noise and high speed.

Additionally, a frequency comparison circuit according to the present invention comprises: a first phase comparison circuit containing a first active element switching a short-circuit between two terminals to which a first differential input signal is applied, the phase comparison circuit, by driving said first active element by a second differential input signal having a predetermined frequency, intermittently switching on a short-circuit between said terminals at a predetermined multiple frequency of said predetermined frequency; a second phase comparison circuit containing a second active element switching a short-circuit between two terminals to which a third differential input signal is applied, the phase comparison circuit, by driving said second active element by said second differential input signal having a predetermined frequency, intermittently switching on a short-circuit between said terminals at a predetermined multiple frequency of said predetermined frequency; and a multiplier multiplying a phase comparison result of said first phase comparison circuit and a phase comparison result of said second phase comparison circuit.

According to the above frequency comparison circuit, when the mixer circuit according to the present invention is used as a phase comparison circuit, it is possible to implement a frequency comparison circuit with a small size and low electric power consumption.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIG. 4is a circuit diagram illustrating a fundamental structure of a mixer circuit according to the present invention.

A mixer fundamental circuit20inFIG. 4contains NMOS transistors21and22serving as active elements having a switching function and capacitors C1and C2. An RF signal is supplied as a differential signal to terminals a+and a−of the mixer fundamental circuit20via a resistor Ri. A local signal having twice the period of a carrier wave is supplied to a gate terminal b+of the NMOS transistor21, and a complementary signal of the local signal supplied to a gate terminal b+is supplied to a gate terminal b−of the NMOS transistor22. The mixer fundamental circuit20outputs a demodulated signal to both terminals of a resistor R1. It is noted that the NMOS transistors21and22are not restricted to NMOSs and may generally be FETs (Field Effect Transistors).

Regarding the mixer fundamental circuit20inFIG. 4, a variation of the RF signal is transmitted to the output terminal either by short-circuiting the differential signal wherein the NMOS transistor21or22switches on in a span in which the gate terminal b+or b−is HIGH or by capacitive coupling with the capacitors C1or C2in the other period.

FIG. 5is a circuit diagram illustrating a circuit equivalent to the mixer fundamental circuit20inFIG. 4.

The equivalent circuit20inFIG. 5, in which a differential input signal is illustrated as a single signal, contains switches S1and S2and a capacitor Cp. An RF signal is supplied to an input terminal a. The switch S1is switched on while a local signal is HIGH. The switch S2is switched on while a complementary signal of the local signal is HIGH.

FIG. 6is a diagram illustrating an operation of the equivalent circuit inFIG. 4.

As is shown inFIG. 6, the RF signal is supplied to the terminal a, and the local signal and the complementary signal of the local signal are supplied to the terminals b+and b−. A threshold Vth at which the switches S1and S2are switched on for the local signal and the complementary signal of the local signal (which corresponds to a threshold of the NMOS transistors21and22inFIG. 4) is illustrated. As is shown inFIG. 6, in a span in which the local signal or its complementary signal is more than the threshold Vth, an output terminal o inFIG. 5is short-circuited at a ground potential through the switch S1or S2that is switched on, whereby an output current io becomes zero. In the other period, since the switch S1or S2is switched off, a voltage variation of the input RF signal is output as the output current io. As a result, a direct demodulation is realized.

An orthogonal digital modulation signal Re[(I+jQ)ejωct]=I cos ωct−Q sin ωct is set as an input signal. A local signal having twice the period of a carrier signal with a frequency ωcbecomes HIGH once every two periods, whereby the corresponding switch S1is switched on. On the other hand, the complementary signal of the local signal becomes HIGH once every two periods of the carrier signal, whereby the corresponding switch S2is switched on. Since the switches S1and S2are alternately switched on, a signal line transmitting the input signal is short-circuited to the ground in a predetermined interval for each period of the carrier signal.

If a load resistance R1is assumed to be enough small, an instantaneous current I0-inst generated on the output terminal o in a state of not being switched on is as follows;
I0-inst=Cp(dVa/dt).
From the instantaneous current, the output current i0per a unit time is solved as follows;

Thus, if φ0=0 and the value Δφis enough smaller than 2π, the current per unit time is given as
i0=(CpQΔφ)/2π.

If φ0=π/2 and the value Δφis enough smaller than 2π, the current per unit time is given as
i0=(CpIΔφ)/2π.

As mentioned above, when the start time when the switch begins to be switched on is set to a zero phase and a π/2 phase, it is possible to obtain an orthogonal baseband signal.

FIG. 7is a circuit diagram illustrating the first embodiment of the mixer circuit according to the present invention.

A mixer circuit30contains two mixer fundamental circuits20as shown inFIG. 4, NMOS transistors31through38, bipolar transistors39through44, capacitors45and46, and resistors47and48. The two mixer fundamental circuits20receive an RF signal.

FIG. 8is a diagram illustrating a phase relationship of local signals being supplied to the mixer fundamental circuits20and the NMOS transistors31through38.

As is seen inFIG. 7andFIG. 8, one of the mixer fundamental circuits20receives a local signal lo0and its complementary signal lo5, and the other receives a local signal lo3and its complementary signal lo7. From the viewpoint of frequencies of the local signals, the local signal lo0is misaligned to the local signal lo3by a phase of 90°. Thus, from the viewpoint of the frequency of the carrier signal RF, it is concluded that there is a discrepancy corresponding to a phase of 180°. If such two mixer fundamental circuits20are provided, one mixer circuit20is constructed to output a demodulation signal from the other mixer fundamental circuit20while two outputs of one of the two mixer fundamental circuits are short-circuited. In such a construction, it is possible to balance the entire circuit and prevent noise from generating.

Signals from the mixer fundamental circuits20are supplied to a current sense amplifier formed with bipolar transistors39through44being at the next stage via the NMOS transistors31through38. Here, the NMOS transistors31through38are provided to deliver the outputs from the mixer fundamental circuits20to the current sense amplifier while a demodulation signal is supplied as an output in the individual mixer fundamental circuits20, that is, the switches in individual mixer fundamental circuits20are not short-circuited.

The current sense amplifier formed with the bipolar transistors39through44has a small enough input impedance. The current sense amplifier receives the demodulation signals from the mixer fundamental circuits20as current signals, amplifies the received signals and outputs the amplified signal as voltage signals. The output voltage signals are integrated with the capacitors45and46, thereby obtaining baseband signals BB+ and BB−.

Here, in order to realize an orthogonal demodulation, a circuit having the same structure as the circuit shown inFIG. 7is added as a second circuit. The second circuit receives local signals lo2, lo6, lo4and lo8whose phases are misaligned by 90° to those of the local signals lo1, lo5, lo3and lo7supplied to the first circuit, respectively. As a result, it is possible to obtain demodulation signals whose phases are misaligned by 90° with respect to the first circuit and realize an orthogonal demodulation.

FIG. 9is a circuit diagram illustrating a second embodiment of the mixer circuit according to the present invention.

An RF signal is supplied to bases of the transistors55through58, and amplified current signals are supplied to two switches formed with the NMOS transistors51and52and the NMOS transistors53and54separately. The switch circuits correspond to those of the mixer fundamental circuit20formed with the NMOS transistors21and22. In the second embodiment shown inFIG. 9, a current signal corresponding to the RF signal is directly supplied to the switch circuits without any capacitive coupling via a capacitor.

The local signals lo1and lo5are supplied as switch control signals to the switch circuit formed with the NMOS transistors51and52. The local signals lo3and lo7are supplied as switch control signals to the switch circuit formed with the NMOS transistors53and54. Phase relationship of the local signals is similar to the phase relationship shown inFIG. 8. The individual switch circuits are short-circuited once a period of the carrier signal, and there arises a phase difference of 180° with respect to a short-circuit between two of the switch circuits.

A cascode circuit formed with the transistors59and60receives an output in an RF+ side of one switch circuit and an output in an RF− side of the other switch circuit. A cascode circuit formed with the transistors61and62receives an output in an RF− side of one switch circuit and an output in an RF+ side of the other switch circuit. The capacitors63and64integrate a voltage output from the cascode circuits to obtain baseband signals BB+ and BB−.

As mentioned above, one of the cascode circuits and a baseband signal output circuit part receive the output in the RF+ side of the one of the switch circuits and the output in the RF− side of the other of the switch circuits. Thus, even if one of the switch circuits is short-circuited, a demodulation signal is supplied from the other of the switch circuits.

The baseband signals BB+ and BB− are supplied to the common mode feedback circuit73via the NMOS transistors67and68and the resistors69and70. Based upon a feedback signal, the common mode feedback circuit73adjusts the voltage of the RF signal via the resistors71and72, thereby operating the mixer circuit inFIG. 9at an operating point of an appropriate voltage. As a result, it is possible to guarantee an appropriate operation even when the voltage of the current source varies, a characteristic of the device is changed, and noise is mixed with the voltage of the current source.

FIG. 10shows a characteristic of a mixer circuit according to the second embodiment of the present invention with respect to the conversion gain. WhenFIG. 10is compared toFIG. 2, it is observed that the mixer circuit according to the present invention achieves a positive gain, that is, a far larger conversion gain than a conventional subharmonic mixer.

FIG. 11shows a characteristic of the mixer circuit according to the second embodiment shown inFIG. 9with respect to a noise index. WhenFIG. 11is compared toFIG. 3, it is observed that the mixer circuit according to the present invention achieves the minimal noise index of about 10 dB, that is, a far better noise characteristic than a conventional subharmonic mixer.

FIG. 12is a diagram illustrating a WCDMA transceiver employing a mixer circuit according to the present invention.

A signal received by the antenna81is divided by the duplexer82into a signal to be delivered to the transmitter94and a signal to be bandwidth-filtered. The low-noise amplifiers83through85amplify received signals from the duplexer82and adjust gains. The mixers86and87according to the present invention receive amplified and gain-adjusted signals. Each of the two mixers86and87receives four phase local signals from eight phase local signals produced by the polyphase VCO93and produces orthogonal demodulated baseband signals. The baseband signals whose out-of-band signals are suppressed with the channel selection filters88and89are converted into digital signals with the AD converters90and91. The DSP92performs a filtering process, a rake process, a code despreading process and so on for the digital signals to retrieve a signal sent to the WCDMA transceiver80.

When the mixer circuit according to the present invention is accommodated in the above WCDMA transceiver80, it is possible to not only suppress a DC offset caused by a local leak signal but also achieve a large gain and a good noise characteristic even if a direct demodulation circuit having properties of small size, low cost and low electric power consumption is used.

FIG. 13is a circuit diagram illustrating an optical receiver circuit to which a mixer circuit according to the present invention is applied.

The optical receiver circuit inFIG. 13contains a photoelectric converter OE, a limiter101, a mixer fundamental circuit102, a VCO103, a current sense amplifier104, a loop filter105, a differential amplifier106, D flip-flops107and108, and an output circuit109. Resistors R1through R6and inductors L1and L2are provided as components serving to connect between the circuits.

An optical input is converted into an electric signal by the photoelectric converter OE and then its waveform is shaped for the mixer fundamental circuit102according to the present invention. As long as the optical signal has a property of a NRZ (nonreturn-to-zero), the optical signal, which may be a binary burst signal or a random pattern, has a fundamental wave component formed of less than half of a clock signal.

In the optical receiver, it is necessary to extract the clock signal from the received signal and incorporate the received signal by using the extracted clock signal. In the construction shown inFIG. 13, a shaped received signal is used as a switch control signal of the mixer circuit according to the present invention. As mentioned above, since the mixer circuit according to the present invention employs a local signal whose frequency is half of the frequency of a carrier signal, it is possible to detect a phase of the frequency of the carrier signal. Similarly, it is possible to detect a phase of the clock signal in the construction shown inFIG. 13by controlling the switches based upon a fundamental wave whose frequency is half of the clock signal included in the received signal.

Namely, the mixer fundamental circuit102detects a phase of a frequency component corresponding to the clock signal of the received signal from output signals of the VCO103by using the shaped received signal as a local signal, and then the detected phase is produced as a current signal. The operation regarding the mixer fundamental circuit corresponds to a phase comparison operation. Low-frequency components of the output current signal are supplied to the current sense amplifier104via the inductors L1and L2. Voltage outputs of the signals amplified by the current sense amplifier104are returned as feedbacks to the VCO103via the loop filter105and the differential amplifier106. The feedback signals control an oscillation of the VCO103and form a phase synchronous loop.

Outputs of the VCO103controlled by the loop become clock signals synchronous to the clock signals included in the received signal, and then the resulting clock signals are supplied to the D flip-flop107and108. The D flip-flops107and108form a master-slave latch circuit and latch the received data in synch with the clocks. The latched signals are output as output signals D+ and D− via the output circuit109.

The phase comparison unit for a received PLL of an optical fiber and binary communication is basically equivalent to a phase detector. Thus, a conventional multiplying circuit such as Gilbert cell is used as a mixer circuit in an RF receiver together with a phase comparison unit in an optical receiver. In general, since the RF signal is weak, the RF mixer should have certain properties: low noise, low phase noise, high conversion gain and low distortion. Since the PLL of the optical receiver receives a VCO output and a received data signal sufficiently amplified and clipped as its input signals, it is necessary to execute an operation at high speed instead of the requirement of high performance for noise and conversion gain. Furthermore, there might be another requirement for small size and low electric power consumption. If the mixer circuit according to the present invention is used, it is possible to implement a small and low electric power consuming optical receiver achieving high speed and low phase noise.

FIG. 14is a diagram illustrating a structure of a frequency detector to which the mixer circuit according to the present invention is applied. The frequency detector is formed by integrating structures inFIG. 14AthroughFIG. 14D.

In the construction inFIG. 13, although the phase synchronous loop is used to synchronize a phase with a clock signal component in a received signal, it is not actually sufficient to simply adjust the phase. It is necessary to detect a frequency and control an oscillating frequency of the VCO so as to cause the detected frequency to coincide with the frequency of the clock signal in the received signal.FIG. 14is a circuit diagram illustrating the frequency detector for the purpose.

FIG. 14Ashows a structure in which four distribution circuits121through124are connected to outputs of the mixer fundamental circuits20. For example, the two mixer fundamental circuits20receive data signals dp and dn corresponding to the signals D+ and D− detected by the circuit shown inFIG. 13as local signals to control switches. One of the mixer fundamental circuits receives an in-phase clock c1and its complementary clock c5, and the other of the mixer fundamental circuits receives an orthogonal phase clock c3and its complementary clock c7. As a result, an orthogonal phase signal is generated by detecting a phase at the time t1.

FIG. 14Bshows a demultiplexer130to which the data signal dp and dn are given as its inputs. The demultiplexer130four-way demultiplexes the data signals dp and dn in order to generate four demultiplexed data signals d1through d4each of which has a different phase from the others. Also, the demultiplexer130generates high voltage level demultiplexed data signals ud1through ud4whose voltages are set at a higher level and whose logic is the same as the demultiplexed data signals d1through d4.FIG. 15shows the data signals dp and dn and the demultiplexed data signals d1through d4.

InFIG. 14A, when the demultiplexed data signals d1through d4are supplied to the quarter-split circuits121through124, a detected clock signal, which is an orthogonal phase signal, is interleaved in four divisions.FIG. 16shows an example of a circuit structure of the quarter-split circuits121through124.

FIG. 14Cshows current sense amplifiers131through138and level shifters139through146. The current sense amplifiers131through138receive and amplify four-divided interleave signals c11–c14, c51–c54, c31–c34and c71–c74generated by the quarter-split circuits121through124. The amplified voltage signals are reset by the NMOS transistors to which interleaving clocks udl1through udl4of an approximate 1 data width are gate-input. The voltage levels of the amplified voltage signals are decreased by level shifters139through146, and are output as orthogonal phase signals s11–s14, s31–s34, s51–s54and s71–s74.FIG. 17shows an example of a circuit structure of the current sense amplifiers131through138.FIG. 18shows an example of a circuit structure of the level shifters139through146.

FIG. 14Dshows the mixer fundamental circuits20, quarter-split circuits151through154, multipliers155through158and current sense amplifiers159and160, respectively. The mixer fundamental circuits20and the quarter-split circuits151through154generate orthogonal phase signals demodulated by data signals at the time t2with respect to the clock signals c1and c5and the clock signals c3and c7. The multipliers155through158multiply the orthogonal phase signals being at the time t2and orthogonal signals s11–s14, s31–s34, s51–s54and s71–s74being at the time t1, and the outputs are supplied to the current sense amplifiers159and160. Frequency error signals “op” and “on” are obtained by adding outputs of the current sense amplifiers159and160.FIG. 19shows an example of a circuit structure of the multipliers155through158.

A frequency error fe is given by the following formula;
fe=dφ/dt≈(φt2−φt1)/ΔT.

In the vicinity of zero, the value φt2−φt1is approximated to;
exp[j(φt2−φt1)]≈1+j(φt2−φt1).

From the above approximation, since an imaginary part of the formula exp [j(φt2−φt1)]/ΔT is an approximate value, the frequency error fe is represented by the following formula;
fe≈(sin φt2cos φt1−cos φt2sin φt1)/ΔT,
where ΔT is a data transition interval. In the construction shown inFIG. 14, the frequency error fe is computed in accordance with the above formula. Although the data transition interval ΔT is not included in the outputs of the construction, the construction can be easily implemented. Since only the polarity of the frequency error fe is important in most cases, the construction inFIG. 14can be used in practice. Here, when burst data formed of 0/1 alternating codes are used, ΔT becomes a data transfer period. Even if a random pattern is used, the signal is attenuated in the range ΔT>>2π√RC, where R and C are a resistor component and a capacitance component of an output impedance of the current sense amplifier, respectively. Thus, it is possible to avoid the problem that the value sin(φt2−φt1) in the above formula is valid only in the range from −π to π.

FIG. 20shows a frequency lock loop having a structure simplified from the structure inFIG. 14.FIG. 20Ashows a mixer fundamental circuit and quarter-split circuits (qs) corresponding toFIG. 14C.FIG. 20Cshows a photoelectric converter OE generating the corresponding electric signal from a received optical signal and a limiter LM shaping a waveform of the electric signal.FIG. 20Dshows a mixer fundamental circuit, quarter-split circuits (qs), multipliers (qxs) and a current sense amplifier corresponding to the structure inFIG. 14D. InFIG. 20E, frequency error signals f+ and f− detected in the structures ofFIG. 20Athrough FIG.20D are set as inputs, and clock signals c1, c3, c5and c7are generated by controlling the frequencies of the inputs according to the inputs. When the generated clock signals c1, c3, c5and c7are given as their inputs to the structures ofFIG. 20AthroughFIG. 20D, a frequency lock loop is formed by receiving a feedback.FIG. 20Fshows a demultiplexer generating demultiplexed signals from the clock signals c1, c3, c5and c7. The demultiplexed signals are supplied to individual quarter-split circuits.

In the following, a subharmonic steering mixer and its variations will be described.

FIG. 21is a circuit diagram illustrating a circuit structure of the subharmonic steering mixer.

The circuit structure inFIG. 21such that a signal is transmitted therein as a differential local signal is known as a conventional technique. In this case, since a signal having half of the frequency of a carrier signal is used, it is possible to prevent a DC offset caused by self demodulation of a leaked local signal. However, since a signal having the frequency of the carrier signal is generated in a common emitter node by Cbe of a transistor to which a local signal is input, it is impossible to prevent Ccb of an RF input transistor from leaking to an RF signal. Namely, measures for the local leak are incomplete, and it is impossible to use the circuit structure for a WCDMA receiver wherein an accurate direct demodulation is required.

FIG. 22is a circuit diagram illustrating a circuit structure of a subharmonic transfer mixer in which a Gilbert cell is formed with an orthogonal differential local signal whose local frequency is equal to a carrier frequency.

As is shown inFIG. 22, when a circuit is provided to have the structure in which a Gilbert cell is formed with an orthogonal differential local signal whose local frequency is equal to a carrier frequency, the equation 2Fcarrier=4Flo is satisfied with respect to a local higher harmonic generated in a collector of an RF input transistor. Thus, the problem is solved. InFIG. 22, CMFB, LPF, CMP and FILT represent a common mode feedback circuit, a LOW pass filter, a comparator and a digital filter, respectively. A current source controlled by LPF, CMP, FILT and a FILT output form a ΣΔA/D converter and a receiver filter or a portion of the receiver filter.

In an embodiment of the above-mentioned mixer circuit and an embodiment of its application, a polyphase oscillation circuit is used to obtain a local signal. In the following, a description will be given of an example of a structure of such a polyphase oscillation circuit. In order to ensure the performance of individual circuits, it is preferred to use a polyphase oscillation circuit as shown in the following to provide a local signal of low-phase noise or a clock of a low-timing jitter to the above mixer circuit or a phase/frequency detector. However, the present invention is not limited to the use of the polyphase oscillation circuit.

FIG. 23is a diagram illustrating an example of a structure of a polyphase oscillation circuit. In the structure example, a polyphase oscillator with more than or equal to three phases is provided to combine individual oscillators of Colpitts circuits.

A polyphase oscillator200contains a plurality of Colpitts circuits201-1through201-nand a plurality of capacitors Cp. The Colpitts circuits201-1through201-nare connected in a chain via the capacitors Cp and are formed as a loop by connecting both ends of the chain. The Colpitts circuits201-1through201-nhave identical structures. The individual Colpitts circuit contains an NMOS transistor202, a resistor203, a constant current source204, an inductor L and a capacitor Cf.

When the phase difference between adjacent Colpitts circuits is small, the amplitude is enlarged by the overlap of waves. As a result, the individual Colpitts circuits work to restrict the amplitude. When an oscillation phase difference between adjacent Colpitts circuits is larger than 2π/n, there arises a portion having a small phase difference at another place in the loop, thereby restricting the amplitude at that place. As for the above process, an oscillation is attenuated through the above amplitude restriction if the oscillation has an oscillation phase difference larger than 2π/n or smaller than 2π/n. Consequently, only a wave whose oscillation phase difference between adjacent Colpitts circuits is just equal to 2π/n can grow in amplitude in collaboration with the adjacent Colpitts circuits, thereby obtaining the stable oscillation.

FIG. 24shows another example of a structure of a polyphase oscillation circuit. In the example, a polyphase oscillator with n phases is formed by base-grounding Colpitts circuits.

A polyphase oscillator210inFIG. 24contains a plurality of Colpitts circuits211-1through211-nand a plurality of capacitors Cp. The Colpitts circuits211-1through211-nare connected in a chain through the capacitors Cp and are formed in a loop by connecting both ends of the chain. The Colpitts circuits211-1through211-nhave identical structures. The individual Colpitts circuit contains a bipolar transistor212, a resistor213, an inductor L and a capacitor Cf. Here, the bipolar transistor212may be an NMOS transistor.

Bias voltages Vb1through Vbn are applied to bases, which are control ends of the transistors (gates in NMOS transistors). At the starting time of an oscillation, when bias voltages are sequentially applied to a part or all of the bias voltages Vb1through Vbn, it is possible to determine a sign of the oscillation phase difference between adjacent Colpitts circuits. Outputs of the oscillator inFIG. 23are obtained through collectors or emitters of the bipolar transistors212.

Among some types of polyphase oscillation circuits, an orthogonal (four phase) oscillator is the most commonly used. The orthogonal (four phase) oscillator is formed by providing four Colpitts circuits to the oscillation circuit shown inFIG. 24. Since the orthogonal (four phase) oscillator works poorly to correct a phase between adjacent Colpitts circuits in a steady oscillation state, there is a possibility that the phase error becomes large depending on device selections and the circuit structure. Thus, it is necessary to take measures for solving the problem.

FIG. 25shows an example of a structure of an orthogonal oscillation circuit with Colpitts circuits. InFIG. 25, the same parts as those parts inFIG. 24are referred to as the same numerals. The orthogonal oscillator inFIG. 25is provided with a resistor Rd to realize a stable oscillation.

An orthogonal oscillation circuit210A inFIG. 25contains a plurality of Colpitts circuits211-1through211-4, a plurality of capacitors Cp and the resistor Rd. One end of the resistor Rd is connected to bases of transistors of individual Coplitts circuits211-1through211-4, and a bias voltage Vb is applied to the other end.

When the bias voltage Vb is supplied to the bases of the individual transistors via the resistor Rd, there arises a voltage drop in the resistor Rd in proportion of the flow amount of the current, thereby decreasing the voltage in the bases. Some cases are supposed with respect to the above problem, for example, the case in which two of four oscillation circuits oscillate together around a phase of 0° because the two oscillation circuits have close oscillation phases and the other two oscillation circuits also oscillate together around a phase of 180° because the other two oscillation circuits also have close oscillation phases. In this case, since there are base current flows corresponding to two transistors around the phases of 0° and 180°, the voltage drop in the resistor Rd becomes large, thereby considerably decreasing the base voltages. Conversely, since there are little base current flows around phases of 90° and 270°, the voltage drop does not arise in the resistor Rd, thereby rarely decreasing the base voltages. Thus, transistors are difficult to switch on around the phases of 0° and 180°, in other words, a suppression function works around the phases of 0° and 180°, whereas transistors are easy to switch on around phases of 90° and 270°, in other words, an induction function works around phases of 90° and 270°.

As mentioned above, the resistor Rd not only prevents a plurality of transistors from simultaneously switching on but also serves to avoid a phase condition in which no transistor switches on. As a result, it is possible to realize a condition in which individual Colpitts oscillation circuits separately oscillate in different phases and the base currents become constant in a whole range of phases from 0° to 360°. Namely, it is possible to realize a steady state in which four Colpitts oscillation circuits sequentially oscillate at phase differences of 90°.

The resistor Rd may be set in a range from a few ohms to scores of ohms, or an internal resistance in the source of the bias voltage Vb may function instead of the resistor Rd.

In the following, a description will be given of an oscillation frequency of a polyphase oscillation circuit with Colpitts circuits.

When adjacent Colpitts circuits in an n-phase oscillation circuit operate in a phase difference 2π/n, an oscillation phasor of the adjacent Colpitts circuits is given by the formula;
Aei(θ±2π/n),
where Aeiθis an oscillation phasor of the Colpitts circuit of interest. Using a capacitance Cp for a connection between the adjacent Colpitts circuits, a resonant capacitance Cr is represented as follows;
Cr=Cp|2eiθ−ei(θ+2π/n)−ei(θ−2π/n)|.
The resonant capacitance Cr contributes to a charge transmission between the adjacent Colpitts circuits. Accordingly, an oscillation angle frequency is given by the following formula;
ωosc=((Cf+Cr)/(LCfCr))1/2,
whereby the oscillation angle frequency is valid in the range n≧2.

FIG. 26shows another example illustrating a structure of an orthogonal oscillation circuit with Colpitts circuits. InFIG. 26, the same parts as those inFIG. 25are referred to as the same numerals and the description thereof will be omitted.

An orthogonal oscillation circuit210B inFIG. 26contains oscillation circuits221-1and221-2integrally formed with two Colpitts circuits whose phases are opposite to each other, a plurality of capacitors Cp and a resistor Rd. As is shown in FTG.26, the oscillation circuits221-1and221-2have identical structures. The oscillation circuit221-1contains two Colpitts circuits211-1and211-3whose phases are opposite to each other and capacitors C1and C3. Inductors of the Colpitts circuits211-1and211-3are trans-connected to be reverse-wound to each other and form a symmetrically connected inductor Lsym. The symmetrically connected inductor Lsym controls the Colpitts circuits211-1and211-3so that the Colpitts circuits211-1and211-3can oscillate in a reversed phase. In the construction, it is possible to realize a state in which the resistor Rd works similarly to the circuit structure inFIG. 25, individual Colpitts circuits separately oscillate in different phases, and the base current remains constant in a whole phase range from 0° to 360°.

FIG. 27is a circuit diagram illustrating a structure of a RTW oscillator based upon Colpitts circuits. The RTW (Rotary Traveling-Wave) oscillator is a polyphase oscillator formed by connecting a large number of multi-vibrators in a differential transmission line and is designed to distribute clocks of a digital circuit. By replacing the multi-vibrators in a conventional RTW oscillator with Colpitts circuits, a Colpitts RTW oscillator230is formed as shown inFIG. 27.

A Colpitts RTW oscillator230shown inFIG. 27Ais formed by connecting a plurality of Colpitts circuit oscillators240. The Colpitts RTW oscillator230has a structure such that outputs of Colpitts circuit oscillators240being at the final stage are crossed and connected to inputs of Colpitts circuit oscillators240being at the first stage. The Colpitts circuit oscillators240have identical structures.FIG. 27Bshows a circuit structure of the Colpitts circuit oscillator240. The Colpitts circuit oscillator240contains an inductor231, resistors232and233, a capacitor234, an NMOS transistor235, a resistor236, a capacitor237, an inductor241, resistors242and243, a capacitor244, an NMOS transistor245, a resistor246, a capacitor247, and a resistor250. The Colpitts circuit oscillator240is formed to have a structure in which two Colpitts circuits are aligned in parallel.

In the RTW oscillator, an inductance Luper a unit length of a differential transmission line is given as follows;
Lu=(μ0/π)Log(πs/(ω+tc)+1),
where s, w and tcare an interval, a width and a thickness of the differential transmission line, respectively. Also, a phase speed vp is represented as follows;
vp=1/(LuCl/SegLen)1/2,
where Cl is a capacitance between the transmission lines, and SegLen is a unit cell length. According to the above formula, an oscillation frequency fc is given as follows;
fc=vp/(2·RingLen),
where RingLen is a circumference. By deleting the circumference RingLen from the above formula, the following formula is obatained;
fc=1/(2·(LtotCtot)1/2,
where Ltotis a total inductance including mutual inductances (which is an inductance corresponding to two entire loops because of the double loops) and Ctotis a total capacitance between differential transmission lines (which is a quarter of a total stray capacitance corresponding to the two entire loops by considering conversion from a parallel to GND into differential and serial), respectively. By representing the inductors231and241as Lc, capacitances of the capacitors234and244as Cf, and capacitances of the capacitors237and247as Cr inFIG. 27B, the oscillation frequency of the circuit inFIG. 27Ais given by the following formula;
fc=¼n·((Cf+Cr)/LcCfCr)1/2.

In the following, a description will be given of an optical receiver circuit wherein the mixer circuit according to the present invention is applied and the above-mentioned orthogonal (four-phase) oscillator is used.

FIG. 28Ais a circuit diagram illustrating an optical receiver circuit to which the mixer circuit according to the present invention is applied. An orthogonal (four phase) oscillator shown inFIG. 25andFIG. 26is used in the circuit.

The optical receiver circuit inFIG. 28Acontains a photoelectric converter OE, a limiter301, mixer fundamental circuits302-1through302-4, an orthogonal oscillator VCO303, current sense amplifiers304-1through304-4, a loop filter305, a differential amplifier306, and output determination circuits307-1through307-4.

The current sense amplifiers304-1through304-4have identical circuit structures, and the circuit structure is illustrated inFIG. 28B. The current sense amplifier contains NMOS transistors321through326.

The output determination circuits307-1through307-4have identical structures, and the circuit structure is illustrated inFIG. 28C. The output determination circuit contains a first high-speed D flip-flop331and a second high-speed D flip-flop332. The high-speed D flip-flops331and332have identical circuit structures, and the circuit structure is illustrated inFIG. 28D. The high-speed D flip-flop contains NMOS transistors341through348and resistors349through351. Input signals ip and in are followed while a signal t is HIGH and are latched when a signal1becomes HIGH. The latched signals are output as op and on. Regarding the first high-speed D flip-flop331, the signals op and on are supplied to the flip-flop332being at the next stage. Regarding the second high-speed D flip-flop332, the signals op and on are outputs of the optical receiver.

In detail, while the signal t is HIGH, the input signals ip and in are read into gate capacitances of the NMOS transistors345and346and an offset cancel is provided by a negative-feedback via the NMOS transistors343and344. While the signal1is HIGH, the input signals ip and in are discarded and a positive-feedback is provided in the D latch circuit. These operations realize a high-speed read.

InFIG. 28A, an optical input is converted into an electric signal by the photoelectric converter OE, and then is wave-shaped by the limiter301. The resulting signal is supplied to the mixer fundamental circuits302-1through302-4according to the present invention. Here, the optical input may be a binary burst signal or a random pattern as long as the optical input satisfies a NRZ (nonreturn-to-zero). The optical input has a fundamental wave component less than half of a clock signal.

In the optical receiver, it is necessary to extract a clock signal from a received signal and retrieve the received signal by using the clock signal. As mentioned above, the mixer circuit according to the present invention makes it possible to detect a phase of a frequency of a carrier signal by using a local signal with half of the carrier frequency. In the structure shown inFIG. 28A, the orthogonal oscillator VCO303produces four phase clock signals, and the four mixer circuits302-1through302-4compare a clock signal in the individual phases with the input signal. At this time, in the individual mixer circuits302-1through302-4, the NMOS transistor400short-circuits between signals in sync with a reversed phase clock. As a result, the phase comparison result of only an in-phase clock is provided to the current sense amplifiers304-1through304-4. A voltage output amplified by the current sense amplifier104is returned as a feedback to the VCO303via the loop filter305and the differential amplifier306. The feedback signal makes it possible to control the transmission of the VCO303and form a phase synchronous loop.

Since each of the four mixer circuits supplies the phase comparison results of individual in-phase clocks to the current sense amplifier, it is possible to realize a 4 way-interleaved operation. For example, when a received optical signal has 40 GHz, individual four phase clock signals generated by the VCO303have 10 GHz. As a result, it is possible to detect a data signal at 40 GHz by using the output determination circuits307-1through307-4that operate at 10 GHz.

Loop-controlled clock outputs of the VCO303are supplied to the output determination circuits307-1through307-4. As mentioned above, the output determination circuits307-1through307-4formed with two high-speed D flip-flops maintain received data as their output signals through the latch operation synchronous with the clock.

In the above-mentioned embodiments, when the mixer circuit according to the present invention is used, it is possible to realize an optical receiver having the properties of being small sized, low electric power consuming, high speed and with a low phase noise. Furthermore, when the orthogonal (four-phase) oscillation circuit is used as a VCO formed in a phase synchronous loop, it is possible to receive and detect a high-speed optical signal reliably and easily. As is described with reference toFIG. 13andFIG. 28, a conventional subharmonic mixer such as APDP may be used to detect a phase of an NRZ/NRZi signal. As a result, it is possible to realize a less electric power consuming and higher-speed optical receiver whose interleave structure is easy.

It is possible to realize a mixer circuit according to the present invention with a small size, low cost and low electric power consumption. In addition, since a signal having half of a frequency to be detected (carrier frequency) is used as the local signal, it is possible to achieve a high gain and a good noise characteristic because a DC offset caused by a local leak signal is not generated and a signal path always has an impedance less than a constant value.

Additionally, when the mixer circuit according to the present invention is used as a phase comparison circuit in a receiver circuit, it is possible to realize a small-sized, low-noise and high-speed phase comparison circuit.

The present application is based on Japanese priority applications No. 2001-323843 filed Oct. 22, 2001 and No. 2002-158275 filed May 30, 2002, the entire contents of which are hereby incorporated by reference.