Apparatus and method for masking audio signals in a signal distribution system

An apparatus and method is provided for transmitting "masked" stereo audio signals in a signal distribution network such as a cable television system within the bandwidth constraints of conventional frequency allocation schemes. In various embodiments, left and right stereo audio signal components (L+R and L-R) are used to amplitude modulate a carrier using independent sideband (ISB) modulation without the transmission of a pilot signal. A jamming signal or low bandwidth data signal may be transmitted in place of the normal pilot tone. Various aspects of the invention contemplate the use of Hilbert transforms to perform the ISB modulation and demodulation.

BACKGROUND OF THE INVENTION
 1. Technical Field
 This invention relates generally to an apparatus and method for
 transmitting "masked" stereo audio signals in a signal distribution system
 such as a cable TV network. More particularly, the invention prevents the
 unauthorized reception of narrowband signals while minimizing bandwidth
 requirements needed for masking such signals.
 2. Related Information
 FIG. 1 shows the frequency spectrum of a conventional BTSC (Broadcast
 Television Standards Committee) encoded audio signal. Stereo signals
 consist of a left and a right speaker component, each of which is
 band-limited between 50 Hz and 14.5 Khz. The BTSC signal transmits the sum
 of the two components (L+R) 101 at baseband. The difference of the
 components (L-R) is amplitude modulated onto a suppressed carrier signal
 at twice the horizontal line frequency (2f.sub.h, where the line frequency
 is 15.73 KHz) thus producing two sidebands 103 and 104. The L-R signal is
 usually DBX encoded before AM modulation in order to provide greater
 immunity to noise introduced by FM demodulation at the receiver. A pilot
 tone signal 102, which is phase-locked to the carrier signal, is
 transmitted at the horizontal line frequency (f.sub.h).
 At the receiving end, the L-R signal is AM demodulated and recombined with
 the L+R signal to generate the L and R audio signals. Conventional BTSC
 audio decoders lock on to the pilot tone 102 and re-generate the
 suppressed carrier signal; this pilot tone also indicates the presence of
 a stereo signal to BTSC decoders. In the absence of the pilot tone, BTSC
 decoders assume that only a monaural signal is transmitted.
 It is often desirable in a cable television system to scramble or otherwise
 render unintelligible portions of the transmitted signal, including the
 audio signal, in order to prevent unauthorized reception. Various
 techniques for providing secure access to the audio portion of the
 transmitted signal have been devised. As one example, U.S. Pat. No.
 4,956,862 to Robbins et al. describes a technique wherein the audio
 portion of the television signal is modulated at a frequency that is
 offset from the standard carrier frequency (e.g., 4.75 MHZ instead of 4.5
 MHZ). Unfortunately, this technique has the disadvantage of increasing the
 amount of bandwidth required to carry the signal, and suffers from other
 side effects as well.
 Other techniques, such as those exemplified by U.S. Pat. No. 5,159,631 to
 Quan et al., vary the frequency of the modulation carrier in a pseudo
 random fashion or use other means of scrambling the audio signal. Other
 methods make use of digital encryption techniques. However, such
 conventional approaches undesirably increase the overall bandwidth
 required to transmit the audio signal and require complicated circuitry.
 Accordingly, conventional techniques have been found to be unsatisfactory.
 SUMMARY OF THE INVENTION
 In certain embodiments of the present invention, an audio "masking"
 operation is performed at a headend along with video scrambling. The
 scrambled video signal is AM-VSB modulated, and the masked audio signal is
 FM-modulated and combined with the TV signal at IF. The TV IF signal is
 then upconverted to a desired channel for transmission across a network.
 At a customer's premises, a settop terminal descrambles the video signal
 and "demasks" the stereo audio signal.
 At the transmitting end, the BTSC signal is split into a baseband L+R
 signal and a baseband (DBX encoded) L-R signal. These two signals are used
 to form an independent sideband (ISB) modulated signal containing
 information from the L+R signal on the lower sideband and information from
 the L-R signal on the upper sideband. The ISB modulated signal is combined
 with a video signal at a headend and transmitted over a network such as a
 cable TV distribution system. At the receiving end, the "masked" audio
 signal is demodulated and provided as an output to a user's television.
 It is thus one object of the present invention to provide for the secure
 transmission and reception of BTSC encoded stereo audio signals over a
 cable TV network or other distribution system using a minimum of
 bandwidth.
 It is a further object of the present invention to transmit narrow
 bandwidth signals in a secure manner within the constraints of the NTSC
 frequency allocation scheme.
 In various aspects of the invention, a Hilbert transformation operation is
 used to perform the independent sideband modulation. A jamming or low
 bandwidth data signal may also be transmitted at a frequency location
 normally used for transmitting a pilot signal.
 Other advantages and benefits of the present invention will become apparent
 through the following detailed description, figures, and the appended
 claims.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
 FIG. 2 shows a frequency plan incorporating various principles of the
 present invention. In various embodiments, the L+R and L-R signals are
 separated from the conventional BTSC encoded signal and used to amplitude
 modulate a carrier signal using independent sideband modulation (ISB-AM),
 thus creating independent sidebands 206 and 208 in the space normally
 occupied by the L-R sidebands in the conventional BTSC encoded signal (see
 FIG. 1). A carrier signal 207 may be transmitted at 3f.sub.H.
 Additionally, a jamming signal 205 may be transmitted in the location
 conventionally reserved to transmit the L+R signal. In various
 embodiments, no pilot signal is transmitted (note absence of pilot signal
 102 of FIG. 1).
 In accordance with the signal plan shown in FIG. 2, conventional BTSC
 stereo decoders will not be able to decode the audio signal because, among
 other things, no pilot signal is transmitted, and further because the L+R
 and L-R components are not located in their normal location and are not
 modulated according to the conventional BTSC approach. Advantageously,
 however, the signal plan shown in FIG. 2 fits within the same bandwidth
 space required by the conventional BTSC signal and, accordingly, does not
 result in an increase in bandwidth. The scheme shown in FIG. 2 and
 variations thereof will be referred to generally as "masked audio".
 FIG. 3 shows a system employing audio masking in accordance with various
 aspects of the present invention. A headend 301 includes two channel
 processing modules 301a and 301b each of which modulates and scrambles
 video and audio signals. Each module includes a video scrambler and an
 audio masking circuit which respectively receive baseband video and BTSC
 audio signals. The scrambled video and masked audio are fed to a vestigial
 sideband modulator followed by an upconverter. As is conventional,
 multiple programming channels are combined together for transmission
 across a cable system 302.
 At the receiving end, settop terminals 303 and 304 include signal
 processing components which demodulate, descramble and "demask" the
 signals for presentation on televisions 305 and 306. The present invention
 provides a novel and useful means for masking the BTSC stereo audio signal
 so that unauthorized persons cannot hear the audio signal.
 Stereo Signal Recovery
 FIG. 4 is a simplified block diagram of an audio masking circuit which
 employs various principles of the present invention. The first step in
 various embodiments is the recovery of the separate L+R and L-R stereo
 signals from the BTSC encoded audio signal. A BTSC encoded audio signal
 (see FIG. 1) is input to a lowpass filter 401, a bandpass filter 402, and
 a clock synthesizer 404. Lowpass filter 401 recovers the L+R signal using
 a stop band of about 14.5 Khz. These types of filters are common in BTSC
 receivers.
 The AM modulated L-R signal is isolated in bandpass filter 402 (passband:
 17 Khz to 46 Khz). Both BPF 402 and LPF 401 preferably have less than 0.5
 dB ripple in the passbands. After the AM-modulated component has been
 isolated, it is demodulated by mixing it with a 2f.sub.H carrier in mixer
 403. The mixer output is then lowpass filtered in LPF 405 to isolate the
 L-R component (LPF 405 also has a passband edge at 14.5 KHz). Preferably,
 LPF 405 provides a gain of 0.5 to compensate for the gain introduced in
 the mixing process. The L-R signal preferably retains DBX encoding for
 better noise immunity.
 The recovered L-R and L+R signals are digitized using A/D 406 and A/D 407
 respectively (LPFs 401 and 402 also serve as anti-aliasing filters). The
 sampling clock for the A/Ds is 4f.sub.H. The sampling clock is
 phase-locked with the incoming pilot tone. The sampled L+R and L-R signals
 are then "masked" in digital processing circuit 408 which includes an ISB
 modulation function. Following signal processing in circuit 408, the
 resulting signal is converted back to analog in D/A 409 and low-pass
 filtered in LPF 410. The D/A sampling clock can be 16 times the f.sub.H
 tone. The output of LPF 410 is a masked audio signal (see FIG. 2) which
 can be modulated onto the audio subcarrier and added to the (VSB
 modulated) TV signal at IF (see FIG. 3).
 The clock synthesizer 404 shown in FIG. 4 locks onto the f.sub.H pilot tone
 and generates phase and frequency locked clocks for the system. In
 addition to the A/D clock and the D/A clocks, it generates a high speed
 (50 MHZ or so) digital clock for digital processing circuit 408, which
 need not be phase-locked with the f.sub.H pilot tone, and may be derived
 from a crystal oscillator. The clock synthesizer also generates a 2f.sub.H
 signal which can be used to demodulate the L-R component.
 The process of separating the L+R and L-R signals could be carried out
 entirely in the digital domain by digitizing the BTSC signal at the input
 of the system and carrying out the filtering and mixing operations in the
 discrete-time domain. However, partitioning the system as illustrated
 reduces the amount of digital processing required, making it possible to
 carry out all of the digital processing on a single DSP.
 Field-programmable gate-arrays (FPGAs) or high complexity programmable
 logic devices (HCPLDs) can also be used to carry out the digital
 processing.
 FIG. 5 shows one possible circuit for clock synthesizer 404. The circuit of
 FIG. 5 comprises a dual-loop PLL. The circuit includes two voltage
 controlled oscillators 505 and 511, and a phase detector 512. Any scheme
 for clock generation can be used. In practice, many harmonics of the pilot
 tone and several high speed clocks are readily available in headend
 modulators.
 The L+R and L-R signals are sampled at 4f.sub.H, which is greater than
 twice the minimum sampling rate required to prevent aliasing. Because
 these signals are preferably over-sampled, the anti-aliaing filter
 requirements (e.g., LPFs 401 and 405 in FIG. 4) can be simplified. The
 sampled signals are lowpass filtered to filter out the residual noise in
 the stop-band. Because the analog anti-aliasing filters preferably have a
 stop-band attenuation of greater than 55 dB, the digital lowpass filters
 can be simplified. The combination of analog anti-aliasing filters and
 digital lowpass filters should provide 60 dB of attenuation in the signal
 stop-band (approximately 1/4 the sampling frequency and higher). The
 outputs of the lowpass filter are decimated to reduce the processing
 requirements for the elements which follow them.
 Audio Masking Operation
 A detailed description of the processing performed in signal processing
 circuit 408 of FIG. 4, including independent sideband modulation using
 Hilbert transforms, will now be provided with reference to FIG. 6 and
 FIGS. 7A to 7J. A brief theoretical background explanation will first be
 provided to aid in understanding various principles of the invention.
 The Fourier transform of an arbitrary real signal is complex. The real part
 of the complex Fourier transform is symmetrical around zero-frequency. The
 imaginary part of such a Fourier transform is anti-symmetrical about
 zero-frequency. The Fourier transform of an even, real function is
 symmetrical about the f=0 axis while the frequency spectrum of an odd,
 real function is anti-symmetrical about the same axis.
 It is possible to isolate either the positive frequency component or the
 negative frequency component of a real signal by using Hilbert transforms.
 A Hilbert transform has the following frequency response:
EQU H.sub.Hilbert (.OMEGA.)=j sgn(.OMEGA.).OMEGA. (1)
 The function sgn(.OMEGA.) represents the sign of .OMEGA. in equation (1).
 The impulse response of a Hilbert-transformer is given by:
EQU h.sub.Hilbert (t)=1/.pi.t (1b)
 Since the impulse response of a Hilbert transformer has an infinite
 discontinuity at t=0, analog realizations of a Hilbert transformer are
 difficult to realize. The discrete-time Hilbert transformer has a periodic
 frequency response with a period 2.pi.. The impulse response of the
 discrete-time Hilbert transformer is given by:
EQU h.sub.Hilbert [n]=sin.sup.2 (.pi.n/2)/(.pi.n/2) (1c)
 The discrete-time Hilbert transform is zero for n=0, and has a continuously
 decreasing amplitude for n.notident.0. Discrete-time Hilbert transformers
 can be realized using finite-impulse-response (FIR) filters.
 The effect of filtering a signal with a Hilbert transformer is to multiply
 the positive frequency component of the input signal by "-j" and the
 negative frequency component by "j".
 The frequency spectrum of any signal can be represented by a positive
 frequency component and a negative frequency component. Let X.sub.p
 (.OMEGA.) and X.sub.n (.OMEGA.) be the Fourier transforms of the positive
 frequency component and the negative frequency component of a signal x(t).
 The Fourier transform of x(t), X(.OMEGA.), can be represented as:
EQU X(.OMEGA.)=X.sub.p (.OMEGA.)+X.sub.n (.OMEGA.) (2)
 The frequency spectrum of a Hilbert-transformed signal, X.sub.h (.OMEGA.)
 is given by:
EQU X.sub.h (.OMEGA.)=-jX.sub.p (.OMEGA.)+jX.sub.n (.OMEGA.) (3)
 The positive frequency component of a signal can be isolated by adding to
 the signal a "Hilbert-transformed" version of itself, multiplied by "j".
EQU X(.OMEGA.)+j Xh(.OMEGA.)=(X.sub.p (.OMEGA.)+X.sub.n (.OMEGA.))+j(-jX.sub.p
 (.OMEGA.)+jX.sub.n (.OMEGA.))=2X.sub.p (4)
 In an analogous manner, the negative frequency component can be isolated by
 subtracting j X.sub.h (.OMEGA.) from X(.OMEGA.). Signals which contain
 only positive frequency components, or only negative frequency components
 are called "analytic signals".
 For convenience, the L+R signal will be referred to as signal A and the L-R
 signal will be referred to as signal B. An analytic version of signal A,
 which contains only negative frequency components, can be formed using the
 following Hilbert transform relationships. The notation A.sub.n will be
 used to refer to the analytic version of the signal A (i.e., the version
 containing only negative frequency components). A.sub.n can be expressed
 as:
EQU A.sub.n =A-jA.sub.h (5)
 where A.sub.h is the Hilbert-transformed version of A.
 Similarly, the positive frequency components of signal B, which will be
 referred to as B.sub.p can be expressed using the following relationship:
EQU B.sub.p =B+jB.sub.h (6)
 where B.sub.h is the Hilbert-transformed version of B.
 The two analytic signals can be combined to form a complex independent
 sideband signal which contains information from L+R on the lower sideband
 and L-R on the upper sideband. This signal will be referred to as C:
EQU C=A.sub.n +B.sub.p =(A-jA.sub.h)+(B+jB.sub.h) (7a)
EQU =(A+B)+j(B.sub.h -A.sub.h) (7b)
EQU =(A+B)+j(B-A).sub.h =I+jQ (7c)
 Going from equation 7b to 7c takes advantage of the fact that a Hilbert
 transform operation is a linear one. The real and imaginary components of
 the complex signal C have been relabelled I and Q, respectively. Q can be
 formed by taking the Hilbert transform of the difference of L-R and L+R,
 and I can be formed by taking the sum of L+R and L-R.
 FIG. 6 shows a circuit which can be used to perform audio masking in
 accordance with various aspects of the present invention. FIGS. 7A to 7J
 show spectra for signals taken at points A through J in the circuit of
 FIG. 6. In the figures, .omega..sub.c is used to denote the cutoff
 frequency of a filter and .omega..sub.p is used to denote the passband
 edge of the filters. Unless otherwise indicated, filter gain is assumed to
 be unity and the stop band is assumed to be at least 60 dB below the
 passband. FIR filters are preferred for stability and phase linearity. For
 some filters, the form of the filter is specified by indicating the length
 of the filter. For example, the notation L=2M+1 where M is an integer
 constrains the length of the filter to be an odd integer. FIR filters of
 this type have a group delay of M samples. Elements labeled Z.sup.-M
 introduce a delay of M samples.
 Beginning with the left side of FIG. 6, the recovered L+R signal (FIG. 7A)
 is input to a low pass filter 601 ((.omega..sub.p =0.46.pi.,
 .omega.c=0.5.pi.), followed by decimation by 2 in decimator 602 (see FIG.
 7B). Similarly, the recovered L-R signal is input to a low pass filter 609
 having the same characteristics, followed by decimation in decimator 610
 (see FIG. 7C). The decimated filtered signals are provided to respective
 summers 603 and 611. The L+R component is subtracted in summer 603 while
 the others are added.
 The output of summer 603 is provided to a Hilbert transformer 604
 (.omega..sub.p 0.003.pi., 0.92.pi.) preferably having a length of 2M1+1
 (group delay of M1 samples) which produces a quadrature component which is
 delayed by M1 samples in relation to its inputs. The real component must,
 therefore, also be delayed by M1 samples, as shown by delay element 612.
 Because the passband of the Hilbert transformer 604 extends from 50 Hz to
 14.5 Khz, the order of the Hilbert transformer is relatively high. A
 length 301 Hilbert transformer can be implemented using a least-squares
 approach which satisfies the requirements of the audio signal. The
 spectrum output from Hilbert transformer 604 is shown by D1 in FIG. 7D,
 and the spectrum output from delay element 612 is shown by D2 in FIG. 7D.
 The independent sideband signal C is interpolated by two by inserting zeros
 between each sample (elements 605 and 613) and filtering the result in low
 pass filters 606 and 614, respectively (.omega..sub.p =0.46.omega.,
 .omega..sub.c =0.5.pi., gain=2). The interpolating filter is of a
 relatively high order since the passband edge and Nyquist frequency of the
 independent sideband signal are relatively close. Even so, a half-band
 filter can be implemented which uses 26 fourteen bit coefficients to
 accomplish the interpolation. Further optimization of the half-band filter
 is possible. The expected frequency response of the interpolated
 independent sideband signal is shown in FIG. 7E.
 The interpolated signal is then multiplied by cos(.pi.n) by generator 608
 and multipliers 607 and 615. The resulting spectrum is shown in FIG. 7F.
 The interpolated, mixed independent sideband signal (I2+jQ2) is then rate
 expanded by two (elements 616 and 618), resulting in the frequency
 spectrum shown in FIG. 7G. Note that the upper sideband of this signal is
 centered around .omega.=.pi./2, which corresponds to 2f.sub.H at the new
 sample rate.
 The upper sideband of (I3+jQ3) can be isolated using Hilbert transforms.
 Using C3 to designate I3+jQ3, the upper sideband C.sub.u can be formed by:
EQU C.sub.u =C3+jC3.sub.h, (8a)
 where C3.sub.h is the Hilbert transformed version of C3
EQU =(I3+jQ3)+j(I3.sub.h +jQ3.sub.h) (8b)
EQU =(I3=Q3.sub.h)+j(Q3+I3.sub.h) (8c)
 The real part of C.sub.u can be used to form a real signal with symmetrical
 frequency response. The entire operation of first multiplying by
 cos(.pi.n), rate-expanding and Hilbert transformation is equivalent to
 interpolation by two and mixing with e.sup.i.pi.n. The advantage of using
 this approach is that the Hilbert transform approach is less
 computationally demanding. The interpolation and mixing operations can be
 performed using only a length 13 Hilbert transformer 619 (L=2M2+1,
 .omega..sub.p =0.25.pi., 0.75.pi.) which can be implemented using only
 three 12 bit coefficients. The frequency response of the interpolated,
 modulated signal is shown in FIG. 7H.
 The foregoing description explains how the independent sideband modulated
 signal can be formed using Hilbert transforms. A baseband jamming or audio
 barker signal (620) and a reference pilot tone (element 622) is then added
 to the resulting signal, producing the masked signal shown in FIG. 7I. The
 pilot tone is placed at .pi.=3.pi./4 which corresponds to 3f.sub.h after
 D/A conversion. The pilot tone could also be placed at 2f.sub.H instead,
 but placing the tone at 3f.sub.H simplifies the design of the
 clock-recovery circuit at the receiver.
 Finally, the audio signal is interpolated by two (element 624) and passed
 through a low pass filter 625 (.omega..sub.p =0.375.pi.,.omega..sub.c
 =O.625.pi., gain=2) . There are two reasons for this. First, all DACs
 exhibit a sin(x)/x amplitude distortion on the reconstructed spectrum. The
 distortion is fairly small at frequencies less than 1/4 the sampling rate.
 The interpolation filter also predistorts the signal by giving it a
 x/sin(x) shape. The combination of the predistortion and oversampling
 virtually eliminates any DAC artifacts. The second reason is that the
 analog reconstruction filter (following DAC 409 in FIG. 4) is simplified
 greatly because of the oversampling. A fifth order elliptic filter is
 sufficient for this purpose. The frequency response of the signal at the
 output of the digital processing circuit is shown in FIG. 7J. As explained
 above, all of the operations may be carried out using a single floating
 point or fixed point DSP processor.
 Audio Demasking Operation
 The audio demasking operation can be carried out using a monolithic ASIC
 with very few external components. FIG. 8 shows a circuit which can be
 used to perform the audio demasking operation, and FIGS. 9A to 9L show
 frequency spectra for various labelled portions of the circuit in FIG. 8.
 The signal input to the audio demasking circuit of FIG. 8 is assumed to be
 the same as the output of the masking circuit (e.g., the headend of a
 cable TV distribution system). The signal is first passed through a low
 pass filter 801 (passband 47 Khz, stop-band edge 63 Khz) to band limit it
 to below 50 Khz. It is also assumed that the signal has been gain adjusted
 so that the dynamic range of the A/D 810 can be fully utilized. The signal
 is sampled in A/D 810 using for example 12 bits operating at 8 f.sub.H
 (i.e., 126 KS/sec). A/D 810 should provide at least 10 effective bits of
 resolution at this sampling rate. There is an integral and nonlinearity
 requirement of 1/2 LSBs or better. All of the spurious signals introduced
 by A/D 810 should be 55 dBC or lower. The spectrum at point A in the
 circuit is shown in FIG. 9A.
 The A/D sampling clock is assumed to be phase locked to the 3f.sub.H pilot
 tone in the input signal. The sampling clock can be generated using a PLL
 as explained with reference to FIG. 5. Once phase and frequency lock with
 the pilot tone is achieved, the masked signal can be demodulated by mixing
 it with e.sup.-j.pi./2. The spectrums of the signal before and after the
 mixing operation are shown in FIGS. 9A and 9B, respectively.
 The complex signal resulting from the mixing operation is filtered using
 lowpass filters 804 and 814 (.omega..sub.p =0.23.pi., .omega..sub.c
 =0.25.pi.), which corresponds to the location of the 3f.sub.H pilot tone.
 The spectrum of the resulting signal is shown in FIG. 9C. Note that the
 independent sideband signal has been isolated in FIG. 9C.
 Because the signal at the input is oversampled, it is decimated by a factor
 of 4 in decimators 805 and 815, thereby reducing some computational burden
 in the elements that follow. The spectrum after decimation is shown in
 FIG. 9D.
 The two sidebands of the ISB signal can then be isolated using Hilbert
 transformers. Let C represent the independent sideband signal, and I and Q
 represent the real and imaginary parts of C, respectively. Let C.sub.n
 represent the lower sideband of C, and let C.sub.p represent the upper
 sideband of C. Both C.sub.n and C.sub.p are analytic signals which can be
 separated using the following relationships:
EQU C.sub.n =C-jC.sub.h =(I+jQ)-j(I.sub.h +jQ.sub.h) (9a)
EQU =(I+Q.sub.h)+j(Q-I.sub.h) (9b)
EQU C.sub.p =C+jC.sub.h =(I+jQ)+j(I.sub.h +jQ.sub.h) (9c)
EQU =(I-Q.sub.h)+j(Q+I.sub.h) (9d)
 I.sub.h and Q.sub.h are Hilbert transformed versions of I and Q,
 respectively. The L+R signal can be recovered by taking the real part of
 C.sub.n. Similarly, the L-R signal can be recovered by taking the real
 part of C.sub.p. The frequency response of the recovered L+R signal is
 shown in FIG. 9F. The frequency response of the recovered L-R signal is
 shown in FIG. 9E. The Hilbert transformer 816 shown in the circuit of FIG.
 8 is the same as those in the circuit of FIG. 6, as is delay element 806.
 The L+R and L-R signals are then processed to generate a BTSC signal. The
 first part of this processing involves interpolation by two. The L+R
 signal is interpolated by inserting zeros between each sample (rate
 expander 808) followed by lowpass filtering in LPF 809 which is preferably
 of a high order because of the narrow transition band. The frequency
 response of the signal at the output of LPF 809 is shown in FIG. 9G. The
 interpolated L+R signal is again interpolated by two in expander 821 and
 filtered in LPF 822 (.omega.p=0.23.pi., .omega.=0.75.pi., gain=2). The
 resulting frequency response is shown in FIG. 9J.
 The L-R signal is interpolated and modulated by cos(.pi.n) in the same
 step. First, zeros are inserted between samples of the L-R signal
 (expander 818), and the resulting signal is high pass filtered in HPF 819
 (.omega..sub.c =0.5.pi.,.omega..sub.p =0.46.pi., gain=2). The resulting
 signal is rate expanded in element 820, and the expanded signal is delayed
 by M3 samples in delay element 829, where M3 is the group delay of the
 interpolation filter used to filter the L+R signal. The resulting spectrum
 is shown in FIG. 9I. Alternatively, the L-R signal could have been
 interpolated by 4, just as with the L+R signal, and then mixed with
 cos(.pi./2n).
 The interpolated L+R signal and the interpolated and AM modulated L-R
 signal are added to each other in summer 823, and a pilot tone at f.sub.H
 is added. The BTSC signal is thus regenerated. The frequency response is
 shown in FIG. 9K.
 The next step, as in the masking circuit, is interpolation and
 predistortion. A 12 bit D/A converter 827 can be used to convert the
 signal back into the continuous time domain. The LPF 828 after D/A 827
 serves as a reconstruction filter. Like the A/D clock, the D/A clock
 should be phase-locked with the 3f.sub.H carrier in the input signal.
 FIG. 10 is a block diagram for an ASIC which performs the audio demasking
 operation under microprocessor control. The ASIC includes analog
 anti-aliasing and reconstruction filters and a dual-loop clock synthesizer
 for generating all the clocks needed for on-chip operation. A 12 bit
 successive approximation 130 KS/sec A/D converter and a 12 bit 260 KS/sec
 D/A converter are also included. The digital portion of the ASIC takes
 advantage of a high speed clock (50 MHZ) and a DSP engine which can be
 time-shared between different filtering operations for optimal
 performance. Filter coefficients can be stored in an integrated ROM.
 It is apparent that many modifications and variations of the present
 invention are possible, and references to specific values are by example
 only. For example, various method steps of the invention may be practiced
 in a different ordered sequence from that illustrated without departing
 from the scope of the invention. Moreover, it will be appreciated that
 references to L+R and L-R signals, where designated "first" and "second"
 audio components in the claims, can of course be reversed in designation
 to achieve the same effect and, therefore, references to "first" and
 "second" should not be deemed to limit the scope of the claims. It is,
 therefore, to be understood that within the scope of the appended claims
 the invention may be practiced otherwise than as specifically described.