Power module

A power module includes a current sensing circuit in which a transistor includes an emitter connected to a sense emitter of a current sense element of an IGBT and a base connected to ground, a current sensing resistor including one end thereof connected to a collector of the transistor and the other end thereof connected to a common connection portion. The power module detects, as a current sensing voltage, a potential difference generated by the current sensing resistor based on the common connection portion as a reference, compares the current sensing voltage with a predetermined threshold voltage, and determines whether or not an overcurrent flows through the IGBT according to a magnitude relation therebetween.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a power module used for controlling a motor of an industrial or consumer apparatus.

2. Description of the Background Art

A power switching semiconductor device such as an IGBT (Insulated Gate Bipolar Transistor) or a power MOSFET (Metal-Oxide Semiconductor Field-Effect Transistor) adopts the following method to detect an overcurrent. The power switching semiconductor device is configured of a main element through which a main current flows and a current sense element through which a part of the main current flows; a sense current outputted from an output terminal (sense emitter) of the current sense element is converted, by a resistor (current sensing resistor), into a voltage to serve as a detection voltage; and the detection voltage is compared with a predetermined reference voltage to determine whether the detection voltage is normal or abnormal (overcurrent level).

Here, the current sense element is structured of a collector (drain) which is shared by the main element, and an emitter (source) having an area which is arranged to have a predetermined area ratio to an area of an emitter (source) of the main element so that a sense current at a predetermined division ratio to the main current flows therethrough.

For example, in the case where an emitter area ratio of the current sense element with respect to the main element is 1/10000, a current which is 1/10000 of the current of the main element flows through the current sense element. This makes it possible to detect a current by a resistor having a relatively smaller resistance.

Here, when the current sensing resistor is connected to the current sense element, a difference between voltages of gates of the main element and the current sense element is caused, which results in a change of a current division ratio. Since this division ratio changes largely when a resistance of the current sensing resistor is large, it is necessary to use a smaller resistance for sensing.

However, when a smaller resistance is used for sensing, a threshold voltage (reference voltage) for determining an overcurrent becomes smaller. This causes a factor for a malfunction (false detection).

Japanese Patent Application Laid-Open No. 10-322185 (1998) discloses, inFIG. 1, a configuration in which a sense current is not directly sensed by a resistor, but is received by a current mirror circuit configured of an N-channel MOS transistor, and a mirror current (current I4) obtained by the current mirror circuit is converted into a voltage by a current sensing resistor (resistor R1) connected to a power source (voltage V3) of the current mirror circuit to thereby form a detection voltage (voltage V1).

According to this configuration, the detection voltage V1is expressed as V1=V3−(I4×R1). Since the detection voltage V1depends on the voltage V3of the power source, the detection voltage V1varies according to a change of the voltage V3, and this may lead to a drop in accuracy of current sensing.

A similar problem is also caused in a technique disclosed in Japanese Patent Application Laid-Open No. 1-193909 (1989) (FIGS. 1 and 2) in which a current mirror circuit for receiving a sense current and generating a mirror current is combined with a current mirror circuit for generating a reference current as a mirror current, and the presence or absence of an overcurrent is determined according to a magnitude relation between the mirror current of the sense current and the reference current. Also, in this case, when a source voltage of the current mirror circuit changes, the mirror current changes, and a drop in accuracy of current sensing may be caused.

As described above, according to a conventional configuration of a power switching semiconductor device for sensing an overcurrent, in the case where a sense current is sensed by a resistor, a variation thereof becomes large when a current sensing resistance is large, and a false detection tends to be caused when the current sensing resistance is small. Also, in the case where the sense current is sensed by using a current mirror circuit, a drop in accuracy of current sensing may be caused with fluctuations in power source.

SUMMARY OF THE INVENTION

It is an object of the present invention to provide a power module that does not cause a false detection and a drop in accuracy of current sensing even in the case where a sense current is sensed by a resistor or a current mirror circuit.

An electronic device according to an aspect of the present invention includes a power switching semiconductor device provided with a main element through which a main current flows, and a current sense element which is configured to allow a part of the main current to flow therethrough and includes an output terminal from which a sense current is outputted, a current sensing circuit provided with a first transistor including a first main electrode connected to the output terminal of the current sense element, and a current sensing resistor including one end thereof connected to a second main electrode of the first transistor and other end thereof connected to a common connection portion, the first transistor including a control electrode connected to a first reference potential, an overcurrent determination circuit that detects, as a current sensing voltage, a potential difference generated by the current sensing resistor based on the common connection portion as a reference, compares the potential difference with a predetermined threshold voltage, and determines whether or not an overcurrent flows through the power switching semiconductor device according to a magnitude relation between the potential difference and the predetermined threshold voltage, and a drive circuit that generates a control signal applied to a control electrode of the power switching semiconductor device.

According to an aspect of the power module, when the on-voltage of the power switching semiconductor device is low, and the current sensing resistor is connected to the current sense element, a difference is caused between the voltages applied to the control electrodes of the main element and the current sense element, and a current division ratio changes. As a result, an accurate sense current cannot be obtained. However, since the first transistor is connected to the output terminal of the current sense element, the voltage change at the output terminal of the current sense element is minimized to a voltage drop caused by the on-resistance of the first transistor, which is, for example, about 0.7 V. As a result of this, a voltage difference between voltages applied to the control electrodes of the main element and the current sense element is reduced to about 0.7 V and becomes stable, so that the accuracy in detecting the sense current improves. Since the voltage difference between the voltages applied to the control electrodes of the main element and the current sense element is reduced to about 0.7 V, there is no need to consider that the current division ratio varies between the main element and the current sense element, and the resistance of the current sensing resistor can be arbitrarily set. Therefore, the false detection can be prevented by increasing the resistance of the current sensing resistor.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

Hereinafter, although a description will be given of an IGBT as an example of a power switching semiconductor device, the present invention can be applied to other types of power switching semiconductor devices such as those configured of a MOSFET or a bipolar transistor. Although the conductivity type of the power switching semiconductor device will be described as an N-channel type, it is needless to say that a P-channel type can also serve the purpose.

First Embodiment

FIG. 1is a circuit diagram illustrating a configuration of a power module100according to the first preferred embodiment of the present invention. As illustrated inFIG. 1, the power module100is provided with a drive control circuit10that drives an IGBT1to turn into an on state or an off state by controlling a voltage (gate voltage) supplied between a gate and an emitter of the IGBT1.

The drive control circuit10includes a drive circuit C1with a DC power source V1as a drive power source, an overcurrent determination circuit C2, and a current sensing circuit C3.

A free wheel diode2is connected in reverse direction in parallel between a collector C and an emitter E of the IGBT1. A return current from a main circuit flows through the free wheel diode2when the IGBT1is in the off state.

In the power module100illustrated inFIG. 1, a main power source potential VCC is applied to the collector C of the IGBT1, and a ground potential GND (first reference potential) is applied to the emitter E because the emitter E is grounded. Then, an output of a driver DR of the drive circuit C1is fed to the gate G of the IGBT1.

The driver DR receives the DC power source V1as a drive power source, and a cathode of the DC power source V1is connected to a common connection portion BP of the drive control circuit10so as to provide a reference potential of the drive control circuit10. Two input portions of the driver DR are connected to an anode of the DC power source V1and the common connection portion BP, respectively.

The IGBT1, for detection of an overcurrent, includes a main element through which a main current flows and a current sense element configured to allow a part of the main current to flow therethrough, and is configured so that a sense current is outputted from an output terminal (sense emitter) of the current sense element.

While the current sense element has the collector (drain) thereof shared with the main element, an area of the emitter (source) thereof is arranged to have a predetermined area ratio to an area of the emitter (source) of the main element so that a sense current at a predetermined division ratio to the main current flows therethrough.

The current sensing circuit C3includes a PNP transistor Q5having an emitter thereof connected to an sense emitter SE of the IGBT1and a base thereof that is grounded, and a current sensing resistor SR having one end thereof connected to a collector of the PNP transistor Q5and other end thereof connected to the common connection portion BP. Here, a potential difference between a potential generated by the current sensing resistor SR and a potential of the common connection portion BP serving as a reference is assumed as a current sensing voltage Vs.

The overcurrent determination circuit C2includes a comparator CP receiving the potential of the common connection portion BP as a reference potential and operating with the power source potential Vc. One input of the comparator CP is connected to a connection node between the collector of the PNP transistor Q5and the one end of the current sensing resistor SR, the other input of the comparator CP is connected to an anode of a DC power source V3for supplying an arbitrary threshold voltage, and a cathode of the DC power source V3is connected to the common connection portion BP.

The comparator CP compares the current sensing voltage Vs with the threshold voltage, determines, according to the magnitude relation there between, whether an overcurrent is flowing or not, and feeds a result thereof to the drive circuit C1for use in controlling the driver DR. When the current sensing voltage Vs indicates an overcurrent level, the drive circuit C1performs control such as turning off the IGBT1. However, since this is remotely related to the present invention, any further description will not be given.

An anode of a DC power source V2included in the drive circuit C1is grounded, and a cathode thereof is connected to the common connection portion BP. Here, the anode of the DC power source V2is grounded together with the emitter E of the IGBT1. It is also possible to use a P-channel MOSFET transistor instead of the PNP transistor Q5. In that case, transistors other than the IGBT1can also be configured of MOSFETs.

Next, current sensing operation of the power module100will be described. In the power module100, the drive control circuit10includes its own common connection portion BP that receives a negative bias applied from the DC power source V2and serves as a drive circuit reference potential (second reference potential). Then, since the DC power source V1drives the driver DR while making the drive circuit reference potential as a reference, the gate of the IGBT1is configured to receive, as a control signal, a positive bias and a negative bias. Here, since the DC power source V2sets a negative potential, it will be sometimes referred to as “potential setting means”.

A result of the simulation of the current sensing operation of the power module100when the positive and negative biases are applied to the gate of the IGBT1in this way is illustrated inFIG. 3. Here,FIG. 2is a circuit diagram illustrating a simulation condition set therein while elements of the current sensing circuit C3and the drive circuit C1for performing the simulation are specified. It should be noted that portions identical with those inFIG. 1are identified with identical reference numerals or symbols, and descriptions thereof will not be repeated.

Referring toFIG. 2, the IGBT1is divided into a main element MT and a current sense element ST, a gate-emitter voltage (gate voltage) of the main element MT is expressed as Vge, and a collector-emitter voltage is expressed as Vice. Further, a current flowing through the IGBT1in its entirety is expressed as a main current Ic, and a current flowing through the current sense element ST is expressed as a sense current Is. At the same time, a current flowing through the PNP transistor Q5is expressed as a current Ie.

In the drive circuit C1, the driver DR includes an NPN transistor Q1having a collector thereof connected to the anode of the DC power source V1and an emitter thereof connected to the gate of the current sense element ST through a resistor R1; a PNP transistor Q2having a collector thereof connected to the common connection portion BP and an emitter thereof connected to the gate of the current sense element ST; and a pulse signal source VP that applies a pulse signal having a height of 0 to 20 V to the bases of the NPN transistor Q1and the PNP transistor Q2. The pulse signal source VP is connected to the common connection portion BP and receives the drive circuit reference potential as a reference. Here, the resistor R1is a resistor for setting a switching speed of the IGBT1when the IGBT1is on, and the resistor R2is a resistor for setting a switching speed of the IGBT1when the IGBT1is off.

The DC power source V1is a power source to generate 20 V as a potential B, and the DC power source V2is a power source to generate −5 V as a potential A. Here, the main power source potential VCC is set at 200 V.

An inductance L1of a load between the collector of the IGBT1and the anode of the main power source PW is set at 500 μH, and a resistance of the current sensing resistor SR is set at 12 ohms.

Among the results of the simulation performed under the simulation condition depicted inFIG. 2, a waveform of the gate voltage Vge is illustrated in (a) portion ofFIG. 3, waveforms of the collector-emitter voltage Vice and the main current Ic are illustrated in (b) portion ofFIG. 3, a waveform of the sense current Is is illustrated in (c) portion ofFIG. 3, a waveform of the current flowing through the PNP transistor Q5is illustrated in (d) portion ofFIG. 3, and a waveform of the current sensing voltage Vs is illustrated in (e) portion ofFIG. 3.

In accordance with rising and falling of the gate voltage Vge which is a pulse signal illustrated in (a) portion ofFIG. 3, the IGBT1turns off and on. When the IGBT1turns on, the main current Ic flows through as illustrated in (b) portion ofFIG. 3, and, at the same time, the sense current Is flows through as illustrated in (c) portion ofFIG. 3. In a same manner as in the case of the sense current Is, the current Ie flows through the PNP transistor Q5as illustrated in (d) portion ofFIG. 3, and, accordingly, the current sensing voltage Vs can be obtained as illustrated in (e) portion ofFIG. 3.

Here, as illustrated in (a) portion ofFIG. 3, the gate voltage Vge is formed of not only a positive bias ranging from 0 V to 15 V but also a negative bias ranging from 0V to −5 V. In this way, it is possible to securely perform the off operation of the IGBT1by using a pulse signal formed of the positive and negative biases as a gate voltage.

In other words, it is possible to turn off the IGBT if the gate-emitter voltage becomes equal to or lower than a threshold value of the IGBT even with application of a pulse formed only of a positive bias. However, with application of a pulse signal formed by including a negative bias, the IGBT can be more securely turned off.

In addition, in the case where a pulse signal formed of the positive and negative biases is used as the gate voltage, as compared with the case where the pulse signal is formed only of the positive bias, there is an advantage in that, even when an on-voltage of the power switching device such as the IGBT is low, a chance of false operation is minimized.

Further, when the on-voltage of the power switching device is low, and the current sensing resistor is connected to the current sense element, a potential difference (ΔVge) for an amount of voltage drop in the current sensing resistance is caused in a voltage applied to the gates of the main element and the current sense element. Since the voltage drop in the current sensing resistor becomes larger as a current flowing therethrough increases, the voltage drop becomes particularly large when an overcurrent is detected, which causes the potential difference ΔVge to increase and changes the current division ratio. As a result, an accurate sense current cannot be obtained. However, since the PNP transistor Q5is connected to the sense emitter SE of the IGBT1, the voltage change at the sense emitter SE is minimized to a voltage drop caused by the on-resistance of the PNP transistor Q5, which is, for example, about 0.7 V. As a result of this, the potential difference ΔVge is restricted down to about 0.7 V and becomes stable, so that the accuracy in detecting the sense current improves.

Since the potential difference ΔVge can be lowered to about 0.7 V regardless of the resistance of the current sensing resistor SR, there is no need to consider that the current division ratio varies between the main element MT and the current sense element ST, and the resistance of the current sensing resistor can be arbitrarily set. Therefore, the false detection can be prevented by increasing the resistance of the current sensing resistor SR.

Further, the overcurrent determination circuit C2determines the state of the overcurrent by comparing the current sensing voltage Vs with an arbitrary threshold voltage by the comparator CP. However, since the threshold voltage is generated based on the drive circuit reference potential, as a reference, which is also the potential of the common connection portion BP which provides the lowest potential (negative potential), the drive circuit reference potential does not change even if the DC power source V1changes, which makes accurate detection of current possible. Here, when the voltage of the DC power source V2changes, the drive circuit reference potential also changes. However, when the drive circuit reference potential changes, not only the potential of the DC power source V3but also the reference potentials of all the other circuits change in a similar way. This results in a zero relative change, and highly accurate current sensing can be maintained.

Second Preferred Embodiment

FIG. 4is a circuit diagram illustrating a configuration of a power module200according to a second preferred embodiment of the present invention. As illustrated inFIG. 4, the power module200is provided with a drive control circuit20that drives an IGBT1to turn into an on state or an off state by controlling a voltage (gate voltage) supplied between a gate and an emitter of the IGBT1. Here, portions identical with those of the power module100illustrated inFIG. 1are identified with identical reference numerals or symbols, and descriptions thereof will not be repeated.

The drive control circuit20includes a drive circuit C1with a DC power source V1as a drive power source, an overcurrent determination circuit C2, and a current sensing circuit C4. A difference from the drive control circuit10illustrated inFIG. 1is found in the current sensing circuit C4.

The current sensing circuit C4includes PNP transistors Q3and Q4having emitters thereof connected to a sense emitter SE of the IGBT1, and a current sensing resistor SR having one end thereof connected to a collector of the PNP transistor Q4and the other end thereof connected to a common connection portion BP. Bases of the PNP transistors Q3and Q4are connected together to a collector of the PNP transistor Q3, and the PNP transistors Q3and Q4constitute a current mirror circuit.

The collector of the PNP transistor Q3is grounded, a cathode of a DC power source V2is connected to the common connection portion BP, and an anode of the DC power source V2is grounded together with an emitter E of the IGBT1.

A connection node ND between the collector of the PNP transistor Q4and one end of the current sensing resistor SR is connected to one input of the comparator CP.

Next, current sensing operation of the power module200will be described. In the power module200, the drive control circuit20includes its own common connection portion BP that receives a negative bias applied from the DC power source V2and serves as a drive circuit reference potential. Then, since the DC power source V1drives a driver DR while making the drive circuit reference potential as a reference, the gate of the IGBT1is configured to receive, as a control signal, a positive bias and a negative bias.

A result of the simulation of the current sensing operation of the power module200when the positive and negative biases are applied to the gate of the IGBT1in this way is illustrated inFIG. 6.

Here,FIG. 5is a circuit diagram illustrating a simulation condition set therein while elements of the current sensing circuit C4and the drive circuit C1for performing the simulation are specified. It should be noted that portions identical with those inFIG. 1are identified with identical reference numerals or symbols, and are assumed to have the same simulation condition. Therefore, descriptions thereof will not be repeated.

Referring toFIG. 5, a current flowing through the IGBT1in its entirety is expressed as a main current Ic, a current flowing through the current sense element ST is expressed as a sense current Is, and currents flowing through the PNP transistors Q3and Q4are individually expressed as currents Ie. Here, the transistor characteristics of the PNP transistors Q3and Q4are identical with each other, and the current Ie is a half of the sense current Is.

Among the results of the simulation performed under the simulation condition depicted inFIG. 5, a waveform of the gate voltage Vge is illustrated in (a) portion ofFIG. 6, waveforms of the collector-emitter voltage Vice and the main current Ic are illustrated in (b) portion ofFIG. 6, a waveform of the sense current Is is illustrated in (c) portion ofFIG. 6, a waveform of the current flowing through the PNP transistor Q4is illustrated in (d) portion ofFIG. 6, and a waveform of the current sensing voltage Vs is illustrated in (e) portion ofFIG. 6.

In accordance with rising and falling of the gate voltage Vge which is a pulse signal illustrated in (a) portion ofFIG. 6, the IGBT1turns off and on. When the IGBT1turns on, the main current Ic flows through as illustrated in (b) portion ofFIG. 6, and, at the same time, the sense current Is flows through as illustrated in (c) portion ofFIG. 6. Then, as illustrated in (d) portion ofFIG. 6, the current Ie which is about a half of the sense current Is flows through the PNP transistor Q4, and, accordingly, the current sensing voltage Vs can be obtained as illustrated in (e) portion ofFIG. 6.

In this way, since the drive control circuit20is configured to receive the output from the sense emitter SE of the IGBT1by the current mirror circuit, and allows to flow the current Ie which is about a half of the sense current Is through the current sensing resistor SR, a power consumption by the current sensing resistor SR can be reduced.

For example, assuming that it is an overcurrent if the current sensing voltage Vs is 0.5 V when the main current Ic is 100 A, and the division ratio of the current sense element to the main element is 1/10000, in the drive control circuit10of the first preferred embodiment, the power consumption by the current sensing resistor SR is expressed as Vs×Is=0.5×(100/10000)=5 mW. In addition, in the drive control circuit20, the power consumption by the current sensing resistor SR is expressed as Vs×(½)Is=0.5×( 50/10000)=2.5 mW.

In this way, it is possible to arbitrarily change the current flowing through the current sensing resistor SR by providing a configuration in which the output of the sense emitter of the IGBT1is received by the current mirror circuit, and by changing a size (size ratio) of the transistor of the current mirror circuit or providing a plurality of transistors that generate the mirror current.

For example, if it is assumed that the size ratio of the PNP transistor Q4to the PNP transistor Q3is 10 to 1, the current Ie which is about 1/10 of the sense current Is flows through the PNP transistor Q4.

In the first and second preferred embodiments described above, the drive circuit reference potential is generated by applying a negative bias to the common connection portion BP from the DC power source V2. However, instead of using the DC power source V2, the negative bias may be obtained by dividing the potential B of the DC power source V1by resistors, or the negative bias may be obtained by using a Zener diode.

FIG. 7illustrates a configuration for obtaining the negative bias by a resistor divider, andFIG. 8illustrates a configuration for obtaining the negative bias by using a resistor and a Zener diode. It should be noted that, inFIGS. 7 and 8, portions identical with those illustrated inFIGS. 2 and 5are identified with identical reference numerals or symbols, and descriptions thereof will not be repeated.

In a power module100A illustrated inFIG. 7, resistors R4and R5are inserted, in series, between an anode of a DC power source V1and a common connection portion BP in order of the resistors R4and R5. A connection node between the resistors R4and R5is connected to a base of a PNP transistor Q5and is grounded together with an emitter E of an IGBT1.

This configuration has an advantage in that a negative bias determined by a resistor dividing ratio, for example, −5 V, can be applied to the common connection portion BP while the potential A serves as a reference, and the DC power source V2is not required. Here, since the resistors R4and R5can set the negative potential, these will be referred to as “potential setting means PS”.

In a power module100B illustrated inFIG. 8, a resistors R4and a Zener diode Z1are inserted, in series, between an anode of a DC power source V1and a common connection portion BP in order of the resistor R4and the Zener diode Z1. An anode of the Zener diode Z1is connected to the common connection portion BP, a cathode of the Zener diode Z1is connected to the resistor R4, and a connection node therebetween is connected to a base of a PNP transistor Q5and grounded together with an emitter E of an IGBT1.

This configuration has an advantage in that a negative bias determined by a Zener voltage, for example, −5 V, can be applied to the common connection portion BP while a potential A serves as a reference, and the DC power source V2is not required.

Since the negative bias is defined by the Zener voltage of the Zener diode Z1, the negative bias can be easily set by using a Zener diode having a desired Zener voltage. Here, since the resistor R4and the Zener diode Z1can set the negative potential, these will be referred to as “potential setting means PS”.

FIGS. 7 and 8illustrate variations based on the power module100. However, the same may also applied to the power module200.

In the power module100according to the first preferred embodiment, a configuration in which the anode of the DC power source V2is connected to the base of the PNP transistor Q5is described. In the power module200according to the second preferred embodiment, a configuration in which the anode of the DC power source V2is connected to the collector of the PNP transistor Q3is described. However, as illustrated inFIGS. 2 and 5, the gate voltage (gate-emitter voltage) of the current sense element ST becomes lower than the gate voltage (gate-emitter voltage) of the main element MT by a base-emitter voltage, i.e., 0.7 V, of the PNP transistor Q5and the PNP transistor Q4. For this reason, it may be possible that the current division ratio of the current sense element ST to the main element MT changes, and this may lead to a drop in accuracy of current sensing.

To avoid this, it is also possible to adopt the configurations illustrated inFIGS. 9 and 10. Specifically,FIG. 9illustrates a configuration of a power module100C in which a potential D resulted from reducing a predetermined potential from a potential A of the DC power source V2is applied to a base of a PNP transistor Q5.FIG. 10illustrates a configuration of a power module200A in which a potential D resulted from reducing a predetermined potential from a potential A of the DC power source V2is applied to a base of a PNP transistor Q4. It should be noted that, inFIGS. 9 and 10, portions identical with those illustrated inFIGS. 2 and 5are identified with identical reference numerals or symbols, and descriptions thereof will not be repeated.

In the power module100C illustrated inFIG. 9, a diode D2and a resistor R6are inserted, in series, between an anode of a DC power source V2and a common connection portion BP in order of the diode D2and the resistor R6. A connection node between the diode D2and the resistor R6is connected to a base of a PNP transistor Q5.

The diode D2is connected to the DC power source V2in the forward direction and capable of generating a potential D which is resulted from reducing a built-in voltage (p-n voltage) of the diode from the potential A, which is about 0.7 V. Applying this to the base of the PNP transistor Q5offsets an amount of a voltage drop of the gate voltage of the current sense element ST, and makes it possible to bring the gate voltages (gate-emitter voltages) of the main element MT and the current sense element ST to coincide with each other. As a result, it is possible to prevent the current division ratio of the current sense element ST to the main element MT from changing and achieve more accurate current sensing.

In the power module200A illustrated inFIG. 10, a diode D2and a resistor R6are inserted, in series, between an anode of a DC power source V2and a common connection portion BP in order of the diode D2and the resistor R6. A connection node between the diode D2and the resistor R6is connected to a base of a PNP transistor Q4.

The diode D2is connected to the DC power source V2in the forward direction and capable of generating a potential D which is resulted from reducing a built-in voltage (p-n voltage) of the diode from the potential A, which is about 0.7 V. Applying this to the base of the PNP transistor Q4offsets an amount of a voltage drop of the gate voltage of the current sense element ST, and makes it possible to bring the gate voltages (gate-emitter voltages) of the main element MT and the current sense element ST to coincide with each other. As a result, it is possible to prevent the current division ratio of the current sense element ST to the main element MT from changing and achieve more accurate current sensing.

Further,FIGS. 9 and 10illustrate the configurations in which the diode D2and the resistor R6are inserted, in series, between the anode of the DC power source V2and the common connection portion BP. Alternatively, a diode-connected transistor may be used instead of the diode D2as illustrated inFIGS. 11 and 12.

In a power module100D illustrated inFIG. 11, a PNP transistor Q6and a resistor R6are inserted, in series, between an anode of a DC power source V2and a common connection portion BP in order of the PNP transistor Q6and the resistor R6. An emitter of the PNP transistor Q6is connected to a base thereof, and the PNP transistor Q6functions as a diode. A connection node between the PNP transistor Q6and the resistor R6is connected to a base of a PNP transistor Q5.

In a power module200B illustrated inFIG. 12, a PNP transistor Q6and a resistor R6are inserted, in series, between an anode of a DC power source V2and a common connection portion BP in order of the PNP transistor Q6and the resistor R6. An emitter of the PNP transistor Q6is connected to a base thereof, and the PNP transistor Q6functions as a diode. A connection node between the PNP transistor Q6and the resistor R6is connected to a base of a PNP transistor Q4.

With this configuration, it is possible to generate a potential D resulted from reducing a built-in voltage from the potential A as in the case of using the diode. In addition to this, since the same transistor (if possible, the transistor in an identical production lot) as used for the PNP transistor Q5and the PNP transistor Q4is used for the PNP transistor Q6, individual differences caused by temperature characteristics and variations in the process are minimized among the transistors. As a result, a voltage drop in the PNP transistor Q6can be made the same as those in the PNP transistor Q5and the PNP transistor Q4, and further accurate current sensing is made possible.

In each of the power modules100and200described in the first and second preferred embodiments, the drive control circuits10and20are configured by excluding the IGBT1, the free wheel diode2, the power source applying the main power source potential VCC, and the DC power source V1. However, a whole or a part of the drive control circuit10or20may be built into a control IC.

A package integrating therein such a control IC, the IGBT1, and the free wheel diode2will be referred to as an intelligent power module (IPM).

By integrating the drive control circuits10and20into individual ICs, it is possible to reduce the circuit size and miniaturize the power modules100and200in their entirety.

In addition, by integrating the drive circuits10and20in their entirety into IC, the power module is configured of the IGBT1, the free wheel diode2, and the drive control circuit10or20. This arrangement reduces the number of components, and individual differences among the components, and lowers the rejection rate.

Since the number of components is reduced, mistakes in assembling are reduced, a probability in causing a fault during assembling is reduced, and the rejection rate is reduced.

Further, since the number of components is reduced, it becomes easier to manage the components and assemble the components together, which can lower the production cost.

With reduced individual differences among the components resulted from the reduced number of components, it is possible to increase the accuracy in current sensing.

As an example of integrating a part of the drive control circuits10and20into individual ICs involves a case of integrating the NPN transistor Q1and the PNP transistor Q2that constitute the driver DR into an IC, and a case of integrating also the resistors R1and R2in addition to the NPN transistor Q1and the PNP transistor Q2into an IC.

Integrating the NPN transistor Q1, the PNP transistor Q2, and the DC power source V1into an IC also involves a case of integrating also the resistors R1and R2in addition to the NPN transistor Q1, the PNP transistor Q2, and the DC power source V1into an IC. The DC power source V1is incorporated as a regulator into the IC.

Further, the case also involves integrating the drive control circuits10and20excluding the current sensing resistor SR into individual ICs. Since the current sensing resistor SR requires to have its resistance value to be exactly set for performing highly accurate sensing, and the resistance cannot be changed once it is integrated into an IC, it is preferable that the current sensing resistor SR be configured separately.

The same is also applied to the resistors R1and R2for setting the switching speed of the IGBT1. Therefore, there are sometimes cases in which the resistors R1and R2are separately configured in preparation for the case in which the switching speed is changed differently for each product.

[Use of Semiconductor Having a Wide Band Gap]

In the power modules100and200according to the first and second preferred embodiments, no descriptions have been given of materials used for the IGBT1and the free wheel diode2. However, the IGBT1and the free wheel diode2may be arranged as a silicon semiconductor device formed on a silicon (Si) substrate. Alternatively, the IGBT1may be arranged as a silicon semiconductor device, and the free wheel diode2may be arranged as a silicon carbide semiconductor device formed on a silicon carbide (SiC) substrate or as a gallium nitride semiconductor device formed on a substrate that is made of a gallium nitride (GaN) based material.

SiC and GaN are wide band gap semiconductors. Since the semiconductor device formed of the wide band gap semiconductor has a high withstand voltage and allowable current density, it is possible to miniaturize the device as compared with the silicon semiconductor device. Using such a miniaturized semiconductor device makes it possible to miniaturize the power module incorporating therein such a device.

Since the wide band gap semiconductor has a high thermal resistance, it is possible to miniaturize a radiation fin of a heatsink and use air cooling instead of water cooling. Accordingly, further miniaturization of the power module is possible.

Further, since the device can be made smaller than the silicon semiconductor device, the drive control circuits10and20can be miniaturized if they have the same rating.

Contrary to this, the free wheel diode2may be formed as a silicon semiconductor device, and the switching device (including a bipolar transistor and a MOSFET) such as the IGBT1may be formed as a wide band gap semiconductor device such as a silicon carbide semiconductor device or a gallium nitride semiconductor device. In this case, the same effect as the foregoing can also be obtained.

In the case where the switching device is a silicon semiconductor device, the current division ratio tends to change by the voltage difference (ΔVge) applied between the gates of the main element and the current sense element because the on-voltage is low. However, when the wide band gap semiconductor device is used as the switching device, the on-voltage increases, and the change in the current division ratio caused by the voltage difference ΔVge is suppressed. Therefore, an improvement in the accuracy of current sensing can be expected.

It is needless to say that both the IGBT1and the free wheel diode2may be formed by the wide band gap semiconductor device.

In the power modules100and200described with reference to the first and second preferred embodiments, a configuration in which the free wheel diode2is connected in reverse direction in parallel to the IGBT1is described. Alternatively, instead of the IGBT1and the free wheel diode2, an RC-IGBT (Reverse Conducting Insulated Gate Bipolar Transistor) including an IGBT and a diode connected in reverse direction in parallel thereto in an integrated manner may be used.

Here, referring toFIG. 13, the structure of the RC-IGBT will be described.FIG. 13is a cross sectional view of a semiconductor chip31incorporating therein an IGBT and a diode. The semiconductor chip31is formed by using an n substrate32. An n-type impurity layer33containing n-type impurities is provided on the n−substrate32, and a p base layer34containing p-type impurities is selectively provided thereon.

An emitter region35containing highly concentrated n-type impurities is selectively provided on the p base layer34. A groove36is formed from the emitter region35, while penetrating through the p base layer34and the n-type impurity layer33, to the n−substrate32. A gate insulating film37is formed on an inner wall of the groove36, and a gate electrode38of polysilicon is formed further inside.

An interlayer dielectric film39is provided on the emitter region35. An emitter electrode40is provided so as to make contact with parts of the emitter region35and the p base layer34. An n+cathode layer41and a p+collector layer42are provided on a reverse side of the n−substrate32. Further on the reverse side of these layers, a collector electrode43is formed. According to this structure, in a region where the n+cathode layer41is present, the diode is formed, and, in a region where the p+collector layer42is present, the IGBT is formed. In this way, the IGBT and the diode connected in reverse direction in parallel thereto are formed in an identical chip to thereby constitute the RC-IGBT.

The diode of the semiconductor chip31illustrated inFIG. 13turns on when a voltage between the p base layer34and the n-type impurity layer33exceeds a built-in potential of the p-n junction. When the gate of the IGBT is turned on, the n-type impurity layer33and the emitter region35become conductive and have an identical potential. However, since the emitter region35shares a common contact area with the p base layer34, it becomes difficult to apply a voltage to the p-n junction formed by the p base layer34and the n-type impurity layer33by turning the gate on. Accordingly, it becomes difficult for the holes to be injected into the p-n junction, and, as a result, a forward voltage drop (Vf) increases.

In this way, by using the RC-IGBT in which the IGBT and the diode are formed within an identical chip, the number of components further decreases as compared with the case where the IGBT and the diode are individually used, and easiness to produce the power module increases.

The RC-IGBT may be formed as a silicon semiconductor device, or, alternatively, may be formed as a silicon carbide semiconductor device or a gallium nitride semiconductor device.