Electric power conversion equipment

A two level inverter having a switching element the switching frequency of which is higher than that of GTO, and providing a continuous control substantially over a full range of output voltages wherein the two level inverter comprises as its pulse modes an asynchronous bipolar modulation mode, a multi-pulse mode of an overmodulation mode, and a one pulse mode, and wherein advantageously the multi-pulse mode and the one pulse mode are arranged to be changed, thereby to enable a continuous control of its output voltage and reduce tone changes in the magnetic noise as well.

BACKGROUND OF THE INVENTION 
This invention relates to power conversion equipment which converts a 
direct current into an alternating current or vice versa, and in 
particular it relates to an apparatus for controlling a pulse width 
modulation (PWM) inverter. 
An example of prior art inverter modulation methods has been disclosed in 
an article entitled "Evaluating the Recent Inverter Control Methods" which 
appeared in the "Denkisha no Kagaku", pp 14, FIG. 1, April 1993, published 
by the Denkisha Kenkyuukai. 
As illustrated in FIG. 2 herein, in a traction motor inverter a constant 
control is provided to maintain a constant ratio between the magnitude and 
the fundamental frequency of an output voltage when the fundamental 
frequency of the output voltage is low (which region will be referred to 
as a variable-voltage variable-frequency region), whereas a frequency 
control is provided while maintaining a maximum voltage value in a region 
in which the output voltage fundamental frequency increases such that the 
magnitude of its output voltage becomes a maximum (which region will be 
referred to as a constant-voltage variable-frequency region). Since in the 
variable-voltage variable-frequency region the output voltage is regulated 
by pulse width modulation control, a multi-pulse mode is utilized in which 
a half cycle of its output voltage comprises a plurality of voltage 
pulses. On the other hand, in the constant voltage variable-frequency 
region, in order to maximize its voltage usage rate and to minimize the 
size of the equipment, a one pulse mode is utilized in which a half cycle 
of its output voltage comprises a single broad pulse. 
A prior art inverter which uses GTO thyristors as its switching elements 
(referred to as a GTO inverter hereinafter), has adopted a multi-pulse 
mode according to a pulse count switching method which decreases the 
number of pulses to be included in one cycle thereof with increasing 
output voltage fundamental frequency, as shown in FIG. 3. This is due to 
an upper limitation of several hundred Hz imposed on a switching frequency 
in the GTO thyristor. In this method, however, a magnetic noise 
accompanies tone variation, thereby generating a cacophonic noise due to 
discontinuity between switching frequencies at the time the pulse count 
switching is employed. 
Further, there is another problem associated with the GTO thyristor in that 
a voltage jump of approximately 10% occurs between a three-pulse mode 
output voltage (containing three pulses in a half cycle of its output 
voltage and a one-pulse mode output voltage) containing one pulse, 
depending on the limitation of a minimum OFF time in the GTO thyristor, 
thereby causing a fluctuation to take place in the torque generated by the 
traction motor at the time of switching from the three-pulse mode to the 
one-pulse mode. 
SUMMARY OF THE INVENTION 
A principal object of the invention is to provide a two-level inverter 
apparatus which is capable of controlling the magnitude of its output 
voltage from zero to a maximum value through a combination of a 
multi-pulse mode and a one-pulse mode, which substantially eliminates the 
discontinuity between the switching frequencies, thereby reducing the 
cacophonic magnetic noise accompanying tone variations, and which enables 
continuous control over substantially the entire region of the output 
voltage, by minimizing a gap in the output voltages between the 
multi-pulse mode and the one-pulse mode. 
The aforementioned main object of the invention is achieved by the inverter 
apparatus according to the invention, which comprises: a multi-pulse 
generator for generating a gate control signal for outputting a bipolar 
modulation voltage which is controlled by pulse width modulation having a 
uniform pulse generation cycle over any cycle of its output voltage 
fundamental wave, and a gate control signal for outputting an 
overmodulation voltage the pulse width of which is widened in the vicinity 
of a peak of the output voltage fundamental wave relative to that in the 
vicinity of a zero cross point thereof; a one-pulse generator for 
generating a single pulse having the same polarity as that of the output 
voltage fundamental wave; and a selection circuit for selecting the output 
from either the multi-pulse generation means or the one pulse generation 
means, dependent on a condition of the number of pulses contained in one 
cycle of the output fundamental wave, a magnitude of an output voltage, a 
modulation ratio, or an output voltage fundamental frequency. 
The multi-pulse generator is arranged so that a pulse generation cycle of a 
pulse width modulated portion on an output voltage waveform can be set 
independently of the output voltage fundamental frequency. Thus, its 
switching frequency at the time of bipolar modulation can be maintained 
constant, and its switching frequency at the time of overmodulation can be 
caused gradually to approach a predetermined switching frequency set for 
at the time of one-pulse, thereby eliminating discontinuity between 
switching frequencies. 
Further, by controlling both the output voltage at which changes between 
the multi-pulse mode and the one-pulse mode are enabled, and the phase 
relative to the output voltage fundamental wave, it is possible to achieve 
the change between the modes smoothly, without causing fluctuations in the 
current and torque that the motor generates. 
Other objects, advantages and novel features of the present invention will 
become apparent from the following detailed description of the invention 
when considered in conjunction with the accompanying drawings.

PREFERRED EMBODIMENTS 
Preferred embodiments of the present invention will be described in the 
following with reference to FIGS. 1 to 17. 
A diagram which illustrates a PWM mode of inverter operation according to 
the invention is shown in FIG. 4. The inverter is operated in a bipolar 
mode in its low output voltage range, in an overmodulation mode in its 
high output voltage range, and in a one pulse mode in its maximum output 
voltage range respectively. 
FIG. 1 is a schematic diagram of one embodiment of the invention, wherein 
numeral 6 is an induction motor, 5 is a two-level three-phase PWM inverter 
which drives the induction motor, 9 is an inverter power source from an 
aerial direct current supply, and 7, 8 denote respectively a filter 
reactor and a capacitor provided on the DC input side of the inverter. 
A multi-pulse generator 2, a one-pulse generator 3 and a PWM mode select 
means 4 respectively generate inverter control signals Su, Sv, Sw (the 
subscript x representing any phase of u, v, w) in response to an output 
voltage command E.sup.* and a frequency command Fi.sup.*. The frequency 
command Fi.sup.* is integrated in integrator unit 1, to generate an output 
voltage fundamental wave phase .theta.x of each phase. The inverter 
control signals S1.sub.x, S2.sub.x and S.sub.x are characterized by a 
switching function which is defined to take 1 when the positive branch of 
the inverter is ON, and take 0 when the negative branch thereof is ON. 
An example of the multi-pulse generator 2 is shown in FIG. 5 (which 
indicates only one phase component of its contents). Here, switching 
functions in the bipolar mode and the overmodulation mode are generated by 
the same generator. An output voltage command-to-modulation ratio 
convertor 21 generates a signal indicative of a modulation ratio A, (that 
is, the amplitude of a modulation wave relative to that of the carrier), 
from an output voltage command E*. Assuming a carrier amplitude of 1, the 
ratio takes on a value of 0.ltoreq.A.ltoreq.1 in the bipolar mode, and 
that A&gt;1 in the overmodulation mode. So that the magnitude of the output 
voltage fundamental wave coincides with the voltage command the 
relationship between E* and A is defined by equation 1 in the bipolar 
mode, and by equation 2 in the overmodulation mode. 
##EQU1## 
When the sinusoidal wave generator 22 generates an output y=sin (x), 
sin.theta.x of the output voltage fundamental wave phase .theta.x 
(equivalent to modulation phase) is obtained. The latter is multiplied by 
the modulation ratio A to produce a modulation wave a.sub.x, which is 
input to a switching function computer 24, together with a carrier 
frequency Fc (which is equivalent to the switching frequency in the 
bipolar mode). The switching function computer 24 generates a carrier of 
triangular pulses having an amplitude of 1 and a frequency of Fc, which is 
compared with the value of the modulation wave a.sub.x to generate a 
switching function S1.sub.x. Alternatively, the switching function may be 
obtained without using such triangular pulses, through computation of the 
modulation wave ax and its pulse intervals. 
FIGS. 6 and 7 show examples of waveforms of the switching functions 
according to the bipolar mode and the overmodulation mode, obtained 
through comparison with the triangular pulses. 
In the inverter equipment according to the present invention, devices such 
as IGBTs, large capacity transistors or the like which can be operated at 
switching frequencies of several kHz are utilized as a switching element 
(hereinafter referred to as an IGBT inverter in general), and the 
modulation wave and the carrier are arranged to be asynchronous in the 
multi-pulse mode of operation. Thereby, the switching frequency is 
maintained constant in the bipolar mode, and gradually approaches a 
predetermined switching frequency in the one pulse mode which will be 
described later. 
Since the modulation wave and the carrier wave are asynchronous in the 
multi-pulse mode, it is necessary for the carrier to have a substantially 
higher frequency (preferably approximately ten times higher, based on 
experience) than the modulation frequency. 
An example of a switching function waveform which is generated by the one 
pulse generator of FIG. 1 is shown in FIG. 8. When the sign of the 
fundamental wave of output voltage is positive, irrespective of its 
amplitude, the value of switching function S2.sub.x is 1, and when the 
sign is negative the value of S2.sub.x is 0. 
Next, an arrangement for combining the multi-pulse mode and the one pulse 
mode so that the high output voltage range can be controlled smoothly will 
be described in the following. 
A prior art overmodulation method is discussed in a publication entitled 
"Overmodulation control method for voltage-type three-phase PWM inverters" 
which appeared in the National Convention Proceedings No. 106, Heisei 3, 
Industrial Application Division, the Japan Society of Electric Engineers. 
According to this prior art, a six-step inverter operation is overextended 
to increase the modulation ratio in the overmodulation mode, thereby 
covering one pulse mode operation. However, when the one pulse mode is 
achieved by extending the overmodulation mode (that is, the overmodulation 
is realized by excessively increasing the modulation ratio) certain 
problems occur as follows. 
Once problem is that the point at which the overmodulation mode is changed 
to the one pulse mode depends on its switching frequency. Thus, it cannot 
be set at discretion. 
Another problem is that in the case where a modulation wave and a carrier 
wave are asynchronous in the overmodulation mode, pulses of the modulation 
wave in the vicinity of the zero-cross point may or may not be generated 
in the vicinity of a change between the overmodulation mode, and the 
one-pulse mode due to the influence of turn-on and turn-off time of the 
devices. As a result, the positive and the negative output voltages are 
balanced, thereby causing a beat phenomenon to be introduced by which a 
low frequency pulsation is superimposed on a load current of the inverter. 
A third problem is that with reference to FIG. 7, an output voltage 
waveform (which is equivalent to the waveform of a switching function to 
be described later) can be divided into two portions: one having a uniform 
pulse interval (equidistance pulse), that is, having a uniform pulse 
generation cycle in the vicinity of the zero crossing of the modulation 
wave; and the other portion having a single broad pulse width component 
with its center corresponding to a peak of the modulation wave. Therefore, 
a switch-over from the overmodulation mode to the one-pulse mode may occur 
in any portion having an equidistance pulse in the overmodulation mode. In 
this instance, load current in the inverter fluctuates, which may result 
in a breakdown of switching elements due to overcurrents or a substantial 
fluctuation in motor torque. 
In order to solve these problems, it is necessary to specify a particular 
shift voltage (hereinafter referred to as a shift voltage) at which a 
shift between the overmodulation and the one-pulse mode is allowed, and a 
particular phase (hereinafter referred to as a shift phase) in the output 
voltage fundamental wave at which the shift should be executed. 
First, determination of the shift voltage will be described below. 
It is desirable for the shift voltage to be set at a value in close 
proximity to the output voltage in the one-pulse mode operation (that is, 
close to 100%), because fluctuation in motor torque during mode shifting 
is minimized when the difference between its output voltage and its 
maximum value available is minimum. 
In the asynchronous PWM, however, individual voltage pulses contained in 
one cycle of the output voltage fundamental wave have a different pulse 
width according to each cycle, and when the number of pulses decreases in 
the vicinity of the zero crossing point of the output voltage fundamental 
wave as the output voltage in the overmodulation mode approaches 100%, its 
adverse effect becomes significant, causing an unbalance between the 
positive and negative polarities of output voltages, and eventually 
causing a beat phenomenon in the load current of the inverter. An example 
of such is shown in FIG. 9. 
FIG. 10 shows an example of the relationship between the average pulse 
number in the vicinity of the zero crossing point of the output voltage 
fundamental wave and current pulsation due to the beat phenomenon. A 
portion of the modulation wave where its absolute value is under 1.0 
corresponds to an equidistance pulse region as shown in FIG. 7, thereby, 
an average pulse number can be obtained by equation 3. Further, the 
current pulsation ratio is defined by equation 4. As can been seen from 
FIG. 10 unless at least one pulse is secured in the vicinity of zero cross 
point of the output voltage fundamental wave, the low frequency pulsation 
in the inverter load current due to the beat phenomenon becomes extremely 
great. 
##EQU2## 
Thereby, an appropriate shift voltage is preferably set at a value which 
ensures at least one voltage pulse in the vicinity of the zero cross point 
of the output voltage fundamental wave. Since this value depends on the 
output fundamental frequency Fi* and the carrier frequency Fc in the 
multi-pulse mode, it may be derived from calculation of these values. 
Alternatively, this value may be calculated and preset beforehand from an 
upper limit of the output voltage fundamental frequency Fi*. 
Next, the manner in which the shift phase is controlled will be described. 
Dependent on the phase of the output voltage fundamental wave at the time 
when a change between the overmodulation mode and the one-pulse mode 
occurs, the manner in which transient fluctuations appear in a load 
current in the inverter and in a torque generated immediately after the 
mode change differs. An example of such current fluctuations is shown in 
FIG. 11. FIG. 11(a) shows a case where all the three phases were changed 
in batch at the 0.degree. point of the U-phase of the output voltage 
fundamental wave (see FIG. 12), and a transient current fluctuation was 
observed immediately after the mode change. In contrast, FIG. 11(b) shows 
another case where all the three phases were changed in batch at 
90.degree. of the U-phase of the output voltage fundamental wave (FIG. 
13), and there was observed almost no transient current fluctuation. 
FIG. 14, illustrates the relationships between the shift phase (U-phase 
taken as a reference) of the output voltage fundamental wave when shifting 
all the three phases in batch from the overmodulation mode to the 
one-pulse mode and transient current fluctuation, where the current 
fluctuation rate is defined by equation 5. 
EQU (current fluctuation rate)=(transient peak current at mode change)-(peak 
steady current in one-pulse mode)!/(peak steady current in one-pulse 
mode)!.times.100(%) eq. 5 
In FIG. 14, at every 60.degree. of phase of the output voltage fundamental 
wave a large current fluctuation rate appears. This occurs where a mode 
change between the overmodulation and the one-pulse modes takes place 
while one of the three phases is in the equidistance pulse region in the 
overmodulation mode, thereby allowing mixed presence of both the modes, 
which in turn causes an increased transient unbalancing of the output 
voltages among three phases, and a resultant large transient current 
fluctuation. Therefore, according to the invention, such transient 
fluctuations in the current and torque can be successfully suppressed by 
assuring that the shift phase occurs in a portion where all the phases are 
in a wide pulse region and in the overmodulation mode as illustrated in 
FIG. 15. 
In order to provide for batch mode changes of all three phases from the 
overmodulation mode to the one-pulse mode, it is necessary to secure a 
region where the output voltages in the overmodulation mode for all the 
three phases coincide with a wide pulse region. For this purpose, it must 
be arranged that at an intersection of modulation waves of any two of the 
three phases (that is, at the 30.degree., 90.degree., 150.degree., 
210.degree., 270.degree., or 330.degree. points of the reference U-phase 
modulation wave), the absolute value of the modulation wave is greater 
than 1.0. For example, in respect of 30.degree., where au=Asin30.degree.&gt;1 
(and thus A&gt;2), and where a correlation between the modulation ratio A and 
the output voltage E* in the overmodulation mode can be obtained by 
equation 2, it must hold that E*&gt;95.6%. Therefore, in order for a batch 
mode change for the three phases to be executed between the overmodulation 
and the one-pulse mode, the shift voltage must be set at a value which is 
greater than 95.6%, and has at least one voltage pulse in the vicinity of 
the zero dross point of the output voltage fundamental wave in the 
overmodulation mode. 
An example of a PWM mode select means 4 for implementing the 
above-mentioned shift voltage and shift phase controls of the invention is 
shown in FIG. 16. A mode select command generation means 42 compares a 
shift voltage Ec (set in a shift voltage setting unit 41) and a voltage 
command E*, and outputs a mode select command Mc indicating of which of 
the multipulse mode and the one-pulse mode is to be selected. 
The mode select command Mc has been described hereinabove as being obtained 
on the basis of the output voltage command E*. However, since the output 
voltage command E* corresponds uniquely to the modulation ratio A, a 
particular modulation ratio Ac corresponding to the shift voltage may be 
preset, which is then compared with the modulation ratio A to generate the 
mode select command Mc. 
Further, also in the variable voltage variable frequency region, since the 
output voltage command and the output voltage fundamental frequency 
correspond uniquely with each other, it may be arranged such that a 
particular output voltage fundamental frequency Fic which corresponds to 
the shift voltage may be preset beforehand. This output voltage 
fundamental frequency is then compared with the frequency command Fi* to 
generate the mode select command Mc. 
A shift phase control means 44 reads Mc, and in case a mode change is 
required, compares phase .theta.x of the output voltage fundamental wave 
and a shift phase .theta.c set in a shift phase set means 43; then if 
.theta.x=.theta.c, it changes its mode select signal M. Dependent on the 
mode select signal M, mode select switches 45, 46, 47 select either one of 
output S1.sub.x of the multipulse generation means and output S2.sub.x of 
the one-pulse generation means, then determine its switching function 
S.sub.x. 
The shift phase may also be controlled by taking absolute values of the 
modulation wave at respective phases. If all exceed 1.0 (which indicates 
that all the three phases lie in a wide pulse region in the overmodulation 
mode), a change of output from between the multipulse generation means and 
the one-pulse generation means is enabled at this instance. 
Thereby, a two-level inverter according to the invention has been provided 
which features the advantage that a gap in the output voltages at the time 
of mode change between the multipulse mode and the one-pulse mode can be 
reduced as small as 1-2% in comparison with approximately 10% in the 
conventional GTO inverter, thereby allowing continuous control of the 
output voltage from zero to its maximum value. In addition, a smooth 
change can be ensured between the multipulse mode and the one-pulse mode 
operation without causing any fluctuations in the current and the 
generated torque. 
FIG. 17 illustrates the relationship between the output voltage fundamental 
frequency and the switching frequency, where there exist no such large 
discontinuities such as observed in FIG. 3 in respect of the conventional 
inverter modulation method. Thus, discontinuous tone change occurring due 
to the magnetic noise is eliminated. 
In the inverter equipment which is capable of controlling its output 
voltage from zero to a maximum voltage by use of the multi-pulse mode and 
the one-pulse mode in conjunction according to the invention, 
discontinuous changes in the magnetically-induced noise can be eliminated, 
and a substantially continuous control over the full range of its output 
voltage can be accomplished as well. 
Although the invention has been described and illustrated in detail, it is 
to be clearly understood that the same is by way of illustration and 
example, and is not to be taken by way of limitation. The spirit and scope 
of the present invention are to be limited only by the terms of the 
appended claims.