Extended range boost converter circuit

A power converter comprises a power transformer having a primary winding and a secondary winding flux coupled to the primary winding, input terminals for receiving an input voltage, and output terminals for providing an output voltage. A power switching arrangement comprising two pairs of switching devices arranged in a bridge configuration has a corresponding duty cycle which is selectable so as to cause the power converter to manifest different input output transfer characteristics corresponding to buck and boost modes of operation. An inductor having a single winding is coupled between the input terminals and the primary winding through the switching devices. Secondary winding rectification and filtering provides the power supply output. A reset circuit coupled to the output terminal of the single winding inductor and connected to the input terminals provides a current path for discharging the inductor during a predetermined time interval corresponding to that portion of the duty cycle when all of the switching devices are off to enable operation of the power supply in a buck mode of operation.

FIELD OF THE INVENTION
 The present invention relates to electrical power converters and, more
 particularly, to an electric power converter that can operate with an
 input voltage varying over a wide range.
 BACKGROUND OF THE INVENTION
 Converter circuits, such as DC-to-DC converters, are often used in
 electronic systems of the type, such as avionics systems and the like,
 where an electronic-regulated power supply is required to operate even
 though energized with an input voltage which varies over a very wide input
 voltage range. One such regulated power supply is commonly known as a
 boost converter. In general, a boost converter circuit operates to boost
 the input voltage to generate a higher output voltage. A conventional
 boost converter circuit 10 is depicted in FIG. 1 (Prior Art), wherein a DC
 input voltage Vin is applied at an input terminal 10a with reference to a
 common terminal 10c. An output voltage Vout is developed at an output
 terminal 10b with reference to common terminal 10c (e.g. ground
 potential), and thus appears across a capacitor 18. An inductor 12 has a
 first terminal 12a coupled to input terminal 10a and a second terminal 12b
 coupled to both the anode of a rectifier diode 14 and a drain element of a
 switching device 16. As one skilled in the art understands, switch 16
 (which is coupled between the output side of the boost inductor 12 and
 ground terminal 10c) is switched on and off responsive to the switching
 device gate electrode drive signal, which has a duty cycle (i.e. ratio of
 `on` or `off` portion to an entire on-off cycle) D, which is never greater
 than 1. In each switching cycle or duty cycle D, energy is stored in the
 inductor 12 when the switch is closed or conducting (ON period) and
 released to output terminal 10b via diode rectifier 14 when the switch is
 opened or non-conductive (OFF period). Thus, energy is stored in inductor
 12 such that energy output from the inductor upon discharge is added to
 the input voltage Vin to produce an output voltage Vout that is greater
 than the input.
 FIGS. 2A and 3A illustrate conventional enhancements to the basic boost
 configuration shown in FIG. 1. FIG. 2A shows a converter 10' with a
 conventional transformer 20 forming a push-pull transformer-coupled boost
 converter operated in a boost mode (greater than 50% duty cycle). The duty
 cycles DQ1 and DQ2 associated with the switching devices 16-1 and 16-2 for
 this circuit are shown in FIG. 2B. FIG. 3A shows a conventional
 full-bridge transformer-coupled boost converter 10" operated in boost mode
 (greater than 50% duty cycle) with duty cycles DQ1 and DQ2 associated with
 the respective switching devices Q1, Q3 and Q2, Q4 driven by the switching
 waveforms as shown in FIG. 3B. Each of the converters shown produces an
 output voltage according to the equation Vo=N*Vin/(1-D) where D is the
 duty cycle of the circuit and N is the secondary winding-to-primary
 winding turns ratio of the transformer 20 (N=1 if no transformer, as in
 converter 10 of FIG. 1).
 From the foregoing, one can ascertain that, in any of the circuits depicted
 in these Figures, the output voltage has a range between Vin and an
 extremely large value. That is, the output voltage cannot be less than the
 product of the input voltage and the turns ratio. Since the boost circuit
 only stores energy in excess of the input voltage, such a circuit is
 inherently higher efficiency than a circuit that must store the entire
 output energy, such as a conventional flyback or buck-boost converter
 system. However, the inability to control the output voltage to a value
 less than the input voltage can produce significant problems, even when
 normal operation requires an output voltage greater than the voltage at
 the input. For instance, at startup, the output voltage is zero while the
 input voltage, when applied, is usually non-zero. This can lead to a very
 large current applied to raise the output voltage from zero to the input
 voltage. In addition, an abnormal condition such as a fault or short
 circuit at the output may also produce a condition where the output
 voltage may be less than the input voltage. Under both of these
 conditions, a boost converter is uncontrolled and the currents produced
 are not controllable. To permit operation under these conditions, it is
 customary to add a second switch in series with the boost inductor, and a
 flyback diode, so as to operate the boost converter as a buck-mode
 converter. This, however, results in energy loss associated with the
 additional switch, even when that switch is not in use. In addition, in
 applications where a rectified alternating-current (AC) waveform, such as
 a rectified sine wave, is used as the input source, it may be desirable to
 operate at a voltage that is less than the peak voltage of the input.
 Conventional transformer-isolated boost converter circuits, such as those
 depicted in Prior Art FIGS. 2A and 3A, include additional switches that
 operate to open connections between the input and the output terminals in
 order to steer the transformer flux as well as control large currents
 caused by the above-described conditions. Opening of these switches,
 however, has the undesirable effect of interrupting the current flowing in
 the boost inductor. Since the energy stored in the boost inductor no
 longer has a path through which to flow, it will discharge through
 whatever element it can, thereby destroying the device. Thus, for
 conventional boost converters, operation in a buck mode (where the
 switches are off for a given time interval) is not permissible. Adding an
 additional winding to the boost inductor as disclosed in commonly assigned
 U.S. Pat. No. 5,654,881, entitled "Extended Range DC--DC Power Converter
 Circuit" issued Aug. 5, 1997 to Albrecht et al, the subject matter of
 which is herein incorporated by reference, allows the flux in the inductor
 to be continuous and produce a buck operating range where the output can
 be less than the input. However, use of additional windings and associated
 circuitry to provide an extended range converter proves to be quite costly
 in most applications. Furthermore, the voltage on the switches when the
 inductor is discharged may be less than optimal. Still further, it is
 known that boost converters suffer from parasitic losses such as loss due
 to leakage inductance, resulting in undesirable energy loss and circuit
 inefficiency. Accordingly, a power converter which overcomes these
 problems and which obviates the need for additional windings to operate
 over an extended range of voltages, is highly desired.
 SUMMARY OF THE INVENTION
 In accordance with the invention, a power converter comprises: a power
 transformer having a primary winding and a secondary winding with a
 secondary winding flux coupled to the primary winding; input terminals for
 receiving an input voltage; and output terminals for providing an output
 voltage. A power switching arrangement comprising two pairs of switching
 devices arranged in a bridge configuration has a corresponding duty cycle
 which is selectable so as to cause the power converter to manifest
 different input-output transfer characteristics corresponding to buck and
 boost modes of operation. An inductor having a single winding is coupled
 between the input terminal and the primary winding through the switching
 devices. Secondary winding rectification and filtering provides the power
 supply output. A reset operating circuit coupled to the output terminal of
 the single winding inductor and connected to the input terminals provides
 a current path for discharging the inductor during a predetermined time
 interval corresponding to that portion of the duty cycle when all of the
 switching devices are off, to enable operation of the power supply in a
 buck mode of operation.
 A buck-boost converter can comprise a power transformer having a primary
 winding and a secondary winding with a secondary winding flux coupled to
 the primary winding, input terminals for receiving an input voltage,
 output terminals for providing an output voltage, a single winding
 inductor coupled between the input and the primary winding, a switching
 arrangement comprising a plurality of switches to be turned on and off
 according to a duty cycle for controllably causing a flow of current
 through the primary winding, with a plurality of unidirectional conduction
 devices coupled to the transformer secondary winding for rectifying flux
 coupled energy to provide an output voltage Vo to the output terminals,
 and a reset operating circuit coupled to an output terminal of the
 inductor and operable in a first mode for providing a current path for
 discharging the inductor during a predetermined time interval associated
 with a portion of the duty cycle of the switching arrangement when the
 plurality of switches are each non-conducting ("off") to enable the boost
 converter to operate in a first mode where the voltage Vo is lower than
 the input voltage, and operable in a second mode for providing a path for
 discharging energy associated with leakage inductance reflected to the
 output terminal of said single winding inductor when the boost converter
 is operated in a second mode wherein the plurality of switches are each
 conducting ("on") during a same portion of the duty cycle.

DETAILED DESCRIPTION OF THE INVENTION
 Referring now to FIG. 4, there is shown a circuit schematic of a boost
 power converter 40 according to my present invention, which is operable in
 both boost and buck modes according to an aspect of the present invention.
 A DC input voltage Vin is applied between input terminal 40a and a common
 terminal 40c; a DC output voltage Vo at an output current Io is provided
 at an output terminal 40b, with respect to an output common terminal 40d.
 The output voltage Vo appears across a filter capacitance 48.
 A boost inductor 42 is a single-winding inductive element having a first
 terminal 42a coupled to input terminal 40a and a second terminal 42b
 connected both to a switching arrangement 60 and to an anode terminal 44a
 of a diode 44 which is indicative of a unidirectional current element.
 Switching arrangement 60 comprises first, second, third and fourth current
 conductive switching elements 62, 64, 66 and 68, respectively, connected
 in a conventional bridge configuration. Each of the conductive switching
 elements is preferably an active switching device such as a power MOSFET
 and responsive to control signals G1 and G2 at gate terminals thereof. In
 the preferred embodiment, conductive switching elements 62 and 64 are
 responsive to control signal G1 at a first (e.g. "high") level to turn
 "ON" or conduct, and at a second (e.g. "low") level to turn "OFF" or
 non-conduct, with a duty cycle DQ1 as shown in FIG. 5A. In similar
 fashion, conductive switching elements 66 and 68 are responsive to control
 signal G2 so as to turn ON and OFF with a duty cycle DQ2 as shown in FIG.
 5A. A pulse-width modulator (PWM) and control means 52 has an input 52a
 monitoring the magnitude of the output voltage V.sub.o and another input
 52c receiving a periodic clock CLK signal. The clock signal establishes
 the operating frequency of the Boost/Buck converter; an operating
 frequency in excess of 1 KHz is generally desirable. Controller means 52
 has respective first and second outputs 52b-1 and 52b-2 at which the first
 and second switching device gating, or control, signals G1 and G2
 respectively are provided to turn respective device pairs 62, 64 and 66,
 68 into the conductive, or ON, condition or into the non-conductive, or
 OFF, condition. It should be understood that each illustrated device may
 be a single power-switching device, of semiconductive or other form, or
 may be plural devices (as necessary to properly switch the required
 current and/or voltage) controlled in unitary fashion. It should be
 further understood that the a push-pull topology is also contemplated
 wherein the pairs of switching devices 62, 64 and 66, 68 may be replaced
 with two single switching devices and corresponding transformer coupling
 analogous to that illustrated in FIG. 2A. Controller means 52 further
 includes third output 52b-3 at which gating control signal G3 is provided
 to another switching device 46 to enable the device to turn ON or conduct,
 or to turn off (i.e. disable the device) so that the switch is in a
 non-conductive state.
 In operation, means 52 input 52a monitors the voltage at output 40b and, by
 any of various well-known means, compares the actual output voltage
 V.sub.o to a selected output value; means 52 then controls the converter
 duty cycle, responsive to this determination, to regulate and maintain
 V.sub.o at the selected value. Means 52 thus determines, at start-up, if
 the output voltage is greater than, or less than the desired value; this
 can also be thought of as determining if the input voltage V.sub.IN is
 greater, or less, than N*V.sub.o. In the case where the output voltage
 V.sub.o is less than the input voltage, means 52 adjusts the outputs 52b
 to control operation of converter 40 in the Buck mode; the converter is
 operated in the Boost mode when the input voltage V.sub.o is less than the
 selected output voltage value. For Buck mode operation, if the duty-cycle
 is less than 50% (i.e., the ratio of switch ON conduction time to the
 total of switch ON and switch OFF, or non-conduction times in one ON/OFF
 cycle), then the Buck mode is being used. Buck mode utilizes operating
 cycle segments (of duration T) during which both switch pairs are
 non-conductive of OFF (see the DQ1, DQ2 waveforms in FIG. 5B). Also, in
 Buck mode, a switch pair is conductive, or ON, during a time segment
 different from the time segment during which the other switch pair is ON.
 Diode 50 forms part of reset operating circuit 54 which includes a
 capacitor 62, another inductor 64, switching element 46 and diode 50.
 Diode 44 has its anode electrode connected to terminal 12b of the boost
 inductor and its cathode terminal connected to a controlled current
 circuit such as a drain-source circuit of switching device 46, and to a
 first terminal of capacitor 62. Capacitor 62 has a second terminal
 connected to first boost inductor terminal 12a and to a first terminal 64a
 of second inductor 34. Second inductor second terminal 64b is connected to
 the cathode electrode of diode 50 and to switching device 46. The anode
 electrode of diode 50 is coupled to common reference potential 40c.
 A transformer 70 has a primary winding 70p coupled to switching arrangement
 60 and a secondary winding 70s coupled to output terminal 40b through
 unidirectionally-conducting elements 72 and 74, such as semiconductor
 diodes. In particular, the primary winding has a first end 70pa connected
 to the controlled currents of first and fourth switching devices 62 and
 68, and a second end 70pb connected to the controlled currents of second
 and third switching devices 64 and 66. Secondary winding 70s includes a
 first portion 70s1 and a second portion 70s2. Each secondary winding
 portion has first end 70sa and 70sb connected to an anode electrode of an
 associated one of the like-poled diodes 72 and 74 whose cathodes are
 connected in common to capacitor 18 and output terminal 10b. The second
 ends are connected to secondary winding center tap 70sc. The transformer
 secondary windings have essentially equal turns coupled to a core so that
 a first secondary voltage Vs1 of a first polarity appears at a first
 secondary winding end, while a second secondary voltage Vs2 appears at the
 second secondary winding end. As shown in FIG. 4, the relationship between
 primary and secondary windings is 1:N with the windings phased as
 indicated by the illustrated phasing dots.
 In describing normal boost mode operation of the converter 40, reference is
 made to the waveforms illustrated in FIGS. 5A, 5B and 5C. At time to,
 switching devices 62 and 64 are off and 66 and 68 are on. Therefore,
 current Ix passes from the input through boost inductor 42. Current
 conductive devices 66 and 68 enable the current to pass through the
 primary winding of transformer 70 and back to the input terminals. During
 this interval, the inductor is discharging. At time t1, switching devices
 62 and 64 are turned on so that all four switching transistors 62, 64, 66,
 68 are on. This time interval is considered the "ON" time for boost duty
 cycle calculations. When all four transistors are ON, a short circuit
 current exists across the primary winding of transformer 70. Diodes 72 and
 74 operate to block the output voltage Vo so that it is not applied back
 to the primary winding of transformer 70, and the voltage across the
 switching arrangement 60 is pulled down to zero. Thus, the voltage at
 terminal 42b of the boost inductor is clamped to zero volts when all four
 switching devices are ON. The entire input voltage is thus applied across
 boost inductor 42, causing the current to ramp up such that energy is
 stored in the inductor. At time t2, switching devices 66 and 68 are turned
 off while 62 and 64 remain on. The voltage is then applied across the
 primary winding 70p in the opposite direction and the energy stored in the
 inductor 42 now discharges to the output terminal. For normal boost mode
 operation, the switching devices 62, 64, 66, 68 in the bridge circuit
 arrangement 60 operate in an overlap condition as illustrated in FIG. 5A.
 The ratio of the time that all four of the switches are ON (corresponding
 to when energy is being stored in inductor 12) to the time that two of the
 four switches are OFF (corresponding to when energy is being discharged
 from the inductor) is the effective duty cycle D of the circuit. The time
 when only the diagonal switches (62, 64 or 66, 68) of bridge circuit
 arrangement 60 are ON (or when only one switch of a push-pull switch
 arrangement) the voltage at terminal 42b is given as the output voltage Vo
 divided by the turns ratio N of the transformer. As the boost duty cycle
 approaches zero boost (corresponding to DQ1 and DQ2 each at 50% duty cycle
 and 180 degrees out of phase) the output voltage Vo approaches N times the
 input voltage Vin and the voltage across inductor 42 approaches zero. FIG.
 5B illustrate the timing relationships associated with the switching
 arrangement for DQ1 and DQ2 under these conditions.
 When the switches are controlled in a manner so that the boost duty cycle
 goes below zero boost (corresponding to less than 50% switching ON time),
 then there exists a time period where all of the switches are OFF (i.e.
 periods t2-t1, t4-t3). Under these conditions, depicted in FIG. 5C, the
 inductor current Ix is maintained by diode 44 and capacitor 62 by
 providing an electrical communication path 44A for the inductor current to
 flow. That is, the inductor current Ix flows through diode 44 into
 capacitor 62 to terminal 42a of boost inductor 42 and circulates through
 the boost inductor. The input current from the source then goes to zero.
 Because all four switches are OFF during the aforementioned time periods,
 the current in the switching arrangement 60 also falls to zero. However,
 the inductor current (which cannot drop to zero) flows through diode 44
 and capacitor 62. In this manner energy is transferred from inductor 42 to
 capacitor 62 so as to develop a charge (i.e. voltage Vc) across capacitor
 62. The voltage Vc across capacitor 62 is thus the discharge voltage of
 inductor 42. Controllable switch 46 is then turned ON via controller means
 52 through control signal G3 to cause capacitor 62 to discharge through
 second inductor 64 via electrical communication path 44b causing energy to
 be stored in second inductor 64. When switching device 46 is then turned
 off, the current I.sub.L in second inductor 64 will continue to flow and
 cause the voltage across second inductor 64 to reverse polarity. This
 causes diode 50 to become forward biased, causing the current from the
 inductor to flow back into the input and thereby reducing the average
 current drawn from the source. In the preferred embodiment, switching
 device 46 is switched in synchronism with the OFF condition of DQ1 and
 DQ2. In the general case, however, it is to be understood that the
 switching of device 46 can occur at a fixed duty cycle K proportional to
 the time that all four switching devices 62, 64, 66, and 68 are OFF.
 Therefore, by connecting reset operating circuit 54 to the output of the
 single winding boost inductor 42 as described, a path is provided for the
 energy stored in boost inductor 42 to flow when the switches in power
 switch arrangement 60 are off and current in the output is interrupted.
 This allows the boost converter having a single winding boost inductor to
 function over a wider operating range than possible with a conventional
 boost converter circuit. Note also that the switching of MOSFET transistor
 device 46 to enable electrical communication between second inductor 64
 and diode 50 operates to invert and scale the voltage across capacitor 62
 to the level of the input. As one can ascertain, the magnitude of this
 voltage may be controlled by both the duty cycle of the switches in bridge
 switching arrangement 60 and of reset switch 46.
 The integral of the voltage across the inductor must be zero over a full
 cycle in steady state operation. Note that in order to maintain a zero
 integral of voltage on the inductor 42, this voltage must be at least
 (Vin-Vo)*.alpha. where .alpha. is the buck mode duty-cycle, defined as the
 time either switch pair (62,64 or 66, 68) is ON divided by the total
 ON-and-OFF time period T. As shown in FIG. 6, operation of the circuit in
 this mode produces a transfer function 90 different from that of a normal
 buck mode (reference numeral 92), while permitting the voltage to drop
 below the input for one ratio of voltage to the ratio of duty cycle on the
 buck reset switch. In this manner the voltage to which the inductor 42 can
 flyback can be controlled by the duty cycle of the reset switch device 46.
 This enables one to maintain an OFF state voltage on the switching
 arrangement switches that is substantially less than would be produced by
 resetting the inductor with an additional winding, as is commonly used.
 While the transfer function becomes more nonlinear, proper operation is
 still maintained and the voltage on the switching devices is reduced. FIG.
 7 shows the effects on the transfer functions of varying the reset voltage
 ratio, i.e. the ratio of ON time to OFF time, for the reset switch circuit
 54.
 Another advantageous development of operating circuit 54 is realized when
 operating the converter 40 in normal boost mode. While operating circuit
 54 is not essential for operation in boost mode, switching of switch
 device 46 at a duty cycle such that the combination of the reset voltage
 and input voltage is the same as the output voltage permits the circuit to
 recover certain energy loss. As is understood by one skilled in the art, a
 voltage spike on switching circuitry 60 may develop due to parasitic
 inductance or leakage inductance in the transformer or inductance in the
 output, when the current is zero. When operating in overlap boost mode
 (see FIG. 5A) such that only two of the four switches are on and current
 flows through inductor 42 through the operating switches to the
 transformer primary winding, any inductance in the transformer primary has
 zero current. Therefore, current ramp-up results in a voltage spike
 developing across the switches caused by parasitic or leakage inductance.
 If switch 46 is switched at a duty cycle such that the reset voltage plus
 the input voltage is substantially equal to the output voltage of the
 buck, then no current will flow in the reset operating circuit 54 since
 the load will absorb all of the current. Any energy reflected back to the
 output of the buck boost inductor 42 will attempt to increase this voltage
 over the output voltage. This increased voltage at terminal 42b causes the
 operating circuit 54 to conduct the energy back to the input. More
 particularly, the energy associated with the leakage inductance is applied
 to capacitor 62 via diode 44 and circulated back to the input through
 operation of switch 46 so that energy stored in the leakage inductance can
 be recovered. In this manner, the reset circuit acts to snub the voltage
 spike due to the leakage inductance. Therefore, the reset circuit, in
 addition to permitting operation where the output is less than the input
 (buck mode), may also be used as a loss recovery circuit during normal
 boost mode.
 While one presently preferred topology has been described herein, other
 converter topologies such as half bridge and push-pull topologies, can be
 equally utilized as well. It is our intent therefore, to be limited only
 by the scope of the appended claims and not by the specific embodiments or
 details described herein.