Analog dereverberation system

Room reverberation characteristic of monaural systems is removed, in accordance with the principles of this invention, by employing two microphones at the sound source and by manipulating the signals of the two microphones to develop a single nonreverberant signal. Both early echoes and late echoes in the signal received by each microphone are removed by manipulating the signals of the two microphones in separate frequency bands of the signal. Corresponding frequency bands of the two signals are co-phased and added and the magnitude of each resulting frequency band is modified in accordance with a computed phase difference average between the corresponding frequency bands. The modified frequency bands are combined, thereby forming the nonreverberant signal.

BACKGROUND OF THE INVENTION 
1. Field of the Invention 
This invention relates to signal processing systems and, more particularly, 
to systems for reducing room reverberation effects in audio systems such 
as those employed in "hands free telephony." 
2. Description of the Prior Art 
It is well known that room reverberation can significantly reduce the 
perceived quality of sounds transmitted by a monaural microphone to a 
monaural loudspeaker. This quality reduction is particularly disturbing in 
conference telephony where the nature of the room used is not generally 
well controlled and where, therefore, room reverberation is a factor. 
Room reverberations have been heuristically separated into two catergories: 
early echoes, which are perceived as spectral distortion and their effect 
is known as "coloration," and longer term reverberations, also known as 
late reflections or late echoes, which contribute time-domain noise-like 
perceptions to speech signals. An excellent discussion of room 
reverberation principles and of the methods used in the art to reduce the 
effects of such reverberation is presented in "Seeking the Ideal in 
`Hands-Free` Telephony," Berkley et al., Bell Labs Record, November 1974, 
page 318, et seq. Therein, the distinction between early echo distortion 
and late reflection distortion is discussed, together with some of the 
methods used for removing the different types of distortion. Some of the 
methods described in this article, and other methods which are pertinent 
to this disclosure, are organized and discussed below in accordance with 
the principles employed. 
U.S. Pat. No. 3,786,188, issued Jan. 15, 1974, described a system for 
synthesizing speech from a reverberant signal. In that system, the vocal 
tract transfer function of the speaker is continuously approximated from 
the reverberant signal, developing thereby a reverberant excitation 
function. The reverberant excitation function is analyzed to determine 
certain of the speakers's parameters (such as whether the speaker's 
function is voiced or unvoiced), and a nonreverberant speech signal is 
synthesized from the derived parameters. This synthesis approach 
necessarily makes approximations in the derived parameters, and those 
approximations, coupled with the small number of parameters, cause some 
fidelity to be lost. 
In "Signal Processing to Reduce Multipath Distortion in Small Rooms," The 
Journal of the Acoustics Society of America, Vol. 47, No. 6 (Part I), 
1970, pages 1475 et seq. J. L. Flanagan et al describe a system for 
reducing early echo effects by combining the signals from two or more 
microphones to produce a single output signal. In accordance with the 
described system, the output signal of each microphone if filtered through 
a number of bandpass signals occupying contiguous frequency ranges, and 
the microphone receiving greatest average power in a given frequency band 
is selected to contribute that signal band to the output. The term 
"contiguous bands" as used in the art and in the context of this 
disclosure refers to nonoverlapping bands. This method is effective only 
for reducing early echoes. 
In U.S. Pat. No. 3,794,766, issued Feb. 26, 1974, Cox et al. describe a 
system employing a multiplicity of microphones. Signal improvement is 
realized by equalizing the signal delay in the paths of the various 
microphones, and the necessary delay for equalization is determined by 
time-domain correlation techniques. This system operates in the time 
domain and does not account for different delays at different frequency 
bands. 
In U.S. Pat. No. 3,662,108, issued on May 9, 1972, to J. L. Flanagan, a 
system employing cepstrum analyzers responsive to a plurality of 
microphones is described. By summing the output signals of the analyzers, 
the portions of the cepstrum signals representing the undistorted acoustic 
signal cohere, while the portions of the cepstrum signals representing the 
multipath distorted transmitted signals do not. Selective clipping of the 
summed cepstrum signals eliminates the distortion components, and inverse 
transformation of the summed and clipped cepstrum signals yields a replica 
of the original nonreverberant acoustic signal. In this system, again, 
only early echoes are corrected. 
Lastly, in U.S. Pat. No. 3,440,350, issued Apr. 22, 1969, J. L. Flangan 
describes a system for reducing the reverberation impairment of signals by 
employing a plurality of microphones, with each microphone being connected 
to a phase vocoder. The phase vocoder of each microphone develops a pair 
of narrow band signals in each of a plurality of contiguous narrow 
analyzing bands, with one signal representing the magnitude of the 
short-time fourier transform, and the other signal representing the phase 
angle derivative of the short-time Fourier transform. The plurality of 
phase vocoder signals are averaged to develop composite amplitude and 
phase signals, and the composite control signals of the plurality of phase 
vocoders are utilized to synthesize a replica of the nonreverberant 
acoustic signal. Again, in this system only early echoes are corrected. 
In all of the techniques described above, the treatment of early echoes and 
late echoes is separate, with the bulk of the systems attempting to remove 
mostly the early echoes. What is needed, then, is a simple approach for 
removing both early and late echoes. 
Such a simple approach is disclosed in a copending application Ser. No. 
791,418 entitled "A Method and Apparatus for Removing Room Reverberation," 
filed by J. B. Allen. In accordance with the Allen disclosure, room 
reverberation characteristic of monaural systems is removed by employing 
two microphones at the sound source and by manipulating the signals of the 
two microphones to develop a single nonreverberant signal. Both early 
echoes and late echoes in the signal received by each microphone are 
removed by manipulating the signals of the two microphones in the 
frequency domain. Corresponding frequency samples of the two signals are 
co-phased and added and the magnitude of each resulting frequency sample 
is modified in accordance with the computed cross-correlation between the 
corresponding frequency samples. The modified frequency samples are 
combined and transformed to form the nonreverberant signal. 
This Allen approach is simplified somewhat, and the apparatus embodying 
some of the principles in the Allen approach is made less expensive by the 
improvements of this invention. 
SUMMARY OF THE INVENTION 
In accordance with the principles of this invention, room reverberation is 
removed by employing two microphones at the sound source and by 
manipulating the signals of the two microphones to develop a single 
nonreverberant signal. Both early echoes and late echoes are removed by 
performing a co-phase and add operation in separate frequency bands of the 
signal and by modifying each frequency band resulting from the co-phase 
and add operation in accordance with a computed phase difference average 
between the corresponding frequency bands. The modified frequency bands 
are combined to form the nonreverberant signal.

DETAILED DESCRIPTION 
In accordance with the Allen disclosure, room reverberation can be reduced 
by applying the equation 
EQU S(.omega.) = [Y(.omega.) + A(.omega.)X(.omega.)]G(.omega.) (1) 
to the signals of two microphones situated in a reverberant room, where 
X(.omega.) is the spectrum of the signal x(t) of a first microphone, 
Y(.omega.) is the spectrum of the signal y(t) of a second microphone, 
A(.omega.) is a frequency dependent phasor, and G(.omega.) is a frequency 
dependent gain control factor. This S(.omega.) signal, transformed into 
the time domain, is approximated in accordance with the principles of this 
invention by operating on a plurality of frequency bands rather than on 
the spectrum signals. Thus, an approximation to s(t), s'(t), which is the 
inverse function of S(.omega.) may be realized by 
##EQU1## 
In equation (2), x.sub.i (t) is the first microphone's signal filtered 
through bandpass filter i, y.sub.i (t) is the second microphone's signal 
filtered through bandpass filter i, a.sub.i is the delay imposed on the 
signal x.sub.i (t), and g.sub.i (t) is a control signal developed from the 
signals x.sub.i (t+a.sub.i) and y.sub.i (t) which relates to the phase 
difference variations between the component signals. The various filters 
(i = 1,2,. . . N) may or may not be of equal bandwidth but they do cover 
the full frequency bandwidth of the signals developed in the two 
microphones. 
One system embodiment realizing the approximation defined by equation (2) 
is shown in FIG. 1, where the signal of a first microphone is applied 
through terminal 10 to filter/delay element 20 and the signal of a second 
microphone is applied through terminal 11 to filter/delay element 30. 
Filter/delay element 20 provides a variable delay in response to control 
signals on bus 60, while filter/delay element 30 provides a fixed delay. 
The fixed delay developed by element 30 permits both positive and negative 
delay in the signal output of element 20 with respect to the signal output 
of element 30. 
Filter/delay element 20 separates the spectrum of the signal applied to 
terminal 10 into a plurality of bands covering the entire spectrum of the 
signal and applies a delay to the signal of each band. The bands may 
overlap and the delay in each band may be related or unrelated to the 
delay in other bands. Filter/delay element 30 performs the same function 
as filter delay element 20 and is of the same construction, except insofar 
as it employs no means for developing a variable delay. 
The plurality of bandpass signals developed by elements 20 and 30 are 
applied to signal combiner circuit 40 which develops an equal plurality of 
equalized bandpass signals. By "equalized" it is meant that corresponding 
bandpass signals (covering the same frequency band) developed in elements 
20 and 30, forming pairs of bandpass signals, are co-phased and added, and 
the magnitude of the co-phased and added signal is modified; all in 
accordance with equation (2). This is illustrated in FIG. 1 by the drawing 
of correlator element 40-1 which is one of N identical elements in signal 
combiner circuit 40, where N is the number of bandpass signals developed 
by filter/delay element 20. 
In accordance with one embodiment for correlator element 40-1, as shown in 
FIG. 1, the first pair of bandpass signals are applied to the first 
correlator circuit, wherein a measure of the correlation between the two 
applied signals is obtained by measuring the phase difference between the 
two signals. When two signals are delayed from one another but are 
otherwise well correlated, the correlation function (as a function of the 
classical variable .tau.) is maximum at some value of .tau. other than 
zero (which corresponds to the delay between the two signals). To 
determine whether two signals are correlated, it is not important to know 
the value of .tau. where the correlation function is maximum but rather, 
the variability of .tau.. 
The measure of phase difference between two signals behaves as the 
correlation function. When the signals are highly uncorrelated, the phase 
difference is a rapidly varying signal and when the signals are perfectly 
correlated, the phase difference is an unvarying signal. Thus, a measure 
of the variability in the phase difference signal provides a measure of 
the correlation between the two measured signals and, accordingly, a 
measure of the energy in some preselected frequency band of the phase 
difference signal indicates the measure of correlation. 
Therefore in accordance with the principles of this invention, a phase 
difference signal is developed in phase detector 41-1 and is passed 
through bandpass filter 42-1. The power in the signal at the output of 
bandpass filter 42-1 is determined with rectifier circuit 43-1 followed by 
a low-pass filter circuit 44-1 and that developed power serves as the 
measure of correlation between the applied signals. 
Filter 42-1 is a bandpass filter, e.g., 1 Hz to 100 Hz, rather than a 
low-pass filter because fixed or very slow changes in phase (e.g., 
movements in the speaker's head) provide only a measure of the value of 
.tau. where the correlation function is maximum but do not provide a 
measure of the variability of .mu..tau.. Thus, such slow phase changes (or 
fixed dc phase differences) are not really of interest. Precisely because 
of those reasons, the output signal of phase detector 41-1, passed through 
a low-pass filter 47-1 with a 1 Hz cutoff frequency, serves as one control 
signal of bus 60 which is fed back to and affects filter/delay element 20 
to control the delay applied to the signal of terminal 10. By operation of 
this control signal, the bandpass signals applied to correlator element 
40-1 are co-phased. The addition of the co-phased signals applied to 
correlator 40-1 is accomplished in adder circuit 45-1 which is responsive 
to the two applied bandpass signals. The co-phased and added output signal 
of adder 45-1 is modified to reduce the effects of reverberation by 
effectively multiplying the co-phased and added signal by the output 
signal of low-pass filter 44-1 which, as indicated above, provides a 
measure of the correlation in the signals applied to the correlator 
element. Thus, when correlation is high, the multiplier factor in the 
multiplication process is large and the output is large thereby 
accentuating nonreverberant signals, and when correlation is low, the 
multiplier factor is small and the output is small thereby attenuating 
reverberant signals. The multiplication process is achieved in element 
46-1 which is responsive to element 44-1 and 45-1. Element 46-1 may be an 
analog multiplier circuit, a variolosser, or the like. 
The output signal of element 46-1, which is also the output signal of 
correlator 40-1, is a bandpass signal occupying essentially the same band 
that its component signals occupy. The output signal of correlator 40-1 
comprises one of the elements in the summation equation of equation 2, and 
comprises one of the output signals of signal combiner 40. The other 
output signals of correlator elements 40-2, . . . 40-N, are also bandpass 
signals occupying essentially the same bands as their component signals 
and, together, the signals from the correlation elements form the desired 
nonreverberant signal. The addition of these signals is performed in 
summer circuit 50. Summer 50 may be a conventional summing circuit 
comprising, for example, a plurality of resistors each connected to a 
different output signal of signal combiner 40 and to an input terminal of 
an operational amplifier of preselected gain. 
One simple implementation for phase detector circuit 41-1 is shown in FIG. 
2. Therein, amplifier 411 (providing positive gain) is responsive to one 
of the signals applied to phase detector 41-1 and amplifier 412 (providing 
a negative gain) is responsive to the other signal applied to detector 
41-1. The signals applied to each amplifier are amplified to the point of 
amplifier saturation, causing the amplifier output signals to be square 
waves. The square wave signals of amplifiers 411 and 412 are 
differentiated in elements 413 and 414 and applied to the set and reset 
leads, respectively, of flip-flop 415. The circuit of FIG. 2 develops a 
pulse train output signal whose average is a sawtooth function of the 
phase difference between the applied signals. The sawtooth nature of the 
transfer function may be recognized when it is observed that two applied 
signals separated by a phase A from each other develop square waves which 
are similarly separated by phase A at the outputs of amplifiers 411 and 
412. The resulting output of flip-flop 415 is a pulse train with each 
pulse having a width that is a fraction of the pulse train period. The 
pulse width fraction is equal to the amount by which the phase angle A is 
removed from -180.degree.. The average signal of that pulse train provides 
a measure of the phase angle A and that average increases linearly from 
-180.degree. through 0.degree. and up to +180.degree., thus appearing as a 
sawtooth function. 
Filter/delay element 20 may be implemented in a number of ways. Most 
generally, the delay of each bandpass signal developed by element 20 may 
be totally unrelated to the delay of any of the other bandpass signals. To 
achieve such independence of delay, it is most advantageous to first 
separate the signal of terminal 10 into individual bandpass signals and to 
then delay each signal. Such capability is offered by the circuit of FIG. 
3 where bandpass filters 22-1, 22-2, . . . 22-N are connected to terminal 
10, developing a plurality of bandpass signals. The output signal of each 
bandpass filter 22-i is connected to a shift register 23-i (i=1,2 . . . N) 
and the output signals of the shift registers comprise the output signals 
of element 20. Shift registers are selected for implementation of the 
variable delay in the FIG. 3 circuit because delay is easily controlled in 
shift registers by controlling the length of the shift register and the 
frequency of the signal clocking the shift register. It is more 
convenient, generally, to dynamically control the frequency of the clock 
rather than the length of a shift register. Thus, FIG. 3 depicts voltage 
controlled oscillators (VCOs) 24-1, 24-2, . . . 24-N which are responsive 
to control signals applied to element 20 and which develop clock signals 
that are respectively applied to shift registers 23-1, 23-2, . . . 23-N. 
The control signals affecting the above-described VCOs are the control 
signals of bus 60 which emanate from signal combiner element 40. 
It may be noted at this point that all circuits in the FIG. 1 system are 
depicted as being analog but that, at first blush, shift registers 22-i 
require digital circuits. In fact, registers 22-i may be implemented with 
CCDs (Charge Coupled Devices) which can sustain pulsed signals of 
different analog amplitudes. Therefore, all that is required is a sampling 
of the analog signals at the input of the shift registers and a low-pass 
filter at the output of shift registers. 
In situations where the direct signal is much stronger than the reflected 
signals, there is a strong coherence between the signal of one microphone 
and the delayed signal of the other microphone. In such circumstances, a 
single delay element applied directly to the signal of terminal 10 may be 
employed. Such a filter/delay element 20 is shown in FIG. 4 where a shift 
register 25 is interposed between terminal 10 and the bank of bandpass 
filters 22-1, 22-2, . . . 22-N. Shift register 25 is also controlled with 
a VCO (27), but the control signal affecting the VCO 27 is a signal which 
is proportional to the average of the control signals present on bus 60. 
This average signal is developed in averaging circuit 26 which is 
interposed between bus 60 and VCO 27. 
It was indicated above that to obtain a good measure of the correlation 
between the signals applied to correlator circuit 40-1, the fixed and 
slowly varying phase differences must be removed and that, therefore, 
bandpass filter (42-1) is required. The need for such a bandpass filter 
and the accompanying rectifier and low-pass filter circuits (43-1 and 
44-1) is eliminated with the phase detector circuit illustrated in FIG. 5. 
In FIG. 5, the signals applied to correlator circuit 41-1 are applied to a 
sawtooth phase detector (such as shown in FIG. 2) 401 and to a triangular 
phase detector 402. Triangular phase detector 402, as shown in FIG. 5, is 
implemented by amplifying both input signals to the point of amplifier 
saturation (developing thereby square waves) and by applying the resulting 
square waves to an Exclusive OR gate. The output signal of the Exclusive 
OR gate is a pulse train whose average is maximum when the phase angle A 
between the signals is zero, is linearly reduced with increased (or 
decreased) phase angle, and reaches a minimum when the phase angle A is 
.+-.180.degree.. The required averages of the signals developed by 
detectors 401 and 402 are obtained with low-pass filters 403 and 404, 
respectively. The magnitude of low-pass filter 403 is developed in 
full-wave rectifier circuit 405 and the output signals of rectifier 
circuit 405 and of low-pass filter 404 are combined in adder circuit 406. 
Adder circuit 406 develops the output signal which is applied to elements 
46-1; while the output signal of sawtooth phase detector 401, as in the 
circuit of FIG. 2, is applied to low-pass filter 47-1 to develop the 
required control signal. A perusal of the output signal of the FIG. 5 
circuit reveals that the output signal is insensitive to slow variations 
in angle A and that in the presence of only fixed or slowly varying phase 
differences, the output of adder 406 is maximum. When reverberant signals 
are present, however, the average signals of both phase detectors (401 and 
402) is zero and, therefore, the output signal of adder 406 is also zero.