Digital echo canceller comprising a double-talk detector

A digital echo canceller has a receive path (2) and a send path (3) and comprises combining apparatus (10) for forming a send output signal [r(k)] as the difference between the signal [z(k)] applied to the send input (SI) and a replica signal [e(t)] used for cancelling an additive echo signal [e(k)] at the send input (SI) that has developed in response to a receive input signal [x(k)] applied to the receive input (RI), which echo canceller at least includes transforming apparatus (13, 15) for transforming the receive input signal [x(k)] and the send output signal [r(k)]; transforming apparatus (14) for transforming the replica signal [e(k)]; a digital adaptive filter (9) which has a number of filter coefficients for generating the replica signal [e(t)] in response to the receive input signal [x(k)] and the send output signal [r(k)], adaptation apparatus (12) for determining for each block m, adaptation components [A(p;m)] for each of the filter coefficients [W(p;m)] in response to the receive input signal [x(k)] and the send output signal [r(k)]; controllable gate means (17) for selectively passing the adaptation components [A(p;m)] to the adaptive digital filter (9); control apparatus (25-28, 21-24, 16) for determining respective levels of the send output signal [r(k)] and the receive input signal [x(k)] and for generating a control signal for the gate apparatus (17) in response to the levels thus determined, which control signals depend in a predetermined manner on the difference between the levels concerned.

The invention relates to a digital echo canceller which has a receive path 
between a receive input and a receive output and a send path between a 
send input and a send output, which echo canceller comprises combining 
means for forming a send output signal as the difference between the 
signal applied to the send input and a replica signal used for cancelling 
an additive echo signal at the send input that has developed in response 
to a receive input signal applied to the receive input, this echo 
canceller at least comprising: 
first transforming means for performing an N'-point Discrete Orthogonal 
Transform (DOT) of each block m of N' time-domain points of the receive 
input signal; 
second transforming means for performing an N'-point DOT of each block m of 
N' time-domain points of the send output signal; 
third transforming means for performing an N'-point Inverse Discrete 
Orthogonal Transform (IDOT) of each block m of N' frequency-domain points 
of the replica signal; 
a digital frequency-domain block-adaptive filter having a block length of 
N' components, in which for each signal block m a number of N' 
frequency-domain filter coefficients W(p;m) is available with p=0, 1, 2, . 
. . , N'-1 for generating the replica signal as an estimate of the echo 
signal in response to the receive input signal and the send output signal; 
adaptation means for determining for each block m adaptation components for 
each of the filter coefficients in response to the receive input signal 
and the send output signal; 
controllable gate means for selectively passing the adaptation components 
to the adaptive digital filter; 
control means for determining respective levels of the send output signal 
and a second signal and for generating a control signal for the gate means 
in response to the levels thus determined, which control signal depends in 
a predetermined manner on the difference between the levels concerned. 
An echo canceller of such a structure is known from European Patent 
Application EP-A-0 301 627. The echo canceller described in that Patent 
Application is especially arranged for preventing the disturbing influence 
of double-talk on the echo canceller adjustment. Double-talk occurs when a 
desired signal to be transmitted and an echo signal are simultaneously 
applied to the send input. The super-positioning of these signals then 
entails that the adjustment of the echo canceller for cancelling the echo 
signal can be deranged considerably by the desired signal to be 
transmitted that is also present. This means that the replica produced by 
the echo canceller no longer sufficiently cancels the current echo signal. 
In the above Patent Application a robust solution is given for the problem 
of a possible derangement of the echo canceller caused by double-talk. For 
this purpose a combination is used consisting of a time-domain digital 
filter comprising a programmable filter coefficient memory, which filter 
forms the proper echo cancelling signal, and a digital frequency-domain 
block-adaptive filter (FDAF). These two filters each generate a replica 
signal and as long as the replica signal generated by the frequency-domain 
block-adaptive filter is a better estimate of the echo signal than the 
replica generated by the programmable filter, the filter coefficients of 
the frequency-domain block-adaptive filter are transferred to the 
programmable filter. During double-talk the adjustment of the 
frequency-domain block-adaptive filter is disturbed and the transfer of 
the filter coefficients is interrupted by the gate means. In this way the 
adjustment of the frequency-domain block-adaptive filter does not disturb 
the operation of the programmable filter for proper echo cancellation 
during double-talk. 
In some cases it may be inconvenient that the structure of the prior-art 
echo canceller is rather complex and for this reason it is an object of 
the present invention to provide for these cases an echo canceller whose 
echo-cancelling properties are better than those of the prior-art echo 
canceller but which nevertheless has an essentially simpler structure. 
Therefore, a digital echo canceller according to the invention is 
characterized in that the second signal is the receive input signal, in 
that the control and gate means operate in the frequency domain and in 
that a separate control signal is determined for each of the N' 
frequency-domain points in dependence on the respective levels of the 
receive input signal and the send output signal for the frequency-domain 
point concerned. 
The measures according to the invention result in that the adjustment of 
the frequency-domain block-adaptive filter is blocked only in the sections 
of the echo canceller which represent a frequency band in which a 
disturbance occurs due to double-talk, whereas the adjustment for 
obtaining a minimum error signal still proceeds in the remaining sections. 
This is advantageous in that, when double-talk has terminated, only 
several filter coefficients will show a distinct deviation from the value 
desired for an optimal error correction at that moment, whereas the 
remaining filter coefficients of the sections in which the adjustment has 
not been blocked already have the desired value. Unlike the case with the 
prior-art echo canceller, it is in this case not necessary to provide 
means for storing the filter coefficients valid immediately prior to the 
occurrence of double-talk in order to use them as new filter coefficients 
after double-talk has ended. This leads to an essential simplification of 
the echo canceller, inter alia because no use is made of a time-domain 
programmable filter. 
It should be observed that the article entitled: "Kompensation akustischer 
Echos in Frequenzteilbander" by W. Kellermann, in Frequenz, Vol. 39 
(1985), No. 7/8, pp. 209-215, describes a transversal filter intended for 
echo cancellation in a telephone system, in which filter the signals to be 
processed are subdivided into a number of frequency bands which have each 
their own transversal adaptive filter. This article does not mention at 
all the problems occurring with double-talk. The principle forming the 
underlying thought of the present invention can also be advantageously 
used in an echo canceller of the type described in above article. 
Therefore, the invention also provides a digital echo canceller which has a 
receive path between a receive input and a receive output and a send path 
between a send input and a send output, which echo canceller comprises 
combining means for forming a send output signal as the difference between 
the signal applied to the send input and a replica signal used for 
cancelling an additive echo signal at the send input that has developed in 
response to a receive input signal applied to the receive input, this echo 
canceller at least comprising: 
first filter means for transforming the frequency band of the receive input 
signal into Q consecutive frequency bands; 
second filter means for transforming the frequency band of the send output 
signal into Q consecutive frequency bands; 
third filter means for assembling a single frequency band from the Q 
consecutive frequency bands of the replica signal; 
a digital transversal adaptive filter comprising Q filter sections having a 
number of filter coefficients for each frequency band, for generating for 
each frequency band a replica signal that is an estimate of the echo 
signal for the frequency band concerned in response to the receive input 
signal and the send output signal; 
adaptation means for determining adaptation components for the filter 
coefficients of each section on the basis of the receive input signal and 
the send output signal; 
controllable gate means for selectively passing adaptation components to 
the adaptive digital filter; 
control means for determining respective levels of the send output signal 
and a second signal and for generating in response to the levels thus 
determined a control signal for the gate means which control signal 
depends in a predetermined manner on the difference between the levels 
concerned, characterized in that the second signal is the receive input 
signal and in that the control means and gate means determine a separate 
control signal for each of the Q frequency bands in dependence on the 
respective levels of the receive input signal and the send output signal 
for the frequency band concerned. 
In a similar fashion, for each of the Q frequency bands, a time-domain 
comparison may be effected between the normalized power of the receive 
input signal and that of the send output signal, the adaptation of the 
filter coefficients in the adaptive filter section belonging to a specific 
frequency band being interrupted once it has appeared that the power of 
the send output signal increases relative to that of the receive input 
signal, which is an indication, as extensively discussed hereinbefore, 
that double-talk occurs in the frequency band concerned.

In the drawing Figures corresponding elements are denoted by the same 
reference numerals. 
FIG. 1 shows a simplified block diagram of an echo canceller used in a 
telephone set with loudspeaker reproduction of a received speech signal. 
Such an echo canceller 1 has a receive path 2 with a receive input RI and 
a receive output RO, as well as a send path 3 with a send input SI and a 
send output SO. A receive input signal x(t), to be referred to as far-end 
signal in the following, is applied to receive input RI and transferred 
over receive path 2 to receive output RO which is connected to a 
loudspeaker 5 by means of a receive amplifier 4. In the absence of a 
far-end signal, a microphone 6 generates a desired signal to be 
transmitted which is applied as a send input signal s(t), to be referred 
to as near-end signal in the following, to send input SI through a send 
amplifier 7. This near-end signal s(t) is transmitted over send path 3 to 
send output SO. Between loudspeaker 5 and microphone 6 there is an 
acoustic echo path, symbolically represented in FIG. 1 by an arrow 8. Over 
this acoustic echo path 8 a far-end signal x(t) at receive output RO, if 
present, may introduce an undesired additive echo signal e(t) at send 
input SI through the microphone 6, so that a sum signal z(t)=s(t)+e(t) is 
applied to send input SI. Echo canceller 1 has now for its task to cancel 
in the best possible way this undesired echo signal e(t). For this 
purpose, echo canceller 1 comprises a filter 9 which in response to 
far-end signal x(t) on receive path 2 generates a signal e(t) that is an 
estimate of the undesired echo signal e(t). By means of a combining 
circuit 10 this signal e(t) is subtracted from the sum signal 
z(t)=s(t)+e(t) at send input SI for forming a send output signal r(t) 
which can be written as: 
EQU r(t)=s(t)+[e(t)-e(t)] 
From this expression it appears that signal r(t) at send output SO 
represents the desired signal s(t) to be transmitted when replica signal 
e(t) is a reliable estimate of echo signal e(t), because in that case the 
second term of the right-hand side of this expression will practically be 
equal to zero. In general the transfer characteristic of the echo path 
between receive output RO and send input SI will vary with time, and 
especially acoustic echo path 8 may show large variations. Since echo 
signal e(t), in a good approximation, may be considered to be the linear 
convolution of far-end signal x(t) with the impulse response h(t) of the 
echo path between receive output RO and send input SI, the shape of 
time-varying impulse response h(t) will lead to corresponding variations 
of echo signal e(t) at send input SI. Filter 9 in echo canceller 1 is 
therefore arranged as an adaptive filter which has for its task to make 
its impulse response w(t) substantially equal to impulse response h(t) of 
echo path RO-SI. The adaptive adjustment of this filter 9 is controlled by 
signal r(t) at the output of combining circuit 10. This adaptive 
adjustment is continued as long as there is a correlation between control 
signal r(t) and far-end signal x(t). When only far-end signal x(t) is 
present (and thus, near-end signal s(t)=0), adaptive filter 9 will 
generate a replica signal e(t) which is a reliable estimate of echo signal 
e(t). However, when both far-end signal x(t) and near-end signal s(t) are 
present, a situation will arise which is commonly referred to as 
double-talk. If no appropriate measures are taken, the adjustment of 
adaptive filter 9 may be considerably deranged during double-talk due to 
the presence of near-end signal s(t) as a disturbing term in control 
signal r(t). This misadjustment of adaptive filter 9 leads to a replica 
signal e(t) which is no longer a reliable estimate of echo signal e(t) so 
that at send output SO a signal r(t) will occur which is disturbed to an 
annoying degree by an echo signal which has been cancelled insufficiently 
or even incorrectly. 
Since the present invention relates to a digital echo canceller, a 
discrete-time modelling will be utilized in the following description. The 
simplest manner in which such a modelling may be obtained is to assume 
that in the diagram of FIG. 1 the signals x(t) and z(t) are applied to 
receive input RI and send input SI via analog-to-digital converters (not 
shown), the signals x(t) and r(t) at receive output RO and send output SO 
are derived by means of digital-to-analog converters (not shown) and to 
assume that all further relevant signals in echo canceller 1 are digital 
signals. These digital signals are denoted in the conventional manner so 
that, for example, x(k) is a quantized sample of continuous-time signal 
x(t) at instant t=kT, where 1/T is the sampling frequency. 
If an echo canceller is used for cancelling strongly auto-correlated 
signals, such as speech, adaptive frequency-domain filters are 
advantageous in that the convergence properties may be improved 
considerably. By transforming to the frequency-domain, the gain factor for 
each of the substantially orthogonal frequency-domain components can then 
be normalized in a simple manner in accordance with the power of the 
frequency-domain component concerned. It is also possible in the 
frequency-domain to considerably reduce the complexity of long filters by 
performing efficient Fourier transforms. Consequently, adaptive 
frequency-domain filters are extremely attractive to use in acoustic echo 
cancellers, because in these cancellers it is necessary to copy impulse 
responses of great length. Therefore, the digital echo canceller with 
double-talk detection according to the invention will be assumed to be a 
digital echo canceller comprising an adaptive frequency-domain filter in 
the following. 
FIG. 2 shows in a diagrammatic manner the general structure of a 
frequency-domain block-adaptive filter 9. In FIGS. 2 and 3 double-line 
signal paths denote paths in the frequency-domain over which paths blocks 
of frequency-domain points are transmitted, and single-line signal paths 
denote paths in the time-domain. Transformations from the time-domain to 
the frequency-domain and vice versa take place by means of Discrete 
Orthogonal Transforms (DOT's) or their Inverses (IDOT's). An illustrative 
example of such transformation is the Discrete Fourier Transform (DFT) and 
its inverse (IDFT), which are widely utilized and, for example, 
extensively discussed in an article entitled "A Unified Approach to Time- 
and Frequency-Domain Realization of FIR Adaptive Digital Filters" by G. A. 
Clark et al., published in IEEE Transactions on Acoustics, Speech and 
Signal Processing, Vol. ASSP-31, No. 5, October 1983, pp. 1073-1083 and in 
an article entitled "Unconstrained Frequency-Domain Adaptive Filter" by D. 
Mansour et al., published in IEEE Transactions on Acoustics, Speech and 
Signal Processing, vol. ASSP-30, No. 5, October 1982, pp. 726-734. From 
practical considerations of computational complexity and permissible 
signal delays these DOT's have a finite block length N' and in the 
literature such transformations are known as N'-point DOT's, where "point" 
may refer to a discrete time-domain component and to a discrete 
frequency-domain component. With respect to the block length N' the 
following observation is made. FDAF 9 has to generate a replica signal 
e(k) which is a good estimate of echo signal e(k). Echo signal e(k), in 
its turn, may be considered to be a linear convolution of far-end signal 
x(k) with impulse response h(i) of echo path 8 with i=0, 1, 2, . . . , 
N-1. It need not be further clarified that FDAF 9 then also has to present 
an impulse response of a length N for generating replica signal e(k) as a 
linear convolution of far-end signal x(k) with the impulse response of 
FDAF 9. The operations necessary for this purpose are performed in FDAF 9 
on blocks of N' points in the frequency-domain, and it is well known that 
these operations correspond with a circular convolution in the 
time-domain, in which the period is equal to the block length N'. The 
desired linear convolution can then be obtained by suitably sectioning of 
the time-domain signals involved in the N' -point DOT's, while the most 
current segmenting procedures are the overlap-save method and the 
overlap-add method. The above implies that, generally, the block length N' 
of the DOT's is larger than the desired length N of the impulse response 
of FDAF 9. In the above article by Clark et al. it is stated that for the 
most efficient implementation of FDAF 9 with an impulse response of length 
N, DFT's having a block length of N'=2N are used and time-domain signals 
are sectioned into blocks of N'=2N points, where each block overlaps the 
previous block by N points. For large values of N, for example, N=1000 to 
N=2000 in the present case of acoustic echo paths 8, the computational 
complexity can yet be reduced considerably by utilizing efficient 
implementations of the DFT's known as "Fast Fourier Transform" (FFT), due 
to which the number of computational operations per N points of replica 
signal e(k) are of the order of N log N. Such computationally efficient 
implementations are also known for other types of DOT's than the DFT, but 
for simplicity it will always be assumed in the following that the 
N'-point DOT is an N'-point DFT with N'=2N. Furthermore, frequency-domain 
signals will be denoted by upper-case letters in order to distinguish in a 
simple manner between frequency-domain and time-domain signals, the 
time-domain signals being denoted by lower-case letters as has been done 
in the foregoing. Finally, the further description is aimed at using the 
overlap-save method as a procedure of sectioning the time-domain signals. 
The FDAF 9 shown in FIG. 2 comprises a filter section 11 and an adaptation 
processor 12. Filter section 11 and adaptation processor 12 operate in the 
frequency domain so that three domain transformations are to be performed, 
that is to say: 
by transforming means 13 and associated sectioning means: a 2N-point DOT 
for transforming each block of 2N time-domain points of far-end signal 
x(k) into a block of 2N frequency-domain points, which are denoted X(p;m) 
with p=0, 1, 2, . . . , 2N-1 for a block having block number m; 
by transforming means 14 and associated sectioning means: a 2N-point IDOT 
for transforming each block of 2N frequency-domain points E(p;m) into a 
block of N time-domain points of replica signal e(k); 
by transforming means 15 and associated sectioning means: a 2N-point DOT 
for transforming each block of N time-domain points of error signal r(k), 
after it has been augmented to a block of 2N Time-domain points, into a 
block of 2N frequency-domain points R(p;m). 
The details of the overlap-save method used for the sectioning procedure 
will further be explained with reference to FIG. 3. Filter section 11 of 
FDAF 9 comprises a memory 11(1) for storing the 2N frequency-domain filter 
coefficients W(p;m) of block m, a combining circuit 11(2) for adding 
together the output signal of memory 11(1) and the output signal of the 
adaptation processor 12, and a circuit 11(3) for multiplying each 
frequency-domain point X(p;m) by an associated frequency-domain filter 
coefficient W(p;m) in order to form products X(p;m)W(p;m) which represent 
the 2N frequency-domain points E(p;m). Adaptation processor 12 is arranged 
for producing block-by-block adaptation signals for the frequency-domain 
filter coefficients W(p;m) in response to the 2N frequency-domain points 
X(p;m) and R(p;m), which adapted filter coefficients W(p;m) are stored in 
memory 11(1). 
As described hereinbefore, the adjustment of the filter coefficients in the 
filter coefficient memory 11(1) may be considerably disturbed in the case 
of double-talk and, therefore, the supply of the adaptation signal from 
adaptation processor 12 to filter section 11 is always interrupted by gate 
means 17 once a situation of double-talk for a frequency-domain component 
has been detected by means of a comparator 16. According to the invention 
the double-talk detector comprising gate means 17 and the comparator 16 is 
also implemented in the frequency-domain so that each frequency-domain 
component has its own double-talk detector. Consequently, it may be 
provided that during double-talk the adaptation of the filter coefficients 
is interrupted only for the filter coeffients which belong to 
frequency-domain points at which double-talk occurs. 
Since speech has a discrete line spectrum for most voiced sounds, probably 
only a restricted number of frequency-domain points will be disturbed as a 
result of double-talk. Therefore, double-talk detection per 
frequency-domain point according to the invention is advantageous in that, 
when the situation of double-talk has terminated in course of time, only a 
restricted number of filter coefficients in the filter coefficient memory 
11(1) is not adaptively adjusted during the period of double-talk, so that 
for this restricted number of filter coefficients a relatively large 
adaptive adjustment may be necessary. The remaining filter coefficients 
will be adaptively adjusted in a normal fashion for the whole period of 
double-talk, or at least for part of the period, so that the disturbance 
of the replica signal due to double-talk is minimized. Another advantage 
is, as will be explained hereinbelow, that when double-talk is detected 
according to the invention, the power control already present in the FDAF 
may be used efficiently. 
In order to avoid that various filter coefficients are already deranged due 
to double-talk before the gate means 17 block a further derangement, it is 
desirable that the comparator 16 can react fast to a sudden large change 
in the send output signal which, as has appeared, is a certain indication 
of the occurrence of double-talk. 
FIG. 3 is a more detailed representation of the structure of the adaptation 
processor 12 and the circuit for generating the input signals for the 
comparator 16. FIG. 3 also shows explicitly that the N'-point DOT is an 
efficient implementation of an N'-point DFT, with N'=2N, known as 2N-point 
FFT, and in which the sectioning means are shown separated from the 
transforming means so as to properly describe their function. 
The structure of the adaptation processor is, in essence, equal to that 
described in aforementioned European Patent Application EP-A-0 301 627. 
In FIG. 3 a far-end signal x(k) is applied to sectioning means 13(1) to be 
subdivided into blocks of 2N points by means of a serial-to-parallel 
conversion, each block overlapping its predecessor by N points, as is 
shown symbolically in the Figure. The points of a block having block 
number m are denoted x(i;m) with i=1, 2, . . . , 2N-1. With the aid of 
transforming means 13 for performing a 2N-point FFT the 2N time-domain 
points x(i;m) are transformed into 2N points X(p;m) with p=1, 2, . . . , 
2N-1 in frequency-domain. In multiplier 11(3) each point X(p;m) is 
multiplied by an associated filter coefficient W(p;m) from memory 11(1) so 
as to form products X(p;m)W(p;m) which represent 2N points E(p;m). With 
the aid of transforming means 14 for performing a 2N-point IFFT these 2N 
points E(p;m) are transformed into 2N points e(i;m) in the time domain. 
Since the filter coefficients W(p;m) may be considered points of a 
2N-point DFT performed on time-domain filter coefficients w(i;m) which 
represent values of impulse response w(i) during block m, the 
multiplication in circuit 11(3) corresponds with a time-domain circular 
convolution of far-end signal vector x(m) during block m with impulse 
response w(i) during block m. The desired replica signal vector e(m), 
however, is the linear convolution of far-end signal x(k) with impulse 
response w(i). In accordance with the overlap-save method this desired 
replica signal e(k) is now obtained by applying the 2N points e(i;m) of 
this circular convolution for each block m to sectioning means 14(1) in 
which by means of a parallel-to-serial conversion the first N points 
e(i;m) with i=0, 1, 2, . . . , N-1 are discarded and the last N points 
e(i;m) with i=N, N+1, N+2, . . . , 2N-1 are transferred as replica signal 
e(k), as is shown symbolically in FIG. 3. 
For the block-by-block adaptation of the frequency-domain filter 
coefficients W(p;m), a known adaptation algorithm is utilized, for 
example, a complex Least Mean Square (complex LMS) algorithm. In 
accordance with the latter algorithm the filter coefficients W(p;m) are 
adapted as long as a correlation occurs between far-end signal x(k) and 
error signal r(k). Since adaptation processor 12 operates in the 
frequency-domain, according to the overlap-save method this error signal 
r(k) is applied to sectioning means 15(1) to be subdivided by means of 
serial-to-parallel conversion into blocks of 2N points, each block 
overlapping its predecessor by N points and the value of zero being forced 
onto the first N points r(i;m), with i=0, 1, 2, . . . , N-1, as is 
symbolically shown in FIG. 3. With the aid transforming means 15 for 
performing a 2N-point FFT these 2N points r(i;m) are transformed into the 
frequency-domain in 2N points R(p;m). 
Each of the 2N points R(p;m), is multiplied in a multiplier 20 by a factor 
of 2.mu.(p;m), which determines the gain factor in the adaptation 
algorithm, so that a product 2.mu.(p;m)R(p;m) is formed. The 2N points 
X(p;m) of each block m are applied to conjugating means 18 for forming the 
complex conjugate value X*(p;m) of each point X(p;m). In a multiplier 19 
each conjugate point X*(p;m) is multiplied by the output signal of 
multiplier 20 for the associated point R(p;m) so as to form the products: 
EQU A(p;m)=2.mu.(p;m)X*(p;m)R(p;m) 
which correspond with the time-domain circular correlation between far-end 
signal x(k) and error signal r(k) during block m; the product A(p;m) 
determines the modification of filter coefficient W(p;m). If the gate 
means 17 are not activated which, as will be explained in the following, 
is the case as long as double-talk does not occur for the frequency-domain 
component concerned, these modifications A(p;m) are applied to the 
(accumulator) circuit formed by memory 11(1) for storing the filter 
coefficients W(p;m) of block m and an adder 11(2) for forming the sum of 
each coefficient W(p;m) and its associated modification A(p;m), which sum 
is stored in memory 11(1) to provide the filter coefficients W(p;m+1) for 
the next block (m+1). The adaptation algorithm can thus be written as: 
EQU W(p;m+1)=W(p;m)+2.mu.(p;m)X*(p;m)R(p;m). 
The 2N filter coefficients W(p;m) in memory 11(1) are thus available for 
the multiplications in circuit 11(3). 
If the far-end signals x(k) are uncorrelated or only slightly correlated, 
gain factor .mu.(p;m) may have a same constant value .alpha. for each 
filter coefficient W(p;m), which value .alpha. is independent of the block 
number m (this constant .alpha. is known as the adaptation factor of the 
algorithm). For strongly (auto)correlated far-end signals x(k), such as 
speech, the convergence speed of the FDAF may be considerably increased in 
a simple manner by decorrelating the far-end signals x(k) which, as is 
known per se, may be effected by normalizing their power spectrum; cf., 
for example, page 36 of the article entitled "Echo Cancellation 
Algorithms" by C. W. K. Gritton and D. W. Lin, published in IEEE ASSP 
Magazine, April 1984, pp. 30-38. Since the frequency-domain components 
X(p;m) are already available in the FDAF, such a normalization can be 
simply realized with the aid of normalizing means 23 in which the 
adaptation factor .alpha. is divided by the power [X(p;m)].sup.2 of point 
X(p;m), which power is formed in squaring circuit 21 and is smoothed 
blockwise by means of a simple low-pass filter 22 whose bandwidth is 
selected so that the convergence behaviour of the whole system is in 
essence determined by the time constant of the adaptive filter. The output 
signal of the normalizing means 23 may then be used as a gain factor 
.mu.(p;m) for the multiplication by 2.mu.(p;m) in circuit 20. 
For detecting double-talk for each of the 2N frequency components p, a gain 
factor v(p;m), which is normalized in accordance with the power in the 
frequency component R(p;m), is determined in an identical manner. This is 
effected by normalizing means 27 in which the adaptation factor .alpha. is 
divided by the power [R(p;m)].sup.2 of point R(p;m) formed in squaring 
circuit 25 and smoothed blockwise by means of a simple low-pass filter 26. 
The output signal of the normalizing means 27 can then be used as the gain 
factor v(p;m) for the multiplication by the output signal [R(p;m)].sup.2 
of circuit 25 in multiplier 28. 
The output signal v(p;m).[R(p;m)].sup.2 of multiplier 28 is compared to the 
output signal .mu.(p;m).[X(p;m)].sup.2 of multiplier 24 in a comparator 
circuit 16. For, if the frequency component R(p;m) is disturbed in the 
case of double-talk, the power in R(p;m) will increase and the following 
will hold: 
EQU v(p;m).[R(p;m)].sup.2 &gt;.mu.(p;m).[X(p;m)].sup.2. 
If this condition is satisfied, circuit 16 will produce an output signal 
for activating the gate means 17, so that these means will interrupt the 
transfer of adaptation components A(p;m) from multiplier 19 to summing 
circuit 11(2). 
In the manner described above, the adjustment of the adaptation factors is 
interrupted only in a restricted number of the 2N adaptation factor 
control circuits, that is to say, only in the control circuits belonging 
to a frequency component for which double-talk is detected. 
On reading the aforementioned and the article by W. Kellermann, those 
skilled in the art will readily recognize in what manner the principle of 
the present invention may be implemented in an echo canceller of the type 
described in said article. 
It should further be observed that the above embodiment for transformation 
to the frequency-domain may advantageously be combined with the echo 
canceller known from European Patent Application EP-A-0 301 627 and 
wherein a combination of a FDAF and a TDAF is used. The echo canceller 
formed by such a combination then has the advantage that both a perfect 
echo cancellation in case of double-talk is provided and the delay 
inherent in the presence of a FDAF is avoided.