Spatial position determination system

A system is disclosed that determines a spatial position of a tracker device relative to an object sending a return signal to the tracker. Such a system advantageously maintains phase accuracy between a forward signal from the tracker device and the return signal generator from the object. The system can include, as part of a tracker device, a reference signal generator, a transmitter, a receiver, and a spatial position computer. The reference signal generator is responsive to and phase-stabilized by a broadcast signal, e.g., a signal received from a commercial AM broadcast transmitter. The transmitter and receiver are both coupled to and phase-stabilized by the tracker reference signal generator. Variations and methods with different advantageous features are also described.

BACKGROUND OF THE INVENTION

The problem of determining the spatial position of objects is an ancient one. Perhaps the simplest and oldest known solution is to pace off a distance to a visible object by walking toward it along a straight path. More accurate and recent techniques include triangulating the location of a hidden object based on estimated distances or azimuthal angles to the object.

Measurement of azimuthal angle to a given object tends to be less accurate than measurement of distance to that object. Extremely precise instruments have been developed for distance measurement. For example, an optical instrument disclosed in U.S. Pat. No. 5,430,537 to Liessner et al. purports to have accuracy around the 1-10 micron resolution of light wavelengths. This instrument is based on phase changes between a light beam sent to a passive reflector and another light beam returned from the reflector.

Less precise instruments for phase-based distance measurement can provide benefits in particular applications. For example, R. S. Trenam, “Automatic Animal Tracking on a Limited Budget,” in The Collection and Processing of Field Data (1967) (pp. 273-82), discloses tracking of sheep to 20-yard accuracy using RF phase measurements.

In any system relying on phase differences between forward and return signals, frequency stability of the signals is critical to maintaining accuracy of distance measurement. Slight frequency deviations in the forward and return signals can cause significant phase deviations, especially when the distance to be measured includes a large number of wavelengths. Such phase deviations interfere with those expected from changes in distance and can significantly degrade accuracy.

SUMMARY OF THE INVENTION

A spatial position determination system according to various aspects of the present invention determines a spatial position of a tracker device relative to an object sending a return signal to the tracker. Such a system advantageously maintains phase accuracy between a forward signal from the tracker device and the return signal from the object.

A system according to particularly advantageous aspects of the invention includes, as part of a tracker device, a reference signal generator, a transmitter, a receiver, and a spatial position computer. The reference signal generator is responsive to and phase-stabilized by a broadcasted signal, e.g., a signal received from a commercial AM broadcast transmitter. The transmitter and receiver are both coupled to and phase-stabilized by the tracker reference signal generator. The spatial position computer is coupled to the receiver and (1) the tracker reference signal generator or (2) the tracker transmitter, or (3) both. The spatial position computer is responsive to indicia of a phase relationship between an output signal from the tracker transmitter and an input signal to the tracker receiver. Based on that indicia, the spatial position computer determines the spatial position of the tracker relative to the input signal source.

A spatial position can be expressed in a number of ways. It can be expressed as a stationary position, i.e., a point in space. Alternatively, it can be expressed as a differential position, e.g., a velocity or an offset from a previous spatial position. In addition, a spatial position can be expressed as a physical measure of distance, or as a proportion of a wavelength of the input signal.

A spatial position determination system according to particular aspects of the invention advantageously includes a transponder coupled via field radiation to the tracker and triggered by it to produce a return signal. The transponder includes a transmitter and a receiver, which are coupled via field radiation to the tracker receiver and transmitter, respectively. A tracker's spatial position computer in such a system is responsive to indicia of a phase relationship between an output signal from the tracker transmitter and an input signal received from the transponder transmitter. Based on that indicia, the spatial position computer determines the spatial position of the tracker relative to the transponder.

A transponder in a system according to further aspects of the invention includes its own reference signal generator, which is responsive to and phase-stabilized by a broadcasted signal. The receiver and transmitter in such a transponder are coupled to and phase-stabilized by the transponder reference signal generator. In such a system, the tracker reference signal generator and the transponder reference signal generator can both be responsive to the same broadcasted signal.

Phase-stabilized and phase-stabilizing signal generators according to further aspects of the invention include a stabilizing DDS (direct digital synthesis) module having a phase accumulator that is clocked responsive to sync pulses, and an output DDS module. The output DDS module is coupled to the stabilizing DDS module and has a phase accumulator that is clocked by system clock pulses but forced to the accumulated phase of the first DDS module upon occurrence of a qualified sync pulse.

DESCRIPTION OF PREFERRED EXEMPLARY EMBODIMENTS

A spatial position determination system according to various aspects of the present invention provides numerous benefits, including permitting highly accurate phase-based determination of distance without the need to include a high-stability internal oscillator. An example of such a system100including a tracker device110and a transponder150attached to a hidden object (here, a lost dog155) may be better understood with reference toFIGS. 1-2.

In operation of exemplary system100, a person holding tracker110can determine the spatial position of tracker110relative to transponder150and, based on repeated updates to that position determination, locate dog155. By maintaining phase stability responsive to a broadcasted signal from broadcast transmitter105, both tracker110and transponder150cooperate to permit highly accurate (e.g., about 4% of a wavelength) distance determination while omitting the expense and bulk of high-stability oscillators.

FIG. 2schematically depicts functional modules that tracker110and transponder150implement. These functional modules can be suitably implemented by hardware, software, or both. Functional modules can interact via any suitable routes of interconnection, including hardware (e.g., a bus, dedicated signal lines, etc.), access to shared storage media (e.g., arguments and returned values of function calls in RAM media, dual-access RAM, files residing on hard disk media, etc.), and combinations of hardware and shared media access.

Tracker110implements functional modules including: a reference signal generator112; a transmitter114and a receiver116, both coupled to generator112; a spatial position computer118coupled to receiver116and generator112; and an I/O module120coupled to computer118and to a suitable user interface not shown in FIG.2. Tracker110can also include a GPS (Global Positioning System) module122, which can advantageously cooperate with spatial position computer118as discussed below.

Transponder150implements functional modules including: a reference signal generator152; a receiver154; and a transmitter156. Receiver154and transmitter156are both coupled to generator152. They are also coupled to each other such that output of receiver154controls transmitter156.

Tracker110and transponder150include some of the same types of functional modules. Both devices include reference signal generators, transmitters, and receivers. These functional modules can be implemented by similar or identical hardware in both devices, with different software for causing them to operate appropriately for tracker110or transponder150. For example, transmitter114and receiver116in tracker110do not couple to each other. Thus, software in tracker110need not cause transmitter114to control receiver116.

A reference signal generator according to various aspects of the invention includes any hardware or software, or combination of both, that is phase-stabilized by a broadcasted signal. With its consequent phase stability, such a generator can phase-stabilize other functional modules. A functional module or device is phase-stabilized by a broadcasted signal when the phase stability of its internal operations and output signal(s) is not substantially worse than if controlled by an internal clock having phase stability as good as that of the broadcasted signal. In other words, the output of a device phase-stabilized by a broadcasted signal has phase stability nearly as good as that of the broadcasted signal itself. If a broadcasted signal's phase stability were such that it has phase noise of only −100 dBc at +/−100 Hz, for example, an output signal from a device phase-stabilized by the signal would not be expected to have phase noise of −50 dBc at +/−100 Hz, even if that were the phase stability of the device's internal oscillator. As another example, if the broadcasted signal's phase stability were such that the signal's frequency always remained within 0.01 ppm of average, the phase-stabilized device would not be expected to produce an output signal deviating 10 ppm from average.

Reference signal generators112and152of devices110and150are both phase-stabilized by a broadcasted signal from broadcast transmitter105. In the example ofFIGS. 1 and 2, transmitter105is a commercial AM broadcast transmitter operating at a frequency between about 500 kHz and about 1600 kHz. Such transmitters have operating ranges of at least several miles, and their signal quality tends to degrade gradually rather than abruptly. Consequently, dog155ofFIG. 1(with transponder150) and tracker device110(typically carried by the dog's owner) are both likely to be within the coverage zone of transmitter105.

In a particularly advantageous configuration according to various aspects of the invention, a reference signal generator phase-stabilizes other functional modules to a degree of stability greater than the generator's system clock granularity would conventionally permit. For example, conventionally performing a phase adjustment to a direct digital synthesis (DDS) module by skipping or effectively doubling a cycle of a 24 MHz system clock (one of many possible frequencies) induces a phase granularity of {fraction (1/24)} MHz, or 42 nanoseconds. This equates to 203 degrees of a 13.56 MHz transmit/receive frequency cycle (again, one of many possibilities), which is clearly unacceptable for a phase-based distance measurement system.

Advantageously, a reference signal generator according to various aspects of the invention can achieve its high level of performance without the need for analog phase locking. As discussed below with respect to a specific embodiment and with reference toFIGS. 4-6and10-15, this phase stabilization is performed using a combination of sync pulses and qualified sync pulses.

A transmitter according to various aspects of the invention includes any hardware, software, or combination of both capable of transmitting an output signal as field radiation, via a suitable coupling device. Any suitable type of radiation in any suitable field can be employed, including sound waves below, within, or above the range of human hearing in air or water, and electromagnetic radiation in the RF, infrared, or visible light spectrum. Suitable coupling devices include antennas (e.g., loops, whips, directional arrays, etc.) for coupling to an RF field and piezoelectric transducers for coupling to an acoustic field.

The many different types of transmitters suitable for various embodiments of the invention operate with widely varying output levels and signal frequencies. For example, an acoustic transmitter for use in the ocean (e.g., to track movements of marine mammals) may have a relatively high output level and a particular frequency selected for accuracy, range, avoiding interference with other tracking systems, and minimal adverse impact to the mammal. An optical transmitter aimed from the earth to its moon may operate at a very high output power to overcome the significant path loss between the two distant bodies.

Government regulations may severely restrict the output levels and signal frequencies that RF transmitters can employ in determining positions of terrestrial objects. In a preferred implementation of exemplary system100(FIG.2), transmitters114and156advantageously operate with an output level of less than one microwatt, at a frequency of 13.56 MHz, with the only modulation being periodic four-second bursts of a carrier wave. Harmonics are attenuated by 30 dB. This operating arrangement complies with regulations promulgated by the Federal Communications Commission, Part 15 (Section 15.209) for unlicensed transmitter operation. T. Warnagiris, “Legal Unlicensed Transmitting,” in Applied Microwave & Wireless, Spring 1996, pp. 32-54, which is incorporated herein by reference, provides guidance for implementation of other unlicensed embodiments.

In the preferred implementation of system100, calculated field strength at 30 meters from transmitters114and156is 140 microvolts per meter. Even at this very low power level, system100can achieve high accuracy, for example detecting three-foot (4% at 13.56 MHz) distance changes at 500 feet of separation and distance changes of about 1.5 foot (2%) at 50 feet of separation. Longer integration times than the four seconds employed in exemplary system100can yield even better accuracy or range at this low transmit power level.

A variation employing licensed transmitters can operate with significantly higher power levels for greater range and reliability. In such a variation or others, the transmit and receive frequencies of tracker110can be randomly assigned within a narrow band to reduce the probability of interference with other systems operating in the vicinity. In a variation employing direct sequence spread spectrum transmission (DSSS), pseudo random sequence codes can be randomly assigned to likewise avoid interference. Depending on local regulations, it may be possible to employ higher transmit power levels in an unlicensed DSSS variation because the spread spectrum transmission interference in a given narrow frequency range is lower than narrowband transmission interference within such a range.

A receiver according to various aspects of the invention includes any hardware, software, or combination of both capable of receiving an input signal coupled to it via field radiation and a coupling device of any suitable type, e.g., whip antenna112of FIG.1. As with a transmitter, any suitable type of radiation in any suitable field can be employed. Preferably, a transmitter and receiver employ the same coupling device. The transmitter and receiver can be suitably isolated from each other by conventional hardware such as a high-Q resonant device (in variations where the transmitter and receiver operate at different frequencies), unidirectional (e.g., ferromagnetic) circuitry, or a single-pole double-throw switch.

By coupling tracker110and transponder150together via RF electromagnetic field radiation, system100permits determination of a spatial position of tracker110relative to transponder150. This spatial position can be expressed as a physical measure of distance, depicted with arrow “d” in FIG.2. In advantageous variations, the spatial position can be expressed as an offset from a previous spatial position or as an azimuthal angle from tracker ˜110to transponder150. For example, a particular spatial position may be expressed as an azimuthal angle of 90 degrees, in which case transponder150is directly east of tracker110.

Transmitter114in tracker110transmits a forward signal15to receiver154in transponder150. Transmitter156in transponder150replies with a return signal51, which is received by receiver116in tracker110. Both reference signal generator152and receiver154couple to and control the output phase of transmitter156. In exemplary transponder150, receiver154sets the output phase of transmitter156to the phase it receives of forward signal15. (Of course, all such phases are relative to respective phase offsets induced by intervening signal processing.) Thus, changes in phase of return signal51are substantially determined by changes in distance “d,” which successive measurements can detect. A given change in distance “d” results in a phase change in return signal51(when received at receiver116) that is proportional to twice the change in distance.

Receiver154and transmitter156are both phase-stabilized by reference signal generator152. Thus, additional phase instability of return signal51over that of forward signal15substantially corresponds to phase instability of the broadcasted signal from transmitter105.

An exemplary device300that implements a reference signal generator400and a transmitter500and receiver600, phase-stabilized by generator400to a broadcasted signal332according to various aspects of the invention, may be better understood with reference to FIG.3. Operation of device300and its generator400, transmitter500, receiver600, and other components are discussed in the context of an advantageous embodiment that illustrates benefits of various aspects of the invention when such aspects are employed. However, certain aspects can provide benefits even when various other aspects are omitted. In addition, device300need not be employed as tracker110or transponder150of system100(FIGS.1-2), though such is presently preferred. Thus, neither this nor any other example provided herein should be considered as limiting the scope of the invention in any way; that limiting function is reserved exclusively for the issued claims.

Device300includes various analog and digital components mounted on a printed circuit board. These components are conventionally arranged and are omitted fromFIG. 3for clarity. Digital components include: a microcontroller (e.g., a P1C16C73); an FPGA (e.g., a XILINX XCS30XL); and miscellaneous support circuits (e.g., a 27C256 EPROM coupled to a 4040 counter to provide clock signals, etc.). Analog components include: a crystal (e.g., 24 MHz) coupled to the microcontroller to provide a main system clock; voltage regulators (e.g., separate TPS76950 5-volt regulators for digital and analog components, a 78L033 3.3-volt regulator); decoupling capacitors; and RF circuitry for implementing generator400, transmitter500, and receiver600.

For implementation of reference signal generator400, RF circuitry of exemplary device300suitably includes: an MMBF4416 FET and TK1235 RF transformer, coupled together (gate to transformer via coupling capacitor) along with associated RLC (resistor, inductor, capacitor) components for amplification of broadcasted signal332; an NE602 mixer and associated RLC components; a 455 kHz ceramic filter; an LM7131 op-amp with associated RLC components to serve as an IF amplifier; and an LM311 op amp/comparator with associated components, including a resistor-capacitor “L-network” coupled to the LM311's inverting input for comparator hysteresis.

For implementation of receiver500, RF circuitry of device300suitably includes circuitry similar to that implementing generator400. The TK1235 RF transformer is preferably replaced with a TK1237 version, and RLC component values suitably adjusted, to account for the different frequency of operation employed in receiver500.

For implementation of transmitter600, RF circuitry of device300suitably includes: an MMBF4416 FET and TK1235 RF transformer, coupled together with associated RLC (resistor, inductor, capacitor) components for amplification of an output signal to be transmitted; and a pair of ZC2811E diodes separated by an LM7131 op-amp stage (inverting, with DC bias of 2.5 volts) and selectably biased for on/off control of the transmitter output signal.

Device300(FIG. 3) further includes a control module310coupled to reference signal generator400, receiver600, and transmitter500via a number of data lines, represented as a group inFIG. 3by a bus320. Some of these data lines are illustrated inFIGS. 4-6. These include: a main system clock line410; a transmit control line510; a phase integrator output line610; and a phase control line520, which may be omitted if device300is used as a tracker in system100. Data lines in bus320need not be physically grouped together or have any particular physical form. In a variation where large portions of generator400, receiver600, and transmitter500are implemented by software in a DSP, particular “data lines” may be implemented by an intangible passing of arguments from one software function to another.

Exemplary reference signal generator400produces two output signals of sync pulses (unqualified and qualified types), which are represented as a group inFIGS. 3-6by a bus340. Generator400phase-stabilizes the signals in bus340to broadcasted signal332, which may be received from any suitable broadcast source like transmitter105ofFIGS. 1-2. Generator400receives signal332via a broadcasted signal line330(FIGS.3-4).

The internal operation of exemplary reference signal generator400may be better understood with reference to FIG.4. Generator400includes an analog signal processing subsystem having an RF amplifier420, a mixer422, a bandpass filter424, and a comparator426. Signal line330couples broadcasted signal332to RF amplifier420, which couples the signal to mixer422with frequency selectivity (e.g., for image rejection) and suitable amplitude. Mixer422frequency translates the selectively amplified signal to an IF (intermediate frequency), which is 455 kHz in exemplary generator400. Bandpass filter424rejects spurious output signals from mixer422and defines selectivity of reference signal generator400to one, and only one, broadcasted signal. (In variations discussed below, a reference signal generator is made responsive to multiple broadcasted signals.) Comparator426acts as a 1-bit A/D converter, providing a logic high signal on line427when the output signal from filter424exceeds a predetermined threshold and providing a logic low signal on line427otherwise.

Reference signal generator400further includes a digital signal processing (DSP) subsystem405. This subsystem is preferably implemented within a single FPGA with DSP subsystem505of transmitter500and DSP subsystem605of receiver600.

Signal line427enters DSP subsystem405and couples to a second mixer430implemented in subsystem405. Mixer430is controlled by the same local oscillator signal as mixer422. Consequently, mixer430frequency translates the filtered, 1-bit signal on line427to the same frequency as broadcasted signal332on line330. The local oscillator signal controlling both mixers comes from local oscillator (LO) generator432.

Generator432is a direct digital synthesis (DDS) module, commonly called a frequency synthesizer. In accordance with various aspects of the invention, such a module includes any functional module implemented by any suitable hardware, software, or combination of both that produces a periodic output signal by adding a predetermined increment to a phase accumulator. A DDS module typically achieves a periodic output by modulo-adding the increment with a predetermined modulus. Generator432produces a 1-bit output, which drives digital mixer430and, through an output port of the FPGA implementing DSP subsystem405, analog mixer422.

Generator432is not phase-stabilized to any broadcasted signal, and consequently its phase (and frequency) varies with variations in the frequency of the clock (not shown) controlling DSP subsystem405. Because mixers422and430perform complementary frequency translations, however, the output of first mixer430is phase-synchronous with the input of second mixer422.

The output of mixer430, a 1-bit facsimile of the input to mixer422, couples to a bandpass filter434for translation into a sinusoid that that very closely approximates a carrier of the selected broadcasted signal332on line330. In exemplary reference signal generator400, filter434is an IIR bandpass filter having a single pole pair (i.e., second-order) operating with 24-bit precision in its coefficients and output signal.

Generator432facilitates selection of a broadcasted signal for phase stabilization from a plurality of available broadcasted signals by making filter434digitally tunable. (Other techniques, many of them less convenient than that of generator432, can be employed, e.g., local oscillator adjustment.) When device300is to operate in the San Francisco Bay area, for example, coefficients of filter432can be predetermined such that filter432has a center frequency at 810 kHz, the operating frequency of radio station KGO. The coefficients are selected such that filter434has exceptionally high Q, e.g., a bandwidth of about 60-80 Hz at the exemplary center frequency of 810 kHz. Coefficients can be generated “on the fly” using conventional filter design equations. Alternatively, a lookup table can be provided containing coefficients for all available broadcasted signals, e.g., all of the approximately 100 AM broadcast channels available in the U.S.

Reference signal generator400requires no analog phase locking for its operation. Surprisingly, generator400can still phase stabilize transmitter500and receiver600to a degree of stability greater than that which the 24 MHz clock of system300would conventionally permit. This advantageous phase stabilization is performed using a combination of sync pulses (on signal line450) and qualified sync pulses (on line460).

In exemplary generator400, negative-to-positive transition detector436derives sync pulses from zero crossing transitions of a highly filtered broadcasted signal (output from filter432). With any clocked digital system, such transitions can only be determined to within a particular window of time uncertainty, which is proportional to the finite period of the generator's system clock. For example, an observed zero crossing transition may occur just before it is observed (upon transition of one system clock cycle), or it may occur nearly one full system clock cycle earlier, just after the previous system clock cycle has halted observation.

Qualified sync pulses are selected sync pulses that occur during conditions meeting one or more predetermined criteria. Magnitude detector438and “AND” gate440cooperatively produce qualified sync pulses when detector436has produced a corresponding sync pulse within a predetermined time before or after an actual zero crossing. In other words, qualified sync pulses are produced when the time uncertainty induced by clock granularity (e.g., {fraction (1/24)} MHz clock period) randomly permits a close match between the (unqualified) sync pulse and the actual event triggering it. This close match will typically occur when the trigger event occurs very shortly before or after transition of the generator's system clock.

Generator400employs zero crossing transitions as a trigger event. This configuration is advantageous in that it allows signal magnitude around the zero crossing to be employed as a qualifying criterion for qualified sync pulses. Magnitude detector438determines whether the zero crossing transition was observed close enough to the actual zero crossing that the signal level was less than or equal to a predetermined threshold (e.g., {fraction (1/64)} its maximum value) at the observation time. If it was, the sync pulse (on line450) resulting from the observation is considered close enough to being an accurate observation of its trigger event to be accompanied by a qualified sync pulse, on line460.

In variations, any predictable point along the cycle of a periodic signal (to a resolution limited by system clock granularity) can be employed as a trigger for a sync pulse and an associated qualifying criterion for a qualified sync pulse. For example, the point at which a signal reaches a predetermined threshold with a differential of a given sign (plus or minus) can be employed as a trigger. The differential between the signal level and the threshold can be employed as a qualifying criterion.

Sync pulses on line450and qualified sync pulses on line460are represented together inFIGS. 3-6by bus340. Again, this is only an illustrative grouping; no particular physical bus structure is required. As discussed below, the sync pulse on line450and qualified sync pulse on line460, in accordance with various aspects of the invention, form a powerful team of signals that facilitate completely digital stabilization of phase well beyond the resolution of the system clock period.

Reference signal generator400couples sync pulses and qualified sync pulses to various DDS modules in transmitter500and receiver600for phase stabilization, via bus340. In accordance with various aspects of the invention, these and other types of devices can be phase-stabilized by including (1) a first DDS module whose phase accumulator is clocked by sync pulses, and a (2) second DDS module whose phase accumulator is clocked by system clock pulses but forced to the accumulated phase of the first DDS module when a qualified sync pulse occurs. The occurrence of a qualified sync pulse indicates that the first DDS module was last clocked by a sync pulse that was produced suitably close to a predetermined trigger event. That event is a predictable, consistent point along a periodic cycle of the broadcasted signal being employed for phase stabilization.

When the serendipitously accurate clocking occurs, the output of the first DDS module accurately represents the phase that the second DDS module should have at that instant to achieve a desired phase stability and/or deviation. (As discussed below with reference toFIG. 6, a predetermined phase deviation can actually be induced in a DDS module to impart overall phase stability.) As discussed below, a less accurate system clock, i.e., a clock operating further from its expected frequency, causes the accumulated phase of the second DDS module to be further deviated from the accumulated phase of the first DDS module. In such cases, the second DDS module's accumulated phase undergoes more significant correction when qualified sync pulses occur.

An example of phase stabilization according to various aspects of the invention may be better understood with reference to the simulated signal plots ofFIGS. 10-15. A computer program listing below provides code that was executed with the Octave numerical language environment (similar to MATLAB) to produce these plots.

FIGS. 10 and 11illustrate multiple signals produced during the simulation with (1) a fairly accurate system clock frequency, and (2) a less accurate clock frequency. The signals depicted are: a system clock having segments1010(FIG. 10) and1110(FIG.11); a broadcasted signal with segments1020,1120on which phase stabilization is based; sequences1030,1130of regular (unqualified) sync pulses; sequences1040,1140of qualified sync pulses; a sinusoid-transformed rendition of a phase stabilization signal having segments1050,1150; an unstabilized output signal with segments1060,1160(illustrated for comparison); and a stabilized output signal having segments1070,1170.

In the simulation, the nominal clock frequency is modeled at 24 MHz. The frequency of the broadcasted signal (segments1020and1120ofFIGS. 10 and 11) is modeled at 5 MHz, and the desired frequency of the output signal is modeled at 3.5 MHz. The 24 MHz clock frequency is sometimes employed in FPGA devices currently available, and 5 MHz is one of the frequencies of NISI broadcast transmitter WWV. However, these are only exemplary signals and frequency values, which do not in any way limit the possible signals employed during implementation of the invention.

The simulation generated 4096 data points for each signal, with each point representing approximately {fraction (1/20)} of a system clock cycle at its expected frequency of 24 MHz. The points are too close together to be individually identifiable in the plots ofFIGS. 10-15, but they provide a convenient reference frame on the X-axis (i.e., the time axis) of the plots ofFIGS. 10-13.

FIG. 10depicts the simulated signals in the interval from points0001through1024. During this interval, five qualified sync pulses occur, in sequence1040. The simulation generated these qualified sync pulses when it had generated a corresponding sync pulse in sequence1030sufficiently close to a zero crossing of the r1broadcasted signal in segment1020. The simulation qualified this closeness to the zero crossing using the criterion that the broadcasted signal have an amplitude less than 15% of its maximum. (For clarity of illustration, this simulation criterion was set much higher than the {fraction (1/64)}=1.5% criterion of reference signal generator400ofFIG. 4.) This portion of the simulation, and the method it exemplifies, may be better understood with reference to lines52-65of the program listing below.

During its segment1010within this interval, the system clock was fairly close to its expected frequency. See lines10,26, and30-31of the program listing for a better understanding of how the simulation modeled a time-varying deviation in system clock frequency.

In their respective segments1060and1070within the 0001-1024 point interval ofFIG. 10, the unstabilized output signal and the stabilized output signal look very similar. This similarity exists because the accumulated phase of the stabilized DDS module (simulated at lines100-115of the program listing) did not undergo a very dramatic correction when qualified sync pulses of sequence1040occurred, e.g., at time T1.

At time T1, system clock transition1012caused the simulated system to observe a zero crossing transition1022and generate a sync pulse1032shortly after the transition actually occurred. (To keep the plots compact, both positive and negative clock transitions were recognized.) The observation was accurate enough that the broadcasted signal in segment1020was unable to reach 15% of its maximum amplitude by the time the observation was made. Consequently, the simulated system generated a qualified sync pulse1042in sequence1040.

Qualified sync pulse1042forced the phase accumulator of the stabilized DDS module to the accumulated phase of a stabilizing DDS module, which was simulated at lines76-85of the program listing. Portion1052of the DDS module's phase stabilization signal is at the accumulated phase (transformed to a sinusoid inFIG. 10) to which qualified sync pulse1042forced the stabilized DDS module. This forcing caused the stabilized output signal in segment1070to reach a very slightly different amplitude at portion1072, upon transition of the system clock, from what it would have without forcing. Because the system clock in segment1010is close to its expected frequency, the forcing is not visually apparent in segment1070of the stabilized output signal.

In the interval from about points3084through4096, depicted inFIG. 11, the effect of phase stabilization according to various aspects of the invention is much more apparent. During its segment1110within this interval, the system clock was significantly deviated from its expected frequency. A visual comparison shows that the clock frequency in segment1110was significantly higher than in segment1010.

The broadcasted signal had the same frequency in both segment1120and segment1130, which is consistent with the broadcasted signal being from a source that is phase stable. Although the invention does not require a broadcasted signal to have any minimum phase stability, it does not make much sense in typical implementations to phase stabilize an output to a highly unstable signal.

At time T2, as at time T1ofFIG. 10, the occurrence of a qualified sync pulse1142forces the output DDS module's phase accumulator to the value of the phase stabilization signal in sinusoid-transformed segment1150, at portion1152. Here, this forcing causes the stabilized output signal in segment1170to make a dramatic transition at portion1172. This transition results in a significant phase change, visibly stretching the negative half-cycle of the stabilized output signal.

FIGS. 12 and 13are time-domain plots of the unstabilized and stabilized output signals, respectively, after they passed through a filter having a narrow passband at the expected output frequency. (See lines135-155of the program listing.) These plots, which span the entire4096simulation points, illustrate how the frequency of the unstabilized output signal varied with system clock frequency and how the frequency of the stabilized signal resisted such variation.

At the beginning of the simulation, points500-1000(all references to simulation data points are approximate), the filtered outputs gradually rose to maximum amplitude, a phenomenon resulting from the filter impulse response rather than the output signal themselves. Between points500-1000, the system clock remained within about 4% of its expected frequency (program listing, line26), and both signals remained substantially within the narrow passband of the simulation's bandpass filter.

Between points1000-2000, the system clock rose from about 4% greater than its expected value to a positive deviation of about 9%. In this interval, the difference in frequency between the unstabilized and stabilized output signals is visually noticeable inFIGS. 12 and 13. The unstabilized output signal (FIG. 12) steadily decreased in amplitude as its frequency drifted outside the filter passband. The stabilized signal (FIG. 13) remained within a 3 dB amplitude range as its primary frequency component remained substantially within the filter passband.

As the dock frequency continued to increase beyond point2000, toward its maximum positive deviation of 25%, the unstabilized signal ofFIG. 12continued to drift further away from the simulation filter passband. The stabilized signal ofFIG. 13increased and decreased in amplitude, though a 3.5 MHz frequency component clearly remained within the passband of the simulation filter at various times. The simulation exemplifies that, even with the 10-25% deviation from an expected clock frequency, with the dramatic phase corrections illustrated inFIG. 11, phase stabilization according to various aspects of the invention can still operate under certain circumstances. In typical implementations, however, clock frequency deviations are likely to be measured in the parts per million, and maintaining consistent performance with such dramatic clock frequency deviations is then unnecessary.

Perhaps the clearest depiction of performance in the simulation of phase stabilization according to various aspects of the invention is found in the X-Y plotsFIGS. 14 and 15.FIG. 14depicts phase differences between two intervals of the unstabilized filter output signal, with the amplitude of points500-900plotted on the X-axis and the amplitude of points1500-1900plotted on the Y-axis. Phase increases of the signal in one region clearly outpaced those of the signal in the other region, and a frequency difference between the signal in the two regions is thus clearly apparent.

FIG. 15depicts the same types of phase differences in the same regions (points500-900vs. points1500-1900), but for the stabilized filter output signal. The clean ellipse ofFIG. 15is a clear illustration of the advantageous result of phase stabilization performed according to various aspects of the invention. With this simulated phase stabilization, the stabilized output signal maintained a relatively constant phase (and frequency) relationship even with clock frequencies varying between about 2% (simulation point500) and about 8% positive deviation.

Exemplary device300ofFIG. 3includes a transmitter500and receiver600that are phase-stabilized in accordance with the various aspects of the invention discussed above. Functional modules of transmitter500are implemented mostly in a DSP subsystem505. As illustrated inFIG. 5, exemplary transmitter500includes just one analog signal processing component: a selectable output amplifier590. Control module310can enable or disable operation of amplifier590via transmit control line510, which is part of bus320of FIG.3.

DSP subsystem505includes a transmit DDS module530, which is phase-stabilized to broadcasted signal332by a stabilizing DDS module532. DDS modules530and532act cooperatively under control of sync pulses and qualified sync pulses from bus340to produce a transmit signal595that is phase-stabilized to broadcasted signal332. Transmit DDS module530includes a phase accumulator (none are shown) that produces transmit signal595by having its value increased by a predetermined increment with each system clock cycle. The increment is set to the transmit frequency divided by the nominal system clock frequency.

Stabilizing DDS module532includes a phase accumulator that produces a phase stabilization signal by having its value increased by a predetermined increment with each sync pulse on line450. The increment for stabilizing DDS module532is set to the transmit frequency divided by the broadcasted signal frequency. The increment is computed with a suitable modulus for phase increments beyond 360 degrees per sync pulse, i.e., when several transmit signal cycles are expected between each cycle of the broadcasted signal.

Qualified sync pulses appearing on line460force the phase accumulator of transmit DDS module530to the value of the phase stabilization signal from module532. A similar process in another example is described above with reference to simulated signal plots ofFIGS. 10-15. The phase-stabilized transmit signal is suitably amplified by amplifier590(when it is enabled by transmit control line510) and the signal595is transmitted via a suitable coupling device, e.g., a whip or loop antenna, a piezoelectric transducer, etc.

Exemplary receiver600ofFIG. 6receives a signal at line602via a suitable coupling device, preferably the same device from which the transmitter500ofFIG. 5transmits signal595. Receiver600includes an analog signal processing subsystem having a bandpass filter618, an RF amplifier620, a mixer622, a bandpass filter624, and a comparator626. Signal line602couples the received signal to bandpass filter618, which performs image rejection and protects RF amplifier620and mixer622from high-amplitude extraneous signals. Bandpass filter618couples the filter signal to RF amplifier620, which amplifies it and overcomes the noise figure of mixer622. Mixer622frequency translates the filtered and amplified signal to an IF (intermediate frequency), which is 455 kHz in exemplary receiver600. Bandpass filter624rejects spurious output signals from mixer622and largely defines selectivity of receiver600. Comparator626acts as a 1-bit A/D converter, providing a logic high signal on line627when the output signal from filter624exceeds a predetermined threshold and providing a logic low signal on line627otherwise.

The digital signal on line627enters a DSP subsystem605of receiver600and couples to a second mixer630implemented in subsystem605. Mixer430frequency translates the filtered, 1-bit signal on line627to a baseband signal, which is integrated by a summing module640.

The local oscillator signal (a 1-bit signal from an FPGA output line) controlling analog mixer622comes from a first local oscillator DDS module650, which is unstabilized. The local oscillator signal controlling digital mixer630comes from a second local oscillator DDS module660, which is phase-controlled by a stability compensating DDS module662.

DDS modules660and662cooperatively form a phase-stabilizing signal generator. A phase-stabilizing signal generator according to various aspects of the invention includes any hardware, software, or combination of both producing an output with phase that varies in a useful, predictable manner with respect to a reference signal, e.g., a broadcasted signal. Such variation can be configured to be opposite the expected variation of an unstabilized signal generator. In receiver600, the outputs of the phase-stabilizing generator formed by DDS modules662and660and unstabilized signal generator650are applied to successive mixers622and630.

Stability compensating DDS module662causes mixer630to frequency translate the first IF signal at line627with an induced phase instability. This phase instability is opposite that of stabilized DDS module650, and opposite the phase instability that mixer622consequently imparts to the received signal. Advantageously, the phase instabilities cancel each other out. The signal integrated by summing module640(e.g., for four seconds or about 100×106samples clocked at 24 MHz) is substantially phase stable with respect to sync pulses on bus340, and with respect to the broadcasted signal on line330that generates them.

The output of summing module640, on line610, varies with the phase of the received signal. Line610couples via bus320to control module310, where device300can implement functions of a spatial position computer. When device300is employed as tracker110of system100, for example, the spatial position computer it implements determines a spatial position of transmitter114and receiver116(which is typically but not necessarily the same as the position of tracker110itself) relative to transponder150.

As mentioned above, the configuration discussed with reference toFIGS. 3-6is merely exemplary. Again, a tracker, transponder, reference signal generator, transmitter, and receiver according to various aspects of the invention can include any suitable hardware, software, or combination of both for performing the respective functions of those devices.

Spatial position determination according to various aspects of the invention may be better understood with reference to an exemplary method700of FIG.7. Method700begins at process710, at which the tracker transmits a forward signal having phase φA. This phase represents the unknown, non-referenced phase of a transmitted signal after passing through various stages of signal processing. Method700continues at process720, at which a transponder receives the signal with a phase α. As shown inFIG. 7, phase a is directly proportional to the distance between tracker and transponder. Phase a also includes an unknown additive term θAthat results from signal processing phase shifts in the transponder receiver.

At process730, the transponder transmits a return signal at the received phase α. The return signal, as transmitted at the transponder, is thus also directly proportional to the distance between tracker and transponder. When received at the tracker, at process740, the return signal phase φBis directly proportional to twice this distance. Phase φBalso includes an unknown additive term φBthat results from signal processing phase shifts in the tracker receiver.

Method700concludes at process750, at which a spatial position computer (preferably in the tracker device itself) determines distance (including some unknown additive term) based on stored indicia of wavelength(s), here a common wavelength λ, of the forward and return signals.

A method800for determining an offset from a previous spatial position (here, change in distance) may be better understood with reference to FIG.8. In addition, a method900for determining spatial position expressed as an azimuthal angle may be better understood with reference to FIG.9.

Method800involves movement to three locations, by processes810,820, and830. At these locations, three distance measurements d1, d2and d3are obtained, by processes812,822, and832, respectively. The additive terms θAand θBprevent determination of an absolute spatial position (here, distance) based on a single measurement. Thus, the measured distance values d1, d2, and d3are proportional values that all include some unknown additive term.

Process840determines an offset Δd1from d2and the previous spatial position d1. Similarly, process850determines an offset Δd2from d3and the then-previous spatial position d2. These offsets (i.e., changes in distance) are output to a user by process860. ThoughFIG. 8depicts three distance measurements, this is only exemplary. As few as two can give meaningful results, and many more measurements are likely to be made during a typical search using a system according to various aspects of the invention.

Method900ofFIG. 9also involves movement to three locations, by processes910,920, and930. Unlike method800, method900includes processes914,924, and934for determining tracker position at these three locations, expressed as relative X and Y coordinates (X1,Y1), (X2,Y2), and (X3,Y3). The locations can be determined based on instructions to a user. For example, a user may be instructed to “take a measurement, move three paces north, take a measurement, then move three paces east and take a measurement.” In an advantageous variation, processes914,924, and934can employ a position determination device (e.g., optional GPS module122of tracker110) to determine positional coordinates for multiple locations as the user moves about in search of a lost object.

Processes912,922, and932obtain three distance measurements d1, d2, and d3at the respective known locations. Preferably, these processes, and processes812,822, and832of method800, each perform an instance of method700of FIG.7. Based on the known coordinates (X1,Y1), (X2,Y2), and (X3,Y3) and associated distance measurements d1, d2, and d3, process940determines spatial position, which can be expressed and displayed as an azimuthal angle from tracker to transponder.

Process940can employ any suitable technique for such position determination. For example, the azimuthal angle can be computed from the equation φ=tan−1(Δd1/Δd2). Given enough measurements, process940may also compute a rough near/far approximation of absolute distance between tracker and target.

PUBLIC NOTICE REGARDING THE SCOPE OF THE INVENTION AND CLAIMS

The inventor considers various elements of the aspects and methods recited in the claims filed with the application as advantageous, perhaps even critical to certain implementations of his invention. However, the inventor regards no particular element as being “essential,” except as set forth expressly in any particular claim. The following are various systems, devices, and methods contemplated by the inventor that omit various advantageous but non-essential elements discussed above.

A spatial position determination system, which omits transponder150ofFIG. 2, includes a tracker reference signal generator that is coupled to and phase-stabilized by a broadcasted signal. The system further includes a tracker transmitter and tracker receiver that are both coupled to and phase-stabilized by the tracker reference signal generator. The system further includes a spatial position computer that is coupled to the tracker receiver and at least one of the tracker reference signal generator and the tracker transmitter. The spatial position computer is responsive to indicia of a phase relationship between an output signal from the tracker transmitter and an input signal to the tracker receiver. Thus, the spatial position computer can determine a spatial position of the tracker transmitter and tracker receiver relative to a source of the input signal to the receiver.

An active reflector contemplated by the inventor, which omits tracker110ofFIG. 2, includes a receiver and a transmitter that is phase-controlled by the receiver. Such a device advantageously transmits a signal having a phase determined by the phase of the signal receives. Consequently, the device provides an “echo” of a signal for phase-based distance measurement without the need to overcome path loss for both the forward and return trip, as well as passive reflection loss and phase uncertainty induced by an irregular reflecting surface.

A phase-stabilized or phase-stabilizing signal generator, which can be advantageously employed in any device requiring an output phase-stabilized to an input, includes (1) a first DDS module having a phase accumulator that is clocked responsive to sync pulses, and (2) a second DDS module, coupled to the stabilizing frequency synthesizer and having a phase accumulator that is clocked by system clock pulses but forced to the accumulated phase of the first DDS module upon occurrence of a qualified sync pulse.

A phase-stabilized signal generator produces an output of substantially the same phase stability as the source of sync pulses and qualified sync pulses. A phase-stabilizing signal generator produces an output with phase that varies in a useful, predictable manner with respect to the source of pulses. As discussed above, such variation can be configured to be opposite the expected variation of an unstabilized signal generator. When the outputs of the phase-stabilizing and unstabilized signal generators are applied to successive mixers in a signal processing chain, the opposite variations cancel each other out. A signal that is frequency translated by the signal processing chain can thus avoid phase instability from the unstabilized signal generator.

While the invention is described herein in terms of preferred embodiments and generally associated methods, the inventor contemplates that alterations and permutations of the preferred embodiments and methods will become apparent to those skilled in the art upon a reading of the specification and a study of the drawings. Below is a listing of some examples of variations contemplated by the inventor and falling within the scope of the claims unless excluded by specific claim language.

EXAMPLE A

Instead of transmitting an unmodulated carrier as described above, tracker110can transmit a carrier phase modulated by a pseudo-random sequence. The chip time (i.e., duration of each phase modulation symbol) can be determined by the time base shared by the tracker and the target. The chip rate can be a predetermined fraction of the carrier frequency, or a predetermined fraction of the broadcasted signal's frequency.

Transponder150performs a correlation maximization search to determine the start time of the pseudo-random sequence. This correlation needs to be performed only during the initial synchronization acquisition process. All subsequent transmissions can remain synchronized due to the common time base.

In this variation, system100relies on the ability of both tracker110and transponder150to measure the phase of the signals they each receive. The DSSS phase modulation does not interfere with this measurement. In each DSSS receiver the incoming signal is multiplied by the pseudo-random phase sequence to yield a constant phase received signal.

The phase-modulated DSSS signal has a spectral bandwidth proportional to the chip rate. An equalization filter can be applied to reduce the effect of variable group delay across the band. This filter does not necessarily interfere with the phase measurement of the incoming signal. For example, without loss of generality, the equalization filter can be chosen to have zero phase at the carrier frequency.

EXAMPLE B

In some implementations, it may be desirable to phase stabilize to an FM broadcast station. However, FM signals do not have constant phase. Variations of tracker110and transponder150may overcome this issue by both computing an averaged reference signal in the same way. For example, each unit may compute the instantaneous broadcast frequency to be107divided by the time elapsed during the previous 107 cycles. Continuous computation would require a circular buffer.

EXAMPLE C

Phase stabilization can be to subcarriers of a broadcasted signal rather than carrier of the signal. For example, variations of reference signal generators112and152can phase stabilize other components to the color-burst frequency from TV stations, or to commercial subcarriers from FM broadcast stations.

EXAMPLE D

Frequency synthesis can be performed in the optical domain. Nonlinear optical media can be used to generate a transmitted signal or signals from a broadcasted coherent light signal. For example, if light generated by a single laser is dispersed over an area including a tracker and transponder, nonlinear optical media in each unit can be employed to generate the same phase-coherent type of light, of a wavelength different from that of the dispersed laser light.

EXAMPLE E

In typical implementations, the broadcasted signal is received by both a tracker and transponder from an external transmitter. With a suitably stable self-contained oscillator in either unit, however, such a signal can be broadcasted from that unit, received by the other, and employed by both for phase stabilization.

EXAMPLE F

Systems not requiring the benefits of digital phase stabilization can employ one or more conventional phase-locked loops (e.g., with a VCXO) for phase stabilization.

EXAMPLE G

An advantageous use of system100is the location of buried avalanche victims and missing skiers, hikers, firefighters, etc. Many ski resorts have poor broadcasted signal reception due to remoteness and surrounding mountains. To overcome this issue, a variation of broadcast transmitter105can be a local broadcast beacon having transmission coverage over an area that includes potential avalanche sites.

EXAMPLE H

In a variation, the transponder can transform the received forward signal with a nonlinear transfer function. The frequency scaled signal in such a variation is a harmonic of the received forward signal. As defined herein and in Barry Truax, ed., Handbook For Acoustic Ecology (1999), a harmonic is an integer multiple of a fundamental and a subharmonic is an integer submultiple or fraction of a fundamental.

EXAMPLE I

In another variation, the transponder can digitally frequency divide the received forward signal, whereby the frequency scaled signal is a subharmonic of the received forward signal.

EXAMPLE J

In another variation, the transponder can digitally synthesize the frequency scaled signal responsive to the received forward signal, whereby the frequency scaled signal is a subharmonic of the received forward signal.

EXAMPLE K

A loop antenna could be substituted for the whip antenna112of FIG.1.

EXAMPLE L

The antenna for transponder150can be a ferrite loopstick, or any suitable alternative. One such alternative is including a conductive loop in the collar of dog155that carries transponder150. Preferably, the loop is resonated with suitable capacitive tuning of the loop, with optional resistive Q dampening and an at least partially horizontal loop orientation (as illustrated inFIG. 1) to avoid directional nulls.

EXAMPLE M

Phase-stabilizing tracker110and transponder150to a single AM broadcast station (one of many options) requires both devices to choose the same station. In a variation, each unit can phase stabilize to numerous receivable AM stations in an aggregate, weighting each station's influence by the receive strength for that station to yield a reference signal whose phase varies slowly with the geographical position of the unit.

Accordingly, neither the above description of preferred exemplary embodiments, nor the code listing of a merely illustrative simulated embodiment below, nor the abstract defines or constrains the invention. Rather, the issued claims variously define the invention. Each variation of the invention is limited only by the recited limitations of its respective claim, and equivalents thereof, without limitation by other terms not present in the claim. For example, claims that do not recite any specific components of a spatial position computer read on methods that include, and exclude, advantageous components recited in other claims, such as memory cells including indicia of a plurality of previous spatial positions. As another example, claims not reciting limitations regarding components of a transponder read on devices and methods that include, and exclude, advantageous components such as a transponder reference signal generator.

In addition, aspects of the invention are particularly pointed out in the claims using terminology that the inventor regards as having its broadest reasonable interpretation; the more specific interpretations of 35 U.S.C. § 112(6) are only intended in those instances where the terms “means” or “steps” are actually recited. The words “comprising,” “including,” and “having” are intended as open-ended terminology, with the same meaning as if the phrase “at least” were appended after each instance thereof.