Low power low voltage differential signal receiver with improved skew and jitter performance

A folder common cascode circuit with symmetric parallel signal paths from the differential inputs to the differential outputs provides a low skew, low jitter, low power differential amplifier. The signal paths on either side of the differential amplifier are made equal with equal loads along each path. Pairs of complementary NMOS and PMOS transistor pairs with parallel complementary biasing stacks on the output cascode circuitry maintain symmetrical parallel signal paths, amplification and impedance loading from differential input to differential output. Output voltage translating inverters provide a higher voltage level output signal while maintaining low skew and jitter.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to interface circuits, and more particularly to translating interface circuits and receivers, and even more particularly to high speed interface receivers with low voltage differential input signals (LVDS) and complementary, low skew CMOS differential output signals.

2. Background Information

Low voltage differential signals are common for high speed signal transmission. Saturation effects are avoided and power is limited thereby encouraging the use of LVDS, and, since low voltage signals are prone to noise, differential signals, where the noise is common to both signals, generally will overcome this problem.

However, since common mode signals will occur, LVDS circuits are designed to accept differential signals that ride on a common mode level that, in the best case will range from the low power, or ground, rail to the Vcc power rail.

Designs for such translators strive for low power dissipation, increased speed, reduced edge delays, reduced jitter, reduced skew, and for efficient in the use of die area. Often the output of the receiver requires a TTL or CMOS logic level, so such designs also strive for high to low rail to rail outputs.

U.S. Pat. Nos. 6,275,073 B1, 6,252,432 B1 and 6,236,269 B1 set out a high speed differential input circuits that operates over a wide input common mode range but each provides only a single ended output. In these patents the differential input signals are each input to the gates of both an NMOS and a PMOS transistor so that NMOS will handle common mode voltages up to within about 200 millivolts of the high power rail and the PMOS down to within about 200 millivolts of the ground.

Notice that in the above prior art patents, the circuitry shown is not symmetrical. That is the differential inputs signal paths to the outputs is not the same for the two inputs. This lack of symmetry adds to the skew and jitter of these circuits.

There is a need to provide in many applications both a signal and its complement. The prior art circuits require a series inverter delay to complement the signal. In such cases the complement signal is offset from the output signal by the gate delay on each signal edge. The complement will have an inherent skew and non-symmetry due to the need for a series inverter.

There is a need for a differential low voltage receiver with a wide common mode range that provides complementary outputs with improved skew and jitter performance.

It will be appreciated by those skilled in the art that although the following Detailed Description will proceed with reference being made to illustrative embodiments, the drawings, and methods of use, the present invention is not intended to be limited to these embodiments and methods of use. Rather, the present invention is of broad scope and is intended to be defined as only set forth in the accompanying claims.

SUMMARY OF THE INVENTION

In view of the foregoing background discussion, the present invention provides a high speed, low power, differential receiver that provides complementary TTL and CMOS compatible differential outputs, and that demonstrates lower skew and jitter, and that accepts rail to rail common mode inputs while providing rail to rail complementary outputs.

Use of precise symmetry, duplicated circuitry and therefore loads along each of the differential input signal paths and full differential circuitry throughout the invention provides for the objective operational improvements of the present invention. Inspection of a preferred embodiment in FIG. 3 clearly shows this symmetry.

DETAILED DESCRIPTION OF AN ILLUSTRATIVE EMBODIMENT

FIG. 1 shows a cascode circuit with a differential input composed of IN and IN connected to the gates of NMOS's N 0 and N 1 , respectively. The drains of these NMOS transistors are fed into the sources of PMOS's P 4 and P 5 , respectively, as in a typical cascode circuit. Cascode circuits had the beneficial characteristic of a differential voltage signal input to high impedance gates (or bases, in bipolar circuits) of the NMOS transistors causing a output differential current from the high impedance drains that are fed into low impedances. If there were a resistance or other such high impedance loads on the drains, an inverted voltage signal would appear at the drains magnifying the Miller capacitance from gate to drain and, thereby, limiting the circuit's frequency range. Since the load on the drains is the low impedance sources (or emitters) little or no voltage signal appears at the drains of the NMOS's and the Miller capacitance is not magnified. However, a preferred embodiment of the present invention, as in FIGS. 1 and 2 shows a topology of the cascode common source load transistors shown folded to the right and current sources, I 9 and I 10 in FIG. 1 and I 5 and I 6 in FIG. 2 , making vertical connections to the Vdd and ground, respectively. This circuit configuration lends its name to the circuit as folded cascode. Designs of Analog CMOS Integrated Circuits by Behzad Razavi published by McGraw Hill is a good reference describing folded cascode circuitry and is hereby incorporated herein by reference.

The differential output voltage signal (Vdif , Vdif ), in FIG. 1 is taken between the drains of P 4 and P 5 . In FIG. 1 the current sources I 0 , I 9 and I 10 are used to bias the transistors. In a preferred embodiment I 9 and I 10 provide half the bias current, I 0 , for the input NMOS's with the other half going to the P 4 and P 5 . Also, as practitioners in the art will understand, the current sources are MOS devices configured as mirrored current sources.

FIG. 2 is a parallel folded cascode circuit to that of FIG. 1 but with the opposite polarity using a PMOS input pair. The input transistors are PMOS P 1 and P 2 with their drains connected to current sources 15 and 16 , respectively, and to low impedance sources of N 3 and N 4 , respectively.

The input NMOS pair of FIG. 1 allows the common mode input voltage to run from at least Vdd/2 to Vdd, and the PMOS pair of FIG. 2 allows the common mode input voltage to run from at least Vdd/2 to gnd. This parallel combination of differential input circuits allows the input common mode voltage to traverse from ground to Vdd. This can be seen from FIG. 1 where if IN and IN were at both at Vdd, the current sources I 9 and I 10 would still operate supplying half their currents to I 0 and the other half to P 4 and P 5 . As it evident from the schematic, if there were a differential input voltage the currents from I 9 and I 10 would unequally supply half their total currents for I 0 and the remaining currents to P 4 and P 5 in proportion to the differential input signal value. Similar operation exists for the PMOS circuit of FIG. 2 , with the resulting operation allowing the full rail to rail input common mode operation.

Notice that in both FIG. 1 and FIG. 2 the outputs are Vdif and Vdif . In fact N 3 and N 4 provide an active load for P 4 and P 5 with N 0 and N 1 as the operating differential input stage. Correspondingly, P 4 and P 5 are active loads for N 3 and N 4 with P 0 and P 1 as the operating differential input stage. When input common mode goes up to less than Vdd (Vtp Vdsat), P 1 and P 2 stop working but N 1 and N 2 still operate and amplify the input. Similarly when input common mode goes to less than gnd (Vtn Vdsat), N 1 and N 2 stop working but P 1 and P 2 still operate and amplify the input.

FIG. 3 circuit shows a more complete schematic drawn to show the parallel symmetrical nature of the circuitry and biasing. P 4 and P 5 and N 3 and N 4 , respectively, form low impedance loads with their respective sources connected the drains of the input tranistors, and active loads on each other with their drains connected to each other. As can be seen from FIGS. 3 and 4 , there is a parallel symmetrical nature to the entire circuit.

From inspection of FIG. 3 , the transistor pair P 3 and N 2 is symmetrical to P 6 and N 5 . P 3 and N 2 providing biasing to P 4 and N 3 . Similarly P 6 and N 5 provides biasing to P 5 and N 4 . Moreover, the pair P 4 /N 3 is a symmetrical duplicate of the P 5 /N 4 biasing pair. The symmetric parallel nature of this circuitry is verified by inspection.

The overall circuit symmetry provides low skew and low jitter performance with the differential output Vdif /Vdif being a true differential output. These differential outputs can be passed through identical inverters, as in FIG. 5 , to buffer the outputs while maintaining the low skew and jitter. Here symmetrical refers to the parallel circuitry in either the IN or the IN circuits being identical mirrors of each other including signal path lengths. That is, N 0 is an identical mirror of N 1 , and P 4 of P 5 ,and I 9 of I 10 , and N 3 of N 4 , as so on, and the signal path length from IN through to OUT is arranged to be equal to that from IN to OUT .

Still referring to FIG. 3 , the left side outside tree composed of I 8 , P 3 , N 2 , and I 4 and the right side composed of I 11 , P 6 , N 5 , and I 7 are arranged as symmetrical bias voltage providers for the single handling inner trees. These outside biasing trees are designed for lower current, 0.25 ma, to save power. The inner signal trees are, on the left, I 9 , P 4 , N 3 , and I 5 and on the right, I 10 , P 5 , N 4 and I 6 . P 4 and P 5 are running at (with no differential input) at 0.5 ma each. The other 0.5 ma from I 9 and I 10 run through N 0 and N 1 . These currents and the sizes of the components are only one preferred example and other sizes, current ranges and at other proportions may be used to advantage in the present invention.

Expanding on the symmetry discussion, in FIG. 3 , following the circuitry from the input through to the output the circuitry is identical and symmetrical so that the loading on each transistor is equal to its mirror transistor. Moreover, the layout signal path lengths are designed to be virtually identical. This means that the transistor sizes and parameters, the capacitances, resistances and layout path lengths are all identical along both the IN and the IN signal paths. This designed symmetry leads to minimized skew.

Jitter is an apparent signal riding on an actual differential signal causing variations in its period. One factor causing jitter is noise. One common source of noise is the power rail, Vdd. The circuit topology and component parameters are made symmetrical so that any signal riding on the power rail appears equally on both sides of the circuitry or in common mode and therefore its contribute to the differential signal is minimized. This factor is called the power supply rejection ratio, which is the signal gain (output signal value divided by the input signal value divided by a valued calculated from an output signal value due to a signal on the power rail basically the gain from the power rail to the output).

Back to FIG. 3 , it is instructive to follow a signal through the schematic. Here a differential signal at a common mode voltage of greater than Vdd/2 will be described traveling through the NMOS transistors, N 0 and N 1 . There will be a corresponding equivalent signal path through P 1 and P 2 for common mode levels below Vdd/2.

For description purposes consider both inputs at Vdd/2 and both outputs at Vdd/2 also. This state will typically only exist transiently during switching but the distribution of currents will be clear from the circuitry and from there differential signals will be evident. In this condition, one half of I 9 and I 10 travel through N 0 and N 1 to match I 0 . And one half of I 5 and I 6 travel through P 1 and P 2 to match I 1 . The remaining one half of I 9 supplies the one half of the remaining I 5 , and the remaining one half of I 10 supplies the remaining one half of I 6 . I 8 runs to I 4 and I 11 to I 7 .

Now, from FIG. 1 , if IN is higher than IN at some common mode voltage, more of I 0 travels through N 0 and less through N 1 . Therefore less of I 9 will travel through P 4 and more of I 10 will travel through P 5 . With a large enough differential all of I 0 travels through N 0 and which is supplied from I 9 leaving no current for P 4 , and correspondingly all on I 10 travels through P 5 , pulling Vdif high. In this case, with matched current source, all of I 6 accepts all of I 10 . In this case, also, all of I 8 runs through P 3 and N 2 to I 4 . Under this condition, N 3 is on and pulled low by I 5 and so Vdiff is lowered.

If the extreme case is not considered, the in the above analysis if IN is slightly higher than IN the resultant analysis will drive more I 9 current through N 0 leaving less for P 4 , and more of I 10 will travel through P 5 and less through N 1 . The result of more current through P 5 and N 4 and less through P 4 and N 3 , resulting in a higher Vdiff compared the Vdif

An advantage of the input structure shown in FIGS. 1 and 2 is the available input common mode operation. Up this point the discussion considered a differential input and a differential output. But, each input signal is at some common voltage with respect to ground (gnd) and Vdd. So, for example if IN were at 1.25 volts (with respect to gnd) and if IN were at 1.15 volts, the differential signal would be 100 millivolts, and the common mode voltage would be 1.2 volts. It is advantageous to design circuits which will operate over wide ranges of common mode voltages. A differential circuit with to power rail of 3.3 volts and gnd operates properly with an input common mode signal that could run from gnd to 3.3 volts would constitute a flexible circuit design. The use of the NMOS input circuitry of FIG. 1 and the PMOS circuitry of FIG. 2 allows this extensive input common mode.

It should be understood that above-described embodiments are being presented herein as examples and that many variations and alternatives thereof are possible. Accordingly, the present invention should be viewed broadly as being defined only as set forth in the hereinafter appended claims.