Method and apparatus for high-speed data transfer employing self-synchronizing quadrature amplitude modulation

A Quadrature Amplitude Modulation (QAM) method and apparatus including a QAM transmit modulator with at least one unbalanced mixer, which creates an asymmetric two-dimensional (2-D) QAM symbol constellation. The asymmetrical symbol constellation provides baseband symbol clock signal leakage sufficient to facilitate quick and simple baseband symbol clock recovery and signal channel compensation at the QAM receiver without significantly degrading the system bit-error rate (BER). While slightly degrading static BER, overall system performance is improved when considering baseband symbol clock recovery and received signal compensation for an imperfect signal channel. This allows QAM to be deployed in systems where QAM is otherwise prohibitively expensive and improves overall system performance for any existing QAM system application without additional bandwidth, cost or complexity.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates generally to electronic information transfer systems and more particularly to a communication system for transferring data in a pipe inspection system.

2. Description of the Related Art

Analog and digital Quadrature Amplitude Modulation (QAM) methods for amplitude modulating two symbol clocks phase-locked in quadrature have been known and used since the early days of signal processing and are widely used today. For example, analog QAM is used to transfer the chroma component information in the 1953 National Television System Committee (NTSC) and the 1963 Phase Alternating Line (PAL) standard television signals and a 1977 Compatible QAM variation (C-QUAM) is still used to transfer the stereo difference information in some AM stereo radio signals. More recently, a variety of digital QAM schemes (quantized QAM) were adapted for widespread use in cellular systems and for other wireless applications, including the WiMAX and Wi-Fi 802.11 standards.

Advantageously, digital QAM may be configured with Amplitude-Shift Keying (ASK) to provide many data bits per symbol and thereby increase data transfer rates in a channel without increasing Inter-Symbol Interference (ISI). Amplitude modulating two symbol clocks in quadrature (QAM) can be equivalently viewed as both amplitude modulating and phase modulating a single symbol clock and each such modulation value (amplitude and phase) can be represented as a single point (symbol) on the phase plane diagram, as is well-known in the art. For example, by using two distinct amplitudes and four phase shift states for each of these amplitudes, a single symbol clock cycle can serve to carry one symbol having eight states; equivalent to three bits of information. In this example, a 5 MHz channel baseband can transfer data at 15 Mb/s at the expense of requiring a more robust method for reducing the impact of noise and increasing the Signal-to-Noise Ratio (SNR) to permit recovery of the significantly higher number of discrete signal amplitudes involved in each symbol clock cycle.

Proper separation of the I(t) and Q(t) quadrature components of a digital or analog QAM signal requires the coherent demodulator signal phase at the receiver to be exactly in phase with the received QAM signal carrier. Even a small demodulating phase error introduces crosstalk between the I(t) and Q(t) quadrature components recovered from a digital or analog QAM signal. Both symbol clock and carrier recovery systems in a receiver attempt to derive information about timing from the received signal, often in a similar manner. While carrier recovery is only necessary in a coherent demodulation system, symbol clock recovery is required in all schemes, and accurate clock recovery is essential for reliable data transmission. Confusion often exists between clock and carrier recovery. Clock recovery attempts to synchronize the receiver clock with the baseband symbol rate transmitter clock, whereas carrier recovery attempts to align the receiver local oscillator with the transmitted carrier frequency.

Thus, symbol clock synchronization at the receiver must be handled somehow in any QAM system. Any phase and frequency variations introduced by the channel must be removed at the receiver by properly tuning the sine and cosine components of the local QAM demodulator, which requires a local symbol clock phase reference that is typically provided by some useful version of a local Phase-Locked Loop (PLL). But this local phase reference must somehow be synchronized with the received QAM signal symbol clock. For example, early analog QAM television systems transmit a burst of the color subcarrier after each horizontal synchronization pulse for local clock phase reference synchronization.

The QAM art has evolved in various ways to increase throughput and reliability. A typical QAM data communication system includes a transmitter, a receiver, and an unknown time-invariant channel in which a complex-valued sequence of input data representing a series of symbols selected from a complex symbol alphabet (also denominated a “constellation” on the complex I-Q plane or “phase plane”) are sent through the channel to be interpreted by the receiver. Conventional QAM systems assume that channel noise is independent of input data and relatively stationary. Some distortion of the transmitted signal is typical of non-ideal channel media including wired and wireless connections.

The QAM demodulator is by far the most complex element of the QAM system. The demodulator must detect the phase and amplitude of the received signal, decode each symbol based on the phase and amplitude of the baseband symbol clock and then finally convert the symbol data back to a serial stream. The baseband symbol clock must be recovered to complete the symbol demodulation. Clock recovery is a recurring problem with any digital signal processing system.

The QAM art is replete with improvements intended to increase channel data transfer capacity while reducing receiver cost and complexity. There is an undesirable level of complexity and overhead in conventional QAM receivers for filtering signals and recovering baseband symbol clock synchronization. In applications where channel bandwidth is limited, such as pipe inspection system channels with a handful of hard-wired conductors, additional problems include correcting for a variable-length copper channel and limiting camera-end hardware complexity to facilitate the small package size necessary for movement inside pipes.

Practitioners in the art have proposed a wide variety of methods simplifying the QAM carrier and clock recovery problem. For example, in U.S. Publ. Appl. No. 2009/0,147,839 A1, Grenabo discloses an improved phase error detector for a QAM receiver but neither considers nor suggests any symbol constellation adjustments. Similarly, in U.S. Pat. No. 7,283,599 B1, Herbig discloses an improved phase error detector for a QAM receiver suitable for improving phase locking characteristics but neither considers nor suggests using an asymmetric symbol constellation. And, in U.S. Pat. No. 4,987,375, Wu et al. disclose a carrier lock detector for a QAM system employing symbol detection ratios and useful for improved reliability at low SNR but neither consider nor suggest any symbol constellation adjustments.

Practitioners in the art have also proposed a wide variety of methods for improving QAM system performance through manipulation of the symbol constellations. For example, in U.S. Publ. Appl. No. 2008/0,317,168 A1, Yang et al. disclose an integer spreading rotation technique for shaping symmetric QAM symbol constellations to enhance signal space diversity but neither consider nor suggest techniques for improving baseband symbol clock recovery at the receiver. These practitioners appear to firmly believe that the QAM symbol constellation must be as symmetric as possible about the phase plane origin to minimize the system Bit-Error Rate (BER).

Some practitioners have found certain slight asymmetries in the QAM symbol constellation to have some utility but have neither taught nor suggested using changes to the symbol constellation to improve baseband symbol clock recovery in QAM system receivers. For example, O'Hara et al. (“Orthogonal-Coded Selective Mapping (OCSM) For OFDM Peak-To-Average Power Reduction Without Side Information,”Proceeding of the SDR04Technical Conference and Product Exposition.2004) propose a selective mapping (SM) method for reducing peak-to-average power (PAP) in Orthogonal Frequency Division Multiplexing (OFDM) systems that is achieved by introducing a very small asymmetry to the QAM subcarrier constellations before scrambling. But O'Hara et al. take pains to point out that this does not mean that the QAM subcarrier constellations are no longer zero-mean over time because the subsequent antipodal scrambling process returns the subcarrier symbol constellations to zero-mean symmetry again before transmission.

Other practitioners have suggested using a pilot tone in a QAM channel to improve channel estimation. For example, Tariq et al. (“Efficient Implementation Of Pilot-Aided 32 QAM For Fixed Wireless And Mobile ISDN Applications,”Vehicle Tech. Conf. Proc.,2000,VTC2000-Spring Tokyo.2000 IEEE 51st, Vol. 1, pp. 680-684) discloses an improved QAM system where a gap is created in the center of the information bearing signal spectrum and a pilot tone inserted therein before transmission. Tariq et al. neither teach nor suggest that their pilot tone has any relationship to the QAM baseband symbol clock; in fact, they teach using the pilot tone at the receiver only for the purpose of channel estimation and compensation. In U.S. Pat. No. 3,813,598, Stuart discloses a pilot-tone aided QAM carrier recovery system that adds a pilot tone to the QAM transmission either above or below the QAM modulator output spectrum, which may be recovered and used to deduce channel distortion effects at the receiver, but Stuart neither considers nor suggests any manipulation of the symmetric QAM symbol constellation for baseband symbol clock recovery. In U.S. Pat. No. 6,493,490 B1, Lin et al. disclose an improved phase detector for carrier recovery in a dual-mode QAM/VSB (Vestigial Sideband) receiver system. Lin et al. discuss creating a pilot-tone aided Offset-QAM signal by first delaying the Q component by one half of a symbol, thereby offsetting the Q rail, in time, from information on the I rail, but neither consider nor suggest using an asymmetric QAM symbol constellation. Hyun et al., (“Interleaved 5820 Code For Insertion Of Carrier And Clock Pilots In 64-QAM Systems,”IEEE Electronics Letters, Vol. 27, No. 18, pp. 1635-6, 29 Aug. 1991) disclose a method for selecting symbols from a symmetric diamond-shaped symbol constellation to introduce a spectral null at the Nyquist frequency, thereby permitting the detection of a low-power clock pilot signal inserted at the null frequency, but neither consider nor suggest using an asymmetric QAM symbol constellation.

SUMMARY OF THE INVENTION

This invention arises from the unexpectedly advantageous observation that operating a Quadrature Amplitude Modulation (QAM) transmitter modulator with at least one unbalanced mixer, which creates an asymmetric two-dimensional (2-D) QAM symbol constellation, provides baseband symbol clock signal leakage sufficient to facilitate quick and simple baseband symbol clock recovery at the QAM receiver without significantly degrading the system Bit-Error Rate (BER). In fact, the QAM method of this invention flattens the system BER curve to reduce the Signal-to-Noise Ratio (SNR) required to provide lower BERs by as much as several decibels (dB). This is a profound and completely unexpected observation that has advantageous applications in many QAM systems, including (without limitation) pipe inspection systems, cell phone systems, commercial broadcast systems, Wi-Fi systems and many others.

It is a purpose of this invention to provide QAM channel baseband symbol clock recovery that reduces the system BER, complexity and computational load in certain SNR regions.

It is an advantage of this invention that it may be extended to any system generally relying on QAM methods to encode a transmitted signal. More specifically, the QAM method of this invention may be adapted to improve the lower functional layers (the physical transmission, reception, media correction and timing recovery elements) in certain SNR regions of any data transmission and reception system using a variant of QAM or any of its derivatives that employ two-dimensional (2-D) symbol constellations, such as Orthogonal Frequency-Division Multiplexing (OFDM), Quotient Quadrature Amplitude Modulation (QQAM), etc. Except for the improved BER in certain SNR regions, the QAM method of this invention does not affect the higher QAM system functional layers known in the art, such as forward error correction coding, symbol scrambling, symbol mapping, etc.

It is an advantage of this invention that the effects of the QAM channel characteristics can be automatically corrected at the receiver without additional receiver complexity or cost.

It is an advantage of this invention that, in a pipe inspection system with limited camera-transmitter space, the processing complexity is constrained to the QAM receiver, reducing space and complexity requirements for the camera-transmitter.

In one aspect, the invention is a method for transferring data through the signal channel including the steps of encoding the data to produce a first baseband modulating signal I(t) and a second baseband modulating signal Q(t) whose amplitudes together represent a time series of complex symbols (I, Q) each selected from a two-dimensional (2-D) constellation of symbols distributed on the phase plane about the origin such that at least one of the baseband modulating signals has a substantially non-zero mean amplitude; multiplying the first baseband modulating signal I(t) by a first baseband symbol clock signal to produce a first modulation product signal and multiplying the second baseband modulating signal Q(t) by a second baseband symbol clock signal to produce a second modulation product signal, where the phases of the first and second baseband symbol clock signals are generally fixed in quadrature; summing the first and second modulation product signals to produce a transmitter output signal; coupling the transmitter output signal through the signal channel to the data receiver; and demodulating the first and second modulation product signals at the data receiver to recover the series of complex symbols (I, Q).

In another aspect, the invention is a communication system including a data transmitter having an input for accepting data, a Quadrature Amplitude Modulation (QAM) encoder coupled to the data input for producing, responsive to the data, a first baseband modulating signal I(t) and a second baseband modulating signal Q(t) whose amplitudes together represent a time series of complex symbols (I, Q) each selected from a two-dimensional (2-D) constellation of symbols distributed on the phase plane about the origin such that at least one of the baseband modulating signals has a substantially non-zero mean amplitude, a QAM modulator coupled to the QAM encoder for multiplying the first baseband modulating signal I(t) by a first baseband symbol clock signal to produce a first modulation product signal and multiplying the second baseband modulating signal Q(t) by a second baseband symbol clock signal to produce a second modulation product signal, where the phases of the first and second baseband symbol clock signals are generally fixed in quadrature, and for summing the first and second modulation product signals to produce a transmitter output signal, and an output for coupling the transmitter output signal to a signal channel; and a data receiver having a signal input coupled to the signal channel for accepting the transmitter output signal, and a QAM demodulator coupled to the signal input for recovering the series of complex symbols (I, Q) from the first and second modulation product signals.

In yet another aspect, the invention is a data modulator for a video transmitter including an input for accepting data; a Quadrature Amplitude Modulation (QAM) encoder coupled to the data input for producing, responsive to the data, a first baseband modulating signal I(t) and a second baseband modulating signal Q(t) whose amplitudes together represent a time series of complex symbols (I, Q) each selected from a two-dimensional (2-D) constellation of symbols distributed on the phase plane about the origin such that at least one of the baseband modulating signals has a substantially non-zero mean amplitude; and a QAM modulator coupled to the QAM encoder for multiplying the first baseband modulating signal I(t) by a first baseband symbol clock signal to produce a first modulation product signal and multiplying the second baseband modulating signal Q(t) by a second baseband symbol clock signal to produce a second modulation product signal, where the phases of the first and second baseband symbol clock signals are generally fixed in quadrature, and for summing the first and second modulation product signals to produce a transmitter output signal.

In one embodiment, the invention is a pipe inspection system including a video transmitter having a video camera adapted to produce video data, and a QAM modulator coupled to the video camera, including a symbol encoder for producing, responsive to the video data, a first baseband modulating signal IT(t) and a second baseband modulating signal QT(t) whose amplitudes together represent a time series of complex transmitter symbols (IT, QT) each selected from a two-dimensional (2-D) constellation of symbols distributed on the phase plane about the origin such that at least one of the baseband modulating signals has a substantially non-zero mean amplitude, a baseband symbol clock oscillator for producing first and second baseband symbol clock signals generally fixed in quadrature, a dual multiplier coupled to the symbol encoder and baseband symbol clock oscillator for multiplying the first baseband modulating signal IT(t) by the first baseband symbol clock signal to produce a first modulation product signal and for multiplying the second baseband modulating signal QT(t) by the second baseband symbol clock signal to produce a second modulation product signal, a summer coupled to the dual multiplier for summing the first and second modulation product signals to produce a transmitter output signal, and a filter coupled to the summer for producing a filtered transmitter output signal; a mechanical cable assembly coupled to the video transmitter for urging the video transmitter through a pipe under inspection and including an electrical conductor coupled to the QAM modulator for accepting the filtered transmitter output signal; and a video receiver having a signal conditioner coupled to the electrical conductor for producing a baseband receiver input signal representing the filtered transmitter output signal, a QAM demodulator coupled to the signal conditioner, including a baseband symbol clock detector for detecting the first baseband symbol clock signal from the receiver input signal, a baseband symbol clock recovery oscillator coupled to the baseband symbol clock detector for producing a first recovered baseband symbol clock signal generally synchronized with the first baseband symbol clock signal and for producing a second recovered baseband symbol clock signal generally fixed in quadrature with the first recovered baseband symbol clock signal, a dual multiplier coupled to the baseband symbol clock recovery oscillator for multiplying the baseband receiver input signal by the first and second recovered baseband symbol clock signals to produce first and second demodulation product signals, respectively, a dual filter coupled to the dual multiplier for producing, responsive to the first and second demodulation product signals respectively, first and second baseband demodulated signals, IR(t) and QR(t), whose amplitudes together represent a time series of complex receiver symbols (IR, QR), and a decoder coupled to the QAM demodulator for recovering the video data from the first and second demodulated signals, IR(t) and QR(t), and a video display coupled to the QAM demodulator for producing images responsive to the video data.

The foregoing, together with other objects, features and advantages of this invention, can be better appreciated with reference to the following specification, claims and the accompanying drawing.

DETAILED DESCRIPTION OF THE EMBODIMENTS

Digital Quadrature Amplitude Modulation (QAM) schemes may be better understood with reference to the well-known two-dimensional (2-D) QAM symbol constellation diagram showing the QAM symbol states each represented as two (I and Q) amplitudes mapped as points on a complex I-Q plane (herein also denominated “the phase plane”). These 2-D symbol constellation mappings may also be represented as a radius amplitude and a phase angle measured from the phase plane origin, for example, but are generally understood to represent two amplitudes measured from the phase plane origin along the respective orthogonal I and Q axes. In QAM systems, the 2-D symbol constellation states are often arranged in a symmetrical square grid with equal vertical and horizontal spacing, although many other symmetrical configurations are known to be useful (e.g., Cross-QAM). As digital data are usually binary, the number of states (points or symbols) in the constellation is usually a power of two. Because the digital QAM symbol constellation is usually square, the common grids are numbered in powers of four; providing us with 16-QAM, 64-QAM, and 256-QAM systems, etc. These well-known square QAM symbol constellations go as high as 4096-QAM, which provides 4 kb/symbol with 64 different amplitude levels in both I and Q. With a higher-order constellation, the QAM system can transmit more bits per symbol but the points are more closely spaced for the same mean constellation energy and are thus more susceptible to noise and other corruption, producing higher bit error rates. Thus, higher-order QAM delivers more data less reliably than lower-order QAM for a given mean constellation energy.

These square symbol constellations are also denominated Type III QAM constellations. A Type I QAM symbol constellation has states arranged symmetrically about the phase plane origin along equally-spaced radial lines extending out from the phase plane origin with the same number of states in each of several concentric circles. A Type II QAM symbol constellation is similar to the Type I but reduces the number of states on the inner concentric circles (because phase angles detection is less accurate at lower amplitudes) while retaining symmetry about the phase plane origin. Type III QAM symbol constellations are square and centered on the phase plane origin. Each state is a 2-D value (I, Q) representing one of “n” amplitudes in I-space and one of “n” amplitudes in Q-space. It graphically represents each QAM symbol with amplitudes alone and the implicit phase angle defined on the phase plane by arctan(I/Q) arises only because of the 2-D representation of the amplitude pair (I, Q).

FIG. 1is a schematic diagram illustrating a typical 256-QAM Type III constellation100from the prior art. In this “square” constellation, the I-space values are represented as sixteen amplitudes ranging from −7.5 units to +7.5 units spaced along the I-axis102and the Q-space values are represented as the same sixteen amplitude values spaced along the Q-axis104. The I-space values are equally spaced by 1.0 unit as exemplified by the spacing106and the Q-space values are equally spaced by the same amount as exemplified by the spacing108. I-axis102and Q-axis104cross orthogonally at the phase plane origin110. Each symbol state is represented as a finite region about a point (I, Q), as exemplified by the symbol state112, which represents eight bits of data; four bits encoded in each of the sixteen amplitude values reserved for I and four bits encoded in each of the sixteen amplitude values reserved for Q. Constellation100is disposed with the I-space and Q-space ranges symmetrically centered about phase plane origin110such that any I-space baseband modulating signal I(t) and Q-space baseband modulating signal Q(t) together representing a random time-series of (I, Q) symbols will both have zero-mean amplitudes (no DC components) to eliminate clock leakage in the manner well-known in the art.

FIG. 2is a graph of a typical 256-QAM signal spectrum114from a typical 256-QAM data transmission system from the prior art (e.g.,FIG. 3) using symbol constellation100(FIG. 1) while transmitting pseudorandom data at two million symbols per second (using a baseband symbol clock frequency of 2 MHz). As seen inFIG. 2, there is no evidence of any additional signal at the baseband symbol clock frequency116or anything else sufficiently “obvious” to allow reconstruction of the baseband symbol clock timing at the receiver. Thus, the QAM reception problem remains complex and unreliable, as is well-known. Various timing recovery tricks are known in the art, ranging from “early-late” guesses (using a predictor-corrector method) to a combined phase-frequency detector, for example. Note that channel estimation and compensation also must be accomplished using the only available information, which is limited to wide-band or narrowband power estimation. Various channel estimation solutions known in the art include injecting separate “pilot” signals and other similarly complex techniques, for example. These constraints add unwanted complexity to any QAM receiver.

FIG. 3is a schematic diagram illustrating a typical QAM communication system200from the prior art, including a QAM data transmitter202and a QAM data receiver204coupled by a signal channel206. InFIG. 3, a data input208accepts a stream of incoming data210for processing and transmission. Incoming data210is routed to the encoder212for randomizing, interleaving, error-correction and other high-level encoding, for example. The randomized data214is then routed to the QAM encoder216, which separates the data into the two baseband modulating signals, IT(t)218and QT(t)220, which together represent a time-series of complex transmitter symbols (IT, QT) (not shown) selected according to the mapping of each pair of four-bit sequences of randomized data214onto constellation100(FIG. 1). This mapping is important because each complex transmitter symbol (IT, QT) represents eight bits in this example. A channel deficiency is most likely to cause a symbol error by incorrectly assessing the corresponding received symbol as the one immediately adjacent the transmitted symbol in constellation100(FIG. 1), so encoder212must encode incoming data210to minimize the overall bit error rate arising from simple symbol errors. Simple binary coding, for example, is not very robust against bit errors (e.g., a single symbol step from 01111111 to 10000000 collects eight bit errors) so the symbol mapping strategy must be chosen carefully, as is well known. A “Gray” code is useful and commonly used.

The two baseband modulating signals, IT(t)218and QT(t)220, are accepted by a QAM modulator222that includes an I-modulator224and a Q-modulator226embodied as a dual multiplier. I-modulator224modulates a zero-degree-phase baseband symbol clock signal228from the baseband symbol clock oscillator229by multiplying it with baseband modulating signal IT(t)218to produce an I-modulation product signal230and Q-modulator226modulates a ninety-degree-phase baseband symbol clock signal232by multiplying it with baseband modulating signal QT(t)220to produce a Q-modulation product signal234. A summer236then adds I-modulation product signal230and Q-modulation product signal234in the usual manner to produce a transmitter output signal238, which, in this example, is filtered and conditioned at the filter and driver assembly240to produce a filtered transmitter output signal242that is conditioned for transfer through the physical transmission medium in signal channel206to QAM receiver204. Zero-degree-phase and ninety-degree-phase baseband symbol clock signals228and232are said to be generally fixed in quadrature because they are phase-locked to one another with a 90-degree phase difference in the usual manner. Signal channel206may include conductive wiring, optical fiber, modulated radio frequency or optical signals in free space, or any other useful channel means known in the art, for example. Filter and driver assembly240may include an additional modulator(s) (not shown) for reconditioning transmitter output signal238as a modulation product of another carrier signal more suited to the signal channel medium, for example.

Continuing withFIG. 3, a signal conditioner244in QAM data receiver204accepts from signal channel206a channel signal246that represents filtered transmitter output signal242in some manner, depending on particular channel characteristics, added noise, and the like. Signal conditioner244may include an additional demodulator(s) (not shown) for recovering the baseband component of channel signal246when using another carrier signal more suited to the signal channel medium, for example. Signal conditioner244restores the signal level and provides any additional (usually analog) reconditioning necessary to produce a baseband receiver input signal248. From here, receiver input signal248takes two paths; the first taking it to a baseband symbol clock detector250for baseband symbol clock timing recovery and the second taking it to an equalization and correction circuit252for any additional processing necessary to correct for noise, intersymbol interference (ISI) and other unwanted effects of the trip through signal channel206. Baseband symbol clock detector250includes a baseband symbol clock recovery oscillator254that produces a zero-degree-phase recovered baseband symbol clock signal256and a ninety-degree-phase recovered baseband symbol clock signal258, which are generally fixed in quadrature and respectively synchronized with baseband symbol clock signals228and232above. Equalization and correction circuit252produces a baseband receiver input signal260that (as much as possible) represents the recovery of transmitter output signal238.

The baseband receiver input signal260from equalization and correction circuit252is routed to the QAM demodulator262for recovery of the two baseband demodulated signals, IR(t)264and QR(t)266, together representing a time series of complex receiver symbols (IR, QR) that (as much as possible) represent the recovery of the initial time-series of complex transmitter symbols (IT, QT) discussed above. This is accomplished by an I-demodulator268and a Q-demodulator270embodied as a dual multiplier. I-demodulator268demodulates baseband receiver input signal260by multiplying it with zero-degree-phase recovered baseband symbol clock signal256to produce an I-demodulation product signal272and Q-demodulator270demodulates baseband receiver input signal260by multiplying it with ninety-degree-phase recovered baseband symbol clock signal258to produce a Q-demodulation product signal274. I-demodulation product signal272is passed through a first low-pass filter276to recover baseband demodulated signal IR(t)264and Q-demodulation product signal274is passed through a second low-pass filter278to recover baseband demodulated signal QR(t)266in the usual manner. From QAM demodulator262, both baseband demodulated signals, IR(t)264and QR(t)266are presented to the QAM decoder280for reversal of the 2-D constellation mapping process performed in QAM encoder216and discussed above to produce the recovered randomized data282. Finally, in the decoder284, the randomizing, interleaving, error-correction and other high-level encoding processing performed in encoder212and discussed above is reversed to produce a stream of output data286corrected for errors where possible and timed according to a bit rate clock signal288from baseband symbol clock detector250. A feedback line290to equalization and correction circuit252permits recovery optimization by adjusting the conditioning of receiver input signal248to minimize errors detected and corrected in recovered randomized data282by decoder284, for example.

To appreciate the detailed operation of QAM communication system200(FIG. 3) consider a simple QAM encoding example based on 256-QAM Type III constellation100(FIG. 1). Referring toFIG. 3, consider the details of passing several complex transmitter symbols (IT, QT) through QAM communication system200starting with QAM modulator222and assuming that transmitter output signal238passes through signal channel206to QAM data receiver204with perfect fidelity.

As QAM operates with quantized amplitudes, assume that the I-axis102and Q-axis104range from −7.5 units to 7.5 units, in 1.0 unit steps. For example, the units may represent volts or any other physical denomination suitable to the application. This arrangement thereby provides sixteen amplitudes along each axis that may be conveniently mapped (in any sequence) to the sixteen available four-bit binary sequences ranging from 0000 to 1111, consistent with the above discussion. Assume for this illustration that the stream of incoming data210is sixteen bits long and may be mapped by constellation100to the following two exemplary complex transmitter symbols (IT, QT) over two complete four-part baseband symbol clock cycles (using logical amplitude units):Complex transmitter symbols (IT, QT): (+1.5, −6.5) and (−3.5, +5.5)

Assuming that, in QAM modulator222, baseband symbol clock signal228is a square wave with either a 0 or 1 logical amplitude, the following symbol clock signal values describe the two complete four-part symbol clock cycles mapping onto these two complex transmitter symbols (IT, QT):Zero-degree-phase baseband symbol clock signal228(I-clock): +1,+1,−1,−1,+1,+1,−1,−1Ninety-degree-phase baseband symbol clock signal232(Q-clock): −1,+1,+1,−1,−1,+1,+1,−1

After the multiplications in I-modulator224and Q-modulator226, the resulting modulation product signal amplitudes over the two four-part baseband symbol clock cycles are:I-modulation product signal230: +1.5,+1.5,−1.5,−1.5,−3.5,−3.5,+3.5,+3.5Q-modulation product signal234: +6.5,−6.5,−6.5,+6.5,−5.5,+5.5,+5.5,−5.5

When added together at summer236, the amplitude of transmitter output signal238over the two four-part baseband symbol clock cycles is:Transmitter output signal238: +8.0,−5.0,−8.0,+5.0,−9.0,+2.0,+9.0,−2.0

In this example, transmitter output signal238is also the receiver input signal248arriving at QAM data receiver204from which two complex receiver symbols (IR, QR) must be recovered and decoded to recover the stream of incoming data210without error if possible.Receiver input signal248: +8.0,−5.0,−8.0,+5.0,−9.0,+2.0,+9.0,−2.0

Assuming that zero-degree-phase recovered baseband symbol clock signal256can be precisely synchronized with zero-degree-phase baseband symbol clock signal228in QAM data transmitter202, then baseband symbol clock recovery oscillator254provides the following logical amplitudes over two complete four-part recovered baseband symbol clock cycles:Zero-degree-phase recovered baseband symbol clock signal256: +1,+1,−1,−1,+1,+1,−1,−1Ninety-degree-phase recovered baseband symbol clock signal258: −1,+1,+1,−1,−1,+1,+1,−1

Thus, after the multiplications in I-demodulator268and Q-demodulator270, the following two demodulation product signals are produced complete four-part recovered baseband symbol clock cycles:I-demodulation product signal272: +8.0,−5.0,+8.0,−5.0,−9.0,+2.0,−9.0,+2.0Q-demodulation product signal274: −8.0,−5.0,−8.0,−5.0,+9.0,+2.0,+9.0,+2.0

Passing each of these two product signals through their respective low-pass filters276and278can be assumed to produce a average value over each full baseband symbol clock cycle, thereby producing the following logical amplitude averages for the two baseband demodulated signals, IR(t)264and QR(t)266over two complete recovered baseband symbol clock cycles:First baseband demodulated signal IR(t)264: 6.0/4=+1.5, −14.0/4=−3.5Second baseband demodulated signal QR(t)266: −26.0/4=−6.5, 22.0/4=+5.5Complex receiver symbols (IR, QR): (+1.5, −6.5) and (−3.5, +5.5)

Finally, in QAM decoder280and decoder284, the two complex receiver symbols (IR, QR) are decoded with reference to constellation100(FIG. 1) to obtain the stream of output data286that represents (ideally without error) original data stream210. In this example, two symbols at the channel symbol rate serves to transmit and correctly receive sixteen bits of information. In a practical application, assuming adequate timing recovery means, the received signal may be sampled four times during the baseband symbol clock cycle to retrieve the data correctly.

Notice that some form of timing recovery must be performed in baseband symbol clock detector250to recover baseband symbol clock signals256and258as well as bit rate clock signal288. The QAM receiver clock recovery function is expensive in terms of computing (and electrical) power and parts cost. The reason for this may be appreciated with reference toFIG. 2. Note that 256-QAM signal spectrum114inFIG. 2is nulled at 0 Hz (DC) and 4 MHz (twice the 2 MHz symbol clock rate), but has no prominent component at the symbol clock rate116so that baseband symbol clock recovery is feasible only by applying exotic statistical methods to receiver input signal248. But these exotic computational components are expensive. So, although QAM data communication system200provides some utility and QAM data transmitter202alone is relatively inexpensive, QAM data receiver204can be too complex and expensive for simple applications, such as pipe inspection systems, for example.

FIG. 4is a schematic diagram illustrating an exemplary QAM data communication system embodiment300of this invention, including a QAM data transmitter302and a QAM data receiver304coupled through a signal channel306. In this embodiment, many of the transmitter functions in QAM data transmitter302are embodied as software (or firmware) programs in a Digital Signal Processor (DSP)307with programming adapted to accept a stream of incoming data310for processing and transmission. Incoming data310is routed to the encoder312for randomizing, interleaving, error-correction and other high-level encoding, for example. The randomized data314is then routed to the QAM encoder316, which separates the data into the two baseband modulating signals, IT(t)318and QT(t)320, which together represent a time-series of complex transmitter symbols (IT, QT) (not shown) selected according to the mapping of each pair of four-bit sequences of randomized data314onto an asymmetric symbol constellation of this invention exemplified by the 256-QAM asymmetrical symbol constellation400shown inFIG. 5Aand by the 256-QAM asymmetrical symbol constellation500shown inFIG. 5B.

FIG. 5Ais a schematic diagram illustrating an exemplary 256-QAM asymmetrical symbol constellation400of this invention suitable for use in QAM communication system300(FIG. 4). In this “asymmetric” constellation, the I-space values are represented as sixteen amplitudes ranging from −2.5 units to +12.5 units spaced along the I-axis402and the Q-space values are represented as the same sixteen amplitudes ranging from −7.5 units to +7.5 units spaced along the Q-axis404. The I-space values are equally spaced by 1.0 units as exemplified by the spacing406and the Q-space values are equally spaced by the same amount as exemplified by the spacing408. I-axis402and Q-axis404cross orthogonally at the phase plane origin410. Each symbol state is represented as a finite region about a point (I, Q), as exemplified by the symbol state412, which represents eight bits of data; four bits encoded in each of the sixteen amplitude values reserved for I and four bits encoded in each of the sixteen amplitude values reserved for Q. Constellation400is disposed with the I-space and Q-space ranges asymmetrically about phase plane origin410such that any I-space baseband modulating signal I(t) and Q-space baseband modulating signal Q(t) together representing a random time-series of (I, Q) symbols will have mean amplitudes (DC components) that are substantially non-zero (+5.0 units in I-space for constellation400) for I-space baseband modulating signal I(t) and substantially zero for Q-space baseband modulating signal Q(t).FIG. 5Bis a schematic diagram illustrating an alternative 256-QAM asymmetrical symbol constellation500suitable for use in QAM communication system300(FIG. 4). In this “asymmetric” constellation, the I-space values are represented as sixteen amplitudes ranging from +0.5 units to +15.5 units spaced along the I-axis502and the Q-space values are represented as the same sixteen amplitudes ranging from −7.5 units to +7.5 units spaced along the Q-axis504. The I-space values are equally spaced by 1.0 units as exemplified by the spacing506and the Q-space values are equally spaced by the same amount as exemplified by the spacing508. I-axis502and Q-axis504cross orthogonally at the phase plane origin510. Each symbol state is represented as a finite region about a point (I, Q), as exemplified by the symbol state512, which represents eight bits of data; four bits encoded in each of the sixteen amplitude values reserved for I and four bits encoded in each of the sixteen amplitude values reserved for Q. Constellation500is disposed with the I-space and Q-space ranges asymmetrically about phase plane origin510such that any I-space baseband modulating signal I(t) and Q-space baseband modulating signal Q(t) together representing a random time-series of (I, Q) symbols will have mean amplitudes (DC components) that are substantially non-zero (+8.0 units in I-space for constellation500) for I-space baseband modulating signal I(t) and substantially zero for Q-space baseband modulating signal Q(t). The non-zero DC bias of at least one of the two baseband modulating signals is an important element of the system of this invention and either or both of the two baseband modulating signals may be biased to create a suitable asymmetric constellation in accordance with these teachings.

Returning toFIG. 4, a constellation bias signal321is shown as an input to QAM encoder316to illustrate the method for shifting the symbol constellation exemplified by constellation400, which may be thought of as adding a DC bias to either or both baseband modulating signals, IT(t)318and QT(t)320during the encoding process in QAM encoder316, for example. This facilitates using constellation bias signal321to adjust the asymmetric constellation exemplified by constellation400in response to channel type and conditions or for other purposes, for example.

The two baseband modulating signals, IT(t)318and QT(t)320, are accepted by a QAM modulator322that includes an I-modulator324and a Q-modulator326embodied as a dual multiplier. I-modulator324modulates a zero-degree-phase baseband symbol clock signal328from the baseband symbol clock oscillator329by multiplying it with baseband modulating signal IT(t)318to produce an I-modulation product signal330and Q-modulator326modulates a ninety-degree-phase baseband symbol clock signal332from baseband symbol clock oscillator329by multiplying it with baseband modulating signal QT(t)320to produce a Q-modulation product signal334. A summer336then adds I-modulation product signal330and Q-modulation product signal334in the usual manner to produce a digital transmitter output signal337, which is then converted to an analog transmitter output signal338by the digital-to-analog converter339. Transmitter output signal338is filtered and conditioned at the filter and driver assembly340to produce a filtered transmitter output signal342that is conditioned for transfer through the physical transmission medium in signal channel306to QAM receiver304. Zero-degree-phase and ninety-degree-phase baseband symbol clock signals328and332are said to be generally fixed in quadrature because they are phase-locked to one another with a 90-degree phase difference in the usual manner. Signal channel306may include conductive wiring, optical fiber, modulated radio frequency or optical signals in free space, or any other useful channel means known in the art, for example. Filter and driver assembly340may include an additional modulator(s) (not shown) for reconditioning transmitter output signal338as a modulation product of another carrier signal more suited to the signal channel medium, for example.

Continuing withFIG. 4, a signal conditioner344in QAM data receiver304accepts from signal channel306a channel signal346that represents filtered transmitter output signal342in some manner, depending on particular channel characteristics, added noise, and the like. Signal conditioner344may include an additional demodulator(s) (not shown) for recovering the baseband component of channel signal346when using another carrier signal more suited to the signal channel medium, for example. Signal conditioner344restores the signal level and provides any additional (usually analog) reconditioning necessary to produce a baseband receiver input signal348, which may now be converted back to a digital receiver input signal349by means of an analog-to-digital converter351, which may be synchronized with baseband symbol clock detector350substantially as shown. In this embodiment, many of the receiver functions in QAM data receiver304are embodied as software (or firmware) programs in a Digital Signal Processor (DSP)359with programming adapted to accept the digital receiver input signal349for decoding and processing. In addition to the functional elements shownFIG. 4, DSP359may also embrace portions of baseband symbol clock detector350. Most remaining complexity in QAM data receiver304is found in signal conditioner344and the remainder of baseband symbol clock detector350. But baseband symbol clock detector350may now be implemented as a “simple” Phase Locked Loop (PLL) circuit, for example, because of the asymmetrical symbol constellation400used in QAM transmitter302, for the reasons discussed herein below in connection withFIGS. 15-16. Notice that using DSP359in QAM data receiver304and the simple PLL implementation of provides a simple and cost effective embodiment of the receiving element of this invention, thereby meeting the primary purpose of the system of this invention. This allows QAM techniques to be applied in a much more cost effective manner than previously known, making QAM feasible for applications for which it was previously cost prohibitive.

Continuing with the remainder ofFIG. 4, from signal conditioner344, receiver input signal348takes two paths; the first taking it to a baseband symbol clock detector350for baseband symbol clock timing recovery and the second taking it to analog-to-digital converter351for digitization to produce digital receiver input signal349, which is presented to an equalization and correction circuit352for any additional processing necessary to correct for noise, intersymbol interference (ISI) and other unwanted effects of the trip through signal channel306. Baseband symbol clock detector350includes a baseband symbol clock recovery oscillator354that produces a zero-degree-phase recovered baseband symbol clock signal356and a ninety-degree-phase recovered baseband symbol clock signal358, which are generally fixed in quadrature and respectively synchronized with baseband symbol clock signals328and332above.

Equalization and correction circuit352produces a baseband receiver input signal360that (as much as possible) represents the recovery of transmitter output signal338. The baseband receiver input signal360from equalization and correction circuit352is routed to the QAM demodulator362for recovery of the two baseband demodulated signals, IR(t)364and QR(t)366, together representing a time series of complex receiver symbols (IR, QR) that (as much as possible) represent the recovery of the initial time-series of complex transmitter symbols (IT, QT) discussed above. This is accomplished by an I-demodulator368and a Q-demodulator370embodied as a dual multiplier. I-demodulator368demodulates baseband receiver input signal360by multiplying it with zero-degree-phase recovered baseband symbol clock signal356to produce an I-demodulation product signal372and Q-demodulator370demodulates baseband receiver input signal360by multiplying it with ninety-degree-phase recovered baseband symbol clock signal358to produce a Q-demodulation product signal374. I-demodulation product signal372is passed through a first low-pass filter376to recover baseband demodulated signal IR(t)364and Q-demodulation product signal374is passed through a second low-pass filter378to recover baseband demodulated signal QR(t)366in the usual manner. From QAM demodulator362, both baseband demodulated signals, IR(t)364and QR(t)366are presented to the QAM decoder380for reversal of the 2-D constellation mapping process performed in QAM encoder316and discussed above to produce the recovered randomized data382. Finally, in the decoder384, the randomizing, interleaving, error-correction and other high-level encoding processing performed in encoder312and discussed above is reversed to produce a stream of output data386corrected for errors where possible and timed according to a bit rate clock signal388from baseband symbol clock detector350. A feedback line390to equalization and correction circuit352permits recovery optimization by adjusting the conditioning of receiver input signal348to minimize errors detected and corrected in recovered randomized data382by decoder384, for example.

FIG. 6is a flowchart illustrating an exemplary method600of this invention for transferring data through signal channel306. Method600begins at the step602by first selecting a two-dimensional (2-D) constellation of symbols distributed on the phase plane asymmetrically about the origin, such as constellation400or constellation500discussed above in connection withFIGS. 5A-B, for example. Next, at the step604, the incoming data are encoded as complex symbols (I, Q) selected from the 2-D constellation, and, in the step606, first and second baseband modulating signals I(t) and Q(t) are produced, whose amplitudes together represent the time series of complex symbols (I, Q) and at least one of the baseband modulating signals has a substantially non-zero mean amplitude. Then, in the step608, the first baseband modulating signal I(t) is multiplied by an in-phase baseband symbol clock signal to produce a first modulation product signal as, in the step610, the second baseband modulating signal Q(t) is multiplied by a quadrature baseband symbol clock signal to produce a second modulation product signal. In the step612, the first and second modulation product signals are summed to produce a transmitter output signal, which is coupled through the signal channel to the data receiver in the step614. Finally, in the step616, the two modulation product signals are demodulated at the data receiver to recover the series of complex symbols (I, Q), thereby facilitating recovery of the incoming data (not shown).

Improving QAM Bit Error Rate (BER) Performance:

The Type III (square) 2-D symbol constellation known in the art and exemplified by constellation100(FIG. 1), is disposed so that the modulating signal amplitudes are symmetrical around zero (phase plane origin110), as are all other 2-D QAM symbol constellations of any type. This is a well-known QAM system requirement arising from the universal and well-founded belief that QAM communication system BER performance is diminished when any power is “wasted” in a carrier (baseband symbol clock) signal. As is known in the art, adding sufficiently exotic (and expensive) timing recovery means to the QAM receiver can overcome much of the timing recovery problem arising from the complete suppression of the carrier (baseband symbol clock) signal and thereby avoid most of the BER performance penalty arising from baseband symbol clock recovery error. This situation, and the unexpectedly advantageous observation leading to the method of this invention, may be better appreciated with reference to the following discussion of the effects of various system abnormalities on theoretical QAM system BER.

FIG. 7provides a graph700illustrating the theoretical BER under various operating conditions for several 256-QAM communications system embodiments from the prior art. The BER curve702provides the predicted BER of the ideal theoretical QAM modulator and demodulator embodiment810shown inFIG. 8A. As shown inFIG. 8A, embodiment810includes a QAM modulator812coupled to a QAM demodulator814through an ideal signal channel816. No actual channel or baseband symbol clock apparatus is shown because theoretically ideal demodulation is assumed for the purposes of predicting BER curve702.

InFIG. 7, the BER curve704provides the predicted BER of the theoretical QAM modulator and demodulator embodiment830shown inFIG. 8B. As shown inFIG. 8B, embodiment830includes a QAM modulator832coupled to a QAM demodulator834through an ideal signal channel836. The original baseband symbol clock signal838is assumed to be available to QAM demodulator834with neither distortion nor delay other than the addition of Additive White Gaussian Noise (AWGN). The 3 dB reduction in performance in BER curve704compared to the ideal baseline BER curve702is understandable because the four samples per symbol clock cycle assumed for these predictions implies a loss of information otherwise available by integrating out the effects of AWGN.

Returning toFIG. 7, the BER curve706provides the predicted BER of the QAM modulator and demodulator embodiment850shown inFIG. 8C. As shown inFIG. 8C, embodiment850includes a QAM modulator852coupled to a QAM demodulator854through a cable and preamplifier signal channel856. The baseband symbol clock timing is recovered at QAM demodulator854by means of a simple PLL858. The performance shown by BER curve706is dismal because the complete suppression of the baseband symbol clock signal from QAM modulator852makes the reliance on a “simple” PLL858for clock recovery an unrealistic solution to the clock recovery problem.

InFIG. 7, the BER curve708provides the predicted BER of the QAM modulator and demodulator embodiment870shown inFIG. 8D. As shown inFIG. 8D, embodiment870includes a QAM modulator872coupled to a QAM demodulator874through a cable and preamplifier signal channel876. The baseband symbol clock timing is recovered at QAM demodulator874by means of a complex “exotic” clock recovery means878. By using any sufficiently sophisticated baseband symbol clock timing recovery mechanism known in the art for the exotic recovery means878, BER curve708provides a performance that is no worse than BER curve704at higher BER values and no more than 5-6 dB worse at lower BER values. This variation between BER curves704and708is related to timing and equalization error degradation and is accepted in the art as a performance sacrifice made to avoid the undesirable performance reduction from “carrier leakage” in QAM systems (FIG. 9).

The effects on BER of an asymmetric QAM constellation may be appreciated with reference toFIG. 9.FIG. 9provides a graph900illustrating the theoretical BER under various operating conditions for two 256-QAM communications system embodiments using the exemplary asymmetric symbol constellations400and500discussed above (FIGS. 5A-5B). The BER curve902provides the predicted BER of QAM modulator and demodulator embodiment830discussed above (FIG. 8B) and is identical to BER curve704inFIG. 7. The BER curves904and906provide the predicted BER of the QAM modulator and demodulator embodiment1010shown inFIG. 10Aunder two different conditions. As shown inFIG. 10A, embodiment1010includes a QAM modulator1012coupled to a QAM demodulator1014through an ideal signal channel1016. The original baseband symbol clock signal1018is assumed to be provided to QAM demodulator1014with neither distortion nor delay for the purposes of predicting BER curves904and906. A baseband symbol constellation offset1020is provided to move the 2-D baseband symbol constellation (not shown) with respect to one of the phase plane axes and thereby insert a “power wasting” baseband symbol clock signal in accordance with the method and system of this invention. For BER curve904, offset1020is set to +5.0 units to create symbol constellation400(FIG. 5A) and, for BER curve906, offset1020is set to +8.0 units to create symbol constellation500(FIG. 5B).

InFIG. 9, note that offsetting the symbol amplitudes by 5.0 units along the I-axis of the phase plane (FIG. 5A) provides the BER curve904, which shows a BER performance reduction of 2-3 dB with respect to BER curve902. Offsetting the symbol amplitudes by another 3.0 units along the I-axis of the phase plane (FIG. 5B) provides the BER curve906, which shows a BER performance reduction of an additional 2-3 dB with respect to BER curve904. This “power-wasting” penalty is the well-known reason why (until now) all 2-D symbol constellations are forced into symmetry about the phase plane origin. Also, this BER performance loss is consistent with the relative root mean square (RMS) powers contained in the respective time-domain waveforms, as may be appreciated with reference to the following discussion ofFIGS. 11-12.

FIG. 11is a graph illustrating the baseband transmitter output signal1100in the time domain from 256-QAM system embodiment200(FIG. 3) using symmetrical symbol constellation100(FIG. 1).FIG. 12is a graph illustrating the baseband transmitter output signal1200in the time domain from 256-QAM system embodiment300(FIG. 4) using asymmetrical symbol constellation400(FIG. 5A). Note that some additional (“wasted”) RMS power is clearly evident in baseband transmitter output signal1200when compared with baseband transmitter output signal1100.

But examining these same two baseband transmitter output signals1100and1200in the frequency domain provides additional useful insight into the baseband symbol clock recovery problem and the method of this invention.FIG. 13provides a baseband transmitter output spectrum1300illustrating baseband transmitter output signal1100(FIG. 11) in the spectral domain andFIG. 14provides a baseband transmitter output spectrum1400illustrating baseband transmitter output signal1200(FIG. 12) in the spectral domain. Even though system performance is degraded by 2 dB because of the 2 dB increase in RMS power in baseband transmitter output signal1200over the RMS power in baseband transmitter output signal1100, the power at the baseband symbol clock frequency1402in baseband transmitter output spectrum1400now rises above the remainder of the spectrum by about 18 dB compared to the power at the baseband symbol clock frequency1302in baseband transmitter output spectrum1300. This is more than adequate to facilitate a very simple means for symbol clock timing recovery in the manner now discussed. Note that the two baseband transmitter output spectra1300and1400are substantially identical except for the 18 dB spike at the baseband symbol clock frequency1402(FIG. 14).

FIG. 15is a graph1500illustrating the theoretical BER under various operating conditions for 256-QAM communications system embodiments of this invention using asymmetric symbol constellation400(FIG. 5A). The BER curve1502provides the predicted BER of QAM modulator and demodulator embodiment1010with offset1020set to +5.0 units and is identical to curve904fromFIG. 9.

InFIG. 15, the BER curve1504provides the predicted BER of the QAM modulator and demodulator embodiment1030shown inFIG. 10B. As shown inFIG. 10B, embodiment1030includes a QAM modulator1032coupled to a QAM demodulator1034through a cable and preamplifier signal channel1036. The original baseband symbol clock signal1038is assumed to be provided to QAM demodulator1034with delay only and no distortion. The performance of BER curve1504is not significantly different from BER curve1502because the delayed but otherwise unaffected baseband symbol clock signal is also available at QAM demodulator1034.

InFIG. 15, the BER curve1506provides the predicted BER of the QAM modulator and demodulator embodiment1050shown inFIG. 10Cand is generally indistinguishable from BER curve1502because of the advantageous effects of the asymmetric symbol constellation400(FIG. 5A) used in accordance with the method of this invention. As shown inFIG. 10C, embodiment1050includes a QAM modulator1052coupled to a QAM demodulator1054through a cable and preamplifier signal channel1056. The baseband symbol clock timing is recovered at QAM demodulator1054by means of a simple PLL1058. Curve1504BER performance is not significantly different from curve1502because the 18 dB spike at the baseband symbol clock frequency1402(FIG. 14) permits the reliance on a “simple” PLL1058for effective clock recovery, for the first time.

Note that the advantages of the method of this invention may be appreciated by comparing BER curve706(FIG. 7) to BER curve1506(FIG. 15). Although both examples use simple PLL baseband symbol clock recovery, the performance of BER curve1506demonstrates that there is no additional timing recovery penalty. Timing can be recovered without appreciable performance loss using the simple and inexpensive recovery means exemplified by PLL1058(FIG. 10C).

And there are additional benefits as well, including the availability of the large single frequency spike at the baseband symbol clock frequency1402(FIG. 14) for predicting abnormalities in signal channel306(FIG. 4). Referring toFIG. 4, this channel prediction capability facilitates the simplification of signal conditioner344, which represents the only remaining element of QAM communications system300having any significant complexity or expense. Recall that the remainder of baseband symbol clock detector350and all other remaining complexity in QAM data receiver304are embodied within the simple and inexpensive DSP359.

This asymmetric symbol constellation technique differs significantly from and avoids several disadvantages (e.g., increased signal envelope fluctuation and spectral spreading) of a concept for inserting a separate tone in the transmitted signal to facilitate measurement of signal channel characteristics that is sometimes denominated Transparent-Tone-In-Band (TTIB) modulation. The TTIB concept neither considers nor suggests using a simple offset signal to shift the baseband symbol clock constellation about the phase plane as described above. TTIB requires the creation of a separate tone and insertion into the channel in the communications band. The separate tone must then be removed somehow from the received signal before attempting demodulation and decoding. This adds complexity and expense to the communications system rather than reducing complexity. The TTIB modulation may be characterized as offsetting the baseband symbol clock signal in time instead of offsetting the baseband symbol constellation in amplitude on the phase plane and results in generating overlapping sidebands, thereby altering the frequency spectrum and bandwidth of the transmitted signal. This introduces additional well-known problems that may be appreciated with reference to, for example, McGeehan et al. [“Phase-Locked Transparent Tone In Band (TIIB): A new spectrum configuration particularly suited to the transmission of data over SSB mobile radio networks,”IEEE Transactions on Communications, vol COM32, 1984] and Hanzo et al. [“Quadrature Amplitude Modulation,” Second Edition,IEEE Press,2004, John Wiley].

Finally, the utility and advantage of the method of this invention may be best appreciated with reference toFIG. 16, which is a graph1600comparing BER curves704and708fromFIG. 15to BER curves1502and1506fromFIG. 7. Recall that BER curve704provides the system BER performance assuming perfect recovery of the original baseband symbol clock signal838at QAM demodulator834(FIG. 8B). And BER curve708provides just about the best system BER performance known in the QAM art for a real signal channel and is obtained only by using any sophisticated baseband symbol clock timing recovery mechanism known in the art for exotic recovery means878(FIG. 8D). Note that, compared to BER curve708, BER curves1502and1506both show superior BER performance below the BER value represented by a crossover point1602(about 2E-04 to 3E-04 in this example) and falls only 1-2 dB behind BER curve708at the BER values well above crossover point1602. In other words, the method and system of this invention improves BER performance over the QAM prior art in any application operating beyond crossover point1602(SNR=about 26 dB in this example) and does this with substantially less complexity and expense.

By offsetting the 2-D baseband symbol constellation with respect to the phase plane origin, symbol clock leakage is inserted into the transmitted QAM signal. While this slightly degrades static BER performance alone, this discussion discloses for the first time that the asymmetrical constellation actually improves overall system performance when considering baseband symbol clock recovery and received signal compensation for an imperfect signal channel. This improvement, for the first time, allows QAM to be deployed in systems where QAM is otherwise prohibitively expensive. This improvement, for the first time, also allows overall system per-tem performance to be improved for any existing QAM system without additional bandwidth, cost or complexity.

A Pipe Inspection System Embodiment

Advantageously, the QAM system and method of this invention may be embodied in a video transmitter to send high definition video signal up a pipe-inspection system cable to a video receiver. This QAM video signal does not interfere with data link and other cable uses in the pipe-inspection system. For example, the QAM video signal does not use bandwidth near 32 kHz or 512 Hz, so it does not suffer from interference from the system's sonde (512 Hz) or tracer frequency (32,768 Hz). This embodiment provides performance superior to a standard NTSC signal, which is degraded by the cable, offers less picture quality, and interferes with sonde and/or tracer operation.

FIG. 17is a perspective diagram illustrating an exemplary pipe inspection system embodiment1701incorporating the data transfer system and method of this invention. Referring toFIG. 17, a pipe inspection system1701includes a camera head1713operatively connected to the distal end of a push-cable1709. The proximal end of the push-cable1709is operatively connected to a cable-counter and user interface panel1705through a slip-ring assembly1707. Examples of suitable constructions for the camera head1713are disclosed in U.S. Pat. No. 6,831,679 entitled “Video Camera Head with Thermal Feedback Control,” granted to Mark S. Olsson et al. on Dec. 14, 2004, and in U.S. patent application Ser. No. 10/858,628 entitled “Self-Leveling Camera Head,” of Mark S. Olsson filed Jun. 1, 2004, the entire disclosures of which are hereby incorporated by reference. Push-cable constructions and termination assemblies suitable for use in connecting the proximal and distal ends of a push-cable are disclosed in U.S. Pat. No. 5,939,679 entitled “Video Push Cable” granted Aug. 17, 1999 to Mark S. Olsson, U.S. Pat. No. 6,958,767 entitled “Video Pipe Inspection System Employing Non-Rotating Cable” granted Oct. 25, 2005, to Mark S. Olsson et al., U.S. patent application Ser. No. 12/371,540 filed Feb. 13, 2009 entitled “Push-Cable for Pipe Inspection System,” and U.S. Patent Application Ser. No. 61/152,947 filed Feb. 16, 2009 by Mark S, Olsson et al. entitled “Pipe Inspection System with Replaceable Cable Storage Drum,” the entire disclosures of which are hereby incorporated by reference. InFIG. 17, a reel1703holds coils of the push-cable1709. The push-cable1709is paid out from reel1703to force camera head1713down pipe1711. Examples of a suitable reel1703and push-cable1709are disclosed in the aforementioned U.S. Pat. No. 6,958,767. Within the reel1703, a slip-ring assembly1707provides rotary signals to an associated circuit board (not shown) which enables them to be translated into digital measurements of distance traversed by the push-cable1709based on the rotation of the drum. One example of a suitable slip ring assembly is disclosed in U.S. Pat. No. 6,908,310 entitled “Slip Ring Assembly with Integral Position Encoder,” granted Jun. 21, 2005, to Mark S. Olsson et al., the entire disclosure of which is hereby incorporated by reference. The camera head1713with its on-board circuitry transmits image information through embedded conductors such as wires in the push-cable1709. A display unit1715shows the updated field of view (FOV) image from the camera head1713with an overlay indicating the distance down-pipe and the direction of travel based on the values transmitted from the slip ring assembly1707. Circuit boards within the user-interface assembly1705provide memory and processing, user information display and input controls.

Turning now toFIG. 18, the electronic portion1800of pipe inspection system1701includes a central processor1802associated with a volatile memory1818, which receives input data1819from a user interface1806, a slip-ring counter1808, a remote video camera1804including a video transmitter1825, which incorporates the elements of QAM data transmitter302substantially as shown inFIG. 4and operating substantially as discussed above. Video transmitter1825providing a video signal1823representing image data passing through a signal channel1827to a video receiver1824, which incorporates the elements of QAM data receiver304substantially as shown inFIG. 4and operating substantially as discussed above. Signal channel1827is embodied as one or more electrical conductors disposed within push-cable1709(FIG. 17). Central processor1802is also associated with camera control circuitry1814, a system graphical user interface (GUI)1826, and a keyboard1820. The central processor1802sends output signals to the camera control1814, volatile memory1818, SD card storage1810, USB portable (thumb drive) storage1812, and the user GUI1826with its associated display1828, which also displays images1829responsive to video signal1823arriving at video receiver1824upon proper user or software command. The transfer of image and other data may be automated through firmware programming or initiated from the GUI1826using on-board key presses, or by means of the keyboard1820. Algorithmic options in the firmware may permit parameters such as distance interval between image captures, for example, to be set to default values in automatic operation or to be set to user selected values using menu options exercised through UI1826or keyboard1820.

FIG. 19is a flowchart illustrating an exemplary method1900of this invention for transferring video signal1823through signal channel1827in electronic portion1800of pipe inspection system1701. Method1900begins at the step1902by first selecting a two-dimensional (2-D) constellation of symbols distributed on the phase plane asymmetrically about the origin, such as constellation400or constellation500discussed above in connection withFIGS. 5A-B, for example. Next, at the step1904, the video signal data are encoded as complex symbols (I, Q) selected from the 2-D constellation, and, in the step1906, first and second baseband modulating signals I(t) and Q(t) are produced, whose amplitudes together represent the time series of complex symbols (I, Q) and at least one of the baseband modulating signals has a substantially non-zero mean amplitude. Then, in the step1908, the first baseband modulating signal I(t) is multiplied by an in-phase baseband symbol clock signal to produce a first modulation product signal as, in the step1910, the second baseband modulating signal Q(t) is multiplied by a quadrature baseband symbol clock signal to produce a second modulation product signal. In the step1912, the first and second modulation product signals are summed to produce a transmitter output signal, which is coupled through the signal channel to the data receiver in the step1914. Finally, in the step1916, the two modulation product signals are demodulated at the data receiver to recover the series of complex symbols (I, Q), thereby facilitating recovery of the video signal data (not shown).