Simplified d.c. to d.c. converter

A d-c to d-c converter which can start with a shorted output and which regulates output voltage and current is disclosed. Voltage controlled switches directed current through the primary of a transformer the secondary of which includes virtual reactance. The switching frequency of the switches is appropriately varied to increase the voltage drop across the virtual reactance in the secondary winding to which there is connected a low impedance load. A starting circuit suitable for voltage switching devices is provided.

DESCRIPTION 
1. Technical Field 
This invention relates to electrical power supply and is directed more 
particularly to circuits for converting direct current at one voltage to 
either a higher or lower voltage. 
Many different types of D.C. to D.C. converters are known. In general, 
state of the art D.C. to D.C. converters incorporate solid state switching 
devices and saturable transformers. Depending upon the application for 
which a D.C. to D.C. converter is to be used, other ancillary circuits are 
incorporated in such converters. Examples of such circuits are active 
voltage limiting circuits, closed-loop current control circuits, output 
short-circuit current limiting circuits, pre-regulation circuits, and 
in-rush current limiting circuits. All of these circuits increase the 
cost, complexity, weight and probability of malfunction. 
It is highly desirable that the foregoing auxiliary or ancillary circuits, 
be eliminated from D.C. to D.C. converters where weight and reliability 
are important considerations. D.C. to D.C. converters used in space 
vehicles would be an example of a use depending on these factors. 
It is contemplated that some future space vehicles will use ion thrusters 
(ion engines) for maneuvering as well as for traveling. The electrical 
power for the ion thrusters will be obtained from solar cell arrays. The 
voltages from the solar cell arrays will be converted to respective 
different voltages, as required by the various circuits in the ion 
thrusters. Ion engines in general require several D.C. to D.C. converters. 
Two of these D.C. to D.C. converters have unique requirements. One powers 
what is known as a neutralizer keeper while the other powers what is known 
as a cathode keeper. 
Both of these D.C. to D.C. converters must meet certain requirements such 
as galvanic isolation of the input and output circuits, a short circuit 
output current which is substantially constant over a 2:1 input voltage 
range and a first-order correction of any increase in output current due 
to increased input voltage. In addition, such converters should have low 
weight, a minimum number of parts and high reliability. 
2. Background Art 
U.S. Pat. Nos. 3,249,894 and 3,297,959, both to R. P. Massey, disclose D.C. 
to D.C. converters employing asymmetric oscillators controlled by feedback 
circuits connected to the output. Output voltage is controlled by varying 
the duty cycle of the oscillator power transistor. Because of the feedback 
circuits, the output is not isolated from the oscillator portion of the 
converter. 
U.S. Pat. No. 3,327,244 to C. W. Fay et al discloses a D.C. to D.C. 
converter wherein the unidirectional output current is directed through a 
winding on the switching transformer of the oscillator to control its 
frequency at a predetermined value. 
U.S. Pat. No. 3,377,540 to K. H. Meyer shows a D.C. to D.C. converter 
having a circuit connected between an output terminal and the oscillator 
section of the converter to reduce the magnitude of the oscillations in 
the oscillator circuit under overload conditions. 
U.S. Pat. No. 3,611,205 to Ogawa discloses a saturable core oscillator 
having an extra winding on the transformer, which winding is connected in 
a full wave bridge rectifying circuit arrangement to a zener diode. This 
arrangement insures a constant frequency output from the oscillator. 
U.S. Pat. No. 2,852,730 to Magnuski shows a D.C. to D.C. converter in which 
the oscillator section comprises a pair of transistor switches controlled 
from the feeback winding carried on a portion of the core of the switching 
transformer which is of reduced cross section and saturates to cause 
switching action. 
U.S. Pat. No. 3,590,362 to Kakalec discloses an inverter circuit which 
drives a ferro resonant regulator and utilizes a 
core-saturation-simulating circuit in the output to vary the oscillator 
frequency such that the ferroresonant circuit will maintain the A.C. 
output voltage at a relatively constant value. 
U.S. Pat. No. 3,777,248 to Vermolen shows a D.C. to D.C. converter having a 
saturable core oscillator and employing a saturable choke coil in one of 
the D.C. output leads to insure that the oscillator will start. Other U.S. 
Pat. Nos. covering D.C. to D.C. converters are: 3,241,032 to Firestone; 
3,586,957 to Case; 3,996,506 to Kichak; and 4,061,957 to Vader. These four 
patents all employ circuitry wherein a portion of the output voltage is 
fed back to the oscillator section to regulate input or output current or 
output voltage. 
DISCLOSURE OF THE INVENTION 
In accordance with the objects of the present invention, there is provided 
a D.C. to D.C. converter comprised of a transformer having a primary 
winding through which current is directed in alternate directions by metal 
oxide semiconductor field effect transistors (MOSFET) connected between 
the primary winding and a D.C. source or battery. A second winding of the 
transformer is connected to a rectifying and filter circuit to provide 
unidirectional output current. The primary and secondary windings of the 
transformer are carried on the respective outer legs of an E-core with the 
center leg of the core providing a leakage reactance. This leakage 
reactance has the same effect as placing an inductor in series with the 
rectifiers in the output circuit. 
The transformer core may be of the saturable type to effect switching of 
the transistors or, alternatively, an additional winding on the 
transformer may be connected to the primary of a small saturable 
transformer whose secondary is connected to the gates of the transistors. 
Starting circuits comprised of resistors, capacitors and diodes connected 
in particular configurations are also utilized.

BEST MODE FOR CARRYING OUT THE INVENTION 
Referring now to FIG. 1, there is shown in accordance with the invention a 
D.C. to D.C. converter comprising an oscillator section 10, a power output 
section 11 and a start section 12. Oscillator section 10 is comprised of 
MOSFETs Q1 and Q2 the drains of which are connected to respective outer 
ends of a center tapped winding 13 which is carried on a core 14 of a 
transformer T1. The sources of MOSFETs Q1 and Q2 are both connected via a 
lead 15 to a negative power input terminal 16. The center of winding 13 is 
connected through a lead 17 to a positive input terminal 18, a battery 19 
being connected between the input terminals 16 and 18. A capacitor C1 is 
connected between leads 15 and 17 to provide a low impedance path for 
alternating currents to prevent voltage spikes that could damage MOSFETs 
Q1 and Q2. 
Also carried on the core 14 of transformer T1 is a center tapped secondary 
winding 20 and a tertiary winding 21. The core 14 comprises a pair of 
E-shaped ferrite cores arranged such that there is a gap between the ends 
of the center legs. A primary winding 13 of transformer T1 is wound on one 
of the outer legs while the secondary winding 20 is wound on the other. 
The tertiary winding 21 is wound on the same outer leg as primary winding 
13. The air gap in the center legs of core 14 provides a leakage reactance 
which effectively places virtual inductors L1 and L2 in series with the 
respective outer ends of center tapped secondary winding 20. FIG. 3 
illustrates the primary winding 20 and the tertiary winding 21 carried on 
a first outer leg of E-core transformer T1 while the secondary winding 13 
is carried on a second outer leg. As shown, the center leg of T1 includes 
an air gap as discussed previously. 
To the end that MOSFETs Q1 and Q2 will operate in a switching mode, their 
gate electrodes are connected through respective resistors R1 and R2 to 
the respective outer ends of a center tapped winding 22 carried with a 
primary winding 23 on the saturable core 24 of a switching transformer 22. 
The primary winding 23 is connected to the tertiary winding 21 of 
transformer T1 with a resistor R4 being disposed in one of the 
connections. With this arrangement, saturation of core 24 will cause the 
conducting MOSFET to lose drive whereupon the polarities on primary 
winding 13 reverse. This is reflected from winding 21 to the primary 
winding 23 of transformer T2 causing the MOSFET which had been 
non-conducting to turn on. Upon the next saturation of core 24, the MOSFET 
will again switch conducting states. The intrinsic anti-parallel diodes 
contained in the MOSFETS Q1 and Q2 carry reverse currents caused by L1 and 
L2. 
Although it is possible that MOSFETs Q1 and Q2 will begin their switching 
action when DC power is applied across input terminals 16 and 18, there is 
no guarantee that switching action will start under all operating 
conditions. To the end that the switching action will indeed occur, 
starting circuit 12 is provided. This circuit includes a resistor R3 
connected from the positive input lead 17 through a lead 25 to the center 
tap of winding 22 of transformer T2. A capacitor C3 and a zener diode CR1 
are connected in parallel between the lead 25 and the negative input lead 
15. 
When power is applied to input terminals 16 and 18, positive voltage is 
applied to the gate electrodes of MOSFETs Q1 and Q2 through resistor R3 to 
respective halves of winding 22 and resistors R1 and R2, respectively. The 
MOSFET with the lowest threshhold voltage will conduct first. A voltage 
reflected to transformer T2 from the tertiary winding 21 of transformer T1 
prevents the other MOSFET from conducting. When the core of transistor T2 
saturates, the MOSFET will switch conducting states. 
Zener diode CR.sub.1 limits the gate co source for wide changes in 
converter input voltages. C.sub.3 provides a low impedance path for 
currents during the switching interval and improves performance slightly. 
To provide the desired direct current output at terminals 26 and 27, the 
center tap of winding 20 is connected via a lead 28 to terminal 27 while 
the outer ends of winding 20 are connected through respective diodes D1 
and D2 to positive terminal 26. A filter capacitor C2 is connected across 
the output terminals 26 and 27 to average the pulsating direct current 
from D.sub.1 and D.sub.2 thereby providing a more constant output current. 
At no load the voltage across the capacitor can increase due to low energy 
switching transients. A zener diode CR2 can be connected in parallel with 
the capacitor C2 to limit the output voltage at no load caused by the 
switching transients. This allows capacitor C2 to have a lower voltage 
rating as wll as providing a safe path for externally induced transient 
currents due to possible ion thruster arcs or short circuits. 
Due to the inductors L1 and L2 of the power output circuit 11, the load 
reflected to the oscillator circuit 10 is inductive. Consequently, 
oscillator 10 starts easily even with a short circuit between output 
terminals 26 and 27. Further, inductors L1 and L2 help to regulate output 
voltage and to limit output current. These advantages are obtained because 
the frequency of operation of oscillator 10 increases linearly with an 
increase in input voltage. Thus, if the frequency of oscillator 10 
increases due to an increase in input voltage, the impedance of inductors 
L1 and L2 increases to absorb some of the voltage increase which would 
otherwise appear between output terminals 26 and 27. 
One particular use of the D.C. to D.C. converter of FIG. 1 is to provide 
electrical power to the resistive heaters of an ion thruster. In ion 
thrusters the electrical power to the heaters can be modulated, being 
supplied in a form of pulses. The circuit of FIG. 1 can provide a pulsed 
D.C. output by providing D.C. power to the input terminals 16 and 18 in 
the form of pulses produced by mechanical or transistor switches. Pulsed 
D.C. output from the circuit of FIG. 1 can also be more efficiently 
obtained by connecting a commercial available OPTO coupler between the 
negative input lead 15 and the cathodes of a pair of clamping diodes, the 
anodes of which are connected to respective gates of MOSFETs Q1 and Q2. 
The on-off control signal is applied to a light emitting diode in the OPTO 
coupler to provide the desired pulsed output at terminals 26 and 27. 
Referring now to FIG. 2, there is shown a D.C. to D.C. converter which is 
similar to that of FIG. 1 and includes an oscillator section 10, a power 
output section 11 and a start circuit section 12. Where components in FIG. 
2 correspond to those in FIG. 1, like numerals are used for 
identification. 
The output circuit 11 of FIG. 2 is identical to that of FIG. 1. However, 
oscillator section 10 utilizes MOSFETs Q3 through Q6 in a bridge connected 
arrangement wherein a primary winding 30 is carried on one outer leg of 
core 31 along with a start circuit winding 32 and gate drive windings 33 
and 34 for MOSFETS Q3 and Q4. The secondary winding 20 is carried on the 
other outer leg of core 31. 
As shown in FIG. 2, MOSFETs Q3 and Q5 are serially connected between 
negative lead 15 and positive lead 17. Similarly, MOSFETs Q4 and Q6 are 
serially connected between negative lead 15 and positive lead 17. The 
primary winding 30 of T3 is connected between the source electrodes of 
MOSFETs Q3 and Q4. 
In order to have MOSFETs Q4 and Q5 conduct while MOSFETs Q3 and Q6 are 
nonconducting, feedback windings 33 and 34 carried on core 31 of 
transformer T3 are connected between the source electrodes and the gate 
electrodes of those transistors through respective resistors R5 and R6, 
while feedback windings 35 and 36 carried on a saturable core 37 of 
transformer T4 are connected with resistors R7 and R8 between the sorce 
electrodes and the gate electrodes of MOSFETs Q5 and Q6, respectively. A 
primary winding 38 of transformer T4 is connected across a winding 39 
carried on core 31 of transformer T3 in circuit with a resistance R9. 
The black dots shown at the end of the windings in FIG. 2 indicate points 
of like voltage polarity at any given instant of time. Accordingly, it 
will be seen that when the gate electrodes of MOSFETs Q4 and Q5 are 
subjected to a positive voltage, the gate electrodes of MOSFETs Q3 and Q6 
are negative. Upon saturation of core 37 of transformer T4, the voltages 
applied to the gate electrodes of MOSFETs Q4 and Q5 will begin to decrease 
causing reduced conduction of these MOSFETs. When this occurs, the 
polarities will reverse causing MOSFETs Q3 and Q6 to begin conducting 
while MOSFETs Q4 and Q5 become nonconducting. 
Like the D.C. to D.C. converter shown in FIG. 1, the frequency of operation 
of the oscillator section 10 is directly proportional to the input 
voltage. Therefore, if the voltage between input terminals 16 and 18 
decreases, the frequency of the oscillator will decrease. As a result, the 
impedance of virtual inductors L1 and L2 decreases, thereby minimizing a 
decrease in the output voltage between terminals 26 and 27. Of course, if 
the input voltage increases, the frequency of oscillator 10 will increase. 
In this case, the impedance of virtual inductors L1 and L2 will increase 
to minimize any increase of output voltage. 
The starting circuit 12 for the converter of FIG. 2 includes a resistor 
R11, a diode D4, a commercial momentary switch SW1 and a winding 32 
carried on core 31 of transformer T3, all of these components being 
connected serially as shown between positive lead 17 and negative lead 15. 
A resistor R10 and a diode D3 serially connected from a junction common to 
the source of MOSFET Q3 and the drain of MOSFET Q5 to a point between 
resistor R11 and diode D4 are also part of starting circuit 12. A zener 
diode CR3 and a capacitor C4 are connected in parallel across the series 
combination of diode D4, switch SW1 and winding 32. The capacitor C4 
charges to the zener voltage to provide energy for the starting pulse. 
To initiate oscillations in oscillator section 10, switch SW1 is 
momentarily closed causing current to flow through winding 32. The current 
flow through winding 32 induces voltage on winding 39 which is applied to 
the primary winding 38 of the saturable transformer T4. This, in turn, 
induces voltage on the feedback windings 35 and 36 to apply the desired 
voltages to the gates of transistors Q5 and Q6. At the same time winding 
32 induces voltage on feedback windings 33 and 34 such that MOSFETs Q3 and 
Q4 are appropriately biased. 
The converter can be started automatically without the need for the manual 
momentary switch SW1. SW1 and D4 are replaced by a diac 9 or four-layer 
diode and CR3 is eliminated from the circuit, as shown in FIG. 4. 
Capacitor C4 then charges up until the firing voltage of the diac or 
four-layer diode is reached. A positive going pulse is applied to winding 
32 on core 31 and the circuit starts as previously described. Resistor R10 
and diode D3 keep C4 from being recharged while the converter is in 
operation. 
Because of the winding 32 and the voltage it induces on the other windings, 
MOSFETs Q4 and Q5 will always conduct first with MOSFETs Q3 and Q6 being 
nonconducting. By reversing the connections to winding 32 or by reversing 
the direction of its winding, MOSFETs Q3 and Q6 could be made to conduct 
initially. This result could also be achieved by reversing the connections 
or directions of windings of the feedback windings 33, 34, 35 and 36. 
It will be understood that changes and modifications may be made to the 
above described circuits by those skilled in the art without departing 
from the spirit and scope of the invention as set forth in the claims 
appended hereto.