Ultrasonic power supply

An ultrasonic power supply for driving a piezoelectric transducer at its parallel resonant frequency includes a clamped-mode resonant converter for converting direct current to alternating current and a demodulator, a loop filter and a voltage controlled oscillator in a phase locked loop configuration. The power supply also includes a control circuit for adjustably setting and controlling the amplitude of vibration of the transducer, and providing during start-up a ramp start. The power supply, further, includes a control circuit for limiting the maximum permissible current flow to the transducer and the reverse current flow from the transducer resulting from stored mechanical energy. Both control circuits provide feedback signals which control the pulse width modulated operation of the clamped-mode resonant converter and thereby control the voltage amplitude of the alternating current output from the converter. The phase locked loop circuit causes the frequency of the alternating current output from the converter to track the parallel resonant frequency of the transducer.

BACKGROUND OF THE INVENTION 
This invention concerns an electronic power supply for driving an 
electroacoustic transducer at its resonant ultrasonic frequency, typically 
a predetermined frequency in the range between 18 kHz and 60 kHz. More 
specifically, this invention refers to a power supply for driving a 
piezoelectric transducer provided with a horn at its parallel resonant 
frequency, such transducer and horn combination being used most frequently 
for welding thermoplastic parts, requiring power from several hundred 
watts to several kilowatts for time intervals ranging from several 
milliseconds to a few seconds. A typical prior art power supply of this 
type is shown in U.S. Pat. No. 3,432,691 issued to A. Shoh, dated Mar. 11, 
1969, entitled "Oscillatory Circuit for Electroacoustic Converter". The 
power supply disclosed hereafter incorporates improvements and novel 
features not present in the prior art supply, such features being 
necessitated by the requirement that ultrasonic welding apparatus be 
operable by computer control at high speed and under conditions of high 
precision and repeatability involving varying workpieces and production 
runs. 
SUMMARY OF FEATURES OF THE INVENTION 
The present invention discloses a power supply for driving a piezoelectric 
transducer provided with a horn at its parallel resonant frequency, 
typically 20 kHz, wherein the amplitude of mechanical vibrations manifest 
at the output surface of the horn can be adjusted and retained constant at 
the adjusted value, wherein the starting sequence, that is bringing the 
transducer with horn from standing still to its full vibrational 
amplitude, is effected in a fast and stepless manner, wherein the resonant 
frequency of the transducer and horn combination is tracked and used as a 
control signal to adjust the frequency of the power supply, wherein the 
flow of current to and from the transducer is limited to preset values, 
and, finally, including means for accomodating higher rates of operation 
(weld cycles) than has been possible with the prior art devices. 
The following description will provide an overview of the novel features 
indicated above. 
AMPLITUDE ADJUSTMENT 
Ultrasonic transducers for power applications normally are operated in 
their parallel resonant mode. By the use of a compensation inductor, the 
power supply will act as a voltage source. With suitable tuning, the 
amplitude of mechanical vibration manifest at the transducer output 
surface is then proportional to the drive voltage (motional voltage) and 
the value of the current is proportional to the power demanded by the 
workpiece. In this manner, the amplitude of vibration can more easily be 
regulated or adjusted despite wide variations in power requirements. 
Most known power supply designs utilize a two part circuit to accomplish 
vibration amplitude adjustment. (This is apart from the use of mechanical 
booster horns or horns of different mechanical gain coupled to the 
transducer assembly for amplitude adjustment). The major component of the 
power supply is a converter circuit employing electrical switching devices 
for converting direct current voltage to an alternating current voltage 
whose frequency is dictated by the mechanical resonance of the ultrasonic 
transducer. The amplitude of the alternating current voltage is governed 
by the value of the direct current voltage supplied to the switching 
devices and hence, the amplitude of the vibration is a function of the 
direct current voltage amplitude. For changing the amplitude of vibration, 
the value of the direct current voltage must be changed. The direct 
current voltage is derived by rectifying and filtering the alternating 
current line voltage. To make the alternating current voltage variable, 
either a variable autotransformer or a switching regulator has been used. 
This technique is inefficient and expensive as the power is processed 
twice and usually the response to a change in amplitude adjustment is 
slow. 
In contrast, the power supply disclosed hereafter uses a direct current 
power source furnishing a constant amplitude voltage and a converter 
operating in a pulse width modulation mode for providing an alternating 
current voltage of suitable frequency for driving the transducer. The 
alternating current line voltage is rectified, filtered, and retained at a 
fixed value. The converter then generates the operating frequency voltage. 
The operating voltage is electronically controlled using the same 
switching devices of the converter. The result is a low cost, light weight 
and very fast response circuit. The circuit allows the amplitude of 
vibration to be adjusted during the weld process rather than being fixed 
at a preset value. 
MECHANICAL AMPLITUDE REGULATION 
By using pulse width modulation in the direct current to alternating 
current converter, the amplitude of vibration of the transducer and horn 
can easily be electronically controlled. A compensation derived feedback 
circuit is used also in the output network of the power supply. This 
circuit provides a signal which is commensurate with the motional voltage 
or motional amplitude manifest at the transducer output surface. The 
signal is electronically processed and fed back to the pulse width 
modulation circuit. This technique allows the vibrational amplitude to be 
regulated with respect to line voltage variations and load variations. 
Hence, a control voltage signal can be used to set or vary the vibrational 
amplitude of the transducer with little or no influence from line voltage 
or load fluctuations. The fast response allows in-process control of the 
amplitude of vibrations. 
STARTING SEQUENCE 
The pulse width modulation technique described above is used also to start 
the ultrasonic transducer vibrations from rest. A ramp voltage is used to 
increase the motional voltage in a linear manner from zero to a regulated 
value. This is an improvement over the step-start method used previously 
as shown in U.S. Pat. No. 3,469,211 dated Sept. 23, 1969, issued to A. 
Shoh et al, entitled "Oscillatory Circuit for Electro-Acoustic Converter 
with Starting Means". The control is continuous, linear and is self 
adapting. 
Different transducer assemblies require differing time periods to attain a 
set amplitude of vibration because of inertia and stored mechanical 
energy. In the present arrangement, a circuit is used to monitor the 
available current which the power supply can deliver during the start-up 
cycle. This signal is also fed back to the pulse width modulation circuit 
and is used to modify the ramp signal. With this technique, the power 
supply will self-adjust the output power provided to the transducer and 
bring the transducer to the set vibrational amplitude in the shortest 
practical time. 
FREQUENCY TRACKING 
The natural operating frequency (resonance) of an ultrasonic transducer 
will vary somewhat with the operating conditions. Among these conditions 
are mechanical wear of the horn assembly, temperature and mechanical 
loading. Also, there is an inherent variance between individual transducer 
assemblies. In the present invention, a phase-locked loop circuit is used 
for sensing the fundamental frequency current and voltage phase 
relationship in the power circuit and the operating frequency is adjusted 
as a function of the resonant frequency of the transducer at which 
frequency the phase shift is zero. This arrangement results in the maximum 
forward power transfer and operating efficiency with the least amount of 
stress manifest on the switching components. This circuit is active during 
the entire weld cycle, both during start-up and during the power transfer 
interval. 
RATE OF OPERATION 
Most prior art power supplies operate on a pulsed time basis. That is, the 
power supply and transducer intially are at rest. A weld command starts 
the power supply, causing it to deliver power to the transducer and a 
workpiece in contact therewith for a period of time, and thereafter the 
power supply and transducer return to the rest condition. The rate at 
which this sequence can occur is limited by several factors, one of which 
is the power dissipated by the system, and another one is the reaction 
time of the circuits and of the transducer assembly. Currently available 
power supplies are limited to about one hundred operations per minute. The 
power dissipated in the start-up sequence becomes a limiting factor as 
well as the time response of the control circuit. 
By the use of a ramp start and a switch mode control of the power circuit, 
power dissipation is kept to a minimum. The aforementioned control circuit 
is designed to operate at a faster rate and the direct current to 
alternating current converter using pulse width modulation provides for 
much better control of the forward and reverse power to the transducer, 
reverse power being the power generated by the transducer as a result of 
stored mechanical energy when the power to the transducer is rapidly 
decreased. As a result, a power supply, in accordance with the 
improvements indicated hereinabove, can operate at two hundred operations 
per minute, an important improvement needed for high speed, computer 
controlled production runs.

DETAILED DESCRIPTION 
Referring now to the figures and FIG. 1 in particular, a simplified block 
diagram is shown for explaining, in a general manner, the architecture of 
the new and improved ultrasonic power supply. A direct current power 
supply, numeral 10, using conventional means, provides rectified and 
filtered direct current power via conductors 12 and 14 to a direct current 
to altenating current converter unit 100. The converter unit 100 comprises 
semiconductor switches for converting the direct current voltage from the 
power supply 10 to an ultrasonic frequency, typically 20 kHz, a common 
frequency used for operating high power ultrasonic welding apparatus. The 
switching devices in the converter unit 100 are operated in a switch mode 
manner (non-linear) to provide both the frequency generation (20 kHz) and 
control of the output voltage using a pulse width modulation technique. 
The output network 200 via conductors 102 and 104 receives the ultrasonic 
frequency output voltage from the converter unit 100 and provides via 
conductor 16 the driving voltage and current to the ultrasonic transducer 
assembly 18. The output network 200 transforms and matches the output 
impedance of the converter unit to the impedance of the transducer 
assembly. The output network 200 comprises electrical components forming a 
resonant circuit in conjunction with the transducer assembly 18. The 
output network also provides input signals to certain control circuits. 
The transducer assembly 18 comprises a stack of piezoelectric discs clamped 
between metal masses, and a horn coupled thereto for coupling the 
vibrations produced by the piezoelectric discs, responsive to applied 
electrical energy, to a workpiece to be welded. The transducer assembly is 
of conventional construction and is well known in the art. 
The voltage controlled oscillator 300 is the main frequency and timing 
generator for the entire power supply and its control circuits. It 
comprises a voltage controlled oscillator which operates at a harmonic 
frequency of the fundamental ultrasonic frequency, 20 kHz in the present 
embodiment, and a digital frequency divider to derive system reference 
signals. 
A modulator and driver circuit 400 receives three input signals from the 
voltage controlled oscillator 300, designated by numerals 302, 304 and 306 
corresponding to two times the fundamental frequency (2f.sub.p), two times 
the fundamental frequency 180 degrees phase shifted (2f.sub.p), and the 
fundamental frequency (f.sub.p). The modulator and driver circuit includes 
linear and digital circuits for generating signals along conductors 402, 
404, 406 and 408 for controlling the operation of the direct current to 
alternating current converter 100. A bi-phase signal is generated by the 
modulator which provides for the pulse width modulation of the converter 
unit 100. The frequency of operation is responsive to the input from the 
voltage controlled oscillator 300 and is controlled in time by an analog 
input signal from the voltage control circuit 500 and the current control 
circuit 600. The output signals from the modulator are amplified by drive 
stages and then used for controlling the switching devices of the 
converter unit 100. 
The current filter circuit 700 is an electronic filter used for obtaining 
the fundamental current signal (f.sub.p). An input signal responsive to 
the operating current is derived from the output network 200 via 
conductors 202 and 204. The input signal contains the fundamental current 
frequency (f.sub.p) and the odd harmonics of the fundamental frequency. 
The filter is unique in that it is a bandpass filter for a range of 
frequencies around the fundamental frequency, but attenuates the harmonic 
frequencies. At the same time, the filter is substantially phase 
transparent for the fundamental frequency within the bandpass range, that 
is, substantially no phase distortion occurs for the signal passing 
through the filter. 
The current demodulator 800 is a synchronous type analog switch or ring 
demodulator. The filtered signal from the current filter 700 along 
conductor 702 is chopped by digital reference signals from the voltage 
controlled oscillator 300, conductors 308, 310. The resultant output 
signals are proportional to the real and the imaginary components 
contained in the original current signal. These signals convey information 
as to the value of and the phase relation of the current components 
relative to the fundamental frequency drive voltage. The real component 
signal is applied as an input signal, conductor 802, to the current 
control circuit 600, while the imaginary or reactive component signal, 
conductor 804, is applied as input signal to the current control circuit 
600 and to a loop filter circuit 900. 
The loop filter 900 is a low pass type filter. The reactive current signal 
from conductor 804 is fed to the input of the loop filter. The output 
signal from the filter, conductor 902, is used as the phase related 
control or feedback voltage for the voltage controlled oscillator 300 for 
adjusting its frequency. In this manner, a phase locked loop is created 
which tends to keep the voltage and current in phase with each other at 
the switches of the direct current to alternating current converter 100. 
This is accomplished by seeking the frequency for which the reactive 
current is at a minimum. Maximum power transfer occurs when minimum stress 
is applied to the switching devices of the converter unit 100. The filter 
is an integral part of the phase locked loop. Its parameters dictate the 
overall rate of frequency compensation and control the stability of the 
loop. 
The combination of circuits, that is a voltage controlled oscillator 300, 
demodulator 800 and loop filter 900, forming what is known as a phase 
locked loop (PLL) has been disclosed broadly in the art heretofore, see 
for instance patent publication DE 2,726,249, published Dec. 14, 1978, 
assigned to Otto Siebeck GmbH, or U.S. Pat. No. 4,642,581, dated Feb. 10, 
1987, issued to J.J. Erickson, entitled "Ultrasonic Transducer Drive 
Circuit". 
The voltage control circuit 500 is used to regulate the overall motional 
voltage supplied to the transducer 18. A signal corresponding to the 
motional voltage is derived in the output network 200. This motional 
voltage signal is applied to the voltage control circuit 500 via conductor 
206, amplified, and compared with a control voltage. The result of this 
comparison is the input signal to the modulator and driver circuit 400 via 
combiner 20, conductors 502 and 22. In this manner, the conduction angle 
of the switching devices in the converter circuit 100 is controlled for 
maintaining a constant motional voltage to the converter. 
During the start cycle, a ramp voltage is generated in the voltage control 
circuit 500 so as to increase the output amplitude of the signal to the 
modulator and driver unit 400 at a controlled linear rate from zero to a 
maximum limit dictated by a voltage control setting for causing the direct 
current voltage from the converter unit 100 to increase also from zero to 
a maximum. 
The current control circuit 600 is used for regulating the maximum amount 
of current which the power supply delivers during the start and the run 
cycles. Both the reactive and the real current components are received as 
input signals from conductors 802 and 804, and combined in a manner to 
protect the power supply in various modes of operation. The circuit 
includes a set of differential amplifiers for limiting respectively the 
forward current and the reverse current to predetermined values. The 
output signals, conductors 602 and 604, also are supplied as a feedback 
signal to the modulator and driver circuit 400 via the combiner 20 for 
controlling the conduction angle of the switching devices in the converter 
100 and, hence, to control the motional voltage to the transducer. During 
the start-up cycle, the circuit may modify the start ramp voltage signal 
as a function of the transducer characteristics. 
Having described now the circuits in broad terms, the following description 
will more closely discuss the individual circuits. 
FIG. 2 is a schematic circuit diagram of the direct current to alternating 
current converter. It comprises essentially a full wave bridge rectifier 
and gate driving networks for controlling the operation of the respective 
rectifier switches. The circuit comprises four semiconductor switching 
devices 106, 108, 110 and 112, each including a power MOSFET device 114 
for switching the power and a Schottky diode 116 connected in series with 
the drain lead for preventing the body diode in the device 114 from 
conducting current in the reverse direction. A high speed diode 118 is 
coupled across the series connection of the MOSFET device 114 and diode 
116 to conduct reverse current appearing at the switching device. 
The switching devices are driven by respective drive stages 120, 122 
responsive to the output signals from the modulator and driver circuit 
400. The resulting alternating current output appears at conductors 102 
and 104. The switching circuit essentially is a modulator wherein the 
output voltage is a function of the pulse width modulation. At any time, 
two switches will be conductive and two switches will be non-conductive. A 
circuit of this configuration termed "Clamped-Mode Resonant Converter" is 
described in detail in the article entitled "Constant-Frequency 
Clamped-Mode Resonant Converters" by F. Tsai et al, IEEE Transactions on 
Power Electronics, volume 3, number 4, Oct. 1988, pp. 460-473, FIG. 2, p. 
462. 
As seen, the switches are separated into two half bridge circuits and each 
half bridge circuit shares a common gate drive network 120, 122. Each gate 
drive network includes circuit components for electrically isolating each 
switch and to provide proper level generation and waveform timing to 
prevent cross-conduction during transitions. The waveforms are generated 
by the modulator and driver circuit 400. 
The advantage of this converter circuit resides in the fact that amplitude 
changes of the output voltage can be made rapidly without disturbing the 
phase relations existing in the phase locked loop which is used to control 
the frequency of operation of the converter circuit and, hence, the 
frequency of the alternating current output. Also, the circuit enables the 
resonant load to be driven at a much greater efficiency. Still further, 
power can be controlled in both directions, to and from the load. Finally, 
energy management is good in that power delivered to the output network 
and transducer during an ON period is continuous. 
FIG. 3 is a simplified schematic circuit diagram of the output network. 
Since output networks, also known as impedance matching networks, are well 
known and have been used heretofore, the circuit will be described only 
briefly. The network 200 receives via conductors 102, 104 the alternating 
current for driving the transducer 18. The output transformer 208 matches 
the voltage and current levels between the converter circuit 100 and the 
transducer assembly 18. The components in series with the primary winding 
of the transformer are selected to cause the primary side together with 
the secondary side to which the transducer 18 is connected to be 
electrically resonant at the parallel resonant frequency of the 
transducer. A current transformer 210 coupled to the primary side of the 
network provides across resistor 212 a signal commensurate with the 
current flowing to the transducer. This signal is both linear and 
substantially phase transparent with the current in the primary side and 
this signal, evident as a voltage across conductors 202 and 204, in turn, 
is used to control the current supplied to the transducer 18 and for 
frequency tuning. 
From the secondary winding of transformer 208 an output signal is 
developed, conductor 206, which signal is commensurate with the motional 
voltage driving the transducer 18, i.e. the voltage proportional to the 
amplitude of vibration. This voltage is fed to the voltage control circuit 
500 and used, in turn, for regulating the motional output amplitude of the 
transducer 18. 
FIG. 4 is a simplified schematic circuit diagram of the current filter 700. 
The filter is an electrical bandpass filter which will attentuate 
frequencies falling outside the band frequencies, but which will pass 
signals within the selected frequency band. One special feature of the 
filter resides in the characteristic that for the frequency range within 
the pass band, the phase shift of a signal from input to output will be 
minimal, i.e. the filter is phase transparent. 
The pulsating output voltage from the direct current to alternating current 
converter is naturally filtered by the output network 200 and transducer 
18. The output signal contains various frequency components, namely the 
fundamental frequency (20 kHz) and odd harmonics of the fundamental 
frequency. This gives rise to similar and related current signals. Of 
prime interest is the fundamental frequency current signal. At resonance, 
the fundamental frequency current and voltage will be in phase with one 
another. Hence, the first object of the filter is to pass the fundamental 
frequency current signal but to attenuate the harmonic signals. 
The current signal commensurate with the current flow between the network 
200 and transducer 18 is supplied via conductors 202, 204 to the current 
filter circuit 700. The circuit, see FIG. 4, comprises two stagger tuned 
parallel resonant circuits. The operating frequency and Q (quality factor) 
of each tank circuit 704, 706 is selected to have equal and opposite phase 
slope within the frequency band of interest. For instance, for a 
fundamental frequency of 20 kHz, tank circuit 704 may be tuned for 19 kHz 
and circuit 706 for 21 kHz, providing a pass range and phase transparency 
for the fundmental frequency of 20 kHz. The signals from the tank circuits 
are then summed in a summing circuit 708 to provide an output signal along 
conductor 702 containing only information with regard to the fundamental 
frequency as harmonic frequencies above or below the selected band width 
were attenuated by the resonant tank circuits. The signal along conductor 
702, therefore, represents a signal corresponding to the amplitude and 
phase of the fundamental frequency of the current flowing between the 
converter 100 and the transducer 18. 
FIG. 5 is a schematic circuit diagram of the demodulator 800. In order to 
maintain the transducer 18 at its resonant operating point, the current 
and voltage output from the converter 100 must be in phase. The current 
signal, filtered in circuit 700, is applied via conductor 702 to the 
demodulator 800 to obtain both the magnitude and phase information of the 
input signal. 
The demodulator comprises a set of synchronous type analog switches. The 
analog output signal 702 from the filter 700 is applied to both analog 
switches 806 and 808, which are commercially available integrated circuit 
devices. Switch 806 also receives a digital signal f'.sub.p via conductor 
308 commensurate with the fundamental operating frequency, but phase 
shifted by ninety degrees. Similarly, switch 808 receives a digital signal 
f.sub.p via conductor 310 commensurate with the fundamental operating 
frequency, but without phase shift. Therefore, the signal provided by 
conductor 804 will represent the imaginary or reactive current component 
flowing to the transducer 18, while the signal provided by conductor 802 
will be commensurate with the real component value of such current. With 
no imaginary current component present in the current flowing to the 
transducer 18, the output signal of conductor 804 will be zero. If an 
imaginary component is present, the output signal along conductor 804 will 
be a plus or minus voltage signal. 
FIG. 6 is a schematic electrical circuit diagram of the loop filter. The 
filter essentially is a low pass filter and is used to process the 
imaginary current component signal from the demodulator 800. The harmonic 
frequency content is blocked and an average direct current error voltage 
is obtained and sent as a correction (control) signal to the voltage 
controlled oscillator input for adjusting the frequency of the oscillator. 
The filter comprises an integrating circuit 904 which receives the output 
signal, via conductor 804, from the demodulator as described above. 
The integrator 904 has controlled time constants which govern the overall 
response of the phase locked loop and which are selected for stability and 
rate considerations. A phase and current shift will cause the integrating 
circuit capacitor 906 to charge or discharge. The resultant voltage will 
cause the voltage controlled oscillator frequency to change in a manner to 
cause a reduction in phase shift. The voltage at the integrator output, 
conductor 902, will settle and become stable when the phase shift 
approaches zero (tuned condition). Changes in phase shift will cause a 
change in output signal which is fed to the oscillator to cause a 
corresponding frequency correction. The output voltage from the loop 
filter, therefore, is a voltage signal representative of the phase 
relation between the current and the voltage applied to the transducer 18 
and such voltage signal will be a constant direct current voltage when a 
substantially zero phase shift condition prevails, that is, the voltage 
controlled oscillator provides the proper frequency for precise parallel 
resonant operation of the transducer. 
FIG. 7 is a shcematic electrical circuit diagram of the voltage controlled 
oscillator. A specific circuit is shown, but other arrangements could be 
used for providing the same function. The oscillator includes a commercial 
oscillatory timer circuit 312, such as Texas Instruments No. 555, arranged 
to operate in an astable mode. The components 314, 316, 318 and 320 are 
selected to cause the oscillator to operate at a frequency of four times 
the parallel resonant frequency 4f.sub.p of the transducer. The frequency 
of operation of the timer 312 is also a function of the value of the 
direct current voltage signal supplied to its input control by conductor 
902, the signal from the loop filter. Resistors 322, 324, 326 and 328 form 
a voltage shifting and scaling network which couples the control or 
feedback voltage input to the control signal pin on timer 312. As the 
voltage at the input pin of timer 312 is made more positive, the frequency 
of the timer decreases and, conversely, a more negative control voltage 
will cause the frequency to increase. 
Variable resistor 326 is used to set the amount of frequency deviation 
which the control voltage will effect. In this manner, a limit is set for 
the range of frequencies (bandwidth) within which the power supply will 
operate. Variable resistor 314 is used to adjust the center frequency. 
The output from the voltage controlled oscillator 312, operating at four 
times the fundamental frequency, is sent to a D-type flip/flop 330 
connected to form a divide by two circuit. The frequency signal 4f.sub.p 
from the oscillator is thus divided by a factor of two to yield two 
signals, namely 2f.sub.p conductor 302, and its complementary, 180 degrees 
shifted, signal 2f.sub.p, conductor 304. 
Two additional D-type flip/flop circuits 332 and 334 are used to generate 
reference signals. Flip/flop 332 again is connected as a divide by two 
circuit, the signal 2f.sub.p being connected to its input. The outputs 
from this flip/flop 332 are the fundamental frequency signal f.sub.p 
apparent at conductor 306 and conductor 310 and the complementary signal 
f.sub.p. These latter signals are 180 degrees out of phase with one 
another. The signal 2f.sub.p acts as a clocking signal for the flip/flop 
334 and the signal f.sub.p acts as the data input. The result is that 
flip/flop 334 produces an output signal f'.sub.p conductor 308, which 
signal is the fundamental frequency, but shifted by ninety degrees. The 
signals in digital form along conductors 308 and 310 are coupled to the 
demodulator 800 as previously described, whereas the signals of conductors 
302, 304 and 306 are coupled to the modulator and driver circuit 400. 
The voltage (amplitude) control circuit is shown in FIG. 8. A parameter of 
prime importance to an application of this type of ultrasonic apparatus is 
the amplitude of mechanical vibration provided by the transducer and horn. 
As described above, a motional amplitude responsive signal, conductor 206, 
FIG. 3, is derived in the output network, which signal is proportional to 
the driving voltage applied to the transducer. This voltage is known also 
as "motional voltage". The motional voltage is scaled and rectified. It is 
then summed with a reference voltage to produce an error signal. The error 
signal is amplified and fed to a combiner and to the modulator and driver 
circuit input. This feedback loop has the purpose of maintaining a desired 
motional amplitude setting. Because the motional voltage is sensed, the 
circuit arrangement is such as to maintain a set amplitude irrespective of 
line voltage variations and of loading effects reflected on the power 
supply. 
Provisions are made to vary the reference voltage either internal or 
external to the power supply. In this manner, the amplitude of vibration 
may be set by a control potentiometer or by an external signal, such as a 
process derived signal. The system has a fast response time so that an 
amplitude variation can be effected even during a particular weld cycle. 
During the start-up period, the reference voltage signal is modified by a 
ramp function generator. The result is that the motional voltage and the 
resultant amplitude of vibration starts from a rest condition and 
increases at a set linear rate until the preset point of regulation is 
reached. Therefore, the transducer is caused to increase its amplitude of 
vibration in an orderly manner at a linear rate, rather than in a stepwise 
fashion. 
The motional voltage signal, conductor 206, is rectified by rectifier 504 
and filtered by capacitor 506. The resultant direct current signal is sent 
to an integrating amplifier 508. At the amplifier 508, the direct current 
signal responsive to the driving voltage applied to the transducer 18 is 
compared with a reference voltage setting. The reference voltage value may 
be a set point signal from an amplitude adjusting potentiometer 518, or a 
variable signal, such as a ramp signal originating at the ramp voltage 
generator 510 comprising an amplifier 512 and capacitor 514 in conjunction 
with a voltage source and series connected switch 516 having a "STOP" 
position and a "RUN" position. 
If the amplitude responsive signal is lower than the reference voltage, the 
output signal from the amplifier 508 will increase the signal level to the 
combiner 20 and to the modulator and driver circuit 400 for causing the 
output voltage provided by the converter to increase. If the amplitude 
responsive signal is greater than the reference voltage, the output from 
the amplifier 508 will decrease and cause the motional voltage applied to 
the transducer to decrease. 
For starting the power supply from rest, the ramp generator 510 is used to 
modify the reference voltage from potentiometer 518. At rest, switch 516 
is in the "STOP" setting as shown. The ramp generator clamps the reference 
voltage to a zero value. When the switch is moved to the "RUN" position, 
the ramp generator output slowly rises at a linear rate and allows the 
reference voltage to rise also. This occurs until the clamp diode 520 is 
no longer conductive, at which condition the power supply is running at a 
steady state condition. It will be understood, of course, that the switch 
516 will be an electronic switch. 
FIG. 9 is a circuit diagram of the current control circuit 600. The current 
control circuit components regulate and limit the normal output current 
levels produced by the power supply. During a normal sequence of power 
supply operation, there exist various conditions in which these circuit 
components come into operation. 
During the operate or run time, the power supply may be required to deliver 
more power than it can safely provide. The operating current level is 
sensed at the output network 200, processed by the current filter 700 and 
by the real current component circuit of the demodulator 800. The 
resultant signal is compared in the current control circuit 600 with a 
reference signal defining maximum current. Any resulting difference or 
error signal corresponding to excessive current is used to control the 
modulator and driver circuit 400 after having passed through the combiner 
20. The action is to reduce the alternating current voltage amplitude 
output from the converter 100 and, hence reduce and adjust the current to 
its preset maximum value. This control is linear in its characteristic. 
The transducer 18, moreover, is a mechanically resonant device and will 
store energy. The device is bidirectional in that it will both use and 
generate energy, that is electrical current. In order to control the 
amplitude of vibration of the transducer at a fast rate, the power supply 
must be able to receive as well as generate a maximum current. The output 
from the real current component portion of the demodulator, conductor 802, 
will change its polarity and level corresponding to the amount of current 
and its direction to or from the transducer. The output from the current 
control circuit will control, via the combiner 20, the modulator and 
driver circuit 400 to either decrease or increase the effective 
alternating current voltage output from the converter 100 depending upon 
the direction of current flow to or from the transducer. 
During the start-up cycle, large reactive current components may be present 
in the current path from the converter to the transducer. In this event, a 
sample of the imaginary or reactive component signal from the demodulator, 
conductor 804, is combined with the real component current signal. The 
result is a load line shaping or modification of the current level set 
point to better protect the switching devices from failure caused by 
excessive energy switching during periods of load transitions. 
Also, during start-up, large mechanical horns may require excessive energy 
to reach a set amplitude of vibration. In this case, if the current 
requirements are too great for causing the transducer with horn to reach 
its set amplitude during an allotted time interval, the current control 
circuit will modify the start cycle time by automatically reducing the 
signal to the modulator and driver circuit 400. This feature will lengthen 
the start up time and keep the power supply from reaching an overload 
condition. 
The real component of the current signal, conductor 802, from the 
demodulator 800 will have a value dependent on the current flow direction. 
Its polarity will be positive if the current flow is toward the transducer 
18 and will be negative if the current flow is in the direction from the 
transducer. This real current component signal is sent to two integrating 
amplifiers 605 and 606 together with a respective reference signal. A 
signal corresponding to the maximum permissible forward current is 
generated by potentiometer 608 and the signal corresponding to the maximum 
permissible reverse current is generated by potentiometer 610. 
If the actual forward current signal exceeds the reference forward signal 
value, the amplifier 605 will produce an output voltage signal to the 
combiner 20 and to the modulator and driver circuit, conductor 602, to 
decrease the output of the power supply by reducing the alternating 
current voltage from the converter. This condition will either limit the 
rate of rise of the current to the transducer or reduce the current to a 
predetermined safe level. 
If, however, the reverse current signal exceeds the reference reverse 
signal value, the amplifier 606 will produce an output voltage signal to 
the modulator, conductor 604, for causing an increase of current flow from 
the power supply, i.e. raising the alternating current voltage from the 
converter. This action will limit the rate of decrease of current flow 
from the transducer 18 to a safe level. 
A portion of the imaginary or reactive current component received from the 
demodulator via conductor 804 is summed together with the real current 
component at junction 612. This summing action results in that the total 
amount of forward current during start up of the power supply is 
controlled if a mistuned condition prevails. 
FIG. 10 depicts the combiner circuit which combines, at junction 24, the 
output signal from the voltage control circuit 502, the voltage control 
signal, with output signals from the current control circuit provided 
either by conductor 602 or conductor 604 to produce via buffer amplifier 
26 a combined control signal at conductor 22 to the modulator and driver 
circuit 400. This control signal serves as a composite control signal to 
regulate the output voltage provided by the converter 100. If the 
transducer operates within predetermined levels of current flow, only the 
amplitude control signal 502 will be effective as output from the combiner 
22. If the current flow to the transducer or from the transducer is above 
the desired level, the voltage control signal is modified by the current 
responsive signal as described. 
FIG. 11 is a schematic circuit diagram of the modulator and driver circuit 
which receives the frequency responsive signals from the voltage 
controlled oscillator and the combined voltage control signal from the 
voltage control circuit and the current control circuit. Therefore, the 
modulator and driver circuit 400 operates on a voltage control signal and 
the signals generated by the voltage controlled oscillator providing 
output signals for suitably controlling the operation of the direct 
current to alternating current converter 100. 
The timing signal from the conductor 302 (double frequency signal 2f.sub.p) 
coming from the voltage controlled oscillator 300 is sent to an 
integrating amplifier circuit 410 which causes a triangular shaped output 
signal with equal slopes. This signal, in turn, is applied to a comparator 
circuit 412. The comparator circuit 412 also receives, via conductor 22, 
the steady state composite signal from the combiner 22, representing a 
voltage control signal. The comparator is used to compare the control 
voltage with the triangularly shaped signal. The output from the 
comparator 412 is fed to a NAND gate 414 and an AND gage 416. The NAND 
gate 414 also receives the timing signal 2f.sub.p from conductor 302. The 
AND gate 416 receives as its second input signal the signal from conductor 
304 representing the signal of double frequency of the imaginary current 
component 2f.sub.p. The output from the NAND gate 414 and the output from 
the AND gate 416 are applied as inputs to a respective flip/flop circuit 
418 and 420, each of which receives also a signal f.sub.p from conductor 
306, representing a timing signal. 
The output signals from flip/flop 418 and flip/flop 420 exhibit a variable 
phase relationship with each other, varying from a minimum of zero 
degrees, which will be minimum output voltage from the converter 100, to a 
maximum of 180 degrees, providing maximum output voltage. The buffer 
amplifiers 422 form driving stages. The respective 180 degrees shifted 
output signals appearing across conductors 402 and 404, and conductors 406 
and 408 are coupled to the driving stage transformers 120 and 122 of the 
converter circuit, see FIG. 2. Therefore, the converter 100 is caused to 
provide by pulse modulation a feedback controlled alternating current 
output voltage, accurately controlled in respect to frequency, amplitude 
of motional voltage and maximum current flow. Hence, the power supply has 
all the desired attributes set forth at the beginning of the 
specification. 
While there has been described and illustrated a preferred embodiment of 
the invention, it will be apparent to those skilled in the art that 
various changes and modifications may be made therein without departing 
from the broad principle of this invention, which shall be limited only by 
the scope of the appended claims.