Directional microphone system

Full directional pickup coverage is realized by employing a pickup arrangement which provides a plurality of audio polar directivity patterns, i.e., directional beams. These polar directivity patterns are formed in a unique embodiment of the invention by generating a plurality of frequency independent time-delayed versions of a corresponding plurality of spatially sampled signals and by combining each of the plurality of spatially sampled signals with one or more selected ones of the time delayed versions to generate at least a similar plurality of polar directivity patterns. More specifically, the spatially sampled signals are combined with the delayed versions in such a manner that a greater number of polar directivity patterns can be considered than the number of spatially sampled signals. In a specific embodiment, the spatially sampled signals are acoustic (audio) and a plurality of microphones arranged in a predetermined spatial configuration is employed to obtain them.

CROSS REFERENCE TO RELATED APPLICATIONS 
U.S. patent applications Ser. No. 08/268,463 and Ser. No. 08/258,464 were 
filed concurrently herewith. 
TECHNICAL FIELD 
This invention relates to microphone systems and, more particularly, to 
directional microphone systems. 
BACKGROUND OF THE INVENTION 
In certain audio communications systems it is desirable to have full room 
audio (acoustic) pickup. One solution to realize full room coverage is to 
use a single omni-directional microphone. Use of such an omni-directional 
microphone, however, has several limitations, namely, the pickup of sound 
echoes or reverberation as well as noise from the room. Moreover, in 
two-way communications systems using, for example, a speakerphone, the 
acoustic coupling between the receiving loudspeaker and microphone leads 
to objectionable echoes and/or annoying switching transients because of 
the required use of switched loss in the speakerphone. 
The limitations of the omni-directional microphone lead to the 
consideration of using directional microphones in such communications 
system. Directional gradient type microphone elements using internal 
acoustic subtraction are commercially available. However, use of the 
directional gradient type microphone in an apparatus requires a prior 
knowledge of the location of a talker relative to the apparatus. 
Consequently, to obtain full room coverage, a plurality of such 
directional gradient type microphones would be required. This solution, 
however, is complex and expensive. 
SUMMARY OF THE INVENTION 
Full directional pickup coverage is realized by employing a pickup 
arrangement which provides a plurality of polar directivity patterns, 
i.e., a plurality of directional beams. These polar directivity patterns 
are formed in a unique embodiment of the invention by generating a 
plurality of frequency independent time-delayed versions of a 
corresponding plurality of spatially sampled signals and by combining each 
of the plurality of spatially sampled signals with one or more selected 
ones of the time delayed versions to generate at least a similar plurality 
of polar directivity patterns. More specifically, the spatially sampled 
signals are combined with the delayed versions in such a manner that a 
greater number of polar directivity patterns can be considered than the 
number of spatially sampled signals. 
In another embodiment, the spatially sampled signals are also combined with 
each other in such a manner to form additional polar directivity patterns. 
In a specific embodiment, the spatially sampled signals are acoustic 
(audio) and a plurality of microphones arranged in a predetermined spatial 
configuration is employed to obtain them. 
A technical advantage of the invention is that the number of polar 
directivity patterns generated to handle the full directional, e.g., room, 
coverage pickup is greater than the number of microphone inputs required. 
Another technical advantage is the ability to alter the shape of the audio 
polar directivity patterns solely through changing the software code.

DETAILED DESCRIPTION 
FIG. 1 illustrates in simplified form a signal flow diagram for signal 
channels associated with three microphone elements employing one 
embodiment of the invention. It is noted that the signal flow diagram of 
FIG. 1 illustrates the signal flow processing algorithm which may be 
employed in a digital signal processor (DSP) to realize the invention. It 
is noted, however, although the preferred embodiment of the invention is 
to implement it on such a digital signal processor, that the invention may 
also be implemented as an integrated circuit or the like. Such digital 
signal processors are commercially available, for example, the DSP 1600 
family of processors available from AT&T. 
Shown in FIG. 1 are microphone elements 101, 102 and 103, which in this 
embodiment, are arranged in an equilateral triangle as shown in FIG. 2. As 
shown in FIG. 2, microphone elements 101,102 and 103 are placed at the 
vertices of the equilateral triangle with a predetermined spacing "d" 
between the vertices. In this example, the spacing d between the vertices 
is approximately 0.85 inches. An output signal from microphone element 101 
is supplied via amplifier 104 and Codec 105 to DSP 106 and therein to 
balance network 107. DSP 106 includes the digital signal flow processing 
to realize the invention. Also shown is microphone element 102 whose 
output is supplied via amplifier 108 and Codec 109 to DSP 106 and therein 
to balance network 107. Finally, an output signal from microphone element 
103 is supplied via amplifier 110 and Codec 111 to DSP 106 and therein to 
balance network 107. In one example, employing the invention, microphone 
elements 101, 102 and 103 are so-called omni-directional microphones of 
the well-known electret-type. Although other types of microphone elements 
may be utilized in the invention, it is the electret type that are the 
preferred ones because of their low cost. Codecs 105, 109 and 111 are also 
well known in the art. One example of a Codec that can advantageously be 
employed in the invention is the T7513B Codec, also commercially available 
from AT&T. In this example, the digital signal outputs from Codecs 105, 
109 and 111 are encoded in the well-known mu-law PCM format, which in DSP 
106 must be converted into a linear PCM format. This mu-law-to-linear PCM 
conversion is well known. Balance network 107 is employed to balance, 
i.e., match, the long term average broad band gain of the signal channels 
associated with microphone elements 101, 102 and 103 to one another. In 
this example, the long term average broad band gain of the signal channels 
associated with microphone elements 101 and 103 are balanced to the signal 
channel associated with microphone element 102. Details of balance network 
107 are shown in FIG. 3 and described below. 
More specifically, DSP 106 first forms a plurality of polar directivity 
patterns to provide full pick up coverage of a particular space, for 
example, a room, stage, arena, area or the like and then vote on the polar 
directivity pattern (or patterns) that has the best signal-to-noise ratio, 
thus picking up the desired signal source. In this example, the polar 
directivity patterns are acoustic (audio) and are in predetermined spatial 
orientation relative to each other in order to provide full 360.degree. 
coverage of the particular space. To this end the balanced microphone 
signal channel outputs A, B and C corresponding to microphones 101,102 and 
103, respectively, from balance network 107 are delayed by delay units 
112, 113 and 114, respectively. In this example, each of delay units 112, 
113 and 114 provides a time delay interval equivalent to the time that 
sound takes to travel the distance d from one of the microphone pick up 
locations to another to yield frequency independent time delayed versions 
A', B' and C' respectively. The delayed signal outputs A', B' and C' from 
delay units 112, 113 and 114 are then algebraically combined with the 
non-delayed versions A, B and C, respectively, from balance network 107 
via algebraic summing units 121 through 126 to generate six signals 
representing cardioid polar directivity patterns. Alternatively, for 
distance d being twice the above noted value, and the time delay interval 
being equivalent to one-third the time it takes sound to travel the new 
distance, hypercardioid polar directivity patterns will be generated for 
the six polar directivity patterns. FIG. 5A shows the relationship of a 
cardioid polar directivity pattern (solid outline) and a hypercardioid 
polar directivity pattern (dashed outline). Note that by further changing 
the delay interval of each of delay units 112, 113 and 114 and/or the 
spacing "d", the resulting polar directivity patterns can be changed, as 
desired. Changing this delay interval is readily realized simply by 
reprogramming DSP 106. 
FIG. 5 illustrates the relationship of the equilateral triangle 
configuration of microphones 101, 102 and 103 and the resulting six 
cardioid polar directivity patterns, as well as, the resulting three "FIG. 
8" polar directivity patterns which will be discussed below. The six 
cardioid polar directivity patterns result from the algebraic summing of 
the delayed versions of the balanced channel signals A', B' and C' with 
the non-delayed balanced channel signals A, B and C, respectively. Thus, 
summing unit 121 yields at circuit point 131 a signal (B-A') 
representative of a cardioid polar directivity pattern having its null in 
the direction of microphone 101 and having its maximum sensitivity in the 
direction of microphone 102 (shown in dashed outline in FIG. 5 from 
direction 2 to direction 5). Summing unit 122 provides at circuit point 
132 a signal (C-A') representative of a cardioid polar directivity pattern 
having its null also in the direction of microphone 101 and having its 
maximum sensitivity in the direction of microphone 103 (shown in dashed 
outline in FIG. 5 from direction 3 to direction 6). Summing unit 123 
yields at circuit point 133 a signal (A-B') representative of a cardioid 
polar directivity pattern having its null in the direction of microphone 
102 and having its maximum sensitivity in the direction of microphone 101 
(shown in solid outline in FIG. 5 from direction 5 to direction 2). 
Summing unit 124 yields at circuit point 134 a signal (C-B') 
representative of a cardioid polar directivity pattern having its null in 
the direction of microphone 102 and having its maximum sensitivity in the 
direction of microphone 103 (shown in solid outline in FIG. 5 from 
direction 4 to direction 1). Summing unit 125 yields at circuit point 135 
a signal (A-C') representative of a cardioid polar directivity pattern 
having its null in the direction of microphone 103 and having its maximum 
sensitivity in the direction of microphone 101 (shown in solid outline in 
FIG. 5 from direction 6 to direction 3). Summing unit 126 yields at 
circuit point 136 a signal (B-C') representative of a cardioid polar 
directivity pattern having its null in the direction of microphone 103 and 
having its maximum sensitivity in the direction of microphone 102 (shown 
in dashed outline in FIG. 5 from direction 1 to direction 4). The signals 
at circuit points 131 through 136, representative of the cardioid polar 
directivity patterns, are supplied to voting unit 140 and to multiplier 
units 141 through 146, respectively. The purpose of the cardioid polar 
directivity patterns generated by summing units 121 through 126 is to pick 
up single acoustic sources, for example, single talkers. In this example, 
the six cardioid polar directivity patterns are pointing in predetermined 
fixed directions and are spaced 60.degree. apart from each other. 
Algebraic summing units 127, 128 and 129 are employed to derive so-called 
FIG. 8 polar directivity patterns capable of picking up acoustic sources 
on opposite sides of the pickup system which are operating simultaneously, 
for example, two simultaneous talkers. Summing unit 127 provides a signal 
(A-B) at circuit point 137 representative of a FIG. 8 polar directivity 
pattern that is sensitive, in this example, to talkers at the ends of a 
directional line passing through microphones 101 and microphone 102 (shown 
in FIG. 5 as a FIG. 8 for directions 2 and 5). Summing unit 128 provides a 
signal (B-C) at circuit point 138 representative of a FIG. 8 polar 
directivity pattern that picks up, in this example, talkers at the ends of 
a directional line passing through microphone 102 and microphone 103 
(shown in FIG. 5 as a FIG. 8 for directions 1 and 4). Summing unit 129 
provides a signal (A-C) representative at circuit point 139 of a FIG. 8 
polar directivity pattern that picks up, in this example, talkers at the 
ends of a directional line passing through microphone 101 and microphone 
103 (shown in FIG. 5 as a FIG. 8 for directions 3 and 6). The signals at 
circuit points 137, 138 and 139 are also supplied to voting unit 140 and 
to multiplier units 147, 148 and 149, respectively. 
Voting unit 140 determines the optimum weighting provided by each of the 
signal channels 131 through 139 at outputs 151 through 159, respectively. 
Details of voting unit 140 are shown in FIG. 4 and described below. The 
signals representative of these weightings from outputs 151 through 159 
are also supplied to multipliers 141 through 149 respectively, to weight 
each channel in accordance with its desirability to be represented in the 
output. Algebraic summing unit 160 algebraically combines the weighted 
output signals from each of multipliers 141 through 149. Then, Codec 161 
converts the summed output signal into an analog form. The output of Codec 
161 is then transmitted as desired. 
FIG. 3 shows in simplified form a signal diagram illustrating the operation 
of balance network 107. The mu-law PCM output from each of Codecs 105, 109 
and 111 is converted to linear PCM format (not shown) in DSP 106. Then, 
the linear PCM representations of the outputs from Codec 105 and Codec 111 
are supplied to gain differential correction factor generation units 301 
and 302, respectively. Because the long term average broad band gain of 
the microphone signal channels corresponding to microphones 101 and 103 
are being matched to the signal channel of microphone 102, in this 
example, the linear PCM format output of Codec 109 does not need to be 
adjusted. Since each of gain differential correction factor generation 
units 301 and 302 is identical and operates the same, only gain 
differential correction factor generation unit 301 will be described in 
detail. To this end, the elements of each of gain differential correction 
factor generation units 301 and 302 have been labeled with identical 
numbers. 
The matching, i.e., balancing, of the long term average broad band gain of 
the signal channels corresponding to microphone elements 101 and 102 is 
realized by matching the signal channel level corresponding to microphone 
element 101 to that of microphone element 102. To this end, the linear PCM 
versions of the signal from Codec 105 is supplied to multiplier 303. 
Multiplier 303 employs a gain differential correction factor 315 to adjust 
the gain of the linear PCM version of the signal from Codec 105 to obtain 
an adjusted output signal 316, i.e., A, for microphone 101. As indicated 
above, the linear PCM version of the signal from Codec 109 does not need 
to be adjusted and this signal is output B from balance network 107. The 
adjusted output C of balance network 107 is from gain differential 
correction factor generation unit 302. 
The gain differential correction factor 315 is generated in the following 
manner: adjusted microphone output signal 316 is squared via multiplier 
304 to generate an energy estimate value 305. Likewise, the linear PCM 
version of the output signal from Codec 109 is squared via multiplier 307 
to generate energy estimate value 308. Energy estimate values 305 and 308 
are algebraically subtracted from one another via algebraic summing unit 
306, thereby obtaining a difference value 309. The sign of the difference 
value 309 is obtained using the signum function 310, in well known 
fashion, to obtain signal 311. Signal 311 will be either minus one (-1) or 
plus one (+1) indicating which microphone signal channel had the highest 
instantaneous energy. Minus one (-1) represents microphone 101, and plus 
one (+1) represents microphone 102. Multiplier 312 multiplies signal 311 
by a constant K to yield signal 313 which is a scaled version of signal 
311. In one example, not to be construed as limiting the scope of the 
invention, K typically would have a value of 10.sup.-5 for a 22.5 ks/s 
(kilosample per second) sampling rate. Integrator 314 integrates signal 
313 to provide the current gain differential correction factor 315. The 
integration is simply the sum of all past values. In another example, 
constant K would have a value of 5.times.10.sup.-6 for an 8 ks/s sampling 
rate. Value K is the so-called "slew" rate of integrator 314. 
FIG. 4 shows, in simplified block diagram form, details of voting unit 140. 
Specifically, shown are so-called talker signal-to-noise estimation units 
401 through 409. It is noted that each of talker signal-to-noise ratio 
estimate units 401 through 409 are identical to each other. Consequently, 
only talker signal-to-noise ratio estimation unit 401 will be described in 
detail. A signal representative of the cardioid polar directivity pattern 
generated by summing unit 121 is supplied via 131 to talker 
signal-to-noise ratio estimation unit 401 and therein to absolute value 
generator unit 410. The absolute value of the signal supplied via 131 is 
obtained and is then applied to peak detector 411 in order to obtain its 
peak value over a predetermined window interval, in this example, 8 ms. 
The obtained peak value is supplied to decimation unit 412 which obtains 
the generated peak value every 8 ms, in this example, clearing the peak 
detector 411 and supplies the obtained peak value to short term filter 413 
and long term filter 414. Filters 413 and 414 provide noise guarding of 
signals from stationary noise sources. Short term filter 413, in this 
example, is a non-linear first order low pass filter having a 
predetermined rise time constant, for example, of 8 ms and a fall time, 
for example, of 800 ms. The purpose of filter 413 is to generally follow 
the envelope of the detected wave form. Long term filter 414 is also a 
non-linear first order low pass filter having, in this example, a rise 
time of 8 seconds and a fall time of 80 ms. The purpose of filter 414 is 
to track the level of background interference. Ten times the logarithm of 
the filtered output signal from short term filter 413 is obtained via 
logarithm (LOG) unit 415 and supplied to one input of algebraic summing 
unit 417. Similarly, ten times the logarithm of the filtered output signal 
from long term filter 414 is obtained via LOG unit 416 and supplied to 
another input of algebraic summing unit 417. The LOG values from LOG units 
415 and 416 are algebraically subtracted in algebraic summing unit 417. 
The resulting difference signal is supplied to maximum (MAX) detector 418. 
Similarly, the outputs from talker signal-to-noise estimation units 402 
through 409 are also supplied to MAX detector 418. MAX detector 418 
provides a true output, i.e., a logical 1, for the corresponding talker 
signal-to-noise estimation unit output having the largest value output 
during the sampling window, in this example, 8 ms. MAX detector 418 also 
provides a false, i.e., logical 0, output for the signal channels 
corresponding to the other talker signal-to-noise estimation units. 
Additionally, MAX detector 418 provides an output only when a difference 
between the logarithm of the maximum signal-to-noise ratio value minus the 
logarithm of the minimum signal-to-noise ratio value obtained during the 8 
ms window is greater than a predetermined value, in this example, 3 dB, 
and when the logarithm of the maximum signal-to-noise ratio value is 
greater than a second predetermined value, in this example, 15 dB. The 
outputs from MAX detector 418 are supplied to up/down (U/D) counters 421 
through 429. Each of U/D counters 421 through 429 increase their count 
value by a predetermined value, in this example, 0.05, each time the 
signal supplied from MAX detector 418 is true up to a predetermined 
maximum value of, in this example, one (1). Likewise, if the signal 
supplied from MAX detector 418 to U/D counters 421 through 429 is false, 
the counters count down by the predetermined value of, in this example, 
0.05 to another predetermined value of, in this example, zero (0). Each of 
counters 421 through 429 count either up or down once every window 
interval of 8 ms, in this example. When the above noted conditions 
regarding the values of the logarithm of the maximum and minimum 
signal-to-noise ratios are not met, all of counters 421 through 429 
maintain their present count. The outputs from U/D counters 421 through 
429 are the outputs 151 through 159, respectively, of voting unit 140. 
FIG. 6 illustrates, in simplified form, a flow diagram for signal channels 
associated with microphone elements 101, 102 and 103 employing another 
embodiment of the invention. The spatial configuration of microphone 
elements 101, 102 and 103 in this embodiment, includes two legs extending 
from a single point at a right angle and having one of the microphones at 
each end of the legs and at the single point. Thus, as shown in FIG. 7 
microphone element 101 is at one end of one of the legs, microphone 
element 102 is at the single point and microphone element 103 is at the 
end of the other leg of the right angle. As shown in FIG. 7, the spacing 
between the microphones is "d". It is noted that the signal flow diagram 
of FIG. 6 employs some of the elements of the signal flow diagram shown in 
FIG. 1. The elements which are similar have been similarly numbered and 
since their operation is identical to that of FIG. 1 they will not be 
described again in detail. It is noted, however, that instead of employing 
nine summing units, six of which generated the cardioid polar directivity 
patterns and three of which generated the FIG. 8 polar directivity 
patterns in the embodiment of FIG. 1, the embodiment of FIG. 6 employs 
algebraic summing units 121, 123, 124 and 126 to generate four cardioid 
polar directivity patterns and algebraic summing units 127 and 128 to 
generate two FIG. 8 polar directivity patterns. Voting unit 140 generates 
the weighted signal-to-noise ratio values only for the signals supplied at 
circuit points 131, 133, 134, 136, 137 and 138 from their associated 
algebraic summing units. Thus, only six signal channels are being voted on 
and similarly only those six signal channels are being weighted via 
multipliers 141, 143, 144, 146, 147 and 148 via weighted outputs 151, 153, 
154, 156, 157 and 158, respectively, from voting unit 140. Algebraic 
summing unit 160 algebraically sums the weighted outputs from multipliers 
from 141, 143, 144, 146, 147 and 148 to obtain the desired digital output. 
This digital output is supplied to Codec 161 which converts it to audio 
form for further transmission as desired. 
FIG. 8 illustrates the relationship of the right triangle configuration of 
microphones 101, 102 and 103 and the resulting four cardioid polar 
directivity patterns as well as the resulting two FIG. 8 polar directivity 
patterns. The four cardioid polar directivity patterns result from the 
algebraic summing of the delayed versions of the balanced channel signals, 
A', B' and C' with the non-delayed balanced channel signals A, B and C, 
respectively. Thus, summing unit 121 yields, at circuit point 131, a 
signal (B-A') representative of a cardioid polar directivity pattern 
having its null in the direction of microphone 101 and having its maximum 
sensitivity in the direction of microphone 102 (shown in FIG. 8 from 
direction 2 to direction 4). Summing unit 123 provides, at circuit point 
133, a signal (A-B') representative of a cardioid polar directivity 
pattern having its null in the direction of microphone 102 and having its 
maximum sensitivity in the direction of microphone 101 (shown in FIG. 8 
from direction 4 to direction 2). Summing unit 124 yields, at circuit 
point 134, a signal (C-B') representative of a cardioid polar directivity 
pattern having its null also in the direction of microphone 102 and having 
its maximum sensitivity in the direction of microphone 103 (shown in FIG. 
8 from direction 3 to direction 1). Summing unit 126 yields, at circuit 
point 136, a signal (B-C') representative of a cardioid polar directivity 
pattern having its null in the direction of microphone 103 and having its 
maximum sensitivity in the direction of microphone 102 (shown in FIG. 8 
from direction 1 to direction 3). Again, the signals at circuit points 
131, 133, 134 and 136 are supplied to voting unit 140 and to multiplier 
units 141,143, 144 and 146, respectively. The purpose of the cardioid 
polar directivity patterns generated by summing units 121,123, 124 and 126 
is also to pick up single acoustic sources. Algebraic summing units 127 
and 128 are employed to derive so-called FIG. 8 polar directivity patterns 
capable of picking up acoustic sources on opposite sides of the pick up 
system which are operating simultaneously, for example, two simultaneous 
talkers. Summing unit 127 provides a signal (A-B) at circuit point 137 
representative of a FIG. 8 polar directivity pattern that is sensitive, in 
this example, to talkers at the ends of a directional line passing through 
microphones 101 and 102 shown in FIG. 8 as a FIG. 8 for directions 2 and 
4. Summing unit 128 provides a signal (B-C) at circuit point 138 
representative of a FIG. 8 polar directivity pattern that picks up, in 
this example, talkers at the ends of a directional line passing through 
microphone 102 and microphone 103 shown in FIG. 8 as a FIG. 8 for 
directions 1 and 3. 
Although the embodiments of the invention have been described in the 
context of picking up acoustic (audio) signals, it will be apparent to 
those skilled in the art that the invention can also be employed to pick 
up other energy sources; for example, those which radiate radio frequency 
waves, ultrasonic waves, or other acoustic waves in liquids and solids or 
the like.