High voltage dc-dc converter with dynamic voltage regulation and decoupling during load-generated arcs

A high-power power supply produces a controllable, constant high voltage output under varying and arcing loads. The power supply includes a voltage regulator, an inductor, an inverter for producing a high frequency square wave current of alternating polarity, an improved inverter voltage clamping circuit, a step up transformer, an output rectifier for producing a dc voltage at the output of each module, and a current sensor for sensing output current. The power supply also provides dynamic response to varying loads by controlling the voltage regulator duty cycle and circuitry is provided for sensing incipient arc currents at the output of the power supply to simultaneously decouple the power supply circuitry from the arcing load. The power supply includes a plurality of discrete switching type dc--dc converter modules.

BACKGROUND OF THE INVENTION 
1. Field Of The Invention 
The field of the present invention generally relates to high power solid 
state power supplies. More particularly, the field of the present 
invention relates to a high power solid state power supply for producing a 
controllable, constant high voltage output under varying and arcing loads 
suitable for powering an ion source, such as an electron beam gun in a 
vacuum furnace, or an electron beam gun used in the vaporizer of a laser 
isotope separation system, or a plasma sputtering device or the like. 
2. The Prior Art 
An electron beam gun is used in a vacuum furnace system, or the like, for 
providing a high intensity beam of electrons to bombard a target material. 
The electron gun is typically disposed in an evacuated chamber together 
with the target material. The electron gun or E-beam gun usually includes 
a source of electrons, such as a heated cathode or filament, and a 
grounded accelerating anode. The cathode is maintained at a high negative 
potential with respect to the anode to establish a high electrostatic 
field for accelerating the electrons. A magnetic field may typically be 
provided for directing the electrons onto the target material. 
During bombardment of the target material by the electron beam, various 
ionized materials are emitted. The presence of such materials often 
effects a substantial decrease in the voltage withstand capability between 
the various parts of the electron beam gun and other elements. This may 
result in arcing between the electron gun parts and other structures. 
Arcing causes a substantial increase in the electron gun current and may 
result in damage to the electron gun structure and surrounding elements. 
Arcing may also cause damage to the power circuitry driving the electron 
gun. 
In high power and high performance applications, such as the vaporizer in a 
laser isotope separation system, physical spacing between the E-beam gun, 
surrounding components, and target materials is relatively small. As a 
result, the E-beam gun may arc to ground frequently. To avoid damage and 
to achieve long lifetimes, it is essential that the energy stored in the 
power supply output capacitance be small and that the so called power 
supply let through energy during arcing be small. In addition, the close 
physical spacing causes a greater chance for the electron beam to impinge 
on adjacent components and structures during steady, non-arcing operation. 
To avoid this, it is important that the power supply output voltage be 
accurately controllable with low ripple content. 
Conventional thyristor controlled power supplies are inadequate for high 
power and high performance electron beam gun applications. Thyristor 
controlled power supplies generally operate at 60 Hz line frequency and 
generate significant output voltage ripple or require substantial output 
capacitance to reduce the ripple to acceptable levels. If gun arcs are 
frequent, the output capacitance may result in excessive accumulated 
energy discharge into the gun or surrounding components and result in a 
short lifetime. Thyristor controlled power supplies also have a relatively 
slow dynamic response which results in further energy let through to the 
gun during the arc and slow ramp up after the arc is extinguished. 
Thyristor controlled power supplies have a relatively poor input power 
factor and generate high input harmonics. This causes substantial cost 
increases in the 60 Hz utility power system in large power applications. 
Thyristor controlled power supplies are also physically large because the 
transformer and filter components operate at 60 Hz and the lower harmonics 
of 60 Hz. This is an important factor in capital equipment costs where 
large numbers of power supplies are used. 
Conventional power supplies utilizing series pass tetrode vacuum tubes 
eliminate many of the deficiencies of the thyristor controlled power 
supply. The regulating characteristics of the tetrode vacuum tube can be 
used to produce very low output ripple voltages without requiring 
significant output capacitance. The regulating characteristics also permit 
a diode rectifier front end to be used which greatly raises the input 
power factor and reduces the input line harmonics. The current limiting 
tube characteristics, the high speed control capability of the tetrode 
grid, and the low output capacitance provide excellent response to gun 
arcs resulting in low energy into the gun and fast recovery after the arc 
extinguishes. 
However, conventional tetrode vacuum tube E-beam power supplies have 
serious deficiencies of their own. The efficiency of this type of power 
supply is 80% or less compared to approximately 95% for thyristor 
controlled power supplies. This is because the tetrode must drop 
substantial voltage continuously for it to regulate properly. Tetrode 
vacuum tubes also wear out due to the filament breaking and to the 
chemical breakdown of the coating on the cathode which causes the cathode 
to lose its ability to emit electrons. As a result, the tetrode vacuum 
tube is a substantial maintenance expense item having to be replaced at 
least every 10,000 hours. 
Power supplies which use switch-mode dc--dc converters operating at 10 kHz 
and above have the potential to eliminate the deficiencies of the 
conventional thyristor controlled and series-pass tetrode type power 
supplies. Power supplies which employ switch-mode dc--dc converters are 
compact because of smaller transformer and filter components, operate with 
a diode rectifier input for high input power factor, are efficient because 
they do not operate as linear regulators, require low maintenance because 
they are all solid state, and can have good dynamic response because they 
operate at high frequency. 
One type of switching dc--dc converter useful for high power applications 
above 10 KHz with arcing loads is the series resonant type. Power supplies 
which use series resonant type dc--dc converters have an input rectifier 
and filter to produce a dc voltage, an inverter consisting of thyristors 
and a resonant network to produce high frequency current, a transformer 
for producing the desired output voltage level, and a rectifier and filter 
to produce dc for application to the load. This is a well known type of 
power supply which has been applied to E-beam guns (U.S. Pat. No. 
3,544,913, issued Dec. 1, 1970). The major deficiencies in this type of 
power supply for high performance E-beam applications are the amount of 
energy stored in the output filter capacitance and the inability to turn 
off power to the load until the resonant network reverses polarity. The 
output current of the inverter is sinusoidal and a substantial capacitance 
is required after rectification to obtain satisfactory output voltage 
ripple even though the inverter operates above 10 KHz. The dc--dc 
converter also continues to provide current to the load after an arc 
occurs until the resonant network commutates the thyristors. Although 
superior to the conventional 60 Hz thyristor type power supply with 
respect to energy dissipated into the gun during arcing, it is inferior to 
the series pass tetrode type power supply, and is not adequate for high 
performance power supplies for E-beam guns. 
Another type of switching dc--dc converter useful for low power 
applications up to a few kilowatts and arcing loads is the current source, 
pulse-width-modulated type. This dc--dc converter consists of a voltage 
regulator, inductor, non-regulating inverter, transformer, output 
rectifier, and output filter capacitor as described in U.S. Pat. No. 
3,737,755, issued Jun. 5, 1973. The inductor and inverter described in 
this referenced patent produce a square current waveform to the output 
rectifier and filter which allows a small output filter capacitance to be 
used and therefore low energy to the load during load arcs. However, the 
inverter voltage clamping means is inadequate for high power applications. 
This is because the inverter is relatively distant from the input filter 
capacitor in high power applications which results in substantial 
inductance in the clamping network and excessive voltage spikes across the 
inverter transistors. 
As described above, problems exist with conventional power supplies and 
with switching power supplies for high power and high performance ion 
sources and specifically, electron beam guns. In summary, conventional 
thyristor power supplies have high output capacitance, slow dynamic 
response to arcing, and poor input power factor. Series pass tetrode 
regulator type power supplies have relatively low efficiency and 
substantial maintenance expense related to the vacuum tube. Both of these 
conventional types of power supplies are also physically large. Switching 
power supplies using series resonant type dc--dc converters solve many of 
the problems associated with conventional power supplies but still have 
excessive output capacitance and too slow a response to arcing. Switching 
power supplies using current source, pulse-width-modulated type dc--dc 
converters as described in the prior art potentially meet the E-beam gun 
power supply requirements but do not operate at high power levels. 
Accordingly, it is an object of the present invention to provide an 
improved current source, pulse-width-modulated type dc--dc converter 
suitable for operation at 100 kW or more. 
It is another object of the present invention to provide a power supply 
which is modular with one or more dc--dc converter modules of identical 
design rated at 100 kW or more used to achieve the required output power. 
It is also an object of the present invention to provide a power supply 
which has tight voltage regulation and low output voltage ripple for 
precise beam control as well as small output capacitance for small energy 
into the load during load arcs. 
A further object of the present invention is to provide a power supply 
which is current limited during an arc, which cuts back power to zero 
within a few micro-seconds or less after an arc is initiated, and which 
then ramps power back on in several milli-seconds after the cutback 
interval. 
A still further object of the present invention is to provide a power 
supply which operates without excessive voltage transients or cable 
reflections with cable lengths between the power supply and the load of 
100 feet or more during and after load arcing. 
SUMMARY OF THE INVENTION 
In accordance with the present invention, a solid state switching power 
supply is provided for converting a utility supplied ac input voltage into 
a high voltage dc output suitable for driving an electron gun for a high 
power and high performance application such as in an atomic vapor laser 
isotope separation process or the like. The present invention also may be 
useful for applications such as radar systems, which may behave as an 
arcing load. 
The present power supply comprises an input diode rectifier and filter for 
converting the ac input voltage to an unregulated dc voltage. A plurality 
of switching type dc--dc converter modules, each rated at 100 kW or more, 
are connected with their inputs in parallel and their outputs in series 
and used to convert this unregulated dc voltage into a regulated high dc 
voltage output. 
The dc--dc converter modules are an improved version of the well known 
current source, pulse-width-modulated type. Each converter consists of an 
input decoupling network, an input capacitor, a voltage regulator, a 
square wave inverter, a step up transformer, and an output rectifier and 
filter. The input decoupling network functions together with the input 
capacitor to eliminate interaction between modules and to prevent 
significant high frequency current in the cables from the input rectifier 
and filter. This feature permits parallel operation of the modules from 
one dc source. Insulated gate bipolar transistors (IGBTs) are used as the 
switches in the voltage regulator and inverter because of their high power 
capability and fast switching speeds. An improved inverter clamping 
network minimizes stray inductive loops and permits tight control of the 
inverter IGBT voltages during inverter output polarity transitions. Both 
of these features permit module operation with inverter switching 
frequencies of 10 kHz and above and output power levels of 100 kW and 
above. 
Each dc--dc converter module has a control scheme to permit operation with 
widely varying and arcing loads. A feedback loop regulates the module 
output voltage by controlling the on/off ratio of the voltage regulator 
switches. A feedback signal representing the output voltage is generated 
using an output simulator circuit fed from a one turn sense winding on the 
step up transformer and a dc current sensor on the module output lead. 
This feature permits generation of a feedback signal at ground level, 
isolated from the high voltage, which reproduces the dynamics of the 
module output power circuitry and the e-beam gun load. The control scheme 
also blocks power flow to the load during arcing. Output currents above 
normal operating levels are detected by a comparator circuit which 
triggers a cutback timer circuit. This cutback timer simultaneously turns 
off both regulator switches and turns on all four inverter switches for a 
predetermined time interval. This feature permits rapid extinguishing of 
the arcing in the load. 
The overall power supply output is formed by connecting the outputs of the 
modules in series and then connecting to an output decoupling network and 
a transmission line matching impedance before connecting to the 
transmission line to the load. The output decoupling network prevents the 
module output filter capacitors from shorting out the line matching 
impedance. This feature keeps overvoltage transients to a minimum at the 
power supply output and at the load during load arcing. 
The overall power supply includes control circuitry for phasing the timing 
of the switching of the voltage regulator and inverter in each module with 
respect to adjacent modules by 360.degree./n for n total modules. This 
feature reduces the output capacitance needed to achieve a low output 
voltage ripple and therefore reduces the energy delivered to the load 
during load arcing. Circuitry is also included for controlling the power 
supply output voltage. An overall voltage feedback loop produces an error 
signal which serves as the voltage setpoint to the individual module 
feedback loops. A ramp generator controls the rate of rise of the output 
voltage during turn on and after cutbacks.

DETAILED DESCRIPTION 
As shown in FIG. 1, an ac voltage, typically 480 volts, three phase, is 
applied to a rectifier/filter circuit 100. Rectifier/filter circuit 100 
provides a means for converting the three phase ac input voltage to a 
filtered dc output voltage on positive node 101 and reference node 102, 
respectively. The rectifier/filter consists of six diodes 103 arranged in 
a full wave bridge circuit. The output of the full wave bridge circuit 
connects to inductor 104 to provide smoothing of the current ripple. The 
inductor 104 then connects to a series connected network formed by damping 
resistor 107 and filter capacitor 108. For an output rating of 400 kW, the 
inductor 104 is typically 500 microhenries (.mu.H), the damping resistor 
107 is typically 0.5 ohms, and the capacitor 108 is typically 8,000 
microfarads (.mu.F). The rectification of the ac voltage and the filtering 
of the voltage ripple are accomplished in a known manner. 
In accordance with one aspect of the present power supply, a plurality of 
modular dc--dc converter circuits or modules 112-1, 112-2, 112-3, . . . 
112-n, are connected in parallel to the output leads, nodes 101 and 102, 
respectively of the rectifier/filter 100. Each dc--dc converter module 
112-1 through 112-n includes a voltage regulator circuit means 114 for 
producing a source of pulsating voltage of controlled duty cycle; an 
inductor 128 for converting the pulsating voltage into a direct current, 
an inverter circuit means 118 for generating a high frequency square wave 
of alternating polarity from said direct current, a transformer means 150 
for isolation and step up of the inverter output, and an output 
rectifier/filter circuit means 160 for rectifying and filtering the high 
frequency square wave current to produce a high dc output voltage. 
Each voltage regulator circuit 114 is associated with a corresponding 
decoupling network 116 and input capacitor 125. The decoupling network 
comprises a 0.25 ohm resistor 120 and a 40 microhenry (.mu.H) inductor 122 
connected in parallel and having a connection with the positive input lead 
101 from the rectifier 100. The input capacitor 125 is typically 200 
microfarads (.mu.F) and one side connects to the node formed by the 
decoupling network and the positive inverter bus 141a. The other side of 
input capacitor 125 connects to the negative input lead 102 from the 
rectifier/filter 100. The decoupling network, inductor 122 and resistor 
120, provides an impedance at high frequency which forces the current 
surges drawn by regulator 114 to pass through capacitor 125 and not 
through the rectifier/filter 100 or the parallel connected dc--dc 
converter modules. 
Each voltage regulator circuit 114 comprises two insulated gate bipolar 
transistors (IGBTs) 124 in parallel having their emitters connected with 
the negative lead of input capacitor 125 and having their collectors 
connected with the anode of a free wheeling diode 126 and with a first 
lead of a 500 microhenry (.mu.H) inductor 128, respectively. The present 
device is not limited to IGBTs. Any gate controlled switching means of 
suitable current and voltage ratings and switching speed may be used. For 
simplicity, such devices will be referred to as IGBTs. IGBTs 124a, 124b 
are activated alternately by control signals applied to their gates. Both 
IGBTs 124a and 124b are activated at 10 kHz but phase shifted 180.degree. 
with respect to each other. This results in a net 20 kHz switching 
frequency for the pair. This technique eliminates current sharing 
difficulties which might occur if both IGBTs 124a and 124b were activated 
simultaneously. The activation of IGBTs 124a, 124b enable current to build 
up and to decay through inductor 128 in such a manner that the output 
voltage of the dc--dc converter module 112 is controlled. The activation 
of IGBTs 124a, 124b causes a rectangular dc voltage waveform of from 0 to 
650 volts to appear across diode 126. The pulsating dc voltage from 0 to 
650 volts across free wheeling diode 126 is smoothed out to a low ripple 
direct current by inductor 128. The above values stated for the decoupling 
network 116 components, input capacitor 125, and inductor 128 are for a 
100 kW module output rating and for a switching frequency of 10 kHz for 
each voltage regulator IGBT 124a and 124b. 
The dc output voltage from the voltage regulator 114 is passed through 
inductor 128 to produce a substantially smooth direct current which is 
then applied to an inverter means 118 for converting the dc to a high 
frequency square wave current of alternating polarity, that is, a high 
frequency ac. Current is supplied from the node formed by the input 
decoupling network 116 and input capacitor 125 to the inverter positive 
bus 141a and returned from the inverter negative bus 141b through inductor 
128 to the voltage regulator 114. Inverter 118 comprises a plurality of 
insulated gate bipolar transistors (IGBTs) connected in series and 
configured in two branches which are in turn connected in parallel, 
forming a bridge circuit. Here also, any gate controlled switching means 
suitable for high current applications may be substituted for IGBTs. 
Preferably, four IGBTs 140 are used, two series connected IGBTs in each 
parallel branch, which are activated in diagonal pairs. There are two 
IGBTs for each ac line. It will be appreciated by those skilled in the art 
that two ac lines are needed to drive an associated single phase 
transformer. 
The IGBTs 140 are activated in alternate, diagonal pairs by control signals 
applied to their enable leads in a manner well known to those skilled in 
the art, so as to produce a square wave of alternating polarity. The high 
frequency square wave current is applied through the primary winding of an 
associated transformer 150. The IGBTs are able to develop the needed high 
frequency power due to their high switching speeds, high current 
capability, and high breakdown voltage. The high frequency current through 
the primary is inductively coupled to the secondary of the transformer 
150. A transformer 150 is part of each corresponding dc--dc converter 
module 112-1 through 124-n, and provides a means for stepping up the high 
frequency ac voltage applied to the primary in order to produce the high 
voltage needed for powering the electron gun. The transformer 150 also 
provides electrical isolation for the output of each dc--dc converter 
module which permits the outputs to be connected in series. The leads of 
the secondary of transformer 150 are then applied to an output 
rectifier/filter circuit means 160 for converting the high frequency 
alternating current to a smooth high dc voltage. The output 
rectifier/filter circuit 160 consists of a single phase, full wave bridge 
rectifier which connects to a parallel 0.05 microfarad capacitor 161 and a 
parallel damping network formed by a 2400 ohm resistor 163 in series with 
a 0.15 microfarad capacitor 162. The above values are for a four module 
power supply rated at 400 kW and 50 kV output with a .+-.0.5% peak to peak 
output voltage ripple, and each inverter operating at 10 kHz. The output 
rectifier/filter circuit means 160 operates in a well known manner. 
Referring to FIG. 1, an output decoupling network 170 comprises a 100 ohm 
resistor 172 in parallel with a 500 microhenry (.mu.H) inductor 174. One 
side of the output decoupling network 170 is connected to the high voltage 
side of the series connected dc--dc converter module outputs. The other 
side of the decoupling network is connected to the node formed by the high 
voltage transmission line to the load, the line matching network 
consisting of a 50 ohm resistor 175 and 0.01 microfarad capacitor 176 
connected in series, and the high voltage side of the voltage feedback 
divider 180. For the sub-microsecond times in which a load arc is 
initiated and transmission line reflections occur, the impedance of the 
decoupling network is much larger than the impedance of the line matching 
network. As a result, during arcing the transmission line is matched by 
its characteristic impedance and is not shorted out by the output filter 
capacitance of the dc--dc converter modules. This results in small 
over-voltage transients at the load and power supply. The typical 
component values listed above are for an output of 50 kV at 400 kW and 
with a transmission line length of 100 feet. 
It will be apparent to those skilled in the art that the output decoupling 
network 170 can be divided evenly and distributed into each dc--dc 
converter module. The function of the decoupling network will not be 
changed. In some applications this is the preferred embodiment. 
It will also be appreciated by those skilled in the art that the outputs of 
the dc--dc converters can be connected in parallel instead of in series. 
This requires the output decoupling network 170 to be placed in the output 
of each converter module to avoid interactions between the outputs of each 
module. 
In accordance with the present device, the inverter IGBTs 140a, b, c, d 
have their on times synchronized with the on times of the IGBTs or 
switching means 124a, b of each voltage regulator circuit 114 in each 
dc--dc converter module 112. The transition of inverter output current 
polarity occurs when one diagonal IGBT pair in the inverter bridge turns 
off and the opposite diagonal IGBT pair conducts. This time occurs at the 
end of conduction of either IGBT 124a or 124b in the voltage regulator 
circuit. This synchronization permits the phasing of the modules described 
below to be realized. 
It will be appreciated that the dc--dc converter modules, shown as sections 
112-1, 112-2, 112-3, . . . 112-n in FIG. 1, comprise identical modular 
power supply circuits which are interchangeable. The inputs to each dc--dc 
converter module 112 are linked in parallel and the outputs are linked in 
series with an adjacent dc--dc converter module 112. 
The IGBTs 124a, b of each voltage regulator circuit 114 and IGBTs 140a, b, 
c, d of inverter circuit 118 in each separate dc--dc converter circuit 
module 112-1 are switched on in a phased relationship with respect to an 
adjacent module 112-2 . . . 112-n. For a plurality of n modules, the phase 
relationship is equal to 360.degree./n. 
In the case of four modular dc--dc converter sections as shown in FIG. 1, 
the activation of each section or module 112-1 . . . 112-4 precedes the 
activation of a successive module by 90.degree.. A 4-phase oscillator 135 
generates the clock pulses separated by 90.degree. to the modules. The 
phased activation of the separate modules 112-1, 112-2, 112-3 and 112-4 
results in a frequency of the overall power supply output voltage ripple 
which is four times higher than the frequency of the module output voltage 
ripple. This permits a four times lower value of capacitance for the 
output filter capacitors 161 and 162 than would be necessary without 
phased activation. 
A feedback control circuit shown generally at 130 in FIG. 1, controls the 
output voltage of the overall power supply in accordance with well known 
techniques. An analog voltage is generated by control circuit 130 based on 
the difference between the desired output voltage, or power supply voltage 
setpoint, and the actual output voltage as measured by voltage divider 
180. This analog voltage becomes the voltage setpoint to another feedback 
control circuit in each dc--dc converter module. As will be explained more 
fully with reference to FIG. 2, this module feedback control circuit 
generates an error voltage which depends on the difference between its 
setpoint and the derived output voltage of each module. This error voltage 
is converted in the module control circuit to an enable signal having a 
variable duty cycle as a function of the error voltage. This enable signal 
is then applied by the control circuit to the enable leads of the IGBT 
transistors 124a, b. The varying of the on and off times of IGBTs 124a, b 
maintains the output voltage from the dc--dc converter 112-1 at a 
controlled level under varying conditions of load. 
A ramp generation circuit 131 is part of the overall power supply feedback 
control circuit 130. The ramp generator slows down the voltage setpoint 
applied to the feedback control summing means to avoid overshoots of the 
power supply output voltage. The ramp generator is needed during step 
increases in power supply voltage setpoint or after the end of the cutback 
interval which occurs because of load arcing. The ramp time is on the 
order of 10 ms for an output power rating of 400 kW. 
The operation of a single modular section, for example, 112-1 is described 
with reference to FIG. 2. After rectification and filtering by the three 
phase input rectifier and filter 100 shown in FIG. 1, a dc voltage of 650 
volts appears across input lines 201 and 202, respectively, as shown in 
FIG. 2. Lines 201 and 202 correspond to nodes 101 and 102, respectively of 
FIG. 1. An input decoupling circuit 216 comprises an inductor 222 in 
parallel with a resistor 220. The input decoupling circuit 216 provides a 
means for maintaining a substantially smooth current with low ripple from 
the output of the 650 volt source connected across nodes 201 and 202. The 
input decoupling circuit 216 also provides a means for preventing 
interactions between the modules 112 which are connected in parallel 
across the 650 volt source. The input decoupling circuit 216 presents a 
relatively high impedance compared to the impedance of the input capacitor 
225 at the regulator switching frequency and above. As a result, it 
effectively forces the high frequency current pulses drawn by the 
regulator to flow from capacitor 225 and not from the input 
rectifier/filter 100 of FIG. 1. This substantially reduces electromagnetic 
interference caused by the power supply since the currents in the cables 
between the input rectifier/filter and the dc--dc converter modules are 
smooth with low ripple. 
The input decoupling circuit 216 also damps out ringing. It will be 
appreciated that the input decoupling circuit 216 permits parallel 
operation of the modules 112-1 . . . 112-4 of FIG. 1 without large 
interactions. The input decoupling circuit 216 provides a means for 
damping the resonance between cables and capacitors of different modules. 
The resistor 220 provides damping to the series resonant network formed by 
the inductance of the cables between modules and capacitance of the input 
capacitors in the different modules. 
Inductor 222 is connected in parallel with the resistor 220 which provides 
a damping means for suppressing oscillation in current in the two circuit 
input lines 201 and 202. The inductor 222 also provides a means for 
limiting the rate of change of the current. Inductor 222 has an input end 
and output end inserted in the circuit line from 201. A capacitor 225 
provides a charge storage means connected across the two circuit lines 201 
and 202 on the output side of the inductor 222 toward the switching means 
224a, 224b. This configuration insures that the line current on the input 
end of the current rate of change limiting inductor increases at a small 
rate from a first value to a high value when the switching means 224a, 
224b of the voltage regulator conduct current. The line current on the 
input end of the current rate of change limiting inductor 222 decreases 
back to a first level at a small rate when the switching means 224a, 224b 
of the regulator are nonconductive. The capacitor 225 absorbs all current 
flow through the current rate of change limiting inductor 222 when the 
switching means 224a, 224b are nonconductive. The capacitor 225 also adds 
to the current through the current rate of change limiting inductor 222 to 
meet the current requirements of the voltage regulator switching means 
when the switching means 224a, 224b are conducting. 
Each dc--dc convertor module also includes a voltage regulator circuit 
means 214 for providing a pulsating voltage of controlled duty cycle from 
an unregulated dc input voltage on lines depending from nodes 201 and 202 
respectively. The voltage regulator circuit means 214 includes a free 
wheeling diode 226, voltage regulator switching means 224a, 224b each 
having a control lead, and emitter lead connected to a first output 
terminal of the unregulated dc input voltage and having a collector lead 
connected to the anode of free wheeling diode 226. The cathode of free 
wheeling diode 226 is connected through the input decoupling network 216 
to a second terminal of the unregulated dc input. 
Inductor 228 acts as a means for smoothing current having a first end 
connected to the common node of the voltage regulator switching means 224a 
and 224b and to the anode of free wheeling diode 226. The inductor means 
228 provides a means for filtering the pulsating voltage created by the 
switching means 224 and creates a smooth direct current for application to 
the inverter circuit means 218. 
Referring to FIG. 2, the inverter circuit 218 provides a means for 
converting the direct current from inductor 228 into a high frequency 
square wave current in the primary 252 of transformer 250. The inverter 
circuit 218 in a preferred embodiment comprises first and second parallel 
circuit branches. The first circuit branch comprises IGBTs 240a and 240b, 
respectively, connected in series. A second branch comprises IGBTs 240c 
and 240d, connected in series. Both circuit branches are connected in 
parallel to form a bridge network. The IGBTs act as a switching means, 
each having a collector lead, a control lead and an emitter lead. The 
IGBTs 240a and 240c each have their collectors connected to the positive 
inverter bus 241a which supplies a source of direct current. The emitter 
leads of IGBTs 240b and 240d are each connected to the negative inverter 
bus 241b which returns the direct current to inductor 228. 
Also connected in parallel with the inverter circuit 218 between positive 
bus 241a and negative bus 241b are the series connected capacitor 243 and 
diode 242 which, combined with resistor 244, form the inverter clamping 
means which is explained later. The primary winding 252 of transformer 250 
is connected across the output leads of the inverter. The four IGBTs are 
alternately activated in diagonal pairs (IGBTs 240a, 240d and IGBTs 240b, 
240c) by voltages applied to their control leads such that each output 
lead of the inverter is alternately connected to positive and negative 
buses 241a, 241b of the inverter circuit 218. A short overlap time of 
approximately two microseconds occurs at each polarity transition where 
all four IGBTs are conducting. A substantially square wave alternating 
current is developed through the primary 252 of the transformer 250. 
Referring to FIG. 3, the inverter clamping network formed by a 2 microfarad 
(.mu.F) capacitor 343, diode 342, and 8 ohm resistor 344 functions to 
limit the voltage between the positive and negative buses which feed the 
inverter, and thereby limit the voltage across the IGBT switches, during 
change in polarity of the inverter output. FIG. 3 shows a portion of the 
inverter and regulator power circuit driving an equivalent representation 
of the step up transformer load. Inductor 355 represents the transformer 
leakage inductance and voltage source 356 represents the secondary voltage 
reflected to the primary. The magnitude of voltage source 356 is 
substantially equal to the magnitude of the module dc output voltage 
divided by the transformer turns ratio. The polarity of voltage source 356 
is as shown in FIG. 3 when the direction of current i.sub.p is as 
indicated in FIG. 3. It changes to the opposite polarity when the current 
i.sub.p changes to the opposite direction. 
The principles of the inverter clamping network are best described by 
referring to the waveforms during transition of polarity of the inverter 
output as shown in FIG. 4. For an inverter output frequency of 10 kHz, 
these waveforms occur every 100 microseconds during transition from 
positive to negative inverter output polarity. The same waveforms but of 
opposite polarity also occur every 100 microseconds during transition from 
negative to positive inverter output. The transitions are spaced 50 
microseconds apart to create the substantially square current waveforms 
out of the inverter. 
Referring to FIG. 4, prior to time t.sub.a IGBT switches 340a and 340d are 
on and conducting and transistor switches 340b and 340c are off and 
nonconducting. The current I.sub.L in inductor 328 flows through switches 
340a and 340d and through the equivalent load formed by the inductor 355 
and voltage source 356. At time t.sub.a, switches 340b and 340c are also 
turned on which results in zero voltage across the equivalent load and the 
beginning of the decay of current i.sub.p to zero which occurs at time 
t.sub.b. Between times t.sub.b and t.sub.c, all four switches 340a, 340b, 
340c, and 340d are conducting current of magnitude I.sub.L /2. At time 
t.sub.c, switches 340a and 340d are turned off and they become 
nonconducting. The current I.sub.L in inductor 328 is now forced to flow 
through the capacitor 343, which is precharged to 650 V through resistor 
344, and diode 342. This applies 650 V across the equivalent load which 
results in the buildup of current i.sub.p in the negative direction. The 
voltage across capacitor 343 and the current i.sub.p through the 
equivalent load continue to increase until the magnitude of i.sub.p 
reaches the value I.sub.L at time t.sub.d. At this time, t.sub.d, the 
diode 342 becomes reverse biased and capacitor 343 begins discharging back 
to 650 V through resistor 344 in preparation for the next polarity 
transition one half cycle later. 
It will be appreciated that an alternate current path is provided for 
current flow as current builds up from zero in the inductive load 355. A 
capacitor 343 is connected from one of the two input circuit lines and one 
lead of a voltage source capacitor 325 to a circuit node between a 
resistor 344 and diode 342, so that as current builds up from zero to a 
maximum value, excess current flow is from one of the two input circuit 
lines through the capacitor 343 and diode 342 to the other of the two 
input circuit lines. Voltage across the two circuit lines is clamped to 
approximately the voltage of the voltage source 325. This inverter 
clamping network consisting of capacitor 343, diode 342, and resistor 344 
is superior to known prior art because the fast varying currents occur 
only within capacitor 343, diode 342, and the inverter IGBT switches 340a, 
340b, 340c, and 340d. Capacitor 343 and diode 342 are located very close 
to the inverter IGBTs which minimizes the stray inductance of the loop 
formed by the capacitor, diode, and IGBTs. As a result, the voltage 
transient spikes appearing across the IGBTs, which are created by this 
inductance and the fast change of currents, are minimized. 
Referring again to FIG. 2, the inverter circuit also includes a means for 
simultaneously turning on all of the IGBT switch means 240a-240d for 
shorting and thereby isolating the primary 252 of transformer 250 when an 
incipient gun arc is sensed. This minimizes the amount of power supply 
current which passes through the load during load arcs. 
It will be apparent to those skilled in the art that snubber networks, each 
consisting of a resistor, diode, and capacitor, may be needed in 
conjunction with each IGBT in the voltage regulator 214 and inverter 218. 
These snubber networks are not shown in FIGS. 1, 2, and 3 for reasons of 
clarity. Designs of snubber networks for IGBTs are well known. 
A one turn voltage sense winding 255 is wound on the same transformer core 
as primary 252 and secondary 254. Voltage sense winding 255 provides a 
means for sensing an ac voltage proportional to the ac voltage across the 
secondary 254 of transformer 250. The ac voltage induced in sense winding 
255 is then rectified and filtered by an output simulator means 290. In 
addition, the output current of the module is sensed by the dc current 
sensor 280 and used to control a current source in parallel with the 
filter capacitor 293 in the output simulator 290. The current source 
represents the current source characteristic of an ion source, and 
specifically an electron beam gun. The capacitance value of filter 
capacitor 293 and the value and range of current source 292 are scaled so 
that the output of the output simulator 290 accurately represents the 
voltage level and circuit dynamics of the output of the dc--dc converter 
module and is consistent with the voltage levels used in the feedback 
control circuit. Implementation of the output simulator 290 is done with 
well known electronic circuit techniques. Use of the one turn sense 
winding 255 and the output simulator permit generation of the module 
output voltage feedback signal without requiring high voltage isolation 
circuitry which would be needed if the module output voltage was measured 
directly. 
The output simulator 290 comprises a diode rectifier means 291 responsive 
to the ac voltage produced by the sense winding 255 to produce a dc 
voltage on two output lines. This dc voltage is proportional to the dc 
output voltage of the power supply module 112. The output simulator means 
290 also includes a filter capacitor 293 connected across the two output 
lines from the rectifier means 291. A controlled current source means 292 
is also connected across the two output lines for applying the output 
voltage of the current sensing means 280 to produce a current proportional 
to the output current of the power supply module. The controlled current 
source means 292 draws current out of the two output lines to create an 
accurate simulation of power supply module output voltage. 
The output of the output simulator 290 is then applied to a voltage summing 
means 266. In the summing means, voltage from the output simulator 290 is 
subtracted from a set point voltage. The set point voltage is a reference 
voltage for controlling the amplitude of the dc--dc converter output 
voltage. The set point voltage enables the amplitude of the converter 
output voltage to be controlled to a desired level. The output of the 
voltage summing means 266 is a difference voltage, that is, the difference 
between the set point voltage and the voltage output simulator. 
The difference voltage is amplified and filtered in accordance with known 
techniques in the compensator 268. The amplified voltage from the 
compensator 268 is then compared to a clock generated sawtooth voltage in 
a comparator 270. The sawtooth voltage is generated by a sawtooth 
generator means 272 and is synchronized to the clock pulse of the system. 
The output of the comparator is a logic level enable signal (+5 V for 
example) which has an enable and a disable state. The output of the 
comparator 270 and the clock logic signal feed the regulator logic and 
driver circuit 274 which distributes the enable signal to the regulator 
IGBTs 224a and 224b. Logic and driver circuit 274 also provides the means 
for alternately enabling IGBTs 224a and 224b. This logic and driver 
circuit 274 is implemented using well known electronic techniques. The 
enable signal has a variable ratio of the time during which the voltage is 
high to the time during which the voltage is low as a function of the 
error voltage output of compensator 268. The enable signal is then applied 
to the enable leads of the IGBT switch means 224a and 224b of the voltage 
regulator circuit 214. This provides dynamic regulation of the dc voltage 
on the output of the power supply in accordance with varying loads being 
sensed by the output simulator 290 and sense winding 255. 
In accordance with another aspect of the present power supply, a dc current 
sensor means 280 is provided on the negative output lead of the power 
supply. There, the sensed output current is compared with a predetermined 
threshold indicative of current conditions during an incipient gun arc. 
The sensed current and current threshold are compared in a voltage 
comparator 282. The voltage comparator produces an output signal when 
current above a predetermined threshold is sensed and an incipient gun arc 
is present. This in turn sends a signal to a cutback timer 284 which 
generates a pulse ranging from 50 ms to 200 ms and which in turn activates 
the inverter logic and driver circuit 286. When a gun arc is sensed, the 
inverter logic and driver circuit 286 simultaneously activates all four 
IGBTs 240a-240d which shorts out the primary 252 and thereby terminates 
current let through from the primary side to the load side. The cutback 
timer 284 also simultaneously sends a signal to the regulator logic and 
driver circuit 274 which turns off the regulator switch means 24a and 
224b. 
While the invention has been described in connection with what is presently 
considered to be the most practical and preferred embodiments, it is to be 
understood that the invention is not limited to the disclosed embodiment, 
but on the contrary, is intended to cover various modifications and 
equivalent arrangements included within the spirit and scope of the 
appended claims.