Signal-to-noise improving system

A signal-to-noise improving system is described which comprises a circuit input for incoming noisy analogue signals and a circuit output for digitally stored input signals which have an improved signal-to-noise ratio provided by the system and which have been reconverted to analogue form, PA0 said circuit input and said circuit output being connected to inputs of an analogue comparator arranged to give an output which signifies that the stored signal is either higher or lower in magnitude than the incoming signal or that the incoming signal is either higher or lower in magnitude than the stored signal, PA0 said comparator output being connected to a signal incrementor which is arranged to give a signal output which is the stored digital signal incremented higher or lower by a number digitally in response to either a higher or lower signal output from said comparator, PA0 a store for storing in digital from the so incremented input signals, the store output being connected to a digital to analogue converter 1 the output of which is connected to said circuit output, PA0 said comparator, said incrementor, said store and said digital to analogue converter 1 being operative cyclically to compare the incoming noisy signals with the stored analogue output signals and to up date the stored signals to new stored signals determined by adding or subtracting a number digitally from the stored signals in accordance with whether said comparator, comparing the analogue input and output signals gives a higher or lower output whereby to eventually store signals representative of the incoming signals with enhanced signal-to-noise ratio so that said circuit output can provide an output signal of those enhanced stored signals. Preferably there is provided an incrementor controller, for controlling the incrementation of said incrementor, and wherein, in use, said incrementor initially, increments in a series of increments which are similar to those of a successive approximation analogue to digital converter whereby to provide for rapid convergence to a signal value near the mean value of the analogue input signal.

BACKGROUND OF THE INVENTION 
1. Field of the Invention 
This invention relates to a signal store with a signal-to-noise improving 
system which has particular, but not exclusive application, where still 
picture television video frame signals are to have the signal-to-noise 
ratio improved. A preferred embodiment of the invention has particular 
application in scientific environments, wherein a video signal involving 
still pictures having an inherently noisy nature can be improved. 
2. Description of Prior Art 
In scientific applications noise has hitherto been reduced in video picture 
frame signals by either a summing technique involving averaging a number 
of T.V. frames in order to suppress non-coherent signal components. The 
improvment in the signal-to-noise is proportional to the square root of 
the total number of frames involved in the averaging process. The 
necessary electronic hardware used to perform this method, if a 
considerable signal-to-noise improvement is to be obtained, requires that 
the memory be large and thus the resulting cost of the equipment is 
generally prohibitive. For example in a system using a gray-scale 
resolution of 8 bits, a signal-to-noise enhancement of 40 dB would require 
a memory size based on at least 21 bits per picture element. 
A further method of reducing the signal-to-noise ratio has been by 
exponential smoothing: where an exponentially weighted moving average of A 
frames yield an ultimate signal-to-noise ratio improvement of .sqroot.2A-1 
and allows a normalized image to be displayed while the signal averaging 
is progressing. This is referred to in "smoothing, forecasting and 
prediction of Discrete Time Series, by R. G. Brown, Prentice Hall, (1963) 
chaps. 7 and 8". 
Theoretically the best result (in terms of both signal-to-noise improvement 
rate and ultimate value) which can be expected in any form of filtering 
technique is given by the summing algorithm i.e. enhancement=.sqroot..eta. 
where .eta. is the number of the frames. 
For a system operating according to the summing technique it is a simple 
matter to calculate the memory size (i.e. bits per pixel) required to meet 
specific performance criteria. For example if a (voltage) enhancement 
factor of (say) 90 is required and the system gray-scale resolution is 8 
bits then the size of the memory will be based on 21 bits per pixel and 
the total accumulation time (625/50 system) is approx. 6 minutes. If, in 
addition, a digital signal normalizer is required to produce a continuous 
display during the signal averaging process then the total system's 
hardware complexity would be considerable. 
STATEMENT OF THE INVENTION 
Accordingly, we have devised a system to attempt to overcome these 
problems. 
A preferred embodiment of the invention does not use an expensive video 
signal analogue to digital converter and it provides a continuous 
normalized video output during the process. Further it achieves a 
(voltage) enhancement ratio equal to 
.sqroot.2/.pi..sqroot..eta..apprxeq.0.8.sqroot..eta. and makes more 
efficient use of memory in terms of the ultimate enhancement ratio (per 
memory bit). 
Therefore in accordance with one broad aspect of the present invention 
there may be provided a signal-to-noise improving system comprising, a 
circuit input for incoming noisy analogue signals and a circuit output for 
digitally stored input signals which have an improved signal-to-noise 
ratio provided by the system and which have been reconverted to analogue 
form, 
said circuit input and said circuit output being connected to inputs of an 
analogue comparator arranged to give an output which signifies that the 
stored signal is either higher or lower in magnitude than the incoming 
signal or that the incoming signal is either higher or lower in magnitude 
than the stored signal, 
said comparator output being connected to a signal incrementor which is 
arranged to give a signal output which is the stored digital signal 
incremented higher or lower by a number digitally in response to either a 
higher or lower signal output from said comparator, 
a store for storing in digital form the so incremented input signals, the 
store output being connected to a digital to analogue converted the output 
of which is connected to said circuit output. said comparator, said 
incrementor, said store and said digital to analogue converter being 
operative cyclically to compare the incoming noisy signals with the 
analogue output signals generated from the stored digital signals and to 
up date the stored signals to new stored signals determined by adding to 
or subtracting from the stored signals a digital number in accordance with 
whether said comparator comparing the analogue input and output signals 
gives a higher or lower output whereby to eventually store signals 
representative of the incoming signals with enhanced signal-to-noise ratio 
so that said circuit output can provide an output signal of those enhanced 
stored signals. 
It is preferred that the system components are of a size digitally to 
process each of the picture element signals in a frame of a video picture 
image and wherein each of the said picture element signals is assigned 
with a respective N-bit digital word by said incrementor. 
It is also preferred that said incrementor will, in use, increment by small 
levels, such that eventually the stored signal will `hunt` about a mean 
value of the analogue input signal. 
It is also preferred that there be an incrementor controller, for 
controlling the incrementation of said incrementor, and wherein, in use, 
said incrementor initially, increments in a series of increments which are 
similar to those of a successive approximation analogue to digital 
converter whereby to provide for rapid convergence to a signal value near 
the mean value of the analogue input signal. 
It is also preferred that said incrementor, in use, is caused to increment 
initially by a value corresponding to the most significant bit and then 
increments successively to the least significant bit. 
It is also preferred that said small level of incrementation is 
controllable by said incrementor controller according to a predetermined 
sequence based on a prior knowledge of the signal-to-noise ratio 
contamination such that during commencement of said small level of 
incrementation, the incrementation will be approximately equal to 
.sqroot.2/.pi..times.R.M.S. input noise voltage, and will be reduced to 
approximately .sqroot.2/.pi./A where A is the number of times the picture 
element is incremented. 
It is also preferred that the digital number of the N-bit word is greater 
than the gray-scale resolution of each picture element of the frame and is 
also greater than the resolution of said digital to analogue converter. 
It is also preferred that each of the picture elements is incremented by 
the same amount during that frame.

DESCRIPTION OF PREFERRED EMBODIMENT 
The preferred system includes an incrementor controller in order to 
accelerate the convergence of the stored signal to a substantially noise 
free signal. 
The preferred circuit comprises (a) An N-bit digital to analogue converter 
1 (b) A comparator 2 having two analogue inputs and a 1 bit (0 or 1) 
digital output (c) An incrementer 3 which generates the sum or difference 
S of two M-bit input words (L and .DELTA.) according to a binary "sign 
bit" input signal. It should be noted that M will always be greater than 
N) (d) A digital memory 4 representing the frame store and whose capacity 
in bits is given by M times the (total number of picture elements) and (e) 
An incrementor controller 5 which presets the magnitude of the 
incrementation .DELTA. at the beginning of each T.V. frame according to a 
predetermined algorithm. The increment .DELTA. is the same for all words 
in a single frame. 
In use, the digital frame store 4 is read at the T.V. scan rate and the N 
most significant bits converted to analogue form in the analogue converter 
1 to form a restored analogue output signal. This analogue output signal 
is also presented to one input of the comparator 2 whose other input is 
the incoming (noisy) T.V. video signal. In this way a sign bit of the 
difference between the stored picture element value and the corresponding 
picture value of the input signal is formed at a frame clock period for 
incrementing each picture element. This sign bit (SGN) is then used to 
determine the sense in which the stored picture element value L is updated 
by the increment value .DELTA. to form (S)--the overall effect being to 
converge each stored picture element value digitally towards the 
corresponding input picture element value but with reduced noise. It 
should be noted that all stored picture values are updated once each frame 
period. During the process, in order to achieve a rapid convergence of the 
stored signal to a close replica of the input signal but with reduced 
noise, the increment size .DELTA. can be stepped down successively after 
each frame period from the most significant bit--the highest number stored 
divided by 2. In this process each M-bit picture element cell of the frame 
store may be likened to the register of a successive approximation 
analogue to digital converter which is clocked at the frame rate. During 
the final stage of the process, smaller values of .DELTA. will apply over 
many frame periods. It can therefore be seen that the size of word 
increment .DELTA. is reduced so the influence of any random noise 
contaminating the input signal will be reduced. Thus any random noise 
contaminating the input signal will progressively have a less and less 
effect on the value of the stored signal. The maximum degree of noise 
reduction is finally obtained when the value of .DELTA. is equal to one 
least significant bit of stored number in the store 4. Therefore it can be 
seen that the process has an effective integrating function on the value 
of each stored picture element. Therefore, when the value of .DELTA. 
reaches a low value the stored picture element number in the store will 
`hunt` about a corresponding noise free input picture element value. 
Ultimately, when the desired degree of image enhancement has been achieved 
(for example by observing the output signal on a picture monitor) the 
incrementing process is terminated and the contents of the frame store 
frozen. 
In practice for a high quality imaging system based on the 625/50 T.V. 
standard, a raster of (typically) 512.times.512 picture elements is 
required for the frame store and 256levels (N=8) are necessary for an 
accurate gray-scale rendition. 
A mathematic analysis of a quantitative model of the pixel incrementing 
mechanism for input signals contaminated by stationary gaussian noise will 
now be provided. The mathematical treatment is intended to be plausible 
rather than rigorous, and it is to yield expressions which quantify the 
behaviour of the system in the engineering sense. 
Referring again in FIG. 1 consider a (still) video input signal comprising 
a wanted signal component masked by a noise component whose RMS value is 
.nu..sub.in. Although the real time signal sequence formed by looking 
along a T.V. line at spacially adjacent pixels may be a Markoff process 
due to band limiting or frequency weighting of the input noise signal it 
can confidently be said that the signal sequence formed by looking along 
temporally adjacent values at any particular pixel will be purely random 
(i.e. in all practical cases the autocorrelation function of the 
contaminating noise will be assumed zero for times equal to or greater 
than the T.V. frame period). Let V.sub.out (t) be the RMS value of the 
output (stored signal) at any time t. 
For the moment let us consider the case N=M. 
Two distinct phases of the convergence process for the ensemble of pixels 
constituting the T.V. frame will be given: 
(i) Outer convergence. 
This is the initial phase of the convergence process during which each 
pixel cell of the frame store behaves like the register of a successive 
approximation analogue-to-digital converter (i.e. during the first frame 
the value of .DELTA. is set to the value of the most significant bit; 
during the second frame the value of .DELTA. is set to the value of the 
2nd most significant bit and so on). The aim of this phase is to allow the 
stored signal to achieve a rapid approximation to the input signal. 
Obviously in the absence of noise (.nu..sub.in =0) the outer convergence 
process alone would be sufficient to give us the desired result. For 
.nu..sub.in .noteq.0 we shall assume that the value of .nu..sub.out at the 
completion of the outer convergence process shall be approximately equal 
to .nu..sub.in. An accurate treatment of this phase is difficult because 
the initial conditions as determined by arbitrary picture content are 
difficult to define for the ensemble of pixels that constitute a complete 
T.V. frame. In practice the above approximation was found to be 
conservative as the typical result for a wide range of picture contents 
gave (empirically): 
##EQU1## 
(where .tau.=T.V. frame period) 
(ii) Inner convergence. 
In this second phase of the convergence process the smaller values of 
.DELTA. (culminating in .DELTA. equal to one least significant bit) are 
used to allow each pixel value to approach the desired average value. The 
initial conditions for inner convergence shall be assumed to be: 
##EQU2## 
The total time required to process a noisy T.V. signal is therefore the sum 
of the times taken for the outer and inner convergence processes i.e. 
##EQU3## 
In a typical case we may be seeking a noise improvement factor of greater 
than 10. Also, the pixel word size will be typically between 6 and 12. We 
know that the shortest total convergence time will be greater than that 
given by the theoretical limit as set by the summing algorithm .sqroot.N 
##EQU4## 
From this it is evident that the magnitude of N dominates the total 
convergence time and for practical purposes we may make the approximation: 
EQU T.sub.TOTAL .apprxeq.T.sub.INNER 
In order to characterize the inner convergence behaviour the purpose of the 
following analysis shall serve to establish: 
(i) The algorithm giving the magnitude of .DELTA. (as a function of time) 
to ensure the fastest convergence. 
(ii) The enhancement ratio as a function of time using the above algorithm 
for .DELTA.. 
(iii) The ultimate (limiting value) of enhancement ratio. 
Considering, again, the behaviour at any particular pixel cell. Let V be 
the mean value of the input signal at that pixel (for still pictures the 
value of V is a constant). Let m.DELTA. (m is an integer) be the deviation 
of the stored pixel value from V at time t=n.tau. (i.e. after n frame 
periods). Note that the value .DELTA. is quantized due to its digital 
origin. Let .nu..sub.o be the stored pixel value at an arbitary time 
origin n=0. Once each frame period a decision is made by the comparator 
causing the stored pixel value to be increased or decreased by the 
increment .DELTA.. Clearly this behaviour constitutes a Markoff process 
for which the probability of moving either higher or lower depends on the 
deviation of the output voltage from V at the time of decision i.e. on the 
value of m.DELTA.. For m=0 the probability of an up movement is equal to 
that of a down movement, while for m.noteq.0 the probabilities are 
weighted to favour a movement towards m=0. More precisely, the probability 
of a down movment is given by 
##EQU5## 
and the probability of an upward movement is given by 
##EQU6## 
We may expect that the statistical properties of m.DELTA. are completely 
described by the second order conditional probability function P.sub.2 
(s.DELTA./m.DELTA.;n.tau.) (i.e. the probability that the pixel voltage 
takes on a value m.DELTA. after n frame periods given that m=s at time 
n=0). 
The RMS output noise voltage .nu..sub.out (t) is obtained by scanning an 
ensemble (constituting a complete T.V. frame) of such pixel voltages each 
satisfying these statistics .nu..sub.out.sup.2 (t) may therefore be 
equated with the variance of m.DELTA.. 
Unfortunately we were unable to find an exact solution to this problem due 
to the non-linear nature of .rho. and . However, an approximation 
applicable to the particular situation enables us to find the "engineering 
solution" we seek. 
Bearing in mind the initial condition for the inner convergence process 
i.e. .nu..sub.out (0)=.nu..sub.in and letting the inequality 
.DELTA.&lt;&lt;.nu..sub.in apply, we shall assume that the excursions of the 
ensemble of voltages m.DELTA. are confined to the essentially linear 
region of the probability function for .rho. given by (1) near m=0 (see 
FIG. 2). If a solution can now be found then such a solution will in 
itself be a test for the validity of the above approximation. The 
proportionality between .rho. and m.DELTA. near m=0 is found from (1) to 
be: 
##EQU7## 
Hence the probability of a down movement at m.DELTA. may now be 
approximated by: 
##EQU8## 
and the probability of an upward movement by 
##EQU9## 
The approximation has thus achieved a simplified problem formulation which 
is now seen to be identical to that of a discrete one dimensional random 
walk of an elastically bound particle. A detailed solution of this problem 
is given in M. Kac "Random Walk & Theory of Brownian Motion" Am. Math. 
Monthly, 14:369 (1947). According to this formulation the probabilities of 
the voltage moving down or up at each decision instant are 
##EQU10## 
respectively. This leads to a difference equation for the conditional 
probability whose solution is shown to be: 
##EQU11## 
Although this is the correct solution to our discrete random walk model 
involving quantized voltage levels the format of (6) does not readily lend 
itself to an interpretation of the behaviour of the variance of m.DELTA.. 
A more convenient form of the solution is the continuous case which may be 
derived from (6) by letting 
##EQU12## 
.nu. is now a continuous approximation to the discrete variable m.DELTA. 
and for which the second order probability density function is found to be 
(see Kac, Supra and an introduction to statistical Communication Theory, 
D. Middleton, McGraw Hill (1960) pp 438-466): 
##EQU13## 
the validity of (10) being subject in our situation to the conditions: 
##EQU14## 
Thus, at any pixel, the deviation of the output voltage from the mean input 
voltage (V) at that pixel is seen to have a gaussian distribution whose 
variance approaches a final value D/.gamma. with a time constant 
1/2.gamma.. The initial .sigma. value of zero corresponds to our knowledge 
that .nu.(t=0)=0 with a probability of 1. 
The identical result (10) is obtained by solving the first order Langevin 
equation which describes the physical process governing the behaviour of 
the ensemble of pixel voltages all subject to identical statistics and all 
subject to the initial condition .nu.(t=0)=0 
##EQU15## 
In the analysis that follows, the convergence process shall be analysed in 
terms of the power ratio: 
##EQU16## 
where .nu..sub.in is constant. 
Notice from equation (10) that the value of .sigma..sup.2 diverges from an 
initial value zero to its final (stationary) value 
.sigma..sub..infin..sup.2. In the convergent situation the initial value 
of .sigma..sup.2 (=.sigma..sub.o.sup.2) will be greater than 
.sigma..sub..infin..sup.2. Because the stochastic differential equation 
(12) for the process is linear we can expect .sigma..sup.2 to approach 
.sigma..sub..infin..sup.2 with the same time constant (1/2.gamma.). 
Thus in the convergent case: 
##EQU17## 
Combining (3), (5) and (14) and observing the proportionality between 
.sigma..sup.2 and .rho. we find: 
##EQU18## 
Furthermore if we apply the initial condition .rho..sub.0 =1 for inner 
convergence we have: 
##EQU19## 
Notice that the ultimate value of Voltage enhancement ratio for a given 
value of .DELTA. is: 
##EQU20## 
Let us examine the convergence behaviour in more detail with the help of 
equation (18). A conflict in the choice of .DELTA. is immediately 
apparent. If we make .DELTA. as small as possible in order to achieve a 
good ultimate reduction ratio .rho.(t.fwdarw..infin.) the convergence rate 
is slow. Conversely, if we aim for a faster convergence rate by choosing a 
higher value of .DELTA. the ultimaate power reduction ratio suffers. 
Intuitively we may anticipate an optimum performance 
.rho.(t)=.rho..sub.opt (t) by continuously (within quantization 
constraint) reducing the value of .DELTA. according to some predetermined 
algorithm .DELTA..sub.opt (t). 
The function .DELTA..sub.opt (.rho.) may be found by determining the values 
of .DELTA. which will give the steepest slope at all points along the 
curve .rho.(t). We know that at any time t.sub.1, the slope is equal to 
the gradient of the function (cf (15)); 
##EQU21## 
and the maximum gradient at t=t.sub.1 occurs when (differentiating (21) 
w.r.t..DELTA.) 
##EQU22## 
but (22) is true for all values of t.sub.1, 
##EQU23## 
Substituting (23) in (21) we obtain: 
##EQU24## 
whose solution when subject to the initial condition 
##EQU25## 
Alternatively, expressed in terms of the number of frames (n) processed 
(25) becomes: 
##EQU26## 
The voltage enhancement ratio is then: 
##EQU27## 
which compares favourably with the theoretical limit as set by the summing 
algorithm: 
##EQU28## 
Notice that according to equations (25), (26) and (27) the enhancement 
ratio would increase ad infinitum with increasing time. Obviously the 
maximum value of voltage enhancement ratio corresponding to the smallest 
value of .DELTA. is determined by the smallest quantizing step (as 
determined by the value of M). From (19) this asymptotic value is seen to 
be: 
##EQU29## 
In summary the optimum convergence process in terms of voltage enhancement 
ratio is seen initiaally to follow a quadratic law according to (27) until 
a "breakpoint" value is reached and thence to asymptote to a value defined 
by (29). The number of frames taken to reach the breakpoint value may be 
considered as an index for the conversion rate for a particular value of 
.nu..sub.in and is given by: 
##EQU30## 
So far we have considered only systems for which all bits of the pixel word 
are converted to an analog signal to close the feedback loop at the 
comparator input. The function of the Digital-to-Analogue converter is one 
of ensuring a proportionality between the stored pixel values and the 
feedback component. The speed requirements on the Digital-to-Analogue 
converter are quite stringent and some hardware simplification may be 
achieved by using a converter of reduced resolution (i.e. N&lt;M) provided 
that N is sufficiently large for the gray-scale requirements of the system 
to be met. 
The effect of truncating the stored pixel word by omitting some of the less 
significant bits in the conversion process is discussed below. 
Obviously for input noise levels less than the smallest resolvable step of 
the converter (M-N).DELTA..sub.min we would expect no enhancement 
whatsoever. On the other hand for input noise levels much greater than 
(M-N).DELTA..sub.min we would expect little performance degradation due to 
the coarser conversion quantising as long as the residual output noise 
level was much larger than (M-N).DELTA..sub.min. Intuitively it would seem 
that the degradation in enhancement would not be seriously affected 
(irrespective of the input level) until the residual output level was 
comparable with (M-N).DELTA..sub.min. Most practicle situations (as 
discussed hereinafter) allow the value N to be determined solely by the 
gray-scale resoltuion requirements of the system. In such cases it has 
been empirically found that the lower limit on absolute output noise level 
is comparable in magnitude to the quantising noise for an N bit system. 
The function of the incrementer controller 5 is to generate the appropriate 
sequence of .DELTA. values for correct outer convergence and optimum inner 
convergence. It will be remembered that for the outer convergence the 
sequence for .DELTA. is: 1st frame--MSB, 2nd frame--2nd MSB and so on. 
The optimum inner convergence process commences with a .DELTA. value (see 
equation (23)) equal to 
##EQU31## 
Thereafter the value of .DELTA. must be varied according to equation (23), 
##EQU32## 
It has been found that a quite coarse discrete approximation 
.DELTA..sub.opt (n) to (32) exists which represents a considerable 
hardware saving while at the same time causing negligible impairment to 
the convergence rate. 
Bearing in mind that the outer convergence process involves preferred 
values of .DELTA. corresponding to discrete bit magnitudes, the 
convenience of using the same preferred values of .DELTA. for the inner 
convergence process is apparent. Using this approach the inner convergence 
behaviour for .rho.(t) would take the form of a discrete sequence of 
exponential decays. 
The discrete sequence .DELTA..sub.opt (n) approximating the curve 
.DELTA..sub.opt (n) may be tabulated thus: 
##EQU33## 
Substituting these values for .DELTA. in (15) and remembering that 
t=n.tau. we obtain the relationship between the power ratio at the 
beginning (.rho..sub.Q) and the end (.rho..sub.Q+ 1) of the Q th 
exponential decay section: 
##EQU34## 
where the frame number n=2.sup.Q -1 It can be easily shown that the 
convergence process as defined by equation (34) is a good approximation to 
.rho..sub.opt (n) as per equation (26), the error being less than 5% over 
the range of interest. The asymptotes are of course the same in both cases 
(being defined by (29)). 
It is evident from equation (23) that the choice of the optimum convergence 
algorithm depends on the input noise level as this determines the initial 
conditions of the inner convergence process. The apparatus desirably 
therefore has a control for selecting the best algorithm to span a wide 
range of input noise levels. With the algorithm as given by .rho..sub.opt 
(n) (26) such a control would be continuous and thereby allow an optimum 
matching of the algorithm to the input noise level. 
With the algorithm as given by .rho..sub.Q (34) we no longer have a 
continuous control due to the preferred fixed values of .DELTA..sub.1. The 
control in this case is a geometric series with adjacent settings 
differing by a facter of 2. In practice this does not lead to significant 
performance degradations as the signal-to-noise ratios in typical 
operational situations are not accurately known anyway. Ideally the chosen 
setting would put .DELTA..sub.1 as close as possible to the value 
##EQU35## 
of the particular signal to be processed. A preferred filter is shown in 
FIG. 3 and follows the block schematic of FIG. 1. 
Such filter is suitable for the 625/50 T.V. system. The instrument is 
designed primarily for the scientific market and comprises a square 
512.times.512 pixel frame store matrix. 
A design goal requiring noise voltage enhancement ratios in excess of 100 
(for appropriately large input noise levels) dictates a pixel word depth 
of 12 (M=12), whereas a gray-scale resolution of 8 bits per pixel word was 
considered adequate to meet the needs of most applications. 
A pixel word duration of 69 ns places some quite critical performance 
criteria on the pixel incrementing circuitry. 
Each 12-bit pixel word has to be retrieved from the memory, D/A converted, 
compared with the incoming video signal and modified by .DELTA.(another 12 
bit word) within a 69 ns time-slot. No sufficiently fast 12-bit D/A 
converter was available at the time of design and the choice thus fell on 
the Motorola chip MC10318--an 8-bit (i.e. N=8) device with a settling time 
of around 10 ns. A suitable voltage comparator was found in the AMD 685--a 
6 ns latched device. All digital operations associated with the 
incrementor have been designed in ECL logic. 
The only cost effective type of memory device was the 16k dynamic RAM whose 
read--modify--write cycle time is typically 375 ns. The data rate 
commensurate with a 69 ns pixel duration has been achieved with a 
stagger-phased combination of 8 such memory chips. The incrementor 
controller is implemented according to the .rho..sub.Q algorithm (see 
equation (34)) with .DELTA..sub.1, values selectable from MSB down to LSB 
(the latter giving purely an outer convergence process). The corresponding 
values of video input signal-to-noise ratio catered for (in terms of 
optimum convergence times) thus range from 4 dB in 3 dB steps up to 37 dB. 
Signal-to-noise ratios less than 8 dB are of course also capable of being 
processed but with sub-optimum convergence times. 
The temporal filter has been designed with a front-end video signal 
processor capable of providing a large range of gain and level shifts. 
The detailed circuit description is shown in FIGS. 3 to 7 and and are as 
follows. The overall circuit is shown in FIG. 3. 
The circuit has the following features: 
Instrument controls have been provided to give the equipment the following 
facilities: 
1. "Integration Mode" (switch S1 FIG. 6A) controls the incrementer 3 such 
that in the "Peak" mode, only positive increments are recognised and 
processed thus allowing an irreversible build up of brightness of an image 
being processed. The normal position of this switch is the "Mean" position 
whereby the incrementer 3 operates as has been described so far. 
2. "Step Size" (switch S2 FIG. 6B) allows the selection of a particular 
increment size. Also the last position of the switch enables one of eleven 
fast convergence algorithms according to the setting of S3 (see item 3 
which follows. 
3. "Integration Time" (switch S3 FIG. 6B) sets the initial increment size 
according to the a priori knowledge of the input signal-to-noise ratio and 
thereby determines the total time taken to complete the convergence 
algorithm. 
4. "Video Polarity" (switch S4 FIG. 6A) enables the stored video signal to 
be inverted to achieve a "negative" display effect. 
5. "Display Mode" (switch S5 FIG. 6A) allows the selection of the output 
video between input only (Direct) stored only (Stored) and stored blended 
into input (Insert). 
6. "Field Select" (switch S6 FIG. 6C) allows the selection of each TV field 
(i.e. half the total memory) for display as a complete TV frame. 
7. "Input Set-Up" (switch S7 FIG. 6A) affects a selectable DC shift of the 
input processor. 
8. "Input Gain" (switch S8 FIG. 6A) affects a selectable gain of the input 
processor. 
9. "Reset" (momentary switch K3 FIG. 6C) sets all pixel locations of the 
memory to black level. 
10. "Start"(momentary switch K2 FIG. 6C) initiates the incrementing 
process. 
11. "Hold" (momentary switch K1 FIG. 6C) terminates the incrementing 
process and holds the memory contents unchanged until the activation for 
either "Start" or "Reset". The I.C.'s used are identifable as follows: 
__________________________________________________________________________ 
IC IDENTIFICATION 
__________________________________________________________________________ 
U1 LM3086 
U21 DM7407 
U41 DM74157 
U66 DM74300 
U2 LM3086 
U22 DM7404 
U42 DM74157 
U67 CD4069 
U3 LM3086 
U23 DM74123 U68 74C221 
U4 DM4011 
U24 DM7402 U69 DM74304 
U5 74C221 
U25 DM74123 
U50 DM74LS374 
U70 LM7805 
U26 DM74123 
U51 DM74LS374 
U71 LM7812 
U7 LM3086 
U27 DM74LS374 
U52 DM74LS374 
U72 LM7812 
U8 LM3086 
U28 DM74LS374 
U53 DM74LS374 
U73 LM7812 
U9 CD4069 
U29 DM74LS374 
U54 DM74LS374 
U74 LM7812 
U10 
CD4013 
U30 DM74LS374 
U55 DM74LS374 
U75 LM7805 
U11 
74C221 
U31 DM7430 
U56 DM74LS374 
U76 LM7812 
U12 
CD4011 
U32 DM74191 
U57 DM74LS374 
U77 LM7812 
U13 
CD4013 
U33 DM74123 
U58 DM745153 
U14 
74C221 
U34 DM7430 
U59 DM745153 
U15 
CD4080 
U35 DM74191 
U60 DM745153 
U16 
CD4013 
U36 DM7404 
U61 DM745153 
U17 
DM74504 
U37 DM7430 
U62 DM745153 
U18 
DM7474 
U38 DM74191 
U63 DM745153 
U19 
AMD685 
U39 DM7474 
U64 DM7474 
U20 
DM74500 
U40 DM74191 
U65 DM74574 
__________________________________________________________________________ 
It should be noted that in FIGS. 3A-3K, 4A 4B, 6A-6C, the individual A,B,C, 
etc figures are combinable to produce an overall diagram of the respective 
part of the circuit designated by the individual figure numbers 3, 4 and 
6. 
It should also be noted that where a circuit line leaves one figure, say 
FIG. 4A, it will be designated say (BB)/B. This in turn means that it 
connects with circuit line (BB)/A in FIG. 4B. 
In all cases the letter in the denominator designates which of the FIGS. 
A-L in the case of FIGS. 3, that it connects with. 
Similar consideration applies to each of the lines in FIGS. 4 & 6. Where BB 
is repeated several times it is designated as follows B2B, B3B, B4B etc. 
Similar considerations apply for each of the letters C, D E etc. 
INCREMENTING OPERATION 
The video input signal is passed through a buffer amplifier and clamp (U1 
and U2 FIG. 1) and thence via a low pass filter to a variable gain and 
level processing stage (Q4 to Q8 FIG. 3). The processed video signal is 
now converted from an unbalanced to balanced format (Q1, Q2, Q3 FIG. 4) 
before being presented to the input of a voltage comparator (pins 3 and 4 
of IC U21 FIG. 4). This comparator corresponds to the functional block 2 
of FIG. 1. The Digital-to-Analogue converter U5 output (pins 14 and 15) 
forms a balanced drive to the cascode stage (Q4 and Q5) whose balanced 
current source output is subtracted from the balanced video signal 
(representative of the video input) at the comparator input (pins 3 and 
4). In this way the sign of the difference between the video input signal 
and the D to A converter output signal is generated at the complementary 
output (pins 11 and 12 of the U21) of the comparator. This one bit word is 
stored within the comparator (The AMD685 has a latching capability) in 
response to the latch enable command which appears at pixel rate at pin 6 
of the comparator. The complementary binary signal at the comparator 
output (pins 11 and 12) corresponds to the SGN parameter of FIG. 1. 
The digital memory of the instrument is made up to 192 16K dynamic RAM 
chips (The industry standard 4116) whose storage capacity forms a raster 
matrix of 512.times.512 picture elements each of which constitutes a 12 
bit word. 
During the incrementing process these memory chips are operated in the 
"read-modify-write" mode whereby a picture element word is extracted from 
the memory, modified in the incrementor and written back into the same 
memory location. When incrementing ceases the memories are operated in the 
read mode. The data bits are accessed at pin 14 of the memory chips and 
are written back into the memory by presenting the modified bits at pin 2. 
In order to achieve data read and write rates commensurate with the 
picture element rate of incoming video signal the memory bank is divided 
in to 8 groups of chips per T.V. field. The members of the 8 groups are 
addressed cyclicly out of phase in order to achieve a high data rate. The 
circuit diagram of the memory shown in FIG. 5. constitutes one quarter of 
the total memory bank of the instrument. 
Four identical circuit boards make up the complete memory bank A, B, C and 
D. Note that each circuit board contains rows of chips 1 to 4. The 
addressing sequence may now be described thus: ODD T.V. FIELDS: A1, B1, 
C1, D1, A3, B3, C3, D3, A1, B1, etc. EVEN T.V. FIELDS: A2, B2, C2, D2, A4, 
B4, C4, D4, A2, B2, etc. 
Addressing of the memory chips is accomplished in the normal way according 
to the row and column address multiplexing method. In order to comply with 
dynamic RAM refresh requirements the addressing pattern has been chosen 
such that all row address locations are cycled in less than 2 ms. In this 
way the need for a separate refresh cycle disappears. 
A timing diagram of the 8-phase clocking cycle is shown in FIG. 7B. Here we 
follow the event sequences pertaining to the memory group A1. It should be 
noted that the event sequences for each of the other memory groups are 
identical except for a time shaft. The generation of the multiplexed 
address word pattern is realised in U31, U32, U34, U35, U37, U38, U39, 
U40, U41, U42 FIG. The address pattern is then passed through an 8 stage 
shift register bank (U50 to U57 FIG. 3) to achieve the desired 8-phase 
format as fed to the memory bank via connectors J7 to J10. The row address 
strobe pulse, column address strobe pulse and write enable pulse are 
generated in U23 and U26 (FIG. 3) and are presented in the required 
8-phase format to the memories via U27 to U30 (FIG. 3). 
The stored picture element word (12-bit) stream appears in serial format at 
the outputs (pins 7 and 9) of the multiplexing IC's (U58 to U63 FIG. 3) 
and corresponds to the quantity L of FIG. 1. The picture element word 
stream (TTL format) is translated to ECL format within the incrementer 
(U7, U8 and U9) and latched by means of type D flip flops (U4 and U6). 
The 8 most significant bits of the picture element word stream (L) are fed 
to the Digital-to-Analogue converter (MC 10318--U21 pins 1 to 8). All 12 
bits of the picture element word stream are fed to the input of a 12-bit 
adder/subtractor (corresponding to functional block 3 of FIG. 1) as 
implemented by means of three ALU chips of type MC 10181 (U10, U11 and U12 
pins 10, 16, 18 and 21). The other (12-bit) input word corresponding to 
the quantity D of FIG. 1. to this adder/subtractor is derived from the 
Incrementor controller FIG. 6 and is generated according to the algorithem 
for D.sub.opt. D appears at the pins 9, 11, 19 and 20 of the three ALU 
chips that make up the adder/subtractor. The 12-bit output of the 
adder/subtractor (pins 2, 3, 6 and 7) corresponds to the quantity S of 
FIG. 1 and is equal to either the sum quantity L+D or the difference 
quantity L-D according to the sense of the SGN parameter as present in 
complementary form at pins 11 and 12 of the voltage comparator U21. The 
value S thus derived is in accordance with the incrementing algorithm and 
must be written into the same memory location as the picture element word 
L from which it was derived. The 12 type D flip-flops (IC's 17 and 18) 
serve to hold the 12 bits of S for the most optimum time slot available 
for writing back into the memory. The interface chips 16, 19 and 20 
translate the ECL format into the TTL format as required by the memory 
chips. See FIG. 7B for timing details of the incrementing process. 
The balanced analog output of the Digital-to-Analogue converter serves also 
as a basis for the derivation of the output signal. Transistors Q7 to Q10 
(FIG. 3) form a balanced to unbalanced buffer stage with selectable signal 
inversion. The analog signal thus generated contains only picture 
information and is devoid of synchronising pulses. The mixing amplifier 
(U7 and U8 FIG. 3) serves to blend the stored (analog) signal into the 
input video signal and thereby restoring synchronising information. At the 
same time this amplifier provides the facility of additive mixing of the 
inverted stored signal with the incoming signal for comparison 
measurements. Finally the output signal is presented in 1.nu..sub.pp 
(75.OMEGA.) format via Q5 FIG. 3. 
TIMING PULSE GENERATION 
All timing pulses, as required by the memory PCB;3 s the Incrementor and 
the Incrementor Controller are generated on the main circuit (FIG. 3) and 
are locked to the synchronising pulses of the incoming video signal. 
Separation of the synchronising pulses from the input video signal is 
performed by U3. A negative polarity composite synchronising pulse in CMOS 
format is available at U4 pin 4. The origin of this signal is selectable 
between video input and external composite sync input by means of a sync 
selector switch on the rear panel. U11 (pin 1 and 4) is a one-shot timed 
to suppress the twice line frequency components associated with the 
equalising and serration content of the composite sync pulse stream. Thus, 
the pulse stream at pin 4 of U11 will be at T.V. line rate only. The 
purpose of the field pulse detector (U10 pins 8 to 13) and the gate (U12) 
back to back one-shot arrangement (U14) is to provide missing line pulses 
when the input video contains a non-standard industrial sync. format. The 
two halves of U14 form a self-sustained oscillation capable of 
"fly-wheeling" over broad vertical pulses and thereby providing the 
missing line pulses. The importance of this is to maintain clock 
continuity to the memory bank during the vertical block. 
U17 (pins 1 and 2) serves to level shift the line reference pulse into TTL 
format. 
U19, U20 (pins 11, 12,13) and U21 (pins 8, 9, 10) form a gated oscillator 
operating at the picture element rate of 14.5 MHz and which is locked to 
the T.V. line reference pulse. The incrementor clock pulse is derived 
directly from the 14.5 MHz via the pulse former (C46, C47, R141) and gate 
(U66 pins 11, 12, 13). Also the clock pulse for the 8 phase shift 
registers (U50 to 57) is derived from the 14.5 MHz via a phase shift 
network (U22 pins 11 to 13) A further phase shift (U69 inverter 
propagation delays) derives the clock pulse for the generation of the 8 
phase memory write enable pulse. The 14.5 MHz clock pulse for the 
generation of both 8 phase row and column address strobe pulses are taken 
directly from U22 pin 12. 
The drive to the memory address word generator is taken via U18 pins 3, 6 
(which performs a frequency halving) and U21 pins 1, 2, 3, 11, 12, 13 
which performs the gating function for correct positioning of the clock 
pulse to the divide by 32 counters (U32, 35). U39 (pins 1 to 6) provides a 
clock gating drive to prevent address word overflow at the termination of 
the count for each T.V. line. 
U18 supplies a complementary drive to the video switch U7 and U8. Switching 
points are defined by the trailing edges of the one-shots U33. Vernier 
control of the commencement of stored signal blend-in boundary is achieved 
by R181. Vernier control over the end of the blend-in boundary is achieved 
by R172. U20 (pins 4, 5, 6, 8, 9, 10) allow manual overide of the window 
blend-in drive by the front-panel "diaplay" control to obtain either 
"direct" (i.e. input signal) only or "stored" signal only. 
The one-shots of U25 define the position and width of the X-Y enable 
command--relevant only when the application of the instrument is extended 
to X-Y (as well as T.V.) scan. 
The flip-flops U64 and U65 provide a coherent two-bit drive to the 4-way 
multiplexers U58 to U63. The function of these multiplexers is to convert 
the 12-bit parallel data stream as accessed in the memory banks into a 
12-bit serial stream as required by the incrementor. 
Separation of the field pulse from the incoming composite sync component 
pulse is achieved by means of the integrating network R132, C32 and 
subsequent schmitt trigger U67 (pins 1, 2, 13, 13). U68 and U10 (pins 1 to 
6) constitute the frame pulse discriminator. U16 in conjunction with U22 
(pins 1,2,5,6) and U24 (pins 1 to 6) define the field alternating drive to 
the two halves of the memory bank corresponding to the two T.V. fields. 
Selection of either field (front panel control) is made possible by means 
of a reset or set command to U16 via U12 (pins 4, 5, 6) or U9 (pins 8,9) 
respectively. The one-shot U11 (pins 9,12) is set to approximately 90% of 
the field period and by inhibiting the generation of the field pulse in 
U68 (pins 10, 5) improves the systems noise immunity by reducing the 
probability of interference from false field pulses. 
The vertical position of the video blend-in command is determined by U13 
(pins 1 to 6), U15 and U16 (pins 1 to 6). This command is mixed with the 
line switching command by means of an over-riding clear operation in U33 
(pin 3). 
GENERATING THE INCREMENTS (D) 
The quantity designated D is generated by the Incrementor Controller. FIG. 
6. 
During the initial stage of the picture acquisition process (outer 
convergence) the magnitude of D is halved after each frame period 
commencing with a value equal to half the dynamic range of L (i.e. by 
stepping down the value of D by one bit level after each frame period). 
This process is allowed to continue until the value of D is comparable 
with the value 
##EQU36## 
At this point (say D=Do) the rate of halving of the value D will be 
reduced in such a way as to give the following approximation to a 
hyperbolic function of time: 
______________________________________ 
VALUE OF D DURATION (NO. OF FRAMES) 
______________________________________ 
Do 1 
Do/2 2 
Do/4 4 
Do/8 8 
Do/2.sup.N 2.sup.N 
______________________________________ 
The transition point between the "outer" and "inner" convergence processes 
is preset (on the front panel of the instrument) by a prior knowledge of 
the input noise level. Convergence is complete after the value of D has 
been held at one least significant bit (of M) for a sufficiently long 
period for the residual noise level to assymptote to its final value. 
With reference to the circuit diagram of the incrementor controller (FIG. 
6) 11 to 1 are the bits constituting the value D. 
Each bit is generated at one output of a chain of D type flip-flops (U13, 
U14) which forms a 12 stage shift register. Prior to the commencement of a 
convergence cycle all shift register outputs are set to zero by means of a 
reset command on pin 1 of U13 and U14 ("step size" selector on "Auto"). 
Thus the initial value of D is 011111111111. Let us assume for the moment 
that a high logic level (corresponding to the 12 bit ripple counter (U10) 
set to all zeros) at pin 12 of U12 allows a frame rate pulse to drive the 
shift register (U13, U14) via the clock inputs (pin 1). On initiation of a 
convergence cycle the reset command is removed from the shift register and 
the high logic level on the input stage (pin 11, U13) is allowed to 
propagate through the register. The sequence of D values generated in this 
way may be tabulated thus: 
__________________________________________________________________________ 
D VALUE Most signifi- 
TIME .DELTA..sub.12 
.DELTA..sub.11 
.DELTA..sub.10 
.DELTA..sub.9 
.DELTA..sub.8 
.DELTA..sub.7 
.DELTA..sub.6 
.DELTA..sub.5 
.DELTA..sub.4 
.DELTA..sub.3 
.DELTA..sub.2 
.DELTA..sub.1 
cant bit 
__________________________________________________________________________ 
1st Frame 
0 1 1 1 1 1 1 1 1 1 1 1 
2nd Frame 
0 0 1 1 1 1 1 1 1 1 1 1 
3rd Frame 
0 0 0 1 1 1 1 1 1 1 1 1 
4th Frame 
0 0 0 0 1 1 1 1 1 1 1 1 
5th Frame 
0 0 0 0 0 1 1 1 1 1 1 1 
6th Frame 
0 0 0 0 0 0 1 1 1 1 1 1 
7th Frame 
0 0 0 0 0 0 0 1 1 1 1 1 
8th Frame 
0 0 0 0 0 0 0 0 1 1 1 1 
9th Frame 
0 0 0 0 0 0 0 0 0 1 1 1 
10th Frame 
0 0 0 0 0 0 0 0 0 0 1 1 
Least signifi- 
11th Frame 
0 0 0 0 0 0 0 0 0 0 0 1 cant bit 
__________________________________________________________________________ 
This sequence will generate a purely "outer" convergence cycle with a 
convergence time equal to 11/25 Th of a second and is obtained with the 
input signal-to-noise ratio selector on the 2nd lowest setting. 
With the input signal-to-noise ratio selector switch set for higher input 
noise levels the above sequence is modified below the appropriate bit 
level by a progressive reduction in the number of clock pulses allowed to 
reach the shift register via the gate U12 (pins 11, 12, 13). This 
progressive reduction is defined by the counter U10 and the combinational 
network U1 to U9. For example with the input signal-to-noise ratio 
selector on pin 7 of U13 the following sequence of D values is obtained: 
__________________________________________________________________________ 
TIME D VALUE 
__________________________________________________________________________ 
1st Frame 
0 1 1 1 1 1 1 1 1 1 1 1 .uparw. 
2nd Frame 
0 0 1 1 1 1 1 1 1 1 1 1 .uparw. 
3rd Frame 
0 0 0 1 1 1 1 1 1 1 1 1 OUTER 
4th Frame 
0 0 0 0 1 1 1 1 1 1 1 1 CONVERGENCE 
5th Frame 
0 0 0 0 0 1 1 1 1 1 1 1 .dwnarw. 
6th Frame 
0 0 0 0 0 0 1 1 1 1 1 1 .dwnarw. 
7th Frame 
0 0 0 0 0 0 0 1 1 1 1 1 .uparw. 
8th Frame 
0 0 0 0 0 0 0 1 1 1 1 1 .uparw. 
9th Frame 
0 0 0 0 0 0 0 0 1 1 1 1 .uparw. 
to .uparw. 
12th Frame 
0 0 0 0 0 0 0 0 1 1 1 1 INNER 
13th Frame 
0 0 0 0 0 0 0 0 0 1 1 1 CONVERGENCE 
to .dwnarw. 
20th Frame 
0 0 0 0 0 0 0 0 0 1 1 1 .dwnarw. 
21st Frame 
0 0 0 0 0 0 0 0 0 0 1 1 .dwnarw. 
to .dwnarw. 
36th Frame 
0 0 0 0 0 0 0 0 0 0 1 1 .dwnarw. 
37th Frame 
0 0 0 0 0 0 0 0 0 0 0 1 .dwnarw. 
to .dwnarw. 
End of .dwnarw. 
Convergence 
0 0 0 0 0 0 0 0 0 0 0 1 .dwnarw. 
__________________________________________________________________________ 
The above algorithm may be overidden by means of the step size selector 
which provides a means of manual step size selection. U11 and U18 serve to 
synchronise all command transitions to the T.V. frame pulse to ensure that 
all processing occurs for an integral number of frames.