A four-way power splitter/combiner apparatus comprises five unbalanced ports, configured as a common port and four other ports, with power introduced into the common port, over a relative wide frequency band, being substantially equally divided between the four other ports, and vice versa. Each of the first and second ports of each pair of the four other ports has a substantially in-phase signal thereat, with the first in-phase pair of ports having a substantially 180.degree. signal phase difference from the substantially in-phase signals at the other pair of ports. Each of the balun and 2-to-1 transformers of the push-pull 4-port apparatus can be fabricated of a magnetically-decoupled coaxial cable assembly, wherein the cables are all of the same characteristic impedance as the load impedance magnitude at each of the five ports.

BACKGROUND OF THE INVENTION 
The present invention relates to passive radio-frequency circuitry and, 
more particularly, to a novel radio-frequency (RF) four-way power 
splitter/combiner apparatus operating in a push-pull mode. 
In many high-power RF applications, such as magnetic resonance imaging and 
the like, vacuum tube amplifiers are utilized to provide the relatively 
high RF excitation energy needed. However, solid-state devices capable of 
generating RF power outputs of the required magnitude are available at 
lower RF frequencies, and are expected to be shortly available at RF 
frequencies in the lower VHF region; use of solid-state power amplifiers 
is highly desirable, to provide the desired excitation signal: with larger 
bandwidth; over a longer lifetime (higher reliability); require simpler 
cooling systems; and cost less. At present, providing even a 
high-frequency RF power amplifier with significant, e.g. greater than 
several hundred watts, power output requires that the output of several 
individual power amplifier modules, each capable of providing only a 
portion of the required output power, be combined. The input power for the 
plurality of amplifier modules must itself be provided by splitting a 
common input signal. Accordingly, it is highly desirable to provide 
apparatus having a common terminal and four other terminals, a first pair 
of which is at a 180.degree. phase difference from a second pair of other 
terminals, and with the four-way apparatus being utilizable for either a 
power-splitting function, i.e. wherein the common port receives input 
power and the four ports each provide approximately one-fourth of the 
input power as driving power to an associated input of a different power 
amplifier, or as a power combiner, i.e. wherein the output power received 
at each of the four other ports is combined into a signal at the common 
output port. 
It is well known in the prior art to provide four-way splitter/combiner 
apparatus utilizing broadband transformers in which coaxial lines of 
severel different characteristic impedances, e.g. 25 and 50 ohm lines, are 
utilized with 4:1 impedance transformers, to provide four other port 
signals all having the same phase angle responsive to a single signal at 
the common port. Typical prior art configurations can be found in 
Application Note AN-749, entitled "Broadband Transformers and Power 
Combining Techniques for RF", in the Motorola RF Data Manual (1980) at 
pages 2-67 and 2-68. 
BRIEF SUMMARY OF THE INVENTION 
In accordance with the invention, a four-way power splitter/combiner 
apparatus comprises five coaxial (unbalanced) ports, configured as a 
common port and four other ports, with power introduced into the common 
port, over a relatively wide frequency band, being substantially equally 
divided between the four other ports, with each of first and second pairs 
of ports having 180.degree. phase difference between the pairs and with a 
first one of each pair of the other ports having a 0.degree. signal phase 
difference from the remaining one of that port pair. 
In presently preferred embodiments, coaxial cables of only the normal 
system impedance, e.g. 50 ohms, are utilized; three or five ferrite-loaded 
transformation sections are used to provide each push-pull four-way power 
splitter/combiner apparatus. 
Accordingly, it is one object of the present invention to provide a novel 
five-port four-way power splitter/combiner apparatus having each of two 
pairs of other ports separated by substantially 180.degree. phase 
difference and operating in the push-pull mode. 
This and other objects of the present invention will become apparent upon 
reading of the following detailed description of the invention, when 
considered in conjunction with the drawings.

DETAILED DESCRIPTION OF THE INVENTION 
Referring initially to FIG. 1, an illustrative high-power RF amplifier 10 
contains a plurality of amplifier modules (labelled with a reference 
designation beginning with the number 11) and a plurality of power 
splitting/combining means (labelled with a reference designation beginning 
with the number 12), for amplifying the power output of an RF source 14, 
and providing the amplified power signal to a load 16. In the particular 
illustrated configuration, four power modules 11-1 through 11-4 are 
utilized, along with a pair of power splitting/combining means 12-1 and 
12-2. Each of amplifier modules 11-1 through 11-4 has an associated RF 
signal input 11-1a through 11-4a and an associated RF signal output 11-1b 
through 11-4b. Each of signal inputs 11-1a through 11-4a is connected to 
one of a like number of a plurality of splitting/combining means ports 
12-1a through 12-1d, respectively, of first means 12-1, herein operating 
as a power splitting means. Similarly, each respective power amplifier 
module output 11-1b through 11-4b is individually connected to an 
associated one of the respective four ports 12-2a through 12-2d of the 
second power means 12-2, herein operating as a power combining means. The 
four ports 12-1a through 12-1d of power splitting means 12-1 can be 
considered as outputs A-D (or OUT A through OUT D), while the four ports 
12-2a through 12-2d of power combining means 12-2 can be considered as 
inputs A-D (or IN A through IN D). Each of means 12 has a common RF port, 
e.g. common port 12-1e of the power splitting means and common port 12-2e 
of the power combining means. For purposes of description, each of ports 
12-1a through 12-1d or 12-2a through 12-2d will be hereinafter referred to 
as a 4-port, and ports 12-1e and 12-2e will be referred to as a common 
port. 
In accordance with one aspect of the present invention, the 4-ports are 
paired, with each pair of 4-ports, e.g. the pair of 4-ports port A and B 
or the pair of 4-ports C and D, having identical phase, e.g. a phase of 
about 0.degree., or about 180.degree., respectively, with respect to a 
signal at the common port; each pair of 4-ports thus has a phase 
difference of about 180.degree. with respect to the other pair of 4-ports. 
Each 4-port provides a splitting/combining ratio of about 0 dB. with 
respect to the other 4-port of that pair; each pair has about a 0 dB. 
ratio with respect to the other pair. Thus, one pair of 4-ports, e.g. 
ports A and B, forms a push-pull configuration with the other pair of 
4-ports, e.g. ports C and D. This provides the additional beneficial 
features normally obtained from push-pull amplifier operation, such as 
suppression of even harmonics of the operating RF frequency, and the like. 
Referring now to FIGS. 2A and 2B, a first presently preferred embodiment of 
a four-way power splitting/combining means 12, having two pair of 4-ports, 
with each pair in phase opposition to the other pair, is illustrated. 
Four-way apparatus 12' comprises first and second three-port hybrid 
circuits 18-1 and 18-2 each comprised of a pair of transmission line 
transformers 20-1a and 20-1b, or 20-2a and 20-2b, respectively, and an 
isolation resistance element 22-1 or 22-2, respectively. Coaxial 
balanced-to-unbalanced, or balun, transformer 24 is connected between the 
pair of three-port hybrid circuits 18-1 and 18-2 and the apparatus common 
port 12'-1e. Each of the apparatus 4-ports 12'-1A through 12'-1D and the 
common port 12'-1E are designed for optimum operation with the same load 
resistance R.sub.L, which is typically equal to the magnitude of the 
impedance of coaxial cables utilized within the system, e.g. 50 ohms and 
the like. In addition to the pair of hybrid circuits 18-1 and 18-2 and the 
transmission line balun 24, a variable capacitance (labelled with a 
reference designation beginning with the number 26) is typically provided 
in shunt with each port, to allow any residual inductive impedance 
component to be compensated for. Thus, the 0.degree. 4-ports 12'-1A and 
12'-1B of the first hybrid circuit 18-1 are respectively provided with 
variable capacitive elements 26-1 and 26-2, settable to capacitive values 
C.sub.A and C.sub.B, respectively. The 180.degree. 4-ports 12'-1C and 
12'-1D of the second hybrid circuit 18-2 are respectively provided with 
variable capacitive elements 26-3 and 26-4, settable to capacitive values 
C.sub.C and C.sub.D respectively. The output of each hybrid circuit 18-1 
or 18-2, at the terminals labeled Z.sub.AB and Z.sub.CD respectively, has 
an effective unbalanced impedance substantially equal, in magnitude, to 
one-half the characteristic impedance, i.e. equal to (R.sub.L /2), with 
respect to circuit ground potential. Each of outputs Z.sub.AB and Z.sub.CD 
has a phase of 180.degree. with respect to the other. Therefore, the 
impedances add and the total impedance between terminals 24a and 24b is a 
balanced impedance of R.sub.L ohms. This balanced impedance is converted, 
by transmission line balun 24, to an unbalanced impedance of magnitude 
R.sub.L, at common port 12'-1E. Any residual inductive impedance component 
at this common port is compensated for by variable capacitance 26-5, 
having an output capacitor value C.sub.o . 
Each of transmission line transformers 20-1a, 20-1b, 20-2a and 20-2b is 
constructed of a length of a coaxial cable 28, having an impedance Z.sub.O 
equal to the splitter/combiner apparatus load impedance R.sub.L value. 
Each coaxial transmission line 28 has a center conductor, having a first 
end 29a connected to the associated one of the four 4-ports, and having an 
opposite end 29b, connected to one end of the associated isolation 
resistor 22-1 or 22-2. A cylindrical sheath 30 of insulation dielectric 
material surrounds each center conductor, and is itself surrounded by a 
coaxial outer conductor. The coaxial outer conductor ends 31a, closest to 
the inner conductor ends connected to the associated 4-port, are connected 
together to form the respective transmission line transformer output 
terminal Z.sub.AB and Z.sub.CD. The opposite end 31b of each coaxial cable 
is cross-connected to the center conductor end 29b of the opposite coaxial 
cable of that transmission line transformer 20 and, therefore, to an end 
of the associated isolation resistance 22. A plurality of magnetic 
decoupling elements 33, such as ferrite beads, cores, tubes and the like, 
generally having a toroidal shape, are placed about the outer conductor 31 
of each transmission line assembly 20. Each of the transmission lines 28 
is of a length, between opposite center conductor ends 29a and 29b, which 
is less than one-quarter of an effective wavelength at the maximum 
frequency of apparatus operation. Each of the ferrite decoupling elements 
33 is chosen such that, when the desired number of such elements 33 are 
"strung" upon the associated coaxial element 28, the impedance measured 
between at least the outer conductor input and output ends 31a and 31b is 
relatively high in comparison to, e.g. at least an order of magnitude 
greater than, the characteristic impedance Z.sub.O of the coaxial cable 28 
at the lowest frequency of desired operation. The decoupling elements, in 
each transformer described herein, act to minimize common-mode RF currents 
in the associated transformer. 
The balun transformer 24 is fabricated of a length of coaxial cable 34 
having a characteristic impedance Z.sub.O equal to the apparatus port 
impedance R.sub.L. Coaxial cable 34 has a center conductor, having an 
input end 35a connected to the output of the first hybrid circuit means 
18-1 and having an opposite end 35b connected to common connector 12-1E. A 
generally cylindrical member 36 of dielectric insulation material 
surrounds the center conductor, and is itself surrounded by a cylindrical 
outer conductor. That outer conductive sheath end 37a closest to center 
conductor end 35a is connected to the output of the second hybrid circuit 
means 18-2, while the opposite outer conductor end 37b is connected to 
system RF common potential at the common port end. The coaxial cable 34 of 
the balun transformer 24 is placed through the central apertures of 
another plurality of magnetic decoupling (ferrite) elements 39. Cable 34 
can be run in a generally linear fashion through the plurality of ferrite 
elements 39, or may be wound about a single large ferrite element. Cables 
28 can similarly be run through, or wound about, element 33. At power 
levels on the order of 1-10 kilowatts, at frequencies in the low VHF 
region (e.g. frequencies from about 30 MHz. to about 100 MHz.), the 
ferrite elements 33 and/or 39 may exhibit a rise in temperature due to 
dissipation of RF power caused by unavoidable common mode currents and 
other unsymmetrical parameters. As approximately twice the power in each 
of hybrid circuit means 18 is present in transmission line balun 24, I 
generally prefer to make transmission balun line 24 of about twice the 
length of each of the transformer lines 20, and to use about twice as many 
ferrite elements 39 as elements 33, so that the losses per ferrite element 
are not increased in the transmission line balun. It will be understood 
that the actual form of coaxial cables 28 and 34, the particular materials 
and forms of elements 33 and 39, and the power ratings and configurations 
for isolation resistances 22, can all be selected from a wide range of 
readily-available, custom-available or custom-built components as 
necessary for the particular impedance and/or power levels and frequencies 
involved in a particular application of my novel 4-port push-pull power 
splitter/combiner apparatus. 
Referring now to FIG. 2C, the common port impedance vs. frequency response, 
of one splitter/combiner apparatus of this embodiment type, is 
illustrated. The magnitude of the 50 ohm-nominal common impedance is 
plotted along ordinate 41, for values between 40 ohms and 60 ohms, while 
the 0.degree.-nominal phase, is plotted along auxiliary ordinate 42, for 
values between -10.degree. and +10.degree., with respect to frequencies 
between about 2 MHz. and about 110 MHz. plotted along abscissa 43. Solid 
curve 45 is the impedance magnitude curve, read in conjunction with 
ordinate 41, while broken-line curve 46 is the common port phase curve, 
read with respect to auxiliary ordinate 42. Both curves are plotted for 
the condition wherein each of the four ports is terminated with a 
low-reflectivity load of characteristic impedance, e.g. about 50.+-.jO 
ohms. It will be seen that a common port VSWR of less than 1.1:1 and a 
total phase change of less than 6.degree. occurs over the entire range 
between about 2 MHz. and about 110 MHz. 
Referring now to FIG. 2D, a 4-port voltage ratio, which I call a V.sub.x 
/V.sub.y ratio, in decibels dB., is plotted along ordinate 51 and a 4-port 
pair-to-pair phase difference .PHI. (which is "normalized" to zero by the 
addition of a 180.degree. offset to the actual phase difference) is 
plotted along auxiliary ordinate 52, for frequencies plotted along 
abscissa 53. Curve 54, to be read with respect to ordinate 51, illustrates 
the change in the ratio of the voltage at 4-port A with respect to that at 
in-phase 4-port B (the ratio between the 0.degree. pair of 4-ports). Curve 
55, also read with respect to ordinate 51, is the ratio of the voltage at 
one of the 180.degree. 4-ports (4-port C) to that at one of the 0.degree. 
4-ports (4-port B). It will be seen that the in-phase 4-port signal 
amplitudes are within one-half of a decibel of each other over an 
extremely large frequency range, and that the push-pull voltage ratio 
(curve 55) is within about .+-.0.3 dB. over a frequency range from less 
than 2 MHz. to in excess of 200 MHz. The phase between the 0.degree. 
4-ports A and B is, as shown by curve 57, within less than .+-.2.degree. 
over the full frequency range, while the phase difference .PHI. (again 
normalized to zero by the addition of 180.degree.) between 4-ports B and C 
is, as shown by curve 58, within about .+-.3.degree. over a relatively 
wide frequency range (e.g. from about 20 MHz. to greater than 200 MHz.) 
and is within .+-.5.degree. over an even larger frequency range. 
Referring now to FIGS. 3A and 3B, the five-cable embodiments 12 and 12' can 
be replaced by another presently preferred embodiment 12", which requires 
only three coaxial cable assemblies. Splitter/combiner apparatus 12" 
utilizes a first coaxial transformer means 60-1 for combining the in-phase 
power provided at first and second input 4-ports 12"-1A and 12"-1B each 
providing, and operating with, a terminating impedance R.sub.L equal to 
the characteristic impedance of the coaxial members forming the various 
transformation means. A second transformer means 60-2 combines the 
in-phase power provided to another pair of 4-ports 12"-1C and 12"-1D. Each 
of transformer means 60-1 and 60-2 utilizes at least one magnetic member 
61 through which, or about which, are wound transformer means windings 
63-1 or 63-2, respectively. A resistive element 65-1 or 65-2, 
respectively, for absorbing unbalanced dissipative powers, is respectively 
connected between 4-ports 12"-1A and 12"-1B, or 12"-1C and 12"-1D; each of 
balance resistances 65-1 and 65-2 are of resistance substantially equal to 
twice the port termination resistance R.sub.L. The transformation means 
60-1 and 60-2 provide respective unbalanced output terminal 66-1 and 66-2 
impedances Z'.sub.AB and Z'.sub.CD ; each output impedance is of a 
magnitude of one-half the common port termination value, i.e. a magnitude 
R.sub.L /2, with respect to circuit RF common potential, and is 
180.degree. out-of-phase with the other output terminal. Thus, a balanced 
impedance is provided between transformation means output terminals 66-1 
and 66-2. The balanced impedance is of magnitude substantially equal to 
R.sub.L, and is present at the input of a balun transformer means 67. 
Transformer means 67 includes "windings" 69 about magnetic means 71. The 
unbalanced side of balun transformation means 67 is connected between 
common port 12"-1E and RF ground potential. Preferably, a variable 
capacitance element 73-1 through 73-5 is provided in parallel with each of 
apparatus ports 12"-1A through 12"-1E, to provide an associated 
capacitance of magnitude C.sub.A through C.sub.D and C.sub.o, 
respectively, to substantially compensate residual inductive reactance at 
the associated apparatus port. 
As seen in FIG. 3B, I prefer to realize the pair of 2-to-1 impedance 
transformer means 60-1 and 60-2 with each transformer means having a 
single coaxial cable 63-1 or 63-2. In the first transformation means 60-1, 
the first end 74a of a center conductor is connected to 4-port 12"-1A and 
one end of resistance 65-1, while the other center conductor end 74b is 
connected to the transformation means output 66-1. The center conductor is 
surrounded by insulative dielectric material 75 and has an outer 
conductive sheath having a first end 76a, adjacent to first center 
conductor end 74a, which is connected to second center conductor end 74b 
and the output 66-1. The opposite outer conductor end 76b, adjacent to 
second center conductor end 74b, is connected to second 4-port 12"-1B and 
the remaining end of resistance 65-1. In the second transformation means 
60-2, the first end 74c of the center conductor is connected to output 
66-2. The center conductor second end 74d is connected to second 4-port 
12" -1D and the one end of resistance 65-2. The center conductor is 
surrounded by insulative dielectric material 75 and has an outer 
conductive sheath having a first end 76c, adjacent to first center 
conductor end 74c, which is connected to the remaining 4-port 12"-1c and 
the remaining end of resistance 65-2. The second outer conductor end 76d, 
adjacent to second center conductor end 74d, is connected to first center 
conductor end 74c and the second transformer output 66-2. A plurality of 
ferrite elements 61 enclose each of cable 63-1 and 63-2. 
Balun transformer means 67 utilizes a single coaxial cable 69, having a 
first center conductor end 77a connected to the output 66-1 of the first 
impedance transformer means 60-1 and having the center conductor second 
end 77b connected to common output connection means 12"-1E. The center 
conductor is supported by insulative dielectric material 78 within a 
tubular outer conductor having a first end 79a connected to the second 
transformation means output 66-2 and a second end 79b connected to RF 
common potential adjacent to common port connection means 12"-1E. A 
plurality of ferrite decoupling elements 71 encloses the coaxial cable 69. 
Referring now to FIG. 3C, the common port impedance vs. frequency response 
of one splitter/combiner apparatus of this embodiment type is illustrated. 
The magnitude of the 50 ohm-nominal common impedance is plotted along 
ordinate 80, for values between 40 ohms and 60 ohms, while the 
0.degree.-nominal phase is plotted along auxiliary ordinate 81, for values 
between -10.degree. and =10.degree., with respect to frequencies between 
about 2 MHz. and about 90 MHz. plotted along abscissa 82. Upper curve 84 
is the impedance magnitude curve, read in conjunction with ordinate 80, 
while lower curve 86 is the common-port phase curve, read with respect to 
auxiliary ordinate 81. Both curves are plotted for the condition wherein 
each of the four other ports is terminated with a low-reflectivity load of 
characteristic impedance, e.g. about 50.+-.jO ohms. It will be seen that a 
common port VSWR of less than about 1.1:1 and a total phase change of less 
than about 10.degree. occurs over the multi-octave frequency range between 
about 5 MHz. and about 80 MHz., with a somewhat greater total phase change 
occurring over the entire range between about 2 MHz. and about 90 MHz. 
Referring now to FIG. 3D, a 4-port voltage V.sub.x /V.sub.y ratio, in 
decibels dB., is plotted along ordinate 90 and a 4-port pair-to-pair phase 
difference .PHI. (+180.degree.) is plotted along auxiliary ordinate 91, 
for frequencies plotted along abscissa 92. Curve 94, to be read with 
respect to ordinate 90, illustrates the change in the ratio of the voltage 
at 4-port D with respect to that at the out-of-phase 4-port A (the ratio 
between a 180.degree. pair of 4 -ports). It will be seen that the 
push-pull voltage ratio, of curve 94, is within about .+-.0.3 dB. over a 
frequency range from less than 2 MHz. to at least 200 MHz. The phase 
difference (normalized to zero by the addition of +180.degree.) is 
represented by the quantity .phi. between out-of-phase 4-ports B and C is, 
as shown by curve 95, within about .+-.3.degree. over a relatively wide 
frequency range (e.g. from about 10 MHz. to at least 200 MHz.) and is 
within .+-.8.degree. over an even larger frequency range. 
Accordingly, a wide-band 4-port splitter/combiner apparatus can be provided 
with a single balanced transformer and a pair of 2-port combining 
transformation means, to realize a relatively simple and physically 
compact 4-way power splitting/combining apparatus having first and second 
pairs of ports, with the two ports in each pair being substantially 
in-phase with one another and with each port pair being substantially 
180.degree. out-of-phase with the opposite port pair, i.e. operating in a 
push-pull mode. 
While several presently preferred embodiments of my novel push-pull 4-port 
power splitting/combining apparatus are described in detail herein, many 
variations and modifications will now become apparent to those skilled in 
the art. It is my intent, therefore, to be limited only by the scope of 
the impending claims and not by way of the specific details and 
instrumentalities provided herein as illustrative of presently preferred 
embodiments.