Hand-held microwave spectrum analyzer with operation range from 9 KHz to over 20 GHz

A spectrum analyzer is provided that includes components to achieve from below 9 kHz to above 20 GHz operation range while remaining hand-held. Components of the spectrum analyzer include an integrated precision stand-alone step attenuator that does not rely on printed circuit board (PCB) mounted circuit elements within the signal path. Further, a PIN diplexing switch separates signals into different base-band and highband paths. The baseband path includes a pre-amplifier for low frequency signals, while the higher frequency bands may not necessarily include a pre-amplifier. The baseband path further provides improved broadband termination of its 1st mixer IF port by incorporating a new quadrature-coupled directional (QCD) filter that includes a ring resonator. An inexpensive air dielectric multi-cavity baseband filter is also used to suppress 2nd mixer IF images. The highband path incorporates multi-throw MMIC PIN diode switches to selectively filter different bands of input signals. At least three total 1st mixers are used to increase operation bandwidth. A phase locked loop providing a 1st LO to the 1st mixers is created that uses a divide-by-two frequency divider in cascade with a sampler-type frequency downconverter. The output of the 1st LO is frequency doubled and filtered to increase the frequency range of the highband signal path.

BACKGROUND

1. Technical Field

The present invention relates to a handheld spectrum analyzer, and more particularly over components to enable the spectrum analyzer to operate over a wide bandwidth.

2. Related Art

Currently available hand-held microwave spectrum analyzers have an input frequency range of up to 7.1 GHz. Examples include the Anritsu MS2721B (7.1 GHz), and the Rohde+Schwarz FSH-6 (6 GHz). An external frequency converter can be connected to downconvert a received input signal to the spectrum analyzer and effectively boost the frequency range of the handheld spectrum analyzer. But adding the external frequency converter may create a device that is no longer handheld. Further, to preserve measurement accuracy the cost of the external frequency converter can exceed the value of the spectrum analyzer.

The upper frequency limit of previous handheld spectrum analyzers was constrained largely by the perception that achievement of higher frequency capability would result in unacceptable measurement performance or cost. The selection of inexpensive surface-mount (SMT) switches, amplifiers, mixers, and other elements used to construct current low-cost small size spectrum analyzers has been limited for designs operating at frequencies greater than 6 GHz. A simple extension of prior-art designs using these circuit elements would result in a spectrum analyzer with excessive input noise, signal distortion, and susceptibility to damage from large signals and electrostatic discharge.

The operation range of components tested using a spectrum analyzer, including telecommunication and computing devices, is increasing beyond the 7 GHz limit. Accordingly, it is desirable to find ways to increase the frequency range of a hand-held spectrum analyzer while still providing a low-cost small sized device.

SUMMARY

According to embodiments of the present invention an improved low-cost hand-held microwave spectrum analyzer is provided that includes components enabling it to operate at frequencies well above 7.0 GHz. The purpose of this spectrum analyzer is to measure and display or record the power vs. frequency characteristics of electrical signals. It can also serve to analyze signal quality and to demodulate and decode information-bearing signals.

In one embodiment, the spectrum analyzer is designed to achieve 9 kHz to 20 GHz useful input frequency range with high dynamic range and hand-portability. The spectrum analyzer remains “hand-held,” meaning that it can be powered from an internal battery, and a person can comfortably carry it in one or both hands while operating its controls.

Components of the spectrum analyzer initially include an electromechanical step attenuator that does not rely on printed circuit board (PCB) mounted circuit elements within its RF signal path. The step attenuator is a stand-alone precision moving transmission line type device that incorporates relays in an integrated package. The step attenuator achieves lower signal loss, lower SWR, less signal distortion, and greater immunity to electrostatic discharge than can be achieved by a design that relies on PCB-mounted semiconductor switches.

Embodiments of the present invention further include PIN diode diplexing switches that selectively direct signals to either base-band or highband signal paths. The separate base-band path incorporates circuitry to allow operation from the low KHz region up to approximately 5.5 GHz, while the highband path allows operation from 5.5 GHz to 20 GHz or higher.

The baseband path initially provides a pre-amplifier for signals below approximately 4 GHz. Low-cost components are available for the pre-amplifier at this frequency, while at frequencies above 4 GHz in the baseband and highband paths, no pre-amplifier is used since it would require more costly components. To provide broadband termination of the 1stmixer IF port, the baseband path incorporates a new quadrature-coupled directional (QCD) filter that incorporates a ring resonator to provide a narrow passband. Further, an inexpensive air dielectric multi-cavity bandpass filter is used to pass a 1stmixer IF signal to the 2ndmixer input while suppressing signals at the 2ndmixer image frequency. The air dielectric 1stIF filter operates at higher frequencies than would be practical for ceramic filters commonly used in this type of application.

The highband path is broken into highband and midband paths. In the midband path, multi-throw MMIC PIN diode switches are used to direct the signal through a bank of bandpass filters to selectively filter different bands of signals. The PIN diode MMIC dice are integrated into surface-mount packages that enhance MMIC compatibility with the PCB and improve switch performance. The highband, midband and baseband signals are downconverted using three separate 1stmixers, such that the midband and highband paths each operate over an octave of frequency.

A 1stLO signal is created that is selectively provided by a multi-way switch to the 1stmixer of each of the baseband, midband and highband. The 1stLO oscillator frequency is controlled and stabilized by a phase lock loop (PLL) circuit. The PLL incorporates a divide-by-two frequency divider in conjunction with a frequency sampler in the feedback path from VCO to a phase detector. The inclusion of a frequency divider between the VCO and sampler enables the use of a low-cost sampler that has been optimized for lower frequencies to serve with a higher frequency VCO. The output of the 1stLO is frequency doubled to provide a signal to the highband 1stmixer to extend the frequency range of the highband path significantly, while a selectable filter reduces the spurious subharmonics that result from the frequency doubling.

DETAILED DESCRIPTION

FIG. 1shows a simplified block diagram of components of a spectrum analyzer according to embodiments of the present invention. The following description along with subsequent figures describes the function of the interconnected blocks ofFIG. 1, as well as additional details about components shown in block diagram inFIG. 1.

I. Input Path to Spectrum Analyzer

An electrical signal to be analyzed enters the spectrum analyzer at port2through a coaxial connector. For purposes of illustration, the input is shown from 9 kHz-20 GHz, although an alternative input frequency range can be used. The signal passes from input2through to a step attenuator4. The attenuator4for the example shown can provide 0 to 65 dB of attenuation, settable in 5 dB increments. The attenuator4is used to adjust signal level to within the spectrum analyzer's useful input amplitude range.

A. Precision Stand Alone Step Attenuator

The attenuator4is an electromechanical step attenuator that does not rely on printed circuit board (PCB) mounted circuit elements within the signal path. Instead, the step attenuator4is of a moving-transmission-line type commonly found in relatively non-portable laboratory test equipment. An example of the step attenuator4is the Anritsu 6372B 65 dB step attenuator. This attenuator includes multiple attenuators internally that selectively are connected by electromechanical relays integrated in a precision package. The Anritsu 6372B is a stand alone microwave component with a coaxial cable input and output and a separate low frequency control line connection cable.

Previous step attenuators used in lower frequency hand-held spectrum analyzers similarly used relays, but the relays were soldered onto a PCB to select the individual attenuators. The Anritsu 6372B, or similar precision integrated stand alone step attenuator, achieves better performance across the entire 20 GHz range than similar PCB-mounted switches and attenuation elements. In particular, compared to step attenuators that rely on semiconductor switches, it has lower signal loss, lower SWR, produces less signal distortion, and has greater immunity to electrostatic discharge. These advantages serve to improve the present spectrum analyzer's measurement accuracy, dynamic range, and durability.

Output from the step attenuator is directed to either a “baseband” path or “highbands” path by a PIN diplexing switch6.FIGS. 2A and 3Aprovide circuit diagrams illustrating components of the diplexing switch6in different control states. The control state ofFIG. 2Aswitches inputs to a high band path, while the control state ofFIG. 3Aswitches to a baseband path. A graph of the attenuation provided between the ports in the configuration ofFIG. 2Ais shown inFIG. 2B, while a graph showing attenuation between ports inFIG. 3Ais shown inFIG. 3B.

In the switch circuits shown inFIGS. 2A and 3A, a means is provided to split a signal path from Port1into separate, selectable paths for high frequency (Port2) and low frequencies (Port3). For the high frequency path control state ofFIG. 2A, high frequency signals are provided from port1to port2with low attenuation when switch A (202) remains open. The port1to port2path exhibits a highpass filter characteristic controlled by capacitor206and inductors210and208. Switch B (204) grounds one end of inductor210to complete the highpass filter network, and to attenuate high band signals exiting through Port3.FIG. 2Bshows the attenuation from port1to2is significantly lower than the low pass path of port1to port3with the switch control state shown inFIG. 2A.

For the low frequency path control state ofFIG. 3A, low frequency signals are provided from port1to port3with low attenuation when switch B (204) remains open. The port1to port3path exhibits a lowpass filter characteristic controlled by inductor210and capacitors206and212. Switch A (202) grounds one end of capacitor206to complete the lowpass filter network, and to attenuate low band signals exiting through Port2.FIG. 3Bshows the attenuation from port1to3is significantly lower than the high pass path from port1to port2with the switch control state ofFIG. 3A.

The switch circuitry ofFIGS. 2A and 3Aincludes reactive elements chosen such that the cutoff frequency of the low frequency path is significantly greater than that of the high-frequency path. The switch control state is changed at a “band-switch” frequency that is between the lowpass and highpass cutoff frequencies. Proper selection of element values yields a switch having low loss from port1to port3for frequencies from DC to the band-switch frequency, and low loss from port1to port2for frequencies greater than or equal to the band switch frequency.

The diplexing switch ofFIGS. 2A and 3Ahas performance advantages compared to alternative band selection technologies such as diplexing filters and traditional semiconductor switches. Namely, the alternative band selection diplexing filter provides a gradual transition from low-band to highband as a function of frequency. So, near “crossover” the band selection diplexing filter has high insertion loss (about 3 dB), high reflection (about −3 dB), and only a few dB isolation from port2to port3. By comparison, the diplexing switch used in embodiments of the present invention achieves much lower attenuation in the “on” path (less than 1 dB), much greater attenuation in the “off” path at the crossover (band-switch) frequency, and lower in-band reflection. A traditional wide-band semiconductor SPDT switch has semiconductor elements in series with the signal path, and therefore produces more signal distortion than the diplexing switch, particularly at low frequencies. Because the diplexing switch according to embodiments of the present invention does not have switch elements in series with the signal path, it can be realized with PIN diodes as the switch elements202and204without compromising low-frequency performance. The diplexing switch, made with PIN diodes is much less prone to damage from transient voltages than are GaAs MMIC switches.

The PIN diplexing switch ofFIGS. 2A and 3Aprovides low loss, low SWR, low signal distortion, and high tolerance to ESD. Common frequency diplexers and other types of 20 GHz-capable electronic switches do not possess all of these benefits. So, use of the PIN diplexing switch benefits the spectrum analyzer's dynamic range and/or durability.

The baseband path of the lower half ofFIG. 1includes some features carried over from conventional hand-held spectrum analyzers that operated below 7 GHz, but also includes new features that enable a combined operation with higher frequency input bands sharing a common RF input port.

The baseband path further includes switches that selectively connect a pre-amplifier (pre-amp)8or a through line10. The pre-amplifier8can be switched into the low frequency signal path to reduce system input noise figure. The pre-amplifier8implementation is innovative in the spectrum analyzer circuit ofFIG. 1in that the pre-amplifier8serves only the baseband path. In one embodiment, the pre-amplifier8is switched in with frequencies below 4 GHz, while other baseband signals above 4 GHz are switched around the pre-amplifier using the through line10. By restricting function of pre-amp8to lower frequency baseband signals only, the pre-amp8is realized with inexpensive SMT parts, including the amplifier and supporting GaAs RF switches.

Signals in the baseband path are lowpass-filtered using filter12to remove frequency components that would cause unwanted conversion products in the baseband 1st mixer16, and then mixed with a 1stlocal oscillator (LO) signal from LO14in the baseband mixer16to produce a 1stintermediate frequency (IF) signal that is greater than the cutoff frequency of the baseband input lowpass filter12. The output of LO14is provided in the switch position of switch18to the mixer16for baseband signals. The switch18provides for connection of the LO14to the highband 1stmixers34and54described subsequently as well. A unique configuration of circuitry for the LO14enables a single LO to be used to drive all of the baseband and highband signals.

B. Sampler Based LO PLL with Frequency Divider

FIG. 4shows circuitry used to form the 1stLO14, which is a sampler based phase locked loop (PLL) circuit. At least one prior-art 7 GHz hand-held spectrum analyzer and various non-handheld higher-frequency spectrum analyzers have also used a sampler in the 1stLO PLL circuit. In these prior-art cases, the sampler is used to convert the 1stlocal oscillator frequency to a DC or low-frequency IF for the purpose of phase locking the oscillator. The circuitry ofFIG. 4also uses a sampler in the 1stLO PLL circuit for essentially the same purpose, but with the addition of a simple frequency divider406to reduce the sampler input frequency. The sampler therefore can be optimized to operate at a lower input frequency than would be possible without a divider. As a result, a low-cost sampler is realized from inexpensive surface-mount parts.

In summary, the PLL circuit ofFIG. 4includes a voltage controlled oscillator400providing the output of the 1stLO. A small portion of the signal from the VCO400is split from the VCO output path by coupler402, amplified through404, and then applied to a divide-by-two frequency divider406. The output of the frequency divider406is amplified through408, then selectively provided through one of two filters410, depending on the frequency of the LO signal used, to the “RF” input port of sampler412. The sampler412is effectively a harmonic mixer with a low-frequency “LO” input provided at a frequency F1. The sampler mixes “RF” from410with harmonics of F1to produce an “IF” output. The sampler IF output is filtered by lowpass filter414to reject unwanted mixing products, amplified at416and applied to the “feedback” input of phase/frequency detector418. Frequency F2is applied to the “reference” input of phase/frequency detector (PFD)418. The PFD418produces a DC output that is amplified by loop amplifier420then applied to the tuning port of 1stLO VCO400. The closed loop made up of the described circuit elements causes the sampler IF signal to equal the reference signal F2in frequency and phase. In one example, the 1stLO VCO frequency is within a range of 6 to 12 GHz, signal F1is provided by a low-noise frequency synthesizer having a small fractional tuning range centered at approximately 200 MHz, and F2is provided by a frequency synthesizer having a small fractional tuning range centered at approximately 25 MHz.

The circuitry ofFIG. 4is particularly innovative in its use of the monolithic microwave integrated circuit (MMIC) frequency divider406in the input path to the sampler412. The frequency divider406allows the use of a pre-existing, proven RF synthesizer and sampler412to phase-lock the VCO400at frequencies that are N=2 times the design frequency of the sampler412. Although the frequency division number for divider406is set at two, other values of N could also be practical. In the circuit ofFIG. 4, the frequency divider406divides the frequency of VCO400by N before sampling occurs. The sampler412can be optimized to down-convert frequency F/N with minimal conversion loss. The prescaler (frequency divider406) and sampler412combination confers much of the performance benefit of a sampling downconverter while permitting the use of a sampler412that is simpler to design or that costs less than would a sampler that is optimized to operate at a higher, undivided frequency. As compared to a traditional divider-less design in which the sampler must operate at the VCO frequency, the divider/sampler combination ofFIG. 4allows the sampler412to be optimized for a lower input frequency. The lower-frequency design is more tolerant of parasitic circuit elements in the parts that make up the sampler412, and therefore better suited to inexpensive surface-mount construction. Although the traditional divider-less approach offers a theoretical performance advantage when used in a PLL, that advantage may not be fully realized as compared to a divider/sampler combination due to the difficulty in achieving ideal sampler behavior at higher frequencies.

The PLL ofFIG. 4operates essentially as follows: a portion of the output of VCO400is applied to a GaAs MMIC prescaler frequency divider406which divides the output frequency of VCO400by 2. Output from the prescaler406is down-converted by the sampler412to produce a low-frequency sampler IF output. LO drive for the sampler412(F1) is provided from a programmable RF synthesizer, not shown. The IF from sampler412is compared in the PFD418against a reference signal F2. The PFD418produces a DC output that minimizes when the PFD418inputs are synchronous and aligned in phase. The DC output from the PFD418is amplified and used to tune the VCO400. The closed-loop circuit ofFIG. 4continuously adjusts the frequency of VCO400such that the two inputs to PFD418align in frequency and phase. As a result, the frequency of VCO400(FVCO) stabilizes at a frequency that is FVCO=2*(N*(F1)+(P)*F2), where N=an integer defining the frequency division, F1=frequency of the RF synthesizer, P=polarity of the PFD, and F2=PFD reference frequency.

Referring again toFIG. 1, the 1st IF signal from mixer16passes through a directional bandpass filter20. This filter20passes the IF signal with minimal loss, but absorbs signals that are outside of its narrow passband. The directional filter20improves 1stmixer inter-modulation performance by absorbing the unused 1:1 mixing product. Some embodiments of the present invention introduce a new topology for the directional filter20that provide improved performance and manufacturing advantages that are described to follow.

Prior to describing the new topology of directional filter20, reference is made toFIG. 5that shows an alternative directional filter that can be used for filter20ofFIG. 1that offers some performance advantages. The circuit ofFIG. 5is a traveling wave directional filter that uses a ring resonator. The circuit was described generally in: “Traveling Wave Directional Filter” by F. S. Coale (October 1956 IRE Transactions on Microwave Theory and Techniques). The two port non-reflective bandpass filter ofFIG. 5is commonly implemented as a stripline or microstrip circuit, with conductors printed on a planar dielectric substrate.

The resonator of the circuit ofFIG. 5is a transmission line loop500with an effective electrical path length around the loop of one wavelength at its fundamental resonant frequency. An input directional coupler507formed by transmission line502as a primary and leg504of loop500as a secondary introduces a signal into the loop500, launching the signal predominantly in one direction. At resonance, a circulating “traveling wave” builds within the loop500, synchronously reinforced by the coupled input signal. An output directional coupler509formed by transmission line506as a primary with leg508as a secondary is located on the side of the loop opposite the input coupler507, and couples a signal out of the loop500. The circuit ofFIG. 5passes a signal from input to output with low loss at signal frequencies for which electrical length of the loop is one wavelength. The coupling factor of the input and output of couplers507and509largely determines spectral width of the passband. At frequencies sufficiently “off resonance,” the input signal does not couple effectively to the loop resonator500, and instead is absorbed by the input coupler termination505. Out-of-band signals are therefore largely absorbed, and so their reflection is suppressed. The circuit is reciprocal: its input and output connections can be exchanged without affecting its behavior.

The circuit ofFIG. 5has some disadvantages. First, the microstrip implementations are not generally suitable for fractional bandwidth greater than a few percent. Further, the higher fractional bandwidth requires the couplers507and509to be made with a very small gap512between primary and coupled arms, which make circuit behavior very sensitive to fabrication tolerance. Because larger fractional bandwidths are not practical, the type of filter shown inFIG. 5is commonly made with a fractional bandwidth of a few percent or less. But for such narrow filters, center frequency tolerance can be significant vs. bandwidth. As a result, a further drawback is that center frequency tuning may be required to avoid excessive insertion loss. Another drawback is that the microstrip implementation of the filter ofFIG. 5is prone to having an undesirable secondary transmission peak due to the excitation of resonant modes that are close in frequency to the fundamental resonance of the traveling wave. Suppression of the spurious modes is highly dependent upon circuit fabrication tolerance.

FIG. 6shows the basic layout of components of the QCD filter according to embodiments of the present invention, the QCD filter being usable as filter20ofFIG. 1. The QCD Filter is a two-port electrical bandpass filter that absorbs the signals it does not pass. It is considered an improvement to the directional filter circuit ofFIG. 5in that the QCD Filter ofFIG. 6achieves greater fractional bandwidth, lower sensitivity to fabrication tolerance, improved suppression of a spurious resonance mode, and reduced reflections. Although the QCD filter is contemplated for use in other applications than a spectrum analyzer, it is described herein with application within the frequency converter section of a spectrum analyzer ofFIG. 1, where the “QCD Filter” passes a desired mixing product while absorbing other mixing products.

The QCD filter can be implemented in either stripline or microstrip. The impedance of both input Port1(601) and output Port2(602) is assumed to be Z0to match a connecting impedance of Z0for purposes of this description. In practice, circuit element dimensions and values can be adjusted to optimize port match, and the circuit can be made to present unequal impedances at Port1(601) and Port2(602). Signal frequency is the design passband center frequency for purposes of this description, unless stated otherwise.

Like the circuit ofFIG. 5, the QCD filter ofFIG. 6includes a ring resonator602that is a closed loop of transmission line having an electrical circumference of one wavelength at the design passband center frequency. Four directional couplers,610,620,630and640, are formed using the ring600and respective coupling transmission lines604,605,606,607such that the ring metal acts as one entire branch of each coupler. The couplers are spaced equally about the circumference of the ring such that the path length along the ring between the centers of adjacent couplers is ¼ wavelength. Ideally, each coupler is ¼ wavelength long, although in typical applications, the coupled sections are made less than ¼ wavelength to allow space for terminations and transmission line bends. Given the simplifying case of port impedance being Z0at both Port1and Port2, all four couplers610,620,630and640have an identical length and coupling gap (illustrated by gap608), and are designed to have coupler port impedances of 2Z0. Impedance of the ring transmission line segments that connect between couplers is also, ideally, 2Z0.

Signal power incident to Port1is split onto the two transmission line paths604and605, each transitioning to a characteristic impedance 2Z0. These transmission lines deliver half of the input power to coupler610and half to coupler620. Couplers610and620are adjacent on the ring600, and are configured such that both couplers launch signal into the ring600in the same direction. Signals injected into the ring by coupler610will travel ¼ wavelength in the ring before reaching the electrical center of coupler620. Transmission line length of line605from the input splitter is made ¼ wavelength longer than the length of line604from the splitter so that the traveling wave introduced into the ring by coupler620will be aligned in phase with the traveling wave introduced by coupler610. The traveling waves introduced into the ring by the two couplers610and620therefore add constructively. Because of their ¼ wavelength spacing about the ring, and the 90 degree relative phase of their inputs, the two couplers610and620can be said to act “in quadrature” to reinforce the circulating traveling wave. The lines604and605are each terminated with real impedance 2Z0to absorb input power that does not couple into the ring.

A portion of the power circulating within the ring is coupled out by couplers630and640, and exits these couplers through lines606and607respectively. Signals exiting coupler640are delayed by ¼ wavelength relative to the signals exiting coupler630due to the circulating wave's direction of travel in ring resonator600and the relative position of coupler630and640around the ring. The transmission lines606and607have a characteristic impedance of 2Z0and join to form a single output602at impedance Z0at Port2. Transmission line606is made ¼ wavelength longer than the transmission line607so the coupled outputs will sum in-phase at the combiner port602. Combined output impedance presented at Port2is Z0. Coupler port lines606and607are each terminated with real impedance 2Z0to absorb power incident to Port2that does not couple into the ring.

The QCD filter circuit ofFIG. 6offers several advantages over previous filter circuits, including the ring resonator circuit ofFIG. 5. First, the “QCD Filter” couples signal into and out of the ring resonator using couplers that have twice the port impedance “2Z0” of the single couplers ofFIG. 5. For stripline and microstrip implementations, the higher impedance translates to greater coupling gap608for a given filter bandwidth, as opposed to the gap512ofFIG. 5. For a given bandwidth, the larger coupling gap reduces filter sensitivity to coupling gap tolerance. So, where minimum coupling gap tolerance is constrained by limits of lithography or other manufacturing processes, the QCD filter ofFIG. 6can yield filters with greater passband bandwidth and lower sensitivity to gap tolerance compared to filter circuitry such as shown inFIG. 5. Second, the QCD filter ofFIG. 6can yield practical microstrip and stripline filters with passband bandwidth great enough to eliminate the need for precise center frequency tuning. Third, the QCD filter ofFIG. 6reduces the magnitude of and sensitivity to spurious resonance modes that are excited by the presence of traveling waves circulating about the ring600in opposite directions as compared toFIG. 5. As such, it reduces the degree to which spurious resonance modes can affect passband shape (transmission vs. frequency characteristic). Finally, the QCD filter achieves lower signal reflection than does the prior art, particularly in the transition regions bordering the 3 dB passband.

Referring again back toFIG. 1, the output of the directional filter20, which can be the QCD filter ofFIG. 6, is amplified using amplifier22and then passed through a 1stIF bandpass filter24that is used mainly to suppress 1stIF signal power at the 1:1 image frequency of the baseband 2ndmixer26. The baseband 2ndmixer26combines the first IF signal output from the baseband 1stmixer16with a LO signal from the 2ndLO25to convert the 1stIF signal to a comparatively low 2ndIF frequency. Baseband 2ndmixer IF output is provided to switch28that will select either the 2ndIF signal from the baseband 2ndmixer26or from the highbands 2ndmixer42for application to the 2ndIF bandpass filter72.

The baseband filter24can be an air-dielectric multi-cavity filter shown in perspective view ofFIG. 7making it unique in hand-held spectrum analyzers. Compared to other types of bandpass filters commonly used for this purpose, the air-cavity filter has lower passband loss for a given amount of image rejection. High-Q and frequency precision of the air-cavity filter give sufficient selectivity to enable the use of a low-cost, low frequency 2ndIF, but with loss low enough to achieve an acceptable system input noise figure. The low loss of the air-cavity 1stIF filter, thus, improves the spectrum analyzer's overall noise figure. Previously, the use of such air-dielectric filters was limited to expensive, relatively non-portable laboratory equipment.

The air-dielectric multi-cavity filter ofFIG. 7includes a metal filter body700that is preferably a good conductor, or plated with a good conductor such as silver. The cavities are in the form of a cylindrical hollow with coaxial center posts702formed by machining out areas in the body700. The machined areas, thus, are air-dielectric filled. At resonance each post702acts like an inductor and the gap from the top of the post to the lid704acts like a capacitor. Taken together, the inductance and capacitance act like an LC tank circuit and set the resonant frequency. The machined cavities, if suitably excited, will resonate at a particular frequency that is determined by the physical dimensions of the cavity, cavity center post, and air gap from post to cavity cover704.

To excite the filter formed in the body700, a PCB trace706transmits a signal through a coupling element (not shown) into a first cavity of the body700. This coupling element is essentially a small antenna within the cavity. In the multi-cavity filter, the separate coupling elements transmit a signal from cavity to cavity. An aperture in the wall between cavities (not shown) can be used as the cavity-to-cavity coupling element. An output coupling element transfers signal power from the final cavity to a load, such as a PCB trace708. The filter body700may be mounted on a PCB substrate710as shown. The coupled cavities form a bandpass filter that passes a single frequency with low attenuation, and attenuates signals at other frequencies. The cavity resonators are effectively ganged together by being formed in the same metal body700. Prior art filters commonly used in handheld spectrum analyzers were formed by ganging together quarter-wave coaxial resonators having a ceramic dielectric. But such a ceramic filter can have insufficient selectivity or too great a signal loss for application within the present spectrum analyzer. The present spectrum analyzer has an unusually high ratio of 1stIF to 2ndIF frequency, which requires a low fractional bandwidth of the 1stIF filter24shown inFIG. 7, and which requires a 1stIF filter24to be fabricated with low-loss, high-selectivity resonators, such as air cavities. The filter ofFIG. 7is not to scale, in that the filter will be very small as realized for the 6.5 GHz 1stIF application.

Referring back toFIG. 1, the PIN diplex switch6has a second output separate from the baseband path described above to connect to the highband path. The highband path from PIN diplex switch6then is directed by switch30into either highband or midband paths.

A. PIN Diode Band Switched Filters

The “midbands” signal path is directed through midband RF bandpass filters32. A block diagram of the switches and filters making up bandpass filter32is shown inFIG. 8. The filters32include single-pole-three-throw (SP3T) PIN diode-type switches illustrated as switches800ofFIG. 8that selectively directs the midband signal through one of five filters804, each of the filters804covering a different portion of the midband. A similar switching circuit802selects one of the five filter outputs for application to the midbands 1stmixer. The use of multi-throw MMIC PIN switches for band switching is unique among hand-held spectrum analyzers. Prior-art hand-held spectrum analyzers did not use PIN MMICs for band switching because inexpensive SMT GaAs MMIC multi-throw switches are available for use at frequencies below 8 GHz. But multi-throw GaAs switches for use above 8 GHz were not commercially available at the time of this design.

To provide the multi throw switches800and802a cascade of the two SP3T PIN switches were used as illustrated inFIG. 9. The two SP3T switches900and902realize the SP5T switch800function with fewer series switching elements and fewer switch packages than would be possible using the more obvious alternative of packaged SPDT GaAs MMIC devices. The two SP3T cascade has lower insertion loss and material cost, thus reducing spectrum analyzer noise figure and cost.

In addition to the SP3T switch configuration, an innovative surface mount technology (SMT) package is preferably used to better facilitate use of MMIC PIN switch dies that make up the SP3T switches, such as the illustrated switches900and902inFIG. 9. Each individual SP3T switch die is inside a special SMT package that is attached to the present spectrum analyzer PCB. The filters interconnected by the SP3T switches are separately printed on the PCB to which the SMT switch packages are soldered. For example, the switch arrangement shown inFIG. 9would be done using two SMT packages, each containing a SP3T PIN switch MMIC. The switch902would most likely be fed by position1or3of switch900. Although described with individual SP3T switch packages for simplicity, higher levels of integration in the MMIC packaging are possible. The switch dies are mounted to a metal or ceramic substrate within the SMT package. The package provides a comparatively large ground pad for the dies, a microstrip interface from the wire bond pads to the substrate edges, and a protective cover.

The package provides a reliable, high-performance RF interface from MMIC to a PCB. The surface mount package facilitates mounting of the PIN switch dies to the spectrum analyzer PCB. The package provides a much larger ground patch and more widely spaced connections to the PCB traces than would be practical with chip-on-board assembly techniques. The package provides controlled-impedance lead-outs from the wire bond pads to the PCB connections at the package edges. The package protects the die and wire bonds, and facilitates assembly to a PCB using standard surface mount processes. The package improves switch isolation by reducing ground inductance and by increasing separation of PCB traces. The use of a multi-throw PIN switch, packaged in this manner, is unique among hand-held spectrum analyzers, and provides cost benefits compared to SMT GaAs SPDT switches in multi-throw switch applications.

Referring again back toFIG. 1, the filtered “midbands” signal from filters32combines with 1stLO signal from LO14in the midband 1stmixer34to produce a 1stIF frequency that is lower than the midbands RF input frequencies provided from the output of midband filters32. The IF signal from mixer34is filtered by a simple LC lowpass filter36that has a diplexed input termination to absorb high frequency mixing products The signal is then amplified by amplifier38and then filtered again in a ceramic coaxial resonator bandpass filter which rejects primarily signal power at the 2ndmixer421:1 image frequency. With the midband having a higher ratio input frequency to the 1stmixer34vs. 1stIF frequency output from mixer34, the more complex QCD filter20of the baseband path is not required. Further, the bandpass filter40can be a less complex ceramic filter than the air-dielectric multi-cavity filter24of the baseband. The switch41allows the midband and highband paths to both use the common 2ndmixer42and 2ndLO44. Switch28selects the 2ndIF output from either the highbands 2ndmixer42or the baseband 2ndmixer26for application to the 2ndIF amplifier70. Output of the 2ndIF amplifier is bandpass filtered by 2ndIF filter72. The 1stand 2ndLO frequencies can be chosen such that a single 2ndIF filter72can be used in conjunction with both the highbands 2ndmixer42and lowbands 2ndmixer26. Signal output from 2ndIF filter72can be converted to a lower frequency then digitized and analyzed, or can be digitized directly without further downconversion, then analyzed.

Referring back to the switch30, the highband signals are provided from the switch30to highband RF filters52. In the case of highband filter52, only two filters are desired as opposed to the five filters ofFIG. 32shown inFIG. 8, so the switches can be single pole double throw MMIC switches. Signal output from the selected highband filter52is applied to the highband 1stmixer54. Each of the filters in highband filters52suppresses primarily the 1:1 image frequencies of the 1stmixer54.

The “highbands” 1stmixer combines the filtered “highbands” RF signal with a frequency-doubled signal from the 1stLO14to produce a comparatively low 1stIF frequency. The 1st IF signal from mixer54is filtered using a simple LC filter56, that has a diplexed input termination to absorb high frequency mixing products, amplified by amplifier58, and then filtered by a ceramic coaxial resonator bandpass filter58. As with the midband path, in the highband path the QCD filter20and the air-dielectric multi-cavity filter24of the bandpass path are not required. An electronic switch41then directs the filtered 1stIF signal into the 2ndmixer42, where it is combined with 2ndLO44for conversion to a comparatively low 2ndIF frequency. The 2ndIF frequency will have a range matching that of the baseband path output to apply through switch28to the 2ndIF amplifier70. Output from 2ndIF amplifier is bandpass filtered by 2ndIF filter72. The signal output from 2ndIF filter72can then be converted to a lower frequency then digitized and analyzed, or can be digitized directly without further downconversion, then analyzed.

D. LO Frequency Doubler and Filters for HighBand Mixer

Embodiments of the present invention are unique in that they incorporate switch-selected multiple bandpass filters62in the doubled LO path applied to the high band mixer54. The frequency of the 1stLO14is doubled by frequency multiplier64, with the doubled LO output provided to highband LO filters62. Previous handheld spectrum analyzers that have a doubled 1stLO used only a single bandpass filter in the doubler output path. For a given attenuation of VCO fundamental and 3rdharmonic, the multiple switched filters62can collectively pass a greater range of the doubled VCO frequency than would a single bandpass filter. The switched filters62can have a configuration similar toFIG. 8, and can use MMIC SPDT switches. The multiple filters62collectively pass a wider range of doubler output frequency than would be possible with a single bandpass filter that provides similar suppression of the fundamental and 3rdharmonic of the doubler input frequency. The bandpass filters62suppress primarily the fundamental and 3rdharmonic of the VCO frequency so as to reduce associated spurious mixing products in the “highbands” 1stmixer. Suppression of output frequencies of the doubler64other than 2*(doubler input frequency) is necessary to minimize spurious frequency products in the mixer54that uses this frequency-doubled signal as it's LO. By incorporating multiple switch-selected filters62in its 1stLO frequency output that is doubled by doubler64, the system is able to achieve greater useful frequency range from its highbands” 1stmixer54than would otherwise be practical while still suppressing spurious outputs of the doubler64. RF input frequency range for the highband path mixer54is correspondingly increased by the greater LO range, without compromise to the spectrum analyzer's RF-related spurious specification.

Embodiments of the present invention are further unique among hand-held microwave spectrum analyzers in that they use three 1stmixers16,34and54. The midbands and highbands 1stmixers34and54each operate across an input frequency range of approximately one octave, resulting in low conversion loss. Splitting the spectrum analyzer input frequency range among the three 1stmixers16,34and54allows each mixer to process a significantly narrower frequency range than would be possible with fewer mixers. Reducing mixer frequency range in this way results in reduced conversion loss, and therefore reduced system noise figure. By comparison, a prior-art two-mixer design, simply scaled in frequency, would require much greater frequency range from one or both of its 1stmixers, resulting in higher conversion loss, and therefore higher system noise figure.

IV. Spectrum Analyzer Output with Selectable 2ndIF Bands

The 2ndIF output from either the “baseband” or “highbands” 2ndmixer42is directed to a 2ndIF amplifier70, the output of which is then filtered by a bank of 3 selectable SAW bandpass filters72. The filters can be connected using SP3T switches in the configuration shown inFIG. 8. The SAW filters72suppress the 1:1 image and higher order products of a 3rdmixer (not shown) that can be used for downconversion of the 2ndIF signal to a signal frequency acceptable for analysis by a digital signal processor. The SAW filters72can also serve to limit input bandwidth to the 3rdIF mixer. In the case where there is not a 3rdfrequency conversion prior to signal digitization, the selectable SAW 2ndIF filters72serve to suppress unwanted image products in the digitizer and to limit the bandwidth of signals incident to the digitizer. The multiple filters72allow more than two selectable 2ndIF bandwidths. The multiple filter selections allow 2ndIF bandwidth to be optimized for any of several digital modulation formats, including UMTS, DVB-T, and WIMAX. Bandwidth can be selected to admit one signal channel of interest while blocking much of the power from other signal channels that are spectrally close-by. By blocking much of the unwanted spectrum, the selected filter reduces inter-modulation distortion produced in the 3rdmixer, 3rdIF amplifiers, and 3rdIF digitizer. This improves system ACLR (adjacent channel leakage ratio) when measuring a signal in a spectrum of closely spaced signals. The performance advantage realized by incorporation of 3 or more selectable 2ndIF bandwidths also applies to variants of embodiments of invention in which the 2ndIF signal is digitized without an intermediate 3rdmixer stage.

Although the present invention has been described above with particularity, this was merely to teach one of ordinary skill in the art how to make and use the invention. Many additional modifications will fall within the scope of the invention, as that scope is defined by the following claims.