High resolution, high rate X-ray spectrometer

A pulse processing system (10) for use in an X-ray spectrometer in which a ain channel pulse shaper (12) and a fast channel pulse shaper (13) each produce a substantially symmetrical triangular pulse (f, p) for each event detected by the spectrometer, with the pulse width of the pulses being substantially independent of the magnitude of the detected event and with the pulse width of the fast pulses (p) being substantially shorter than the pulse width of the main channel pulses (f). A pile-up rejector circuit (19) allows output pulses to be generated, with amplitudes linearly related to the magnitude of the detected events, whenever the peak of a main channel pulse (f) is not affected by a preceding or succeeding main channel pulse, while inhibiting output pulses wherein peak magnitudes of main channel pulses are affected by adjacent pulses. The substantially symmetrical triangular main channel pulses (f) are generated by the weighted addition (27-31) of successive RC integrations (24, 25, 26) of an RC differentiated step wave (23). The substantially symmetrical triangular fast channel pulses (p) are generated by the RC integration ( 43) of a bipolar pulse (o) in which the amplitude of the second half is 1/e that of the first half, with the RC time constant of integration being equal to one-half the width of the bipolar pulse.

BACKGROUND OF THE INVENTION 
The present invention relates generally to high resolution, high rate x-ray 
spectrometers and more particularly to systems for processing the pulses 
which are generated in response to detected radiation. The United States 
Government has rights in this invention pursuant to Contract No. 
DE-ACO3-76SF-00098 between the U.S. Department of Energy and the 
University of California. 
The pulse processing system of the present invention was designed in 
response to the need for an x-ray spectrometer capable of use in plasma 
diagnostics in a fusion reactor. High temperature plasmas produced in such 
reactors emit considerable black body radiation in the x-ray energy level 
up to several tens of kilovolts. Measurement of this flux is important in 
determining the plasma temperatures and in detecting the presence of 
impurities which produce characteristic x-ray lines. 
In such diagnosis, spatial and temporal variations in the radiant energy 
must be measured. Since temporal variations during the plasma pulse must 
be observed on a millisecond time scale, the solid angle of the detectors 
observing the plasma must be large enough to give counting rates adequate 
for statistically meaningful spectra to be accumulated on a time scale of 
a few milliseconds. 
The combination of very high counting rates and the excellent energy 
resolution requirement for the observation and separation of impurity 
spectral lines presents very difficult design problems. The final 
usefulness of the spectrometer requires that a high rate of analyzable 
pulses be passed by the signal processing system. Rejection of detected 
events must be minimized and a high throughput must be realized. 
As a further complication, since the whole purpose of a fusion reactor is 
to produce thermonuclear reactions, the spectrometer must cope with 
significant production of 14 MeV neutrons. The estimated maximum flux of 
such neutrons through the spectrometer detectors is in the order of 
5000/second. Such neutrons will interact by colliding with the silicon or 
germanium nuclei in the semiconductor detectors, with resulting signals 
ranging up to a few MeV. The signal processing system must be able to 
handle the large signals and recover very quickly to process the x-ray 
signals in the energy range from 1 keV to 50 keV. 
Typically, the semiconductor detectors will detect single radiation events 
(photons or charged particles) and produce impulses of current into a 
preamplifier which will output detected signals as step waveforms. These 
signals will then be processed by pulse shaping circuits to provide pulses 
having an optimal signal-to-noise ratio and having amplitudes which are 
linearly relates to the energy absorbed by the detectors due to the 
detected events. To cope with high counting rates, the total width of the 
pulses must be minimized. 
In the environment for which the present invention is intended, the very 
short pulse widths involved make series (delta) noise, i.e. the "shot" 
noise of the input amplifier, the dominant noise in the pulse shaping 
system. 
It is known that series (delta) noise is dependent on the rate of change of 
the step response in the shaping system. It is also known that a 
symmetrical triangular pulse will have a minimum and constant slope for a 
given amplitude and will thus have an optimum shape for increasing the 
signal-to-noise ratio. Any other pulse shape, such as the Gaussian shape 
commonly produced in existing pulse shaping circuits will have a poorer 
signal-to-noise ratio since a pulse with a lower slope along some portion 
of its length must have a higher (and noiser) slope along another portion 
if it is to reach to the same maximum amplitude in the allotted time. 
Although it has been known that symmetrical triangle pulse shaping would be 
beneficial in reducing series (delta) noise, suitable apparatus for 
producing such pulses of the character necessary for use in the present 
invention has not been heretofore devised. 
Symmetrical triangle pulse generation has been achieved by proper 
integration of a symmetrical biphase delay line pulse, but such delay 
lines are bulky, are not easily varied in their time scale, and cause 
severe sensitivity of gain to temperature variations. As a consequence, 
they are not practical circuit elements for the main channel of a pulse 
processing system. 
All modern spectrometer pulse processors use a method to prevent the 
analysis of pulses whose amplitude is subject to interference by other 
signals in close time proximity. Generally speaking, this function is 
achieved by a "pile-up rejector" containing four elements: (a) a gate at 
the output of the main pulse processing channel; (b) a parallel fast 
inspection channel where a short duration signal is generated in response 
to each detected event; (c) a fast discriminator which produces a logic 
signal having an output width corresponding to the time that the fast 
inspection channel pulses exceed the fast discriminator threshold level; 
and (d) a pile-up detector which examines the fast discriminator logic 
signals, and, by measuring the time between these signals, senses whether 
two main pulses may distort the signal amplitude of each other--if not, 
both of the main pulses are gated through to an output; if so, one or both 
pulses are not gated through. 
The accuracy of the pulse processor will depend to a large extent on the 
"resolving time" of the system, i.e. on the width of the logic signal 
outputted by the fast discriminator, which width is determined by the 
length of time that the fast channel pulses exceed the fast discriminatory 
threshold level. If two detected events occur within this resolving time, 
the fast inspection channel cannot recognize them as separate signals and 
both main channel pulses will be gated through together to the output. 
This will result in a number of output pulses whose amplitude is the sum 
of two (or more) separate signals, causing the output spectrum to contain 
"sum" peaks. In the present environment, these sum peaks will distort the 
thermal black body spectrum seen from hot plasma discharges because some 
counts that should appear in the intense low energy part of the spectrum 
will be shifted into the weak high energy part of the spectrum. Such 
distortion of the black body radiation will affect the measurement of the 
plasma temperature. 
If the resolving time were constant and reasonably well known, an 
approximate correction could be applied to the continuum spectrum to 
compensate for this type of pulse pileup. Unfortunately, existing 
spectrometers fall far short of meeting the criteria of providing a well 
determined and consistent resolving time. Typically the fast channel 
pulses are produced by integration or with a single delay line or simple 
RC pulse shaping. As a consequence there will be a long exponential tail 
on the back edge of the fast channel signal waveform, causing the 
resolving time to be very dependent on pulse amplitude. Such shapes of 
existing fast channel pulses are thus not desirable in environments where 
a wide and unknown dynamic range of pulse amplitudes is to be measured. 
In addition, most existing systems are disadvantageous in that they provide 
fast channel pulse shaping having a poor signal-to-noise ratio, requiring 
the fast discriminator level to be set at a high level in order to reduce 
noise triggering. 
In order to achieve maximum throughput, the system should operate to gate 
all main channel pulses through which do not actually distort each other. 
At times a second main channel pulse will begin during the time that the 
preceding main channel pulse is decreasing from its peak, with the peak of 
the second pulse occurring after the first pulse has ended. Desirably the 
peaks of both of these main channel pulses should be gated through to the 
output since neither peak is affected by the other. However, existing 
systems do not permit this, generally because of the fact that the main 
channel pulse stretcher waits until the tail of the first pulse reaches a 
low threshold level before permitting the stretching of a normal pulse. 
SUMMARY OF THE INVENTION 
It is an object of the invention to provide a pulse processing system for 
use with detected signals of a wide dynamic range which is capable of very 
high counting rates, with high throughput, with excellent energy 
resolution and a high signal-to-noise ratio. 
It is a further object of the invention to provide a pulse processing 
system wherein the fast channel resolving time is quite short and 
substantially independent of the energy of the detected signals. 
Another object of the invention is to provide a pulse processing system 
having a pile-up rejector circuit which will allow the maximum number of 
non-interfering pulses to be passed to the output. 
It is also an object of the invention to provide new methods for generating 
substantially symmetrically triangular pulses for use in both the main and 
fast channels of a pulse processing system. 
Additional objects, advantages and novel features of the invention will be 
set forth in the description which follows, and in part will become 
apparent to those skilled in the art upon examination of the following or 
may be learned by practice of the invention. The objects and advantages of 
the invention may be realized and attained by means of the 
instrumentalities and combinations particularly pointed out in the 
appended claims. 
To achieve the foregoing and other objects and in accordance with the 
present invention, as embodied and broadly described herein, a pulse 
processing system is provided, such system having a first pulse generator 
for generating substantially symmetrical triangular main pulses in 
response to detected signals, the triangular pulses having amplitudes 
proportional to the magnitudes of the detected signals and substantially 
constant total pulse widths, a second pulse generator for generating 
substantially symmetrical triangular fast channel pulses in response to 
the detected signals with the pulse widths of the fast channel pulses 
being substantially constant, substantially shorter than the main channel 
pulses, and substantially independent of the magnitude of the detected 
signals, and a pulse rejector system which will allow the peaks of 
successive main channel pulses through whenever the time duration between 
fast channel pulses is greater than a time which is equal to the longest 
of the rise or fall of the main channel pulses. 
In order to generate substantially symmetrical triangular pulses in the 
main channel of the pulse processing system, a differentiated waveform 
generated in respone to a step waveform of a detected signal is 
successively integrated by active RC integrators, and a weighted addition 
of three successive integrated waveforms is used. 
In order to generate substantially symmetrical triangular pulses in the 
fast channel of the pulse processing system in response to step function 
waveforms from detected signals, a single rectangular pulse is produced 
for each step function waveform, a bipolar pulse is then produced with the 
amplitude of the second half of the pulse being approximately 1/e times 
that of the first half of the pulse, and the bipolar pulse is than 
integrated with the RC integration time being equal to one-half the width 
of the bipolar pulse.

DETAILED DESCRIPTION OF THE INVENTION 
Referring now to the drawings, which illustrate a preferred embodiment of 
the invention, the pulse processing system 10 of the present invention is 
shown in conjunction with a sensor and preamplifier 11 of an x-ray 
spectrometer. The sensor and preamplifier 11 is shown only in block for 
since the details thereof are not part of the present invention. 
Typically, for plasma diagnostics, the sensor will include semiconductor 
crystals of germanium and silicon. Because of its great mass, the 
germanium will be effective in the absorption of high energy radiation, 
with the use of silicon being more advantageous at lower energy levels. 
The output of the preamplifier will be a series of step waveform signals 
(a) whose amplitude is linearly related to the amount of energy absorbed 
due to each detected event. 
Each signal (a) is then applied to the main channel triangular pulse shaper 
12 and the fast channel triangular pulse shaper 13. As will be brought out 
in more detail below, the main channel triangular pulse shaper will 
generate a substantially symmetrical triangular pulse (f) for each 
detected signal (a), with each pulse (f) having a substantially constant 
pulse width and an amplitude linearly related to the magnitude of the 
detected signal (a). 
At the same time, each detected signal (a) is also applied to the fast 
channel triangular pulse shaper 13 which generates a substantially 
symmetrical triangular pulse (p) for each detected signal, the pulses (p) 
having a substantially constant pulse width which is essentially 
independent of the magnitudes of the detected signals (a) and which is 
much shorter (e.g. in the order of 1/50 th) than the pulse width of the 
main channel pulses (f). 
The main channel pulses f are applied to a "track-hold" circuit 14 which 
functions to follow the pulses (f) and hold the peak value thereof. This 
held value is then passed through a linear gate 16 and amplified by output 
amplifier 17 so that an output pulse is produced at output 18, with the 
amplitude of the output pulse being linearly related to the energy level 
of the detected event. The output pulses will then be sent to a 
pulse-height analyzer (not shown) where the amplitude spectrum of the 
output pulses will be determined. 
The fast channel pulses (p) are applied to a pile-up detector circuit 19 
which controls the track-hold and linear gate circuits 14 and 16 so that 
only non-interfering main channel pulse peaks will be passed through to 
the output 18. 
The main channel triangular pulse shaper 12 functions to produce a 
substantially symmetrical triangular pulse by a weighted mixing of the 
outputs of stages that are already present in shaping amplifiers used to 
produce Gaussian pulse shapes, and thus uses convenient active integrators 
whose behavior is suitable for stable amplifiers (unlike delay lines). 
In particular, the step waveform (a) is first applied to the capacitor 21 
and resistor 22 of the operational amplifier 23 which functions as a 
differentiator to produce the differentiated waveform (b). The 
differentiated waveform is then successively integrated by operational 
amplifier integrating stages 24, 25 and 26. The integrating stage 24 
functions as an inverting stage while stages 25 and 26 are of the 
non-inverting type. Analytically, the stages are analogs of each other, 
all having their poles about 27.3.degree. off the real axis. 
The waveforms (c), (d) and (e) from the outputs of the integrator stages 
24, 25 and 26 are then applied through weighting resistors 27, 28 and 29 
to adder 31. Preferably, the resistors 27, 28 and 29 are chosen in value 
so that the outputs of the three integrator stages are mixed in the ratio 
of 0.324:0.168:1. FIG. 5 shows the three component waveforms (c), (d) and 
(e), with amplitudes corresponding to the weighting, and the resulting 
substantially symmetrical triangular wave (f) produced by adding these 
three components and normalized to unity amplitude. A true symmetrical 
triangle (g), closely matching the sides of waveform (f) is also shown in 
FIG. 5. 
As is indicated in FIG. 5, the actual output waveform (f) departs from a 
true symmetrical triangle (g) in two significant respects. First, the peak 
is rounded. This factor increases the parallel (step) noise slightly but 
is also a necessary feature for good operation of the pulse stretcher used 
at the output of the main channel of the system. Since parallel noise is 
negligible in the present system, virtually no loss of performance results 
from the rounded peak. 
Secondly, a slightly rounded tail occurs on the output pulse (f). This is 
undesirable since it increases slightly the pulse pile-up effects, but is 
unavoidable using passive (i.e. time-invariant) networks. A choice of the 
integrator and differentiator stages 23, 24, 25 and 26 is based on 
minimizing this effect. 
The effect of the present invention on noise performance can be shown by 
comparing the waveform (f) of the present system with the performance of a 
system producing the quasi-Gaussian waveform (e) which is typical of the 
pulse shapes of many present spectroscopy systems, bearing in mind that 
the integral of the slope is the important factor is series (delta) noise. 
There is little difference between waveforms (e) and (f) on the falling 
part of those waveforms, but the rising slope of waveform (f) is 
essentially constant and much less than the steepest part of the rising 
slope of waveform (e). This avoidance of excess slope in the fundamental 
reason for the improved delta noise of the present invention. 
Approximately 8% improvement in delta noise can be obtained. 
In the fast channel 13, each step signal (a) is applied through a diode 
clamp 36 to the inputs of operational amplifier 37. With delay line 38 in 
the negative input, amplifier 37 will perform a subtractive operation 
between the input step function waveform (m) from the diode clamp 36 and a 
delayed version of that input. The resulting output from amplifier 37 is a 
single phase rectangular pulse (n). This pulse is than applied to the 
inputs of operational amplifier 39, with delay line 41 and variable 
resistor 42 functioning so that the amplifier 39 performs a subtraction 
between waveform (n) and a delayed and adjustable amplitude version of 
waveform (n) which is delayed by its width. The output of amplifier 39 is 
a bipolar pulse (o), whose second half amplitude is 1/e times that of its 
first half, "e" being the base of the natural logarithmic system. 
The bipolar pulse (o) is then applied to integrator stage 43 having values 
of resistor 44 and capacitor 45 such that the RC time constant is equal to 
one-half the width of bipolar pulse (o). The resulting output of 
integrator 43 is a waveform (p) having a substantially symmetrical 
triangular shape. 
As illustrated in FIGS. 3 and 4, the shape of the fast channel pulses (p) 
will enable to system to have a very short and substantially constant 
resolving time as compared with existing spectroscopy systems. 
Typically, existing systems use inherent integration (usually in the 
preamplifier) and single delay line or simple RC pulse shaping wherein a 
single phase pulse 46 is integrated to form a fast pulse 47 having a long 
exponential tail on the back edge of the waveform. As mentioned before, 
the resolving time of the fast channel is the time when the amplitude of 
the fast channel pulses exceeds the discriminator threshold level. In FIG. 
3, the resolving time for the fast channel pulse 47 is indicated by 
T.sub.R1. If the single phase pulse 48 has a greater amplitude (because of 
a greater magnitude of detected event), the corresponding fast pulse 49 
will have a greater amplitude, causing a significantly longer time for 
exponential decay to the fast discriminator threshold level and resulting 
in a substantially greater resolving time T.sub.R2. 
As is seen in FIG. 4, in the present invention the magnitude of the first 
halves of the bipolar pulses 51 and 52 will likewise cause the magnitude 
of the corresponding fast pulses 53 and 54 to vary in accordance with the 
magnitude of the detected event. However, the negative second halves of 
the bipolar pulses 51 and 52 will force a quicker decay of the fast pulses 
53 and 54. As shown in FIG. 4, the fast pulses 53 and 54 both decay to the 
zero baseline at a time equal to the total bipolar pulse width, causing 
the resolving times T.sub.R1 and T.sub.R2 to be substantially the same for 
greatly varying energy levels in the detected signals. As may be seen, 
such resolving times are considerably less than the resolving times shown 
in FIG. 3. 
Also, as described in connection with the main channel 12, the symmetrical 
triangular shape of the fast pulses will give the best signal-to-noise 
ratio where series (delta) noise is dominant, as is always the case in the 
fast channel 13. The reduction in noise is advantageous since it enables 
the fast discriminator threshold level to be maintained at a relatively 
low level without noise triggering, thereby enabling the system to be used 
with relatively weak levels of detected energy. 
The pile-up rejector portion 19 of the present invention is shown in more 
detail in FIG. 2. As mentioned before, the symmetrical triangular 
waveforms (f) from adder 31 are applied to amplifier 61 of track-hold 
circuit 14. Normally, switch 62 maintains the hold capacitor 63 connected 
to the output of amplifier 61 so that the voltage across the hold 
capacitor 63 will follow the rising and falling voltage of the waveform 
(f). 
When a detected event occurs, a fast channel pulse (p) will be applied 
through the base line restore circuit 66 to the input of the fast 
discriminator 67, causing an output pulse (q) therefrom having a pulse 
width equal to the time that the magnitude of the fast signal (p) exceeds 
the fast discriminator threshold set by variable resistor 68. The base 
line restore circuit 66 functions in a conventional manner to maintain the 
baseline of the fast pulses at zero. The trailing edge of the fast 
discriminatior pulse (q) will trigger the T.sub.2 one-shot 69 to generate 
a single pulse lasting for a time T.sub.2, which time is chosen to be 
equal to the longest of the rise or fall times of a slow channel 
triangular pulse (f). 
Normally, the T.sub.2 one-shot 69 holds the pile-up flip-flop 71 in reset 
condition so that its output applies a high input to AND gate 72. When the 
T.sub.2 one-shot pulses, the reset voltage is removed from flip-flop 71. 
If there is another fast discriminator pulse (q) during the T.sub.2 
one-shot pulse, the leading edge of that fast discriminator pulse can 
clock the flip-flop, causing the output thereof to go low. 
When not pulsing, the T.sub.2 one-shot 69 prevents the zero-cross 
discrimination 73 from operating so that it cannot respond to noise, but 
enables such operation during the time duration of the T.sub.2 pulse. 
The output from integrator stage 25 of the main channel is applied through 
an RC differentiator (capacitor 74 and resistor 75) to the input of the 
zero-cross discriminator 73 which produces a peak sensing signal (z) at 
the time when the main channel waveform (f) reaches peak value. The peak 
sensing signal (z) triggers the 0.1 microsecond one-shot 76, which in turn 
applies a high output to AND gate 72. A high output of AND gate 72 will 
trigger the two-microsecond stretch one-shot 77 whose output is then 
applied for two microseconds through OR gate 78 to open switch 62. The 
opening of switch 62 occurs when the slow channel waveform (f) peaks, and 
this peak voltage will then be held by capacitor 63 until switch 62 is 
later closed. 
The two microsecond output pulse from the stretch one-shot 77 will 
sequentially trigger the 0.5 microsecond delay one-shot 79, the 0.1 
microsecond one-shot 80, and the 0.75 microsecond one-shot 81. The outputs 
of the latter one-shot are connected to linear gate 16 so that 0.5 
microseconds after the main channel pulse (f) has peaked, linear gate 16 
will operate for 0.75 microseconds to pass the peak voltage held by 
capacitor 63 through to the output amplifier 17. The pulse from the 0.75 
microsecond one-shot 81 is also applied through the OR gate 78 to ensure 
that switch 62 is held open during operation of linear gate 16. 
FIG. 6 illustrates the manner in which the present system will inhibit 
interfering output pulse peaks while allowing the maximum of 
non-interfering pulse peaks to pass through. In particular, FIG. 6 
illustrates the occurrence of a first main channel pulse (f.sub.1) and a 
second main channel pulse (f.sub.2) occurring at four different times 
after the first main channel pulse. To simplify the diagram, an 
asymmetrical triangular waveform is shown, with a rise time of T.sub.1 and 
a somewhat longer fall time of T.sub.2. 
Case A represents a situation wherein the start of the second main channels 
pulse (f.sub.2) occurs during the rise time of the first pulse (f.sub.1) 
and the peak of the second pulse occurs during the fall time of the second 
pulse. It is clear that the peak amplitudes of both the first and second 
pulse (f.sub.1) and (f.sub.2) are each affected by the other pulse, so 
that both pulse peaks must be rejected to prevent pulse interferences. 
As described above, the fast channel pulse (p) corresponding to the first 
main channel pulse will cause the fast discriminator 67 to generate pulse 
(q.sub.1), to trigger T.sub.2 one-shot 69. The pulse from one-shot 69 
enables the zero-cross discriminator 73 and removes the reset voltage from 
pile-up flip-flop 71. With the next detected event ocurring during the 
rise of the first main channel pulse (f.sub.1), a second fast channel 
pulse will be generated and a second fast discriminative pulse (q.sub.2A) 
will be produced. Since this second fast discriminator pulse (q.sub.2A) 
occurs during the T.sub.2 one-shot pulse, the front edge of the second 
fast discriminator pulse (q.sub.2A) will clock the flip-flop 71, causing 
it to output a low to AND gate 72. This low will be maintained until 
flip-flop 71 is again reset. The rear edge of the second fast 
discriminator pulse (q.sub.2A) will retrigger the T.sub.2 one-shot 69. 
In due course the zero-cross discriminator will generate pulse (z.sub.1), 
when the first main signal (f.sub.1) peaks. However, with the pile-up 
flip-flop 71 holding AND gate 72 closed, the two microsecond stretch 
one-shot 77 will not be triggered. Likewise, linear gate 16 will not be 
enabled. 
During the retriggered operation of the T.sub.2 one-shot 69, the zero-cross 
discriminator will generate pulse (z.sub.2A) at the time that the second 
main channel pulse (f.sub.2A) peaks. However, since this will occur during 
T.sub.2 time, the clocked flip-flop 71 will prevent the stretch one-shot 
77 from being triggered by the second zero-cross discriminator pulse 
(z.sub.2A). 
In due course, if no further detected signal occurs, the T.sub.2 one-shot 
will time out to reset the pile-up flip-flop 71, and thereby enable the 
AND gate 72. 
Case B represents a situation wherein the second main channel pulse 
(f.sub.2B) starts after the peak of the first pulse (f.sub.1) and peaks 
during fall of the first pulse. In such case, the peak of the first pulse 
will not be affected by the second pulse, but the peak of the second pulse 
will be affected by the first pulse. In this case it is desired to have an 
output pulse corresponding to the peak of the first main pulse (f.sub.1) 
while inhibiting an output pulse corresponding to the peak of the second 
main pulse (f.sub.2B). 
The first fast pulse will cause a first fast discriminator pulse (q.sub.1), 
the trailing edge of which triggers the T.sub.2 one-shot 69. In due 
course, the zero-cross discriminator 73 will generate pulse (z.sub.1) when 
the first main signal (f.sub.1) peaks. Since the pile-up flip-flop 71 has 
not yet been clocked by the second fast discriminator pulse (q.sub.2B), 
the zero cross discriminator pulse (z.sub.1) will trigger the stretch 
one-shot 77 and cause the peak of the first signal (f.sub.1) to be gated 
through to the output amplifier 17. 
The second fast discriminator pulse (q.sub.2B), coming during the T.sub.2 
one-shot pulse, will clock the pile-up flip-flop 71 and retrigger the 
T.sub.2 one-shot pulse. Accordingly, the AND gate 72 will be closed 
against the second zero-cross discriminator pulse z.sub.2B and no output 
pulse will be produced by the second main channel pulse (f.sub.2B). 
In case C, the second main channel pulse (f.sub.2C) starts after the peak 
of the first pulse (f.sub.1) and peaks after the first pulse ends. In this 
case, neither pulse interferes with the peak of the other pulse and output 
signals corresponding to both peaks should be produced. 
In the present system, the zero-cross discriminator pulse (z.sub.1) for the 
first main pulse occurs during time T.sub.2 and before the second first 
discriminator pulse (q.sub.2C). As a consequence, the pile-up flip-flop 71 
will hold AND gate 72 high so that the first zero-cross discriminator 
pulse (z.sub.1) will trigger the stretch one-shot 77 and cause the linear 
gate 16 to operate. 
The second fast discriminator pulse (z.sub.2C) will occur after the T.sub.2 
one shot 69 has timed out (which in turn caused reset voltage to be 
applied to flip-flop 71). With flip-flop 71 now held in reset condition, 
it cannot be clocked by the second fast discriminator pulse (q.sub.2C). 
Instead, the trailing edge of the second fast discrimination pulse 
triggers the T.sub.2 one-shot 69. The zero-cross discriminator is again 
enabled to generate pulse (q.sub.2C) at the peak of the second main 
channel pulse (f.sub.2C) and to trigger the stretch one-shot 77 so that an 
output pulse will be produced with an amplitude corresponding to the 
magnitude of the second main channel pulse (f.sub.2C). 
Case D is a situation wherein the second main pulse (f.sub.2D) does not 
begin until after the first main channel pulse ends. Since neither pulse 
affects the peak of the other, separate output pulses should be allowed 
through. 
Since the second fast discriminator pulse (q.sub.2D) does not occur until 
after the first pulse of the T.sub.2 one-shot 69, then the system will 
operate in the same manner as described in connection with case C. 
Thus, with the present invention, separate output pulses will be produced 
whenever there is more than T.sub.2 time between the peaks of the two 
pulses. With T.sub.2 being slightly more than the rise time of the main 
pulses, then the throughput of the system can be as much as 40% greater 
than previous systems which allowed separate pulses through only if the 
second main pulse began after the first main pulse ended. 
The present system also utilizes feedback limiters in the amplifying 
stages, one of the integrator stages and in the final output stage. These 
feedback limiters play an important role in minimizing the system recovery 
time following large overload pulses produced by fast neutrons in the 
detectors. 
The foregoing description of a preferred embodiment has been presented for 
purposes of illustration and description. It is not intended to be 
exhaustive or to limit the invention to the precise form described, and 
obviously many modifications and variations are possible in light of the 
above teaching. The embodiment was chosen and described in order to best 
explain the principles of the invention and its practical application to 
thereby enable others in the art to best utilize the invention in various 
embodiments and with various modifications as are suited to the particular 
use contemplated. The above description has been directed to a pulse 
processing system in an X-ray spectrometer for measuring 
plasma-temperatures and detecting impurities in a high temperature fusion 
test reactor, and in this intended use the system meets the requirements 
of high counting rate and increased throughput, high signal-to-noise 
ratio, large dynamic range, and excellent resolution in the presence of a 
high energy neutron background. However, the systems of the present 
invention can also be used in other instruments wherein some or all of 
these advantages are needed. It is intended that the scope of the 
invention be defined by the claims appended hereto.