Power factor controller

A power factor controller for a motor is of the type having a bidirectional triode thyristor and a circuit for sensing the voltage across the thyristor to determine the time by which the current lags the line voltage and for generating a signal voltage across an integrating capacitor that is inversely proportional to the current lag time. In addition, a voltage ramp generator produces a ramp voltage having a slope that increases as the lag time decreases. A sum of the signal voltage and ramp voltage is applied to the input of a threshold detector that triggers firing of the thyristor when a predetermined threshold is reached. The entire controller employs only five integrated switches or gates and two individual transistors, and power dissipation, size and cost are thereby minimized.

BACKGROUND OF THE INVENTION 
The present invention relates to a power factor control system for AC 
induction motors and more particularly to such a system that 
electronically varies the duration of each cycle of AC power that is 
applied to an induction motor winding inversely with the phase lag angle 
of the winding current with respect to the applied voltage. 
Such a system is described by F. Nola in the patent U.S. Pat. No. 4,052,648 
issued Oct. 4, 1977. A high power resistor is placed in series with the 
motor winding to develop a voltage that is exactly in phase with the motor 
winding current. This resistor voltage and the line voltage are compared 
in an electronic circuit, e.g. including 8 operational amplifiers, that 
controls the firing of a bidirectional triode thyristor or TRIAC 
(Trademark of General Electric Company, Syracuse, N.Y.) that in turn 
controls delivery of the AC power to the motor winding. The above noted 
lag angle of the motor current is sensed and maintained essentially 
constant by this controller. 
In a NASA publication entitled, Improved Power-Factor Controller, Brief No. 
MFS-23280, summer 1980, there is described a lower cost design than that 
in the patent. Three transformers and the high power resistor of the 
patent are eliminated and fewer components are used. However, two DC power 
supplies at +15 V and -15 V, respectively, operate from the AC power line 
and require two large filter capacitors. Also, only six operational 
amplifiers are employed. 
These controller circuits have been adapted for insertion between a motor 
driven appliance (e.g. refrigerator, freezer, fan, etc.) and the home 
power outlet. They typically sell for $35. For a typical refrigerator 
operating at 10% duty factor, the use period required for the dollar 
savings in electrical energy to equal the cost of the power factor 
controller is at $0.08/Kwh almost 3 years. The installation of such a 
power factor controller with each of the several billion electric motors 
presently in use and with each of the approximately 50 million motors 
manufactured each year is clearly desirable and consistent with public 
policy for conserving energy, but is generally not yet cost effective in 
the home. 
It is an object of the present invention to provide a power factor 
controller that is substantially simpler and less costly. 
It is a further object of the present invention to provide such a 
controller that itself dissipates less energy and may be realized almost 
entirely in integrated circuit form. 
It is yet a further object of the present invention to provide such a 
controller that is capable of being formed as an integral part of a power 
cord for an electrical appliance. 
It is also an object of the present invention to provide such a controller 
that is suitable for permanent incorporation in a motor housing. 
SUMMARY OF THE INVENTION 
A power factor controller, intended for interposing between an induction 
motor and an AC power line has a thyristor switching means for connecting 
the motor to the line at each occurrence of a trigger voltage and 
disconnecting the motor at each zero crossing of the motor current. 
The trigger voltage is produced by a threshold detector means when the 
input voltage applied thereto exceeds a predetermined value. 
The controller further includes an integrating capacitor, an integrating 
means for generating a voltage across the integrating capacitor which 
voltage has a magnitude that varies inversely with the lag time by which 
each zero-crossing of the motor current lags the previous zero-crossing of 
the power line voltage. Also included is a ramping capacitor and resistor 
forming a series circuit which circuit is connected across the integrating 
capacitor so that the integrating capacitor voltage determines the slope 
of the ramp voltage developed across the ramping capacitor. A resetting 
means is for periodically discharging the ramping capacitor at each 
zero-crossing of the AC line voltage and the integrating capacitor voltage 
is the above-noted input voltage of the threshold detector. Thus, when the 
current lag time decreases, both the integrating capacitor voltage and the 
slope of the ramp voltage increases and for both reasons the threshold 
detector turns on earlier. This high gain relationship makes possible a 
considerable economy in needed circuit components. Further toward economy 
of size and of manufacturing cost, no linear amplifiers are needed. The 
preferred embodiment employs only two transistor switches and five 
integrated circuit gates or switches. 
Operation of power factor controllers of this and the foregoing type 
depends upon the fact that the power factor of a lightly loaded induction 
motor may be improved by decreasing the applied voltage. Lightly loaded 
induction motors normally have a very low power factor (the current lag 
angle is large or from another view point the time lag by which the 
zero-crossing of the motor current relative to the preceding zero-crossing 
of the applied voltage is large). At full voltage and at the full rated 
mechanical load of the motor, the power factor will be high, e.g. 0.95. As 
the mechanical load is increased, the power factor often reaches a maximum 
and for most motors from there declines. Eventually the motor stalls, 
i.e., is in a locked rotor condition. It is important to provide full 
voltage under locked rotor conditions to achieve maximum stall and 
starting torque. As is further explained herein the power factor 
controller of this invention is relatively very simple, requires 
substantially less power itself than prior art power factor controllers 
and is capable of providing substantially the same high power savings when 
used with lightly loaded induction motors and provides essentially full 
power to a stalled motor as will the prior art controllers. Thus, the 
present invention leads to a power factor controller of substantially 
lower cost and size.

DESCRIPTION OF THE PREFERRED EMBODIMENTS 
The power factor controller 10 of FIG. 1 includes a bidirectional triode 
thyristor 12. Controller 10 is capable of controlling the voltage applied 
to a motor 14 such that the lag angle between motor current and applied 
voltage decreases as the mechanical load (not shown) increases. 
A plug 16 is designed to be plugged into a standard 110 volt outlet. Thus 
with respect to the reference buss 18, the 110 volts (r.m.s.) line voltage 
appears at line 20 as the sine wave 20w in FIG. 2a. In the following 
explanation of the operation of the controller 10, this waveform 2a will 
be used as the time reference wherein zero-crossings of line voltage 20w 
occur at times 0, .pi., 2.pi., 3.pi., etc., radians or more generally at 
n.pi.where n is an integer. 
A DC voltage of +10 volts appears at Vcc buss 21 derived from the circuit 
made up of dropping resistor 22, diode rectifier 24, smoothing capacitors 
26 and 27, series resistor 28 and zener diode 30. Four C-MOS exclusive OR 
gates 32, 34, 36 and 38 are powered from this Vcc buss 21, whereas the 
operational amplifier 40 is supplied about +25 DC volts from line 42. 
The bidirectional thyristor 12 is turned on each time, T.sub.f, that 
thyristor 44 is triggered on by a positive voltage that appears on line 
46. How this trigger voltage 46w of FIG. 2b is generated is explained as 
follows. 
When thyristor 12 turns on at a time T.sub.f, in a positive interval, e.g. 
between 0 and .pi., the voltage 47w, FIG. 2c, across thyristor 12 becomes 
about +0.6 volts. The motor current having a waveform 48w as shown in FIG. 
2d, flows in the positive direction (in interval prior to the .pi. zero 
crossing), as indicated by the arrow 48 of FIG. 1. The motor 14 is 
inductive and causes current 48w to continue the flow after each zero 
voltage crossing, i.e. at a time corresponding to n.pi. radians. 
Similarly, when thyristor 12 turns on at a time T.sub.f in a negative 
voltage interval, e.g. between .pi. and 2.pi., the voltage 47w across 
thyristor 12 becomes about -0.6 volts and is so maintained beyond the 
subsequent zero-crossing, e.g. 2.pi.. In each half cycle, the thyristor 
voltage 47w changes polarity abruptly at the instant (T.sub.0) that 
thyristor 12 ceases to conduct. The times T.sub.0 of this abrupt change 
therefore corresponds to those at which the motor current 48w falls to 
zero. The transistor switch 50 has connected at the base a bias network 
made up of resistors 52 and 53. This network provides a positive bias 
voltage of about 1.0 volts added to the thyristor voltage 47w. This sum 
voltage appears at the base of transistor 50. Thus transistor 50 is on 
when a positive motor current 48w flows, abruptly turning off at time 
T.sub.0 when the current 48w terminates. But transistor 50 is off when a 
negative motor current 48w flows, abruptly turning on at the time T.sub.0. 
Thus at the collector 54 of transistor 50 a voltage waveform 54w appears 
as in FIG. 2e. This voltage changes state at each instant T.sub.0, and 
only then. 
The transistor switch 56 is turned on during positive half cycles and 
turned off during negative half cycles of the power line (waveform 20w). 
The bias network made up of resistors 58 and 59 provide a positive bias 
voltage of about one volt at the base of transistor 56 to compensate for 
the V.sub.BE threshold thereof and to cause more perfect synchronism 
between the turning on and off of transistor 56 with the times (n.pi.) of 
zero crossing of the power line voltage 20w. The square wave voltage 60w 
(FIG. 2f) at the collector 60 of transistor 56 is thus .pi. radians out of 
phase with the power line voltage. This voltage 60w (FIG. 2f) is applied 
to one of the inputs of the inverting gate 32 which presents a square wave 
to an input of the exclusive OR gate 36 that is in phase with the power 
line voltage. 
The other signal input to gate 36 is the voltage 54w (FIG. 2e) from the 
collector 54 of transistor 50. Thus at the output 62 of exclusive OR gate 
36 there is generated a positive voltage 62w (FIG. 2g) that at every zero 
crossing n.pi. of the power line voltage 20w goes to zero and remains zero 
until the next time T.sub.0 at which motor current 48w has dropped to 
zero. 
The circuit node 64 can be considered a summing point for the signals 
generated at the outputs of the C-MOS exclusive OR gates 36 and 38. The 
voltage waveform 64w is illustrated in FIG. 2h. The output voltage 62w of 
gate 36 is near zero from the time of each zero crossing n.pi. to the time 
T.sub.0 that the lagging motor current drops to zero. At T.sub.0, voltage 
62w jumps to the +10 volts of buss 21. Time delay capacitor 66 and series 
resistor 68 have a time constant of about 2 milliseconds while the much 
larger capacitor 70 charges through resistor 72 and resistor 68 at a 
characteristic time constant of about 100 milliseconds, corresponding to 
12 half cycles of the power line voltage. Resistor 72 has a relatively 
small value. Capacitor 66 is not essential but adds stability of the 
circuit when transients occur. The magnitude of the voltage 64w is thus an 
inverse function of the lag time T.sub.0 and in this embodiment, an 
integrating means is comprised of the components 50, 52, 53, 54, 55, 56, 
58, 59, 61, 32, 36, 68, 66, 72 and 70. 
Exclusive OR gate 34 has at one input the voltage 60w (FIG. 2f) that is 
2.pi. radius out of phase with the power line voltage 20w. The other input 
74 to gate 34 is a waveform 74w as in FIG. 2k that is in phase with the 
power line voltage but slightly (about 100 .mu.sec) delayed by the network 
made up of resistor 76 and capacitor 78. Thus the output waveform 80w, 
FIG. 2m at the output 80 of gate 34 is at +10 volts except for 100 .mu.sec 
at each zero crossing (n.pi.) when it is at zero volts. During that short 
time, positive charge accumulated on timing capacitor 82 is drained off 
through diode 84. 
Thereafter, capacitor 82 charges, as in the ramp voltage waveform 82w of 
FIG. 2n toward the voltage appearing at node 64 through resistors 86 and 
88. These components may be designated ramping capacitor 82 and resistors 
86 and 88, respectively. When, at Tw, the voltage 90w, FIG. 2p at the 
input 90 of switching amplifier 40 exceeds that of the threshold voltage 
at its input 91, amplifier 40 turns on, producing a positive step voltage 
at line 46 to turn on the bidirectional triode thyristor 44 and in turn 
thyristor 12. The threshold bias voltage at input 91 is about 6 volts 
established by resistors 92 and 94, capacitor 96 and diode 98. In this way 
the delayed thyristor 12 firing time, T.sub.f, taken with reference to the 
corresponding (preceding) zero crossing n.pi. of the power line voltage, 
is inversely related to the voltage at node 64. 
Furthermore, the ramp slope of voltage waveform 82w becomes greater when 
the voltage at node 64 becomes greater, which occurs when over many cycles 
the lag time T.sub.0 of the motor current 48w becomes smaller. A steeper 
ramp voltage 82w effects the turning on of thyristor 12 at an even earlier 
time (T.sub.f), which may be more generally viewed as a means for 
increasing the "gain" of firing time T.sub.f to lag time T.sub.0 and 
enabling considerable simplification of the circuit. 
A positive feedback circuit is employed in the controller of this 
invention, namely the CMOS gate 38 and resistor 39 that are connected 
around amplifier 40. The output from gate 38 is a step of current through 
resistor 39 to node 64 (e.g. 20 v/180 K.OMEGA..perspectiveto.0.10 ma.) 
that occurs from Tw to the next n.pi.. This positive feedback embues the 
threshold detecting amplifier 40 with a hysteresis characteristic whereby 
the step of current charges capacitor 70 through resistor 72 to produce a 
step of voltage thereacross and to lock amplifier 40 in the "on" state. 
The presence or absence of positive feedback components 38 and 39 will not 
change the T.sub.f /T.sub.0 gain or the initial threshold of the amplifier 
40. It only changes the width of the input hysteresis characteristic of 
the detector portion of the circuit (including components 92, 94, 40, 38 
and 39). The greater the positive feedback, e.g. achieved by reducing the 
value of resistor 39, the greater will be the width of the hysteresis, 
i.e., the lag time T.sub.0 at which the detector will turn off is 
increased. 
An optimum adjustment of the amount of positive feedback can be made for a 
particular motor, specifically that adjustment for which full voltage is 
reliably (without oscillation or hunting) applied to the motor in the 
locked rotor condition. This adjustment should be made after an optimum 
adjustment of detector threshold is made, e.g. by varying the value of 
resistor 92 and thus the DC voltage at input 91 so that the detector 
threshold is set at a value just above that for which hunting of the 
system tends to occur for a medium to heavy range of mechanical loads on 
the motor. The tendency for hunting in the system will be much less for 
some induction motors than in others for which the characteristic power 
factor versus mechanical load for such motors peaks as load increases and 
drops severely when stalling (locked rotor condition) is approached. A 
fixed conserative setting may be made of threshold and feedback in the 
power factor controller so that a wide variety of motors can be powered 
from it without risk of instability. However, such a fixed setting will 
always provide less power savings at such a conservative setting. 
The very simple circuit of the present invention is capable of being 
produced with such a conserative fixed setting providing about as great a 
power savings with any particular motor as will a similarly adjusted but 
much more complex power factor controller of the prior art. This is in 
part due to the novel variable slope ramp feature of this invention that 
for light to medium heavy loads produces an almost constant current lag 
angle. 
A power factor controller circuit was assembled that including Triacs, 
Triac heat sinks, and all components of FIG. 1, that measures 
1.5.times.2.times.10.75 inches. This controller was connected in turn to 
six fractional horsepower motors. Optimum adjustments having been made for 
connections to the motor most prone to hunt, a Dayton 5K280A rated at 1/4 
horsepower, the power delivered from the AC line was measured at light to 
heavy loads with and without the controller and the results are briefly 
shown in Table II. 
TABLE II 
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INPUT WATTS 
with without 
P.F.C. 
P.F.C. 
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65 165 
500 600 
50 990 
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The power consumption of this experimental controller is 0.85 watts 
excluding the Triacs while that of the prior art controllers is 
approximately five times as much. The low power consumption and simple 
circuitry of the controller of this invention make realization in silicon 
integrated circuit form practical and costs will be a small fraction of 
those known heretofore. 
The component values are given in Table I 
TABLE I 
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Resistors 
Value Rating Value Rating 
No. (K ohms) (watts) No. (.mu.fd) 
(volts) 
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Capacitors 
22 6.8 1 26 47 25 
28 10 1/4 27 0.1 50 
37 470 1/4 66 0.1 50 
39 130 1/4 70 4.7 16 
41 2.7 1/4 
42 100 1/4 
45 1 1/4 78 0.001 50 
52 100 1/4 82 0.1 50 
53 1000 1/4 96 4.7 16 
55 100 1/4 
58 220 1/4 Transistors 
59 2200 1/4 50 2N3904 
61 100 1/4 56 2N3904 
68 22 1/4 
72 3.9 1/4 Diodes 
76 100 1/4 24 1N4004 
86 22 1.4 84 1N914 
88 100 1/4 98 1N914 
92 100 1/4 
94 68 1/4 Zener 
30 1N5856B 
I.C.s 
32 CD4070 
34 1/4 CD4070 
36 1/4 CD4070 
38 1/4 CD4070 
40 .mu.A741 
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