Dual-band monolithic microwave IC (MMIC) power amplifier

A dual-band MMIC power amplifier and method of operation to amplify frequencies in different RF bands while only requiring input drive signals at frequencies f1 and f2 in a narrow RF input band. This allows for the use of a conventional narrowband RF IC to drive the MMIC and does not require additional circuitry (e.g., a LO) on the MMIC power amplifier. The matching network of the last amplification stage is modified to pass f1 (or a harmonic thereof), reflect f2, pass a Pth harmonic of f2 where P is 2 or 3 and to reflect any unused 1st, 2nd or 3rd order harmonics of f1 or f2 back into the MMIC. In response to an input signal at f1, the MMIC power amplifier amplifies and outputs a signal at f1 (or a harmonic thereof). In response to an input signal at f2 at sufficient RF power, the last amplification stage operates in compression such that the MMIC power amplifier generates the harmonics, selects the Pth harmonic and outputs an amplified RF signal at P*f2.

BACKGROUND OF THE INVENTION

Field of the Invention

This invention relates to Monolithic Microwave Integrated Circuit (MMIC) Power Amplifiers, and more particularly to dual-band MMIC Power Amplifiers.

Description of the Related Art

A typical RF transmitter may include an RF IC, a MMIC power amplifier and an antenna. The RF IC modulates a data signal onto an RF carrier signal. The data signal is typically at a relatively low rate e.g. ˜1 GHz or ˜10% of the center frequency of the RF carrier signal. Typically the RF IC can vary the RF carrier signal over a narrow frequency band e.g., 12-15 GHz or 42-45 GHz. The MMIC amplifies the modulated RF carrier signal to drive the antenna, which transmits the RF signal over the air.

A MMIC power amplifier is a distributed amplifier that is fabricated on a single chip using, for example, GaN, GaAs or SiGe (bipolar transistors or RF CMOS). The distributed amplifier includes a plurality of amplification stages connected in a chain. Each amplification stage includes a transistor biased to provide gain and a matching network (e.g. a lumped element inductor (L)/capacitor (C) or distributed circuit) to allow the amplified RF signal to flow from one amplification stage to the next. The peak-to-peak voltage (a proxy for RF power assuming current is constant) of the signal driven into each amplification stage determines whether that stage operates in its linear or compressed regions. Linear operation provides less distortion of the input signal but less amplified power.

In certain applications, it may be required that the RF transmitter have the capability to selectively transmit in multiple, typically dual, RF bands e.g. 16-18 GHz (Kuband) and 33-50 GHz (Q-band). As shown inFIG.1, a network of satellites10communicate with each other in, for example, a 45 GHz band12and with ground data links 14 in, for example, a 12 GHz band16. The lower frequency Ku-band being used to communicate through the atmosphere. The higher frequency Q-band providing higher bandwidth utilized for inter-satellite communication. In such an application the satellites require an RF transmitter that can amplify and transmit RF signals in both the Ku and Q bands.

One approach to realizing a dual band MMIC power amplifier is to have two channels, one MMIC channel that operates in the lower RF band and another MMIC channel that operate in the upper RF band to drive a dual-band antenna. Another approach is to design extremely wideband MMIC power amplifier that spans both the lower and upper bands.

Another approach is to have a MMIC that operates at the lower band and integrate a diode ring and a broadband local oscillator (LO) on the MMIC power amplifier. With this approach the data signal (with a waveform frequency denoted F1) is mixed with the LO (with a waveform frequency denoted F2) generating harmonics using a process called heterodyning. The mixing process generates high and low frequencies F1−F2and F1+F2(major lowest order mixing components). In most RF devices the low frequency mixed signal F1−F2is filtered. The high frequency mixed signal F1+F2is then passed to the amplifier and is eventually routed to the antenna where the waveform is radiated into space as the transmission frequency. To vary the transmission frequency F1+F2the LO frequency F2is altered. The frequency of the local oscillator is typically varied by tuning a voltage in the local oscillator denoted VLO. Consequently, the output frequency is a function of a voltage in the local oscillator denoted F2(VLO). Since it is possible to vary the input voltage into the local oscillator VLOthe output MMIC frequency can be varied thereby effectively creating a dual (or multi) band power amplifier MMIC by the following formula F1+F2(VLO).

SUMMARY OF THE INVENTION

The present invention provides a dual-band MMIC power amplifier configured to receive RF input signals within a narrow input frequency band, amplify and upconvert at least one of the frequencies via a compressed non-linear response of the last amplification stage to output amplified signals in two different frequency bands. This allows for the use of a conventional narrowband RF IC to drive the MMIC and does not require additional circuitry (e.g., a local oscillator) on the MMIC power amplifier.

This is accomplished by modifying the matching network of the last amplification stage and controlling the RF frequencies f1and f2and amplitude of the input signals that are driven into the MMIC power amplifier. The matching networks of the first L amplification stages are configured to pass f1and f2and to reflect at least the 2ndand 3rdharmonics thereof. The matching network of the last (Mth) amplification stage is configured to pass f1(or a harmonic thereof, namely N*f1where N=1, 2 or 3) and block f2, to pass a Pthharmonic of f2where P is typically 2 or 3 and to reflect any unused 1st, 2ndor 3rdorder harmonics of f1or f2. When the RF IC generates an input signal at f1, the MMIC power amplifier amplifies and outputs a signal at f1(or a harmonic thereof). When the RF IC generates an input signal at f2at sufficient RF power, the last amplification stage operates in compression. When a MMIC power amplifier operates in compression it has a non-linear transfer function. This results in the generation of higher-order harmonics of the input signal (f2+2f2+3f2+ . . . ). The amplifier's matching network is designed such that only one harmonic, denoted the Pthharmonic with a frequency Pf2is amplified and is the signal output from the MMIC. To summarize, in the example above when the MMIC is driven at a frequency f1, an amplified f1signal is produced by the MMIC. When the MMIC is driven at a frequency f2, an amplified Pf2is produced by the MMIC.

The RF IC is configured to generate the RF input signal at frequencies f1and f2that span a narrow input frequency band. The bandwidth choice is dependent upon the value of the frequencies chosen for amplification and the design of the matching network. At a minimum however, the bandwidth for f1denoted 2Δf1(see: f1−Δf1≤f1≤f1+Δf1) and bandwidth of f2denoted 2Δf2(see: f2−Δf2≤f2≤f2+Δf2) needs to be such that f1+Δf1<f2−Δf2. If the previous inequality is not satisfied, then signal leakage will occur across the bands and the matching networks in the device will attenuate or multiply undesired frequencies resulting in signal distortion. Moreover, the choice of frequencies f1and f2and associated bandwidths 2Δf1and 2Δf2need to be selected so the harmonics correctly pass through the matching networks.

The RF IC may generate the RF input signal at frequency f1either at low RF power such that the last amplification stage (and all preceding stages) operate in the linear region or at high RF power such that the last amplification stage (and possibly some or all preceding stages) operate in the compressed region. If operating in the linear region, the MMIC power amplifier will output the N=1 harmonic of f1(i.e., the fundamental frequency). If operating in the compressed region, the MMIC power amplifier may be configured to output N*f1where N=1, 2 or 3. In this later case, the unused harmonics are reflected back into the transistor to improve power added efficiency.

The matching network of the final amplification stage may be further configured to pass a Qthharmonic of a third input frequency f3where Q is 2 or 3. The frequencies f1, f2and f3are selected and the passbands of the matching network configured such that any unused 1st, 2ndor 3rdorder harmonics of f1, f2, or f3are reflected back into the amplification stage. 4thor higher order harmonics are naturally attenuated to a point that they do not have to be reflected and add little to the recycled power.

The matching network of the next to last (M−1th) amplification stage may be configured to pass f1(or a harmonic thereof), reflect f2and pass an Rthharmonic of f2where R is 2 or 3 and to reflect any unused 1st, 2ndor 3rdorder harmonics of f1or f2in order to cascade the higher order harmonics of the M−1hand Mthamplification stages to output an amplified RF signal at a frequency of R*P*f2. Cascading of two or more stages allows the MIMIC power amplifier to reach much higher frequencies.

DETAILED DESCRIPTION OF THE INVENTION

The present invention provides a dual-band MMIC power amplifier configured to receive input signals within a narrow input frequency band, amplify and upconvert at least one of the frequencies via a compressed non-linear response of the last amplification stage to output amplified signals in two different frequency bands. This allows for the use of a conventional narrowband RF IC to drive the MMIC and does not require additional circuitry (e.g., a LO) on the MMIC power amplifier. For example, an RF IC that can generate signals in the Kuband can be configured to drive the dual-band MMIC power amplifier to generate amplified signals in both the Kuand Q bands without modifying the RF IC or adding circuitry to the MMIC.

The difference between the linear and the compressed region depends on the relationship between the input and output voltage from the transistor amplifier. If the relationship between input and output voltage is linear, namely Vout=GVinwhere G is the amplifier gain, then the frequency input is the same as the frequency output if Vin=V0sin(2πfint). Now if the relationship between the input and output voltage is nonlinear, namely expressed as a finite or infinite polynomial series Vout=G1Vin+G2Vin2+G3Vin3+ . . . then passing in a sinusoidal signal through the device as Vin=V0sin(2πfint) and using trigonometric product identities (ex: sin2(α)=−(½)cos(2α)) results in Voutcomposed of a sum of sinusoidal function with frequencies that are integer multiples of the input frequency fin, 2fin, 3fin, . . . . The power in each of the tones (fin, 2fin, 3fin, . . . ) is dependent on the values of the kthpower coefficients Gk. For most real transistors the Gkcoefficients decay in magnitude for higher order tones. Typically 4finand higher order tones are not significant contributors to the total output waveform power. Therefore, driving a MMIC with a single tone waveform at frequency finin the non-linear region generates an output signal that is composed of a sum of tones that are integer multiples of the input drive frequency fin.

For most transistors driven at small RF input powers the voltage input to output response is linear. As the input RF power becomes progressively larger the voltage input to output response curves transition from a linear to a non-linear response (creating what is known as harmonic distortion). This process continues until in the transistor the compression is so extreme that no more gain is possible and the transistor cannot output any more power. Power amplifiers operate at maximum power added efficiency in compression, where some harmonic distortion is generated.

As used herein, when a transistor is described as being driven in the “linear region” it implies that the voltage input to output relationship is approximately linear as the transistor is driven at low power and no spurious tones are generated. When a transistor is described as being driven in “compression” it means the transistor is being driven at high input RF power and the resulting voltage input to output relationship is non-linear. This non-linear input to output voltage relationship generates multiple integer multiples of the input signal frequency in the output signal.

Referring now toFIGS.2and3, in an embodiment an RF transmitter20includes an RF IC22, a dual-band MMIC power amplifier24and a dual-band antenna26.

A conventional off-the-shelf RF IC22(e.g, SiGe or SiGe CMOS) includes a frequency mixer28that modulates a carrier signal Vcs(fcs)30with a data signal Vdata(fd)32to produce a modulated carrier signal Vs(fcs)34referred to as the RF input signal. A filter35removes unwanted frequency products generated by mixer28. The frequency fdof the data signal is much lower than the frequency of the carrier signal fcs. A control voltage Vcis applied to a voltage controlled oscillator (VCO)36to generate carrier signal Vcs(fcs)30. By varying control voltage Vc, the carrier frequency fcscan be varied over an input frequency band to produce different but closely spaced frequencies f1and f2.

The RF IC is configured to generate the RF input signal at frequencies f1and f2that span a narrow input frequency band. The bandwidth choice is dependent upon the value of the frequencies chosen for amplification and the design of the matching network. At a minimum however, the bandwidth for f1denoted 2Δf1(see: f1−Δf1≤f1≤f1+Δf1) and bandwidth of f2denoted 2Δf2(see: f2−Δf2≤f2≤f2+Δf2) needs to be such that f1+Δf1≤f2−Δf2. If the previous inequality is not satisfied, then signal leakage will occur across the bands and the matching networks in the device will attenuate or multiply undesired frequencies resulting in signal distortion. Moreover, the choice of frequencies f1and f2and associated bandwidths 2Δf1and 2Δf2need to be selected so the harmonics correctly pass through the matching networks.

Dual-band MMIC power amplifier24(e.g. GaN, GaAs or SiGe (bipolar junction or RF CMOS devices)) is a distributed amplifier that includes a plurality of M amplification stages40operatively coupled in a chain to amplify the RF input signal34. Each amplification stage40includes a transistor42biased to provide gain and a matching network44to allow the amplified RF input signal to flow from one amplification stage to the next. The matching network is shown as a simple LC circuit but is actually a plurality of lumped parallel-connected LC circuits or distributed matching networks designed to provide certain passband and rejection band characteristics.

The matching networks44of the first L<M amplification stages are configured with a passband46at the RF input frequency band and a rejection band48to reflect at least 2ndand 3rdorder harmonics of the RF input signal. Accordingly, the RF input signal at either f1or f2is amplified and flows from one amplification stage to the next. If these intermediate stages are operated in compression mode, the higher order harmonics are reflected back into each stage and the power recycled to increase the power at the fundamental frequency f1or f2thereby boosting MMIC power added efficiency.

In accordance with the present invention, the matching network of the Mth(last) amplification stage is configured with a first passband50at an Nthharmonic of f1where N is 1, 2 or 3 that rejects f2, a second passband52at a Pthharmonic of f2where P is 2 or 3, and a rejection band54to reflect any unused 1st, 2ndor 3rdorder harmonics of f1or f2. It is critical that frequencies f1and f2are different frequencies with a certain minimum spacing to avoid overlap of the filter/matching network passbands. f1and f2are suitably closely spaced so that they can be generated by an off-the-shelf (OTS) RF IC.

Dual-band antenna26suitably includes first and second antenna elements configure to transmit RF signals56at N*f1and P*f2. For example, the antenna may include a pair of patch antennas configured to resonate at N*f1and P*f2.

For dual-band operation of the RF transmitter, the RF IC22selectively generates the RF input signal34at frequencies f1and f2within the input frequency band.

At frequency f1, the RF input signal34is amplified by the transistor42at each amplification stage and flows from one stage to the next via matched network44through passband46. At the last or Mthamplification stage, the RF input signal is amplified, passes through passband50and is output at the Nthharmonic N*f1. If the peak-to-peak voltage of the RF input signal applied to the last stage is such that the transistor operates in the linear region than N=1. If the peak-to-peak voltage of the RF input signal (e.g., the RF power) applied to the last stage is such that the transistor operates in the compressed region, the transistor will generate multiple higher order harmonics60of the input signal such that N may equal 1, 2 or 3. The MMIC may be configured and driven to operate in compression to maximize the RF transmit power at f1(i.e. operate at the point of maximum power added efficiency). Alternately, the MMIC may be configured and driven to operate in the linear region to minimize signal distortion. Either is possible at drive frequency f1.

At frequency f2, the RF input signal34is amplified by the transistor42at each amplification stage and flows from one stage to the next via matched network44through passband46. At the last or Mthamplification stage, the RF input signal must have sufficient peak-to-peak voltage (RF power) to compress the transistor and generate the higher order harmonics60. One of the higher order harmonics (P=2 or 3) passes through passband52and is output at the Pthharmonic P*f2. The fundamental frequency f2and the other unused harmonic are reflected by rejection band54. For the case of N=1, passband50is narrower than passband46in the preceding stages such that f1is passed but f2is rejected.

When driven into compression, the transistor generates the harmonics60of the RF input signal. For most transistors used in power amplification, the power in each successive harmonic exhibits a natural decay that is dependent on the transfer function of the specific transistor. To optimize RF output power, it is important that the matching networks are designed to reflect the power of any of the unused harmonics (including the fundamental) back into the transistor. Some of that power will recombine and increase the overall power added efficiency of the MMIC thereby increasing the overall output power of the selected fundamental or harmonic frequency. Any remaining power will be waste heat. However, the natural decay of the power spectrum means reflecting the 4thharmonic and higher order tones yields a minimal improvement in device power added efficiency (as there simply isn't that much power in these tones). Therefore, the MMIC power amplifier is limited to generating the fundamental and 2ndor 3rdharmonics. Simply put, 4thand higher harmonics have so little power that engineering the matching network to reflect these harmonics is not necessary.

The dual-band MMIC power amplifier as described is a fixed or ‘dumb’ device, it merely acts on the RF input signals according to their frequency and power. The design of the dual-band MMIC power amplifier, and particularly the matching network for the last stage, is intimately tied to the selection of the specific input frequencies f1and f2. Given the complex matching network requirements of the dual-band MMIC, compared to single-band power amplifier MMIC, a narrower input frequency bandwidth around f1and f2will be a design compromise. To illustrate how this dual band MMIC architecture will result in reduced bandwidth around f1and f2consider what happens as f1and f2are varied slightly about their respective center frequencies. Varying f1may cause the fundamental to be rejected by the last stage instead of amplified and output or may place an unwanted harmonic of f1in the second passband resulting in distortion of the output signal. Varying f2may cause the desired Pthharmonic to lie outside the second passband resulting in a rejection of the fundamental and harmonics and no output signal. Depending on the tightness of specifications for the passbands and rejection band some small variation in f1and f2may be tolerated. Therefore, the design of the matching/filter networks will place constraints on the allowable bandwidth around f1and f2(in addition to the natural constraints associated with the performance of the transistors themselves).

Referring now toFIGS.4A-4C and5A-B, a dual-band MMIC power amplifier100is configured to amplify an RF input signal at frequency f1and to amplify and upconvert an RF input signal at frequency f2to 3*f2using the non-linear properties of the compressed final amplification stage. The matching networks for the first M−1 amplification stages are designed to exhibit a passband102that passes f1and f2within the input frequency band. The matching network for the last or Mthamplification stage is designed to exhibit a passband104that passes f1while rejecting f2and a passband106that passes 3*f2.

As shown inFIG.4B, for frequency f1, if the RF input power is sufficient to compress the final stage, harmonics at f1, 2f1and 3f1are generated. f1is passed through passband104while 2f1and 3fAare rejected and recycled. It is critical that passband106does not pass either of these unwanted harmonics. This is why f2must be different than f1and the frequencies carefully selected to avoid overlap and unwanted filtering and reflection of various frequencies.

As shown inFIG.4C, for frequency f1, if the RF input power operates the final stage in its linear region than only the fundamental f1is generated and passed through passband104.

As shown inFIGS.5A-5B, for frequency f2, the RF input power must be sufficient to compress the final stage and generate harmonics at f2, 2f2, 3f2. . . . Although f2passed through the passband102in the preceding M−1 amplification stages it lies outside passband104in the final stage and is recycled along with the 2ndharmonic 2*f2. The 3rdharmonic 3*f3passes through passband106and is output as the amplified RF signal. Although the 3rdharmonic is naturally attenuated, the transitor gain is increased due to the recycled power from the fundamental and second harmonics.

Consider an example using real frequencies: if the input frequency band is the Kuband from 12-18 GHz the RF IC may be configured to generate f1=12 GHz and f2=15 GHz. Passband102in the first M−1 stages is configured to pass frequencies between 11 Ghz to 16 GHz. In the final stage passband104is configured to pass frequencies from 11 GHz to 13 GHz and passband106is configured to pass frequencies from 44 GHz to 46 GHz. Assuming the final stage is compressed for both f1and f2, f1will generate harmonics at 12, 24, 36 and 48 GHz and f2will generate harmonics at 15, 30, 45 and 60 GHz. At f1, the RF input signal is amplified through all M stages producing an amplified RF signal at 12 GHz. At f2, the RF input signal is amplified at 12 GHz through the first M−1 stages. At the final stage, f2is rejected and recycled into the final stage while the 3rdharmonic is amplified and produces an amplified RF signal at 45 GHz.

With only a modification to the matching network of the final amplification stage and careful selection of f1and f2in the Kuband, the dual-band MMIC power amplifier can generate an amplified signal in the Kuband and the V band. This is achieved without requiring duplicative channels at the Kuand V band, without requiring an RF IC that supports frequencies across both bands and without additional circuitry on the MMIC.

Referring now toFIG.6, the RF IC and dual-band MMIC power amplifier are configured to amplify and output RF signals at 2*f1and 3*f2. As shown, both f1and f2pass through a passband200in the first M−1 amplification stages. The matching network for the final amplification stage includes a passband202at the 2ndharmonic of f1(2*f1) and a passband204at the 3rdharmonic of f2(3*f2). In both cases, the RF power of the input signal must be sufficient to compress the transistor in the final amplification stage to generate the higher order harmonics. Furthermore frequencies f1and f2must be selected such that the fundamental frequencies and, more particularly, the unused harmonics do not overlap and conflict with passband204. The unused fundamental and 2ndor 3rdorder harmonics are reflected and recycled to increase the output power of the transistor in the final stage at the desired harmonic of either f1or f2.

For example, if the input frequency band is the Kuband from 12-18 GHz the RF IC may be configured to generate f1=12 GHz and f2=15 GHz. Passband200in the first M−1 stages is configured to pass frequencies between 11 Ghz to 16 GHz. In the final stage passband202is configured to pass frequencies from 23 GHz to 25 GHz and passband204is configured to pass frequencies from 44 GHz to 46 GHz. The final stage is compressed for both f1and f2. f1will generate harmonics at 12, 24, 36 and 48 GHz and f2will generate harmonics at 15, 30, 45 and 60 GHz. At f1, the RF input signal at 12 GHz is amplified through the first M−1 stages. At the final stage, the fundamental at 12 GHz is reflected (and thus recycled) while the 2ndharmonic is amplified and pass through passband202producing an amplified RF signal at 24 GHz. At f2, the RF input signal is amplified at 12 GHz through the first M−1 stages. At the final stage, f2is reflected and recycled into the final stage while the 3rdharmonic is amplified and produces an amplified RF signal at 45 GHz. In this configurations, the fundamental frequencies f1, f2, the 3rdharmonic of f1and the 2ndharmonic of f2are reflected and recycled.

Referring now toFIG.7, the RF IC and dual-band MMIC power amplifier are configured to amplify and output RF signals at f1, 2*f2and 3*f3. As shown, each of f1, f2and f3pass through a passband300in the preceding M−1 amplification stages. The matching network for the final amplification stage includes a passband302at f1that rejects f2and f3, a passband304at the 2ndharmonic of f2(2*f2) and a passband306at the 3rdharmonic of f3(3*f3). The last amplification stage must operate in compression for f2and f3. Furthermore, frequencies f1, f2and f3must be selected such that the fundamental frequencies and, more particularly, the unused harmonics do not overlap and conflict with passband304and306. The unused fundamental frequencies and 2ndor 3rdorder harmonics are reflected and recycled to increase the output power of the transistor in the final stage at the desired fundamental or harmonic of either f1, f2or f3. The addition of a 3rdoutput frequency makes selection of the input frequencies f1, f2and f3and the placement/characteristics of the passbands and bandwidth available for the 3 drive tones more demanding than the dual band case.

To illustrate a 3 band MMIC consider it being driven by the RF IC at the following frequencies in Kuband: f1=13 GHz, f2=15 GHz and f3=17 GHz. Passband300in the first M−1 stages is configured to pass frequencies between 12 Ghz to 18 GHz.

In the final stage passband302is configured to pass frequencies from 12 GHz to 14 GHz, passband304is configured to pass frequencies from 29 GHz to 32 GHz and passband306is configured to pass frequencies from 50 GHz to 52 GHz. The final stage is compressed for at least f2and f3. If the MMIC is in compression, f1will generate harmonics at 13, 26, 39 and 52 GHz, f2will generate harmonics at 15, 30, 45 and 60 GHz and f3will generate harmonics at 17, 34, 51 and 68 GHz. At f1, the RF input signal at 13 GHz is amplified through all M stages to output an amplified RF signal at 13 GHz. At f2, the RF input signal is amplified at 15 GHz through the first M−1 stages. At the final stage, f2is reflected and recycled into the final stage while the 2nd harmonic is amplified and produces an amplified RF signal at 30 GHz. At f3, the RF input signal is amplified at 17 GHz through the first M−1 stages. At the final stage, f3is reflected and recycled into the final stage, while the 3rdharmonic is amplified producing 51 GHz RF signal. Note, the 4thharmonic of f1is at 52 GHz, which may partial pass through passband306. However, the 4thharmonic is naturally attenuated to such an extent that this is unlikely to be a problem. If it is a problem, f1could be operated in the linear region to avoid the generation of the overlapping 4thharmonic.

Referring now toFIG.8, the RF IC and dual-band MMIC power amplifier400are configured to amplify and output RF signals at f1and R*P*f2where R=2 and P=3. The matching network402in the first M−2 stages is configured to exhibit a passband404that spans both input frequencies f1and f2. The matching network402for the M−1 stage is configured to exhibit a passband406that passes f1and reflects f2and a passband408that pass R*f2where R=2. The matching network402for the final Mthstage is configured to exhibit a passband410that passes f1and reflects f2and a passband412that passes R*P*f2where P=3. At f1, the RF input signal is amplified at each stage flowing from one stage to the next through passbands404,406and410until being output as the amplified RF signal at f1. At f2, the RF input signal is amplified at f2by the first M−2 stages. The compressed M−1 stage generates harmonics414, of which the 2ndharmonic is passed through passband408and input to the final stage. The compressed final Mthstage generates harmonics416, of which the 3rdharmonic at 6*f2is passed through passband410and output as the amplified RF signal at 6*f2. Cascading the non-linear effects of the last two stages greatly extends the reach of the second band of the dual-band MMIC power amplifier. The natural attenuation of cascading two harmonics will reduce the available output power, a portion of which can be recaptured by recycling the unused harmonics.

For example, if the input frequency band is the C and X band from 6-10 GHz the RF IC may be configured to generate f1=6 GHz and f2=10 GHz. Passband400in the first M−2 stages is configured to pass frequencies between 6 Ghz to 10 GHz. In the next to last stage, passband406is configured to pass frequencies between 6 GHz and 9 GHz and passband408is configured to pass frequencies from 19-21 GHz. In the final stage, passband410is configured to pass frequencies between 6-9 GHz and passband412is configured to pass frequencies between 59-61 GHz. The final two stages are both compressed for f2. f2will generate harmonics at 10, 20, 30 and 40 GHz at the M−1 stage and 20, 40, 60 and 80 GHz at the final Mthstage. At the M−1 stage, the 2ndharmonic at 20 GHz passes through passband408to drive the final stage. In turn, the 3rdharmonic at 60 GHz passes through passband412and is output as the amplified RF signal at 60 GHz.