Tracing electrical conductors by high-frequency loading and improved signal detection

Tracing and identifying electrical conductors in a power distribution network is achieved by use of a transmitter operating on a duty cycle of delivering or absorbing current pulses at a predetermined frequency from the power distribution network, and by use of a remotely located receiver which detects the electromagnetic field signals resulting from the current pulses in a predetermined cyclic manner of operation defined by a sample period and a reset period. During the sample period the receiver supplies an indication of the maximum received strength of the transmitter signal. The relationship of the duty cycle of the transmitter and the sampling and reset periods of the receiver reduce the potentially adverse influences of spurious signals on the receiver. The transmitter may include means for delivering constant energy content pulses to the power distribution network, thereby rendering the signals detected by the receiver insensitive to variations in voltage on the power distribution network. Improved signal filtering is achieved in the receiver by connecting a plurality of relatively low Q filters in series to reduce the effects of ringing and increase the damping factor, or by employing a digital capacitive filter switching technique for precise filtering by phase discrimination.

BACKGROUND 
This invention relates to new and improved apparatus and methods for 
tracing electrical conductors of both alternating and direct current 
electrical power, and more specifically it pertains to identifying circuit 
breakers, fuses, switches and other current conducting or handling devices 
connected to a source of electrical power. 
It is oftentimes necessary to trace and identify particular circuits and 
electrical devices in a power distribution network, such as circuit 
breakers or fuses. Identification has typically been accomplished by 
practicing one or two manual techniques. One technique is to selectively 
disrupt power by opening the circuit breakers one at a time. When power is 
no longer present at the circuit, electrical device or feeder conductor in 
question, the opened circuit breaker identifies the item in question. The 
disadvantage to this technique is that electrical power is temporarily 
disconnected from each of the circuits and branch conductors in the course 
of the search, and it may be critical to maintain power to some of these 
circuits and branch conductors. Critical circuits include those which 
supply power to hospital equipment, computers, and many other types of 
sensitive electronic equipment. Another disadvantage is that a 
considerable amount of time is consumed in selectively and individually 
opening each of the circuit breakers. The second manual technique of 
identifying a circuit breaker is to introduce a sufficiently high 
electrical current load on the particular branch conductor to trip the 
circuit breaker or open the fuse. This technique is typically achieved by 
introducing an intentional short circuit to the branch conductor. The 
disadvantage of this technique is that the power will then be totally 
disrupted, creating the detrimental consequences previously mentioned. The 
increased current drawn by the short circuit can create dangerous 
momentary overheating or fire conditions or can cause larger trunk or 
distribution breakers to trip open at the same time the branch circuit 
breaker is tripped open. Of course, once a distribution breaker trips 
open, a large number of branch and distribution conductors will be 
disconnected from the source of electrical power. 
A variety of test instruments are also available for testing and 
determining a variety of different electrical conditions including tracing 
and identifying feeder conductors, circuit breakers and other current 
conducting devices as well as tracing and identifying short circuited 
conductors. Certain of these prior art devices require interruption of 
power to the conductors in order to accomplish the tracing and 
identification. Other types of prior art devices employ means which 
cyclically create a current load on a particular conductor of sufficient 
magnitude to allow the increased current load, and hence the electrical 
device, to be identified with a conventional ammeter or impedence 
measuring device. Still other types of prior art devices introduce a 
relatively high-frequency signal on the conductor while conventional power 
is maintained and high-frequency signal is inductively detected. The 
high-frequency signal detection apparatus offers the best potential for 
reliable and simple circuit identification and detection, but such prior 
art devices are typically subject to adverse and detrimental influences, 
such as false signals resulting from spurious signals from transients and 
switching currents, reduced sensitivity for detecting and identifying the 
desired feeder conductors through panel enclosures and tubular conduits, 
and a somewhat limited specificity in isolating one particular conductor 
from a number of other conductors located in close proximity. Other 
disadvantages of such prior art systems and methods are known to those 
appreciating this particular field and its problems, and will be made more 
particularly apparent with comprehension of the desirable features of the 
present invention. 
SUMMARY 
It is the general objective of this invention to provide a new and improved 
apparatus and method for tracing and identifying electrical conductors 
which exhibits a relatively high immunity to adverse influences from 
spurious signals such as transients and switching currents, which does not 
require the interruption of applied power during the testing and tracing 
procedures, which exhibits a relatively high sensitivy and selectivity for 
more precisely identifying the particular electrical circuit conductor out 
of a closely assembled group without disassembly of various housings and 
enclosures containing the feeder conductor, and which is rendered 
substantially insensitive to different magnitudes of voltage applied over 
the power distribution network. 
In accordance with certain summary aspects, the present invention comprises 
a transmitter means and a receiver means. The transmitter means 
operatively conducts a predetermined waveform of electrical current 
through the conductors and electrical devices of the electrical power 
distribution network. The predetermined current waveform is defined by a 
duty cycle having a first predetermined on-time period during which a 
plurality of pulses of current are conducted at a predetermined frequency, 
and a second predetermined off-time period during which no current is 
conducted. The on-time period is considerably less in duration than the 
off-time period. The receiver means operatively detects the predetermined 
frequency characteristic of the electromagnetic field signal induced in 
the conductors of the power distribution network by the predetermined 
waveform from the transmitter means. The receiver means includes a 
frequency filter for passing the inductively received signals of 
frequencies related to or the same as the predetermined frequency of the 
transmitter pulses. The receiver also includes means for periodically 
determining the magnitude of the detected signals on a cyclic basis and 
for indicating the relative magnitude of the detected signals. Each cycle 
of operation of the receiver includes a predetermined sampling time period 
and a predetermined reset time period. Each period of the cyclic operation 
of the receiver is less than each period of the duty cycle of the 
transmitter. By delivering the pulses from the transmitter over a limited 
time period of the duty cycle, the circuit elements of the transmitter are 
operated over a limited time and do not experience adverse heating. As a 
result, a very strong signal of low time duration is available. By 
operating the receiver on a cyclic basis by first sampling and then 
resetting, the adverse influences from spurious signals are minimized 
because such signals would have no effect on the transmitter if they occur 
during the reset time period. The relationship of the on and off periods 
of the transmitter duty cycle and the sample and reset periods of the 
cyclic operation of the receiver is such that a transmitter signal will be 
reliably detected on a sufficiently consistent basis to provide a reliable 
indication to the user, while the effects of spurious signals and other 
signals causing inaccuracies in detection are minimized or reduced as a 
result of the lack of signals during the off time period of the 
transmitter and the reset time period of the receiver. 
In accordance with another significant summary aspect of the present 
invention, the transmitter includes means for delivering constant energy 
content pulses. When utilizing the transmitter and receiver in an 
alternating current power distribution network, it is appreciated that the 
voltage over the power network is constantly changing. Adjusting the time 
duration of the pulses in accordance with a hyperbolic relationship to the 
voltage magnitude obtains constant energy content pulses. A resonant 
circuit of a transducer of the receiver is influenced by the constant 
energy pulses to supply an approximately constant output signal. As a 
result, the instantaneous voltage magnitude over the power distribution 
network at the point in time when each transmitter pulse is delivered will 
have no effect on varying the relative indication signals obtained by the 
receiver, since the receiver will obtain a relatively constant output 
signal by detecting the constant energy content pulses. 
The invention itself is more precisely defined by the appended claims. The 
improvements and concepts from the present invention are more specifically 
described in the accompanying description of the preferred embodiments 
taken in conjunction with the drawings.

PREFERRED EMBODIMENTS 
The two basic components of the present invention are a transmitter 10 and 
a receiver 12 shown in FIG. 1. The transmitter 10 is electrically 
connected into a power distribution network through which electrical power 
is supplied, as by connecting a plug prong 14 of the transmitter 10 into a 
convenience outlet 16, for example. The convenience outlet 16 is 
electrically connected to a branch line or conductor 18 extending from a 
load center 20. Other branch lines, e.g. 22 and 24, also extend from the 
load center 20 and are connected with other elements such as additional 
convenience outlets 16a and 16b. Electrical power is supplied to the 
branch conductors 18, 22 and 24 through branch circuit breakers 26, 28 and 
30, respectively, which are located within the load center 20. Neutral 
conductors 32 return current flow from the branch conductors 18, 22 and 24 
to the load center at a neutral bus 34. Electrical power is supplied to 
the load center 20 from a secondary line transformer 36, a plurality of 
feeder conductors 38, a plurality of main distribution breakers 40 and a 
plurality of distribution conductors 42 of the power distribution network, 
as is typical. A ground conductor 44 also extends through the power 
distribution network between the secondary line transformer 36 and each 
neutral bus 34. The branch circuit breakers 26, 28 and 30 protect the 
branch conductors 18, 22 and 24, respectively, from current overload 
conditions by tripping open upon the occurrence of an increased current 
drawn through any one of the branch lines. Similarly, the main 
distribution breakers 40 protect the distribution conductors 42 from 
current overload conditions. 
The transmitter 10 receives electrical power from one of the branch 
conductors, e.g. 18, and draws or absorbs current from that particular 
branch line in a predetermined pattern or waveform. The predetermined 
pattern or current waveform drawn from the particular branch conductor to 
which the transmitter is connected induces a predetermined electromagnetic 
field directly related to the current waveform. The electromagnetic field 
extends along all of the branch conductors and each of the electrical 
devices operatively connected with these conductors. For example, when the 
transmitter 10 is connected into the branch conductor 18, the 
electromagnetic field is present at the branch conductor 18, the branch 
circuit breaker 26, and at one of the distribution conductors 42 and main 
distribution circuit breakers 40 and feeder conductors 38 electrically 
connected to the branch line circuit breaker 26 and over the neutral 
conductor 32. 
The receiver 12 includes an electromagnetic field transducer means 
positioned at the distal end of a probe 46. The receiver 12 operatively 
senses the strength of the predetermined electromagnetic field, rejects 
substantially all other signals and operatively indicates the strength of 
the electromagnetic field by lighting one of a plurality of indicators 48. 
In this manner, branch conductors, electrical devices connected to the 
branch conductors, branch circuit breakers, distribution conductors, main 
distribution breakers, the feeder conductors and the neutral conductors 
can be traced and identified without interrupting the supply of power from 
the power distribution network. 
Identification proceeds by placing the transducer means of the probe 46 
adjacent each of the electrical devices in question and noting the field 
strength indication on the indicators 48 with respect to each. The 
particular electrical device exhibiting the greatest field strength 
indication is the device identified. As will become more apparent, the 
nature and operation of each transmitter 10 and receiver 12 provide an 
improved capability for tracing and identifying electrical devices. 
One embodiment 10' of the transmitter 10 is better understood by reference 
to FIGS. 2 and 3A to 3I. As shown in FIG. 2, the transmitter 10' comprises 
a power supply circuit 50, a duty cycle controller 52, a gated oscillator 
54, a driver circuit 56, and an output circuit 58. The power supply 
circuit 50 generally draws applied AC or DC power from the power 
distribution network through the plug prongs 14. Alternating current power 
is rectified by a full wave rectifier 60 and pulsating direct current 
power (FIG. 3G) is applied on conductor 62 to a conventional integrating 
and regulating circuit comprising a transistor 64, a Zener diode 66, a 
capacitor 68, and the resistors 70 and 72. Regulated DC power is thereby 
present on terminal 74 and is conducted to other active components of the 
elements 52, 54, 56 and 58 of the transmitter 10'. 
In the duty cycle controller 52, a pair of capacitors 76 and 78 are 
electrically connected in series between the terminal 74 and a ground 
reference 80. The input terminal to an inverting Schmitt trigger 82 is 
connected to the terminal 3A between capacitors 76 and 78. The output 
terminal 3B of the Schmitt trigger 82 is connected to the input terminal 
3A through a parallel-branch feedback network comprising resistors 84 and 
86 and a diode 88. The capacitors 76 and 78, the resistors 84 and 86, and 
the diode 88, in conjunction with the Schmitt trigger 82, form a resistive 
capacitive timing network of the duty cycle controller. Capacitors 76 and 
78 charge and discharge at terminal 3A in accordance with the waveform 
diagram shown in FIG. 3A. The output waveform from the Schmitt trigger 82 
at terminal 3B is illustrated in FIG. 3B. Accordingly, the two time 
periods established by the timing network, as shown in FIGS. 3A and 3B: a 
considerably longer off-time period 89, for example approximately 900 
milliseconds, and a considerably shorter on-time period 91, for example 
approximately 45 milliseconds. The ratio of the on-time period 91 to the 
off-time period 89 is controlled by the ratio of the sum of the 
resistances of resistors 84 and 86 to the resistance of the resistor 84. 
The on-time period 91 and the off-time period 89 are controlled by the 
values of the resistors 84 and 86 in relation to the values of capacitors 
76 and 78. The diode 88 operatively connects the resistor 86 in the 
feedback path of Schmitt trigger 82 during the on-time period 91 but 
eliminates the resistor 86 from the feedback path during the off-time 
period 89. 
The gated oscillator 54 is operative during the on-time period 91 when a 
diode 90 connected between terminal 3B and an input terminal 3C of an 
inverting Schmitt trigger 92 is not conductive. A capacitor 94 is 
connected between terminal 3C and the ground reference 80. A resistive 
feedback path defined by resistor 96 and potentiometer 98 is connected 
between an output terminal 3D of the Schmitt trigger 92 and its input 
terminal 3C. When the potential of the signal at terminal 3B attains its 
minimum level during the off-time period 89, the diode 90 is conductive. 
The voltage increases on capacitor 94 until the trip point of the Schmitt 
trigger 92 is attained. At the trip point, the resistive feedback path of 
the resistor 96 and potentiometer 98 in conjunction with the capacitor 94 
causes the Schmitt trigger 92 to oscillate at a frequency established by 
the values of the elements 94, 96 and 98, for example about 6 kHz. The 
frequency of oscillation can be adjusted by varying the resistance of the 
potentiometer 98. The oscillation occurring during the on-time period 91 
is illustrated in FIGS. 3C and 3D, with FIG. 3C illustrating the input 
signal at terminal 3C and FIG. 3D illustrating the output signal at 
terminal 3D from the Schmitt trigger 92. 
Four parallel-connected, inverting Schmitt triggers 100, 102, 104 and 106 
primarily define the driver circuit 56. The Schmitt triggers 100, 102, 104 
and 106 receive as an input signal the signal at terminal 3D shown in FIG. 
3D. The Schmitt triggers of the driver circuit 56 provide added current to 
drive a transistor 108 of the output circuit 58. The waveform at the 
common output terminal 3E of the Schmitt triggers 100, 102, 104 and 106 is 
shown in FIG. 3E. 
The output circuit 58 draws high freqency alternating current from the 
power distribution network when the transistor 108 is conductive. Of 
course, the transistor 108 conducts in accordance with the alternating 
high frequency current established during the on-time period 91 of the 
waveform, shown in FIG. 3E. When conductive, the transistor 108 
operatively connects resistors 110 and 112 between the conductor 62 and 
the ground reference 80. Since the transistor 108 conducts in accordance 
with its input signal (FIG. 3E) during the on-time period 91, a high 
frequency current is conducted through resistors 110 and 112 as shown by 
the current waveform in FIG. 3H, which is better illustrated by virtue of 
the time expansion shown by FIG. 5A. FIG. 3G illustrates the voltage 
waveform present on conductor 62 and at terminal 3G. A gas indicator bulb 
114 lights during the time periods that the transistor 108 is conductive 
and indicates the operation of the transmitter 10. 
The high frequency rectified current loading shown in FIG. 3H is conducted 
from the full wave rectifier 60 over conductors 115 and 116 to the pronged 
plug 14, as the alternating waveform shown in FIG. 3I. A varistor 118, is 
connected between conductors 115 and 116 to protect the transmitter 10 
from overvoltage conditions due to voltage transients, lightning and 
inductive spikes and the like and from possible improper use. A fuse 120 
is connected in the conductor 116 to protect the transmitter from 
excessive currents. 
One of the significant advantages of conducting a high frequency current 
loading signal from the power distribution network in the duty cycle 
established by the on and off time periods 91 and 89 respectively, is that 
the high-frequency current-conducting transistor 108 does not experience 
excessive heating. The extent of heating of the transistor is related to 
the square of the voltage at the plug prong 14 during the time when the 
transmitter is conductive. In a practical embodiment, the transistor 108 
can conduct as much as one amp of current which, during the time the 
transistor is conductive, results in significant heat creation. However, 
by operating with a duty cycle having a significantly long off-time period 
89 as compared to the on-time period 91, the average effect of the heating 
is greatly reduced. The relatively large current conducted during the 
on-time period creates an electromagnetic field of sufficient strength to 
be reliably detected at significantly remote locations along the 
conductors within the power distribution network. By not operating with a 
duty cycle characteristic, the strength of the field would be 
substantially reduced, or relatively expensive and additional components 
would be required to obtain comparable field strength. Accordingly, the 
number and cost of the elements in the transmitter is reduced, the life of 
the transistor 108 is prolonged and the reliability of the transmitter is 
enhanced. 
One embodiment 12' of the receiver 12 is better understood by reference to 
FIG. 4. The receiver 12' includes a transducer means 122, a filter section 
124, a variable gain section 126, an indicator section 128, a peak 
detector 130, a level indicator 132, and a reset section 134 operatively 
connected together. In addition, the receiver 12' includes a ground 
reference section 136 for maintaining the voltage reference levels of the 
elements of the receiver 12'. 
The receiver 12' receives energy from self-contained batteries 140 and 142 
of the ground reference section 136. The battery 140 is electrically 
connected between terminals 146 and 148 to operatively establish a 
positive voltage level on terminal 146 and a negative voltage level on 
terminal 148. The positive and negative voltage levels at terminals 146 
and 148, respectively, are equally spaced above and below ground reference 
144 and are maintained in the equally spaced relationship by the 
operational amplifier 150, the resistors 152 and 154 and the capacitor 156 
connected in a known operative arrangement. 
The transducer 122 is of the inductive type and utilizes an inductor or 
coil 160. A capacitor 162 is connected in parallel relationship with the 
inductor 160, and the capacitor-inductor combination is a tuned or 
resonant circuit with a resonant frequency equal to the frequency of the 
current loading pulses delivered by the transmitter 10 during the on-time 
period 91. A resistor 164 is connected between the coil 160 and capacitor 
162 and the inverting input terminal of an operational amplifier 166. The 
resistor 164 reduces the effects of ringing in the tuned circuit 160 and 
162 which may occur as a result of high-frequency transients that appear 
randomly and spuriously on the conductors of the power distribution 
network. A feedback loop defined by a capacitor 168 and resistor 170 is 
connected between the output terminal of the operational amplifier 166 and 
its input terminal. The values of elements 164, 168 and 170 establish the 
operational amplifier 166 as a low gain, low pass frequency amplifier. 
Accordingly, the signal inductively received by the tuned circuit 160 and 
162 is amplified to a magnitude well within the range between the positive 
and the negative voltage supply levels established by the ground reference 
section 136. The values of the elements 164, 166, 168 and 170 also 
operatively establish a roll-off frequency point, for example about 35 
kHz, at a frequency substantially higher than the high-frequency signal 
from the transmitter 10 but substantially less than the major 
high-frequency components of voltage transients and spikes. A very flat 
gain response is thereby obtained between the frequency of the transmitter 
signal and the power line frequency, typically 60 Hz. Consequently, the 
power line frequency will not be amplified more than the transmitter 
signal frequency, and the high frequency transients and spikes will be 
attenuated. 
The filter section 124 basically comprises three serially connected filter 
means, 172, 174 and 176. Each of the filter means 172, 174 and 176 is an 
identical Sallen-Key band-pass filter of well-known circuit configuration. 
The component values of each Sallen-Key band-pass filter are selected to 
provide a low Q for each individual filter, for example approximately two. 
As is well known, a Q is one measure of the ability of a band-pass filter 
to pass a particular range of frequencies. One definition of Q is the 
center frequency of the band pass filter divided by the frequency band 
width which the filter passes. Band-pass filters with high Q's are more 
susceptible to ringing than band-pass filters of lower Q's. During ringing 
a band-pass filter is rendered inoperative for its intended purpose. By 
placing three relatively low Q filters in series, the total Q of the 
filter section 124 for band-pass purposes is the sum of the Qs of each 
filter. For example, if each of the filters 172, 174 and 176 has an 
individual Q of two, the total Q of the filter section 124 is 
approximately six with respect to filtering the desired signal. The 
ability of the filter section 124 to withstand the effects of ringing from 
high-frequency transients is not related to its total Q, as it would be in 
the case of a single high Q filter. Instead, the ability to withstand 
ringing in the filter section 124 is related to the ability of one of the 
filters 172, 174 or 176 to withstand the potential for ringing. As an 
important result, the filter section 124 is highly selective in passing 
only the transmitter signal but is not highly susceptible to ringing. 
Accordingly, the signal present on conductor 178 is essentially a signal 
directed related to the high-frequency current loading created by the 
transmitter 10. The alternating power frequency has been selectively 
removed by the operational effects of the Sallen-Key filter means 172, 174 
and 176. In addition, the effects of high-frequencies have been 
significantly attenuated by the coil 160 and capacitor 162, and by the 
resistive capacitive network of elements 164, 168 and 170. The three 
Sallen-Key band-pass filter means 172, 174 and 176 even further isolate 
and supply the signal created by the transmitter 10. 
The signal available on conductor 178 is generally referenced positively 
with respect to ground 144. The positive reference results because each of 
the operational amplifiers of the Sallen-Key band-pass filters 172, 174 
and 176 are referenced directly to ground 144 without current offsets. 
It is the general function of the variable gain section 126 to amplify the 
signal on conductor 178 and to reference that signal midway between the 
positive and negative voltage levels at terminals 146 and 148. The signal 
on conductor 178 is applied through a potentiometer 180 and a resistor 182 
to the inverting input of an operational amplifier 184. A feedback network 
comprising resistors 186, 188 and 190 and a multiposition switch 192 is 
provided to adjust the gain of the operational amplifier 184. By 
positioning the switch 192 in one of its three positions, one or more of 
the resistors 186, 188 and 190 is connected in the feedback network to 
control the gain. The various stages of gain provided by the feedback 
network accommodate different strengths of signals detected. It is 
apparent that the high-frequency current drawn by the transmitter 10 
induces a signal strength in the conductors which diminishes in accordance 
with the length over which the signal is conducted, the number of elements 
through which the signal must be conducted and the presence of an exterior 
shielding enclosure. The highest gain available is when the switch 192 is 
open and all resistors 186, 188 and 190 are connected in the feedback 
loop. The highest level of gain is desirable for identifying current 
carrying devices in panel boxes or distribution or feeder conductors. In 
the medium gain setting, where resistors 186 and 190 are connected in the 
feedback loop, the gain is generally sufficient for tracing and 
identifying circuit breakers and switches. The low stage of gain, when 
switch 192 connects only resistor 186 in the feedback loop, is useful for 
detecting accessible branch conductors, for example. 
With the appropriate level of gain established, the signal supplied from 
the output terminal of the operational amplifier 184 is supplied to an 
inverting input terminal of an operational amplifier 194 of the indicator 
section 128. The current supplied from the operational amplifier 194 
drives a piezoelectric speaker 196. The speaker 196 supplies an audio 
signal at the frequency of the current loading signal from the transmitter 
10, for example 6 kHz. This frequency of the transmitter signal is easily 
audibly perceived by the user as an assurance of proper identification and 
use of the transmitter 10 and receiver 12. 
Feedback from the output terminal of the operational amplifier 194 is 
conducted from the audio indicator section 128 through a resistor 198 to 
the noninverting input of the operational amplifier 184 of the variable 
gain section 126. A capacitor 200 charges to the center level of the 
signal delivered from the operation amplifier 194. Since the operational 
amplifiers 184 and 194 each invert their input signals, the signal present 
on capacitor 200 is essentially of the same polarity as the average center 
level of the signal supplied to the inverting input terminal of the 
operational amplifier 184. Accordingly, the signal supplied at the output 
terminal of the operational amplifier 184 is centered with respect to the 
ground reference 144 and this signal is applied on conductor 202 to the 
peak detector section 130. 
In the peak detector 130, the signal on conductor 202 passes through 
capacitor 204 which, in conjunction with resistor 206, changes the level 
of the signal on conductor 202 from being referenced to the ground 
reference 144 to being referenced to the negative voltage level at 
terminal 148. A resistor 208 is connected between the noninverting input 
of an operational amplifier 210 and the capacitor 204. The resistor 208 
prevents current from being drawn from the capacitor 204 when the voltage 
on the capacitor 204 swings below the negative voltage level at terminal 
148 on every other half cycle of the signal on conductor 202. Output 
current delivered from the output terminal of the operational amplifier 
210 is conducted through a resistor 212, a diode 214 to a capacitor 216. 
As will become more apparent, the capacitor 216 is normally maintained in 
a discharged condition with respect to the negative voltage level on 
terminal 148. During the off-time period 89 of the transmitter signal, no 
signal is present on conductor 202, and the voltage signal applied to the 
noninverting input of the operation amplifier 210 is essentially at the 
negative power supply voltage at terminal 148. The output terminal of the 
operational amplifier 210 is also held at the negative power supply 
voltage of terminal 148 due to the discharged condition of the capacitor 
216. The feedback resistor 218 assures that the output terminal of the 
operational amplifier 210 is maintained at the negative supply voltage by 
balancing the offset voltages produced by the operational amplifier 210 
and the resistor 208. 
Upon the detection of a current loading signal from the transmitter 10, the 
operational amplifier 210 delivers pulses of current to the diode 214 and 
creates a voltage level on conductor 220 which is somewhat positive with 
respect to the negative supply voltage 148. The resulting signal on 
conductor 220 creates effects in the level indicator 132 and the reset 
section 134 which allow the capacitor 216 to charge to a level 
representative of the maximum level of the current loading frequency 
signal with respect to the negative supply voltage. Stated another way, 
the voltage on capacitor 216 will be allowed to increase to one half of 
the maximum peak to peak voltage of the signal present at the noninverting 
input terminal of the operational amplifier 210. The resistor 212 allows 
the voltage level on the capacitor 216 to increase at a predetermined rate 
and causes the voltage on capacitor 216 to reach its maximum value only 
after a predetermined number of complete cycles of the current loading 
transmitter signals have been conducted through the operational amplifier 
210. 
The voltage level on conductor 220 is supplied to a level detector means 
222 of the level indicator section 132. The level detector 222 is a bar 
and dot graph integrated circuit marketed under the designation LM3914N. 
The level detector 222 has connected thereto a plurality of ten 
light-emitting diodes, each of which is referenced 48. The diodes 48 are 
arranged in a predetermined order along a predetermined scale. Depending 
upon the voltage level on the conductor 220, one of the diodes 48 will be 
energized. A higher voltage level on the conductor 220 will energize a 
light-emitting diode toward one end of the predetermined scale, and a 
lower voltage level will energize the light-emitting diode toward the 
other end of the predetermined scale. By noting the position of the 
light-emitting diode 48 which is energized, the relative strength of the 
transmitter signal is indicated. The user can determine which of the 
various electrical devices in close proximity to the probe 46 (FIG. 1) 
containing the transducer 122 is conducting the current loading signal 
created by the transmitter 10. 
The reset section 134 operates in conjunction with the level detector 132 
to periodically allow the capacitor 216 to charge to a predetermined 
maximum voltage, to hold the maximum voltage for a predetermined period of 
time, and to thereafter discharge to a condition ready for reception of 
another current loading signal supplied by the transmitter. The reset 
section 134 includes a transistor 224 which is rendered conductive when 
one of the light-emitting diodes 48 is energized by the level detector 
222. When transistor 224 becomes conductive, it triggers transistor 226 
into conduction. Normally, transistor 226 is not conductive and capacitor 
228 has charged through resistors 230 and 232 to a voltage level present 
between the positive supply voltage 146 and the negative voltage 148. A 
terminal 234 of the capacitor 228 thereby achieves a voltage approximating 
the negative supply voltage at terminal 148. The terminal 234 is connected 
to the gate terminal of a field effect transistor 236. The source terminal 
of the transistor 236 is also connected to the negative supply voltage 
148. When the gate terminal voltage and the source terminal voltage are 
approximately equal, the transistor 236 becomes conductive to discharge 
the capacitor 216 through resistor 238. The transistor 236 remains 
conductive only during the time period that a signal is not present on 
conductor 220, i.e. when transistors 224 and 226 are not conductive. 
However, when transistor 224 becomes conductive under the condition of a 
signal being applied to conductor 220 and one of the light-emitting diodes 
48 becomes conductive, the terminal 240 of capacitor 228 is operatively 
connected to the negative supply voltage at terminal 148. The voltage at 
terminal 234 immediately goes to a level substantially below the negative 
supply voltage at terminal 148 which causes the transistor 236 to become 
nonconductive. The capacitor 216 commences charging and continues to 
charge so long as the voltage at terminal 234 remains below the voltage at 
the negative supply voltage at terminal 148. This condition exists for a 
sampling time period 239, shown in FIG. 5B, the length of which is 
determined by the discharge period established by the values of the 
resistor 232 and capacitor 228. The time period 239 is substantially 
greater than the on-time period 91 of the transmitter signal but less than 
the off-time period 89 of the transmitter signal (FIGS. 3E and 8J). 
Accordingly, the capacitor 216 is in condition to charge to its maximum 
level during the time period 89 that the transmitter 10 creates the 
current loading signal. For false spurious signals of short duration and 
of frequency comparable to the transmitter frequency, resistor 212 is 
conductive only momentarily and capacitor 216 does not attain a 
significant level to operatively result in a discernably intelligible 
indication at the light-emitting diodes 48 before the false signal 
dissipates. 
The maximum voltage level to which capacitor 216 is charged is maintained 
during the sampling time period 239. The maximum charge level is 
maintained on conductor 220 for a sufficient period of time 239 to allow 
the level detector 222 to energize the appropriate light-emitting diode 48 
and indicate the maximum attained transmitter signal strength. By holding 
the voltage level on conductor 220 for the sample time period 239, a 
constant indication is available from one of the diodes 48 for an amount 
of time sufficient for intelligent observation. After the sampling time 
period 239 ends, the transistor 236 again becomes conductive and the 
capacitor 216 is immediately discharged through resistor 238. In the 
discharged condition during a reset time period 241, the receiver awaits 
the reception of another current loading transmitter signal during the 
on-time period 89. Once the first cycle of the transmitter signal is 
conducted through the receiver 12' in the manner described, the capacitor 
216 again starts charging to the maximum level during the sample period 
239. After the appropriate light-emitting diode 48 has been energized to 
indicate the maximum attained transmitter signal strength, represented by 
the voltage level on conductor 220 and across capacitor 216, the capacitor 
216 is again discharged during the reset time period 241. During the reset 
time period 241, the capacitor 228 recharges to the voltage level between 
the positive and negative supply terminals 146 and 148, respectively. The 
length of the reset time period 241 is established by the values of the 
capacitor 228 and the resistors 230 and 232. 
From the foregoing description, it is apparent that the receiver 12' 
operates during the sample time period 239 to indicate the presence and 
strength of the transmitter signal. The detected signal is effectively 
filtered by an improved filtering arrangement to eliminate or reduce the 
influence of spurious signals such as voltage transients. Occasional 
spurious signals which may be coupled through the receiver to the diodes 
48 remain only for a short period of time due to the lack of significant 
effects from false signals and/or the relatively short sample and reset 
periods of operation provided by the reset section 134. Any false or 
spurious signals are quickly eliminated from consideration because they do 
not continually cause the repeated energization of the same or 
approximately the same light-emitting diode, as would occur upon detection 
of a constantly applied transmitter signal of the same signal strength. 
Accordingly, not only does the receiver 12' utilize improved filtering 
techniques to eliminate many of the adverse effects of spurious signals, 
but its indication of the strength of the transmitter signal has the 
effect of substantially further eliminating various adverse effects. By 
causing the sample period 239 to be considerably longer than the on-time 
period 91 of the transmitter signal, a sufficient time frame is 
established whereby one group of current loading transmitter pulses will 
be detected and their signal strength established. By making the sample 
period 239 less than the off-time period 89 of the transmitter signal, 
only one signal group from the transmitter will have an operative effect 
on the receiver 12. By making each period of receiver operation (the sum 
of periods 239 and 241) less than the period of the transmitter duty cycle 
(the sum of periods 89 and 91), the receiver will be in condition to 
respond to each new signal supplied by the transmitter. Accordingly, the 
improved filtering and sampling effects of the receiver assure high 
transmitter signal sensitivity and improved immunity to the effects from 
spurious signals, to a degree which has heretofore been unavailable in the 
field of tracing and identifying electrical conducting devices. 
Another embodiment of an improved receiver can be understood by reference 
to FIG. 6. The elements illustrated in FIG. 6 are an alternative to the 
Sallen-Key band-pass filter means 172, 174 and 176 employed in the filter 
section 124 of the receiver 12' shown in FIG. 4. The function of the 
elements shown in FIG. 6 is to provide more improved filtering than that 
available from the technique of serially connecting a plurality of 
relatively low Q band-pass filters. An even more improved and enhanced 
sensitivity and ability to reliably detect the transmitter signal results. 
The improved filter section 124' shown in FIG. 6 includes a prefilter means 
242, a digital switching filter means 243, and impedence converter means 
244 and a terminal filter means 246. The prefilter 242 takes the form of a 
typical Sallen-Key band-pass filter which employs resistive and capacitive 
component values selected to primarily reduce the typical alternating 
current power frequency, i.e. 60 Hz, and other low frequencies to an 
acceptable level for preventing unnecessary influences on the digital 
switching filter 243. The digital switching filter 243 has the capability 
of allowing only signals of a very narrow preselected main band-pass 
frequency to remain on conductor 248, as well as very low frequencies and 
harmonics of the main band-pass frequency. The other signals on conductor 
248 are in essence coupled to ground 144 and are not passed to the 
impedence converter 244. Signals of a spurious or random nature are 
therefore essentially coupled to ground since such signals typically do 
not fall within the low frequency range or the precise narrow primary 
band-pass frequency range or harmonics of the primary band-pass frequency. 
In essence, the digital switching filter 243 will pass signals having a 
consistent repetitious phase angle relative to the phase angle of the 
signals of the primary band-pass frequency. 
The digital switching filter 242 comprises a plurality of capacitors 250, 
252, 254, 256, 258, 260, 262 and 264 which are connected between the 
conductor 248 and the eight output terminals of a one-of-eight input 
select switch 268. The switch 268 functions to connect one of its output 
terminals to which the capacitors 250-264 are connected, to the ground 
reference 144. The one of the output terminals which is connected to 
ground is selected by a binary signal supplied on conductors 270. While 
the selected input is connected to ground reference 144, the remaining 
other inputs are disconnected from ground reference and are allowed to 
float, thereby not providing a conduction path through those other inputs 
to the ground reference. A binary counter 272 supplies signals on the 
conductors 270. A clock signal is supplied to the clock terminal of the 
binary counter 272 from an oscillator 273, which comprises an inverting 
operational amplifier 274, a crystal 276, capacitors 278 and 280 and a 
resistor 282. The predetermined operational frequency of the oscillator 
273 is established and is very precisely regulated by the characteristics 
of the crystal 276. The oscillator frequency supplied to the clock input 
terminal of the binary counter 272 is an exact predetermined multiple of 
the predetermined narrow band-pass frequency of the digital switching 
filter 243, with the predetermined multiple being equal to the number of 
output terminals of the switch 268 to which capacitors are connected. For 
example, if the primary band-pass frequency of the digital switching 
filter is 6 kHz, the frequency of the clock pulses supplied by the 
oscillator 273 to the clock terminal of the binary counter 272 is 48 kHz. 
For signals at the primary band-pass frequency of the digital switching 
filter 243, each of the capacitors 250-264 will charge or integrate over 
one eighth of each cycle of the signal. Each capacitor will eventually 
charge to a level equal to an average applied signal level during its 
conduction interval. Thereafter, when a current conduction path exists 
through each of the capacitors 250-264 during its conduction interval, the 
voltage level previously established on that capacitor is essentially 
equal to the voltage level present on conductor 248 during that time 
interval. The voltage level or signal on conductor 248 is thereby 
unaffected since the switching of the capacitor 250-264 to ground 
reference 144 does not adversely shunt the signal level on conductor 248 
to ground. However, for signals which are not in phase with the primary 
band-pass frequency or its harmonic multiples, each of the capacitors 
250-264 will charge to randomly different levels during the intervals when 
they are connected individually to the ground reference 144. Accordingly, 
since there will be no similarity of the signal levels on the capacitors 
250-264 relative to the applied signal on conductor 266 during the 
subsequent conduction intervals, a substantial portion of the signal on 
conductor 248 will be shunted to ground 144 or will be smoothed by the 
effect of the capacitors 250-264 discharging to or from the conductor 248. 
The end result is that all signals other than the primary band-pass 
frequency or its exact multiples are substantially attenuated on conductor 
248. 
The impedence converter 244 essentially buffers the impedence of the 
digital switching filter 243 with respect to the terminal filter 246. The 
impedence converter 244 also provides a desired amount of gain established 
by its feedback resistor 286. The signals which are allowed to pass from 
the digital switching filter 243 on the conductor 248 are passed through 
the impedence converter 244 on conductor 288 to the terminal filter 246. 
The terminal filter 246 essentially comprises another typical Sallen-Key 
band-pass filter, the primary function of which is to attenuate any 
switching noise included with the signal on conductor 288. The terminal 
filter 246 also attenuates any harmonic frequency components that may be 
included with the signal. The terminal filter supplies its signal on 
conductor 178, to the other elements of the receiver 12' shown in FIG. 4. 
The receiver 12' otherwise functions in the manner previously described. 
Due to the precise frequency passage characteristics of the digital 
switching filter 243, a highly reliable means for filtering or attenuating 
all spurious signals except the predetermined transmitter signal is 
achieved. Operation of the receiver 12 is thereby rendered even more 
immune to spurious signals, transients and potential ringing. The receiver 
12 becomes even more reliable in identifying and tracing electrical 
devices which conduct the predetermined transmitter signal. 
A transmitter 10" shown in FIG. 7 includes circuit elements functioning to 
deliver constant energy content current loading pulses to the power 
distribution network. The desirable result available from transmitting 
constant energy content current loading signal pulses is that the detected 
electromagnetic signals have approximately the same signal effect on the 
receiver 12 irrespective of the magnitude of the instantaneous voltage on 
the power distribution network. The end result is that the receiver 12 
will exhibit approximately the same sensitivity for signals from the 
transmitter 10", regardless of the point in time in the alternating power 
supply cycle that the signal pulses are delivered, and even in power 
distribution networks with different voltages. 
The energy content of a pulse is determined by its magnitude or voltage and 
its time width or duration. Specifically, the energy content is equal to 
the integral of the magnitude of the pulse over its time duration. FIG. 8L 
indicates the hyperbolic relationship of magnitude and duration to obtain 
a pulse of predetermined constant energy content. It is apparent that if 
the magnitude varies, for example, the duration must be adjusted 
accordingly to obtain the constant energy pulse. In the specific 
application described herein, the voltage magnitude of the transmitter 
pulse will vary continuously due to the alternating nature of the applied 
voltage over the power distribution network, and because the transmitter 
and receiver may be utilized in conjunction with power distribution 
networks in which different voltage levels are present. It is one of the 
functions of the transmitter 10" to modulate the time duration of the 
pulses in accordance with the constant energy hyperbolic relationship by 
utilizing the instantaneous voltage at approximately the time when the 
pulse is initiated as an independent variable. 
Constant energy current loading pulses create beneficial effects at the 
resonant or tuned circuit defined by the inductor 160 and capacitor 162 of 
the transducer 122 of the receiver 12 (FIG. 4). It has been discovered 
that a tuned circuit will integrate signals applied to it, and the output 
signal of the tuned circuit is the product of integrating the signal level 
with time, so long as the duration of the applied pulse signal does not 
exceed the time period of a half wave at the resonant frequency. Upon 
applying constant energy pulses, the output signal from the tuned circuit 
is of constant magnitude at the resonant frequency. The constant output 
signal obtains a more reliable, accurate and sensitive indication from the 
receiver. 
The output signal from a tuned circuit will essentially be a sine wave even 
though the input signal is a pulse wave. Assuming a positive going input 
pulse, the sine wave supplied from the resonant circuit will be in a phase 
relationship with the input pulse such that the center point (ninety 
degree phase position) of the positive going half wave of the sine wave 
output will be centered about the center point of the input pulse. Since 
the transmitter 10" will be delivering constant energy pulses of different 
magnitude and pulse width durations, it is important to consistently 
deliver the current loading pulses from the transmitter 10" at consistent 
time intervals equal to the period of the transmitter frequency to 
consistently reinforce the sine wave established in the resonant circuit 
in a consistent phase relationship. Since the current loading pulses will 
be pulse width modulated, the initial leading edges of these pulses from 
the transmitter 10" will vary slightly with respect to the time center 
point of each pulse delivered, and of course, the time center point of 
each delivered pulse will be desireably positioned at the equal intervals 
of the transmitter frequency. 
The hyperbolic relationship set forth in FIG. 8L also describes in 
proportional terms the time at which the leading edge of a pulse is to be 
initiated relative to a center reference at the center of the time 
duration of pulses of various widths. This proportionality relationship 
exists because the first half of the pulse will also possess constant 
energy characteristics, and the width of the first half of the pulse is 
established by the leading edge of the pulse relative to the pulse center 
point. 
The transmitter 10", shown in FIG. 7, includes circuit elements for 
achieving the above discussed considerations. The transmitter 10" 
comprises a clock 290, a hyperbolic generator 292, a start pulse generator 
294, a duty cycle controller 296, an output section 298 and a termination 
pulse generator 300. The clock 290 generally delivers frequency and timing 
signals for achieving proper operation of the transmitter 10". The 
hyperbolic generator 292 supplies a curve approximating a hyperbolic 
function, and the hyperbolic curve is utilized in establishing the point 
at which the leading edge of the current loading pulse will be delivered 
from the transmitter 10" so as to result in the centering of each current 
loading pulse at time intervals equal to the period of the transmitter 
frequency. The start pulse generator 292 delivers a signal on conductor 
300 for initiating the leading edge of the current loading pulse. The duty 
cycle controller 296 achieves two important functions. One function is to 
limit the duration of each current loading pulse to a time not exceeding 
one half of the period of the transmitter frequency. A safety signal 
delivered over conductor 302 achieves this function. The second function 
is to establish the overall duty cycle of the transmitter 10". A signal 
delivered over conductor 304 establishes the duty cycle from the 
transmitter 10". Upon occurrence of the signals on conductors 300, 304 and 
306, the output section 298 begins delivering current loading signal 
pulses to the power distribution network by absorbing power through the 
plug prong 14. The current loading pulses are of constant energy content, 
limited in pulse width duration to no greater than one half the period of 
the transmitter frequency, and are limited to the predetermined duty cycle 
established by the on-time 91 and off-time 89 (FIGS. 3A and 8J). After the 
initiation of each individual pulse, the termination pulse generator 300 
integrates the magnitude of the voltage of the particular pulse width 
time, and delivers a termination pulse on conductor 306 when the constant 
energy content is attained. The termination pulse on conductor 306 
operatively terminates the pulse from the output section 298. Thereafter, 
the transmitter 10" delivers the next constant energy pulse which is time 
centered with respect to the next time interval equal to the period of the 
transmitter frequency. 
The clock 290 comprises a pair of inverters 308 and 310, a crystal 312 and 
capacitors and resistors connected in a manner to obtain a square wave 
clock signal (FIG. 8A) of predetermined frequency. The clock signal is 
supplied from the clock 290 on the conductors leading from terminal 8A. 
The frequency of the clock signal is a predetermined even multiple of the 
transmitter frequency. In the exemplary arrangement shown in FIG. 7, the 
clock frequency is eight times the desired transmitter frequency. 
The hyperbolic generator 292 includes a binary counter 314 which receives 
the clock signal. The first three divide-by-two output terminals 8B, 8C 
and 8D of the counter 314 divide the clock signal and supply the pulse 
waveforms shown in FIGS. 8B, 8C and 8D, respectively. The signal on 
terminal 8D (FIG. 8D) establishes the transmitter frequency. The clock 
signal on terminal 8A and the signals on terminals 8B, 8C and 8D are 
applied through resistors to an integrating capacitor 316. When the 
signals on terminals 8A, 8B, 8C and 8D are high, the current is delivered 
to the capacitor 316 and the voltage thereacross is integrated. The 
resulting voltage across capacitor 316, present at terminal 8E, 
approximates a hyperbolic curve and is illustrated in FIG. 8E. The 
hyperbolic curve shown in FIG. 8E is established once during each initial 
half wave of the transmitter frequency square wave present at terminal 8D. 
The start pulse generator 294 includes an operational amplifier 318 which 
receives a signal representative of the hyperbolic curve shown in FIG. 8E 
on its noninverting input terminal 8E. On the inverting input terminal, 
the comparitor 318 receives a signal on conductor 320 representative of 
the instantaneous voltage on the power distribution network. The signal on 
conductor 320 is obtained from the full wave rectifying bridge 60 which is 
connected through the plug prong 14 to the power distribution network. The 
resistors 322 and 324 reduce the magnitude of the applied power signal on 
conductor 320 to a level generally commensurate with and proportional to 
the magnitude of the hyperbolic curve signal. When the voltage of the 
hyperbolic curve signal on terminal 8E slightly exceeds the voltage on 
conductor 320, the output of the operational amplifier 318 goes high and 
the start signal is applied on conductor 300. Because the power 
distribution voltage is compared to the hyperbolic curve signal, the time 
relationship of the leading edge of the current loading pulse relative to 
the magnitude of the applied voltage is established, as previously 
described in conjunction with FIG. 8L. The current loading pulse is 
initiated at a time whereby the resulting center point of the pulse will 
occur coincidentally with the same reference or phase point in each cycle 
of the transmitter signal, and that pulse center point will occur 
approximately at the positive-going edge of the pulses shown in FIG. 8D. 
In this manner the transmitter pulses, even though variable in duration, 
are centered in time with respect to a consistent reference point or phase 
angle of each cycle of the transmitter signal. The result is that the 
timed circuit of the receiver transducer detects the constant energy pulse 
electromagnetic fields at consistent points in time to best reinforce and 
continue the resonance established by the prior pulses supplied by the 
transmitter. 
The duty cycle controller 296 includes binary counters 326 and 328. The 
binary counter 326 is positive edge triggered while the counters 314 and 
328 are negative edge triggered. The signal on terminal 8C is applied to 
the enable input of the counter 326. The signal on terminal 8C is a square 
wave having a frequency twice that of the transmitter frequency, as can be 
seen by comparing FIGS. 8C and 8D. Because the counter 326 is positive 
edge triggered, an output signal from the first divide-by-two output 
terminal connected to conductor 302 is a signal shown in FIG. 8F, and that 
signal is approximately ninety degrees delayed in phase from the 
transmitter frequency signal shown in FIG. 8D. The signal on conductor 302 
is a safety signal and is operatively utilized for limiting the maximum 
pulse duration of the current loading pulses delivered by the transmitter 
10' to no greater than one half of the period of the transmitter frequency 
period. The signal on conductor 302 is applied as one input to the AND 
gate 330. 
The function of the binary counter 328 of the duty cycle controller 296 is 
to establish the overall transmission duty cycle for the transmitter 10". 
Input signals to the counter 328 are taken from the twelfth divide-by-two 
output terminal 8G of the binary counter 314 and are shown in FIG. 8G. The 
on-time period of the square wave shown in FIG. 8G establishes the on-time 
period 91 of the transmitter duty cycle shown in FIG. 8J. Output signals 
from the counter 328 are supplied on the first divide-by-two output 
terminal 8H and the eighth divide-by-two output terminal 8I. The signal on 
terminal 8I, shown in FIG. 8I, is operative to establish the overall time 
period for each duty cycle from the transmitter 10". 
The signals on terminals 8F, 8G, 8H and 8I are supplied as inputs to the 
AND gate 330. A high signal from the output terminal 8J of the AND gate 
330 is present on conductor 304 upon the coincidence of four high input 
signals to the AND gate 330, and this signal is shown in FIG. 8J. During 
the on-time period 91, the transmitter is allowed to deliver constant 
energy content current loading pulses to the power distribution network, 
as shown in FIGS. 8K and 8K'. During the off-time period 89 of the signal 
shown in FIG. 8J, the transmitter does not deliver signals. This 
arrangement, as described previously, greatly increases the power 
conducting capability, without detrimental heating. 
An AND gate 332 receives the signals present on conductors 300, 304 and 
306. As will be understood subsequently, the signal on conductor 306 is a 
normally high signal. The signal on conductor 300 is the start signal for 
initiation of the current loading pulse. The signal on conductor 304 is a 
signal which limits the time duration of current loading pulses. Upon 
receipt of three high signals on the conductors 300, 304 and 306, the AND 
gate 332 delivers a high trigger signal to the amplifier 334. The high 
signal, applied on the terminal 8K, biases a V-mos power transistor 336 to 
conduct current through a power resistor 338 to the ground reference. The 
current conducted through the power resistor 336 is reflected through the 
full wave bridge 60 to the power distribution network, in the manner 
previously described. 
In order to terminate the current loading pulse after a sufficient time 
duration has lapsed in accordance with the magnitude of the pulse to 
obtain a constant energy content, the termination pulse generator 330 
becomes operative to reduce the signal level on conductor 306 from a 
previous high level to a low level, thereby operatively terminating the 
bias signal to the transistor 336. When transistor 336 is conducting, 
resistor 340 applies approximately one half of the power network voltage 
to resistor 342. Capacitor 344 is charged through resistor 342. The 
resistor 342 and capacitor 344 act as an integrator and the voltage at 
terminal 346 is proportional to the energy content of each current loading 
pulse on an instantaneous basis. The voltage at terminal 346 is applied to 
the noninverting input terminal of an operational amplifier comparitor 
348. The voltage level of a reference 350 is applied to the inverting 
input terminal of the operational amplifier 348. The voltage established 
by the reference 350 is of a magnitude equal to the voltage across 
capacitor 344 at terminal 346 when the predetermined constant energy 
content of each current loading pulse is attained. When the voltage at 
terminal 346 reaches a level below that of the reference 350, the signal 
on conductor 306 goes low, thereby terminating the pulse when the 
predetermined constant energy content has been achieved. The signal on 
conductor 302 biases the transistor 352 to prevent integration by the 
capacitor 344 except during times when the safety signal (FIG. 8F) is 
high. The maximum pulse width duration of the pulse is limited to a 
half-wave of the transmitter frequency, or fifty percent, thereby keeping 
within the limitation over which the resonant circuit will be effective 
for integration. 
The transmitter 10" thereby delivers constant energy content pulses over 
the power distribution network irrespective of the voltage applied to the 
power distribution network. The pulses occur over time periods which are 
centered at regularly occurring reference intervals equal to the period of 
the transmitter frequency. The duration of each pulse is limited to a 
predetermined portion of each cycle of the transmitter frequency, and the 
transmission of signals from the transmitter is limited to a predetermined 
duty cycle. The effect at the receiver 12 is that an approximately 
constant level signal is available from the transducer and variations in 
the voltage of the power distribution network will not create false 
signals which might otherwise falsely indicate an incorrect electrical 
device sought to be identified. Although the tramitter 10" has been 
described primarily in conjunction with a use for tracing and identifying 
electrical conductors in a power distribution network, the previously 
described advantageous concepts of the transmitter 10" may be utilized in 
other applications as well. 
The significant advantages and improvements available from the embodiments 
of the transmitter and receiver of the present invention have been 
described. The specificity of description has, however, been made by way 
of example. The invention itself is defined by the scope of the appended 
claims.