A complementary push-pull emitter follower amplifier is coupled between the output of a high voltage driver amplifier and the cathode of a kinescope for reducing the effective capacitance presented to the driver amplifier that is attributable to the kinescope cathode, socket, spark gaps and related stray capacitances. A secondary undesired capacitance loading of the amplifier attributable to the collector to base capacitances of the follower amplifier is effectively reduced by regulating the collector emitter voltages of the push-pull follower output transistors at respective substantially constant values thereby improving parameters such as the slew rate and bandwidth of the overall video display system.

FIELD OF THE INVENTION 
This invention relates to amplifiers generally and particularly to 
apparatus for providing amplification of a video signal for driving the 
cathode electrode of a kinescope. 
BACKGROUND OF THE INVENTION 
In television apparatus employing direct view or projection kinescopes as 
display devices, it is desirable that the amplifier which drives the 
kinescope cathode provide a relatively high voltage drive signal having a 
wide bandwidth and a high slew rate. Typically, drive voltages may be on 
the order of 200 volts or so, bandwidths may be 5 MHz or higher and slew 
rates may be substantially more than 100 volts/micro-second. 
To facilitate high voltage operation it is common to employ a cascode 
configuration of a common emitter input stage driving a common base output 
stage. Such a configuration requires only one high voltage transistor (the 
output stage) and since it is connected in a common base configuration the 
Miller effect is suppressed and very wide bandwidth operation is thus 
possible. In practice, the actual bandwidth and slew rate which may be 
achieved in a cascode amplifier depends, to a great extent, on the 
effective load capacitance presented to the output stage and the available 
output current. 
In general one may either increase the amplifier operating current or 
decrease the effective load capacitance to maximize the bandwidth and slew 
rate of the amplifier. However, since increasing the current necessarily 
implies increasing the amplifier power dissipation, it is preferable to 
take steps to reduce the effective load capacitance for improved 
performance rather than resort to increases in operating power. 
In kinescope driver applications, the "effective" load capacitance 
presented to the amplifier is principally that of the kinescope cathode 
and stray capacitances associated with the socket, spark gaps, wiring and 
the like. An effective approach to reducing the effective capacitance 
loading is to couple the amplifier to the cathode by means of a push-pull 
complementary emitter follower amplifier. Such an amplifier effectively 
"isolates" the load capacitance approximately in proportion to the 
reciprocal of the transistor current gain ("beta"). The additional current 
provided by the follower amplifier provides faster charging and 
discharging of the load capacitance and thus enhances slew rate and 
bandwidth. To avoid substantially increasing the quiescent power 
dissipation, it is customary practice to operate the follower amplifier in 
a "class-B" mode in which the push-pull transistors are biased to avoid 
simultaneous conduction. 
It has been recognized by John H. Furrey, in U.S. Pat. No. 4,860,107 
entitled VIDEO DISPLAY DRIVER APATUS which issued 22 Aug., 1989, that 
one may obtain improved capacitance reduction by use of a "series form" of 
push-pull complementary emitter follower rather than the more ordinary 
"parallel form" of push-pull complementary emitter follower and Furrey 
developed such an amplifier having the desired class-B conduction 
operation. 
In more detail, by definition as used herein a "parallel form" of 
complementary emitter (or source) follower is one in which the inputs 
(base or gate electrodes) of a pair of complementary transistors (bipolar 
or field effect) are connected in parallel for receiving an input signal 
from an amplifier and the outputs (emitters or sources) are connected in 
parallel for driving a load. The term "series form" of complementary 
emitter or source follower is one in which the complementary emitter (or 
source) followers are connected in series to form a cascade connection 
between the output of an amplifier and a load and which includes diodes 
for by-passing the series transistor that is not driving the load. 
In the Furrey series form of complementary "push-pull" emitter follower, 
diodes are provided for each transistor to by-pass the transistor that is 
biased off. Specifically, a diode is connected across the base emitter 
junction of each transistor and each diode is poled for forward current 
conduction in a direction opposite to that of the associated base emitter 
junction. This significantly reduces the effective load capacitance of the 
display (kinescope load and stray capacitances) thereby improving positive 
and negative video signal transient response. 
SUMMARY OF THE INVENTION 
The use of complementary push-pull emitter follower (or, more generally, 
"voltage follower") amplifiers as described is effective in decreasing the 
capacitance presented to the driver amplifier attributable to the 
capacitance associated with the kinescope cathode. However, it is herein 
recognized that still further improvement in the driver amplifier 
performance may be obtained when using either type (i.e., the "series" 
type or the "parallel" type) of push-pull complementary emitter follower 
amplifier for coupling the output of the driver to the cathode electrode 
load. The present invention is directed to meeting this need. Also, the 
principles of the invention may be extended to single-ended driver 
applications as will be explained. 
Kinescope driver apparatus embodying the invention comprises a video 
amplifier having an output coupled to a kinescope cathode electrode via a 
voltage follower. The voltage follower comprises a transistor having a 
conduction path and a control electrode for controlling the conduction of 
the path, the control electrode being coupled to receive a video signal 
from the video amplifier, a first end of the conduction path being coupled 
to a point of reference potential via a current source and being coupled 
to the kinescope cathode, the second end of the conduction path being 
coupled to a source of supply voltage. A feedback circuit is provided for 
applying a positive feedback voltage to the second end of the conduction 
path of the voltage follower transistor for maintaining a substantially 
constant voltage across the conduction path, the substantially constant 
voltage being independent of variations in the video signal applied to the 
control electrode.

DETAILED DESCRIPTION 
Before considering the details of the television system of FIG. 1, it is 
helpful first to consider in more detail the problem of using conventional 
push-pull emitter follower buffer amplifiers for isolating the kinescope 
cathode capacitance from the output of the kinescope driver amplifier. As 
previously explained, the use of a follower amplifier is effective in 
reducing the capacitance attributable to the cathode (and associated 
strays) that is presented to the output of the high voltage video driver 
amplifier. However, it is herein recognized that the follower amplifier 
itself may introduce a capacitance loading effect on the driver amplifier 
and this will tend to limit the overall system performance. 
In more detail, it has been found that the main source of the undesirable 
capacitance loading effects in kinescope driver systems of the type using 
push-pull followers is attributable to the collector to base capacitances 
of the follower output transistors. Typically, these capacitances are 
smaller than the kinescope cathode capacitance and so isolating the 
cathode by a follower amplifier does provide an overall capacitance 
reduction and an improvement in slew rate and bandwidth as compared with 
direct coupled systems. However, to achieve the maximum benefit from the 
use of emitter follower isolation, it is herein recognized as being 
desirable to reduce the effective capacitance of the follower amplifier 
itself. 
To achieve an effective reduction in follower capacitance, in accordance 
with the present invention, feedback is used in such a manner as to reduce 
the flow of current under dynamic signal conditions in the collector to 
base capacitances of the follower transistors. This is achieved, as will 
be explained in more detail later, by applying the feedback in such as 
manner as to maintain a substantially constant collector to emitter 
voltage for the follower transistors. This, in turn, tends to maintain the 
collector to base voltage constant. As a result, under dynamic signal 
conditions there will be little or no charging or discharging of the 
collector to base capacitances as the signal voltage varies. 
The effective reduction in follower input capacitance attributable to the 
transistor collector to base capacitances is a function of the percentage 
of feedback applied to regulating the collector to emitter voltage. If, 
for example, the feedback percentage is selected such that collector to 
emitter voltage variations are reduced by fifty percent, then the reactive 
currents charging and discharging the collector to base capacitances of 
the follower amplifier will be also reduced by fifty percent and so the 
"effective" capacitance loading will be cut in half. Greater reductions in 
follower capacitance may be achieved as the feedback percentage is 
increased towards unity. In the examples of the invention which follow the 
percentage of positive feedback applied approaches one hundred percent. 
For purposes of circuit stability, provisions are made to ensure that the 
feedback gain can not equal or exceed unity. In the illustrated examples, 
this is achieved by connecting all "active" semiconductor devices in the 
feedback paths in voltage or "emitter" follower configurations. 
The foregoing overview of the principles of the invention will now be 
discussed in detail with respect to the example of FIG. 1 which 
illustrates a television display system including a video signal source 10 
for supplying a video signal S1 to a kinescope cathode 16 for display. To 
simplify the drawing, the details of the kinescope and the signal source 
are not shown. It will be appreciated that for a color system there would 
be three driver amplifiers. 
As an overview, to amplify the video signal to the high voltage levels 
required at cathode 16 the system includes a cascode type of high voltage 
amplifier 20 (outlined in phantom). To isolate the output of the high 
voltage amplifier 20 from the capacitance of the kinescope cathode 16 the 
output of amplifier 20 (collector of transistor Q3) is coupled to cathode 
16 via a push pull complementary emitter follower amplifier 30 (outlined 
in phantom). To protect the driver amplifier from kinescope arcs, the 
follower output terminal 15 is coupled to the cathode 16 by means of a 
kinescope arc protection resistor R15 and inductor L1. To provide for 
automatic kinescope bias (AKB) operation, a cathode current sensing 
circuit 40 ("Ik sense", outlined in phantom) is provided which senses the 
collector current of a PNP transistor (Q7) in the push-pull emitter 
follower amplifier 30 to generate an AKB output signal at an output 
terminal 18 proportional to the cathode current, Ik, of the kinescope 
cathode 16. This feature is optional and may be omitted as discussed 
later. 
Finally, to reduce the effective capacitance presented to the high voltage 
amplifier which is attributable to the collector to base capacitances of 
the complementary emitter follower 30, the system includes a feedback 
control circuit 50 (outlined in phantom) which maintains a substantially 
constant collector emitter voltage for the NPN transistor Q4 of the 
follower 30 and another feedback control circuit 60 (outlined in phantom) 
which maintains a substantially constant collector emitter voltage for the 
PNP transistor Q7 of follower 30. As previously noted, and explained in 
more detail later, the operation of the follower transistors at constant 
values of collector to emitter voltage tends also to regulate the 
collector to base voltage at a nearly constant value and this, in turn, 
tends to reduce the magnitude of the charging and discharging currents of 
the collector to base capacitances of the follower transistors. The 
beneficial result is that, since the driver amplifier 20 does not have to 
supply charging and discharging currents for these "parasitic" 
capacitances, the overall slew rate, bandwidth and transient response 
characteristics are improved. 
Consideration will now be given to circuit details and further operational 
features of the video display system of FIG. 1. The signal source 10 may 
be of conventional design including a tuner, IF amplifier and video 
detector as well as baseband processing providing hue and saturation 
control, brightness and contrast control and matrixing to component (e.g., 
RGB) for display. The kinescope may be of monochrome form or it may be of 
the color type (direct view or projection). For such color video 
applications three of the kinescope driver systems will be needed, one for 
each cathode to be driven. High voltage power (e.g., 200 volts or so) for 
operation of the amplifier 20 and the feedback or regulator circuits 50 
and 60 is provided by high voltage (H.V.) supply terminal 20. Decoupling 
of the high voltage supply (20) is provided by a decoupling network or low 
pass filter comprising resistor R20 and capacitor C20. A low voltage 
(L.V.) supply terminal 21 provides a relatively low voltage (e.g., 12 
volts or so) for biasing the input and cascode stages (transistors Q1-Q3) 
of the high voltage video driver amplifier 20. This supply input is also 
decoupled by means of an RC network comprising resistor R21 and capacitor 
C21. 
The high voltage driver amplifier 20 comprises an NPN common emitter 
connected input transistor Q2 connected in cascode with a common base 
connected NPN output transistor Q3. A fixed base bias voltage for the 
cascode output transistor Q3 is provided by the low voltage (e.g., +12 
volts) decoupling network (R21, C21). A lower potential for operation of 
the emitter load resistor R6 of the input transistor Q2 is provided by a 
Zener diode regulator comprising resistor R5 and Zener diode CR1 coupled 
between the base of transistor Q3 and ground. Illustratively, the Zener 
voltage may be 5 or 6 volts which establishes a DC reference for the load 
resistor R6 of the cascode input transistor as well as a DC reference for 
the AKB sense amplifier 40. The emitter electrode of the input transistor 
Q2 is also coupled to ground via a high frequency peaking network 
comprising resistor R7 and capacitor C2 which are coupled in series. 
The video input signal to be amplified, provided by source 10, is applied 
to the base of the cascode input transistor via an emitter follower input 
stage comprising PNP transistor Q1 which is connected at the collector 
thereof to ground and coupled at the base thereof to the video input 
terminal 12 via an input resistor R3. The emitter of transistor Q1 is 
coupled to the base of transistor Q2 and to the low voltage supply 21 via 
an emitter resistor R4. Additional high frequency peaking is provided by a 
further peaking network comprising series connected resistor R1 and 
capacitor C1 coupled in parallel with the input resistor R3. 
The collector load for the cascade amplifier 20 is provided by resistor R8 
which is coupled from the high voltage supply 10 to the collector of the 
cascode output transistor Q3. A diode CR3 is interposed between the load 
resistor R8 and the collector of transistor Q3 to provided a small offset 
voltage for reducing cross-over distortion in the complementary emitter 
follower amplifier 30. 
In operation of the cascode amplifier 20, the open loop gain is directly 
proportional to the value of the load resistor R8 and inversely 
proportional to the impedance of the emitter network R6, C2 and R7 as 
previously discussed. The open loop gain, bandwidth and slew rate is also 
a function of the capacitive loading of the output of amplifier 20 (i.e., 
the capacitance presented to the collector of transistor Q3). This is 
reduced, as explained in detail later, by operating the push-pull 
transistors of the complementary emitter follower amplifier 30 at constant 
values of collector to emitter voltage. The closed loop gain, assuming 
that the open loop gain is adequate, is directly proportional to the value 
of the feedback resistor R2 and inversely proportional to the impedance of 
the input network R1, R3 and C1. 
Considering now the details of the push-pull complementary emitter follower 
amplifier 30, this amplifier includes a pair of complementary transistors 
Q4 and Q7 coupled at the base electrodes thereof to the output (collector 
of Q3) of amplifier 20 and coupled at the emitters thereof to an output 
terminal 15 via respective emitter resistors R9 and R12. The output 15 of 
follower 30 is coupled, as previously noted, to the cathode 16 via a 
kinescope arc suppression network comprising the series connection of 
inductor L1 and resistor R15. Supply voltage (collector potentials) for 
the follower transistors Q4 and Q7 are provided by respective feedback 
circuits 50 and 60. 
Circuit 50 provides the function of regulating the collector to emitter 
voltage of follower transistor Q4 at a fixed value. To this end the 
circuit 50 includes a voltage regulator transistor Q6 connected at the 
collector thereof to supply 20 and at the emitter thereof to the collector 
of transistor Q4. The input (base) of the voltage regulator transistor Q6 
is coupled to the emitter electrode of the follower transistor Q4 via a 
capacitor C3 in parallel with a threshold conduction device (i.e., a Zener 
diode) CR3. This positive feedback path establishes a substantially 
constant collector to emitter offset voltage for follower transistor Q4 
equal to the Zener voltage. To provide an operating current for the Zener 
diode, the cathode thereof is coupled to the high voltage source 20 via a 
resistor R11. To minimize loading of the emitter circuit of transistor Q4, 
the emitter is coupled to the capacitor C3 and Zener diode CR3 via an 
emitter follower transistor Q5. Specifically, transistor Q5 is a PNP 
transistor coupled at the base thereof to the emitter of the follower 
transistor Q4 via a resistor R10. The collector-emitter path of follower 
transistor Q5 is coupled between the junction of capacitor C3 and Zener 
diode CR3 and ground. In certain applications transistor Q5 may be omitted 
as will be shown and described in a later example of the invention. 
Circuit 60 is similar to circuit 50 and provides the function of regulating 
the collector to emitter voltage of follower transistor Q7 at a fixed 
value. To this end the circuit 60 includes a voltage regulator transistor 
Q9 connected at the collector thereof to a supply input of the sense 
amplifier 40 and at the emitter thereof to the collector of transistor Q7. 
The input of the voltage regulator transistor Q9 is coupled to the emitter 
electrode of the follower transistor Q7 via a capacitor C4 in parallel 
with a threshold conduction device (i.e., a Zener diode) CR4. This 
feedback path regulates the collector emitter voltage of the follower 
transistor Q7 at the Zener voltage. To provide an operating current for 
the Zener diode, the anode thereof is coupled to ground via a resistor 
R14. To minimize loading of the emitter circuit of transistor Q7, the 
emitter is coupled to the capacitor C4 and Zener diode CR4 via an emitter 
follower transistor Q8. Specifically, transistor Q8 is a NPN transistor 
coupled at the base thereof to the emitter of the follower transistor Q7 
via a resistor R13. The collector-emitter path of transistor Q8 is coupled 
between the junction of capacitor C4 and Zener diode CR4 and the high 
voltage supply 20. 
The sense amplifier 40 is provided for use in video display systems of the 
type featuring automatic kinescope bias (AKB) circuitry and thus requires 
sensing of the kinescope cathode current "Ik". Amplifier 40 comprises a 
cathode current sensing transistor Q10 connected at the emitter thereof to 
the collector of the voltage regulator transistor Q9. A reference 
potential for the base of transistor Q10 is provided by the Zener diode 
CR1. Capacitor C5, in parallel with diode CR1 provides filtering of the 
regulated Zener voltage. An output voltage, proportional to the cathode 
current Ik is developed at output terminal 18 across the load resistor R16 
coupled between the collector of transistor Q10 and ground. In 
applications not requiring AKB operation the sense amplifier may be 
omitted. If so, as shown in a later example, the collector of voltage 
regulator transistor Q9 should be coupled to ground or another suitable 
low voltage reference potential. 
To summarize the operation described above, the cascode amplifier 20 
amplifies the video signal provided by source 10 as previously described. 
To minimize the capacitive loading on load resistor R8 that is 
attributable to the capacitance associated with the kinescope 16, its 
socket and spark arrestors (not shown) and other stray capacitances, the 
output (collector of transistor Q3) of the cascode amplifier 20 is coupled 
to the kinescope cathode electrode via a push-pull complementary emitter 
follower amplifier 30. This particular follower amplifier is of the 
"parallel" type in which the base electrodes are in parallel for receiving 
the amplified video signal and the emitters are in parallel for driving 
the cathode. 
The inclusion of the follower amplifier 30, as recognized herein, does 
provide a reduction in cathode capacitance presented to the amplifier 20 
but introduces a secondary capacitance effect. Namely, the collector to 
base capacitances of follower transistors Q4 and Q7. To effectively reduce 
the values of these unwanted capacitances, the reactive charging and 
discharging currents supplied to these capacitances are reduced. This 
feature is provided by the two positive feedback regulators 50 and 60 
which maintain the collector to emitter voltages for the follower 
transistors at constant values. 
As an example, if the output voltage of amplifier 20 increases, then the 
emitter voltage of the follower transistor Q4 will increase but the Zener 
diode CR3 and the regulator transistor Q6 will increase the collector 
voltage of the follower transistor Q4. Similarly, for a decreasing output 
voltage of amplifier 20, the emitter voltage of follower transistor Q4 
will decrease and the Zener diode CR3 and the regulator transistor Q6 will 
cause a decrease in the collector voltage of the follower transistor Q4. 
Illustratively, for a Zener voltage of 10 Volts, the collector emitter 
voltage of transistor Q4 will equal the Zener voltage minus the 
base-emitter junction voltages (Vbe) of transistors Q5 and Q6. For the 
assumed Zener voltage of 10 volts, the resultant collector-emitter voltage 
of transistor Q4 will thus be about equal to 8.8 Volts (assuming a Vbe 
value of 0.6 volts). 
Thus, whether the follower input voltage is increasing or decreasing, the 
voltage across the follower transistor from the collector to the emitter 
is constant. As the input signal goes through points of inflection, the 
base voltage will vary by a few hundred millivolts relative to the emitter 
as the follower transistor is biased on and off (push-pull operation). 
However, it has been found that the base emitter voltage variations are 
relatively minor as compared with the regulated collector emitter voltage 
(e.g., a Zener voltage of 10 volts or so). As a result one may consider 
that the collector to base voltage variations are "substantially" constant 
and so there can be little charging and discharging of the collector to 
base capacitance under dynamic signal conditions. Since such reactive 
currents are suppressed, in accordance with the invention, the effective 
collector to base capacitances are reduced for the follower amplifier. 
As described above, the feedback for regulating the collector emitter 
voltages for the follower transistors is nearly one hundred percent. It 
can never exactly equal one hundred percent because the gains of 
transistors Q5 and Q6, for example, can not equal unity since that would 
require infinite current gains. In other words, transistors Q5 and Q6 are 
both connected as emitter followers and the gain of an emitter follower 
may be very close to unity but never equal to unity. Accordingly, even 
though the feedback is positive, the circuit is stable. Lesser amounts of 
feedback, e.g., 50% may be used if desired in a given application. It will 
be noted that the actual Zener voltage is not a critical parameter of the 
circuit. The Zener by-pass capacitor (C3 or C4) provides a desirable 
reduction in AC impedance of the voltage regulator to further facilitate 
wideband operation. 
The example of FIG. 1 may be modified as shown in FIG. 2. In this example 
feedback control of the gain of the cascode amplifier has been replaced by 
feedforward control and the AKB sense amplifier 40 has been deleted. 
Additionally, the voltage regulators 50 and 60 have been simplified. 
In more detail, in the high voltage cascode amplifier 20 of FIG. 2 the 
feedback resistor R2 has been removed as well as the input peaking 
components resistor R1 and capacitor C1. The gain, as thus modified, is 
determined by the load resistor R8 and the emitter impedance of input 
transistor Q2 (i.e., emitter resistor R6 and the peaking network 
comprising capacitor C2 and resistor R7. Aside from these modifications, 
operation is otherwise the same as in the example of FIG. 1. 
Omission of the AKB sense amplifier 40, as previously explained, requires a 
source of relatively low potential for the collector of the positive 
feedback voltage regulator transistor Q9. The collector could be connected 
any suitable potential near ground. Here it is connected directly to 
ground. 
Simplification of the positive feedback voltage regulator circuits 50A and 
50B comprises removing transistors Q5 and Q8 and removing resistors R10 
and R13. In the previous examples, these elements providing coupling of 
the emitters of the follower transistors to the respective threshold 
conduction devices and capacitors. In this example, the emitter of 
follower transistor Q4 is coupled to capacitor C3 and Zener diode CR3 by 
connecting these elements directly to the output terminal 15. The same is 
done for capacitor C4 and Zener diode CR4. 
In operation, resistor R11 supplies current from the high voltage supply 20 
through Zener diode CR3 to the output terminal 15. This establishes a 
regulated voltage at the base of regulator transistor Q6 that equals the 
emitter voltage of transistor Q4 less the drop across resistor R9 plus the 
Zener voltage of diode CR3. Resistor R9 is provided primarily to provide 
protection against simultaneous conduction of transistors Q4 and Q7 and so 
may be of a relatively small value (e.g., 30 Ohms or so). Accordingly, the 
voltage drop across resistor R9 is negligible and the transistor Q4 
operates at a substantially constant collector emitter voltage. Operation 
of the modified feedback regulator 60A is the same as for 50A, except for 
the transistor polarities and directions of current flow. 
FIG. 3 illustrates a modification of the example of FIG. 1 in which the 
"parallel" form of complementary push-pull emitter follower 30 is replaced 
by a "series" form of complementary push-pull emitter follower 30B. The 
modified follower comprises an NPN transistor Q302 having the base-emitter 
path thereof coupled in series with that of a PNP transistor Q306 between 
an input terminal 301 and an output terminal 308. Respective diodes CR300 
and CR304 are coupled across the base-emitter junctions of the transistors 
Q302 and Q306 and poled opposite to the poling of the associated junction. 
Accordingly, diode CR300 is conductive when transistor Q302 is biased off 
and vice versa. Similarly, diode CR304 is rendered conductive which 
transistor Q306 is biased off. 
The collector to emitter voltage of transistor Q302 is regulated at about 
the value of the Zener voltage of diode CR3 by connecting resistor R10 to 
the emitter of follower transistor Q302 to sense the emitter voltage and 
connecting the emitter of the voltage regulator transistor Q6 to the 
collector of follower transistor Q302. This provides positive feedback for 
regulating the collector voltage of transistor Q302 at a value offset from 
the emitter voltage and proportional to the Zener voltage of diode CR3. 
Similarly, the collector to emitter voltage of transistor Q306 is regulated 
at about the value of the Zener voltage of diode CR4 by connecting 
resistor R13 to the emitter of follower transistor Q306 to sense the 
emitter voltage and connecting the emitter of the voltage regulator 
transistor Q9 to the collector of follower transistor Q306. This provides 
positive feedback for regulating the collector voltage of transistor Q302 
at a value offset from the emitter voltage and proportional to the Zener 
voltage of diode CR4. 
Since diode CR2 is not needed in the modified circuit, the load resistor R8 
for the cascode amplifier 20 is connected directly to the collector of the 
cascode output transistor Q3 and this point is connected directly to the 
input 301 of follower 30B. In operation, an increasing video signal 
voltage at input 301 will forward bias transistor Q302 to supply drive 
current via diode CR304 to the kinescope cathode 16 and regulator 50 will 
maintain the collector emitter voltage of transistor Q302 constant. A 
decreasing video signal voltage at the input 301 will forward bias 
transistor Q306 to withdraw drive current via diode CR300 from the 
kinescope cathode and regulator 60 will maintain the collector emitter 
voltage of transistor Q306 at a substantially constant value. For purposes 
of AKB sensing the collector current of the regulator transistor Q9 is 
applied to the sense amplifier circuit 40 the operation of which is as 
previously described. 
FIG. 4 illustrates a modification of the example of FIG. 1 in which the 
"parallel" form of complementary push-pull emitter follower 30 is replaced 
by a "series" form of complementary push-pull emitter follower 30C. The 
modified follower comprises an NPN transistor Q400 having the base-emitter 
path thereof coupled in series with that of a PNP transistor Q402 between 
an input terminal 401 and an output terminal 409. Respective diodes CR404 
and CR406 are coupled across the base-emitter junctions of the transistors 
Q400 and Q402 and poled opposite to the poling of the associated junction. 
Accordingly, diode CR404 is conductive when transistor Q400 is biased off 
and vice versa. Similarly, diode CR406 is rendered conductive which 
transistor Q402 is biased off. 
The collector to emitter voltage of transistor Q400 is regulated at about 
the value of the Zener voltage of diode CR3 by connecting the emitter of 
transistor Q6 to the collector of transistor Q400 and coupling the emitter 
of transistor Q6 to the output 409 via capacitor C3 and Zener diode CR3. 
This provides positive feedback for regulating the collector voltage of 
transistor Q400 at a value offset from the emitter voltage and 
proportional to the Zener voltage of diode CR3. 
Similarly, the collector to emitter voltage of transistor Q402 is regulated 
at about the value of the Zener voltage of diode CR4 by connecting the 
emitter of regulator transistor Q9 to the collector of transistor Q402 and 
coupling the base of transistor Q9 to the output terminal 409 via 
capacitor C4 and Zener diode CR4. This provides positive feedback for 
regulating the collector voltage of transistor Q302 at a value offset from 
the emitter voltage and proportional to the Zener voltage of diode CR4. 
Since diode CR2 is not needed in the modified circuit of FIG. 4, the load 
resistor R8 for the cascode amplifier 20 is connected directly to the 
collector of the cascode output transistor Q3 and this point is connected 
directly to the input 401 of follower 30C. In operation, an increasing 
video signal voltage at input 401 will forward bias transistor Q400 to 
supply drive current via diode CR406 to the kinescope cathode 16 and 
regulator 50 will maintain the collector emitter voltage of transistor 
Q400 constant. A decreasing video signal voltage at the input 401 will 
forward bias transistor Q402 to withdraw drive current via diode CR404 
from the kinescope cathode and regulator 60 will maintain the collector 
emitter voltage of transistor Q402 at a substantially constant value. 
Since AKB sensing is not required, the collector of the voltage regulator 
transistor Q9 is coupled of a source of relatively low voltage (ground). 
FIG. 5 illustrates a modification of the example of FIG. 2 for providing 
single-ended voltage follower operation. The term "voltage" follower, as 
used herein refers to emitter followers (which employ bipolar transistors) 
and to source followers (which employ field effect transistors). In this 
example of the invention the voltage follower operates in a Class A mode 
in which the follower transistor is conductive all the time. This 
eliminates crossover distortion which may occur in complementary followers 
in which the transistors operate in a Class B mode with limited 
conduction. On the other hand, class-B or push-pull operation is preferred 
from a power dissipation standpoint as the efficiency is much higher than 
with single ended followers. 
As a brief overview, in this example of the invention a video amplifier 
(20A) is coupled to a kinescope cathode electrode 16 via a voltage 
follower 500. The voltage follower comprises a transistor having a 
conduction path and a control electrode for controlling the conduction of 
the path. In this case the voltage follower is an emitter follower and the 
follower transistor is a bipolar transistor Q502. The control electrode 
(e.g., the base of transistor Q502) is coupled to receive a video signal 
from the video amplifier. A first end of said conduction path (e.g., the 
emitter of Q502) is coupled to a point of reference potential (here, 
ground) via a current source 504 and is coupled to the kinescope cathode 
16. The second end (i.e., the emitter of Q502) of the conduction path 
being coupled to a source of supply voltage (20). A feedback circuit 50A 
is coupled to the first end of the conduction path for applying a positive 
feedback voltage to the second end of the conduction path of the voltage 
follower transistor (Q502) for maintaining a substantially constant 
voltage across the conduction path that is independent of variations in 
the video signal applied to said control electrode. 
In more detail, in FIG. 5 the output of cascode amplifier 20A is coupled to 
the kinescope cathode 16 via a single ended emitter follower amplifier 500 
comprising an emitter follower transistor Q502 connected at the base 
electrode thereof to the output (collector) of transistor Q3 in the 
cascode amplifier 20A. In this case the collector load resistor R8 is 
connected directly to the collector of the output transistor Q3. The 
emitter of transistor Q502 is coupled to an output terminal 506 which is 
coupled to ground via a current source 504 that provides a constant 
current drive to the output terminal 506. The cathode electrode 16 is 
coupled to the output terminal 506 via the kinescope arc arresting network 
comprising the series connection of resistor R15 and inductor L1. 
For regulating the collector to emitter voltage of the emitter follower 
transistor at a constant value, the emitter is connected to capacitor C3 
and Zener diode CR3 of the positive feedback voltage regulator circuit 
50A. The output of this regulator is the emitter of regulator transistor 
Q6 which is coupled to the collector electrode of the emitter follower 
transistor Q502. 
Operation of the emitter follower is similar to operation of the 
corresponding transistors previously described except with regard to the 
efficiency and cross over effects discussed above and the method of 
providing a pull down current. Specifically, for decreasing values of the 
video signal the reduction of the cathode voltage is provided by the 
current source 504. While this source may comprise a passive element such 
as a resistor, an active device may be preferable in certain applications, 
e.g., where a faster negative going slew rate is desired at low output 
voltage levels. A constant current source, such as a suitable biased 
bipolar or field effect transistor is suitable for this purpose. 
To reiterate the overall operation, when the amplified video signal 
provided by amplifier 20A is increasing in voltage the emitter voltage of 
transistor 502 will increase also thus boosting the base potential of the 
feedback regulator transistor Q6 and so maintaining the collector to 
emitter voltage of the emitter follower transistor constant. Since this 
voltage does not change significantly, there is no charging of the 
collector to base capacitance of transistor Q502 and so the effective 
capacitance presented to the output of amplifier 20A is reduced over that 
of a conventional emitter follower amplifier. Conversely as the base 
voltage falls, so does the emitter voltage and transistor Q6, being offset 
from the emitter voltage by the Zener voltage of diode CR3, decreases the 
collector voltage of the follower transistor Q502 so as to maintain a 
constant collector to emitter voltage. In this latter case there is no 
active pull-down of the video output voltage but this function is provided 
by the current source 504. 
FIG. 6 illustrates a modification of the example of FIG. 1 with regard to 
the manner of providing base drive current for transistor Q8. 
Specifically, in FIG. 1 the base of transistor Q8 was connected to the 
emitter of the follower transistor Q7 via a resistor whereas in FIG. 6 the 
base of transistor Q8 is coupled via a diode CR600 and a capacitor C6 to 
the emitter of transistor Q8 and is coupled via a resistor R600 to the 
emitter of transistor Q5. 
The purpose of the foregoing changes is to reduce potential cathode current 
(Ik) sensing errors by DC biasing the base of transistor Q8 from the 
emitter of transistor Q5. This eliminates the DC base current demand for 
transistor Q8 from the emitter of transistor Q7 which conducts the cathode 
current Ik. The added capacitor C6 provides AC coupling of the emitter of 
transistor Q7 to the base of transistor Q8 and so the high frequency 
operation is the same as in the previous example. The added diode CR600 
provides a correction for video signal conditions involving high 
frequencies and high duty cycles. Specifically, this diode provides a DC 
path around the AC coupling capacitor for high duty cycle, high frequency 
signal conditions to prevent a reduction in base bias for transistor Q8 
under high duty cycle, high frequencie, signal conditions. Briefly, diode 
CR600 prevents capacitor C6 from developing a significant average charge 
that would otherwise tend to reduce the base bias of transistor Q8 for 
video signals Of high frequency and high duty cycle. 
In more detail, it has been discovered that trader certain conditions in 
the example of FIG. 1 the current demand due to the base current of 
transistor Q8 may introduce an undesired error in measurement of the 
cathode current Ik by the sense amplifier 40. At the point where Ik is 
measured for AKB purposes, the cathode is near cutoff (a high voltage 
level) and thus the current through the base circuit of transistor Q8 is 
relatively high and so may cause a significant error in the cathode 
current Ik measurement. The modifications described above ensure that for 
DC and low frequencies the base current of transistor Q8 comes from the 
emitter of transistor Q5 thus reducing the Ik measurement error. However, 
for optimum high frequency response during active video intervals it is 
not desirable just to drive the base of transistor Q8 just from the 
emitter of transistor Q5. For such a case more drive is required, that is, 
for active video signals (i.e., displayed video as compared with video 
measurement levels in AKB operation) transistor Q8 should receive the high 
current push-pull and reduced phase shift benefits that AC coupling from 
the emitter of transistor Q7 provides. The function of the added diode, 
CR600, in the base drive circuit for transistor Q8 is to provide for those 
occassions when the driver is subjected to high duty cycle, large 
amplitude high frequency signals which would otherwise result in 
transistor Q8 shifting is bias point. For such transient conditions, the 
added diode CR600 provides a by-pass around the AC coupling capacitor C6. 
It will be apparent that various other changes may be made to the examples 
of the invention herein shown and described. For example the cascode 
amplifier 20 may be provided with an active collector load rather than the 
passive (resistor) load shown. A suitable active load would be a 
transistor biased for operation as a current source. Another modification 
to the cascode amplifier load impedance would be to couple an inductor in 
series with resistor R8. Another alternative would be to couple a small 
capacitance from the output of the push-pull amplifier to a "center tap" 
on load resistor R8 to optimize the overall performance. To facilite the 
center tap, resistor R8 may be fromed from two smaller valued resistor 
connected in series with the common connection there used for the tap 
point.