Method of forming the radiation pattern of a high efficiency active antenna for an electronically-scanned radar, and an antenna implementing the method

A method of forming the radiation pattern of a high efficiency active antenna for electronically-scanned radar, wherein its illumination laws, and thus its radiation patterns are dissociated in transmission and in reception; and equal amplitude illumination is provided in transmission in order to maximize the efficiency of the transmit amplifiers which are all identical, thereby minimizing their DC energy consumption and their dissipation. The invention is particularly suitable for use in space radars.

BACKGROUND OF THE INVENTION 
The invention relates to a method of forming the radiation pattern of a 
high efficiency active antenna for electronically-scanned radar, and to an 
antenna implementing the method. 
Electronic scanning greatly increases the performance of radars, by virtue 
of its flexibility (number of possible operating modes) and its speed 
(quasi instantaneous beam shifting). 
However the main drawback thereof is the very high number of phase control 
circuits required and often the very high number, of amplitude control 
circuits also required, thereby giving rise to losses, expense, mass, and 
power consumption which are often prohibitive. 
Drawbacks relating to losses, expense, mass, and bulk have been overcome by 
mass-producing monolithic microwave integrated circuits (MMIC) on gallium 
arsenide. It is thus possible to make active transmit-receive (TR) modules 
very compactly which integrate the functions of phase-shifting, switching, 
transmission and reception, and amplification. 
However, transmission power amplifiers made in MMIC technology have 
relatively low efficiency, and in addition their efficiency falls off if 
the output power level is varied. 
In conventional designs, such variation in power level is essential both: 
as a function of position to form a radiation pattern having low side lobe 
level (SLL); and 
as a function of time in order to modulate lobe width to adapt it to the 
mission. 
As a result, the power consumption of this type of active radar antenna is 
prohibitive. 
There exist several documents in the state of the art and in particular: 
parts I and II of the article entitled "Array radars: an update", by Eli 
Brookner published in Microwave Journal (Feb. and Mar. 1987); 
the article entitled "Applicability, availability, and affordability of 
GaAs MMICs in military systems" by Eugene H. Gregory, published in 
Microwave Journal (March 1987); and 
the article entitled "Affordable MMIC designs for phased arrays" by Ronald 
J. Naster, Anthony W. Jacomb-Hood, and Mark R. Lang, published in 
Microwave Journal (March 1987). 
The first prior art electronically-scanned radars used diode or ferrite 
phase shifters for controlling beam depointing: 
the major drawback of diode phase shifters is significant losses (several 
dB for 4/5-bit phase shifters), thereby requiring the already-critical 
power of the amplifiers to be increased; and 
although ferrite phase shifters have losses of less than 1 dB, they suffer 
from significant mass and bulk. These parameters become critical with 
airborne radars, and they prevent such radars being mounted on satellites. 
An important advance was made when monolithic microwave integrated circuits 
(MMICs) on gallium arsenide started to be mass produced. This technology 
makes it possible to manufacture various types of microwave circuit having 
very low mass and bulk, at relatively low cost, and in mass production 
quantities, and in particular it can be used for manufacturing 
controllable attenuators and phase shifters. 
The major drawback of MMIC phase shifters relates to significant losses 
(more than 5 dB for a 0.degree.-360.degree. C. phase shifter having analog 
control or 4/5-bit digital control). However this aspect is secondary when 
these phase shifters are associated with amplifiers: 
either high power amplifiers (HPAs) situated downstream from transmit phase 
shifters, since the losses then take place at low level and have no effect 
on limiting the output power of the amplifiers, it merely being necessary 
to increase the gain of the output amplifiers a little; 
or low noise amplifiers (LNAs) situated upstream from the receive phase 
shifters, since providing the gain of these amplifiers is adequate (20 dB 
to 30 dB), then the losses inherent to the phase shifters have 
substantially no deleterious effect on the noise factor of the receiver. 
Transmit-receive (TR) modules are generally manufactured on a common 
(alumina) substrate by connecting together a plurality of gallium arsenide 
chips each performing an elementary function. These chips are themselves 
mass produced using doping (diffusion or ion implanting), masking, 
oxidizing, . . ., techniques based on those used for making logic 
integrated circuits on silicon. Silicon ICs have shown their capacity for 
reducing cost enormously, without loosing reliability. 
By connecting together several hundreds or thousands of such MMIC-TR 
modules in an active radar antenna (called "active" because it includes 
active devices in the form of amplifiers), it is possible to reconcile the 
requirements of electronic scanning with cost, mass, and bulk, which are 
critical parameters for airborne radars and even more important for space 
radars. 
The final critical parameter for such active radar antennas is their DC 
power consumption. 
The added power efficiency of HPA amplifiers: 
EQU .eta.a=(Pout-Pin)/PDC 
is much lower in MMIC technology (by 15% to 20%) than in travelling wave 
vacuum tube technology which lies in the range 30% to 60% depending on the 
microwave waveband. 
Efficiency is particularly poor when using class A (linear) HPAs while 
varying the input power and thus the output power: power consumption is 
determined by the bias currents and voltages which are set for the maximum 
Pout to be delivered. The same amount of power is consumed when Pin is 
reduced to reduce Pout. 
An alternative consists in reducing bias voltages when a lower Pout is 
required. Power consumption is thus reduced, but considerably less than 
power output (in percentage or in dB). Efficiency .eta.a is thus 
significantly reduced. 
However if high performance radiation patterns are to be formed, it is 
necessary: 
at least to have one different Pout per TR module, so as to obtain the 
weighting required for illumination taper; and 
in some cases where the mission requires a lobe of variable width, it is 
also necessary to vary the illumination taper amplitude law as a function 
of time, thus requiring the Pout of the HPAs to be varied. 
As a result, active radar antennas have hitherto confronted the following 
dilemma: a radiation pattern having low side lobe levels, and preferably 
also being capable of being modulated, can only be obtained by reducing 
the efficiency of the distributed HPAs. 
The resulting increase in power consumption has so far restricted the 
generalization of active antennas for airborne radar applications, and 
even more for space radar applications where the available power is very 
limited. 
An object of the present invention is to escape from this performance/power 
consumption dilemma by providing a method of forming the radiation pattern 
of a radar antenna which is particularly well suited to active antennas 
(i.e. antennas having distributed modules including transmit and receive 
amplifiers). 
SUMMARY OF THE INVENTION 
To this end, the present invention provides a method of forming the 
radiation pattern of a radar antenna optimized for active antennas having 
amplifiers distributed over the antenna immediately behind the radiating 
elements, which method makes it possible: 
to dissociate the illumination laws and thus the radiation patterns in 
transmission and in reception; 
to provide equal amplitude illumination in transmission to maximize the 
efficiency of transmit amplifiers which are all identical and to minimize 
their DC power consumption and dissipation, both of which are critical 
points for active antennas (operation in class B or in class AB); 
to optimize the illumination law in reception by adjusting the gain in the 
receive path (using adjustable attenuators or variable gain last stages in 
the LNAs); 
to obtain an identical phase law in transmission or in reception, thereby 
enabling reciprocal phase shifters to be used while being controlled at a 
moderate rate; 
to synthesize radiation patterns of variable width by applying an 
appropriate phase law while changing neither the power of the transmit 
amplifiers nor the gain of the receive paths; and 
to control receive path gain during transmission so as to form radiation 
patterns of greater width and having sharper cutoffs, thereby improving 
the discriminating power of the radar. 
Thus, the gain on the receive path of each TR type module is controlled so 
as to form a receive radiation pattern which is adapted to the transmit 
pattern, i.e. which has sensitivity lows where the transmit pattern has 
interferring side lobes: 
the performance of a radar depends on the product Ge.times.Gr (transmit 
gain.times.receive gain) in a given direction specified but its angles 
(.theta., .phi.) in spherical coordinates; 
reciprocal passive antennas have the same radiation pattern in transmission 
and in reception, i.e.: 
EQU Ge(.theta., .phi.)=Gr(.theta., .phi.); whereas 
an active antenna of the invention having different radiation patterns 
provides the same performance as a reciprocal antenna having a radiation 
pattern Ger(.theta., .phi.) given by: 
EQU Ger(.theta., .phi.)=.sqroot.Ge(.theta., .phi.).times.Gr(.theta., .phi.) 
Ger is called the equivalent transmit-receive pattern. 
When compared with the pattern obtained by the conventional method where Ge 
is equal to Gr, it can be seen that better performance is obtained. 
The invention thus relates to a method of synthesizing radiation patterns 
of variable width and having low side lobe levels by varying the gain of 
its TR modules in reception only, while operating in transmission at a 
level which is constant over the area of the antenna and over time. 
By adapting the receive pattern to the transmit pattern, very good 
transmit/receive performance is obtained, and the invention provides the 
fundamental advantage of retaining acceptable efficiency in the 
distributed transmit amplifiers. Power consumption becomes significantly 
lower both in airborne radars and in space radars. 
The fundamental advantage of the invention is that gain adjustment in the 
TR modules takes place only on reception, thereby having no effect on the 
power consumption of the antenna. The additional power dissipated on the 
receive paths is negligible compared with the power dissipated by the HPAs 
since the level of the received signals (echoes of the transmitted pulses) 
is at least 100 dB below the level of the transmitted signal. 
By using active modules that are all identical (e.g. in MMIC technology), 
costs are reduced by the mass production effect since the modules can be 
dimensioned for a power level that is lower than the maximum level that 
would be required by amplifiers of different gains or of variable gain. By 
causing all of the HPAs to operate at the same output power, their 
efficiency is optimized, thereby reducing power consumption, and this is a 
critical point for active antennas. 
Beam width can be varied by controlling the phase shifters only, and by 
controlling them in the same manner both for transmission and for 
reception (thereby limiting the reconfiguration rate). 
This thus has the additional advantage of limiting the number of control 
signals. 
Advantageously, an antenna may be provided having active modules which are 
distributed non-uniformly in one of its dimensions, thereby making it 
possible to reduce the number of modules considerably. When the electronic 
scanning takes place in a single plane only, the method of the invention 
is applied to said single plane, in which case amplitude weighting in the 
other plane is identical both in transmission and in reception in the 
other plane, generated by the physical distribution of the active modules. 
Overall, an active antenna is provided in which the transmit amplifiers 
all operate at the same level, and in which the illumination law and the 
radiation patterns in the two main planes of the antenna (azimuth, 
elevation) are "separable".

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
The structure of the active transmit/receive (TR) modules is based on the 
conventional TR module structure shown in FIG. 1 which comprises a 
transmit input E connected to an active element of the radar and a 
transmit output S connected to a radiating element of the antenna, a 
digital phase shifter 10 and a head-to-tail parallel connection between 
first and second switches 11 and 12 of a high power amplifier (HPA) 13 for 
transmission and a low noise amplifier (LNA) 14 for reception. Gain 
control is provided in the receive path of this structure. This gain 
control can be provided in two different ways: 
either by adding a variable attenuator 15 after the LNA as shown in FIG. 2; 
or else by controlling the gain of the last stage of the LNA by means of a 
two-gate field effect transistor (FET) 16 having one gate serving 
conventionally as an input port and having its other gate modulating its 
gain when the bias voltage applied thereto is varied (see FIG. 3). 
All of the HPAs operate continuously at output power. This can thus be 
maximized and efficiency can be optimized by making them operate in their 
compression region or in their saturation region. It is thus possible to 
use class B or class AB amplifiers whose DC power consumption and whose 
dissipation are considerably lower (for constant Pout) than in amplifiers 
used in their linear region (class A). Class A amplifiers would be 
essential if any attempt were to be made to vary transmit power. 
The output switch 12 is a double-pole double-throw (DPDT) output switch 
which simultaneously provides transmit-receive switching and two ports 
leading to the radiating elements (two polarizations: horizontal 
polarization H and vertical polarization V). It is well adapted to being 
implemented as an integrated circuit (MMIC technology) being very small in 
size and in mass. 
However, in the prior art such a configuration presents technical problems 
for very high power HPAs. Under such circumstances a solution is adopted 
(not shown in the figures) which includes a circulator followed by a 
single-pole double-throw output power switch. The LNA then needs to be 
protected by a limiter. 
Although the method of the invention is applicable to two dimensional 
synthesis where the amplitudes Amn and the phases .phi.mn of all of the 
radiating elements are independent, the results described relate to a 
simplified case for a rectangular antenna having separable illumination. 
In fact, this is the technique which is applied in most radars, and in 
particular in airborne or space radars operating as synthetic aperture 
radars (SAR): 
the radiation pattern in the main elevation plane (the vertical plane 
including the normal to the antenna) is synthesized independently from the 
radiation pattern in the main azimuth plane (the horizontal plane 
including the normal to the antenna); 
the elevation pattern G.sup.el (v) with v=(H/.lambda.) sin El and where El 
is the angle from the normal in the elevation plane, is related to the 
illumination law E.sup.el (y) on the vertical axis of an antenna of height 
H operating at a frequency where its wavelength in vacuo is .lambda.as 
follows: 
EQU G.sup.el (v)=.vertline.F.sup.el (v).vertline..sup.2, where 
F.sup.el (v) is the complex Fourier transform of E.sup.el (y), with the 
modulus of El representing the source excitation amplitude at Y-coordinate 
y, and the argument of El representing the phase of said excitation; 
similarly, the azimuth radiation pattern G.sup.az (u), where u=(L/.lambda.) 
sin Az, Az being the angle from the normal in the azimuth plane, is 
related to the illumination law E.sup.az (x) on the horizontal axis of an 
antenna of length L, by: 
EQU G.sup.az (v)=.vertline.F.sup.az (v).vertline..sup.2, where 
F.sup.az (v) is the complex Fourier transform of E.sup.az (x), representing 
the amplitude and phase of the excitation for the source of X-coordinate 
x. 
All amplitudes are normalized to a maximum of 1 (or 0 dB) and all phases 
are taken to within a constant. 
The source (or radiating element) excitation at position (x, y) is thus 
E.sup.az (x) x E.sup.el (y), and the radiation pattern in the direction 
.theta., .phi. is then: 
EQU G(.theta., .phi.)=.vertline.F(.theta., .phi.).vertline..sup.2, where 
F(.theta., .phi.)=F.sup.az (u).times.F.sup.el (v) u=(L/.lambda.) sin 
.theta..multidot.cos .phi. 
G(.theta.,.phi.)=G.sup.az (u).times.G.sup.az (v) v=(L/.lambda.) sin 
.theta..multidot.cos .phi. 
where .theta. and .phi. are the conventional Euler spherical coordinate 
angles, .theta. being taken relative to the normal Os to the antenna and 
.phi. being taken relative to the horizontal axis Ox, as shown in FIG. 4. 
Thus, in each of the main planes xOz (azimuth) and yOz (elevation), this 
reduces to synthesizing a one-dimensional radiation pattern generated by 
an alignment of sources (a linear array). 
For a radar antenna operating in transmission, and then in reception, the 
radiation pattern is applied twice over: 
on transmission, it concentrates the transmitted energy in the desired 
direction while simultaneously avoiding exceeding a fixed side lobe level 
in all other directions (e.g. SLL=-20 dB); and on reception, a signal 
coming from a direction other than the aiming direction is again weighted 
by the radiation pattern. Overall, the discriminating power of the radar 
depends on the product Ge(.theta., .phi.).times.Gr(.theta., 
.phi.),=transmission pattern.times.reception pattern. In dB, the effects 
of the two patterns are summed. If the side lobe level is -20 dB on 
transmission and -20 dB on reception, then attenuation in directions other 
than the aiming direction is 40 dB. 
However in order to be able to compare different patterns Ge and Gr with a 
conventional pattern where Ge is equal to Gr (conventional antennas 
without amplifiers are reciprocal and their transmit and receive patterns 
are identical), an equivalent "transmit-receive" pattern is drawn: 
EQU Ger(.theta., .phi.)=.sqroot.Ge(.theta., .phi.).multidot.Gr(.theta., .phi.) 
i.e. the single equivalent pattern which, if used both on transmission and 
on reception, would give rise to the same overall result in a radar 
application (i.e. the same product Ge.Gr). 
Consider, by way of example, synthesizing a narrow lobe using 128 sources, 
with the distance between sources being 0.57.lambda., at each elementary 
source being omnidirectional. 
FIG. 5A shows the transmit pattern obtained using uniform illumination as 
shown in FIG. 5B (using 128 patch lines) i.e. equal amplitude illumination 
(all of the HPAs are operating at the same level) and equal phase 
illumination (in order to obtain a narrow lobe). 
FIG. 6A shows the receive pattern: the illumination shown in FIG. 6B is 
still equiphase but adapted amplitude weighting serves to lower the first 
side lobes relative to the following side lobes. 
FIG. 7 shows the equivalent transmit-receive pattern, whose performance is 
excellent: 
the secondary side lobe level is close to -20 dB which was the target level 
in this particular case; 
the base of the main lobe is narrower than for the patterns obtained by 
synthesis in one direction only; this is a fundamental quality, 
particularly for SAR radiation patterns which are required to limit 
ambiguity as much as possible, i.e. to attenuate close echoes as much as 
possible both in elevation and in azimuth; 
directivity performance is specified by "illumination efficiency" 
.andgate.i which represents the loss of directivity due to the amplitude 
and phase law compared with a uniform law (where a uniform law gives a 
directivity of (2L/.lambda.) for an alignment of length L). 
The loss in dB on the equivalent transmit-receive pattern is 
.eta.i=1/2(.eta..sub.i.sup.e +.eta..sub.i.sup.r), where .eta..sub.i.sup.e 
and .eta..sub.i.sup.r are the illumination efficiencies of the transmit 
and receive laws, respectively. 
The resulting efficiency .eta.i (-0.16 dB) is as good as that which can be 
obtained for the best (so-called "Taylor") laws which provide the same 
SLL.apprxeq.-20 dB) in one direction only. 
Curves 80 and 81 shown in the A portions of the figures are respectively 
the outer and the inner limit characteristics, both of which are design 
requirements. 
If gain quantification is used on reception, the amount of deterioration is 
small. 
For practical reasons, it is preferable to simplify gain adjustment in the 
TR modules on reception and thus to control the amplitude law by 
quantizing the gain to a limited number of levels. 
FIGS. 8A, 9A, and 10 are respectively the receive, the transmit, and the 
transmit-receive radiation patterns. They show that a suitable 
quantization scheme using eight levels (and thus 3 bits) suffices to have 
a receive illumination law close to that of "continuous control" (FIG. 6), 
and to obtain a radiation pattern of similar quality. Using 128 
omni-directional sources separated by 0.57.lambda., the overall 
illumination efficiency is just as good (-0.16 dB). 
A narrow lobe may also be synthesized using only 17 amplitude controls. 
FIGS. 11A, 12A and 13 are respective transmit, receive, and 
transmit-receive radiation patterns showing that even if the number of 
feed points to the antenna (and thus amplitude control points) is greatly 
reduced, and if quantization is kept to 8 levels only, the above-described 
method produces an equivalent transmit-receive radiation pattern of 
similar quality. This applies to an active antenna for space SAR, the 
antenna having a length of 8.16 meters, and being split up into 17 
subpanels each having a length of 48 centimeters, with the subpanels being 
controlled in amplitude only, thereby making it possible to use identical 
distributors within each of the subpanels. The overall illumination 
efficiency that is obtained is nearly as good (-0.18 dB). 
The method may be generalized to an arbitrary number of active modules 
above a minimum number of about ten in the antenna dimension under 
consideration. 
It is also possible to synthesize a lobe which is widened by using the 
phase shifters only while retaining the same equal-amplitude transmission 
law (same output power HPAs) and the discovered weighing law in reception 
(quantized as shown in FIG. 8). For example, a parabolic type phase law 
may be added if the mission requires a broad lobe (e.g. in side looking 
airborne radar (SLAR) mode for imaging the ground by generating pixels of 
brightness proportional to the intensity of the echo received from each 
revolution cell or SAR) if it is desired to maintain a ground swath of 
constant width. The corresponding radiation patterns are shown in FIGS. 
14, 15, and 16: these are respectively receive, transmit, and 
transmit-receive diagrams. With 128 omnidirectional sources spaced apart 
at 0.57.lambda., an overall illumination efficiency of -4.83 dB is 
obtained. 
This method limits the number of control signals to be transmitted to the 
active modules to the number needed reconfiguring the phase shifters only: 
this is required in any case for depointing the beam, which is generally 
related to beam widening. Since the widening phase is identical in 
transmission and in reception, the control rate as applied to the phase 
shifters is moderate. 
The attenuators situated on the receive paths of the active modules (or the 
variable gain LNAs) do not need to be controlled. They are preadjusted to 
a desired value and there is thus no risk of disturbing phase. 
If it is desired to obtain widened radiation patterns that are better 
formed, an additional degree of freedom may be used which consists in 
controlling the gain of the TR modules on reception and as a function of 
time. This makes it possible to adapt the receive illumination law better 
in amplitude and in phase to each intended lobe width. 
Thus, FIGS. 17 to 19 and 20 to 22 which are successive sets of transit, 
receive, and transmit-receive radiation patterns, show two lobes of very 
different widths (1.4.degree. and 8.8.degree. at 3 dB from the maximum) as 
generated by the same active array of 48 sources spaced apart by 
0.822.lambda.. 
Transmit illumination continues to be equal amplitude so that the HPAs all 
operate at identical level. 
However, by changing the receive illumination law both in amplitude and in 
phase, it is possible to generate either a very fine lobe having an 
SLL.apprxeq.-20 dB (FIG. 19) or else a wide lobe having very steep sides 
and a similar SLL (FIG. 22). 
By using this additional degree of freedom in control (amplitude on 
reception), the wide lobe pattern has much steeper sides than the pattern 
of FIG. 16, thereby improving the discriminating power of the radar. 
The only drawback of this variant is that the gain of the receive portions 
of the TR modules must be varied over time. This increases the control 
rate and means that additional drivers must be provided in the TR modules. 
Two options are available for gain control: 
either a maximum dynamic range is accepted (6.5 dB in the present example), 
thereby avoiding any need to change insertion phase in the receive path of 
a TR module; 
or else, if a larger dynamic range is necessary to obtain a better pattern, 
then the resulting phase changes are determined and account is taken of 
them when controlling the phase shifters. This has the drawback of 
requiring them to be reconfigured very quickly between transmission and 
reception. 
Some radars need to scan electronically in one plane only, e.g. in the 
elevation plane for side-looking airborne radar or for synthetic aperture 
radar. 
In the horizontal plane of the antenna there is no need for numerous 
amplitude and phase control operations. To simplify the antenna, a small 
number of active modules are distributed over its length. 
In this case, there are not enough amplitude and phase controls to be able 
to apply the above method to synthesizing the azimuth radiation pattern. 
However, the larger spacing of the TR modules makes it possible to apply a 
different principle: 
this time an identical azimuth pattern is provided both in transmission and 
in reception; 
the TR modules are distributed in non-uniform manner over the length of the 
antenna to provide the required amplitude weighting with all of the HPAs 
operating at the same output power, so as to optimize their efficiency as 
in the basic version. 
An example of the above principle is described below. 
FIG. 23 shows a large active antenna (8.32 meters by 1.91 meters) for use 
as a space observation radar operating in SAR mode. In order to enable it 
to be folded beneath the nose cone of the launcher, the antenna is split 
up into three panels 25, 26, and 27. It includes 88 identical horizontal 
transmission lines 28, as shown in FIG. 24. 
Five TR modules 29 are disposed on each horizontal line 28 of the radiating 
elements (in this case superposed slotted waveguides 32 and 33 radiating 
respectively in horizontal polarization and in vertical polarization). 
They are nonuniformly distributed (closer together in the center) and this 
distribution is associated with the conductances of the radiating slots 
being adjusted to obtain an amplitude law which varies linearly in dB from 
one end to the other, thereby enabling a good quality pattern to be 
obtained (FIG. 25). 
Simultaneously, only two types of radiating waveguide are required: 
waveguides 30 of length 60.4 cm having an angled illumination law; and 
waveguides 31 of length 53.7 cm having a uniform illumination law. 
This greatly simplifies industrial manufacture of the antenna. However, for 
the elevation pattern, the method described above is applied. 
Since electronic scanning is important in this plane, each subpanel is 
phased controlled (where a subpanel comprises a set of two waveguides 32 
and 33 which radiate with H and V polarization), i.e. phase control is 
provided at a pitch close to 0.7 times the wavelength; 
there are therefore 88 active modules spread uniformly over the height of 
the antenna, with the height of the radiating waveguide always being the 
same; 
on transmission, all of the HPAs have the same output power, thereby 
providing uniform illumination as shown in FIG. 26; 
on reception, attenuators situated behind the LNAs are adjusted to provide 
weighting, thereby providing the pattern shown in FIG. 26; and 
the equivalent transmit-receive diagram as shown in FIG. 28 has the same 
qualities as described above. 
Naturally, the present invention has been described and shown merely by way 
of preferred example and its component parts could be replaced by 
equivalents without thereby going beyond the scope of the invention.