LED driver circuit

A semiconductor chip includes an LED driver circuit operably coupled to at least one LED and configured to supply a load current to the at least one LED such that an average load current matches a desired current level defined by a drive signal. A temperature measurement circuit is thermally coupled to the LED driver circuit or the LED(s) or both, and is configured to generate, as drive signal, a temperature dependent signal in such a manner that the drive signal is approximately at a higher constant level for temperatures below a first temperature, is approximately at a lower constant level for temperatures above a second temperature but below a maximum temperature, and continuously drops from the higher constant level to the lower constant level for temperatures rising from the first temperature to the second temperature.

TECHNICAL FIELD

The present description relates to circuits and methods for driving light emitting diodes (LEDs), particularly to circuits and methods for driving LEDs including an over temperature protection.

BACKGROUND

Light emitting diodes (LEDs) are becoming increasingly popular as energy-saving substitute for incandescent lamps in various applications. Unlike incandescent lamps LEDs are current-driven components and as such require driver circuits including a load current regulation. In order to reduce power dissipation within the driver circuits switched mode power supplies are usually employed to supply a LED or a series circuit of several LEDs (also referred to as LED chain) with a well-defined load current. Generally, the resulting luminous intensity (usually measured in candela) is directly proportional to the load current. The power dissipation within the driver circuit (even when including a switching converter) may, however, still become a problem which—if no security mechanism is included—may result in a thermal destruction of the driver circuit, particularly of the power stages included therein. Not only the power stages of the LED driver but also the LEDs themselves are at risk to overheat.

For this purpose many LED driver devices (including an integrated driver circuit) include a sense terminal (i.e., a chip pin) to which an external temperature sensor may be attached (usually as an option). For example, the high power white LED driver STCF02 of STM (see STMicroelectronics, data sheet STCF02, February 2007) provides a chip pin for connecting an NTC temperature sensor which is a temperature dependent resistor (thermistor) having a negative temperature coefficient (NTC). The external temperature sensor is usually used to trigger a shut-down of the device when a critical temperature has been detected.

However, in security relevant applications (e.g., the illumination of emergency exits, escape routes, emergency shut-down switches, etc.) a simple shut-down of the LED driver is insufficient as maintaining the illumination is essential. Furthermore, also in non-security related applications reliability (even in hot environments or where sufficient cooling is problematic) may also be a desired feature of an illumination device including a LED driver and respective LEDs. Finally, it is desirable to reduce the required external components necessary to operate the LED driver and to protect the driver as well as the LEDs. The still required external components should be inexpensive and easy in integrate into an illumination device.

Thus there is a need for improved LED driver circuits that are easy to use and include an intelligent over-temperature protection.

SUMMARY OF THE INVENTION

A semiconductor chip including integrated circuitry for driving LEDs is described. In accordance with one example of the invention the circuit comprises a LED driver circuit operably coupled to at least one LED and configured to supply a load current to the at least one LED such that an average load current matches a desired current level determined by a drive signal. A temperature measurement circuit is thermally coupled to the LED driver circuit and configured to generate, as drive signal, a temperature dependent signal in such a manner that the drive signal is approximately at a higher constant level for temperatures below a first temperature, approximately at a lower constant level for temperatures above a second temperature but below a maximum temperature, and continuously drops from the higher constant level to the lower constant level for temperatures rising from the first temperature to the second temperature.

DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS

FIG. 1, which includesFIGS. 1a-1c, illustrates difference examples of LED driver circuits. In the example ofFIG. 1athe driver circuit includes a switching converter (precisely, a buck converter) whereas, in the example ofFIG. 1b, the driver circuit includes a modulator MOD to provide a modulated load current to the LED. The modulator MOD may be any common on/off-modulator such as a pulse width modulator (PWM), a pulse frequency modulator (PFM), a sigma-delta modulator or the like.

The circuit ofFIG. 1aincludes a first semiconductor switch, which is implemented as a MOS transistor M1, and a second semiconductor switch, which is implemented as a silicon diode D1. The MOS transistor M1and the diode D1are connected in series between a first supply terminal supplied with a first supply potential VBand a second supply terminal GND supplied with a second supply potential, e.g., ground potential VGND. The MOS transistor M1and the diode D1form a kind of a half bridge wherein the common circuit node of the transistor M1and the diode D1is the half-bridge output node at which the load current iL is provided. The LED is connected to that half-bridge output node via an inductor L1. As such a first inductor terminal is connected to the half-bridge output node whereas a second inductor terminal is connected to the anode of the LED. The cathode of the LED is coupled to the second supply terminal GND via a current sensing resistor RS such that LED, inductor L1and resistor RS form a series circuit. The voltage drop VSacross the resistor RS is representative of (in the present example proportional to) the load current iL passing through the LED. A comparator K1with hysteresis receives the a temperature dependent drive signal VDRIVE(T) and the voltage drop VSrepresenting the load current iL. The output of the comparator K1is coupled to the gate of the MOS transistor M1, e.g., via a designated gate driver circuit (not shown).

When voltage VS=RS·iLfalls below the lower threshold VDRIVE-ΔV, the output of the comparator K1drives the MOS transistor M1into an on-state in which the load current iLpasses from the first supply terminal to the second supply terminal GND via the MOS transistor M1, the inductor L1, the LED, and the sense resistor RS. In this case the diode D1is reverse biased. When the voltage VS=RSiLexceeds the higher threshold VDRIVE+ΔV, the output of the comparator K1drives the MOS transistor M1into an off-state in which—due to the self-inductance of the inductor L1—the load current iLpasses from the second supply terminal GND via the diode D1(which is then forward biased), the inductor L1, the LED, and the sense resistor RS back to the second supply terminal GND. As a result, the average load current iAVGcorresponds to VDRIVE(i.e., VAVG=VDRIVE/RS) whereas the peak-to-peak value of the ripple current is 2·ΔV. It should be noted that the LED driver circuit illustrated inFIG. 1ahas to be regarded as an example. The MOS transistor M1may be replaced by any other type of transistor, the diode D1may be substituted by an adequately driven transistor. The LED is coupled to the low side of the circuit. However, the LED may also be placed in a high-side configuration.

FIG. 1billustrates another exemplary driver circuit which does not require an inductor. In the present example the LED is connected in series with the load current path of a transistor M1(e.g., the drain-source current path in case of a MOSFET) and a current sense resistor RS. The total supply voltage (VB−VGND) is applied to this series circuit. In the present example the load current iL passes from the first supply terminal (which is supplied with the first supply potential VB) via the LED, the transistor's load current path, and the resistor RS to the second supply terminal GND which is supplied with a second supply potential VB, e.g., ground potential. The instantaneous load current value is dependent on the conduction state of the transistor M1. As in the previous example, the voltage drop VS(sense signal) across the sense resistor RS represents the load current iL wherein the voltage drop VSequals RSiL. In the current example, the transistor M1is driven by an operational amplifier whose output is coupled to the gate of the transistor M1(e.g., via a designated gate driver, not shown). The operational amplifier OP1is supplied with the sense signal VSand a corresponding reference signal VM. It operates as a P-regulator which regulates the load current iL(by appropriately controlling the conductance of the transistor M1) such that the sense signal VSapproximately equals the reference signal VM, which is tantamount to iL=VM/RS. That is, the load current is regulated to a value VM/RScorresponding to the reference signal VM.

The reference voltage is usually an on/off-modulated signal having an amplitude and a variable duty cycle D, wherein Dε[0, 1]. As a result, the load current iLpassing through the LED will be correspondingly on/off-modulated. The average load current iAVG(which determines the perceivable luminous intensity of the LED) is then iAVG=iLON·D wherein iLONis the on-value of the load current iLwhereas its off-value is zero. The on/off-modulated signal VM is usually generated by a common analog or digital modulator which is configured to generate the on/off-modulated signal VMand to set the duty cycle D to a value corresponding to a drive signal VDRIVE. As in the previous example, the drive signal VDRIVEis temperature dependent and indirectly determines the average load current iAVGpassing through the LED.

The general concept is summarized below with reference toFIG. 1c. A LED driver10is coupled to a LED (or a series circuit of LEDs) and configured to provide a load current iLto the LEDs. The LED driver10generates the load current iLin accordance with a drive signal VDRIVEsuch that the average load current iAVGmatches the drive signal. Thus, the drive signal indirectly determines the average load current iAVGand thus the luminous intensity of the LED. The drive signal is provided by a temperature measurement circuit20which generates the drive signal VDRIVEsuch that it depends on temperature. The temperature dependency of the drive signal VDRIVEfollows some specific characteristic curve which is described further below with reference toFIGS. 2 and 3. The temperature measurement circuit20, the LED driver circuit may be in close thermal contact. For example, both circuits10,20may be included in one integrated circuit (IC) placed in one single chip package. A detailed example of the circuit20will be described further below with reference toFIG. 4. The circuit20usually includes an integrated temperature sensor such as, for example, a diode.

FIG. 2illustrates a specific example of how the drive signal VDRIVEdepends on the temperature T. The diagram shown inFIG. 2illustrates the drive voltage in percent of a maximum drive voltage level VDRIVEmaxwhich is provided at low temperatures, e.g., below 70° C. When a specific first temperature (further referred to as temperature T1) is exceeded, the drive voltage VDRIVEis reduced. The decrease of the drive voltage VDRIVEcontinues as the temperature continues rising. The maximum drive voltage level VDRIVEmaxand the rate of the mentioned decrease (in volts per Kelvin) may be set by appropriate circuit design. When a specific second temperature (further referred to as temperature T2) is exceeded, the drive voltage remains approximately constant or is further reduced at a much lower rate. In the present example, the drive voltage VDRIVEstays at approximately 40 percent of the maximum level VDRIVEmaxfor temperatures above 108° C. However, when the temperature still rises and exceeds a maximum temperature TMAXthen a thermal shut-down is initiated. In the present example TMAXis approximately 160° C. The maximum temperature TMAXmay also be set by appropriate circuit design. The temperature measurement circuit20(seeFIG. 1c) may be configured to allow the adjustment of the first temperature T1and the second temperature T2using an external component such as an external resistor. This allows integrating the temperature measurement circuit20and the driver circuit10(seeFIG. 1c) into one single chip package and to allow the user to configure the temperature characteristic of the drive voltage VDRIVEby attaching a single external resistor to one specific pin of the chip package.

FIG. 3illustrates the temperature characteristic of the drive voltage on a more abstract level. The solid line illustrates one specific characteristic curve describing the behavior of the circuit20, which provides the temperature dependent drive voltage VDRIVE(T). Below a first temperature T1the drive voltage VDRIVEapproximately equals the maximum drive voltage level VDRIVEmax. Above a second temperature T2the drive voltage VDRIVEapproximately equals the low drive voltage level VDRIVElowprovided that, however, the temperature remains below the maximum temperature TMAX(TMAX>T2). A temperature equal to or higher than TMAXtriggers an over-current shut-down of the driver circuit. Between the first temperature T1and the second temperature T2the drive voltage drops approximately linearly. However, any other smooth or continuous transition between VDRIVEmaxand VDRIVElowwould be appropriate.

Reducing the drive voltage VDRIVEat elevated temperatures (above T1) entails a lower average load current passing through the LED resulting in a lower power dissipation in both, the driver circuit10as well as the LED(s). The lower power dissipation counteracts a further increase in temperature and may lead to a cooling-down of the LED and the driver circuit. However, the flat portion of the curve for temperatures T lower than T1ensures that the load current iLand thus the perceivable luminous intensity is maintained on a constant desired level during normal operation in a pre-definable temperature range T<T1. The gradual decrease of the drive voltage helps to reduce the dissipated power and thus reduces the risk of overheating. However, the perceivable luminous intensity is also reduced. The flat portion of the characteristic curve for high temperatures T>T2is provided to maintain a defined minimum luminous intensity (corresponding to a minimum drive voltage VDRIVEmin), which is advantageous in security relevant applications such as illumination of emergency exits, emergency shut-off switches or the like. To avoid a thermal destruction of the driver circuit, the circuit is deactivated when the temperature exceeds a maximum temperature TMAX. ASlong as the temperature remains lower than the maximum temperature TMAXa thermal equilibrium may occur at any point on the curve shown inFIG. 3, dependent on the actual temperature of the driver circuit and the ambient temperature.

The parameters T1and T2fully determine the characteristic curves. According to one example of the invention these parameters may be set by adjusting the resistance on one external resistor connected to the measurement circuit. As such the curve defined by the temperatures T1′ and T2′, T1″ and T2″, T1′″ and T2′″, and T1″″ may be chosen (the temperature T2″″ corresponding to T1″″ would be higher than TMAXand thus ineffective).

One exemplary measurement circuit that allows an efficient implementation of the measurement circuit is illustrated inFIG. 4. The circuit ofFIG. 4is supplied with a supply voltage VSwith respect to a reference potential referred to as ground potential GND in the present circuit. The circuit ofFIG. 4is further provided with an input voltage VIN(corresponds to VDRIVEmaxinFIG. 2) that which sets the maximum output voltage VDRIVE(T). Several reference current sources Q1, Q2, Q3, Q4, and Q5are used in the circuit. All these current sources provide fixed multiples of a reference current iREFwhich is essentially temperature independent. For this purpose a band-gap reference circuit may be used to generate a temperature independent reference current, and all current sources may derive the sourced current from the stable output current of the band-gap reference circuit.

In the present example the temperature dependent forward voltage VBEof a two silicon diodes D1and D2are used to provide the middle portion of the characteristic curve (between temperatures T1and T2) depicted inFIG. 3. The forward voltage VBEof a diode (this is also valid for the base-emitter-diode of a bipolar transistor) has a temperature coefficient of about −2 mV/° C., that is the voltage VBEdrops for about 2 mV as the temperature rises by one degree Celsius. The two diodes D1and D2are connected in series to a first current source Q1, which provides a current iREF. The diodes D1and D2are connected between the supply node at which the supply potential VS is provided and the current source Q1. The voltage drop 2·VBEacross the diodes D1, D2is converted into a temperature dependent current iSLOPEwhich approximately equals VBE/R1. For this purpose a bipolar transistor T1(pnp type) is provided. The emitter of the transistor T1is connected so the supply node via the resistor R1(emitter resistor) and the base of the transistor T1is connected to the common circuit node of current source Q1and diode D1. As a consequence, the voltage drop across the emitter resistor R1is approximately VBE(assuming the base-emitter voltage of transistor T1is also VBE) and thus the collector current of the transistor T1(denoted as iSLOPE) equals VBE/R1(assuming the base current of the transistor T1is negligible). Therefore the current iSLOPEexhibits the same temperature dependency as the diode forward voltage VBE. In essence the transistor T1and the resistor R1can be regarded as voltage-to-current converter which converts the temperature dependent forward voltage VBEinto a corresponding current iSLOPE.

The current iSLOPEadds to the emitter current iET2of a second bipolar transistor T2(npn type) and the sum current iSLOPE+IET2is directed through the resistor R3to the ground node, at which the ground potential GND is provided. That is, the resistor R3is connected between the emitter of transistor T2and ground. The base of the transistor T2is supplied with a base voltage of 2·iREF·R2+VBE, whereby the current 2·iREFis provided by the second current source Q2, the voltage VBEis the forward voltage of a further diode D3. The resistor R2is connected in series with the diode D3and the current source Q2such that the sourced current 2·iREFis mainly (i.e., neglecting the base current of transistor T2) directed through the diode D3and the resistor R2. The transistor T2essentially operates as an emitter follower and thus the emitter voltage V3of the transistor T2follows essentially the base voltage minus the forward voltage of the base-emitter diode. That is, the emitter voltage V3equals approximately the voltage drop across the resistor R2and thus V3=2·iREF·R2. As a result the emitter current iET2of the transistor T2can be calculated as iET2=2·iREF·R2/R3−iSLOPE. This emitter current iET2is copied and magnified by a factor 10 using the current mirror CM1. That is, the current mirror output current at the circuit node N equals 20·iREF·(R2/R3)−10·iSLOPE. The capacitor C1coupled to the current mirror output node (node N) is used to suppress transient current spikes. In essence, the current mirror CM1in combination with the transistor T2(and the circuitry for biasing the base of the transistor T2) and the resistor R3can be regarded as subtracting circuit configured to subtract the current iSLOPEfrom a pre-defined constant current (2·iREF·R2/R3).

The first break of slope of the characteristic curve ofFIG. 3at temperature T1(temperature threshold) may be set by appropriately choosing the values of the resistors R1, R2, and R3, wherein the steepness of the slope between the temperatures T1and T2is mainly determined by the value of resistor R1. The characteristic curve ofFIG. 3may be shifted to the right as illustrated inFIG. 3by means of the resistors R4, R5, and REXT, which is an external component placed outside the chip, the MOS transistor M1, the current source Q4, and the operational amplifier OA1, particularly by adjusting the resistance of the external resistor REXT. Accordingly, the current source Q4sources a current 5·iREFwhich is directed through the resistors R5and REXTwhich are connected in series between the current source Q4and the ground node GND. Furthermore, the resistor R4is connected between the ground node GND and the source electrode of the MOS transistor M1, which has a gate electrode that is driven by the output of the operational amplifier OA1. The operational amplifier OA1controls the MOS transistor such that the voltage drops across the resistor REXTand the resistor R4are approximately equal. The resulting drain current passing through the MOS transistor (n-channel type) is denoted as iM1. As such, the terminals of the resistors REXTand R4not connected to ground are connected to the inverting and non-inverting inputs of the operational amplifier OA1, respectively. As the voltage iM1·R4=5·iREF·REXT, it follows that the current iM1equals 5·iREF·REXT/R4. The current iM1is copied and downscaled to the output of the current mirror output branch of current mirror CM2. The respective mirror current 0.5·iM1=5·iREF·REXT/R4is also supplied to the circuit node N. As compared to the mirror current (10·iET2) at the output of the first current mirror CM1the mirror current (0.5·iM1) does not significantly depend on temperature. In essence the current mirror CM2in combination with the circuitry providing the input current to the current mirror CM2can be regarded as current source providing an offset current (i.e., the mirror output current 2·iM1) that can be set using the external resistor REXT.

The minimum drive voltage VDRIVEmin(seeFIG. 3) may be set my appropriately choosing the resistors R6and R7which are used in combination with the third current mirror CM3, the MOS transistor M2(n-channel type), the current source Q5, and the operationally amplifier OA2. The input branch sinks the residual current iRESfrom circuit node N, whereby another current 2.5·iREFis sunk from node N using current source Q3. That is, iREScalculates as iRES=10·iET2+0.5·iM1−2.5·iREF. This residual current iRESis copied and downscaled to the output branch of the current mirror CM3. A series circuit of current source Q5(sourcing a current of 2·iREF), MOS transistor M2and resistor R7is connected between the supply node (supply voltage VS) and the ground node, wherein the MOS transistor is connected between the resistor R7and the current source Q5, and the resistor R7is connected between the MOS transistor M2and the ground node. The gate of MOS transistor M2is controlled by the operational amplifier OA2, which receives the input voltage VIN(corresponds to VDRIVEmax) at its non-inverting input and the voltage across resistor R7at its inverting input. The output branch of the current mirror CM3is connected to the drain of the MOS transistor M2via resistor R6. That is, the resulting drain current of the MOS transistor M2is the current 2·iREFprovided by the current source Q5minus the (mirrored and downscaled) residual current 0.5·iRESwhich is sunk by the current mirror CM3via resistor R6. Thereby the voltage drop across the resistor R6is R6·iRES.

At low temperatures, the current 0.5·iRESsunk by the current mirror CM3is low and thus the operational amplifier may regulate the output voltage (drive voltage VDRIVE) to equal the input voltage VIN, while the current source Q5operates as a high-impedance active load. As the temperature rises, the current 0.5·iRESsunk by the current mirror CM3also rises and the operational amplifier saturates and the MOS transistor M2becomes fully conductive with a low drain-source voltage drop. In this operational state the drive voltage VDRIVEwill follow the voltage drop across the resistor R6which is temperature dependent. This voltage drop across the resistor R6will not exceed the value 0.5·iREF·R6(as the current source Q5will not deliver more). Thus, the value of R6determines the minimum drive voltage VDRIVEmin.

Finally, the comparator K1in combination with the further MOS transistor M3may be used to deactivate the drive voltage VDRIVEwhen a maximum temperature TMAXis exceeded (seeFIG. 3). The comparator is configured to compare the voltage VS−2·VBEwith a reference voltage representing the maximum temperature. In case the voltage VS−2·VBEdrops below the reference voltage VREF(at a temperature TMAX) then the MOS transistor, which is controlled by the comparator output, will clamp the output voltage VDRIVEto zero volts.