Acoustic admittance measuring apparatus with wide dynamic range and logarithmic output

An electrical circuit system for measuring the acoustic admittance of the ear cavity, for use in single and multiple tone tympanometry and acoustic reflex response testing, in which the probe tone applied to the ear cavity can be varied over a wide frequency range and a wide range of admittance variation can be measured. The signal output of the system, derived from the output signal of a microphone located in the ear cavity, is a DC voltage accurately proportional to the logarithm of the measured admittance of the ear cavity. An RMS to DC converter, an error integrator and an exponential element provide a closed control loop system for a variable gain amplifier so as to maintain the microphone output signal level constant regardless of admittance variations in the ear cavity. The level of the signal applied to the probe tone driver to accomplish this is a measure of the cavity admittance.

BACKGROUND OF THE INVENTION 
The present invention is directed to a circuit apparatus for measuring the 
acoustic admittance of the ear. 
In the practice of tympanometry, measurements of the electrical impedance 
of the tympanic membrane of the inner ear (actually the admittance 
thereof, which is the reciprocal of the impedance) are made as ranges of 
air pressure or vacuum levels are applied thereto in the presence of tone 
signals of single, or more recently differing, frequency. The positive and 
negative pressure levels are applied to the eardrum via a probe inserted 
into the ear which also serves to apply the acoustic tone levels and act 
as a microphone to measure the admittance of the tympanic membrane in 
response to the tone stimulus as pressure changes are made. A number of 
tympanometry testing instruments are commercially available to the 
practitioner; including those made by Amplaid Spa, of Milan, Italy, 
Grason-Stadler, of Littleton, Mass., and Madsen Electronics, of Oakville, 
Ontario and Buffalo, N.Y. However, at present, the existing tympanometric 
instrumentation in the market allows only a few frequency tones to be 
applied to the eardrum, typically the standard frequency tones of 226 and 
678Hz. To expand the range of diagnostic evaluation in this field it is 
desirable to perform multiple frequency tympanometric measurements and 
thus to be able to apply to the ear cavity via a probe selected tones at 
various frequencies over the mid-sonic range. During tympanometry, it is 
desirable to have a fast response to changes in the ear cavity's 
admittance in order to reduce the total measurement time and to track the 
changes in admittance. On the other hand, in another type of test, when a 
stimulus tone signal is present together with a standard probe tone level 
signal in measuring the acoustic reflex response of the ear, it is 
desirable that the bandpass filter have a narrow bandwidth in order to 
better reject large amplitude stimulus signal which may be up to 40 dB 
above the probe tone level. 
BRIEF SUMMARY OF THE INVENTION 
The system of the present invention enables, in an acoustic admittance 
measuring system, selected probe tones over the frequency range of about 
150 to about 2.5 KHz to be applied as a stimulus to the ear cavity via an 
acoustic probe, and then to measure the response of the tympanic membrane, 
as represented by its electrical admittance and changes therein. 
(Limitations in the mechanics of the probe design may reduce the effective 
upper frequency limit for measuring response characteristics to about 1 
KHz.) The herein disclosed circuitry provides a simple and inexpensive 
means for accurately measuring a wide range of admittance variation (in 
excess of 40 dB) in response to probe tones varied within the 
aforedescribed range. 
The innovative advantages and features of the acoustic admittance measuring 
system of the present invention include: 
The probe tone level within the ear cavity is maintained constant 
regardless of ear size and/or the static pressure level. 
The system is insensitive both to interference from noise and to the 
presence of much larger signals at frequencies different from the probe 
tone, such as stimulus tones. 
The system's dynamic response to rapidly varying admittance is independent 
of the mean admittance level. 
The signal output of the system is a DC voltage accurately proportional to 
the logarithm of the admittance (in decibels). 
A noise-free, constant amplitude version of the probe tone within the ear 
cavity is provided by the system for use in phase determination, if 
desired. 
An electronically-tunable bandpass filter is used to reject interfering 
signals and to permit rapid electronic control of the probe tone 
frequency. 
An integrating error amplifier is provided to maintain the cavity probe 
tone level at a constant value. 
An exponential element is employed, inside the level control loop for the 
circuit, to provide a DC output proportional to the cavity's admittance 
while maintaining constant loop gain and loop dynamics regardless of 
cavity admittance.

DETAILED DESCRIPTION OF THE INVENTION 
The functional components of the system shown in the block diagram of FIG. 
1 are arranged in a closed control loop which automatically adjusts the 
output of the Probe Tone Driver 30 within the Acoustic Cavity 10 so that 
the Microphone Output Signal Vp from the Microphone 35 is constant 
regardless of cavity admittance variations. The level of the signal Vd 
applied to the driver to accomplish this is a measure of the cavity's 
admittance. 
The Microphone Output Signal Vp is amplified to a suitable level by the 
Microphone Preamplifier 40, and applied to the Bandpass Filter 50. The 
filter's passband is centered about the probe tone frequency Fp. The 
filter is built from a switched capacitor filter element, such as the MF10 
marketed commercially by National Semiconductor and other suppliers, which 
allows the center frequency to be controlled over a wide range by the 
frequency Fc of the Filter Clock 60. The Bandpass Filter 50 has a 
relatively narrow bandwidth (approximately 10% to 25% of the center 
frequency Fp), and sharply increasing attenuation outside the passband, 
which prevents interfering signals from being passed to the rest of the 
system. 
The filtered, normalized probe signal Vpn is passed to the RMS to DC 
Converter 70 which outputs a DC voltage Ap proportional to the RMS value 
of the probe tone. Since in operation the signal level at the converter 
input is constant, the accuracy of the converter is not important, only 
its stability. The filtered probe tone signal Vpn is also at a suitable 
level for use as an output in a phase measuring subsystem (not shown). 
The RMS to DC Converter output Ap is subtracted (by current summing at 
summer 80) from the Probe Level Reference signal Apr. The difference is 
the Probe Signal Level Error Ver, and it is applied to the Error 
Integrator 90 whose output is the time integral of the error. The 
integrated error signal Ay is input to an Exponential Element 100 whose 
output (a current level) Agc is accurately proportional to the exponential 
function (e.sup.x) of its input. 
The probe tone Driver Reference Signal Vdr is a constant amplitude sinewave 
at the probe frequency Fp. It is applied to the signal input of a Variable 
Gain Amplifier 110. The amplifier's gain, and therefore also its output 
Yd, are proportional to the Gain Control Signal Agc. Assuming that the 
gain of the Probe Tone Driver 30 is also fixed, the Gain Control Signal 
and the Probe Driver's Tone acoustic output are proportional. 
The control loop is closed by the sound level at the Microphone 35, which 
determines the Microphone Output Signal Vp. The feedback around the loop 
is negative, so that an increase (or decrease) in microphone output causes 
a decrease (or increase) in driver output. 
The use of an integrating error amplifier 90 means that, on average, the 
RMS to DC Converter output Ap must exactly equal the Probe Level Reference 
Apr. Otherwise, the Error Integrator's output continues to change and, via 
the Exponential Element and variable Gain Amplifier, changes the Driver 
Signal Vd until equality is established. Therefore, the probe tone sound 
level at the Microphone is also accurately constant since it is related to 
the RMS to DC Converter's output by the fixed gain of the 
preamplifier/filter/converter chain. Given a constant cavity sound level, 
the amplitude Vd of the Driver Signal required to accomplish this level is 
proportional to the cavity's acoustic admittance. The input signal Ay to 
the Exponential Element 100 is proportional to the logarithm of its output 
Agc. Since this output is proportional to the Driver Signal Vd, the input 
(which is also the Error Integrator's output) is proportional to the 
logarithm of the Acoustic Cavity's admittance. 
The aforesaid logarithmic relationship provided as an output signal Ag by 
the operation of the Exponential Element 100 permits the admittance to 
vary widely while the output per percent admittance change remains 
constant. A relatively low resolution and inexpensive analog-to-digital 
converter (not shown) can then be provided to convert this signal output 
while still maintaining constant percentage resolution. The accuracy and 
acceptable acoustic admittance range of this portion of the system depends 
on the accuracy of the Exponential Element. This is preferably implemented 
by using the exponential relationship existing between the collector 
current and base-to-emitter voltage of a silicon bipolar transistor which, 
in modern transistors, is very accurate over several decades of current 
levels. 
An additional result of using the Exponential Element inside the feedback 
loop is that the incremental loop gain (expressed, for example, as the 
percent change in driver output per percent change in microphone input) is 
constant, regardless of the Driver Signal level Vd. Therefore, the static 
and dynamic response of the system to slow and fast cavity admittance 
changes is independent of the cavity admittance--a very desirable 
characteristic. 
In the system of FIG. 1 there is shown therein a means for accomplishing a 
rapid change in the cavity's admittance, and thus the response speed of 
the system, by electronically increasing (or, alternatively, decreasing) 
the bandwidth of the Bandpass Filter 50, to which the signal Vp from the 
Microphone 35 is inputted, in response to the Control Signal Vb. The 
bandwidth of the Error Integrator 90 is also increased (or decreased) in 
the same proportion as the Bandpass Filter's bandwidth in order to 
maintain a clean dynamic system step response, with fast settling to the 
final value and minimum overshoot and ringing. 
An exemplary circuit for implementing the above-described acoustic 
admittance measuring system is shown in the schematic diagram of FIG. 2. 
The Variable Gain Amplifier 110 is U101 and associated components; the 
Microphone Preamp 40 is U111 and associated components; the Bandpass 
Filter 50 is U121 and associated components; and the RMS to DC Converter 
70 is U141 and associated components. The Probe Level Reference Apr is the 
current through R143, and the RMS to DC Converter output Ap is the current 
through R142 The Error Integrator 90 is U161 and associated components; 
and the Exponential Element 100 is Q171, and associated components, which 
is driven by a low impedance buffer forming a part of U141. 
Bandpass Filter 50 is built from an MF10 switched capacitor filter element 
U121. In narrow bandwidth operation FET (field effect transistor) switches 
Q121 and Q131 are set to OFF (with the gates at approximately -11V) and 
the bandwidth is determined by R124 and R134. To increase the bandwidth, 
without changing the center frequency of the filter these resistors must 
be reduced proportionately. However, the gain at center frequency Fp 
depends on the resistor ratios R124/R122 and R134/R132. In the exemplary 
embodiment of the invention shown, the required switching to wide 
bandwidth is accomplished by turning ON FET switches Q121 and Q131 through 
the medium of switching the gates to zero volts. This step connects R123, 
R133, respectively, in parallel with R124, R134, respectively, to increase 
bandwidth, and also connects R121, R131, respectively, in parallel with 
R122, R132, respectively, to maintain constant the gain at center 
frequency of the bandpass Filter. 
The Error Integrator 90 is built from element U161 and its gain .times. 
bandwidth product is determined by 1/(C161.times.error signal source 
resistance). For a narrow bandwidth, FET switch Q161 is set to OFF (i.e., 
gate at approximately .times.11V),a nd the source resistance is then R161 
in series with the combination of R142 in parallel with R143. However, for 
wide bandwidth operation, Q161 is turned to ON (gate to zero volts) with 
the result that the signal source resistance is now merely R142 in 
parallel with R143 (plus the negligibly small resistance of the FET when 
in the ON condition), and the bandwidth of the Error Integrator is 
increased as required. 
Amplifier U151 and associated components are used as a level translator for 
converting the Bandwidth Control Signal Vb to the -11V or zero volts 
signal as required to control the FET switches Q121, Q131 and Q161 in the 
Bandpass Filter and error Integrator, respectively. 
The terms and expressions which have been employed int eh foregoing 
specification are used therein as terms of description and not of 
limitation, and there is no intention, in the use of such terms and 
expressions, of excluding equivalents of the features shown and described 
or portions thereof, it being recognized that the scope of the invention 
is defined and limited only by the claims which follow.