Magnetic recording system and method

Magnetic recording circuitry supplies to the recording head in addition to the signal to be recorded a train of constant amplitude pulses of extremely short duration and of low duty cycle; for example, pedestal pulses of polarity corresponding to the signal are superimposed on the signal for higher positive and negative signal amplitudes, while alternating polarity pulses are supplied during periods of low signal amplitude, such that the transducer system exhibits a substantially linear transfer function.

BACKGROUND OF THE INVENTION 
In the magnetic recording field, there has been a continuing development 
effort toward producing magnetic transducer heads of higher and higher 
resolution. As the effective length measured across the transducing gap of 
the head is reduced, with a consequent reduction in gap reluctance, higher 
input signals to the head are generally required to produce a given level 
of recorded magnetization. The requirement for the use of a superimposed 
high frequency bias signal can result in undue heating of the head, 
particularly as the coercivity of the record medium is increased for the 
sake of higher resolution. The undue heating is a problem in recording 
heads as well as erasing heads, not only because of possible damage to the 
record medium, but also because the record medium coating material may be 
softened and accumulate on the head if the head is operated at relatively 
high temperatures and slow scanning speeds. Also the magnetic properties 
of the head core and of the record medium may be adversely affected. 
While magnetic records can be monitored shortly after being made, such 
monitoring has typically been for the purpose of verifying that fact of a 
recording being made without the availability of immediate correction of 
the record in the event of an inaccuracy in the recorded signal. 
SUMMARY OF THE INVENTION 
Accordingly, it is an object of the present invention to mitigate the 
foregoing problems and to provide a transducer head of higher resolution 
capability and a head capable of more accurate recording or erasing while 
avoiding the problem of high operating temperature. 
In accordance with one aspect of the invention, a transducer head 
configuration is utilized which is operable by means of recording pulses 
of extremely short duration and of low duty cycle; further, provision may 
be made for reading the recorded magnetization on the record medium 
between recording pulses and correcting the recorded magnetization is 
necessary in a subsequent recording pulse prior to substantial movement of 
the record medium relative to the transducing gap. 
By the use of such recording pulses in such a way as to provide a required 
recording bias in conjunction with the signal to be recorded, or a 
required erasing field, much less energy is needed and the head is able to 
operate at a much lower temperature, thus also reducing any danger of 
harming the tape or of gathering tape coating material at record medium 
contacting surfaces of the head. A turn-in-gap spacer can be made of 
thinner material having higher resistance without excessive heating. With 
audio frequency recordings the duty cycle can be made exceptionally small 
while assuring the fidelity of the recorded magnetization. For example, 
the interval between pulses can be about twelve microseconds and the pulse 
width can be in a range from about twelve nanoseconds to four nanoseconds 
for a duty cycle of 0.001 to 0.00033, or even shorter. Where desired, the 
short duration of the pulses allows modification of the field at the gap 
before the record medium has moved appreciably. Pulse bias or pulse 
erasing is especially effective with thin film gap spacers (e.g. used as 
head windings) since angstrom magnitude film thicknesses have high 
resistance and prohibitive losses when used in a conventional manner. 
The fine particle (or thin film) structure of the record tape magnetizable 
layer provides response to nanosecond pulses. Such pulse type magnetic 
fields are sufficient to set the magnetic domains of the recording layer 
of the record medium. It is not necessary to maintain the magnetic field 
once the domains are set, so there is a great savings in turning off the 
applied magnetic field between the pulses. Further the off interval may be 
used to advantage (e.g. for monitoring or linearization of the recording). 
To read the recorded magnetization in such off intervals, a flux sensitive 
pickup such as a Hall element can be utilized although transducers based 
on magnetoresistance or magnetic modulator effects can also be used. 
Feeding back the playback signal through a comparator with a slight delay 
or memory to modify the subsequent recording pulse (when the initial 
magnetization deviates from the input signal) allows recordings to be made 
on the record medium which are corrected for variations in tape-head 
contact, dropout effects, biasless recording, etc. 
For successful alternating polarity pulse bias, a discontinuous pulse 
waveform is used with relatively short duration pulses of alternate 
polarity and with substantially zero amplitude intervals of relatively 
long duration between the successive pulses of alternate polarity. 
Pedestal type pulses of constant amplitude and of polarity corresponding to 
the instantaneous polarity of the signal information waveform may be 
superimposed on the signal during recording, and such pedestal pulses may 
be interrupted when the signal has negligible amplitude, e.g. during 
polarity transitions of the signal waveform. Alternatively during polarity 
transitions of the signal waveform, the single polarity pedestal pulses 
may be replaced by alternating polarity low duty cycle pulses which insure 
linear recording of low amplitude signals and a zero magnetization of the 
record medium for a zero signal amplitude. The pulse repetition rate of 
the constant amplitude pulses preferably corresponds to a fundamental 
frequency which is more than twice the maximum frequency which can be 
recorded by the magnetic transducer head. 
In Camras U.S. Pat. No. 2,900,443, the teaching in one embodiment was to 
record video signal information in the form of successive pulses where the 
amplitude of each pulse represented the amplitude of a sample of the 
analog signal. Where precise linearity was desired, a high frequency bias 
field was to be superimposed, e.g. by means of an auxiliary bias head. 
Thus, the early Camras patent directs the art away from the present 
invention and toward the actual recording on a record medium of individual 
pulses which constitute the signal information. The teachings of the 
present invention of applying pulses to the recording head at a repetition 
rate at least twice the head resolution is totally contrary to the 
teaching of the early Camras patent

DETAILED DESCRIPTION 
FIG. 1 is a side view of a transducer head useful in carrying out the 
recording and playback operations of the present invention. The transducer 
head comprises a ring type core 10 with a pair of core pieces 11 and 12 
separated by a gap conductor element 14 in the transducing region of the 
core, and separated by a Hall element assembly 15 at a gap region remote 
from the transducing region. A tape record medium travels successively 
over the pole pieces 16 and 17 of the transducer head along a tape path 
indicated at 18. 
FIG. 2 is an end view of the head assembly 10 and shows the gap conductor 
14 as having respective terminals 21 and 22 which are connected by means 
of suitable electrical conductors with terminals 21a and 22a of a driver 
circuit 24. FIG. 2 also shows an energizing circuit 26 associated with the 
Hall element 15. In particular, terminals 31 and 32 in FIGS. 1 and 2 are 
connected with terminals 31a and 32a of a shielded cable 33, leading to a 
monitor amplifier 34. FIG. 1 may be taken as illustrating the playback of 
a previously recorded tape whereby a direct current source 35 provides a 
unidirectional current longitudinally of the Hall element 15 (as viewed in 
FIG. 2), so that terminals 31 and 32 supply a continuous output analog 
signal in accordance with the signal flux threading the loop magnetic core 
10 and thus traversing the Hall element 15 at the region of the back gap 
between core pieces 11 and 12. 
During recording operation with the system of FIG. 2, a square wave 
generator 40 has its rectangular waveform output differentiated by means 
of capacitor 41 and resistor 42 so as to generate opposite polarity pulses 
as indicated in FIG. 3 at transformer 44. The secondary windings of 
transformer 44 drive respective power field effect transistors 45 and 46 
so as to provide alternating pulse bias at terminals 21a, 22a as in FIG. 
3. As indicated respective secondary windings 44a and 44b of transformer 
44 are oppositely poled so as to alternately supply pulses of positive 
polarity via the respective diodes 48 and 49, the gate biasing circuits 
being completed by respective resistors 51 and 52 which are shunted by 
respective zener diodes 53 and 54 (built into the transistors). The zener 
diodes serve to protect the gates of the field effect transistors 45 and 
46. A fifty volt power supply is connected to terminals 56 and 57, 
terminal 56 receiving plus twenty-five volts relative to ground potential 
and terminal 57 receiving minus twenty-five volts relative to ground 
potential. Resistors 58 and 59 are connected in series between terminals 
56 and 57 to act as a voltage divider for the common power supply and have 
a common circuit point 60. Capacitors 61 and 62 are also connected across 
terminals 56 and 57 and have a common circuit point 63 connected with 
terminal 22a. 
A signal to be recorded is supplied from any suitable recording signal 
source 70 via amplifier 71 and recording signal driver 72 to a transformer 
73 having its secondary winding 74 connected between circuit points 60 and 
63. In this way, where the recording signal has a waveform as indicated in 
FIG. 5, a current waveform as indicated in FIG. 4 is actually supplied to 
the gap conductor 14 via terminals 21a, 22a. 
As indicated in FIG. 6, gates 81 and 82 may be placed in open 
(transmissive) condition during time intervals such as represented at 82a, 
82b, 82c, etc. after time delays such as indicated at 83a, 83b, 83c, etc. 
The time delays are such that a signal once recorded on the record medium 
at tape path 18 is immediately reproduced by means of Hall element 15 
during an interval such as 82a. Delays 85 and 86 are such that the monitor 
amplifier 34 supplies to comparator 88 only a reproduction of the recorded 
signal (without any influence from the recording signal field 
corresponding to the recording current indicated in FIG. 4). Thus, the 
signal is completely recorded during an interval such as 83a, and playback 
of the recorded signal at 82a is dependent only on the actually recorded 
signal. The delay of delay element 86 may be somewhat greater than the 
delay provided by delay element 85 to eliminate transients resulting from 
the operation of the Hall element exciter gate 81. The gates 81 and 82 may 
be six diode switches or other standard types of gating circuits. Delay 
element 86 may be a monostable multivibrator that holds the gate 82 open 
during the "playback" interval such as indicated at 82a. Thus the logical 
one output level from delay element 86 may have a duration corresponding 
to the time interval 82a of FIG. 6, so as to switch gate 82 to the 
nonconducting mode before the next pulse of the sequence of FIG. 3 or FIG. 
4. Alternatively, delay 86 may comprise a monostable multivibrator which 
may turn off the gate 82 during intervals such as indicated at 83a, the 
monostable multivibrator being actuated after an appropriate delay from 
the previous recording pulse so that it turns off the gate 82 just prior 
to the recording interval and turns on the gate 82 after the recording 
interval. 
In the comparator 88, the monitored signal magnetization from the record 
tape during a transmissive interval of gate 82 is compared with the 
desired signal from recording signal source 70. Any error between the 
desired and actual recorded signal thus results in an error signal at 
output 91 of comparator 88. Such error signal may be supplied to a gain 
control circuit 92 which controls the amplification of amplifier 71 so 
that during the next recording interval, the recorded magnetization is 
modified so as to reduce the error signal at output 91 of comparator 88. 
Thus, the monitored and gated playback signal which is present during 
interval 82a, FIG. 6, is utilized during recording interval 83b to apply 
any necessary correction to the signal recorded during previous interval 
83a. This is possible because of the high repetition rate of the recording 
intervals 83a, 83b, in comparison to the time required for the tape to 
traverse the recording region of the recording head, and in comparison to 
the highest frequency component of the signal to be recorded. 
In order to use the system of FIG. 2 for erasing, recording signal driver 
72 may be disconnected from winding 73 so that the waveform of the current 
applied to the turn-in gap conductor 14 is as represented in FIG. 3. The 
current pulses of FIG. 3 would be supplied to the conductor 14 at a 
sufficiently high repetition rate in comparison with tape velocity such 
that a tape element does not move away from the influence of the head gap 
in the time between successive magnetic field pulses. Thus, each element 
on the tape magnetizable layer is subjected to a series of alternately 
opposite magnetic field pulses of progressively decreasing intensity. 
During recording operation, compensation for any residual magnetization of 
the magnetic core 11, 12 may be effected so that the gated playback 
signals at conductor 94, FIG. 2, are a function only of the tape 
magnetization and not of the magnetization of the magnetic core. This may 
be effected by reading the core magnetization via the Hall element 15 in 
the absence of a tape at the recording gap of the head 10 and introducing 
a corresponding compensation in the circuit of the Hall element 15 or at 
the comparator 88, for example. The effect of core magnetization at the 
turn-in gap conductor 14 is reduced by the provision of a relatively large 
additional non-magnetic gap in the magnetic circuit of the core. The Hall 
element 15, by increasing the dimension of the back gap between core parts 
11 and 12, is beneficial in this respect. 
Examples of magnetic core configurations which reduce the effect of core 
magnetization are found in Camras U.S. Pat. No. 3,591,729 issued July 6, 
1971. 
The waveforms of FIG. 3 and FIG. 7 can be utilized for effecting erasure of 
a prerecorded magnetic record tape or may be utilized in conjunction with 
the self correcting recording system as described in conjunction with FIG. 
2. Preferably, each of the pulse waveforms has a very low duty cycle, 
preferably one percent or less with very high frequency components of the 
pulses or damped waves producing magnetic fields of corresponding 
waveform. Thus, each of the waveforms of FIG. 3 and FIG. 7 has the 
advantage of giving higher intensity magnetic fields than can be obtained 
otherwise without excessive heating, such high field intensities being 
useful for very high coercivity tapes and/or for thick tape magnetizable 
layers. Such pulse waveforms use less energy than other methods and the 
transducer heads can be very small, for example with microdeposited core 
configurations where heat dissipation and conductor size are otherwise 
severe problems. The long intervals between pulses or wave trains in 
relation to the intervals of the pulses or wave trains themselves allows 
sampling of the recorded magnetization as shown by the circuit diagram of 
FIG. 2 between recording (or erasing) intervals. Thus feedback type 
correction acting on subsequent recording impulses as shown in FIG. 2, for 
example, may be carried out. The higher magnetic fields penetrate deeper 
into a recording magnetizable layer allowing thicker layers to be used. 
Eddy currents in the gap spacer are more effective in sharpening the 
spacial field because of the high pulse train frequency. A quieter erasing 
or biasing its obtained by the stronger fields and higher frequencies 
possible with these waveforms. Though the duty cycle is low, the 
repetition rate is so high that the tape has not moved appreciably between 
pulses. 
FIG. 3 is an example of a suitable waveform where the pulse duration is 
about twenty nanoseconds, while the time between pulses is two thousand 
nanoseconds, for example, corresponding to a fundamental frequency of 250 
kilohertz. In an audio recorder where twenty kilohertz is the maximum 
signal frequency, such a signal would have at least 12.5 pairs of pulses 
as in FIG. 3 to magnetically imprint each cycle of the highest audio 
frequency. If the tape speed is 3.75 inches per second then the recorded 
wavelength for the highest audio frequency is 0.0001875 inch. With a head 
gap of one ten-thousandth inch which is typical, and with the tape moving 
a distance between pulses of 7.5 microinches, an element of tape is 
subjected to thirteen pulses while moving across the head gap, confirming 
that even the shortest wavelengths are imprinted by the pulses with fine 
gradations. A waveform as in FIG. 3 of magnetic field intensity sufficient 
to magnetize the tape to saturation may be used to erase the tape if 
symmetrical as shown. A similar wave of equal or less field intensity but 
with pulses unsymmetrical, e.g. with positive pulses progressively 
increasing in amplitude as the alternating negative pulses progressively 
decrease in intensity and vice versa at the highest frequency to be 
recorded, faithful recording will be obtained on the tape according to the 
envelope of dissymmetry. Similarly if a wave such as indicated in FIG. 4 
is used where the pulses are superimposed on a lower frequency wave to be 
recorded, the result will be a magnetization of the tape record medium 
such as represented in FIG. 5. A waveform such as shown in FIG. 3 can be 
generated by differentiating a square wave as by means of a circuit such 
as indicated at 41, 42 in FIG. 2. 
By way of example and not of limitation, the circuit components of FIG. 2 
may be implemented as follows: 
capacitor 41 (500 picofarads); 
resistor 42 (100 ohms); 
resistors 51 and 52 (560 ohms each); 
power field effect transistors 45 and 46 (type 2N3660); 
capacitors 61 and 62 (0.1 microfarad); 
resistors 58 and 59 (750 ohms each). 
The capacitor bypassing secondary winding 74 has a value such that terminal 
22a and circuit point 63 are essentially grounded with respect to the 
pulse signal of FIG. 3, for example. Output monitor 95 may comprise an 
audio transducer and/or a video display unit, or a data signal monitor, or 
the like. 
A different kind of "imprinting" waveform as shown in FIG. 7 comprises a 
damped wave train of extremely high frequency and such a waveform may be 
obtained by shock exciting a resonant circuit. The resonant circuit may 
comprise the head inductance itself with inherent capacitance of its 
wiring or with added capacitance or inductance. The duty cycle here also 
is preferably quite low. A symmetrical (unmodulated) waveform as 
illustrated in FIG. 7 (preferably initiated alternately with positive 
going and negative going half cycles) is advantageous for erasing 
purposes. Offsetting the waveform, for example so that positive cycles 
have higher amplitude, or by superimposing a signal waveform by analogy 
with FIG. 4 will record a magnetization on the tape. The recording 
waveforms such as indicated in FIG. 7 may have only a single cycle, i.e. a 
half cycle positive and a half cycle negative during each recording 
interval. Such double pulses can be obtained from a differentiating 
circuit by exciting the input with pulses such as those in FIG. 3 instead 
of with the square wave. In other words, such a waveform would be the 
second derivative with respect to time of a square wave signal. 
The recording waveform of FIG. 8 is advantageous in comparison to the 
waveform of FIG. 4 since it provides no net magnetic field in the core 10 
during playback. The waveform of FIG. 8 may be generated by the circuit of 
FIG. 2 if the gate electrode of each of the transistors 45, 46 is 
negatively biased enough so that the signal to be recorded (e.g. FIG. 5) 
alone does not produce conduction, but so that the transducers become 
conductive when the pulses of FIG. 3 are superimposed (via transformer 
44). The transistors 45, 46 are normally cut off during operation 
according to FIG. 4 without positive forward bias, and in addition, the 
diodes 48, 49 need about 0.7 volt positive bias before they conduct, so 
the diodes 48, 49 will give operation according to FIG. 8 even without 
negative bias supply when the amplitude of the signal to be recorded (e.g. 
FIG. 5) is kept low enough. 
In general for erasing a record medium with a given saturation level, the 
peak amplitude of each pulse (e.g. for the waveform of FIG. 3) or the peak 
amplitude of the initial half cycle of each pulse in FIG. 7 would provide 
an erasing field reaching or exceeding such saturation level. 
For recording bias, such peak amplitudes in FIGS. 3 and 7 should produce 
bias magnetic fields in the record medium magnetizable layer exceeding the 
coercivity of such magnetizable layer. 
It will be apparent that many modifications and variations may be effected 
without departing from the scope of the novel concepts and teachings of 
the present invention. 
SUPPLEMENTAL DISCUSSION 
In a signal recording operation for FIGS. 1 and 2, without the signal 
correction feature, components 88, 91 and 92 may be omitted. A signal to 
be recorded may comprise an audio frequency signal representing music, 
speech or an analog instrumentation signal or the like having a maximum 
signal frequency component such as indicated in FIG. 5. 
For the case of an alternating polarity pulse recording signal as shown in 
FIG. 8, a maximum signal frequency component such as indicated in FIG. 5 
would produce a first sequence of alternating polarity pulses where the 
positive polarity pulses exceed the negative polarity pulses in amplitude 
in proportion to successive amplitude samples of the positive amplitude 
half cycle of the waveform of FIG. 5. In the region of the zero crossing 
at 5-1 in FIG. 5, successive pulses such as 8-1 and 8-2 would have 
substantially equal amplitude, after which the negative polarity pulses 
would exceed the positive polarity pulses in amplitude in proportion to 
successive amplitude samples of the negative amplitude half cycle of the 
waveform of FIG. 5. For example, the successive pulses at 8-3 (zero 
amplitude) and at 8-4 (maximum negative amplitude) would represent a 
sample at the negative peak region 5-2 of the waveform of FIG. 5. 
As previously explained, the waveform of FIG. 8 may be produced by the 
circuit of FIG. 2, and thus FIG. 8 may represent the alternating polarity 
pulse waveform which is supplied to output terminals 21a, 22a, FIG. 2. The 
head 10 may have an effective frequency response extending to the third 
harmonic of the pulse repetition rate, and preferably to substantially 
higher odd harmonics than the third harmonic, such that the current flow 
through the gap spacer conductive path 14 is represented by a 
discontinuous pulse waveform with relatively short duration pulses of 
alternate polarity. Thus if generator 40, FIG. 2, operates at 250 
kilohertz, the head 10 will have an effective frequency response extending 
at least to 750 kilohertz, and preferably substantially higher, e.g. 
including the seventh harmonic at 1.75 megahertz. The time interval 
between successive pulses of the actual current flow in the gap spacer 
conductive path 14 will substantially exceed the duration of the 
individual pulses of such actual current flow. As previously described the 
duty cycle of the driving waveform at the output terminals 21a, 22a, FIG. 
2, may have a duty cycle of one percent or higher. Where the pulse 
repetition rate is 250 kilohertz, the maximum signal frequency component 
of FIG. 5 may have a frequency of twenty kilohertz, for example. 
DESCRIPTION OF FIGS. 9, 10 AND 11 
In digital recording, it is customary to record a NRZ (non-return to zero) 
signal as shown in FIG. 9. The signal at 9-1 in FIG. 9 may have a duration 
of two clock intervals and represent two binary digits, e.g. two binary 
zeros. Then the signal at 9-2 with a duration of four clock intervals 
would represent four binary ones, and so on. Where FIG. 9 represents the 
amplitude of current flow in a conventional recording head (between a 
positive maximum I+ and a negative maximum I-), FIG. 10 may represent the 
actual recorded magnetization on a channel of a tape record medium. The 
rounded edges as at 10-1 of the waveform are due to resolution limitations 
of the conventional record-playback system. 
FIG. 11 illustrates a pulse method of recording the signal of FIG. 9 
according to the teachings of the present invention. In FIG. 11, the 
pulses 11-1A and 11-1B represent the information at 9-1 in FIG. 9, and the 
pulses 11-2A through 11-2D represent the information at 9-2 in FIG. 9, and 
so on. The pulses such as 11-1A and 11-1B, and 11-2A through 11-2D are 
each of extremely brief duration in comparison to a clock pulse interval 
of the signal of FIG. 9. The pulses of the waveform of FIG. 11 are of such 
short duration that the resistive power loss (I.sup.2 R) in the head is 
only a fraction of that which results from the waveform of FIG. 9. Thus 
the duty cycle of the pulses in FIG. 11 may be a small fraction such as 
one-tenth or less the clock pulse interval of the waveform of FIG. 9. 
In the system of FIG. 11, the record medium moves appreciably between the 
occurrence of successive pulses such as 11-1A and 11-1B, and therefore 
more than one pulse is preferred to represent plural successive binary 
digits of the same value. Thus constant polarity portions of the waveform 
of FIG. 9 of greater than one clock cycle in duration would produce plural 
pulses of the waveform of FIG. 11. 
For high repetition frequencies, efficient heads with close coupling of the 
winding conductor to the record medium magnetizable surface are 
recommended, (such as the heads shown in FIGS. 1, 2, 14 and 15 herein). 
By way of example, a computer with a suitable high clock rate could be 
programmed to produce one pulse (or more) per bit of a word to be recorded 
and to change the polarity of the pulse each time the binary value 
changes. Using separate transformers to activate the transistors 45 and 
46, FIG. 2, the pulses of one polarity could activate a primary winding 
coupled with a secondary winding corresponding to winding 44a in FIG. 2, 
while pulses of opposite polarity would activate a separate primary 
winding coupled with winding corresponding with the winding 44b in FIG. 2. 
For digital recording of this type, components 70, 71, 72, and 73 in FIG. 
2 would be omitted, and components 40, 41, 42 and 44 would be replaced by 
the source of pulses for producing the waveform of FIG. 11 at terminals 21 
and 22, FIG. 2. 
For successful operation with alternating polarity recording and erasing 
waveforms as herein described, the head core must be of low loss 
construction capable of delivering a high magnetic field (exceeding the 
coercivity of the magnetic record medium in amplitude) at frequencies many 
times higher (e.g. at least three times higher) than the pulse repetition 
rate (i.e. the repetition rate of pulses of one polarity). A single 
conductor excitation path, avoiding multiple winding turns, gives a very 
low inductance of the order of fractions of microhenries (i.e. equal to or 
less than one microhenry and for extremely short pulses about one 
nanohenry or even less), and this enables enough current to flow through 
the single conductor path to properly record variable amplitude analog 
signals with substantially a linear transfer function, and to record pulse 
and frequency modulation analog signals and digital signals with an 
amplitude of recorded magnetization approaching the maximum available 
magnetization of the record medium; and to effectively erase higher 
coercivity record media (e.g. having a coercivity in the range from about 
700 oersteds to 100 oersteds or more). The exemplary values for inductance 
and the specification of a head response to frequencies at least three 
times higher than the pulse repetition rate of the excitation pulse 
waveform, are applicable to the record and erase embodiments disclosed 
herein, which utilize the exciting waveforms of FIGS. 3, 4, 8 and 11, 
and/or which utilize head configurations such as shown in FIGS. 1, 2, 14 
and 15. 
An operable waveform of the type shown in FIGS. 3, 4, 8, 11, may have 
pulses with an active time duration of less than one fifth of the time 
period of the maximum signal frequency, and the quiescent time interval 
between successive pulses of opposite polarity may be at least twice the 
time duration of each pulse. 
Certain audio frequency systems using relatively low frequency pulses, may 
have transducer heads with inductances up to about one hundred 
microhenries. 
DESCRIPTION OF FIG. 12 
FIG. 12 illustrates a recording system wherein source 12-1 supplies a NRZ 
signal to be recorded such as that represented in FIG. 9. A clock pulse 
source 12-2 may supply clock pulses at 12-3 to gates 12-4 and 12-5 such 
that gate 12-4 transmits a positive clock pulse when the NRZ waveform is 
at its positive value, and gate 12-5 transmits a positive clock pulse when 
the NRZ waveform is at its negative value. In correspondence with the 
description of FIG. 11, the clock pulses at 12-3 may be occur at the clock 
pulse rate of the NRZ signal and be timed such that two clock pulses occur 
during the time of signal 9-1, FIG. 9, four clock pulses occur during the 
time of signal 9-2, and so on. Where a master sync signal occurs at twice 
the clock rate, one set of clock rate pulses at 12-6 may coincide with 
polarity transitions such as 9-3 and 9-4 in FIG. 9, while an alternate set 
of clock rate pulses may be supplied at 12-3, FIG. 12. 
Positive and negative pulses as shown in FIG. 11 may be generated by 
generator 12-7 which may include two transformers in place of the single 
transformer of FIG. 2, with separate primary windings coupled with 
respective secondary windings such as 44a and 44b, FIG. 2. Components 40, 
41, 42, 44 and 70, 71, 72 and 73 of FIG. 2 would not be required for such 
a generator 12-7. The magnetic recording head 12-8 may correspond with the 
head 10 of FIGS. 1 and 2 or with the head of FIG. 14 or FIG. 15. The 
pulses from the gate 12-4 would be supplied to a primary winding of 
polarity to activate transistor 45, FIG. 2, while pulses from the gate 
12-5 would be supplied to an oppositely wound primary coupled so as to 
trigger transistor 46 via secondary winding 44b. 
DESCRIPTION OF FIG. 13 
FIG. 13 illustrates an erasing system for producing an alternating polarity 
erasing pulse waveform such as shown in FIG. 3 or FIG. 7. Generator 13-1 
may correspond with square wave generator 40 in FIG. 2 and together with 
circuit 41, 42 may produce a pulse rate such that the pulse interval 
between successive pulses corresponds with movement of the record medium 
over a distance small in comparison with the effective longitudinal gap 
dimension of the erasing head 13-2. For example if the effective erase gap 
dimension is ten microns (one micron equals 10.sup.-6 meter), the record 
medium channel may move relative to the erase head a distance of about one 
micron or less in the interval between successive pulses. 
The circuit 13-3 may be formed by the circuit of FIG. 2, with parts 70, 71, 
72 and 73 omitted. 
For high frequencies of the pulses of FIG. 3, the erasing head 13-2 should 
comprise an efficient head configuration with close coupling of the 
winding conductor to the magnetizable surface of the record medium. Such 
head configurations are shown in FIGS. 1, 2, 14 and 15. 
DESCRIPTION OF FIGS. 14 AND 15 
FIGS. 14 and 15 illustrate low inductance head configurations which can be 
used in producing effective pulse erase and record fields corresponding to 
head driving signals such as indicated in FIGS. 3, 4, 7, 8 and 11. 
FIG. 14 shows a head configuration wherein a ring type magnetic core 14-1 
has an electrically conductive winding 14-2 with a first conductor portion 
14-2A extending through a gap in the magnetic core adjacent a record 
medium path 14-3. The configuration of the magnetic core 14-1 at the gap 
or coupling region is similar to that of the magnetic core 10 of FIGS. 1 
and 2. The core 14-1 may comprise two C-shaped core parts which are in 
butting contact at their ends remote from the coupling gap, the ends of 
the core parts defining the coupling gap being located at opposite sides 
of the portion 14-2A of the winding 14-2. The conductor portions 14-2A and 
14-2B of conductor 14-2 are arranged for minimum flux linkage with the 
magnetic core 14-1 and so as to enclose a minimum area at 14-4 between 
portions 14-2A and 14-2B. 
FIG. 15 shows a ring type magnetic core 15-1 corresponding to core 14-1 and 
having a gap region receiving an electrically conductive winding 15-2 of 
"hairpin" configuration adjacent a record medium path 15-3. In this 
embodiment conductor portions 15-2A and 15-2B are both on the same side of 
the record medium path 15-3 as the magnetic core 15-1. The conductor 
portions 15-2A and 15-2B have a minimum spacing therebetween in the plane 
of the coupling gap for minimum flux linkage with the magnetic core. 
By way of example, gap conductors 14, 14-2 and 15-2 may have thicknesses, 
corresponding to the gap dimension, in the range from one-tenth micron to 
about ten microns. Erase gaps generally range up to one hundred microns, 
for example. 
DESCRIPTION OF FIGS. 16, 16A, 16D AND 17 
In an embodiment according to FIGS. 16, 16A, 16D and 17, an analog signal 
to be recorded is supplied to a recording amplifier 16-1 via an input 
terminal 16-2 and produces an amplified signal at A1. A component 16-3 is 
utilized to sense the polarity of the analog signal at input 16-2 and, for 
example, at each pulse of a clock signal C to supply a positive pulse 
signal at output P1 when the analog signal is positive and to supply a 
negative pulse signal at output P2 when the analog signal is of 
instantaneous negative polarity. The component 16-3 may have an input high 
gain amplifier 16-4 which drives an amplitude comparator circuit 16-5. The 
sensitivity of the comparator circuit may be such that it senses the 
polarity of the analog signal with negligible error; for example in the 
case of speech or music signals, the component 16-3 will switch from 
transmitting positive pulses of a predetermined amplitude at output P1 to 
transmitting negative pulses of such predetermined amplitude at output P2 
as the signal amplitude at input 16-2 goes from a positive amplitude value 
which is 0.1% of the maximum positive signal amplitude of the analog 
signal at input 16-2 to a corresponding negative amplitude value. 
Alternatively, the sensitivity can be sufficient to reverse the polarity 
of pulses from component 16-3 in a time corresponding to one clock pulse 
interval of clock signal C, when the analog signal reverses polarity. The 
input amplifier 16-4 of component 16-3 may be overloaded by the input 
analog signal so that its output is compressed or flat as a function of 
time over most of the analog signal duration, becoming sensitive near the 
time when the analog signal is reversing its polarity. 
In one specific embodiment according to FIG. 16A and 17, the comparator 
16-5 may be a window comparator and may control two analog switches of a 
sensitive pulse train generator 16-6, one controlling supply of positive 
pulses to output P1 and the other controlling supply of negative pulses to 
output P2. In the case of an audio input signal at 16-2, with a highest 
frequency component to be recorded of twenty kilohertz, the pulse 
repetition rate of each pulse train should be more than twice this 
frequency as for example 48 kilohertz or higher, and as a preferred 
example, may be 250 kilohertz. The positive and negative pulses may be 
rectangular pulses each with a pulse duration of about forty nanoseconds 
or less, and an interval between pulses of 4000 nanoseconds. The 
synchronizing clock pulse signal C may control the production of the 
positive and negative pulse trains of component 16-3 such that the 
respective positive and negative pulses at the inputs to the respective 
analog switches are time coincident. Alternatively the positive and 
negative pulse trains at the respective analog switch inputs may be 180 
degrees out of phase, with each negative pulse occurring at half intervals 
midway between successive positive pulses at the input side of the analog 
switches; for this example the clock pulses C have an interval of 2000 
nanoseconds. 
At essentially zero signal input level at input 16-2, corresponding to a 
comparator input signal level algebraically greater than Vref-2 and less 
than Vref-1, both analog switches may be blocked so that mixer circuit 
16-10, FIG. 16A, which superimposes the signal from recording amplifier 
16-1 onto the output pulse train of component 16-3 will provide 
essentially a zero output level for a zero input signal at 16-2. Thus, 
transducer head 16-12 of FIG. 16D, which may be connected with output B1, 
FIG. 16A, may receive zero signal current and the corresponding portion of 
the magnetic record medium channel 16-13 remains in a demagnetized state. 
By way of example, the record medium channel 16-13 may move at constant 
speed in the direction of arrow 16-14 successively across longitudinally 
spaced poles of a ring type magnetic core forming part of the head 16-12. 
In one specific embodiment of FIGS. 16, 16A, 16D and 17 previously referred 
to, component 16-3 supplies a series of positive pulses of constant 
amplitude and constant pulse interval during positive polarity portions of 
the input waveform, and these may be superimposed on the instantaneous 
value of the input analog signal by circuit 16-10, for example. Similarly 
during negative excursions of the input analog signal at input 16-2, the 
circuit 16-10 may superimpose constant amplitude constant rate pulses of 
negative polarity on the negative signal portions. The resultant waveform 
at B1 at the output of the circuit 16-10 is indicated in FIG. 17, where 
polarity transition regions such as 17-1 and 17-2 represent only the input 
signal without any superimposed pulses during transition time intervals 
each with a duration of at least two clock intervals of pulses of a given 
polarity, e.g. at least during a transition time interval of 8000 
nanoseconds for the example previously given. 
DESCRIPTION OF FIGS. 16, 16B, 18A, 18B AND 18C 
In a second embodiment according to FIGS. 16, 16B, 18A, 18B and 18C, the 
selective pulse train generator 16-6 may have three active outputs P1, P2 
and P0 for supplying positive pulses, negative pulses and alternating 
polarity pulses, respectively. In this case, the clock input signal C may 
generate an alternating polarity pulse train which is supplied to the 
input of a third analog switch of component 16-6, the output P0 receiving 
this alternating pulse train when the input analog signal at 16-2 has an 
amplitude within a transition region corresponding to the comparator 
amplitude range between Vref-1 and Vref-2. FIG. 18A shows a portion of the 
amplified analog waveform 18-A supplied by recording amplifier 16-1 (on a 
magnified scale) and shows at 18-1 and 18-2 signal amplitude levels 
corresponding to exemplary comparator reference inputs Vref-1 and Vref-2. 
When the amplified analog waveform 18-A has a negative amplitude 
algebraically less than the amplitude value at 18-2, component 16-3 
supplies a negative pulse train at P2 to produce a resultant waveform at 
B2 in FIG. 16B as shown at 18-3 in FIG. 18C. When the amplitude of 
waveform 18-A is between the amplitudes at 18-1 and 18-2, the third analog 
switch becomes transmissive to supply the alternating polarity pulse train 
at P0, FIG. 16, and to produce a resultant waveform a indicated at 18-4 in 
FIG. 18C. Finally, when the amplitude of waveform 18-A exceeds the 
amplitude at 18-1, the positive pulse train is supplied at P1 to produce a 
resultant waveform as shown at 18-5 in FIG. 18C. 
The clock signal C in FIG. 16 may actuate a bistable toggle circuit to 
generate a square wave signal. The square wave signal may be 
differentiated to produce alternate positive and negative pulses which are 
then shaped to the desired alternating polarity rectangular waveform and 
adjusted to desired equal positive and negative amplitude values. The 
positive pulse train for output P1 and the negative pulse train for output 
P2 may be derived from the alternating polarity pulse train by 
rectification to transmit only the positive polarity pulses to the input 
of the analog switch controlling output P1 and to transmit only the 
negative polarity pulses to the input of the analog switch controlling 
output P2. 
In one implementation of component 16-3, FIG. 16, the amplitude comparator 
16-5 may correspond with the window comparator circuit shown in FIG. 3.25 
at page 116 of the text Linear and Interface Circuit Applications by the 
Engineering Staff of Texas Instruments Incorporated, 1974, by way of 
example but not of limitation. In this type of circuit, a first NAND gate 
(C) supplies a zero logic level (ground potential) when the input analog 
signal level (Vout) is equal to or greater than 2.4 volts (corresponding 
to Vref-1 in FIG. 16). In FIG. 3.25, the second reference level is 0.4 
volts, but if this level is set at minus 2.4 volts, then a second NAND 
gate (D) will supply a zero logic level (ground potential) when the input 
signal level (Vout) is less than minus 2.4 volts. Further when the outputs 
of NAND gates (C and D) are both at a high logic level, the output of a 
third NAND gate (E) will be at a zero logic level signifying that the 
input analog signal level (Vout) is in the range between plus and minus 
2.4 volts. In the one specific embodiment with operation as illustrated in 
FIGS. 16A and 17, the first and second NAND gates (C and D) would control 
respective analog switches and the output of the third NAND gate (E) would 
not be used. In operation according to FIGS. 16B and 18C, the outputs of 
the three NAND gates would control three respective analog switches. In 
each case the amplification of the input high gain amplifier 16-4 and/or 
the reference levels Vref-1 and Vref-2 would be adjustable to 
correspondingly select the amplitudes of the polarity transition switching 
points such as indicated at 18-1 and 18-2 in FIG. 18A. 
In a simplified comparator circuit for component 16-3, a pair of matched 
NOR gate circuits which switch e.g. at 2.5 volts could receive 
respectively the output from the input high gain amplifier, and an 
inverted version thereof, so that one NOR gate would switch at plus 2.5 
volts and the other NOR gate would switch with minus 2.5 volts supplied to 
the inverter input. The two NOR gates could control respective inputs of a 
NAND gate corresponding to the third NAND gate (E) previously mentioned so 
as to provide three selective logical zero outputs for controlling three 
analog switches as in the previously described implementation. 
DESCRIPTION OF FIGS. 16, 16C, 19A, 19B AND 19C 
FIG. 16 also illustrates a system for recording wherein an analog signal is 
supplied to input 16-2 but a component 16-21 which may be in series with 
amplifier 16-1 includes a signal sampler circuit operating at the clock 
rate of clock signal C so that the analog signal is converted to a sampled 
signal 19-A where successive pulses such as 19-1, FIG. 19A, accurately 
represent by their amplitude and polarity successive samples of the 
amplified analog waveform such as indicated at 19-2. Component 16-21 also 
includes an input low-pass filter to prevent aliasing by the sampler of 
signal components outside the band of interest. 
The clock pulse signal C supplied to component 16-21, FIG. 16, may control 
the sampling rate of the analog waveform at input 16-2 and may operate at 
a pulse repetition rate more than two times higher than the highest 
frequency component of the analog waveform (corresponding to waveform 
19-2) which is to be recorded. The sampler circuit of component 16-21 may 
control the amplitude of the pulses supplied by a pulse generator for 
example by modulating the amplitude of the pulses supplied by the pulse 
generator linearly in accordance with the amplitude and polarity of the 
instantaneous sample of an amplified analog waveform 19-2. 
As indicated by FIGS. 19A and 19B, where component 16-21 supplies a signal 
pulse waveform 19-A during a polarity transition of the input analog 
waveform at input 16-2, and component 16-3 supplies a composite pulse 
train as shown in FIG. 19B with a positive polarity section 19-B1, an 
alternating polarity section 19-B0 and a negative polarity section 19-B2, 
the resultant signal to be recorded will have a waveform as shown in FIG. 
19C. Thus, for a signal sample more negative than an amplitude level 19-3 
set by threshold input Vref-2, e.g. for signal samples such as 19-1, the 
component 16-3 supplies negative pulses such as 19-6 to generate resultant 
head drive pulses such as 19.9. For signal samples at 19-10 and 19-11 in 
the illustration of FIG. 19A, the amplitude of the signal samples is below 
the threshold 19-3 selected by component 16-3, and the component 16-3 
supplies an alternating polarity pulse group including a negative polarity 
pulse 19-12, and a positive polarity pulse 19-13 which are superimposed on 
signal values as indicated at 19-10 and 19-11; the resultant pulses are 
displaced by the amount of superimposed signal sample as indicated by the 
resultant pulses at 19-16 and 19-17. Where a signal sample in the 
transition region between levels 19-3 and 19-18 has negative polarity but 
coincides with a superimposed pulse of positive polarity, the net 
amplitude of a resultant pulse (e.g. at 19-19 at the time of superimposed 
positive pulse 19-20) may be positive but of a relatively low absolute 
value in comparison to the net negative pulse amplitude of a following 
resultant pulse such as 19-21 at the time of a negative pulse 19-22 from 
component 16-3. The pulse interval between pulses such as 19-19 and 19-21 
is preferably such that the transducer system does not resolve these 
individual pulses. For example the magnetic record medium 16-13 may move 
at such a speed that the net magnetic field pulses due to resultant pulses 
19-21 and 19-23 modify the magnetization produced by resultant pulse 19-19 
to leave a net magnetization substantially linearly proportional to the 
signal value e.g. at 19-24 in the transition region. 
In the next pulse sequence comprising positive polarity pulse 19-30 and 
negative polarity pulse 19-31 with a superimposed signal value as sampled 
at 19-32 and 19-33, the averaging effect on the magnetic record medium 
continues so as to leave a residual magnetization essentially linearly 
proportional to the signal value. Thus, for the pulse sequence 19-30 and 
19-31, the signal value may be essentially zero, and the residual 
magnetization produced by the resultant pulses 19-23 and 19-35 is 
substantially zero. 
Similarly for alternating polarity pulses following pulse 19-31, where a 
positive signal value is superimposed, a residual magnetization linearly 
proportional to the instantaneous signal value is produced on the record 
medium 16-13. A plot of the residual magnetization B.sub.R as a function 
of the successive signal magnetization values resulting from the signal 
samples in the transition region between levels 19-3 and 19-18 would 
substantially conform with the plot of signal waveform at 18-A or 19-2 
during its transition from negative polarity of positive polarity. 
The magnitudes of the negative polarity pedestal pulses e.g. as at 19-6 and 
of the positive polarity pedestal pulses at 19-45, etc., are preferably 
selected at a value such that the analog waveform is recorded with optimum 
fidelity; that is the plot of residual magnetization B.sub.R along the 
record medium channel would conform essentially identically to the sample 
values of FIG. 19A at the corresponding sampling intervals. As an 
alternative, component 16-3 can supply a signal I for blocking the output 
from component 16-21 during amplitude intervals of waveform 19-2 within 
the transition range between negative threshold level 19-3 and positive 
threshold level 19-18, so that only alternating polarity pulses as shown 
at 19-B0 are supplied via output B3, FIG. 16C, to the head 16-12. 
Under some circumstances pedestal pulses from component 16-3 may be omitted 
e.g. by permanently blocking the output from the three analog switches of 
component 16-6. This could be the case, for example, where the recorded 
magnetization after each sampling interval is compared with the input 
signal, and corrected as needed during the next sampling interval to 
provide a linear recording in the high amplitude ranges below negative 
threshold 19-3 and above positive threshold 19-18. In this alternative the 
output at I could disable the sampler circuit of component 16-21 at low 
signal amplitudes corresponding to transition regions as selected by 
component 16-5 so that the analog signal represented by waveform 19-2 for 
example is supplied by circuit 16-1 to the head 16-12 during signal 
polarity transitions (rather than supplying sampled values of the analog 
continuous wave signal). 
Where a pulse generator is omitted in component 16-21, a sampler circuit 
may provide a non-linear amplification of the pulse waveform 19-A such 
that the record medium 16-13 receives residual magnetization linearly 
proportional to each sample of the waveform to be recorded, the high 
magnitude sample pulses being recorded without any superimposed pedestal 
pulse from component 16-3, but the low magnitude sample pulses being 
superimposed on the alternating polarity pulse sequences from component 
16-3 as in the previous examples. 
In certain instances the sampler and pulse generator of component 16-21 may 
be operated at a submultiple of the repetition of the clock signal C. For 
example, if the highest frequency component of signal 19-2 to be recorded 
is twenty kilohertz, the sampling frequency of sampler circuit of 
component 16-21 may be one hundred kilohertz while the clock signal C may 
have a rate of 400 kilohertz or 500 kilohertz such that the time interval 
between successive pulses of alternate polarity is two microseconds or 2.5 
microseconds, the longitudinal coupling gap of head 16-12 having a 
physical dimension of 0.0001 inch and the longitudinal scanning speed 
being 3.75 inches per second as in an example previously given. Thus 
during movement of the record medium a distance equal to the longitudinal 
gap dimension, there might occur 2.67 large signal amplitude sample 
pulses, but more than ten pulses of the alternating polarity pulse train 
during low signal amplitudes, e.g. four or five times as many pulses from 
component 16-3. The rapid alternate polarity pulse train thus sets the 
average value of residual magnetization at intervals when the signal 
amplitude is low, but is absent during high amplitude intervals where it 
is not necessary and would increase power dissipation and heating if 
present. In such an example, the resonant frequency of the head 16-12 
might be at a frequency of six hundred kilohertz or higher. The head 
inductance would preferably be about one microhenry or less. Such very low 
inductance (and impedance) allows the brief duration pulses to fully 
magnetize the record medium efficiently with low power and low heating, 
allowing transducer heads and systems that would be impractical otherwise 
because of heat dissipation problems. 
In one modification of FIG. 16C, analog switches may control the outputs A1 
and A2 and may be controlled by the I signal suppled by component 16-5. 
With higher amplitude signal levels, the I signal may be at a logical one 
level and control the analog switches so that output A2 is supplied to a 
mixer component similar to that of FIG. 16C along with pulse train P1 or 
P2. When, however, the analog input signal at 16-2 is within a low 
amplitude range as defined by reference levels Vref-1 and Vref-2, then the 
analog switch at output A2 may be rendered nontransmissive by a logical 
zero signal level of the I signal, while the output A1 may have its analog 
switch in a transmissive state so that the analog signal is supplied to a 
fifth, A1 input of the mixer. Where the reference levels Vref-1 and Vref-2 
are such that the I signal is at logical zero level only for signal 
amplitudes which are negligible, then the analog switch at output A1 may 
be permanently deenergized, so that all signal information is prevented 
from reaching the mixer when the I signal is at a logical zero level, and 
only pulse train P0 is active. 
In another modification of FIG. 16C, the component 16-1 may include a 
signal sampler and a non-linear amplifier supplying a non-linearly 
amaplified signal to an output (A3, not shown). This non-linear amplifier 
pre-distorts the signal so that it is linearly recorded by the transducer 
system of FIG. 16D without the superposition of pedestal pulses P1 and P2. 
Where an I signal is at a logical zero for negligible amplitudes of the 
input signal at input 16-2, such I signal can control an analog switch at 
the signal output (A3) such that no signal information is transmitted 
while the P0 pulses are being applied to the transducer system of FIG. 
16D. 
With a signal waveform as shown in FIG. 18A or FIG. 19A, the reference 
levels Vref-1 and Vref-2 may be set at relatively high amplitude levels 
such that signal amplitudes outside the window range can be recorded 
without pedestal pulses P1 and P2. In these embodiments, the mixer 
receives the pulse train P0 when low signal amplitudes are present 
requiring superimposed alternating polarity pulses, but outputs P1 and P2 
are not connected to the mixer. In one such embodiment, the inputs to the 
mixer are P0 and A1, while in the other embodiment the inputs to the mixer 
are P0 and A2. In a third embodiment an output (A3) from a non-linear 
amplifier may predistort higher amplitudes for linear recording of such 
higher amplitudes, while the alternating polarity pulse train P0 is 
superimposed on lower amplitude portions of the signal waveform.