Multiple output decoupled synchronous generator and electrical system employing same

A multiple output synchronous generator in accordance with the instant invention comprises a stator having a first stator winding having a number of poles p and a second stator winding having a number of poles 2*p*n where p and n are any whole numbers, the first and second stator windings being wound utilizing a 60.degree. phase belt winding configuration such that the first stator winding is magnetically decoupled from the second stator winding preventing interaction therebetween. The generator additionally comprises a rotor rotatably disposed within the stator defining an airgap therebetween. The rotor has a first field winding of a first polarity and a second field winding having a second number of windings wound thereon for generating a flux across the airgap such that the flux induces a voltage in the first and second stator windings. These first and second field windings are configured such that the flux generated in the airgap comprises a fundamental component and an even harmonic component of substantially equal magnitude. The winding configuration of the stator and rotor insure that only the fundamental flux component and odd harmonics induce a voltage in the first stator winding, and only the even harmonic component and even harmonics induce a voltage in the second stator winding. The magnetic decoupling of the first and second stator windings ensures that harmonic distortion induced in the second output by external equipment powered thereby is not reflected in the first output, thereby maintaining a "clean-bus" for use by utilization equipment sensitive to such distortion.

FIELD OF THE INVENTION 
The instant invention relates to hybrid electric power generating systems, 
and more particularly to a multiple output, decoupled synchronous 
generator providing variable frequency regulated AC and rectified DC power 
outputs from separate armature windings. 
BACKGROUND ART 
As the size and complexity of modern aircraft continue to increase, so to 
the complexity and power demands of the utilization equipment installed 
therein. Conventional electrical power generating systems typically supply 
constant frequency AC power to all of the main distribution buses for use 
by various loads, and rely on separate downstream regulated or unregulated 
transformer/rectifier units (TRUs) to convert a portion of this power to 
DC for use by certain other loads. Since the generators which produce the 
constant frequency AC power are typically driven by the aircraft engines, 
which may be rotating a various speeds depending on the particular portion 
of the flight leg, hydromechanical conversion of the variable engine speed 
to a constant speed to drive the generator or electronic frequency 
conversion of the generator output power itself is required to maintain a 
constant output frequency. These conversion apparatuses, along with the 
downstream TRUs, add weight and complexity to the aircraft, but have been 
necessary to supply the utilization equipment with the constant frequency 
AC, and DC power they demand. 
Future power systems, however, are characterized by an increased amount of 
loads utilizing active power electronics for their power inputs. This 
requires a substantial amount of filtering on the electrical system to 
remove harmonics induced by these loads to meet present power quality 
requirements. To minimize this trend's impact on the electrical power 
generating system, aircraft manufacturers are looking to generation and 
distribution architectures which will allow segregated loads to be 
supplied by variable frequency (VF) power as illustrated in FIG. 1. 
However, with the perspective of low, or no, bleed air from future 
aircraft engines, more electrical power will be required to power loads 
such as the environmental control system (ECS), which will increase still 
further the non-linear loads' percentage of the total installed load. This 
will impact the electrical generating system power quality still further. 
Maintaining VF power quality at the same level as the current constant 
frequency (CF) systems will lead to increased size, weight, and cost, and 
lower reliability for any filtering technique utilized, whether active or 
passive. 
One of the architectures that is being considered for new electric power 
generating systems for future more electric airplanes (MEAs) is the dual 
"Clean-bus"/"Dirty-bus" architecture as shown in FIG. 2. The "clean-bus" 
is a voltage regulated variable frequency bus targeted for power quality 
sensitive loads. The "dirty-bus", on the other hand, is a nonregulated bus 
providing power to the non-linear load(s) on the system. The "dirty-bus" 
can be either dedicated to a single load or can provide power to a series 
of compatible loads. The conventional single output generator is not very 
compatible with this bus architecture because it requires substantial 
amounts of filtering to meet power quality requirements similar to 
constant frequency systems on the "clean-bus" due to the induced harmonics 
from the nonlinear loads. 
To reduce the amount of filtering, a conventional dual output generator 
with a split armature, such as described by R.F. Schiferl in "Six Phase 
Synchronous Machine with AC and DC Stator Connections" IEEE Transactions 
on Power Apparatus and Systems, Vol. 102, No. 8, August 1983, can be used. 
The two outputs of this conventional generator are magnetically coupled 
because they share the same fundamental flux. This configuration, however, 
still requires substantial filtering to remove the harmonics induced on 
the "clean-bus" winding from the active electronic rectification of the 
"dirty-bus" winding's output because of the magnetic coupling 
therebetween. Furthermore, with a single control loop on the "Clean-bus", 
the "dirty-bus" voltage range is much wider which then requires the 
utilization equipment to provide appropriate compensation, leading to 
increased cost and complexity downstream of the electrical system. 
The instant invention is directed at overcoming one or more of the above 
problems present in these conventional modern systems. 
SUMMARY OF THE INVENTION 
It is a principle objective of the instant invention to provide a new and 
improved dual output/dual frequency generator. More specifically it is the 
object of the instant invention to provide a dual output/dual frequency 
which will exhibit characteristics useful for a dual-bus architecture to 
reduce or eliminate filtering requirements. 
In a preferred embodiment of the instant invention, the multiple output 
synchronous generator comprises a stator having a first and a second 
stator winding wound thereon utilizing a pattern such that the first 
stator winding is magnetically decoupled from the second stator winding 
preventing interaction therebetween. The first stator winding has a first 
number of poles p and the second stator winding has a second number of 
poles 2*p*n, where p and n are whole numbers. In a preferred embodiment 
the first stator winding has 4 poles and the second has 8 poles. The 
rotor, which is rotatably disposed within the stator, defines an airgap 
between its outer periphery and the stator. The rotor has a first and a 
second field winding wound thereon for generating a flux across this 
airgap such that it induces a voltage in the first and second stator 
windings. 
In a highly preferred embodiment of the instant invention, the winding 
pattern of the stator windings is a 60.degree. phase belt winding 
configuration. The first and said second field windings of the rotor are 
configured such that the flux generated in the airgap comprises a 
fundamental component and a first even harmonic component of substantially 
equal magnitude. Based on the relationship of the pole number and the 
first stator winding configuration, only the fundamental flux component 
and the odd harmonics induce a voltage in the first stator winding. This 
same relationship ensures that only even harmonic component and even 
harmonics induce a voltage in the second stator winding. 
In a preferred embodiment of the instant invention, the first field winding 
and the second field winding are energized by a single source of 
electrical energy controlled by a controller to regulate the first 
generator output. Alternatively, the first field winding and the second 
field winding are energized by independent sources of electrical energy. 
The regulation of these sources of electrical energy may be controlled by 
a single or multiple controllers and allow for independent regulation of 
the output of the first and the second stator windings.

DESCRIPTION OF THE PREFERRED EMBODIMENTS 
In a preferred embodiment of the instant invention, as illustrated in FIG. 
3, the dual output/dual frequency generator 100 comprises an armature 102 
and a main field 104 having 2 sets of windings 106a-c and 108, and 110x-z 
and 112, with a different number of poles. The two sets of main field 
windings 108 and 112 are supplied from the same source of DC current, as 
illustrated in FIG. 3, a rotating rectifier 118 which rectifies the output 
from the rotating exciter armature windings 120a-c when the exciter field 
winding 122 is energized. This section of the generator 100 is illustrated 
as being separate only because its specific structure is not important to 
the preferred embodiment of the instant invention. Other structures which 
provide DC current for excitation of the two main field windings 106a-c 
and 110x-z are applicable as may be appropriate. 
In an alternate embodiment of the instant invention, as illustrated in FIG. 
4, the dual output/dual frequency generator 100 comprises an armature 102 
and a main field 104 having 2 sets of windings 106a-c and 108, and 110x-z 
and 112, with a different number of poles. The two sets of main field 
windings 108 and 112 are supplied from different sources of DC current, as 
illustrated in FIG. 4. The first source comprises a rotating rectifier 129 
which rectifies the output from the rotating exciter armature windings 
125a-c when the exciter field winding 121 is energized. The second source 
comprises a rotating rectifier 131 which rectifies the output from the 
rotating exciter armature windings 127a-c when the exciter field winding 
123 is energized. This section of the generator 100 is illustrated as 
being separate only because its specific structure is not important to the 
preferred embodiment of the instant invention. Other structures which 
provide DC current for excitation of the two main field windings 106a-c 
and 110x are applicable as may be appropriate. 
The preferred embodiment of the instant invention combines judicious 
selection of the relationship between the pole numbers (p vs 2*p*n where p 
and n are any numbers) combined with an adequate selection of the main 
stator winding, 106a-c and 110x-z, and rotor winding, 108 and 112, 
configuration, which leads to a reduction or elimination of harmonic 
coupling between the low frequency outputs (hereinafter LF) 114a-c and 
114n, and high frequency outputs (hereinafter HF) 116x-z of the generator 
100. The Table below shows a list of preferred pole combinations: 
______________________________________ 
Low frequency output (LF) 
Possible High frequency output (HF) number 
Number of poles (P) 
of poles (2 .times. P .times. n) 
______________________________________ 
2 4, 8, 12, 16 . . . 
4 8, 16, 24 . . . 
6 12, 24 . . . 
8 16 . . . 
10 20 . . . 
12 24 . . . 
______________________________________ 
In addition and alternate to the preferred pole combinations listed above, 
the number of poles of the low frequency output and the number of poles of 
the high frequency output need not be related by the preferred p vs. 2*n*p 
relationship, however such alternate relationships may suffer from 
increased interaction between the windings. Such other interactions can be 
compensated and may not present a problem for specific applications. 
The preferred embodiment of the instant invention utilizes a number of 
poles of 4 for the low frequency windings 106a-c, and a number of poles of 
8 for the high frequency windings 110x-z. As used in an electric power 
generation system, the preferred embodiment further comprises a means 150 
for actively controlling the output magnitude of the low frequency 
windings 106a-c by sensing the output 114a-c and adjusting the excitation 
of the exciter field 122 to maintain the output magnitude at a 
predetermined level, regardless of connected load. This means of 
controlling the output magnitude may be a conventional pulse width 
modulated (PWM) exciter field control utilizing feedback control, or may 
be a conventional series voltage regulator controller, or other topology 
voltage regulator which allows control of the output magnitude by 
adjusting the excitation level of the exciter field. In the preferred 
system, the output 116x-z is unregulated. An alternate embodiment 
comprises a means 151 for actively controlling the output magnitude of the 
low frequency (LF) windings 106a-c by sensing the output 114a-c and 
adjusting the excitation of the exciter field 121 to maintain the output 
magnitude at a predetermined level, regardless of connected load. 
Additionally, the means 151 actively controls the output magnitude of the 
high frequency (HF) windings 110x-z by sensing the output 16x-z and 
adjusting the excitation of the exciter field 123 to maintain the output 
magnitude at a predetermined level, regardless of connected load. 
The main field lamination and winding is designed to create a flux 
distribution in the airgap that has a fundamental component (p-pole 
component) and its first even (2*n) harmonic (2*n pole component) of 
substantially the same magnitude. The p-pole component generates voltage 
on the low frequency output 114a-c, 114n and the 2*n*p-pole component 
generates voltage in the high frequency output 116x-z. In the preferred 
embodiment, a cylindrical rotor configuration with distributed field 
windings is utilized to generate this complex and unusual flux 
distribution. 
This configuration for the preferred embodiment corresponds to the magnetic 
axis of the 8-pole and 4-pole being aligned as illustrated in FIG. 5, 
which also illustrates the rotor slots and winding distributions. As can 
be seen from FIG. 5, the resultant air gap flux can be generated with a 
cylindrical rotor configuration with no geometric saliency. The flux 
distribution shows an apparent saliency that is created by the 4-pole MMFs 
124 and the 8-pole MMF 126 canceling each other. With a smooth airgap, the 
resultant flux distribution 128 is similar to a field MMF distribution 
having its fundamental and second harmonic of substantially the same 
magnitude. 
Assuming a smooth airgap, the inverse of the air-gap length g.sup.-1 as a 
function of the mechanical angle .PSI. is constant: 
EQU g.sup.-1 (.theta..sub.r,.psi.)=1/g 
Where: 
.PSI. the mechanical angle in the stator reference. 
.theta..sub.r angular position of the rotor relative to a stator reference 
g constant airgap. 
Neglecting saturation, the airgap flux density established by the field 
winding MMF at the space angle .PSI. can be resolved into a Fourier series 
as follows: 
##EQU1## 
Where: p is the number of poles of the LF output. 
B.sub.v is magnitude of v th harmonic. 
The predominant terms of this Fourier series are the fundamental and its 
second harmonic. 
The winding functions of phase a of the LF output (N.sub.a) 106a and phase 
x of the HF output (N.sub.x) 110x respectively can be resolved into 
Fourier series as follows: 
##EQU2## 
Where N.sub.a,v and N.sub.x,v are proportional to the harmonic winding 
factors K.sub.p K.sub.d,v. 
The flux linkage in phase a 106a due to the airgap flux density 
distribution, can be written as follows: 
##EQU3## 
Where: D.sub.r is the armature inside diameter 
L is the generator effective stack length 
Once finctions N.sub.a and B.sub.g are replaced by their Fourier series and 
the expression developed, the only remaining terms are the ones with the 
same pulsation (v.P/2): 
##EQU4## 
As can be seen from the above expression, for phase a voltage to be 
unaffected by the even harmonic fluxes generated by the 2*n*p-pole field 
winding 112, it is necessary that N.sub.a,v =0 with v an even harmonic. 
Therefore, for the LF output 114a-c to be uncoupled from the HF output 
116x-z field flux, an integral slot per pole as well as a 60.degree. phase 
belt winding are required. 
On the other hand, the flux linkage in phase x 110x can be written as 
follows: 
##EQU5## 
Once the above expression is developed, the only remaining terms are the 
ones that verify the expression: 
EQU v.multidot.p/2=h.multidot.n.multidot.p 
Or 
EQU v=2.multidot.h.multidot.n 
Therefore, 
##EQU6## 
From the above equation, it can be seen that only the even harmonics 
generate voltage in the 2xpxn pole windings 110x-z. 
When balanced currents are drawn from the armature windings 106a-c and 
110x-z, fluxes are generated by the distributed armature MMFs. These 
armature reaction fluxes interact with the field windings 108 and 112. If 
a balanced set of sine-wave currents of frequency w is drawn from the LF 
output 114a-c and a set of balanced currents of frequency 2*n*.omega. is 
drawn from the HF output 116x-z, the total armature MMFs are as follows: 
##EQU7## 
where v is an odd harmonic. 
Neglecting magnetic saturation, the airgap flux density waves resulting 
from the above MMF waves can be found as follows: 
EQU kB.sub.abc (.psi.)=.mu..sub.o .multidot.g.sup.-1 (.psi.).multidot.F.sub.abc 
(.psi.) 
and 
EQU BK.sub.xyz (.psi.)=.mu..sub.o .multidot.g.sup.-1 (.psi.).multidot.F.sub.xyz 
(.psi.) 
The total flux density at a space angle .PSI. is: 
EQU B(.psi.)=B.sub.g (.psi.)+B.sub.abc (.psi.)+B.sub.xyz (.psi.) 
The flux linkage in phase .alpha. 106a of the LF output due to this airgap 
flux distribution can be expressed as: 
##EQU8## 
Or when developed: 
##EQU9## 
The flux linkage in phase a 106a due to currents in the HF output phases 
110x-z is expressed in the last term of the above expression. This term 
can be developed as follows: 
##EQU10## 
Once the above integral expression is developed, the only remaining terms 
are the ones that verify the relation: 
EQU .nu..multidot.p/2=.gamma..multidot.n.multidot.p 
or 
EQU .nu.=2.multidot.n.multidot..cuberoot. 
Which is not possible because .nu. is an odd harmonic. Therefore, if the 
generator has an integral slot per pole per phase and the low frequency 
output winding is 60.degree. phase belt, the above integral is zero. 
Consequently, no coupling through the airgap flux will take place due to 
these high frequency output fundamental currents. 
If a set of symmetrical harmonic currents of frequency 2*n*.omega.*k are 
drawn from the high frequency output 116x-z, the total armature MMF can be 
expressed as follows: 
##EQU11## 
Neglecting saturation, the airgap flux density wave resulting from the 
above MMF wave is as follows: 
EQU B.sub.xyz (.psi.)=.mu..sub.o .multidot.g.sup.-1 (.psi.).multidot.F.sub.xyz 
(.psi.) 
The flux linkage in phase a 106a of the low frequency output can be 
expressed as follows: 
##EQU12## 
The flux linkage in a-phase 106a due to the harmonic currents in the high 
frequency output 116x-z is expressed in the last term of the above 
expression. This term can be developed as follows: 
##EQU13## 
As for the fundamental currents, once the above integral expression is 
developed, the only remaining terms are the ones that verify the relation: 
EQU .nu.=2.multidot.n.multidot..gamma. 
Which is not possible because .nu. is an odd harmonic. Therefore, as for 
the fundamental, the last integral term is zero. Consequently, no harmonic 
coupling through the airgap flux takes place due to harmonic current drawn 
from the high frequency output 116x-z. 
In the above analysis, a smooth airgap was assumed, which is a good 
approximation for a cylindrical rotor geometry. In reality, the rotor slot 
openings will introduce harmonics in the airgap permeance wave that may 
affect the interaction between the windings under load conditions. 
Introduction of the airgap permeance variation complicates the theoretical 
development considerably. Therefore, to present a clearer description of 
the effect of space harmonics, only the fundamental winding functions 
combined with a discrete inverse gap function will be considered. With the 
rotor lamination design 130 of the preferred embodiment (4-pole/8-pole for 
example) as illustrated in FIG. 6, the inverse gap function is assumed 
constant at the pole face 132 and having a lower value at the slot opening 
134. 
The inverse airgap function can be written as follows: 
##EQU14## 
Where g.sub.2 is constant airgap over pole face 132. 
g.sub.z is related to the size of the slot opening 134 and rotor slot tip 
saturation level (g.sub.2 =g.sub.o x Carter Coefficient). 
n=0, 1, 2 . . . (p-1) 
The coupling between the two sets of windings 108 and 112 is expressed in 
the mutual inductance expressed as: 
##EQU15## 
Taking only the fundamental terms of the winding functions, this 
expression becomes: 
##EQU16## 
Once the above expression is fully developed, the mutual inductance can be 
written as: 
##EQU17## 
It can be shown, that for any n value (n=1 for 4/8 pole configuration), 
the cosine sums are all zero. This is due to the fact that the terms: 
##EQU18## 
are non even numbers. Therefore, rotor slotting will have no effect on the 
mutual inductance between the two output windings through the modulation 
of the airgap useful flux by the airgap permeance. 
The two output windings abc 106a-c and xyz 110x-z share the same slots 138 
on the stator 136. One of the windings 110 occupies the top of the slot, 
the other winding 106 occupies the bottom of the slot as illustrated in 
FIG. 7. Magnetic coupling can occur when magnetic flux which does not 
cross the machine airgap 140 couples the two stator windings of different 
three phase sets. The resultant mutual coupling can be characterized by 
leakage flux linkage .lambda..sub.1 which, for phase a, can be expressed 
as: 
EQU .lambda..sub.1,a =L.sub.1,ax .multidot.i.sub.x +L.sub.1,ay .multidot..sub.y 
+L.sub.1,az .multidot.i.sub.z 
Where L.sub.1,ax, L.sub.1,ay, L.sub.1,az are the mutual leakage inductances 
between phase a 106a and phases x, y, and z 110x-z respectively. 
These three mutual inductances are function of winding configuration and 
winding pitch. For the two outputs abc 114a-c, and xyz 116x-z to be 
uncoupled, the leakage flux linkages .lambda..sub.1,a,b,c,x,y,z! must be 
zero. Assuming a 4 layer winding as illustrated in FIG. 7, with arbitrary 
phase displacement between the two sets 106 and 110 and different winding 
pitches, the mutual leakage fluxes may not cancel out. The slot reactance 
of this 4 layer winding are characterized by 10 individual components as: 
X.sub.1, X.sub.2, X.sub.3, X.sub.4 
X.sub.12 /X.sub.21 
X.sub.13 /X.sub.31 
X.sub.14 /X.sub.41 
X.sub.23 /X.sub.32 
X.sub.24 /X.sub.42 
X.sub.34 /X.sub.43 
Where 
X.sub.n{1,2,3,4} =Slot leakage reactance due to coil side in position 1, 2, 
3 or 4 respectively 
X.sub.mn =Mutual slot leakage reactances between layers 1, 2, 3 or 4 
FIG. 8a illustrates a 24 slot armature (1 slot/pole/phase for 8 pole and 2 
slots/pole/phase for the 4-pole) utilizing a 60.degree. phase belt full 
pitch winding distribution. For this armature, the mutual leakage flux 
linkage in phase a 106a is: 
Slots 1/2 & 13/14: 2.(X.sub.13 +X.sub.14 +X.sub.23 +X.sub.24).i.sub.x 
2.(X.sub.13 +X.sub.14 +X.sub.23 +X.sub.24).i.sub.y 
Slots 7/8 & 19/20: -2.(X.sub.13 +X.sub.14 +X.sub.23 +X.sub.24).i.sub.x 
+2.(X.sub.13 +X.sub.14 +X.sub.23 +X.sub.24).i.sub.y 
The total mutual leakage is the sum of the above two expressions. It is 
clear, therefore, that the mutual flux linkage in phase a 106a cancels out 
over 2 pair poles. The same result applies to phases b 106b and c 106c. 
If the above abc winding 106 is pitched one slot as illustrated in FIG. 8b, 
the mutual leakage flux linkage in phase a 106a becomes: 
Slots 1/2/23/24 & 11/12/13/14: 
EQU 2.(X.sub.14 +X.sub.24 -X.sub.13 -X.sub.23).i.sub.y -2.(X.sub.14 
+X.sub.24)i.sub.z +2.(X.sub.13 +X.sub.23),i.sub.x 
Slots 5/6/7/8 & 17/18/19/20: 
EQU 2.(X.sub.13 +X.sub.23 -X.sub.14 -X.sub.24).i.sub.y +2.(X.sub.14 
+X.sub.24).i.sub.z =2.(X.sub.13 +X.sub.23).i.sub.x 
As in the full pitch case, the mutual flux linkage in phase a 106a cancels 
out. The same result applies to phases b 106b and c 106c. 
The same result as above applies if an arbitrary phase shift is introduced 
between the phase sets as illustrated in FIG. 8c and FIG. 8d. 
The 4 pole winding 106 requires that the machine slot number be a multiple 
of 12 or: 
Generator number of slots=k.times.12 with k=2, 3 . . . 
The slot/pole/phase (SPP) for the 8 pole winding 110 are then: 
EQU SPP.sub.SP =k.times.12/3/8=k/2. 
Therefore, the allowed SPPs for the 8-pole winding 110 are: 
EQU 1, 1.5, 2., 2.5 . . . or 24,36,48,60,72,84 . . . slots 
When the same approach as above is applied to the next allowed 
configuration, 36 slots, the same result is obtained both at full and 2/3 
pitches as illustrated in FIG. 8e and FIG. 8f The mutual leakage fluxes 
seen by phases a, b, c 106a-c cancel out. 
The mutual leakage flux linkage taking place within the slot in this 4 
layer winding configuration of the preferred embodiment is zero. 
Consequently, there is no mutual coupling between the two sets of windings 
106 and 110 due to the slot related leakage flux. This result can be 
extrapolated toppole and 2*n*p-pole generator configuration. Additionally, 
layer winding configurations other than the preferred 4 layer winding 
configuration may be appropriate in a particular application, the 
particulars of which are apparent to one skilled in the art from the 
foregoing. 
The non-slot related leakage mutual fluxes including end-turns, zig-zag and 
belt leakages can be reduced by keeping the stator turns to a minimum, 
and/ or increase the airgap size. The reduced magnetic coupling between 
the low frequency and high frequency outputs is valid as long as a 
symmetry between the poles and coils location is maintained. With any 
non-symmetrical features such as slot location and airgap, additional 
harmonics can appear in the winding functions and airgap permeance and 
some coupling may take place. 
The fluxes generating voltages on the two winding sets share the same 
magnetic circuit. Saturation of the magnetic circuit changes the apparent 
reluctance of the circuit affecting the magnitudes and shape of the airgap 
permeance and consequently, the airgap flux distribution. Therefore, 
change in magnitude and harmonic content of both output voltages can be 
expected if saturation is present. 
Numerous modifications and alternative embodiments of the invention will be 
apparent to those skilled in the art in view of the foregoing description. 
Accordingly, this description is to be construed as illustrative only and 
is for the purpose of teaching those skilled in the art the best mode of 
carrying out the invention. The details of the structure may be varied 
substantially without departing from the spirit of the invention, and the 
exclusive use of all modifications which come within the scope of the 
appended claims is reserved.