Radio receiver for demodulating both wideband and narrowband modulated signals

A radio receiver is disclosed for demodulating both wideband and narrowband frequency-modulated signals. The radio receiver improves the capacity of a system by reducing the bandwidth occupancy of analog frequency modulation systems. The radio receiver comprises a superheterodyne frequency device which converts the received frequency-modulated signal to a fixed intermediate frequency signal. A filter then filters the intermediate frequency signal with a first filter bandwidth adapted to a wideband frequency-modulated signal. The filtered wideband frequency-modulated signal is demodulated in a demodulator using a frequency discriminator. The intermediate frequency signal is then filtered using a second and third filter bandwidth adapted to a narrowband frequency-modulated signal. Finally, the further-filtered intermediate frequency signal is demodulated using a frequency discriminator.

FIELD OF THE INVENTION 
The present invention relates to adaptive analog frequency modulation (FM) 
for transmitting speech in radiotelephone communication systems, and more 
particularly to analog FM cellular receivers having bandwidths adapted so 
as to optimize average demodulated voice quality. 
BACKGROUND OF THE INVENTION 
At the present time, cellular mobile telephone systems use analog frequency 
modulation to transmit speech. The three principal cellular mobile 
communication system standards are the AMPS system used in the United 
States which uses wideband frequency modulation with a spacing between 
channels of 30 KHz, the TACS system used in the United Kingdom which uses 
25 KHz channel spacings, and the NMT system used in Scandinavia which uses 
narrowband frequency modulation with 12.5 KHz channel spacings. 
In an effort to alleviate the capacity restrictions of the current analog 
frequency modulation systems, digital transmission has been standardized 
for future systems in Europe, the U.S.A., and Japan. Nevertheless, digital 
transmission standards are complicated and not suitable for use 
everywhere. As a result, it is an object of the present invention to 
implement capacity improvements by reducing the bandwidth occupancy of 
analog frequency modulation systems. 
Another approach to increasing system capacity by reducing bandwidth 
requirements is disclosed in a narrowband FM system according to the NAMPS 
specification. In the NAMPS system, a channel spacing of 10 KHz is 
achieved by splitting each 30-KHz channel of AMPS system into three 
sections. To accommodate the signal within the reduced bandwidth, both the 
frequency deviation, or modulation index, of the transmitted signal and 
the receiver's bandwidth are reduced. 
Various aspects of frequency modulation communication systems are set forth 
in the prior art, including H. Taub et al., Principles of Communication 
Systems, chapt. 4, McGraw-Hill Book Co., New York (1971). For a sinusoidal 
modulating signal, the bandwidth B required to transmit or receive an FM 
signal with at least 98% power transmission is given by the following 
expression of Carson's rule: 
EQU B=2(.DELTA.f+f.sub.m) 
where .DELTA.f is the maximum frequency deviation of the instantaneous 
frequency of the FM signal from the carrier frequency and f.sub.m is the 
frequency of the sinusoidal modulating frequency. The modulation index 
.beta. is related to .DELTA.f and f.sub.m by the following expression: 
EQU .beta.=.DELTA.f/f.sub.m 
When the modulation index is dominant, i.e., when .DELTA.ff.sub.m, the 
bandwidth according to Carson's rule is reduced proportional to the 
frequency deviation .DELTA.f. Thus, reducing the frequency deviation 
results in a commensurate reduction in the bandwidth. 
For narrowband FM systems in which the modulation index is small, i.e., 
when .DELTA.f.ltoreq.f.sub.m, reductions in the frequency deviation 
.DELTA.f do not result in commensurate reductions in the bandwidth B. 
Thus, the level of the desired modulating signal transported by the 
modulation system decreases faster than the noise passed by the receiver 
as the frequency deviation decreases, and a worsening signal-to-noise 
ratio results. 
To overcome this problem, bandwidths smaller than the Carson's-rule 
bandwidth can be used when the noise level is high. A smaller bandwidth, 
however, causes distortion in the demodulated signal since more energy in 
the sidebands of the FM signal is being discarded, but this is preferable 
to passing more noise when the noise level is high. When the noise level 
is low, however, the demodulated signal quality is limited by the 
distortion components, and it is desirable to increase the bandwidth. 
Since, in practice, signal levels received by a mobile phone fade up and 
down due to motion and other effects, it is apparent that the bandwidth of 
a narrowband FM receiver preferably would adapt to those effects by 
varying continuously. 
In addition, when signals on adjacent channels are strong, a narrower 
bandwidth that suppresses the adjacent-channel signals more may be better 
than a wider bandwidth that avoids distortion. When adjacent channel 
signals are weak or absent, however, a wider bandwidth that avoids 
distortion is preferable. Because both the levels of the desired signal 
and the adjacent-channel signals fade up and down in an uncorrelated 
fashion, their ratio can vary over a wide range, again indicating that an 
adaptive bandwidth can be advantageous. 
U.S. Pat. No. 4,352,208 to Schroeder describes a microprocessor-controlled 
radio receiver system for automatically switching the bandwidth of an 
intermediate frequency (IF) stage between narrow and wide values. The 
narrow bandwidth is used in a normal mode of operation in which the 
receiver scans several channels. The wide bandwidth is used in a second 
mode in which a channel has been selected and there are no interfering 
signals on adjacent channels. Upon selection of a channel in the second 
mode, the microprocessor periodically causes a frequency synthesizer to 
scan up one channel and down one channel to determine if any interfering 
signal is present and then to return to the selected channel. If no 
interfering signal is present on an adjacent channel, the IF stage is 
switched from a narrow bandwidth to a wide bandwidth to improve the 
quality of reception. If an adjacent channel signal of sufficient IF 
energy is present, the IF stage remains in its narrow-bandwidth mode. 
The system described in the Schroeder patent is unusable in a communication 
system, such as a mobile radio telephone system, in which interference 
situations are continuously changing. In such systems, it is undesirable, 
if not impossible, to permit even a temporary loss of the signal on the 
selected channel while the receiver checks the adjacent channels for 
interference. The present invention provides continuous comparison of 
in-channel and out-of-channel energy without continuously retuning the 
receiver. 
U.S. Pat. No. 4,124,817 to Takahashi discloses a bandwidth switching 
circuit for an intermediate frequency amplifier stage in an FM receiver 
that insures clear reception of desired signals by automatically switching 
the intermediate frequency amplifier stage between the wide and narrow 
bandwidths according to the radio field conditions. The bandwidth 
switching circuit includes a detector for detecting beat components due to 
interference contained in received signals and a change-over switch for 
switching the bandwidth switching circuit according to signals detected by 
the detector, whereby the bandwidth of the intermediate frequency 
amplifier stage is automatically switched depending on whether the beat 
components are present or not. 
SUMMARY OF THE INVENTION 
It is an object of the present invention to provide improved capacity by 
reducing the bandwidth occupancy of analog frequency modulation systems. 
In particular, it is an object of the present invention to provide an 
implementation of a receiver that gives improved performance using 10 KHz 
channel-spaced narrowband frequency modulation. 
The present invention concerns the provision of dynamically variable 
bandwidth in either AMPS mode or NAMPS mode in order to obtain the best 
compromise between distortion and noise or interference suppression. A 
preferred embodiment of the present invention allows the bandwidth to be 
dynamically varied between the 30 KHz value needed for AMPS and the 10 KHz 
value needed for NAMPS, thus realizing a dual-mode receiver. 
In the present invention, a radio receiver is disclosed for demodulating 
both wideband and narrowband frequency-modulated signals. The radio 
receiver comprises a superheterodyne frequency means which converts the 
received frequency-modulated signal to a fixed intermediate frequency 
signal. A filter then filters the intermediate frequency signal with a 
first filter bandwidth adapted to a wideband frequency-modulated signal. 
The filtered wideband frequency-modulated signal is then demodulated in a 
demodulator using a frequency discriminator. The intermediate frequency 
signal is then filtered using a second and third filter bandwidth adapted 
to a narrowband frequency-modulated signal. Finally, the further-filtered 
intermediate frequency signal is demodulated using a frequency 
discriminator.

DETAILED DESCRIPTION 
FIG. 1 illustrates a block diagram of a conventional FM receiver 5 suitable 
for conforming to either the AMPS or NAMPS cellular mobile radiotelephone 
specifications, with the appropriate choice of intermediate frequency (IF) 
bandwidth of IF filters 50. 
The received signal passes from the antenna through a transmit/receive 
duplex filter 10, a low-noise amplifier 20, an image rejection filter 30 
and downconverter 40, such as a superheterodyne frequency device, where 
the received signal is converted to a suitable intermediate frequency 
(IF). The IF filters 50 impose the main channel bandwidth restriction to 
just less than 30 KHz for the AMPS specification, or around 10 KHz for the 
NAMPS specification. The IF amplifiers 60 provide most of the 
amplification and also generate a received signal strength indication 
(RSSI). It is common practice for all of the IF amplifiers 60 to be 
contained in a single integrated circuit, and for a second frequency 
downconversion to be employed part way through the amplification process 
ahead of the second IF filter 50, to permit the use of small, low-cost, 
ceramic filters. In addition, it is also easier to avoid unwanted 
oscillations due to stray feedback if the total amplification is split 
between two frequencies. 
The frequency discriminator 70 operates at the final intermediate frequency 
(IF.sub.final) and produces a signal output proportional to the 
instantaneous frequency deviation of the radio signal from its nominal 
center frequency, which is a facsimile of the speech signal used to 
frequency modulate the transmitter. The discriminator output signal is 
band-limited to the 300 Hz to 3.4 KHz speech frequency range in a 
de-emphasis filter 80 to exclude as much noise as possible. It is also 
well-known that the noise output of a frequency discriminator increases 
across the audio frequency band, so that the use of pre-emphasis at the 
transmitter with compensating de-emphasis in the de-emphasis filter 80 at 
the receiver 5 improves the signal-to-noise ratio by applying most of the 
attenuation to the highest noise components. It is also well-known that 
the perceived quality of speech has more to do with the background noise 
during silent or quiet periods than the signal-to-noise ratio during loud 
periods, so that the use of decompanding or expansion in a decompander 90 
at the receiver 5, which magnifies the disparity between loud and quiet 
signals, with compensating companding at the transmitter, leaves the voice 
signal unchanged while reducing receiver noise in the quiet periods. The 
output of decompander 90 is then fed to the telephone earpiece. 
In the conventional, known FM receiver illustrated in FIG. 1, the 
bandwidths of the IF filter 50 are fixed, since available analog filter 
technology does not easily permit the construction of filters of 
dynamically adjustable bandwidth. To construct a receiver suitable for 
operation either in the AMPS or NAMPS systems, a so-called dual-mode 
receiver would need to contain selectable 30 KHz wide AMPS filters and 10 
KHz wide NAMPS filters. Both of these filters however would, in the 
conventional solution, have constant bandwidths. 
A preferred implementation of the dynamically variable bandwidth receiver 
of the present invention is illustrated in FIGS. 2 and 3. The dynamically 
variable bandwidth receiver begins with a conventional downconversion to a 
suitable first intermediate frequency (IF) as in FIG. 1, using a 
transmit/receive duplex filter 10, a low-noise amplifier 20, an image 
rejection filter 30, and a downconverter 40. Filters 100 and 110 impose a 
channel bandwidth in the 30 KHz region for the AMPS system. The 
intermediate frequency amplifier circuit 120 is a silicon integrated 
circuit which also contains a second downconversion at a downconverter 45. 
Instead of a frequency discriminator, the hardlimited output from the final 
IF amplifier 60 is fed to a phase digitizing circuit that produces samples 
of the instantaneous phase of the signal. The received signal strength 
indicator (RSSI) signal generated by the IF amplifier circuit 120 is fed 
to an A-to-D converter 140 which produces a digitized value of signal 
strength. The RSSI signal is preferably linearly proportional to the 
logarithm of the signal amplitude. When digitized to 8-bit accuracy, the 
8-bit value represents signals over, for example, a 128 dB range in steps 
of 0.5 dB, or over a 64 dB range in steps of 0.25 dB. The phase digitizer 
130 compares signal transitions or edges on the hardlimited IF signal with 
those of a stable reference clock 135 to quantize the signal transitions 
to 6-bit phase accuracy. The 6-bit modulo-64 value maps exactly to the 
phase angle range 0 to 2 .pi. when both are viewed in the circular domain. 
Approximating the circle with a regular 64-gon, the mapping is given by: 
##EQU1## 
where .phi..sub.64 .tbd..phi..sub.0 because of circular periodicity. 
The digitized log-amplitude and digitized phase together form a complex 
digital number in logpolar form, which is disclosed in U.S. Pat. No. 
5,084,669 which is commonly assigned, the specification of which is hereby 
incorporated by reference. The stream of complex numbers at a suitably 
high sample rate, for example 240,000 samples per second (240 kilosamples 
per second or 240 kS/s), are fed to a digital signal processor (DSP) 150 
where logpolar-to-cartesian conversion takes place resulting in complex 
numbers of the form, X+jY, where j=.sqroot. -1. Automatic scaling takes 
place during this conversion so that the cartesian components fit within 
the fixed-point wordlength of the DSP 150. 
Inside the DSP 150, the X+jY sample rate is first reduced by downsampling, 
that is, adding up blocks of adjacent samples over a moving window, to 80 
kS/s to reduce the amount of subsequent arithmetic. 
When the receiver is required to operate in the AMPS mode, the 80 kS/s 
stream is then submitted to a numerical frequency discrimination 
algorithm. The preferred algorithm is a digital phase lock loop, the 
details of which are beyond the scope of this disclosure but can be 
devised by anyone with ordinary skill in the art of numerical radio signal 
processing. In the alternative, the following algorithm can be employed: 
1) Perform discrimination: 
EQU F.sub.n =X.sub.n Y.sub.n-1 -Y.sub.n X.sub.n-1 ; n=1,2 , . . . , N-1 
2) Perform square amplitude calculation: 
EQU R.sub.sq =X.sub.n X.sub.n +Y.sub.n Y.sub.n ; n=0,1,2 , . . . , N-1 
3) Rescale post-hardlimiting: 
##EQU2## 
where X.sub.n.sup.R and Y.sub.n.sup.R are rescaled values, and N is the 
number of samples. 
Another alternative is to submit the 240,000 samples per second (240 kS/s) 
of phase directly to a digital phase lock loop which calculates an 
instantaneous frequency at the desired 80 KHz downsampled rate. These 
calculations at an elevated rate may be performed with the aid of special 
digital logic in order to relieve the programmable DSP 150 of some of the 
load. 
Once 80 kS/s of instantaneous frequency samples have been calculated, they 
are submitted to digital post-discriminator filtering, de-emphasis, 
downsampling and decompanding according to known digital filter and 
digital signal processing techniques to yield 8000 samples per second (8 
kS/s) of digitized speech which is converted to an analog signal in D-to-A 
converter 160. The DSP 150 may also contain numerical calculation programs 
for other purposes, such as the decoding of the Manchester coded 
signalling data used in the AMPS system for control purposes. 
When the receiver is desired to operate in the NAMPS mode, the X+iY stream 
is, after a first downsampling to 80 kS/s, submitted to a 64-tap finite 
impulse response (FIR) filter, such as the FIR filter 190 shown in FIG. 3, 
according to known theory, which reduces the bandwidth to that needed by 
NAMPS system, namely, a bandwidth that is in the 8 KHz to 12 KHz range. 
Furthermore, the output of the FIR filter 190 is further downsampled to 16 
kS/s in order to reduce the amount of processing needed. A numerical 
frequency discriminator algorithm as outlined above is then applied to 
frequency demodulate the FIR filter 190 output and the usual audio 
filtering, de-emphasis and decompanding are also applied numerically, as 
shown by audio filter/de-emphasizer 210, decompander 220, and audio 
filter/downsampler 230 in FIG. 3. 
The 64-tap filter coefficients may be calculated by means of a 64-point 
Fourier transform of an ideal rectangular filter frequency response having 
the desired bandwidth, and then multiplying the result with a raised 
cosine windowing function to reduce unwanted sidelobes. According to the 
present invention, at least two sets of these coefficients are 
precalculated which corresponds to at least two alternate NAMPS 
bandwidths, for example 8.75 KHz and 11.25 KHz. These correspond 
respectively to 64-point rectangular frequency responses respectively 7 or 
9 points wide, that is, 
EQU 0000000000000000000000000000011111110000000000000000000000000000 
or 
EQU 00000000000000000000000000111111111000000000000000000000000000 
since 
##EQU3## 
Only one set of coefficients is used at a time. Normally, the coefficients 
corresponding to 11.25 KHz bandwidth are used if adjacent channel 
interference levels are low. The amount of adjacent channel interference 
is determined by comparing the total signal power in the 80 kS/s stream 
before the FIR filter 190 with the total signal power in the 16 kS/s 
stream after the FIR filter 190. The former represents the power in a 30 
KHz bandwidth embracing both adjacent channels while the latter represents 
the power in the desired or wanted channel. By subtracting the latter from 
the former, the power in the two adjacent channels is obtained. If this 
exceeds the power in the desired or wanted channel by more than a first 
threshold, an alternate, narrow bandwidth FIR filter is used. 
When more than one alternate bandwidth is available, an even narrower 
alternate bandwidth may be selected when the adjacent channel-to-inband 
(in-channel) power ratio exceeds a second threshold. If the adjacent 
channel-to-inband (in-channel) power ratio falls below a third threshold, 
the bandwidth is widened again. A deliberate difference between the 
thresholds for narrowing and widening the bandwidths is employed in 
conjunction with the time constant .tau. for measuring the average signal 
powers in order to provide hysteresis and prevent excessive frequent 
bandwidth switching, which could otherwise cause audio noise. 
Typically, the second threshold power ratio for switching from a first 
bandwidth of 11.25 KHz to a narrower bandwidth of 8.75 KHz would be 128 
and the third threshold for switching back to 11.25 KHz would be 32. The 
time constant .tau. for determination of the desired or wanted channel 
power and the adjacent channel power is approximately 10 milliseconds (10 
ms). Thus, the desired or wanted channel power is determined as the sum of 
squares of 16 KHz samples over a 160-sample moving window, since (16 
kS/s)(10 ms)=160 samples, and the adjacent channel power is determined as 
the sum of squares of 80 KHz samples over an 800-sample moving window, 
since (80 kS/s)(10 ms)=800 samples. 
The above-described digital signal processing is further illustrated in 
FIG. 3. The logpolar phase and RSSI values enter at 240,000 samples per 
second (240 kS/s) into a logpolar-to-cartesian conversion routine 170 that 
uses a COSINE/SINE table to calculate cos.phi. and sin.phi. and an 
ANTILOGARITHM table to convert the received signal strength indicator 
(RSSI) signal to an amplitude, having first subtracted a scaling value 
from the RSSI signal. The scaled amplitude A so obtained multiplies the 
cos.phi.+isin.phi. values to obtain X+jY values, where X=Acos.phi., and 
Y=Asin.phi.. 
The scaling value is determined such that the moving average of X.sup.2 
+Y.sup.2 over 800 samples is in a desired range without risk of overflow 
or underflow. To ensure this, the moving average is supplied to the 
scaling routine which increases the scaling value if the moving average is 
too high, and lowers the scaling value if the moving average is too low. 
The scaled X+jY values at 240,000 samples per second (240 kS/s) are then 
downsampled to 80,000 samples per second (80 kS/s) in downsampler 180 by 
calculating the moving average over three consecutive samples and then 
adding up three consecutive values of the moving average to obtain each 80 
kS/s sample. The 80 kS/s X+jY values are then squared and summed over an 
800-sample moving window. This involves adding the newest sum-of-squares 
and subtracting the sum-of-squares calculated 800 samples ago, requiring a 
delay memory 800 samples long. To reduce the use of memory in the digital 
signal processor (DSP) 150, the moving average can be calculated with 
exponential rather than rectangular weighting of the past history of past 
samples. In exponential weighting, the new average Ai is calculated from 
the old average A(i-1) and the new sample Si as shown below 
EQU Ai=A(i-1)+d(Si-A(i-1)) 
where d is a small value, e.g., 1/800 
EQU Ai=(1-d)A(i-1)+dSi 
EQU Ai=(799/800)A(i-1)+Si/800 
Thus, all of the old samples contained in the old average get progressively 
reduced by the value (1-d)=799/800 which is slightly less than 1 at each 
iteration, so that a value of Si that was used N steps ago (Si-N) has been 
deweighted by (1-d).sup.N. The average is thus 
Ai=Si+(1-d)S(i-1)+(1-d).sup.2 S(i-2) . . . ect with increasing powers of 
(1-d). The advantage of this system is that no old sample values like 
S(i-799) are needed, but rather only the previous average and the new 
sample. In that case, the 160-sample moving average should also use 
exponential weighting of the past history of past samples. 
The 64-tap FIR filters 190 also operate on the X+jY values using 
coefficients C.sub.1, C.sub.2, . . . , C.sub.64 which are selected from 
one of two alternative coefficient stores corresponding to a 8.75 KHz or 
11.25 KHz bandwidth, respectively. The FIR-filtered X+jY values are output 
from the FIR filters 190 at a rate downsampled to 16 kS/s and are then 
processed in a digital phase lock loop frequency discriminator 200 to 
demodulate the frequency modulation. The frequency discriminator 200 
output is then audio filtered and de-emphasized according to the NAMPS 
specification in an audio filter/de-emphasizer 210, decompanded in a 
decompander 220, and finally further audio filtered and downsampled to 8 
kS/s in an audio filter/downsampler 230 before being output to a D-to-A 
converter 160, as shown in FIG. 2. 
The bandwidth determiner 240 uses an algorithm for determining whether the 
wide or narrow bandwidth should be used. The moving average and/or the 
moving sum of the FIR-filtered signal power is scaled by a factor of 5 in 
a scaler 250 to compensate for the difference between the different 
numbers of squares summed, or, alternatively, by another number 
appropriate to compensate as well for a non-unity scaling through the 
FIR-filters 190. The scaling of the FIR-filter coefficients C.sub.1, 
C.sub.2, . . . , C.sub.64 can in fact be deliberately chosen so that the 
required scaling for the 160-sample moving average is a power of two. The 
scaled, 160-sample moving average is then subtracted from the 800-sample 
moving average in a subtracter 260 to obtain the adjacent channel power. 
Then the adjacent channel power is compared in a comparator 270 with 
either 128 or 32 times the inband (in-channel) signal power, supplied by a 
scaler 255 (corresponding to either the second or third threshold power 
ratio, respectively), in order to determine whether the bandwidth should 
be decreased or increased, respectively. 
It will be appreciated by those of ordinary skill in the art that the 
present invention can be embodied in other specific forms without 
departing from the spirit or essential character thereof. The presently 
disclosed embodiments are therefore considered in all respects to be 
illustrative and not restrictive. The scope of the invention is indicated 
by the appended claims rather than the foregoing description, and all 
changes which come within the meaning and range of equivalents thereof are 
intended to be embraced therein.