Programmable highly temperature and supply independent oscillator

An oscillator circuit generates an output frequency that is substantially independent of power supply and temperature variations. The oscillator circuit can be implemented using conventional complementary metal-oxide-semiconductor technology. The oscillator circuit is suitable for use as an internal oscillator for generating a stable reference frequency in telecommunication receiver modules.

BACKGROUND OF THE INVENTION 
The present invention relates in general to integrated circuits, and in 
particular to a robust oscillator circuit that is insensitive to 
variations in temperature and power supply voltage level. 
In various types of telecommunication systems a reference frequency is 
commonly used as an acquisition aid in the receiver module. In those 
applications, such as ATM and SONET, that require reference frequencies of 
35 Mhz or higher, generating a stable reference signal at such high 
references places a high cost and size burden on the sub-system or module. 
Crystal oscillators or tank circuits that are commonly used to generate 
the reference frequency take up additional board space and can be 
expensive. Thus, a receiver that does not require an externally generated 
reference frequency is very desirable. Internally generated reference 
frequencies, however, tend to be less accurate than their external 
counterparts. This is because on-chip oscillator circuitry is more 
susceptible to variations in temperature and power supply voltage. There 
have been various types of oscillator circuits developed that offer some 
degree of temperature and power supply independence. However, most require 
complex circuitry often relying on the temperature behavior of bipolar 
transistors. These types of circuits thus require an optimized bipolar 
process or a combination of bipolar and complementary 
metal-oxide-semiconductor (CMOS) process. 
There is a need for a cost-effective oscillator circuit that generates a 
well controlled and stable reference frequency that is insensitive to 
temperature and power supply variations. 
SUMMARY OF THE INVENTION 
The present invention provides a fully integrated, temperature and supply 
independent oscillator circuit that provides a stable and accurate 
reference frequency. Broadly, the circuit of the present invention uses a 
relaxation type oscillator that operates by supplying a charge current to 
a capacitor that is charged and discharged based on the level of a 
threshold voltage. The invention further includes circuitry that generates 
the charge current and the threshold voltage such that the temperature 
dependent parameters in the charging current substantially cancel those in 
the threshold voltage, thereby yielding a substantially temperature 
independent output frequency. In one embodiment, the oscillator circuit is 
made digitally programmable so it can generate different frequencies 
required for different circuit applications. The oscillator circuit of the 
present invention can be implemented using conventional CMOS technology. 
Accordingly, in one embodiment, the present invention provides an 
oscillator circuit including a relaxation oscillator having a current 
input adapted to receive a charge current, a voltage input adapted to 
receive a threshold voltage, and a frequency output, a current generating 
circuit having an output coupled to the current input of the relaxation 
oscillator, and a voltage generating circuit having an output coupled to 
the voltage input of the relaxation oscillator, wherein the current 
generating circuit and the voltage generating circuit are configured to 
generate a charge current and a threshold voltage such that their 
temperature dependent parameters substantially cancel to yield a 
substantially temperature independent output frequency. 
In another embodiment, the present invention provides a voltage controlled 
oscillator including a reference circuit having a reference voltage output 
and a reference current output, a resistor circuit coupled to receive the 
reference current and configured to generate a voltage output, a voltage 
to current converter coupled to receive the voltage output of the resistor 
circuit and configured to generate a current output in response thereto, a 
gain and buffer circuit coupled to receive the reference voltage output 
and configured to generate a threshold voltage, and a relaxation 
oscillator having a current input coupled to the current output of the 
voltage to current converter, and a voltage input coupled to receive the 
threshold voltage. The gain and buffer circuit is designed such that the 
threshold voltage substantially replicates temperature-dependent 
parameters in the current output of the voltage to current converter, 
resulting in an output frequency that is substantially temperature 
independent. 
The following detailed description and the accompanying drawings provide a 
better understanding of the nature and advantages of the voltage 
controlled oscillator according to the present invention.

DESCRIPTION OF THE SPECIFIC EMBODIMENTS 
Referring to FIG. 1, there is shown a simplified block diagram of the 
oscillator circuit according to one embodiment of the present invention. 
The oscillator circuit includes a relaxation oscillator 100 that has a 
charge current I(ch) input, a threshold voltage Vth input and a frequency 
output FRQ.sub.-- OUT. The VCO further includes a current generating 
circuit 102 that receives a current output I(bg) of a band-gap circuit 104 
and generates a current output I(ch) for the charge current input of 
relaxation oscillator 100. A voltage generating circuit 106 receives a 
voltage output V(bg) of band-gap circuit 104 and generates a voltage Vth 
for the threshold voltage input of relaxation oscillator 100. The band-gap 
voltage output V(bg) is substantially power supply and temperature 
independent, while its current output I(bg) varies with temperature. With 
the output frequency (FRQ.sub.-- OUT) of relaxation oscillator 100 being 
proportional to the ratio I(ch)/Vth, current and voltage generating 
circuits 102 and 106 are designed such that the temperature dependent 
parameters in the I(ch) expression substantially equal, and therefore 
cancel, the temperature dependent parameters in the Vth expression. By 
thus canceling the temperature dependent parameters from the expression 
defining output frequency FRQ.sub.-- OUT, the oscillator circuit of the 
present invention generates a highly stable and temperature insensitive 
reference frequency. 
In the embodiment shown, the desired relationship between I(ch) and Vth is 
realized by carefully selecting the values of resistors R1 and R2 in 
current generating circuit 102 that produce temperature dependent voltage 
inputs V1(t) and V2(t) for a voltage to current converter 108. A buffer 
110, resistor R3 and current source 112 in voltage generating circuit 106 
are then designed with mirrored currents and tracking transistors to 
generate the desired Vth value. The operation of the present invention 
will be described in greater detail with reference to an exemplary CMOS 
circuit implementation shown in FIG. 2. 
Referring to FIG. 2, internal circuitry for the various blocks of FIG. 1 is 
shown according to an exemplary embodiment of the present invention. 
Band-gap reference circuit 104, implemented in CMOS technology, uses PNP 
transistors that are found inherent in conventional CMOS processes. 
Temperature dependent current output I(bg) of band-gap circuit 104 is 
mirrored by transistors 200 and 202 and flows through resistors R1 and R2. 
This generates temperature dependent voltages V1=I(bg).times.R1 and 
V2=I(ch).times.R2 at nodes N1 and N2, respectively. Voltage V1 is applied 
to one input of an operational amplifier 204, and voltage V2 is applied to 
the gate terminal of an n-channel field effect transistor M1. Operational 
amplifier 204 is configured as a source follower having its negative input 
coupled to the source terminal of transistor 206 (and drain of M1), and 
its output connected to the gate of transistor 206. The current I(M1) 
through transistors 206 and M1 is mirrored and applied to the I(ch) input 
of relaxation oscillator 100. 
Band-gap circuit 100 also generates a very stable temperature independent 
output voltage V(bg) that is applied to circuit 106. Voltage generating 
circuit 106 includes an amplifier 208 that amplifies the level of V(bg) 
and generates voltage V3 at its output. The signal V3 is applied to an 
n-channel diode-connected filed effect transistor M2 followed by resistor 
R3 and current source 210. Current source 210 is designed to draw an 
amount of current substantially equal to I(bg). The current through R3 
sets up the threshold voltage Vth at node N3 that connects to the 
threshold voltage input Vth of relaxation oscillator 100. 
In operation, the oscillation frequency FRQ.sub.-- OUT of relaxation 
oscillator 100 is determined by the value of charge current I(ch), 
threshold voltage Vth and an internal capacitance. FIG. 3 shows one 
example of a relaxation oscillator for use in the oscillator circuit of 
the present invention. As the operation of the relaxation oscillator is 
well understood, a detailed description of the circuit and its operation 
is not provided. Current I(ch) charges and discharges timing capacitors C 
(implemented by source-drain connected NMOS transistors), in response to 
the comparator outputs that toggle with Vth as the trip point. This 
results in an oscillating signal at the output. The frequency of 
oscillation of the output signal is thus given by the expression: 
EQU FRQ.sub.-- OUT=I(ch)/(2*A*Coxn*Vth) (1) 
Where: A=Area of each timing capacitor C, and 
Coxn=unit capacitance of NMOS gate capacitance. 
In order to realize a temperature and supply independent FRQ.sub.-- OUT, 
I(ch), Coxn, and Vth are designed, according to the present invention, 
such that their temperature variations substantially cancel each other 
while each parameter maintains insensitivity to power supply variations. 
Referring back to FIG. 1, the current output I(bg) of band-gap circuit 100 
depends on the base-emitter voltage of the bipolar transistors and the 
value of RBG1, and is thus directly proportional to absolute temperature 
(PTAT). Two temperature dependent voltages V1 and V2 are generated by 
mirroring current I(bg) through resistors R2 and R3, respectively. The 
voltages V1 and V2 are give by: 
EQU V1=(R2/RBG1)*ln8*VT; (2) 
EQU V2=(R3/RBG1)*ln8*VT (3) 
Where, VT is the thermal voltage (constant) and the natural log of 8 (or 
ln8) is the result of the size ratio of the bipolar transistors in 
band-gap circuit 100. The particular ratio of 8:1 for the band-gap bipolar 
device is arbitrary and is used herein as an example for illustrative 
purposes only. Note that V1 and V2 are dependent upon temperature and 
ratio of resistors only, and that they are not dependent upon absolute 
resistor values. These two voltages are designed to bias transistor M1 
into its linear region of operation. Voltage V2 is connected to the gate 
terminal of transistor M1, while V1 is coupled (via source follower 
204/206) to the drain terminal of transistor M1. By selecting proper 
values for resistors R2 and R3, the value of V1 can be made smaller than 
(V2-Vtn), where Vtn is the threshold voltage of transistor M1. The 
relation V1&lt;(V2-Vtn) ensures the operation of transistor M1 in linear 
region. The current I(M1) flowing through a linearly operating transistor 
M1 can thus be given by the following equation: 
EQU I(M1)=.mu.n.times.Coxn.times.(W/L)m1.times.[V2-Vtn-(1/2).times.V1].times.V1 
(4) 
Where: .mu.n=mobility of NMOS transistor, 
Coxn=unit capacitance of NMOS gate capacitance, and 
(W/L)m1=ratio of channel width to length for M1. 
This current is mirrored and supplied to the current input I(ch) of 
relaxation oscillator 100. 
Referring back to FIG. 2, band-gap voltage output V(bg) generates voltage 
V3 by passing through a gain amplifier 208. V3 is connected to a 
preferably large diode-connected NMOS transistor M2 and resistor R3 in 
series, with the series combination biased by current source 210. The 
value of R3 is set to one half of R2 (i.e., R3=1/2R2), and the magnitude 
of current through 210 is set to equal I(bg). Thus, assuming M2 is large 
enough such that its gate-to-source voltageVgs(m2) is very close to its 
threshold voltage Vtn at the drain current level of I(bg), this circuit 
produces a voltage Vth at node 3 that is given by: 
EQU Vth=V3-Vtn-(1/2).times.V1 (5) 
Note that equation (5) defining the threshold voltage Vth for the 
relaxation oscillator is similar to the bracketed portion of equation (4) 
defining the relaxation oscillator charge current I(ch). Combining 
equations (2), (4) and (5) into equation (1), results in the following 
expression for FRQ.sub.-- OUT: 
EQU FRQ.sub.-- 
OUT={.mu.n.times.Coxn.times.(W/L)m1.times.[V2-Vtn-(1/2).times.V1].times.V1 
}/{2A.times.Coxn.times.[V3-Vtn-(1/2).times.V1]} 
Simplifying the above equation yields: 
EQU FRQ.sub.-- 
OUT={.mu.n.times.(W/L)m1.times.[V2-Vtn-(1/2).times.V1].times.V1}/{2A.times 
.[V3-Vtn-(1/2).times.V1]} (6) 
Thus, by setting V2 and V3 to an equal voltage, and assuming a 
well-tracking n-channel threshold voltage Vtn for transistors M1 and M2 
that are preferably identical in size, equation (6) above can be further 
simplified to: 
EQU FRQ.sub.-- OUT=[.mu.n.times.(W/L)m1.times.V1]/2A (7) 
In the resulting expression for FRQ.sub.-- OUT, most temperature dependent 
parameters are thus canceled. A more thorough temperature analysis of this 
equation reveals the advantages of the present invention. Since V1 shown 
in equation (2) above is directly proportional to absolute temperature 
(PTAT), it can compensate a first order dependence of mobility on 
T.sup.-1. However mobility is almost a second order function of 
temperature, as shown below: 
EQU .mu.n(T)=.mu.o.times.(T/To).sup.(-n) where, 1.5&lt;n&lt;2. (8) 
Considering that V2 and V1 are both PTAT and V3 is temperature insensitive, 
and performing a partial derivative of the equation (6) defining 
FRQ.sub.-- OUT with respect to absolute temperature, shows the overall 
temperature dependence of the output frequency. Since V3 is a multiple of 
the band-gap output voltage V(bg), d(V3)/dT=0. Therefore: 
##EQU1## 
where, k is the temperature coefficient of Vtn. 
Note that V1, V2 and V3 can be adjusted to exactly compensate the mobility 
variation for a given temperature. It can be shown that for a temperature 
range from, for example, -20 C. to 150 C., the maximum frequency change of 
FRQ.sub.-- OUT for the circuit of FIG. 2 is about .+-.2.5%. This exemplary 
number represents 294PPM/C in average, and is sustained at all process 
corners. It can also be shown that a temperature coefficient of zero may 
be obtained around 60 to 100 C. for the circuit of FIG. 2. 
In one embodiment, the present invention allows for the output frequency 
FRQ.sub.-- OUT to be digitally programmable. As explained above, 
transistor M1 operates in linear mode, and sets the magnitude of charge 
current I(ch) for relaxation oscillator 100. Referring back to equation 
(7) which defines the value of FRQ.sub.-- OUT, it can be seen that 
FRQ.sub.-- OUT can be linearly scaled by adjusting the size (W/L) of 
transistor M1. This can be accomplished by connecting one or more 
transistors in parallel to transistor M1 via digitally programmable 
switches. FIG. 2 shows (in phantom) one programmable transistor PM1 that 
is connected in parallel to transistor M1 via a programmable switch S1. By 
closing switch S1, transistor PM1 is connected in parallel to transistor 
M1, and the combined transistors have an increased W/L ratio, causing an 
increase in I(M1) and I(ch). This in turn increases the frequency of 
FRQ.sub.-- OUT. Thus, the linear scaling of the size of transistor M1 
allows the VCO of the present invention to be used in telecommunication 
applications requiring different acquisition frequencies. For example, the 
VCO frequency can be switched to the reference frequency required by T3, 
or E3 or STS1. 
In conclusion, the present invention provides an oscillator circuit that 
generates a temperature and supply independent output frequency. The 
highly stable oscillator circuit can be fabricated using conventional CMOS 
technology, and does not rely on optimized bipolar technology. While the 
above is a complete description of the preferred embodiment of the present 
invention, it is possible to use various alternatives, modifications and 
equivalents. Therefore, the scope of the present invention should be 
determined not with reference to the above description but should, 
instead, be determined with reference to the appended claims, along with 
their full scope of equivalents.