Video signal adjusting apparatus, display using the apparatus, and method of adjusting the display

A color corrector receiving a video signal as an input signal and producing an output signal representing a function of power of the input signal includes a logarithmic converter for achieving a logarithmic conversion of the input signal and outputting a resultant signal, a variable gain amplifier for altering amplitude of the signal from the logarithmic converter and outputting an obtained signal, and an antilogarithmic converter connected to the amplifier for outputting a signal representing an exponential function of the signal from the amplifier.

BACKGROUND OF THE INVENTION 
The present invention relates to a color corrector, a signal converter, a 
video display using the corrector and the converter, and a method of 
adjusting the display suitable for a gamma correction and a white balance 
adjustment related to a video signal processing in a video apparatus such 
as a video display, a television receiver, a printer, or a camera. 
In this connection, the term "color corrector" adopted in this 
specification is defined to include apparatuses which conduct, in addition 
to the color correction of signals of three primary colors including red, 
green, and blue, the video signal correction and gradation correction for 
video displays handling monochrome and black-and-white video signals. 
There has been known a gamma correction circuit for a display, a video 
camera, or the like, for example, as shown in FIG. 2 of the Japanese 
Utility Model application publication No. JP-U-63-47113. In the gamma 
correction circuit of this diagram, assuming a relationship Va&lt;Vb&lt;Vc&lt;Vd 
between potential values Va, Vb, Vc, and Vd respectively of the power 
sources 30, 33, 36, and 39, a conductive state is respectively established 
in diodes 32, 35, 38, and 41 as a voltage applied on an input terminal 1 
is increased. 
Consequently, when a voltage signal having a waveform 42 of FIG. 3 of the 
utility model publication above is applied to the input terminal 1, there 
is attained from an output terminal 5 an inverse output signal having a 
waveform 43 which can be approximated to a .gamma.-th power of the input 
signal. In consequence, using a gamma correction circuit of FIG. 2, there 
can be approximately obtained a proportional relationship between an input 
light quantity and an output light quantity respectively of the 
transmission and reception systems. 
However, according to the gamma correcting circuit described in the 
JP-U-63-47113, the .gamma.-th power transfer characteristic is implemented 
as indicated by a broken-line approximation and hence there cannot be 
sufficiently attained a satisfactory correction precision. This makes it 
difficult to develop a high-fidelity reproduction of colors between the 
video sending and receiving systems. In addition, when it is desired to 
cope with changes in the parameter .gamma. associated with correction due 
to variation in devices or the like, there is required alteration in 
values of a large number of constituent elements. Namely, this is 
impossible in practices. 
SUMMARY OF THE INVENTION 
It is therefore a first object of the present invention to provide a color 
corrector in which the correction precision of a gamma correction circuit 
is improved and the gamma parameter of the circuit can be easily changed. 
Moreover, a second object of the present invention is to apply the color 
corrector to various video apparatuses, thereby improving performance and 
functions of the apparatuses. 
In an embodiment of the color corrector according to the present invention, 
an input signal is applied to an input terminal of a logarithmic 
converter, an output terminal of the logarithmic converter is linked to an 
input terminal of a variable gain amplifier, and an output terminal of the 
amplifier is connected to an input terminal of an exponential or 
antilogarithmic converter. 
Moreover, the color corrector according to the present invention is applied 
to the following video apparatuses and systems using such apparatuses. The 
video apparatuses include, for example, video displays employing display 
elements such as a Braun tube, a liquid-crystal display panel, a plasma 
display panel, and a light-emission-diode (LED) panel; video input devices 
such as a video camera and a scanner, video output devices such as a 
printer and an electronic camera; and a computer system, a broadcast and 
communication system, and the like using above apparatuses and/or devices. 
In the embodiment of the color corrector according to the present 
invention, the logarithmic converter achieves a function to replace a 
computation of power with a multiplication for easily computation. 
Furthermore, the amplifier has a function to vary the value of the .gamma. 
parameter. In addition, the antilogarithmic converter carries out a 
function to restore the multiplication to the computation of power, 
thereby implementing the color corrector. 
Each of the apparatuses and systems to which the color corrector of the 
present invention is applicable includes an internal circuit having a 
nonlinear characteristic to be corrected.

DESCRIPTION OF TEE PREFERRED EMBODIMENTS 
FIG. 1 shows in a block diagram a color corrector in a first embodiment 
according to the present invention. As can be seen from FIG. 1, the color 
corrector of the present invention includes a series connection of a 
logarithmic converter 2, a variable gain amplifier 3, and an 
antilogarithmic converter 4. This converter is operable in two signal 
modes for the voltage and current signal operations. In FIG. 1, an input 
signal S.sub.IN supplied to an input terminal 1 of the color corrector is 
converted by the logarithmic converter 2 into 1n(S.sub.IN) to be fed to 
the amplifier 3 of the next stage. The converter 2 replaces a variable 
power operation via the variable gain amplifier 3 into a variable 
multiplication for easy computation, which is an essential purpose of the 
present invention. In consequence, the amplifier 3 is used to alter the 
gamma parameter. In this connection, there may be used a variable 
attenuator as the amplifier 3. Assume that the amplifier 3 has a gain 
.gamma.. In the antilogarithmic converter 4 of the subsequent state, there 
is accomplished a signal conversion from .gamma.1n(S.sub.IN) into 
S.sub.IN.gamma.. The resultant signal is obtained from an output terminal 
5. In this way, there is developed a .gamma.-th power curve characteristic 
unlike the characteristic of broken-line approximation of the prior art. 
Consequently, according to the embodiment, the correction precision of the 
color corrector is remarkably increased. 
FIG. 4 shows a first concrete example of the circuit of the color corrector 
shown in FIG. 1. According to an aspect of this example, the system is 
configured in a simple circuit structure. In operation of the corrector of 
FIG. 4, a signal supplied from an input terminal 1 is converted into a 
logarithmic voltage by a logarithmic converter 2 including transistors 12 
to 16 to be then attenuated through a variable attenuator (employed as the 
variable gain amplifier 3 having the gain less than one) including a 
low-priced variable resistor 18. Thereafter, the attained signal is fed to 
an antilogarithmic converter 4 including transistors 25 to 27 to be 
converted into a signal current having a .gamma.-th power characteristic. 
The resultant signal is obtained from an output terminal 5. In the 
operation, the input signal may be in either one of the voltage and 
current forms. As shown in FIG. 4, connecting a resistor 28 between a 
collector of the transistor 27 and a grounding point with respect to the 
alternate current, there can be attained the output signal in the voltage 
format. In the logarithmic converter 2, a signal current Ic attained from 
a grounded-emitter circuit including the transistor 16 and a resistor 17 
is converted into a logarithmic voltage 4V.sub.T 1n(Ic/i.sub.s) between 
the base and the emitter through a 4-stage circuit including the 
transistors 12 to 15, where V.sub.T and i.sub.s are respectively the 
thermal voltage and the saturation current in the p-n junction. Assume 
that the respective transistors have substantially an identical thermal 
voltage V.sub.T and substantially an identical saturation current i.sub.s. 
Furthermore, a bias circuit including a constant current circuit 24 
including a constant current diode and the like, the transistors 20 to 23, 
and the resistor 19 compensates for the thermal drifts respectively of the 
thermal voltage V.sub.T and the saturation current i.sub.s of the 
transistors constituting the logarithmic converter 2 and the 
antilogarithmic converter 4. In the connecting structure, the number of 
base-emitter pn junctions of the transistors 20 to 23 is selected to be 
equal to that of each of the logarithmic converter 2 and the 
antilogarithmic converter 4. The bias circuit above moreover functions to 
set a fixed operating point which is independent of the parameter .gamma. 
of the power characteristic of the color corrector. When the value of the 
collector current of the transistor 16 is equal to that of the constant 
current circuit 24, the relationship between the input and output signal 
levels is fixed regardless of the value of the parameter .gamma.. In 
consequence, it is possible to use the input signal level as a reference 
for the color correction. Assume that the signal voltage obtained from a 
slider terminal of the variable resistor 18 disposed between the 
logarithmic converter 2 and the bias circuit is expressed as 
K.multidot.{4V.sub.T 1n(Ic/i.sub.s)}. The value of K can be altered in a 
range from 0 to 1. In this situation, setting the resistance value of the 
variable resistor 18 to an appropriate value, the correction precision of 
the color corrector can be maximized. That is, the resistance value is set 
to a high value such that the maximum value of the current flowing into 
the resistor 18 is sufficiently smaller than that of the constant current 
circuit 24. At the same time, the resistance value is favorably set to a 
low value such that the voltage drop of the resistor 18 due to the base 
current of the transistor 27 is negligible. When the signal voltage is 
supplied to the antilogarithmic converter 4, according to the exponential 
or antilogarithmic conversion developed by the two-stage base-emitter pn 
junctions of the transistors 26 and 27, there is obtained from the 
collector of the transistor 27 an output current which is the (2.gamma.)th 
power of Ic. Consequently, in the color corrector of this specific 
example, there is obtained a characteristic ranging from 0-th power to 
second power. In a case, like the circuit of FIG. 4, where the color 
corrector includes a linear circuit having a gain K between the circuits 
each including series connection of pn-junction elements, when there are 
included M stages of pn junctions for the logarithmic conversion and N 
stages of pn junctions for the antilogarithmic conversion, there are 
attained a power characteristic of (KM/N)th power. In addition, in the 
circuit of FIG. 4, there is utilized an exponential or antilogarithmic 
characteristic of the current flowing into the pn junction. However, the 
color corrector of the specific example can be naturally adopted in any 
devices and circuits which develop an antilogarithmic characteristic like 
a field-effect-transistor (FET) and a diode having an antilogarithmic 
characteristic, for example, in a tail region in the vicinity of the 
rising edge of the drain current. In addition, to improve precision of the 
temperature compensation, it is only necessary to use a pair transistor 
configuration at least for each of the combinations of transistors 12 and 
21, 13 and 21, 14, 22, and 26, and 15, 23, and 27. The transistor 25 
functions to set an identical collector-emitter voltage for the transistor 
26 and 22, thereby suppressing the antilogarithmic conversion error due to 
the influence of the grounding voltage. 
FIG. 5 shows the circuit configuration of a second specific example of the 
color corrector shown in FIG. 1. According to an aspect of this example, 
the input and output signals are handled in a differential mode to expand 
the application range of the circuit. In the color correction of FIG. 5, 
an input signal V.sub.IN applied across input terminals 44 and 45 is 
converted, by a logarithmic converter 2 including a differential circuit 
including transistors 48, 49, 51, and 52, a variable gain amplifier 3 
including transistors 57 and 58 and a variable resistor 62, and an 
antilogarithmic converter 4 including a differential circuit configured 
with transistors 65 and 66, into an output signal V.sub.OUT between output 
terminals 46 and 47 having a characteristic of power. Thanks to the common 
mode rejection of the logarithmic converter 2 including the differential 
circuit, it is possible to connect either one of the input terminals 44 
and 45 to the ground in the sense of the alternating current so as to 
input the signal from only one input terminal. The gain in the logarithmic 
conversion may be altered by changing the pertinent values of the 
resistors 481 and 491, respectively. However, when the gain increases, 
there appears increase in the distortion associated with the logarithmic 
characteristic of the base-emitter voltage of each of the transistors 48 
and 49. In this situation, to prevent the distortion due to the influence 
of the logarithmic conversion, it is also possible to use current sources 
53 and 54. Utilizing the current sources 53 and 54, there is established a 
bypass for the direct-current (dc) component to increase the ratio of the 
signal current component in the current flowing through each of the 
transistors 51 and 52, thereby attaining a high gain without deteriorating 
the linearity. The current sources 53 and 54 may be respectively 
substituted for resistors. The resistors need not have an identical 
resistance value in this case. In addition, in order to increase the gain, 
to the emitter of each of the transistors 51 and 52 of which the 
respective bases are biased to a fixed voltage, there may be connected in 
series such devices having a logarithmic voltage characteristic as diodes. 
The output signal from the logarithmic converter 2 including the 
differential circuit is fed via the variable gain amplifier 3 to the 
antilogarithmic converter including the differential circuit. In the 
operation, manually adjusting the value of the variable resistor 62 to 
alter the gain of the amplifier 3, there can be attained an output having 
an arbitrary power characteristic from the antilogarithmic converter 4 
including the differential circuit. 
The gain of the variable gain amplifier 3 shown in FIG. 5 can be altered 
through the manual adjustment of the resistance value of the variable 
resistor 62. However, the gain can also be changed by replacing the 
variable resistor 62 with an electronically controllable impedance element 
such as an FET, a diode, or a photocoupler. In this fashion, the 
electronic control of the gain of the variable gain amplifier enables an 
automatic control of the color corrector. Moreover, the variable gain 
amplifier may include a multiplier to develop the electronic control of 
the color corrector. A specific circuit example of the amplifier will be 
shown in FIG. 6. This is a circuit diagram showing an example of a 
variable gain amplifier of an electronic control which may be replaced 
with the variable gain amplifier of FIG. 5. Namely, the variable gain 
amplifier of the electronic control type shown in FIG. 6 can be used in 
place of the variable gain amplifier of FIG. 5 by connecting input 
terminals a and b and output terminals c and d to the terminals 
respectively assigned with the same reference codes in FIG. 5. The 
amplifier of FIG. 6 includes a 4-quadrant multiplier circuit including 
transistors 70, 72, 74, and 77 and transistors 80, 88, 85, and 86 
controlling the multiplier circuit. As a voltage V.gamma. of a control 
voltage source 90 is increased, the gain of the amplifier becomes larger. 
In addition, using pnp-type transistors like the transistors 74 to 77 for 
the transistors 85 and 86, there is guaranteed integrity in the circuit 
integration. Moreover, the base current of the transistors 74 to 77 is 
prevented from flowing into the side of the control circuit, thereby 
keeping an appropriate control range. 
Furthermore, there is shown in FIG. 7 a specific circuit example of a 
variable gain amplifier including a diode as an impedance element so as to 
electronically controlling the gain thereof. FIG. 7 is a circuit diagram 
showing another example of a variable gain amplifier of an electronic 
control which may be used in placed of the variable gain amplifier of FIG. 
5. In FIG. 7, the operational resistance value of each of the transistors 
95 and 96 respectively connected to diodes is controlled by adjusting the 
bias current supplied to the diodes. The voltage V.gamma. of the control 
voltage source 107 disposed to alter the gain is applied via transistors 
106 and 104 to a resistor 105 to conduct the voltage-to-current conversion 
while suppressing the influence of the temperature drift of the 
base-emitter voltage of the transistor. The converted current biases the 
transistors 95 and 96 thus connected to the diodes via the current mirror 
circuit including transistors 102 and 99. As the voltage V.gamma. of the 
control voltage source 107 become higher, the bias current is increased. 
In consequence, the operational resistance value of each of the 
transistors 95 and 96 is lowered and the gain of the variable gain 
amplifier is increased. 
FIG. 8 is a circuit diagram showing a third specific example of the color 
corrector shown in FIG. 1. This example has an aspect in which the 
integrity of the paired transistors is advantageously employed to possibly 
suppress the temperature dependence. The color corrector of FIG. 8 
primarily includes a logarithmic converter 2 including a pair of 
transistors 400 and 402, a variable gain amplifier 3 including a 
differential amplifier 407, and an antilogarithmic converter 4 including a 
pair of transistors 403 and 404. A current I.sub.IN from an input signal 
current source 188 flows into an emitter of the transistor 400 to be 
subjected to a .gamma.-th power conversion, thereby producing an output 
current I.sub.OUT from an output terminal 5. The conversion having a 
favorable stability with respect to temperature is obtained by the 
integrity of each of the pairs of transistors 400 and 402 as well as 403 
and 404. A current I.sub.K from a variable current source 401 flows into 
an emitter of the transistor 402. Consequently, an intra-emitter voltage 
.DELTA.V.sub.E between the transistors 400 and 402 becomes independent of 
saturation currents of the transistors. Namely, the voltage takes a value 
proportional to a ratio between the currents I.sub.IN and I.sub.K and to 
the thermal voltage. Moreover, the emitter potential of the transistor 400 
becomes equal to the base potential of the transistor 403 to which a 
negative feedback is effected, i.e., the current from the constant current 
source 406 is supplied to the corrector of the transistor 403. In this 
case, the negative feedback loop ranges from the collector of the 
transistor 403 via the base and emitter of the transistor 400 to the base 
of the transistor 403. Furthermore, the voltage .DELTA.V.sub.E is 
multiplied by .gamma. in the differential amplifier 407 to be applied to 
the base of the transistor 404 having the same emitter potential as the 
transistor 403. Influences from the thermal voltage is consequently 
cancelled out and there is obtained an output current I.sub.OUT undergone 
the .gamma.-th power conversion. In this situation, thanks to the 
integrity of the transistors 403 and 404, there is attained a 
voltage-current conversion characteristic which can be approximated to 
depend only on the thermal voltage and the voltage .DELTA.V.sub.E. 
However, since the thermal voltage is proportional to the absolute 
temperature of the emitter junction, the difference between the pairs is 
ordinarily decreased and hence the temperature dependence is considerably 
lowered. When the paired transistors are manufactured in an identical 
chip, the temperature dependence is much more improved. In addition, the 
gain .gamma. of the differential amplifier 407 is readily altered. Also in 
a case of the variable gain .gamma., there is retained the relationship 
that the output current becomes identical to the current I.sub.C when the 
input current is equal to the current I.sub.K. Furthermore, setting the 
common emitter potential V.sub.E to 0 volt or a negative voltage, the 
output dynamic range can also be expanded. In consequence, when the gain 
.gamma., the currents I.sub.K and I.sub.C, and the emitter voltage V.sub.E 
become variable in the color corrector of the specific example, the 
conversion characteristic and the output dynamic range thereof can be 
arbitrarily selected. Moreover, connecting the output terminal 5 via an 
impedance element to a voltage source, a voltage output can be obtained 
from the color corrector of the specific example. The transistors 400, 
402, 403, and 404 may be replaced with elements, for example, diodes and 
FETs, which have a nonlinear characteristic, namely, a voltage-current 
characteristic representable by an antilogarithmic function or a 
logarithmic function. 
FIG. 9 is a circuit diagram showing a fourth specific example of the color 
corrector shown in FIG. 1. This example has an aspect that there is used 
the integrity of the paired transistors constituting the differential 
pairs to possibly suppress the temperature dependence. The color corrector 
of FIG. 9 mainly includes a logarithmic converter 2 including a pair of 
transistors 16 and 409, a variable gain amplifier 3 including a 
differential amplifier 407, and an antilogarithmic converter 4 including 
pairs of transistors 403 and 404 as well as 415 and 416. Furthermore, a 
loop ranging from a collector of the transistor 403 via transistors 409 to 
411 to a base of the transistor 403 forms a negative feedback loop to set 
the collector current of the transistor 403 to Ic. An input voltage 
V.sub.IN from an input terminal 1 is transformed into a current through 
the transistor 16 and the impedance 17 to be fed to a common emitter 
terminal of the differential pair of the transistors 409 and 410. The 
obtained signal is then subjected to the .gamma.-th power conversion 
through the paired transistors 403 and 404 to be delivered therefrom as an 
output current I.sub.OUT. The principle of cancelling out influences from 
the thermal voltage and thereby guaranteeing the stability with respect to 
temperature is the same as for the specific example shown in FIG. 8. 
Moreover, in a case where the parameter .gamma. is variable in the color 
corrector of the specific example, when the magnitude of the current 
transformed from the input voltage V.sub.IN becomes equal to twice that of 
the constant current source 408 deciding the corrector current of the 
transistor 410, there is kept the relationship that the output current is 
identical to the current Ic. Furthermore, since the emitters respectively 
of the paired transistors 403 and 404 are connected via the paired 
transistors 415 and 416 to the terminal 405 having the common emitter 
potential V.sub.E, the voltage source impedance can be set to a high value 
in the operation to establish the potential V.sub.E, thereby increasing 
the degree of freedom in the circuit designing process. The gain of the 
differential amplifier 407 has been set to 2.gamma. because the 
alternate-current (ac) grounding with respect to the common emitter 
potential is effected via the paired transistors 403 and 404 as well as 
415 and 416. In addition, the circuit system is driven by an emitter 
follower or a common collector amplifier including the transistors 411 and 
412, a relatively high output power is attained from the paired 
transistors 403 and 404. The transistors 409, 410, 403, and 404 may be 
substituted for elements, for example, diodes and FETs, which have a 
nonlinear characteristic, namely, a voltage-current characteristic 
representable by an antilogarithmic function or a logarithmic function. 
FIG. 10 is a circuit diagram showing a fifth concrete example of the color 
corrector of FIG. 1. According to an aspect of this example in which the 
integrity of paired transistors is employed to suppress the temperature 
dependence, a signal path is not included in the negative feedback loop 
disposed to set the output reference current so as to increase the 
operation speed. The color corrector of FIG. 10 includes a logarithmic 
converter 2 including a pair of transistors 418 and 419, a variable gain 
amplifier 3 including transistors 420 and 421 and a variable resistor 422, 
and an antilogarithmic converter including paired transistors 403 and 404 
as well as 415 and 416. An Input signal voltage V.sub.IN is subjected to a 
logarithmic conversion through the paired transistors 418 and 419 to be 
multiplied by .gamma. in the amplifier 3 including the differential 
circuit constituted with the transistors 420 and 421 so as to supply the 
resultant signal to the transistor 404. In this configuration, the 
negative feedback loop which sets the collector current of the transistor 
403 includes a loop ranging from the collector of the transistor 403 via 
transistors 430 and 433 and a resistor 429 to the base of the transistor 
403. Namely, the loop is separated from a signal path including the 
transistors 420 and 404 and the resistor 423. With provision of the 
circuit construction, the signal frequency band and the response speed can 
be satisfactorily guaranteed without any restriction from the stability 
condition of the negative feedback loop. In this regard, to improve 
precision of setting collector currents of the transistors 403 and 404, 
the constant current sources 424 and 426 and the resistors 423 and 429 are 
set to an identical current value and an identical resistance value, 
respectively. It is to be understood that the gain of the variable gain 
amplifier 3 can be altered according to the resistance value of the 
variable resistor 422. 
FIG. 11 is a circuit diagram of a sixth specific example of the color 
corrector shown in FIG. 1. According to an aspect of this example in which 
the integrity of paired transistors is utilized to suppress the 
temperature dependence and a signal path is separated from the negative 
feedback loop disposed to set the output reference current so as to 
increase the operation speed. The characteristic of the circuit system can 
be controlled according to the voltage and current. The color corrector of 
FIG. 11 includes a logarithmic converter 2 including a pair of transistors 
449 and 450, a variable gain amplifier 3 including transistors 451 and 
452, paired transistors 418 and 419 as well as 435 and 436, and a variable 
current source 437, and an antilogarithmic converter including transistors 
403 and 404. An Input signal voltage V.sub.IN is subjected to a 
logarithmic compression through the paired transistors 449 and 450 to be 
multiplied by .gamma. in the amplifier 3 including the differential 
circuit constituted with the transistors 451 and 452, the paired 
transistors 418 and 419 as well as 435 and 436, and the variable current 
source 437. The resultant signal is fed to the transistor 404. In this 
construction, in operation of the variable gain amplifier, the ratio 
between emitter currents respectively of the paired transistors 418 and 
419 is equal to that between the collector currents respectively of the 
paired transistors 435 and 436. In consequence, the gain takes a larger 
value when the current I.sub.E of the variable current source 437 is 
increased. Moreover, the signal current flowing into the collector of the 
transistor 436 is fed via a grounded-base transistor 439 to an impedance 
441 so as to be transformed into a voltage signal. The obtained signal is 
applied to the transistor 404. In this operation, to improve precision of 
setting collector currents of the transistors 439 and 440, a constant 
current sources 438 connected to the emitter of a transistor 440 is set to 
I.sub.E /2 in cooperation with the variable current source 437. The 
negative feedback loop which sets the collector current of the transistor 
403 to Ic includes a loop ranging from the collector of the transistor 403 
via a transistor 448, a resistor 4447, and the transistor 440 to the base 
of the transistor 403. Namely, the loop is separated from a signal path 
including the 436 and 439 and the resistor 445. Moreover, also connected 
to the collector of the transistor 435 are a transistor 444 and a resistor 
446 to cancel out a small signal current flowing into the emitter of the 
transistor 448. This further guarantees separation between the negative 
feedback loop and the signal path. It is to be appreciated that the 
section of the variable gain amplifier 3 can be replaced with any one of 
the variable gain amplifiers of the electronic control type shown in FIGS. 
6 and 7, respectively. 
In addition, each of the resistors 445 to 447 respectively connected to the 
transistors 439, 444, and 440 of FIG. 11 can be substituted for a variable 
current source circuit supplying a one-directional current. Thanks to the 
variable current source circuit, it is possible to suppress distortion due 
to the operational resistance of the emitter of each of the transistors 
439, 444, and 440. Namely, according to the current division ratio of the 
operational resistance of each of the transistors 439, 444, and 440, 
portions of the signal current are leaked out. In the operation, the 
values of the operational resistance respectively of the emitters of the 
transistors 439, 444, and 440 are changed in accordance with the values 
respectively of the emitter current values respectively of the transistors 
439, 444, and 440. Consequently, the current division ratio is also varied 
according to the values respectively of the emitter current values 
respectively of the transistors 439, 444, and 440, which leads to the 
distortion. In this situation, when the resistors 445 to 447 are replaced 
respectively with variable current source circuits, the current division 
ratio of signal currents to the emitters of the transistors 439, 444, and 
440 becomes substantially equal to one, thereby suppressing the 
distortion. FIG. 12 shows a specific circuit example in which the variable 
current source circuits are adopted in place of the resistors. Namely, the 
configuration of FIG. 12 is obtained by substituting the resistors 445 to 
447 of the color corrector of FIG. 11 for variable current source 
circuits, respectively. That is, transistors 4391, 4441, and 4401 form a 
current source circuit such that the current value thereof is controlled 
by a grounded-emitter transistor 4471. In this circuit structure, the 
control characteristic of the current value of the current source circuit 
is decided by a voltage bias source 4483 and resistors 4481, 4482, 4471, 
4461, and 4451. In this regard, there may be formed a short circuit to 
delete the voltage bias source 4483. 
FIG. 13 shows in a block diagram a color corrector in a second embodiment 
according to the present invention in which a differential transmission of 
signals is employed to develop a high-speed wide-band operation with a 
high precision in a minimized configuration. As can be seen from FIG. 13, 
the color corrector of this embodiment includes a series connection of a 
logarithmic converter 700, a variable gain amplifier 701, and an 
antilogarithmic converter 702. The corrector can be operated with voltage 
and current signals. In FIG. 13, an input signal SIN applied to an input 
terminal 1 of the corrector is subjected to a logarithmic conversion 
through the logarithmic converter 700 to be outputted in a differential 
manner as a differential input signal to the amplifier 701. The signal is 
multiplied by .gamma. therein to be outputted in a differential manner as 
a differential input signal to the antilogarithmic converter 702. The 
signal is then subjected to an antilogarithmic conversion and then to a 
.gamma.-th power conversion to be outputted from an output terminal 5. 
Thanks to the differential signal transmission, it is possible to remove 
the noise associated with the in-phase superimposition of signals on the 
signal line so as to further improve precision of the conversion 
characteristic. Moreover, there can be unnecessitated the negative 
feedback loop which has been required for a high-precision transmission of 
a signal having a small amplitude after the logarithmic conversion. Thanks 
to absence of the negative feedback loop, the configuration of the circuit 
can be minimized; moreover, restriction imposed due to the stability of 
feedback on the operation speed and the operation band can be removed to 
develop a high-speed wide-band operation. In addition, it is to be 
appreciated that the gain .gamma. of the variable gain amplifier 701 need 
not be variable to obtain the advantageous effect of the embodiment only 
if the gain .gamma. is fixed to the value specified by the designer. 
Furthermore, the present invention is not restricted by the color 
corrector but is naturally applicable to a converter having a power 
characteristic which can be broadly used in the general signal processing. 
FIG. 14A shows in a circuit diagram a first specific embodiment of the 
color corrector shown in FIG. 13. According to an aspect of the 
embodiment, a signal transmission path from a logarithmic converter via a 
variable gain amplifier to an antilogarithmic converter is configured in a 
differential structure to reduce the circuit configuration and to conduct 
a high-speed wideband operation. Moreover, it is possible to transfer a 
small signal with a high stability in a range from an output of the 
logarithmic converter to an input of the antilogarithmic converter. The 
color corrector of FIG. 14A primarily includes a logarithmic converter 700 
including transistors 532 and 533, a variable gain amplifier 701 including 
transistors 504 and 505 and a variable resistor 514, and an 
antilogarithmic converter 702 including transistors 506 and 508 as well as 
507 and 509. An input voltage VIN from an input terminal 1 is transformed 
into a current through a transistor 16 and an impedance element 17 to be 
subjected to a logarithmic conversion through the transistors 532 and 533, 
thereby producing a differential voltage between emitters respectively of 
the transistors 532 and 533. In the variable gain amplifier including the 
transistors 504 and 505, the voltage is multiplied by a differential gain 
2.gamma. determined by resistance values respectively of the resistors 518 
and 519 and the variable resistor 514. The resultant signal is applied 
between bases respectively of the transistors 506 and 508. Through the 
transistors 506 and 508 as well as 507 and 509, the amplified voltage of 
the differential signal is subjected to an antilogarithmic conversion into 
a signal current, which flows into a collector of the transistor 509. The 
signal current is transformed into a voltage through an impedance 516 so 
as to obtain from an output terminal 5 an output voltage V.sub.OUT 
undergone a .gamma.-th power conversion. In this way, thanks to the 
differential configuration of the signal transmission path from the output 
of the logarithmic converter to the input of the antilogarithmic 
converter, it is possible, for example, to remove the negative feedback 
loop deciding the output reference current shown in FIGS. 10 and 11. 
Accordingly, there can be guaranteed a sufficient signal frequency band 
and a high response speed. Moreover, it is possible to transfer a small 
signal with a high stability in a range from the output of the logarithmic 
converter to the input of the antilogarithmic converter. In addition, when 
a current I.sub.D from a variable current source 515 is variable, there 
can be attained a variable amplitude for the output voltage V.sub.OUT. 
That is, the embodiment leads to a drive adjustment, which will be 
described later in conjunction with FIG. 29. Moreover, it is to be 
understood that when pnp-type transistors are employed as the transistors 
504 and 505 constituting the variable gain amplifier, the similar 
advantageous effect is obtained also in a circuit configuration in which 
the polarity of each of the current sources 512 and 513 and the voltage 
source is reversed. Furthermore, additionally connecting a grounded-base 
transistor to each of the transistors 504 and 505, the variable gain 
amplifier is configured as a differential circuit in a cascode structure, 
thereby much more increasing the frequency band and the response speed. It 
is to be appreciated that according to the resistance value of the 
variable resistor 514, the gain of the variable gain amplifier can be 
altered to obtain a desired value of the parameter .gamma.. In this 
regard, the section of the variable gain amplifier can be replaced by use 
of the terminals a to d with the variable gain amplifier of the electronic 
control type shown in FIGS. 6 and 7. Furthermore, using pairs of 
transistors having integrity for the paired transistors 532 and 533, 504 
and 505, 506 and 508, and 507 and 509, there can be suppressed the dc 
offset voltage appearing after the logarithmic conversion. Consequently, 
according to the current ID set in the variable current source 515, it is 
possible to set an appropriate output amplitude so as to improve precision 
of the .gamma.-th power conversion. In addition, when each or some of the 
paired transistors is or are fabricated in an identical chip or arranged 
to be tightly adjacent to each other, an equal pn junction temperature is 
assumed in each of the paired transistors. In consequence, the drift of 
the conversion characteristic with respect to temperature is much more 
suppressed and the precision is further improved. It is accordingly 
natural that the embodiment is suitable for an integrated circuit in which 
all transistors are fabricated in a chip. Moreover, according to the 
embodiment, the current sources 536 and 515 are employed as the current 
sources of the reference currents for the logarithmic and antilogarithmic 
converters, respectively. However, when the deviation of the voltage 
applied to each of these current sources is within a desired range of 
precision, these current sources can be naturally substituted for 
resistors. Also, it is to be understood that the variable current sources 
515 and 536 may be replaced with variable resistors or fixed resistors 
having resistance values stipulated in the designing process. Similarly, 
the variable resistor 514 and the current sources 512 and 513 constituting 
the variable gain amplifier may be naturally replaced with fixed 
resistors. 
The configuration of the logarithmic converter of FIG. 14A may be 
substituted for the circuit configuration of FIG. 14B. In FIG. 14B, in 
place of a transistor 532' connected to a reference current source 536' 
there may be employed an element having a polarity opposite to that of the 
transistor 533. 
FIG. 15 shows in a circuit diagram of a second specific embodiment of the 
color corrector shown in FIG. 13. According to an aspect of the embodiment 
having the aspect of the embodiment of FIG. 14A, even when the 
differential gain .gamma. of the variable gain amplifier, namely, the 
power .gamma. of the characteristic is set as .gamma..ltoreq.1, a 
satisfactory dynamic range can be guaranteed for the output signal. The 
color corrector of FIG. 15 mainly includes a logarithmic converter 
including transistor pairs 500 and 502 as well as 501 and 503, a variable 
gain amplifier including transistors 504 and 505 and a variable resistor 
514, and an antilogarithmic converter including transistor pairs 506 and 
508 as well as 507 and 509. An input voltage V.sub.IN from an input 
terminal 1 is converted into a current through a transistor 16 and an 
impedance 17 to be subjected to a logarithmic conversion through the 
transistors 500 and 501. The attained signal is then subjected to a level 
shift through the transistors 502 and 503 to be developed as a 
differential voltage between bases respectively of the transistors 500 and 
502. In the variable gain amplifier including a differential circuit. 
constituted with the transistors 504 and 505, the resultant voltage signal 
is amplified by the differential gain .gamma. determined by the resistance 
values of the resistors 518 and 519 and the variable resistor 514 to be 
applied between bases respectively of the transistors 506 and 508. The 
differential signal voltage thus amplified is subjected to an 
antilogarithmic conversion through the transistors 506 and 508 as well as 
507 and 509 to be a signal current flowing into a collector of the 
transistor 509. The signal is then converted by an impedance 516 into a 
signal to be delivered from an output terminal 5 as an output voltage 
V.sub.OUT undergone the .gamma.-th power conversion. According to the 
configuration, when the collector current of the transistor 16 is small, 
namely, the current flowing through each of the transistors 500 and 501 is 
small, the base-emitter current of the transistors 500 and 501 approaches 
exactly to zero, thereby enhancing blockage or interruption of the 
transistors 500 and 501. As a result, according to the embodiment, even 
when the parameter .gamma. of the power characteristic of the conversion 
is set as .gamma..ltoreq.1, there can be suppressed the reduction in the 
output amplitude from the logarithmic converter due to the base current of 
the input transistors 504 and 505 of the variable gain amplifier. 
Consequently, a sufficient dynamic range can be reserved for each output 
signal appearing after the output port of the logarithmic converter. In 
this constitution, the voltage source 510 can be replaced with a 
resistance-type potential divider. Thanks to the internal impedance of the 
voltage source 510, even when the base voltage of the transistor 500 is 
varied, there occurs in a subsequent stage a common mode rejection in the 
differential variable gain amplifier or the antilogarithmic converter of 
the differential input type, and hence the .gamma.-th power conversion 
characteristic can be guaranteed. In addition, the similar advantageous 
effect is attainable by a circuit configuration developed by using, in the 
color corrector of FIG. 14, pnp-type transistors for the transistors 504 
and 505 constituting the variable gain amplifier and the polarity is 
reversed for the current sources 512 and 513 and the power source. That 
is, the base current of the transistor 504 flows in a direction to enhance 
interruption of the transistor 533, which consequently prevents the 
reduction in the output amplitude of the logarithmic converter. Moreover, 
using a pair of transistors having integrity with respect to 
characteristics thereof for the transistors 500 and 502, 501 and 503, 504 
and 505, 506 and 508, and 507 and 509, there can be exactly set an output 
amplitude according to adjustment of the current ID of the variable 
current source 515, thereby improving precision of the .gamma.-th power 
conversion. Furthermore, when each or some of the pairs of transistors is 
or are manufactured on an identical chip or arranged to be tightly 
adjacent to each other, there can be attained substantially an identical 
pn junction temperature for the paired transistors. This suppresses the 
drift in the conversion characteristic due to temperature and hence much 
more improves the conversion precision. It is consequently to be 
understood that the embodiment is suitable for an integrated circuit in 
which all transistors are fabricated in a chip. 
FIG. 16 is a circuit diagram of a color corrector in a third embodiment 
according to the present invention in which the transistors are kept 
protected even when a signal having a large amplitude is supplied as the 
input voltage V.sub.IN. The color corrector of FIG. 16 is constructed on 
the basis of the embodiment shown in FIG. 15. Namely, only a diode 541 or 
a series connection of the diode 541 and a resistor 542 is connected as a 
protection circuit between the base terminals respectively of the 
transistors 501 and 500. In the color corrector of FIG. 15, when a signal 
having a large amplitude is supplied to the input terminal 1 and the 
transistor is resultantly interrupted, the base current of the transistor 
503 flows into a connecting region between the base and the collector of 
the transistor 501. This increases the base terminal voltage of the 
transistor 501 to a value similar to the power source voltage Vcc and 
hence may possibly destroy the transistor 500. To cope with the 
difficulty, according to the color corrector of FIG. 16, a protective 
circuit including only a diode 541 or a series connection of the diode 541 
and a resistor 542 is connected in parallel to a point between the base 
terminals respectively of the transistors 501 and 500. Accordingly, when 
the base terminal voltage of the transistor 503 becomes higher than that 
of the transistor 500, the diode 536 is set to a conductive sate, thereby 
preventing a voltage equal to or more than the reverse breakdown voltage 
from being applied between the base and the emitter of at least one of the 
transistors 500 and 501. In this regard, due to the diode 541, the 
parasitic capacitance of the collector terminal of the transistor 16 is 
increased and hence the time constant becomes greater in the pertinent 
section. Consequently, there is disposed the resistor 542 to prevent the 
increase and thereby to keep the characteristic of the high-speed 
wide-range conversion. Moreover, it is to be appreciated that an area 
saving element in an integration circuit such as a Schottky barrier diode 
or a Zenner diode can be used for the diode 541. 
FIG. 17 is a circuit diagram showing a fourth specific embodiment of the 
color corrector of FIG. 13. According to an aspect of the embodiment 
having the aspect of that shown in FIG. 15, there is suppressed a 
deviation in the output voltage at a dc operation point which may take 
place when the gain .gamma. of the variable gain amplifier, namely, the 
.gamma.-th power of the conversion characteristic for the conversion is 
changed. The color corrector of FIG. 17 includes a logarithmic converter 
700 including transistors 520 and 521 as well as 522 and 523, a variable 
gain amplifier including transistors 504 and 505 and a variable resistor 
514, and an antilogarithmic converter including transistors 506 and 508 as 
well as 507 and 509. An input voltage VIN from an input terminal 1 is 
transformed into a current through the transistors 522 and 520. The 
current signal is passed via the transistors 523 and 521 to be obtained as 
a differential voltage between the bases respectively of the transistors 
522 and 523. In the variable gain amplifier including a differential 
circuit configured with the transistors 504 and 505, the signal is 
amplified by a differential gain .gamma. decided by the resistance values 
of resistors 518 and 519 and the variable resistor 514 to be applied 
between the bases of the transistors 506 and 508, respectively. Through 
the transistors 506 and 508 as well as 507 and 509, the amplified voltage 
of the differential signal is subjected to an antilogarithmic conversion 
to be a collector current of the transistor 509. The current signal is 
transformed by an impedance 516 into a voltage to be outputted from an 
output terminal 5 as an output voltage V.sub.OUT undergone the .gamma.-th 
power conversion. When the base-collector voltage of the transistor 522 
set by diodes 529 and 530 and a transistor 524 is substantially equal to 
that of the transistor 523 set by a transistor 525, the collector loss of 
the transistor 522 becomes similar to that of the transistor 523, thereby 
suppressing the variation in the output voltage at the dc operation point 
which possibly appears when the gain .gamma. of the variable gain 
amplifier, namely, the .gamma.-th power of the conversion characteristic 
for the conversion is changed. The variation is caused as follows. Since 
the collector loss considerably varies between the transistors 
constituting the logarithmic converter, there occurs a dc offset in the 
differential output of the logarithmic converter. The offset changes in 
relation to the variation in the gain .gamma. of the variable gain 
amplifier, which leads to the amplitude variation after the 
antilogarithmic conversion. The amplitude variation leads to an alteration 
of the dc operation point existing at a point, regardless of the value of 
.gamma., in association with the output from the antilogarithmic 
converter. Furthermore, employing a pair of transistors, namely, paired 
transistors for the transistors 524 and 525, 522 and 523, 520 and 521, 504 
and 505, 506 and 508, and 507 and 509, the output amplitude can be exactly 
established through adjustment of a current ID of a variable current 
source 515, which improves precision of the .gamma.-th power conversion. 
In addition, when each or a plurality of the pairs of transistors is or 
are formed in a chip or arranged to be closely adjacent to each other, 
there can be developed substantially the same pn junction temperature for 
the paired transistors, thereby suppressing the drift in the conversion 
characteristic due to temperature and further improving the conversion 
precision. Namely, it is to be appreciated that the embodiment is suitable 
for an integrated circuit. Moreover, according to the embodiment, to 
establish the base-collector voltage of the transistor 522, there are 
adopted the diodes 529 and 530 and the transistor 524. However, in place 
of the diodes 529 and 530, there may naturally be utilized resistors. 
Furthermore, since the transistor 525 configures an emitter follower 
circuit, there is attained an advantage that even when there is required a 
long connection wire to the variable gain amplifier in the subsequent 
stage, the output impedance is lowered to thereby send signals with a 
satisfactory stability. 
FIG. 18 shows a fifth specific embodiment of the color corrector in which 
the stability of the output voltage at the dc operation point is more 
improved in a case where the gain .gamma. of the variable gain amplifier, 
namely, the .gamma.-th power of the conversion characteristic for the 
conversion is changed. In the structure of the color corrector of FIG. 18, 
diodes 539 and 540 are connected between a base terminal of a transistor 
523 and an emitter terminal of a transistor 525, and a collector terminal 
of a transistor 521 is connected to an emitter terminal of a transistor 
537. Thanks to the constitution, even when a current source 527 has a 
small current value, discrepancy between collector losses respectively of 
transistors 522 and 523 can be minimized by setting a collector-emitter 
voltage of the transistor 523 to be higher than the voltage of the 
transistor 523. Moreover, an emitter-collector voltage of the transistor 
521 can be lowered by setting an emitter potential of a transistor 534 to 
an appropriate value to reduce discrepancy between collector losses of the 
transistors 520 and 521, respectively. This further improves the stability 
of the output voltage at the dc operation point when the gain .gamma. of 
the variable gain amplifier, namely, the .gamma.-th power of the 
conversion characteristic for the conversion is changed. In this 
embodiment, the diodes 539 and 540 are connected between the base terminal 
of the transistor 523 and the emitter terminal of the transistor 525. 
However, the similar advantage is naturally attained when resistors are 
used in place of the diodes. Furthermore, to obtain the same advantageous 
effect, in place of the transistors 537 connected to the collector 
terminal of the transistor 521, there may be employed a resistor with its 
another terminal connected to a low-impedance point such as the grounding 
point. 
FIG. 19 shows in a circuit diagram a sixth embodiment of the 
antilogarithmic converter shown in FIGS. 14A and 14B and FIGS. 15 to 18 in 
which the frequency band and the response speed are increased. In FIG. 19, 
the logarithmic converter and the variable gain amplifier are the same as 
those of FIG. 14 and hence the operation principle is identical to that 
described above. A signal multiplied by the differential gain .gamma. in 
the variable gain amplifier is applied between bases respectively of 
transistors 506 and 508. Through the transistors 506 and 508 and 543 and 
509, the amplified voltage of the differential signal is subjected to an 
antilogarithmic conversion to be a collector current of the transistor 
509. The signal is then transformed by an impedance element 516 into a 
voltage to be delivered from an output terminal 5 as an output signal 
V.sub.OUT undergone the .gamma.-th power conversion. Thanks to an action 
of a negative feedback loop ranging from a collector terminal of the 
transistor 543 via a base terminal of a transistor 544 and an emitter 
terminal thereof to a base terminal of the transistor 543 and a buffering 
action of the transistor 544 constituting an emitter-follower circuit, it 
is possible to minimize the actual time constant appearing in the base 
terminal of the transistor 544. This resultantly expands the frequency 
band and the response speed of the antilogarithmic converter. 
In this connection, it is to be appreciated that bipolar transistors 
constituting the logarithmic and antilogarithmic converters in each of the 
embodiments shown in FIGS. 4 to 19 can be substituted for devices such as 
MOSFETs (sub-threshold region) having an operational region developing a 
similar power and logarithm characteristics in relation to the 
voltage-current conversion. 
In each color corrector described above, when the input signal S.sub.IN 
takes a value "1" in the input port of the logarithmic converter 2, there 
is obtained a fixed output which is not influenced from the parameter 
.gamma.. In a system including the color corrector, the input/output 
operation point independent of the parameter .gamma. is effectively used 
as another characteristic adjusting point. In consequence, to sufficiently 
expand the application field of the color corrector, a fixed output 
independent of the parameter .gamma. is required to be obtained for any 
value of the input signal S.sub.IN. In this connection, description will 
be given of an embodiment of the color corrector. In this configuration, 
to convert the signal amplitude into an appropriate value, there is 
arranged an amplifier, an attenuator, or a variable gain amplifier in a 
first stage. This enables the color corrector to be applied to video 
apparatuses in a wide range. FIG. 20 shows in a block diagram a color 
corrector in a third embodiment according to the present invention. The 
configuration of FIG. 20 includes, in addition to the constitution of FIG. 
1, a signal amplitude adjuster 6 in a first stage thereof. Description 
will now be given of operation of the color corrector of the embodiment. 
FIG. 21 is a graph showing input-output characteristics of the color 
corrector shown in FIG. 20. In the graph of FIG. 21, the input S.sub.IN 
and the output S.sub.OUT of the color corrector are respectively 
represented by the ordinate and the abscissa and the adjusting parameter 
is expressed as .gamma.. In the case of the embodiment, assuming the gain 
of the signal amplification adjuster 6 to be K, when KS.sub.IN as the 
input signal to the logarithmic converter 2 takes a value "1" there is 
attained a fixed output 1.gamma. independent of the parameter .gamma.. 
Consequently, as shown in FIG. 21, when the input S.sub.IN is 1/K, the 
output value S.sub.OUT is fixed independently of the parameter .gamma.. 
Like in the circuit of FIG. 5 capable of processing input and output 
signals in the differential format, the input and output operation points 
are similarly fixed when the input S.sub.IN is -1/K as shown in FIG. 21. 
Next, description will be given of an embodiment of the color corrector 
applicable to an arbitrary video apparatus having input and output 
characteristics to be approximated to two kinds of power characteristic. 
FIG. 22 shows in a block diagram a color corrector in a fourth embodiment 
according to the present invention. Shifting the dc level of an input 
signal to the color corrector having the characteristics of input and 
output signals in the differential format shown in FIG. 21, there can be 
selectively attained two different input/output characteristics to be 
approximated to two kinds of power characteristics such as a 
characteristic of S shape and a characteristic of inverse S shape for an 
arbitrary input signal. For example, in the circuit of FIG. 5, while 
supplying a signal from a terminal 44, a dc voltage for the dc shift can 
be inputted to a terminal 45. However, as can be clear from a fact that 
each of the curves of input/ output characteristics is symmetric with 
respect to the origin as shown in FIG. 21, namely, according to the sign 
of the signal value, only two kinds of symmetric power characteristics are 
realized by shifting the dc level of the input signal. Namely, two kinds 
of arbitrary power characteristics cannot be obtained. To overcome this 
difficulty, in the color corrector of the embodiment shown in FIG. 22, at 
least one of the circuit blocks including a signal amplitude adjuster 7, a 
logarithmic converter 8, a variable gain amplifier 9, and the 
antilogarithmic converter 10 has an asymmetric input/output characteristic 
in which the characteristic varies depending on the polarity of the 
signal. Providing each circuit block with directivity or orientation 
according to the signal polarity as above, there can be independently 
established two kinds of .gamma. parameters and two kinds of input and 
output operational points fixed independently of the .gamma. parameters. 
In consequence, using the embodiment, there can be provided a 
high-precision color corrector suitable for a circuit to drive a liquid 
crystal panel or the like in which an asymmetric inverse S characteristic 
is required as the input/output characteristic of the video signal system. 
FIG. 23 shows in a circuit diagram the signal amplitude adjuster 7 and the 
logarithmic converter 8 in the first specific example of the color 
corrector shown in FIG. 22. In this example, the adjuster 7 and the 
converter 8 are assigned with mutually different input/output 
characteristics according to the the signal polarity as above. In the 
color corrector of FIG. 23, a signal delivered to an input terminal 1 of 
the signal amplitude adjuster 7 is subjected to an amplitude variation 
according to a level of the signal to be fed to bases respectively of 
transistors 134 and 145 of the logarithmic converter 8 in the subsequent 
stage. Moreover, the signal is subjected to a logarithmic conversion 
according to the signal level to be outputted from terminals 150 and 151 
so as to be inputted to the variable gain amplifier 9 in the next stage 
shown in FIG. 22. In this situation, when the input signal level of the 
adjuster 7 is higher than the voltage at a connecting point between diodes 
123 and 124, transistors 116 and 130 turn on to a conductive state such 
that the voltage of the input signal is converted into a current via a 
resistor 117. Conversely, the input signal level is less than the voltage 
at a connecting point between diodes 123 and 124, transistors 113 and 128 
turn on such that the voltage of the input signal is converted into a 
current via a resistor 129. The signal current is fed through transistors 
122 and 119 and output resistors 125 and 127 to be restored to a voltage 
signal so as to be fed to the logarithmic converter 8 in the following 
stage. In this operation, changing the resistance value of a variable 
resistor 1201, the voltage developed at a connecting point between diodes 
123 and 124 can be controlled. Moreover, setting the resistors 117 and 119 
to appropriate resistance values, there can be realized a desired 
input/output characteristic. Each of the diodes 111, 112, 123, and 124 
compensates for a temperature characteristic of the base-emitter voltage 
of a transistor connected to a base thereof. In addition, although current 
sources 118 and 121 function to guide signal currents to emitters 
respectively of the transistors 119 and 122, these sources 118 and 121 can 
be replaced with resistors. For the logarithmic converter 8 in the 
succeeding stage, a different input/output characteristic can be attained 
in a circuit configuration .thereof similar to that of the signal 
amplitude adjuster 7. When the base voltage of the transistor 134 is 
higher than that of the transistor 145, the transistors 134 and 147 are 
set to be conductive. Conversely, when the base voltage of the transistor 
134 is lower than that of the transistor 145, the transistors 145 and 133 
become conductive. According to the cases above, the voltage of the signal 
is converted into a current respectively via a resistor 146 or 136 to be 
delivered from a diode 137 or 139 as a voltage signal undergone a 
logarithmic conversion. A voltage source 138 sets the dc level of the 
output signal. Grounded-base transistors 140 and 143 guide, like the 
transistors 119 and 122, the signal current to the output side. Diodes 131 
and 132 as well as 148 and 149 compensate, like the diodes 134, for the 
temperature characteristic of transistors respectively connected thereto. 
Current sources 152 and 153 to bias the diodes connected thereto may 
possibly be replaced with resistors. It is to be understood that the above 
configuration having variable input/output characteristics is applicable 
also to the other circuit blocks. 
Subsequently, description will be given of an embodiment capable of easily 
developing, by use of a nonlinear portion of the input/output 
characteristic of an amplifier, an input/output characteristic approximate 
to the characteristic of power. FIG. 24 shows in a block diagram a color 
corrector in a fifth embodiment according to the present invention. In 
FIG. 24, an input signal S.sub.IN (+) supplied to an input terminal 1 is 
subjected to a nonlinear amplification through a differential amplifier 
155 to be an output S'.sub.OUT. The signal is then fed to a dc regenerator 
156 to be shifted to a desired dc level, thereby delivering an output 
signal from a terminal 157. To an inverse input terminal of the 
differential amplifier 155, there is connected a dc signal source 154 with 
a signal level S.sub.IN (-) to set a signal dynamic range. FIG. 25 shows 
an input/output characteristic of a general differential amplifier. In 
FIG. 25, the abscissa and the ordinate represent a differential input 
S.sub.IN (+)-S.sub.IN (-) and the output S'.sub.OUT, respectively. A 
characteristic curve 158 includes nonlinear sections enclosed with a 
broken line 160 and a dot-and-dash line 159, respectively. In consequence, 
changing a signal level of the dc signal source 154, a desired approximate 
power characteristic can be obtained according to an area of the nonlinear 
sections. To suppress the shift in the output operational point caused by 
the variation in the signal level of the dc signal source, the dc 
regenerator 156 is arranged in a stage following the differential 
amplifier 155. In addition, it is to be understood that the circuit system 
can be operated with the voltage and current signals and an inverse 
amplification can also be conducted in the signal amplification. Moreover, 
since ordinary amplifiers also have nonlinear regions in the end portions 
of the signal dynamic range thereof, the differential amplifier 155 can be 
substituted for an arbitrary amplifier. In such a case, an adder is 
disposed in a stage preceding the amplifier such that a signal produced 
from the dc signal source 154 is added by the adder to an input signal 
received from the input terminal 1 so as to deliver the resultant signal 
to the amplifier. 
FIG. 26 shows in a circuit diagram a first specific example of the color 
corrector shown in FIG. 24. In this example, a dc regenerator 156 includes 
a clamping circuit. In the color corrector of FIG. 26, an output signal 
from a differential amplifier 155 including transistors 161 and 162 is 
passed through a clamping circuit (namely, a dc regenerator) including a 
coupling capacitor 169, a clamping voltage source 171, and a switching 
circuit 170 to be fed to an output terminal 157. Operation of the color 
corrector will be described by reference to a signal waveform shown in 
FIG. 19. When a signal having a waveform 172 is supplied to an input 
terminal 1, a signal having a waveform 173 resultant from a nonlinear 
conversion appears from a collector of the transistor 161. For example, in 
a case where the switching circuit 170 is closed in synchronism with a 
peak voltage level of the signal waveform 173, there is attained from the 
output terminal 157, as indicated by a waveform 174, a signal of which the 
peak level is clamped to a voltage Vcc of the clamping voltage source 171. 
In this connection, the switching circuit 170 can be replaced with a diode 
175 or a switching transistor 176 by using terminals a and b. When the 
diode 175 is employed, the peak level of the output signal is 
automatically clamped to the voltage Vcc of the clamping voltage source 
171. When the switching transistor 176 is adopted, the clamping operation 
is carried out at a timing synchronized with a clamping pulse source 178. 
In a case where another output terminal 168 of the differential amplifier 
155 is connected to the clamping circuit, the output signal waveform is 
naturally reversed. 
FIG. 27 is a circuit diagram showing a second specific example of the color 
corrector shown in FIG. 24. In this example, the dc regeneration is 
accomplished according to a reverse output of the nonlinear output signal 
delivered from the differential amplifier circuit 155. In the color 
corrector of FIG. 27, a nonlinear signal is obtained from a non-reverse 
output terminal 157 of the differential amplifier 155 including 
transistors 179 and 181. Moreover, a dc regenerator 156 includes a 
resistor 185, a capacitor 186, a transistor 187, and a resistor 184 to 
attain a mean dc level of an inverse output signal from a low-pass filter 
including the resistor 185 and the capacitor 186. The mean dc level is 
applied via the transistor 187 and the resistor 184 to the output terminal 
157, thereby achieving the dc regeneration. In this situation, the mean dc 
level of the inverse output signal attained from the low-pass filter 
including the resistor 185 and the capacitor 186 is changed in association 
with a variation in the signal level of the dc signal source 154 to set a 
signal dynamic range. However, since the voltage drop appearing across the 
resistor 184 having a resistance value equal to that of the resistor 185 
is also varied to cancel out the variation in the mean dc level, the dc 
level of the output terminal 157 is fixed. In addition, changing the 
signal level of the dc signal 154, the nonlinear characteristic is also 
varied. However, the characteristic can be altered by changing the current 
values of bias current sources 182 and 183. For example, increasing the 
current value of at least one of the current sources 182 and 183, the 
value of the parameter .gamma. approaches one. 
Subsequently, description will be given of an embodiment in which a current 
signal is transmitted in a differential circuit configuration as above to 
achieve a high-speed signal conversion of an approximated power 
characteristic. FIGS. 28A and 28B respectively show the circuit 
configuration of a color corrector in a sixth embodiment according to the 
present invention and a graph of the input/output characteristic of the 
color corrector. In the color corrector of FIG. 28A, the dc regenerator 
156 is removed from the constitution of the embodiment shown in FIG. 17. 
In FIG. 28A, a signal current I.sub.IN is supplied via impedance elements 
189 and 190 including resistors to a common emitter terminal of paired 
differential transistors 192 and 193 so as to obtain an output signal 
current I.sub.OUT via a collector of the transistor 193. In the operation, 
since the signal is transmitted in the form of a current, it is possible 
to neglect influence from the time constant of each circuit, thereby 
achieving a high-speed transmission. As the input/output characteristic of 
the color corrector of the embodiment, there is obtained, a nonlinear 
conversion characteristic representatively indicated by characteristic 
curves 194 to 196 according to magnitudes respectively of a control 
voltage V.sub.32 and a bias voltage V.sub.32 as shown in FIG. 28B. 
Each color corrector described above is suitable for color correction in 
various video apparatuses having the nonlinear characteristic and/or 
systems including the apparatuses. For example, the color corrector is 
applicable to displays employing a Braun tube, a liquid crystal panel, a 
plasma display panel, an LED panel, or the like and video display 
apparatuses for a television receiver or the like. Moreover, the color 
corrector can be applied to video input devices such as a video camera and 
a scanner, video output devices including a printer, an electronic 
photograph, or the like, and/or a computer system, a broadcast and 
communication system, and the like using above devices. 
In addition, the color corrector according to the present invention can be 
configured in the form of a small-sized analog circuit and hence can be 
fabricated as a module or an integrated circuit for a function block of a 
signal processor. For example, setting the parameter .gamma. of the power 
characteristic to a value less than one, it is possible to compress the 
signal amplitude. When there is employed a combinational structure for 
compression and expansion of the signal amplitude as shown in FIG. 29, the 
signal dynamic range can be efficiently used in a signal processor A1 such 
as an amplifier and a high signal-to-noise ratio can be guaranteed. In 
FIG. 29, A2 indicates an amplifier compressor in which the parameter 
.gamma. of the power characteristic according to the present invention is 
set to a value less than one and A3 stands for an amplitude expander in 
which the parameter .gamma. is set to a value greater than one or to 
1/.gamma.. 
Moreover, in this regard, senses of sight and hearing have a logarithmic 
characteristic. Using a signal converter in which the parameter .gamma. of 
the power characteristic is set to a value less than one according to the 
present invention, there can be attained approximated signals associated 
with magnitudes of stimulation of these senses. FIG. 30 shows a sense 
simulation system having the configuration above. In FIG. 30, B1 denotes a 
sensor to obtain a magnitude of sense and B2 indicates a signal converter. 
Description will now be given of an embodiment in which the color corrector 
is applied to a television receiver adopting a color Braun tube which is a 
video display as one of the video apparatuses. FIG. 31 is a circuit 
diagram showing a television receiver in a seventh embodiment according to 
the present invention. In the television receiver 197 of FIG. 31, video 
signals of primary colors respectively corresponding to red, green, and 
blue are received from terminals 1R, 1G, and 1B and are then amplified to 
drive a color Braun tube 217. The video signals thus inputted are first 
subjected to such control operations common to the three primary colors as 
a contrast control and a brightness control. Namely, in variable gain 
amplifiers 198 to 200 each for conducting contrast adjustment, the video 
signals are amplified with an identical gain. Next, in adder circuits 201 
to 203 each accomplishing brightness adjustment, a dc shift associated 
with a voltage of a voltage source 207 is applied to the video signals 
according to an identical level. Subsequently, each of the video signals 
is subjected to a gamma correction, a drive control, and a cutoff control, 
which are to be individually achieved for each color video signal. That 
is, in a gamma correction circuit 204, 205, or 206 associated with a color 
video signal supplied thereto, there is compensated for a nonlinearity 
between the video signal and a luminance of light emission of the Braun 
tube in response to the video signal, thereby accomplishing a fine 
adjustment of white balance. In variable gain amplifiers 208 to 210 for 
drive adjustment and adder circuits 214 to 216 for cutoff adjustment, 
deviations between the drive voltages of the Braun tune 21 respectively 
associated with three primary colors are coarsely compensated for through 
approximation to a linear characteristic so as to adjust the white 
balance. In this regard, although the gamma correction, the drive control, 
and the cutoff control may be executed in an arbitrary order, the gamma 
correction process is first conducted in the circuit configuration of FIG. 
31. 
Description will now be given in detail of the method of adjusting the 
white balance through the gamma correction, the drive control, and the 
cutoff control described above. FIG. 32 is a characteristic graph showing 
relationships (input/output characteristics) between the input signal 
level of the gamma correction circuit and the drive voltage level of the 
color Braun tube of FIG. 31. In FIG. 32, the abscissa stands for the input 
signal level V.sub.IN Of either one of the gamma correction circuits 204 
to 206 and the ordinate represents the drive voltage level V.sub.OUT (the 
voltage is reversed if this is a cathode voltage) of the Braun tube 217. 
Assume that the input/output characteristic before the white balance 
adjustment including the gamma correction is represented by a curve 218 
drawn with a broken line and the cutoff, gamma, and drive adjusting points 
associated with the input signal level V.sub.IN are denoted as V.sub.CON, 
V.sub..gamma., and V.sub.DR, respectively. First, like in the conventional 
process of white balance adjustment, there are conducted a cutoff 
adjustment and a drive adjustment designated by arrows 221 and 222, 
respectively. Next, achieving a gamma correction designated by an arrow 
223, an objective characteristic curve 220 is obtained. In the operation, 
setting a fixed operation point independent of the parameter .gamma. to 
the drive adjusting point V.sub.DR, it is guaranteed that the gamma and 
drive adjustments are carried out independently of each other. Moreover, 
at the cutoff adjustment point, influences from the parameter .gamma. is 
inherently suppressed as shown in FIG. 23. However, to develop a higher 
precision for the white balance adjustment, the cutoff adjustment point 
V.sub.CON is set to a lower value and/or the adjustment is repeatedly 
effected. 
The construction of FIG. 31 is also applicable to other video display 
apparatuses including, for example, a display and a television receiver 
utilizing as a display element thereof a liquid crystal panel, plasma 
display panel, and/or an LED panel. For example, when a liquid crystal 
panel is adopted as the display element, the color Braun tube 217 of FIG. 
31 is substituted for a liquid crystal panel. Moreover, as the gamma 
correction circuits 204 to 206, there are disposed color correctors 
developing an input/output characteristic to be approximated to two kinds 
of power characteristics as shown in FIG. 22. Description will be now 
given of a method of adjusting the white balance through the gamma 
correction, the drive control, and the cutoff control when a liquid 
crystal panel is employed as the display device. FIG. 33 is a 
characteristic graph showing relationships (input/output characteristics) 
between the input signal level of the gamma correction circuit and the 
drive voltage level of the liquid crystal panel in a case where the liquid 
crystal panel is used as the display element. First, in a stage in which 
the amplification gain is set to a minimum value through the drive 
control, a cutoff control is accomplished to set the drive voltage level 
V.sub.OUT to a standard level V.sub.COUT to set an inflection point of a 
target characteristic curve 226 (arrow 227). Subsequently, the input 
reference level V.sub.CIN is set to the standard inflection point to 
thereby attain a characteristic 224 draw with a dot-and-dash line (arrow 
234). In this connection, for example, when the specific example of FIG. 
23 is used as the color corrector of FIG. 22, the input reference level 
V.sub.CIN can be established as described above by adjusting the variable 
resistor 1201 and the like of FIG. 23. Next, to guarantee independence of 
the gamma correction at the cutoff adjusting point, there is adjusted the 
gain K.sub.1 on the negative polarity side of the signal amplitude 
adjuster 7 having directivity according to the signal polarity as shown in 
FIG. 23 (arrow 228). Similarly, to retain independence of the gamma 
correction at the drive adjusting point, there is adjusted the gain 
K.sub.2 on the positive polarity side of the signal amplitude adjuster 7 
having directivity according to the signal polarity as shown in FIG. 23 
(arrow 229). However, in the designing process, it may also be possible to 
determine, in association with the signal dynamic range of the circuit, 
the setting of the input reference level V.sub.CIN and the adjusting 
process to reserve independence of the gamma correction at the cutoff and 
drive adjusting points, thereby unnecessitating the setting and the 
adjusting process. Subsequently, the cutoff and drive adjusting steps are 
effected as indicated by arrows 230 and 231 to realize a characteristic 
curve 255 drawn with a broken line. Finally, gamma corrections are 
achieved as indicated by arrows 232 and 233 to obtain a target 
characteristic 226, thereby completing the white balance adjustment. 
Next, description will be given of an embodiment capable of achieving 
automatic adjustment of the gamma correction, drive adjustment, and cutoff 
adjustment. According to the embodiment, in a television receiver 
employing a color Braun tube, the gamma correction, the drive control, and 
the cutoff control are automatically accomplished. FIG. 34 shows a circuit 
diagram of a television receiver in a eighth embodiment according to the 
present invention. In FIG. 4, there are missing the common circuit 
sections related to the contrast and brightness control operations. In the 
television receiver of FIG. 34, video signals of three primary colors 
inputted thereto are respectively fed through signal switching circuits 
235 to 237 respectively to gamma correction circuits 239 to 241 each 
including the color corrector above. Thereafter, the video signals are 
respectively passed through variable gain amplifiers 208 to 210 for drive 
adjustment, adder circuits 214 to 216 for cutoff adjustment, and beam 
current sensor 251 to 253 to a cathode of a color Braun tube 217. 
Incidentally, the deflection modulator 267 is not necessarily required to 
be disposed in a television receiver of an ordinary over-scan scheme, 
which will be described later. Next, description will be given of 
operation of the automatic adjustment by reference to FIG. 35. FIG. 35 is 
a signal timing chart showing timings of primary signals and operations of 
FIG. 34. In FIG. 35, a waveform (a) represents an input video signal of a 
primary color, whereas a pulse waveform (b) stands for a blanking signal. 
First, when signal switching circuits 235 to 237 are respectively set to 
connecting points 1 as can be seen from a waveform (c) in a range of time 
from immediately after a vertical blanking period to immediately before a 
display period. As a result, a reference signal S1 outputted from a 
reference signal source 238 as indicated by a waveform (g) is delivered 
via the signal switching circuits 235 to 237 to the gamma correction 
circuits 239 to 241. The sensor circuits 251 to 253 then sense resultant 
beam currents and supply sensed values to comparators 255 to 257, 
respectively. On the other hand, a comparison signal source 354 creates a 
comparison signal $2 as denoted by a waveform (h) at a timing synchronized 
with the reference signal S1 to feed the signal $2 to the comparators 255 
to 257. In the comparators 255 to 257, the inputted sense signals are 
respectively compared with the signal $2 such that control signals 
representing the respective comparison results are delivered to switching 
circuits 258 to 260, 261 to 263, and 264 to 266, respectively. In 
synchronism with the reference signal S1, the cutoff adjusting switches 
258, 261, and 264 are turned on at a timing indicated by a waveform (d), 
the drive adjusting switches 259, 262, and 265 are turned on at a timing 
indicated by a waveform (e), and the gamma correction switches 260, 263, 
and 266 are turned on at a timing indicated by a waveform (f). In 
consequence, according to the timings above, the respective control 
signals are sequentially supplied from the gamma correction circuits 239 
to 241, the drive adjusting amplifiers 208 to 210, and the cutoff 
adjusting adders 214 to 216 such that for the respective control 
operations, the cutoff adjustment, the drive adjustment, and the gamma 
correction are repeatedly achieved in this order. Furthermore, the control 
signals obtained in the cutoff adjustment, those attained in the drive 
adjustment, and those developed in the gamma correction are kept in the 
capacitors 248 to 250, 245 to 247, and 242 to 244, respectively for one 
vertical synchronization period. During the automatic adjusting operation 
above, the respective adjustments are repeatedly effected. Consequently, 
only when the circuit system is configured in conformity with a 
convergence condition, it is not necessary to guarantee the independence 
of the gamma correction of the cutoff and drive adjustments. 
In the operation, as shown in the waveform (c) of FIG. 35, the beam current 
is to be supplied according to the reference signal S1 during a 
non-display period. Consequently, this operation cannot be directly 
applied to video display apparatuses such as a computer display and a 
monitor system of a broadcast station using an under-scan scheme in which 
the overall scanning screen is displayed in the effective display area of 
the Braun tube for the following reasons. When the beam current flows in 
response to the reference signal S1, there occurs light emission in the 
effective display screen to resultantly gives an uncomfortable feeling to 
the user; moreover, there may possibly take place deterioration of 
fluorescent materials of the Braun tube. Description will now be given of 
the configuration also applicable to the video display apparatus of the 
under-scan scheme. Namely, in the configuration of FIG. 32, there is 
additionally disposed a deflection modulator 267 to be connected via 
control signal lines 268H and 268V to a deflection yoke 269 of the color 
Braun tube. The deflection modulator 267 modulates a deflection current 
supplied to a vertical or horizontal deflection coil of the deflection 
yoke 269 to remove the electron beam caused by the reference signal S1 
from the effective display screen, thereby preventing the light emission. 
This operation will now be described by reference to FIG. 36. FIG. 36 is a 
signal timing chart showing timings of the primary signals and sections of 
FIG. 34 including the deflection modulator 267. Waveforms (a) to (c) of 
FIG. 36 are equal to waveforms (a) to (c) of FIG. 33, whereas a waveform 
(d) of FIG. 36 represents a signal timing of the vertical deflection 
current. That is, modulating the vertical deflection current by the 
modulator 267, the vertical deflection current is temporarily increased as 
denoted by a broken like 271 relative to the state indicated by a solid 
line 270. Resultantly, like in the video display apparatus of an over-scan 
scheme employed in an ordinary television receiver or the like, it is 
possible to delete the electron beam produced in response to the reference 
signal S1 from the effective display screen of the Braun tube. Moreover, 
it is to be appreciated that the embodiment coping with the under-scan 
scheme is applicable to a video receiver of the over-scan scheme. 
FIG. 37 shows in a circuit diagram an example of the configuration of the 
deflection modulator 267. In FIG. 37, on receiving a vertical retrace 
signal, an additional deflection signal generator C1 produces a signal 
V.sub.A. The additional deflection signal V.sub.A is added by an adder C2 
or C3 to the deflection signal to resultantly obtain a vertical deflection 
current indicated by the waveform (d) of FIG. 36. 
Next, description will be given of an embodiment capable of preventing the 
light emission occurring in response to the reference signal S1 like in 
the case above without using the deflection modulator 267 of FIG. 34. FIG. 
38 shows in a circuit diagram of a video receiver such as a computer 
display in a ninth embodiment according to the present invention. Also in 
FIG. 38, the common control circuits including the contrast and brightness 
control sections are missing. The video receiver of FIG. 38 includes a 
digital control section including a bus line 277 and a central processing 
unit (CPU) 274, an input/output (I/O) interface 276, a memory 273, and a 
timing pulse generator 275 which are connected to each other via the bus 
line 277. In addition, to connecting pints of the capacitors 242 to 250 
keeping the control signals attained in the respective adjustments in FIG. 
34, there are respectively connected digital-to-analog (D/A) converters 
281 to 289. Furthermore, outputs from the comparators 255 to 257 are 
connected to A/D converters 280 to 292, respectively. Description will now 
be given of operation of the video receiver of FIG. 38. First, in a 
pre-adjusting stage prior to an ordinary operation of the video receiver, 
like in the video receiver of FIG. 34, switching circuits 235 to 237 are 
respectively set to connection points 1 to supply the reference signal S1 
from a reference signal source 238 to gamma correction circuits 239 to 
241, thereby accomplishing the cutoff adjustment, the drive adjustment, 
and the gamma correction. In this operation, control signals produced from 
the comparators 255 to 257 are fed via the A/D converters 290 to 292, the 
signal line 279, and the I/O interface 276 to the digital control section 
so as to write the signals in a memory 273 of the digital controller in 
synchronism with the scanning period. In this regard, the signal switching 
circuits 235 to 237 and switches 258 to 266 are controlled according to 
timing pulses from the timing pulse generator 275. Thereafter, in the 
ordinary operation of the video receiver, the switching circuits 235 to 
237 are respectively set to connection points 0 to prevent the reference 
signal S1 from being delivered from the reference signal source 238 to the 
gamma correction circuits 239 to 241. At a timing synchronized with the 
scanning period, the control signals beforehand obtained in the 
pre-adjustment are read from the memory 273 to be supplied via the I/O 
interface 276, a signal line 280, the D/A converters 281 to 289 
sequentially to the gamma correction circuits 29 to 241, the drive 
adjusting amplifiers 208 to 210, and the cutoff adjusting adders 214 to 
216, thereby achieving the respective adjustments. Consequently, in an 
ordinary use of the video receiver, it is only necessary to read the 
control signals beforehand obtained in the pre-adjustment from the memory 
273. Namely, since the operation to supply the beam current in response to 
the reference signal S1 during the non-display period is unnecessitated, 
there does not occur the light emission due to the reference signal S1 in 
the effective display screen. In addition, using the control signals of 
the respective adjustments stored in the memory 273, it is possible to 
compensate for unevenness in the brightness as well as colors (color 
shading). Moreover, setting in advance data of historical or secular 
changes in the form of coefficients, the control signals of the respective 
adjustments can be altered according to the accumulated utilization period 
of the video receiver, thereby compensating for the secular variation in 
characteristics of the video receiver. 
In this connection, for the compensation and adjustment of the nonuniform 
brightness and colors, the correction is required to be achieved in 
association with a display position of each pixel on the display screen. 
Description will now be given of an embodiment capable of achieving the 
correction in association with a display position of each pixel on the 
display screen. FIG. 39 is a configuration diagram showing a video 
receiver in a tenth embodiment according to the present invention. In FIG. 
39, a circuit block 293 is connected between terminals 1R, 1G, and 1B and 
the color Braun tube of FIG. 38. In the video receiver of FIG. 38, when 
conducting the pre-adjustment before an ordinary operation thereof, the 
beam current is sensed by the sensor circuits 251 to 253 to thereby sense 
colors on the display screen. In contrast thereto, according to the video 
receiver of FIG. 39, the correction is effected in association with the 
display position of each pixel. Consequently, in the pre-adjustment, a 
video camera 295 or a photodetector 294 is adopted in place of the beam 
current sensors 251 to sense colors, namely, luminance of the light 
emission on the display screen. That is, when using the video camera 295 
including an imaging tube or a charge-coupled device (CCD) sensor, the 
sense output is supplied to the comparators 255 to 257 of FIG. 38 in 
synchronism with the image scanning operation of the video camera 295. 
Control signals obtained as a result of the above operation are delivered 
to the digital control section to enhance an automatic high-speed 
compensation of the nonuniform luminance and the color shading. Moreover, 
when using the photodetector 294 representatively a tintometer or a 
photometer such as a spectrophotometer, the sense output signal is 
inputted to the comparators 255 to 257 correspondingly to the sense 
region. Resultant control signals are sent to the digital control section 
to achieve a high-precision compensation of the nonuniform luminance and 
colors. As the sense outputs from the video camera 295 and the 
photodetector 294, in addition to the primary color signals obtained from 
signal lines 296R, 296G, and 296B, there can be considered luminance 
signals, color signals, and signals corresponding to color stimulation 
values X, Y, and Z attained from signal lines 297Y and 297C. It is to be 
appreciated that the circuit block 293 includes an interface section for 
the sense output signals. Moreover, using the video camera 295 and 
photodetector 294, it is also possible to conduct a purity compensation in 
association with influences from the convergence and terrestrial 
magnetism. In addition, imaging an object printed on a printing form by a 
printer by the video camera, a scanner, or the like in place of display 
pixels of the Braun tube or the like, it is possible to conduct a color 
compensation for other imaging device. Alternatively, using a reference 
video camera 295, imaging results of a plurality of video cameras can be 
compared with each other for correction, thereby conducting the color 
compensation for other imaging device. 
FIGS. 40A and 40B show examples of adjusting display patterns to be 
presented on the color Braun tube in the pre-adjustment of the video 
receiver of FIG. 39. In the adjusting pattern of FIG. 40A, there is 
displayed in an effective display area 298 a drive adjusting 
high-luminance window 299, a gamma adjusting middle-luminance window 300, 
and a cutoff adjusting low-luminance window 301. The window patterns can 
be displayed, in addition to the vertical arrangement as shown in FIG. 
40A, in an arrangement that the patterns are sequentially displayed in the 
same location, thereby suppressing influences from the nonuniform 
luminance and colors. The patterns can be displayed in an arbitrary color 
such as white, a primary color, or the like; moreover, these colors can be 
sequentially displayed. Measuring the window patterns by a photodetector 
or the like, it is possible to conduct the color correction such as the 
white balance adjustment including the gamma correction. Moreover, in the 
adjusting pattern of FIG. 40B, window patterns 302 are displayed in a 
central location and in four corners of the effective display area 298. 
These window patterns may be displayed in an arbitrary color such as white 
or a primary color and is suitable for detection of the nonuniform 
luminance and colors. Furthermore, sequentially displaying the window 
patterns, the nonuniform luminance and colors can be compensated for by a 
single photodetector or the like. 
Referring now the color reproduction space shown in FIG. 41, description 
will be given of an application in which the color corrector is employed 
in a video display apparatus, a video input device, and/or a video output 
device. In FIG. 41, the plane defined by the x and .gamma. axes represents 
a chromaticity plane, whereas the ordinate stands for a display brightness 
at a display position for the video display apparatus, brightness of an 
obtained image for a video input device such as a scanner or an imaging 
device, and a printing lightness for a video output device such as a 
printer. In general, the color reproduction space of a video apparatus, 
the range of chromaticity is narrowed according to increase in brightness 
or the like as representatively indicated by a solid line 307. When 
conducting a color adjustment in a video apparatus according to a 
conventional technology, there are employed as two adjusting target points 
chromaticity points 311 and 312 in chromaticity regions 308 and 309 
related to a low luminance BCO and a high luminance BDR. Consequently, the 
color reproduction characteristic after the color adjustment becomes 
disadvantageously apart from an appropriate chromaticity point 313 in a 
middle luminance By as indicated by a curve 314 drawn by a 
two-dots-and-dash line. In accordance with the present invention, also in 
the chromaticity region 310 of the middle luminance B.gamma., the color 
adjustment can be achieved as indicated by an arrow 3141 with the 
chromaticity point 313 set as a third target adjusting point. As a result, 
through the color adjustment, there can be obtain a high-precision 
characteristic of color reproduction as indicated by a solid line 315. 
Subsequently, influences of external lights upon the video apparatus will 
be described by reference to a chromaticity diagram in an x-y plane of 
FIG. 42. In the diagram of FIG. 42, a triangular region 317 in an actual 
color area 316 represents a color reproduction range in a state free of 
external lights. A reference white chromaticity point 320 of the video 
apparatus in the absence of external lights can be substantially retained 
even when there exists a strong external light only if the external light 
has a color which can be approximated to a reference white color. However, 
due to influences from the external light, the color reproduction range is 
minimized to a region enclosed by a broken line 319. In a case of an 
external light primarily including components of green, although the color 
reproduction range is slightly reduced to an area enclosed by a 
dot-and-dash line 318, a chromaticity point 321 of the white display is 
considerably moved toward green. In a conventional video display, 
typically, a television receiver or a computer display, to overcome this 
difficulty, there are employed countermeasures, for example, to increase 
the brightness control value and the contrast control value against 
influences from illumination of external lights. However, influences from 
chromaticity of external lights cannot be compensated for. 
Next, description will be given of an embodiment capable of compensating 
for also influences from chromaticity of external lights. FIG. 43 shows in 
a perspective view a video display as a 11th embodiment - according to the 
present invention. As can be seen from FIG. 43, the video display 322 
using the color corrector above includes a display screen 322, external 
light sensors 330 and 331 for measuring chromaticity and brightness of 
external lights, a mouse 327, a keyboard 326, a computer 325, and a user 
324 operating the computer 325 via the mouse 327 and the keyboard 326. In 
addition, as shown in FIG. 43, the sensors 330 and 331 are arranged at 
positions respectively over and below the display screen. Thanks to the 
horizontally shifted arrangement of the sensors 330 and 331, it is also 
possible to estimate distribution of external lights on the display screen 
323. In other words, the color correction can also be achieved on 
assumption that differences between measurement results from a plurality 
of external light sensors are uniformly distributed between the 
measurement points. In such a case, like in the compensation for the 
nonuniform brightness and color unevenness or shading, it is possible to 
conduct the correction in association with the display position. 
Furthermore, description will be given of another embodiment capable of 
compensating for influences from external lights. FIG. 44 is a perspective 
view showing a video display as an 12th embodiment according to the 
present invention. As shown in FIG. 44, the video display 322 to which the 
color corrector is applied includes external light sensors 335 and 336 on 
a filter 332 to prevent external light from reflecting on the display 
screen and for prevention of possible injuries to health of the user due 
to static electricity from the display screen. Signals from the sensors 
335 and 336 are delivered via a signal line to a connector 339 on a tilt 
display-stand 338 so as to be sensed by an appropriate sensor. Recently, a 
filter (332) of this type is mounted on the video display 322 by retainers 
333 and 334 in many cases. Consequently, disposing on the filter 332 a 
plurality of external light sensors in the positional relationships like 
those of the sensors 335 and 356, the correction can be conducted for each 
display position like in the the embodiment of FIG. 43. In this 
connection, the sensors 334 and 336 may be fixed, for example, with pasted 
onto a surface of the filter 332 or buried therein. Furthermore, similarly 
arranging photodetectors on a display side 322 of the filter 332, the 
display color can be detected for correction thereof. Also in the case of 
display color sensors, like in the external light sensors, there may be 
disposed a plurality of display color sensors in consideration of the 
positional relationships to achieve the correction for each display 
position. Moreover, signals attained from the color sensors may be sent to 
the display 322 via a signal line integrally disposed in the retainers 333 
and 334 or may be sensed by a wireless system employing a radio wave, an 
infrared ray, or the like. 
Considering distance between the video display and the user, the quantity 
of light observed by the user is minimized as the distance increases and 
hence it is necessary to set the contrast control to a higher value, 
thereby increasing the display brightness. Description will now be given 
of means for measuring the distance between the user and the video display 
of FIG. 43. The user 324 operates the mouse 327 and the keyboard 326. 
Consequently, distances respectively between the user 324 and the computer 
325 and the display 322 can be obtained by measuring lengths of cords or 
wirings of the mouse 327 and keyboard 326, respectively. In short, assume 
that the cords respectively of the mouse 327, the keyboard 326, and the 
computer 325 are accommodated respectively therein in a rolled state. In 
operation, when the cords are unrolled, the lengths respectively of the 
unrolled portions thereof need only be determined according to the number 
of rotations or divisions or marks beforehand disposed on the cords, 
thereby measuring the distances above. Alternatively, infrared ray sensors 
may be arranged at the positions of the external light sensors 330 and 331 
and on the filter 332 of FIG. 44 to measure the distances. 
Next, description will be given of a case in which a video display using 
the color corrector is applied to a multimedia system. FIG. 45 shows in an 
explanatory diagram an example of a display screen presenting images 
according to a multiwindow scheme on a video display employed in a 
multimedia system. In a multimedia system, it is necessary to process 
image or video information items received from a plurality of video media. 
Consequently, as shown in FIG. 45, there is conducted in many cases a 
multiwindow image presentation in which video display windows 304 to 342 
are arranged in an effective display area 298 of the video display. 
However, since the color setting may occasionally vary between the windows 
340 to 342, there possible occurs destruction of the color specification 
in windows of the conventional video display depending on cases. That is, 
the color temperature of white display is set to, for example, 9300K in 
conventional computer displays in most cases such that the images are 
usually processed also according to the color temperature in the computer. 
However, for television images conforming to the NTSC and MUSE, the color 
temperature of white display is set to 6500K. This consequently leads to a 
problem that when the television images according to the NTSC and MUSE are 
displayed on the computer display, there is resultantly developed a bluish 
hue. To overcome this difficulty, there can be considered a countermeasure 
to control the color setting of the computer display together with the 
display image. However, in the case of the multiwindow image display as 
shown in FIG. 45, a plurality of color setting screens possibly exist in 
an identical screen and hence it is impossible to correct the color 
setting for all image display windows. 
Description will now be given of an embodiment capable of solving the 
problem of the multiwindow image display. FIG. 46 is a configuration 
diagram showing a video receiver in a 13th embodiment according to the 
present invention. In FIG. 46, the video display 343 includes a color 
Braun tube 217 and a color corrector 344 of the type described above. A 
video signal source 345 such as a computer is disposed at a position 
external with respect to the display 343 as indicated by a dot-and-dash 
line 346. Alternatively, the source 345 may be arranged therein as denoted 
by a portion of the diagram in which the line 346 is missing. Video 
signals sent from the signal source 345 are delivered via signal lines 348 
and 350 to a color corrector 344 to be subjected to a color setting 
correction, thereby supplying the corrected signals to the Braun tube 217. 
The color setting operation in this situation is accomplished in response 
to a control signal sent from the signal source 345 via a signal line 347 
to the color corrector 344. In this connection, the control signal for the 
correction is transmitted in synchronism with a display position of the 
video signals. In the operation, the color setting may be corrected by 
controlling the amplitude ratios between the primary color signals in the 
video signal source 345. However, when the amplitude ratio between the 
primary color signals is thus adjusted, it is necessary to reserve a 
follow-up range for the color setting. Consequently, the signal input 
dynamic range of the video display 343 cannot be efficiently utilized and 
the display brightness is inevitably lowered. In contrast thereto, when 
the color setting correction is accomplished in the display 343 as 
described above, the dynamic range can be effectively used to guarantee 
the maximum display brightness. As a result, it is possible to correct the 
color setting for all video display windows. 
Description will be given of another embodiment capable of removing the 
problem of the multiwindow video presentation. FIG. 47 shows the 
configuration of a video receiver in a 14th embodiment according to the 
present invention. In FIG. 47, a mixed output terminal 357 of a video and 
control signal source 352 including a video signal source 345, a control 
signal source 356 for color setting, and a mixer 355 is connected to an 
input terminal 358 of a video display 351 including a control signal 
separator 359, a color corrector 362, and a color Braun tube 217. The 
video signal source produces a video signal, whereas the control signal 
source 356 outputs a control signal for color setting correction. In the 
mixer 355, for example, a carrier of which a frequency varies according to 
the color setting is modified on the basis of the video signal or the 
control signal having a dc value associated with the color setting is 
added to the video signal to obtain a mixed output signal. The resultant 
signal is delivered from the output terminal 357. In the control signal 
separator 359, the mixed output signal received from the input terminal 
358 is modified on the basis of a demodulation frequency according to the 
color setting and the added dc value is sensed, thereby separating the 
control signal for color setting correction from the video signal. The 
obtained control signal and video signals are sent respectively via signal 
lines 360 and 361 to the color corrector 362. The signal undergone the 
color setting correction is passed to the color Braun tube 217. According 
to the embodiment, the color setting can be corrected for all video 
display windows without increasing the number of signal lines to transmit 
control signals for color setting correction between the video and control 
signal source 352 and the video display 351. 
According to the present invention, there can be provided a color corrector 
in which the correction precision of the gamma correction circuit is 
increased and the parameter .gamma. of the correction circuit can be 
easily changed. In consequence, applying the color correction of the 
present invention, the high-fidelity color reproduction is enabled between 
the transmission and reception systems, thereby improving performance and 
functions of various video apparatuses. 
While the present invention has been described with reference to the 
particular illustrative embodiments, it is not to be restricted by those 
embodiments but only by the appended claims. It is to be appreciated that 
those skilled in the art can change or modify the embodiments without 
departing from the scope and spirit of the present invention.