Method and apparatus for RF input coupling for inductive output tubes and other emission gated devices

An input circuit of a microwave amplification tube achieves improved instantaneous bandwidth. By directly coupling the transmission line carrying a modulating radio frequency signal to a control grid, a low-Q input circuit is created that increases the fractional bandwidth of the system. A resonant cavity may be used to generate a voltage across the gap between the cathode and the control grid. Alternative geometries are presented whereby the electron beam is emitted from a cathode connected either to the center conductor of the transmission line or to the outer conductor of the transmission line. Alternatively, the electric field of the radio-frequency signal propagating through the transmission line may be used to create a voltage across the gap between the cathode and the control grid without using a resonant cavity. Likewise, alternative geometries are presented by which the electron beam is emitted from a cathode connected either to the center conductor or to the outer conductor of the transmission line.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to microwave amplification tubes, such as an inductive output tube (IOT), and, more particularly, to an input circuit for an IOT or other emission-gated device providing improved instantaneous bandwidth.

2. Description of Related Art

It is well known in the art to utilize a linear beam device, such as a klystron or traveling wave tube amplifier, to generate or amplify a high-frequency RF signal. Such a device generally includes an electron-emitting cathode and an anode spaced therefrom. The anode includes a central aperture, and by applying a high voltage potential between the cathode and anode, electrons may be drawn from the cathode surface and directed into a high-power beam that passes through the anode aperture. One class of linear beam device, referred to as an inductive output tube (IOT), further includes a grid disposed in the inter-electrode region defined between the cathode and anode. The electron beam may thus be density modulated by applying a radio-frequency (RF) signal to the grid relative to the cathode. After the anode accelerates the density-modulated beam, the beam propagates across a gap provided downstream within the IOT, and RF fields are thereby induced into a cavity coupled to the gap. The RF fields may then be extracted from the output cavity in the form of a high-power, modulated RF signal.

More particularly, an IOT, as well as other emission-gated microwave amplifiers, use density modulation to establish an AC current Jbon the electron beam directly at the cathode surface. This current is subsequently converted to RF energy through the Jb·Ecinteraction with the output circuit field, Ec. Density-modulated amplifiers are highly efficient, even when operated in the linear region. Direct modulation of the beam at the cathode also enables compact device size.

In most density-modulated devices, RF gating of the electron emission is accomplished via an input cavity structure with a high-electric-field region situated between the cathode surface and a control grid. Energy from the signal generator is coupled into the input circuit, modulating the electron beam at the grid-to-cathode (g-k) gap. The basic elements of the input circuit are a resonant cavity, a coupled transmission line and a DC block. The gain-bandwidth product is limited by the interaction impedance R/Q·Q, where R/Q is the shunt impedance across the g-k gap, primarily determined by the gap geometry, and Q is the quality factor. The Q, proportional to the ratio of stored energy to dissipated power, determines the bandwidth of interaction between the drive signal and the electron beam. The power is dissipated by cavity ohmic losses, beam loading and external loading. The total Q is thus the parallel combination of the ohmic quality factor Q0, the beam loading quality factor Qband the external quality factor Qext. When heavily loaded by the generator impedance through the transmission line, the cavity is strongly coupled and has a correspondingly low Qext. This reduces the total Q, which increases the bandwidth.

The input resonant cavity can be modeled as a parallel RLC circuit. The beam is included as a shunt impedance and the connection to the drive line is represented by a transformer with a turns ratio of N. The Qextis related to the turns ratio by:
N2Z0=R/Q·Qext,
where Z0is the characteristic impedance of the input transmission line. Driven at its resonant frequency ω0, the cavity presents a purely resistive load of magnitude R/Q·Q to the signal generator, where R/Q is the shunt impedance across the g-k gap. As the drive frequency is shifted away from ω0, the load becomes increasingly reactive, and the resistive component decreases. At a small offset Δω from the center frequency, the load impedance is given by:

When the real component of the load impedance has dropped to half of its value at resonance, or R/2, the power delivered by the generator will be halved. This occurs when Δω/ω0=1/(2Q). Hence, the fractional bandwidth of a resonant cavity, defined as the distance between the two half-power points divided by ω0, is given by the reciprocal of the total quality factor (1/Q).

The coupling transformer connecting the signal generator to the resonant cavity is typically implemented using an inductive loop to transfer power from the signal generator to the cavity. The degree of coupling is proportional to the ratio of the magnetic flux enclosed by the inductive loop to the total flux in the cavity. A resonant cavity is formed around the electron gun in the IOT, with the g-k gap supporting the electric fields that modulate the electron beam. The electron beam passing through the grid is bunched at the frequency of the input signal. Electrons are accelerated towards a positively biased anode before their energy is extracted by the output circuit. For existing IOT applications, such as UHF television broadcast, loop coupling provides adequate bandwidth of a few percent. Practical limits on the loop size prevent substantially larger bandwidths from being achieved. Hence, if a wide-bandwidth IOT were possible, the compactness and linearity of this device would make it an attractive option for many other applications.

Accordingly, it is highly desirable to improve the instantaneous bandwidth of the input circuit of an IOT or other density-modulated device.

SUMMARY OF THE INVENTION

The instantaneous bandwidth achievable in an IOT or other density-modulated device is increased by employing an input circuit that directly couples the radio frequency signal carried by an input coaxial transmission line to the control grid. Such a directly coupled system comprises a coaxial transmission line with one conductor connected directly to the cathode and the other connected directly to the control grid, DC isolation being provided by an appropriately located DC block. Intermediate coupling methods, such as inductive loops or capacitive probes, are not used. Several methods exist for implementing the directly coupled system. One class of implementations utilizes a resonant cavity to generate a voltage between the cathode and the control grid. In its most basic topology, the center conductor of the transmission line is connected to the cathode, while the outer conductor of the transmission line is connected to the outside wall of the resonant cavity, the outside wall also serving to support the control grid and to provide an electrical connection between the outer conductor and the control grid. In another topology employing a resonant cavity, the cathode takes the form of an annular ring supported by an annular cathode support structure within the resonant cavity. The outer conductor of the coaxial transmission line is connected to the cathode support structure. The center conductor of the coaxial transmission line extends through the center of the resonant cavity and connects to the top of the cavity, which also serves as a grid support structure, holding an annular control grid in place in close proximity to the cathode and providing an electrical connection between the grid and the center conductor of the transmission line.

In both of these topologies, the impedance mismatch between the coaxial transmission line and the resonant cavity can be tuned by employing several techniques. First, an iris can be positioned at the location where the outer conductor of the coaxial transmission line joins the resonant cavity. The iris has an opening with a diameter that is smaller than that of the outer conductor of the transmission line but larger than the diameter of the center conductor, allowing the center conductor to pass through the iris. The effect of the iris is to change the magnitude of the capacitive discontinuity that appears at the transition from the coaxial transmission line to the resonant cavity. Second, various transmission line filters, well known to those skilled in the art, may be employed to change the impedance of the coaxial transmission line. For example, a slug tuner, or a parallel- or series-connected coaxial filter, such as a quarter-wave tuning stub, may be employed on the coaxial transmission line.

Another class of implementations support a voltage between the cathode and the control grid without the use of a resonant cavity. In this class of implementations, the electric field propagating in the coaxial transmission line directly generates a time-varying voltage across the grid-to-cathode gap. In one non-resonant topology, the cathode is connected to the center conductor of the coaxial transmission line while the grid is connected to the outer conductor in such a way that it is positioned in close proximity to the cathode. The center conductor may terminate in a right circular cylinder, or may be shaped to affect the impedance of the transmission line and the position of the cathode attached to it.

In another non-resonant topology, the cathode is connected to the outer conductor of the transmission line while the grid connects to the center conductor. To implement this, the coaxial transmission line transitions to a radial transmission line and the cathode takes the form of an annular ring connected to the bottom conductor of the radial transmission line. The control grid also takes on an annular form and is supported by the upper conductor of the radial transmission line, which also provides an electrical connection to the center conductor of the coaxial transmission line.

In both of these topologies, the impedance of the coaxial transmission line can be tuned by employing slug tuners or coaxial transmission line filters as described above. Furthermore, the transmission line can be terminated by the electron beam alone or in combination with a resistive termination disposed between the cathode and the control grid.

A more complete understanding of the directly coupled system providing increased operating bandwidth to IOTs and other density-modulated electron beam devices will be afforded to those skilled in the art, as well as a realization of additional advantages and objects thereof, by consideration of the following detailed description of the preferred embodiment. Reference will be made to the appended sheets of drawings which will first be described briefly.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

The invention provides improved instantaneous bandwidth of the input circuit of an IOT or other density-modulated device. In the detailed description that follows, like numbers are used to describe like elements illustrated in one or more of the figures.

FIG. 1is a schematic drawing of an exemplary IOT, typical of the prior art. The IOT includes three major sections, including an electron gun150, a tube body160, and a collector170. The electron gun150, shown in more detail inFIG. 3, provides an axially directed electron beam that is density modulated by an RF signal. Now returning toFIG. 1, the electron beam passes through a first drift tube230and a second drift tube232and then passes into an inner structure234inside the collector170that collects the spent electron beam. The electron gun further includes a cathode206with a closely spaced control grid204. The cathode is disposed at the end of a cathode support structure208that includes an internal heater coil220coupled to a heater voltage source224. The control grid204is positioned closely adjacent to the surface of the cathode206, and is coupled to a bias voltage source to maintain a DC bias voltage relative to the cathode. A resonant input cavity202receives an RF input signal via a coaxial transmission line210. The RF signal is coupled between the control grid204and cathode206to density modulate the electron beam emitted from the cathode206. The control grid is physically held in place by a grid support structure226. An example of an input cavity for an inductive output tube is provided by U.S. Pat. No. 6,133,786, the subject matter of which is incorporated in its entirety by reference herein.

FIG. 2depicts a parallel RLC circuit model of a conventional input circuit of the prior art. The electron beam is modeled as a shunt impedance112of the beam impedance Zb, and the resonant cavity is modeled as a parallel combination of a resistor106, an inductor108, and a capacitor110. The input transmission line102, with a characteristic impedance of Z0, is coupled to the resonant cavity via an inductive loop, modeled as a transformer104, with an effective turns ratio of N. As discussed previously, this results in a load impedance presented to the input transmission line due to the cavity of

Zload=R1+2⁢j⁢⁢Q⁢⁢Δω/ω0,
where Δω represents a small offset from the cavity resonant frequency ω0. Using this expression to calculate the half power points, the fractional bandwidth of the system is obtained as 1/Q, where Q is the quality factor.

FIG. 3represents an exemplary physical layout of the conventional prior-art input circuit modeled inFIG. 2. The coupling transformer is implemented as an inductive loop212that couples energy from the input coaxial transmission line210into the resonant cavity202. The cathode206is situated atop a cathode support structure208to place it in close proximity to a control grid204that permits passage of the electron beam emitted by the cathode206. The cavity geometry places practical limitations on loop size, and as a consequence, limits the fraction of the magnetic flux that is intercepted, restricting this technique to applications requiring relatively narrow bandwidths.

The invention described herein discloses a method for coupling to the input circuit of an IOT or other emission-gated device that allows for a substantially lower Qextthat is able to achieve substantially greater bandwidths. This is achieved by providing a coaxial transmission line that directly couples to the cavity surrounding the grid-to-cathode interaction region. This direct coupling results in a relatively low external quality factor (Qext) that reduces the total Q, increasing the bandwidth of the input circuit.

Several implementations of the directly coupled input circuit are possible. The most basic embodiment of the invention is shown inFIGS. 4(a) and4(b).FIG. 4(a) presents a three-dimensional view of the input circuit, andFIG. 4(b) presents an axial cross-sectional view of the input circuit. Like numbers are used to refer to corresponding structures between the two figures. In this embodiment, the center conductor316of the coaxial input transmission line transitions to the cathode support structure312, and the outer conductor318is connected to the outside wall of the cavity308. A control grid306is connected to the wall of the cavity308and held in close proximity to the cathode310, which is situated at the top of the cathode support structure312. A DC block is located between the outer conductor318and the grid306to enable a DC bias to be maintained between the grid306and the cathode310while permitting direct coupling of the RF signal from the transmission line to the grid. An optional iris314, in the form of an annular ring, may be disposed at the location where the outer conductor318of the transmission line joins the cavity wall308. The diameter of the opening of the iris314is larger than the diameter of the cathode support structure312, but smaller than the diameter of the outer conductor318. In the discussion that follows, the radius of the iris opening having diameter322is represented by ra. The radius of the resonant cavity having diameter320is represented by rc. The inner radius of the outer conductor having diameter324and the radius of the center (inner) conductor having diameter326of the transmission line are represented by roand ri, respectively. ThoughFIGS. 4(a) and4(b) depict a center conductor that is a right circular cylinder in shape, the center conductor may be stepped or tapered, such as the center conductor depicted inFIG. 7(c), in order to modify the impedance of the coaxial transmission line.

FIG. 4(c) depicts an alternative embodiment having a center conductor316supporting a cathode330, wherein the center conductor316has a taper332.FIG. 4(d) depicts an alternative embodiment having a center conductor316supporting a cathode334, wherein the center conductor316has a step336. In bothFIGS. 4(c)and4(d), a control grid306is connected to the cavity wall308. Dimension arrows324indicate the inner diameter of the outer conductor318, and dimension arrows326indicate the diameter of the center conductor316prior to the taper332or the step336shown inFIGS. 4(c) and4(d), respectively. An iris314may be disposed at the location where the outer conductor318joins the cavity wall308.

The geometry represented inFIGS. 4(a) and4(b) can be modeled by the equivalent circuit shown inFIG. 5. The beam impedance, Zb, is modeled as a shunt element412. The cavity is modeled as a parallel RLC circuit including a resistor406, an inductor408, and a capacitor410. The coupling of the coaxial transmission line402to the cavity is modeled as a transformer404as well as a shunt capacitance414, called the discontinuity capacitance, Cd, to account for the higher order modes excited at the impedance step that results from the change in diameter as a signal leaves the coaxial transmission line and enters the resonant cavity. The turns ratio of the transformer, N, is approximately
N2≈Zcp/Ztl.
The cavity port impedance, Zcp, and the transmission line impedance, Ztl, are given by
Zcp=[(μ/∈)1/2/2π]ln(rc/ri), and
Ztl=[(μ/∈)1/2/2π]ln(r0/ri),
where r0and rcare the radii of the outer conductor318of the coaxial transmission line and the resonant cavity308respectively, and riis the radius of the center (inner) conductor316. The calculation of the discontinuity capacitance, Cd, requires a full field solution. The Qextof the cavity is defined as Qext=ω0U/Pi, where U is the energy stored in the cavity and Piis the power dissipated in the transmission line load.

This power, defined as PI=½I2R, requires calculation of the current, I, flowing out of the cavity into the transmission line. The shunt capacitance in parallel with this load acts as a current divider. The fraction of the current that flows through the transmission line load is 1/(α2+1), where α=N2Ztlω0Cd. Since Q is inversely proportional to I2, the reduction in current modifies the Qextdefined above, resulting in:

For a typical design at L-band, the discontinuity capacitance is on the order of 0.1 picofarads, resulting in α≈0.1, and hence Qext≈ZcpI R/Q. Depending on the specific geometry, very low Qext, approaching unity, can be achieved.

If an iris314is included, where ra<r0, the discontinuity capacitance is increased, shunting a larger portion of the current and increasing the Qextwithout changing the cavity or transmission line geometry. A tapered or stepped transmission line or other impedance transformer may be used in place of, or in conjunction with, the iris to change the transmission line impedance presented to the cavity. Placement of a filter network in the transmission line offers further control of the bandwidth. An example of this, well known to those skilled in the art, is a coaxial impedance transformer, such as a slug tuner, on the center of the transmission line.FIG. 4(a) depicts a dielectric slug tuner328used to tune the impedance of the input line. Another example of such a filter network is a transmission line resonant cavity, connected either in series or in parallel, such as the tuning stub528depicted inFIG. 6(b).

FIGS. 6(a) and6(b) illustrate a second embodiment of the direct coupling system. A three dimensional view is depicted inFIG. 6(a), and a cross-sectional view is presented inFIG. 6(b). Like numbers are used to refer to corresponding structures. A ring cathode510is mounted on an annular support structure512, and this support structure is connected to the outer conductor518of the transmission line. The center conductor514of the transmission line extends through the cavity and is connected to a grid support structure520that supports an annular control grid506and further provides an electrical connection between the center conductor514and the control grid506. A DC block is located between the outer conductor518and the grid506to enable a DC bias to be maintained between the grid506and the cathode510while permitting direct coupling of the RF signal from the transmission line to the grid. An optional iris516may be used to alter the magnitude of the discontinuity capacitance between the coaxial transmission line and the cavity508. An optional stub tuner528, shown inFIG. 6(b), may likewise be used to tune the impedance of the coaxial transmission line to alter the magnitude of the discontinuity capacitance. Using a coaxial impedance transformer, a cold test model of this embodiment has been fabricated and tested, and has achieved an instantaneous bandwidth in excess of twenty percent.

FIG. 6(c) depicts an alternative embodiment having a center conductor530that has a taper532.FIG. 6(d) depicts an alternative embodiment having a center conductor534that has a step536. In bothFIGS. 6(c) and6(d), an annular control grid506is supported by grid support structure520. An iris516may be located where the outer conductor518joins the cavity508. An annular cathode support structure512supports the annular cathode510. A stub tuner528may be used to tune the impedance of the transmission line.

The voltage across the grid-to-cathode gap need not be provided by a resonant cavity. Instead, the electric field of the transmission line mode may be used to generate the voltage in a non-resonant directly coupled system. A portion of the power carried by the transmission line is coupled into the electron beam. Termination of the transmission line in its characteristic impedance results in maximum bandwidth. The termination can be provided by the beam as illustrated inFIGS. 7(a)-7(f), by a resistive load located after the beam as illustrated inFIGS. 8(a)-8(f), or by some combination of the two. A transmission line transformer, such as a slug tuner or resonant cavity filter, may be used to facilitate the match.

FIGS. 7(a)-7(f) show three possible embodiments of the non-resonant direct coupling system.FIG. 7(a) represents a three-dimensional view andFIG. 7(b) represents a cross-sectional view of a cylindrical non-resonant directly coupled system. The cathode608is disposed at the end of the center conductor610of the input coaxial transmission line. The outer conductor612of the transmission line is connected to the control grid606. The voltage across the grid-to-cathode gap, between the cathode608and grid606, is provided by the electric field of the electromagnetic wave traveling in the coaxial transmission line. The termination of the transmission line is provided by the electron beam itself.

FIGS. 7(c) and7(d) show an alternative embodiment in which the center conductor628is tapered. The cathode626surrounds the tapered end of the center conductor628and is held in close proximity to the control grid624that is situated around the tapered center conductor. The outer conductor630is connected to the control grid624. Varying the geometry of the tapered center conductor will change the impedance of the transmission line, which is terminated by the electron beam itself.

FIGS. 7(e) and7(f) depict an alternative embodiment of the non-resonant directly coupled system. In this embodiment, the coaxial transmission line comprising a center conductor658and an outer conductor660, transitions to a radial transmission line. The center conductor658attaches to the annular control grid654. The annular cathode656is attached to the lower wall of the radial transmission line and connected directly to the outer conductor660of the coaxial transmission line. In this embodiment, as well, the transmission line is terminated by the electron beam.

FIGS. 8(a)-8(f) present the same embodiments of the non-resonant direct coupling system shown inFIGS. 7(a)-7(f), except that here the termination is provided by a resistive load rather than solely by the electron beam. InFIGS. 8(a) and8(b), the resistive load714is situated between the cathode708and the control grid706, which are in turn attached to the center conductor710and the outer conductor712, respectively. Similarly inFIGS. 8(c) and8(d), the resistive load732is placed between the center conductor728, which supports the cathode726, and the control grid724that is connected to the outer conductor730. Finally, inFIGS. 8(e) and8(f), the resistive load762is situated around the outside of the radial transmission line between the cathode756, visible inFIG. 8(f), connected to the outer conductor760, and the grid754, connected to the center conductor758. It should be noted that the beam can be emitted from a cathode connected either to the center conductor, as shown inFIGS. 7(a)-7(d) and8(a)-8(d), or to the outer conductor, as shown inFIGS. 7(e)-7(f) and8(e)-8(f).

It should be appreciated that the above-described geometries are not meant to be comprehensive but are representative embodiments of the present invention that utilize direct coupling of a transmission line to achieve wideband coupling from the transmission line to the electron beam. By employing the direct coupling system, this invention enables inductive output devices to be adapted for service in wide-instantaneous-bandwidth applications. The method is also likely to spur the development of other novel emission-gated devices, employing thermionic and non-thermionic cathodes.

Having thus described a preferred embodiment of a novel input circuit that provides improved instantaneous bandwidth for an inductive output tube or other emission-gated device, it is apparent to those skilled in the art that certain advantages of such systems have been achieved. It should also be appreciated that various modifications, adaptations, and alternative embodiments thereof may be made within the scope and spirit of the present invention. The invention is further defined by the following claims.