Data transmitting and receiving apparatus

A data transmitting and receiving apparatus is disclosed. The receiving apparatus receives a spread spectrum signal prepared by modulating a carrier wave with data to produce a data modulated signal and multiplying the data modulated signal with a spreading signal which has a bandwidth greater than that of the data modulated signal. The receiving apparatus may include three bandpass filters for passing three different band portions of a full band of the spread spectrum signal and producing an intermediate signal, and a detector for detecting data from the intermediate signal and for producing a detector output. The apparatus may also include a reception state evaluation device for detecting the current reception state of the detector and for producing a poor reception signal, when the detector is in a poor reception state. The poor reception signal is used for changing from a current band to another band.

BACKGROUND OF THE INVENTION 
1. Field of the Invention 
The present invention relates to a transmitting and receiving apparatus for 
data communications using a spread spectrum signal. 
2. Description of the Prior Art 
Spread spectrum communications has been identified as a suitable method for 
local area wireless data communications systems (e.g., wireless LANs) and 
power-line carrier (PLC) data communications because of its good 
transmission characteristics in multipath environments and excellent 
ability to eliminate interference signals. The major frequency band 
allocated to wireless LAN systems is the ISM band used by industry, 
science, and medicine. The ISM band is the frequency band used by devices 
using electromagnetic power waves, such as microwave ovens, and 
transmitters and receivers used in wireless LAN systems must be able to 
maintain normal data communications even under extremely high interference 
conditions. 
A variety of methods not requiring spreading signal synchronization has 
been proposed as a means of simplifying the data receiver in spread 
spectrum communications. One of these methods is the spread spectrum 
differential detection method whereby the period of the spreading signal 
is synchronized to the data symbol period. An example of this method is 
described in Japanese patent laid-open number 1987-257224. The 
configuration and operation of a spread spectrum communications apparatus 
applying this spread spectrum differential detection method is described 
below with reference to accompanying figures. 
FIG. 26 is a block diagram of a spread spectrum differential detection 
method transmitter and receiver. FIGS. 27a and 27b show waveform diagrams 
of the signals processed by various transmitter and receiver components. 
As shown in FIG. 26, the transmitter 10 comprises a differential encoder 
11, PSK (phase shift keying) modulator 12, spreading signal generator 13, 
multiplier 14, and symbol clock generator 15. The clock generator 15 
supplies the symbol clock CK of period T to the differential encoder 11, 
PSK modulator 12, and spreading signal generator 13. 
The receiver 20' comprises a differential detector 22 and a decoder 23. The 
differential detector 22 further comprises a delay 221, multiplier 222, 
and low-pass filter (LPF) 223. 
The bit stream data (binary data of value .+-.1) is read synchronized to 
the symbol clock CK, and is differential coded by the differential encoder 
11. The PSK modulator 12 modulates the carrier wave with the differential 
coded data to obtain data modulated signal p, which is a binary PSK signal 
of symbol cycle period T. As a result, the data modulated signal p is the 
same phase as the previous symbol when data d is 1, and is opposite phase 
to the previous symbol when data d is -1. The spreading signal generator 
13 generates the spreading signal q synchronized to and with the same 
period as the symbol clock CK. The spreading signal q is a constant 
amplitude, pseudorandom pulse wave generated from pseudorandom series. The 
multiplier 14 multiplies the data modulated signal p and spreading signal 
q to output the spread spectrum signal a. 
FIG. 27a shows the time-based waveforms of the data modulated signal p, 
spreading signal q, and spread spectrum signal a. The baseband waves of 
the data modulated signal p and spread spectrum signal a are shown for 
convenience. 
The spread spectrum signal a thus obtained is input through the 
transmission path to the receiver 20'. The differential detector 22 
multiplies the received spread spectrum signal a by delayed signal a.sub.d 
(which is the spread spectrum signal a delayed symbol cycle period T by 
the delay 221) using the multiplier 222, and removes the high frequency 
component of the product using the LPF 223 to obtain detector output c. 
Because multiplying spreading signal components will always result in a 
constant value, only the data modulation component will appear in the 
detector output c. As with the differential detection output to the normal 
differential PSK signal, the detector output c is therefore a positive 
value when there is no phase change from the previous symbol, and is a 
negative value when opposite phase to the previous symbol. The decoder 23 
outputs the decoded data d' as a value of +1 when the detection output c 
is positive, and -1 when negative. 
FIG. 27b shows the time-based waveforms of the received spread spectrum 
signal a, delayed signal a.sub.d, and detector output c. As in FIG. 27a, 
the baseband waves of the spread spectrum signal a and delayed signal 
a.sub.d are shown. It is to be noted that the normally received spread 
spectrum signal a has jamming, interference, or distortion components 
added in the transmission path. The effects of such jamming, etc., are 
removed from the waveforms shown in FIG. 27b. 
By means of this configuration, a transmitter and receiver of relatively 
simple construction not requiring complex means for spreading signal 
synchronization and other functions can be obtained while retaining the 
jamming elimination capability and multipath transmission performance 
characteristic of spread spectrum communications. 
When extremely strong interference components are added to the spread 
spectrum signal band, however, this transmitter and receiver is incapable 
of signal reception when the band of the interference component overlaps 
only part of the signal band. In addition, a wide band delay having a 
constant delay characteristic across the complete spread spectrum signal 
band must be used, and such delays are difficult to achieve. 
SUMMARY OF THE INVENTION 
Therefore, an object of the present invention is to provide a data 
transmitting and receiving apparatus that does not require a wide band 
delay and yet can reliably transmit and receive data signals even when 
there is an extremely strong interference component superimposed to the 
spread spectrum signal band. 
To achieve this object, a data receiving apparatus for receiving a spread 
spectrum signal prepared by modulating data with a carrier wave to produce 
a data modulated signal and multiplying the data modulated signal with a 
spreading signal which has a bandwidth greater than that of the data 
modulated signal, comprises bandpass means for passing a portion of a full 
band of said spread spectrum signal and for producing a bandpass signal; 
and detection means for detecting an intermediate signal from said 
bandpass signal and for producing a detection signal. 
Note that the spread spectrum signal is preferably a chirp signal. 
The data modulated signal is also preferably a differential PSK signal, and 
the detection means is preferably a differential detector. When 
differential detection is used, the bandwidth of the signal processed by 
the differential detector is smaller than the bandwidth of the spread 
spectrum signal, and a narrow band delay can therefore be used in the 
differential detector. 
It is further preferred to selectively use a band not containing 
interference components by changing the bandpass frequency of the bandpass 
means, or by selecting one of plural bandpass frequencies. In addition, 
the quality of the detection output is preferably improved by synthesizing 
plural detection outputs. 
The bandpass means can be achieved by means of a frequency mixer, local 
oscillator, and bandpass filter. In this case, the bandpass frequency can 
be changed by changing the frequency of the local oscillation signal 
output by the local oscillator. When differential detection is used, the 
frequency change of the local oscillation signal is set to an integer 
multiple of 1/symbol period. 
An alternative embodiment of a data transmitting and receiving apparatus 
according to the present invention comprises a transmitter and a receiver. 
The transmitter transmits a burst-type spread spectrum signal which is 
formed by dividing the transmission data into blocks containing a 
predetermined number of bits to generate data packets containing a unique 
word and an error detection bit. The data modulated signal, which is 
obtained by data modulation of the carrier wave with this data packet, is 
multiplied with a spreading signal, which has a bandwidth greater than 
that of the data modulated signal, to obtain and produce the burst-type 
spread spectrum signal. 
The receiver for demodulating the spread spectrum signal to output the 
decoded data comprises plural channels each comprising a bandpass means, 
detector, clock generator, decoder, unique word detector, packet 
extractor, and error detector for extracting and demodulating only the 
mutually different partial bandwidth signal components in the spread 
spectrum signal band, and a selector for selecting an output from a 
channel not containing any bit errors based on the error detection means 
output to generate the decoded data. 
A data transmitting and receiving apparatus thus comprised detects 
intermediate signals by extracting one or plural partial band signal 
components in the spread spectrum signal bandwidth, and selectively 
obtains the outputs of one or more decoders not containing bit errors as 
the decoded data. As a result, the effects of signal deterioration caused 
by high strength interference waves present in the signal band and 
frequency-selective distortion resulting from multipath transmission can 
be avoided, and signal components extracted from the frequency band with 
good reception can be selectively used. As a result, deterioration of the 
error rate due to strong interference or frequency-selective distortion 
can be reduced.

DESCRIPTION OF PREFERRED EMBODIMENTS 
FIG. 1 is a block diagram of a data transmitting and receiving apparatus 
according to the first embodiment of the invention. FIGS. 3a and 3b show 
waveform diagrams of the signals observed at major points in the apparatus 
shown in FIG. 1. Note that the baseband wave is shown for convenience. 
FIGS. 4a and 4b are simplified spectrum diagrams of the signals at 
selected points. 
As shown in FIG. 1, the transmitter 10 comprises differential encoder 11, 
PSK modulator 12, multiplier 14, clock generator 15, and spreading signal 
generator 13, and outputs the spread spectrum signal a(t). The receiver 20 
comprises bandpass device 21, differential detector 22, decoder 23, and 
reception state evaluation device 24. 
The operation of the transmitter 10 shown in FIG. 1 is similar to that 
described with reference to FIG. 26 in the discussion of the prior art 
above. Specifically, the m.sup.th data item d.sub.m (a binary value of 
plus or minus 1) is obtained synchronized to the symbol clock CK of period 
T and differential coded by the differential encoder 11, and is modulated 
by the PSK modulator 12 to obtain the data modulated signal p(t), a binary 
PSK (phase shift keying) modulated signal of symbol cycle period T. The 
spreading signal generator 13 generates a spreading signal q(t) 
synchronized to and with the same period as the symbol clock CK. This 
spreading signal q(t) may be, for example, a constant amplitude 
pseudorandom pulse wave generated from pseudorandom series. The multiplier 
14 multiplies the data modulated signal p(t) and spreading signal q(t) to 
output the spread spectrum signal a(t). 
This differential coded data can be expressed as 
EQU d.sup.m =.delta..sub.m .times..delta..sub.m-1 1! 
assuming .delta..sub.m is a binary value of +1 and -1. Therefore, if the 
frequency of the carrier wave is f.sub.c and the value Re . . . ! is a 
real number, the spread spectrum signal a(t) transmitted during the symbol 
period corresponding to .delta..sub.m can be expressed as 
EQU a(t)=Re .delta..sub.m .times.q(t).times.exp{2.pi.f.sub.c t}! 2! 
The signal waves at the selected transmitter components shown in FIG. 3a 
are simulated. 
The spread spectrum signal a(t) input through the transmission path to the 
receiver 20 is first bandwidth limited by the bandpass device 21 to obtain 
intermediate signal b(t). 
As shown in the bandpass device 21a in FIG. 2, this bandpass device 21 
comprises first, second, and third bandpass filters, identified as BPF1 
211, BPF2 212, and BPF3 213 below, and selects one of these bandpass 
filters for operation. Bandpass filters BPF1 211, BPF2 212, and BPF3 213 
selectively pass three different pass bands B1, B2, and B3. The bandpass 
device 21a also has a counter 209 for counting from one to three 
repeatedly, and incremented each time a band selection signal e, which 
will be described later, from reception state evaluation device 24 is 
produced. When the counter 209 is counted to 1, the switch is connected in 
a manner shown in FIG. 2. Thus, the first bandpass filter BPF1 is selected 
to process the input signal a(t) in the first bandpass filter BPF1 and 
send out the filtered signal b(t). When the counter 209 is counted to 2, 
the switch is so connected to select the second bandpass filter BPF2, and 
when the counter 209 is counted to 3, the switch is so connected to select 
the third bandpass filter BPF3. 
Thus, when the bandpass device 21 receives the band selection signal e from 
the reception state evaluation device 24, a filter different from the one 
currently used is sequentially selected by looping through the bandpass 
filters in a predetermined sequence, e.g., BPF1 .fwdarw. BPF2 .fwdarw. 
BPF3 .fwdarw. BPF1. The spectrum of the received spread spectrum signal a, 
and the pass bands B1-B3 of the bandpass filters are illustrated in FIG. 
4a. The spectrum of the bandwidth limited output (intermediate signal b) 
is shown in FIG. 4b. When BPF1, BPF2, or BPF3 is selected, the band of the 
intermediate signal b(t) is b1, b2, or b3, respectively, as shown in FIG. 
4b. 
The differential detector 22 delays and detects the intermediate signal 
b(t) to obtain detector output c(t). 
The baseband waves of the signals output by selected receiver components 
are simulated in FIG. 3b. In each symbol period, the symbol of the 
baseband wave of the spread spectrum signal a(t) is the same as the data 
modulated signal when the phases match, but is the reverse symbol when the 
phases are opposite. 
The intermediate signal b(t) can be expressed as 
EQU b(t)=Re .delta..sub.m .times.q'(t).times.exp{2.pi.f.sub.c t}! 3! 
where .delta..sub.m =.+-.1. When the intermediate signal band is greater 
than the symbol repeat frequency, q'(t) in equation 3! is approximately 
equal to the signal obtained by bandwidth limiting spreading signal q(t). 
Because q(t) is thus replaced by q'(t) through bandwidth limiting, 
intermediate signal b(t) is a waveform significantly different from that 
of spread spectrum signal a(t). The phase relationship, however, remains 
the same, i.e., the waveforms are approximately equal when the phase 
matches the data modulated signal in each symbol period, but the symbols 
are opposite when the phase does not match the data modulated signal 
phase. This is because the symbol of the intermediate signal b(t) is 
reversed by the symbol of .delta..sub.m as indicated by equation 3!. More 
specifically, the waves are not precisely identical and intersymbol 
interference occurs in the region near the adjacent symbol because the 
signal is affected by the adjacent symbol. However, when the intermediate 
signal band is greater than the symbol repeat frequency, intersymbol 
interference is small and there is no real problem. 
The differential detector 22 first delays the intermediate signal b(t) one 
symbol period T by means of the delay 221, obtaining delayed intermediate 
signal b.sub.d (t). Noting that the spreading signal q(t) is a repeated 
wave of period T and q'(t) will also be approximately a repeated wave of 
period T, 
##EQU1## 
By precisely controlling T or adjusting the phase of the delay 221 output 
signal so the statement 
EQU exp{-2.pi.f.sub.c T}=1 5! 
is true, the delayed intermediate signal b.sub.d (t) becomes 
EQU b.sub.d (t)=Re .delta..sub.m-1 .times.q'(t).times.exp{2.pi.f.sub.c t}!. 6! 
The detector output c(t) is the low frequency component of the multiplier 
222 output extracted by the LPF 223. Thus, by multiplying equations 3! 
and 6!, eliminating the item of the high frequency component 
exp{4.pi.f.sub.c t}, and using equation 1!, 
EQU c(t)=.delta..sub.m .times..delta..sub.m-1 {Re q'(t)!}.sup.2 =d.sub.m 
{Re q'(t)!}.sup.2 7! 
is obtained. By evaluating the polarity of the detector output c(t) from 
equation 7!, the data can be decoded. 
FIG. 3b simulates the wave detection process. Specifically, multiplied same 
(or similar) wave pulses result in a positive pulse when there is no phase 
change from the preceding wave symbol, but a negative pulse when the phase 
is reversed from that of the previous wave symbol because opposite-symbol 
pulses are multiplied. As a result, detector output c(t) will become a 
positive or negative pulse depending upon whether the waves are of same or 
opposite phase. The decoder 23 evaluates the detector output c(t) pulse 
and outputs decoded data d'.sub.m as a value of 1 or -1 when the detector 
output c(t) is positive or negative, respectively. 
Because the bandwidth of the intermediate signal b(t) handled by the 
differential detector 22 is narrower than that of the original spread 
spectrum signal a(t), the delay 221 is sufficiently precise if it operates 
within the bandwidth of the intermediate signal b(t), and it is 
specifically not necessary for the delay 221 to maintain high precision 
throughout the entire bandwidth of the spread spectrum signal a(t). 
The reception state evaluation device 24 monitors the detector output c(t) 
level to estimate whether reception is currently good. If reception is 
determined to not be good, the band selection signal e is output to the 
bandpass device 21. The reception state evaluation device 24 has a 
comparator for comparing the detector output c(t) level with a 
predetermined threshold level and produces a band selection signal e when 
the detector output c(t) level is lower than the predetermined threshold 
level. For example, when the jamming j is present in the bands B2 and B3, 
as shown in FIG. 4a, and if the currently used bandpass filter is BPF2 
212, the reception state evaluation device 24 produces the band selection 
signal e to switch the bandpass device 21a from BPF2 212 to BPF3 213. 
Since the jamming j is also covering the band B3, reception state 
evaluation device 24 again produces the band selection signal e to switch 
the bandpass device 21a from BPF3 213 to BPF1 211. Since there is no 
jamming present in the band B1, the currently selected state using the 
bandpass filter BPF3 is maintained. 
According to the above embodiment, the bandpass filters BPF1 211, BPF2 212, 
and BPF3 213 are explained to have three different pass bands B1, B2, and 
B3 shown in FIG. 4a. Alternatively, it is possible to arrange such that 
the bandpass filter BPF1 211 has a wide pass band, such as B1+B2+B3 in 
FIG. 4a, the bandpass filter BPF2 212 has an intermediate width pass band, 
such as B1+B2 in FIG. 4a, and the bandpass filter BPF3 213 has a narrow 
pass band, such as B1 in FIG. 4a. 
Referring to FIG. 5 a bandpass device 21 is shown which is a modification 
of the same shown in FIG. 1. The bandpass device 21b comprises a bandpass 
filter BPF 214, frequency mixer 215, and local oscillator 216. The input 
signal is converted by the frequency mixer 215 to the frequency band 
obtained as the difference between the input signal and the local 
oscillation signal output from the local oscillator 216, and is then 
bandwidth limited by the BPF 214 to extract a partial frequency component 
of the frequency-converted spread spectrum signal a(t). The partial 
frequency component is output as intermediate signal b(t). The local 
oscillator 216 is normally a phase locked loop (PLL) synthesizer, and can 
change the frequency of the local oscillation signal at an interval that 
is an integer multiple of symbol rate 1/T. This configuration equivalently 
extracts the intermediate signal b(t) as a component of a different 
frequency part of the spread spectrum signal a(t). Note that if the local 
oscillation signal frequency is set to the middle frequency of the pass 
band, a low bandpass filter can be used in place of BPF 214. 
FIG. 6a shows the pass bands (pass band 1 B1, pass band 2 B2, and pass band 
3 B3) when the frequency of the local oscillation signal is varied three 
ways. FIG. 6b shows the spectrum b of the frequency-converted, 
bandwidth-limited intermediate signal. As shown in FIG. 6a, each of the 
pass bands B1-B3 is set to a different frequency. When the band selection 
signal e is input from the reception state evaluation device 24, counter 
209 counts up to switch the local oscillator 216 and to change the 
frequency of the local oscillation signal, and sequentially changes the 
pass band to a band different from that currently in use by looping 
through the pass bands in a predetermined sequence, e.g., B1 .fwdarw. B2 
.fwdarw. B3 .fwdarw. B1. Thus, the bandpass device 21b operates 
equivalently to the bandpass device 21a shown in FIG. 2. Compared with the 
configuration shown in FIG. 2, however, this bandpass device 21b can be 
achieved using a single bandpass filter. In addition, because the band of 
the intermediate signal b(t) is fixed, a relatively narrow band 
differential detector can be used downstream, and the overall 
configuration is thus simplified. 
It is to be noted, however, that the delay/detection operations of the 
bandpass device shown in FIG. 5 will only function normally when the 
change in the local oscillation signal frequency is an integer multiple of 
1/T as will be described below. If the frequency of the local oscillation 
signal is f.sub.L, the intermediate signal b(t) can be expressed as 
EQU b(t)=Re .delta..sub.m .times.q'(t).times.exp{2.pi.(f.sub.c -f.sub.L)t}! 3'! 
and the delayed intermediate signal b.sub.d (t) as 
##EQU2## 
By precisely controlling T or adjusting the phase of the delay 221 output 
signal so the statement 
EQU exp{-2.pi.(f.sub.c -f.sub.L)T}=1 5'! 
is true, the delayed intermediate signal b.sub.d (t) becomes 
EQU b.sub.d (t)=Re .delta..sub.m-1 .times.q'(t).times.exp{2.pi.(f.sub.c 
-f.sub.L)t}!. 6'! 
By multiplying equations 3'! and 6'!, eliminating the item of the high 
frequency component exp{4.pi.(f.sub.c -f.sub.L)t}, and using equation 1!, 
the detector output c(t) is obtained as 
EQU c(t)=.delta..sub.m .times..delta..sub.m-1 {Re q'(t)!}.sup.2 =d.sub.m 
{Re q'(t)!}.sup.2 7'! 
By evaluating the polarity of the detector output c(t) from equation 7'!, 
the data can be decoded. By modifying equation 5'! obtained from the 
above process and defining k as an integer, 
EQU f.sub.L =f.sub.c -k.times.(1/T) 8! 
is obtained, and we know that the frequency of the local oscillation signal 
must have a frequency interval of an integer multiple of symbol rate 1/T. 
This embodiment is described in further detail below with reference to the 
receiver 20 operation when a jamming j as shown in FIG. 4a is applied to 
the transmission path. Normal reception by the conventional apparatus 
shown in FIG. 26 is not possible because most of the jamming j energy is 
detected by the differential detector. With the apparatus according to the 
present embodiment as shown in FIG. 1, however, the bandwidth of the 
received signal is limited by providing a bandpass device 21 before the 
differential detector 22. As a result, when the bandpass device 21 selects 
pass band 1 B1, the intermediate signal b is not affected by the jamming 
j, and normal reception is possible. If the bandpass device 21 selects 
either pass band 2 B2 or pass band 3 B3, however, the intermediate signal 
b is greatly affected by the jamming, and normal reception is not 
possible. If this state persists, the reception state evaluation device 24 
determines that current reception is poor (not good) and outputs the band 
selection signal e to change the selected pass band of the bandpass device 
21. Band selection continues until pass band 1 B1 is selected and a good 
reception state is restored. The reception state evaluation device 24 
stops band selection signal output when normal reception is restored (pass 
band 1 B1 is selected in this case), and stable data transmission 
unaffected by jammings can be maintained. 
Note that while the above embodiment has been described using binary PSK 
(phase shift keying) modulation as the modulation method of the data 
modulated signal p(t), 4-ary PSK, 8-ary PSK, or other multilevel PSK 
modulation method can be used. The above embodiment will be modified in 
this case as follows: the differential detector 22 must be a two channel 
design with an orthogonal axis added, and the decoder 23 outputs bit 
series decoded data d'.sub.m by parallel-serial conversion after the 
detector output c is evaluated and the evaluated symbol data is input to 
the decoder 23 (cf., William R. Bennet and James R. Davey, "DATA 
TRANSMISSION" 1965 by McGraw-Hill Book Co., New York). 
It should also be noted that the invention shall not be limited to PSK 
modulation, and other modulation methods, including amplitude shift keying 
(ASK) modulation can be used. 
In addition, while differential detection is applied by the differential 
detector 22, other detection methods can be used. For example, envelope 
detection using the data modulated signal p as an ASK signal can be used. 
When a detection method other than differential detection is used, e.g., 
envelope detection, the spreading signal q period can be set irrespective 
of the symbol period T of the data modulated signal p. In addition, the 
frequency of the local oscillation signal in the bandpass device 21b shown 
in FIG. 5 can be changed to any other frequency without being limited to 
an interval that is an integer multiple of 1/T. 
The spreading signal q(t) is also not limited to the constant amplitude, 
pseudorandom pulse wave generated by the pseudorandom series as described 
above, and another noise-like signal or chirp signal as described with 
reference to the fourth embodiment below can be used. When a chirp signal 
is used for the spreading signal q(t), the energy of a specific frequency 
component tends to concentrate in a specific position in each symbol 
period. As a result, the intermediate signal b(t) bandwidth limited by the 
bandpass device 21 becomes a pulse with a high peak at a specific position 
in each symbol period as shown in FIG. 12, and intersymbol interference is 
not as easily generated. 
The spreading signal q(t) period is also described as equal to the symbol 
period T of the data modulated signal p(t) above, but a period that is 1/n 
(where n is a natural number) of the symbol period T can be used. 
Alternatively, the spreading signal q(t) period can be n times (where n is 
a natural number) the symbol period T if the delay 221 has a delay time 
that is also n times the symbol period T. 
The bandpass device 21 is furthermore not limited to the configurations 
shown in FIGS. 2 and 5, and the pass band characteristics thereof can be 
changed. For example, the pass band characteristics can be changed by 
changing the parameters of a single filter. Alternatively, plural local 
oscillators each generating a local oscillation signal at a different 
frequency can be provided and selected rather than changing the 
oscillation frequency of the one local oscillator shown in FIG. 5. 
While this bandpass device 21 selects from three pass bands, a number other 
than three pass bands can also be used. 
The reception state evaluation device 24 is described as monitoring the 
detector output c level, but may monitor any other parameter(s) enabling 
evaluation of the reception state. For example, the aperture of the eye 
pattern of the detector output or the error rate of the decoded data 
d'.sub.m can be monitored. 
The transmitter 10 is also not limited to that shown in FIG. 1, and a 
transmitter 10a comprising a shift register 51, waveform memory 52, D/A 
converter 53, carrier wave oscillator 54, and modulator 55 as shown in 
FIG. 7 can be used. The operation of the transmitter shown in FIG. 7 is 
described below. 
The bit stream data d is input to a k stage (where k is a natural number) 
shift register 51, and supplied as k-bit serial data to the address input 
of the waveform memory 52. The waveform memory 52 is a read-only memory 
(ROM) device storing as waveform data the precalculated baseband waveforms 
of the spread spectrum signals determined by the patterns of all k-bits of 
data d. This waveform data is stored to the address expressed as a k-bit 
pattern, and is output from the address specified by the shift register 51 
output. The D/A converter 53 converts the waveform data to an analog wave, 
and outputs the result as the spread spectrum signal baseband wave. The 
carrier wave oscillator 54 oscillates and outputs the carrier wave. The 
modulator 55 product modulates the carrier wave using the spread spectrum 
signal baseband wave to obtain the spread spectrum signal a(t). This 
transmitter 10a generates and transmits the same spread spectrum signal 
a(t) as that obtained by the transmitter 10 shown in FIG. 1. 
FIG. 8 is a block diagram of a receiver in a data transmitting and 
receiving apparatus according to a second embodiment of the invention. 
Further description of the transmitter 10 in this embodiment is omitted 
below because it is the same as that of the first embodiment (FIG. 1). The 
receiver 20a in this embodiment is essentially identical to the receiver 
20 of the first embodiment except that this receiver 20a does not have a 
reception state evaluation device, but has a manually operable switch SW 
from which the band selection signal e is externally applied to the 
bandpass device 21. 
Specifically, decoding data is processed by an external device connected to 
the receiver and using the decoding data thereof. The result returned by 
the external device is used to automatically supply an effective band 
selection signal, or band selection is manually controlled for more 
precise band selection. 
FIG. 9 is a block diagram of a receiver in a data transmitting and 
receiving apparatus according to a third embodiment of the invention. 
Further description of the transmitter 10 in this embodiment is omitted 
below because it is the same as that of the first embodiment (FIG. 1). In 
the receiver 20b shown in FIG. 9, the first, second, and third bandpass 
device 210a, 210b, and 210c are the same as bandpass filters 211, 212, 213 
in the first embodiment, respectively. The spread spectrum signal a is 
bandwidth limited to pass band 1 B1, pass band 2 B2, and pass band 3 B3, 
respectively, resulting in first, second, and third intermediate signals 
b1, b2, and b3. The spectra of these intermediate signals are as shown in 
FIG. 4b with reference to the first embodiment above. The first, second, 
and third differential detectors 22a, 22b, and 22c delay detect the first, 
second, and third intermediate signals b1, b2, and b3, respectively, to 
output corresponding first, second, and third detector outputs c1, c2, and 
c3. The configuration and operation of each of the differential detectors 
22a-22c are the same as that of the differential detector 22 shown in FIG. 
1 with reference to the first embodiment. 
These detector outputs c1, c2, and c3 are input to the optimum band 
evaluation device 34, which monitors the level of each input to determine 
which of the detector outputs c1-c3 is in the best reception state. The 
optimum band evaluation device 34 is formed by a maximum detector 34 for 
detecting the maximum level of the three outputs c1-c3. The result is 
output to the detector output selector 35. Based on this reception state 
signal, the detector output selector 35 selects one of the detector 
outputs c1-c3 which has the maximum output level, and outputs the selected 
signal as the detector output c to the decoder 23. The decoder 23 
evaluates the data as described in the first embodiment above, and outputs 
the decoded data d'.sub.m. 
In this embodiment, the three intermediate signals b1-b3 obtained by the 
three bandpass device 210a-210c are simultaneously detected to obtain 
three detector outputs c1-c3, of which the best signal is selected. The 
time required for sequential bandwidth selection in the first embodiment 
is therefore not needed in this embodiment. In addition, when the 
reception state varies over time it is possible to select another 
bandwidth before reception becomes impossible due to a deteriorated 
reception state, and interruption of data receiving can be prevented. 
The three bandpass device 210a-210c of this embodiment may be configured to 
change the pass band by means of a construction similar to the bandpass 
device 21a shown in FIG. 2 or the bandpass device 21b shown in FIG. 5 and 
described in the first embodiment above. Furthermore, if the bandpass 
device 210a-210c comprise a frequency mixer, local oscillator, and 
bandpass filter identically to the bandpass device 21b shown in FIG. 5, 
the frequency interval of each local oscillator is an integer multiple of 
1/T, and the center frequency of the intermediate signals b1-b3 is 
identically selected, identical structures can be used for the same 
differential detectors 22a-22c. 
FIG. 10 is a block diagram of a receiver in a data transmitting and 
receiving apparatus according to a fourth embodiment of the invention. The 
transmitter 10 in this embodiment is the same as that of the first 
embodiment (FIG. 1). FIG. 11 is a waveform diagram of selected signals in 
the transmitter, and FIG. 12 shows a spectrum diagram of the intermediate 
signals and waveform diagram of selected signals in the receiver of the 
fourth embodiment. 
The modulation process of this embodiment is essentially identical to that 
of the first embodiment described with reference to FIG. 3a. As shown in 
FIG. 11, however, the spreading signal q output by the spreading signal 
generator 13 is a chirp signal obtained by frequency modulation of a sine 
wave signal. The period of this signal is equal to the symbol period T of 
the data modulated signal p. 
The first, second, and third bandpass device 210a, 210b, and 210c of the 
receiver 20 shown in FIG. 10 are identical to the bandpass device of the 
third embodiment shown in FIG. 9; the spread spectrum signal a is 
bandwidth limited to pass bands B1-B3, and the first, second, and third 
intermediate signals b1, b2, and b3 shown in FIG. 12 are respectively 
output. The first, second, and third differential detectors 22a, 22b, and 
22c are identical to the differential detectors of the third embodiment 
shown in FIG. 9; the first, second, and third intermediate signals b1, b2, 
and b3 are delay detected, resulting in corresponding first, second, and 
third detector outputs c1, c2, and c3, respectively. 
As shown in FIG. 12, the first part of each symbol period is a low 
frequency component with the frequency gradually increasing to a high 
frequency component at the end of each symbol period because the spread 
spectrum signal a is a chirp signal. Because intermediate signal b1 is 
obtained by extracting the low frequency component of the original spread 
spectrum signal a, intermediate signal b1 starts as a high amplitude 
signal early in each symbol period and diminishes to a low amplitude 
signal at the end of each symbol period. Conversely, intermediate signal 
b3 is obtained by extracting the high frequency component of the original 
spread spectrum signal a, and therefore starts each symbol period with a 
low amplitude and ends each period as a high amplitude signal. 
Intermediate signal b2 has a high amplitude in the middle of each symbol 
period and a low amplitude at the beginning and end of each period. As 
with the intermediate signal b of the first embodiment above, however, the 
shape of each signal is essentially the same in each symbol period and the 
symbol inverts according to the data modulated signal, thus enabling 
demodulation by differential detection. As shown in FIG. 12, the detector 
outputs c1-c3 are pulse streams with a peak at a specific position in each 
symbol period, and the pulse peaks are located at the beginning, middle, 
and end of each symbol period. The peak positions are determined by the 
spreading signal q frequency sweep parameters and the characteristics of 
each bandpass device. 
The detector output synthesizer 41 comprises first, second, and third 
detector output delay device 411, 412, and 413. The first detector output 
c1 is input to the first detector output delay device 411 and delayed time 
t1, the second detector output c2 is input to the second detector output 
delay device 412 and delayed time t2, and the third detector output c3 is 
input to the third detector output delay device 413 and delayed time t3. 
Note that times t1-t3 are equal to the difference between the detector 
output c1-c3 pulse peak positions and the evaluation timing as shown in 
FIG. 12. After thus adjusting the peak position of each detector output to 
the evaluation timing, the signals are added by the adder 414 to obtain 
synthesized detector output c. The decoder 23 decodes the data based on 
the symbol of the detector output at this evaluation timing, and outputs 
decoded data d'.sub.m. 
All signal components contained in each intermediate signal can be used in 
this embodiment because all plural intermediate signals are detected and 
synthesized. A higher signal/noise (S/N) ratio can therefore be obtained 
in the detector output, and reliable transmission is possible even with 
high noise levels. The energy of each intermediate signal is also 
concentrated at a specific position in each symbol period, and intersymbol 
interference can be effectively reduced by using a chirp signal for the 
spreading signal. The detector outputs can also be efficiently synthesized 
because each of the detector outputs c1-c3 has a high peak at a specific 
position in each symbol period and the peak positions of each detector 
output input to the detector output synthesizer 41 are aligned. 
Note that if the pulse peak position of the third detection signal c3 is 
adjusted to the evaluation timing, time t3 can be defined as zero (0) and 
the third detector output delay device 413 can be eliminated in this 
fourth embodiment. While the adder 414 of the detector output synthesizer 
41 simply adds the inputs, the inputs can also be weighted according to 
the reception state of each detector output, thereby further improving the 
S/N ratio of the synthesized detector output c'. 
As in the third embodiment, the three bandpass device 210a-210c of this 
embodiment may be configured similarly to the bandpass device 21a shown in 
FIG. 2 or the bandpass device 21b shown in FIG. 5. Furthermore, if the 
bandpass device 210a-210c are identical to the bandpass device 21b shown 
in FIG. 5, the frequency interval of each local oscillator is an integer 
multiple of 1/T, and the center frequency of the intermediate signals 
b1-b3 is identically selected, identical structures can be used for the 
differential detectors 22a-22c. 
FIGS. 13a and 13b taken together show a block diagram of a data 
transmitting and receiving apparatus according to a fifth embodiment of 
the invention. As shown in FIG. 13, the transmitter 10b comprises a packet 
assembler 16, a data modulator 17, multiplier 14, clock generator 15, and 
spreading signal generator 13 which outputs the spread spectrum signal a. 
The receiver 20d comprises bandpass devices 21A, 21B and 21C, differential 
detectors 22A, 22B and 22C, decoders 23A, 23B and 23C, clock regenerators 
25A, 25B and 25C, unique word detectors 26A, 26B and 26C, packet 
extractors 27A, 27B and 27C, error detectors 29A, 29B and 29C, and 
evaluator/selector 240. 
Note that the data modulator 17 in the transmitter 10b comprises both the 
differential encoder 11 and PSK modulator 12 provided separately in the 
transmitter 10 of the first embodiment shown in FIG. 1. Also, the receiver 
20d in FIG. 13 has three detection units for three different channels A, B 
and C, the first, second and third detection units are defined by elements 
with suffixes "A", "B" and "C", respectively. The bandpass devices 21A, 
21B and 21C are arranged to selectively pass three different pass bands 
B1, B2, and B3, respectively, which are also referred to as channels A, B 
and C. 
The operation of this embodiment is described below with further reference 
to FIGS. 14, 15a, 15b, 16 and 17a, in which FIG. 14 shows the code format 
of the data packets output from the packet assembler 16, FIGS. 15a and 15b 
show the waveform diagrams of various signals (baseband waves shown for 
simplicity) from the circuit of FIG. 13, FIG. 16 shows the data packets 
observed in the evaluated data stream output from the decoders 23A, 23B 
and 23C, and FIG. 17a show the wave spectrum diagram. 
The configuration and operation of the transmitter 10b shown in FIG. 13 is 
essentially identical to that of the first embodiment transmitter 10 shown 
in FIG. 1, but differs in that a packet assembler 16 for generating 
transmission data packets is further provided. Also, according to the 
transmitter 10b of FIG. 13, the transmission data is presented in packets, 
and the spread spectrum signal a corresponding to each packet is in a form 
of a burst signal. More specifically, the transmission data is first 
divided into data blocks of information bits 93 (FIG. 14), each block 
containing a predetermined number of bits. A preamble 91, unique word 92, 
and error detection bits 94 are then added to each data block of 
information bits 93 to form the data packets 61-64. These data packets 
61-64 are input to the data modulator 17, which outputs the data modulated 
signals in bursts corresponding to each packet. A (differential) PSK 
method using binary, 4-ary, or 8-ary PSK (or another number of phases) is 
used for the data modulation method. The basic configuration and operation 
of the data modulator 17 are identical to the configuration and operation 
of the differential encoder 11 and PSK modulator 12 shown in FIG. 1 of the 
first embodiment. Note that the data modulator 17 may also add a ramp wave 
with a smooth envelope at the beginning and end of each burst because the 
sudden rise and fall of each burst expands the width of the transmission 
signal spectrum. As in the first embodiment, the data modulated signal p 
is multiplied by the spreading signal q, and the spread spectrum signal a 
is output from the transmitter 10b in bursts corresponding to each packet. 
As will be described below, the unique word 92 is a fixed bit pattern 
string inserted to identify the corresponding data block of information 
bits 93 during the decoding operation of the receiver 20d. The error 
detection bits 94 are variable bit patterns inserted for the receiver 20d 
to determine whether there is a bit error in the information bits 93 or in 
the error detection bits 94 itself. In practice, a parity code or CRC 
(cyclic redundancy check) code is used for the error detection bits 94. 
The operation of this embodiment is further described below starting with 
the transmitter 10b and using binary PSK modulation as the data modulation 
method. 
The m.sup.th data dp.sub.m (a binary value of .+-.1) bit in the data 
packets 61-64 output from the packet assembler 16 is read synchronously to 
the symbol clock CK of period T output from the clock generator 15, and 
differentially coded and then modulated by the differential encoder 11 and 
PSK modulator 12 of the data modulator 17. The PSK modulator 12 output is 
thus a binary PSK modulated signal of symbol period T, and is input to the 
multiplier 14 as the data modulated signal p. The spreading signal 
generator 13 generates a spreading signal q synchronized to and with the 
same period as the symbol clock CK. The spreading signal q is, for 
example, a constant amplitude, pseudorandom pulse wave generated from 
pseudorandom series. The multiplier 14 multiplies the data modulated 
signal p and spreading signal q to output the spread spectrum signal a. 
The spread spectrum signal a is input through the transmission path to the 
receiver 20d, and bandwidth limited by the bandpass devices 21A, 21B or 
21C to obtain the intermediate signal b. FIG. 17a shows the spectrum of 
the received spread spectrum signal a and the pass bands B1, B2 and B3 of 
the bandpass devices 21A, 21B and 21C, respectively. Note that the number 
of pass bands shall not be limited to three as shown in FIG. 17a, and two 
or more plural bands can be used. Any plural number of bandpass device 21A 
can also be used. 
The intermediate signals b are then detected by the differential detectors 
22A, 22B and 22C, obtaining detector outputs c. These differential 
detectors 22A, 22B and 22C are the same, for example, as the differential 
detector 22 of the first embodiment. The process whereby the intermediate 
signals are detected by the corresponding differential detectors is the 
same as that of the first embodiment, and the data is decoded by 
evaluating the polarity of the detector output c. 
This detection process is illustrated in FIG. 15b. Specifically, same (or 
similar) wave pulses are multiplied resulting in a positive pulse when 
there is no phase change from the preceding wave symbol, but a negative 
pulse results when the phase is reversed from that of the previous wave 
symbol because opposite-symbol pulses are multiplied. As a result, 
detector output c will become a positive or negative pulse depending upon 
whether the waves are of same or opposite phase. The clock regenerators 
25A, 25B and 25C produce the regenerated symbol clock from the detector 
output c. Using this timing, the decoders 23A, 23B and 23C sequentially 
sample/recognize the detector output c, determine the polarity of the 
symbol at the sampling point, and output the evaluated data dp'.sub.m as a 
value of 1 or -1 when the detector output c is positive or negative, 
respectively. 
Note that while the above embodiment has been described using the binary 
PSK modulation, 4-ary PSK, 8-ary PSK, or other multilevel phase modulation 
method can be used. 
The evaluated data stream contains data packets 61'-64' (FIG. 16) 
corresponding to the data packets 61-64 shown in FIG. 14 and identically 
formatted. The unique word detectors 26A, 26B and 26C are previously 
stored with the unique word and compare the unique word detected from the 
evaluated data dp'.sub.m with the previously stored unique word. When the 
detected and the stored unique words match, the unique word detector 
produces a frame signal to the corresponding packet extractor 27A, 27B or 
27C. Based on the timing of this frame signal, the packet extractors 27A, 
27B and 27C extract the data packet 95' comprising a data block of 
information bits 93' and error detection bits 94', and forward the packet 
to the error detectors 29A, 29B and 29B. Based on the corresponding error 
detection bits 94', the error detectors 29A, 29B and 29C detect errors, if 
any, in the decoded data packet 95', and output the detection results 
(error signal e if error is detected) and the data blocks of information 
bits 93' in the decoded data packets 95' to the evaluator/selector 240. 
The evaluator/selector 240 selects the data block 93' only for the 
channels in which no error signal e is produced, and outputs the data 
blocks 93' as the final decoded data of the receiver 20d. 
In this embodiment, there are two possible cases in which each error 
detector will produce an error signal e. One is when the unique word 
matching is not successful in the unique word detector, and the other is 
when the unique word matching is successful in the unique word detector, 
but some errors are detected in the decoded data packet 95'. 
This embodiment is described in further detail below with reference to the 
operation when a jamming j as shown in FIG. 17a is applied to the 
transmission path. Normal reception by the conventional apparatus shown in 
FIG. 26 is not possible because most of the jamming j energy is detected 
by the differential detector. With the apparatus according to the present 
embodiment as shown in FIG. 13, however, bandpass device 21A, 21B and 21C 
pass only pass bands B1, B2 and B3, respectively, of the transmitted 
spread spectrum signal a. In the example shown in FIG. 17a, since the 
bandpass device 21A is set to pass band B1, the intermediate signal b 
input to the detector 22A for that channel A will not be affected by the 
jamming j, and normal reception is possible and the error detector 29A in 
that channel A will not detect bit error. The signals in the other 
channels B and C which are set to pass bands B2 and B3, will be affected 
by the jamming j causing disturbance in the receiving condition. Thus, in 
these other channels B and C, the error detectors 29B and 29C will detect 
bit errors. Thus, the evaluator/selector 240 will select the data block 
93' from the bit error-free channel A and output this data block 93' as 
the decoded data. 
FIG. 17b is a block diagram of a data transmitting and receiving apparatus 
according to a modification of the fifth embodiment of the invention. 
In this modification, instead of three detection units, only one detection 
unit is provided. Also, the bandpass device 21 is the same as the one used 
in the first embodiment so that the bandpass device 21 has three bandpass 
filters which are sequentially selected by the error signal e. 
Furthermore, no evaluator/selector 240 is provided. 
FIG. 18 is a block diagram of a data transmitting and receiving apparatus 
according to a sixth embodiment of the invention. The transmitter 10b in 
this embodiment is identical to the transmitter 10b of the fifth 
embodiment shown in FIG. 13. The receiver 20e has two or more channels 
(two channels are shown in FIG. 18). The configuration and operation of 
the receiver 20e are essentially identical to those of the receiver 20d in 
the fifth embodiment, but differs in that the bandpass device 21A and 21B 
each has the same structure as the bandpass device of the first 
embodiment. Thus, each bandpass device 21A, 21B has a plurality of, such 
as three bandpass filters for passing different bands, e.g., B1, B2 and 
B3, or B1+B2+B3, B1+B2 and B1. Specifically, when any one of the error 
detectors 29A and 29B detects a bit error and produces an error signal e', 
the corresponding bandpass device is switched so that the passing band is 
changed or narrowed, or the center frequency of the passing band is 
changed, or the passing band is changed or narrowed and at the same time, 
the center frequency of the passing band is changed. 
If the total bandwidth of the combined individual bandwidths of the 
bandpass means 21A and 21B is only part of the bandwidth of the 
transmitted spread spectrum signal a, efficient jamming prevention is made 
possible by changing the pass band of the corresponding bandpass means to 
a band not used for reception when noise is detected using the bit error 
detection of the error detectors 29A and 29B. For example, even if there 
are plural (3 or more) pass bands and there are only two reception 
channels from the bandpass device 21A and 21B to the error detectors 29A 
and 29B assigned to two of these plural pass bands, the probability that 
jamming will interfere with both reception channels simultaneously is low. 
In addition, when jamming interferes with reception on one channel, that 
channel can be assigned to the unused band, and efficient jamming 
prevention can be achieved without greatly increasing the scale of the 
hardware. Note that the bandpass device 21A and 21B shown in FIG. 18 are 
configured similarly to the bandpass device 21a shown in FIG. 2 for the 
first embodiment, for example, and the pass band is changed by selectively 
choosing from the plural bandpass filters. Note that part or all of the 
bandpass filters can be shared by part or all of the bandpass device. As 
in the first embodiment, the bandpass device 21A and 21B can be configured 
as shown in FIG. 5 to vary the center frequency of the pass bands by 
varying the frequency of the local oscillation signal. 
Note that as the pass band width of the bandpass device 21A and 21B 
increases, the usable bandwidth of the transmitted spread spectrum signal 
a increases, and reception sensitivity improves. On the other hand, 
jamming interference also increases as the pass band width increases. To 
handle this, the present embodiment can be configured to narrow the pass 
band width of the corresponding bandpass device when interference is 
detected using the bit error detection of the error detectors 29A and 29B. 
With this design, sensitivity can be emphasized when there is no 
interference, jamming prevention can be emphasized when there is 
interference, and reception characteristics with good overall balance can 
be achieved. Note that once the pass band width is narrowed, the noise or 
jamming source is determined to have disappeared if no bit errors are 
detected for a predetermined time, and the pass band width is then 
restored to the full band width. 
In this embodiment, it is possible to arrange the bandpass devices 21A and 
21B such that bandpass device 21A is provided with an up counter 209a and 
bandpass device 21B is provided with a down counter 209b so as to avoid 
selecting the same band simultaneously in both bandpass devices 21A and 
21B. If the same band is selected simultaneously in both bandpass devices 
21A and 21B, it is possible to advance the band selection in one of the 
bandpass devices. This can be done by monitoring the counters 209a and 
209b. 
Alternatively, it is possible to arrange the bandpass devices 21A and 21B 
such that bandpass device 21A narrows the bandwidth in the steps of 
B1+B2+B3 .fwdarw. B1+B2 .fwdarw. B1, and bandpass device 21B narrows the 
bandwidth in the steps of B1+B2+B3 .fwdarw. B2+B3 .fwdarw. B3. Another 
detection unit may be provided so that its bandpass device narrows the 
bandwidth in the steps of B1+B2+B3 .fwdarw. B1+B3 .fwdarw. B2. 
FIG. 19 is a block diagram of a data transmitting and receiving apparatus 
according to a seventh embodiment of the invention. The transmitter 10b in 
this embodiment is identical to the transmitter 10b of the fifth 
embodiment shown in FIG. 13. The configuration and operation of the 
receiver 20f are essentially identical to those of the receiver 20d in the 
fifth embodiment or the receiver 20e in the sixth embodiment. This 
receiver 20f differs in that frame error detectors 28A and 28B are added, 
and the frame error signals output therefrom are used to switch the 
corresponding bandpass device so that the passing band is changed or 
narrowed, or the center frequency of the passing band is changed, or the 
passing band is changed or narrowed and at the same time, the center 
frequency of the passing band is changed. 
The operation of the receiver 20f of the embodiment shown in FIG. 19 is 
described below with further reference to FIG. 20. The complete frame 
signals output from the unique word detectors 26A and 26B are input to the 
frame error detectors 28A and 28B, respectively. The frame error detectors 
28A and 28B evaluate unique word detection failures in the respective 
systems, and output a frame error signal accordingly. This operation is 
illustrated in FIG. 20. Referring to FIG. 20, the regenerated clocks A and 
B are output by the clock regenerators 25A and 25B, the evaluated data A 
and B are output by decoders 23A and 23B, and frame signals A and B are 
output by the unique word detectors 26A and 26B, respectively. The frame 
error signal B is output from frame error detector 28B. 
If it is assumed that the unique word detector 26A finds the end of the 
unique word 92' and outputs the corresponding frame signal A at a given 
time as shown in FIG. 20, output of the frame signals from the other 
channels is monitored for a predetermined monitoring period. If frame 
signal B is output (the dotted line) during this monitoring period as 
shown in FIG. 20, the frame error signal B is not output. If, however, the 
frame signal B is not output (solid line) during this monitoring period, 
the frame error signal B is output at the end of the monitoring period. 
The monitoring period is used to avoid evaluation errors caused by 
transmission path/signal processing delay characteristics or jitter in the 
regenerated clocks, and must be at least as long as approximately one 
symbol period. FIGS. 19 and 20 also illustrate the use of two reception 
channels, but the above description also applies to three or more 
reception channels. In this case, the monitoring period starts at the 
timing of the earliest frame signal output by one of the other reception 
channels. 
If a frame error is detected, the corresponding bandpass device is switched 
so that the passing band is changed or narrowed, or the center frequency 
of the passing band is changed, or the passing band is changed or narrowed 
and at the same time, the center frequency of the passing band is changed 
based on the frame error signal using the same means as described in the 
sixth embodiment above. When jamming interferes with reception, unique 
word detection fails and a frame error is generated. As a result, 
efficient jamming prevention can be achieved without greatly increasing 
the scale of the hardware, or reception characteristics balancing 
reception sensitivity with jamming prevention can be obtained as in the 
sixth embodiment. Furthermore, while the next decoding data packet may 
also be lost because the error detectors 29A and 29B must wait for the end 
of the decoding data packet before bit error detection and pass band or 
pass band width adjustment occurs after that, evaluation is completed at 
an early point in the decoding data packet in the present embodiment (see 
FIG. 16), and this type of problem thus does not occur in the present 
embodiment. 
FIG. 21 is a block diagram of a data transmitting and receiving apparatus 
according to an eighth embodiment of the invention. The transmitter 10b in 
this embodiment is identical to the transmitter 10b of the fifth 
embodiment shown in FIG. 13. The configuration and operation of the 
receiver 20g are essentially identical to those of the receiver 20f in the 
seventh embodiment. This receiver 20g differs in that the packet 
extractors 27A' and 27B' can be triggered to extract the data packet not 
only by the frame signal from its own detection unit (same channel), but 
also by the frame error signal from the other detection unit (other 
channel). If the unique word detector of the same channel fails to detect 
the unique word to produce a frame signal, the packet extractors 27A' and 
27B' may not be triggered to extract the data packet, but is triggered to 
extract the data packet by the frame error signals output by the frame 
error detectors 28A and 28B. Thus, in such a case, the packet extractors 
27A' and 27B' will extract the decoded data packet based on the timing of 
the frame error signal from the other channel. 
The operation of the receiver 20g of the embodiment shown in FIG. 21 is 
described below with further reference to FIG. 22. Referring to FIG. 21, 
the frame error signals output by the frame error detectors 28A and 28B 
are input to the corresponding packet extractors 27A' and 27B'. The 
operation of the frame error detectors 28A and 28B illustrated in FIG. 22 
is identical to that of the seventh embodiment shown in FIG. 20, and 
further description is therefore omitted. What differs from the seventh 
embodiment is that a decoded data packet extraction function is added to 
the packet extractors 27A' and 27B' so as to extract the decoded data 
packet in response to the frame error signal of the other channel. 
Specifically, when the frame signal B of a given channel (channel B in 
FIG. 22) is not output, the frame error signal B is substituted for the 
frame signal B, and an equivalent operation is performed on the delayed 
evaluated data B (to which a known correction delay is applied) and the 
regenerated clock B' (of which the timing is adjusted to the delayed 
evaluated data B) using the frame error signal B to extract the decoded 
data packet. 
When the length of the unique word is set sufficiently long, the 
probability of false unique word detection is extremely low but the 
probability of not detecting unique words increases greatly. In 
particular, when burst transmission is used as in this embodiment, unique 
word bit errors caused by the AGC or tracking errors in the 
synchronization systems occur frequently even when bit errors are not 
present in the data block or error detection bits because the unique words 
are found at the beginning of each burst as shown in FIG. 14. As a result, 
unique words are undetected, extraction of decoded data packets fails, and 
data is lost. However, while the timing of the evaluated data of each 
reception channel may be affected by propagation delay differences, signal 
processing time differences, and cross time differences comparable to 
regenerated clock jitter, the duration of these factors is generally less 
than half of the symbol period, and is sufficiently short. As a result, 
the data block can be decoded in these cases even if the unique word is 
undetected by extracting the decoded data packet using the detection 
timing of another channel, and reception quality can be improved. 
Because the frame error signal B is to begin with the frame signal A from a 
channel on which the unique word was detected delayed by the monitoring 
period as shown in FIG. 22, decoded data packet extraction is possible 
using the detection timing from another channel even if the unique word is 
not detected by setting the correction delay time equivalent to the 
monitoring period. While the correction delay time can be applied to 
evaluated data B (delayed evaluated data B) and the regenerated clock B 
simultaneously delayed by an equal amount (regenerated clock B'), it is 
sufficient to adjust the timing by the difference between an integer 
multiple of the repeat period and the correction delay time because the 
regenerated clock is a repeating wave. Note that because the monitoring 
period and correction delay time shown in FIG. 22 are one regenerated 
clock period, delay of the regenerated clock B is not necessary. 
Note also that while FIG. 21 shows a receiver 20g with two reception 
channels, the receiver 20g can be expanded to three or more reception 
channels as in the seventh embodiment, in which case the description 
provided in the seventh embodiment also applies. 
When 4-ary PSK or greater multilevel transmission is used, parallel/serial 
converters are provided in the decoders 23A and 23B as described in the 
fifth embodiment above. Because these parallel/serial converters output 
the evaluated data A and B and corresponding regenerated clocks A and B 
(the unique word detectors 26A and 26B compare bit strings in this case), 
the regenerated clocks and evaluated data (FIG. 22) can in this case be 
thought of as bit clocks and evaluation bit data (in a binary PSK channel 
the bit strings and symbol strings match). However, if there is a time 
difference between reception channels (of a maximum 0.5 symbol period), 
and the same-channel timing is estimated using the timing of another 
channel based on a bit clock with a short period 1/2 (in a 4-ary PSK 
system) or 1/3 (in an 8-ary PSK system) of the symbol clock period), bit 
shifting occurs and decoded data packet detection failures increase. To 
compensate during multilevel transmission with a 4-ary PSK or greater 
system in the present embodiment, the decoders 23A and 23B output 
evaluated data A and B (which are symbol strings) and the corresponding 
regenerated symbol clocks A and B (the unique word detectors 26A and 26B 
compare symbol strings in this case), and the regenerated clocks and 
evaluated data (FIG. 22) preferably express the symbol clock and 
evaluation symbol data. Conversion from a symbol string to a bit string is 
possible in this case by providing a parallel/serial converter immediately 
before the decoded data packet output of the packet extractors 27A' and 
27B', or in the error detectors 29A and 29B. Alternatively, the last 
decoded data can be output as a symbol string. 
When the spreading signal q is a chirp signal as described in the fourth 
embodiment with reference to FIG. 12, the peak positions of the detector 
outputs differ and are determined by the pass band characteristics of the 
bandpass device 21A and 21B and the characteristics of the spreading 
signal q. The above correction delay time should therefore also add timing 
correction determined by the pass band characteristics of the bandpass 
device 21A and 21B and the characteristics of the spreading signal q as 
described below. This timing correction is illustrated in FIG. 23, a 
waveform diagram of the detection process as shown in FIG. 15 where a 
chirp signal is used for the spreading signal q. The pass bands B1-B3 of 
the bandpass device 21A and 21B are shown in FIG. 23a. The intermediate 
signals b1-b3 and detector outputs c1-c3 corresponding to the pass bands 
B1-B3 are shown in FIG. 23b according to the detection process. The 
detection operation is the same as in the first embodiment, and further 
description is omitted. 
As shown in FIG. 23b, time t13 is the difference between the pulse peak 
position of detector output c1 and the pulse peak position of detector 
output c3, and time t23 is the difference between the pulse peak positions 
of detector outputs c2 and c3. These timing periods are clearly determined 
by the parameters of the chirp signal and the characteristics of the pass 
bands B1-B3. When the decoded data packets are extracted using the timing 
provided by the frame signal from another channel as in this embodiment, 
it is preferable to provide timing correction equivalent to the offset of 
the peak positions as described above (i.e., times t23 and t13 in FIG. 
23). More specifically, the correction delay time shown in FIG. 22 is 
preferably the sum of the monitoring period and this timing correction 
period (the delay between the process channel and the frame signal 
reference channel). For example, when extracting the decoded data packet 
of the c1 detector output channel based on the frame signal from the c3 
detector output channel, the correction delay time is the sum of the 
monitoring period and time t13 (which is a negative value reducing the 
delay). 
FIG. 24 is a block diagram of a data transmitting and receiving apparatus 
according to a ninth embodiment of the invention. The transmitter 10b in 
this embodiment is identical to the transmitter 10b of the fifth 
embodiment shown in FIG. 13. The configuration and operation of the 
receiver 20b are essentially identical to those of the receiver 20g in the 
eighth embodiment. This receiver 20h further has selectors 251A and 251B 
controlled by the outputs of the frame error detectors 28A and 28B, timing 
adjusters 252A and 252B for adjusting the timing of the regenerated 
clocks, and decoders 23A' and 23B' for referencing the regenerated clock 
of the other channel through the timing adjusters 252A and 252B and 
outputting the evaluated data. 
When one unique word detector fails the unique word detection and the 
corresponding frame error detector outputs the frame error signal in this 
receiver 20h (FIG. 24), the decoded data packet is extracted from the 
evaluated data decoded using the regenerated clock output from the clock 
regenerator on the other channel. 
The operation of the embodiment shown in FIG. 24 is described below with 
reference to FIG. 25. The timing adjusters 252A and 252B (FIG. 24) delay 
the regenerated clocks A and B output by the clock regenerators 25A and 
25B by a predetermined period to adjust the timing, and output the 
regenerated clocks A' and B' to the other-channel decoders 23B' and 23A', 
respectively. The added decoders 23A' and 23B' output the evaluated data 
A' and B' based on the other-channel regenerated clocks B' and A', 
respectively. When the frame error signals are input from the 
corresponding frame error detectors 28A and 28B, the selectors 251A and 
251B switch from the evaluated data A and B output from the normal 
decoders 23A and 23B (described in the fifth embodiment above) to the 
above evaluated data A' and B'. While various reasons can be offered to 
explain unique word detection failures, if the cause is a regenerated 
clock tracking failure, there is a high probability of bit errors being 
contained in the extracted decoded data even when using the unique word 
timing from the other channel as described in the eighth embodiment. An 
improvement in reception quality can be expected in the present 
embodiment, however, because the regenerated clock is also supplied from 
the other channel. 
Operation when there is a unique word detection failure on channel B is 
illustrated in FIG. 25. Specifically, FIG. 25 shows the timing at the edge 
of the eye pattern (the diamond shaped area in the detector output B) 
resulting from the regenerated clock B output from the same-channel clock 
regenerator 25B not accurately tracking the detector output B of the 
differential detector 22B. As a result there is a high probability of bit 
errors being contained in the evaluated data B output by the decoder 23B. 
On the other hand, there is a low probability of bit errors being 
contained in the evaluated data B' output by the decoder 23B' using the 
regenerated clock A' obtained through the timing adjuster 252A from the 
regenerated clock A output by the other-channel clock regenerator 25A. If 
bit errors occur in the evaluated data B, unique word detection fails, and 
the frame error signal B is output as described in the seventh embodiment. 
The selector 251B therefore supplies evaluated data B' to the packet 
extractor 27B', which substitutes the frame error signal B for the frame 
signal and begins the decoded data packet extraction operation as 
described in the eighth embodiment. An improvement in reception quality 
even greater than that obtained with the eighth embodiment can therefore 
be expected because evaluated data B' having a high probability of good 
data is selected. 
Note that the adjustment time of the timing adjusters 252A and 252B is 
normally equivalent to the signal processing delay, or can be completely 
eliminated, but it is necessary to add a delay period determined by the 
frequency sweep parameters of the spreading signal q and the 
characteristics of each bandpass device (as shown by t23 and t13 in FIG. 
23 in the eighth embodiment) when the spread spectrum signal a is a chirp 
signal within each symbol period. For the correction delay time, however, 
it is sufficient to delay the monitoring period equivalent by the 
regenerated clock repeat period. Unlike the eighth embodiment, however, 
this correction is applied not to the input side of the packet extractors 
27A' and 27B' but to the detector output c on the input side of the 
decoders 23A' and 23B' or the evaluated data on the output side. 
Note also that, as in the eighth embodiment, only two reception channels 
are shown in FIG. 24, but the receiver 20h may be expanded to three or 
more channels, in which case the description provided in the eighth 
embodiment applies. 
In addition, during 4-ary PSK or greater multilevel transmission, the 
decoders 23A, 23B and 23A', 23B' may comprise an internal parallel/serial 
converter and output a bit clock and evaluation bit data, but it is 
preferable to output the symbol clock and evaluation symbol data to reduce 
the occurrence of decoded data packet extraction failures due to bit 
shifting. 
The invention being thus described, it will be obvious that the same may be 
varied in many ways. Such variations are not to be regarded as a departure 
from the spirit and scope of the invention, and all such modifications as 
would be obvious to one skilled in the art are intended to be included 
within the scope of the following claims.