Dual conversion FM receiver using phase locked direct conversion IF

A receiver for frequency modulated signals, having down-conversion to a baseband, zero intermediate frequency for selectivity, followed by up-conversion to a non-zero IF for amplification, limiting, and demodulation. A phase-lock loop locks the frequency of the down-conversion source to the center frequency of the signal coupled to the baseband IF, converting the signal to precisely zero frequency. This avoids the beat note often found in direct conversion systems. The phase-lock loop additionally provides inherent demodulation of the FM signal. The received FM signal is coupled to the baseband IF through a radio frequency amplifier for improved sensitivity and local oscillator isolation, or through a first IF comprising a mixer, local oscillator, filter, and amplifier.

BACKGROUND OF THE INVENTION 
This invention relates generally to radio receivers for frequency modulated 
(FM) signals and, more particularly, to FM receivers incorporating direct 
conversion to baseband. 
DESCRIPTION OF THE PRIOR ART 
Several approaches to FM receiver design are known in the art. The commonly 
used superheterodyne receiver converts an incoming radio signal to one or 
more intermediate frequencies at which amplification and frequency 
selection can more readily be performed than at the frequency of the 
received signal. Conversion to an intermediate frequency (IF) is 
accomplished by mixing the received signal with a locally generated 
oscillator (LO). After filtering and amplification at a first IF, the 
signal may be immediately demodulated in a detector circuit, which 
constitutes a "single conversion" receiver. Alternatively, dual and triple 
conversion designs are known having successive conversion to additional 
IF's followed by demodulation of the signal. 
An undesirable feature of conventional superheterodyne receivers has been 
the difficulty in applying microelectronic techniques to achieve 
miniaturization. To fabricate amplifier circuits in monolithic form has 
long been known, but it has been difficult to integrate the high-Q crystal 
or ceramic bandpass filters generally used for selectivity at the 
intermediate frequencies. A solution known in the art is to convert an 
incoming signal directly to baseband in what is known as a "direct 
conversion" or "zero-IF" receiver. The local oscillator frequency is made 
equal to the carrier frequency of the received signal, and the spectrum 
occupied by the modulation is translated directly to baseband. The 
necessary sharp selectivity is then achieved through lowpass, rather than 
bandpass, filtering. Low frequency lowpass filters are readily fabricated 
in monolithic form, allowing a much greater degree of miniaturization. 
Direct conversion receivers for FM signals may include a technique used for 
single-sideband (SSB) reception known as the "third-" or "Weaver-method," 
disclosed in Proc. IRE 44 (1956), pages 1703-1705, and the subject of U.S. 
Pat. No. 2,928,055. As applied to FM detection, the purpose of the 
technique is to distinguish the modulation information carried by positive 
and negative frequency excursions about the carrier. When an FM signal is 
mixed with a down-conversion oscillator to translate it to baseband, equal 
positive and negative frequency excursions about the carrier result in the 
same difference frequency, and the polarity of the modulation can no 
longer be determined without some phase reference. The Weaver method 
provides two substantially identical paths in which the signal is 
down-converted to baseband, lowpass filtered to remove the sum products of 
mixing as well as undesired adjacent channel signals, and up-converted to 
an output frequency. The down-and up-conversion oscillators for one path 
are in phase quadrature with their counterparts in the other path. The 
process of two successive frequency conversions produces phase inversions 
between the sidebands of the signals in the two paths. When the outputs of 
the two paths are added, cancellation of the sidebands occurs in such a 
manner that the modulation polarity of the original input signal is 
retained, though translated to a new, predetermined output frequency. In 
effect, the received signal is translated from an incoming frequency to 
baseband, filtered to remove interfering adjacent channel signals, and 
retranslated to an output frequency at which conventional FM demodulation 
can take place. Such a circuit following the Weaver method may be termed a 
"translating bandpass filter" (TBPF). 
Prior art FM receivers following the direct conversion approach suffer from 
a number of shortcomings. For example, in a direct conversion receiver, 
the LO frequency equals the received signal carrier frequency, and if no 
RF preamplifier is used, there is very little reverse isolation to prevent 
LO energy from reaching the antenna and causing interference with other 
receivers. Furthermore, noise and DC offsets make it difficult at baseband 
to achieve the low noise amplification and high gain required for adequate 
sensitivity. 
In a superheterodyne FM receiver, amplitude limiting is generally used to 
reduce noise and to improve signal capture. However, to maintain fidelity 
of the modulation in the baseband zero-IF application, both instantaneous 
amplitude and phase must be preserved. Limiting is not practical. Strong 
adjacent channel signals that overload the IF can cause distortion if the 
circuits are not designed for wide dynamic range. 
Finally, if there is an offset in frequency between the incoming signal and 
the down-conversion oscillator in the zero-IF, several undesired results 
may occur. If the baseband paths are imperfectly matched, the cancellation 
of mixing products will be incomplete, and a beat-note will oocur. The 
beat-note has a primary component at twice the offset frequency but can 
have distortion products at other harmonics of the offset frequency. If 
the beat note is in the audible range, it can interfere with demodulated 
audio output. Furthermore, the lowpass filter bandwidth necessary to pass 
an FM modulated signal increases by the amount of the offset, and it 
becomes difficult to obtain narrow selectivity with other than negligible 
offsets. It is known in the design of FM receivers for broadcast 
reception, for example, that a low-level beat-note slightly above the 
audible band (approx. 20 kHz), caused by a 10 kHz offset, does not 
interfere with reception. That is because the baseband lowpass filters 
already have cutoff frequencies on the order of 100 kHz to pass a 
broadcast FM signal, and they can accommodate a 10 kHz offset without 
significantly distorting the modulation. However, in other applications, 
say, for example, the land-mobile service in which channel spacing may be 
as close as 12.5 kHz, the baseband filtering must be narrower than 
one-half the spacing, or about 6 kHz, in order to separate the adjacent 
channel signal from the desired. A frequency offset higher than a few tens 
of Hertz would require that the baseband filters be widened to accommodate 
the modulation swing plus offset, and this would degrade selectivity. 
Frequency offset must therefore be tightly controlled 
SUMMARY OF THE INVENTION 
It is an object of the present invention to provide a direct conversion FM 
receiver that nevertheless overcomes the foregoing deficiencies. 
It is a more particular object to provide a receiver having direct 
conversion to baseband to enable the selectivity function to be performed 
with low frequency lowpass filtering and to thereby facilitate 
miniaturization through microelectronic techniques. 
It is a further object of the invention to provide a zero-IF system with 
sharp selectivity to be used to receive FM signals transmitted in a narrow 
channel spacing environment. 
It is also an object to avoid the detrimental effects of mismatch in a 
zero-IF system and to avoid a beat note by providing for precise frequency 
control of the frequency conversion sources in the IF. 
A further object is to include in a direct conversion FM receiver the 
advantageous placement of the amplification and limiting functions at a 
non-zero intermediate frequency as in a superheterodyne receiver. 
The invention lies in the use of a baseband zero-IF for providing the 
ultimate selectivity, followed by up-conversion to an intermediate 
frequency at which amplification and limiting take place. Frequency 
control by phase lock technique is used to precisely center the zero-IF 
signal. The phase lock loop also demodulates the received signal.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
Referring now to the drawings, an FM receiver 10 is shown in FIG. 1, which 
receiver has been constructed in accordance with the present invention. An 
incoming signal 12 is amplified in preamplifier 14, a major function of 
which is to provide low noise amplification that establishes the 
sensitivity of the receiver. The preamplifier 14 additionally provides 
reverse isolation that reduces the escape of radiation from voltage 
controlled oscillator (VCO) 40. The output of preamplifier 14 feeds a 
translating bandpass filter 30 (TBPF). Filter 30 may be of a type known in 
the art for accomplishing direct conversion that is frequently referred to 
as a "Weaver" circuit. A detailed description of this prior art circuit 
may be found in Weaver, D. K. Jr. "A Third Method of Generation and 
Detection of Single-Sideband Signals," Proc. IRE, vol. 44, pp. 1703-1705, 
December 1956. This circuit translates a signal from a known input 
frequency to a predetermined output frequency while simultaneously 
effecting appropriate bandpass filtering. A more detailed description of 
filter 30 is set forth in FIG. 2 and will be described hereinafter. 
The input signal 31 from preamplifier 14 is applied to each of two 
parallel, substantially identical, paths 32a-33a-34a and 32b-33b-34b. 
Elements 32a and 32b are like down-conversion mixers that translate the 
incoming signal to essentially baseband. A down-conversion frequency is 
supplied at line 35 in quadrature to both mixers 32a and 32b using phase 
shifter 36 or the equivalent to provide two signals in phase quadrature. 
The outputs of mixers 32a and 32b are fed to two identical lowpass 
filters, 33a and 33b, having a cutoff frequency on the order of one-half 
the bandwidth of the desired passband. The design of these lowpass filters 
affects the frequency response of the demodulated signal and will be 
treated below in discussion of the phase lock loop components. The 
respective outputs of these lowpass filters are coupled to up-conversion 
mixers 34a and 34b, which are likewise supplied in phase quadrature with 
frequency f.sub.up, at 37, through phase-shifter 38 or similar apparatus 
for providing quadrature signals. The up-converted signals from mixers 34a 
and 34b are summed in element 39 to provide a bandpass filtered output at 
a predetermined frequency. The circuit elements of the TBPF require good 
matching, and it is advantageous to fabricate them in monolithic form. 
Referring again to FIG. 1, it may be seen that the incoming signal is 
translated from frequency f.sub.down to frequency f.sub.up while being 
filtered to a desired selection bandwidth. The output 41 of TBPF 30 is 
then applied in sequence to filter 60, amplifier 70, and then to limiter 
80. The purpose of filter 60 is to attenuate harmonics of f.sub.up and 
their sidebands, which may be generated in mixers 34a and 34b in the TBPF. 
The harmonics must be suppressed so that limiter 80 does not respond to 
them. It should be noted that filter 60 in fact is unnecessary with 
certain types of up-conversion mixers. Amplifier 70 and limiter 80 provide 
the bulk of the receiver gain. Gain in the zero-IF path is intentionally 
avoided so as to prevent overload, which would result in distortion of the 
demodulated signal. Limiter 80 functions both to reject amplitude noise 
variations and to maintain a constant amplitude signal for phase detector 
90. This keeps the phase detector gain constant, which is necessary for 
control of the overall loop gain and closed loop frequency response. 
Up-conversion oscillator 50, phase detector 90, loop filter 100, and 
voltage controlled oscillator (VCO) 40 form a phase lock loop (PLL) that 
controls the zero-IF down-conversion frequency. The use of a phase lock 
loop for frequency control and the components constituting the loop are 
well known in the prior art. For example, see Gardner, F. M., Phaselock 
Techniques, New York: Wiley, 2d ed. 1979, or Blanchard, A., Phase Locked 
Loops, New York: Wiley, 1976. 
The output of limiter 80 is applied to phase detector 90, for which it is 
advantageous to have a wide range phase capability (.+-.2Pi radians). The 
other input to the phase detector is the in-phase component of f.sub.up, 
37. The phase detector output is applied through loop filter 100 to the 
control line of oscillator 40, which may be a voltage controlled 
oscillator (VCO) or voltage controlled crystal oscillator (VCXO). Loop 
filter 100 is a lowpass filter used to prevent spurious harmonics of the 
phase detector reference frequency from modulating the controlled 
oscillator. If a low feedthrough sample-and-hold phase detector is used, 
filter 100 may be unnecessary. The signal at the oscillator control line, 
110, is the demodulated audio of the incoming signal 
When the loop locks, both inputs to the phase detector are at the same 
frequency, f.sub.up. This happens when the frequency of VCO 40 exactly 
equals the carrier frequency of signal 31 applied to the TBPF 30. Then, 
the signals in the quadrature baseband paths are precisely centered about 
zero frequency, and the up-converted output signal 41 equals f.sub.up. 
Under these conditions, no beat-note will exist in the baseband paths. 
Several design details of the components comprising the translating 
bandpass filter and the phase lock loop are noteworthy. The phase lock 
loop not only controls the down-conversion frequency, but it also recovers 
the modulation of the input signal. The amplitude versus modulation 
frequency response of the demodulated output is determined by the closed 
loop response of the oscillator control signal versus deviation frequency. 
The closed loop frequency response depends on the open loop pole locations 
and the overall loop gain. To obtain a desired closed loop response, one 
design technique is to work back to find the appropriate open loop filter 
response and open loop gain to give the closed loop response. 
The dynamic elements of the PLL include lowpass filters 33A and 33B, 
bandpass filter 60, and loop filter 100. With so many poles of filtering 
in the loop, it is difficult to maintain stability unless the loop gain 
falls below unity at a frequency below which 180.degree. of phase shift is 
reached. It is generally found that the loop gain necessary to satisfy the 
dual requirements of stability and frequency response is low compared with 
typical loop gains in phase lock loop demodulators having equivalent 
bandwidth. This occurs because of the greater number of dynamic filtering 
elements in the system here. If the phase detector is designed for low 
spurious output, filtering of its output by loop filter 100 may be 
avoided, reducing the overall number of poles in the loop and simplifying 
the design. 
Further considerations in the choice of loop components relate to adjacent 
channel selectivity. Adjacent channel selectivity depends primarily on the 
open-loop attenuation characteristics of the loop. For best selectivity, 
the cutoff frequency of the open loop response should be as low as 
possible. The loop is designed to track the deviation of the input signal 
up to a predetermined modulation cutoff frequency. For the loop to 
properly track the input signal, the phase error between the two signal 
inputs to the phase detector must not exceed the phase difference 
capability of the detector. The maximum phase error depends on the loop 
gain, the peak deviation and maximum modulation frequency of the input 
signal, and the cutoff frequency of the open loop response. Because the 
loop gain is otherwise made low for stability reasons, and the loop cutoff 
frequency is low for selectivity reasons, the phase error may be large. It 
is therefore advantageous to design the phase detector for a wide range 
phase difference capability. 
Because the signals in the translating bandpass filter 30 are centered 
about zero frequency, DC offsets and carrier feedthrough can affect 
receiver performance. The chief problem of carrier feedthrough at f.sub.up 
is that it would tend to capture the limiter under weak- or no-signal 
conditions, causing self-quieting and degrading sensitivity. Feedthrough 
may be caused by poor carrier suppression in up-conversion mixers 34A and 
34B and by DC offsets in the baseband circuits. DC offsets may be 
minimized by the use of differential circuits in the zero-IF. 
The embodiment illustrated in FIG. 1 accomplishes the above mentioned 
objects of the invention-while providing a particular realization of 
direct conversion FM receiver. The translating bandpass filter 30 provides 
a zero-IF system that is adaptable to microelectronic techniques. Through 
precise frequency control as provided by the phase lock loop, the zero-IF 
signals are centered at baseband, and no beat note will arise, even with 
imperfect matching of the quadrature paths. Narrow bandpass filtering is 
achieved through the use of low frequency lowpass filter elements, which 
are readily fabricated in monolithic form. Signal amplification is 
primarily achieved in a non-zero IF, which may be operated in limiting as 
is known in the FM receiver art to be advantageous for good noise and 
capture performance. Furthermore, the problem of achieving low noise gain 
at baseband is avoided through the use of a low noise radio frequency 
preamplifier that dominates the noise performance of the entire system. It 
is the cooperation of circuit elements, in particular precise frequency 
control along with up-conversion to a non-zero IF, that makes possible 
achieving these objects 
FIG. 3 shows an embodiment of the invention that illustrates how the 
teachings of the invention may be used with a first, non-zero IF stage 
instead of an RF preamplifier. This arrangement can offer improved 
performance in an environment of strong signals that may cause 
intermodulation distortion or in which the frequency band of signals to be 
received is such that low noise amplification is difficult or uneconomical 
to achieve. 
Whereas FIG. 1 illustrates a receiver 10 in which the incoming signal 12 is 
coupled through preamplifier 14 to the TBPF 30, FIG. 3 depicts receiver 
10' used in conjunction with a first IF stage 14' for coupling the 
incoming signal 12 to the TBPF 30. Stage 14' includes a preselector 201, a 
first local oscillator 202 and its injection filter 203, mixer 204, first 
IF filter 205, and IF amplifier 206. The preselector 201 is a bandpass 
filter that blocks all but a selected band of radio channels from 
appearing at mixer 204, protecting the mixer from overload from undesired 
signals and preventing signals at the so-called "image" frequency from 
being converted to IF. The oscillator 202 provides the frequency by which 
incoming signals will be translated in mixer 204. Injection filter 203 
prevents extraneous noise energy from oscillator 202 from degrading the 
mixer noise performance. The output of mixer 204, which is at a 
predetermined intermediate frequency, is filtered in first IF filter 205 
to remove the undesired products of the frequency conversion process and 
to couple the desired signal to first IF amplifier 206. The output of 
amplifier 206 is coupled to receiver 20 as described earlier in the 
description of FIG. 1. 
In addition to providing traditional functions of an IF stage, the elements 
of block 14' also protect the TBPF circuits from overload. Signal levels 
are chosen so that in the presence of excessively strong on-channel 
signals the circuits in stage 14' will saturate and thereby provide a 
predetermined maximum output amplitude below the overload point of the 
zero-IF circuits. Protection from limiting on strong off-channel signals 
is provided by IF filter 205. 
FIG. 4 illustrates an embodiment 10", constructed in accordance with the 
invention, which includes alternative means for controlling the 
down-conversion oscillator frequency. Instead of being converted to 
exactly zero frequency, the translated input signal is provided with a 
small offset from zero. The purpose of the frequency offset is to avoid 
the problems of DC offsets yet retain the precise frequency control and 
inherent demodulation capabilities of the phase lock loop, as in the 
receiver of FIG. 1. The two input signals to phase detector 90 are now the 
output signal from limiter 80 and a reference frequency 52 from oscillator 
51. Reference 52 differs from up-conversion frequency 37 by a frequency 
lower than the modulation frequencies to be received. The phase lock loop 
locks with the output of limiter 80 equal to the frequency of reference 
oscillator 51. For this to occur, the translated carrier frequency of the 
signals in the baseband paths of translating filter 30 must equal the 
difference between frequencies 37 and 52. This occurs when VCO 40 differs 
from input 31 by the same offset amount. This offset frequency may be set 
on the order of 10 to 100 Hz. It should be noted that there will be an 
image response, that is, that input 31 may be offset above or below the 
VCO. However, the separation between images will be much less than the 
channel spacings, and no interfering signals should be present. 
Because an unmodulated carrier at 31 produces no DC signals in the baseband 
paths, DC coupling and the need to maintain low offsets may be avoided. 
Any beat note that may arise because of imperfect matching will be below 
the lowest modulation frequency and may be blocked from the demodulated 
output signal by high-pass coupling techniques. 
In all other respects, the receiver of FIG. 4 is identical with that of 
FIG. 1 and is designed according to the same principles. 
Although the present invention has been disclosed in connection with the 
embodiments herein, it is understood that the novelty lies in the 
particular combination of translating bandpass filter, up-conversion to an 
intermediate frequency at which amplification and limiting are performed, 
and precise automatic control of the translation frequencies. 
Modifications and additional applications of the invention apparent to 
those skilled in the art are included within the scope of the invention.