Amplitude modulated telephone carrier system and terminal equipment therefor

A plural channel amplitude modulated frequency division multiplexed station carrier system wherein the receivers of the subscriber and central office channel terminal circuits are each equipped with a bandpass filter which is tuned to the carrier frequency to be received and with a circuit for synchronously detecting the carrier signal to which the bandpass filter is tuned. In the subscriber channel terminal circuits the bandpass filters are of the first order type, and in the central office channel terminal circuits the order of each bandpass filter is no greater than two. Carrier signal level adjustment is provided at the subscribers' end of the carrier system's transmission line for the purpose of causing all of the central office transmitted carrier signals to arrive at the subscriber channel terminal receivers at or at least approximately at a common level and for the additional purpose of causing all of the subscriber transmitted carrier signals to arrive at the central office channel terminal receivers at or at least approximately at a common level.

FIELD OF INVENTION 
This invention relates to improvements in amplitude modulated, frequency 
division multiplexed telephone carrier systems and is particularly 
concerned with, though not limited to, central office and subscriber 
terminal equipment for plural channel station carrier systems. 
The designations "station carrier" and "subscriber carrier" are used 
interchangeably herein to refer to those telephone carrier systems which 
are employed to establish communication between a central office or 
central station and the remote telephone stations of individual 
subscribers. 
Trunk carrier systems, on the other hand, are employed between central 
office exchanges (major switching points). The designations "telephone 
carrier" and "telephone carrier system" are used herein to refer to both 
trunk and station carrier systems. 
BACKGROUND OF INVENTION 
In a plural channel AM station carrier system a plurality of central office 
transmitting and receiving terminals (also referred to as central office 
channel terminal units or circuits) are connected by a single transmission 
line to a corresponding number of subscriber transmitting and receiving 
terminals (also called subscriber channel terminal units or circuits) 
which are located remotely from the central office. The central office and 
subscriber terminal units each have a transmitter equipped to transmit a 
carrier signal of pre-selected frequency and a receiver tuned to receive a 
carrier signal of a different pre-selected frequency. The carrier signals 
transmitted over the transmission line from the central office and 
subscriber channel terminal units are frequency division multiplexed. 
The receivers in the central office and subscriber channel terminal units 
each comprise some sort of frequency selective means for tuning it to 
receive just one of the incoming carrier frequencies so that each receiver 
receives a different pre-selected carrier frequency. In this manner each 
subscriber channel terminal unit is paired with a different central office 
channel terminal unit to make up a carrier derived circuit or two-way 
transmission channel. 
Prior to this invention it has been the general custom to use bandpass 
filters as the means for achieving carrier frequency selectively in the 
receivers of the central office and subscriber channel terminal units. It 
has also been the general practice to equip the receivers with envelope 
detectors for effecting carrier signal detection. 
In a typical telephone carrier system receiver the frequency selective 
bandpass filter (hereinafter called a channel or receive bandpass filter) 
is usually of the LC type and is tuned to a pre-selected carrier frequency 
so that it effectively rejects all but the desired carrier signal and its 
sidebands. The envelope detector is connected in the receiver to detect 
the received carrier signal, and the frequency components resulting from 
the detection are applied to a low pass filter which has an upper cutoff 
frequency of about 3000 Hz for separating the voice frequency detection 
components from the detected carrier frequency component and the other 
components having frequencies higher than the upper limit of 3000 Hz. In 
this way voice frequency signals present at the output of the detector 
will be passed by the low pass filter for transmission to a subscriber's 
telephone. 
Envelope detectors are advantageous because of their low cost, simplicity 
and reliability in operation. They do, however, have a major drawback in 
that they create a potential crosstalk problem. The crosstalk problem can 
develop because the envelope detector will produce the sum and difference 
frequency components for all combinations of the applied carrier and 
sideband frequencies. Since any practical channel bandpass filter design 
will not completely attenuate the unwanted carrier frequencies and their 
sidebands, they will be applid in partially attenuated form to the 
envelope detector. As a result, the side frequencies of each unwanted 
carrier frequency will be demodulated at the envelope detector as 
frequencies that are in the 300 Hz to 3000 Hz audio range. These 
demodulated voice frequency signal components from the disturbing channel 
are therefore within the passband of the receiver's low pass filter and 
will unavoidably be passed in unattenuated form by the low pass filter. 
They consequently represent disturbing crosstalk signals that can be heard 
at the subscriber's telephone. 
In the past it has been the general practice to reduce the power in these 
crosstalk signals to an acceptable level by achieving a more effective 
rejection of the unwanted carrier frequencies and their sidebands at the 
receiver's channel bandpass filer. This is done by increasing the order of 
or the number of pole pairs in the bandpass filter to establish a faster 
rolloff. 
In a carrier system using 8 kHz spacing between adjacent carrier 
frequencies for example, a three or four pole channel bandpass filter is 
normally required to reduce the crosstalk to a level that is acceptable 
under Rural Electrification Administration (REA) standards. For this 
reason the receive bandpass filters used in prior carrier system receivers 
are relatively expensive and represent a significant percentage of the 
cost of the signal receiving equipment. 
In an IEEE technical paper published in 1975 and entitled "FDM Subscriber 
Carrier: Expansion of Electronics In Telephone Plant Technology", 
synchronous detection of the desired incoming carrier signal was proposed 
in lieu of envelope detection for the cost-saving purpose of eliminating 
the channel bandpass filters in a multi-channel AM station carrier system. 
This IEEE paper discloses a carrier system receiver in which all of the 
incoming carrier signals are applied to a synchronous detector without any 
pre-detection bandpass filtering and in which a phase locked loop is used 
for generating the desired synchronous carrier signal frequency to drive 
the synchronous detector. 
Unlike an envelope detector, a synchronous detector produces difference 
frequency components which are only the difference between the zero beat 
frequency (i.e., the frequency of the desired carrier signal) and all of 
the other carrier and sideband frequencies that pass into the receiver to 
the input side of the detector. For a carrier frequency allocation having 
an 8 kHz spacing between adjacent carriers, the closest side frequency in 
each adjacent channel will synchronously demodulate as a 5 kHz signal 
component rather than a 3 kHz signal as is the case with an envelope 
detector. 
Of all the unwanted components of detection at the output of the 
synchronous detector, 5 kHz will be the lowest frequency from a 
neighboring channel for the 8 kHz carrier frequency spacing example given 
above. While this 5 kHz signal component is within a person's hearing 
range, it is above the 3000 Hz cutoff of the receiver's low pass filter 
and therefore can be attenuated sufficiently by making the filter's 
rolloff fast enough. 
Thus by using the synchronous detector to cause the unwanted signal 
components to have frequencies above the 3000 Hz cutoff and by making the 
low pass filter's rolloff fast enough, the crosstalk problem may be 
avoided. 
While advantageously eliminating the channel bandpass filters, the system 
described in the above-mentioned IEEE paper is subject to a number of 
troublesome conditions such as the tendency of the phase locked loop to 
fall out of lock during operation, the tendency of the phase locked loop 
to acquire lock with a signal of improper frequency, and the tendency of 
the phase locked loop to become unstable and break into oscillation under 
certain conditions. To eliminate these problems additional circuitry is 
required, thus offsetting to an extent the cost advantage derived from 
eliminating the bandpass filters. 
SUMMARY AND OBJECTS OF INVENTION 
The present invention avoids the use of expensive bandpass filters of 
relatively high orders and yet does not have the troublesome conditions or 
expensive circuitry associated with the synchronous detection arrangement 
disclosed in the IEEE paper mentioned above. Additionally, it keeps 
crosstalk down to an acceptable level and at the same time permits the use 
of a low pass filter that is more simplified and less expensive than the 
one that would normally be required for the receiver shown in the IEEE 
paper. 
These advantageous, cost-reducing features are achieved in the present 
invention by equipping the communications receiver with a low cost first 
or second order channel bandpass filter and by driving a synchronous 
detector in the receiver with an inexpensive zero crossover detector. The 
first or second order bandpass filter is more simplified and much less 
expensive than the third or fourth order bandpass filters which is 
required for envelope detectors. 
Although the first or second order filter does not attenuate the unwanted 
frequencies as much as the higher order bandpass filters, the combined 
operation of the low order channel bandpass filter and the synchronous 
detection in this invention has the surprising and unique effect of 
reducing the power in the VF crosstalk signals to or below an acceptable 
level without creating any troublesome conditions. This novel receiver 
circuit design additionally permits the use of an inexpensive single coil 
low pass filter. 
In the illustrated embodiment the foregoing features of the present 
invention are incorporated into, but are not limited to, a station carrier 
system in which the subscriber channel terminal units are connected by way 
of a four-wire subscriber group terminal unit to the subscribers' end of 
the transmission line and in which the central office channel terminal 
units are connected by way of a further four-wire group terminal (referred 
to as the central office group terminal unit) to the central office end of 
the transmission line. 
A group automatic gain and slope control or equalization circuit is 
advantageously included in the receive section of the subscriber group 
terminal. This group automatic gain and slope control circuit acts on the 
complete composite of incoming, central office-transmitted carrier signals 
in such a way that the incoming carrier signals are adjusted to or closely 
to a common pre-selected level regardless of the transmission line length 
between termination points. By adjusting the strengths of incoming 
carriers to a common level, no one carrier signal will be significantly 
stronger than the others to cause objectionable interference in the 
receivers of neighboring channels at the subscriber terminal equipment. 
In addition to the foregoing, the transmit section of the subscriber group 
terminal unit is advantageously equipped with an automatic carrier level 
coordination control circuit which adjusts the levels of the outgoing 
subscriber-transmitted carrier signals as a group. 
Carrier level coordination control involves the coordination of carrier 
levels between two or more carrier systems operating at corresponding 
frequencies and having their transmission lines in the same 
multi-conductor cable or in otherwise close proximity to each other. The 
carrier signal power levels in each system are so adjusted that at any 
point along the cable, carrier signals of like frequencies will be at 
least approximately at the same power level to minimize crosstalk due to 
power differentials. 
In the illustrated embodiment, the carrier level coordinating adjustment or 
control of the carrier signal power levels is automatically accomplished 
(a) by developing a single d.c. control signal which is a measure of 
transmission line length over which the carrier signals are to be 
transmitted and (b) by controlling the transmit power level of each of the 
carrier signals as a function of the above-mentioned d.c. control signal 
and also as a function of the carrier signal's own frequency. The d.c. 
control signal may be derived from the automatic gain and slope control 
circuit in the receive section of the subscriber group terminal unit. 
In addition to controlling the carrier signal levels to provide level 
coordination between two or more individual telephone carrier systems, the 
group automatic carrier coordination control circuit also advantageously 
adjusts the levels of the subscriber-transmitted carrier signals in each 
carrier system in such a way that they arrive at the central office or 
other termination point at or at least approximately at the same level. By 
this slope adjustment no one carrier signal arriving at the central office 
terminal equipment is significantly stronger than any of the other 
subscriber-transmitted carrier signals to cause objectionable interference 
in the receivers of neighboring channels at the central office terminal 
equipment. 
With the foregoing in mind a major object of this invention resides in the 
provision of a novel plural channel AM telephone carrier system whose cost 
is reduced without compromising performance. 
Another important object of this invention is to provide a novel carrier 
system communication receiver in which the filter and detection equipment 
is less expensive than prior equipment using a carrier-detecting envelope 
detector or a phase locked loop in a synchronous detection circuit. 
Other important objects of this invention are the provision of a novel 
telephone carrier system communication receiver in which: 
1. only a one or two pole pair channel bandpass filter is used in 
conjunction with a synchronous detector to keep crosstalk at an acceptable 
level in the voice frequency portion of the receiver; 
2. a zero crossover detector is used to supply the synchronous signal for 
driving the synchronous detector in the receiver; and 
3. a one-coil low pass filter is used to separate voice frequency signals 
from those components of detection having frequencies in excess of 
approximately 3000 Hz. 
Further objects of this invention will appear as the description proceeds 
in connection with the appended claims and the below-described drawings.

DETAILED DESCRIPTION 
Referring to FIG. 1, one embodiment of a plural channel AM station carrier 
system (indicated at 20 in the drawings) incorporating the principles of 
this invention is shown to comprise a single two-conductor transmission 
line 22, a selected number of central office transmitting and receiving 
channel terminal circuits or units, and a corresponding number of 
subscriber transmitting and receiving channel circuits or units. Any 
suitable number of central office and subscriber channel terminal circuits 
may be employed depending upon the number of channels desired. For 
example, eight central office channel terminal circuits and eight 
subscriber channel terminal circuits may be employed as shown to make up 
an eight-channel carrier system. The eight central office transmitting and 
receiving terminal circuits are indicated at COT1, COT2, COT3, COT4, COT5, 
COT6, COT7 and COT8, and the eight subscriber channel terminal circuits 
are indicated at STU1, STU2, STU3, STU4, STU5, STU6, STU7 and STU8. 
Since the central office terminal circuits COT1--COT8 are alike, only three 
(namely COT1, COT2 and COT8) are illustrated in any detail in FIG. 1. For 
the same reason only three of the subscriber terminal circuits, namely 
STU1, STU2 and STU8, are detailed in FIG. 1. 
As shown in FIG. 1, each of the central office terminal circuits COT1-COT8 
comprises a communication transmitter 24 for transmitting a carrier signal 
of pre-selected frequency and a communication receiver 26 tuned to receive 
a carrier signal from only a pre-selected one of the subscriber terminal 
circuits. Likewise, the subscriber terminal circuits STU1-STU8 each 
includes a communication receiver 28 tuned to receive a carrier signal 
from only a pre-selected one of the central office circuits and a 
transmitter 30 for transmitting a carrier signal of pre-selected 
frequency. 
Each of the subscriber terminal circuits STU1-STU8 is paired with a 
different one of the central office terminal circuits COT1-COT8 to provide 
eight two-way transmission channels. It is understood that each of these 
transmission channels has two different allocated frequency bands to 
provide service for a subscriber, one band being for transmission in one 
direction from the central office terminal equipment to the subscriber 
terminal equipment, and the other band being for transmission in the 
opposite direction from the subscriber terminal equipment to the central 
office terminal equipment. In this example, the subscriber terminal 
circuits STU1-STU8 are paired with the central office terminal circuits 
COT1-COT8, respectively. The transmission channels are designated by the 
reference characters CH1-CH8. The frequency spacing between adjacent 
carrier signals transmitted in either direction may be 8 kHz. 
The central office terminal circuits COT1-COT8 form a part of the central 
office terminal equipment and are located at a central office or central 
office station which is generally indicated at 33 in FIG. 1. The 
subscriber channel terminal circuits STU1-STU8 form a part of the 
subscriber terminal equipment and are located remotely from central office 
33 at the subscriber's end of the transmission line 22. 
The transmitters and receivers in each of the terminal circuits COT1-COT8 
are connected to line 22 by way of a central office group terminal unit or 
circuit 34 which is also located at the central office. The transmitters 
and receivers in each of the subscriber terminal circuits STU1-STU8 are 
connected to line 22 remotely from central office 33 by way of a 
subscriber group terminal unit or circuit 36. 
The subscriber channel terminal circuits STU1-STU8 are separately connected 
to the telephones (indicated at 40 in FIG. 1) of eight different 
subscribers by suitable means such as subscriber drops 42. At the central 
office, the central office terminal circuits COT1-COT8 are separately 
connected by central office drops 44 to appropriate terminals in the 
central office exchange equipment which is indicated at 46 in FIG. 1. 
As is customary in telephone carrier systems, each of the central office 
terminal circuits COT1-COT8 transmits at a pre-selected carrier frequency 
that is different from the transmission carrier frequencies allocated to 
the remaining central office channel terminal circuits and also different 
from the carrier frequencies that are transmitted up the transmission line 
22 in the opposite direction from the subscriber channel terminal circuits 
STU1-STU8. Likewise, the subscriber terminal circuits STU1-STU8 transmit 
at pre-selected carrier frequencies that are different from each other and 
different from the transmit frequencies assigned to the central office 
terminal circuits COT1-COT8. The allocation of different carrier 
frequencies for the carriers on transmission line 22 is referred to and 
designated as frequency division multiplexing (FDM). 
A typical FDM allocation scheme of carrier frequencies is shown in FIG. 1 
for the eight channels in the carrier system. According to this allocation 
scheme the carrier frequencies transmitted from the central channel 
terminal circuits COT1-COT8 are all contained in a frequency band (104 
kHz-160 kHz) that is higher than the band (8 kHz-64 kHz) containing the 
carrier frequencies that are transmitted from the subscriber channel 
terminal circuits STU1-STU8. 
As shown in FIG. 1, all of the central office terminal circuits COT1-COT8 
and unit 34 may advantageously be grouped together in a single central 
office terminal. Similarly, all of the subscriber terminal circuits 
STU1-STU8 and unit 36 may advantageously be grouped together in a single 
subscriber terminal. 
As shown in FIG. 1 the central office group terminal unit 34 is a four-wire 
circuit providing separate transmit and receive signal paths or sections 
80 and 82 which are coupled by a transformer 84 to the central office end 
of transmission line 22. The transmit signal path 80 feeds the central 
office-transmitted carrier signals from the transmitters of the central 
office channel terminal circuit COT1-COT8 to transmission line 22. The 
receive signal path feeds the arriving subscriber transmitted carrier 
signals from transmission line 22 to the receivers in the central office 
channel terminal circuits COT1-COT8. 
As shown in FIG. 2 the transmit section of group terminal unit 34 includes 
a transformer 85, an amplifier 86 and a high pass directional filter 88. 
The receive section of the central office group terminal unit includes a 
low pass directional filter 90, a fixed gain amplifier 92, an automatic 
gain control circuit 94 and an attenuator 95 or other variable gain 
circuit. 
All of the carrier signals transmitted from the transmitters 24 of the 
central office channel terminal circuits COT1-COT8 are summed into 
transformer 85 which couples the resulting composite to amplifier 86. 
Amplifier 86 amplifies the outgoing composite of the central 
office-transmitted carrier signals, and the amplified composite is fed to 
the high pass filter 88. Filter 88 passes the outgoing central 
office-transmitted carrier frequencies which are in the high frequency 
group (104 kHz-160 kHz) while rejecting the low group of 
subscriber-transmitted carrier frequencies (8 kHz-64 kHz) to keep the low 
group of carrier frequencies out of the transmit section of unit 34. From 
filter 88 the central office-transmitted carrier signals are coupled by 
transformer 84 to transmission line 22 for transmission to the remote 
subscriber terminal equipment. 
Filter 90, which is in the receive path of group terminal unit 34, is 
connected between transformer 84 and amplifier 92 and passes the low group 
of subscriber-transmitted carrier frequencies, while rejecting the high 
group of central office-transmitted carrier frequencies to keep the high 
group of carrier frequencies out of the receive section of unit 34. The 
subscriber-transmitted carrier frequencies passed by filter 90 are 
attenuated by attenuator 95 under the control of the AGC circuit 94 and 
are then amplified by amplifier 92. From the output of amplifier 92 the 
amplified carrier signals are fed to the receivers 26 in central office 
terminal circuits COT1-COT8 and also to the AGC circuit 94 in unit 34. As 
shown, attenuator 95 is connected to the signal path between filter 90 and 
amplifier 92. 
The group AGC circuit 94 provides a feedback loop and may be of any 
suitable circuit design for causing attenuator 95 to adjust the levels of 
the arriving carrier signals in such a way that the carrier signals are 
stabilized at a predetermined level despite variations in the levels in 
the incoming carrier signals at the input of filter 90. 
Similar to group terminal unit 34, unit 36 is a four-wire circuit providing 
separate transmit and receive signal paths or sections 98 and 100 which 
are coupled by a transformer 102 to the end of transmission line 22 remote 
from central office 33. The receive signal path in group terminal unit 36 
feeds the carrier signals arriving from the central office to the 
receivers in the subscriber channel terminal circuits STU1-STU8. The 
transmit signal path in unit 36 feeds the carrier signals from the 
transmitters in the subscriber channel terminal circuits STU1-STU8 to 
transmission line 22 for transmission up the line to the central office 
terminal equipment. 
As shown in FIG. 3 the receive section 100 of the subscriber group terminal 
unit 36 is equipped with a high pass directional filter 106, group 
aromatic gain control and slope equalization or level control circuit 108 
and a fixed gain amplifier 109. Circuit 108 comprises a pair of individual 
circuits, namely an AGC (automatic gain control) circuit 110 and a carrier 
level adjusting circuit 112. The level adjusting circuit 112 is under the 
control of the AGC circuit 110 which provides a feedback loop as shown. 
The transmit section of the subscriber group terminal unit 36 is equipped 
with a transformer 118, an automatic carrier level coordination control 
circuit 120, a fixed gain amplifier 122, and a low pass directional filter 
124. 
All of the carrier signals transmitted over transmission line 22 from the 
central office terminal equipment are coupled by transformer 102 to the 
high pass filter 106. Filter 106 passes this incoming composite of carrier 
signals, which are in the high frequency group of 104 kHz to 160 kHz, and 
rejects the low group of subscriber transmitted carrier frequency (8 
kHz-64 kHz) to keep the low group of carrier frequencies out of the 
receive section of unit 36. From filter 106 the incoming group of central 
office carrier signals are applied to the level adjusting circuit 112. 
As will be described in greater detail later on the level adjusting circuit 
112 is controlled by the AGC circuit 110 in such a way that it adjusts the 
amplitudes of all of the incoming carrier signals to or at least 
approximately to a common, pre-selected level for any transmission line 
length within the operating capacity of circuit 108. Following this level 
adjustment the incoming composite of carrier signals is amplified by 
amplifier 109. From amplifier 109 the composite of the incoming carrier 
signals is coupled to the receiver 28 in each of the subscriber terminal 
circuits STU1-STU8. Because of the carrier level adjustment by circuit 112 
and because amplifier 109 has a fixed gain, the incoming carrier signals 
will arrive at the receivers of the subscriber channel terminal circuits 
at or approximately at a common pre-selected level. 
On the transmit side of unit 36, all of the carrier signals transmitted 
from the subscriber channel terminal circuits STU1-STU8 are summed into 
transformer 118 which couples the resulting composite to the level 
adjusting automatic coordination control circuit 120. The automatic 
coordination control circuit 120 is also under the control of the AGC 
circuit 110 to adjust the levels of the outgoing signals in such a manner 
to establish inter-system carrier level coordination control and also to 
establish slope control to cause the outgoing carrier signals to arrive at 
the central office 33 or some other transmission line termination point 
(e.g., a repeater) at or at least approximately at a common, pre-selected 
level for any transmission line length up to some predetermined maximum 
value. 
Following level adjustment by circuit 120, the composite of the 
subscriber-transmitted carrier signals are amplified by amplifier 122 and 
are fed to filter 124. Filter 124 passes the subscriber-transmitted 
carrier signals, which are in the low group of carrier frequencies, and 
rejects the high group of central office-transmitted carrier frequencies 
to keep the high group of carrier frequencies out of the transmit section 
of unit 36. From filter 124 the subscriber-transmitted carrier signals are 
coupled by transformer 102 to transmission line 22 for transmission up the 
line to the central office terminal equipment. 
Since the central office channel terminal circuits COT1-COT8 are alike and 
since the subscriber channel terminal circuits STU1-STU8 are also alike, 
only the channel terminal circuits for channel 1 (namely, channel terminal 
circuits COT1 and STU1) will be described in greater detail. To this end, 
the transmitter of the central office terminal circuit COT1 comprises a 
compressor 50, a low pass filter 52 and a modulator 54 all connected in 
series in the manner shown in FIG. 2. The receiver in the central office 
channel terminal circuit COT1 is equipped with an LC channel bandpass 
filter 62, a synchronous detector 64, a low pass filter 66, a fixed gain 
amplifier 68, an expandor 70, a zero crossover detector 72, a hybrid 73 
and an automatic gain control circuit 74. In this embodiment the AGC 
circuit 74 controls an attenuator 75 or other variable gain circuit in the 
audio portion of receiver 26 to provide AGC action for the voice frequency 
signals that are passed by filter 66. 
The transmitter 30 and receiver 28 in the subscriber channel terminal 
circuit STU1 are the same as the construction thus far described for the 
transmitter and receiver in the central office terminal circuit COT1. To 
the extent that the terminal circuits STU1 and COT1 are the same, like 
reference numerals have been applied to designate the corresponding parts, 
except that the reference numerals applied to the parts of the subscriber 
terminal circuit STU1 have been suffixed by the letter "a" to distinguish 
them from the reference characters that are applied to designate the parts 
of the central office channel terminal circuit COT1. 
Voice frequency intelligence to be transmitted by way of channel 1 from the 
central office exchange equipment 46 to the subscriber's telephone 40 is 
fed by drop 44 and hybrid 73 to compressor 50 in terminal circuit COT1. 
Compressor 50 compresses the dynamic range of the voice signals in the 
usual manner. 
From compressor 50 the compressed voice frequency signals are fed through 
filter 52 to modulator 54 where they amplitude modulate a carrier 
frequency signal from an oscillator 128 to produce a double sideband 
amplitude modulated carrier signal. Being on channel 1, the frequency of 
the sinusoidal wave produced by oscillator 128 will be 88 kHz in 
accordance with the carrier frequency allocation scheme illustrated in 
FIG. 1. The circuit of filter 52 may be of any suitable design for 
rejecting frequencies above approximately 3000 Hz. Filter 52 therefore 
passes only voice frequency information up to 3000 Hz and serves to keep 
the carrier and other high frequencies out of compressor 50. By limiting 
the upper frequency of the VF modulating signal to 3 kHz due to the 
filtering action of filter 52, the upper and lower sidebands of the 
modulated carrier signal will therefore extend only to a maximum of 3 kHz 
from the carrier frequency. 
The modulated carrier signal supplied at the output of modulator 54 is fed 
through the transmit section 80 of the central office group terminal unit 
34 and is coupled to transmission line 22 for transmission down the line 
to the subscriber group terminal unit 36 along with the other carrier 
signals that are transmitted from the central office channel terminal 
circuits COT2-COT8. 
At the subscriber group terminal unit 36 the amplitude modulated 104 kHz 
carrier signal from the central office channel terminal circuit COT1 is 
fed along with the other central office-transmitted carriers through the 
receive section of terminal unit 36 to the channel bandpass filter 62a in 
the subscriber channel terminal circuit STU1 as well as to the 
corresponding channel bandpass filters in the remaining subscriber channel 
terminal circuits STU2-STU8. 
The channel bandpass filters in the subscriber terminal circuits STU1-STU8 
are tuned to the different carrier frequencies that are allocated to their 
associated central office channel terminal circuits COT1-COT8. Each 
channel bandpass filter therefore passes with the least attenuation the 
incoming carrier frequency (together with its sidebands) that is allocated 
to its associated central office channel terminal circuit. Thus, for this 
example, the channel bandpass filter 62a in the subscriber terminal 
circuit STU1 will pass the modulated 104 kHz carrier signal with the least 
attenuation, the 104 kHz carrier being the one transmitted from the 
central office channel terminal circuit COT1. In a similar fashion the 
channel bandpass filter (not shown) in subscriber channel terminal circuit 
STU2 will pass with least attenuation the modulated 112 kHz carrier 
frequency from central office channel terminal circuit COT2, and so on. 
Upon passing through filter 62a, the modulated 104 kHz carrier signal is 
applied to the synchronous detector 64a and also to the zero crossover 
detector 72a in terminal circuit STU1. The zero crossover detector 72a 
operates conventionally to detect or sense the zero crossovers of the 
received 104 kHz carrier signal to generate a local square wave 
synchronous signal having the same frequency as the received 104 kHz 
carrier signal. This square wave synchronous signal is applied to 
synchronous detector 64a to drive the synchronous detector and thereby 
cause the synchronous detection of the received 104 kHz carrier signal 
which is passed through filter 62a. To achieve satisfactory synchronous 
detection of the received carrier signal, the synchronous signal mentioned 
above is also required to be at least approximately in phase or 
approximately 180.degree. out of phase with the received carrier signal. 
Synchronous detection will be achieved even through there is as much as a 
10.degree. phase error (i.e., 0.degree..+-.10.degree. or 
180.degree..+-.10.degree.) or more between the synchronous signal and the 
received carrier signal. 
When the synchronous signal from the zero crossover detector 72a and the 
desired incoming carrier signal (in this case the 104 kHz signal) are 
combined at detector 64a, the resulting beat or different frequency will 
be zero, and the demodulated or detector output will contain an audio 
signal reproduction of the transmitted modulation (i.e., the VF signal 
intelligence which was applied to modulate the carrier at the central 
office channel terminal circuit COT1). 
For example, if a 1 kHz tone is applied to modulate the 104 kHz carrier at 
the central office channel terminal circuit COT1, the upper and lower side 
frequencies of the 104 kHz carrier will be 103 kHz and 105 kHz, 
respectively. Each of these side frequencies will demodulate as a 1 kHz 
signal at the synchronous detector 64a in the subscriber channel terminal 
circuit STU1, and the recovered 1 kHz tone will be passed by filter 66a 
into the audio section of the receiver. 
In this embodiment the maximum side frequency that can be present in each 
of the upper and lower sidebands will be spaced 3 kHz from the carrier 
frequency. This maximum side frequency will synchronously demodulate at a 
3 kHz signal at the output of detector 64a and will therefore be passed by 
filter 66a which has an upper cutoff of about 3000 Hz. The 3 kHz sideband 
maximum is typical of commercially available station carrier systems. 
The beat or difference frequency between the applied synchronous signal and 
each of the remaining unwanted carrier signals which are passed in 
attenuated form by filter 62a will be significantly above the 3000 Hz 
cutoff of low pass filter 66a. Likewise, the beat or difference frequency 
between the applied synchronous signal and each of the upper and lower 
side frequencies of these unwanted carrier signals from channels 2-8 will 
also be signficantly above the 3000 Hz cutoff of filter 66a. As a result, 
these undesired frequency components will be filtered off by filter 66a 
which is connected between the synchronous detector 64a and amplifier 68a 
as shown. 
Considering, for example, the numerical example of allocated carrier 
frequencies shown in FIG. 1, the adjacent 112 kHz carrier (on channel 2) 
will demodulate at detector 64a in STU1 as an 8 kHz signal and is 
consequently easily filtered out by filter 66a. The closest unwanted side 
frequency will demodulate only as a 5 kHz signal, being spaced 5 kHz from 
the received carrier frequency of 104 kHz. This 5 kHz component will also 
be rejected by filter 66a. Filter 66a thus separates the desired voice 
frequency intelligence from the other components of detection. 
The recovered VF signals which are passed by filter 66a are attenuated by 
attenuator 75a under the control of AGC circuit 74a and are then amplified 
by amplifier 68a. From amplifier 68a the VF signals are a.c. coupled to 
expandor 70a. Up to this point in the signal path, the dynamic range of 
the VF signals are still in their compressed state, having been compressed 
by the companion compressor 50 in the central office terminal circuit 
COT1. Expandor 70a operates to restore the VF signals to their original 
dynamic range. From expandor 70a the voice frequency signals are coupled 
to the receiver in telephone 40 by way of hybrid 73a and drop 42. 
As is well known a d.c. component is produced in the output voltage of the 
synchronous detector 64a by the synchronous detection of the received 
carrier signal. None of the other unwanted carrier signals from the other 
transmission channels is effective to develop this d.c. voltage component 
because of the frequency difference that exists between each of these 
unwanted carrier signals and the synchronous signal which is supplied by 
the zero crossover detector 72a. 
The d.c. component in the synchronous detector's output voltage therefore 
varies directly with the strength or peak amplitude of only the received 
104 kHz carrier signal in the subscriber channel terminal circuit STU1. 
This d.c. component will be fed along with the recovered voice frequency 
signals through filter 66a and amplifier 68a to the AGC circuit 74a. The 
AGC circuit 74a compares the d.c. component with a d.c. reference voltage 
to develop an AGC voltage which varies with the difference between the 
level-indicating d.c. component and the d.c. reference. The AGC voltage is 
applied to attenuator 75a to vary the extent of signal attenuation at the 
attenuator in order to stabilize the VF signal level at the output of 
amplifier 68a against input signal variations. In this manner AGC circuit 
74a serves to keep the output sound, which is applied to the receiver of 
telephone 40, at an essentially constant value despite variations that may 
occur in the strength of the incoming signal. For accomplishing the 
foregoing AGC action, AGC circuit 74a and attenuator 75a may be of any 
suitable design. 
The AGC action provided by the AGC circuit 74a in the subscriber channel 
terminal circuit ensures that the audio level will remain essentially 
constant even though there is or may be a day-to-day variation in the 
transmission line attenuation that is not fully compensated for by the 
group AGC circuit 108 in the subscriber group terminal unit 36. 
Considering now the transmission of intelligence from the subscriber 
terminal equipment to the central office terminal equipment, voice 
frequency intelligence signals originating from the subscriber's telephone 
40 on channel 1 are fed by way of hybrid 73a to compressor 50a. Compressor 
50a preforms the same function as compressor 50. 
From compressor 50a the compressed voice frequency signals are fed through 
filter 52a to modulator 54a where they modulate a carrier signal of 
pre-selected frequency from an oscillator 130 to develop a double sideband 
amplitude modulated carrier signal for transmission from the subscriber 
terminal circuit STU1. 
The purpose of filter 52a is the same as that described for filter 52. 
Being on channel 1 the frequency of the sinusoidal waveform produced by 
oscillator 130 will be 8 kHz in accordance with the carrier frequency 
allocation scheme illustrated in FIG. 1. 
The 8 kHz carrier signal transmitted from the subscriber terminal circuit 
STU1 and the carriers transmitted from the other subscriber terminal 
circuits STU2-STU8 are fed through the transmit section of the subscriber 
group terminal unit 36 to the transmission line 22 for transmission to the 
central office group terminal unit 34. 
At the central office group terminal unit 34 the amplitude modulated 8 kHz 
carrier signal from the subscriber terminal circuit STU1 is coupled along 
with the other subscriber-transmitted carriers through the receive section 
of terminal unit 34 to the channel bandpass filter 62 in the central 
office channel terminal circuit COT1 as well as the corresponding channel 
bandpass filters in the remaining central office channel terminal circuits 
COT2-COT8. 
Similar to the channel bandpass filters 62a in the subscriber channel 
terminal circuits, the channel bandpass filters in the central office 
terminal circuits COT1-COT8 are so designed that each filter is tuned to 
and hence passes with the least attenuation the transmit carrier frequency 
and associated sidebands coming in from its associated subscriber channel 
terminal circuit. Thus, for this example, the bandpass filter 62 in the 
central office channel terminal circuit COT1 will pass with the least 
attenuation the modulated 8 kHz carrier frequency from the subscriber 
terminal circuit STU1. The corresponding channel bandpass filter in the 
central office terminal circuit COT2 will pass with the least attenuation 
the modulated 16 kHz carrier frequency from the subscriber channel 
terminal circuit STU2, and so on. 
Upon passing through filter 62 the modulated 8 kHz carrier signal is 
applied to the synchronous detector 64 and also to the zero crossover 
detector 72 in the central office terminal circuit COT1. In the same 
manner described for the subscriber terminal circuit STU1, the zero 
crossover detector 72 operates to detect or sense the zero crossovers of 
the received 8 kHz carrier signal to generate a local square wave 
synchronous signal which is closely matched in phase and frequency with 
the received 8 kHz carrier signal. This square wave synchronous signal is 
applied to synchronous detector 64 to drive the synchronous detector and 
to thereby cause the synchronous detection of the received 8 kHz carrier 
signal which is passed through filter 62. 
This mode of synchronous detection is the same as that described for the 
subscriber terminal circuit STU1. Accordingly, the demodulated or detected 
output of detector 64 will contain an audio signal reproduction of the 
transmitted modulation (i.e., the VF signal intelligence which was applied 
to modulate the 8 kHz carrier at the subscriber channel terminal circuit 
STU1). 
Like the synchronous detection operation described for the subscriber 
channel terminal circuit STU1, the beat or difference frequency between 
the applied synchronous signal and each of the remaining unwanted carrier 
signals, which are passed in attenuated form by filter 62, will be 
significantly above the 3000 Hz cutoff of the low pass filter 66. 
Likewise, the beat or difference frequency between the applied synchronous 
signal and each of the upper and lower side frequencies of these unwanted 
carrier signals from channels 2-8 will also be significantly above the 
3000 Hz cutoff of filter 66. These undesired frequencies components will 
therefore be filtered off by filter 66. 
Considering the numerical example of the allocated carrier frequency shown 
in FIG. 1, the adjacent subscriber-transmit carrier frequency of 16 kHz on 
channel 2 will demodulate at detector 64 in COT1 as an 8 kHz signal and is 
consequently easily filtered out by filter 66. The closest unwanted side 
frequency will demodulate only as a 5 kHz signal, being spaced 5 kHz from 
the received carrier frequency of 8 kHz. This 5 kHz signal will also be 
rejected by filter 66. Filter 66 in performing the same function as filter 
66a separates the desired voice frequency intelligence from the other 
components above 3000 Hz. 
The VF signals passed by filter 66 are attenuated at attenuator 75 under 
the control of the AGC circuit 74 and are then amplified by amplifier 68. 
From amplifier 68 the VF signals are a.c. coupled to expandor 70 which 
restores the VF signals to their original dynamic range in the same manner 
as described for expandor 70a. From expandor 70 the voice frequency 
signals are coupled to the central office exchange equipment 46 by way of 
hybrid 73 and drop 44. 
The synchronous detection of the received 8 kHz carrier signal establishes 
a d.c. voltage component which varies only in accordance with the strength 
or peak amplitude of the 8 kHz carrier signal. This d.c. component is fed 
along with the recovered voice frequency signal components through 
amplifier 68 to the AGC circuit 74. The AGC circuit 74 employs this d.c. 
component to control the extent of signal attenuation at attenuator 75 in 
the same manner described for the AGC circuit 74a in the subscriber 
terminal circuit STU1. The VF signal level is therefore stabilized in the 
same manner described for the subscribier terminal circuit STU1. 
The operation of the remaining central office and subscriber channel 
terminal circuits COT2-COT8 and STU2-STU8 is the same as that just 
described for the terminal circuits COT1 and STU1. 
In accordance with this invention the bandpass filter 62a in the subscriber 
channel terminal circuit has less than 3 poles or pole pairs and is 
advantageously a single pole or single pole pair filter as shown in FIG. 
4. For purposes of this invention any suitable single pole bandpass filter 
can be employed. 
However, the channel bandpass filter used in this invention is 
advantageously a parallel, single pole, LC resonant circuit as illustrated 
in FIG. 4. This resonant filter circuit design is preferred because it has 
a symmetrical response to provide equal attenuation for corresponding 
upper and lower side frequencies above and below the center frequency to 
which the resonant circuit is tuned. It is understood, however, that other 
LC resonant circuit configurations can be used. Also, an active bandpass 
filter could be used instead of the illustrated passive filter 
configuration. 
Regarding the bandpass filter nomenclature, active and passive bandpass 
filters are commonly referred to as an n pole bandpass filter or an n pole 
pair bandpass filter and also, sometimes, as an n order bandpass filter, 
wherein n is some interger that is determined by the filter circuit 
design. All three of these bandpass filter designations are considered to 
be the equivalent of one another and may be used interchangeably to 
identify the same filter. Thus, filter 62a may be identified as a single 
pole pair bandpass filter, a one pole bandpass filter, or a first order 
bandpass filter. 
In this embodiment, the filter 62a is shown in FIG. 4 to comprise a step-up 
transformer 130 and a capacitor 132. A resistor 134 establishes the 
terminating resistance for the filter and has the effect of setting the Q 
of the filter's parallel resonant circuit along with the inductive 
reactance of the transformer's secondary coil 136. The single pole pair of 
filter 62a is established by capacitor 132 and the transformer's secondary 
coil 136. As shown, capacitor 132 is connected across coil 136 in the 
secondary of transformer 130. 
The values of capacitor 132 and coil 136 are selected so that the parallel 
resonant circuit is tuned to the desired incoming carrier frequency (e.g., 
88 kHz for subscriber terminal circuit STU1). This resonant circuit 
provides a single peak response curve typical of an ordinary tank circuit 
response. 
The signal source supplying the incoming carrier signal composite is across 
an input terminal 138 and ground so that the incoming carrier signal 
composite is coupled into the primary of transformer 130 through a 
resistor 140. The selected carrier signal at the resonant frequency is 
coupled by transformer 130 to the synchronous detector 64a and also to the 
zero crossover detector 72a. 
As shown, the transformer's secondary coil 136 is center tapped to earth 
ground at 142 to provide two induced carrier signal voltages that are 
180.degree. out of phase with each other, one being developed across the 
upper secondary terminal 143 and center tap 142, and the other being 
developed across the lower secondary terminal 145 and center tap 142. The 
secondary carrier signal voltage at terminal 143 is indicated at 147 in 
FIG. 6, and the other secondary carrier signal voltage at terminal 145 is 
indicated at 149 in FIG. 6. The two out-of-phase carrier signal voltages 
are used to provide for full wave rectification as will be explained in 
greater detail shortly. 
The synchronous detector 64a comprises at least one and preferably two 
switching devices which are cyclically switched on and off by the 
synchronous switching signal from the zero crossover detector 72a to 
detect the received carrier signal. Any suitable switching device may be 
used for this purpose. 
In the illustrated embodiment, for example, a pair of MOS analog switches 
144 and 146 are employed in detector 64a to provide a full wave 
rectification of the received carrier signal. Alternatively, field effect 
transistors may be used in place of the illustrated analog switches. 
Furthermore, only one switch may be used in place of the two shown to 
provide for the half wave rectification of the received carrier signal. 
Full wave rectification, however, is preferred and provides improved 
performance. 
As shown in FIG. 4, each of the analog switches 144 and 146 has an input 
electrode 148, and an output electrode 149 and a control electrode 150. 
The output electrodes of switches 144 and 146 are connected to a common 
junction 152 which feeds the low pass filter 66a by way of a resistor 154. 
Switch 144 is serially connected in the signal current path between the 
terminal 143 of transformer coil 136 and junction 152. Switch 146 is 
similarly serially connected in the signal current path between the 
secondary terminal 145 and junction 152 as shown. 
In the illustrated embodiment the zero crossover detector 72a comprises a 
pair of comparators 156 and 158. The outputs of comparators 156 and 158 
are respectively connected to the control electrodes of switches 144 and 
146 so that comparator 156 individually controls switch 144 and comparator 
158 individually controls switch 146. 
Still referring to FIG. 4, the non-inverting input terminal of comparator 
158 is connected to terminal 145 of the transformer secondary coil 136, 
and the inverting input terminal of comparator 158 is connected to 
terminal 143 of coil 136. The connections for the input terminals of 
comparator 156 to terminals 143 and 145 are reversed from that just 
described for comparator 158. 
By the foregoing circuit connections the received carrier signal is applied 
to both of the comparators 156 and 158. However, the carrier signal 
voltage at the input of comparator 156 will be 180.degree. out of phase 
with the carrier signal voltage at the input of comparator 158 due to the 
phase relationship of carrier signal voltages that are established at the 
transformer secondary terminals 143 and 145 by the grounded center tap 
142. 
Each of the comparators 156 and 158 is non-inverting. This means that the 
output of each comparator will have the same polarity as the signal 
voltage at the comparator's non-inverting input. As long as the carrier 
signal voltage at the non-inverting input for each of the comparators 156 
and 158 remains positive, the comparator's output voltage will be positive 
at some pre-selected value. When the carrier signal voltage at the 
comparator's non-inverting terminal crosses over zero volts and begins to 
swing negative, the comparator's output voltage switches sharply to a 
pre-selected negative value and remains at the negative value as long as 
the input signal voltage is negative. When the carrier signal input 
voltage begins to swing positive again, the comparator's output voltage 
switches back to its positive value. 
Each of the comparators 156 and 158 will therefore supply a square wave 
output voltage, the square wave output voltage for comparator 156 being 
indicated at 162 in FIG. 6, and the square wave output voltage for 
comparator 158 being indicated at 164 in FIG. 6. The combination of these 
square wave output signal voltages 162 and 164 represent the synchronous 
signal for driving the detector switches 144 and 146. 
In the illustrated embodiment each of the detector switches 144 and 146 
will be switched on to conduct current when the synchronizing switching 
signal voltage at its control electrode is positive at a predetermined 
value. Each of the switched 144 and 146 will turn off to interrupt the 
flow of current between its input and output electrodes when the switching 
voltage on its control electrode becomes negative. 
As a result, switch 144 will be alternately turned on and off by the 
switching signal voltage at the output of comparator 156. Likewise, switch 
146 will also be alternately turned on and off by the switching signal 
voltage at the output of comparator 158. 
As shown in FIG. 6, the switching signal voltage 162 at the output of 
comparator 156 will be matched in phase and frequency with the carrier 
signal voltage at the secondary terminal 143. This carrier signal voltage 
will therefore be synchronously detected and hence rectified by switch 144 
to provide the half wave rectified carrier signal voltage 166 (see FIG. 6) 
at the output electrode of switch 144. 
Similarly, the switching signal voltage 164 at the output of comparator 158 
will be matched in phase and frequency with the carrier signal voltage at 
the transformer secondary terminal 145. This carrier signal voltage will 
therefore be synchronously detected by switch 146 to provide the half wave 
rectified carrier signal voltage 168 (see FIG. 6) at the output electrode 
of switch 146. 
Since the switching signals 162 and 164 are 180.degree. out of phase with 
each other, switch 144 will be on when switch 146 is off and vise versa. 
The half-wave rectified voltages 166 and 168 will therefore be 180.degree. 
out of phase with each other. They consequently combine at junction 152 to 
produce the desired full wave rectified signal voltage as indicated at 170 
in FIG. 6. 
In addition to establishing full wave rectification the dual switch 
arrangement in the synchronous detector 64a is advantageous because it has 
the effect of cancelling objectionable transient spikes that may occur in 
complementary fashion in the course of turning switches 144 and 146 on and 
off. 
Because of the attenuation caused by the bandpass filter 62a and because of 
the frequencies that result from the synchronous demodulation of the 
incoming carrier, the low pass filter 66a may be of the single coil, third 
order elliptic type shown in FIG. 4 without creating noise problems. The 
single coil in filter 66a is indicated at 176 and is used for peaking in 
the 1 kHz to 3 kHz range to compensate for the attenuation that the side 
frequencies of the received carrier signal undergo in the single pole 
bandpass filter 62a. The peaking established by coil 176 complements the 
rolloff in the single pole bandpass filter 62a to make the overall VF 
frequency response of the receiver flat up to 300 Hz which is the cutoff 
for filter 66a. Filter 66a additionally smooths the rectified carrier 
signal pulses to provide the d.c. voltage component that is used by the 
AGC circuit 74a to establish the desired gain control of the audio 
signals. 
The attenuator 75a and the AGC circuit 74a may be of any suitable circuit 
design. The one shown in FIG. 4, however, is advantageous because of its 
low cost. 
In the illustrated example, attenuator 75a is an FET (field effect 
transistor) 179, and circuit 74a comprises a comparator 180 and a feedback 
capacitor 182 for filtering off the a.c. components, leaving the desired 
d.c. voltage component at the input of comparator 180 where it is compared 
against a fixed d.c. reference voltage. The output of comparator 180 
represents the AGC voltage and varies with the difference between the 
level-indicating d.c. voltage component and the fixed reference voltage. 
The AGC voltage at the output of comparator 180 is applied to the gate of 
FET 179 to control the FET's impedence between its drain and source 
terminals. As shown, FET 179 has its drain and source connected between 
the non-inverting input of amplifier 68a and earth ground to establish a 
variable resistance shunt across the amplifier's input for bypassing a 
variable portion of the incoming signal to ground. The variable 
drain-source resistance of FET 179 is determined by the magnitude of the 
AGC voltage at the output of comparator 180 and develops a voltage divider 
effect with the amplifier input resistor 184 which is connected in the 
signal path between filter 66a and the non-inverting input of amplifier 
68a. By controlling the drain-source resistance of FET 179, the AGC 
voltage controls the level of the signal voltage at the non-inverting 
input of amplifier 68a. 
In addition to the order of bandpass filter 62a, the sharpness of the 
filter's resonant response curve requires consideration. If the response 
curve is very sharp to achieve high selectivity, the side frequencies of 
the received carrier signal will be greatly attenuated by widely varying 
amounts, requiring complex and costly equalization circuitry for 
developing a flat VF response in the audio portion of the communication 
receiver. On the other hand, if the bandpass filter's response curve is 
not made sharp enough and is relatively broad, two other troublesome 
conditions occur. 
First, there will be insufficient attenuation of the carrier and side 
frequencies from the adjacent channels, thus leading to objectionably high 
levels of crosstalk in the audio portion of the receiver unless a much 
more expensive and complex low pass filter is used with a fast rolloff. It 
was also found that a relatively broad response curve causes phase jitter 
in the switching signal voltages at the outputs of comparators 156 and 
158. The phase jitter leads to two additional problems. 
First, it causes objectionable variations in the d.c. voltage component 
which is produced by the synchronous detection of the desired carrier 
signal and which is used in the AGC circuit 74a to obtain the AGC action 
in the VF portion of the receiver. If large enough, these variations in 
the d.c. voltage component have the effect of causing unacceptable AGC 
action in the VF portion of the receiver. 
Furthermore, it was found that the above-mentioned phase jitter in the 
comparators' switching signal voltages (indicated at 162 and 164 in FIG. 
6) causes a certain amount of the unwanted frequency components from 
adjacent channels to be synchronously detected. The synchronous detection 
of these unwanted frequencies creates in-band (i.e., 300 to 3000 Hz) 
crosstalk in the receiving channel. This crosstalk is particularly 
troublesome because it is within the low pass filter's passband and is 
therefore transmitted without attenuation to the subscriber's telephone. 
In the present invention the foregoing problem of inadequate selectivity 
and phase jitter are satisfactorily overcome by designing the bandpass 
filter's resonant circuit with selected LC values which cause the unwanted 
carrier signals in the adjacent channels to be attenuated to a level that 
is at least approximately 13 db below the level of the received carrier 
frequency to which the bandpass resonant circuit is tuned. 
To achieve satisfactorily flat VF response in the 300 Hz to 3000 Hz range 
with the single coil low pass filter 66a, it was found that the side 
frequencies of the received carrier signal,--particularly the upper and 
lower 3 kHz side frequencies--could not tolerate an attenuation of more 
than approximately 6 db at the bandpass filter (i.e., attenuation of the 
modulating intelligence signal by an amount not exceeding 6 db more than 
the received carrier frequency is attenuated. 
If the LC values in filter 62a are selected to attenuate the 3 kHz side 
frequencies by the maximum of 6 db, the carrier signals in the adjacent 
channels will be attenuated to a level that is about 15 db down from the 
level of the received carrier signal for the customary 8 kHz spacing 
between adjacent carrier frequencies. For satisfactorily overcoming all of 
the troublesome conditions mentioned above (i.e., the flatness of the 
response, the selectivity and the phase jitter) without resorting to a 
more complex low pass filter or to additional equalization circuitry, the 
sharpness of the bandpass filter's response curve can therefore be 
characterized as one that attenuates the adjacent carrier frequencies by 
an amount in the range extending from about 13 db to about 15 db. 
Preferably, the LC values of the resonant circuit in filter 62a are 
selected to attenuate the adjacent carrier signals (i.e., the carrier 
signals in the adjacent channels) to a level that is 15 db down from the 
level of the received carrier signal to which the resonant circuit is 
tuned. In this manner, the phase jitter is minimized to an extent that it 
does not present any real problem, and the bandpass filter's selectivity 
is maximized to achieve adequate attenuation of frequencies from adjacent 
channels without unduly attenuating the side frequencies of the received 
carrier signal. 
The combined effect of the first order bandpass filter 62a and the 
synchronous detector 64a of this invention has the unique effect of 
reducing the crosstalk power levels in the audio portion of the receiver 
to values that are significantly less than maximum levels permitted under 
REA (Rural Electrification Administration) specifications. In this regard, 
the unwanted frequency component normally presenting the greatest 
crosstalk problem from an adjacent channel will be the closest one to the 
desired carrier signal because it will be attenuated the least by the 
bandpass and low pass filtering in the receiving channel. The closest 
unwanted frequency component will be the 3 kHz side frequency in each 
adjacent channel and will be spaced 5 kHz from the received carrier signal 
for the customary 8 kHz frequency spacing between adjacent carrier 
signals. 
Considering reception of the 104 kHz carrier on channel 1 at the subscriber 
terminal circuit STU1, the closest unwanted frequency component will be at 
109 kHz. This 109 kHz frequency component passes through the channel 1 
bandpass filter 62a where its level is reduced or attenuated only by 
approximately 12 db for the preferred sharpness on the response curve. 
This amount of bandpass attenuation of the 109 kHz component would not be 
enough to keep crosstalk down to an acceptable level under REA standards 
if an envelope detector were used in place of the synchronous detector 
64a. 
In this invention, however, the synchronous detector 64a demodulates the 
109 kHz component as a 5 kHz component rather than a 3 kHz component as is 
the case with an envelope detector, and the unwanted 5 kHz component will 
undergo additional attenuation at low pass filter 66a, being significantly 
above the low pass filter's 3000 Hz cutoff. In the present invention, 
therefore, the unwanted 109 kHz component undergoes 12 db attenuation at 
the bandpass filter 62a and the resulting 5 kHz detection component 
undergoes an additional 30 db attenuation at the low pass filter 66a, 
bringing the aggregate bandpass and low pass filter attenuation to -42 db. 
When this -42 db loss is taken together with all of the other losses 
(which amount to about 53 db) the 5 kHz detection component at the output 
of expandor 70a, as well as all of the other crosstalk components, will be 
significantly below the maximum crosstalk power levels permitted under REA 
specifications, namely +10 BrnC (which is -80 db down from the desired 
received signal at odbm) for intelligable crosstalk and +20 dBrnC for 
non-intelligable crosstalk. 
With the possible exception of the channel bandpass filter configuration, 
the detailed circuit design for the receivers in the central office 
terminal circuits COT1-COT8 is advantageously the same as that just 
described for the subscriber terminal circuit STU1. Depending upon the 
particular design of the station carrier system, however, it may in some 
instances be desirable to make the central office bandpass filter a second 
order filter rather than a first order filter. 
For example, a second order channel bandpass filter may be preferred for 
the central office receiver equipment in a carrier system which provides 
subscriber-to-central office signalling by turning the 
subscriber-transmitted carrier signal on and off. According to this mode 
of signalling, the subscriber channel terminal circuit (STU1-STU8) will 
inhibit the transmission of the subscriber-to-central office carrier 
signal when the subscriber's telephone is on-hook and will effect 
transmission of the subscriber-to-central office carrier signal upon 
sensing the transfer of the subscriber's telephone to its off-hook state, 
thus signalling the central office that the subscriber's telephone has 
come off-hook. In such a signalling operation, the subscriber-to-central 
office carrier signal in one channel could busy one or both adjacent 
channels if it were not adequately attenuated by the bandpass filtering in 
the adjacent channels. To avoid this undesirable condition with a greater 
margin of safety, it is desirable to provide the central office channel 
bandpass filter 62 with an additional pole pair, thus making it a second 
order filter. One suitable second order configuration for filter 62 is 
shown in FIG. 5. 
As shown in FIG. 5, filter 62 is the same as filter 62a except that a 
second coil 190 and a second capacitor 192 have been added to define the 
filter's second pole. To the extent that filters 62 and 62a are alike, 
like reference numerals have been applied to designate like components, 
except that the reference characters used for filters 62 have been 
suffixed by the letter "b" to distinguish them from the reference numerals 
used for filter 62a. 
Coil 190 and capacitor 192 are connected in series in the primary of 
transformer 130b to form a series resonant circuit which is tuned to the 
carrier frequency to be received. 
The circuitry for the synchronous detector 64, the zero crossover detector 
72, the low pass filter 66, and AGC circuit 74 and the attenuator 75 are 
all the same as that shown in FIG. 4 and previously described for the 
subscriber channel terminal circuit. 
The AGC circuit 110 in the subscriber group terminal unit 36 may be of any 
suitable circuit design for developing an AGC current or voltage that 
varies as a function of the composite voltage of incoming carrier signals 
arriving from the central office terminal equipment and hence as a 
function of the length of transmission 22 between its terminal points. One 
example of a suitable design for circuit 110 is shown in FIG. 7 to 
comprise a half wave averaging detector 200 for rectifying the composite 
carrier signal voltage, a filter 202 for filtering the resulting half wave 
rectification out of detector 200 to produce a d.c. voltage which is 
indicative of the average value of the rectified signal, and a comparator 
204 for comparing the average voltage with a fixed d.c. reference 
potential to establish an AGC signal that varies with the difference 
between the above-mentioned d.c. voltage and the reference potential. 
As shown in FIG. 7, detector 200 includes an NPN transistor 206 whose gain 
is set by a collector resistor 210 and an emitter resistor 212. For 
biasing transistor 206, a diode 214 is connected between the junction of 
two resistors 216 and 218 and the negative terminal of a suitable d.c. 
voltage supply source 220. Resistors 216 and 218 are connected in series 
between the positive terminal of source 220 and the base of transistor 
206. Resistor 216 is relatively small, and diode 206 thus sets the base 
biasing voltage of transistor 206 at about +0.6 VDC so that in absence of 
an a.c. signal voltage at the base of transistor 206, the transistor is on 
the verge of turning on. 
The composite of the incoming carrier signals at the output of aomplifier 
109 is coupled by a capacitor 222 to the base of transistor 206. 
Transistor 206 will be driven into conduction only by the positive 
alternations of the applied carrier signal composite. The transistor's 
collector voltage will therefore be driven negatively on the positive 
alternations of the composite carrier signal to establish the half-wave 
rectification of the composite carrier signal. 
Filter 202 includes a capacitor 223 and an RC network comprising a resistor 
224 and an additional capacitor 226 for filtering the half wave rectified 
voltage. The smoothed-out d.c. voltage is indicative of the average value 
of the half wave rectified composite carrier signal and is applied to the 
positive input of comparator 204. Comparator 204 compares the d.c. average 
voltage with the fixed AGC reference which is at ground potential as 
indicated at 228. 
Still referring to FIG. 7, the level adjusting circuit 112 comprises two 
signal attenuators 230 and 231, the former being frequency independent and 
connected to comparator 204 to provide an AGC adjustment, and the latter 
being frequency dependent to establish a slope equalizing adjustment. In 
the illustrated embodiment attenuator 230 comprises two diode strings 232 
and 234 connected in series between the positive terminal of the voltage 
source 220 and the output of comparator 204. Each diode string has a 
plurality of diodes (e.g., four diodes in this embodiment) connected in 
series with each other. 
As shown, the adjacent terminals of diode strings 232 and 234 are 
interconnected by a common junction 236 at the carrier signal path between 
filter 106 and the fixed gain amplification circuitry in the receive 
section of unit 36. In the illustrated embodiment an additional fixed gain 
amplifier 109 may be connected serially between the output and amplifier 
238 to provide additional amplification of the composite carrier signal 
following level adjustment of the incoming carrier signals by the 
level-adjusting circuit 112. 
As shown, the terminals of diode strings 232 and 34 remote from the common 
junction 236 are connected to ground by relatively large capacitors 240 
and 242 and are therefore at a.c. ground. For a.c. signals diode strings 
232 and 234 are therefore effectively connected in parallel between 
junction 236 and a.c. ground for shunting a variable portion of the 
incoming composite carrier signal to ground. This a.c. signal circuit is 
such that the parallel combination of diode strings 232 and 234 cooperates 
with a resistor 244 to establish a voltage divider for dividing down the 
incoming carrier signal composite by an amount depending upon the 
current-dependent impedances of the diodes in strings 232 and 234. The 
a.c. signal output voltage from this voltage divider is developed at 
junction 236 and is coupled by a d.c. blocking capacitor 246 to the input 
of amplifier 238 for fixed gain amplification. 
If the variable impedance of the parallel combination of diode strings 232 
and 234 is represented by Z and the resistance of resistor 244 is 
represented by R then the a.c. divider output voltage will be proportional 
to Z divided by the sum of Z and R. In this way the a.c. voltage divider 
action established by resistor 244 and the two diode strings divides down 
the incoming carrier signal composite at the input end of resistor 244 to 
a value that is determined by the diode impedances in strings 232 and 234. 
In effect, the amount of carrier signal current shunted to ground through 
strings 232 and 234 and thus bypassing amplifier 238 will vary inversely 
with the current-dependent impedances of the diodes in strings 232 and 
234. 
The impedances of diode strings 232 and 234 are determined by the direct 
current drawn from the positive terminal of source 220. The amount of 
direct current drawn from source 220 in turn is determined by the extent 
to which the diodes in strings 232 and 234 are forward biased by the 
voltage at the output of comparator 204. 
When the carrier signal composite is applied to transistor 206 to initiate 
the AGC action, capacitor 226 starts charging toward ground potential from 
some positive value at its upper plate and hence at the positive input of 
comparator 204. When capacitor 226 pulls the positive input of comparator 
204 to a value approaching zero volts, the output voltage of comparator 
204 will begin to switch in a negative direction from the supply voltage 
(e.g., 6VDC) and will continue decreasing to a predetermined value that 
forward biases the diodes in strings 232 and 234. 
The diodes in strings 232 and 234 therefore conduct, and the amount of 
direct current drawn through the diodes will vary as a function of the 
voltage at the output of comparator 204. In particular, the amount of 
direct current conducted serially through the diode strings 232 and 234 
will increase as the voltage at the output of comparator 204 becomes more 
negative with respect to the positive supply voltage furnished by source 
220. 
The impedance of the diodes in strings 232 and 234 varies inversely with 
respect to the amount of direct current conducted through the diodes. 
Thus, the impedance of diode strings 232 and 234 decreases as the voltage 
at the output of comparator 204 becomes more negative with respect to the 
supply voltage 220. 
Due to the voltage divider effect of diode strings 232 and 234 with 
resistor 244, the level of the incoming carrier signal composite at 
junction 236 will therefore be decreased as the direct current through 
diode strings 232 and 234 is increased. The amount of direct current drawn 
through diode strings 232 and 234 will continue to increase under the 
control of the AGC circuit 110 until the voltage at the positive input of 
comparator 204 approaches the fixed reference potential which in this case 
is zero volts. 
The voltage at the positive input of comparator 204 will therefore continue 
to decrease until it reaches a value at which the AGC loop stablizes. When 
this happens, the amount of direct current drawn through the diode strings 
232 and 234 will stabilize at a predetermined value for a given input 
level of the carrier signal composite, and the level of the carrier signal 
composite at junction 236 will therefore stabilize at a predetermined 
value for all values of carrier composite input level to resistor 244. The 
action of AGC circuit 110 thus has the effect of continuously attempting 
to drive the voltage at the positive input of comparator 204 to a value 
equal or approaching the fixed AGC reference potential. In this way each 
of the incoming carrier signals in the composite waveform at junction 235 
will be adjusted by AGC circuit 110 to predetermined value which does not 
change regardless of the length of transmission line 22 up to some 
predetermined maximum length. 
The direct current that AGC circuit 110 draws through diode strings 232 and 
234 will vary proportionately with the level (peak amplitude) of the 
composite carrier signal that is applied to input of circuit 110 at the 
base of transistor 206. The extent of carrier signal attenuation by diode 
strings 232 and 234 will therefore vary with the composite carrier level 
at the input of circuit 110. Preferably, the impedances of diode strings 
232 and 234 are equal. 
In the illustrated embodiment, the central office-to-subscriber carrier 
signals will be transmitted continuously from the central office terminal 
circuits COT1-COT8 at a common level. This is a simplified way of 
coordinating the levels of central office transmitted carrier signal in 
the carrier system shown in FIG. 1 with the levels of central 
office-transmitted carrier signals in one or more other carrier systems 
having their transmission lines in the same cable containing line 22. 
Because of their different frequencies (i.e., 104 kHz to 160 kHz) these 
central office-transmitted carrier signals will be attenuated by different 
amounts by transmission over the transmission line 22 and will 
consequently arrive at the subscriber group terminal unit 36 at different 
levels, the higher frequency carrier signals being attenuated more than 
the lower frequency carrier signals as is well known. 
The diode string attenuator 230 alone will not provide correction for these 
differences in the levels of the carrier signals arriving from the central 
office. Thus, if no further corrective adjustments were made in the levels 
of the carrier signals arriving from the central office, the lower 
frequency carriers would be significantly stronger than the higher 
frequency carriers. This condition could be troublesome because of the use 
of the first order channel bandpass filters 62a in the subscriber terminal 
circuits STU1-STU8. In this regard the levels of the lower frequency 
carrier signal may not be attenuated sufficiently in the higher frequency 
channels (e.g., channel 8 at 160 kHz) to avoid a crosstalk problem in the 
subscriber receivers for the higher frequency channels. 
To avoid the occurrence of this potentially troublesome condition, the 
frequency dependent attenuator 231 cooperates with attenuator 230 to 
adjust the incoming carrier signals to or at least approximately to a 
common value to thereby provide correction for the cable slope (i.e., the 
differences in attenuation due to transmission of the central 
office-transmitted carrier signals over line 22). 
As shown in FIG. 7, the frequency dependent attenuator 231 advantageously 
comprises a parallel resonant circuit 250 and a series resonant circuit 
252. Circuits 250 and 252 are connected between junction 236 and ground. 
In the illustrated embodiment a single coil 254 is common to both the 
parallel resonant circuit 250 and the series resonant circuit 252. Thus, 
the parallel resonant circuit is defined by coil 254, a capacitor 256 and 
a shunt resistor 258 all connected in parallel. The series resonant 
circuit is defined by coil 254, a further capacitor 260 and a resistor 262 
all connected in series as shown. 
Attenuator 231 cooperates with resistor 244 to establish a voltage divider 
for dividing down each of the incoming carrier signals as a function of 
its own carrier frequency. The parallel resonant circuit 250 is preferably 
tuned to a frequency somewhat higher than the highest carrier frequency 
(144 kHz) transmitted down transmission line 22 from the central office 
channel terminal equipment. The series resonant circuit 252, on the other 
hand, is preferably tuned to a frequency somewhat lower than the lowest 
carrier frequency (88 kHz) that is transmitted down line 22 from the 
central office terminal equipment. 
The response of circuit 252 is in the form of a notch at resonance since 
circuit 252 is in parallel with the input carrier signal source. On the 
other hand, the response of the parallel resonant circuit 250 is in the 
form of a peak. With the selection of the series and parallel resonant 
frequencies as indicated above to bracket the central office-transmitted 
carrier frequencies, the central office-transmitted carrier frequencies 
will all fall on one side or skirt of the overall frequency response curve 
between the parallel resonant peak and the series resonant notch and 
preferably in the overall response curve region having a substantially 
uniform slope. 
As is known, the frequency-dependent impedance of the parallel resonant 
circuit 250 increases to maximum at the parallel circuit's resonant 
frequency, while the frequency-dependent impedance of the series resonant 
circuit 242 decreases to a minimum at the series circuit's resonant 
frequency. Thus, the overall frequency-dependent impedance of attenuator 
231 varies with frequency and increases from a low value at the lowest of 
the incoming carrier frequencies to a relatively high value at the highest 
incoming carrier frequencies. The higher carrier frequencies will 
therefore be attenuated less than the lower carrier frequencies to 
compensate for the frequency-dependent attenuation that the carrier 
signals undergo by transmission over transmission line 22. 
It will be noted that the parallel a.c. circuit combination of diode 
strings 232 and 234 acts as a shunt resistance across the parallel 
resonant circuit 240 to load the parallel resonant circuit. Variations in 
the shunt resistance established by diode strings 232 and 234 thus varies 
the circuit Q of the parallel resonant circuit. As this shunt resistance 
decreases, the Q of parallel resonant circuit 250 decreases to make the 
parallel resonant response curve broader (i.e., less sharp). Conversely, 
the circuit Q will increase and the response curve will become sharper as 
the shunt resistance increases. 
Thus, the change in the shunt resistance will have the effect of changing 
the slope of the resonant response curve in the frequency range covering 
the central office-transmitted frequencies. 
For relatively short transmission line lengths, the shunt resistance 
established by diode strings 232 and 234 will be decreased. The slope of 
the resonant response curve's skirt will therefore decrease in the 
frequency band containing the incoming carrier frequencies, thus reducing 
the differences in attenuation between the different carrier frequencies. 
In other words, the difference in attenuation between the highest and 
lowest of the incoming carrier frequencies will be decreased to account 
for the decreased cable slope. Attenuator 231 therefore will have less 
slope correcting effect upon the incoming carriers as the transmission 
line is made shorter. 
Conversely, the diode string shunt resistance will be increased for longer 
transmission line lengths, thus increasing the slope of the overall 
response in the frequency band covering the central office transmitted 
carrier frequencies. As a result, the differences in attenuation for the 
different carrier frequencies will increase to compensate for the 
increased cable slope. 
In this manner attenuator 231 and the diode string shunt resistance combine 
to have the effect of attenuating all of the incoming carrier signals to 
or approximately to a common level regardless of the length of the 
transmission line up to a predetermined maximum limit. The level adjusting 
circuit 112 therefore provides the desired slope correction to achieve a 
flat carrier frequency response for any transmission line length within 
the circuit's operating capability. 
As a result, all of the carrier signals arriving at the bandpass filters 
62a in the subscriber terminal circuits STU1-STU8 will be at or 
approximately at a common level. Thus, no one carrier signal will be 
significantly stronger than another to cause objectionable interference in 
the receivers of neighboring channels. 
The automatic coordination control circuit 120 is used to minimize 
inter-system crosstalk or interference between carrier signals of like 
frequencies in two or more carrier systems which have their transmission 
lines in the same cable or in otherwise close proximity to each other. In 
this regard it is common practice to connect two or more carrier systems 
to the same central office and to place the separate transmission lines 
for the carrier systems in the same cable so that portions of the 
transmission lines extend coextensively from the central office. 
The lengths of these transmission lines, rather than being equal, are quite 
often different, with one line extending a greater distance away from the 
central office than another. If the power levels for carrier signals of 
like frequencies on the separate transmission lines are not adjusted to be 
approximately equal at any given point along the coextensive portions of 
the lines in the common cable, objectionable crosstalk will occur between 
the carrier systems. An example of a plural carrier system facility or 
installation in which inter-system crosstalk can occur is shown in FIG. 8. 
In FIG. 8, the previously described carrier system 20 and one additional 
station carrier system 20' are shown to be both connected to the same 
central office. Carrier system 20' for purposes of this example may be of 
the same design as carrier system 20. Accordingly, like reference 
characters have been applied to designate corresponding parts of systems 
20 and 20' except that the reference characters for system 20' have been 
primed to distinguish them from the reference numerals used for 
identifying the parts of system 20. 
As shown in FIG. 8, the transmission lines 22 and 22' of the two carrier 
systems having portions extending coextensively from the central office in 
a common cable 270. Transmission line 22' is longer than transmission 
lines 22 so that the cable slope for line 22' will be greater than that 
for line 22. Assuming the same carrier frequency allocation scheme for 
both systems, the carrier frequencies transmitted over line 22 will 
correspond to the carrier frequencies that are transmitted over line 22'. 
The inter-system crosstalk problem for the central office-transmitted 
carrier signals is avoided by simply transmitting all of the central 
office carrier signals from the central office terminal equipment in both 
systems at a common level. The automatic coordination control circuit 120 
in system 20 and its counterpart circuit 120' (not shown) in system 20' 
are used to avoid the crosstalk problem for the carrier signals that are 
transmitted in the opposite direction from the subscriber terminal 
equipment to the central office. This is done by coordinating the 
adjustment of the subscriber transmit carrier signal levels in such a way 
that regardless of any difference between the lengths of transmission 
lines 22 and 22' the power of carrier signals of like frequencies on the 
different transmission lines will be at least approximately the same at 
any point along the complete cable or the coextensive portions of the 
transmission lines. 
The expression "automatic carrier level coordination control" is therefore 
understood to mean that at any point along coextensive portions of two or 
more separate transmission paths or transmission lines for different 
carrier systems in a common installation, the power levels of the carrier 
signals of common or like frequency will automatically be at least 
approximately equal, regardless of any differences in the lengths of the 
transmission lines in the installation and for any transmission line 
lengths within the operating range of the automatic coordination control 
system or circuitry. 
To automatically accomplish this adjustment, the automatic coordination 
control circuit 120 is shown in the illustrated embodiment to comprise two 
carrier signal attenuations 272 and 274. The circuit configuration of 
attenuators 272 and 274 are advantageously the same as attenuators 230 and 
231, respectively. Accordingly, like reference numerals have been applied 
to designate corresponding components of attenuators 272 and 230 except 
that the reference numerals applied to attenuator 272 have been suffixed 
with the letter a to distinguish them from the reference numerals used for 
attenuator 230. Likewise, like reference numerals have been used to 
designate corresponding components of attenuators 274 and 231 except that 
the reference characters for attenuator 274 have been suffixed by the 
letter a to distinguish them from the reference numerals used for 
attenuator 231. 
As described in detail below, attenuator 272 is a frequency independent 
network and adjusts the level of the outgoing subscriber carrier signal 
composite as a function of the length of the transmission line 22 between 
its termination points. Attenuator 274, on the other hand, is frequency 
dependent and makes a slope adjustment to compensate for the different 
attenuations that the subscriber carrier signals will undergo due to their 
different frequencies. 
As shown in FIG. 7, the diode strings 232a and 234a in attenuator 272 are 
connected in series between the output of comparator 204 and the positive 
terminal of the d.c. voltage source 220. The common junction between diode 
strings 232a and 234a is indicated at 236a and is connected in the carrier 
signal path between transformer 118 and the fixed gain amplifier circuitry 
which may include further fixed gain amplifier 281 in addition to 
amplifier 122. 
Similar to the circuit design and operation described for diode strings 232 
and 234, diode strings 232a and 234a are effectively in parallel for a.c. 
signals and shunt a variable portion of the outgoing subscriber carrier 
signal composite to ground. Resistor 244a and the a.c. parallel 
combination established by diode strings 232a and 234a form a voltage 
divider 291 which acts in the samer manner as the voltage divider that is 
formed by resistor 244 and diode strings 232 and 234. 
The output voltage from divider 291 is developed at junction 236a, is 
coupled to amplifier 122 by a d.c. blocking capacitor 292 and will vary 
proportionately with the impedance of the diodes in strings 232a and 234a. 
The impedance of diode strings 232a and 234a will, in turn, vary inversely 
with the amount of direct current conducted through the diode strings, and 
the amount of direct current conducted through the diode strings will be 
varied by the voltage at the output of comparator 204. 
Since the voltage at the output of comparator 204 varies proportionately 
with the length of transmission line 22 and is thereby a measure of the 
transmission line length, the direct current drawn through diode strings 
232a and 234a will also vary proportionately with the transmission line 
length. The impedance of diode strings 232a and 234a therefore vary 
inversely with respect to the length of transmission line 22. Thus, the 
longer the transmission line becomes, the greater the composite carrier 
signal voltage at the output of voltage divider 291 and vice versa. The 
composite carrier signal voltage at the output of attenuator 230 will 
therefore vary with the transmission line length, there being less 
attenuation for long transmission lines and more attenuation for short 
lines. 
The parallel resonant circuit 250a is preferably tuned to a frequency 
somewhat higher than the highest subscriber carrier frequency (64 kHz) 
that is transmitted over transmission line 22 to the central office 
terminal equipment. The series resonant circuit 252a is preferably tuned 
to a frequency somewhat lower than the lowest subscriber carrier frequency 
(8 kHz). 
Like the parallel and series resonant circuits in attenuator 231, the 
response of the series resonant circuit 252a is in the form of a notch at 
resonance, while the resonant response of the parallel resonant circuit 
250a is in the form of a peak. The band of subscriber carrier frequencies 
(8 kHz to 64 kHz) therefore lies on one side or skirt of the overall 
frequency response curve between the parallel resonant peak and the series 
resonant notch and preferably in the response curve region having a 
substantially uniform slope. 
The frequency-dependent impedance of the parallel resonant circuit 250a 
increases to maximum at the parallel circuit's resonant frequency, while 
the frequency-dependent impedance of the series resonant circuit 252a 
decreases to a minimum at the series circuit's resonant frequency. The 
overall frequency-dependent impedance of attenuator 274 in the band of the 
subscriber carrier frequencies thus increases from a low value at the 
lowest of the subscriber carrier frequencies (8 kHz) to a relatively high 
value at the highest subscriber carrier frequencies (64 kHz). The higher 
subscriber carrier frequencies will therefore be attenuated less than the 
lower subscriber carrier frequencies to provide the slope correction that 
compensates for the frequency-dependent attenuation that the carrier 
signals undergo by transmission over transmission line 22. 
Similar to the relationship of diode strings 232 and 234 to the resonant 
circuits 250 and 252, the parallel a.c. circuit combination of diode 
strings 232a and 234a acts as a shunt resistance across the parallel 
resonant circuit 250a to load the parallel resonant circuit. Variations in 
the shunt resistance established by diode strings 232a and 234a thus 
varies the circuit Q of the parallel resonant circuit. As thus shunt 
resistance decreases, the Q of parallel resonant circuit 250a decreases to 
decrease the sharpness of the parallel resonant response curve. 
Conversely, the circuit Q will increase and the response curve becomes 
sharper as the shunt resistance increases. 
Thus, a change in the shunt resistance will have the effect of changing the 
slope of the overall resonant response curve in the frequency range 
covering the subscriber carrier frequencies. 
For short transmission line lengths, the shunt resistance established by 
diode strings 232a and 234a will be decreased. The slope of the resonant 
response curve's skirt will therefore decrease in the frequency band 
containing the subscriber carrier frequencies, thus reducing the 
differences in attenuation between the different carrier frequencies. In 
other words, the difference in attenuation between the highest and lowest 
of the subscriber carrier frequencies will be decreased to account for the 
decreased cable slope. Attenuator 231a therefore will have less slope 
correcting effect upon the outgoing subscriber carriers as the 
transmission line is made shorter. 
Conversely, the diode string shunt resistance will be increased for longer 
transmission line lengths, thus increasing the slope of the overall 
response in the frequency band covering the outgoing subscriber carrier 
frequencies. As a result, the differences in attenuation for the different 
subscriber carrier frequencies will increase to compensate for the 
increased cable slope. 
In this manner attenuator 231a and the diode strings 232a and 234a combine 
to have the effect of attenuating each of the outgoing subscriber carrier 
signals as a function of its own carrier frequency and as a function of 
the length of transmission 22. 
As previously explained the subscriber-transmitted carrier signals are in a 
frequency band that is lower than the frequency band containing the 
central office-transmission carrier signals. The subscriber-transmitted 
carrier signals therefore undergo less attenuation than the central office 
carrier signals. Because of this attenuation difference it is desirable to 
attenuate the outgoing subscriber carrier signal composite by an amount 
that is less than the attenuation that the incoming central carrier signal 
composite undergoes at attenuator 230. This difference in attenuation is 
achieved mainly by resistors 294 and 296. 
As shown in FIG. 7, resistor 294 is connected in series with diode strings 
232 and 234 between the circuit point or junction 298 and the output of 
comparator 204. Resistor 296, on the other hand, is connected in series 
with diode strings 232a and 234a between the circuit junction 300 and the 
output of comparator 204. 
Because the voltage drop across the forward biased diodes in strings 232, 
234, 232a and 234a will not be varied to any significant degree by changes 
in the amount of current flowing through the diodes strings, junctions 298 
and 300 each act or appear as a voltage source. Likewise, the output of 
comparator 204 also appears as a voltage source. It also will be noted 
that the sum of the direct current conducted through attenuator 230 and 
the direct current conducted through attenuator 272 is equal to the output 
current of comparator 204. 
Because of this and because the output of comparator 204 and circuit points 
298 and 300 appear as voltage sources, the ratio of I.sub.1 /I.sub.2 will 
be closely equal to the ratio of R.sub.2 /R.sub.1 where I.sub.1 is the 
amount of direct current conducted through diode strings 232a and 234a, 
I.sub.2 is the amount of direct current conducted through diode strings 
232 and 234, R.sub.1 is resistance of resistor 300, and R.sub.2 is the 
resistance of resistor 298. 
Thus the ratio of I.sub.1 /I.sub.2 may be set at a selected value less than 
unity by selecting the ratio of resistors 298 and 300 to compensate for 
the difference in overall attenuation between the high band of central 
office carrier frequencies and the low band of subscriber carrier 
frequencies. 
From the foregoing discussion it will be appreciated that the coordination 
control circuit 120 adjusts the levels of the subscriber transmit carrier 
signals as a group under the control of the AGC circuit 110. The levels of 
each subscriber carrier signal is adjusted by circuit 120 as a function of 
its own carrier frequency and as a function of the transmission line 
length which may and usually does vary for different carrier system 
installations. Each subscriber transmit carrier signal will therefore be 
adjusted in accordance with the attenuation that it undergoes at its 
transmit frequency upon transmission over line 22. In this manner the 
subscriber carrier signals will arrive at the central office group 
terminal unit 34 or some other transmission line terminating point at 
pre-selected levels which remain the same regardless of the length of 
transmission line 22. 
In the illustrated embodiment the preferred slope adjustment of the 
subscriber transmit carrier signals at circuit 120 causes all of the 
subscriber carrier signals to arrive at the central office group terminal 
unit 34 or other transmission line termination at or at least 
approximately at the same pre-selected level. As a result, no one of the 
subscriber transmit carrier signals arriving at the central office 
terminal equipment will be significantly stronger than any other 
subscriber carrier signal to cause objectionable interference in the 
receivers of neighboring channels at the central office channel terminal 
equipment. 
When automatic carrier level coordination control is achieved by adjustment 
of the subscriber carrier transmit levels, it will be appreciated that at 
any given distance measured along transmission line 22 from the central 
office or other remote termination, the level of each subscriber transmit 
carrier signal remains at least approximately the same regardless of the 
transmission line length. 
By equipping the subscriber group terminal 36' in carrier system 20' with 
the automatic coordination control circuitry (120' and 110') corresponding 
to the coordination control and AGC circuits 120 and 110 and by using 
equal AGC reference voltages in the two AGC circuits 110 and 110', then 
each pair of subscriber transmit carrier signals of the same frequency 
(one on line 22 and the other on line 22') will be at or at least 
approximately at the same level at any point along the coextensive 
portions of transmission lines 22 and 22' in cable 270. 
The AGC attenuator 95 at the central office group terminal unit 34 does not 
perform any slope equalization function and is not required to adjust the 
strengths of the incoming carriers to a common level because this is 
effectively accomplished beforehand by the automatic coordination control 
circuit 120 at the subscriber group terminal 36. 
The central office terminal circuits COT1-COT8 and the central office group 
terminal unit 34 may be centralized in a single card cage or card shelf at 
the central office and may be mounted on one or more circuit boards. 
Additionally, the subscriber terminal circuits STU1-STU8 and the subscriber 
group terminal unit 36 may be mounted on one or more circuit boards which 
are housed in a suitable card cage or card shelf to provide a single 
centralized subscriber terminal. With such an arrangement, subscriber 
drops will extend out from the subscriber terminal units to the telephone 
40 wherever the telephones may be located. 
If more than eight channels are needed at the single centralized subscriber 
terminal mentioned above, the carrier installation may be expanded by 
adding the required number of carrier systems. For example, if a 48 
channel installation is desired then six carrier systems corresponding to 
carrier system 20 may be arranged together, and the subscriber channel 
terminal circuits and the subscriber group terminal units for all six 
carrier systems may be grouped together at a single centralized subscriber 
terminal. The central office channel terminal circuits and the central 
office group terminal units for the six carrier systems also may be 
grouped together at a single centralized terminal in the central office. 
The use of the term "unit" in referring to the central office and 
subscriber group terminal units is considered to be the equivalent of and 
interchangeable with the term "circuit". 
The invention may be embodied in other specific forms without departing 
from the spirit or essential characteristics thereof. The present 
embodiments are therefore to be considered in all respects as illustrative 
and not restrictive, the scope of the invention being indicated by the 
appended claims rather than by the foregoing description, and all changes 
which come within the meaning and range of equivalency of the claims are 
therefore intended to be embraced therein.