Sensorless motor driving device

In a sensorless motor driving device for driving a motor by controlling the energization of the coils of individual phases of the motor according to the result of detecting the position of the rotor of the motor without using an external sensor, before the motor is started, which phase to energize first when the motor is started is determined according to the result of detecting the position in which the rotor is at rest by exploiting the fact that the coils of the individual phases have varying apparent inductances according to the position in which the rotor is at rest. This circuit configuration permits the motor to be started always in the same rotation direction, and thus prevents reverse rotation of the motor.

BACKGROUND OF THE INVENTION

Field of the Invention

The present invention relates to a sensorless motor driving device for driving motor by detecting the position of the rotor of the motor without using an external sensor such as a Hall element and controlling the energization of the coils of individual phases of the motor according to the result of the detection.

First, taking up a two-phase half-wave motor driving device as an example, prior art will be described.FIG. 15shows a block diagram of a conventional, common two-phase half-wave motor driving device100′. A Hall element H is arranged so as to face a rotor of a motor M. A comparator101outputs a binary signal that represents the relationship in magnitude between the voltages output from both ends of the Hall element H. A commutation portion102, according to the binary signal output from the comparator portion101, devices which of two transistors T1and T2, which together constitute an output portion104, to turn on, and outputs a logic signal to be fed to the gates of the transistors T1and T2.

The signal output from the commutation portion102is converted, by a pre-drive portion103, to a level high enough to turn on and off the transistors T1and T2, and is then fed to the gates of the transistors T1and T2. In the output portion104, an n-channel MOS field-effect transistor T1is connected between one end of a first-phase coil L1, of which the other end is connected to a drive voltage VMfor the motor M, and ground. Moreover, an n-channel MOS field-effect transistor T2is connected between one end of a second-phase coil L2, of which the other end is connected to the drive voltage VMfor the motor M, and ground.

The signal output from the Hall element H represents the position of the rotor. Thus, with the circuit configuration described above, it is possible to switch which phase to energize with appropriate timing according to the position of the rotor, and thereby rotate the rotor smoothly.

The problem with this conventional, common two-phase half-wave motor driving device is that it requires an external sensor (Hall element) to detect the position of the rotor. This has been hindering cost reduction and miniaturization. The inventor of the present application has once proposed, in another application, a sensorless driving method for driving a motor by detecting the position of the rotor of the motor according to a back electromotive force appearing in the coil of each phase of the motor as the rotor rotates and switching which phase to energize according to the result of the detection. However, in this sensorless driving method, when the rotor is at rest, no back electromotive force appears in the coil of each phase, and therefore the position of the rotor cannot be detected. Thus, simply applying a drive signal to the motor when it is started may cause, quite inconveniently, the motor to start rotating in the reverse direction.

SUMMARY OF THE INVENTION

An object of the present invention is to provide a sensorless motor driving device that prevents reverse rotation of a motor.

To achieve the above object, according to the present invention, in a sensorless motor driving device for driving a motor by controlling the energization of the coils of individual phases of the motor according to the result of detecting the position of the rotor of the motor without using an external sensor, before the motor is started, which phase to energize first when the motor is started is determined according to the result of detecting the position in which the rotor is at rest by exploiting the fact that the coils of the individual phases have varying apparent inductances according to the position in which the rotor is at rest.

By determining, in this way, which phase to energize first when the motor is started according to the position in which the rotor is at rest, it is possible to start the motor always in the same rotation direction and thereby prevent reverse rotation of the motor.

Here, the coils of the individual phases have varying apparent inductances according to the position in which the rotor is at rest, and therefore the waveforms with which currents start flowing through the coils of the individual phases when they start being energized vary from phase to phase. Thus, for example, by energizing the coils of the individual phases to such a degree as not to cause the rotor to start rotating and then comparing the waveforms with which currents start flowing through those coils, it is possible to detect the position in which the rotor is at rest.

DESCRIPTION OF THE PREFERRED EMBODIMENTS

Hereinafter, embodiments of the present invention will be described with reference to the drawings.FIG. 1is a block diagram of a two-phase half-wave fan motor driving device100embodying the invention which is designed for the driving of a two-phase half-wave motor used as a fan motor. In this figure, reference numerals1-1and1-2represent respectively a first and a second comparator portion, reference numeral2represents a commutation portion, reference numeral3represents a pre-drive portion, reference numeral4represents an output portion, reference numeral5represents a starter portion, reference numeral6represents a lock protection portion, reference numeral7represents an overheat protection portion, and reference numeral8represents a rotor rest position detection portion. Reference symbols L1and L2represent respectively a first-phase coil and a second-phase coil of a motor M.

The comparator portion1-1outputs a binary signal BEMF1that represents the relationship in magnitude between the voltage at the output point OUT1for the first phase (i.e. the node between the first-phase coil L1and a transistor T1included in the output portion4) and a driving voltage VMfor the motor M. The comparator portion1-2outputs a binary signal BEMF2that represents the relationship in magnitude between the voltage at the output point OUT2for the second phase (i.e. the node between the second-phase coil L2and a transistor T2included in the output portion4) and the driving voltage VMfor the motor M.

The commutation portion2, according to the signals BEMF1and BEMF2output respectively from the comparator portions1-1and1-2, produces and outputs signals G1and G2by which the transistors T1and T2constituting the output portion4are respectively turned on and off in such a way that the rotor of the motor rotates smoothly.

The pre-drive portion3performs level conversion on the signals G1and G2output from the commutation portion2to make their levels high enough to turn on and off the transistors T1and T2constituting the output portion4, and then feeds those signals to the gates of the transistors T1and T2.

The output portion4is composed of an n-channel MOS field-effect transistor T1connected between one end of the first-phase coil L1, of which the other end is connected to the drive voltage VMfor the motor M, and ground and an n-channel MOS field-effect transistor T2connected between one end of the second-phase coil L2, of which the other end is connected to the drive voltage VMfor the motor M, and ground.

The starter portion5starts the motor M by energizing first the phase determined according to the result of detection by the rotor rest position detection portion8. How this is achieved in practice will be described later. On the basis of the output signals from the comparator portions1-1and1-2and the internal signals within the commutation portion2, the lock protection portion6checks whether the motor is locked or not (for example, when those signals remain unchanged for a predetermined period, the motor is recognized to be locked). If the motor continues being driven in the locked state, the motor and the driving device will be destroyed. To prevent this, when the motor is recognized to be locked, the lock protection portion6de-energizes the coils of both phases of the motor and, a predetermined period thereafter, makes the starter portion5restart the motor. The overheat protection portion7prevents thermal runaway by monitoring the ambient temperature and, when the monitored temperature exceeds a predetermined level, de-energizing the coils of both phases of the motor.

Before the motor M is started, the rotor rest position detection portion8detects the position in which the rotor of the motor is at rest by exploiting the fact that the first-phase coil L1and the second-phase coil L2have varying apparent inductances according to the position in which the rotor is at rest. On the basis of the result of this detection, which of the first-phase and second-phase coils to energize first when the motor M is started is determined.

FIG. 2shows a practical example of the circuit configuration of the comparator portions1-1and1-2. A pnp-type transistor11has its emitter connected to the drive voltage VMfor the motor M, and has its base and collector connected together. A resistor12is connected, at one end, to the collector of the transistor11and, at the other end, to one end of a constant-current circuit13, which is grounded at the other end. A pnp-type transistor14has its base connected to the node between the resistor12and the constant-current circuit13, has its emitter connected to the output point of the corresponding phase (i.e. to the output point OUT1of the first phase in the case of the comparator portion1-1, and to the output point OUT2of the second phase in the case of the comparator portion1-2), and has its collector connected to one end of a resistor15, which is grounded at the other end. The node between the collector of the transistor14and the resistor15serves as the output terminal at which the signal BEMF1or BEMF2appears.

In this circuit configuration, if variations in the base-emitter forward voltage of the transistors11and14are ignored, when the voltages at the output points OUT1and OUT2of the first and second phases are higher than threshold voltages that are lower than the drive voltage VMfor the motor M by the voltage drop across the resistor12, the transistor14turns on, and thus the signals BEMF1and BEMF2respectively turn to a high level. On the other hand, when those voltages are not higher than the threshold voltages, the transistor14turns off, and thus the signals BEMF1and BEMF2respectively turn to a low level. The voltage drop across the resistor12is set to be so small that the threshold voltages are substantially equal to the drive voltage VMfor the motor M.

FIG. 3shows a practical example of the circuit configuration of the commutation portion2. As shown in this figure, the commutation portion2is composed of NOR circuits201,202,203, and204, NAND circuits205and206, NOT circuits (inverter circuits)207and208, and selectors209and210. These are interconnected as follows.

Between the NOR circuits201and202, the output terminal of one is connected to one of the input terminals of the other so as to form an RS flip-flop circuit. The NOR circuit201receives, at the other input terminal, the signal BEMF2output from the comparator portion1-2, and the NOR circuit202receives, at the other input terminal, the signal BEMF1output from the comparator portion1-1.

The NAND circuit205receives, at one input terminal, the signal BEMF2output from the comparator portion1-2and, at the other input terminal, the output of the NOR circuit202. The output terminal of the NAND circuit205is connected to the input terminal of the NOT circuit207.

Between the NOR circuits203and204, the output terminal of one is connected to one of the input terminals of the other so as to form an RS flip-flop circuit. The NOR circuit203receives, at the other input terminal, the signal BEMF1output from the comparator portion1-1, and the NOR circuit204receives, at the other input terminal, the signal BEMF2output from the comparator portion1-2.

The NAND circuit206receives, at one input terminal, the signal BEMF1output from the comparator portion1-1and, at the other input terminal, the output of the NOR circuit204. The output terminal of the NAND circuit206is connected to the input terminal of the NOT circuit208.

As a result, the signals BEMF1and BEMF2output respectively from the comparator portions1-1and1-2and the back electromotive force drive signals B1and B2output respectively from the NOT circuits207and208have a relationship as shown in Table 1; that is, the signals B1and B2are never at a high level at the same time. Thus, the transistors T1and T2of the output portion4are never on together. In Table 1, “1” represents a high level, and “0” represents a low level.

The selector209receives the back electromotive force drive signal B1output from the NOT circuit207and a starting signal S1output from the starter portion5. The selector209selects one of these two signals according to a select signal SEL output from the starter portion5, and outputs the selected signal. Specifically, the selector209outputs the starting signal S1when the select signal SEL is at a low level, and outputs the back electromotive force drive signal B1when the select signal SEL is at a high level.

The selector210receives the back electromotive force drive signal B2output from the NOT circuit208and a starting signal S2output from the starter portion5. The selector210selects one of these two signals according to the select signal SEL output from the starter portion5, and outputs the selected signal. Specifically, the selector210outputs the starting signal S2when the select signal SEL is at a low level, and outputs the back electromotive force drive signal B2when the select signal SEL is at a high level.

The selectors209and210respectively output signals G1and G2, which are fed to the pre-drive portion3, where they are subjected to level conversion to be converted into signals G1′ and G2′, which are fed to the gates of the transistors T1and T2constituting the output portion4.

FIG. 4shows a practical example of the circuit configuration of the rotor rest position detection portion8. In this example, the rotor rest position detection portion8is composed of a switch81of which one end is connected to the output point OUT1of the first phase, a switch82of which one end is connected to the output point OUT2of the second phase, a resistor83of which one end is connected to the other end of both the switches81and82and of which the other end is grounded, a comparator84of which the non-inverting input terminal (+) is connected to a reference voltage Vrefand of which the inverting input terminal (−) is connected to the node P at which the switches81and82and the resistor83are connected together, and an up/down counter85. The resistor83is given a resistance so high that, even when the switches81and82are turned on, the current flowing through the resistor83is so low that the rotor does not rotate.

When instructed by the starter portion5to start counting up, the up/down counter85starts incrementing its output value CNT, starting with zero, by one every time a clock signal CLK generated by a clock generator9rises (this operation will be referred to as the “count-up operation”). When instructed to start counting down, the up/down counter85starts decrementing its output value CNT by one every time the clock signal CLK rises (this operation will be referred to as the “count-down operation”). Once the count-up or count-down operation is started, it is continued until stopped on a trailing edge in a signal A output from the comparator84. The up/down counter85, when its output value becomes equal zero in the count-down operation, thereafter keeps it at zero, ignoring the following rising edges in the clock signal CLK.

Here, it is assumed that the coil of the phase that is being energized produces a magnetic field of the S pole. Then, in general, whichever of the first-phase and second-phase coils L1and L2is closer to the S pole of the rotor has a higher apparent inductance, and therefore the current flowing through that coil varies more gently. Thus, in a case where the coil L2is closer to the S pole of the rotor, the length of time required for the signal A output from the comparator84to turn from a high level to a low level when the switches81and82are turned on individually is longer when the switch82is turned on than when the switch81is turned on. By contrast, in a case where the coil L1is closer to the S pole of the rotor, the aforementioned length of time is longer when the switch81is turned on than when the switch82is turned on.

Before the motor M is started, the starter portion5turns the switch81on and the switch82off, and instructs the up/down counter85to start counting up. Thereafter, when the voltage at the node P becomes higher than the reference voltage Vrefand thus the signal A output from the comparator84falls, the starter portion5turns the switch81off and the switch82on, and instructs the up/down counter85to start counting down.

Then, according to whether the output value CNT of the up/down counter85is equal to (i.e. has reached) zero or not on the next trailing edge in the signal A output from the comparator84, which of the first-phase and second-phase coils L1and L2to energize first to start the motor M is determined. Specifically, when the output value of the up/down counter85is zero, the motor M is started by energizing the second-phase coil L2first; by contrast, when the output of the up/down counter85is not zero (has not reached zero), the motor M is started by energizing the first-phase coil L1first.

In this embodiment, the two-phase half-wave motor used as the target to be driven is so structured that, as shown inFIGS. 5A and 5B, the air gaps between the stators20-1and20-2and the rotor10are made narrower and narrower in the direction of rotation indicated by arrows so that those air gaps have different widths at different points. The rotor10comes to rest in a position where the air gaps are narrowest right at the N and S poles of the magnet of the rotor10. Accordingly, when the rotor10is at rest, it is either in the state shown inFIG. 5Aor in the state shown inFIG. 5B. Moreover, as described earlier, it is assumed that the coil of the phase that is being energized produces a magnetic field of the S pole. That is, when the coil of the phase that is located closer to the S pole of the rotor10is energized first, the rotor10starts rotating in the normal rotation direction indicated by the arrows. It is to be understood that, although the magnet is provided on the part of the rotor and the coils are provided on the part of the stators in this embodiment, it is also possible to provide instead the magnet on the part of a stator and the coils on the part of a rotor.

When the rotor rest position detection portion8configured as shown inFIG. 4is used, relevant signals behave as shown in a timing chart inFIG. 6when the motor M is started. When an internal reset signal RST turns to a low level as a result of power-on resetting, the starter portion5turns the select signal SEL and the starting signals S1and S2to a low level. Then, at a time point pi inFIG. 6, the starter portion5turns the switches81and82of the rotor rest position detection portion8on and off, respectively, and instructs the up/down counter85to start counting up. It is to be noted that, although the output value CNT of the up/down counter85is shown as varying voltage levels inFIG. 6for easy understanding, it instead may be a binary signal that represents either zero or larger than zero.

As a result, the output value CNT of the up/down counter85starts being incremented by one at a time in synchronism with the clock signal CLK, and the input voltage to the inverting input terminal (−) of the comparator84starts rising. When the input voltage to the inverting input terminal (−) of the comparator84becomes higher than the reference voltage Vrefand the signal A output from the comparator84turns from a high level to a low level, the up/down counter85stops operating.

Thereafter, the starter portion5turns the switches81and82off and on, respectively, and instructs the up/down counter85to start counting down (at a time point P2inFIG. 6). As a result, the output value CNT of the up/down counter85starts being decremented by one at a time in synchronism with the clock signal CLK. Meanwhile, the input voltage to the inverting input terminal (−) of the comparator84first falls to the ground level and then starts rising again. When the input voltage to the inverting input terminal (−) of the comparator84becomes higher than the reference voltage Vrefand the signal A output from the comparator84turns from a high level to a low level, the up/down counter85stops operating.

At this point, the count value CNT of the up/down counter85is not equal to zero in a case where the first-phase coil L1is closer to the S pole of the rotor than the second-phase coil L2is, because then the current through the first-phase coil L1rises more slowly than the current through the second-phase coil L2. By contrast, the count value CNT is equal to zero in a case where the second-phase coil L2is closer to the S pole of the rotor than the first-phase coil L1is, because then the current through the second-phase coil L2rises more slowly than the current through the first-phase coil L1.

In the case shown inFIG. 6, the output value CNT of the up/down counter85is not equal to zero, meaning that the first-phase coil L1is closer to the S pole of the rotor than the second-phase coil L2is (the state shown inFIG. 5A). Accordingly, the starter portion5turns the switch82off, simultaneously turns the starting signal S1to a high level (at a time point p3inFIG. 6), and thereafter turns the select signal SEL to a high level (at a time point p4inFIG. 6).

As a result, the transistor T1of the output portion4turns on first, and thus the first-phase coil L1, which is closer to the S pole of the rotor, is energized first, causing the rotor to start rotating in the normal rotation direction. Thereafter, the turning on and off of the transistors T1and T2of the output portion4is controlled respectively by the back electromotive force drive signals B1and B2, and thus which phase to energize is switched every time the voltage at the output point of the phase that is not being energized falls below the drive voltage VMfor the motor M (more precisely, the threshold voltage of the corresponding comparator portion), in other words, substantially at zero-cross points of the back electromotive force appearing in the coil of the phase that is not being energized as the rotor rotates. Thus, the rotation of the motor is maintained.

By contrast, in a case where the output value CNT of the up/down counter85is equal to zero, this means that the second-phase coil L2is closer to the S pole of the rotor than the first-phase coil L1is (the state shown inFIG. 5B). Accordingly, the starter portion5turns the starting signal S2to a high level, and thereafter turns the select signal SEL to a high level.

As a result, the transistor T2of the output portion4turns on first, and thus the second-phase coil L2, which is closer to the S pole of the rotor, is energized first, causing the rotor to start rotating in the normal rotation direction. Thereafter, the turning on and off of the transistors T1and T2of the output portion4is controlled respectively by the back electromotive force drive signals B1and B2, and thus which phase to energize is switched every time the voltage at the output point of the phase that is not being energized falls below the drive voltage VMfor the motor M (more precisely, the threshold voltage of the corresponding comparator portion), in other words, substantially at zero-cross points of the back electromotive force appearing in the coil of the phase that is not being energized as the rotor rotates. Thus, the rotation of the motor is maintained.

After the motor M is started in this way, the operation of the comparator84and the up/down counter85of the rotor rest position detection portion8shown inFIG. 4may be stopped. This helps reduce power consumption. The time difference resulting from the difference between the waveforms with which currents start flowing through the first-phase and second-phase coils L1and L2is in the range from several microseconds to several tens of microseconds, and therefore the frequency of the clock signal CLK generated by the clock generator9needs to be of the order of MHz. However, in cases where no such high-frequency clock is needed once the motor has been started, the frequency of the clock signal CLK generated by the clock generator9may be lowered, or the operation of the clock generator9may be stopped until the clock signal CLK is needed again. This helps further reduce power consumption.

The rotor rest position detection portion8is provided with the up/down counter85, and the lock protection portion6requires a counter to measure time. Whereas the rotor rest position detection portion8has only to operate when the motor is started, the lock protection portion6need not operate when the motor is started. Therefore, a single counter may be shared between the lock protection portion6and the rotor rest position detection portion8. This helps minimize the increase in the circuit scale and the increase in costs.

In practical terms, it is preferable that the rotor rest position detection portion8shown inFIG. 4be provided with a switch connected between the node P and ground so that this switch is turned on as required to ensure that the potential at the node P has fallen to the ground level before the switches81and82are individually turned on. This permits the position in which the rotor is at rest to be detected with higher accuracy.

FIG. 7shows another practical example of the circuit configuration of the rotor rest position detection portion8. In this example, the rotor rest position detection portion8is composed of capacitors C1and C2each grounded at one end, a switch SW1connected between the output point OUT1of the first phase and the other end of the capacitor C1, a switch SW2connected between the output point OUT2of the second phase and the other end of the capacitor C2, a comparator801of which the non-inverting input terminal (+) is connected to the node P1between the switch SW1and the capacitor C1and of which the inverting input terminal (−) is connected to the node P2between the switch SW2and the capacitor C2, an n-channel MOS field-effect transistor T11of which the drain is connected to the node P1between the switch SW1and the capacitor C1and of which the source is grounded, and an n-channel MOS field-effect transistor T12of which the drain is connected to the node P2between the switch SW2and the capacitor C2and of which the source is grounded.

Here, it is assumed that the coil of the phase that is being energized produces a magnetic field of the S pole. Then, whichever of the first-phase and second-phase coils L1and L2is closer to the S pole of the rotor has a higher apparent inductance, and therefore the current flowing through that coil varies more gently. Thus, in a case where the coil L2is closer to the S pole of the rotor, when first the capacitors C1and C2are discharged and then the switches SW1and SW2are turned on simultaneously, the potential at the node P2between the switch SW2and the capacitor C2rises more slowly than the potential at the node P1between the switch SW1and the capacitor C1as shown inFIG. 8. Thus, at any time point before the capacitors C1and C2are fully charged, the potential at the node P2is lower than the potential at the node P1, and thus the signal “a” output from the comparator801is at a high level. It is to be noted that the switches SW1and SW2are kept on only for a period of time so short as not to cause the rotor to change its position.

By contrast, in a case where the coil L1is closer to the S pole of the rotor, when the switches SW1and SW2are turned on simultaneously, the potential at the node P1rises more slowly than the potential at the node P2, and thus the signal “a” output from the comparator801is at a low level.

The turning on and off of the switches SW1and SW2and of the transistors T11and T12is controlled by the starter portion5. The signal “a” output from the comparator801is fed to the starter portion5. Before the motor M is started, the starter portion5turns the transistors T11and T12on momentarily to discharge the capacitors C1and C2respectively, and then turns the switches SW1and SW2on simultaneously.

Then, according to whether the signal “a” output from the comparator801turns to a high or low level, the starter portion5determines which of the first-phase and second-phase coils L1and L2to energize first to start the motor M. Specifically, when the signal “a” turns to a high level, the starter portion5starts the motor M by energizing the second-phase coil L2first; by contrast, when the signal “a” turns to a low level, the starter portion5starts the motor M by energizing the first-phase coil L1first.

When the rotor rest position detection portion8configured as shown inFIG. 7is used, relevant signals behave as shown in a timing chart inFIG. 9when the motor M is started. When an internal reset signal RST rises as a result of power-on resetting, the starter portion5turns the select signal SEL to a low level, and turns the starting signals S1and S2to a low level. In addition, the starter portion5turns on the switches SW1and SW2of the rotor rest position detection portion8simultaneously, and keeps them on for a period of time t1so short as not to cause the rotor of the motor M to rotate.

Now, since the motor to be driven is structured as described earlier, in a case where the first-phase coil L1is closer to the S pole of the rotor than the second-phase coil L2is, the potential at the node P1is lower than the potential at the node P2in the rotor rest position detection portion8, and thus the signal “a” output from the comparator801is at a low level. By contrast, in a case where the second-phase coil L2is closer to the S pole of the rotor than the first-phase coil L1is, the potential at the node P1is higher than the potential at the node P2, and thus the signal “a” output from the comparator801is at a high level.

In the case shown inFIG. 9, the signal “a” output from the comparator801turns to a high level, meaning that the second-phase coil L2is closer to the S pole of the rotor than the first-phase coil L1is (the state shown inFIG. 5B). Thus, the starter portion5first keeps the select signal SEL at a low level, the starting signal S1at a low level, and the starting signal S2at a high level for a predetermined period of time t2, and then turns the select signal SEL to a high level.

As a result, the transistor T2of the output portion4turns on first, and thus the second-phase coil L2, which is closer to the S pole of the rotor, is energized first, causing the rotor to start rotating in the normal rotation direction. Thereafter, the turning on and off of the transistors T1and T2of the output portion4is controlled respectively by the back electromotive force drive signals B1and B2, and thus which phase to energize is switched every time the voltage at the output point of the phase that is not being energized falls below the drive voltage VMfor the motor M (more precisely, the threshold voltage of the corresponding comparator portion), in other words, substantially at zero-cross points of the back electromotive force appearing in the coil of the phase that is not being energized as the rotor rotates. Thus, the rotation of the motor is maintained.

By contrast, in a case where the signal “a” output form the comparator801turns to a low level, this means that the first-phase coil L1is closer to the S pole of the rotor than the second-phase coil L2is (the state shown inFIG. 5A). Thus, the starter portion5first keeps the select signal SEL at a low level, the starting signal S1at a high level, and the starting signal S2at a low level for a predetermined period of time t2, and then turns the select signal SEL to a high level.

As a result, the transistor T1of the output portion4turns on first, and thus the first-phase coil L1, which is closer to the S pole of the rotor, is energized first, causing the rotor to start rotating in the normal rotation direction. Thereafter, the turning on and off of the transistors T1and T2of the output portion4is controlled respectively by the back electromotive force drive signals B1and B2, and thus which phase to energize is switched every time the voltage at the output point of the phase that is not being energized falls below the drive voltage VMfor the motor M (more precisely, the threshold voltage of the corresponding comparator portion), in other words, substantially at zero-cross points of the back electromotive force appearing in the coil of the phase that is not being energized as the rotor rotates. Thus, the rotation of the motor is maintained.

After the motor M is started in this way, the operation of the comparator801of the rotor rest position detection portion8shown inFIG. 7may be stopped. This helps reduce power consumption. In the rotor rest position detection portion8shown inFIG. 7, by inserting resistors respectively between the switch SW1and the capacitor C1and between the switch SW2and the capacitor C2, it is possible to reduce the capacitances of the capacitors C1and C2.

Now, still another practical example of the circuit configuration of the rotor rest position detection portion8will be described with reference toFIG. 10. Resistors R1and R2are connected respectively through switches SW11and SW12to the output point OUT1of the first phase and the output point OUT2of the second phase. The resistors R1and R2are given resistances so high that, even when the switches SW11and SW12are respectively turned on, the currents flowing therethrough are so low that the rotor does not rotate.

An operational amplifier802has its output terminal connected to its inverting input terminal (−) to form a buffer circuit. The non-inverting input terminal (+) of this operational amplifier802is connected to the node P11between the switch SW11and the resistor R1. An operational amplifier803forms a buffer circuit in the same manner as the operational amplifier802, and the non-inverting input terminal (+) of this operational amplifier803is connected to the node P12between the switch SW12and the resistor R2.

The output of the operational amplifier802is subjected to level conversion by being passed through two inverter circuits804and805, and is then fed to the data input terminal D of a D flip-flop circuit808. The output of the operational amplifier803is subjected to level conversion by being passed through two inverter circuits806and807, and is then fed to the clock input terminal CK of the D flip-flop circuit808.

By using comparators instead of the operational amplifiers802and803individually forming buffer circuits, it is possible to omit the inverter circuits804,805,806, and807.

Here, it is assumed that the coil of the phase that is being energized produces a magnetic field of the S pole. Then, whichever of the first-phase and second-phase coils L1and L2is closer to the S pole of the rotor has a higher apparent inductance, and therefore the current flowing through that coil varies more gently. Thus, in a case where the coil L2is closer to the S pole of the rotor, when the switches SW11and SW12are turned on simultaneously, the potential at the node P12rises more slowly than the potential at the node P11as shown inFIG. 11. Thus, in the D flip-flop circuit808, the signal X1fed to the data input terminal D rises from a low level to a high level earlier than the signal X2fed to the clock input terminal CK, and accordingly the signal Y output from the output terminal Q turns to a high level. InFIG. 11, VINVrepresents the threshold voltage of the inverter circuits804,805,806, and807.

By contrast, in a case where the coil L1is closer to the S pole of the rotor, when the switches SW11and SW12are turned on simultaneously, the potential at the node P11rises more slowly than the potential at the node P12. Thus, in the D flip-flop circuit808, the signal X2fed to the clock input terminal CK rises from a low level to a high level earlier than the signal X1fed to the data input terminal D, and accordingly the signal Y output from the output terminal Q turns to a low level.

The turning on and off of the switches SW11and SW12is controlled by the starter portion5. The signal Y output from the D flip-flop circuit808is fed to the starter portion5. Before the motor M is started, the starter portion5turns the switches SW11and SW12on simultaneously, and, according to whether the signal Y output from the D flip-flop circuit808turns to a high or low level at this point, determines which of the first-phase and second-phase coils L1and L2to energize first to start the motor M. Specifically, when the signal Y turns to a high level, the starter portion5starts the motor M by energizing the second-phase coil L2first; by contrast, when the signal Y turns to a low level, the starter portion5starts the motor M by energizing the first-phase coil L1first.

When the rotor rest position detection portion8configured as shown inFIG. 10is used, relevant signals behave as shown in a timing chart inFIG. 12when the motor M is started. When an internal reset signal RST rises as a result of power-on resetting, the starter portion5turns the select signal SEL to a low level, and turns the starting signals S1and S2to a low level. In addition, the starter portion5turns on the switches SW11and SW12of the rotor rest position detection portion8simultaneously.

Now, since the motor to be driven is structured as described earlier, in a case where the first-phase coil L1is closer to the S pole of the rotor than the second-phase coil L2is, in the D flip-flop circuit808of the rotor rest position detection portion8shown inFIG. 8, the signal X2fed to the clock input terminal CK rises from a low level to a high level earlier than the signal X1fed to the data input terminal D, and thus the signal Y output from the output terminal Q turns to a low level. By contrast, in a case where the second-phase coil L2is closer to the S pole of the rotor than the first-phase coil L1is, the signal X1rises from a low level to a high level earlier than the signal X2, and thus the signal Y turns to a high level.

In the case shown inFIG. 12, the signal Y output from the D flip-flop circuit808turns to a high level, meaning that the second-phase coil L2is closer to the S pole of the rotor than the first-phase coil L1is (the state shown inFIG. 5B). Thus, the starter portion5first keeps the select signal SEL at a low level, the starting signal S1at a low level, and the starting signal S2at a high level for a predetermined period of time t3, and then turns the select signal SEL to a high level. The switches SW11and SW12are turned off when both the signals X1and X2have turned to a high level.

As a result, the transistor T2of the output portion4turns on first, and thus the second-phase coil L2, which is closer to the S pole of the rotor, is energized first, causing the rotor to start rotating in the normal rotation direction. Thereafter, the turning on and off of the transistors T1and T2of the output portion4is controlled respectively by the back electromotive force drive signals B1and B2, and thus which phase to energize is switched every time the voltage at the output point of the phase that is not being energized falls below the drive voltage VMfor the motor M (more precisely, the threshold voltage of the corresponding comparator portion), in other words, substantially at zero-cross points of the back electromotive force appearing in the coil of the phase that is not being energized as the rotor rotates. Thus, the rotation of the motor is maintained.

By contrast, in a case where the signal Y output from the D flip-flop circuit808turns to a low level, this means that the first-phase coil L1is closer to the S pole of the rotor than the second-phase coil L2is (the state shown inFIG. 5A). Thus, the starter portion5first keeps the select signal SEL at a low level, the starting signal S1at a high level, and the starting signal S2at a low level for a predetermined period of time t3, and then turns the select signal SEL to a high level.

As a result, the transistor T1of the output portion4turns on first, and thus the first-phase coil L1, which is closer to the S pole of the rotor, is energized first, causing the rotor to start rotating in the normal rotation direction. Thereafter, the turning on and off of the transistors T1and T2of the output portion4is controlled respectively by the back electromotive force drive signals B1and B2, and thus which phase to energize is switched every time the voltage at the output point of the phase that is not being energized falls below the drive voltage VMfor the motor M (more precisely, the threshold voltage of the corresponding comparator portion), in other words, substantially at zero-cross points of the back electromotive force appearing in the coil of the phase that is not being energized as the rotor rotates. Thus, the rotation of the motor is maintained.

After the motor M is started in this way, in the rotor rest position detection portion8shown inFIG. 10, the operation of the operational amplifiers802and803, the inverter circuits804,805,806, and807, and the D flip-flop circuit808may be stopped. This helps reduce power consumption.

In practical terms, it is preferable that the rotor rest position detection portion8shown inFIG. 10be provided with a switch connected between the node P11and ground and a switch connected between the node P12and ground so that these switches are turned on as required to ensure that the potentials at the nodes P11and P12have fallen to the ground level before the switches SW11and SW12are individually turned on. This permits the position in which the rotor is at rest to be detected with higher accuracy.

As described above, in this embodiment, before the motor is started, the position in which the rotor is at rest is detected by exploiting the fact that the coils of individual phases have varying apparent inductances according to the position in which the rotor is at rest; specifically, which of the first-phase and second-phase coils is closer to a predetermined magnetic pole (the same pole as that produced in the coil of the phase that is being energized) is detected. Then, according to the result of this detection, which phase to energize first when the motor is started is determined. This makes it possible to start the motor always in the same rotation direction, and thus to prevent reverse rotation of the motor.

Moreover, in this embodiment, which phase to energize is switched according to the back electromotive force appearing in the coil of each phase as the rotor rotates. This eliminates the need to use a hole element to detect the position of the rotor, and thus helps promote cost reduction and miniaturization.

When the rotor is rotating, a magnet fixed to the rotor moves together, causing the magnetic flux passing through the coil of each phase to vary with time. Thus, a back electromotive force appears in the coil of each phase, and therefore the voltage at the output point of the phase that is not being energized has a waveform as shown inFIG. 13with the back electromotive force superposed on the drive voltage VMfor the motor M. The back electromotive force has a sinusoidal waveform that is synchronous with the rotation of the rotor, becoming equal to zero when the rotor is at an electrically stationary point. Therefore, it is possible to detect the position of the rotor according to the back electromotive force appearing in the coil of each phase. This makes it possible to rotate the rotor as smoothly as when a Hall element is used by, as described above, switching which phase to energize according to the back electromotive force appearing in the coil of each phase as the rotor rotates.

Moreover, in this embodiment, when the voltage at the output point of the phase that is not being energized drops below the drive voltage VMfor the motor (in other words, when the back electromotive force appearing in the coil of the phase that is not being energized drops below zero), this serves as a trigger that causes the coil of the phase that has been energized thus far to stop being energized. As a result, a back electromotive force causes the voltage at the output point of the phase that has been energized to rise above the drive voltage VMfor the motor (in other words, the back electromotive force appearing in the coil of the phase that has been energized rises above zero), and this serves as a trigger that causes the coil of the phase that has not been energized thus far to start being energized. Therefore, the first and second phases are never energized at the same time when which phase to energize is switched. This helps reduce power consumption, and ensures efficient rotation of the motor.

In the voltages at the output points OUT1and OUT2of the first and second phases, spike noise “n” as shown inFIG. 14appears when the coils of the first and second phases are respectively switched from an energized state to a de-energized state. Thus, if the threshold voltages Vthof the comparator portions1-1and1-2are higher than the drive voltage VMfor the motor M, even when the rotor is not rotating, the output signals BEMF1and BEMF2of the comparator portions1-1and1-2may vary, causing entry into the mode in which the back electromotive force drive signals B1and B2control the turning on and off of the transistors T1and T2constituting the output portion4. In this state, as shown inFIG. 14, which phase to energize is switched so fast that the rotor does not rotate.

In this embodiment, however, the comparators1-1and1-2have the circuit configuration shown in.FIG. 2, and therefore, as long as variations in the base-emitter forward voltage of the transistors11and14are within tolerated limits (specifically, within the voltage drop across the resistor12), the threshold voltages of the comparator portions1-1and1-2are never higher than the drive voltage VMfor the motor M. This prevents the problem described above.

Setting the threshold voltage Vthfor the voltage appearing at the output point of the phase that is not being energized in such a way that it is never higher than the drive voltage VMfor the motor M is equivalent, if the situation is put in other words as which phase to energize being switched when the back electromotive force appearing in the coil of the phase that is not being energized crosses a threshold level, to setting this threshold level in such a way that it is never higher than zero.

This embodiment deals with a case where the open end of the coil of the phase that is not being energized is on the current outflow side when this coil is energized. In a case where the open end of the coil of the phase that is not being energized is on the current inflow side when this coil is energized, the threshold level used to switch which phase to energize when the back electromotive force appearing in the coil of the phase that is not being energized crosses the threshold level is set in such a way as to be never lower than zero. This makes it possible to prevent the motor's failure to rotate resulting from spike noise appearing in the coil of each phase when the coil of each phase is switched from an energized state to a de-energized state.

In this embodiment, the comparator portions1-1and1-2use separate threshold voltages; however, they may use a common threshold voltage. The pre-drive portion3may be omitted by incorporating it in the commutation portion2. The output portion4may be composed of npn-type bipolar transistors.

The embodiment described above deals with a case where a two-phase half-wave motor is driven. It is to be understood, however, that the present invention helps prevent reverse rotation of a motor also in cases where the motor is of a single-phase all-wave type, three-phase half-wave type, three-phase all-wave type, or any other type.