Methods and systems to compensate IQ imbalance in zero-IF tuners

Methods and systems to calibrate I/Q imbalance with digital equalizers to compensate frequency dependent IQ imbalance, weighted summation modules to compensate frequency independent IQ imbalance, switch modules to controllably direct calibration signals through selected components and to a digital compensation calculator and to control tuner phases during calibration. The equalizers and summation modules may provide genetic IQ imbalance compensation. Methods and systems disclosed herein may be implemented with respect to relatively wideband systems having non-linear spectrum responses, and other systems.

BACKGROUND

A zero intermediate frequency (ZIF) tuner, also referred to herein as a direct frequency converter and single stage frequency converter, converts between relatively high frequency signal and baseband directly rather than through one or more intermediate stages or intermediate frequencies.

ZIF tuners may introduce imbalance between in-phase and quadrature phase (IQ) paths. IQ imbalance may arise from current leakage, characteristics, and differences between circuit components in I and Q paths. IQ imbalance may include frequency dependent IQ imbalance arising from low pass filters of a ZIF tuner, and frequency independent IQ imbalance arising from mixers of the ZIF tuner.

IQ imbalance may be reduced with stricter tolerances of ZIF tuner components and/or with compensation applied in an analog domain, but at a relatively substantial cost.

Where ZIF low pass filters have a relatively flat spectrum response and a linear phase over a passband, IQ imbalance may be compensated with gain and phase correction in a digital domain. A narrow bandwidth system may have a relatively flat low pass filter spectrum response and a relatively linear phase over a bandwidth. Thus, calibration and compensation may be performed in a digital domain. In wideband systems, however, such as a Multimedia over Coax Alliance (MoCA) standard based system, a ZIF tuner may have relatively high order low-pass filters to provide a wide passband and sharp cut-off. In such a wideband system, a flat passband may be prohibitively expensive and digital gain and phase compensation techniques may not be sufficient to correct IQ imbalance.

In the drawings, the leftmost digit(s) of a reference number identifies the drawing in which the reference number first appears.

DETAILED DESCRIPTION

Disclosed herein are methods and systems to calibrate I/Q imbalance in transceivers, including zero intermediate frequency (ZIF) based transceivers, and including calibration of frequency independent I/Q imbalance generated within frequency converters and frequency dependent I/Q imbalance generated within baseband low pass filters.

Methods and systems disclosed herein may be implemented with respect to relatively wideband systems, narrowband systems, systems having linear spectrum responses, systems having non-linear spectrum responses, and combinations thereof.

Methods and systems disclosed herein include digital equalizers to compensate frequency dependent IQ imbalance, weighted summation modules to compensate frequency independent IQ imbalance, and switch modules to controllably direct calibration signals through selected components and to a digital compensation calculator, and to control tuner phases during calibration.

FIG. 1is a process flowchart of an exemplary method100of calibrating frequency dependent and frequency independent IQ imbalance in transmit and receive paths of a transceiver. Method100is described below with reference toFIGS. 2-11. Method100is not, however, limited to the examples ofFIGS. 2-11. Method100, or portions thereof, may be implemented under control of a state machine.

FIG. 2is a block diagram of an exemplary transceiver200, including a transmit path202and a receive path204. Transmit path202may include I and Q frequency up-converters206and208, and receive path204may include I and Q frequency down-converters210and212. Frequency converters206through212may include zero-intermediate frequency (ZIF) converters, also referred to herein as direct frequency converters and single-stage frequency converters, to convert between a carrier frequency and a relatively low intermediate frequency or baseband.

Frequency up-converters206and208may be configured to mix I and Q reference or carrier signals214and216with corresponding baseband signals It(t) and Qt(t), to generate an image of the baseband signals at the carrier frequency, illustrated here as g(t) at a summation node222.

Frequency down-converters210and212may be configured to mix I and Q reference signals214and216with a received signal r(t) at a node224, to generate corresponding I and Q baseband signals Ir(t) and Qr(t). Baseband signals Ir(t) and Qr(t) may correspond to relatively low frequency I and Q images of signal r(t), or may correspond to I and Q baseband components of signal r(t).

Transceiver200may include I and Q transmit path analog baseband processing components230and232, and I and Q receive path analog baseband processing components234and236. In the example ofFIG. 2, components230through236include corresponding low pass filters (LPFs)238through244.

Transceiver200may include a digital baseband processor (DBP)276, including a transmit portion278and a receive portion280.

Transceiver200may be configured as a relatively wideband system, and may be implemented in accordance with a Multimedia over Coax Alliance (MoCA) standard as promulgated and/or proposed by the MoCA at http://www.mocalliance.org/, such as to facilitate home networking on coaxial cable, which may operate in a bandwidth of 50 MHz or higher, and which may include OFDM modulation. Transceiver200is not, however, limited to wideband systems.

Transceiver200may be subject to one or more of frequency dependent and frequency independent IQ imbalance. For example, receive path baseband analog components234and236may have respective transfer characteristics Hr(s) and Fr(s), which may include filter characteristics associated with I and Q LPFs242and244. Where characteristics Hr(s) and Fr(s) differ from one another, frequency dependent IQ imbalance may be imparted to corresponding signals Ir(t) and Qr(t).

Similarly, transmit path baseband analog components230and232may have respective transfer characteristics Ht(s) and Ft(s), which may include filter characteristics associated with I and Q LPFs238and240. Where characteristics Ht(s) and Ft(s) differ from one another, frequency dependent IQ imbalance may be imparted to corresponding signals It(t) and Qt(t).

As disclosed herein, frequency dependent IQ imbalance may be compensated with a digital equalizer in one of the I and Q baseband receive paths, and a corresponding delay element in the other receive path, and with a digital pre-equalizer in one of the I and Q baseband transmit paths and a corresponding delay element in the other transmit path.

Frequency independent IQ imbalance may arise in transmit path mixers206and208, and in receive path mixers210and212. As disclosed herein, frequency independent IQ imbalance may be compensated in a digital domain with a weighted summation module in each of the transmit and receive paths.

DBP276may include compensation calculator290to determine IQ compensation values from receive path digital baseband data292.

FIG. 3is a block diagram of DBP276, wherein transmit path DBP278includes a frequency dependent compensation module302and a frequency independent compensation module304, and receive path DBP280includes a frequency dependent compensation module306and a frequency independent compensation module308.

In the example ofFIG. 3, transmit path frequency dependent compensation module302includes a pre-equalizer310and a delay line312to compensate imbalance between transfer functions Ht(s) and Ft(s) of transmit path202ofFIG. 2.

Receive path frequency dependent compensation module306includes an equalizer314and a delay line316to compensate imbalance between transfer functions Hr(s) and Fr(s) of receive path204ofFIG. 2.

Transmit path frequency independent compensation module304includes a weighted summation module, including weighted modules320and322and a summation node324. Weighted modules320and322are weighted with corresponding parameters μ and λ.

Receive path frequency independent compensation module308includes a weighted summation module, including weighted modules326and328and a summation node330. Weighted modules326and328are weighed with corresponding parameters α and β.

Calibration of transceiver200may include calibrating one or more of coefficients of pre-equalizer310, coefficients of equalizer314, and weights λ, μ, α, and β.

Transceiver200may include switches and control circuitry to selectively control signal flow within transceiver200and to control tuner phases during a calibration mode. In the example ofFIG. 2, transceiver200includes switches282,284,286, and288, each having corresponding positions A and B.

Referring to method100inFIG. 1, at102, a calibration signal is applied to digital baseband I and Q nodes of a transceiver transmit path. The calibration signal may include a phase of an internally generated modulated carrier signal. InFIG. 2, the calibration signal may be output from transmit DBP278as Ia(n) and Qa(n).

At104, receive path frequency dependent IQ imbalance is calibrated. Calibration at104may include configuring the transceiver in a full loop back mode and directing one of I and Q calibration signals from the transmit path to inputs of both of I and Q receive path LPFs. Calibration at104may further include determining receive path digital baseband equalizer coefficients in response to corresponding receive path I and Q digital baseband signals, and may include adjusting the receive path equalizer coefficients may to balance the receive path I and Q digital baseband signals.

InFIG. 2, switches282and284may be placed in position B to couple outputs of I and Q receive path mixers210and212to corresponding I and Q receive path LPFs242and244. Switch286may be placed in position A to provide a common reference signal, illustrated here as I phase reference signal cos(ωt), to mixers206,208,210, and212. This effectively provides only an in-phase calibration signal to both of I and Q receive path LPFs242and244, which may avoid introducing frequency independent, mixer-based IQ imbalance during calibration of frequency dependent receive path IQ imbalance. Switch288may be placed in position A to configure transceiver200in a full-loop back mode. Switch288may remain in position A throughout a calibration mode to isolate receive path204from external interference.

FIG. 4is a corresponding signal flow diagram during receive path frequency dependent IQ imbalance calibration. In the example ofFIG. 4, transfer functions Hr(z)404and Fr(z)408represent transfer functions Hr(s) and Fr(s) of receive path LPFs242and244inFIG. 2in a digital domain. Equalizer314may be calibrated to compensate IQ imbalance arising from differences between transfer functions Hr(z)404and Fr(z)406.

Frequency-dependent I/Q imbalance caused by Hr(z)404and Fr(z)406is compensated when the convolution of I-channel low-pass filter transfer function Hr(z)404and equalizer Er(z)314is equal to a convolution of Q channel low-pass filter transfer function Fr(z)406and delay line316, such that:
Hr(z)Er(z)=Fr(n)*zN,

where N is a delay of delay line316.

The delay N of delay line316may be approximately equal to one half a length of equalizer314. The length of equalizer may be determined by channel transfer functions Hr(z)404and Fr(z)406.

Since both Ir(n) and Qr(n) are mixed with the same signal214at corresponding mixers210and212, equalizer314and delay line316are such that Ie(n) equals Qe(n), IQ imbalance is compensated. Mathematically:
Ie(z)=Er(z)Id(z)=Er(z)Hr(z)Ir(z); and
Qe(z)=Qd(z−N)=z−NFr(z)Qr(z), which lead to;
Er(z)Hr(z)=z−NFr(z).

FIG. 5is a block diagram of an exemplary configuration to calibrate receive path frequency dependent IQ imbalance.

Receive path digital baseband signal Id(n) and Qd(n) are received at compensation module300, and signals Ie(n) and Qe(n) are output from compensation module300to compensation calculator290. Compensation calculator290may determine a difference between signals Ie(n) and Qe(n) and generate a difference signal or value e(n)502. Compensation calculator290or equalizer314may adjust one or more coefficients of equalizer314in response to difference signal e(n)502.

Equalizer314may be implemented with one or more of a variety of architectures and coefficients of equalizer314may be computed in accordance with one or more of a variety of techniques. For example, and without limitation, equalizer314may include one or more of a finite impulse response (FIR) filter and an infinite impulse response (IIR) filter, and coefficients of equalizer314may be computed adaptively and/or directly.

The example ofFIG. 5may correspond to an adaptive FIR filter, and coefficients of equalizer314may be updated continuously in response to difference signal e(n)502. Updating of the coefficients may be halted when an energy of difference signal e(n)502is below a threshold. Coefficients may be loaded or applied to equalizer314prior to further calibration of system200.

Referring to method100inFIG. 1, at106, transmit path frequency dependent IQ imbalance is calibrated. Calibration at106may be performed subsequent to calibration of receive path frequency dependent IQ imbalance at104. Calibration at106may include directing the calibration signal from outputs of transmit path low pass baseband filters to corresponding inputs of receive path low pass baseband filters, and determining transmit path equalizer coefficients in response to corresponding receive path I and Q digital baseband signals.

InFIG. 2, switches282and284may be placed in position A to direct outputs It(t) and Qt(t) of transmit path analog baseband components230and232, to inputs Ir(t) and Qr(t) of receive path analog baseband components234and236.

FIG. 6is a corresponding signal flow diagram during transmit path frequency dependent IQ imbalance calibration.

In the example ofFIG. 6, digital domain transfer functions Ht(z)602and Ft(z)604represent transfer functions Ht(s) and Ft(s) of transmit path202inFIG. 2. Pre-equalizer310may be calibrated to compensate IQ imbalance arising from differences between transfer functions Ht(z)602and Ft(z)604.

FIG. 7is a block diagram of an exemplary configuration to calibrate transmit path frequency dependent IQ imbalance.

Baseband transmit signals Is(n) and Qs(n) may be set equal to one another, and may be directed through respective ones of transmit path delay line312and pre-equalizer310, through transmit path LPFs230and230, represented inFIG. 7as Ht(z)602and Ft(z)604, through receive path LPFs242and244, represented inFIG. 7as Hr(z)404and Fr(z)406, and through equalizer314and delay line316. Corresponding receive path baseband signals Ie(n) and Qe(n) may be provided to compensation calculator290.

Where receive path equalizer314is previously calibrated with respect to LPFs242and244, any frequency dependent IQ imbalance between baseband signals Ie(n) and Qe(n) may be attributed to transmit path LPFs230and230.

Compensation calculator290may determine a difference between signals Ie(n) and Qe(n) generate a difference signal or value e(n)702. Coefficients of pre-equalizer310may be adjusted to decrease the energy of difference signal e(n)702such that baseband signals Ie(n) and Qe(n) are substantially equal to one another, such as described above with respect toFIG. 5.

Pre-equalizer310may be implemented with one or more of a variety of architectures and coefficients of pre-equalizer310may be computed in accordance with one or more of a variety of techniques. For example, and without limitation, pre-equalizer310may include one or more of a finite impulse response (FIR) filter and an infinite impulse response (IIR) filter, and coefficients of pre-equalizer310may be computed adaptively and/or directly.

The example ofFIG. 7may correspond to an iterative update technique to determine coefficients of pre-equalizer310. Coefficients of the pre-equalizer310may be iteratively updated by a preset or pre-determined increment in response to difference signal e(n)702. An equalization state may correspond to:
Et(z)Ft(z)=z−MHt(z),
where M is a delay of delay line312.

The length of delay line312may be substantially equal to a delay of pre-equalizer310.

Referring to method100inFIG. 1, at108, transmit path frequency independent IQ imbalance is calibrated. Calibration at108may be performed subsequent to receive path and transmit path frequency dependent IQ imbalance calibration at104and106, respectively.

Calibration at108may include configuring the transceiver in a full loop-back mode, operating the transmit and receive path I and Q tuners at corresponding I and Q phases, and determining transmit path summation weights in response to one of the corresponding receive path I and Q digital baseband signals.

InFIG. 2, switches282,284, and286may be placed in position B, and switch288may remain in position A.FIG. 8is a corresponding signal flow diagram to calibrate transmit path frequency independent IQ imbalance. Transfer functions T(z)802and R(z)804represent effects of transmit path and receive paths, respectively.

where φ represents a phase difference to be compensated.

At receive node224, g(t) is down-converted to in-phase and quadrature components, which may be represented as:
Ir(τ)=It(τ)cos(θ)+Qt(τ)sin(φ+θ); and
Qr(τ)=−It(τ)sin(θ)+Qt(τ)cos(φ+θ);

There may not be a complex number a+jb such that:
Ir(τ)+j*Qr(τ)=(a+jb)(It(τ)+j*Qt(τ)),

due to φ≠0.

This is referred to herein as frequency independent IQ imbalance in g(t).

To compensate frequency independent IQ imbalance in g(t), let:
It(τ)=λItc(τ)+μQt(τ).

Applying It(τ) to equations above for Ir(τ) and Qr(τ) provides:
Ir(τ)=λItc(τ)cos(θ)+μQt(τ)cos(θ)+Qt(τ)sin(φ+θ)
and
Qr(τ)=−λItc(τ)sin(θ)−μQt(τ)sin(θ)+Qt(τ)cos(φ+θ)

In other words:
Ir(τ)+j*Qr(τ)=λ(cos(θ)−jsin(θ))(Itc(τ)+j*Qt(τ)).

I and Q components of transmission signal g(t) are thus balanced in term of input:
Itc(τ)+j*Qt(τ).

To determine λ and μ, Ir(t) may be expressed as:
Ir(t)=r(t)cos(ωτ)=(It(t)cos(ωt)+Qt(t)sin(ωt+φ))cos(ωτ).

After passing through the low-pass filter Hr(t), where θ=ω(τ−t), Ir(t) may be expressed as:
Ir(t)=It(t)cos(θ)+Qt(t)sin(φ+θ).

Since Ir(t) is generated in analog circuits, τ≈t. Ir(t) may thus be expressed as:
Ir(t)=It(t)+Qt(t)sin(φ).

A function ƒ(x) may be defined as ƒ(x)=|Ir(t)2|.

When Itc(t)=0 and ƒ(x)=0, or Itc(t) and Qt(t) are uncorrelated and the expectation of ƒ(x) is zero, x=φ. Phase φ may be obtained in one or more of a variety of ways.

For example, Itc(t) may be set to zero and an energy of ƒ(x) may be minimized. Where transmit path LPFs238and240are substantially identical to one another and substantially linear, computation of phase φ may be performed in the digital baseband domain. The output signal from reception channel Ie(n) may be utilized as an error signal to update μ, such as illustrated inFIG. 9. Where μ is implemented on a scale, updating may converge relatively quickly. An initial value of μ may be between −sin(π/30) and sin(π/30), such as where phase φ is less than 5 degrees.

Parameter λ, may be determined as. λ=√{square root over (1−μ2)}

FIG. 10is another block diagram of an exemplary configuration to calibrate transmit path frequency independent IQ imbalance. Transmission signal g(t) may be represented as:
g(t)=Is(t)cos(ωt)+Qs(t)sin(ωt).

Referring to method100inFIG. 1, at110, receive path frequency independent I/Q imbalance is calibrated. Calibration at110may be performed subsequent to receive path and transmit path frequency dependent IQ imbalance calibration at104and106, respectively, and subsequent to transmit path frequency independent IQ imbalance calibration at108.

Calibration at110may include configuring the transceiver in a full loop-back mode, operating the transmit and receive path I and Q tuners at corresponding I and Q phases, and determining transmit path summation weights in response to one or more of the corresponding receive path I and Q digital baseband signals, such as described above with respect to108.

InFIG. 2, switches282,284, and286may be placed in position B, and switch288may remain in position A.

FIG. 11is a corresponding signal flow diagram to calibrate receive path frequency independent IQ imbalance, wherein transfer functions1102and1104represent effects of transmit path202and receive path204, respectively.

I and Q components of the corresponding receive path baseband signal may be represented as:
Ir(t)=Ic(t)cos(θ)+Qc(t)sin(θ); and
Qr(t)=−Ic(t)sin(θ−φ)+Qc(t)cos(θ+φ);
where:
Ic(t)=T(t)Is(t)=T(t)Qs(t), and θ is the signal propagation phase.

The propagation phase or delay may be relatively insignificant during tuner calibration but may be arbitrary in (0, 2π) when the calibration signal is from a remote transmitter.

Qr(t) may be further expanded to:
Qr(t)=−Ic(t)(sin(θ)cos(φ)−cos(θ)sin(φ))+Qc(t)(cos(θ)cos(φ)−sin(θ)sin(φ),
or
(Qr(t)−Ic(t)sin(θ))/cos(φ)=−Ic(t)sin(θ)−Qc(t)cos(θ).

which indicates that Ir(t) and {circumflex over (Q)}r(t) are balanced.

Parameters α and β may be set to or defined as:
α=1/cos(φ); and
β=−sin(φ)/cos(φ).

{circumflex over (Q)}r(t) may then be expressed as {circumflex over (Q)}r(t)=αQr(t)+βIc(t), which may be generated in the digital domain as illustrated inFIG. 11, where Ir(t) and {circumflex over (Q)}r(t) correspond to Ic(n) and Qc(n), respectively.

Parameters)λ=cos(φ) and μ=−sin(φ) may be determined as described above with respect to108, and α and β may be determined as:
α=1/λ; and
β=μ/λ.

Methods and systems are disclosed herein with the aid of functional building blocks illustrating the functions, features, and relationships thereof. At least some of the boundaries of these functional building blocks have been arbitrarily defined herein for the convenience of the description. Alternate boundaries may be defined so long as the specified functions and relationships thereof are appropriately performed.

One or more features disclosed herein may be implemented in hardware, software, firmware, and combinations thereof, including discrete and integrated circuit logic, application specific integrated circuit (ASIC) logic, and microcontrollers, and may be implemented as part of a domain-specific integrated circuit package, or a combination of integrated circuit packages. The term software, as used herein, refers to a computer program product including a computer readable medium having computer program logic stored therein to cause a computer system to perform one or more features and/or combinations of features disclosed herein.

While various embodiments are disclosed herein, it should be understood that they have been presented by way of example only, and not limitation. It will be apparent to persons skilled in the relevant art that various changes in form and detail may be made therein without departing from the spirit and scope of the methods and systems disclosed herein. Thus, the breadth and scope of the claims should not be limited by any of the exemplary embodiments disclosed herein.