High voltage waveform generator

Generally, the present invention provides a high voltage waveform generator for use in an ion mobility spectrometer (IMS) that detects trace concentration level ionic species present in a sample gas stream. The present invention consists of a first electromagnetic transformer having a pair of oscillating circuits that are simultaneously excited by a transformer input winding controlled by a controller such as a power semiconductor device. Each oscillating circuit in the pair includes inductive and capacitive components that generate discrete frequency waveforms corresponding to the fundamental and second Fourier harmonic frequencies of an electric signal that approximates an ideal square wave used in creating a transverse electrical field for transport of ion species through an ion mobility spectrometer. The oscillating circuits are electromagnetically coupled to each other. The extent of this electromagnetic coupling can be varied by an inductance juxtapositioned to the first transformer so as to vary the magnetic field coupling the oscillating circuits. The amount of electromagnetic coupling is adjusted by a phase correction circuit to eliminate phase differences between the fundamental and second Fourier harmonic frequencies to ensure that the electrical signal generated by the present invention is as close an approximation of the ideal square voltage waveform as possible. The amplitudes of the fundamental and second Fourier harmonic frequency components of the output waveform are also adjusted by an amplitude correction circuit in such a way as to maintain a constant ratio between them to ensure that the output waveform is correctly shaped for use in the ion mobility spectrometer.

FIELD OF THE INVENTION 
The present invention relates to a high voltage waveform generator for use 
in generating a periodically varying electrical signal to create a 
periodically varying high voltage electrical field in a field ion mobility 
spectrometer. 
BACKGROUND OF THE INVENTION 
Field ion spectrometry (FIS) offers a new method of detecting species 
present at trace (parts per million to parts per billion) concentration 
levels in a sample gas to be analyzed. U.S. Pat. No. 5,420,424, 
incorporated by reference herein, provides an ion mobility spectrometer 
(IMS) for use in detecting trace concentration level species present in a 
sample gas stream. The IMS disclosed in U.S. Pat. No. 5,420,424 utilizes 
periodic high voltage electrical fields to separate different species of 
ions according to the functional dependence of their mobility with 
electric field strength. Ions generated in the ionization chamber of the 
IMS are guided through an ion filter to an ion detector by an asymmetric 
periodic radio frequency (RF) electric field known as the "dispersion 
voltage" that is created between a pair of closely spaced longitudinal 
electrodes located across the ion filter. The displacement of the ions 
induced by the dispersion voltage is modified or compensated by an 
adjustable second time independent electrical potential that is applied 
between the electrodes to isolate a particular ion species for detection 
as a result of the variance in mobility between particular ion species as 
a function of electric field strength. 
The dispersion voltage waveform must be sufficiently high so that the 
electric field created in the IMS will cause the ion mobility values of 
the species selected for analysis to deviate significantly from their low 
electric field values. For electrode spacing on the order of 1 to 3 
millimeters, this requires a dispersion voltage waveform with peak values 
in the 1 to 6 kilovolt (kV) range. The optimum dispersion voltage waveform 
for obtaining the maximum possible ion detection sensitivity on a per 
cycle basis takes the shape of an asymmetric square wave with a zero 
time-averaged value. The power consumption of a conventional electrical 
waveform generator in generating this type of voltage waveform is in 
excess of 100 watts. The generation of asymmetric periodic high voltage 
waveforms is discussed in The International Journal of Mass Spectrometry 
and Ion Processes, Vol. 128. pp. 143-148 (1993); in Russian Inventor's 
Certificate No. 966583; in Devices and Techniques of Experiment, Vol. 4, 
pp. 114-115 (1994); and in Proceedings: Fourth International Workshop on 
Ion Mobility Spectrometry, Aug. 6-9, 1995. 
In order to reduce the power consumption requirements to a level that will 
allow the incorporation of a high voltage waveform generator into a 
portable IMS, it has become necessary to design a waveform generator 
circuit using inductive and capacitive components to produce an output 
voltage waveform that permits input energy storage and recirculation in 
the inductive and capacitive components of the circuit. The present 
invention provides such a waveform generator which produces an output 
voltage waveform that is a two harmonic Fourier series approximation of 
the ideal dispersion voltage square waveform discussed above. In addition, 
the present invention utilizes a unique configuration for the relative 
physical positioning of the inductive components in the circuit that gives 
rise to a unique dual discrete frequency waveform that approximates the 
ideal dispersion voltage waveform as closely as possible. Finally, the 
invention provides circuitry which ensures phase and amplitude 
stabilization of this dual discrete frequency output voltage waveform. 
Accordingly, the present invention provides a high voltage waveform 
generator that uses inductive and capacitive components to produce an 
oscillating output voltage. 
Preferably, the high voltage waveform generator permits input energy 
storage and recirculation in the inductive and capacitive components of 
the circuit so as to produce an output voltage waveform that is a two 
harmonic Fourier series approximation of an ideal asymmetric periodic high 
voltage square waveformn. 
The present invention also preferably provides a unique physical 
configuration for the positioning of the inductive components in the 
circuit which gives rise to the dual discrete frequencies of the output 
voltage waveform. 
The present invention also preferably provides circuitry which ensures 
phase and amplitude stabilization of the output voltage waveform. 
SUMMARY OF THE INVENTION 
Generally, the present invention provides a high voltage waveform generator 
for use in an ion mobility spectrometer (IMS) that detects trace 
concentration level species present in a sample gas stream. The present 
invention consists of a first electromagnetic transformer having a pair of 
oscillating circuits that are simultaneously excited by a transformer 
input winding controlled by a controller such as a power semiconductor 
device. Each oscillating circuit in the pair includes inductive and 
capacitive components that generate discrete frequency waveforms 
corresponding to the fundamental and second Fourier harmonic frequencies 
of an electric signal that approximates an ideal square wave used in 
creating a transverse electrical field for transport of ion species 
through an ion mobility spectrometer. The oscillating circuits are 
electromagnetically coupled to each other. The extent of this 
electromagnetic coupling can be varied by an inductance juxtapositioned to 
the first transformer so as to vary the magnetic field coupling the 
oscillating circuits. The amount of electromagnetic coupling is adjusted 
by a phase correction circuit to eliminate phase differences between the 
fundamental and second Fourier harmonic frequencies to ensure that the 
electrical signal generated by the present invention is as close an 
approximation of the ideal square voltage waveform as possible. The 
amplitudes of the fundamental and second Fourier harmonic frequency 
components of the output waveform are also adjusted by an amplitude 
correction circuit in such a way as to maintain a constant ratio between 
them to ensure that the output waveform is correctly shaped for use in the 
ion mobility spectrometer. 
Other details, objects, and advantages of the present invention will become 
apparent in the following description of the presently preferred 
embodiment.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
FIG. 1 shows a schematic electrical circuit diagram of a preferred 
embodiment of the present invention. The circuit shown in FIG. 1 
preferably generates a radio frequency (RF) electrical voltage signal 
output corresponding to the periodic waveform Vout(t) shown in FIG. 2B 
across the first and second electrodes 21 and 22 of the ion mobility 
spectrometer described in U.S. Pat. No. 5,420,424, which is incorporated 
by reference herein and shown in FIG. 3. This output voltage signal 
Vout(t) is the periodic asymmetric potential referred to in U.S. Pat. No. 
5,420,424, and it creates a periodically varying electric field across the 
electrical capacitance formed by electrodes 21 and 22 which guides an ion 
species across the analytical gap 25 from the ionization chamber 28 to the 
ion detector 40. The preferred range of output voltages generated by the 
circuit of FIG. 1 is 1 (one) to 6 (six) kilovolts (kV). 
The output voltage waveform Vout(t) shown in FIG. 2B is the fundamental and 
second harmonic Fourier series approximation of the ideal dispersion 
voltage waveforn Vdis(t) shown in FIG. 2A. The ideal dispersion voltage 
waveform Vdis(t) represents the optimum shape of the periodic asymmetric 
potential applied across electrodes 21 and 22 for obtaining the maximum 
possible detection sensitivity of an ion species by the ion detector 40. 
This ideal dispersion voltage waveform Vdis(t) can be expressed 
mathematically by the following characteristics: 
##EQU1## 
Due to the large input power requirements for generating the ideal 
dispersion voltage waveform Vdis(t) shown in FIG. 2A, its shape is 
approximated by the two harmonic Fourier series output voltage waveform 
Vout(t) in a preferred embodiment of the present invention. Vout(t) 
permits input energy storage and recirculation in the inductive and 
capacitive components of the circuit of FIG. 1, drastically reducing the 
input power requirements for generating the desired dispersion voltage. 
The output voltage waveform Vout(t) as shown in FIG. 2B can be 
characterized by the combination of two component waveforms with discrete 
frequencies that obey the following mathematical expression: 
EQU Vout(t)=Vfund,max(cos(wt))+Vharm,max(cos(2wt+.phi.)) (1) 
with a) a fundamental frequency component w having maximum amplitude 
Vfund,max 
b) a second harmonic frequency component 2w having maximum amplitude 
Vharm,max 
c) a phase difference .O slashed. between the fundamental and second 
harmonic frequency components 
The fundamental frequency w and second harmonic frequency 2w of the output 
voltage waveform Vout(t) are respectively set by the electrical inductance 
and capacitance combinations L1/C1 and L2/C2 in the circuit of FIG. 1. 
These inductance/capacitance combinations are preferably series circuit 
connections that form "tank circuits" 1 and 2 that are electromagnetically 
coupled by a principal transformer 10 into a pair of dual resonance 
oscillation circuits that each simultaneously resonate at the frequencies 
given in Equation (1). The entire output voltage waveform Vout(t) appears 
across each Inductance L1 and L2 while capacitance C1 represents the 
capacitance formed by electrodes 21 and 22 in the ion mobility 
spectrometer of FIG. 3. Thus, the output voltage waveform Vout(t) is 
applied across electrodes 21 and 22 by the voltage created across 
inductance L1 to operate the ion mobility spectrometer. 
Inductance/capacitance combination L2/C2 applies the output voltage 
waveform Vout(t) to control circuits that adjust for phase and amplitude 
variations in the waveform as described below. The output voltage waveform 
Vout(t) fundamental and second harmonic frequencies w and 2w, 
respectively, are set according to the following expressions: 
EQU w=(w1.sup.2 +w2.sup.2 -((w1.sup.2 +w2.sup.2).sup.2 -4w1.sup.2 w2.sup.2 
(1-k.sup.2)).sup.1/2)/2(1-k.sup.2) !.sup.1/2 (2) 
EQU 2w=(w1.sup.2 +w2.sup.2 +((w1.sup.2 +w2.sup.2).sup.2 -4w1.sup.2 w2.sup.2 
(1-k.sup.2)).sup.1/2)/2(1-k.sup.2)!.sup.1/2 (2) 
where: 
w1=1/L1*C1!.sup.1/2 
w2=1/L2*C2!.sup.1/2 
As shown in FIG. 4, the tank circuits 1 and 2 are preferably located in 
separate sections of a principal transformer 10, preferably torodially 
shaped, which forms an electromagnetic coupling between the tank circuits 
1 and 2 that can be characterized by a coupling coefficient k. Principal 
transformer 10 preferably has a pot core made of any conventional 
ferrimagnetic material, such as the material 3F3, with a gap 11 to 
separate the sections housing the respective tank circuits 1 and 2. The 
coupling coefficient k is initially set by the physical positioning of 
excitation inductance L0 in relation to L1 and L2 inside the principal 
transformer 10 housing as shown in FIG. 4. The ideal physical positioning 
of L0 relative to L1 and L2 is so as to generate the dual discrete 
fundamental frequency w and second harmonic frequency 2w waveforms, where 
w and 2w are given by Equations (2) and (2a), respectively. If L0 is 
positioned equidistant from L1 and L2, only w will be generated. A 
difference in the relative positioning of L0 with respect to L1 and L2, 
respectively, will generate the dual discrete frequencies given by 
Equations (2) and (2a). The extent of electromagnetic coupling k between 
tank circuits 1 and 2 also determines the amount of phase difference .O 
slashed. existing between the fundamental and second harmonic waveforms w 
and 2w, respectively. This phase shift elimination is determined from 
Equations (2) and (2a) to occur at a coupling coefficient value k=0.6. A 
coupling coefficient k of 0.6 ensures the closest possible approximation 
of Vout(t) to the optimum dispersion voltage waveform Vdis(t). 
Variations in the ambient temperature and self-heating of circuit 
components tend to shift the values of the inductances and thus the extent 
of electromagnetic coupling k between the tank circuits 1 and 2 as the 
circuit is operated. This in turn will give rise to a phase difference .O 
slashed. between the fundamental frequency w and second harmonic 2w 
frequency waveforms making up the output voltage waveform Vout(t). The 
elimination of this phase difference .O slashed. is critical to the proper 
approximation of the ideal dispersion voltage waveform Vdis(t) shown in 
FIG. 2A by the output voltage waveform Vout(t) shown in FIG. 2B. As shown 
in FIGS. 1 and 4, a separate inductive coil 12 surrounding a ferrimagnetic 
material is preferably provided with a feedback inductance of value Ls 
that adjusts (or "fine tunes") the extent of electromagnetic coupling k 
between L1 and L2. This feedback inductor 12 has a flat surface that is 
positioned next to principal transformer 10 such that the center of 
feedback inductor 12 is aligned with the center of the gap 11 in principal 
transformer 10. The amount of current through feedback inductance Ls is 
adjusted to "fine tune" the coupling coefficient k between L1 and L2 to 
eliminate any phase difference .O slashed. created between the fundamental 
frequency w and second harmonic frequency 2w waveforms during operation of 
the circuit. 
The amount of current through feedback inductance Ls is preferably 
controlled by the phase correction circuit 3 shown in FIG. 1. Vout(t) is 
input to the phase correction circuit 3 through a current transformer 13 
which can be connected in series with the inductance/capacitance 
combination of either tank circuit 1 or 2. In FIG. 1, the current 
transformer 13 is connected in series to inductance/capacitance 
combination L2/C2 in tank circuit 2. Reflecting the current flowing 
through tank circuit 2 through current transformer 13 produces a signal 
V'out(t), shown in FIG. 2C, which has a maximum amplitude V'out,max at the 
points where the output voltage signal Vout(t) is changing at a maximum 
rate. 
Referring to FIGS. 1 and 2C, the current transformer 13 electromagnetically 
couples V'out(t) to a pair of peak detector circuits 4 and 5 which detect 
the peak magnitudes of V'out(t) as it oscillates between opposite polarity 
maximum and minimum points. Each peak detector circuit 4 or 5 is 
respectively comprised of a diode D4 or D5 in combination with a commonly 
grounded charging capacitor C4 or C5. Diode D4 or D5 acts as a gate to 
allow charging of its respective capacitor C4 or C5 during successive 
opposite polarities of V'out(t). The accumulated charge on capacitor C4 
will thus be proportional to the maximum positive amplitude of 
V'out(t)=+V'out,max while the accumulated charge on capacitor C5 will be 
proportional to the maximum negative amplitude of V'out(t)=-V'out,max 
during one complete cycle of V'out(t). The net output voltage Vsum from 
the peak detector circuits 4 and 5 is obtained by measuring the combined 
voltage across the commonly grounded capacitors C4 and C5 and will be 
proportional to the net sum of the maximum positive amplitude +V'out,max 
and the maximum negative amplitude -V'out,max in any given cycle of 
V'out(t). As can be seen from FIG. 2C, when no phase difference exists 
between the fundamental frequency w and second harmonic frequency 2w 
components of Vout(t), the net output voltage Vsum of the peak detector 
circuits 4 and 5 will be zero. When a positive phase difference (.O 
slashed.=+30.degree.) exists, the net output voltage Vsum will be 
positive. When a negative phase difference (.O slashed.=-30.degree.) 
exists, the net output voltage Vsum will be negative. 
In either case, the net output voltage Vsum of the peak detector circuits 4 
and 5 is fed through a variable resistance device R1 such as a 
potentiometer or a rheostat to the negative input of a conventional 
operational amplifier 6 that is configured to operate as a summing 
amplifier. The output of operational amplifier 6 is fed back through a 
conventional current amplifying transistor 7 to Ls. R1 provides a means 
for calibrating the input signal Vsum to the operational amplifier 6. The 
feedback signal provided by operational amplifier 6 is a direct current 
(DC) signal that is proportional to the net output voltage Vsum of the 
peak detector circuits 4 and 5. If the net output voltage Vsum is zero 
(indicating a zero phase difference between the fundamental frequency w 
and the second harmonic frequency 2w of Vout(t)) then no feedback signal 
is provided to Ls and as a result no change in the coupling coefficient k 
between L1 and L2 takes place. If the net output voltage Vsum is positive 
(indicating a positive phase difference between the fundamental frequency 
w and the second harmonic frequency 2w of Vout(t)), the feedback signal 
operates to decrease the amount of current through Ls to adjust the 
coupling coefficient k to a higher value thereby increasing the extent of 
electromagnetic coupling between L1 and L2 to eliminate the phase 
difference. If the net output voltage Vsum is negative (indicating a 
negative phase difference between the fundamental frequency w and the 
second harmonic frequency 2w of Vout(t)), the feedback signal operates to 
increase the amount of current through Ls to adjust the coupling 
coefficient k to a lower value thereby decreasing the extent of 
electromagnetic coupling between L1 and L2 to eliminate the phase 
difference. 
In addition to the elimination of phase differences between the fundamental 
frequency w and second harmonic frequency 2w components, variations in the 
ratio between the maximum amplitudes Vfund,max and Vharm,max of the 
fundamental and second harmonic waveforms, respectively, must be 
eliminated to ensure the closest possible approximation of Vout(t) to the 
optimum dispersion voltage waveform Vdis(t). These maximum amplitudes 
Vfund,max and Vharm,max have a ratio that is also initially set by the 
physical positioning of excitation inductance L0 in relation to L1 and L2 
in principal transformer 10. This ratio obeys the following expression: 
EQU Vfund,max/Vharm,max=(sin(a)cos(2a)-2sin(a)cos(2a))/3(a-sin(a)cos(a))(3) 
where: 
a=(*.pi.)/T as shown in FIG. 5. 
The maximum amplitudes Vfund,max and Vharm,max can be made to vary by 
adjusting the amount of current I0 passing through excitation inductance 
L0. The excitation inductance L0 provides input power from voltage source 
Vcc to excite the tank circuits 1 and 2. The amount of current I0 passing 
through L0 is controlled by a controller, preferably a power semiconductor 
8, which activates to allow L0 to excite the tank circuits 1 and 2 and 
which deactivates to cut off input power to L0 and the tank circuits 1 and 
2. Any conventional power semiconductor can be used for this purpose, such 
as a power metal-oxide field effect transistor (MOSFET) or a power 
bipolar-junction transistor (BJT). Power semiconductor 8 is in turn driven 
by a gating inductance Lf, also housed within principal transformer I0 as 
shown in FIG. 4, which applies an activating signal V"out(t) between the 
gate and source of the power semiconductor 8 that mirrors Vout(t). 
As shown in FIG. 5, the activating signal V"out(t) controls the period of 
time during which current I0 passes through excitation inductance L0 by 
controlling the on-time of the power semiconductor 8. The on-time is in 
turn controlled by the gating voltage Vg. Gating voltage Vg is an 
adjustable voltage level that must exceed the intrinsic threshold voltage 
Vthresh of the power semiconductor 8 in order for the power semiconductor 
8 to conduct. Vg is set at a level which will ensure that the on-time of 
the power semiconductor 8 is within a range that will provide a nearly 
constant value for the ratio between the maximum amplitudes Vfund,max and 
Vharm,max of the fundamental w and. second harmonic 2w waveforms given in 
Equation (3). 
The activating signal V"out(t) provided by the gating inductance Lf is 
controlled by the amplitude correction circuit 9 shown in FIG. 1 The 
amplitude correction circuit 9 contains two cascaded operational 
amplifiers 14 and 15 that operate in tandem as a differential amplifier 
having two inputs A and B. The operational amplifier configuration in the 
amplitude correction circuit 9 can consist of one or more than one 
conventional operational amplifiers similar to that used in the phase 
correction circuit 3. The inputs A and B to the amplitude correction 
circuit 9 are taken from the peak detectors 4 and 5. The voltage 
+V'out,max across capacitor C4 is provided to one input A while the 
voltage -V'out,max across capacitor C5 is simultaneously provided to the 
opposite input B. The difference between these two voltages Vdiff is then 
compared to a setpoint value Vset which is adjusted by variable resistance 
device R2 to set the gating voltage Vg of the power semiconductor 8 to the 
desired level. 
As shown in FIGS. 1 and 5, the magnitude of gating voltage Vg relative to 
the threshold voltage Vthresh of the power semiconductor 8 controls the 
amount of current Ids passing through the power semiconductor 8 and thus 
the amount of current I0 passing through excitation inductance L0. If Vg 
is increased, I0 will increase, causing an increase in the activating 
signal V"out(t) to the power semiconductor 8. By virtue of the increased 
current I0 through excitation inductance L0, the amplitudes Vfund,max and 
Vharm,max of the fundamental w and second harmonic 2w waveforms will have 
increased. The peak detectors 4 and 5 will detect this increase, causing 
Vdiff to increase. At the same time, the increase in V"out(t) will cause 
an increased charging of the capacitance Cf in the gating inductance Lf 
circuit. This increased charge on Cf will in turn decrease gating voltage 
Vg, keeping the on-time of power semiconductor 8 and thus the ratio of 
Vfund,max to Vharm,max essentially unchanged. 
Values and models of circuit components used in a preferred embodiment of 
the invention shown in FIG. 1 are as follows: 
TABLE 1 
______________________________________ 
C0 0.1 .mu.F (microfarads) 
C2 50 pF (picofarads) 
C3 0.01 .mu.F 
C4 1000 pF 
C5 1000 pF 
C6 0.1 .mu.F 
C7 0.2 .mu.F 
C8 0.1 .mu.F 
C9 0.1 .mu.F 
C10 0.1 .mu.F 
Cf 0.1 .mu.F 
R1 10 k.OMEGA. (kiloohms) 
R2 20 k.OMEGA. 
R3 100 k.OMEGA. 
R4 10 k.OMEGA. 
R5 10 k.OMEGA. 
R6 10 k.OMEGA. 
R7 750 .OMEGA. (ohms) 
R8 100 .OMEGA. 
R9 2 k.OMEGA. 
R10 100 k.OMEGA. 
R11 1 M.OMEGA. (megohm) 
R12 1 M.OMEGA. 
R13 5 k.OMEGA. 
R14 1 M.OMEGA. 
R15 100 k.OMEGA. 
R16 10 k.OMEGA. 
R17 1 M.OMEGA. 
D4 1N5711 (model number diode) 
D5 1N5711 
D6 1N4148 
Df 1N4148 
L0 2 (number of coil turns) 
L1 250 
L2 250 
Ls 3000 
Lf 1 
6 LF412A (op amp model no.) 
14 LF412A 
15 LF412A 
7 2N3904 (BJT model no.) 
8 RFP2N08 (MOSFET model no.) 
______________________________________ 
While a presently preferred embodiment of practicing the invention has been 
shown and described with particularity in connection with the accompanying 
drawings, the invention may otherwise be embodied within the scope of the 
following claims.