Bandgap voltage reference circuit

A bandgap voltage reference circuit includes a feedback controlled current mirror, a bandgap voltage generator, and a voltage comparator. The current mirror, in response to a feedback control signal from the voltage comparator, generates a controllable reference current. The bandgap voltage generator generates two reference voltages based upon conduction of the reference current from the current mirror through two PN diodes having different emitter areas. The voltage comparator compares the two reference voltages and generates the feedback control signal for the current mirror. Such a bandgap voltage reference circuit simultaneously generates a bandgap voltage reference and a current mirror reference while also being operable over a wide power supply voltage range and down to very low power supply voltage values.

BACKGROUND OF THE INVENTION 
1. Field of the Invention 
The present invention relates to a bandgap voltage reference generator 
circuit which operates over a wide supply voltage range and consumes 
little supply current. 
2. Description of the Related Art 
Bandgap voltage references are well known for obtaining a reference voltage 
that is relatively constant over a substantial temperature range. The 
basic concept is to combine two potentials, one having a positive 
temperature coefficient and one having a negative temperature coefficient. 
The sum of these two potentials is made equal to the semiconductor bandgap 
potential extrapolated to absolute zero temperature. For silicon, this 
value is close to 1.2 volts. 
Typically, the negative temperature coefficient potential is obtained from 
a forward-biased PN junction, i.e., the emitter-base junction in a 
conducting transistor operated at a current that will produce a voltage 
drop of about 600 mV at 300.degree. K. This voltage has a negative 
temperature coefficient of about 2 mV/.degree.C. The positive temperature 
coefficient is obtained from a .DELTA.V.sub.BE -producing circuit that 
develops a 600 mV potential at about 300.degree. K. This voltage has a 
positive temperature coefficient of about 2 mV/.degree.C. Thus, when these 
two voltages are combined at 300.degree. K, a 1.2 V potential is produced 
with close to zero temperature coefficient. 
The .DELTA.V.sub.BE potential is typically produced by operating a pair of 
transistors or diodes at substantially different current densities. This 
can be done by ratioing the transistor or diode areas and passing equal 
currents, or by using matched area devices and ratioing the currents. If 
desired, a combination of transistor size and current ratioing can be 
employed. The low-current-density transistor includes a series resistor. 
The two devices are equivalently connected in parallel so that the 
differential voltage drop (.DELTA.V.sub.BE) appears across the resistor. 
Typically, at 300.degree. K and a current-density ratio of 10, the 
.DELTA.V.sub.BE will be about 60 mV. This value, when multiplied by 10, 
produces a voltage of about 600 mV having a positive temperature 
coefficient. 
SUMMARY OF THE INVENTION 
A bandgap voltage generator in accordance with the present invention 
provides a highly stable temperature-constant bandgap voltage reference 
circuit that simultaneously generates a bandgap voltage reference and a 
current mirror reference and operates over a wide power supply voltage 
range and down to very low power supply voltage values. 
In accordance with one embodiment of the present invention, a bandgap 
voltage generator includes a current source, a current amplifier, a 
voltage generator and a voltage comparator. The current source is 
configured to receive a control signal and in accordance therewith provide 
an input current which is adjustable in accordance with the control 
signal. The current amplifier is coupled to the current source and is 
configured to conduct the input current and in accordance therewith 
conduct an output current which is proportional to the input current. The 
voltage generator is coupled to the current amplifier, includes first and 
second PN junction devices having first and second current densities, and 
is configured to receive the output current and in accordance therewith 
generate first and second voltages. The first and second voltages are 
approximately proportional to the output current and are dependent upon 
the first and second current densities, respectively. The voltage 
comparator is coupled to the voltage generator and the current source and 
is configured to receive and compare the first and second voltages and in 
accordance therewith provide the control signal such that the first and 
second voltages are equal. 
In accordance with another embodiment of the present invention, a bandgap 
voltage generator includes a reference voltage generator, a voltage 
converter and a voltage comparator. The reference voltage generator 
includes a controllable shunt circuit which is configured to receive a 
control signal, generate a shunt current which is adjustable in accordance 
with the control signal and generate a reference voltage which is 
adjustable in accordance with the shunt current. The voltage converter is 
coupled to the reference voltage generator, includes first and second PN 
junction devices having first and second current densities, and is 
configured to receive the reference voltage and generate first and second 
voltages which are dependent upon the first and second current densities, 
respectively, and are adjustable in accordance with the reference voltage. 
The voltage comparator is coupled to the voltage converter and the 
reference voltage generator and is configured to receive and compare the 
first and second voltages and in accordance therewith provide the control 
signal such that the first and second voltages are equal. 
These and other features and advantages of the present invention will be 
understood upon consideration of the following detailed description of the 
invention and the accompanying drawings.

DETAILED DESCRIPTION OF THE INVENTION 
To aid explanation and understanding, where components are similar between 
figures, the same identifying label has been used. Where components of 
different diagrams perform similar functions, but vary in dimensions, they 
are identified with the number of the figure as the most significant digit 
and with identical least significant digits in the identifying label. 
FIG. 1 illustrates a simplified schematic diagram of the .DELTA.V.sub.BE 
portion of a bandgap voltage reference circuit 100 with a high gain 
feedback loop driven by a voltage output comparator 10 in accordance with 
the present invention. In the FIG. 1 circuit, the currents I.sub.a and 
I.sub.b flow in PN diodes 12 and 13, respectively. As shown in FIG. 1, the 
emitter area of diode 13 is 10 times that of diode 12. Voltage V.sub.a is 
developed across diode 12 and appears at circuit node 15. Voltage V.sub.b 
is developed across the series combination of diode 13 and resistor R4 so 
that this voltage appears at circuit node 16. Resistors R2 and R3 function 
primarily to determine the levels of currents I.sub.b and Ia, 
respectively, which, in the preferred embodiment of the invention, are 
made equal. As shown in FIG. 1, and described in detail below, and in 
accordance with the present invention, a voltage output comparator 10 has 
its differential inputs connected to nodes 15 and 16 and its voltage 
output 11 connected to control the current input I.sub.s shown in FIG. 1. 
In FIG. 1, a current reference circuit made up of p-channel reference 
transistor P1, p-channel current mirror transistor P2, n-channel current 
reference transistor N1, n-channel current mirror transistor N2, and 
current reference circuit resistor R1 establish a reference current I. The 
reference current I in transistor P1 is mirrored and ratioed in p-channel 
current mirror transistor P3 to control current I.sub.s which is the 
current source for the resistor and divider network of the bandgap voltage 
reference circuit. Resistor R1 is sized such that the reference current I 
established by the current reference circuit is insufficient to achieve 
balance between voltages V.sub.a and V.sub.b, at circuit nodes 15 and 16 
respectively, which are input to voltage output comparator 10. 
The comparator 10 monitors the voltages V.sub.a and V.sub.b in the bandgap 
circuit and generates feedback control signal 11 which controls the 
conductance of a current shunt transistor N3. Transistor N3, in turn, 
produces a current I.sub.3 which shunts the resistor R1 in the current 
reference. By varying the magnitude of I.sub.3, the current in transistor 
P1 and the mirror current I.sub.s in transistor P3 can be modified which 
controls the currents I.sub.a and I.sub.b in the two branches of the 
bandgap voltage reference. The voltages V.sub.a and V.sub.b vary in 
response to currents I.sub.a and I.sub.b, respectively, to complete the 
feedback loop. The feedback causes the shunt transistor N3 to conduct the 
correct amount of current to balance voltages V.sub.a and V.sub.b at the 
inputs to comparator 10. The high sensitivity of the shunt transistor N3 
to the voltage output 11 from comparator 10 gives the feedback loop very 
high gain. 
The schematic for an embodiment of the bandgap voltage reference circuit 
100 is shown in FIG. 2. In this circuit, the resistors R2, R3 and R4 in 
the bandgap branches are formed by four smaller resistors R200, R202, R203 
and R204 to save die area. A startup transistor N210 is shown that is 
driven by a Startup signal to ensure that current flow is established in 
the current reference circuit. Diode 13 is fabricated as a plurality of 
ten diode-connected transistors of the same size as diode-connected 
transistor 12 connected in parallel to produce the equivalent of a large 
diode with a PN junction area ten times that of diode-connected transistor 
12. 
An embodiment of the voltage comparator 10 is also shown in FIG. 2. Input 
transistors 224 and 226 are native devices connected together at their 
sources and also to the drain of current mirror transistor 228. Current 
mirror transistor 228 has its gate connected to an n-channel current 
reference voltage Nmr which is obtained from the current reference 
circuit, as shown here, or from an external current reference, as 
described below. The input transistors 224 and 226 are connected to load 
transistors 220 and 222 respectively. The load transistors 220 and 222 are 
driven by p-channel current reference voltage Pmr which is obtained from 
the current reference circuit, as shown here, or from an external current 
reference. 
Branching off from the drain of each input transistor 224 and 226 is a 
clamped-active p-channel transistor (230 and 234, respectively) in series 
with an n-channel transistor (232 and 236, respectively). The 
clamped-active transistors 230 and 234 each have their gate connected to 
the ground potential and function to maintain the input transistors in 
their active region by preventing the drain voltage of the input 
transistors from falling below one p-channel threshold voltage. N-channel 
transistor 236 is diode connected and connected to the gate of transistor 
232 which will consequently mirror the current in transistor 236. 
At balance, the current reference voltages will induce a current of 
magnitude I in transistor 228 and in each of load transistors 220 and 222. 
Because transistor 228 can only sink current I, the remaining current 
sourced by load transistors 220 and 222 must be sunk by the branch legs 
off of each input transistor 224 and 226. With the circuit at balance, I/2 
flows through the branch leg including clamped-active transistor 234 and 
diode connected transistor 236. Current mirror transistor 232 mirrors the 
current in transistor 236 so that I/2 flows in clamped-active transistor 
230 and transistor 232. 
When the voltages V.sub.a and V.sub.b are not balanced, then the resulting 
difference in input voltage to transistors 224 and 226 alters the currents 
in the branch legs to produce an output voltage signal 11 at the drain of 
transistor 232. For example, when V.sub.b is greater than V.sub.a, input 
transistor 224 conducts more current than input transistor 226. Because 
current sink transistor 228 maintains a constant current level, the 
additional current passing through transistor 24 must be counterbalanced 
by reduced current flow in input transistor 226. The reduction in current 
through input transistor 226 results in more current flowing into 
transistor 234 and diode connected transistor 236. The current in 
transistor 236 is mirrored by transistor 232 which is driven harder while, 
simultaneously, there is less current flowing through transistor 230 
because of the additional current drawn by input transistor 224. The 
consequence is that the output voltage 11 at the drains of transistors 230 
and 232 drops in proportion to the amount by which V.sub.b exceeds 
V.sub.a. 
Conversely, when V.sub.a is greater than V.sub.b, the relationship between 
the relative currents in the branch legs of the comparator is reversed. 
Input transistor 224 conducts less current resulting in more current 
flowing through input transistor 226. As a consequence, less current flows 
in transistor 234 and diode connected transistor 236 resulting in less 
current draw in mirror transistor 232. At the same time, because input 
transistor 224 is conducting less current, more current is available 
through transistor 230 and the output voltage 11 rises in proportion to 
the difference between V.sub.a and V.sub.b. 
The circuit nodes 15 and 16 of the bandgap voltage reference are connected 
to the input transistors 224 and 226 of the voltage comparator. The 
voltage output control signal 11 of the comparator drives the gate of 
shunt transistor N3 to control the current in transistor N1 and, 
consequently, also the current in transistor P3 to form the high gain 
feedback loop described above. The signal V.sub.ref is the stable voltage 
reference level generated by the circuit 200. 
The circuit of FIG. 2 may be formed using CMOS technology employing the 
following components: 
______________________________________ 
Component Value/Size (W/L in Microns) 
______________________________________ 
Resistor R1 50 K ohms 
Resistor R200 22 K ohms 
Resistors R202 and R203 
44 K ohms 
Resistor R204 10 K ohms 
Transistor N1 20/5 
Transistors 224, 226 
25/5 
Transistors N2, P1, P2, 220, 222, 228 
10/5 
Transistors 230, 234 
3/1 
Transistor N3 5/2 
Transistors 232, 236 
5/5 
Transistor 210 3/5 
Transistor P3 50/5 
______________________________________ 
Transistors 224 and 226 are constructed to have low (about 0.2 volt) 
thresholds. The nominal operating power supply voltage range is 1.5-6.0 
volts. When the circuit is stable (i.e., balance is achieved), the current 
in transistor P3 is 12 microamperes. Hence, the current "I" in transistor 
P1 is 2.4 microamperes (i.e., 10/50 of 12 microamperes). 
In FIG. 3, a simplified schematic of a bandgap voltage reference circuit 
300 with an externally controlled current source S1 is shown. When current 
reference voltages are available from an external source, then the 
reference circuit 300 may be employed which is simplified version of the 
bandgap voltage reference circuit 100 of FIGS. 1 and 2 and requires no 
startup signal. 
The current sink S1 sinks a current I which, as was the case above, induces 
current I.sub.s in transistor P3 that is less than the current necessary 
to balance voltages V.sub.a and V.sub.b in the two legs of the bandgap 
reference circuit. Voltages V.sub.a and V.sub.b are input to comparator 10 
which produces voltage output control 11 that drives shunt transistor N3 
in order to vary the current I.sub.3 and thereby form the high gain 
feedback loop discussed above. 
FIG. 4 is a detailed schematic of a bandgap voltage reference 400 which 
implements the design of FIG. 3. The current source S1 is driven by 
externally supplied n-channel current reference signal Nmr which also 
drives current sink transistor 428 in the voltage comparator circuit. 
Similarly, externally supplied p-channel current reference signal Pmr 
drives load transistor 420 and 422 that source constant currents in the 
two branches of the voltage comparator circuit. The current I in current 
sink S1 combines with the current I.sub.3 in shunt transistor N3 to 
determine the current in transistor P1 and, consequently, control the 
current I.sub.s supplied to the two branches of the bandgap voltage 
reference circuit by transistor P3 and which ultimately determines the 
level of the voltage V.sub.a and V.sub.b input to the comparator. The 
current I.sub.3 in shunt transistor N3 is controlled by the voltage output 
control signal 11 from the comparator circuit to form the high gain 
feedback loop that controls the bandgap voltage circuit 400. The output 
signal V.sub.ref is the stable voltage reference output generated by the 
circuit 400. 
The circuit of FIG. 4 may be formed using CMOS technology employing the 
following components: 
______________________________________ 
Component Value/Size (W/L in Microns) 
______________________________________ 
Resistor R400 22 K ohms 
Resistors R402 and R403 
44 K ohms 
Resistor R404 10 K ohms 
Transistor S1 5/10 
Transistors 424, 426 
25/5 
Transistor P1 10/5 
Transistors 430, 434 
3/1 
Transistor N3 5/2 
Transistors 420, 422, 428, 432, 436 
5/5 
Transistor P3 100/5 
______________________________________ 
Transistors 424 and 426 are constructed to have low (about 0.2 volt) 
thresholds. The nominal operating power supply voltage range and current 
are 1.5-6.0 volts and 15.2 microamperes, respectively. When the circuit is 
stable (i.e., balance is achieved), the current in transistor P3 is 12 
microamperes. Hence, the current in transistor P1 (i.e., the sum of 
currents "I" and "I.sub.3 " in transistors S1 and N3, respectively) is 1.2 
microamperes (i.e., 10/100 of 12 microamperes). 
It should be understood that various alternatives to the embodiments of the 
invention described herein may be employed in practicing the invention. It 
is intended that the following claims define the scope of the invention 
and that methods and circuits within the scope of these claims and their 
equivalents be covered thereby.