Integrated circuitry for exchanging signals between telephone station and central office

The two talking wires of a line loop extending from a subscriber station to a central office, provided with respective blocking capacitors, are connected on the subscriber side of these blocking capacitors to respective signal detectors fed from a common battery and cross-coupled by a common compensating capacitor. The two signal detectors receive outgoing signals from the central-office equipment via a common photoelectric coupler and transmit incoming signals to that equipment by way of respective bistable threshold comparators and individal photoelectric couplers. The two signal detectors with their threshold comparators are constituted in integrated circuity by mutually complementary transistors and diodes together with associated resistors.

FIELD OF THE INVENTION 
My present invention relates to integrated circuitry serving to exchange 
signals between a subscriber station of a telephone system and an 
associated central office. 
BACKGROUND OF THE INVENTION 
In present-day telephone systems, the subscriber stations are generally 
energized by a central battery connected across the talking conductors of 
their respective line loops whereby, upon closure of a line loop by the 
subscriber's hook switch, a direct current circulates over these 
conductors to mark the line busy and to convey switching signals such as 
dial pulses. The switching equipment responding to these signals at the 
central office is connected to the talking conductors, at points lying on 
the subscriber side of respective blocking capacitors or transformers, by 
circuit elements having a low resistance to direct current but a high 
impedance for alternating current in order to prevent the transmission of 
voice frequencies between lines supplied by the same battery. The high a-c 
impedance is normally an inductance which in many instances is constituted 
by the winding of an associated busy relay. 
In contrast to resistances and capacitances, an inductance is not readily 
realizable in integrated circuitry and must therefore generally be 
designed as a coil which, aside from occupying considerable space, has to 
be protected against dust and moisture; such coils, moreover, are 
relatively expensive on account of the considerable quantity of copper 
required for them. 
Efforts have therefore already been made to replace the conventional line 
inductances of telephone systems by electronic components of more or less 
equivalent characteristics. Thus, the use of constant-current generators 
has been proposed (Zurich Seminar 1976 and 1978, Paper Co. C7) as a means 
for suppressing current alternations and providing a direct current of a 
magnitude nearly independent of line length. Drawbacks of this method 
include the difficulty of obtaining a perfect balance between the current 
generators connected to the two line conductors as well as their 
incompatibility with variable-gain electronic telephones in which the 
intensity of voice-frequency currents is adjusted in response to 
direct-current magnitude to compensate for differences in line length. 
According to another proposal (Zurich Seminar 1976, Paper No. C8), 
inductances are simulated by integrated circuits comprising two resistors 
in series. This circuitry is rather complex and consumes considerable 
power (about 0.6 watt). It also requires the insertion of a small 
transformer in the line loop to reduce interference by spurious signals of 
the so-called common-mode type which propagate cophasally along the two 
line conductor to their common ground return but which are not transmitted 
to the secondary of that transformer. 
A transistor circuit has also been described (Zurich Seminar 1976, Paper 
No. C7) which simulates a busy relay as concerns response to line current 
and inductive behavior. Such a circuit, however, has a nonlinear 
voltage/current characteristic especially in the low-voltage range and is 
sensitive to current reductions and noise pulses which may be 
misinterpreted as switching signals. 
OBJECTS OF THE INVENTION 
The general object of my present invention, therefore, is to provide 
improved integrated circuitry for the transmission of signals between a 
subscriber station and central-office equipment with avoidance of the 
above-discussed inconveniences. 
A more particular object is to provide circuitry of this description which 
is readily adaptable to modifications in subscribers' telephone apparatus 
yet can be easily substituted for conventional circuit components in 
existing central offices. 
It is also an object of my invention to provide signal-transmitting 
circuitry for the purpose set forth which has a low impedance for cophasal 
signals of the common-mode type for effectively shunting these signals to 
ground ahead of a pair of blocking capacitors serving to transmit the 
voice currents between the associated subscriber station and another 
station communicating therewith. 
SUMMARY OF THE INVENTION 
I realize these objects, in accordance with my present invention, by 
providing two signal-detecting networks connected between respective 
talking conductors and respective terminals of a direct-current source, 
such as the usual central battery, together with two threshold comparators 
connected to these networks for translating line-voltage changes of 
predetermined minimum magnitudes into incoming d-c signals detectable by 
the switching equipment of the central office, the networks as well as the 
comparators being respectively constituted by sets of first and second 
integrated-circuit components. A common capacitor connected between the 
two networks compensates voice-frequency currents propagating in opposite 
directions over the talking conductors, thereby providing a high network 
impedance for such currents. A common biasing circuit for two 
semiconductor elements of these networks includes a light-responsive 
device juxtaposed with a light-emitting device controlled by the switching 
equipment to translate outgoing d-c signals into voltage changes on the 
talking conductors which are detectable at the subscriber station; the 
incoming signals generated by the threshold comparators are transmitted by 
individual coupler circuits from these comparators to the switching 
equipment. 
Advantageously, pursuant to a more particular feature of my invention, the 
individual coupler circuits referred to are also constituted by 
photoelectric couplers galvanically separating the line loop and its 
signal-transmitting circuitry from the central-office equipment responding 
to the incoming d-c signals. 
In a preferred embodiment more fully described hereinafter, the 
integrated-circuit components of the two signal-detecting networks as well 
as of the associated threshold comparators are symmetrically disposed with 
reference to the talking conductors and include mutually complementary 
transistors. Each comparator may include a pair of such transistors 
interconnected in a bistable circuit which establishes a certain 
hysteresis with distinct thresholds for saturation and desaturation. These 
two transistors are advantageously of opposite conductivity types, as are 
two other transistors which form part of the associated signal-detecting 
network and which include a first transistor with an input electrode 
(base) connected to the common capacitor and a second transistor in series 
with a voltage divider having a tap connected to the threshold comparator, 
this latter transistor constituting the semiconductor element biased by 
the light-responsive device.

SPECIFIC DESCRIPTION 
FIG. 1 shows two talking conductors a and b extending between a central 
office and a subscriber station to form a line loop. A central battery B 
has its negative terminal 1 connected to conductor a via a signal detector 
IS and has its positive terminal 1', assumed to be grounded, connected to 
conductor b via a signal detector IS'. The flow of direct current from 
this battery to the selected stages of the central office is prevented by 
blocking capacitors Ca and Cb respectively inserted in conductors a and b. 
Incoming d-c signals, generated in the usual manner by the closing and 
opening of the line loop at the remote subscriber station, are detected by 
networks IS and IS'; these networks have a high impedance for voice 
frequencies and transients so that only legitimate voltage changes of 
predetermined minimum duration, such as dialing pulses, give rise to 
significant potential differences at their outputs which extend to 
respective threshold comparators CS and CS'. The two comparators are 
essentially bistable so as to be settable and resettable by voltages 
rising above and falling below respective thresholds which are well 
separated from each other. The state of each comparator CS and CS' is 
communicated to the switching equipment of the central office by means of 
respective photoelectric couplers AO and AO' working into signal lines 3 
and 3'. Another signal link 2 carries outgoing signals to be transmitted 
to the subscriber station, this link terminating at a photoelectric 
coupler AO1 delivering pulses of opposite polarities to the two signal 
detectors IS and IS'. 
As further shown in FIG. 1, a common capacitor C1 is connected between the 
two networks IS and IS' for the purpose of compensating rapid current 
alternations, in and above the voice-frequency range, occurring with 
relatively inverted phase at the junctions of conductors a and b with the 
inputs of these networks. It is the presence of this compensating 
capacitor that, as will become clearer from the following description of 
FIG. 2, imparts to networks IS and IS' a simulated inductive 
characteristic. Spurious common-mode pulses propagating cophasally along 
conductors a and b, however, do not pass through capacitor C1 so as to 
find a relatively low resistance in networks IS and IS' which shunt them 
to battery B and ground. 
Details of blocks IS, IS', CS, CS', AO, AO' and AO1 are shown in FIG. 2. As 
will be apparent from this Figure, signal detectors IS and IS' are 
symmetrically identical with reference to loop a, b except for the 
polarities of their diodes and transistors; the same is true of threshold 
comparators CS and CS'. It will therefore be sufficient hereinafter to 
describe the construction and mode of operation of components IS and CS 
which are constituted by transistors, diodes and resistors realized in 
integrated circuitry. Such integration is also possible for the 
photoelectric couplers AO, AO' and AO1 so that only capacitors Ca, Cb and 
C1--because of their size--may have to be designed as discrete components. 
More particularly, network IS includes a PNP transistor T1 and an NPN 
transistor T2, three diodes D1, D2 and D3, and seven resistors R1-R7. 
Comparator CS includes an NPN transistor T3 and a PNP transistor T4, two 
further diodes D4, D5 and six resistors R8-R13. The associated 
photoelectric coupler AO comprises a light-emitting diode (LED) D6 
juxtaposed with a phototransistor T6 connected across signal link 3; the 
common photoelectric coupler AO1 comprises a light-emitting diode D7, 
connected across signal link 2, and a juxtaposed phototransistor T5 whose 
emitter is connected through a resistor R14 to a junction of the anodes of 
diodes D1, D2 in network IS and whose collector is connected to the 
junction of the cathodes of the corresponding diodes D1', D2' in network 
IS'. 
The circuit elements of components IS, CS and AO have counterparts in 
components IS', CS' and AO' designated by the same reference characters 
with the addition of a prime mark. 
Transistor T1 has its base connected via resistor R1 to the collector of 
transistor T2 which in turn is connected to conductor a via resistor R4 
designed to reduce the emitter/collector voltage of this transistor. The 
collector of transistor T1 is tied to negative terminal 1 whereas its 
emitter is connected to ground at terminal 1' via resistor R6 and is 
further connected to the cathode of diode D1 serving to prevent any 
reverse biasing of the input circuit of this transistor. Companion diode 
D2 has its cathode tied to the base of transistor T2 which is connected to 
negative battery by way of resistor R7; the emiter of transistor T2 is 
connected to the negative terminal 1 through a voltage divider consisting 
of resistors R3 and R5 whose junction is connected by way of diode D3 to 
the junction of resistors R8 and R9 in comparator CS constituting another 
voltage divider. Transistor T3 has its emitter tied to negative terminal 1 
and its collector connected to ground by way of LED D6 and a further 
voltage divider formed by resistors R11 and R12 whose junction is tied to 
the base of transistor T4. The base of transistor T3 is connected to its 
emitter through resistor R10 and, via the stacked diodes D4 and D5, to 
resistor R9 as well as to the collector of transistor T4 whose emitter is 
grounded through resistor R13, with voltage divider R8, R9 extending 
between ground and the collector of transistor T4. Thus, the two 
complementary transistors T3 and T4 have their bases and collectors 
cross-connected to form a multivibrator-type bistable circuit. 
The impedance stability of network IS depends on the gain of the active 
element of this network which could be made high through the use of a pair 
of transistors in Darlington configuration; this, however, would introduce 
an elevated response threshold on account of the two cascaded base/emitter 
paths. In the illustrated embodiment, the emitter-follower configuration 
of transistors T1 and T2 assures a high gain with mutual compensation of 
their base/emitter voltages to make the response threshold negligible. 
Similarly, diode D2 compensates for the voltage drop across diode D1 in 
its conductive state. 
A biasing current for the base of main transistors T2 and T2' normally 
flows from ground terminal 1' through resistor R7', diode D2', 
phototransistor T5, resistor R14, diode D2 and resistor R7 to negative 
terminal 1, provided that the phototransistor T5 is sufficiently 
illuminated by LED D7; part of this current will also pass through 
ancillary transistor T1, T1' and diodes D1, D1'. The resulting conduction 
of transistors T2 and T2' connects the line conductors a and b to 
terminals 1 and 1', respectively. If the hook switch at the subscriber 
station is closed, direct current will flow from ground terminal 1' via 
resistors R3'-R5' and transistor T2' to conductor b and from conductor a 
by way of transistor T2 and resistors R3-R5 to negative terminal 1. In 
order to signal the subscriber station (e.g. to release the line when the 
subscriber fails to dial promptly after lifting the receiver, or if 
excessive leakage losses are detected), the central office interrupts the 
excitation current for LED D7 passing along link 2 with resulting blocking 
of the line current. 
As long as the line loop is open, the potential of the junction of 
resistors R3 and R5 differs from that of battery terminal 1 only by a 
relatively small voltage drop across resistors R3 due to current passing 
through that resistor from ground termninal 1' by way of resistor R8 and 
diode D3. Under these conditions the transistor T3 in comparator CS is 
biased to cutoff whereby LED D6 in coupler AO is de-energized and no 
current flows in link 3. 
When the subscriber closes the line loop, the junction of resistors R3 and 
R5 goes sufficiently positive to exceed the conduction threshold of 
transistor T3 defined by the voltage drop across the stacked diodes D4, D5 
(which are permanently traversed by current flowing through resistors 
R8-R10) plus the base/emitter voltage of transistor T3 less the voltage 
drop across diode D3. The resulting conduction of transistor T3 drives the 
junction of resistors R11 and R12 more negative whereby transistor T4 is 
also rendered conductive and further increases the base voltage of 
transistor T3. The cumulative effect causes both transistors to saturate 
and flips the bistable circuit of comparator CS into its alternate state 
in which LED D6 illuminates the phototransistor T6 and lets signal current 
pass along link 3. This conduction threshold can be easily changed by 
varying the number of diodes in series with the base of transistor T3 
and/or the magnitudes of the associated biasing resistors. 
It will be apparent that the saturation of transistors T3 and T4 in the set 
state of the bistable circuit establishes a different threshold for the 
resetting of that circuit by a reduction in the voltage drop across 
resistor R3. When this voltage drop becomes low enough to divert a 
substantial portion of the saturation current of transistor T4 by way of 
resistors R9 and R3, transistor T3 begins to desaturate and initiates the 
cutoff of transistor T4 by a reverse avalanche effect so as to restore the 
original reset state of the circuit with concurrent termination of current 
flow in link 3 by coupler AO. 
In an analogous manner, components IS' and CS' respond to a closure and an 
opening of the line loop by turning on and off the current flow in link 3' 
under the control of photoelectric coupler AO'. 
When talking currents are transmitted over the closed line loop, 
corresponding variations in the collector voltage of transistor T2 are 
transmitted by voltage divider R1, R2 to the base of transistor T1 but are 
compensated by voltage variations of opposite phase transmitted to that 
base from network IS' by way of common capacitor C1. The effect of these 
voice currents upon the conduction of transistors T2 and T2' is therefore 
retarded, as with an inductance. Since the capacitor C1 is ungrounded, 
networks IS and IS' are automatically balanced for alternating current. 
This contrasts with conventional inductance-simulating circuits in which a 
capacitor is connected between a conductively biased based and a grounded 
emitter of a transistor whose collector impedance depends on the gain 
factor as well as on temperature. In the present instance, the resistive 
component of the network impedance is determined by the static 
current/voltage characteristic of main transistor T2 whose slope depends 
on the magnitudes of the associated biasing resistors. 
Common-mode signals arriving cophasally at the terminals of capacitor C1 
will not pass through this capacitor whereby transistors T1, T2 and T1', 
T2' act as two-stage amplifiers shunting these spurious signals to 
terminals 1 and 1', respectively. Such common-mode signals will rarely if 
ever have a duration and a magnitude sufficient to trip the threshold 
comparators CS, CS'; even if this should happen, the switchover would not 
occur simultaneously in both comparators--as it does in the case of normal 
signaling--so that the central-office equipment will readily recognize the 
situation as anomalous.