Acceleration sensor with differential capacitance

In an acceleration sensor, in particular for automobiles, at least two electrodes which are surrounded by a dielectric are arranged, substantially parallel to each other, in a housing and immersed over part of their length in an electrically conductive liquid. Flip-flops output pulses having durations based on capacitance of the sensor. Output signals of the flip-flops are combined arithmetically.

FIELD AND BACKGROUND OF THE INVENTION 
The present invention relates to an acceleration sensor, in particular for 
automobiles. 
For the conversion of acceleration values into electrical signals, 
capacitive acceleration sensors having plate-shaped electrode arrangements 
are known in which by means of a seismic mass under the effect of an 
acceleration, the distance between the plates is changed or the plates are 
rotated or displaced similar to a rotary capacitor. A measurement of the 
capacitance then gives the acceleration which is to be determined. These 
known sensors, however, have very small capacitances, so that the 
evaluation encounters circuit difficulties. Furthermore, a seismic mass is 
required. The temperature sensitivity of the slope of the characteristic 
curve is frequently high. Furthermore, mechanically complex constructions 
are required which do not permit high shock-resistances. Also, in the 
known sensors, the stability of the zero point suffers under the effect of 
temperatures and overloads. Finally, linear characteristic curves can be 
obtained only at relatively high expense. 
SUMMARY OF THE INVENTION 
It is an object of the invention to provide an acceleration sensor which 
operates reliably and accurately and can be manufactured at low cost. The 
acceleration sensor of the invention furthermore is to make possible a 
simple evaluation of the output signal. 
According to the invention, at least two electrodes (1, 2) which are 
surrounded by a dielectric (9, 10) are arranged substantially parallel to 
each other within a housing (4), a part of their length dipping into an 
electrically conductive liquid (3). 
In this connection, it is preferably provided that each of the electrodes 
(1, 2) is connected to a separate device for measuring the capacitance 
present between the electrode and the liquid, and that the devices for 
measuring the capacitance are connected at the output to a subtraction 
circuit. 
The acceleration sensor of the invention does not require a solid seismic 
mass nor any elements to be deformed mechanically such as torsion beams 
with strain gauges or quartzes, which change their properties due to 
changes in temperature, aging or overstressing. The acceleration sensor of 
the invention can be produced as a compact part of small size with high 
shock resistance and high zero-point stability. This part can, on the one 
hand, be produced at low cost. On the other hand, a printed circuit board 
which bears an evaluation circuit can be provided directly with the 
acceleration sensor of the invention. Further advantages of the 
acceleration sensor of the invention consist of its symmetrical 
construction, a high change in capacitance which makes a simple evaluation 
circuit possible, a wide range of operating temperatures, a linear 
characteristic curve, very low temperature dependence and good 
electromagnetic compatibility. 
The above-mentioned advantages of the acceleration sensor of the invention 
make it possible, in particular, to use it in an automobile, in order, for 
instance, to control the damping of the chassis. Its application is, 
however, not limited to this purpose but it can also be used, for 
instance, in aircraft and water vehicles. 
One advantageous development of the acceleration sensor of the invention 
provides two electrodes (1, 2) which extend parallel to each other and are 
covered over about half their length with liquid (3). 
In this connection the housing (4) preferably has the shape of a 
semi-circle and a greater extension in the plane formed by the electrodes 
(1, 2) than perpendicular to that plane. This embodiment makes a 
particularly compact part possible, which is inexpensive, in particular if 
the housing is a commercially available oscillating quartz housing. 
In order to measure the acceleration in several directions, several - 
preferably three or four - parallel electrodes, in accordance with a 
further development of the invention, are provided to form different 
planes in pairs. Also the invention can provide, for the measurement of 
the acceleration in several directions, one acceleration sensor with two 
electrodes for each direction. 
The conducting liquid should have low adhesion forces with respect to the 
inner wall of the housing so that hysteresis influences become negligible. 
It is furthermore advantageous to adapt the cooperation with the cohesive 
forces in such a manner that the liquid level is not influenced by 
capillary action due to the small distance between electrode and housing. 
According to a feature of the invention, the electrodes (1, 2) are coated 
with the dielectric (ceramics, glass, or insulating varnish) in such a 
manner that a high dielectric coefficient and high tightness to the liquid 
result. 
An acceleration sensor in accordance with the invention which has an 
advantageous evaluation circuit is characterized by the fact that two 
monostable flip-flops (14, 15) are provided, that the resistance or 
capacitance values of the acceleration sensor determine the time constants 
of the monostable flip-flops (14, 15), and that in each case one output of 
one monostable flip-flop is connected to a setting input (trigger) input 
of the other monostable flip-flop in such a manner that by passage of in 
each case, one monostable flip-flop (14; 15) into the stable state, the 
corresponding other monostable flip-flop (15; 14) passes into the unstable 
state. The outputs of the monostable flip-flops (14, 15) are preferably 
connected here to the trigger inputs via differentiating circuits (16, 17; 
18, 19). 
This evaluation circuit is characterized by a low expense and it produces a 
binary output signal which can be further processed in simple manner both 
by a microprocessor and by a simple analog circuit. Another advantage is 
that information concerning both sensor variables can be sent out over one 
signal line. However, a symmetric output of the evaluation circuit can 
also be utilized, which results in good electromagnetic compatibility. 
For the further processing of the output signal, an output signal of at 
least monostable flip-flop (14; 15) can, in accordance with a further 
development of the evaluation circuit, be fed to an integrator (22: 23; 
24, 25). Furthermore, the transmission of an analog signal to a 
further-processing circuit is possible in the manner that the output 
signal of a monostable flip-flop can be fed directly to a first integrator 
and, inverted, to a second integrator and that the outputs of the 
integrators are connected to the inputs of a subtraction circuit. In this 
way, a simple asymmetric binary interface is created between the 
evaluation circuit present in the immediate vicinity of the sensor and a 
circuit for the further processing of the output signal of the evaluation 
circuit. 
A symmetric binary interface is formed in the manner that each of the 
output signals of the monostable flip-flop (14, 15) can be fed to a 
separate integrator (22, 23; 24, 25; 33, 34) and that the outputs of the 
integrators are connected to the inputs of a subtraction circuit (35; 39). 
Further according to the invention, the output signal of one monostable 
flip-flip (14) can be fed directly to a first integrator (31, 32) and, 
inverted, to a second integrator (33, 34) and the outputs of the 
integrators are connected to inputs of a subtraction circuit (35). 
Still further according to the invention, the oscillation of at least one 
monostable flip-flop (14, 15) is monitored and, upon absence of the 
oscillation, at least one monostable flip-flop (14, 15) is again placed 
into the unstable state via the corresponding trigger input. 
Still further according to the invention, the oscillation of the circuit is 
monitored by a logical OR connection of the two output signals of the 
monostable flip-flops (14, 15). 
Circuits consisting of two monostable flip-flops which place each other 
into the unstable condition may possibly not oscillate again after a 
disturbance, for instance due to a brief short-circuiting of one of the 
output signals. In order to make dependable restarting of the oscillation 
possible even in such a case, the evaluation circuit can be further 
developed in the manner that the differentiating circuits consist of one 
capacitor (16, 18) each and one resistor (17, 19) each which is connected 
to a terminal (41) for the operating voltage, and that both terminals of 
the capacitors (16, 18) belonging to the differentiating circuits are 
connected via one diode each (51, 52, 55, 56) to a resistor (53) the 
terminal of which facing away from the diodes is acted on by ground 
potential.

Identical parts in the figures have been provided with the same reference 
numbers. 
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
In the diagrammatic showing of FIG. 1, two round, elongated electrodes 1, 2 
which are coated with a dielectric, not shown in FIG. 1, are immersed in 
an electrically conductive liquid 3. An acceleration a acting on the 
housing 4 in the direction of the arrow causes an oblique positioning of 
the liquid level by the angle .alpha.. As a result, the electrode 1 is 
covered by liquid merely up to the level h.sub.1 while the liquid rises to 
h.sub.2 at the electrode 2. The average liquid level is (h.sub.1 
+h.sub.2)/2. As can be noted from the triangle of forces also shown in 
FIG. 1, we have, for the angle .alpha., tan .alpha.=F.sub.m /F.sub.g, in 
which F.sub.m is the force of inertia counteracting the acceleration force 
F.sub.a and F.sub.g is the weight. 
The dielectric on the electrode surface forms a capacitor between each of 
the electrodes 1, 2 and the conductive liquid, the capacitor being shown 
diagrammatically in FIG. 2. The capacitance of such a coaxial capacitor is 
EQU C=[.pi..epsilon..sub.O .epsilon..sub.r /(lnD/d)].multidot.h. 
In this equation, D is the outside diameter of the dielectric, d is the 
diameter of the electrode, and h is the liquid-covered height of the 
electrode in question. 
For the difference in the capacitances, we then have: 
EQU C.sub.2 -C.sub.1 =[2.pi..epsilon..sub.O .epsilon..sub.r 
/(lnD/d)].multidot.(h.sub.2 -h.sub.1) 
As can be noted readily from the triangle of forces shown in FIG. 1, 
EQU h.sub.2 -h.sub.1 =s.multidot.(a/g), 
in which a is the acceleration transverse to the acceleration of gravity g 
and s is the distance between electrodes 1 and 2. There then results, for 
the standardized acceleration the equation 
EQU a/g=(C.sub.2 -C.sub.1).multidot.(lnD/d)/2.pi..epsilon..sub.O 
.epsilon..sub.r s. 
It can be noted in this connection that the value a obtained by the 
measurement of the difference in capacitances is independent of the height 
of filling (h.sub.1 +h.sub.2)/2, but is dependent on .epsilon.. according 
to a further development of the invention, there is, however, the 
possibility, upon the evaluation, of dividing the difference by the sum of 
the capacitances. We then have for the standardized acceleration 
EQU a/g=(C.sub.2 -C.sub.1).multidot.(C.sub.2 +C.sub.1).multidot.(h.sub.1 
+h.sub.2)/s 
The result of the measurement is in this case independent of .epsilon. but 
dependent on the height of filling. This has, for instance, the advantage 
that a temperature dependence of .epsilon. does not enter into the result 
of the measurement. 
In the embodiment shown in FIG. 3, the two electrodes 1 and 2 are held by 
the bottom 5 of a metal housing 6 obtainable for oscillating quartzes. 
Glass lead-throughs 7, 8 are provided for insulation. The housing is 
sealed hermetically by corresponding soldering. The liquid 3 covers about 
one-half of the electrodes 1, 2. Extensions of the electrodes 1, 2 serve 
at the same time as soldering pins 11, 12. A third soldering pin 13 
connected directly to the housing bottom 5 forms the common 
counterelectrode. 
The soldering pins 11 to 13 are arranged in a standard grid so that the 
acceleration sensor can be mounted directly on a printed circuit board. 
The acceleration sensor shown in FIG. 4 can possibly be used with the 
terminals 11 to 13 facing upward. As a result, the region of the passage 
of the electrodes 1, 2 through the housing bottom 5 is not covered by 
liquid, so that the seal between the dielectric and the housing bottom is 
not constantly acted on by the liquid. 
The housing 6 may be formed with a semicircular shape as shown in FIG. 3. 
Suitable liquids, which are furthermore electrically conductive and 
suitably stable chemically, and furthermore do not attack the dielectric 
substance or the housing, are readily available. Thus, water, mercury and 
glycol can be used. A dielectric (9,10) having a wetting-reducing surface 
is preferably used, for instance unsintered PTFE 
(polytetrafluoroethylene). 
In the acceleration sensor of the invention, the insulating layers of the 
dielectric which surround the electrodes can be made relatively thin so 
that high values of capacitance result. In one actually constructed 
acceleration sensor according to the invention, a filling height of 10 
millimeters and a distance between the electrodes of also 10 millimeters 
was selected, which, depending on the type and thickness of the 
dielectric, resulted in capacitance values of 30 pF to 1,000 pF, which can 
be measured with relatively simple circuits. 
FIG. 4 shows an equivalent circuit diagram of the acceleration sensor of 
the invention which consists of two capacitors of which the capacitances 
C.sub.1 and C.sub.2 which, corresponding to the above derivations, are a 
function of the acceleration acting on the acceleration sensor. 
In the circuit shown in FIG. 5 two monostable flip-flops 14, 15 are 
provided which are so designed that the duration of the unstable state is 
proportional to a value X.sub.1 or X.sub.2 present as resistance or 
capacitance value, which values correspond in the present case to the 
capacitances C.sub.1 and C.sub.2 . The output of each monostable flip-flop 
14, 15 is connected via differentiating circuits 16, 17 and 18, 19 
respectively to an inverting setting input (hereinafter called trigger 
input) of the other monostable flip-flop. In this way, one monostable 
flip-flop is in each case placed in the unstable state when the other one 
returns to the stable state. 
Square signals then result at the outputs 20, 21 of the monostable 
flip-flops 14, 15, their course being shown in FIG. 6. The two periods of 
time t.sub.1 and t.sub.2 are in this case always proportional to the input 
variables X.sub.1 and X.sub.2. 
If the difference between the variables X.sub.1 and X.sub.2 is evaluated, 
one obtains a reduction in the temperature sensitivity in the event that 
both variables have the same temperature course, namely, in the event that 
both capacitors drift the same amount with temperature. The reduction in 
temperature occurs because in a differential measurement the temperature 
offsets cancel. However, this is true only for the zero-point stability. 
For a reduction of the temperature course of the slope, the presence of 
inclination of the liquid surface, division by the temperature-dependent 
variables must still be effected. The division is a normalization obtained 
by dividing the differential capacitance (or capacitive reactance) by the 
sum of the capacitances (or capacitive reactances). In the evaluation 
circuit of the invention, such a signal is obtained in simple manner by 
subtracting the mean values as follows: 
EQU U.sub.A1 -U.sub.A2 =U.sub.B .multidot.(X.sub.1 -X.sub.2)/(X.sub.1 
+X.sub.2). 
In most sensors, particularly in those in which high resistances or small 
capacitances are evaluated (as in the acceleration sensor of the 
invention), the evaluation circuit is located close to the resistors or 
capacitors while a device which processes the output signals of the 
evaluation circuit is connected to the evaluation circuit by one or more 
lines. The output signals of the evaluation circuit of FIG. 5 form a good 
basis for transmission to the processing circuit, for instance a control 
device in an automobile. Depending on the specific requirements, the 
transmission from the evaluation circuit to the control device can take 
place in the form of a binary signal or in the form of an analog signal. 
FIG. 7 shows several circuit arrangements for this. In FIG. 7a, one of the 
output signals U.sub.A1 is transmitted in unchanged form, i.e. binary, by 
an evaluation circuit 37. On the receiver side, a digital computer 28 is 
provided by means of which the times t.sub.1 and t.sub.2 are measured, 
whereby the values X.sub.1 and X.sub.2 are recovered. The formation of a 
difference, sum and quotient can then take place in simple manner in the 
digital computer 28 so that a value (X.sub.1 -X.sub.2)/(X.sub.1 +X.sub.2), 
normalization of the differential capacitance, is produced, which is 
directly compensated for by the normalization. The compensation occurs in 
the case of sensors whose output variable is formed by the difference of 
the two variables X.sub.1 and X.sub.2, and wherein X.sub.1 and X.sub.2 are 
subjected to a disturbing effect such as a temperature dependence. 
The binary signal U.sub.A1 is also transmitted in the case of the circuit 
shown in FIG. 7b. The further evaluation, however, takes place by means of 
an analog circuit which consists of an input amplifier 29, an inverter 30, 
an integrator comprising resistor 31 and capacitor 32, and an integrating 
element comprising resistor 33 and capacitor 34, and a difference 
amplifier 35. Due to the integration by means of the integrating member 
31, 32, the mean value of the signal U.sub.A1 is formed, which is 
proportional to X.sub.1 /(X.sub.1 +X.sub.2). The mean value of the 
inverted signal is formed by the integrating member 33, 34 and corresponds 
to X.sub.2 /(X.sub.1 +X.sub.2). The difference amplifier 35 then forms the 
desired result, which is present at the output 36 as analog signal. 
While the circuit shown in FIG. 7b between the output of the evaluation 
circuit and the input of the control device represents an asymmetric 
binary interface, the circuit shown in FIG. 7c has a symmetric binary 
interface. For this purpose, the two outputs of the evaluation circuit are 
connected by one line each to the input amplifiers 29, 38 of the control 
device. The analog signal is formed, as in the case of the circuit 
arrangement of FIG. 7b, by integrating members 31, 32 or 33, 34 and a 
difference amplifier 35. The advantage consists in the freedom from 
interference upon transmission over long lines. 
Finally, FIG. 7d shows a further possibility for signal transmission 
between an evaluation circuit and a control device, in which an analog 
signal is transmitted. For this purpose, an integrator comprising resistor 
22 and capacitor 23 and an integrator comprising resistor 24 and capacitor 
25, and a difference amplifier 39 are arranged in the region of the 
evaluation circuit 37. The control device is connected by a line 40. 
In the circuit arrangement shown in FIG. 8, the two monostable flip-flops 
14, 15 are formed by an integrated module of Type Series 556 (dual clock 
circuit). As in the block diagram of FIG. 5, each of the outputs is 
connected via a differentiating circuit 16, 17 and 18, 19 respectively to 
the inverting trigger input of the other monostable flip-flop. Resistors 
41 and 42 are connected between the terminal 43 for the operating voltage 
UB and the corresponding output and serve as working resistances. Each of 
the inverting trigger inputs is also connected by one diode 44, 45 each to 
the terminal 43 in order to limit the voltage at the trigger inputs. 
The inputs Dis and Thr of the monostable flip-flops 14, 15 are connected to 
corresponding time constant members each of which consists of a resistor 
46, 47 and a capacitor 48, 49 of variable capacitance. The capacitors 48, 
49 are part of the acceleration sensor, in connection with which the 
capacitances are changed in opposite direction as a function of the 
variable to be measured. As already explained in connection with FIG. 5, 
the duration of the unstable state is proportional to the capacitance, 
whereby the output signals U.sub.A1 and U.sub.A2 are produced at the 
outputs of the monostable flip-flops 14, 15 in accordance with the diagram 
shown in FIG. 6. 
The circuit arrangement shown in FIG. 8 starts to oscillate only if the 
speed of rise of the operating voltage exceeds a predetermined value upon 
the connection. In the case of a brief interruption of the oscillation, 
for instance by a short-circuit or by the action of an interference pulse, 
the circuit does not start to oscillate again. In order to make dependable 
starting of the oscillation possible, the circuit arrangement shown in 
FIG. 9 is developed further in advantageous manner as compared with the 
circuit arrangement of FIG. 8. 
For this, the outputs of the monostable flip-flops 14, 15 are connected to 
ground potential via respective diodes 51 and 52 and a common resistor 53. 
The diodes 51, 52 act as OR-connection of the signals U.sub.A1 and 
U.sub.A2. Due to the fact that U.sub.A1 and U.sub.A2 are inverted with 
respect to each other, the voltage U.sub.53 at the resistor 53 amounts to 
U.sub.B -0.7V upon oscillation of the circuit at any time. A capacitor 54 
smooths any peaks which are produced while the flanks (leading and 
trailing edges of a pulse waveform) of U.sub.A1 and U.sub.A2 are produced. 
If there is no oscillation, the-output voltages U.sub.A1 and U.sub.A2 are 
at ground potential and both diodes 51, 52 block. The voltage U.sub.53 and 
the voltages at the trigger inputs are then determined by the resistors 
17, 19 of the diodes 55, 56, which then become conductive, and by the 
resistor 53. The voltages at the trigger inputs thus drop to the value 
U.sub.tr =(U.sub.B -0.7V)/(1+R.sub.17 /R.sub.53)+0.7V. By the selection of 
the values R.sub.17 and R.sub.53 of the resistors 17 and 53, U.sub.tr is 
set below the value specified in the data sheet of the monostable 
flip-flop. As a result, both output signals are again transferred to the 
unstable state and the starting assistance effected by the diodes is 
terminated. 
In addition to their function as starting assistance, the diodes 55, 56 
also serve to limit the voltages fed to the trigger inputs so that they do 
not rise above the operating voltage U.sub.B. The dimensioning of the 
resistors 17 or 19 and 53 respectively takes place in accordance with the 
following equation: 
EQU R.sub.17(19) /R.sub.53 .gtoreq.(U.sub.B -0.7V)/(U.sub.tr -0.7V)-1, 
in which U.sub.B =5V and U.sub.tr .gtoreq.1.26V when using the 556 module. 
From this there results R.sub.17(19) /R.sub.53 .gtoreq.6.68. In one 
circuit arrangement used in actual practice, R.sub.17 =R.sub.19 =47 kOhms 
and R.sub.53 =6.8 kOhms. 
The differentiating circuits 16, 17, 18, 19 have the task of deriving a 
short pulse which characterizes the dropping flank of the instantaneous 
output signal. For this a time constant is required which is substantially 
less than the duration of the unstable conditions of the monostable 
flip-flops. We thus have the following condition: R.sub.17 
.multidot.C.sub.3 &lt;&lt;R.sub.46 .multidot.C.sub.48, which applies by analogy 
also for the elements 18, 19, 47 and 49.