Method and apparatus for dual-band modulation in powerline communication network systems

A novel method and apparatus for modulating in dual operational bands in powerline networking systems is described. A transmitter and a receiver are described wherein the transmitter and receiver are operable in different modulation frequency bands. The present invention can easily switch between operational frequency bands by utilizing a fundamental signal for performing modulations in a first frequency band and by utilizing a first alias signal for performing modulations in a second frequency band. The present inventive method and apparatus can switch operation from a first operational frequency band to a second operational frequency band by modifying two components in existing transmitters and only one component in existing OFDM receivers. Advantageously, therefore, the present invention can be utilized with existing powerline networking technology.

DETAILED DESCRIPTION OF THE INVENTION Throughout this description, the preferred embodiment and examples shown should be considered as exemplars, rather than as limitations on the present invention. The present invention is a method and apparatus for dual-band modulation in powerline networking systems. The present invention can be easily utilized with existing powerline networking technology. The inventive method and apparatus utilizes a transmitter and a receiver that can operate in different modulation frequency bands with little modification. The present invention can easily switch between operating frequency bands by utilizing a fundamental signal for modulating a first frequency band and a first alias signal for modulating a second frequency band. In one embodiment, the fundamental signal modulates frequency bands below 25 MHz (e.g., between 4-25 MHz for the U.S. operating frequency band) while the first alias signal modulates frequency bands above 25 MHz (e.g., greater than 25 MHz for the European frequency band). The present inventive method and apparatus can switch operation from a first frequency band to a second frequency band by slightly modifying two components in existing OFDM transmitters and by modifying only one component in existing OFDM receivers. In one embodiment, designed to operate in frequency bands below 25 MHz, the inventive OFDM transmitter includes a low-pass anti-aliasing filter and a first set of weighting values. The inventive OFDM receiver includes a low-pass anti-aliasing filter. For operating in frequency bands above 25 MHz, the inventive OFDM transmitter includes a band-pass anti-aliasing filter and a second set of weighting values. The inventive OFDM receiver includes a band-pass anti-aliasing filter. One embodiment of the inventive OFDM transmitter for use with the present invention is now described. OFDM Transmitter FIG. 5 a shows a simplified block diagram of one embodiment of an OFDM transmitter made in accordance with the present invention. As shown in FIG. 5 a , the OFDM transmitter 500 comprises a digital data source 502 , a modulation operations stage (comprising the processing blocks 504 - 520 ), and a line driver/power line coupler stage 530 . The digital data source 502 outputs a digital bitstream to the input of a serial-to-parallel converter 504 . As described above with reference to FIG. 3 , the serial to parallel converter 504 converts the digital bitstream into a series of parallel words wherein each parallel word includes complex values. In one embodiment, a QPSK modulation scheme utilizing all frequency tones preferably converts 168-bit blocks of the digital bitstream into single words comprising 84 complex values each. The QPSK modulation scheme, block bit values and word values are not meant to limit the present invention as one skilled in the art shall recognize that different modulation schemes and values can be used without departing from the spirit or the scope of the present invention. In the present invention each complex value ultimately imposes one of four phases on one of the carriers in the OFDM carrier set. The serial-to-parallel converter 504 outputs each parallel word to an input of the weighting stage process 506 . The weighting stage process 506 performs amplitude weighting on the complex values of each parallel word. Weighting techniques are well known in the modulation art, and thus, are not described in more detail herein. Each carrier can potentially be weighted differently. Weighting can be applied for various reasons such as for providing power control (if applied to all values equally). Another motivation for applying weighting is to shape the transmit frequency spectrum. In powerline networking, weighting of the complex values is desirable in order to compensate for the response generated by the digital-to-analog (D/A) converter 514 (described hereinbelow), which in one embodiment produces a sin(x)/x response. The weighting that is used depends upon the frequency band being utilized in the OFDM transmitter 500 because the D/A converter 514 responses are frequency-dependent. Thus, in a dual-band OFDM transmitter, a first set of weighting values is used for operating within a first frequency band, and a second set of weighting values is used for operating within a second frequency band. In one embodiment of the present inventive OFDM transmitter 500 , a first set of weighting values is used for operating within a “low” frequency band, and a second set of weighting values is used for operating within a “high” frequency band. In this embodiment, the low band is defined herein as frequency bands below 25 MHz (e.g., the 4-25 MHz U.S. operating frequency band), and the high band is defined herein as frequency bands above 25 MHz (e.g., the greater than 25 MHz European operating frequency band). Table 1 (shown below) contains exemplary low band and high band weighting values for use with the transmitter 500 of FIG. 5 a. 1 TABLE 1 Weights used for Correction of the D/A Response low band high band tone low band high band tone &num; weight weight &num; weight weight 23 1.01 10.27 65 1.11 3.27 24 1.01 9.81 66 1.12 3.22 25 1.02 9.39 67 1.12 3.17 26 1.02 9.00 68 1.13 3.11 27 1.02 8.64 69 1.13 3.06 28 1.02 8.31 70 1.13 3.01 29 1.02 8.00 71 1.14 2.97 30 1.02 7.71 72 1.14 2.92 31 1.02 7.44 73 1.15 2.88 32 1.03 7.18 74 1.15 2.83 33 1.03 6.95 75 1.16 2.79 34 1.03 6.72 76 1.16 2.75 35 1.03 6.51 77 1.17 2.71 36 1.03 6.31 78 1.17 2.67 37 1.04 6.13 79 1.18 2.63 38 1.04 5.95 80 1.18 2.60 39 1.04 5.78 81 1.19 2.56 40 1.04 5.62 82 1.19 2.53 41 1.04 5.47 83 1.20 2.49 42 1.05 5.33 84 1.20 2.46 43 1.05 5.19 85 1.21 2.43 44 1.05 5.06 86 1.21 2.40 45 1.05 4.94 87 1.22 2.37 46 1.06 4.82 88 1.22 2.34 47 1.06 4.70 89 1.23 2.31 48 1.06 4.59 90 1.24 2.28 49 1.06 4.49 91 1.24 2.25 50 1.07 4.39 92 1.25 2.23 51 1.07 4.29 93 1.26 2.20 52 1.07 4.20 94 1.26 2.17 53 1.07 4.11 95 1.27 2.15 54 1.08 4.03 96 1.28 2.13 55 1.08 3.95 97 1.28 2.10 56 1.08 3.87 98 1.29 2.08 57 1.09 3.79 99 1.30 2.06 58 1.09 3.72 100 1.30 2.03 59 1.09 3.65 101 1.31 2.01 60 1.10 3.58 102 1.32 1.99 61 1.10 3.52 103 1.33 1.97 62 1.10 3.45 104 1.33 1.95 63 1.11 3.39 105 1.34 1.93 64 1.11 3.33 106 1.35 1.91 To facilitate a better understanding of the derived weighting values, a brief description of tone positioning and D/A converter response is now presented. Tone positioning refers to the process of assigning complex values to corresponding tone positions. One method of tone positioning is described above with respect to the IFFT stage 308 ( FIG. 3 ). In one embodiment of the present invention, low band tone positions range from position 0 to position 127 . In this embodiment, high band tone positions range from position 128 to position 256 . As described above, the weighting of complex values depends on the response of the D/A converter 514 . Graphs depicting the D/A converter response for low band and high band operation are now described. FIG. 6 is a graph showing the D/A converter low band response 60 (in decibels), location of high band carrier set tones 62 and a low band correction gain 64 to be applied for weighting purposes. The low band correction gain 64 shows the gain compensation that can be performed by the weighting stage 506 to compensate for the low band response 60 . This weighting can be performed to equalize the power levels of all carriers at the D/A converter output 514 . FIG. 7 is a graph showing the D/A converter high band response 70 (in decibels), location of high band carrier set tones 72 and a high band correction gain 74 to be applied for weighting purposes. The high band correction gain 74 shows the gain compensation that can be performed by the weighting stage 506 to compensate for the high band response 70 . The weighting can be performed to equalize the power levels of all carriers at the D/A converter output 514 . The high band response 70 shows a considerably steeper roll-off than the low band response 60 of FIG. 6 . Thus, the high band correction gain 74 is correspondingly steeper than is the low band correction gain 64 ( FIG. 6 ). The actual weighting of complex values depends on the tone positioning performed during the IFFT stage 508 . When assigning high-band tone positions the set of carriers is replicated from tone position 150 to tone position 233 of the D/A output signal. However, the order of complex values is reversed. Thus, the largest weight is applied to carrier 23 and the smallest weight is applied to carrier 106 during the weighting stage 506 . Those skilled in the art shall recognize that alternative scaling constants may be used for multiplying all of the weights without impacting the desired result of having each carrier have equal power. In one embodiment of the present invention, the weighting values for low band operation and high band operation (Table 1) are derived from the D/A converter responses shown in FIGS. 6 and 7 . In this exemplary embodiment of the present inventive transmitter, the low band weighting values are utilized to weight the complex values corresponding to tone positions 23 to 106 when operating in frequency bands of less than 25 MHz. Similarly, the high band weighting values are utilized to weight the complex values corresponding to tone positions 23 to 106 (in reverse order) when operating in frequency bands greater than 25 MHz. The weighting of complex values is preferably accomplished using weighting multipliers that add weight values to the complex tones. However, one skilled in the art shall recognize that alternative methods can be used without departing from the scope or spirit of the present invention. In an alternative embodiment, well-known shift-and-add operations are used to perform the weighting function of the weighting stage 506 . In an exemplary embodiment, two adders per weight are used for this purpose. In another alternative embodiment, a digital filter is used to perform the weighting function. In this alternative embodiment, the digital filter operates on time domain samples that are output by the IFFT stage. Referring again to FIG. 5 a, the weighting stage 506 outputs the complex and weighted complex values to the input of the inverse fast Fourier transform (IFFT) 508 . The IFFT 508 arranges the complex and weighted complex values within its associated frequency word to ensure that output waveform samples are properly formed. In one embodiment, a frequency word is preferably defined as a set of tone positions. The number of tone positions depends upon the size of the frequency word. In the embodiment shown, each frequency word comprises 256 tone positions. One skilled in the art shall recognize that different values can be used for the number of tone positions without departing from the scope or spirit of the present invention. Different types of data values are preferably assigned to various tone positions. In one embodiment, the complex values assigned to the tone positions from n&equals;0 to 22 inclusive are set to zero. The weighted complex values are placed at the tone positions from n&equals;23 to 106 inclusive. The word positions from n&equals;107 to 128 are preferably filled with zeros (i.e., zero filled). To ensure creation of a real-valued waveform, the complex conjugate of the value at position 256-n is preferably assigned to the word positions from n&equals;128 to 255, inclusive. As is well known in the modulation design art, the complex conjugate of a complex value is created simply by inverting the sign of the imaginary part of the complex value. After arranging the frequency word in this manner, the IFFT stage 508 computes an inverse fast Fourier transform in a well-known manner, and thus, transforms the frequency word into a time-domain waveform having a length of 256 samples. The IFFT stage 508 ( FIG. 5 a ) outputs the time-domain waveform to the input of the add cycle prefix stage 510 ( FIG. 5 a ). As described above with reference to FIG. 3 , the add cycle prefix stage 510 preferably lengthens the time-domain waveform by adding a “cyclic prefix”. As is well known in the modulation art, cyclic prefixes are used to combat the detrimental effects of multi-path interference. The present invention adds a cyclic prefix by taking a number of samples from the end of the time-domain waveform and replicating them at the beginning of the waveform. In one embodiment, the last 164 samples of the time-domain waveform are replicated and placed at the beginning of the waveform. Thus, the total waveform length, including the prefix, is preferably 420 samples (i.e., 256&plus;164). The add cycle prefix stage 510 outputs prefix-added waveforms to the input of the parallel-to-serial converter 512 . The parallel-to-serial converter 512 converts the prefix-added waveforms into a serial waveform. The data rate of the serial waveform is 50 MHz in one embodiment. One skilled in the art shall recognize that different data rates can be used with the present invention without departing from its scope or spirit. The parallel-to-serial converter 512 outputs the serial waveform to the input of the digital-to-analog converter 514 . The digital-to-analog (D/A) converter 514 converts the serial waveform to a serial analog waveform. A well-known phenomenon resulting from the conversion of a digital bitstream (e.g., the serial waveform) to an analog signal (e.g., the serial analog waveform) using a D/A converter is the production of “aliases”. Aliases are defined herein as frequency-shifted copies of the fundamental spectrum of the signal centered at multiples of the D/A sampling frequency. In one embodiment, the D/A converter 514 is designed to hold each sample level for a full sample clock period, and thus, the set of frequency-shifted aliases are weighted by a sin(x)/x response that has its nulls at multiples of the D/A sampling frequency. One skilled in the art shall recognize that different frequency responses will result for different D/A converters. Thus, the weighting of the sin(x)/x response described above is not meant to limit the present invention as different weighting responses can be used without departing from the scope of the invention. As shown in FIG. 5 a , the D/A converter 514 outputs the serial analog waveform (containing the fundamental signal and a first alias signal) to the input of an anti-alias filter 520 . The anti-alias filter 520 is now described. In one embodiment, the first alias of the fundamental signal begins at 29.3 MHz and extends upward to 45.5 MHz. The present inventive method and apparatus advantageously utilizes both the fundamental signal and the first alias signal to permit use of the transmitter in two operating frequency bands. When operating in the low band (i.e., using the fundamental signal) a low-pass anti-aliasing filter is used in the anti-alias filter stage 520 . In one embodiment, the low-pass anti-alias filter only outputs signals below 25 MHz, for example, the fundamental signal (4.5 to 20.7 MHz). Thus, in this embodiment the anti-alias filter stage 520 outputs the fundamental signal to a line driver and power coupler stage 530 . When operating in the high band (i.e., using the first alias signal) a band-pass anti-aliasing filter is used in the anti-alias filter stage 520 . In one embodiment, the band-pass anti-alias filter outputs only signals having frequencies between 25 to 50 MHz, for example, the first alias signal (29.3 to 45.5 MHz). Thus, in this embodiment, the anti-alias filter stage 520 outputs the first alias signal to a line driver and power coupler stage 530 . Depending on the operating mode (low band or high band being used by the present invention), a waveform containing the desired signal (fundamental signal or first alias signal) is output to the input of a line driver and power line coupler stage 530 . The line driver and power coupler stage 530 amplifies the desired signal and couples the signal to a power line. FIG. 5 b shows another embodiment of the present inventive OFDM transmitter 500 ′ made in accordance with the present invention. The embodiment 500 ′ shown in FIG. 5 b is similar to the OFDM transmitter 500 described above with reference to FIG. 5 a . Similar components are therefore not described in more detail below. In the embodiment 500 ′ of FIG. 5 b , switching operation between low band and high band is accomplished using a switching means. The switching means directs a desired signal to be provided as input to a low-pass filter for low-band operation, and to a band-pass filter for high-band operation. As shown in the embodiment of FIG. 5 b , the transmitter includes a switch 522 , a band-pass anti-alias filter 524 and a low-pass anti-alias filter 526 . The D/A converter 514 outputs an analog waveform to the input of the switch 522 . Depending upon the transmitter operating mode, the switch 522 outputs the analog waveform to either the band-pass anti-alias filter 524 or the low-pass anti-alias filter 526 . When operating in low band mode, for example, the switch 522 routes the analog waveform to the input of the low-pass anti-alias filter 526 . The low-pass anti-alias filter 526 produces a fundamental signal and provides input to this signal as the line driver and power line coupler 530 . When operating in high band mode, the switch 522 routes the analog waveform to the input of the band-pass anti-alias filter 524 . The band-pass anti-alias filter 524 produces a first alias frequency signal and provides this signal as input to the line driver and power line coupler stage 530 . Data demodulation is accomplished using an OFDM receiver having an OFDM demodulation operations stage that is selectively detachably coupled to the power line. An embodiment of the inventive OFDM receiver is now described. OFDM Receiver The present inventive receiver switches operation from a low-band mode of operation to a high-band mode of operation by switching between use of a low-pass anti-aliasing filter and a band-pass anti-aliasing filter. Additional modifications to existing receiver designs are not required because an OFDM receiver does not have the same weighting problem as does an OFDM transmitter. Weighting is unnecessary in the receiver because the A/D response in the OFDM receivers is not a rectangular pulse. Furthermore, although the ordering of the tones on the power line wire is reversed when the alias is used, the process of sampling at the receiver automatically removes this reversal. Thus, the existing receivers need very little modification in order to be designed to operate in high-band modes. In one embodiment, when operating in low-band mode (i.e., when operating in frequency bands below 25 MHz), the inventive OFDM receiver includes a low-pass anti-aliasing filter. When operating in the high-band mode (i.e., when operating in frequency bands greater than 25 MHz), the inventive OFDM receiver includes a band-pass anti-aliasing filter. FIG. 8 a is a simplified block diagram of one embodiment of an OFDM powerline receiver 600 made in accordance with the present invention. As shown in FIG. 8 a , the OFDM powerline receiver 600 comprises a power line coupler and AGC (automatic gain control) stage 602 , a demodulation operations stage (comprising processing blocks 610 - 626 ), and a data sink 628 . The power line coupler and AGC stage 602 couples the power line wire (as described above) to the OFDM receiver 600 . The AGC amplifies the input signals across a predetermined dynamic range. Those skilled in the art shall recognize that the AGC is not necessary to practice the present invention. The power line coupler and AGC stage 602 outputs an analog waveform to an anti-alias filter 610 . The anti-alias filter 610 prevents unwanted signal content from being converted by the A/D converter 620 . As described above, during analog to digital conversion, a signal sampled by an A/D converter 620 can produce signal content at each frequency of the sampled signal. The sampled signal content at each frequency contains the sum of the signal content at each frequency in the analog waveform, the signal content of the current frequency and the signal content of all multiples of the sampling rate used by the A/D converter. Usually the signal content of the current frequency and the signal content of all multiples of the sampling rate will produce interference. Thus, to prevent degradation of the desired signal, the anti-alias filter 610 is used to suppress signal energy that might be “folded” (i.e., mix) into the desired band. When operating in the low band (i.e., using the fundamental signal) a low-pass anti-aliasing filter is used in the anti-alias filter stage 610 . When operating in the high band (i.e., using the first alias signal) a band-pass anti-aliasing filter is used in the anti-alias filter stage 610 . The output of the anti-alias filter 610 is input to an analog to digital (A/D) converter 620 as shown. The A/D converter 620 converts the analog waveform to a digital sample stream. The A/D converter 620 outputs the digital sample stream to the input of a serial-to-parallel (S/P) converter 622 . The S/P converter 622 converts the digital sample stream into a parallel set of samples. The S/P converter 622 outputs the parallel set of samples to the input of a fast Fourier Transform (FFT) stage 624 . The FFT stage 624 computes a fast Fourier transform in a well-known manner to obtain frequency domain values. These frequency domain values are output to the input of a parallel-to-serial (P/S) converter 626 . The P/S converter 626 converts the parallel input signals to a serial signal. The P/S converter 626 outputs the received bits in the serial signal to the input of the data sink 628 . FIG. 8 b shows another embodiment of the present inventive OFDM receiver 600 ′ made in accordance with the present invention. The embodiment 600 ′ of the present invention shown in FIG. 8 b is similar to the OFDM receiver 600 described above with reference to FIG. 8 a . Similar components are not described in more detail below. In the embodiment 600 ′ of FIG. 8 b , the switching operation between the low band and high band is accomplished using a switching means. The switching means directs a desired signal to be provided as input to either a low-pass filter (for low-band operations) or a band-pass filter (for high-band operations). As shown in FIG. 8 b , the receiver 600 ′ uses a switch 612 , a band-pass anti-alias filter 614 and a low-pass anti-alias filter 616 . The power line coupler and AGC stage 602 outputs an analog waveform to the input of the switch 612 . Depending upon the operating mode being used by the receiver 600 ′, the switch 612 outputs the analog waveform to the input of either the band-pass anti-alias filter 614 or the low-pass anti-alias filter 616 . When operating in a low band mode, the switch 612 routes the analog waveform to the low-pass anti-alias filter 616 . The low-pass anti-alias filter 616 outputs a filtered signal to the A/D converter 620 . When operating in a high band mode, the switch 616 routes the analog waveform to the band-pass anti-alias filter 614 . The band-pass anti-alias filter 614 outputs a filtered signal to the A/D converter 620 . The OFDM receiver 600 ′ demodulates the filtered signal in a manner described above with reference to FIG. 8 a. Summary In summary, the present invention is a method and apparatus for dual-band modulation in powerline networking systems. The inventive method and apparatus utilizes a transmitter and a receiver that can operate in different modulation frequency bands. The present invention can easily switch between operating frequency bands by using a fundamental signal for modulating in a first frequency band and by using a first alias signal for modulating in a second frequency band. The present inventive method and apparatus can switch between operation in a first frequency band to a second frequency band by slightly modifying only two components of existing OFDM transmitters and by modifying only one component in existing OFDM receivers. Advantageously, therefore, the present invention can be utilized with existing powerline networking technology. A few embodiments of the present invention have been described. Nevertheless, it will be understood that various modifications may be made without departing from the spirit and scope of the invention. For example, the present inventive method and apparatus can weight complex values utilizing weighting multipliers. Alternatively, a shift-and-add operation can be used to weight the complex values without departing from the scope of the present invention.