FM/chirp detector/analyzer and method

A device for determining the frequency range and chirp rate of chirp radars or other sources of frequency-modulated signals includes a compressive receiver (16, 22, 24) for time-compressing single-frequency signals and a discriminator (26) for generating an output that represents the instantaneous frequency of the compressive-receiver output. For narrow-band signals, the frequency-modulated components in the output of the compressive receiver do not last long enough to cause a response from the discriminator (26). When the input of the compressive receiver is a chirp signal, on the other hand, the resultant compressive-receiver output lasts long enough to cause a discriminator response, and its time of occurrence and rate of frequency change are indications of the frequency range and chirp rate of the compressive-receiver input. The discriminator (26) accordingly generates an output whose slope is an indication of the chirp rate of the compressive-receiver input.

BACKGROUND OF THE INVENTION 
The present invention is directed to monitoring for the presence of 
frequency-modulated signals such as those produced by chirp radars. 
In a chirp radar, a radar transmitter transmits a carrier whose frequency 
is swept through a range of frequencies and whose amplitude is modulated 
by pulses that typically are smooth and of relatively long duration. A 
common method of producing this type of signal is to generate a 
short-duration baseband pulse, band-limit it, typically with a Gaussian 
filter to provide a short-duration oscillatory signal, and apply the 
oscillatory signal to a dispersive delay line, which delays different 
frequencies by different amounts and thus spreads the signal in time. The 
spreading of the signal results in lower instantaneous power for a given 
average power. Despite the relatively long duration of the dispersed 
signal, however, a range resolution can be achieved that is approximately 
the same as that possible with the undispersed pulse. This is accomplished 
by using a reverse of the dispersive delay on reception to recompress the 
returned pulses. 
In attempting to detect the presence of such radars and distinguish one 
from another, it is usually desirable to determine the chirp rate, or time 
rate of change of frequency, of the chirp radar as rapidly as possible. It 
has previously been proposed to analyze such signals by observing the 
frequency spectrum that results when they are applied to a compressive 
receiver, a device that responds to an input by generating an output whose 
time of occurrence depends on the frequency of the input. The output of a 
compressive receiver in response to a narrow-band signal is a short burst 
of the compressive-receiver center frequency, whereas the output in 
response to a chirp signal is a longer-duration burst. However, although 
the power spectrum of the output that results from a chirp signal can in 
some cases differ markedly from that resulting from a narrow-band signal, 
the difference is often minimal. Thus, it is often difficult by that 
method even to distinguish between chirp signals and narrow-band signals, 
and it is even more difficult to distinguish between chirp signals of 
different chirp rates. It is accordingly an object of the present 
invention to determine chirp rate automatically and in a reliable manner. 
SUMMARY OF THE INVENTION 
The foregoing and related objects are achieved in an apparatus that 
includes a compressive receiver and a frequency discriminator that 
receives the compressive-receiver output. The compressive receiver 
includes a frequency translator that receives input signals and repeatedly 
translates them in frequency at a constant sweep rate. The output of the 
frequency translator is applied to a linear dispersive delay line, which 
causes different delays for different frequencies. The frequency 
translator converts any single-frequency signal that it receives to an FM 
signal whose frequency is repeatedly swept at a rate that matches the 
delay-versus-frequency relationship of the dispersive delay line. That is, 
the frequency-translator outputs caused by later-arriving portions of the 
single-frequency signal within a given sweep are delayed by less than 
outputs from earlier-arriving portions, and the difference in delay is 
such that all frequency-translator outputs caused by frequency-translator 
inputs of a given frequency arrive at the delay-line output port 
simultaneously. Consequently, any single-frequency signal received by the 
frequency translator results in a compressed pulse in the output of the 
delay line, i.e., a pulse whose duration is a very small fraction of the 
delay-line output sweep time. The time within the sweep at which the 
output pulse occurs indicates the frequency of the signal that gives rise 
to it. 
If the signal received by the frequency translator is a chirp signal, the 
output of the delay line is itself a chirp signal, and we have found that 
the chirp rate of this output chirp is related to the chirp rate of the 
input chirp signal that caused it. By feeding the output of the delay line 
to a frequency discriminator, which provides an output voltage that is 
proportional to the instantaneous frequency of the delay-line output, it 
is possible to determine the chirp rate of the received signal by 
observing the time rate of change of the discriminator output.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
FIG. 1 illustrates a device for monitoring a range of frequencies and 
generating output ramp signals in response to chirp signals. The slope of 
a ramp is an indication of the chirp rate of the signal that gave rise to 
it, and the time of occurrence of the ramp is an indication of the 
frequency range of that signal. It is therefore possible to distinguish 
between chirp radars of different chirp rates by observing the slopes of 
the output ramp signals. 
An antenna 12 feeds its signals to a receiver front end 14, which restricts 
the signals to a pass band of interest and amplifies them. The front-end 
output signals are fed to a frequency translator 16, which includes a 
mixer 18 that multiplies those signals by signals from a local oscillator 
20. The local oscillator 20 is repeatedly swept in frequency; the 
frequency of its output begins at a relatively low value, increases 
linearly with time until it reaches an upper value, and then begins at the 
low value again. That is, a plot of local-oscillator frequency as a 
function of time would have a sawtooth shape. The mixer 18 transmits the 
resultant signal to a Gaussian weighting filter 22, whose output is 
applied to a dispersive delay line 24. The weighting filter 22 eliminates 
the lower sideband produced by the mixing operation, restricts the 
upper-sideband contributions to those within the effective bandwidth of 
the delay line, and weights the frequency components so that ringing, or 
sidelobes, are minimal in the delay-line outputs if the front-end outputs 
are narrow-band signals. 
The delay encounted by a signal in the delay line 24 is a linear function 
of the frequency of the signal. The sweep rate of the frequency translator 
16 is the negative of the ratio of frequency difference to the resultant 
difference in delay. This relationship of the delay-line characteristics 
to the frequency-translator sweep rate causes the delay-line output 
resulting from a single-frequency input to be compressed in time. 
Specifically, if the antenna 12 receives a CW signal (i.e., a 
single-frequency, single-amplitude signal) that lasts throughout a sweep 
of the local oscillator 20, the upper sideband of the resultant 
frequency-translator output is a frequency-modulated signal that lasts 
throughout the sweep period and has an instantaneous frequency (i.e., rate 
of phase advance) that increases linearly with time throughout the sweep. 
The weighting-filter output, on the other hand, lasts through only part of 
the sweep period, and its amplitude envelope is Gaussian. The time within 
the local-oscillator sweep at which that weighting-filter output component 
occurs depends on the frequency of the CW signal that gave rise to it. 
The higher-frequency components that a CW front-end signal causes in the 
delay-line input are launched later than lower-frequency components are. 
Because higher-frequency signals propagate more quickly through the 
dispersive delay line 24 than lower-frequency signals do, though, the 
higher-frequency components tend to catch up with the earlier-arriving 
lower-frequency components. More specifically, because the 
delay-to-frequency relationship in the delay line 24 is the negative of 
the local-oscillator sweep rate, the later-launched higher-frequency 
signals arrive at substantially the same time as that at which the 
earlier-launched lower-frequency signals do. Therefore, all components 
resulting from a CW front-end signal during the same local-oscillator 
sweep occur in the delay-line output at substantially the same time. 
CW signals of different frequencies undergo the same frequency translations 
and filtering; that is, their resultant delay-line inputs are all 
frequency-modulated signals of the same frequency range and envelope 
shape. However, although such delay-line inputs often overlap in time, 
they begin at different times, so the compressed delay-line output pulses 
that they cause occur at different times within the delay-line output 
sweep. Accordingly, the time at which a delay-line output pulse occurs is 
an indication of the frequency of the input signal that gave rise to it. 
The foregoing description is based on the assumption of a CW signal at the 
antenna 12, but it applies to narrow-band front-end signals generally. For 
broader-spectrum front-end signals, however, the operation differs 
somewhat. In the case of a chirp-radar signal, for instance, the chirp 
rate of the radar signal is typically at least a significant fraction of 
the local-oscillator sweep rate, and so the rate of frequency change of 
the resultant frequency-translator output no longer matches the delay-line 
relationship of frequency to delay. As a consequence, signals launched 
into the delay line 24 during different portions of a sweep do not reach 
its output port at substantially the same time. 
It has been proposed in the past to determine the chirp rate of a received 
chirp signal by observing the frequency spectrum that the chirp signal 
causes in the output of the dispersive delay line 24. The theory behind 
this proposal can be understood if one considers a chirp signal whose 
chirp rate is the negative of the frequency-translator sweep rate. For 
such a signal, the output of the frequency translator 16 is a 
constant-frequency signal, as are the outputs of the Gaussian filter 22 
and the dispersive delay line 24. Accordingly, the frequency spectrum of 
the delay-line output that results from such a signal is largely 
concentrated in a very narrow range. Typically, the particular frequency 
range changes from sweep to sweep, but the shape of the spectrum remains 
unchanged, and the chirp signal that causes it can thus be identified 
readily as one whose chirp rate is the negative of the 
frequency-translator sweep rate. Furthermore, the power spectrum of such a 
signal is clearly distinguishable from that caused by a CW signal; the 
delay-line output power spectrum caused by a CW front-end signal is 
broader and matches the curve of the Gaussian filter 22. 
Although such a scheme appears workable in the case of the specific chirp 
rate described above, the results are not so satisfactory in many 
instances. For a signal whose chirp rate is, say, one-quarter that of the 
local-oscillator sweep rate and lasts throughout the local-oscillator 
sweep, it can readily be seen that the output of the frequency translator 
16 could sweep through the same range of instantaneous frequencies as that 
which would result from a CW signal, the only difference being that the 
sweep would be faster so that the output of the delay line 24 would be 
spread in time. The frequency spectrum would be slightly broader than that 
resulting from a CW input signal. Howver, the same broad frequency 
spectrum might also arise from a signal with a lower chirp rate and an 
amplitude envelope of shorter duration. Accordingly, observation of the 
frequency spectrum of the compressive-receiver output has drawbacks as a 
method of distinguishing among different kinds of chirp radars on the 
basis of their chirp rates. 
According to the present invention, chirp radars are distinguished from 
each other on the basis of their chirp rates by observing the 
instantaneous frequency of the delay-line output. The discriminator 26 
generates a voltage whose value is a function of the instantaneous 
frequency of the signal that it receives. Compressive-receiver outputs 
resulting from narrow-band signals last for too short a time to produce a 
discriminator output if the discriminator 26 is appropriately filtered. 
Outputs generated in response to chirp signals, on the other hand, are 
ramps of significant duration, and the chirp rate of the antenna signal 
can be determined from the slope of the ramp. 
It can be shown that the relationship between input chirp rate at the 
antenna and the output chirp rate at the delay-line output port is given 
by the following equation: 
EQU k"=-k-k'/[(k'/k).sup.2 +a.sup.4 .pi..sup.2 k.sup.2 /W.sup.4 ](1) 
where: 
k is the local-oscillator sweep rate; 
k' is the input chirp rate; 
k" is the output chirp rate; 
W is the effective bandwidth of the delay line 24; 
a=(2/.pi.) (ln A).sup.1/2 ; and 
A is the ratio of the weighting-filter attenuation at the ends the 
delay-line pass band to that in the middle of the pass band. 
In FIG. 2, the abscissa represents k'/k, the ratio of the front-end output 
chirp rate to the local-oscillator sweep rate, while the ordinate 
represents k", the delay-line output chirp rate. The plot of FIG. 2 has 
three portions 28, 30, and 32. Portion 28 shows that, for positive input 
chirp rates above a certain minimum, the output chirp rate becomes less 
negative as the input chirp rate increases. Thus, as input chirp rate 
increases, the slope of the ramp signal at the output port of the 
discriminator 26 grows less steep, becoming asymptotic to a value that 
represents the local-oscillator sweep rate. 
For sweep rates that are negative and above a certain minimum magnitude, 
the relationship is described by portion 32, which indicates that, for 
relatively small negative input chirp rates, the output chirp rate has a 
high positive value, so the slope of the discriminator output is 
relatively steep. As the input chirp rate becomes increasingly negative, 
the slope becomes less steep until, when the input chirp rate is the 
negative of the local-oscillator sweep rate, the output chirp rate is zero 
and the discriminator output is a square wave; that is, its slope is zero 
during the portion of the sweep in which it responds to the input chirp 
signal. As the input chirp rate exceeds the negative of the 
local-oscillator sweep rate, the slope of the discriminator output changes 
sign to represent a negative output chirp and again becomes asymptotic to 
a value that represents the local-oscillator sweep rate. 
If only portions 28 and 32 of the curve of FIG. 2 are considered, input 
chirp rate is a single-valued function of output chirp rate; portion 28 
approaches the local-oscillator sweep rate from below, while portion 32 
approaches it from above. With portion 30 included, a second value is 
added, so, in theory, there is an ambiguity. 
But portion 30 represents the results of input signals whose chirp rates, 
for delay lines with reasonably high time-bandwidth products, are lower 
than those of any chirp signals of interest. Such signals result in 
delay-line outputs of very short duration. The discriminator 26 can be 
arranged to respond slowly enough that it does not produce significant 
discriminator outputs in response to such short-duration delay-line 
outputs. Therefore, input chirp rate is effectively a single-valued 
function of output chirp rate. 
A quantitative example will demonstrate the operation of the invention. 
Suppose that the system is intended to monitor the 30 MHz-60 MHz frequency 
band, that the delay line 24 has an effective pass band of 200 MHz to 250 
MHz, and that there is a difference of 50 msec between the delay 
experienced by a 200-MHz signal and that experienced by a 250-MHz signal. 
Suppose further that the attenuation of the weighting filter 22 at the 
ends of the delay-line pass band relative to that at its center is 60 db, 
i.e., that the factor a in equation (1) is approximately 1.67. With such 
parameters, the frequency translator would ordinarily be arranged to have 
a sweep rate of 1000 kHz/msec and a sweep range of 80 MHz, from 140 MHz to 
220 MHz. The time for a single frequency-translator sweep is thus 80 msec, 
and a new sweep might start once every 85 msec, which is known as the 
revisit time. The output sweep time for the delay line--i.e., the 
difference between the times of occurrence of a delay-line output caused 
by a 30-MHz antenna signal and that caused by a 60-MHz antenna signal--is 
30 msec, with a new sweep occurring once in each 85-msec revisit time. 
For such a system, the central portion 30 of FIG. 2 represents only input 
chirp rates on the order of 10 Hz/msec or less in magnitude. These are 
negligible chirp rates, and delay-line output pulses resulting from such 
input chirp pulses are exceedingly short in duration, being virtually 
indistinguishable from pulses caused by CW inputs. Thus, the discriminator 
26 can readily be arranged to respond slowly enough that inputs 
corresponding to the center portion 30 result in no observable 
discriminator output. 
If such a system receives 50-msec chirp pulses that sweep from 40 MHz to 50 
MHz, the input chirp rate is 200 kHz/msec, or one-fifth of the 
local-oscillator sweep rate. If the system of FIG. 1 receives one such 
chirp pulse that starts at the same time as a local-oscillator sweep does, 
the frequency-translator output initially has a frequency of 180 MHz, 
which is below the frequency range of the weighting filter 22. Therefore, 
the front-end chirp signal causes no significant delay-line input at the 
beginning of the local-oscillator sweep. At 16.7 msec into the 80-msec 
sweep, the mixer output frequency moves into the weighting-filter pass 
band, and significant mixer output is launched into the delay line 24. As 
the sweep progresses, the instantaneous frequency of the delay-line input 
increases at a rate, 1.2 MHz/msec, that is equal to the sum of the input 
chirp rate and the local-oscillator sweep rate, and the amplitude of the 
delay-line input varies as a Gaussian function of time, stopping when the 
front-end output does, i.e., at 50 msec into the 80-msec 
frequency-translator sweep period. This occurs before the mixer output 
frequency has reached the 60 db-down frequency of the weighting filter. 
The duration of delay-line input is thus about 33.3 msec. During this 
time, the delay-line input frequency sweeps from 200 MHz to 240 MHz. 
The 240-MHz portion of the signal takes 40 msec less to travel to the 
delay-line output port than the 200-MHz portion does. Since the 240-MHz 
portion arrives at the delay-line input port only 33.3 msec later than the 
200-MHz portion does, it arrives at the delay-line input port 6.7 msec 
earlier, so the output resulting from the chirp signal lasts for 
approximately 6.7 msec of the 30-msec output sweep period and has a chirp 
rate of -6 MHz/msec. Accordingly, the discriminator output is a ramp that 
starts at 10 msec into the delay-line output sweep, lasts for 6.7 msec, 
and has a slope that represents -6 MHz/msec. Appropriate circuitry can 
then be used to measure the slope (e.g., by differentiating the 
discriminator output and measuring the resulting value) and determine the 
input chirp rate in accordance with equation (1), which, for input chirp 
rates on the order of a kilohertz per millisecond or more, is 
approximately equivalent to: 
EQU k"=-k(1+k/k'). 
This implies that 
EQU k'=-k.sup.2 /(k"+k). (2) 
Applying equation (2) for k"=-6 MHz/msec and k=1 MHz/msec yields k'=200 
kHz/msec, which is the correct input chirp rate. 
Two observations should be made at this point concerning the foregoing 
example. The first is that the duration of the discriminator output ramp 
and the times at which it begins and ends depend on the relative timing 
between the input chirp signal and the local-oscillator sweep. This does 
not present a significant problem, however, because the duration of the 
ramp does not affect its slope, which is the feature of most interest. 
Furthermore, the output ramps caused by different relative timings always 
begin and end within the same relatively definite segment of the 
delay-line output sweep. In the example, for instance, the ramp is always 
confined to the portion of the output sweep between 10 msec and 20 msec, 
which corresponds to the input-signal frequency range of 40 MHz-50 MHz. In 
fact, it either begins at 10 msec or ends at 20 msec, so the frequency 
range of the chirp signal, which can also be of interest, can be inferred 
from observation of repeated sweeps. 
The second observation is that the foregoing example is somewhat simplified 
in that it does not take into account the effects of any amplitude 
modulation of the incoming chirp signal, and chirp pulses ordinarily have 
smoothly changing envelopes rather than square-wave envelopes. The effect 
of such modulation is to blur the 10-msec and 20-msec boundaries slightly. 
However, the blurring is not significant, and the frequency range can 
still be determined with reasonable accuracy. 
As a result of the foregoing description, it is apparent that, by employing 
appropriate circuitry for detecting the slope of the discriminator output 
and applying the function represented by the plot of FIG. 2, it is 
possible to identify the frequency range and chirp rate of a received 
chirp signal in an automatic fashion. Thus, the present invention enables 
rapid identification of chirp radars and similar frequency-modulated 
sources.