An ultra-wideband magnetic antenna includes a planar conductor having a first and a second slot about an axis. The slots are substantially leaf-shaped having a varying width along the axis. The slots are interconnected along the axis. A cross polarized antenna system is comprised of an ultra-wideband magnetic antenna and an ultra-wideband dipole antenna. The magnetic antenna and the dipole antenna are positioned substantially close to each other and they create a cross polarized field pattern. The present invention provides isolation between a transmitter and a receiver in an ultra-wideband system. Additionally, the present invention allows isolation among radiating elements in an array antenna system.

BACKGROUND OF THE INVENTION 
1. Field of the Invention 
This invention generally relates to antennas, and more specifically to an 
ultra-wideband magnetic antenna. 
2. Related Art 
Recent advances in communications technology have enabled communication and 
radar systems to provide ultra-wideband channels. Among the numerous 
benefits of ultra-wideband channels are increased channelization, 
resistance to jamming and low probability of detection. 
The benefits of ultra-wideband systems have been demonstrated in part by an 
emerging, revolutionary ultra-wideband technology called impulse radio 
communications systems (hereinafter called impulse radio). Impulse radio 
was first fully described in a series of patents, including U.S. Pat. No. 
4,641,317 (issued Feb. 3, 1987), U.S. Pat. No. 4,813,057 (issued Mar. 14, 
1989) and U.S. Pat. No. 4,979,186 (issued Dec. 18, 1990) and U.S. patent 
application Ser. No. 07/368,831 (filed Jun. 20, 1989) all to Larry W. 
Fullerton. These patent documents are incorporated herein by reference. 
Basic impulse radio transmitters emit short Gaussian monocycle pulses with 
tightly controlled pulse-to-pulse intervals. Impulse radio systems can use 
pulse position modulation, which is a form of time modulation in which the 
value of each instantaneous sample of a modulating signal is caused to 
modulate the position in time of a pulse. 
For impulse radio communications, the pulse-to-pulse interval is varied on 
a pulse-by-pulse basis by two components: an information component and a 
pseudo-random code component. Generally, spread spectrum systems make use 
of pseudo-random codes to spread the normally narrow band information 
signal over a relatively wide band of frequencies. A spread spectrum 
receiver correlates these signals to retrieve the original information 
signal. Unlike spread spectrum systems, the pseudo-random code for impulse 
radio communications is not necessary for energy spreading because the 
monocycle pulses themselves have an inherently wide bandwidth. Instead, 
the pseudo-random code is used for channelization, energy smoothing in the 
frequency domain and jamming resistance. 
The impulse radio receiver is a homodyne receiver with a cross correlator 
front end. The front end coherently converts an electromagnetic pulse 
train of monocycle pulses to a baseband signal in a single stage. The 
baseband signal is the basic information channel for the basic impulse 
radio communications system, and is also referred to as the information 
bandwidth. The data rate of the impulse radio transmission is only a 
fraction of the periodic timing signal used as a time base. Each data bit 
time position modulates many pulses of the periodic timing signal. This 
yields a modulated, coded timing signal that comprises a train of 
identical pulses for each single data bit. The cross correlator of the 
impulse radio receiver integrates multiple pulses to recover the 
transmitted information. 
Ultra-wideband communications systems, such as the impulse radio, poses 
very substantial requirements on antennas. Many antennas are highly 
resonant operating over bandwidths of only a few percent. Such "tuned," 
narrow bandwidth antennas may be entirely satisfactory or even desirable 
for single frequency or narrow band applications. In many situations, 
however, wider bandwidths may be required. 
Traditionally when one made any substantial change in frequency, it became 
necessary to choose a different antenna or an antenna of different 
dimensions. This is not to say that wide band antennas do not, in general, 
exist. The volcano smoke unipole antenna and the twin Alpine horn antenna 
are examples of basic wideband antennas. The gradual, smooth transition 
from coaxial or twin line to a radiating structure can provide an almost 
constant input impedance over wide bandwidths. The high-frequency limit of 
the Alpine horn antenna may be said to occur when the transmission-line 
spacing d&gt;.lambda./10 and the low-frequency limit when the open end 
spacing D&lt;.lambda./2. These antennas, however, fail to meet the obvious 
goal of transmitting sufficiently short bursts, e.g., Gaussian monocycle 
pulses. Also, they are large, and thus impractical for most common uses. 
A broadband antenna, called conformal reverse bicone antenna (hereinafter 
referred to as the bicone antenna) suitable for impulse radio was 
described in U.S. Pat. No. 5,363,108 to Larry Fullerton. FIG. 1 
illustrates a front view of a bicone antenna 100. The bicone antenna 100 
radiates burst signals from impulses having a stepped voltage change 
occurring in one nanosecond or less. The bicone antenna 100 is basically a 
broadband dipole antenna having a pair of triangular shaped elements 104 
and 108 with closely adjacent bases. The base and the height of each 
element is approximately equal to a quarter wavelength (.lambda./4, where 
.lambda. is a wavelength) of an electromagnetic wave having a selected 
frequency. For example, in a bicone antenna designed to have a center 
frequency of 650 MHz, the base of each element is approximately four and a 
half inches (i.e., .lambda./4=four and a half inches) and the height of 
each element is approximately the same. 
Although, the bicone antenna 100 performs satisfactorily for impulse 
radios, further improvement is still desired. One area in which 
improvement is desired is reduction of unbalanced currents on the feed 
cable, e.g., a coaxial type cable, of a wide-band antenna. Generally, 
impulse radios operate at extremely high frequencies, typically at 1 GHz 
or higher. At such high frequencies, currents are excited on the outer 
feed cable because of the fields generated between the center conductor 
and the outside conductor. These currents are unbalanced having poorly 
controlled phase, thereby resulting in distorted ultra-wideband pulses. 
Such distorted ultra-wideband pulses have low frequency emissions that 
degrade detectability and cause problems in terms of frequency allocation. 
Generally, unbalanced currents on feed cables are filtered by balun 
transformers or RF chokes. However, at frequencies of 1 GHz or higher, it 
is extremely difficult to make balun transformers or RF chokes, due to 
degraded performance of ferrite materials. Furthermore, balun transformers 
suitable for use in ultra-wideband systems are difficult to design. As a 
result, unbalanced currents remain a concern in the design of ultra 
wide-band antennas. 
A second area where improvement is desired is the isolation of a 
transmitter from a receiver in an ultra-wideband communications system. 
Because the bicone antenna 100 generates a field pattern that is 
omni-directional in the azimuth, it is difficult to isolate a transmitter 
from a receiver. Additionally, isolation between antennas is desired where 
a plurality of antennas are arranged in an array. In an array system, 
isolation significantly reduces loading of one element by an adjacent 
element. 
For these reasons, many in the ultra-wideband communications environment 
has recognized a need for an improved antenna that provides a significant 
reduction in unbalanced currents in feed cables. There is also a need for 
an antenna suitable for ultra-wideband communication systems that provides 
improved isolation between transmitters and receivers as well as between 
antenna elements in an array system. 
SUMMARY OF THE INVENTION 
The present invention is directed to an ultra-wideband magnetic antenna. 
The antenna includes a planar conductor having a first and a second 
symmetrical slot about an axis. The slots are substantially leaf-shaped 
having a varying width along the axis. The slots are interconnected along 
the axis. A pair of terminals are located about the axis, each terminal 
being on opposite sides of said axis. 
The present invention provides a significant reduction in unbalanced 
currents on the outer feed cables of the antenna, which reduces distorted 
and low frequency emissions. More importantly, reduction of unbalanced 
currents eliminates the need for balun transformers in the outer feed 
cables. 
In one embodiment of the present invention, a cross polarized antenna 
system is comprised of an ultra-wideband magnetic antenna and an 
ultra-wideband regular dipole antenna. The magnetic antenna and the 
regular dipole antenna are positioned substantially close together and 
they create a cross polarized field pattern. 
Furthermore, the present invention provides isolation between a transmitter 
and a receiver in an ultra-wideband system. Additionally, the present 
invention allows isolation among radiating elements in an array antenna 
system. 
Further features and advantages of the present invention, as well as the 
structure and operation of various embodiments of the present invention, 
are described in detail below with reference to the accompanying drawings.

DETAILED DESCRIPTION OF THE EMBODIMENTS 
1. Overview and Discussion of the Invention 
The present invention is directed to an ultra-wideband magnetic antenna. 
Generally, a magnetic antenna is constructed by cutting a slot of the 
shape of an antenna in a conducting plane. The magnetic antenna, also 
known as a complementary antenna, operates under the principle that the 
radiation pattern of an antenna is the same as that of its complementary 
antenna, but that the electric and magnetic fields are interchanged. The 
radiation patterns have the same shape, but the directions of E and H 
fields are interchanged. The relationship between a regular antenna and 
its complementary magnetic antenna is illustrated in FIGS. 2-4. 
FIG. 2 shows a half wave-length dipole antenna 200 of width w being 
energized at the terminals FF as indicated in the figure. The antenna 200 
consists of two resonant .lambda./4 conductors connected to a 2-wire 
transmission line. 
FIG. 3 is a complementary magnetic antenna 300. In this arrangement, a 
.lambda./2 slot of width w is cut in a flat metal sheet. The antenna 300 
is energized at the terminals FF as indicated in FIG. 3. 
The patterns of the antenna 200 and the complementary antenna 300 are 
compared in FIG. 4. FIG. 4A shows the field pattern of the antenna 100 and 
FIG. 4B shows the field pattern of the complementary antenna 300. The flat 
conductor sheet of the complementary antenna is coincident with the xz 
plane, and the long dimension of the slot is in the x direction. The 
dipole is also coincident with the x axis as indicated. The field patterns 
have the same shape, as indicated, but the directions of E and H are 
interchanged. The solid arrows indicate the direction of the electric 
field E and the dashed arrows indicate the direction of the magnetic field 
H. 
2. The Invention 
FIG. 5 illustrates a complementary magnetic antenna 500 in accordance with 
one embodiment of the present invention. The antenna 500 includes a planar 
conductor 504, a pair of leaf-shaped slots 508 and 512, and terminals 516. 
The planar conductor 504 is shown to be rectangular, although other shapes 
are also possible. It is constructed of copper, aluminum or any other 
conductive material. The leaf-shaped slots 508 and 512 are positioned 
symmetrical to a horizontal axis A--A and vertical axis B--B. The slots 
are interconnected at the vertical axis B--B. The terminals 516 are 
located at the vertical axis B--B. The antenna 500 is energized at the 
terminals 516 by a feed cable such as a coaxial cable (not shown). In one 
embodiment of the present invention, the length and width of the planar 
conductor 504 is set at .lambda..sub.c /2 and .lambda..sub.c /4, 
respectively, where .lambda..sub.c is the wavelength of the center 
frequency of a selected bandwidth. Actually, the length and the width of 
the planar conductor 504 should preferrably be at least .lambda..sub.c /2 
and .lambda..sub.c /4 in order to prevent the antenna 500 from becomming a 
resonant antenna. In fact, the greater the length and the width of the 
planar conductor 504, the less resonant the antenna 500 will be. 
The bandwidth of the antenna 500 is primarily determined by the shape of 
the slots 508 and 512 and the thickness of the planar conductor 504 around 
the slot. Both the shape of the slot and the thickness of the planar 
conductor 504 around the slot was experimentally determined by the 
inventor. 
In the past, the inventor has experimented with dipole antennas, such as 
the resistively tapered bowtie antenna 600 shown in FIG. 6. Specifically, 
the antenna 600 comprises radiators 604 and 608, resistor sheet 612, and 
tapered resistive terminators 616 and 620. The tapered resistive 
terminators 616 and 620 create smooth transitions along the edges of the 
antenna 600. 
The resistor sheet 612 helps absorb some of the current flowing to the end 
of the dipole. The resistive loading dampens the signal so that the 
antenna 600 is less resonant and therefore, has a broader band-width. 
There is, however, a disadvantage; the resistive loading causes resistive 
loss which is dissipated as heat. In other words, the bandwidth of the 
antenna 600 is increased by resistive loading, but which also lowers the 
antenna radiation efficiency. The resistive loading results in an 
increasing impedance as the signal approaches the tip of the antenna 600. 
The signal reflects all along the tapered edge and not just the tip. This 
spreads the resonance in much the same manner as a tapered transmission 
line impedance transformer. 
From these experiments, it was recognized that smooth transitions in the 
shape of the dipole is an important factor in minimizing resonance, 
thereby increasing bandwidth. It was also recognized that one way to 
achieve smooth transitions would be to select a function that describes 
the shape of the dipole and its derivative as continuous as possible. 
Using empirical methods, a combination of exponential functions was 
initially selected to describe the shape of the dipole antenna. 
Later, this concept was applied to a complementary magnetic antenna. It was 
hypothesized that creating a smooth and continuous shape of the slot of a 
complementary magnetic antenna would result in an ultra-wideband antenna. 
Since the complement of the tapered bow-tie antenna had an unacceptably 
high input impedance (approximately 170 ohms), other shapes were 
investigated. 
Thereafter, a product of cosine functions were selected which ensured that 
their derivatives are also continuous. The inventor empirically developed 
the equation 
##EQU1## 
where f(l) is the width of the slot and l is the length of the slot. This 
equation provided a symmetric shape of the slot, thus resulting in a 
symmetric field pattern. Moreover, the antenna had an approximately 50 ohm 
impedance that is also the impedance of many coaxial cables, thereby 
eliminating the need for a standard balun transformer that is serving as 
an impedance transformer. Furthermore, the antenna could be easily 
modified to match a 70 ohm impedance by increasing the width of the gap 
slightly. 
The width of the conductor around the slot is determined by several 
factors. An ideal wideband complementary antenna has an infinite conductor 
sheet, while a narrow band loop antenna is constructed from a wire. 
Because an important objective of the present invention was to make the 
overall size of the antenna relatively small, the width of the conductor 
around the slot was reduced until the antenna began to resonate 
unacceptably. It was discovered that these resonances occurred when the 
tip of the slot was less than 1/4 inches from the edge of the conductor 
and the edge of the slot was less than 1 inch from the side of the 
conductor. It was hypothesized that a narrow conductor restricts the flow 
of current such that it performs like a loop radiator. In contrast, a 
broad conductor allows a family of loop currents, each having a distinct 
frequency, to flow around the slot, resulting in a ultra wide-band 
radiator. Based on the foregoing observations, an example embodiment of 
the antenna 500 was constructed having the following dimensions: 
______________________________________ 
length of the conductor plate 500 
5.25 inches 
width of the conductor plate 504 
2.5 inches 
combined length of slots 508 and 
4.6 inches 
512 
maximum width of slots 508 and 
0.62 inches 
512 
______________________________________ 
FIG. 7 shows the direction of surface currents (shown by a series of 
arrows) on the conductor plate 504. As indicated in FIG. 7, the surface 
currents originate at one of the terminals, flow around the slots 508 and 
512 and thereafter terminate at the other terminal. Thus, the surface 
currents form a series of loops around the slots 508 and 512. 
The antenna 500 offers several advantages over existing broad-band 
antennas. As noted previously, impulse radios and other ultra-wideband 
communication systems typically operate at extremely high frequencies, 
e.g., 1 GHz or higher. At such high frequencies, unbalanced currents are 
excited on the outer feed cable because of the fields generated between 
the center conductor and the outside conductor of a coaxial cable. The 
unbalanced currents degrade detectability and frequency allocation. 
In the past, unbalanced currents on feed cables were filtered (i.e., 
attenuated or blocked) by balun transformers or choked by ferrite beads or 
cores (ferrite beads or cores produce high impedance junction around feed 
cables). However, at operating frequencies of 1 GHz or higher, it is 
extremely difficult to make balun transformers or ferrite cores due to the 
performance of ferrite materials at these frequencies. An important 
advantage of the present invention is that the unbalanced currents are 
almost negligible on outer feed cables. 
Generally, in a regular dipole antenna having two radiating elements, the 
first radiating element is driven against the second radiating element 
(the ground side). The first radiating element is isolated from the second 
radiating element by an air gap or some other dielectric medium. This 
produces an electric field in the gap between the inner conductor and the 
outer conductor of the coaxial cable, thereby inducing unbalanced currents 
therein. In contrast, in a magnetic dipole antenna, both the slots are 
electrically connected by the surrounding conductor plate. For example, as 
indicated in FIG. 5, the slots 508 and 512 are electrically connected to 
each other by the surrounding conductor plate 504. Thus, unlike in a 
regular dipole antenna, one element of a magnetic antenna is not driven 
against another element of the magnetic antenna. This reduces unbalanced 
currents to a negligible level, thereby eliminating the need for ferrite 
cores in the outer feed cables. 
Another important feature of the present invention is that it can be used 
to construct a cross polarized antenna system. As noted before, the 
present invention is a magnetic antenna, and thus, its radiation patterns 
have the same shape as the radiation patterns of its complementary dipole 
antenna, but the directions of E and H are interchanged. This allows the 
construction of a cross polarized antenna system by positioning an 
ultra-wideband dipole antenna and a complementary magnetic antenna side by 
side, while keeping the form factor fairly small and their phase centers 
close together. Such a cross polarized system can be used in cross 
polarized feeds for channelization and ground penetrating radars. 
Additionally, a cross polarized antenna system can provide polarization 
diversification. Several embodiments of cross polarized systems are 
briefly described, infra. 
FIG. 8 shows a cross polarized antenna system 800 according to one 
embodiment of the present invention. As indicated in FIG. 8, the cross 
polarized antenna system is comprised of an ultra wide-band magnetic 
antenna 804 and an ultra-wideband dipole antenna 808 positioned end to 
end. Another embodiment of a cross polarized antenna is shown in FIG. 9. 
In this embodiment, an ultra wide-band magnetic antenna 904 and an 
ultra-wideband dipole antenna 908 are positioned side by side. In both 
these embodiments, additional gain can be obtained by placing a back 
reflector. FIG. 10 shows a cross polarized antenna system 1000 having a 
back reflector 1004. The back reflector 1004 also provides improved 
directionality by producing field patterns on only one side of the antenna 
system 800. 
FIG. 11 shows yet another embodiment of a cross polarized antenna system 
1100 in accordance with the present invention. As indicated in FIG. 11, an 
ultra-wideband magnetic antenna 1104 is placed facing an ultra-wideband 
dipole antenna 1108. Since the antenna 1104 comprises a conductor plate, 
it acts as a back reflector to the antenna 1108. The net result is a 
highly compact ultra wide-band cross polarized antenna that can also be 
used to feed a parabolic dish. The spacing between the antennas is based 
on empirical measurements. Specifically, the ultra-wideband antenna 
requires a 0.44 .lambda. gap in order to maximize the peak signal. 
Experimental results have indicated that the cross polarized antenna 
system 1100 performed satisfactorily. Although conventional wisdom would 
indicate that the antenna 1108 would block signals from the antenna 1104, 
it was discovered that the cross polarized antenna system 1100 performed 
satisfactorily. This is attributed to the fact that the polarization of 
both the antennas' 1104 and 1108 are linear even though each antenna has a 
planar structure. 
Yet another feature of the present invention is that it allows isolation of 
a transmitter from a receiver. As noted before, the bicone antenna of FIG. 
1 generates a field pattern that is omni-directional in the azimuth, 
thereby making it difficult to isolate a transmitter from a receiver. 
Since the magnetic antenna 500 according to the present invention produces 
a null in the conductor plate 504, a transmitter and a receiver can be 
appropriately placed so that they are isolated from one another. This 
feature is also useful in array systems where it is often desirable to 
isolate one antenna element from another in order to prevent 
electromagnetic loading by adjacent elements. Because the antenna 500 does 
not radiate from the side (due to the null along the A--A axis in FIG. 5), 
it reduces loading by adjacent elements, thereby significantly improving 
the performance. 
FIG. 12 shows a complementary magnetic antenna 1200 in accordance with the 
present invention constructed from a grid that was used for NEC (numeric 
electromagnetic code) simulation (a moment method simulation). The NEC 
simulation can be used to simulate the field patterns of the antenna 1200. 
FIG. 13 shows the simulated azimuth pattern of the antenna 1200. 
Experimental results of the azimuth pattern indicated that the antenna 
1200 has a peak to trough ratio of approximately 9 dB and HPBW of 
approximately 60 degrees. Thus, the simulation results closely correspond 
to the experimental results. FIG. 14 shows the simulated elevation pattern 
of the antenna 1200 in the x-z plane. Experimental results of the 
elevation pattern indicated that the antenna 1200 has a HPBW of 
approximately 70 degrees that closely corresponds to the simulation 
results. Finally, FIG. 15 shows the simulated elevation pattern of the 
antenna 1200 in the y-z plane. 
While various embodiments of the present invention have been described 
above, it should be understood that they have been presented by way of 
example only, and not limitation. Thus, the breadth and scope of the 
present invention should not be limited by any of the above-described 
exemplary embodiments, but should be defined only in accordance with the 
following claims and their equivalents.