Systems and methods for identifying polarization/quadrature channels in dual-polarization coherent optical transmission

Described are systems and methods for identifying the phase and polarization of independent modulation streams in quadrature channels of a coherent transmission system by using digital code. As a result, phase rotation and polarization of streams that during transmission may have become rotated and swapped around in the channel are correctly labeled and depermuted according to a known and predictable order.

BACKGROUND

The present disclosure relates generally to signal processing in high-speed communication circuits. More particularly, the present invention relates to systems and methods for correctly identifying polarization and quadrature channels in coherent optical transmission schemes.

In the past few decades, telecommunication networks have seen an ever-increasing demand for bandwidth. Large available bandwidth is a major factor in the increasing popularity of high-speed optical communication systems—whether for transferring data chip-to-chip or between Wide Area Network (WAN) fiber-optic links. For example, optical transceivers designed for short-distance (few hundred meters) interconnects over optical fiber are in high demand in data center and campus networks.

Coherent optical links communicate data over different channels that correspond to different phases and polarization sates of an input signal to the optical fiber. For example, in a dual-polarization coherent optical transmission system, X- and Y-polarization channels, ideally, carry independent in-phase (I) and quadrature phases (Q) of the X- and Y-polarizations, conventionally denoted as tributaries XI, XQ, YI and YQ—one for each branch. Using 4-PAM on each branch, yields two bits for every modulation unit interval (UI), and the combination of all four branches yields a total of 8 bits per UI.

However, characteristics that are inherent to the transmitter, receiver, and optical fiber introduce delays that cause phase and polarization in the four channels to arrive at the receiver with unknown phase rotation and polarization state that the receiver cannot identify from the recovered information. While by applying suitable signal processing to the four received streams of information, one may ensure the streams are received with a high degree of orthogonality, i.e., the data streams are successfully unmixed, thus, yielding separate information, a remaining ambiguity lies in the fact that the X- and Y-streams may have been flipped, inverted, and swapped around in arbitrary ways. For example, a 180° rotation in a channel amounts to negating the signal. Because each individual XI, XQ, YI, and YQ stream may have been independently rotated by 90°, 180°, or 270°, there is no easy way to identify the four sets of independent information, even if the data itself is perfectly valid. This results in 1+2+2=5 bits of uncertainty when identifying4output channels at the receiver and, thus, an unwanted permutation of the recovered data is highly likely.

Some existing approaches add a pilot tone at the transmitter to one or more branches to aid in proper identification of individual channels. Yet, such approaches have several shortcomings, including that the added pilot tones can interfere with other signals, such as for example those for lasers that perform signal tracking. Further, the polarity of a pilot tone, typically a sine wave, remains unknown because, unless additional timing information is made available, the pilot tone looks exactly like its negation.

Accordingly, it is desirable to have systems and methods that overcome the shortcomings of existing approaches and provide solutions for properly identifying and descrambling individual modulation streams.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

The use of certain terms in various places in the specification is for illustration and should not be construed as limiting. The terms “include,” “including,” “comprise,” and “comprising” shall be understood to be open terms and any lists the follow are examples and not meant to be limited to the listed items. All documents cited herein are incorporated by reference herein in their entirety.

It is noted that embodiments described herein are given mainly in the context of analog receivers and binary PN codes. Yet, one skilled in the art shall recognize that the teachings of the present disclosure are not limited to analog applications or to any type of code as digital signal processing (DSP) coherent receivers may also be used. Similarly, nonbinary digital codes, such as cyclically repeating ternary code, or M-ary modulation sequences over any arbitrary size of modulation alphabet, M, may equally be used.

FIG. 1is a block diagram of a conventional DP-QAM receiver architecture that is based on analog signal processing. Receiver100is a homodyne receiver driven by an on-channel laser (not shown) that acts as the local oscillator. The arrangement of the polarization beam splitter and 90° hybrids106is designed to provide balanced quadrature light outputs for each of the two orthogonal polarizations, conventionally labeled X and Y, which are incident on eight photodiodes108that are arranged in balanced pairs. This arrangement results in four bipolar photocurrents110that are amplified by respective Trans Impedance Amplifiers (TIAs)112, corresponding to I- and Q-phases of the X- and Y-polarizations, respectively. Thus, four branches of receiver100, i.e., XI, XQ, YI, YQ, are available for further signal processing in the analog domain.

However, imperfections inherent to the transmitter, receiver100, and optical fiber introduce unwanted delays and distortions. One type of distortion that a polarized optical input beam that passes through an optical fiber plant experiences relates to undesirable changes to the state of polarization (SOP) of the signal that occur during transmission. Once polarization and phase in the four channels are mixed into each other, they arrive at the receiver with random phase orientations and unknown phase polarization dimensions with no reference point that would allow one to identify a valid signal. As a result, the receiver may not correctly identify the recovered information, leading to data loss.

In order to avoid having to manipulate polarization states in the DSP domain, some designers have proposed implementing polarization control by using optical modulators. To facilitate this, a pilot or marker tone is added at the transmitter to label and track one of the phases of the two polarizations (e.g., the X-polarization, in-phase signal branch) as a reference, such that a control loop algorithm running in a low-power CPU can monitor and adjust the polarization states to correct for polarization rotations in two or three degrees of freedom.

A pilot tone (e.g., a 50 kHz sinusoidal signal) that has been superimposed onto the XI tributary at the transmitter is used to recover the SPO at the receiver, which low-pass filters the XQ, YI, and YQ signals and synchronously detects these signals in the four branches. Thus, the receiver monitors the amplitudes and signs of these signals, while assuming that carrier phase lock has already been achieved. Low-speed signal processing can then be used to adjust the polarization angles to reduce the unwanted pilot tone amplitudes, such that the receiver can compensate for polarization rotation in the fiber. However, this approach suffers from a number of drawbacks, including that marker tone detection is possible only after carrier phase recovery and that the carrier recovery depends on the polarization states having first been corrected, for example, to ensure that a QPSK constellation is available for detection.

Accordingly, there is a need for systems and methods that allow for reliable identification of polarization channels in coherent optical transmission systems.

FIG. 2shows the magnitude of an autocorrelation function of a common pilot signal utilized in existing designs. As previously mentioned, certain approaches attempt to eliminate the uncertainty in identifying X- and Y-polarization channels by applying a single tone to the XI or XQ phase of the X-polarization channel to provide sufficient information to correctly identify the XI and XQ axes. However, such a single tone is insufficient for identifying the polarity of the transmitted signal since for sinusoidal pilot tones, the following mathematical expression holds true:

Since the absolute phase of the transmitted pilot tone may not be known, the polarity of the received tone cannot be identified with sufficient certainty. Considering a complex pilot signal formed as p(t)=cos(ωt)+i·sin(2ωt), the autocorrelation of p(t) has a unique peak absolute value204over (ωt)=0 . . . 2π. Because peak absolute value204of the autocorrelation function of p(t) is unique over 0 . . . 2π, the argument202of the correlation peak yields the channel rotation without ambiguity. However, as shown inFIG. 2, the correlation sidelobes of the autocorrelation function of the pilot signal are relatively large and only −2.5 dB relative to the peak value.

It is noted that the ideal waveform for detection should have a cyclic autocorrelation that closely approximates an impulse function, e.g., a Dirac delta impulse. However, since this cannot be achieved by using simple pilot tones, in embodiments, relatively long binary PN codes are added to the tributary (e.g., linearly combined by using a summer) to serve as an alignment code or a label, as will be discussed next. Advantageously, due to the length of the sequence, one may correlate1000sof bits to positively identify the tributary and orientation with a high degree of certainty.

FIG. 3illustrates an exemplary PN code generating system according to embodiments of the present disclosure. One skilled in the art will appreciate that PN code generating system300may be implemented, for example, within an optical transmitter. As depicted inFIG. 3, PN code generating system300comprises PN code generator302, and PN phase shift encoder320, which, in embodiments, may comprise a summing element. The PN code generating system illustrated inFIG. 3is not limited to the constructional detail shown there or described in the accompanying text. For example, as those skilled in the art will appreciate, a suitable PN code generator need not be the 4QAM pilot PN code generator depicted inFIG. 3.

In operation, PN code generator302generates one or more PN codes that may be amplified (or attenuated) by gain element310. PN codes312may be sequences of bits that comprise maximal length sequences. These sequences may be generated, for example, by using linear feedback shift registers that have maximal length. In embodiments, PN phase shift encoder320receives PN code312and data signal304, e.g., a 16QAM data signal and, in response, sums PN code312with data signal304, for example, in a predetermined proportion as determined by gain 310, or any other controlled factor, in order to output phase-shift-encoded PN codes330. The phase-shift-encoded PN codes330may thus use different phases of the same PN sequence to label the four independent polarization/phase channels304of a coherent transmission system.

In embodiments, by using the same PN code312(or waveform) at distinct relative phase offsets, zero mutual interference may be obtained due to the “perfect” autocorrelation properties of PN code312. For example, assuming a code length N, phase shift encoder320may encode XI, XQ, YI, and YQ of and data signal304with respective PN phase shifts of

{0,2,3,4}⁢(N+1)8
chips. In embodiments, half of the possible cyclic positions may be left void to enable the beginning of the sequence to be deduced as the first significant correlation peak after gaps of at least (N+1)/2 PN chips.

As discussed below with respect toFIG. 5, in embodiments, signs of correlation peaks may be used to deduce phase rotation, assumed to be a multiple of 90°. In embodiments, the order of correlation peaks indicates whether X- and Y-polarization channels in signal304have been swapped during transmission.

In embodiments, two PN codes312may be generated that have identical length, but different generator polynomials. The codes312may be generated by choosing from predetermined polynomial pairs that have been tabulated for Gold codes, such the maximum cross-correlation for the codes312is limited to |θ(a, b)|≤1+2(n+1)/2for odd n and |θ(a, b)|≤1+2(n+2)/2for even n. In embodiments, the first of a set of predetermined PN codes may be used to label the XI tributary, the other for the YI tributary. For example, for n=10,
p1(z)=Z10+Z3+1;p2(z)=Z10+Z8+Z3+Z2+1 and |θ(a,b)|≤65

The duration of each chip of PN code312should be an integer multiple, m, of the modulation UI. It is noted that the number of UI's per PN chip may be increased to limit the processing speed of the matched filter detector to a manageable value.

In embodiments, PN pilot code312may use an NRZ waveform having an amplitude that is significantly lower than the 4PAM inner symbols, e.g., ± 1/16, where the 4PAM symbols are nominally ±1; ±3, such that a correlation between reference patterns of 255 NRZ bits may be used to uniquely identify the XI and YI streams; and the angle of the correlation coefficient may identify the rotation.

FIG. 4is a flowchart of an illustrative process for generating PN codes for identifying the phase and polarization of independent modulation streams in quadrature channels according to embodiments of the present disclosure. In embodiments, process400begins at step402when, e.g., one or more PN codes are generated, for example, by a 4QAM pilot PN code generator.

In embodiments, at step404, the one or more PN codes may be amplified by a gain. A relatively long (e.g., 1023 bit) binary PN code provides about 30 dB process gain.

At step406, the one or more PN codes may then be combined with a data signal, such as a 16QAM data signal. In embodiments, an encoder may sum the PN code with the 16QAM data signal to obtain phase-shift-encoded PN codes that, at step408, are output by the encoder as labeled modulated data signals that identify signal polarity and identity.

FIG. 5illustrates an exemplary detecting system for identifying the phase and polarization of independent modulation streams in quadrature channels according to embodiments of the present disclosure. One skilled in the art will appreciate that detecting system600may be implemented in an optical receiver. Detecting system500comprises correlation peak detector504that, in embodiments, may be implemented as a set of matched filters, phase rotation detector510, reference pattern detector520, lookup table530, and remapping circuit540. As withFIG. 3, the details of detecting system500depicted inFIG. 5are not intended as a limitation on the scope of the present disclosure.

In operation, in response to receiving quadrature channel data and associated PN codes502, correlation peak detector504generates a correlation function that phase rotation detector510may use to determine signs of correlation peaks and infer therefrom phase rotations in one or more quadrature channels.

PN codes tend to come in a length of 2′−1, where n is the number of bits in the shift register. Assuming a 15-bit shift register, the sequence will have 32,767 bits. Knowing the PN code that is being sent, various embodiments use correlation peak detector to detect a very strong pulse that results from a matching sequence. It is noted that it would be very hard to achieve such a match purely by accident. Advantageously, this creates a very sensitive detector. Further, due to the relatively low magnitude, e.g., at or below noise level that the receiver sees, this may be accomplished without changing the, e.g., 16-QAM data (I and Q) itself and without having to sacrifice actual bits sent through the channel, thus, not affecting the regular operation of the receiver, because the constellation points are disturbed by a relatively insignificant amount that has no practical consequences.

In embodiments, correlation peak detector504may be implemented as a sliding window correlator that uses PN codes502as an input signal and applies, e.g., via a multiplier circuit, a matched filter that may use the PN sequence as a set of tap weights such that, e.g., after passing through a sliding integrator, a peak can be detected once a the PN sequence matches the tap weights. In embodiments, correlation peak detector may utilize an FIR filter as a matched filter peak detector and set tap weights of the FIR filter according to PN codes502to detect a peak (see, e.g.,FIG. 9). Since PN codes502may have a random phase relative to the PN generator, known CDMA receiver techniques may be used to scan and find an appropriate phase that delivers a peak. In embodiments, the “first” phase may be defined relative to an intentional gap in the correlation peaks designed to break the cyclic symmetry.

In embodiments, correlation peak detector504may further apply the correlation function to reference pattern detector520to obtain an order of correlation peaks that may be used to determine whether polarization channels in quadrature channel data502have been swapped and to map the quadrature channel data, e.g., by using remapping circuit540, to correctly detect, identify, and unscramble the XI and YI data streams. In embodiments, remapping circuit540comprises a decision circuit that may use, e.g., threshold detection to perform remapping or depermutation operations according to the information provided by the order of the correlation peaks.

In embodiments, lookup table530may be used to obtain the actual ordering and signs to detect a new ordering or mapping that potentially has occurred. Remapping circuit540, in effect, unscrambles the data streams. In embodiments, remapping circuit540may be implemented as a multiplexer that uses mapping532to ensure that the identified channels are properly (re)mapped to the output channels according to lookup table530. In embodiments, remapping circuit540may comprise a switch matrix that applies the information in lookup table530to the channel data to accomplish remapping. In embodiments, remapping circuit540outputs corrected quadrature channels542that are correctly labeled and depermuted according to a known and predictable order.

FIG. 6is a flowchart of an illustrative process for identifying the phase and polarization of independent modulation streams in quadrature channels according to embodiments of the present disclosure. In embodiments, process600begins at step602when, quadrature channel data and associated PN codes are received by a correlation peak detector, e.g., matched filters.

At step604, the correlation peak detector may apply a correlation function to generate correlation data.

At step606, the correlation data may be applied to a phase rotation detector to obtain phase rotation data.

At step608, the correlation data may be applied to a reference pattern detector to obtain a mapping of the quadrature channel data.

Finally, at step610, the phase rotation and order is used to properly remap the quadrature channel data.

It is noted that certain steps may optionally be performed; steps may not be limited to the specific order set forth herein; certain steps may be performed in different orders; and certain steps may be performed concurrently. It is further noted that embodiments disclosed herein apply to analog and/or digital implementations since digital tuning systems and methods may equally take advantage of the teachings of the present disclosure to identify polarization and quadrature channel data.

FIG. 7illustrates how SNR degradation due to pilot codes affects constellation points in an exemplary 16-QAM constellation according to embodiments of the present disclosure. Each of the 16-QAM constellation points302inFIG. 7is modulated away from its nominal value by the 4-QAM pilot signal (NRZ in I & Q).
PilotC/I=10 log10(2/(10×162)=−31 dB

The SNR including the pilot tone, assuming that the SNR of the receiver is 24 dB, is given by the expression:
−10 log10(10−24/10+2/(10×162))=23.22 dB.∴SNRloss is 0.78 dB.

FIG. 8shows the result of 50 simulations with random polarization using PN-pilot based stream recognition according to embodiments of the present disclosure. Each simulation was run for 21816-QAM symbols. It is noted that the experimental results are provided by way of illustration and were performed under specific conditions using a specific embodiment or embodiments. Accordingly, neither these experiments nor their results shall be used to limit the scope of the disclosure of the current patent document.

Depicted inFIG. 8are example outputs of four matched filters. For each receiver branch, branches 1-4, one output is shown. The order in which correlation peaks801-804occur after a gap may be seen as {2,1,3,4}, and the signs as {−, +, −, −}. This, together with truth tableFIG. 9, provides sufficient information to determine whether X- and Y-polarizations have been swapped during transmission and to determine the rotation of each branch. The simulation results confirm that the method has minimal impact on BER. Scaling and other design parameters may be further optimized to ease hardware design concerns, such as the PN-code chip rate.