Directly mixing receiving system

In a broadcast signal receiving system, which includes a phase control loop composed of a first mixer having two inputs and an output and arranged to provide at its output a signal proportional to the product of the signals applied to its two inputs, one output being supplied with the received high frequency signal, a voltage controlled oscillator connected for supplying to the other input of the first mixer an alternating voltage of substantially constant amplitude, a first lowpass filter connected to the output of the first mixer for providing a filtered output signal representative of the first mixer output signal with the sum frequency component of that signal suppressed, and means for delivering the output signal of the first low-pass filter to the control input of the oscillator, the received high frequency signal is directly demodulated by multiplication with a local oscillator signal synchronized to a substantially fixed difference to the phase or the carrier phase of the received high frequency signal.

BACKGROUND OF THE INVENTION 
The present invention relates in contradistinction to superheterodyne 
receivers to a directly mixing receiving system in which a high frequency 
(HF) received signal is converted directly to the base band by 
multiplication with a synchronized local oscillator signal. The baseband 
is the frequency band seized by the information to be transmitted, for 
example the video band from DC to 6 MHz for television or the audioband 
from 20 Hz to 20 kHz at FM-broadcasting. 
Systems of the type to which the invention is directed employ a phase 
locked loop (PLL) phase control circuit including a mixer connected in 
series with a lowpass filter for suppressing the portion of the mixed 
product at the sum frequency and a voltage controlled oscillator (VCO) 
whose signal frequency or phase can be tuned to or locked on the immediate 
vicinity of the frequency, or the precise frequency, or a fixed phase 
difference with respect to the phase of the received HF signal, or of its 
existing or imaginary carrier. 
Such a receiving system is disclosed in the article "Phase Locked AM Radio 
Receiver" by L. P. Chu in IEEE Transactions on Broadcast and TV Receivers, 
Vol. 15, 1969, pages 300-308. This receiver employs a complicated Costas 
phase control loop including a first mixer in which a local oscillator 
signal in phase with the carrier of the received signal is mixed with the 
received signal, a first lowpass filter with series-connected low 
frequency (LF) amplifier, a second mixer in which the local oscillator 
signal delayed by 90.degree. is mixed with the received signal and which 
is connected in series with a second lowpass filter and a second LF 
amplifier, as well as a phase detector which compares the outputs of the 
two LF amplifiers and feeds the result to a control filter with 
series-connected varactor which controls the local oscillator. Thus, the 
Costas control loop includes two control loops, i.e. an "in-phase channel" 
and a "quadrature channel." The Costas loop is locked in the correct phase 
if a zero signal is produced in the quadrature channel. The demodulated LF 
signal can be obtained at the output of the in-phase channel. The circuit 
is designed to receive double sideband, amplitude modulated, received HF 
signals as defined by the mathematical derivation of the mode of operation 
of this synchronous demodulator at page 301, right-hand column. The 
above-mentioned article does not contain any mention of the usability of 
such a synchronous demodulator in a system for receiving signals modulated 
in another manner. 
SUMMARY OF THE INVENTION 
It is an object of the present invention to adapt a directly mixing 
synchronous receiving system, with its excellent noise properties, to 
other types of modulation. 
A further object of the invention is to provide inexpensive and optimum 
circuits and to construct such circuits for any desired types of 
modulation. 
These and other objects are achieved, according to the present invention, 
in a directly mixing broadcast signal receiving system, in which a 
received modulated high frequency signal is directly demodulated by 
multiplication with a synchronized local oscillator signal, which system 
includes a phase control loop and means connected for delivering the 
received high frequency signal to the input of the phase control loop, 
which is composed of a first mixer having two inputs, the one of which is 
the input of the phase control loop, and an output and arranged to provide 
at its output a signal proportional to the product of the signals applied 
to its two inputs, a voltage controlled oscillator connected for supplying 
to the other input of the first mixer an alternating voltage of 
substantially constant amplitude, a first lowpass filter connected to the 
output of the first mixer for providing a filtered output signal 
representative of the first mixer output signal with the sum frequency 
component of that signal suppressed, and means for delivering the output 
signal of the first lowpass filter to the control input of the oscillator 
for causing the phase of its output signal to be synchronized to a 
difference of exactly or approximately 90.degree. to the phase of the 
receiving signal or to the carrier phase of the high frequency signal. 
The phase shift of approximately 90.degree. between the received high 
frequency signal component and the oscillator signal is an inherent 
property of phase control loops with multiplying mixers (described in: A 
Blanchard, Phase Locked Loops, John Wiley & Sons, New York, 1976, Chapter 
10.1.1.). 
The directly converting receiving system according to the present invention 
makes it possible to utilize, in a simple manner, the many advantages of a 
directly mixing synchronous receiving system also for frequency 
modulation, phase modulation, phase shift keying and single sideband 
amplitude modulation. By eliminating intermediate frequency band filters 
it is possible, for example, to produce, to a much greater degree than 
heretofore, radio and television receivers according to microelectronic, 
integrated techniques.

DESCRIPTION OF THE PREFERRED EMBODIMENTS 
FIG. 1 shows, for comparison purposes, a simple superheterodyne receiver 
including a preamplifier connected to receive the signal supplied by a 
receiving antenna, a local oscillator, a mixer having inputs connected to 
the outputs of the preamplifier and the local oscillator, an intermediate 
frequency amplifier connected to the output of the mixer and a 
conventional demodulator supplied by the IF amplifier. 
FIG. 2, in contrast, illustrates a directly mixing receiver according to 
the invention which also includes a receiving antenna supplying a 
preamplifier, as well as a phase control loop composed of a mixer, a d.c. 
and LF amplifier connected to the mixer output, a control filter connected 
to receive the output signal from the d.c. and LF amplifier, and a voltage 
controlled oscillator (VCO) controlled by the output signal from the 
control filter. A second, LF, amplifier is connected to the output of the 
d.c. and LF amplifier. The mixer inputs are connected to the preamplifier 
and VCO outputs. 
FIG. 3 shows a receiver according to the invention for single sideband 
amplitude modulated signals with vestigial carrier. The undesirable 
sideband of the received HF signal u.sub.E amplified to u.sub.p in a 
preamplifier P, is suppressed, if necessary, in a bandpass filter BF. A 
mixer M1, lowpass filter LP1, control filter RF and voltage controlled 
oscillator VCO are connected to form a phase control circuit PLL which 
settles, or locks, on the vestigial carrier of the input signal. The 
operation of the receiver according to the invention will now be explained 
in detail with the aid of its mathematical derivation. The HF signal is 
assumed to be: 
EQU u.sub.E (t)=u.sub.T sin (.omega..sub.T t+.PHI..sub.T)+u.sub.1 sin 
((.omega..sub.T +.omega..sub.LF)t+.PHI..sub.T +.PHI..sub.LF) (1) 
where 
u.sub.T is the peak amplitude of the vestigial carrier, 
.omega..sub.T is the carrier frequency, 
u.sub.1 is the peak amplitude of the sideband 
.omega..sub.LF is the low frequency of the modulation signal and 
.PHI..sub.T and .PHI..sub.LF are the phase angles of the carrier and 
modulation signals, respectively, at time 
EQU t=0. 
At the output of band filter BF there then exists: 
EQU u.sub.BF (t)=ku.sub.E (t). (2) 
k is the (Frequency dependent) voltage gain of the cascade circuit 
consisting of the preamplifier and the bandpass filter, the bandwidth of 
which is at least the transmission bandwidth of the signal to be received, 
for example 6 MHz for US-television. 
The oscillator output signal u.sub.os (t), in the settled state, is in 
quadrature to the vestigial carrier: 
EQU u.sub.os (t)=u.sub.os cos (.omega..sub.T t+.PHI..sub.T), (3) 
where u.sub.os is the VCO peak output voltage. 
The mixer M1 forms the product 
##EQU1## 
where k.sub.M1 is the mixer voltage gain. 
The lowpass filter LP1 suppresses the product portion of the double carrier 
frequency and the desired information is retained at the output of the 
lowpass filter: 
EQU u.sub.LP1 (t)=1/2k.sub.M1 ku.sub.os u.sub.1 sin (.omega..sub.LF 
t+.PHI..sub.LF). (5) 
In order to tune the receiver to a desired broadcast frequency, an adder is 
interposed between control filter RF and oscillator VCO and a tuning 
voltage U.sub.tuning generated in a conventional manner is supplied to the 
adder, where it is summed with the control filter output. The summed 
signal is supplied, in a conventional manner, to effect tuning of bandpass 
filter BP as well as of oscillator VCO. 
FIG. 4 shows the block circuit diagram of a directly mixing FM receiver. 
The circuit includes a preamplifier P and a broadband PLL composed of a 
mixer M1, lowpass filter LP1 and voltage controlled oscillator VCO. Thus, 
the receiver does not require a limiter amplifier and the complicated 
preselection and after selection heretofore associated therewith is 
eliminated. A further lowpass filter LP3 at the LF side serves to increase 
selectivity. This FM receiver and its operation will also be explained in 
detail with the aid of a mathematical derivation. The following signal is 
considered to be present at the output of preamplifier P: 
##EQU2## 
where u.sub.v is the peak voltage of the preamplified signal, s(.tau.) is 
the low frequency modulation signal, .tau. an integration variable and 
.DELTA..OMEGA. is the maximum frequency deviation from .omega..sub.T, the 
remaining terms being as defined above. 
The oscillator signal may be defined as: 
##EQU3## 
where .omega..sub.os, k.sub.os and .PHI..sub.os are the VCO output 
residual frequency, modulation sensitivity, and phase angle at time t=0, 
respectively. To receive u.sub.p (t), .omega..sub.os has to be tuned to 
.omega..sub.T. In the settled case, .PHI..sub.os becomes approximately 
.PHI..sub.T. The signal 
##EQU4## 
is then present at the output of the mixer M1. 
The lowpass filter LP1 suppresses the voltage component at the sum 
frequency, leaving 
##EQU5## 
In the locked case, the argument of the angle function becomes very small. 
Thus the following applies in good approximation: 
##EQU6## 
By using the abbreviated notation 
EQU .omega..sub.p =1/2k.sub.M1 k.sub.os u.sub.V u.sub.os 
equation (10), when differentiated, provides the following: 
EQU u.sub.LP1 (t)/.omega..sub.p +u.sub.LP1 (t)=.DELTA..OMEGA.s(t)/k.sub.os. 
(11) 
But this is the differential equation of a first order lowpass filter with 
the limit frequency .omega..sub.p /2.pi., which is actuated by the signal 
.DELTA..OMEGA.s(t)/k.sub.os. With a sufficiently high limit frequency 
.omega..sub.p /2.pi., u.sub.LP1 (t) is thus the demodulated low frequency 
information. The lowpass filter LP3 provides increased selectivity. The 
limit frequencies of the lowpass filters LP1 and LP3 and .omega..sub.p 
/2.pi. must be higher than the maximum information frequency to be 
transmitted to not distort the information. The limit frequency of LP3 
should not be much higher than the maximum information frequency. 
A further embodiment of the invention is the receiver circuit shown in FIG. 
5, which can be used to demodulate phase modulated signals which have a 
small maximum phase deviation. It includes a preamplifier P and a narrow 
band PLL composed of mixer M1, lowpass filter LP1, control filter RF and 
voltage controlled oscillator VCO. The lowpass filter LP3 serves the same 
purpose as in the circuit according to FIG. 4. 
The operation of this embodiment will also be explained mathematically. The 
received HF signal is assumed to be 
EQU u.sub.p (t)=u.sub.v sin (.omega..sub.T t+.PHI..sub.T +.PHI.(t)), (12) 
where .PHI.(t) is the phase excursion. 
If the VCO of the narrowband PLL settles on the carrier spectral line of 
the phase modulated signal, the following applies: 
EQU u.sub.os (t)=u.sub.os cos (.omega..sub.T t+.PHI..sub.T). (13) 
Consequently, the voltage 
##EQU7## 
is present at the output of the mixer M1. 
In the lowpass filter LP1 the voltage component at the sum frequency is 
suppressed and the lowpass filter output voltage is then 
EQU u.sub.LP1 (t)=1/2k.sub.M1 u.sub.V u.sub.os sin .PHI.(t). (15) 
For phase excursions .PHI.(t) which are small compared to 1, the following 
applies in good approximation: 
EQU u.sub.LP1 (1)=1/2k.sub.M1 u.sub.V u.sub.os .PHI.(t), (16) 
i.e. the demodulated low frequency information. 
FIG. 12 shows a circuit with Costas loop with which also phase modulated 
signals of low rise with or without carrier spectral lines in the input 
spectrum can be demodulated. This circuit is in contradistinction to the 
circuit of FIG. 5 also well suited for the demodulation of n fold PSK 
(Phase Shift Keying) signals. 
A further embodiment of the invention is shown in FIG. 6 which is well 
suited for the reception of phase modulated signals with larger phase 
excursions. The circuit includes a preamplifier P and a modified 
narrowband PLL containing a mixer which receives at its second input, as 
the mixing signal, the output signal from oscillator VCO which has been 
modulated with the low frequency signal in a phase modulator PM which is 
part of the phase control loop. The demodulated LF signal is available at 
the output of an integrator I connected in series between the lowpass 
filter LP1 and the modulation input of modulator PM. The operation of this 
circuit will now be explained by way of a mathematical derivation. The 
preamplified phase modulated signal is assumed to be 
EQU u.sub.p (t)=u.sub.V sin (.omega..sub.T t+.PHI.(t)+.PHI..sub.T). (17) 
The signal generated by the voltage controlled oscillator VCO is assumed to 
be 
##EQU8## 
where u.sub.RF is the output voltage of the control filter RF which is a 
slowly variable value. The control filter is a lowpass filter with a limit 
frequency, which has to be less than the minimum information frequency to 
be transmitted. 
The oscillator signal is modulated in the phase modulator PM with the 
output voltage u.sub.I (t) of the integrator I, yielding: 
##EQU9## 
where k.sub.p is the modulation sensitivity of the phase modulator PM. At 
the output of the mixer M1 there is then generated the voltage 
##EQU10## 
The lowpass filter LP1 suppresses the voltage component at the sum 
frequency and the following results: 
##EQU11## 
When the control loop locks, the argument of the sine function in equation 
(21) becomes very small compared to 1, i.e. sin .PHI..fwdarw..PHI., and 
moreover, due to the quadrature requirement, the following tendency 
arises: 
##EQU12## 
The following then applies: 
EQU u.sub.LP1 (t)=1/2k.sub.M1 u.sub.PM u.sub.V {.PHI.(t)-k.sub.p u.sub.I (t)}. 
(23) 
The voltage u.sub.I is the result of integration from the voltage u.sub.LP1 
: 
EQU u.sub.LP1 (t)=u.sub.I (t)/k.sub.I (24) 
where k.sub.I is the transfer characteristic of integrator I. 
Using the abbrevation 
EQU .omega..sub.pp =1/2k.sub.p k.sub.M1 k.sub.I u.sub.PM u.sub.V, (25) 
differentiation of equation (23) will then produce 
EQU u.sub.I (t)/.omega..sub.pp +u.sub.I (t)=.PHI.(t)/k.sub.p) (26) 
which corresponds to the differential equation of a lowpass filter at the 
limit frequency .omega..sub.pp/2.pi. and the input signal 
.PHI.(t)/k.sub.p. With a sufficiently high limit frequency, (i.e. higher 
than the maximum information frequency to be transmitted) u.sub.I (t) is 
thus the demodulated signal. If the phase modulator PM has high linearity 
and is poor in harmonics at the output, this demodulation circuit is 
impressive by its high degree of freedom from distortion and noise. 
According to a further embodiment of the invention, as shown in FIG. 7, 
double sideband amplitude modulation can be processed. The circuit 
includes a narrowband phase control loop PLL in which the oscillator 
signal is in quadrature to the carrier, and additionally a second mixer 
which receives the oscillator signal after it has been shifted by 
90.degree. in a phase shifter PH1, i.e. to be in phase with the carrier, 
and the received preamplified HF signal, and at whose series-connected 
lowpass filter LP2 the demodulated LF signal can be obtained. 
The operation of this circuit will be explained below with a mathematical 
derivation. The received signal as amplified in preamplifier P is assumed 
to be 
EQU u.sub.p =u.sub.V (1+m(t)) sin (.omega..sub.T t+.PHI..sub.T), (27) 
where m(t) is the instantaneous modulating function value. From the 
oscillator signal in quadrature to the carrier, 
EQU u.sub.os (t)=u.sub.os cos (.omega..sub.T t+.PHI..sub.T), (28) 
there then results at the phase shifter PH1 output the signal 
EQU u.sub.Q (t)=u.sub.os sin (.omega..sub.T t+.PHI..sub.T). (29) 
This signal is multiplied in mixer M2 with the signal u.sub.p, the result 
being 
##EQU13## 
The lowpass filter LP2 suppresses the mixing product component at the sum 
frequency and the result is: 
EQU u.sub.LP2 (t)=1/2k.sub.M2 u.sub.V u.sub.os (1+m(t)). (31) 
The alternating voltage component of this signal is the desired demodulated 
information. 
A circuit which can demodulate double sideband as well as single sideband 
AM transmissions is shown in FIG. 8. The input signal u.sub.E is amplified 
in a preamplifier P. In the subsequent tunable bandpass filter BF, 
undesired frequencies are suppressed, for example the undesired sideband. 
The band filter output signal u.sub.BF is fed to a Costas loop which 
includes mixers M1, M2 and M3, lowpass filters LP1 and LP2, control filter 
RF, voltage controlled oscillator VCO and 90.degree. phase shifter PH1. In 
the case of a double sideband, amplitude modulated signal, or in the case 
of a single sideband, amplitude modulated signal, where in the frequency 
position of the suppressed sideband a possible interference signal is 
suppressed sufficiently strongly by the bandfilter BF the demodulated 
signal can be obtained from the lowpass filter LP2. In the case of single 
sideband, amplitude modulated signal in which interfering signals are 
still present behind the band filter BF in the image frequency position, a 
compensation of the interference signals must take place. The signal at 
the band filter output is assumed to be 
EQU u.sub.BF (t)=u.sub.1 cos (.omega..sub.T t+.PHI..sub.T)+u.sub.2 cos 
((.omega..sub.T +.omega..sub.LF)t+.PHI..sub.T .PHI..sub.LF)+u.sub.3 cos 
((.omega..sub.T -.omega..sub.st)t+.PHI..sub.T -.PHI..sub.st). (32) 
The following then applies for the settled Costas loop: 
EQU u.sub.os (t)=u.sub.os sin (.omega..sub.T t+.PHI..sub.T) (33) 
EQU u.sub.Q (t)=u.sub.os cos (.omega..sub.T t+.PHI..sub.T). (34) 
With the mixer gains k.sub.M1 and k.sub.M2, there then appears at the mixer 
outputs 
EQU u.sub.M1 (t)=u.sub.BF (t)u.sub.os (t)k.sub.M1 (35) 
EQU u.sub.M2 (t)=u.sub.BF (t)u.sub.Q (t)k.sub.M2. (36) 
With identical mixer gains k.sub.M1 =k.sub.M2 and with suppression of the 
mixed product components at the sum frequencies by means of lowpass 
filters LP1 and LP2, respectively, the following voltages are present at 
the outputs of the lowpass filters: 
EQU u.sub.LP1 (t)=-1/2k.sub.M1 u.sub.os u.sub.2 sin (.omega..sub.LF 
t+.PHI..sub.LF)+1/2k.sub.M1 u.sub.os u.sub.3 sin (.omega..sub.st 
t+.PHI..sub.st) (37) 
EQU u.sub.LP2 (t)=+1/2k.sub.M1 u.sub.os u.sub.2 cos (.omega..sub.LF 
t+.PHI..sub.LF)+1/2k.sub.M1 u.sub.os u.sub.3 cos (.omega..sub.st 
t+.PHI..sub.st)+1/2k.sub.M1 u.sub.os u.sub.1. (38) 
Thus the upper sideband .omega..sub.LF as well as the lower sideband 
.omega..sub.st are generated at both lowpass filter outputs. One of the 
two sidebands, for example the lower sideband .omega..sub.st, can be 
suppressed by compensation due to the fixed phase relationship between the 
voltages u.sub.LP1 and u.sub.LP2. The compensation takes place in the 
compensation block K, for example in the manner that the signals u.sub.LP1 
and u.sub.LP2 are shifted by a further 90.degree. with respect to one 
another. By adding or subtracting these signals, the undesired sideband 
signal is compensated. 
One simple arrangement for effecting such compensation is shown in FIG. 9 
in which the output signal from filter LP2 is shifted in phase by 
90.degree. in a phase shifter PH2 and the shifted signal is added to or 
subtracted from the output signal from filter LP1. This method of 
compensation, however, has the drawback that the entire LF spectrum must 
be shifted in phase over a broad band. 
FIG. 10 shows an embodiment of a compensation block K which avoids this 
drawback and operates with a fixed frequency phase shifter PH2. The 
circuit, which has the lowpass filter outputs connected to its two inputs, 
generates, with mixers M4 and M5, fixed frequency local oscillator LO and 
the 90.degree. phase shifter PH2 and lowpass filters LP4 and LP5, single 
sideband signals which are bound to one another in rigid phase position 
and with the upper sideband suppressed. For this purpose, the lowpass 
filters must be highly selective. Advisably the radian frequency .OMEGA. 
of the local oscillator LO is therefore set only slightly above the 
maximum low frequency. With the local oscillator signal 
EQU u.sub.LO (t)=u.sub.LO cos (.OMEGA.t+.PSI.) (39) 
and the signal at the output of the phase shifter PH2 
EQU u.sub.LQ (t)=u.sub.LO sin (.OMEGA.t+.PSI.) (40) 
which is in quadrature to u.sub.LO (t), the voltages 
EQU u.sub.LP4 (t)=A{u.sub.2 sin 
(.OMEGA.-.omega..sub.LF)t+.PSI.-.PHI..sub.LF)-u.sub.3 sin 
((.OMEGA.-.omega..sub.st)t+.PSI.-.PHI..sub.st)} (41) 
EQU u.sub.LP5 (t)=A{u.sub.2 sin 
((.OMEGA.-.omega..sub.LF)t+.PSI.-.PHI..sub.LF)+u.sub.3 sin 
((.OMEGA.-.omega..sub.st)t+.PSI.-.PHI..sub.st)+2u.sub.1 sin 
(.OMEGA.t+.PSI.)}. (42) 
are obtained at the outputs of the lowpass filters LP4 and LP5, 
respectively, where A=1/4k.sub.M1 k.sub.M4 u.sub.os u.sub.LO is a 
constant. k.sub.M4 is the voltage gain of the mixers M4 and M5. 
By forming the sum or difference of these signals in an arithmetic unit SD, 
the undesirable sideband is compensated. The following applies for the sum 
formation: 
EQU u.sub.SD (t)=2Au.sub.2 sin 
((.OMEGA.-.omega..sub.LF)t+.PSI.-.PHI..sub.LF)+2Au.sub.1 sin 
(.OMEGA.t+.PHI.) (43) 
which constitutes a single sideband signal without interfering signal in 
the image frequency position. 
Demodulation of that signal is then effected in a circuit which operates 
analogously to the circuit of FIG. 3. Since, however, the carrier is 
already available, this portion of the circuit is simplified to a mixer 
M6, behind which there is connected a lowpass filter LP6 at whose output 
the desired information is present without interference. 
A further possible variation of the receiving circuit according to the 
invention is shown in FIG. 11 in which compensation of the interference is 
effected within the Costas loop. The circuit shown in FIG. 11 is 
essentially the circuit of FIG. 8 supplemented by the compensation block 
of FIG. 10, together with a further mixer M7 having its inputs connected 
to the output of arithmetic unit SD and the output of phase shifter PH2, 
and two further lowpass filters LP6 and LP7, each connected to the output 
of a respective one of mixers M6 and M7. In addition, in FIG. 11 the 
inputs to mixer M3 are connected to the outputs of the latter filters LP6 
and LP7. 
The directly mixing receivers according to the present invention are 
characterized by a higher sensitivity and usually better separation than 
superheterodyne receivers. However, this superiority can be utilized only 
if a number of structural and dimensioning rules are adhered to. 
The following criteria apply for the mixer M1 and for the mixer M2, if 
employed: 
1. The local oscillator (VCO) signal must be strongly decoupled from the HF 
input. This requires, as an optimum solution, a design of the mixer in a 
bridge or double bridge structure. 
2. At the output of the mixer only the product of the two signals to be 
mixed should be formed. Signals at other combination frequencies, the 
signals to be mixed themselves and their harmonics should be strongly 
suppressed. Thus a push-pull structure should be provided. 
3. The mixer output must be capable of processing frequencies from direct 
current up to values of at least one channel bandwidth, that is the 
maximum bandwidth of the signal to be received, e.g. 210 kHz for 
FM-broadcasting, without incurring any significant amount of phase 
distortion. A signal level dependent direct voltage value must be avoided. 
4. The mixers must be low in noise. Particular emphasis must be placed on 
low l/f noise on the low frequency side. 
5. The operating points of the mixers must be set so that the output signal 
amplitude depends linearly on the input signal amplitude and linearly or 
at most slightly nonlinearly respectively, on the local oscillator or 
mixed signal amplitude, respectively. When highly frequency selective 
preamplifiers are used, the latter requirement can be weakened. 
The local oscillator (VCO) should satisfy the following requirements: 
1. It should be tunable by means of external signals (DC and LF) over the 
entire frequency band to be received. 
2. It should generate harmonic signals with strongly suppressed harmonics. 
3. It should have an approximately constant local oscillator amplitude over 
the entire frequency band to be received; in particular it should have 
only very slight fluctuations in amplitude within one channel bandwidth. 
4. It is unavoidable, that the control input of a voltage controlled 
oscillator acts as lowpass filter. The limit frequency of this lowpass 
filter must, when the oscillator is used in phase control circuits, be at 
least ten times higher than the lowest limit frequency occurring in the 
control circuit, which will generally require a bridge structure. These 
other limit frequencies are created by the lowpass filters in the control 
circuit. There values depend on the actual application. 
5. It must be highly stable in frequency. 
6. It must have low phase noise. 
For the LF portion of the demodulator the following criteria must be 
observed: 
1. Low noise, particularly low l/f noise. 
2. If LF blocks lie within a control loop, they must, on the one hand, have 
a sufficient gain to permit perfect demodulation and, on the other hand, 
the loop amplification must not become so high that the loop becomes 
unstable. 
3. Selective means within the LF portion increase HF selectivity. This can 
sometimes considerably reduce the selectivity requirements for the HF 
preamplifier, e.g.: high order lowpass filters LP1, LP2 or RF can increase 
the selectivity of a direct mixing FM-receiver to an extent, that no 
selective means in the HF portion of the receiver are necessary. 
The HF preamplifier P should meet the following criteria: 
1. It should be low in reaction, so that the local oscillator signal which 
may reach the output of the preamplifier is suppressed in the reverse 
direction. 
2. It should be low in noise. 
3. It should be high in gain so that the noise factor of the subsequent 
demodulator will not significantly influence the total noise. 
4. It should be highly linear, and particularly the HF signals should not 
be limited within a given range. 
5. For receiving systems without automatic regulation or control of the 
control circuit parameters, automatic selective HF amplitude regulation 
should be considered for higher system demands. 
If these system rules are adhered to, the not insignificant problems 
regarding intermodulation behavior, locking on the own oscillator signal, 
locking on the harmonic of the oscillator signal, locking on signals at 
the harmonics of the input signal frequency and so on should be avoided. 
Directly mixing receiving systems according to the present invention, 
particularly for angle modulated signals, have the advantages that, 
contrary to general opinion, no limiting amplifier is required ahead of 
the demodulator and thus there are no expenditures for preselection and 
postselection. Moreover, the LF output power of the circuits shown in 
FIGS. 4 and 6 even without power regulation is independent of the HF 
signal amplitude if the input signal power does not fall below a minimum 
power P.sub.0. Due to the lowpass filter characteristic , the system 
bandwidth of these two circuits drops below the LF bandwidth if the input 
signal power drops to below P.sub.0. Thus the output signal to noise ratio 
is further improved at the expense of the information bandwidth. This 
adaptive behavior permits communication even if receivers with limiter 
amplifiers have long failed. 
Directly mixing receivers according to the present invention for television 
signals, one example of which is shown in FIG. 13, also have decisive 
advantages due to their simple design and mainly due to their particularly 
simplified matching of the filters which, except for the HF filters, are 
exclusively lowpass filters, while superheterodyne receivers require 
tuning of a complicated interconnected combination of resonant circuits. 
The circuit shown in FIG. 13 includes a single sideband receiver having the 
form of one of the receivers disclosed above, for example that shown in 
FIG. 3. The output of that receiver provides the demodulated combined 
video signal and the sound IF signal. To separate those signals, there are 
provided a lowpass filter LP constructed to pass only the combined video 
signal components and a bandpass filter BF constructed so that the center 
frequency of its passband is tuned to the audio IF carrier frequency and 
the width of its passband assures rejection of the combined video signal 
components. The output of filter BF is connected to an FM demodulator 
which can be of a conventional type or a synchronous demodulator according 
to the invention, e.g. that shown in FIG. 4. 
As a result of the elimination of the intermediate frequency stage, 
receivers according to the present invention also have no problems 
regarding image frequency stability and intermediate frequency stability. 
The proposed systems are easily adaptable to miniaturization according to 
integrated circuit techniques. 
EMBODIMENTS OF BLOCK CIRCUITS ILLUSTRATED IN FIGS. 2-13 
1. Preamplifiers 
As low noise broad band preamplifiers, amplifier modules like optimax 
AH-461 or ADH-559 from Alpha Industries Inc., Woburn, Mass., or different 
models of other manufactures can be used. 
2. Mixers 
Ring modulators are the most preferred mixers for the purposes of the 
circuits given in FIGS. 2-13. For high-frequency application, 
double-balanced bridge mixers with semiconductor diodes like model MD-142 
from Anzac Electronics, Waltham, Mass., or different models of other 
manufactures can be used. For frequencies up to 200 MHz, integrated 
circuit ring modulators like MC 1596 from Motorola Semiconductors Inc., 
Phoenix, Ariz. or different models of other manufactures are applicable. 
FIG. 14 shows an embodiment with MC 1596 designed for frequencies about 
100 MHz. For audio frequency applications, integrated circuits like 
Motorola MC 1594L or MC 1595L or different models of other manufactures 
can be used, typical embodiments of such of multipliers are described in: 
The Semiconductor Data Library, Vol. VI, edited by Motorola Semiconductors 
Inc., in the description of the integrated circuits MC 1594L and MC 1595 
L. 
3. Voltage controlled oscillators 
FIG. 15 shows the circuit diagram of a voltage controlled oscillator 
designed for frequencies of about 5.5 MHz. With the same circuit 
structure, it is possible to construct oscillators up to the UHF-range. 
Other oscillator structures are also applicable. 
4. DC- and LF-amplifiers, integrators, summing amplifiers 
For purposes of DC- and LF-amplification, integration and summation, 
circuits with operational amplifiers can be used for frequencies up to 
about 100 kHz. Circuit structures for these applications are given in the 
literature, e.g. J. Wait, L. Huelsman, G. Korn, Introduction to 
Operational Amplifiers Theory and Applications, McGraw-Hill, New York, 
1975. Low noise operational amplifiers like .mu.A 739 from Fairchild 
Semiconductor, Mountain view, Calif., or amplifiers with similar 
properties should be taken. FIG. 20 shows a typical DC- and LF-amplifier 
with differential input. For video amplifier applications, integrated 
amplifiers with Fairschild .mu.A 733 can be used. 
5. Control filters 
FIGS. 16 and 17 show two typical control filters. Each of them can be used 
to form the control filter block RF of the circuits of FIGS. 3, 5-8, 11, 
12. In the control filter of FIG. 17, an operational amplifier is used. 
One possible realization is the integrated circuit Fairchild .mu.A 739. 
6. Other filters 
As lowpass and bandpass filters, similar constructed filters, as they are 
used in state-of-the-art receivers, are applicable. 
7. 90.degree. phase shifter 
As 90.degree. phase shifter, a quadrature hybrid can be used. A typical 
realization is the hybrid JH-131 from Anzac Electronics, Waltham, Mass. 
The phases of the output signals of quadrature hybrids differ about 
90.degree. at specified input frequencies. FIG. 18 shows a realization, as 
it can be used in the circuits of FIGS. 7-8, 10-12. 
8. Phase modulator 
FIG. 19 shows a circuit diagram of a special phase modulator. It contains a 
voltage controlled oscillator, the residual frequency of which must be 
approximately the same as the residual frequency of the carrier to be 
modulated. Possible embodiments of voltage controlled oscillators are 
described above. The phase modulator of FIG. 19 is useful for frequencies 
up to 80 MHz. Its advantage is a highly linear phase-voltage transfer 
characteristic with maximum phase deviations up to .+-.2.pi.. 
With the aid of frequency multipliers (described in: V. Kroupa, Frequency 
Synthesis, Griffin & Cie., Ltd., London, 1973, chapter 2) it is possible 
to exceed the frequency and phase deviation range of the described phase 
modulator. 
It will be understood that the above description of the present invention 
is susceptible to various modifications, changes and adaptations, and the 
same are intended to be comprehended within the meaning and range of 
equivalents of the appended claims.