Clock and data recovery devices with fractional-N PLL

The present invention relates to data communication and electrical circuits. More specifically, embodiments of the present invention provide a clock and data recovery (CDR) architecture implementation for high data rate wireline communication links. In an embodiment, a CDR device includes a phase detector, a loop filter, and a fractional-N PLL. The fractional-N PLL generates output clock signal based on output of the loop filter. There are other embodiments as well.

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BACKGROUND OF THE INVENTION

The present invention relates to data communication and electrical circuits.

Over the last few decades, the use of communication networks has exploded. In the early days of the Internet, popular applications were limited to emails, bulletin board, and mostly informational and text-based web page surfing, and the amount of data transferred was usually relatively small. Today, Internet and mobile applications demand a huge amount of bandwidth for transferring photo, video, music, and other multimedia files. For example, a social network like Facebook processes more than 500 TB of data daily.

Clock and data recovery (CDR) devices are used in a wide range of applications. For example, to process data received over a communication network, a receiver relies on its CDR to generate a clock signal based on the received data. The performance of the receiver relies on the performance of its CDR and other components. Over the time, there have been many different types of CDR designs and implementations, however, they have been inadequate for the reasons explained below. Therefore, new and improved CDR devices are desired.

BRIEF SUMMARY OF THE INVENTION

The present invention relates to data communication and electrical circuits. More specifically, embodiments of the present invention provide a clock and data recovery (CDR) architecture implementation for high data rate wireline communication links. In an embodiment, a CDR device includes a phase detector, a loop filter, and a fractional-N PLL. The fractional-N PLL generates output clock signal based on output of the loop filter. There are other embodiments as well.

According to an embodiment, the present invention provides A clock data recovery (CDR) device, which includes a first phase detection module configured to receive digital input signal and determine a difference between phases of the digital input signal and an output clock signal. The device additionally includes a loop filter module coupled to the phase detector module and configured to generate a frequency control word. The device also includes a fractional-N phase lock loop (Frac-N PLL) configured to generate the output clock signal based on the frequency control word. The Frac-N PLL includes a second phase detection module coupled to a reference clock signal. The Frac-N PLL also includes a charge pump coupled to the second phase detection module. The Frac-N PLL additionally includes an analog loop filter coupled to the charge pump. The Frac-N PLL also includes a sigma delta modulator configured to process the frequency control word.

According to another embodiment, the present invention provides a clock data recovery (CDR) device that includes a first phase detection module configured to receive digital input signal and determine a difference between phases of the digital input signal and an output clock signal. The device also includes a loop filter module coupled to the phase detector module and configured to generate a frequency control word. The device additionally includes a fractional-N phase lock loop (Frac-N PLL) configured to generate the output clock signal based on the frequency control word, the Frac-N PLL comprising a high-pass filter for processing the frequency control word. The high-pass filter has an inverse filter and an anti-aliasing filter.

According to yet another embodiment, the present invention provides a clock data recovery (CDR) device, which has a first phase detection module configured to receive digital input signal and determine a difference between the digital input signal and an output clock signal. The device also includes a loop filter module coupled to the phase detector module and configured to generate a frequency control word. The device additionally includes a fractional-N phase lock loop (Frac-N PLL) configured to generate the output clock signal based on the frequency control word. The Frac-N PLL has a voltage controlled oscillator configured on a feedforward signal path and a current mode digital-to-analog converter.

It is to be appreciated that embodiments of the present invention provide many advantages over conventional techniques. Among other things, CDR devices implemented according to embodiments of the present invention are smaller and more efficient compared to existing devices (e.g., phase-interpolator or digitally controlled oscillator implementations). For example, compared to CDRs with digitally controlled oscillators, CDRs with fractional-N PLL have better immunity to various types of coupling noise. Additionally, a fractional-N PLL based CDR can provide independent gain suitable for the CDR device.

Embodiments of the present invention can be implemented in conjunction with existing systems and processes. For example, CDR devices implemented with Frac-N PLLs according to the present invention can be used for a wide range of applications and are compatible with existing systems and architectures. Additionally, CDR devices according to the present invention can be manufactured using existing manufacturing processes and equipment. There are other benefits as well.

The present invention achieves these benefits and others in the context of known technology. However, a further understanding of the nature and advantages of the present invention may be realized by reference to the latter portions of the specification and attached drawings.

DETAILED DESCRIPTION OF THE INVENTION

The present invention relates to data communication and electrical circuits. More specifically, embodiments of the present invention provide a clock and data recovery (CDR) architecture implementation for high data rate wireline communication links. In an embodiment, a CDR device includes a phase detector, a loop filter, and a fractional-N PLL. The fractional-N PLL generates output clock signal based on output of the loop filter. There are other embodiments as well.

Low-jitter CDR architectures are often an essential aspect for high data rate wireline receivers. While some of the early CDRs utilized analog voltage-controlled oscillator (VCO) based architectures, phase interpolator (PI)-based clock data recovery (CDR) modules have come to dominate CDR loop implementations due to their relative small-area (as compared to other designs and architectures) and digital control functionality. A shift towards analog-to-digital converter (ADC) based receivers (thus moving from analog to digital) has made digital control necessary for CDR loop implementations. Phase interpolators provide this functionality by converting digital phase control input signal to analog clock phase shift signal at the output. Performance of a PI in a CDR implementation requires high linearity, high output phase resolution, and high-clock-frequency operation. Unfortunately, designing highly linear PIs with high output phase resolution at high clock frequency is difficult.

As a result, CDR architectures utilizing PIs struggle to achieve the low-jitter performance criteria needed for high data rate operation. An alternative to PI-based CDR is a digitally controlled oscillator (DCO) based CDR architecture. The low-jitter requirement mentioned above precludes the use of ring oscillator based DCOs applications that require high data rates. For example, LC-based DCO (LC DCO) can potentially provide low jitter, but they are susceptible to electromagnetic coupling due to low CDR bandwidth achieved by typical high data rate links.

It is thus to be appreciated that embodiments of the present invention provide high-performance CDRs implemented with fractional-N (Frac-N) phase-lock loops (PLL). More specifically, embodiments of the present invention provide CDRs with wide bandwidth Frac-N PLLs to provide the functionalities of digitally controlled oscillators and reduce susceptibility of LC-VCOs to electromagnetic coupling, while providing low jitter digital phase shift capability to CDR loops. In an exemplary embodiment, a CDR architecture is designed to eliminate the need for phase interpolator (PI) by employing fractional-N phase-locked loop (PLL), and it is capable of achieving low-jitter performance critical for high-data rate applications. The present invention also provides methods to ensure CDR loop stability. The present invention additionally provides calibration technique to ensure robust operation. As an example, the following techniques are provided:Use of fractional-N (analog or digital) PLLs in CDR loops for phase and frequency tracking;Digital high-pass filter based technique for compensating adverse effects (such as stability and error transfer function peaking) caused due to introduction of Frac-N PLL in CDR loop;Charge pump feedforward techniques for compensating adverse effects (such as stability and error transfer function peaking) caused due to introduction of Frac-N PLL in CDR loop;A variation of this technique can be used for digital fractional-N PLLs;VCO feedforward techniques for compensating adverse effects (such as stability and error transfer function peaking) caused due to introduction of an analog or digital Frac-N PLL in CDR loop;VCO feedforward gain calibration techniques for in-situ calibration of parallel path gain mismatch in VCO feedforward scheme.

FIG. 1is a simplified diagram illustrating a fractional-N PLL according to an embodiment of the present invention. This diagram is merely an example, which should not unduly limit the scope of the claims. One of ordinary skill in the art would recognize many variations, alternatives, and modifications. The PLL includes a phase detector (PD) or phase and frequency detector (PFD), a charge pump (CP), an analog loop filter (ALF), a voltage-controlled oscillator (VCO), multi-modulus divider (MMD), and a digital ΔΣ modulator (ΔΣ). The input reference clock is denoted as CLKREF, and the output clock is denoted as CLKOUT. A high resolution digital frequency control signal, DCTRL, is converted to an integer division ratio, NDIV, by a digital ΔΣ modulator. The output of the multi-modulus divider (MMD) is divided clock signal denoted as CLKDIV. The output frequency, FOUT, of such a fractional-N PLL depends on the input reference frequency FREFand DCTRL, which is described in Equation 1 below:
FOUT=DCTRLFREFEquation 1:

It is to be noted that DCTRLcan be used to control not only the output frequency, but also the output phase of the Frac-N PLL.FIG. 2provides a small signal model of a multi-modulus divider according to embodiments of the present invention. This diagram is merely an example, which should not unduly limit the scope of the claims. One of ordinary skill in the art would recognize many variations, alternatives, and modifications. It can be shown that the overall output phase of the MMD is given by Equation 2 below:

where ΦDIVis the phase of the divider output, ΦOUTis the phase of the fractional-N PLL output that is input to the MMD, Nnomdenotes the nominal division ratio of the fractional-N PLL, and NDIVdenotes the instantaneous integer division ratio of MMD.

Equation 2 can be expressed as Equation 3:

where STF(z) is the signal transfer function of the digital ΔΣ modulator and DCTRLdenotes the digital frequency control input. If we assume an ideal reference clock for the Frac-N PLL (i.e., ΦDIV=0), the output phase is given by Equation 4 below:

Equation 4 shows that the output phase of the Frac-N PLL can be controlled using digital frequency control signal, DCTRL.

FIG. 3is a simplified diagram illustrating a DCO-based CDR loop. The phase difference between incoming data, DIN, and sampling clock CLKOUTis detected by a digital phase detector (Digital PD). As an example, a proportional-integral path based digital loop filter (Digital LF) is used to generate a digital frequency control word DCTRLfor the DCO. The output frequency FOUTof the DCO depends on DCTRL, and the output frequency (FOUT) of the DCO (characterized by a DCO gain, KDCO) is described in Equation 5 below:
FOUT=DCTRLKDCOEquation 5:

It is to be appreciated that there are similarities between DCO and Frac-N PLL. In particular, the functionality of Frac-N PLL in CDR implementations according to embodiments of the present invention is equivalent to a digitally controlled oscillator (DCO), with the functional characteristics KDCO=FREF. In various embodiments, a Frac-N PLL is used as a functional equivalent of a DCO in a CDR loop. Compared to a stand-alone LC DCO implementation, a wide bandwidth Frac-N PLL provides better immunity to VCO coupling noise. Due to its closed loop operation, it also provides a DCO implementation with a well-defined process independent DCO gain, KDCO.

FIG. 4is a simplified diagram illustrating a Frac-N PLL incorporated inside a CDR loop according to embodiments of the present invention. This diagram is merely an example, which should not unduly limit the scope of the claims. One of ordinary skill in the art would recognize many variations, alternatives, and modifications. The digital frequency control word DCTRL(generated for CDR loop) is used as a frequency control input for a Frac-N PLL. The output clock generated by Frac-N PLL, CLKOUT, is used as a sampling clock for the CDR loop.

As explained above, Frac-N PLL inFIG. 4provides functionalities and advantages of an LC DCO without its penalties as used in a CDR. As an example, Frac-N PLL includes a phase detector (PD) or a phase-frequency detector (PFD) for phase detection, a charge pump for compensation, an analog loop filter (ALF) for filtering. The Frac-N PLL additionally includes an oscillator, an MMD, and a digital ΔΣ modulator. It is to be appreciated that Frac-N PLL can be implemented in other ways as well. It is to be understood CDRs according to the present invention can be implemented using analog Frac-N PLLs or digital Frac-N PLLs.

The incorporation of a Frac-N PLL in a CDR also contributes to certain functional difference from an LC DCO implementation. After all, a Frac-N PLL is not the same as an LC DCO. More specifically, while a Frac-N PLL has well-defined gain at low frequencies, its gain rolls off at high frequencies due to low pass nature of the PLL.FIG. 5is a simplified diagram illustrating small signal phase domain model of a CDR loop with a DCO.

In case of an ideal DCO, the gain is independent of frequency, (i.e., KDCO(s)=KDCO, where s is the reference modulation frequency). The digital loop filter is chosen to achieve appropriate CDR closed loop bandwidth and jitter tolerance. On the other hand, when Frac-N PLL is used (instead of DCO), the gain is given by KDCO(s)=FREFG(s), where G(s) is effective low-pass transfer function of the Frac-N PLL. Due to its low-pass nature, G(s) introduces additional poles in the CDR loop transfer function, which adversely affects the CDR loop stability and creates peaking in the CDR error transfer function E(s), where E(s) is defined by E(s)=ΦE(s)/ΦIN(s), subsequently degrading the jitter tolerance of the CDR loop. Note that jitter tolerance and error transfer function are related as JTOL(s)=1/E(s).

FIG. 6is a simplified diagram illustrating a small signal model of Frac-N PLL functioning as a DCO according to embodiments of the present invention. This diagram is merely an example, which should not unduly limit the scope of the claims. One of ordinary skill in the art would recognize many variations, alternatives, and modifications. The function FOUT(s) represents the output frequency of the Frac-N PLL, while ΦOUT(s) represents its phase. The high-pass shaped quantization noise of the digital ΔΣ modulator is denoted as EQ,NTF(s). The behavior of the Frac-N PLL is described in Equation 6 below:

In Equation 6, LG(s) denotes the open loop gain of the Frac-N PLL. To reduce CDR error transfer function E(s) peaking, the implementation needs to increase the bandwidth of Frac-N PLL and by extending the bandwidth of G(s). This increased bandwidth, however, allows more noise from digital ΔΣ pass through. As a result, there is a trade-off between peaking of E(s) and the output jitter of the Frac-N PLL.FIG. 7is a plot illustrating CDR error transfer functions comparing ideal DCO and actual DCO implementation (e.g., with Frac-N PLL) with a pole that contributes to undesired error transfer function peaking.

FIG. 8provides plots illustrating CDR error transfer function peaking (top) and Frac-N PLL jitter (bottom) as function of Frac-N PLL bandwidth. As can be seen inFIG. 8, the peaking is reduced at higher PLL bandwidth at the cost of worse jitter performance. For optimal system performance, the implementation needs to reduce the bandwidth for FOUT(S)/EQ,NTF(s) while maximizing the bandwidth for FOUT(s)/DCTRL(s).

In various embodiments, the present invention provides feedforward techniques to compensate the bandwidth limitations of Frac-N PLL. The problem discussed in the previous section can be solved by introducing a feedforward path from DCTRL(s) to FOUT(s). Such a feedforward path can decouple the transfer function seen by DCTRL(s) from the transfer function seen by EQ,NTF(s). We propose the following methods for implementing the feedforward path.

According to a specific embodiment, a Frac-N PLL is implemented in a CDR with a digital high-pass filter.FIG. 9is a simplified diagram illustrating a small signal model for a Frac-N PLL according to embodiments of the present invention. This diagram is merely an example, which should not unduly limit the scope of the claims. One of ordinary skill in the art would recognize many variations, alternatives, and modifications. It is to be appreciated that the addition of a digital high-pass filter in the DCTRLpath can effectively counter the low-pass nature of G(s). In this case, the DCO (i.e., DCO equivalent implemented with Frac-N PLL as used in a CDR) transfer function is given by Equation 7 below:

It is to be noted that the transfer function from EQ,NTFto FOUTremains unchanged. For example, for H1(s)=G−1(s), Equation 8 below is obtained:

In practical implementations, a digital finite impulse response (FIR) filter can be used to approximately match the inverse of G(s). Furthermore, there is a possibility of noise in DCTRLsignal folding back in Frac-N PLL in-band frequency region due to up-sampling in the Frac-N PLL. To alleviate this undesirable effect of out of band noise, an anti-aliasing filter is also provided in the DCTRLpath. This anti-aliasing filter can also be implemented as a digital FIR filter.FIG. 10is a simplified diagram illustrating a digital high-pass filter according to embodiments of the present invention. This diagram is merely an example, which should not unduly limit the scope of the claims. One of ordinary skill in the art would recognize many variations, alternatives, and modifications. For example, a digital high pass filter H1(s) includes an inverse filter H1(z) and an anti-aliasing filter AA(z).

FIG. 11provides plots showing simulated CDR error transfer function peaking (top) and output jitter (bottom) as function of Frac-N PLL bandwidth with digital high-pass filter. It can be seen that the CDR error transfer function peaking becomes almost independent of Frac-N PLL bandwidth. This allows for a Frac-N PLL bandwidth that optimizes the jitter performance.

In certain embodiments, a CDR is implemented with Frac-N PLL and a charge pump, where the charge part is configured on the feedforward path.FIG. 12is a simplified diagram illustrating a small signal model for a Frac-N PLL with charge pump according to embodiments of the present invention. This diagram is merely an example, which should not unduly limit the scope of the claims. One of ordinary skill in the art would recognize many variations, alternatives, and modifications. The DCO (i.e., DCO equivalent implemented with Frac-N PLL as used in a CDR) transfer function is provided in Equation 9 below:

where α and ICPdenote phase detector and charge pump gain, respectively. It is to be noted that the transfer function from EQ,NTFto FOUTremains unchanged. If we choose

Equation 10 below is obtained:

This feedforward path implementation requires digital filter as well as a current mode ΔΣ digital-to-analog converter (IDAC).FIG. 13is a simplified diagram illustrating a Frac-N PLL implementation with a charge pump according to embodiments of the present invention. This diagram is merely an example, which should not unduly limit the scope of the claims. One of ordinary skill in the art would recognize many variations, alternatives, and modifications. The IDAC is configured with ΔΣ modulator in the signal path as shown. It is to be understood while PD and CP are shown in the same block inFIG. 13, they are implemented separately.

FIG. 14shows the simulated CDR error transfer function peaking (top) and output jitter (bottom) as function of Frac-N PLL bandwidth with charge pump in the feedforward path. It can be seen that the CDR error transfer function peaking becomes almost independent of Frac-N PLL bandwidth. This allows for choosing Frac-N PLL bandwidth that optimizes the jitter performance.

In various embodiments, a Frac-N PLL is implemented with a feed-forward VCO path as a part of a CDR device.FIG. 15is a simplified small signal model for a Frac-N PLL with a feed-forward VCO path according to embodiments of the present invention. This diagram is merely an example, which should not unduly limit the scope of the claims. One of ordinary skill in the art would recognize many variations, alternatives, and modifications. As shown inFIG. 15, the VCO (with gain KVCO) is configured on the feedforward path. The DCO (i.e., DCO equivalent implemented with Frac-N PLL as used in a CDR) transfer function is described in Equation 11 below:

where HLPF(s) denotes the equivalent low pass transfer function of the analog loop filter connected to charge pump output. It is to be noted that the transfer function from EQ,NTFto FOUTremains unchanged. If we choose

Since H3(s) is a constant, this feedforward path implementation requires only a ΔΣ digital-to-analog converter.FIG. 16is a simplified diagram illustrating a Frac-N PLL implemented with a feed-forward VCO path according to embodiments of the present invention. This diagram is merely an example, which should not unduly limit the scope of the claims. One of ordinary skill in the art would recognize many variations, alternatives, and modifications. For example, the ΔΣ digital-to-analog converter can be implemented as a switching capacitor in the VCO as shown inFIG. 16. For example, a digital gain scaling factor KVFFis utilized to implement H3(s) function.

FIG. 17shows the simulated CDR error transfer function peaking (top) and output jitter (bottom) as function of Frac-N PLL bandwidth with a VCO on the feedforward path. It can be seen that the CDR error transfer function peaking becomes almost independent of Frac-N PLL bandwidth. This allows for choosing a Frac-N PLL bandwidth that optimizes the jitter performance.

It is to be noted that a VCO contributes to Frac-N PLL feedforward gain variation. As discussed before, the feedforward gain must match with Frac-N PLL KVCOto achieve an all-pass transfer function from DCTRLto FOUT. This gain scaling is implemented in digital domain, whereas the KVCOof the analog VCO varies with process, temperature, and supply voltage. As a result, there is a mismatch between the digital gain scaling factor (KVFF) and the Frac-N PLL gain.FIG. 18shows the effect of gain mismatch on CDR error transfer function peaking. It indicates that for optimum performance, the digital path gain KVFFneeds to be calibrated to match the analog gain inside Frac-N PLL.

FIG. 19is a simplified block diagram illustrating feedforward gain calibration for Frac-N PLL implementation of CDR according to embodiments of the present invention. This diagram is merely an example, which should not unduly limit the scope of the claims. One of ordinary skill in the art would recognize many variations, alternatives, and modifications. More specifically, the foreground calibration technique illustrated inFIG. 9involves matching digital feedforward path gain with that of analog path. As discussed above, the gain in the feedforward path (e.g., the HVFF(s) function) is to be matched with the gain Frac-N PLL path (e.g., the HFRAC(S) function).

FIG. 20shows transfer functions for each of the two paths. It is to be noted that signal injected in Frac-N PLL path at frequency F2has the same gain compared to the signal injected at frequency F1in the feedforward path. Variables FCDRand FPLLdenote the bandwidths for CDR and Frac-N PLL respectively. To measure the output because of the injected signal, we utilize the CDR phase detector. Since the CDR phase detector provides digital output, it can be measured in-situ.

FIG. 21is a simplified diagram illustrating a feedforward configuration with injected signals according to embodiments of the present invention. This diagram is merely an example, which should not unduly limit the scope of the claims. One of ordinary skill in the art would recognize many variations, alternatives, and modifications. Digital square wave signals are injected through each path, and they create triangular signals at the CDR phase detector output. By detecting the peak-to-peak value of the triangular waveforms digitally, we can find the gain mismatch and correct for it by digitally adjusting KVFF.FIG. 22provides simulation results of a feedforward implementation according to embodiments of the present invention. As an example, it shows that for an expected ratio10for peak to peak values, KVFFpath gain needs to match with Frac-N PLL path gain. For other scaling factors the ratio of peak-to-peak values deviates from 10.

It is to be appreciated that Frac-N PLLs can be implemented into CDR devices in various ways. For example, the combination and configuration of charge pump, VCO, and/or other components of a Frac-N PLL can be used to allow the Frac-N PLL to satisfy the requirements of CDR devices.