Insulation power factor alarm monitor

The present invention teaches a technique and provides for apparatus eminently useful for the continuous, simultaneous monitoring of the insulation quality of one or more pieces of high voltage ac electrical equipment, while the equipment is in service and connected to its normal high voltage source. The technique or method of the instant invention involves comparing a voltage, developed across a capacitive shunt at the capacitance tap of a transformer or bushing so equipped with a reference voltage taken from a voltage transformer or potential device connected to the same high voltage source. The effecting of the instant technique requires only a minimal amount of relatively inexpensive attendant centralized control equipment and is simply and easily placed into practice by, for example, such a centralized control unit which automatically computes and displays or prints the insulation power factor and/or capacitance, either continuously or at regular intervals for each piece of said ac electrical equipment being monitored by means of the instant invention such that, for example, alarms are sounded and contacts close when the power factor for any such piece being so monitored reaches preset limits. In addition, said central control unit may also make temperature corrections, compute averages, and indicate short-term and long-term trends.

INTRODUCTION 
The present invention relates to a new, novel, and relatively simple and 
inexpensive, as well as highly efficient and reliable system, including 
apparatus means and methods for effecting same for the simultaneous 
monitoring of a plurality of high voltage ac apparatus to determine 
quantitatively the insulating characteristics thereof through the 
mechanism and technique of measuring either the power factor or the 
capacitance of the dielectrics involved. The power factor of electrical 
insulation has long been recognized as a good indication of its quality 
and future serviceability. Almost any form of degradation of the 
insulation will, sooner or later, increase its power factor. Capacitance, 
too, can be an indication of its quality, especially if the insulation is 
in layers with conductive shields therebetween. For instance, one of the 
layers thereof can completely fail and short out, thus increasing the 
capacitance, since the layers are effectively capacitors in series. 
The present invention teaches and describes apparatus and means for 
effecting the utilization thereof which will continuously monitor the 
insulation power factor and capacitance of almost any type of high voltage 
ac electrical equipment, such as bushings of instrument transformers, 
which equipment has a capacitance tap. (As will be appreciated by those 
skilled in this art, a capacitance tap is a connection brought out from 
the shield closest to ground, across the bottom insulation layer, for use 
as a potential tap or for test purposes.) The instant invention and the 
apparatus employed in the conduct thereof effectively compares the 
capacitance tap current with the applied voltage, taken from a voltage 
transformer, and computes power factor and capacitance by adaptations from 
the following formulas: 
##EQU1## 
where v and i are the instantaneous values of reference voltage and 
capacitance tap current, respectively, V and I are the RMS values of the 
same voltage and current, T is one period of line frequency, f is the 
system frequency, and A is the phase angle (expressed in radians) by which 
i leads v. Since A is normally between 1.47 and .pi./2 (i.e., 84 to 90 
degrees, with the power factor 10 percent or less), the approximations are 
valid. 
In the practice of the instant invention the power factor and capacitance 
values of the particular pieces of equipment being monitored thereby are 
automatically computed by the apparatus comprising same and are, either 
continuously (in the analog embodiment) or intermittently in short-time 
sequence (using a digital microcomputer) reported. The analog embodiment 
provides continuous data output and may be utilized to activate alarms for 
certain preset conditions. Operation of the computerized embodiment of the 
present invention may be used to, among other things, print data at 
regular intervals, make temperature corrections, compute averages and 
trends, and determine alarm conditions. 
BACKGROUND OF THE INVENTION 
1.Field of the Invention 
The present invention relates to methods and means for test monitoring 
electrical power equipment--in particular, the insulating values of the 
dielectrics comprising means for confining the course of electric currents 
to prescribed conducting paths in and on high voltage ac electrical power 
equipment. By continuously, accurately, and rapidly monitoring such 
insulation power factor and/or capacitance, the instant invention provides 
for field testing, under a plethora of adverse conditions of external 
electrical stress and the like, for correctly indicating the true 
condition of same to determine whether or not it is safe to leave the 
equipment in service in the immediate future. 
2. Description of the Prior Art 
It has been noted that numerous prior art investigators have discovered, 
taught, and disclosed methods and/or means for determining the immediate 
condition of the insulating materials comprising various parts and 
configurations of ac electrical equipment, including that of the type used 
in applications of relatively high voltage. For example, the teachings of 
U.S. Pat. No. 1,945,263, Doble, Jan. 30, 1934, assigned to Doble 
Engineering Company relates to apparatus useful for measuring power 
factor. Doble's apparatus accurately measures insulation power factor and 
capacitance of electrical equipment, but is taught with the following 
limitations: 
1. The equipment to be tested must be taken out of service and disconnected 
from associated equipment. 
2. The equipment is tested at a voltage that is much lower than its 
operating voltage. This is a consideration of considerable importance when 
it is realized that on some (particularly older) equipment, the insulation 
power factor is significantly higher when in service than can be measured 
by a low voltage test. 
3. The test can be conducted by at one point in time and must therefore be 
repeated periodically to establish trends in insulation quality. In many 
instances this is a formidable drawback since many times, it simply is not 
practical to test the insulation of a given piece of equipment often 
enough to detect deterioration thereof before it leads to rapid failure. 
4. Each test requires hands-on attention by an operator. 
5. Only one piece of equipment can be tested with any one test setup. 
Other prior art investigators have taught and disclosed other methods 
and/or means for determining the immediate condition of such insulating 
materials. U.S. Pat. No. 3,710,242, Povey, Jan. 9, 1973, assigned to Doble 
Engineering Company discloses apparatus which effectively eliminates 
limitations (1) and (2) above. In a manner somewhat similar to the 
apparatus described and used in the present invention, Povey's apparatus 
measures power factor and capacitance by comparing a voltage derived from 
the capacitance tap with a reference potential. Unfortunately his 
advancement of the prior art does not eliminate limitations (3), (4), and 
(5) supra, and (as pointed out in Povey's disclosure) generally requires 
long, shielded leads to a voltage transformer (or other suitable reference 
with a known phase relationship) which may be located a long distance from 
the equipment being tested. For repeated tests, these leads must either be 
left in place indefinitely or rerun for each test. 
The apparatus of the instant invention and the means and methods of 
employment of same effectively eliminates all five of the limitations 
mentioned above by providing automatic, continuous monitoring, with the 
equipment that is to be tested in service and energized at its operating 
voltage, and with minimal human interface. In addition, the practice of 
the present invention provides for apparatus which may be used to compute 
temperature corrections, averages, and short-term and long-term trends. It 
also gives alarms and can be connected to trip equipment out of the 
service under certain preset conditions. In contrast to the inventions of 
the type of the prior art, which test only one piece of equipment at a 
time, the present invention provides for apparatus which may be used to 
monitor any number of pieces of equipment simultaneously. As with Povey's 
equipment, leads must be run to a voltage transformer or other suitable 
reference, but only one (2-conductor, shielded) lead must be run for each 
phase of each voltage on which equipment is to be monitored, regardless of 
the number of pieces of equipment. A separate lead must be run from each 
piece of monitored equipment to the invention apparatus, which is usually 
located in the switchhouse or other central, sheltered location. 
Of course, not all of the investigators of the prior art have directed 
their approaches to solving many of these problems heretofore associated 
with determining the state, or condition of, insulating materials normally 
associated with equipment of the type herein described to the 
determination of power factor and/or capacitance characteristics. One such 
other approach to advancing the state of this art is found in the 
teachings of U.S. Pat. No. 4,293,399, Belanger et al., Oct. 6, 1981, 
assigned to Hydro-Quebec wherein is described apparatus which is employed 
to monitor the hydrogen gas dissolved in insulating oil. This apparatus of 
Belanger may be employed as a continuous monitor, and thus eliminates the 
first four of the five limitations mentioned above for the original Doble 
apparatus. The Belanger et al. invention monitors hydrogen gas only on one 
piece of equipment; no other gasses are monitored and no electrical 
properties are monitored. The device provides alarms, but no hard data 
copies or trend analysis. An output is provided so the user can connect a 
strip-chart recorder if he so desires. Although the teaching of Belanger 
et al. have been reported to advance the state of this art it must be 
realized that, of course, the dissolved hydrogen monitor approach of this 
invention is not applicable to equipment that does not use oil in its 
insulation system. 
Just which type of monitoring system (power factor/capacitance or dissolved 
hydrogen) provides the more desirable indication of insulation quality has 
been the subject of some debate. To this end, TVA has developed data which 
strongly indicates that power factor measurement provides a more reliable 
indication of insulation quality than hydrogen gas content, and usually 
indicates a destructive trend earlier, at least on some equipment which is 
not especially suited for gas monitoring. In any event, it should be noted 
that the apparatus described in the practice of the present invention can 
monitor (and compute averages and trends on) both power factor and 
hydrogen if the hydrogen monitor described above is installed and its 
output is wired into said apparatus. 
SUMMARY OF THE INVENTION 
The instant invention relates to a vastly improved technique, including 
methods and means, for utilizing the capacitance tap of each piece of 
equipment to be monitored by fitting same with a capacitance tap adaptor, 
which capacitance tap adaptor contains a pair of precision, low-loss 
capacitors (the polystyrene type is preferred) to serve as a shunt. The 
values of the two capacitors are selected to produce a voltage drop of 
10-to-140 volts ac to ground. A reference voltage, usually 67 or 120 volts 
ac to ground, is normally taken from a voltage transformer connected to 
the same source as the equipment to be monitored. Each of these voltages 
passes through a differential attenuator/amplifier which scales the 
voltage and reduces common-mode interference. The amplifiers also contain 
protective circuitry and low-pass filters to reduce transients and high 
frequency interference. The amplifier outputs are ac voltages with maximum 
amplitudes of about 5 volts. 
In the first embodiment of the present invention, the amplifier outputs 
feed into an analog circuit that measures the phase angle between them. At 
power factor of 10 percent or less, this phase angle (in radians) is an 
excellent approximation of the power factor, since it is effectively 
subtracted from .pi./2 (90 degrees) by the purely capacitive shunt at the 
capacitance tap. The power factor is read on a digital panel meter; also, 
0-to-10 volt and/or 0-to-1 milliampere dc signals are made available for 
driving a strip-chart recorder. Alarm/trip contacts close when preset 
power factor limits are exceeded. 
In the second embodiment of the instant invention, a microcomputer is used, 
with analog-to-digital converters (ADC), input-output (IO) ports, a 
printer, and other peripherals to interface with the real world. The 
amplifier outputs described above are sampled 32 times each cycle of ac by 
the computer-controlled ADC. A frequency multiplier, which uses a 
phase-locked loop locked to the reference voltage, precisely sets the 
sample rate. The computer, using the discrete numbers derived from the 
samples, computes power factor and capacitance by doing numerical 
integration and other necessary arithmetic. With this method, the power 
factor is accurate over the full theoretical range, rather than only up to 
about 10 percent as in the phase angle procedure of the first embodiment 
supra. These computations of power factor and capacitance are generally 
made once each minute. In addition, temperature corrections, averages, 
trends, and comparisons with limits are also computed. All these data are 
printed at regular intervals and other opportune times. Alarm/trip outputs 
are provided when certain preset limits are exceeded. 
OBJECTS OF THE INVENTION 
It is therefore the principal object of the present invention to develop a 
new method and/or means for easily, quickly, accurately, and continuously 
monitoring the insulation quality of high voltage ac electrical equipment. 
It is another object of the present invention to develop a new method 
and/or means for easily, quickly, accurately, and continuously monitoring 
the insulation quality of high voltage ac electrical equipment wherein 
same is accomplished by providing a continuous readout and/or record of 
power factor and capacitance, along with alarm/trip outputs. 
Still further and more general objects and advantages of the present 
invention will appear from the more detailed description set forth below, 
it being understood, however, that this more detailed description is given 
by way of illustration and explanation only, and not necessarily by way of 
limitation since various changes therein may be made by those skilled in 
the art without departing from the true spirit and scope of the present 
invention.

DETAILED DESCRIPTION OF THE DRAWINGS 
For the sake of clarity and a better understanding of the applicability of 
the illustrations of the various drawings a more detailed description of 
the same is given below. 
Referring now more specifically to FIG. 1, equipment being monitored (1) is 
a high voltage bushing or an instrument transformer. It is represented 
schematically by C.sub.1 and C.sub.2, with capacitance tap (2) bringing 
out the connection between C.sub.1 and C.sub.2. C.sub.1 is actually the 
primary insulation, the quality of which is to be monitored. C.sub.1 is 
usually constructed in layers of oil-impregnated paper (the insulation) 
separated by conductive shields which equalize the electric field. 
Therefore, C.sub.1 is, in effect, several capacitors in series, with the 
impedance of C.sub.1 representing the net impedance of the series string. 
C.sub.2 represents the bottom layer of insulation, which is shorted out 
whenever the capacitance tap is grounded. Those knowledgeable in the field 
of high voltage insulation will be familiar with this construction and 
terminology. 
Ideally, capacitance C.sub.1 is a pure capacitance with no dielectric 
losses. In practice, there will be some loss, which is prepared by 
R.sub.1, shown in parallel with C.sub.1. If the loss is high enough 
(represented by a low value of R.sub.1), significant heating will occur in 
the insulation. This heating is proportional to the watts loss and is 
given by: 
EQU W=V.sup.2 /R.sub.1 
where W is watts loss, V is the system (line-to-ground) voltage, and 
R.sub.1 is the equivalent parallel resistance of the insulation. 
R.sub.1 is an equivalent resistance (rather than an actual physical 
resistor) which is inconvenient to measure. The watts loss can also be 
determined from the capacitance, voltage, and power factor of C.sub.1, as 
follows: 
EQU PF=COS (A) 
EQU C.perspectiveto.I/(2.pi.fV) 
EQU W=VI COS (A) 
EQU W.perspectiveto.2.pi.fCV.sup.2 COS (A) or W.perspectiveto.2.pi.fCV.sup.2 
(PF) 
where PF is the insulation power factor, C is the capacitance (of C.sub.1), 
I is the insulation current, A is the phase angle that V lags I, and f is 
system frequency. Since V and f are known to a reasonable degree of 
accuracy, and PF and C are measured by the central control and monitoring 
unit of the instant invention, hereinafter normally referred to, for the 
sake of simplicity, as the apparatus, the watts loss can be determined. 
(Actually, in most instances, the capacitance of C.sub.1 is also known, 
and need not be measured, except that it also can give an indication of 
insulation quality.) 
The watts loss of the insulation is very significant because the 
temperature rise inside the insulation is directly proportional to this 
loss. High temperature causes further deterioration of the insulation, 
which causes even higher loss, and so on, until the equipment eventually 
fails. 
In practice, the power factor itself, which is a fundamental property of 
the insulation and a widely recognized indicator of insulation quality, is 
more useful than watts loss. It should be recognized, however, that low 
capacitance bushings can tolerate a higher power factor than high 
capacitance bushings. At any rate, the apparatus of the instant invention, 
while accurately monitoring the true power factor, primarily looks for 
increases in power factor beyond what can be expected because of 
temperature fluctuation. 
Capacitance tap (2) is connected to ground through an impedor means, i.e. a 
capacitive shunt, of very low reactance and is illustrated as contained in 
capacitance tap adapter (3). The voltage developed across the capacitive 
shunt is determined almost solely by the current through C.sub.1 and lags 
this current by a phase angle of almost exactly 90 degrees. This voltage 
is passed through 2-conductor, shielded cable (4) to voltage interface 
amplifier (5), which reduces it to a filtered, protected voltage 
compatible with control unit (11) input requirements (approximately 5 
volts). Amplifiers (6) and (7) accept additional inputs, if any. 
Reference source (8) (shown as a magnetic potential transformer connected 
to the same high voltage bus as the equipment being monitored) develops a 
voltage which is a known fraction of the bus voltage and precisely in 
phase with it. This voltage is passed through 2-conductor, shielded cable 
(9) to voltage interface amplifer (10), which reduces it to a voltage 
similar to that developer by voltage interface amplifier (5). 
Potential transformer (8) should be as accurate as possible (0.3 percent 
preferred) and also as lightly loaded as possible. Normal loading is 
acceptable if cable (9) goes all the way to the transformer, thus 
bypassing the cable voltage drop caused by normal metering and/or relaying 
loads. These loads can cause errors as high as 2 percent power factor 
and/or capacitance if their cable is not bypassed. 
A capacitive potential device can be used as a reference source if it is 
carefully tuned and phase tested. A phase error of one degree will cause a 
1.75 percent power factor error. Once tuned, the loading of the potential 
device should not be changed. 
Control unit (11) computes power factor by either of two methods to be 
described later. Capacitance may also be computed. The results are 
displayed continuously or recorded on readout device (12). If certain 
preset conditions are exceeded, or upward unexplainable trends are 
observed, notes or other indications may be printed by (12), and alarm 
circuit (13) may close contacts or provide some other type of alarm 
output. This alarm output may be used only for alarm purposes, or may 
initiate tripping to remove the monitored equipment from service before it 
fails. 
Referring now more specifically to FIG. 2 therein is shown an example of a 
capacitance tap adaptor designed for TVA current transformers. A 
capacitance tap adaptor for other equipment would, in general, take a 
different form; however, certain parameters are necessary, as described 
below. 
The main components of the capacitance tap adaptor are shunt capacitors 
(201) and (202). The value of the capacitors is selected so that, added 
together, they provide a shunt to ground that, with normal insulation 
current flowing through it, develops a voltage of between about 10 and 
about 140 volts RMS. Since the normal voltage of an open circuit 
capacitance tap is usually about 10,000 volts, the voltage developed is 
determined almost solely by the insulation current, with other factors 
having an effect of about 1 percent or less. The type capacitor is also 
important; the dissipation factor (or power factor) of the two capacitors 
should be very low or at least a precisely known, stable value. The 
specified capacitors have a dissipation factor of 0.02 to 0.03 percent at 
60 Hz. For ultimate accuracy, this factor is added to the insulation power 
factor measured by the apparatus. 
Clip (203) is used to connect to the capacitance tap. In this case, the 
connection is to a center pin in the capacitance tap, about 3/8-inch (0.95 
cm) in diameter. The entire adaptor is housed in a 2-inch (5.08 cm) 
diameter pipe nipple 6 inches (15 cm) long (205), which screws into the 
capacitance tap housing using the 21/4 inch, 12 per inch (5.715 cm, 4.724 
per cm) threads (206) machined onto the pipe. The keep flexible lead (204) 
from twisting too much, the adaptor is twisted ccw several turns after 
connecting to the center pin and before starting to screw the pipe nipple 
into the capacitance tap. Thusly, the lead can untwist as the nipple is 
started cw into the capacitance tap. 
It is noted that the principal reason for using two capacitors, instead of 
one of twice the value of capacitance, is to limit the voltage in case of 
capacitor failure. 
For instance, if only one capacitor is used, and if fails open, or a lead 
breaks, the output voltage from the capacitance tap adaptor could increase 
to about 10,000 volts, damaging the apparatus and constituting a safety 
hazard. However, when two capacitors are utilized as shown, the output 
voltage is limited to 20 to 280 volts, unless both should fail 
simultaneously, an eventually considered highly unlikely if the capacitor 
voltage rating is more than twice the expected normal output voltage. An 
open capacitor will normally be detected by the fact that the insulation 
capacitance indicated by the apparatus would be twice the correct value. 
An important consideration which makes the above argument valid is that 
each capacitor's leads are completely independent. Leads (201.1) and 
(202.1) tie to main lead (204.1) at separate connection points. Leads 
(201.2) and (202.2) tie to separate, independent ground lugs (207) and 
(208). Thus, there is no one component or wire that can break that will 
not leave at least one of the capacitors still connected. 
Finally, UHF connector (209) connects to a 2-conductor, shielded cable 
which carriers the voltage to the apparatus. Both the low side of the 
circuit and the shield connect to the shell of the UHF connector (ground). 
If the capacitance tap adaptor is to be left in place for a while and not 
connected to the apparatus, a shorted UHF plug can be connected, shorting 
the capacitance tap to ground as in normal operations, and also protecting 
the UHF connector. 
It is suggested that perhaps a still more desirable embodiment of the 
capacitance tap adaptor would be a design which permitted it to be screwed 
into the capacitance tap without having to first connect to the center 
pin. All connections would be made automatically, with a spring clip 
making the center pin connection as on the capacitance tap cap supplied by 
the equipment manufacturer. It is envisioned that just such a design may 
be utilized in our future planned development of the instant invention. 
Referring now more specifically to FIG. 3 it will be seen that shown 
therein is an example of a voltage interface amplifier utilized in the 
present invention. It should be noted that this particular circuit is used 
in the second embodiment of our invention, but that the first embodiment 
thereof employs a very similar circuit (see for example FIG. 4 infra). 
This circuit was designed for compatibility with the capacitance tap 
adaptor (FIG. 2 supra) used with TVA current transformers, and also with 
the computer used in the second embodiment (see FIG. 7 infra). As noted 
above the circuit parameters would be different in other applications. 
Referring now more specifically to FIG. 3, it will be seen that the high 
side, low side, and shield wires of the cable from the capacitance tap 
adaptor connect to terminals (301, 302, and 303), respectively. Capacitors 
(304) are paralleled to make a single 2 microfarad, 400 volt capacitor, 
but a single 2 microfarad, 200 volt capacitor can be used if available. 
This capacitor completes the reactive differential balance of the 
amplifier, since the high side comes from a 2 microfarad capacitive shunt 
in the capacitance tap adaptor. The low side, on the other hand, is 
grounded directly at the capacitance tap adaptor. We have found that 
capacitors (304) can be omitted with very little degradation of 
performance. 
Resistors (305 and 306), with varistors (309 and 310), form a protective 
transient suppressor circuit. Since the varistors also have inherent 
capacitance, this suppressor circuit combines with resistros (307 and 308) 
and capacitors (311 and 312) to form a two-pole, low-pass filter, with a 
gradual rolloff which is 3 dB down at about 8 kHZ. This filter suppresses 
high frequency interference while causing a phase shift at 60 Hz of only 
about 0.5 degrees. 
Resistors (313, 314, 315, and 316), along with U1B (317), make up a 
differential amplifier with an attenuation factor of 20:1 (26 dB of 
voltage loss). The large voltage reduction allows the circuit to work 
properly with an input voltage of up to 150 volts RMS. Trimpot (318) sets 
the output voltage to the value required by the control unit, which is 
nominally about 2.8 volts RMS. U1A (319) serves as a buffer and impedance 
amplifier. Resistor (320) ensures stability of U1A. 
The differential amplifier, while not an absolute necessity, is very 
effective in reducing interference from nearby high current conductors. 
Such interference can create considerable error on a long cable run, and 
cannot always be reduced sufficiently by shielding alone. The combination 
of the differential amplifier and the shield, which is grounded solidly at 
both ends, eliminates virtually all 60 Hz interference. 
The values of resistors (313, 314, 315, and 316) can be changed to 
accommodate different input and output requirements. The attenuation of 
the circuit is determined by the ratio of (313) divided by (315) or the 
ratio of (314) divided by (316). These ratios must be equal for 
differential balance. 
This same circuit is used to interface with the reference voltage. The 
reference voltage is usually taken from either a magnetic voltage 
transformer or a tuned capacitive potential device. In either case, the 
voltage input to the circuit is usually either 67 or 120 volts RMS, with 
which the circuit is fully compatible. Capacitors (304) are bypassed when 
used on the reference voltage. 
Any difference in phase shift between the capacitance tap amplifier and the 
reference voltage amplifier must be calibrated out of the system. This is 
accomplished by applying the same voltage to both circuits, with the 
voltage to the capacitance tap circuit being applied through a capacitance 
equal to the total of the capacitance tap, the capacitance tap adaptor, 
and the cable. The apparatus is set to read zero (or the capacitance tap 
adaptor dissipation) with this voltage applied as described. 
The next three illustrations, to wit FIGS. 4, 5, and 6 will be referred to 
as a group in that they related to the first embodiment of the apparatus, 
which contains both analog and digital circuits, but no computer. 
Referring now more specifically to FIG. 4, it will be noticed that the 
input circuitry looks very much like that of the voltage interface 
amplifier just described in our treatment of FIG. 3, supra. Varistors 
(401, 402, 403, and 404) provide additional transient protection, and are 
particularly useful to protect calibration switch (405). Unlike the 
computerized second embodiment of the instant invention, this first 
embodiment thereof should be calibrated and zero checked at regular 
intervals. Switches (405 and 406) allow this calibration operation to be 
performed at any time. Switch (405) connects both channels to the 
reference voltage, with capacitor (407) simulating the capacitance tap 
adaptor. With switch (406) in the NORM position, the output is adjusted to 
zero with ZERO potentiometer (408). Calibration can then be checked at 
plus 2 percent power factor and minus 2 percent power factor by setting 
switch (406) appropriately. 
The operation of the transient suppressor circuits, filters, and 
differential attenuator/amplifiers are the same as described for FIG. 3 
supra. There is, however, no output voltage control because this 
embodiment does not monitor capacitance, but only phase angle, making the 
amplitude of the waves unimportant. 
Resistors (409 and 410) and diodes (411, 412, 413, and 414) clip the 
waveforms symmetrically and protect voltage comparators U2A (415) and U4A 
(416). The waveforms are converted to true square waves, with transitions 
at zero and 180 degrees of the original waves, by U2A (415) and U4A (416). 
Resistors (417 and 418) serve as "pull-up" resistors for U2A and U4A, 
which have open collector transistor outputs. 
Referring now more specifically to FIG. 5, it will be appreciated that both 
EXCLUSIVE OR gate U5A (501) and EXCLUSIVE OR gate U5B (502) serve as 
noninverting buffers to make the waveforms fully compatible with the 
remaining CMOS circuitry. EXCLUSIVE OR gate U5C (503) has a high 
(positive) output whenever the C and R waveforms are opposite (one high 
and the other low), and a low (zero) output when they are the same (both 
high or both low). Thus, if the waves are exactly in phase and have the 
same symmetry, EXCLUSIVE OR gate U5C always has a low output except for 
perhaps some extremely fast "glitches" at the transition points. If the 
waves are slightly out of phase, the output of U5C will be a string of 
short pulses, with the length and position of the pulses being the same as 
the time between the zero crossings of the two waves. No distinction has 
yet been made concerning which wave leads. Positive pulses are generated 
at both positive and negative zero crossings, so the repetition rate is 
twice the line frequency. 
Monostable multivibrators U6A, U6B, U7A, and U7B (504-507, respectively) 
help determine which wave is leading, so that the appropriate pulses can 
be passed through to charge storage capacitor (520) to the correct 
polarity. The inappropriate pulses are blocked. 
All the monostable multivibrators have timing networks (200,000 ohms and 
0.022 microfarad) which provide output pulses of about 2.4 milliseconds, 
much longer than the 0.53 millisecond pulse length necessary to give a 
full scale (20 percent) power factor reading. Multivibrator U6A (504) is 
triggered by the positive going zero crossing of the R (reference) wave 
and multivibrator U6B (505) by the negative going zero crossing of the 
same R wave. NAND gate U8A (508) takes its input from the NOT Q outputs of 
multivibrators U6A and U6B, so the output of NAND gate U8A is low most of 
the time; it goes high, however, for 2.4 milliseconds starting at each 
zero crossing (positive or negative) of the R wave. If the R wave is 
leading the C wave, as it should for a positive power factor, then the 
pulse generated by EXCLUSIVE OR gate U5C (503) at each zero crossing 
occurs during the 2.4 millisecond pulse generated by U8A. The two pulses 
drive the output of NAND gate U8D (510) for a time equal to the shorter of 
the two pulses. If the C wave is leading the R wave, then the pulse 
generated by U5C occurs before the 2.4 millisecond pulse, and U8D is not 
driven low. The net result is a negative going pulse at the U8D output 
when, and only when, any zero crossing of the R wave occurs before that of 
the C wave. An additional requirement is that the waves be within 90 
degrees of each other. 
The pulses from U8D go from plus 15 volts to zero volts. These are coupled 
through capacitor (521) to U9 (512), which is a two-stage, noninverting 
buffer operating between minus 15 volts and zero volts. Thus, pulses 
delivered to TP2 and diode (514) are negative going from zero to minus 15 
volts. U10A (518) is an inverting, zero impedance (current summing) 
amplifier, so the negative pulses charge storage capacitor (520) such as 
to fdrive the output of U10A (point M) positive. 
U5C (503), U7A (506), U7B (507), U8B (509), and U8C (511) operate in an 
identical manner to that described above to produce negative going pulses 
at the output of NAND gate U8C when, and only when, any zero crossing of 
the C wave occurs before that of the R wave. These pulses are inverted by 
EXCLUSIVE OR gate U5D (513), and pass through diode (515) to charge the 
storage capacitor (520) such as to produce a negative output at point M. 
Trimpots (516 and 517) adjust the pulse current for calibration of the 
circuit. Resistor (519) is selected so that when the trimpots are properly 
set, a 0.2 radian phase difference produces a voltage of 10 volts at point 
M. 
This circuit reads both positive and negative power factor. Though negative 
power factor cannot actually occur in insulation, the circuit must still 
respond to it, because alternate or random pulses of opposite polarity can 
occur due to waveform distortion, dissymmetry, or noise. Pulses of both 
polarities must be properly averaged together for accuracy near zero power 
factor. 
Referring now more specifically to FIG. 6, it will be seen that the voltage 
developed at point M (as described with FIG. 5 supra, 10 volts dc positive 
for full scale power factor of 20 percent) is passed through resistor 
(605) to the 10 volt output (622). It is also applied to a current pump 
composed of operational amplifier U10B (614) and the associated 10,000 ohm 
resistance bridge (624, 625, 626, and 627). The current pump supplies up 
to 1 ma to the current output (623), which is independent of the external 
resistance up to the voltage compliance limit. The output (615) of 
(current summing) amplifier U10B must always be twice the voltage at the 
current output terminal (623). Since the voltage at (615) is limited to 
about 12 volts, the voltage at (623) is limited to 6 volts. Thus, the 
maximum load resistance is 6,000 ohms if full scale current of 1 milliamp 
is to be delivered. 
One side is grounded on both the voltage and current outputs, so floating 
or differential loads are not required. 
The voltage at point M is also passed to a voltage divider composed of 
resistors (601, 602, and 603). Twenty percent of the voltage is passed to 
digital panel meter (604), which is set for 2 volts full scale. The 
digital panel meter has 31/2 digits, and the decimal point is fixed so 
that a full scale indication of 19.99 is provided. 
The same voltage divider passes 25 percent of the voltage at point M to 
comparator U4B (606), which drives alarm relay (612). Diode (613) protects 
the output transistor of U4B from transients caused by the inductive relay 
coil. The level that trips relay (612) is set by potentiometer (607). It 
and resistor (611) comprise a voltage divider between plus 15 volts dc and 
ground. Hysteresis and positive feedback are provided by resistors 608 and 
609, assuring that the relay picks up completely with a minimum of 
chatter. The amount of hysteresis is about 1.5 percent of full scale or 
about 0.3 percent power factor. If the power factor decreases to 0.3 
percent below the alarm set point, the alarm resets. Capacitor (610) 
guards against the alarm being set off by noise or a transient. 
For better alarm setting accuracy, a switch with precision resistors can be 
substituted for potentiometer (607). 
All circuits are powered by dual 15 volt dc power supply (618). Fuses (616 
and 617) protect against short circuits. Capacitors (619 and 620) are 
located on the circuit board and assure a low source impedance at high 
frequencies. Voltage regulator (621) supplies the digital panel meter. All 
operational amplifiers and voltage comparators require plus 15 volts and 
minus 15 volts. All digital integrated circuits require plus 15 volts and 
common (ground) except U9 (512, FIG. 5 supra) which requires minus 15 
volts and common. 
The following illustrations, to wit, FIGS. 7-9 and 11 are directed to the 
second embodiment of the present invention and the one that we feel is the 
best version of the apparatus because it is more easily expandable and 
provides more information than the first embodiment. FIG. 10 is directed 
to an interface circuit used in the instance of a hydrogen probe. 
The capacitance tap adaptors are identical in the two embodiments, and the 
voltage interface attenuator/amplifiers are similar. Other components are 
different, however, with the main difference being the control unit. A 
Motorola VME microcomputer system, based on the MC 68000 microprocessor, 
is at the heart of the control unit in the second embodiment. Other 
computers may be used in the future as the state of the art changes. 
Referring now more specifically to FIG. 7 therein is shown a diagram of the 
VME computer system with the peripherals used in said second embodiment. 
The computer equipment includes a monoboard microcomputer module (711), 
ADC modules (709) and (710), parallel interface module (708), battery 
backed random access memory and realtime clock (712), and power supply 
module (713). Multiconductor bus (701) is provided to connect modules 
(708), (709), and (710) to microcomputer module (711) and to power supply 
module (713). Multiconductor bus (702) is provided to connect 
microcomputer module (711) to battery backed memory and realtime clock 
module (712). Bus (701) and bus (702) have provisions for connections to 
additional modules. 
Connector (714) and multiconductor cable (719) are provided to connect 
parallel interface module (708) to printer 12. Connector (715) and 
multiconductor cable (720) are provided to connect parallel interface 
module (708) to alarm circuits 13 and to synchronizer circuits (725). 
Connector (715) and cable (721) are used to connect parallel interface 
module (708) to ADC modules (709) and (710). The connection through cable 
(721) is used to provide simultaneous sampling and analog-to-digital 
conversion of an analog input to module (709) and an analog input to 
module (710). Connector (716) and multiconductor cable (722) are provided 
to couple analog voltages to ADC module (709). Connector (717) and 
multiconductor cable (723) are provided to couple analog voltages to ADC 
module (710). Connector (718) and multiconductor cable (724) are provided 
to connect monoboard microcomputer module (711) to a data terminal. The 
data terminal may be used to allow manual control of certain functions and 
to allow modification of certain variables. 
Referring now more specifically to FIG. 8 therein is shown an example of 
the ambient temperature probe used with the second embodiment of the 
apparatus of the present invention. A similar circuit can be used with our 
first embodiment if an additional readout device is provided. The circuit 
shown in FIG. 8 feeds into the control unit computer through an ADC. 
As shown in FIG. 8, a positive 15 volts is fed through resistors (817 and 
815), which drop the voltage to about 10 volts as determined by zener 
diode (801). Resistors (817 and 815), with varistor (816) and capacitor 
(814), provide transient protection and filtering. 
The 10 volts developed across zener diode (801) is applied through resistor 
(802) to zener diode (803), which develops a voltage of about 5 volts to 
ground. This 5 volts is applied to a two (active) terminal temperature 
sensor (804), the current through which is proportional to absolute 
temperature (1 microamp per degree Kelvin). This current passes through 
resistor (805), which develops a voltage of 1 millivolt per degree Kelvin 
(about 0.3 volt at room temperature). This voltage is amplified by 
operational amplifier (806), the gain of which is set at 15 by resistors 
(807 and 808). The amplified voltage passes through a transient protection 
circuit and low-pass filter composed of resistors (809, 811, and 813), 
along with capacitor (810) and varistor (812), to the output (818). 
The active range of this temperature probe is 0.degree. F. to 140.degree. 
F., corresponding to a voltage output of 3.8 to 5.0 volts. The large 
common-mode voltage is subtracted digitally, rather than using a 
drift-prone analog circuit. The temperature sensor (804) is mounted 
snugly, with epoxy, into a hole in the probe case, so that the probe 
becomes a heat sink which assumes ambient temperature. This construction 
provides "peak filtering," with a time constant of about 15 minutes. The 
probe is filled with a potting compound for waterproofing. The entire 
probe is calibrated over the active temperature range, and appropriate 
constants are stored in the computer. Sufficient accuracy could probably 
be attained by calibrating only at room temperature, however. 
An alternative temperature measuring scheme which has also been used in our 
investigations involves utilizing the digital binary coded decimal (BCD) 
output of a commerical digital thermometer. The BCD output is interfaced 
through an input/output (IO) port on the computer. 
Referring now more specifically to FIG. 9 therein is shown an example of a 
transducer interface amplifier used for monitoring temperature, pressure, 
or any other quantity from a transducer with 0-to-5 volt or 4-to-20 ma dc 
output. The amplifier has unity gain, with no provision for zero or offset 
null, which, if needed, is done in other places. 
The value of resistor (901) depends on the transducer output. For a 4-to-20 
milliamp transducer, the value is 250 ohms, so that 1-to-5 volts is passed 
to operational amplifier (906). On a 0-to-5 volt transducer, the value of 
resistor (901) is 1 megohm, which serves only to hold the output (908) 
near zero volts when there is no input connection. 
Resistors (902 and 904), varistor (903), and capacitor (905) provide 
transient protection. Resistor (904) and capacitor (905) also form a 
one-pole, low-pass filter with a 3 dB cutoff frequency of 0.16 Hz. 
Operational amplifier (906) has a field effect transistor input, so its 
input impedance is much higher than the 1 megohm filter impedance. 
Operational amplifier (906) is connected for unity gain, so the 1-to-5 or 
0-to-5 volts across resistor (901), after being filtered, appears at the 
output (908). Resistor (907) assures stability of the operational 
amplifier (906). 
The transducer power supply may be located in the control unit or at the 
transducer. It should be run on separate wires if a 0-to-5 volt transducer 
is used. 
Referring now more specifically to FIG. 10 therein is shown an example of 
an interface circuit used for logging dissolved hydrogen in the oil of an 
oil and paper bushing or instrument transformer. It is designed to convert 
the output of a Syprotec 201R hydrogen monitor (0-to-1 ma) to a 
computer-compatible 0-to-5 volts dc. The 1 ma output of the hydrogen 
monitor is off ground, intended for use with an isolated, ungrounded 
strip-chart recorder. Therefore, a dc differential amplifier was needed. 
As shown in FIG. 10, resistor (1001) is a precision resistor which develops 
a voltage of 0-to-1 volt for a 0-to-1 milliamp signal. This 0-to-1 volt 
differential signal is passed, along with the common-mode voltage, through 
unity-gain operational amplifiers U1B (1007) and U1A (1008) to the 
differential amplifier circuit, operational amplifier (1014) and 
associated circuitry. As described for FIG. 9, resistors (1002 and 1005), 
varistors (1003), and capacitors (1006) provide transient protection and 
filtering. Resistors (1004) hold the output (1018) to near zero volts when 
there is no input connection. Unity-gain operational amplifiers U1B (1007) 
and U1A (1008) provide current gain, transforming the 1 megohm filter 
circuits to less than 100 ohms as required by the differential amplifier. 
The differential amplifier is composed of operational amplifier U2B (1014) 
and resistor (1009, 1010, 1011, and 1012). It has a voltage gain of 6 to 
differential signals and less than 0.1 to common-mode voltages. Thus the 
desired hydrogen information is amplified and the common-mode voltage is 
stripped off. Potentiometer (1013) can be used to zero the circuit and 
offset the hydrogen monitor's zero error, if any. 
The 0-to-6 volts from the differential amplifier is reduced to 0-to-5 volts 
by potentiometer (1015), so as to be compatible with the computer. 
Unity-gain operational amplifier U2A (1016) reduces output impedance, and 
is assured of stable operation by resistor (1017). 
Referring now more specifically to FIG. 11 therein is shown the 
synchronized frequency multiplier circuit used in the second embodiment of 
the instant invention to provide timing pulses for the ADC. It uses a CMOS 
phase-locked loop, CD4046B, which has a phase comparator (phase comparator 
II) that allows phase lock between two waves without allowing any phase 
difference between them. 
As shown in FIG. 11, the input (1100) is usually taken from the voltage 
reference source, and is usually either 67 or 120 volts ac. Resistor 
(1101) and two back-to-back zener diodes (1102) limit the voltage and 
protect voltage comparator (1106), which with resistors (1103 and 1105), 
further squares up the waveform. Transistor (1109), with resistors (1107 
and 1108), converts the wave to a 0-to-5 volt square wave compatible with 
that needed by the phase-locked loop (1110). Transistor (1109) operates 
either fully on (saturated) or fully cut off. 
For the phase-locked loop (1110) to operate as a frequency multiplier, its 
voltage controlled oscillator (VCO) must operate at the desired high 
frequency. Since the input frequency of 60 Hz is to be multiplied by 32, 
the VCO operates at 1920 Hz. Resistors (1112 and 1113) and capacitor 
(1114) set a frequency range to include 1920 Hz. Since the signal input to 
phase comparator (1123) is 60 Hz, the VCO output (1119) of 1920 Hz is 
divided by exactly 32 by counter (1117). The counter output (1126) of 60 
Hz goes to the other phase comparator input (1120). The phase comparator 
output (1129), which is the frequency/phase error signal, is filtered by 
resistors (1111 and 1116) and capacitor (1115), which are selected to 
provide good VCO stability consistent with good locking characteristics. 
The VCO input (1124) fine tunes the VCO frequency to attain and hold 
frequency and phase lock with the input signal (1100, 1123). 
Buffer (1118) passes the 1920 Hz square wave to output (1121) and the 60 Hz 
square wave to output (1122). The waves are precisely synchronous with 
each other and tightly locked to and in phase with the input wave. By 
using these waves to control the ADC, exactly 32 voltage samples can be 
taken of each cycle of each input wave, or one sample every 11.25 degrees. 
Referring now more specifically to FIG. 12 therein is illustrated by means 
of a computer program logic flow chart the method and/or technique 
utilized in the second embodiment of the instant invention. In FIG. 12, 
the computer function steps are indicated within rectangles with the 
stop/start steps indicated within ovals. The logic steps or questions are 
shown within diamondshaped parallelograms. After power on or reset to 
initialize, the program (1202) goes to the startup routine beginning with 
the initialization of peripheral ports for printers, terminals, alarms, 
and analog sampling control interrupts (1211), and then sequences to clear 
to zero all accumulator memory used for summations and clear status and 
control flags (1212). The next function step comprises computing 
correction factors for percent power factor and capacitance computations, 
as well as determining correction factors which are functions of the 
electrical parameters of the interface network that are determined by 
calibration and stored in nonvolatile memory and computing scale factors 
and coefficients for conversion of other analog inputs--temperature, 
pressure, etc.--to desired units (1213). 
The program then sequences to the wait-for-event routine beginning with the 
check time instruction (1221) wherein the computer asks if it is time for 
measurement. If yes, it follows the action instruction set, which directs 
that, subsequent to the determination of measurements and computations, it 
check results of computations and determines action to be taken; i.e., 
alarm, print record, daily averages, monthly averages, etc. and goes to 
the measure subroutine beginning at (1230). If no, it skips the action 
instruction set at (1221) and proceeds to keyboard entry instructions 
(1223) whereupon it checks for operator entries such as system parameters 
entered manually by the computer control operator through a control 
console or terminal and if yes, performs the required task such as 
updating variables, (1225), which are returned to computing correction 
factors function step (1213) or it performs the required task such as 
listing variables, listing command tables, printing record(s), etc. 
(1226). If there are no keyboard instructions, it returns to check time 
instruction set (1221). 
When check time instruction (1221) is yes, the program goes to the measure 
subroutine beginning at (1230) where it sets up for data accumulation, 
clears memory, initializes counters and pointers, enables interrupt 
processing, and sets up arrays for data accumulation. In the next shown 
function step (1231) the program begins analog-to-digital conversion and 
accumulation of data, performs intermediate calculations and continues 
same until all samples for the measurement interval are obtained. The 
program next directs at (1232) the computer to determine the final 
results, i.e., percent power factor, capacitance, temperatures, and other 
desired results including computed internal temperature and temperature 
corrected percent power factor. Subsequently, the program checks for alarm 
conditions (1240) and if yes, enables alarm circuits (1241), if no, it 
checks for results outside specified bounds (1242); and, if yes, outputs 
data to printer (1243), if no, it checks for time to print (1244). If time 
to print is yes, it outputs data to printer (1243); if no, then it checks 
time for end of month averaging (1245). If time for end of month averaging 
is yes, it retrieves summed good data from accumulators (1232), does 
calculating (1246), and outputs data to printer (1247). If not end of 
month, then it checks for midnight (i.e., time for end of day) averaging 
(1248). If yes, it retrieves summed good data from accumulators (1232), 
does calculating (1249) and output data to printer (1250). If no, it 
returns to action instruction set at (1221) of wait-for-event routine. 
Although the measure subroutine has been described in its relationship to 
the operation of this program in this discussion of FIG. 12, it will, of 
course, be appreciated that there are other major subroutines herein 
involved. For a more detailed description of same see the "General 
Microcomputer Software Description" set forth infra, after a description 
of "Embodiment Two" of "The Preferred Embodiments" section. 
Briefly, some of these other major (and minor) subroutines comprise: 
1. Percent Power Factor and Capacitance Computation 
a. Data Accumulation 
b. Accumulated Data Format 
c. RMS Computation--Current and Voltage 
d. Watts Computation 
e. Final Computations 
2. Computation of Temperatures, Pressures, Hydrogen, etc. 
3. Alarm Analysis 
4. Trend Analysis 
5. Daily and Monthly Averages 
6. Compute Estimate of the Internal Temperature of Monitored Equipment 
7. Other (minor) Subroutines Used 
a. Square Root 
b. Binary to Decimal Conversion 
c. Decimal to ASCII Conversion 
d. ASCII to Decimal Conversion 
e. Decimal to Binary Conversion 
f. Double Precision Multiply 
g. Double Precision Addition 
h. Extended Divide 
DESCRIPTION OF THE PREFERRED EMBODIMENTS 
As indicated supra, the instant invention is described both in toto, and in 
two separate embodiments thereof, which two embodiments are as follows. 
EMBODIMENT ONE 
Normally the equipment being monitored by the apparatus in this first 
embodiment of the instant invention is either a high voltage bushing or an 
instrument transformer with a capacitance tap bringing out the connection 
between the primary insulation and the bottom layer of insulation, which 
is shorted out whenever the capacitance tap is grounded. Ideally, the 
capacitance of the primary insulation is "pure" with no dielectric losses. 
In practice, there will be some loss, which can be represented by a 
resistance in parallel with the primary insulation. If the loss is high 
enough (represented by a relatively low resistance value), significant 
heating will occur in the insulation. This heating is proportional to the 
watts loss. Note that this resistance is an equivalent resistance (rather 
than an actual physical resistor) which is inconvenient to measure. The 
watts loss can also be determined from the capacitance, voltage, and power 
factor of the primary insulation, as follows 
EQU PF=COS (A) 
EQU C.perspectiveto.I/(2.pi.fv) 
EQU W=VI COS (A) 
EQU W.perspectiveto.2.pi.fCV.sup.2 COS (A) or W.perspectiveto.2.pi.fCV.sup.2 
(PF) 
where PF is the insulation power factor, C is the capacitance of the 
primary insulation, I is the insulation current, A is the phase angle that 
V lags I, and f is system frequency. Since V and f are known to a 
reasonable degree of accuracy, and PF and C are measured by the apparatus 
being patented, the watts loss can be determined. (Actually, in most 
instances, the capacitance of the primary insulation is also known, and 
need not be measured, except that it also can give an indication of 
insulation quality.) 
The watts loss of the insulation is very significant because the 
temperature rise inside the insulation is directly proportional to this 
loss. High temperature causes further deterioration of the insulation, 
which causes even higher loss, and so on, until the equipment eventually 
fails. 
In practice, the power factor itself is more useful than watts loss. At any 
rate, the apparatus of the instant invention, while accurately monitoring 
the true power factor, primarily looks for increases in power factor 
beyond what can be expected because of temperature fluctuation. 
The capacitance tap is connected to ground through a capacitive shunt that 
is virtually lossless and is contained in a specially designed and 
constructed capacitance tap adapter. The voltage developed across the 
capacitive shunt is determined almost solely by the current through the 
primary insulation and lags this current by a phase angle of almost 
exactly 90 degrees. This voltage is passed through a 2-conductor, shielded 
cable to a voltage interface amplifier which reduces it to a filtered, 
protected voltage compatible with the control unit input requirements 
(approximately 5 volts). 
The reference source, which can be a magnetic voltage transformer connected 
to the same high voltage bus as the equipment being monitored, develops a 
voltage which is a known fraction of the bus voltage and precisely in 
phase with it. This voltage is passed through a similar 2-conductor, 
shielded cable to a similar voltage interface amplifier wherein it is 
reduced to a voltage similar to that developed from the capacitance tap. 
The voltage transformer should be as accurate as possible (0.3 percent 
preferred) and also as lightly loaded as possible; however, normal loading 
is acceptable if said 2-conductor, shielded cable goes all the way to the 
transformer, thus bypassing the cable voltage drop caused by normal 
metering and/or relaying loads. Otherwise, we have found that these loads 
can cause errors as high as 2 percent power factor and/or capacitance if 
their cable is not bypassed. Alternatively a capacitive potential device 
can be used as a reference source if it is carefully tuned and phase 
tested. A phase error of one degree will cause a 1.75 percent power factor 
error. Once tuned, the loading of the potential device should not be 
changed. 
In this first embodiment of the instant invention the control unit of same 
computes power factor by measuring phase angle. The details of such 
computations will be described infra. The results are continuously 
displayed and the alarm circuit may provide some type of alarm output. 
A principal feature of our specially designed capacitance tap adaptor is 
the provision for the use of two shunt capacitors. The value of the 
capacitors is selected so that, added together, they provide a shunt to 
ground that, with normal insulation current flowing through it, develops a 
voltage of between 10 and 140 volts RMS. Since the normal voltage of an 
open circuit capacitance tap is usually about 10,000 volts, the voltage 
developed is determined almost solely by the insulation current, with 
other factors having an effect of about 1 percent or less. The type 
capacitor is also important; the dissipation factor (or power factor) of 
the two capacitors should be very low or at least a precisely known, 
stable value. The specified capacitors have a dissipation factor of 0.02 
to 0.03 percent at 60 Hz. For ultimate accuracy, this factor is added to 
the insulation power factor measured by the apparatus. The reason for 
using two capacitors, instead of one of twice the value of capacitance, is 
to limit the voltage in case of capacitor failure. For instance if only 
one capacitor is used, and it fails open, or a lead breaks, the output 
voltage from the capacitance tap adaptor could increase to about 10,000 
volts, damaging the apparatus and constituting a safety hazard. With two 
capacitors, the output voltage is limited to 20 to 280 volts, unless both 
fail simultaneously, considered highly unlikely if the capacitor voltage 
rating is more than twice the expected normal output voltage. An open 
capacitor will normally be detected by the fact that the insulation 
capacitance indicated by the apparatus would be twice the correct value. 
An important consideration which makes this argument valid is that each 
capacitor's leads are completely independent. 
A voltage interface amplifier is used and designed for compatibility with 
said capacitance tap adaptor with the high side, low side, and shield 
wires of the cable from the capacitance tap adaptor connected thereto. A 
capacitor completes the reactive differential balance of this interface 
amplifier, since the high side comes from a 2 microfared capacitive shunt 
in the capacitance tap adaptor. The low side, on the other hand, is 
grounded directly at the capacitance tap adaptor. A pair of resistors as 
well as a pair of varistors form a protective transient suppressor 
circuit. Since the varistors also have inherent capacitance, this 
resulting suppressor circuit combines with another pair of resistors as 
well as a pair of capacitors to form a two-pole, low-pass filter, with a 
gradual rolloff which is 3 dB down at about 8 kHZ. This filter suppresses 
high frequency interference while causing a phase shift at 60 Hz of only 
about 0.5 degrees. 
Also provided is a differential amplifier with an attenuation factor of 
20:1 (26 dB of voltage loss). The large voltage reduction allows this 
circuit to work properly with an input voltage of up to 150 volts RMS. A 
trimpot and associated capacitor may also be provided to set the phase 
shift of the capacitance tap channel to equal that of the reference 
channel. A unity-gain operational amplifier serves as a buffer and 
impedance amplifier. Said differential amplifier, while not an absolute 
necessity, is very effective in reducing interference from nearby high 
current conductors. Such interference can create considerable error on a 
long cable run, and cannot always be reduced sufficiently by shielding 
alone. The combination of the differential amplifier and the shield, which 
is grounded solidly at both ends, eliminates virtually all 60 Hz 
interference. This same circuit is used to interface with the reference 
voltage. The reference is usually taken from either a magnetic potential 
transformer or a tuned capacitive potential device. In either case, the 
voltage input to the circuit is usually either 67 or 120 volts RMS, with 
which the circuit is fully compatible. 
As has been note supra, any difference in phase shift between the 
capacitance tap amplifier and the reference voltage amplifier must be 
calibrated out of the system. This is done by applying the same voltage to 
both circuits, with the voltage to the capacitance tap circuit being 
applied through a capacitance equal to the total of the capacitance tap, 
capacitance tap adaptor, and cable. The apparatus is set to read zero (or 
the capacitance tap adaptor dissipation) with this voltage applied as 
described. 
Resistors and diodes are employed to clip the waveforms symmetrically and 
to protect two voltage comparators. The waveforms are thereby converted to 
true square waves, with transitions at zero and 180 degrees of the 
original waves. A pair of resistors serve to "pull up" the outputs of the 
voltage comparators. In combination therewith, a pair of EXCLUSIVE OR 
gates serve as noninverting buffers to make these converted true square 
waves fully compatible with the remaining CMOS circuitry. A third 
EXCLUSIVE OR gate in combination with the pair supra has a high (positive) 
output whenever the waveforms are opposite (one high and the other low), 
and a low (zero) output when they are the same (both high or both low). 
Thus, if the waves are exactly in phase and have the same symmetry, said 
third gate should always have a low output. If the waves are slightly out 
of phase, the output of said third gate will be a string of short pulses, 
with the length and position of the pulses being the same as the time 
between the zero crossings of the two waves. Remembering that positive 
pulses are generated at both positive and negative zero crossings, so the 
repetition rate is twice the line frequency, a distinction can be made 
concerning which wave leads so that the appropriate pulses can be passed 
through to charge a storage capacitor to the correct polarity and to block 
the inappropriate pulses. 
To this end, a plurality of monostable multivibrators have timing networks 
(200,000 ohms and 0.022 microfarad) that provide output pulses of about 
2.4 milliseconds, much longer than the 0.53 millisecond pulse length 
necessary to give a full scale (20 percent) power factor reading, are 
employed. A first one is triggered by the positive going zero crossing of 
the R (reference) wave and a second one by the negative going zero 
crossing of the same R wave. A first NAND gate takes its input from the 
NOT Q outputs of said first one and said second one, so the output of said 
NAND gate is low most of the time; it goes high, however, for 2.4 
milliseconds starting at each zero crossing (positive or negative) of the 
R wave. If the R wave is leading the C wave, as it should for a positive 
power factor, then the pulse generated by said third EXCLUSIVE OR gate 
supra at each zero crossing, occurs during the 2.4 millisecond pulse 
generated by said first NAND gate. The two pulses drive the output of a 
second NAND gate low for a time equal to the shorter of the two pulses. If 
the C wave is leading the R wave, then the pulse generated by U5C occurs 
before the 2.4 millisecond pulse, and said second NAND gate is not driven 
low. The net result is a negative going pulse at said second NAND gate 
output when, and only when, any zero crossing of the R wave occurs before 
that of the C wave. An additional requirement is that the waves be within 
90 degrees of each other. 
The pulses from said second NAND gate go from plus 15 volts to zero volts. 
These are capacitance-coupled to a two-stage, noninverting buffer 
operating between minus 15 volts and zero volts. Thus, the delivered 
pulses are negative going from zero to minus 15 volts. Said pulses feed 
into an inverting, zero impedance (current summing) amplifier, so the 
negative pulses charge a storage capacitor such as to drive said amplifier 
output positive (point M, see discussion of FIG. 5 supra). 
Likewise, a mirror bank of circuit components operate in an identical 
manner to that described above to produce negative going pulses at the 
output of still another second NAND gate when, and only when, any zero 
crossing of the C wave occurs before that of the R wave. These pulses are 
subsequently inverted by an EXCLUSIVE OR gate, pass through a diode to 
charge said storage capacitor so as to produce a negative output at said 
point M. 
This circuit reads both positive and negative power factor. Although 
negative power factor cannot actually occur in insulation, the circuit 
must still respond to it, because alternate or random pulses of opposite 
polarity can occur due to waveform distortion, dissymmetry, or noise. 
Pulses of both polarities must be properly averaged together for accuracy 
near zero power factor. 
The resulting voltage developed at point M (10 volts dc positive for full 
scale power factor of 20 percent) is passed through a resistor to a 10 
volt output. It is also applied to a current pump composed of operational 
amplifier and the associated 10,000 ohm resistance bridge. The current 
pump supplies up to 1 ma to the current output, which is independent of 
the external resistance up to the voltage compliance limit. The output of 
the operational amplifier must always be twice the voltage at the current 
output terminal. Since the voltage at the output of the operational 
amplifier is limited to about 12 volts, the voltage at the current output 
is limited to 6 volts. Thus, the maximum load resistance is 6,000 ohms if 
full scale current of 1 milliamp is to be delivered. 
The voltage at said point M is also passed to a voltage divider. Twenty 
percent of the voltage is passed to a digital panel meter, which is set 
for 2 volts full scale. The digital panel meter has 31/2 digits, and the 
decimal point is fixed so that a full scale indication of 19.99 is 
provided. The same voltage divider passes 25 percent of the voltage at 
said point M to a voltage comparator, which drives an alarm relay. A diode 
protects the output transistor of said voltage comparator from transients 
caused by the associated inductive relay coil. The level that trips the 
relay may be set by a potentiometer or, for better alarm setting accuracy, 
a switch with precision resisitors can be substituted therefore. It and a 
resistor comprise a voltage divider between plus 15 volts dc and ground. 
Hysteresis and positive feedback are provided by the use of a pair of 
resistors, assuring that the relay picks up completely with a minimum of 
chatter. The amount of hysteresis is about 1.5 percent of full scale or 
about 0.3 percent power factor. If the power factor decreases to 0.3 
percent below the alarm set point, the alarm resets. A capacitor is 
utilized to guard against the alarm being set off by noise or a transient. 
EMBODIMENT TWO 
The second embodiment of the present invention is the one that we feel is 
the best version of the apparatus because it is more easily expandable and 
provides more information than the first embodiment. The attendant 
computer equipment includes a monoboard microcomputer module, two ADC 
modules, a parallel interface module, a battery backed random access 
memory and realtime clock, and a power supply module. A first 
multiconductor bus is provided to connect the two ADC modules, and the 
parallel interface modules to the microcomputer module and to the power 
supply module. A second multiconductor bus is provided to connect the 
microcomputer module to the battery backed memory and the realtime clock 
module. Both buses have provisions for connections to additional modules. 
An ambient temperature probe is used with this second embodiment of the 
present invention and the output therefrom feeds into the control unit 
computer through an ADC. 
The active range of this temperature probe is 0.degree. F. to 140.degree. 
F., corresponding to a voltage output of 3.8 to 5.0 volts. The large 
common-mode voltage is subtracted digitally, rather than using a 
drift-prone analog circuit. The temperature sensor (804) is mounted 
snugly, with epoxy, into a hole in the probe case, so that the whole probe 
becomes a heat sink which assumes ambient temperature. This construction 
provides "peak filtering," with a time constant of about 15 minutes. The 
probe is filled with a potting compound for waterproofing. The entire 
probe is calibrated over the active temperature range, and appropriate 
constants are stored in the computer. Sufficient accuracy could probably 
be attained by calibrating only at room temperature, however. 
An alternative temperature measuring scheme which has also been used 
involves utilizing the digital BCD output of a commercial digital 
thermometer. The BCD output is interfaced through an input/output (IO) 
port on the computer. 
A transducer interface amplifier is used for monitoring temperature, 
pressure, or any other quantity from a transducer with 0-to-5 volt or 
4-to-20 ma dc output. The amplifier has unity gain, with no provision for 
zero or offset null, which, if needed, is done in other places. 
Depending on the transducer output the value of the resistor between said 
output and ground may be, for a 4-to-20 milliamp transducer, 250 ohms, so 
that 1-to-5 volts is passed to the operational amplifier. On a 0-to-5 volt 
transducer, the value of said resistor is 1 megohm, which serves only to 
hold the output near zero volts when there is no input connection. 
A pair of resistors, a varistor, and a capacitor provide transient 
protection. One of said resistors and said capacitor also form a one-pole, 
low-pass filter with a 3 dB cutoff frequency of 0.16 Hz. The operational 
amplifier has a field effect transistor input, so its input impedance is 
much higher than the 1 megohm filter impedance. Said operational amplifier 
is connected for unity gain, so the 1-to-5 or 0-to-5 volts across the 
resistor between said transducer output and ground, after being filtered, 
appears at the output to the ADC. The transducer power supply may be 
located in the control unit or at the transducer. It should be run on 
separate wires if a 0-to-5 volt transducer is used. 
A synchronized frequency multiplier circuit is used in the second 
embodiment of the instant invention to provide timing pulses for the ADC. 
It uses a CMOS phase-locked loop, CD4046B, which has a phase comparator 
(phase comparator II) that allows phase lock between two waves without 
allowing any phase difference between them. The input is usually taken 
from the voltage reference source, and is usually either 67 or 120 volts 
ac. A resistor and a pair of zener diodes limit the voltage and protect 
the voltage comparator, which with a pair of resistors, further squares up 
the waveform. A transistor, along with another pair of resistors, converts 
the wave to a 0-to-5 volt square wave compatible with that needed by the 
phase-locked loop. 
For the phase-locked loop to operate as a frequency multiplier, its VCO 
must operate at the desired high frequency. Since the input frequency of 
60 Hz is to be multiplied by 32, the VCO operates at 1920 Hz. Still 
another pair of resistors and a capacitor set a frequency range to include 
1920 Hz. Since the signal input to the phase comparator (1123) is 60 Hz, 
the VCO output of 1920 Hz is divided by exactly 32 by a counter. The 
counter output of 60 Hz goes to a second phase comparator input. The phase 
comparator output, which is the frequency/phase error signal, is filtered. 
The VCO input fine tunes the VCO frequency to attain and hold frequency 
and phase lock with the input signal. A buffer passes the 1920 Hz square 
wave to a first output and the 60 Hz square wave to a second output. The 
waves are precisely synchronous with each other and tightly locked to and 
in phase with the input wave. By using these waves to control the ADC, 
exactly 32 voltage samples can be taken of each cycle of each input wave, 
or one sample every 11.25 degrees. 
As noted supra the instant invention and the apparatus employed in the 
conduct thereof effectively compares the capacitance tap current with the 
applied voltage, taken from a voltage transformer, and computes power 
factor and capacitance by adaptations from the following formulas: 
##EQU2## 
where v and i are the instantaneous values of reference voltage and 
capacitance tap current, respectively, V and I are the RMS values of the 
same voltage and current, T is one period of line frequency, f is the 
system frequency, and A is the phase angle (expressed in radians) by which 
i leads v. Since A is normally between 1.47 and .pi./2 (i.e., 84 to 90 
degrees, with the power factor 10 percent or less), the approximations are 
valid. 
In the practice of the instant invention in this "Embodiment Two" the first 
and third formula just supra are utilized, however, they cannot be used 
directly because the computer must deal with discrete points rather than 
continuous functions. The voltage and current waveforms are synchronously 
sampled precisely 32 times each cycle for 16 cycles (about 0.27 seconds). 
Samples taken at the same angle for each of 16 cycles of the wave are 
summed together until a complete set of 32 sums has been acquired. Thus, 
for example, we have a sum of 16 samples made on successive cycles at zero 
degrees, another sum at 11.25 degrees, another at 22.5 degrees, etc., on 
up through the 32nd sum at 348.75 degrees. 
Because a capacitive shunt is used to detect the capacitance tap current, a 
90 degree delay is introduced. To compensate for this 90 degree delay, the 
reference voltage is also delayed 90 degrees. This delay is accomplished 
digitally, by shifting backwards in time, along the reference wave, by 8 
sample sums. Since only 32 sample sums (one equivalent cycle) are saved, 
the 8 sums representing the leading 90 degrees are wrapped back to 
represent the last 8 sums of the previous equivalent cycle. 
The formulas shown above are modified as shown below so as to allow the 
computer and software instructions associated therewith to utilize the 
discrete sampling points and the various correction factors necessary for 
the complete, concise, and correct application of the analog input data 
from the capacitance tap adaptor, reference voltage source, and 
temperature transducer. 
##EQU3## 
where: D is a correction factor, determined during calibration, to 
compensate for phase shift in the analog circuits; 
X and Y are the sample sums of the voltages delivered to the computer from 
the capacitance tap adaptor and the referenced source, respectively; 
T.sub.i is the most recently computed estimated internal temperature in 
degrees Fahrenheit; 
T.sub.i(m) is the estimated internal temperature presently being computed; 
T.sub.i(m-1) is the estimated internal temperature computed one measurement 
period previously; 
T.sub.a(m) is the most recently measured ambient temperature in degrees 
Fahrenheit; 
A is 1/K; 
K is the time constant expressed in units of the measurement period 
(usually minutes); 
B is 1-A; 
H.sub.o, H.sub.1, H.sub.2, H.sub.3 . . . H.sub.8 are the coefficients of 
the best fit eighth order power series, computed by the method of least 
squares, to describe the power factor temperature characteristics 
according to the standard instrument transformer correction curve shown in 
the article "Variations of Power Factor with Temperature" by A. L. Rickley 
and S. H. Osborn, Jr.; 
C.sub.s is the total shunt capacitance, including the capacitance tap, the 
capacitance tap adaptor, and the cable, in picofarads (pf); 
R.sub.p is the potential transformer (pt) ratio; 
G.sub.p is the gain of the voltage reference interface amplifier; and 
G.sub.c is the gain of the capacitance tap interface amplifier; 
All summations in the above formulas are carried out from n=1 to n=32, the 
number of samples per cycle. 
GENERAL MICROCOMPUTER SOFTWARE DESCRIPTION 
1.0 Startup Routine 
1.1 Initialize peripheral ports for printers, terminals, alarms, and analog 
sampling control interrupts. 
1.2 Clear to zero all accumulator memory used for summations. Clear status 
and control flags. 
1.3 Compute correction factors for percent power factor and capacitance 
computations. Correction factors are functions of the electrical 
parameters of the interface network that are determined by calibration and 
stored in nonvolatile memory. Compute scale factors and coefficients for 
conversion of other analog inputs--temperature, pressure, etc.--to desired 
units. 
2.0 Wait for Event Routine 
2.1 Check Time Instruction 
Check time and if time for measurement, go to Measure Subroutine; 
otherwise, skip the Action Instruction Set (2.2) and proceed to Keyboard 
Entry Instructions (2.3). 
2.2 Action Instruction Set 
Following measurements and computations, check results of computations and 
determine action to be taken--alarm, print record, daily averages, monthly 
averages, etc. 
2.3 Keyboard Entry Instructions 
Check for operator entries from terminal and perform required task. Print 
record(s), list variables, list command table, update variables, etc. 
2.4 Return to Check Time Instruction (2.1) 
3.0 Major Subroutines 
3.1 Measure 
3.1.1 Setup for response to analog sampling control interrupts. Setup 
arrays for data accumulation. Begin analog-to-digital conversion and 
accumulation of data. Perform intermediate calculations and continue until 
all samples for the measurement interval are obtained. 
3.1.2 Compute results--percent power factor, capacitance, temperatures, and 
other desired results. Compute internal temperature and temperature 
corrected percent power factor. Check for results within specified bounds 
and sum good data to accumulators for daily and monthly averaging. 
3.1.3 Return to Wait for Event Routine, Action Instruction Set (2.2) 
3.2 Percent Power Factor and Capacitance Computation 
3.2.1 Data Accumulation 
The Measure subroutine has resulted in he summing of up to 256 data sets 
for each piece of monitored equipment, with each data set consisting of 32 
samples each of the current and voltage waveform taken over one cycle of 
the waveform. Data may be taken continuously every 11.25 degrees for up to 
256 cycles, or may be taken in 16 groups of 16 cycles. The current from 
each piece of monitored equipment is sampled in rapid succession, with its 
particular reference voltage being sampled simultaneously. 
3.2.2 Accumulated Data Format 
The resulting format is a combined set of 32 sample sums each of the 
current and voltage waveforms of each piece of monitored equipment. Each 
set appears to be from one cycle, but actually represents a total of many 
cycles. 
3.2.3 RMS Computation--Current and Voltage 
Each of the 32 waveform sample sums is squared. They are then added 
together and this sum is, in effect, divided by 32 times the number of 
original data sets. The square root is then taken. The result is the 
equivalent RMS current and RMS voltage for each piece of equipment 
monitored. 
3.2.4 Watts Computation 
Each voltage is delayed exactly 90 degrees by shifting eight samples back 
(or 24 samples forward) in time. Each current sample sum is then 
multiplied by its new equivalent voltage sample sum and the products are 
added together. This sum is, in effect, divided by 32 times the number of 
original data sets. Expressed mathematically: 
##EQU4## 
Where W is watts, X and Y are the individual current and voltage sample 
sums, respectively, and S is the number of data sets. In the first 90 
degrees of the current wave, when n minus 8 is negative, n plus 24 is 
used. The reason for the 90 degree voltage delay is to compensate for the 
90 degree current delay caused by using a capacitive rather than resistive 
shunt in the capacitance tap adaptor. A resistive shunt was not used 
because it would have increased the high frequency interference and 
harmonic content of the wave being sampled. 
3.2.5 Final Computations 
The power factor and capacitance are computed by the following expressions: 
EQU PF=W/(IV) 
EQU C.perspectiveto.KI/V 
Where PF and C are the power factor and capacitance, respectively, of the 
equipment being monitored; W, I, and V are the watts, RMS current, and RMS 
voltage computed above; and K is a constant which includes 1/2.pi.f, the 
voltage reference ratio, the capacitance of the capacitance tap adaptor, 
and other corrections so that C can be expressed accurately in picofarads. 
The power factor is corrected for phase error in interface equipment and 
multiplied by 100 to be expressed in percent. The formula for capacitance 
is an approximation which ignores the resistive component of I. This 
approximation is excellent with a power factor of 10 percent or less and 
adequate up to 20 percent. 
3.3 Computation of Temperatures, Pressures, Hydrogen, etc. 
Various transducers for measurement of required parameters may be 
incorporated. At the end of each measurement interval, the accumulated 
data is processed and scaled for appropriate units. Scale factors are 
stored in nonvolatile memory and may be changed by an operator if 
required. 
3.4 Alarm Analysis 
There are certain preset alarm levels that are stored in nonvolatile memory 
and may be changed by an operator. The alarm levels may be set for percent 
power factor, internal pressure, parts per million hydrogen, etc. Each 
variable may be independently assigned specified alarm levels including no 
alarm. There are two alarm levels for each variable; a warning level and a 
danger level. After each measurement interval, each variable is checked 
for an alarm condition. If the preset warning alarm level has been 
exceeded for a specified time, the data may be printed with flags for each 
variable. If the preset danger alarm level has been exceeded for a 
specified time, various methods may be employed to alert personnel or 
automatically disconnect the monitored equipment. The data may also be 
printed with appropriate flags. 
3.5 Trend Analysis 
Following each measurement interval, specified variables (percent power 
factor, pressure, hydrogen, etc.) may be checked for specified increases 
over the most recently printed data. When the specified changes are 
detected, a record with appropriate flags is printed. 
3.6 Daily and Monthly Averages 
Specified results of each measurement are accumulated and daily averages of 
temperatures, temperature corrected percent power factor and capacitance 
are computed and stored in nonvolatile memory. At the end of each month, 
monthly averages are computed and stored in nonvolatile memory. Averages 
for the most recent 32 days and for the most recent 12 months are saved 
and may be listed by an operator command. Daily averages include flags to 
indicate exceptional variations (increases) in percent power factor, 
pressure, hydrogen, etc. 
3.7 Compute Estimate of the Internal Temperature of Monitored Equipment 
Computation of percent power factor corrected for temperature is based on 
standard temperature correction curves, using an estimated internal 
temperature which is derived from measured ambient temperature. A step 
change in ambient temperature is assumed to produce an exponential change 
in internal temperature, so that eventually the internal temperature would 
become essentially equal to the new ambient temperature. 
To implement this concept for digital computation, where the ambient 
temperature is measured at discrete time intervals, the expression below 
is used: 
EQU T.sub.i(m) =AT.sub.a(m) +BT.sub.i(m-1) 
Where: 
T.sub.i(m) is the updated internal temperature being computed 
T.sub.a(m) is the most recently measured ambient temperature 
T.sub.i(m-1) is the most recent previously computed estimated internal 
temperature 
A is 1/time constant, K 
B is 1-A 
The time constant, K, is stored in nonvolatile memory and may be changed 
by an operator. It is typically several hours, and is expressed in units 
of the discrete time interval. 
4.0 Other Subroutines Used 
4.1 Square root 
4.2 Binary to decimal conversion 
4.3 Decimal to ASCII conversion 
4.4 ASCII to decimal conversion 
4.5 Decimal to binary conversion 
4.6 Double precision multiply 
4.7 Double precision addition 
4.8 Extended divide 
EXAMPLES 
In order that those skilled in the art may better understand how the 
present invention can be practiced, the following examples are given by 
way of illustration only. 
EXAMPLE I 
In the pursuit of further information gathered for the purpose of more 
clearly defining the parameters affecting the practice of the instant 
invention the investigations herein were made in response to the excessive 
number of violent current transformer (ct) failures at the Raccoon 
Mountain PSP. Accordingly, in our earliest work a computerized ct damage 
monitor was developed in the Central Laboratories and first installed in 
the Raccoon Mountain switchyard. The first pilot model was tested on the 
ct's on breaker 834. Subsequently, an upgraded prototype was installed on 
the ct's on breakers 874 and 878. The ct monitor records power factor and 
capacitance of the ct insulation, along with the amount of hydrogen gas in 
the oil if the ct is equipped with a Syprotec (Manleh) Hydran 201R 
monitor. NOTE: No endorsement of this type of monitor, or for that matter, 
any other type of equipment named herein, is intended, or to be so 
construed. The embodiment of the ct monitor prototype installed at the 
Raccoon Mountain site was designed for up to 12 cts, but only 6 were 
monitored at any one time. Several ct's on breakers other than 874 and 878 
were monitored for a time by substituting them for ct's not showing any 
problems. 
The purpose of the ct monitor is to detect impending ct failures so that 
the ct's can be removed from service before they explode. 
The original prototype monitor at Raccoon Mountain "caught" four ct's that 
almost surely would have failed soon. All four were associated with 
breaker 874. They include the original ct's in the B-phase and C-phase 
positions and the replacement for each, which were of the same vintage. In 
addition, a ct on breaker 888, while not monitored, was deenergized, 
tested, and removed from service because of very high hydrogen content and 
Doble power factor reading. This ct was suspected and tested because it 
had characteristics and history similar to the replacement ct on 874, 
C-phase, which had shown very rapid deterioration on the ct monitor. 
Several unmonitored ct's exploded during the summer, and several others 
were removed from service because periodic sampling revealed high hydrogen 
content in the oil. 
When the ct monitor was first installed, the highest power factor was found 
on C-phase of breaker 874 (874C). The next highest measured power factor 
was found on 874B. Neither of the indicated power factor values were 
considered excessive, but with no experience on what "excessive" was, they 
were monitored very closely. The power factors went up and down with 
temperature fluctuations throughout the day. This was expected because 
heat, in general, degrades insulation. The daily average power factors 
also fluctuated with changes in average temperature from day to day. 
This fluctuation was somewhat surprising since the daily averages are all 
corrected to 68.degree. F. according to the standard instrument 
transformer correction table. It turns out that when the power factor of a 
ct is relatively high, it increases more with a temperature increase than 
the correction table would predict. Good cts, on the other hand, behave 
about like the table indicates they should. 
The extreme temperature sensitivity of the high power factor ct's is 
undoubtedly caused by self heating of the insulation. In these cts, every 
1 percent power factor results in about 60 watts (205 Btu/hr) of heat 
which must be removed from the insulation. Since most of this heat is deep 
in the paper and not readily removed, the temperature must increase to 
well above ambient when the power factor is high. This temperature 
increase causes the power factor to go even higher. The result is 
regenerative amplification, an effect similar to that shown by a 
regenerative brake or an electronic amplifier with positive feedback. 
Though ct's 874C and 874B went to relatively high power factors on warm 
days, they came back down on cool days, and no trends were observed for 
the first few months. Data sheets and graph sheets, compiled through this 
investigation, showed that for all of the daily average power factors for 
the monitored ct's for a period of about three months, three of same, to 
wit, 874C, 874B, and 888C (monitored only for eight weeks) showed similar 
patterns of power factors being about an order of magnitude higher 
(0.8-2.0 vs. 0.1-0.2+) than the rest of the ct's. In addition, these three 
ct's showed substantial fluctuations of observed power factor with daily 
fluctuations of ambient temperatures. The observed power factors of the 
others were much lower, and showed little temperature fluctuation, 
indicating that the standard temperature correction table is about right 
for good cts. It was also observed that the fluctuations of 874C, 874B, 
and 888C tended to lag behind the ambient temperature; in effect, one 
day's average power factor is controlled mostly by the preceding day's 
average temperature. It is postulated that this themal lag is due to the 
10-to-12 hour thermal time constant exhibited by these cts. 
Original 874C (K71911-01 No. 2)--When we compared the power factor of any 
of these ct's with its power factor at the same approximate temperature on 
an earlier date, no significant difference can be seen during this three 
month test period; no trends can be observed. This situation changed 
during the fourth month, however. Toward the end of the fourth month, the 
power factor on 874C started increasing without correspondingly increasing 
temperature. This trend continued until early in the fifth month, when ct 
874C was deenergized to take an oil sample. (During this period, the 
continuous gas monitors were back at the factory being modified.) A 
laboratory analysis of the oil showed only 50 parts per million (ppm) of 
hydrogen, so the ct was returned to service. The power factor continued 
upward until the ct was removed from service permanently one week later. A 
gas analysis then showed 220 ppm hydrogen. 
Replacement 874C (K71911-01 No. 4)--This ct had previously been removed 
from another breaker at Raccoon Mountain because of damage caused by the 
explosion of an adjacent ct. It had been sent to the Power Service Shops 
for repair, then returned to Raccoon Mountain for further service. From 
August 20 through August 27, the power factor of the replacement 874C held 
reasonably steady at about one percent. The upturn on August 28 could be 
attributed to rising temperature. On August 29, however, a definite 
increase was observed, as the replacement 874C rose above 874B (which had 
also started upward by this time). From then on, the power factor of 874C 
rose continuously. The gas was essentially zero until August 30, when it 
rose to an average of 55 parts per million. On August 31, the gas went off 
scale (above 1000 ppm) to stay. The replacement 874C was removed from 
service on September 3, and a lab analysis of an oil sample showed 9720 
ppm of hydrogen. 
888A (K71911-01 No. 5)--This ct was installed, with no monitoring hardware, 
at about the same time the replacement 874C was installed. The two ct's 
had similar histories; both had been damaged and repaired at the Power 
Service Shops. When the power factor and gas on the replacement 874C went 
very high, 888A became a suspect. At the first opportunity, it was 
deenergized, with the intention of installing hardware for monitoring 
power factor. An oil sample was taken, and a lab analysis showed 12,930 
ppm of hydrogen. A Doble power factor test, made a few hours after the ct 
was deenergized (and while still warm) showed 8.3 percent power factor on 
the UST test (between the top of the bushing and the capacitance tap). The 
ct was not returned to service. 
After about five weeks of operation, this ct was unwrapped to search for 
damage. Wrinkles were found in the paper and semi-conductor layers, some 
of which showed apparent partial discharge tracking. A sticky, wax-like 
material was found throughout much of the insulation, indicating 
decomposition of the oil because of high temperature and/or high electric 
field strength. Evidence of high temperature was found, exhibited by the 
discoloration of some of the copper braid and a slight charring of some of 
the paper. The ct was judged to have reached a state that was unfit for 
service; however, the specific cause of the damage could not be 
determined. 
Original 874B (K71911-01 No. 6)--This ct showed the first signs of power 
factor increase at about the time the replacement 874C started its 
irreversible, catastrophic climb toward destruction. The upturn of 874B 
started about a week after the end of a week's outage, and increased very 
slowly at first. Through the next month the power factor appeared to hold 
steady except for daily temperature fluctuations. If we look at the 
graphical depiction of the temperature, however, we see the normal 
downward trend expected this month (September), which is not reflected in 
the power factor of 874B. Also, 874B can be compared with 844A, which has 
a definite downward trend through the month. 
Since the effective power factor increase in 874B was so slow, we had hoped 
it might last through the winter. However, the power factor started up 
rapidly and unmistakably and the ct was taken out of service about a month 
after 874C started toward destruction supra. We noticed that the gas had 
increased through most of the previous month, but was not excessive. When 
the ct was deenergized, a laboratory analysis indicated a relatively high 
gas content, but the continuous gas monitor reading had decreased to a 
very low value. However, while the ct sat idle for a week, the continuous 
gas monitor reading increased to the highest value yet seen on this ct, a 
maximum of 294 ppm. This increase was probably caused by better gas 
diffusion and oil circulation during the week, since it is not likely that 
any new gas would be generated with the ct deenergized. 
Replacement 874B (K71911-03 No. 2)--This ct has a history similar to the 
replacement 874C and 888A. This ct was damaged less severely than they 
were, however, and Power Service Shops personnel were able to repair it 
without removing the windings from the oil. 
From the time it was installed the power factor was high. The power factor 
increased steadily until the ct was removed from service two days later. 
The gas monitor read essentially zero the whole time. 
It is customary for a high power factor to increase during the first day or 
two of operation due to self heating (mentioned earlier). After two days, 
however, it should have leveled off. Since it did not and was already up 
to almost 5 percent, the ct was removed from service. 
EXAMPLE II 
In the pursuit of still further information, gathered for the purpose of 
more clearly defining the parameters affecting the practice of the instant 
invention, the investigations herein were made to determine the response 
of various types of cts via a monitoring system, somewhat similar to that 
used at Raccoon Mountain and described in Example I supra. In the tests 
comprising this Example, the setup was to allow for testing a plurality of 
cts at both 161 kv and 500 kv operating levels and emphasis was directed 
herein to ascertain the reasons underlying ct failure, and also to 
determine the gas and power factor levels that could be determinative of 
alarm conditions for removal of same from service. 
A ct test facility has been constructed at Wilson Substation. Barriers were 
buit in the 500 and 161 kv switchyards so that defective cts could be 
tested, and if they failed, minimal damage would be done. Fire sensors 
were installed to alert operators in case a failure resulted in a fire. 
Four cts, two each at 500 and 161 kv, can be tested simultaneously. 
An updated power factor monitor was installed at Wilson which monitored 
pressure in addition to power factor, capacitance, hydrogen, and 
temperature. In all, three 161 kv cts and two 500 kv cts were tested. Four 
of the five had a history of high power factor and/or gas. 
Two of the 161 kv cts had been removed from Raccoon Mountain the previous 
year because of high power factor and gas. At Wilson, the power factor and 
gas went even higher, confirming the reliability of either type monitor. 
It was also determined that oil pressure, which normally fluxuates with 
temperature, also increases with high power factor (because of 
self-heating) and could be used for monitoring purposes. One of said 161 
kv cts got so bad that (in addition to a power factor well over 10 percent 
in cool weather) its dissolved hydrogen reached more than 30,000 ppm. The 
excessive gassing also caused a sharp pressure increase because when the 
oil became saturated, the undissolved gas formed a blanket at the top of 
the ct. 
Another 161 kv ct was removed from service at a nuclear plant because of 
gassing. It was modified to allow a hydrogen probe to be installed in such 
a way as to maximize response. It was then degassed and monitored at the 
Wilson facility for six months. The power factor was relatively high, but 
did not increase during the period, except for normal fluxuations with 
temperature. No hydrogen was detected and there were no abnormal pressure 
trends. The ct was judged tentatively serviceable, but because of its 
history, will not be returned to service. Power factor and gas increases 
in the past indicates a high probability of further increases in the 
future. 
The two 500 kv cts have also been monitored for six months. Both have shown 
low power factor (generally less than 0.3 percent), low gas (less than 200 
ppm), normal pressure, and no trends. Though judged serviceable, they will 
probably not be returned to service because of past problems. 
EXAMPLE III 
Because of the large number of ct failures and near failures at Raccoon 
Mountain the decision was made to replace all 42 of the 161 kv cts there 
with explosion-proof SF-6 types. A substantial period of time would be 
required for delivery, however. Since more failures could be expected 
during the ensuing hot weather months the prototype ct monitor was further 
improved, updated, and expanded to 24 channels, power factor only (the 
practical limit with existing hardware). This expansion allowed most of 
the remaining original cts to be monitored. 
During said hot weather months, five cts were "caught" and removed from 
service because of sharply increasing power factor. All subsequently 
showed an increase in combustible gas by laboratory analysis (no hydrogen 
monitors were used). Another ct (with no monitor) was caught by routine 
gas sampling. For the first time in several years, thanks to the power 
factor monitor, there were no explosive failures. 
It now has been established that there are at least two viable methods by 
which current transformers can be monitored for impending failure. Power 
factor monitoring appears to provide the quickest indication of trouble, 
but gas monitoring should also provide adequate warning if a continuous 
monitor such as the Hydran 201R can be used. As noted supra, no 
endorsement of any product named herein is intended or to be so construed 
hereby. It appears that periodic sampling of oil will not always provide 
adequate warning since it would be impractical to sample often enough to 
always catch a ct with very rapid deterioration. 
A further benefit of this project is that at least nine ct's were removed 
from service before violent failure. They may or may not have been caught 
by routine oil sampling if the ct monitor had not been in service. 
INVENTION AMETERS 
After sifting and winnowing through the data supra, as well as other 
results and operation of our new, novel, and improved technique, including 
methods and means for the effecting thereof, the operating variables, 
including the acceptable and preferred conditions for carrying out our 
invention are summarized below. 
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Variables Preferred Limits 
Most Preferred Limits 
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Voltage reference 
Magnetic voltage 
Lower accuracy transformer; 
source transformer, 0.3% 
capacitive potential device 
accuracy class 
Voltage reference 
external loading: 
Magnetic voltage 
Light, unchanging 
Normal, unchanging 
transformer 
Capacitive potential 
None Light, unchanging 
device 
Cable connection to 
2-conductor shielded 
Any type, anywhere, but only if 
voltage reference source 
all the way to source 
external loading is very light and 
unchanging 
Capacitance tap adaptor 
2-1 microfarad 
Any stable capacitor, 
polystyrene capaci- 
0.1 microfarad or higher 
tors in parallel 
Voltage drop across 
50 to 100 volts 
10 to 140 volts 
capacitance tap adaptor 
Cable connection to 
2-conductor shielded 
Single conductor shielded, unshielded or 
capacitance tap adaptor 
part of a bundle if no other ac wiring is 
in the bundle or nearby and voltage across 
capacitance tap adaptor is 50 volts or more 
Capacitance tap external 
None Light loading acceptable if unchanging and if 
loading capacitance tap is carefully phase 
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checked. 
While we have shown and described particular embodiments of our invention, 
modifications and variations thereof will occur to those skilled in the 
art. We wish it to be understood therefore that the appended claims are 
intended to cover such modifications and variations which are within the 
true scope and spirit of our invention.