Power conversion apparatus

A power inversion apparatus includes a smoothing capacitor, first and second primary coils, a secondary coil, first to fourth switches of bridge circuit switches, a clamp capacitor, and a switch controller. The switch controller calculates a lower-arm duty ratio of each of the first and second switches using a map or a mathematical expression by feed-forward control based on an input voltage. The switch controller outputs a fixed value that is equal to or greater than a maximum value of the lower-arm duty ratio within a variation range of the input voltage as an upper-arm duty ratio of each of the third and fourth switches. The switch controller generates a pulse width modulation signal based on the calculated lower-arm duty ratio and the fixed value of the upper-arm duty ratio, and outputs the pulse width modulation signal to the bridge circuit switches.

BACKGROUND

Technical Field

The present disclosure relates to a power conversion apparatus.

Related Art

A power conversion apparatus that converts electric power is known. In the power conversion apparatus, the electric power is supplied to a primary side of a transformer by a switching operation. The converted electric power is supplied to a secondary side to which a load is connected.

SUMMARY

The present disclosure provides a power inversion apparatus includes a smoothing capacitor, a first primary coil and a second primary coil, a secondary coil, first to fourth switches of bridge circuit switches, a clamp capacitor, and a switch controller. The switch controller calculates a lower-arm duty ratio that is a duty ratio of each of the first switch and the second switch using a map or a mathematical expression by feed-forward control based on an input voltage. The switch controller outputs a fixed value that is equal to or greater than a maximum value of the lower-arm duty ratio within a variation range of the input voltage as an upper-arm duty ratio that is a duty ratio of each of the third switch and the fourth switch. The switch controller generates a pulse width modulation signal based on the calculated lower-arm duty ratio and the fixed value of the upper-arm duty ratio, and outputs the pulse width modulation signal to the bridge circuit switches.

DESCRIPTION OF THE EMBODIMENTS

For example, in a high-voltage generation apparatus that is described in JP-A-2001-251854, a dust-collecting electrode of an electrostatic air cleaner is connected to the secondary side of a transformer for high-voltage generation. The apparatus detects a current that flows to switching elements (hereafter, switches) that are connected to the primary side of the transformer for high-voltage generation and feeds back the detected current to a control circuit. The control circuit controls a duty ratio of the switches based on a fed-back value of the electrical current. Dust collection performance of the electrostatic air cleaner is thereby kept constant.

When input voltage suddenly changes as a result of an operation state or the like, the current on the transformer primary side is required to be promptly compensated and output on the transformer secondary side is required to be stabilized. However, in the conventional technology in JP-A-2001-251854, because feedback control of the switch current is performed for the sudden change in the input voltage, response is delayed. As a result of the delay in response, overshooting or undershooting of output power, or overcurrent relative to a command value occurs.

In addition, conventionally, a resonant inverter that uses a push-pull circuit is known. A typical conventional push-pull circuit includes a smoothing capacitor and two switches. As a result of the two switches being alternately operated, a transformer primary-side current that flows through first and second primary coils connected to a shared center tap is controlled. A capacitive load is connected to a secondary coil of the transformer. An output current that flows to the load resonates due to an LC component of a secondary circuit. In a resonant inverter such as this, the current that is supplied to the first and second primary coils is primarily taken from the smoothing capacitor. Therefore, the burden placed on the smoothing capacitor is large and ripple current tends to increase.

In this regard, use of an active-clamp push-pull circuit can be considered. As shown inFIG. 1, the active-clamp push-pull circuit includes two lower arm switches Q1and Q2, two upper arm switches Q3and Q4, and a clamp capacitor C2. Source terminals of the upper arm switches Q3and Q4and drain terminals of the lower arm switches Q1and Q2are respectively connected to switch-side end portions23and24of a first primary coil21and a second primary coil22. The clamp capacitor C2is connected between drain terminals of the upper arm switches Q3and Q4and a low-potential input terminal12.

In the active-clamp push-pull circuit, an operation having a period in which the lower arm switch Q1and the upper arm switch Q4are simultaneously turned on and an operation having a period in which the lower arm switch Q2and the upper arm switch Q3are simultaneously turned on are alternately repeated. In this operation, the clamp capacitor C2supports discharge of the smoothing capacitor C1. Consequently, the burden placed on the smoothing capacitor C1can be reduced and ripple current can be reduced.

Here, in light of the issues regarding feedback control in JP-A-2001-251854, a configuration in which feed-forward control of the duty ratio of the lower arm switches Q1and Q2is performed based on the input voltage is used. In this case, when the duty ratio of the upper arm switches Q3and Q4is changed so as to track the duty ratio of the lower arm switches Q1and Q2, a discontinuous mode of the output current may occur. In particular, in a configuration in which switching frequency is changed based on the output power, a plurality of maps are required based on the frequencies because the duty ratio is dependent on the switching frequency in feed-forward control.

It is thus desired to provide a resonant inverter-type power conversion apparatus that uses an active-clamp push-pull circuit, in which the power conversion apparatus suppresses output variations and overcurrent caused by sudden changes in input voltage, reduces ripple current, and prevents occurrence of a discontinuous mode of electrical current.

An exemplary embodiment of the present disclosure provides a power conversion apparatus that includes a smoothing capacitor, a first primary coil, a second primary coil, a secondary coil, first to fourth switches, a clamp capacitor, and a switch controller.

The smoothing capacitor is connected between a high-potential input terminal and a low-potential input terminal to which input voltage of a direct-current power supply is applied. The first primary coil and the second primary coil configure a primary side of a transformer. One of the ends of the first primary coil and one of the ends of the second primary coil are connected to a shared center tap that is connected to the high-potential input terminal. The secondary coil configures a secondary side of the transformer and is connected to a load.

The first switch and the second switch configure a lower arm of a bridge circuit and are alternately operated, at a predetermined switching cycle. Each of the first switch and the second switch has a high-potential side terminal and a low-potential side terminal. In the first switch, the high-potential side terminal is connected to a switch-side end portion that is an end portion of the first primary coil on a side opposite to the center tap, and the low-potential side terminal is connected to the low-potential input terminal. In the second switch, the high-potential side terminal is connected to a switch-side end portion that is an end portion of the second primary coil on a side opposite to the center tap, and the low-potential side terminal is connected to the low-potential input terminal.

The third switch and the fourth switch configure an upper arm of the bridge circuit and are alternately operated at the same switching cycle as that of the first switch and the second switch. Each of the third switch and the fourth switch has terminals. One of the terminals of the third switch is connected to the switch-side end portion of the first primary coil. One of the terminals of the fourth switch is connected to the switch-side end portion of the second primary coil. The clamp capacitor is connected between the other of the terminals of the third switch and the low-potential input terminal and between the other of the terminals of the fourth switch and the low-potential input terminal.

The first switch, the second switch, the third switch, and the fourth switch configure bridge circuit switches. The switch controller calculates a duty ratio that is a ratio of an on-time of each bridge circuit switch relative to the switching cycle, and controls operation of the bridge circuit switches such that at least the fourth switch is turned on during an on-period of the first switch and the third switch is turned on during an on-period of the second switch.

The switch controller may prohibit the first switch and the second switch from being simultaneously turned on and the third switch and the fourth switch from being simultaneously turned on.

The switch controller includes a lower-arm duty ratio calculator, an upper-arm duty ratio calculator, and a pulse width modulation (PWM) generator. The lower-arm duty ratio calculator calculates a lower-arm duty ratio that is a duty ratio of the first switch and the second switch using a map or a mathematical expression by feed-forward control based on the input voltage. The upper-arm duty ratio calculator outputs a fixed value that is equal to or greater than a maximum value of the lower-arm duty ratio within a variation range of the input voltage as an upper-arm duty ratio that is a duty ratio of the third switch and the fourth switch. The PWM generator generates a PWM signal based on output from the lower-arm duty ratio calculator and output from the upper-arm duty ratio calculator, and outputs the PWM signal to the bridge circuit switches.

In the exemplary embodiment, the lower-arm duty ratio calculator calculates the lower-arm duty ratio by feed-forward control based on the input voltage. Therefore, output variations and overcurrent caused by sudden changes in the input voltage can be appropriately suppressed. In addition, instead of a typical push-pull circuit that includes only the smoothing capacitor and the lower arm switches, an active-clamp push-pull circuit that includes the clamp capacitor and the upper arm switches is used. As a result of the clamp capacitor supporting discharge by the smoothing capacitor, the burden placed on the smoothing capacitor can be reduced and ripple current can be reduced.

Furthermore, the upper-arm duty ratio is set to a fixed value that is equal to or greater than the maximum value of the lower-arm duty ratio within the variation range of the input voltage. The upper-arm duty ratio is preferably set to a maximum value that is obtained by a value that is equivalent to dead time being subtracted from 0.5, the dead time being an amount of time ensured between the on-period of the third switch and the on-period of the fourth switch. As a result, the transformer-applied voltage forms a one-pulse waveform. The occurrence of a discontinuous mode of electrical current can be prevented as much as possible. In particular, in a configuration in which switching frequency is changed based on output power, frequency dependency in feed-forward control of the duty ratio can be eliminated. Acquisition and adaptation of maps based on the switching frequencies become unnecessary.

A power conversion apparatus according to a plurality of embodiments will hereinafter be described with reference to the drawings. First and second embodiments are collectively referred to as a present embodiment. The power conversion apparatus according to the present embodiment is a resonant inverter that converts direct-current power that is supplied to a primary side of a transformer by a switching operation of a push-pull circuit and outputs alternating-current power to a secondary side to which a capacitive load is connected. In the resonant inverter, high electric power can be outputted by the switching operation of the push-pull circuit being performed at a frequency that is close to a resonance frequency of an output current.

Configuration and Operations of the Resonant Inverter

First, a configuration and operations of the resonant inverter to which the present embodiment is applied will be described with reference toFIG. 1toFIG. 5. As shown inFIG. 1, a resonant inverter100includes a transformer20that includes two primary coils21and22, and a secondary coil26. Respective one ends of the two primary coils21and22are connected to a shared center tap25. End portions of the first primary coil21and the second primary coil22on the sides opposite to the center tap25are respectively referred to as switch-side end portions23and24. A high-potential input terminal11and a low-potential input terminal12of the resonant inverter100are connected to a positive electrode and a negative electrode of a battery10, and an input voltage Vin of the battery10is applied thereto. The battery10serves as a direct-current power supply. For example, the low-potential input terminal12may be at ground potential, that is, in a grounded state. The center tap25of the transformer20is connected to the high-potential input terminal11.

A smoothing capacitor C1, a first switch Q1, and a second switch Q2are provided on the primary side of the transformer20. The first switch Q1and the second switch Q2configure a basic push-pull circuit. The smoothing capacitor C1is connected between the high-potential input terminal11and the low-potential input terminal21, and smooths the input voltage Vin of the battery10. The smoothing capacitor C1has a high-potential-side electrode17and a low-potential-side electrode18. The smoothing capacitor C1has a relatively high capacitance.

In addition, as a characteristic configuration according to the present embodiment, a clamp capacitor C2, a third switch Q3, and a fourth switch Q4are provided on the primary side of the transformer20. In the present specification, this configuration is referred to as an active-clamp push-pull circuit. The first switch Q1and the second switch Q2configure a lower arm of a bridge circuit. Therefore, the first switch Q1and the second switch Q2are also referred to as lower arm switches Q1and Q2. The third switch Q3and the fourth switch Q4configure an upper arm of the bridge circuit. Therefore, the third switch Q3and the fourth switch Q4are also referred to as upper arm switches Q3and Q4. In addition, the switches of the upper and lower arms are collectively referred to as bridge circuit switches Q1to Q4.

For example, the bridge circuit switches Q1to Q4are configured by metal-oxide-semiconductor field-effect transistors (MOSFETs). When a gate signal is supplied, energization occurs between a drain and a source. In addition, a body diode that allows a current that flows from the source towards the drain is added. Here, an insulated-gate bipolar transistor (IGBT) to which a freewheeling diode is connected in parallel may be used as the switch. In this case, the present disclosure may be interpreted such that the names of the terminals are replaced with collector, emitter, and the like, as appropriate.

In the first switch Q1, a drain terminal is connected to the switch-side end portion23of the first primary coil21. A source terminal is connected to the low-potential input terminal12. In the second switch Q2, the drain terminal is connected to the switch-side end portion24of the second primary coil22. The source terminal is connected to the low-potential input terminal12. The first switch Q1and the second switch Q2are alternately operated at a predetermined switching cycle Ts shown inFIG. 8and the like. As a result, a first current I1and a second current I2that are in opposite directions of each other flow to the first primary coil21and the second primary oil22. In accompaniment, an output current Io of which the direction alternates flows to the secondary side of the transformer20.

In the third switch Q3, the source terminal is connected to the switch-side end portion23of the first primary coil21and the drain terminal of the first switch Q1. In the fourth switch Q4, the source terminal is connected to the switch-side end portion24of the second primary coil22and the drain terminal of the second switch Q2. The third switch Q3and the fourth switch Q4are alternately operated at the same switching cycle Ts as that of the first switch Q1and the second switch Q2. Details of the operation will be described hereafter.

The clamp capacitor C2is connected between the drain terminals of the third switch Q3and the fourth switch Q4, and the low-potential input terminal12. The clamp capacitor C2has a high-potential-side electrode27and a low-potential-side electrode28. The clamp capacitor C2provides a function for supporting discharge performance of the smoothing capacitor C1and reducing ripple current.

On the secondary side of the transformer, electrodes31and32of a capacitive load C3are connected to both ends of the secondary coil26. An end portion of the secondary coil26on the side that is connected to the electrode32is connected to the low-potential input terminal12. As a result of an inductance component of the secondary coil26and a capacitance component of the load C3, resonance is generated in the output current Io that flows through the secondary circuit. When inductance is L and capacitance is C, a resonance frequency thereof is expressed by 1/(2π√LC).

As shown inFIG. 2, for example, the load C3according to the present embodiment is a discharge reactor that is used in an ozone generation apparatus30. In the discharge reactor C3, a plurality of pairs of electrodes31and32are provided along a flow path33. When high-voltage pulse power is supplied between the electrodes31and32, oxygen molecules that pass through the flow path33are decomposed and oxygen radicals are produced. Then, as a result of oxygen radicals (O) reacting with other oxygen molecules (O2), ozone (O3) is produced. For example, the ozone generation apparatus30is mounted of the output current in a vehicle of which an engine is a power source. The ozone generation apparatus30generates ozone for decomposition of unburned CH in exhaust gas. The resonant inverter100adjusts a production amount of ozone by controlling electric power that is outputted to the discharge reactor C3.

Returning toFIG. 1, an input voltage detector15is provided on the battery10side of the resonant inverter100. In addition, at least either of an input power detector16on the primary side of the transformer20and an output power detector36on the secondary side is provided. A switch controller40according to the present embodiment includes a duty ratio calculator50, a power controller60, and a pulse width modulation (PWM) generator70. The duty ratio calculator50performs feed-forward control. The power controller60performs feedback control. The PWM generator70generates a PWM signal and outputs the PWM signal to the gates of the bridge circuit switches Q1to Q4.

The duty ratio calculator50calculates the duty ratio using a map or a mathematical expression by feed-forward control based on the input voltage Vin acquired from the input voltage detector15. Here, the duty ratio is a ratio of an on-time of each of the switches Q1to Q4relative to the switching cycle Ts. Here, the configuration of the duty ratio calculator50shown inFIG. 1is shared with a comparison example that is compared to the present embodiment. Configurations that are characteristic to the present embodiment are shown inFIG. 6, described hereafter.

The power controller60performs feedback control such that actual power P that is acquired from the input power detector16or the output power detector36matches target power Pref. A detailed configuration of the power controller60will also be described hereafter. The PWM generator70generates the PWM signal based on output from the duty ratio calculator50and the power controller60.

Next, an overview of the operations of the active-clamp push-pull circuit will be described with reference toFIG. 3toFIG. 5. InFIG. 1, the current that flows through the first primary coil21is the first current I1. The current that flows through the second primary coil22is the second current I2. The current that flows through the secondary coil26is the output current Io. Regarding the first current I1and the second current I2, a direction from the center tap25towards the switch-side end portions23and24is defined as positive. Regarding the output current Io, a direction from the electrode31of the load C3through the secondary coil26towards the electrode32is defined as positive.

A time chart inFIG. 3shows a relationship between the operations of the switches Q1and Q2, and the changes in the first current I1, the second current I2, and the output current Io. Here, a first period T1during which the first switch Q1and the fourth switch Q4are turned on and a second period T2during which the second switch Q2and the third switch Q3are turned on are alternately switched. Dead time is ignored.

Here, in this example, the first current I1and the second current I2are detected. On/off of each switch is switched at a timing at which the first current I1and the second current I2become equal to a positive switching value ISHIFT. However, the switching timing of the switch is not limited thereto. When the second current I2is greater than the first current I1, the output current Io is positive. When the first current I1is greater than the second current I2, the output current Io is negative.

In the switching cycle Ts, symbols A to F are given to timings at which the first current I1or the second current I2crosses zero, and timings at which the first current I1and the second current I2cross and become equal. At timings A and B during the first period T1, the second current I2respectively crosses zero from positive to negative and from negative to positive. At timing C at which the first period T1transitions to the second period T2, the increasing second current I2and the decreasing first current I1cross. At timings D and E during the second period T2, the first current I1respectively crosses zero from positive to negative and from negative to positive. At timing F at which the second period T2transitions to the first period T1, the increasing first current I1and the decreasing second current I2cross.

FIG. 4A,FIG. 4B,FIG. 5A, andFIG. 5Bshow paths of the first current I1and the second current I2at each timing. In the smoothing capacitor C1and the clamp capacitor C2, an arrow from the low-potential electrodes18and28to the high-potential electrodes17and27indicates discharge. An arrow from the high-potential electrodes17and27to the low-potential electrodes18and28indicates charging. In addition, regarding the direction of the current that flows through the switches Q1to Q4, a direction that flows from the drain to the source is a forward direction and a direction that flows from the source to the drain is a reverse direction.

During a period of timings A to B shown inFIG. 4A, the positive first current I1is discharged from the smoothing capacitor C1, passes from the center tap25through the first primary coil21, and flows through the first switch Q1in the forward direction. The negative second current I2is discharged from the clamp capacitor C2, flows through the fourth switch Q4in the forward direction, passes through the second primary coil22and the center tap25, and charges the smoothing capacitor C1. During this period, the first current I1that is generated as a result of discharge by the smoothing capacitor C1flows through the first primary coil21. In addition, the second current I2that is generated as a result of discharge by the clamp capacitor C2flows through the second primary coil22.

During periods of timings B to C and F to A shown inFIG. 4B, the positive first current I1flows over the same path as that inFIG. 4Ain the same direction as that inFIG. 4A. The positive second current I2flows over the same path as that inFIG. 4Ain a direction that is opposite to that inFIG. 4A. That is, the second current I2is discharged from the smoothing capacitor C1, passes from the center tap25through the second primary coil11, flows through the fourth switch Q4in the reverse direction, and charges the clamp capacitor C2.

During periods of timings C to D and E to F shown inFIG. 5A, the positive second current I2is discharged from the smoothing capacitor C1, passes from the center tap25through the second primary coil22, and flows through the second switch Q2in the forward direction. The positive first current I1is discharged from the smoothing capacitor C1, passes from the center tap25through the first primary coil21, flows through the third switch Q3in the reverse direction, and charges the clamp capacitor C2.

During a period of timings D to E shown inFIG. 5B, the positive second current I2flows over the same path as that inFIG. 5Ain the same direction as that inFIG. 5A. The negative first current I1flows over the same path as that inFIG. 5Ain a direction that is opposite to that inFIG. 5A. That is, the negative first current I1is discharged from the clamp capacitor C2, flows through the third switch Q3in the forward direction, passes through the first primary coil21and the center tap25, and charges the smoothing capacitor C1. During this period, the second current I2that is generated as a result of discharge by the smoothing capacitor C1flows through the second primary coil22. In addition, the first current I1that is generated as a result of discharge by the clamp capacitor C2flows through the first primary coil21.

In the resonant inverter that uses a typical push-pull circuit that is configured by only the smoothing capacitor C1and the lower arm switches Q1and Q2, the current that is supplied to the first primary coil21and the second primary coil22is primarily taken from the smoothing capacitor C1. Therefore, an issue arises in that the burden placed on the smoothing capacitor C1is large, and ripple current tends to be large. In contrast, in the active-clamp push-pull circuit, during the periods of timings A to B and D to E, the current that is generated as a result of discharge by the smoothing capacitor C1and the current that is generated as a result of discharge by the clamp capacitor C2both flow through the primary coils21and22. Consequently, the burden of discharge by the smoothing capacitor C1can be reduced and ripple current can be reduced.

First Embodiment

The configuration of the switch controller40according to the present embodiment in the power conversion apparatus that uses the active-clamp push-pull circuit such as that described above will be described with reference toFIG. 1andFIG. 6toFIG. 10.FIG. 6shows the configuration of the duty ratio calculator50of the switch controller40in more detail than that inFIG. 1. The duty ratio calculator50according to the present embodiment includes a lower-arm duty ratio calculator51and an upper-arm duty ratio calculator53. Hereafter, the duty ratio of the first switch Q1and the second switch Q2that are the lower arm switches is referred to as a lower-arm duty ratio. In addition, the duty ratio of the third switch Q3and the fourth switch Q4that are the upper arm switches is referred to as an upper-arm duty ratio.

The lower-arm duty ratio calculator51calculates the lower-arm duty ratio using a map or a mathematical expression by feed-forward control based on the input voltage Vin detected by the input voltage detector15.FIG. 7shows a map that prescribes a relationship between the input voltage Vin and the duty ratio in feed-forward control. This map indicates a negative correlation in which the duty ratio decreases as the input voltage Vin increases, within a variation range of the input voltage Vin. Therefore, the duty ratio at a lower-limit value Vin_min of the input voltage within the variation range is maximum, and the duty ratio at an upper-limit value Vin_max of the input voltage is minimum. Here, a maximum value of the duty ratio is denoted by [α]. Here, descriptions of (dmax/2), [β], and the like are cited in descriptions hereafter.

Operations by which output is kept constant by duty ratio control will be described with reference toFIG. 8. A relatively low voltage within the variation range of the input voltage Vin is denoted by Vin_L, and a relatively high voltage is denoted by Vin_H. For example, as the voltage of the battery10, Vin_L is assumed to be about 10 V and Vin_H is assumed to be about 16 V.FIG. 8shows an operation in which the first switch Q1is turned on for a first half of the switching cycle Ts and the second switch Q2is turned on for a latter half of the switching cycle Ts. At the low voltage Vin_L that is indicated by a broken line, on-time Ton of the switches Q1and Q2is close to half (Ts/2) of the switching cycle and the duty ratio is close to 0.5. From this state, when the input voltage Vin increases to the high voltage Vin_H that is indicated by a solid line, control is performed such that the on-time Ton of the switches Q1and Q2is short, that is, the duty ratio is small. Consequently, even when the input voltage Vin changes, output is kept constant by the duty ratio control.

Returning toFIG. 6, the upper-arm duty ratio calculator53outputs a fixed value as the upper-arm duty ratio. That is, the feed-forward control of the duty ratio with reference toFIG. 7andFIG. 8is applied only to the lower-arm duty ratio according to the present embodiment and is not reflected in the setting of the upper-arm duty ratio. In this way, according to the present embodiment, the upper-arm duty ratio is outputted as a fixed value. Specific setting of the value of the upper-arm duty ratio and working effects of this configuration will be described hereafter.

The first embodiment and the second embodiment differ in terms of the configuration of the power controller60. The reference numbers of the power controller60according to the first embodiment and the second embodiment are respectively601and602. As shown inFIG. 9, the power controller601according to the first embodiment includes a switching frequency controller65and a PWM frequency generator66. The power controller601performs feedback control of electric power. The switching frequency controller65controls switching frequency such that the detected power P matches the target power Pref. The PWM frequency generator66generates a PWM frequency based on control results of the switching frequency controller65and outputs the PWM frequency to the PWM generator70.

As shown inFIG. 10, the switching frequency and the electric power have a chevron-shaped relationship in which the resonance frequency is the peak. When the input voltage Vin increases from Vin_L on the low voltage side to Vin_H on the high voltage side, the overall chevron-shaped curve shifts to the higher power side. For example, at the input voltage Vin_L, driving at a switching frequency f1at which the target power Pref can be acquired is performed. Subsequently, when the input voltage Vin increases to the input voltage Vin_H, the electric power P corresponding to the switching frequency f1exceeds the target power Pref. Here, the switching frequency controller65changes the switching frequency to f2such that the electric power P matches the target power Pref.

Next, the configuration and working effects of the switch controller40according to the present embodiment will be described mainly with reference toFIG. 11toFIG. 13, in comparison to the configuration and workings of the comparison example shown inFIG. 18toFIG. 20. The duty ratio calculator50inFIG. 1is referenced as the configuration of the duty ratio calculator of the comparison example.FIG. 18shows a driving method for the bridge circuit switches Q1to Q4in the comparison example. Ts is the switching cycle. Ton_L is the on-time of the lower arm switches Q1and Q2. Ton_U is the on-time of the upper arm switches Q3and Q4. A block arrow indicates that, when the input voltage Vin increases from a low state to a high state, the on-time of the switches Q1to Q4is shortened from a state indicated by broken lines to a state indicated by solid lines.

As a premise for driving that is shared between the comparison example and the present embodiment, the first switch Q1and the second switch Q2of the lower arm are equally operated in an alternating manner. The third switch Q3and the fourth switch Q4of the upper arm are equally operated in an alternating manner. In addition, for the workings of discharge support by the clamp capacitor C1shown inFIG. 4andFIG. 5to be achieved, at least the fourth switch Q4is required to be turned on during an on-period of the first switch Q1and the third switch Q3is required to be turned on during the on-period of the second switch Q2. Here, prohibiting the first switch Q1and the third switch Q3, and the second switch Q2and the fourth switch Q4that are upper- and lower-arm pairs from being simultaneously turned on to prevent a short circuit is common technical knowledge.

Furthermore, in the active-clamp push-pull circuit, when the simultaneous-on state of the first switch Q1and the second switch Q2, or the simultaneous-on state of the third switch Q3and the fourth switch Q4occurs, magnetic flux between the primary coils21and22of the transformer20is canceled. As a result, electric power to the secondary side is not outputted. A large current flows on the primary side. To prevent this situation, the switch controller40prohibits the first switch Q1and the second switch Q2from being simultaneously turned on, and the third switch Q3and the fourth switch Q4from being simultaneously turned on. Consequently, an abnormal current can be prevented from flowing to the primary side, and electric power can be appropriately outputted to the load C3on the secondary side.

Under the above-described premise, in the comparison example, the first switch Q1and the fourth switch Q4are simultaneously turned on/off, and the second switch Q2and the third switch Q3are simultaneously turned on/off. That is, the on-time Ton_U of the upper arm switches Q3and Q4is set to be equal to the on-time Ton_L of the lower arm switches Q1and Q2at all times, regardless of the input voltage Vin. Therefore, the upper-arm duty ratio is set to be equal to the lower-arm duty ratio at all times, regardless of the input voltage Vin.

As shown inFIG. 11, according to the present embodiment, the on-time Ton_L of the lower arm switches Q1and Q2is shortened in a manner similar to that in the comparison example, in accompaniment with the increase in the input voltage Vin. However, the on-time Ton_U of the upper arm switches Q3and Q4is fixed relative to the switching cycle Ts, regardless of the changes in the input voltage Vin. That is, the upper-arm duty ratio is set to a fixed value regardless of the changes in the input voltage Vin. The fixed value of the upper-arm duty ratio is set to be equal to or greater than a maximum value of the lower-arm duty ratio, and preferably set to a value obtained by a value equivalent to dead time DT being subtracted from 0.5.

The dead time DT is ensured between the on-period of the third switch Q3and the on-period of the fourth switch Q4. The dead time DT is a minimum amount of time required to prevent a simultaneous-on state. The dead time DT is determined based on element characteristics of the switches Q3and Q4, and manufacturing variations. In general, an amount of time within several % of the switching cycle Ts, such as within 5%, is expected. For example, when the value equivalent to the dead time DT is 5% of the switching cycle Ts, 0.45 that is obtained by 0.05 being subtracted from 0.5 is set as the upper-arm duty ratio.

In a map inFIG. 7, the value obtained by a dead-time equivalent value being subtracted from 0.5 is denoted by [β]. According to the present embodiment, the upper-arm duty ratio is indicated as being a fixed value that is equal to or greater than the maximum duty ratio [α] corresponding to the lower-limit value Vin_min of the input voltage and equal to or less than the value [β] obtained by the dead-time equivalent value being subtracted from 0.5. Preferably, the upper-arm duty ratio is set to a value that is equal to [β] that is the maximum value within this range.

In this manner, the upper-arm duty ratio for realistic driving according to the present embodiment is the value that is obtained by the dead-time equivalent value being subtracted from 0.5. However, when the dead time is idealistically considered to be zero, the driving method according to the present embodiment can be said to be a method in which driving is performed at a full duty ratio in which either of the upper switches Q3and Q4is turned on at all times. Meanwhile, the driving method of the comparison example is a method in which the upper-arm duty ratio tracks the lower-arm duty ratio.

Regarding differences in drive waveform due to such differences in the driving method, refer toFIG. 19of the comparison example andFIG. 12according to the present embodiment.FIG. 19andFIG. 12show the switch current, the transformer primary-side current, the transformer-applied voltage, and the gate commands for the bridge circuit switches Q1to Q4. Regarding the switch current, the current that flows to any of the switches Q1to Q4is shown in a single drawing. The switch through which the current flows is switched at a timing indicated by a vertical line. Regarding the transformer primary-side current, the switch current during periods excluding the switching timing continuously flows.

In the driving in the comparison example inFIG. 19, during a period of time t10to t11that corresponds to about one-fourth of the switching cycle Ts, the first switch Q1and the fourth switch Q4are turned on. During a period of time t12to t13, the second switch Q2and the third switch Q3are turned on. During a period of time t11to t12and a period of time t13to t14, all switches Q1to Q4are turned off. As a result of driving such as this, immediately before transition from the state in which all switches Q1to Q4are turned off to the state in which the first and fourth switches Q1and Q4or the second and third switches Q2and Q3are turned on, a discontinuous mode in which the current changes in a discontinuous manner occurs. During the discontinuous-mode period, a polarity of the transformer-applied voltage is inverted and the waveform is such that a positive-voltage pulse and a negative-voltage pulse are each included twice during the switching cycle Ts.

In contrast, in the driving according to the present embodiment shown inFIG. 12, the fourth switch Q4is turned on during a period of time t0to t2, beyond a period of time t0to t1during which the first switch Q1is turned on. The third switch Q3is turned on during a period of time t2to t4, beyond a period of time t2to t3during which the second switch Q2is turned on. As a result of the full duty ratio driving of the upper arm switches Q3and Q4such as this, the occurrence of a discontinuous mode of electrical current is prevented. In addition, the transformer-applied voltage forms a one-pulse waveform in which the positive-voltage pulse and the negative-voltage pulse are each included once during the switching cycle Ts. The one-pulse waveform means that pulses that are as commanded are applied to the transformer.

In addition, regarding effects of frequency dependency in the duty ratio calculator50, refer toFIG. 20AandFIG. 20Bof the comparison example, andFIG. 13AandFIG. 13according to the present embodiment.FIG. 20AandFIG. 13Ashow maps of the duty ratio that is required for constant electric power to be outputted when the input voltage Vin changes for each switching frequency, determined through simulation.FIG. 20BandFIG. 13Bshow map converted to a relationship of switching frequency to duty ratio for each level of the input voltage Vin, that is, low, medium, and high. Here, the duty ratio on a vertical axis indicates a value that is used for both the lower-arm duty ratio and the upper-arm duty ratio in the comparison example. According to the present embodiment, the duty ratio on the vertical axis indicates the lower-arm duty ratio.

In the comparison example, at the same input voltage Vin, the duty ratio tends to increase as the frequency decreases, and the duty ratio tends to decrease as the frequency increases. That is, the duty ratio is dependent on the switching frequency. Therefore, in the comparison example, a plurality of maps based on the frequencies are required for feed-forward control of the duty ratio. In addition, when the load C3is a discharge reactor or the like, characteristics may change due to variations in load capacitance or the like and temperature change, and the resonance frequency may change. In this case as well, effects caused by changes in the characteristics of the load are received during feed-forward control of the duty ratio.

In contrast, inFIG. 13Aaccording to the present embodiment, maps for a plurality of switching frequencies overlap on a single line. Here, inFIG. 13A, the plurality of lines are intentionally slightly shifted to indicate that a plurality of maps are overlapped. In addition, as shown inFIG. 13B, the duty ratio is not dependent on the switching frequency at each input voltage Vin. Because frequency dependency is not present, even if the characteristics of the load C3changes and the resonance frequency changes, the effects thereof are not received. Therefore, according to the present embodiment, acquisition and adaptation of maps based on the characteristics of the load C3and the switching frequency are unnecessary. Feed-forward control of the duty ratio can be performed with a single map.

Here, the map inFIG. 13Ais acquired through simulation. However, the lower-arm duty ratio calculator51may calculate the lower-arm duty ratio using a mathematical expression in which polynomial approximation of a waveform acquired through simulation or experiment is performed. As a result, an optimal duty ratio can be calculated in advance. In addition, the lower-arm duty ratio calculator51can also derive the duty ratio that is equivalent to the map inFIG. 13Aby a theoretical formula. Next, derivation of the theoretical formula for calculating the duty ratio based on the input voltage Vin will be described.

First, symbols in the expressions will be defined in the following manner.

Vin: input voltage

Vin_min: lower-limit value within the variation range of the input voltage

x: required attenuation amount

Vt: transformer-applied voltage applied to the transformer20

Vt_min: minimum value of the transformer-applied voltage

Vc: clamp capacitor voltage that is an inter-electrode voltage of clamp capacitor C2

Vc_min: minimum value of the clamp capacitor voltage

d: two-fold value of the duty ratio (d=2×duty ratio)

dmax: maximum value of d within the variation range of the input voltage

k: base effective value ratio

As indicated on a vertical axis inFIG. 7, the maximum value of the duty ratio at the input voltage lower-limit value Vin_L is indicated as (dmax/2). The required attenuation amount x is a required amount of attenuation of the duty ratio (=d/2) that is required in accompaniment with the increase in the input value Vin from the input voltage lower-limit value Vin_L. In addition, according to the present embodiment, the d value is defined as shown inFIG. 14based on the transformer-applied voltage Vt forming the one-pulse waveform. That is, when the on-time of the switch is half (2/Ts) of the switching cycle, d=1. In actuality, the d value is set within a range of 0≤d<1.

In the active-clamp circuit, because the transformer-applied voltage Vt is the clamp capacitor voltage Vc, the required attenuation amount x is expressed by expression (1).

In addition, because the primary circuit is a step-up converter, a relationship between the input voltage Vin and the clamp capacitor voltage Vc is expressed by expressions (2) and (3).

When expressions (2) and (3) are substituted into expression (1), the required attenuation amount x is expressed by the input voltage Vin and the d value. When the required attenuation amount x is substituted into expression (4), the d value is calculated. The duty ratio is determined from the d value. As a result, the optimal duty ratio can be calculated from the theoretical expression.

Derivation of expression (4) will be additionally described. When Fourier expansion is performed on the one-pulse waveform inFIG. 14and a fundamental wave component is extracted, expressions (5.1) to (5.3), below, are derived. When expressions (5.1) to (5.3) are arranged, expression (4) is obtained.

Effects According to the Present Embodiment

(1) In the conventional technology in JP-A-2001-251854, the duty ratio is controlled based on a current value that is fed back. Therefore, as a result of response delay that occurs when the input voltage suddenly changes, output variations and overcurrent may occur. In contrast, the lower-arm duty ratio calculator51according to the present embodiment calculates the duty ratio by feed-forward control based on the input voltage Vin. Consequently, output variations and overcurrent caused by sudden changes in the input voltage can be suppressed.

(2) In the resonant inverter that uses the push-pull circuit, the current that is supplied to the first primary coil21and the second primary coil22are primarily taken from the smoothing capacitor C1. Therefore, an issue arises in that the burden placed on the smoothing capacitor C1is large and ripple current tends to be large. According to the present embodiment, through use of the active-clamp push-pull circuit that includes the clamp capacitor C2and the upper arm switches Q3and Q4, the clamp capacitor C2supports discharge by the smoothing capacitor C1. Consequently, the burden placed on the smoothing capacitor C1is reduced. Ripple current can be reduced.

(3) In the driving method of the active-clamp push-pull circuit, when the duty ratio of the upper arm switches Q3and Q4are changed so as to track the duty ratio of the lower arm switches Q1and Q2, a problem arises in that discontinuous mode of the output current occurs. The upper-arm duty ratio calculator53according to the present embodiment outputs a fixed value that is equal to or greater than the maximum value of the lower-arm duty ratio within the variation range of the input voltage Vin as the upper-arm duty ratio. Preferably the upper-arm duty ratio is set to the maximum value from which the value equivalent to the dead time DT has been subtracted. As a result, the transformer-applied voltage forms a one-pulse waveform. The occurrence of discontinuous mode of electrical current can be prevented as much as possible.

(4) The power controller601according to the present embodiment can control electric power to be constant by controlling the switching frequency by feedback control of electric power. In addition, as a result of the transformer-applied voltage forming a one-pulse waveform in this configuration, frequency dependency in feed-forward control of the duty ratio can be eliminated. Consequently, acquisition and adaptation of maps based on the switching frequencies become unnecessary.

Second Embodiment

The power conversion apparatus according to the second embodiment will be described with reference toFIG. 15toFIG. 17B. The second embodiment differs from the first embodiment in terms of the configuration of power feedback control. As shown inFIG. 15, the power controller602according to the second embodiment further includes a burst duty ratio controller67and a burst duty ratio generator68, in addition to the switching frequency controller65and the PWM frequency generator66that are included in the power controller601according to the first embodiment.

The switching frequency controller65and the PWM frequency generator66performs feedback control of the switching frequency such that a detection value of instantaneous power Pinst matches target instantaneous power Prefinst, and outputs the switching frequency to the PWM generator70. The burst duty ratio controller67and the burst duty ratio generator68performs feedback control of a burst duty ratio such that a detection value of average power Pavr matches target average power Prefavr, and outputs the burst duty ratio to the PWM generator70. The PWM generator70generates the PWM signal that intermittently drives the bridge circuit switches Q1to Q4based on the switching frequency and the burst duty ratio generated by the power controller602.

As shown inFIG. 16, the bridge circuit switches Q1to Q4are intermittently driven at a predetermined burst cycle TB that includes a drive period TDRIVE and a stop period TSTOP. That is, the bridge circuit switches Q1to Q4are operated to on/off based on the commanded duty ratio and switching frequency during the drive period TDRIVE. The bridge circuit switches Q1to Q4are all turned off during the stop period TSTOP. The burst duty ratio is a ratio of the drive period TDRIVE relative to the burst cycle TB. For example, when the drive period TDRIVE and the stop period TSTOP are equal, the burst duty ratio is 0.5. Here, burst frequency (Hz) is an inverse of the burst cycle (s). In addition, electric power at output-pulse-on during the drive period TDRIVE is the instantaneous power Pinst. A value obtained by the instantaneous power Pinst being multiplied by the burst duty ratio is the average power Pavr.

All-layer discharge power shown inFIG. 17Arefers to instantaneous power that is required to be provided for discharge to be performed at all layers of the discharge reactor C3, that is, between all electrodes31and32. A lower-limit value of the all-layer discharge power is set as the target instantaneous power Prefinst. The power controller602first controls the instantaneous power Pinst by changing the switching frequency from f4to f3such that the target instantaneous power Prefinst can be obtained on a switching frequency and power characteristics line. As a result, discharge at all layers of the discharge reactor C3can be actualized.

Next, the power controller602generates the burst duty ratio that corresponds to the target average power Prefavr based on a relationship between a burst duty ratio and power characteristics line as shown inFIG. 17B. In other words, proportion of the target average power Prefavr relative to the target instantaneous power Prefinst is generated as the burst duty ratio. In this manner, according to the second embodiment, as a result of both the switching frequency and the burst duty ratio being controlled, the average power Pavr can be controlled while efficient discharge at all layers is maintained. In addition, in a manner similar to that according to the first embodiment, the occurrence of discontinuous mode of the output current can be prevented. Frequency dependency in feed-forward control of the lower-arm duty ratio can be reduced.

Other Embodiments

(a) The power controller602according to the second embodiment controls the average power Pavr by controlling the burst duty ratio upon controlling the instantaneous power Pinst by controlling the switching frequency. In this regard, an aspect in which the switching frequency is, for example, fixed near the resonance frequency of the output current and feedback control of only the average power Pavr is performed through control of the burst duty ratio can be considered. The power controller according to this aspect is merely required to be configured to include only the burst duty ratio controller67and the burst duty ratio generator68. According to this aspect, frequency dependency in feed-forward control of the duty ratio does not become an issue in the first place. Effects similar to those according to the above-described embodiment are achieved regarding prevention of the occurrence of a discontinuous mode of electrical current.

(b) The capacitive load C3that is connected to the secondary coil26of the transformer20is not limited to the discharge reactor that is used in the ozone generation apparatus30and may be another load. In addition, an inductor or the like for adjusting the resonance frequency of the load C3may be connected to the secondary circuit.

(c) For example, when strict control of the output power is not required due to characteristics of the load C3, the switch controller may not perform feedback control of the electric power P. The effects (1) to (3) according to the above-described embodiments can be achieved by at least only the feed-forward control of the duty ratio based on the input voltage Vin being performed.

The present disclosure is not limited in any way to the above-described embodiments. Various aspects are possible without departing from the spirit of the present disclosure.

The present disclosure is described based on the embodiments. However, the present disclosure is not limited to the embodiments and constructions. The present disclosure is intended to cover various modification examples and modifications within the range of equivalency. In addition, various combinations and configurations, and further, other combinations and configurations including more, less, or only a single element thereof are also within the spirit and scope of the present disclosure.