TV signal transmission systems and methods

A television system includes a plurality of complementary transmitter and receiver stages for processing an HDTV signal to minimize co-channel interference to and from NTSC signals, thereby facilitating the use of so-called "taboo" channels for the transmission of HDTV signals. The HDTV signals are encoded for transmission in a format exhibit dominant periodicities equivalent to that of NTSC to further minimize such co-channel interference.

BACKGROUND OF THE INVENTION AND PRIOR ART 
This invention relates generally to television signal transmission systems 
and methods and specifically concerns a novel spectrum compatible High 
Definition Television (HDTV) signal encoding and transmission system. 
Recently, extensive interest has been generated in developing a so-called 
High Definition Television system having higher picture definition than 
existing television systems and improved audio. Such a system should also 
desirably exhibit improved noise performance and may have a wider aspect 
ratio. A difficulty is in the available spectrum space for terrestial 
broadcasting of television signals. It is desirable that HDTV television 
receivers be capable of receiving appropriate video and sound signals 
while the existing population of NTSC receivers continue to receive their 
accustomed NTSC performance without perceptible impairment. 
The proposed HDTV systems may generally be categorized into three groups; 
compatible systems, augmentation systems and non-compatible systems. 
Compatible systems add to or modify an existing NTSC transmission in a 
manner so that it may still be viewed on an existing receiver with NTSC 
resolution, but may be received with higher resolution on a special HDTV 
receiver. The compatible transmission is normally limited to the same 6 
MHz channel used for NTSC transmissions. Augmentation systems, on the 
other hand, transmit augmenting signals in additional spectrum space 
without disturbing the NTSC transmission. Thus, an existing receiver may 
reproduce the televised image with NTSC resolution by tuning the standard 
channel, while a special HDTV receiver may be used to tune both the 
standard channel and the augmenting signals to reproduce an HDTV image. 
Transmissions comprising various combinations of these two techniques are 
also possible. 
It is believed that a non-compatible system will provide the best solution 
to the problem of providing high quality HDTV. Compatible systems have the 
disadvantage of almost inevitably producing undesired artifacts in 
standard NTSC receivers. Augmentation systems have the disadvantage that 
multiple RF bands must be tuned and that different signals must be 
precisely pieced together to form the HDTV image. Non-compatible systems 
provide HDTV transmissions which cannot be received by standard NTSC 
receivers. Since such transmissions must be made over unused spectrum, 
care must be exercised to avoid interference with existing NTSC channels. 
Co-channel and adjacent channel interference are of particular concern in 
this regard. 
With respect to spectrum availability, it is well known that many of the 
designated television signal channels are not used in a particular 
geographical area. This is due to the impossibility of adding new 
transmitted signals in the unassigned positions of the spectrum because of 
a large number of FCC mandated prohibitions (particularly in the UHF 
band), colloquially referred to as "taboos." In most metropolitan areas, 
spectrum utilization is restricted to every second VHF channel and every 
sixth UHF channel. It will be seen that with the transmission system of 
the invention, the transmission format is changed (with corresponding 
changes in receiver requirements) to enable more efficient utilization of 
the existing VHF/UHF spectrum. With the invention, every existing NTSC 
broadcast station will be capable of obtaining a second 6 MHz channel over 
which an HDTV signal can be broadcast simultaneously with the existing 
NTSC program. Thus with the system of the invention, a spectrum compatible 
system with existing NTSC signals is obtained. 
The preferred embodiment of the invention provides special benefits with 
regard to adjacent channel and co-channel interference problems with NTSC 
and other type television signal transmission and receiving systems. The 
improved transmission system permits operation in or adjacent to NTSC 
television signal transmission areas without objectionable adjacent 
channel and co-channel interference, either into or from the NTSC channel. 
An NTSC television signal occupies a 6 MHz bandwidth and imposes 
significant transmission power demands. These power demands are directly 
related to the cost of operating the signal transmitter and reductions 
therein can yield significant economic benefit. Also of great importance 
is the fact that cable television plants, especially those of older 
vintage, are restricted in the number of channels they can handle by the 
signal power handling capabilities of their amplifiers. It would be highly 
desirable to reduce the amount of signal power required to transmit 
television signals, thereby reducing transmitter operating costs and 
permitting a larger number of television channel signals to be handled by 
a cable plant of given power handling capability. The system of the 
invention achieves a marked reduction in the power required to transmit an 
AM television signal without discernible degradation of signal fidelity 
and therefore provides a solution to these needs of the prior art. 
In its FM implementation, the system of the invention enables transmission 
with significantly less bandwidth since the deviation of the FM signal is 
minimized, which will find ready application in Direct Broadcast Satellite 
(DBS) transmission systems. The smaller bandwidth directly improves the 
signal to noise performance of the system, which improvement may be 
translated into smaller receiving antennas. Thus, the system of the 
invention will be seen to solve a longstanding problem in DBS transmission 
systems. 
The inventive transmission system has a number of important aspects. In 
accordance with a fundamental aspect thereof, a television signal is 
configured such that the transmitted signal is a "hybrid," that is; it has 
a coded (digital) portion representing signal components of low picture 
detail and relatively high transmission power demand that may be 
transmitted in a relatively low power utilization format; and an analog 
portion, representing signal components of high picture detail and 
relatively low transmission power demand. The demarcation between the 
analog and digital portions is a function in part of the availability of 
means to transmit the digital data. In accordance with the invention, the 
data is sent in non-active video portions of the transmitted signal. 
The hybrid system is subdivided into a "basic hybrid," in which the removed 
and digitized low frequencies are under about 15 KHz and an "extended 
hybrid" in which the removed and digitized signal frequencies are under 
about 200 KHz. As will be explained in detail below, in the basic hybrid 
form of the invention, the digital part comprises video components below 
the line deflection frequency that are digitally coded and transmitted as 
data during non-active video portions of the television signal. In the 
extended hybrid form of the invention, the digital part comprises video 
components below about 200 KHz that are digitally coded and transmitted as 
data during non-active video portions of the television signal. Since the 
invention may be used with many different television signal formats, the 
non-active video portions of the signal may include either or both the 
horizontal and vertical blanking intervals. 
It has also been found that further benefits are obtained by sequentially 
applying basic hybrid processing and extended hybrid processing (referred 
to as two step processing), with the basic hybrid processing being 
performed for the active video of each horizontal line such that the low 
frequency average of each horizontal line is removed from the analog 
signal. The remaining components below 200 KHz are subsequently removed. 
A further aspect of the inventive system involves "temporal pre-emphasis," 
also referred to as temporal filtering, field processing or frame combing. 
With this approach, transmission power for stationary images is reduced 
while transmission power for moving images is increased. Since the average 
television picture is, relatively speaking, static, the use of temporal 
pre-emphasis is of benefit because the greater interference potential of 
signals corresponding to moving images is outweighed by the fact that 
noise in a moving image is much less noticeable than noise in a stationary 
image. Temporal de-emphasis is applied in the receiver. This aspect of the 
invention has advantages in any "video" transmission system independent of 
the hybrid processing of the signal. This is due to the fact that normally 
there is little change between successive frames of video and emphasizing 
the changes relative to the static portions results in very efficient 
transmission. 
Yet another aspect of the inventive system involves compressing the 
"hybrid" video signal to achieve a large signal to noise ratio for broad, 
flat video areas, where noise is readily discernible, and a low signal to 
noise ratio for narrow video components, representing edges and video 
detail, in which noise is much less discernible. In the receiver, the 
signal is expanded to undo the compression in the transmitter. The 
combination of compression and expansion is referred to as "companding." 
Still another important feature of the inventive system is the use of 
dispersal filtering to reduce the amplitudes of the peak video components 
by distributing their energy among the voids created in the hybrid video 
signal. As will be seen, these voids are the direct result of hybrid 
processing of the video signal whereby low frequency analog components are 
removed, coded and included as data in the non-active video portions of 
the remaining analog high frequency components. 
As those skilled in the art will readily perceive, reduction of the average 
power of the transmitted signal is highly desirable, especially where 
adjacent channel and co-channel interferences are concerned. The reduction 
occurs because of the hybrid processing of the video signal which 
effectively replaces low frequency video signals with "doublets" that 
define the edges of the video image. With temporal filtering, the largest 
signals result from moving video edges which can be compressed even more. 
Companding increases the signal to noise performance for relatively 
stationary edges of video images at the expense of much less observable 
noise associated with moving video image edges. Dispersal filtering 
primarily reduces the amplitudes of the signals above the hybrid 
processing frequency range. 
Other advantages flow from application of the various aspects of the 
inventive transmission system that improve signal to noise performance, 
especially with respect to the ability to operate in an environment of 
adjacent and co-channel NTSC signals. These include; the technique of 
precise carrier frequency offset with respect to co-channel NTSC signals 
to cause "break up" of interfering signals and thereby reduce their 
visibility in the video display; and co-location (locating the hybrid 
signal transmitter of the invention close to the adjacent channel NTSC 
transmitter) to assure that receivers in both reception areas receive 
approximately equal strength signals to enable their respective AGC 
systems to set up properly. Also frame locking the hybrid signal to the 
NTSC signal and incorporating all data in the vertical blanking interval 
of the hybrid signal contributes to the ability to operate in a co-channel 
environment. It will, of course, be clear that the many aspects of the 
inventive system may have benefits that are independent of other aspects 
of the system, and that the use of one or more of the aspects in 
combination with each other produces even greater benefits. 
The encoding format used in the HDTV system of the present invention has 
substantially improved resolution relative to the current NTSC system. 
While being non-compatible with NTSC, it affords a viable solution to the 
search for a high quality HDTV system that will not obsolete the existing 
population of NTSC receivers. The encoding format further minimizes 
adjacent channel and co-channel interference problems by exhibiting 
dominant timing periodicities equal to those characteristic of an 
indigenous television system, such as NTSC in the U.S. 
While the techniques of the invention may be used with non-compatible, as 
well as augmentation-type HDTV systems to allow maximum utilization of 
existing spectrum, they are of particular benefit when used with Zenith 
Electronics Corporation's Spectrum Compatible High Definition Television 
System presently under consideration. 
OBJECTS OF THE INVENTION 
A principal object of the invention is to provide novel television signal 
transmission systems and methods. 
An additional object of the invention is to provide television signal 
transmission systems of improved noise performance. 
A further object of the invention is to provide a novel AM television 
signal transmission system that requires substantially less transmitting 
power. 
A still further object of the invention is to provide a novel FM television 
signal transmission system that requires substantially less bandwidth. 
Another object of the invention is to provide a television transmission 
system that minimizes adjacent channel and co-channel interference. 
Still another object of the invention is to provide an optimal television 
signal transmission system with better noise performance. 
A basic object of the present invention to provide an improved HDTV system. 
It is a further basic object of the invention to provide an improved HDTV 
system of the non-compatible type. 
It is another basic object of the invention to provide an HDTV System in 
which the HDTV transmissions are made over a standard 6 MHz television 
channel in a manner so as to minimize interference with existing NTSC 
transmissions. 
It is yet another basic object of the invention to provide an improved HDTV 
system in which an image is transmitted and reproduced in a manner 
matching the visual performance of the human eye.

DESCRIPTION OF THE PREFERRED EMBODIMENTS 
In the diagram of FIG. 1, a spectrum compatible HDTV system is illustrated 
using an NTSC taboo channel. The full HDTV source is a 30 MHz video signal 
that is encoded to conform to a 6 MHz channel (the HDTV format aspect of 
the invention) and transmitted over an optimal transmission system (the 
transmission aspect of the invention) in a 6 MHz NTSC taboo channel. An 
HDTV receiver (including a receiver, a video decoder and a display) is 
also shown as well as a conventional NTSC transmitter and receiver. The 
ability to transmit a wide-band HDTV signal over a 6 MHz NTSC taboo 
channel results from using the novel HDTV format with the novel 
transmitting system, each of which is separately described and claimed in 
the above-mentioned respective copending applications. 
Transmission System 
The primary function of the transmission system of the invention is to 
provide for the transmission of a wide-band HDTV signal over a 6 MHz NTSC 
taboo channel without causing adjacent or co-channel interference with 
existing NTSC (or other indigenous) transmissions. According to one aspect 
of the invention, a hybrid transmission system is provided to facilitate 
these results. 
The philosophy of the hybrid processing of the invention is to reduce the 
power required to transmit a television signal by extracting low frequency 
video analog components therefrom (that have a high transmission power 
requirement) and transmitting such components in a coded, 
low-power-consumption form along with the remaining high frequency 
components that are transmitted in a conventional manner. As will be 
discussed, this results in a substantial reduction in transmitter power 
reducing the likelihood of interference into an existing NTSC channel. The 
power reduction may also be translated into better low frequency noise 
performance since the transmission power demand is greatest for the low 
frequency components. In FM transmissions, the benefit is in the form of 
reduced bandwidth which also improves noise performance. Hence the hybrid 
television signal transmission systems of the invention have improved 
noise performance over prior art television systems. 
More specifically, in the basic hybrid form of the invention, the baseband 
composite video signal is subjected to "line integration" at the 
transmitter to determine a line averaged value of the active video for 
each horizontal line. In an analog version of the transmitter, the line 
averaged value is passed through an analog-to-digital (A/D) converter 
where it is converted to a digital value which is coded and transmitted 
with the high frequency components of the video signal. These high 
frequency components are obtained by subtracting an analog signal that 
corresponds to the line averaged digital value of the active video portion 
of each horizontal line. To assure that the subtracted analog signal 
corresponds to the appropriate portions of the baseband video signal, the 
baseband composite video signal is subjected to a one horizontal line 
delay. The output of the line integrator may be subtracted directly from 
the baseband video signal to obtain the high frequency components. This 
approach could introduce error since the coded representation of that 
signal, which is used in the receiver to reconstitute the low frequency 
components, may have resolution limitations. Preferably, digital values, 
representing the line averaged video signals, are supplied to a 
digital-to-analog (D/A) converter for developing the analog signals (low 
frequency components), which are subtracted from the baseband video 
signal. This eliminates error due to resolution limitations. Accurate 
reconstruction of the low frequency portions can now be accomplished in 
the receiver because each coded representation truly represents the 
subtracted low frequency portions for that video line. In a digital 
version of the transmitter, the analog video signal is converted to a 
digital signal and a digital average of the active video portion of each 
horizontal line is obtained. 
As will be seen with reference to the extended hybrid form of the 
invention, frequencies below about 200 KHz are removed and sent as coded 
data for even greater power reduction benefits and to yield further 
co-channel interference benefits from temporal pre-emphasis signal 
compression and dispersal. 
Referring in greater detail to the drawings, in FIG. 1A, a source 12 of 
baseband composite video signal supplies a video clamp circuit 14 in 
accordance with conventional techniques for establishing a base line 
reference, generally at blank level, i.e., corresponding to the level of 
the sync signal back porch. The output of video clamp 14 is supplied to a 
one horizontal line (1H) delay circuit 16, to a line integrator 24 and to 
a conventional sync separator circuit 28. The output of sync separator 
circuit 28 supplies sync pulses to a timing and control circuit 30. A data 
source 38 supplies information, in the form of data to be included in the 
transmitted television signal, to timing and control circuit 30. The 
delayed baseband composite video signal output from delay circuit 16 is 
passed through a switch 18 that is operated by timing and control circuit 
30. The output of switch 18 is supplied to a summing network 20 which, in 
turn, supplies a multiplier circuit 22. Line integrator 24 is also coupled 
to, and operated under the control of, timing and control circuit 30 for 
integrating only the active video portion of each horizontal line of the 
baseband composite video signal. Its output is supplied to an A/D 
converter 26 which is coupled over a communication bus 27 to timing and 
control circuit 30. Communication bus 27 is also coupled to a 
digital-to-analog (D/A) converter 32. A control line 29 links A/D 
converter 26 and timing and control circuit 30. A ROM 31 is coupled 
between timing and control circuit 30 and D/A converter 32. ROM 31 
supplies certain fixed reference and identification signals to D/A 
converter 32 as will be explained. 
The output of multiplier 22 is coupled to a low pass filter (LPF) 23 to 
conform the data pulses to channel bandwidth limitations. LPF 23 feeds a 
modulator 34 which, along with multiplier 22, is under control of timing 
and control circuit 30. Modulator 34 is also supplied with an RF carrier 
and, in turn, supplies a summing network 36 that combines an audio signal 
from a source of modulated sound 40 with the modulated video signal of the 
invention for transmission to suitable receivers. Modulator 34 may either 
be an AM or FM modulator, with the FIG. 1A implementation being for the AM 
version. For an FM version, sound source 40 need only be interposed 
between LPF 23 and modulator 34. 
In operation, timing and control circuit 30, under control of the sync 
signals from sync separator 28, sends appropriate timing signals to video 
clamp 14, line integrator 24, switch 18, multiplier 22, modulator 34, A/D 
converter 26, D/A converter 32 and ROM 31. The video clamp 14 maintains 
the sync signal back porch of the composite baseband video signal at a 
predetermined level. The line integrator 24 is operated to independently 
integrate only the active video signal portion of each horizontal line. 
Switch 18 is operated by timing and control circuit 30 to pass active 
video and color burst, but not horizontal sync, if present, to summing 
circuit 20. The line averaged value of video developed by line integrator 
24 for a particular video line is digitized by A/D converter 26 and 
coupled to both timing and control circuit 30 and D/A converter 32. D/A 32 
converts the output of A/D 26 to a corresponding analog signal which is 
subtracted from active video in summing network 20. During the horizontal 
sync signal portion of the composite video signal, ROM 31, in response to 
timing and control circuit 30, couples a digital pedestal signal to D/A 
32, which is converted to a corresponding analog pedestal signal and 
inserted into the signal developed in summing network 20. It will be 
appreciated that some video signal formats may not have horizontal syncs 
or blanking intervals between successive horizontal lines. 
Timing and control circuit 30 develops a data signal comprising positive 
and negative voltage data pulses representing the digitized line averaged 
signals from A/D 26 and applies these data pulses, during the horizontal 
blanking intervals, to multiplier 22. Multiplier 22 multiplies these data 
pulses with the analog pedestal signal previously inserted in the 
horizontal blanking interval to develop positive and negative data pulses 
during the horizontal blanking interval of the signal. As will be 
discussed below, the resultant coded representations of the line averaged 
signals are used to reconstitute the signal in the receiver. 
While the number of data pulses that may be inserted in the horizontal 
blanking interval is dependent upon the data frequency, the inventive 
system envisions that other coded data may be included in the horizontal 
blanking interval if desired, specifically data from data source 38. While 
the coded representation of the digital output of A/D converter 26 for 
each video line preferably comprises three or four bits (3/4), more bits 
can be used depending upon the resolution desired. To assure accuracy of 
the reconstituted signal in the receiver, this line varying 3/4 bit 
digital signal, which represents high (transmission) energy, low frequency 
components of the video signal, is passed through D/A converter 32 to form 
an analog signal that is subtracted from the composite baseband video 
signal. Due to the resolution limitations of the 3/4 bit signal, a small 
residue of low frequency components may remain in the analog video signal 
that is passed by summing network 20 to multiplier 22. However, since the 
low frequency video component added to the high frequency video component 
in the receiver is derived from the same 3/4 bit signal, it will precisely 
match that which was subtracted at the transmitter. 
In FIG. 2, the series of idealized, not-to-scale waveforms labelled A, B, 
C, D and E correspond to those appearing at correspondingly labelled 
portions of FIG. 1A. Waveform A represents the baseband composite video 
signal with negative-going 15.75 KHz horizontal sync pulses 68, a 3.58 MHz 
color burst 70 and a horizontal blanking interval 72. Waveform B is 
indicated as a dashed horizontal straight line and represents the output 
of line integrator 24 which corresponds to the average level of the active 
video signal between successive horizontal blanking intervals 72. Waveform 
C represents the result of the subtraction of the line averaged video from 
the composite video signal and is centered at zero volts. It also includes 
the data pedestal added in summing network 20. Waveform D illustrates the 
result of multiplier 22 multiplying data with the data pedestal to develop 
positive and negative data pulses during the horizontal blanking period. 
While only two such pulses are shown for simplicity, a greater number of 
pulses is contemplated. Modulator 34 modulates an RF television frequency 
carrier with the bandwidth limited video signal, including the data 
pulses, for transmission as illustrated by waveform E. The signal is 
centered about zero carrier and reverses phase each time the envelope 
passes through the zero carrier level. Thus, for example, each half cycle 
of the color burst, as well as each data pulse, reverses phase. Portions 
74 and 76 of the waveform represent RF carrier phase reversals in the 
video signal. 
Referring to FIG. 3, an idealized representative spectrum of power 
distribution of an NTSC type television signal indicates the very 
significant power demand on the transmitter near the carrier frequency. 
FIG. 4 shows a very greatly expanded portion of the curve of FIG. 3 near 
the video carrier frequency. In particular the portion of the spectrum 
between the video carrier frequency and 15 KHz is illustrated. The shaded 
area bounded by the waveform and the dashed line illustrates pictorially 
the power saved with the basic hybrid form of the invention because of the 
subtraction of the low frequency signal components (below 15 KHz) from the 
transmitted signal. These low frequency components are instead transmitted 
in the form of data in a coded low-energy utilization format as explained 
above. This shaded area is estimated to represent approximately 99% of the 
power of a typical television signal. A reduction in transmitted power of 
20dB (100:1) is thus well within that contemplated by the invention. Some 
of this power reduction may, of course, be sacrificed as a tradeoff to 
improve the signal-to-noise ratio of the transmitted signal. It should be 
borne in mind that since the power scales of the curves are logarithmic, 
they do not graphically convey the true magnitude of power reduction 
obtained with the invention. 
A shaded area between 0 and 200 KHz on FIG. 3 will be discussed later in 
connection with the extended hybrid form of the invention in which analog 
components up to about 200 KHz are removed. While the additional 
transmission power savings from removing components below 200 KHz is not 
nearly of the magnitude as with the basic hybrid, the benefits obtained in 
compressing and dispersal of the peak video signals produced are very 
significant. It should also be pointed out that the coded data need not be 
put in the horizontal intervals of the transmitted signal, but may 
advantageously be transmitted in the vertical blanking intervals of the 
transmitted television signal. Indeed, as fully disclosed in the copending 
application above, the transmitted video signal need not have horizontal 
blanking intervals, or even horizontal sync. In general, the data may be 
placed in any non-active video portion of the television signal for 
optimum noise performance. 
In the receiver of FIG. 5, the basic hybrid form of transmitted signal is 
received by a tuner 41 and supplied to a buffer amplifier 42. The output 
of amplifier 42 supplies a sound carrier bandpass filter 44 and a video 
carrier bandpass filter 46. The receiver operates at RF frequencies, 
although operation at IF and baseband frequencies are also contemplated. 
The output of sound bandpass filter 44 is supplied to one input of a 
summing network 66. The output of video bandpass filter 46 is supplied to 
a buffer amplifier 48. Buffer amplifier 48 feeds a multiplier 50 and a 
biphase stable phase locked loop (BPLL) circuit 52. The output of 
multiplier 50 is supplied through a switch 54 to a summing network 58. The 
data output of BPLL 52 is supplied to a controller 56 which, in turn, 
controls operation of multiplier 50 and switch 54. Controller 56 also 
supplies data, including the coded representation of the line averaged 
video, to a D/A converter 60. In the preferred embodiment, BPLL 52 is 
biphase stable and operates to provide recovered data, including the coded 
representation of line averaged video, to controller 56 and a fixed 
amplitude carrier Fo, that is either in phase or 180.degree. out of phase 
with the received signal, to a multiplier 62. BPLL 52 may advantageously 
be constructed in accordance with the teachings of Ser. No. 025,240, filed 
Mar. 12, 1987, entitled COMBINED FPLL AND PSK DATA DETECTOR, in the names 
of R. Citta and G. Sgrignoli, and assigned to Zenith Electronics 
Corporation. That application cites U.S. Pat. Nos. 4,072,909, issued Feb. 
7, 1978, and 4,091,410, issued May 23, 1978, both in the name of R. Citta, 
as examples of biphase stable loops. The copending application and the 
mentioned patents are incorporated herein by reference. 
The received signal at the output of amplifier 48 is either in phase with, 
or 180.degree. out of phase with, Fo. A special identification signal, (to 
be discussed in further detail hereinafter) inserted at the encoder by ROM 
31 (FIG. 1A) into the vertical interval of the television signal, is also 
recovered as part of the data and is interpreted by the controller to 
determine whether the phase of the received signal should be reversed to 
establish the correct phase relationship. Multiplier 50, under control of 
controller 56, multiplies the signal at the output of amplifier 48 by 
either +1 or -1 to assure the correct phase relationship with Fo. Those 
skilled in the art will recognize that, alternatively, the phase of Fo may 
be controlled by appropriate multiplication rather than by controlling the 
phase of the received signal as described. In either case, after any 
necessary corrections, Fo and the received signal will have the same 
phase. It will of course be appreciated by those skilled in the art that 
any other well-known technique may be used in place of that discussed for 
determining the correct phase of Fo. 
Controller 56 develops a number of clock or timing signals from the 
received color burst in a well-known manner. It will be recalled that the 
color burst of the encoded signal changes RF carrier phase every half 
cycle thereby providing a conveniently detectable timing reference. These 
signals include a high frequency clock locked to the color burst and a 
horizontal rate clock derived by counting down therefrom. A low frequency 
clock is developed from an identification signal, to be described. Data is 
removed from the incoming signal by opening switch 54 during time periods 
corresponding to the occurrence of data. Sync information, i.e., a sync 
pulse and pedestal, is regenerated in the controller and applied via D/A 
60 and multiplier 62 to summing network 58. 
Multiplier 62 multiplies Fo with the output of D/A converter 60 to produce 
a carrier signal, the amplitude of which is determined by the coded 
representation of the line averaged low frequency video, for addition to 
the received video signal supplied to summing network 58. The output of 
summing network 58 is therefore the reconstituted video portion of the 
television signal. This signal is supplied to a special AGC circuit 61 and 
to summing network 66 where it is recombined with the sound modulated 
carrier and passed to conventional television signal processing circuitry 
(not shown). The output of AGC circuit 61 controls the gain of amplifier 
42 to assure that the analog value of the digital representation of the 
line averaged video at the receiver matches that in the transmitter since 
the digital data is not altered by transmission attenuation as are analog 
portions of the signal. 
The special AGC circuit 61 includes an RF detector 64, a pair of sample and 
hold (S/H) circuits 63 and 65 and a comparator 67. As will be explained, a 
reference signal is transmitted and portions thereof are sampled in the 
receiver to determine attenuation effects on the analog portions of the 
signal and to compensate the receiver gain accordingly. 
While the receiver of FIG. 5 operates at RF frequencies, in many 
installations, it is desirable that the receiver operate at baseband 
frequencies and FIG. 6 illustrates such a receiver. A tuner/IF 41' 
receives the transmitted signal and applies an IF signal through an 
amplifier 42' to a video IF bandpass filter 46' and to a sound IF bandpass 
filter 44'. Filter 46', in turn, supplies the IF signal to a BPLL 52' and 
to a multiplier 50'. Data is removed by BPLL 52' and applied to a 
controller 56'. BPLL 52' also recovers a pair of quadrature-related IF 
carriers Fo' and Fo'/90.degree., Fo' being applied to a multiplier 58' and 
Fo'/90.degree. to a multiplier 59'. Controller 56' determines on the basis 
of a received reference signal whether the phases of the received signal 
and Fo' are the same and controls multiplier 50' to reverse the phase of 
the signal, if necessary, by multiplying by a +1 or by a -1. Multipliers 
58' and 59' function as synchronous detectors for developing output 
baseband video and 4.5 MHz sound signals, respectively, in response to Fo' 
and Fo'/90.degree.. The 4.5 MHz sound signal is applied to a 4.5 MHz sound 
BPF 44'' and the composite video signal is applied to a switch 54'. Switch 
54' is operated by controller 56' to open during data and horizontal sync 
portions of the received signal. A D/A 60' is operated by controller 56' 
and supplies one input of a summing network 47', the other input being 
supplied by switch 54'. D/A 60' supplies the sync and sync pedestal to 
summing network 47' along with an analog signal corresponding to the coded 
representation of the low frequency components sent in the data recovered 
by BPLL 52', which are added to the baseband video signal developed at the 
output of switch 54'. A reconstituted baseband video signal therefore 
appears at the output of summing network 47' and is applied directly to a 
pair of S/H circuits 63' and 64', which are operated under control of 
controller 56' to sample the reference signal that is transmitted to 
determine the attenuation effects on the analog portions of the 
transmitted signals. Again a comparator 67' supplies any correction 
required to adjust the gain of amplifier 42' to match the analog signal 
portions with the digital representations. The reconstituted video signal 
is also combined with the 4.5 MHz audio signal in a summing network 66' to 
provide an output baseband television signal which may be applied to a 
television monitor or the like for viewing. 
Referring to FIG. 7, a digital implementation of a transmitter constructed 
in accordance with the basic hybrid form of the invention is shown. A 
baseband source of composite video signal 12 is coupled to video clamp 14, 
the output of which supplies an A/D converter 78 and a sync separator 
circuit 28. A timing and control circuit 84 is intercoupled with A/D 
converter 78 and is supplied with the output of sync separator 28. Video 
clamp 14 is operated under control of timing and control circuit 84 to 
clamp the incoming video signal at the back porch level. The source of 
data 38 is coupled to the timing and control circuit 84. The output of A/D 
converter 78 is supplied to a digital averaging circuit 79 and to a RAM 
memory 80. Digital averaging circuit 79 is operated under control of 
timing and control circuit 84 to sample the output of A/D converter 78 
during the active video portions of the signal and to develop an average 
of the digital values for each individual horizontal line. This value is 
supplied back to timing and control circuit 84 and to a summing network 81 
which is also supplied with the output of RAM memory 80. RAM memory 80 
comprises a two video line memory in which one video line is written in as 
the previous video line is read out. This arrangement introduces a one 
line delay to assure that the digital average signal is subtracted from 
the video samples of the appropriate horizontal line. The output of 
summing network 81 is supplied to a multiplexing circuit 82 which is also 
coupled to the output of a ROM 31. ROM 31 supplies the reference and 
identification signals to the multiplexer 82 as will be described further 
below. Data from data source 38 and timing and control circuit 84 is 
applied to a third input of a multiplexer 82 during the horizontal 
blanking intervals of the signal. The data includes a coded representation 
of the line averaged values developed by digital averaging circuit 79. The 
output of multiplexer 82 is coupled to a D/A converter 86 whose output is 
supplied to a low pass channel filter 23 and thence to a modulator 34. 
Summing network 81, multiplexer 82, D/A converter 86 and modulator 34 are 
all operated under control of timing and control circuit 84. Modulator 34 
is supplied with an RF signal and its output is further processed as 
indicated in FIG. 1A. 
Referring back to FIG. 2, it will be seen that the waveform C is obtained 
by subtracting waveform B from waveform A during the active video portions 
of each horizontal line, except during the horizontal blanking interval 
72. It will be appreciated by those skilled in the art that a similar 
result would be obtained by adding a waveform of magnitude B to the 
horizontal blanking interval only and correcting for the resulting change 
in zero level. When considering the digital implementation of the encoder, 
the latter technique involves considerable simplification and is the 
presently preferred method of implementation for this version of the 
invention. 
In FIG. 7, the output of the A/D converter 78 preferably comprises 
approximately 910 samples per horizontal line with about 752 of those 
samples representing the active video portion of the line. Each sample is 
represented by either 8 or 10 bits depending upon the output resolution 
desired. For example, for ordinary commercial type television signals, an 
8 bit resolution is sufficient, whereas for studio level quality and 
transmission applications, a 10 bit resolution is preferred. The number of 
bits selected for the active video portion is preferably divisible by 2 
which greatly simplifies the hardware. As alluded to previously with 
respect to FIG. 2, it may be preferable to add the line averaged value 
(waveform B) to the horizontal blanking interval of the signal rather than 
to subtract the line averaged value from the active video portion. This 
would entail approximately 60 additions as compared with approximately 752 
subtractions and would again materially simplify the operation and 
hardware. However, the result would be the same after correction for the 
zero level and the particular technique utilized should not be considered 
limiting of the invention. The digitally processed signal is then supplied 
from the summing network 81 to the multiplexer 82 along with data from 
timing and control circuit 84 and the fixed identification and reference 
signals from ROM 31. After passage through D/A converter 86, the signal is 
handled in the same manner as described with respect to the transmitter of 
FIG. 1A. 
Because of the nature of the transmitted television signal, that is, a 
hybrid of analog and coded digital information, a system for compensating 
for transmission attenuation experienced by the analog signal (which does 
not alter the digital data) is provided. In order to properly reconstruct 
the received signal, the analog video signal may need to be adjusted to 
maintain the same relationship, between it and the digital data, that 
existed at the transmitter. The invention provides for sending a reference 
signal with a known relationship between the analog and digital data, 
detecting that signal in the decoder and comparing the detected levels to 
determine the amount and polarity of adjustment required, if any. 
Referring to FIG. 8, waveforms A, B and C, depicting two horizontal lines 
of a transmitted signal are shown. Waveform A constitutes a reference 
signal which comprises a white line (indicated as digital level 255) that 
falls to zero or black level (indicated as 0) followed by a second line of 
no video or black level. Waveform B represents an encoded counterpart of 
the reference signal (A) in which the white line has been reduced to a 
digital level of 55 by subtraction of an assumed average level of 200. The 
black level portion of the video line now occupies a level of -200, 
reflecting the subtraction of the average level of 200 therefrom. The 
second line, however, is unchanged in the active video portion since its 
average level is zero. Waveform C represents the decoded (reconstituted) 
signal and also indicates two sample areas identified as sample #1 and 
sample #2. Samples of the levels are taken at the indicated areas and 
stored in the sample and hold circuits of the receiver. Under conditions 
where the analog signal does not experience attenuation, sample #1 will 
reflect that the signal level has been returned precisely to zero level 
and will match sample #2. Should the decoded (reconstituted) signal be 
higher, as indicated by the dashed line portion H, sample #1 will be 
greater than sample #2 and the output of comparators (67 in FIG. 5 and 67' 
in FIG. 6) will generate a correction voltage for application to amplifier 
42 or 42'. If, on the other hand, the decoded signal is at a lower level 
L, sample #1 will be less than sample #2 and an opposite type correction 
will be supplied from the comparator to the amplifier. The provision of 
this reference signal, including one horizontal line with a significant 
analog video portion and a subsequent line with a zero analog video 
portion, provides a built-in standard for determining what has happened to 
the analog signal during transmission and processing. 
In FIG. 9, one form of identification signal is shown that serves the dual 
purpose of providing a start signal for timing purposes and for 
identifying the proper phase relationship between the video carrier signal 
and Fo. A normal encoded line (shown not-to-scale) includes data, 
horizontal pulses 90, a color burst 91 and an active video portion 92, 
which assures a certain number of zero crossings. Detection is based upon 
no zero crossings occurring during a line. An identification line is 
established without zero crossings by removing data pulses and color 
burst. The polarity of the video signal 93 may be used to indicate a 
particular phase relationship between the video carrier and the recovered 
Fo signal. The next line, assumed to be in the vertical blanking interval, 
does not have color burst, but does have data pulses. Thus, it too 
exhibits zero crossings. It will be appreciated that this illustration is 
but one of many arrangements that may be used for an identification 
signal. 
In FIG. 10, a receiver for receiving a basic hybrid processed FM television 
signal is shown. The signal received by IF 13 is supplied to a 
conventional FM demodulator 15 which in turn is coupled to a switch 17, a 
timing and control circuit 25 and a sound bandpass filter 35. Timing and 
control circuit 25 receives data from demodulator 15 and controls 
operation of switch 17 as well as supplying the coded data information to 
a D/A converter 33. The sync signal information is used to develop timing 
signals which are supplied to a D/A 33'. The output of switch 17 feeds a 
summing circuit 19 as does the output of D/A converter 33. The output of 
summing circuit 19 is supplied to another summing circuit 19' which is 
also supplied with the timing signals from D/A 33'. 
The dashed line block 39 labelled LPF 2 is not used in the basic hybrid FM 
receiver, but is used in conjunction with an extended hybrid receiver for 
receiving an extended hybrid processed signal as discussed below. 
The basic hybrid processing circuit described removes low frequency analog 
video signal components (below the horizontal line frequency) by 
subtracting the line averaged value of these components from the analog 
signal. This will be recognized as a specific example of a more general 
hybrid approach to television signal processing in which the video signal 
is divided into low frequency and high frequency components. It will be 
noted that the low frequency video components under 15 KHz account for the 
vast majority of transmission power required and that the removal of 
additional analog video components (under about 200 KHz) does not result 
in significant additional transmission power savings. However, extended 
hybrid processing does yield highly beneficial results in terms of 
developing a television signal of improved signal to noise performance and 
having minimum co-channel and adjacent channel interference potential with 
respect to NTSC signals. Thus, in the extended hybrid processing system of 
the invention, video signal components below about 200 KHz are removed, 
converted into digital form and coded data representative of the removed 
components is sent in a non-active video interval of the analog signal. 
In the extended hybrid transmitter of FIG. 11, the output of video clamp 14 
is supplied to a delay compensation network 49, which in turn is coupled 
to a summing circuit 20. The video signal is also supplied to sync 
separator 28 and to a low pass filter (LPF 1) 37. The filter 37 is 
indicated as passing frequencies up to about 200 KHz. The output of filter 
37 is supplied to A/D converter 26 which in turn supplies D/A converter 
32, the output of which is coupled to a filter 39 (LPF 2) and to a timing 
and control circuit 51. The output of filter 39 is subtracted from the 
full analog video signal in summing circuit 20. A data combiner 55 
receives the output of summing circuit 20 and data and timing signals from 
a timing and control circuit 51. The remainder of the circuit is 
substantially the same as that of FIG. 1A and operation is essentially the 
same. Filter 39 is not required if a full resolution digital coded 
representation is used. In most instances, however, the number of bits 
from A/D converter 26 will be limited, thus limiting the resolution, and 
filter 39 is included to assure that the transmitted signal will match 
that produced in the receiver. 
In FIG. 12, the effects of basic hybrid and extended hybrid processing on 
an idealized video signal consisting of a pulse and a bar are illustrated. 
Curve A represents the video signal having a relatively sharp pulse 
followed by a fairly broad bar. As shown, the average signal level is 
significantly above 0 volts. Curve B illustrates a pulse 57 and a bar 
signal when it is subjected to basic hybrid signal processing in 
accordance with the invention. It will be seen that the average signal 
level has been subtracted, thus reducing the overall magnitude of the 
signal. Curve C shows the signal after extended hybrid processing in 
accordance with the invention. Here it is seen that only high frequency 
signals, such as doublets 43 and 45 and pulse 57, which correspond to 
video edges and video detail, remain since the low frequency components, 
up to about 200 KHz, have been removed. In particular, extended hybrid 
processing produces a significant number of voids, such as that indicated 
at 47, in the transmitted analog signal. As will be discussed below, the 
voids are extremely useful for reducing the peaks in the remaining 
transmitted analog signal by the techniques of compression and dispersal 
filtering. As is well known, peak signals are of greatest importance when 
dealing with co-channel and adjacent channel interference problems. 
FIG. 13 illustrates the effects of hybrid processing of a square wave video 
signal on FM transmission bandwidth. Curve A illustrates a basic hybrid 
processed signal corresponding to a horizontal line of a one-half white 
and one-half black screen. Waveform B illustrates the transmission 
bandwidth centered about a carrier frequency Fo for the signal. In the 
extended hybrid processed waveform illustrated by curve C, only signal 
spikes in the form of doublets remain and the bandwidth shown by curve D 
is very much narrower, with the frequency deviation due to the black and 
white level signals, folding onto each other at the center frequency Fo 
since they each correspond to a zero volt signal. The extended hybrid 
signal FM transmission bandwidth is thus seen to be very much smaller than 
the bandwidth corresponding to the basic hybrid processed signal. This FM 
transmission bandwidth reduction translates into significantly improved 
noise performance in the FM channel and is an outstanding attribute of the 
extended hybrid processing of the invention. It should be noted that while 
signal to noise performance is greatly improved, no information is lost 
(the removed low frequency information is replaced in the receiver) and 
consequently distortion is not increased. 
As mentioned, peak signal magnitude is the major factor in contributing to 
adjacent channel and co-channel interference problems. Such peaks, in the 
form of doublets and pulses, are produced in the extended hybrid 
processing in response to video edge and video detail, i.e. whenever the 
video level changes rapidly. As was also discussed above, the signal level 
during non-video portions of the signal (blanking periods) is arbitrarily 
established in the hybrid processing system. As has been shown in the 
basic hybrid system, the line averaged video is used to set the horizontal 
line signal level at the average signal level for that line. Consequently, 
when the blanking interval ends, the video signal level will in all 
probability be closer to its average value for the line than it would be 
to an arbitrary level that had been established. Therefore, the magnitude 
of video level change (and of doublets produced by extended hybrid 
processing) will be minimized by setting the line to the average video 
level for that line. As mentioned in connection with the copending 
application, the transmitted video signal need not have horizontal 
blanking intervals, nor horizontal sync. It will, of course, have a line 
structure, and there will be transitions from line to line. These 
transitions appear as doublets when hybrid processed, and it would be of 
benefit to minimize their amplitude. Therefore two step processing is 
employed in which the line average of a line is removed with basic hybrid 
processing, prior to performing extended hybrid processing. Thus, two step 
hybrid processing is very desirable since with it, the level of the video 
signal between lines (and in the vertical blanking interval) is set to the 
average video signal level to reduce the magnitude of doublets produced. 
It is also contemplated that the average of adjacent video lines be used 
to reduce the doublet magnitude. 
FIG. 14 shows a transmission system that incorporates two step processing, 
i.e. first basic hybrid and then extended hybrid processing. The basic 
hybrid processing occurs by virtue of line integrator 24, A/D converter 26 
and D/A converter 32, one H delay circuit 16, switch 18 and summing 
circuit 20. The extended hybrid processing occurs by virtue of filter 37, 
A/D converter 26', D/A converter 32', filter 39, delay compensation 
circuit 49 and summing circuit 20'. The basic hybrid processing is 
performed first and is followed by the extended hybrid processing. Timing 
and control circuit 53 supplies appropriate timing and data signals to 
data combiner 55. As mentioned, the data is placed in non-active video 
portions of the signal, which will generally comprise the vertical 
blanking periods or both horizontal and vertical blanking periods. 
FIG. 15 illustrates the basic hybrid receiver of FIG. 6 modified to accept 
a two step hybrid processed signal such as that produced by the 
transmitter of FIG. 14. The receiver includes a D/A 32 and a filter 39 
which are supplied with the extended hybrid data by controller 56''. After 
detection by BPLL 52', the removed basic low frequency information is 
added in summing network 47', and the remainder of the low frequency 
information is added in summing network 69. The frequency response 
characteristic of LPP2 filter 39 is, of course, the same as that of LPF2 
39 in the transmitter, which is indicated by the same reference number 
being used for each. 
The basic hybrid and extended hybrid processing of the invention, when 
combined with certain well known techniques in the art such as temporal 
pre-emphasis or filtering (also referred to as frame combing or field 
processing) signal compression, time dispersal and pre-emphasis, yields a 
television system of optimal signal and noise characteristics. 
In FIG. 16, such an optimal television system with minimum adjacent and 
co-channel interference includes a digital type transmitter having a 
hybrid processing stage (may be two step) to remove and digitally encode 
low frequency video components as data. The remaining signal comprising 
digitized high frequency components is subjected to temporal pre-emphasis 
for emphasizing the changes between successive frames of video and then 
subjected to compression and time dispersion. The dispersed signal is 
subjected to pre-emphasis and channel filtering before modulation on a 
double sideband suppressed carrier that is centered in a 6 MHz frequency 
band. The pre-emphasis and channel filtering may be accomplished by 
applying the time dispersed signal (in digital form) to a digital filter 
and then D/A converting it and applying it to an analog filter which also 
receives the representative data (hybrid) and other data after suitable 
waveshaping. 
The receiver incorporates a true synchronous detector, followed by a signal 
de-emphasis stage and an A/D converter. The digitized signal is subjected 
to inverse time dispersion, expansion and temporal de-emphasis to obtain 
the digitized high frequency component. The removed low frequencies are 
reconstructed from the data and the original signal recovered by adding 
back the removed components. The system has very attractive advantages in 
minimizing adjacent channel and co-channel interference. Further, the 
carrier frequency may be arranged to have a "precise offset" frequency 
relationship with co-channel NTSC (or hybrid) signals. The precise offset 
frequency, as is well known in the art, should be a one-third or one-half 
multiple of the horizontal scan rate and a one-half multiple of the 
vertical scan rate. The effect of this precision offset is to break up 
sections of video, corresponding to DC, into lines of video (at about 10 
KHz) which is visually much less perceptible on a television screen. Thus, 
a co-channel video display would be significantly broken up and thereby 
rendered much less noticeable. The hybrid signal should also be frame 
locked to the NTSC co-channel to assure that the data portions of the 
hybrid processed signal (data is sent in the vertical blanking intervals) 
do not occur during video portions of the NTSC co-channel. In the 
copending application referred to above, the transmitted signal has no 
sync which further reduces the peak signal excursions and aids co-channel 
performance. 
In FIG. 17, temporal pre-emphasis in the transmitter is illustrated. The 
video input signal is subjected to a delay of T (one field delay), 
multiplied by a factor "a" (less than one) and subtracted from the 
undelayed signal. Its impulse response and frequency response are shown. 
The opposite action occurs in the receiver as illustrated in FIG. 18 where 
the input signal is subjected to a delay of T, multiplied by the same 
amplification factor "a" and fed back to the signal. 
Frame combing has been used in the prior art for separation of color 
signals. The benefit of frame combing or temporal filtering has not been 
recognized in encoding video signals for reduced power transmission. Its 
use in a hybrid television signal system reduces transmission power for 
static images and thereby helps to reduce interference of the hybrid 
signal into an NTSC co-channel and also to minimize co-channel 
interference from an NTSC channel into a hybrid signal receiver. The 
temporal pre-emphasis filter response for static images (zero temporal 
frequency) is at a minimum while its response for frequencies equal to T/2 
is at a maximum. As shown, the response for static images is reduced to a 
1-a and at one-half the field rate is increased. These numbers are of 
course a function of the actual filter design. Thus, the video signal 
corresponding to a static image is reduced significantly. Since most 
television pictures are relatively static, the overall interference into 
an NTSC co-channel will be reduced. For video motion, the interference 
into the NTSC co-channel, while not reduced, will be much less noticeable 
because fast moving video images (very small width moving edges) are 
difficult for the human eye to resolve. 
The complementary receiver temporal de-emphasis filter in FIG. 18 has the 
opposite effect and is an infinite impulse response filter with a single 
pole which cancels the zero in the corresponding filter of the 
transmitter. The combined frequency response of the transmitter temporal 
filter and receiver temporal filter is flat. The response of the receiver 
temporal filter at zero temporal frequency (static input) is at a maximum 
while the response for moving images is at a minimum. In conjunction with 
the precision offset mentioned above, the static portions of the NTSC 
co-channel interference can be made to appear as one-half field rate 
temporal frequency components, falling into the troughs of the filter 
response thereby be reduced significantly. 
Signal pre-emphasis and de-emphasis may be used to help improve the noise 
performance of the hybrid signal. Circuitry therefor is not illustrated 
since it involves well known boosting of the high frequencies during 
transmission and reducing or rolling off the boosted high frequencies in 
the receiver. The receiver filter may have a haystack shape and sharply 
discriminate against adjacent channel frequencies. 
A compander compressor is illustrated in FIG. 19 and a compander expander 
is illustrated in FIG. 20. Companding (that is compressing and expanding) 
improves interference performance both to and from an NTSC co-channel. 
This is so, since with hybrid processing, high signal amplitudes only 
occur during transitions, such as on video edges or in video detail. (With 
temporal pre-emphasis, high signal amplitudes only occur on moving edges.) 
As seen in FIG. 19, the compressor has a non-linear transfer 
characteristic that raises the level of low amplitude signals and lowers 
the level of high amplitude signals. By reducing the amplitude of the 
highest amplitude signals, the signal peaks that cause co-channel 
interference are reduced. The low amplitude signals are increased, but 
they are not the peak signals that are responsible for co-channel 
interference. In the receiver, the expander transfer characteristic is 
complementary (see FIG. 20), so that the overall effect on the signal is 
flat. The expander, it will be noted, also serves to reduce interference 
from the NTSC co-channel into the hybrid signal channel. The NTSC 
co-channel will, under the worst conditions, be a low amplitude signal and 
the hybrid signal exhibits high amplitudes only during video movement and 
for video detail. The hybrid receiver will receive a signal that will be a 
sum of the low amplitude co-channel and the hybrid signal. In areas of low 
detail, where the co-channel interference will be most visible, the 
combined signal levels of the co-channel and hybrid signal will still be 
small and the expander will further reduce that by the inverse of the 
amplification factor of the compressor. Thus, the most visible interfering 
signals are reduced. During moving video and video detail in the hybrid 
signal, where the co-channel is least visible, the combined levels of the 
co-channel and hybrid signal is high and will be further increased by the 
expander characteristic. The result is that the co-channel interference is 
shifted from low video detail (flat, stationary) areas to high detail and 
moving areas where it is much less visible. Noise is also processed as 
co-channel interference and therefore the same improvement is obtained 
with respect to noise performance. The compression process may create 
distortion products in the video signal which can be compensated for by 
suitable peaking. 
HDTV System Format 
In accordance with another aspect of the invention, the optimal 
transmission system described above is used for transmitting an HDTV 
signal which is encoded to further minimize the affects of co-channel and 
adjacent channel interference with indigenous (e.g. NTSC) nearby 
transmissions. 
The HDTV system format of the present invention provides an improvement in 
horizontal resolution of about 1.84 relative to NTSC for a receiver having 
a 5:3 aspect ratio, and about 1.73 in the case of a 16:9 aspect ratio 
display. In order to provide an approximate doubling of NTSC resolution, 
the displayed image on the receiver is produced by 720 lines of active 
video progressively scanned at the NTSC vertical field rate of 59.94 Hz 
and at a horizontal scan rate of three times NTSC (47.2 KHz), with each 
line having a horizontal resolution of about 1020 lines per width (lpw), 
or, stated otherwise, about 510 cycles per picture width (cpw). In order 
to support this horizontal resolution, the minimum horizontal bandwidth of 
the video source signal at the transmitter is 28.9 MHz. This source signal 
may be provided at the transmitter by a video source producing a 787.5 
line progressively scanned signal, having a vertical rate of 59.94 Hz and 
a horizontal scan rate of 47.2 KHz. The video source signal is encoded for 
transmission over a 6MHz RF channel, the encoding process converting the 
video source signal into a transmission format having a line and field 
structure equivalent to that used in NTSC to facilitate the reduction of 
co-channel interference between HDTV and NTSC co-channels. In addition, 
the HDTV signal is more easily transcoded into an NTSC signal. 
More specifically, the HDTV source signal is transmitted over a 6 MHz RF 
channel in a format wherein each frame comprises five (5) fields 
transmitted at the NTSC vertical rate of 59.94 Hz. Each field actually 
comprises a pair of sub-fields, each transmitted on a respective 
quadrature component of a suppressed carrier signal approximately centered 
in the RF channel. Within the context of this transmission format, the 
video source signal is encoded according to a scheme whereby low frequency 
horizontal and vertical luma components are transmitted at the NTSC 
vertical rate of 59.94 Hz while the higher frequency luma components and 
color difference components are transmitted at 1/5 this rate (i.e. 
approximately 12 Hz). 
Briefly, the luma component of the video source signal is initially 
separated into three substantially contiguous horizontal frequency bands 
which, together with a pair of color difference signals, are then encoded 
into a series of components collectively comprising 480 lines of video and 
color information every 1/59.94 seconds. Each line has a time duration of 
about 63.56 microseconds (corresponding to an NTSC horizontal line 
including the blanking interval) and has a nominal bandwidth of 2.675 MHz. 
The 480 lines are divided between two sub-fields each including 240 active 
lines of video information. Five pairs of sub-fields comprise a complete 
frame of video information. In addition to the 240 pairs of video lines, 
each sub-field pair also includes a block of audio, timing and 
synchronizing signals. The audio, timing and sync signals occupy the 
equivalent of 22 and 23 NTSC horizontal lines in successive fields, 
corresponding to the vertical blanking interval of an NTSC signal. The 
video lines, which are derived from the lines of the video source signal, 
are selected for transmission in a manner providing an optimal match with 
the human visual system. Thus, the video lines representing the low 
frequency luma component, are transmitted at a high temporal rate (59.94 
Hz) for good motion rendition while the video lines representing the 
higher frequency luma components, together with the color difference 
components, are transmitted at a lower temporal rate (12 Hz). In addition, 
the lines are time multiplexed for transmission such that a predetermined 
number of lines derived for each luma horizontal frequency band is 
transmitted each field. This allows decoding of the transmission to be 
effected using the equivalent of a single frame store. Resolution of edges 
of the video image displaced from true vertical and horizontal is 
optimized by transmitting the low frequency luma band at full vertical 
resolution and reducing the transmitted resolution of the higher luma 
bands in discrete steps. 
The encoding process of the invention is illustrated in detail in FIG. 21. 
As shown, an encoder 110 receives three input signals from an HDTV signal 
source, such as a video camera; an HDTV luma input signal at a terminal 
112 and two HDTV color difference input signals C1 and C2 at terminals 114 
and 116, respectively. The input luma and color difference signals are 
preferably provided in digital form. Also, the HDTV signal source may 
provide RGB output signals, in which case a suitable matrix circuit would 
be used to provide the luma and two color difference input signals. In 
order to achieve the desired resolution, the output of the HDTV signal 
source comprises a progressively scanned 787.5 line image (three times the 
number of lines in an NTSC field), having a vertical rate exactly equal to 
the NTSC vertical rate of 59.94 Hz and a horizontal rate of 47.2 KHz which 
is exactly equal to three times the NTSC horizontal scanning rate. The 
output signal further has a minimum horizontal bandwidth of 28.9 MHz. 
As explained in further detail below, the luma encoding process initially 
separates the luma signal provided at input terminal 112 into three (3) 
horizontal frequency bands of approximately 9.6 MHz as shown in FIG. 22. 
All of the information in the low band (0-9.6 MHz) is transmitted in one 
frame (5 fields) at the full vertical resolution of 720 lines as 
represented by rectangle 111. All of the information in the middle 
(9.6-19.3 MHz) and high (19.3-28.9 MHz) bands are also transmitted in one 
frame but at reduced vertical resolutions of 480 and 240 lines as 
represented by rectangles 121 and 131 respectively. 
If all three horizontal bands were to be transmitted at full vertical 
resolution (720 lines), the spectrum required would be that represented by 
the large rectangle 141. This spectrum may be reduced in half by limiting 
the vertical resolution of each band as defined by the diagonal line 151, 
as has been proposed in the art. However, this technique has a number of 
disadvantages. First, extremely complicated and expensive two-dimensional 
diagonal filters must be used to achieve the illustrated result. Second, 
and perhaps more important, resolution for edges which are not quite 
vertical or horizontal is severely degraded. Vertically tilted or 
displaced edges are represented by angle A in FIG. 22 while horizontally 
tilted or displaced edges are represented by angle B. Such tilted or 
displaced edges occur frequently in video images and preferably should be 
reproduced with maximum resolution. However, as previously mentioned, in 
the case of the diagonally filtered spectrum, such edges are severely 
degraded. In particular, since the maximum vertical resolution is defined 
by diagonal 151, as angle A or B begins to increase (due to an edge being 
tilted from vertical or horizontal respectively) resolution is immediately 
reduced, the reduction increasing linearly along diagonal 151. In fact, 
full vertical or horizontal resolution is only achieved for perfectly 
vertical or horizontal edges. 
These limitations are overcome according to the invention by providing full 
vertical resolution for the entire low frequency band 111, and reducing 
the resolution in discrete steps for the middle and high bands 121 and 131 
as shown. Full vertical resolution is therefore achieved for edges 
displaced from true vertical by as much as the angle A. Similarly, full 
horizontal resolution is achieved for edges displaced from true horizontal 
by as much as the angle B. In addition, the spectrum can be realized using 
only relatively straight-forward vertical and horizontal filters rather 
than the much more complex two-dimensional diagonal filters. The only 
slight disadvantage is that there is only about a 33% spectrum utilization 
reduction (rather than 50%) relative to the full spectrum represented by 
rectangle 141. 
The luma encoding process actually separates the luma signal provided at 
input terminal 112 of encoder 110 into four components, two components 
representing the low horizontal frequency band and one component for each 
of the middle and high bands. These four components are shown in FIG. 23 
where they are labeled LL, LD, MH and HH. Together, the LL and LD 
components represent the lower one-third (0-9.6 MHz) of the horizontal 
frequencies of the HDTV luma input signal with full 720 line vertical 
resolution. The MH component represents the middle one-third (9.6-19.3 
MHz) of the horizontal frequencies with only 480 lines of vertical 
resolution and the HH component represents the upper one-third (19.3-28.9 
MHz) of the horizontal frequencies with 240 lines of vertical resolution. 
The LL component is transmitted at the rate of 59.94 Hz for providing a 
low detail video image at a relatively high temporal rate while all of the 
remaining components are time multiplexed for transmission at 1/5 this 
rate (approximately 12 Hz) for updating the high detail video information 
at a relatively low temporal rate. 
With further reference to FIG. 21, the input luma signal at terminal 112 is 
initially applied to three input filters 118, 120 and 122 which separate 
the input luma signal into the three 9.6 MHz bands illustrated in FIGS. 22 
and 23. Filter 118 is a low-pass filter which passes only the lower 
one-third of the horizontal frequencies of the input luma signal, while 
filters 120 and 122 are bandpass filters which respectively pass the 
middle and upper one-third of the horizontal frequencies. 
The 0-9.6 MHz signal developed at the output of filter 118 is coupled to a 
first vertical low-pass filter 124 and therefrom to a first vertical 
resampler 126. It will be recalled that the video source provided a signal 
having 787.5 progressively scanned horizontal lines at a vertical rate of 
59.94 Hz. Of these 787.5 lines, 720 represent active video. Circuit 126 
resamples the 720 lines of active video to 96 lines for developing 
component LL for transmission at a 59.94 Hz rate. Filter 124 is provided 
to smooth the vertical transitions of the output of horizontal filter 118 
so as to provide a signal compatible with the resampling rate of circuit 
126. That is, by selecting a cut-off frequency for vertical low-pass 
filter 124 corresponding to approximately 96/2 or 48 cycles per picture 
height, no aliasing products will be generated by circuit 126 during the 
resampling process. Prior to transmission, the 96 lines provided at the 
output of vertical resampler 126 are time expanded by a factor of 3.6:1 in 
a first time expander circuit 128. This, at the same time, reduces the 
horizontal bandwidth of each line by a factor of 3.6:1 to approximately 
0-2.7 MHz. This time expansion factor expands each line to a duration of 
approximately 63.56 microseconds. The entire line may be used to transmit 
video information or, alternatively, a small portion of each line may be 
reserved for an appropriate blanking signal. The output of circuit 128 is 
therefore a 0-2.7 MHz horizontal bandwidth component LL (see FIG. 23) 
which represents the lower one-third of the horizontal frequencies of the 
input luma signal with a vertical resolution of 96 lines, each line having 
a duration of 63.56 microseconds (corresponding to an NTSC horizontal 
line). The 96 lines of component LL are transmitted at a vertical rate of 
59.94 Hz. This component, which therefore represents the low horizontal 
and vertical luma information, carries the bulk of the motion information 
which can be seen by the human eye, and is therefore sent at a relatively 
fast update rate. 
In the receiver, to be described in detail hereinafter, the 96 lines of 
component LL received each field are processed by a vertical filter to 
provide a low frequency luma image on all 720 active lines of the display. 
This low frequency luma image is, in effect, obtained by interpolating the 
96 lines of component LL received each field into 720 lines on the 
display. The remainder of the vertical detail in the 0-9.6 MHz horizontal 
band is transmitted as lines of a difference component LD. The transmitted 
lines of difference component LD are used to update the vertical detail in 
the low frequency luma image produced in response to component LL on a 
line-by-line basis during each field of the transmitted five field frame. 
The lines of difference component LD are derived in encoder 110 by using a 
vertical filter 132 identical to the vertical interpolating filter used in 
the receiver. The input to vertical filter 132 is derived from the output 
of vertical resampler 126 and thus comprises the non-time-base-expanded 96 
lines per field of component LL. Vertical filter 132 interpolates this 
signal into the same 720 lines per field produced in the receiver for 
generating the low frequency luma image, and couples these 720 lines to 
the negative input of a summer 134. An interpolation algorithm is 
preferably employed wherein weighted representations of the 96 lines are 
used in deriving the interpolated lines. The positive input of summer 134 
is derived from the output of 0-9.6 MHz horizontal filter 118. This signal 
represents the lower one-third of the horizontal frequencies of the HDTV 
luma signal, but with full 720 line vertical detail. A delay compensation 
circuit 136 is interposed between filter 118 and summer 134 to compensate 
for any delays produced by vertical filter 124 and vertical resampler 126. 
Summer 134 therefore operates to subtract corresponding lines of the 
interpolated signal from the full vertical detail signal to produce 720 
difference lines per field. Each line represents the difference between 
the full vertical detail line in the low horizontal frequency band and the 
corresponding interpolated line developed in the receiver for producing 
the low frequency luma image. As explained below, the difference lines are 
transmitted as component LD to periodically update the vertical resolution 
of the image produced in the receiver in response to the lines of 
component LL. 
The 720 difference lines per field developed at the output of summer 134 
are applied to a temporal low pass filter 138 which is used to reduce 
aliasing components in the difference lines during motion. The output of 
filter 138 is, in turn, applied to a 1 of 5 multiplexer 140 and therefrom 
to a 3.6:1 time expander circuit 142. Multiplexer 140 passes 1/5 or 144 of 
the 720 difference lines developed each field for transmission in an 
interlaced pattern, 720 of such lines therefore being transmitted in 1/12 
second. The interlace pattern for five successive fields may be as shown 
below in Table 1. 
TABLE 1 
______________________________________ 
Fields Difference Lines Transmitted 
______________________________________ 
1 1, 6, 11, 16 . . . 716 
2 3, 8, 13, 18 . . . 718 
3 5, 10, 15, 20 . . . 720 
4 2, 7, 12, 17 . . . 717 
5 4, 9, 14, 19 . . . 719 
______________________________________ 
Time expander 142 is similar to time expander 128 and time expands each 
line provided by multiplexer 140 by a factor of 3.6:1 to develop component 
LD. At the same time, circuit 142 reduces the horizontal bandwidth of the 
difference lines by a factor of 3.6:1 to approximately 0-2.7 MHz. The 
output of circuit 142 is therefore a 0-2.7 MHz horizontal bandwidth 
difference component LD (see FIG. 23) comprising 144 difference lines per 
field (or 720 lines every 1/12 second), each line having a duration of 
63.56 microseconds. Each field of 144 difference lines of component LD 
will be used to update the vertical resolution of the low frequency luma 
image produced in the receiver in response to component LL. 
The middle third of the horizontal frequencies, represented by component MH 
in FIG. 23, are coupled by a second vertical low pass filter 146 to a 
second vertical resampler circuit 148. Resampler 148 resamples the 720 
lines of active video provided each field by filter 146 to 480 lines. 
Vertical low pass filter 146 has a cut-off frequency corresponding to 
approximately 240 cycles per picture height for inhibiting the production 
of aliasing products by circuit 148 during the resampling process. 
The 480 lines per field developed at the output of resampler 148, 
representing the middle third of the horizontal frequencies of the HDTV 
luma component, are frequency shifted in circuit 150 to 0-9.6 MHz. The 
frequency shifted lines are then coupled to a second 1 of 5 multiplexer 
152 by a second temporal low pass filter 154. As in the case of filter 
138, temporal filter 154 is used to reduce aliasing of the MH component 
during motion. Multiplexer 152 passes 1/5 or 96 of the 480 MH component 
lines developed by resampler 148 for transmission during each field. The 
96 lines of the MH component are also interlaced for transmission in a 
manner similar to that previously described. 
The output of multiplexer 152 is coupled to a time expander circuit 156 
which is similar to expanders 128 and 142 and which time expands each line 
provided by multiplexer 152 by a factor of 3.6:1 (and reduces its 
horizontal bandwidth by a corresponding factor) to develop the lines of 
component MH for transmission. The output of expander 156 therefore 
comprises a 0-2.7 MHz horizontal bandwidth component MH (see FIG. 23) of 
96 lines per field (or 480 lines every 1/12 second) representing the 
vertically filtered middle third of the horizontal frequencies of the HDTV 
luma component. Due to the vertical filtering and resampling, the static 
vertical resolution of component MH will be 2/3 that of the low horizontal 
frequency components LL and LD. Each transmitted field of 96 lines of 
component MH will be used to update the horizontal and vertical resolution 
of the low frequency luma image produced in the receiver. 
The high third of the horizontal frequencies of the HDTV luma component 
provided by bandpass filter 122 are processed in a similar manner to 
produce component HH (see FIG. 23) for transmission. The output of filter 
122 is coupled by a third vertical low pass filter 158 to a vertical 
resampler 160. Circuit 160 resamples the 720 active video lines to 240 
lines which are frequency shifted to 0-9.6 MHz by a frequency shifting 
circuit 162. The frequency shifted lines are coupled by a third temporal 
low pass filter 164 to a third 1 of 5 multiplexer 166 which passes 1/5 or 
48 of the 240 lines developed each field in a manner similar to that 
previously described with respect to the lines of component MH. The 48 
lines of component HH will also be interlaced for transmission as 
previously described. The output of multiplexer 166 is finally time 
expanded and reduced in horizontal bandwidth by factors of 3.6:1 in 
expander circuit 168 to develop the lines of component HH for 
transmission. The horizontal frequencies represented by the 48 lines of 
component HH transmitted each field will only have 1/3 the static vertical 
resolution of the low horizontal frequency components. 
The net effect of the foregoing is to allow for the transmission of a HDTV 
luma signal having a 28.9 MHz horizontal bandwidth over a 6 MHz RF 
channel. This is made possible to a large extent by the transmission of 
the various components of the luma signal in a temporal manner as 
described above. The vertical resolution of the luma signal is, however, 
different for each of the horizontal frequency bands as shown in FIGS. 22 
and 23, the low horizontal frequencies having a vertical resolution of 720 
lines, the middle frequencies 480 lines and the high frequencies 240 
lines. The net effect of these differences is a reduction in diagonal 
resolution as previously described. 
In addition, the transmission is effected such that a predetermined number 
of lines of each of the components LL, LD, MH and HH will be transmitted 
during each field. In the case of component LL, all 96 lines are 
transmitted while a reduced number of lines of components LD (144 of 720), 
MH (96 of 480) and HH (48 of 240) are transmitted each field. This 
arrangement allows an updated image to be reproduced by the receiver each 
field, with the low frequency video information being fully updated each 
field and the higher frequency information being updated on a partial 
basis each field and fully updated in a complete frame. Also, since the 
image reproduced by the receiver is updated on a line-by-line basis each 
field, the receiver requires the eqivalent of only a single frame store. 
The color difference components C1 and C2 provided at terminals 114 and 116 
respectively of encoder 10 are processed using techniques similar to those 
used for processing the luma components. The C1 and C2 signals are 
initially bandlimited to 9.6 MHz by low pass filters 172 and 174 
respectively. The bandlimited C1 and C2 signals are then vertically 
filtered by vertical low pass filters 176 and 178 respectively before 
being vertically resampled by resamplers 180 and 182. Both signals are 
resampled from 720 vertical lines to 240 lines and are then filtered by 
temporal low pass filters 184 and 186 prior to being decimated by a factor 
of five by one of five multiplexers 188 and 190. Thus, each of the 
multiplexers 188, 190 passes 48 of the 240 vertically resampled lines 
produced each field in a manner similar to that previously described with 
respect to components MH and HH. The outputs of multiplexers 188 and 190 
are then applied to a pair of 3.6:1 time expanders 192 and 194. As will be 
explained in further detail hereinafter, the 480 (240.times.2) lines of 
color difference components are transmitted in an interlaced pattern and 
will be interpolated at the receiver into 720 active lines to provide a 
smooth color presentation. The color difference resolution will be one 
third the horizontal and vertical resolution of the transmitted luma 
components. In comparison, in the horizontal direction, this is 4.48 times 
NTSC chroma resolution and 0.5 times NTSC chroma resolution in the 
vertical direction. 
As a result of the foregoing encoding, the low horizontal and low vertical 
frequency luma component LL will be updated every field (i.e. every 
1/59.94 seconds) to provide very good motion reproduction where the human 
eye is most sensitive to movement. Updates for the higher frequency luma 
components LD, MH and HH, as well as for the color difference components 
C1 and C2, for which the human eye is less sensitive to movement, will 
occur on a partial basis every field, but will require an entire frame 
(five fields or approximately 1/12 second) for a complete update. 
The encoding process described herein results in the development of 480 
luma and color difference lines which must be transmitted every field 
(1/59.94 seconds), each line of which has a horizontal bandwidth of 2.7 
MHz and a duration of 63.56 microseconds. In particular, 96 lines of 
component LL, 144 lines of component LD, 96 lines of component MH, 48 
lines of component HH and 96 lines of components C1 and C2 must be 
transmitted every field. In order to maintain compatibility with the 
existing NTSC line transmission structure, the lines are paired and 
transmitted on respective quadrature components of a suppressed video 
carrier approximately centered in a 6 MHz RF channel. Each field can 
therefore be considered to comprise two sub-fields, each sub-field 
corresponding to a respective quadrature component of the video carrier. 
Each sub-field is therefore used to transmit 240 of the 480 HDTV lines, 
corresponding to 240 active NTSC video lines. On the average, this leaves 
221/2 NTSC equivalent lines every 1/59.94 seconds for the transmission of 
audio, timing and sync information during each sub-field. 
The structure of the sub-fields is illustrated in FIG. 24. The quadrature 
components of the video carrier on which the respective sub-fields are 
modulated are represented by a first column labeled I channel modulation 
and a second column labeled Q channel modulation. Sub-field 172 of the I 
channel and corresponding sub-field 174 of the Q channel together 
represent a first HDTV transmission field pair having a time duration of 
1/59.94 seconds. Sub-field 176, transmitted immediately after sub-field 
172 on the I channel and sub-field 178 transmitted immediately after 
sub-field 174 on the Q channel, represent a second HDTV transmission field 
pair. In addition to the two sets shown, three similar sets of sub-field 
pairs are sequentially transmitted on the I and Q channels to complete the 
HDTV transmission frame. It will be observed that the line structure of 
each sub-field is equivalent to that of an NTSC field, each sub-field 
comprising 240 lines of luma and color difference HDTV components and, and 
on the average, 221/2 additional lines corresponding to the NTSC vertical 
blanking interval (VBI). Actually, alternate sub-fields include 22 and 23 
VBI lines as illustrated. As mentioned previously, this equivalency 
facilitates the reduction of interference between NTSC and HDTV 
co-channels and also makes it easier to transcode an HDTV signal into NTSC 
format. 
The pairing of HDTV lines in corresponding sub-fields is chosen to minimize 
sensitivity to crosstalk which might occur between the quadrature channels 
under non-ideal conditions. In a preferred embodiment of the invention, 
the five line pairing sequence shown in Table 2 is repeated 48 times in 
each sub-field for a total of 240 lines. It will be observed that this 
sequence provides for the transmission of 96 lines of component LL, 144 
lines of component LD, 96 lines of component MH, 48 lines of component HH 
and 48 lines of each of components C1 and C2 each field. 
TABLE 2 
______________________________________ 
Line # I Channel Q Channel 
______________________________________ 
1 LL LL 
2 LD LD 
3 LD MH 
4 MH HH 
5 C1 C2 
______________________________________ 
Since no synchronizing signals are provided in the transmitted lines of the 
HDTV components, synchronizing and timing information is provided during 
the transmitted VBI lines. Thus, two lines of the VBI of each sub-field 
are reserved for the transmission of a high frequency data clock 
(340.times.15.734 KHz), along with a vertical chirp and a horizontal 
chirp. The data clock establishes the basic timing for the encoder 110 and 
the decoder in the receiver, while the two chirp signals identify the 
phases of the clock signal required for deriving appropriate horizontal 
and vertical deflection signals in the receiver. 
FIG. 25 is a block diagram illustrating the manner by which the luma and 
color components generated by encoder 110, as well as the related audio, 
timing and sync information is transmitted according to the invention. As 
previously described, these signals are transmitted as suppressed carrier 
amplitude modulation of quadrature components of a picture carrier located 
in the center of a 6 MHz RF channel. This technique makes optimum use of 
the 6 MHz channel in an environment subject to interference of various 
forms including interference caused by NTSC co-channels. The audio, timing 
and sync information is preferably transmitted during the vertical 
blanking interval lines of each sub-field (see FIG. 24). Since the line 
structure of the HDTV transmission format described herein is equivalent 
to that used for NTSC transmissions, the HDTV vertical blanking interval 
can be frame locked to coincide with the vertical blanking interval of an 
existing NTSC channel which is likely to receive interference from the 
HDTV channel. Since data would be the most visible interference into the 
NTSC co-channel, frame locking assures that the data will be hidden in the 
vertical blanking time of the NTSC channel. 
With further reference to FIG. 25, the luma components LL, LD, MH and HH 
together with the color difference components C1 and C2 generated by 
encoder 110 are coupled to a formatter 200. Formatter 200 appropriately 
formats the HDTV components into sub-fields as illustrated in FIG. 24 (see 
also Tables 1 and 2) for providing an I channel modulation output and a Q 
channel modulation output. The two outputs of formatter 200 are converted 
to analog signals by D/A converters 202 and 204 and then coupled to inputs 
of respective summers 206 and 208. Both summers also receive inputs from a 
timing and synch source 210 and from a digital audio source 212 for 
inclusion in the VBI lines of the respective sub-fields. The outputs of 
the summers are coupled to an I modulator 214 and a Q modulator 216 for 
transmission. An oscillator 218 provides an in-phase carrier to I 
modulator 214 and a quadrature carrier to Q modulator 216. Quadrature 
modulators 214 and 216 provide RF outputs at the same picture carrier 
frequency, but 90 degrees different in phase. The two RF outputs are 
combined in a summer 220 and passed to a bandpass filter 222 for filtering 
and waveshapping prior to transmission. 
The resulting quadrature modulation of the 6 MHz RF channel is illustrated 
in FIG. 26, where f0 is the frequency of the suppressed video carrier in 
the center of the channel. The overall channel shape is shown in FIG. 27. 
It will be observed that the channel is flat for 2.35 MHz on either side 
of the carrier frequency and then drops off with a Nyquist slope centered 
about 2.675 MHz (170.times.15.734 KHz) from the carrier frequency. The 
Nyquist slope is therefore centered about one-half the bit clock rate of 
340.times.15.734 KHz. This overall channel shape, which must have a linear 
phase response, provides good transition regions while maximizing the data 
rate and minimizing intersymbol interference. The overall bandwidth, 
including the Nyquist transition regions, is plus or minus 3.0 MHz from 
the carrier frequency. In order to achieve the channel shape illustrated 
in FIG. 27, bandshaping is preferably split between the transmitter and 
the receiver such that the receiver can have a "haystack" type response 
centered about the video carrier. The "haystack" receiver response allows 
the simplest and most economical filter design in the receiver with 
excellent adjacent channel rejection. The transmitter bandshaping is 
provided by bandpass filter 222 shown in FIG. 25. 
The spectrum of the HDTV channel in relation to the NTSC channels around it 
is shown in FIG. 28. It will be seen that the placement of the HDTV 
carrier is offset slightly from the center of the channel in order to 
avoid the co-channel sound carrier. This offset permits the HDTV receiver 
to incorporate a sound trap to eliminate the interference. Collocation 
with adjacent NTSC channels is preferred to avoid adjacent channel 
overload. 
FIGS. 29 and 30 illustrate an HDTV receiver for receiving, decoding and 
displaying the HDTV signal transmitted as previously described. Referring 
initially to FIG. 29, the received HDTV signal is applied to a tuner 250 
which selects and translates the received signal to a predetermined 
intermediate frequency (IF) signal. The IF output signal of tuner 250 is 
coupled by an amplifier 252 to the inputs of a pair of IF filters 254 and 
256. The output of IF filter 256 is applied to a frequency and phase 
locked loop (FPLL) 258, which is preferably of the type disclosed in U.S. 
Pat. No. 4,072,909. FPLL 258 develops a pair of output carrier signals at 
the intermediate frequency, one output comprising an in-phase component 
and the other a quadrature component. The in-phase component is coupled to 
one input of a first multiplier 260 and the quadrature component is 
coupled to one input of a second multiplier 262. The output of IF filter 
254 is applied to the other inputs of multipliers 260 and 262, which 
function to demodulate and thereby recover the I and Q channel modulation 
respectively of the received signal. 
The I and Q channel modulation signals recovered at the outputs of 
multipliers 260 and 262 are applied to a controller circuit 264. 
Controller 264 recovers the digital information transmitted in the VBI of 
each sub-field for developing an output timing signal, an output synch 
signal and an output audio signal. It will be recalled that the VBI of 
each transmitted sub-field includes a high frequency data clock, a 
vertical chirp and a horizontal chirp. The data clock establishes the 
basic timing for operation of the receiver, with the two chirp signals 
identifying the clock phases required for deriving appropriate vertical 
and horizontal synch signals. The synch signals are thus derived by 
counting the data clock beginning with the clock phases identified by the 
respective chirp signals. As will be described in further detail, the 
derived horizontal and vertical synch signals are coupled to a CRT 266 
(see FIG. 30) for controlling the deflection of one or more electron beams 
for facilitating reproduction of the received HDTV image. As previously 
described, that derived synch signals will establish a vertical deflection 
rate of 59.94 Hz (identical to NTSC) and a horizontal deflection rate of 
47.2 KHz (three times NTSC). The audio output developed by controller 264 
is applied to an audio processor 268 for processing the audio information 
for reproduction by a suitable speaker system 270. 
The I and Q channel modulation signals recovered at the outputs of 
multipliers 260 and 262 respectively are also coupled to a pair of 
analog-to-digital converters 272 and 274 which convert the received analog 
lines of video information to corresponding digital signals. Each of the 
converters 272 and 274 therefore provides 240 lines of digital video 
information at a rate of 59.94 Hz. This digital video information is 
applied to a demultiplexer 276 which, in response to a timing signal from 
controller 264, separates the luma and color difference lines received 
each field and, provides the separated lines of video information at 
corresponding outputs LL, LD, MH, HH, C1 and C2. These outputs, together 
with the timing and synch signals provided by controller 164, are coupled 
to the video decoder of FIG. 30 for reproducing the transmitted HDTV video 
image. 
Referring to FIG. 30, the 96 lines per field of video component LL 
developed at the output of demultiplexer 276 are coupled to a vertical 
filter 280 which is identical to vertical filter 132 in encoder 110. 
Filter 280, in response to a timing signal from controller 264, converts 
the 96 lines of component LL received each field to 720 lines using the 
same conversion algorithm used by filter 132. At the same time, the 720 
lines are time compressed by a factor of 3.6:1. This allows for 
appropriate retrace blanking levels to be inserted in each of the lines 
consistent with the deflection signals applied to CRT 266. As previously 
mentioned, the conversion algorithm may employ a technique whereby 
weighted representations of the received lines are employed in deriving 
the interpolated lines. The output of vertical filter 280, which comprises 
720 lines of component LL at a rate of 59.94 Hz, is applied to a first 
input of a summer 292. 
The lines of video information comprising components LD, MH, HH, C1 and C2 
developed at the output of demultiplexer 276 are coupled to respective 
memories 282, 284, 286, 288 and 290. Cumulatively, memories 282-290 
provide sufficient memory for storing one complete frame of the received 
HDTV signal. In particular, memory 282 comprises a 720 line memory capable 
of storing the 720 interlaced lines of component LD received each 1/12 
second. As previously described, 144 of these lines are received each 
field of the transmitted signal (159.94 seconds) in an interlaced pattern. 
The received video lines are stored in corresponding lines of the memory 
with 1/5 of the memory being updated every 1/59.94 seconds and the entire 
memory every 1/12 second. While the lines of component LD are written into 
memory 282 at the rate of 144 lines/59.94 seconds, they are read out of 
the memory in a non-destructive manner at the rate of 720 lines/59.94 
seconds. On an individual basis, each line is read out of memory 282 3.6 
times faster than it is read into the memory. As before, this compresses 
each line by an appropriate factor to allow for insertion of retrace 
blanking signals consistent with the horizontal and vertical deflection 
rates at which CRT 266 is operated. The 720 time compressed lines read out 
of memory 282 are coupled to a second input of summer 292. 
Memory 284 comprises a 480 line memory capable of storing the 480 
interlaced lines of component MH received each 1/12 second. As previously 
described, 96 of these lines are received each field of the transmitted 
signal in an interlaced pattern. The received video lines are stored in 
corresponding lines of the memory with 1/5 of the memory being updated 
every 1/59.94 seconds and the entire memory every 1/12 second. The stored 
video information is non-destructively read out of memory 284 at the rate 
of 480 lines/59.94 seconds, with each individual line being read out 3.6 
times faster than it is read into the memory. As previously explained, 
each video line read out of the memory is therefore time compressed by a 
factor of 3.6:1 to allow for insertion of blanking levels consistent with 
the deflection rate at which CRT 266 is operated. 
The time compressed lines of video information read out of memory 284 are 
coupled to a frequency shifting network 294 and therefrom to a vertical 
filter 296. Frequency shifting network 294 translates the video 
information to the frequency band 9.6-19.3 MHz corresponding to the middle 
one-third of the luma horizontal frequencies. The frequency shifted lines 
of component MH are then filtered in vertical filter 296 to provide 720 
lines/59.94 seconds, which are coupled to a third input of summer 292. 
The video lines comprising component HH are processed in a similar manner. 
Thus, the 48 lines received each field are stored in 240 line memory 286 
with 1/5 of the memory being updated every 1/59.94 seconds and the entire 
memory every 1/12 second. The stored video information is 
non-destructively read out of the memory at a rate of 240 lines/59.94 
seconds, each individual line being read out 3.6 times faster than it is 
read into the memory and providing a corresponding time compression. The 
output of the memory is coupled to a second frequency shifting network 298 
which translates the video information to the frequency band 19.3-28.9 MHz 
corresponding to the high one-third of the luma horizontal frequencies. 
The frequency shifted lines of component HH are then filtered in a 
vertical filter 300 to provide 720 lines/59.94 seconds, which are coupled 
to a fourth input of summer 292. 
Summer 292 thus serves to reconstruct the HDTV luma signal by summing 
corresponding ones of the 720 lines of each of the LL, LD, MH and HH 
components applied thereto every 1/59.94 seconds. The reconstructed luma 
signal therefore represents all of the horizontal frequencies of the HDTV 
luma source signal provided at input terminal 112 of encoder 110 with 
reduced diagonal resolution. The signal includes 720 lines of active video 
presented for display at a frame refresh rate of 59.94 Hz. In order to 
provide appropriate retrace blanking intervals, the reconstructed luma 
signal is applied to a blanking signal insert circuit 302, which also 
receives a timing input from controller 264. Blanking insert circuit 302 
inserts appropriate horizontal and vertical retrace blanking levels into 
the reconstructed luma signal. Thus, a blanking level occupying about 1/6 
of each line is provided for horizontal retrace, vertical retrace being 
accommodated by providing a blanking level for the 67.5 VBI lines. The 
output of blanking circuit 302 is then applied to a matrix 304, which also 
receives the reconstructed color difference signals C1 and C2 to provide 
output R, G and B signals. The R, G and B signals are converted to an 
analog form by a D/A converter 306 and then coupled to CRT 266 for 
display. As previously mentioned, the display will comprise 720 lines of 
active video reproduced at a horizontal deflection rate of 47.2 KHz (three 
times NTSC) and at a frame rate of 59.94 Hz (equal to the NTSC field 
rate). The frequencies represented by the LL component of the luma signal 
are updated each display frame while 1/5 of the remaining luma frequencies 
are updated at the display frame rate and fully over five successive 
display frames. 
The lines of color difference components C1 and C2 are reconstructed in a 
manner similar to that described in connection with the luma components. 
Thus, the 48 lines of each of components C1 and C2 received each field in 
an interlaced pattern are stored in respective memories 288 and 290, with 
1/5 of each memory being updated every 1/59.94 seconds and the entire 
memories every 1/12 second. The stored color information is 
non-destructively read out of the memories at a rate of 240 lines/59.94 
seconds, each individual line again being read out of memory 3.6 times 
faster than it is read in to provide a corresponding time compression. The 
outputs of memories 288 and 290 are then filtered by respective vertical 
filters 308 and 310, each providing 720 lines of color difference signals 
every 1/59.94 second. The vertically filtered color difference lines are 
finally coupled to matrix 304 through respective "0" level insert circuits 
312 and 314. Circuits 312 and 314 insert "0" level signals into the color 
difference lines corresponding to the blanking levels inserted in the 
reconstructed luma signal. 
The encoding and transmission systems described above may be conveniently 
combined to form an integrated television system. This combination may, 
for example, take the form illustrated by the dotted line elements shown 
in FIGS. 21, 25, 29 and 30. Referring initially to FIG. 21, respective low 
frequency removal units 400-410 are provided for processing each of the 
components LL, LD, MH, HH, C1 and C2 developed by encoder 110. Each of the 
low frequency removal units is constructed as previously described (see 
FIGS. 1A-15) and provides a data output representing the low frequency 
portion of the associated component. Preferably, the lower 200 KHz portion 
of the LL component is removed by unit 400 while only the lower 15 KHz 
portions of the remaining components are removed by units 402-410. The 
data signals from the low frequency removal units are coupled to summers 
206 and 208 of FIG. 25 for transmission. 
The output of each low frequency removal unit is coupled to respective 
temporal filters 412-422 constructed as shown in FIG. 17. The value of "a" 
for the amplifier of the temporal filter 412 associated with component LL 
preferably has a value of 0.75 while the values of "a" of the amplifiers 
for the remaining filters 414-422 are all 0.50. In an alternative 
embodiment, the temporal filters 414-422 associated with components LD, 
MH, HH, C1 and C2 may be combined with respective temporal LPF's 138, 154, 
164, 184 and 186. 
Referring to FIG. 25, each of the I and Q channel modulation outputs of 
formatter 200 includes a processing unit 424, 426 for effecting the 
compression, time dispersion and pre-emphasis functions as illustrated in 
the transmitter portion of FIG. 16. Corresponding processing units 428, 
430 are shown in the receiver of FIG. 29 connected to the outputs of A/D 
converters 172 and 174. These processing units effect the complementary 
de-emphasis, time dispersion and expansion functions as illustrated in the 
receiver portion of FIG. 16. A data recovery circuit 432 is also provided 
in the receiver of FIG. 29 for recovering the low frequency data provided 
by the low frequency removal units of FIG. 21. 
Referring now to FIG. 30, the recovered low frequency data is applied to a 
plurality of low frequency restoration units 434-444, one such unit being 
provided for each video component. The low frequency restoration units 
restore the low frequencies of each component as previously described. 
FIG. 30 also includes the necessary temporal de-emphasis filters 
constructed as shown in FIG. 18, one such filter being provided for each 
component. The delay for each filter is provided by a respective memory 
280-290, each filter further including a respective amplifier 446-456 and 
a respective summer 458-468 connected as shown. The amplifier 446 for the 
filter associated with component LL has a coefficient ("a") of 0.75 while 
the remaining amplifiers 448-456 have coefficients "a") of 0.50. 
The transmission systems and methods and encoding method described are not 
to be considered limiting of the broad aspects of the invention. It is 
recognized that numerous modifications in the described embodiments of the 
invention may be made by those skilled in the art without departure from 
its true spirit and scope. The invention is to be limited only as defined 
in the claims.