Dual cancellation interferometric AMTI radar

An AMTI radar employs a dual cancellation format to cancel the clutter in the radar returns received by the apertures of an interferometric radar antenna. The three-aperture antenna presents the radar returns to three receivers which demodulate the returns to a complex in-phase signal I and a quadrature signal Q which are sampled at a pulse repetition interval at all ranges of interest. After conventional motion compensation four data sets are derived from the returns of the three apertures, L(t), C(t-.tau.), C(t) and R(T-.tau.) which represents samples taken over multiple pulse repetition intervals for each range interval or bin of interest. Fast Fourier transforms change the data sets into the frequency domain and phase compensations for the data sets are calculated in a triple interferometric signal calibration unit. Compensation phase is further adjusted in a clutter phase calculator allowing a much reduced phase quantity to be applied to the delayed data sets as the second part of the dual cancellation signal. Following subtraction of doppler filter outputs of the delayed data sets from the undelayed data sets in the cancellation unit, maximum clutter cancellation is achieved. The resulting clutter-cancelled signals are presented to a detection and validation processor to determine range, doppler, amplitude and angle measurements to the moving targets with a high degree accuracy.

CROSS REFERENCE TO RELATED APPLICATIONS 
The subject matter of this patent application is related to that disclosed 
in U.S. patent application Ser. No. 325,523 filed Nov. 27, 1981 by J. 
DiDomizio for MAXIMIZED/MINIMIZED PHASE CALCULATOR FOR AN INTERFEROMETRIC 
AMTI RADAR and to U.S. patent application Ser. No. 325,521 filed Nov. 27, 
1981 by J. Alimena and R. Briones for CHANNEL SWITCHING INTERFEROFMETRIC 
AMTI RADAR, and to U.S. patent application Ser. No. 325,524 filed Nov. 27, 
1981 by John A. DiDomizio for LOW TARGET VELOCITY INTERFERROMETERIC AMTI 
RADAR, all of which are assigned to the same assignee as the present case. 
TECHNICAL FIELD 
This invention relates to an aircraft-mounted synthetic aperture radar 
system and, more particularly, to an AMTI radar system that utilizes an 
interferometric processing technique to enhance target identification in 
the radar returns. 
BACKGROUND ART 
An airborne moving target indicator (AMTI) radar is generally known and is 
the type of radar that has the capability to reject or cancel signals from 
fixed, or unwanted targets (non-movers), such as buildings, hills, etc. At 
the same time, such radars typically highlight or enhance the radar return 
signals from any moving targets (movers) such as aircraft, vehicles, or 
the like. One technique used in AMTI radar of the coherent type involves 
utilizing the doppler shift imparted to the reflected radar signals by a 
moving target as a part of a processing scheme to distinguish a mover from 
a non-mover. This doppler shift appears as a change in the phase of the 
received signals between consecutive illuminating radar pulses. 
There are a number of problems which must be considered in the processing 
of radar returns where the AMTI radar is mounted in an aircraft. Because 
the aircraft is moving with respect to both the fixed and moving targets 
the radar returns from both target and clutter experience a frequency 
shift which can be corrected by known motion compensation techniques. 
Synthetic-aperture radars are also generally known and such systems 
generally use a multiaperture antenna together with the movement of the 
platform on which the antenna is mounted as additional inputs into the 
processing of return signals in an AMTI radar. While this adds 
significantly to the complexity of the processing of the radar return 
signals, clutter cancellation to identify the movers can be significantly 
enhanced. 
One well-known method of compensating for the effects of aircraft motion is 
known as displaced phase center technique and involves electronically 
displacing the antenna phase center along the flight path of the aircraft. 
Briefly, the technique involves the transmission and reception of radar 
returns by the antenna of the radar system having its phase center at a 
first known location. A second illuminating pulse is then transmitted and 
the return stored while the antenna has its phase center at a second known 
location. The phase centers of the first and second returns are separated 
by a precisely known distance related to the movement of the aircraft 
during the interpulse period and, knowing this information, the phase 
centers can electrically be changed to essentially coincide in time. At 
that point, the signals received by the multiaperture antenna from 
clutter, or stationary objects, will have properties suitable to 
cancellation leaving only the movers to be detected. 
One technique for clutter cancellation is described in U.S. Pat. No. 
4,093,950 issued Jun. 6, 1978 to T. ap Rhys for MOTION-COMPENSATION 
ARRANGEMENTS FOR. MTI RADARS. The clutter suppression technique described 
in this patent is not limited to two pulses at a time but may be applied 
to a number of pulses. Phase and amplitude adjustments are also made to 
minimize the effects of antenna construction errors. The antenna subarray 
have phase centers which are separated by 2 VT. The sum and difference 
signals from each two adjacent subarray are taken to produce a sum channel 
and a difference channel for each. group of subarrays. After adjustment of 
the difference channel signal in phase and amplitude, the latest return is 
added to a delayed return to produce a correction signal. That correction 
signal is then added to a delayed signal in the corresponding sum channel 
to provide a signal that is synchronized in time and phase with the most 
recent signal in the sum channel. 
U.S. Pat. No. 3,735,400 issued May 22, 1973 to C. Sletten and F. S. Holt 
for AMTI RADAR CLUTTER CANCELING METHOD AND APATUS describes a 
three-aperture simultaneous mode clutter canceller. This clutter 
cancellation technique is based on the premise that the return signals 
from stationary targets on the ground arrive at two antenna apertures with 
a unique and nearly linearly related phase delay as a function of doppler 
frequency if the antennas are displaced laterally along the aircraft 
flight path. Ground clutter cancellation can be achieved by a filter that 
separates the doppler spectrums into narrow channels and applies a given 
phase shift or delay to the returns in the narrow bandpass filter. Three 
channels of signal information from a three-aperture antenna are reduced 
to two clutter cancel channels. Range integrations and phase comparisons 
are performed on each channel of information to provide target detection 
and angle measurement. One of the limiting characteristics of this 
processing technique is that the antennas must be in a line coincident 
with the velocity vector of the aircraft. Each of the antenna apertures 
are spaced apart from the adjacent aperture by a fraction of a wavelength, 
in this particular case one quarter of a wave-length. Another limitation 
is that the transmit aperture, this being one of the three receive 
apertures, is the same aperture as one of the receive apertures and so the 
transmit and receive antenna beamwidths are identical. This is significant 
because the resultant doppler spectra in each of the channels is not 
highly influenced by each beamwidth pattern of the individual receive 
apertures. Still another limitation of this approach is that the aircraft 
veloctiy must be sufficiently large to provide a clutter spectrum of 50 
channels or more. Yet another limitation of this technique is that the 
bandpass filters are controlled by information from the aircraft 
navigation sensor which inherently has potential errors which should be 
considered. Also, this processing technique utilizes only three doppler 
filtering processes which necessarily provides less information compared 
to a system which incorporates a larger number of filtering processes. And 
finally, although this disclosed technique has a means for compensating 
for antenna calibration errors, it does not include any compensation by 
the receive signals to correct for velocity and/or boresite errors. 
DISCLOSURE OF INVENTION 
It is an object of the present invention to provide an airborne moving 
target indicator (AMTI) which effectively detects and measures angles to 
slowly moving targets in main beam clutter. 
A feature of the dual cancellation interferometric AMTI radar according to 
the present invention is that the disclosed technique is not limited to a 
side-looking radar. Stated differently, it is not necessary that the 
antenna apertures be oriented along a line which is coincident with the 
velocity vector of the aircraft. This means that the antenna apertures can 
be mechanically rotated to any desired direction. 
According to another feature of the dual cancellation interferometric AMTI 
radar of the present invention, the receive apertures of the radar antenna 
are spaced apart from adjacent apertures by a number of wavelengths. This 
is significant because it allows the apertures to be much larger in size 
and also allows the phase centers to be further apart for more angle 
accuracy. 
Another feature of the dual cancellation interferometric AMTI radar 
according to the present invention is that the three receive apertures are 
used together as a transmit aperture so that the transmit antenna 
beamwidth is one-third as narrow as each of the receive antenna 
beamwidths. This is meaningful because the resulting doppler spectra in 
each of the receive channels is not highly influenced by the beamwidth 
pattern of the individual receive apertures. 
An advantage of the dual cancellation interferometric AMTI radar according 
to the present invention is that the operation is not constrained by low 
platform velocity. In other words, it is not necessary to have the 
aircraft move at a sufficiently high speed to obtain an adequate clutter 
spectrum for the doppler samples. 
Another advantage of the dual cancellation interferometric AMTI radar of 
the present invention is that it includes a compensation means based on 
the data itself to correct for errors which might be associated with 
reference signals from a navigation system. 
Still another feature of the dual cancellation interferometric AMTI radar 
of the present invention is that four doppler filtering processes are 
performed on the data in the three receive channels to generate precise 
phase compensation for the radar return signals in the clutter spectrum. 
Yet another feature of the dual cancellation interferometric AMTI radar 
according to the present invention is that it compensates the receive 
signals for inaccuracies associated with velocity and/or boresite errors. 
According to the AMTI radar system of the present invention a dual 
cancellation format is utilized in the process of clutter returns received 
by three apertures of an interferometric antenna. The three-aperture 
antenna provides complex in-phase and quadrature signals which are sampled 
each pulse repetition interval (PRI) in an A/D converter at all ranges of 
interest. A particular advantage is that it eliminates the normal 
restriction that the radar pulse repetition frequency (PRF) required to 
achieve a high degree of clutter cancellation be related to the aircraft 
velocity and interarray spacing. This increases the radar capability by 
allowing variable PRF radar transmission. In turn this provides an 
increase in unambiguous target velocities and eliminates low velocity 
blind speeds. An added advantage is the increase in average power that can 
be transmitted with the use of a higher PRF, hence improving the range at 
which moving targets can be detected. 
Four data sets are generated from the motion compensated signals received 
from the three apertures. The sets L(t), C(t-.tau.), C(t) and R(t-.tau.) 
represent samples taken over multiple pulse repetition intervals for each 
range interval or bin of interest. The time .tau. is roughly equal to d/2v 
and is used to provide the first phase correction of the dual cancellation 
scheme. 
From the Fast Fourier Transformed data amplitude and phase compensations 
are calculated in a triple interferometric signal calibration unit. 
Compensation phase is further adjusted in a clutter phase calculator 
allowing a much reduced phase quantity to be applied to the delayed data 
sets as the second part of the dual cancellation signal. Following 
subtraction of doppler filter outputs of the delayed data sets from the 
undelayed data sets in the cancellation unit maximum clutter cancellation 
has been achieved in the clutter region. The two-range doppler maps that 
result after subtraction of doppler filter outputs are presented to a 
detection and validation processor to determine range, doppler, amplitude 
and angle measurements to the moving targets with a high degree of 
accuracy. 
In the clutter-free region, the FFT outputs (one per aperature) are 
noncoherently integrated for detection and the outer two are used for 
angle measurements in a known manner. 
The foregoing and other objects, features and advantages of the present 
invention will become more apparent from the following description of 
preferred embodiments and accompanying drawings.

BEST MODE FOR CARRYING OUT THE INVENTION 
Referring initially to FIG. 1, there is seen one embodiment of a dual 
cancellation interferometric AMTI radar according to the present 
invention. This technique is particularly well suited to suppressing 
clutter in a radar system whose performance is limited by the effects of 
platform motion. The present invention is utilized in processing signals 
acquired by a radar 10 of the coherent type which is known in the art. The 
radar system includes a left aperture 12, a center aperture 14 and a right 
aperture 16 that are provided by an array antenna or the like. Each 
aperture provides a signal indicative of radar returns to a receiver 18, a 
receiver 20 and a receiver 22. 
A particular feature of the present invention is that the antenna 
beamwidths of the left center and right apertures are substantially 
identical and are broader than the illuminating aperture beamwidth, (the 
illuminating aperture beam is more narrow by a factor of 3). As is known, 
each of the receivers typically converts the radar signals received at RF 
frequency range first to an IF frequency and then to a video signal of two 
different channels that are shifted in phase by 90.degree.. In other 
words, the receiver 18 for the left aperture has as outputs an in-phase 
component (I.sub.L) 24 and a quadrature component (Q.sub.L) 25, the 
receiver 20 for the center aperture 14 has as outputs an in-phase 
component (I.sub.C) 26 and a quadrature component (Q.sub.C) 27 and the 
receiver 22 for the right aperture has as outputs an in-phase component 
(I.sub.R) 28 and a quadrature component (Q.sub.R) 29. 
The I and Q video signals from the receivers 18, 20 and 22 are provided to 
analog-to-digital (A/D) converters 34, 36 and 38, respectively, so that 
the signals on the lines 44 and 45, the lines 46 and 47 and the lines 48 
and 49, are digital representations of the magnitudes of in-phase and 
quadrature components of each signal received by the radar apertures. The 
remainder of the disclosure is digital in nature and for simplicity the 
representations in FIG. 1 are shown with individual signal lines rather 
than trunk lines to accommodate binary signals. 
It also should be understood that all signals described hereinafter are the 
actual radar return signals or they are signals upon which modifications 
or changes have been performed in accordance with the described process. 
All such processes are performed with the requisite degree of fineness so 
that there is no significant error that arises as a result of quantization 
noise. The coherent radar 10 includes a pulse repetition frequency (PRF) 
clock 51 to provide timing signals on the line 52 related to each 
transmission or illuminating pulse from the radar system. A particular 
feature of the present invention is that the PRF utilized in this 
configuration is not restricted to any relationship in which aircraft 
motion perpendicular to the antenna orientation in a pulse repetition 
interval (PRI) must be a fixed factor of antenna array spacing. In 
addition a range clock 56 provides a high frequency timing signal on the 
line 57 defining each of the range bins associated with the return radar 
signal and also providing sampling strobes for each of the A/D converters. 
The I and Q signals, in digital form, from the left aperture 12, the center 
aperture 14, and the right aperture 16 are fed to a motion compensation 
unit 60. An inertial navigation system, (not shown), or other similar 
system on the radar platform provides digital signals indicating platform 
motion to the motion compensation unit. The received radar signals are 
complex multiplied by the signals received from the inertial navigation 
system. A complex multiplier circuit is generally known in the art and the 
purpose of the complex multiplier is to rotate the incoming radar data in 
any given range bin during a given PRI by another signal that represents 
the platform motion with respect to the range bin of interest. This 
results in the compensation of the data for platform motion, thus 
correcting for the doppler frequency at the center of the radar beam. The 
signals from the motion compensation unit 60 are on the lines 64 through 
69 and these signals represent the in-phase component I and quadrature 
component Q of the motion compensated signals received by the left, center 
and right apertures, respectively. 
Four bulk memory devices are used for array storage; memory 70, memory 72, 
memory 74 and memory 76. Each memory device is of sufficient size to 
record N.sub.d PRIs of data for each of N.sub.r range bins. The memories 
72 and 74 accept data from the center aperture receiver, the memory 70 
accepts data from the left aperture and the memory 76 accepts data from 
the right aperture. A particular feature of the present invention is that 
the data in these bulk memories is read in at different times. In other 
words, there are two separate time periods in which data is recorded, one 
for the left and a center device together and the other for the right and 
the other center device together. The data gate 52 that controls the left 
and the center allows the storage of the first N.sub.d samples taken 
(i.e., samples 1through N.sub.d) whereas the data gate 59 that controls 
the center and the right allows the storage of N.sub.d samples delayed in 
time by N PRIs (i.e., samples 1+N through N.sub.d +N) through the multiple 
PRI delay unit 58. The data gate delay is controlled by line 54 from the 
delay calculation read only memory (ROM) 53. 
One feature of the present invention is that the multiple PRI delay for the 
optimum clutter cancellation is calculated in a delay calculation ROM 53. 
The delay calculation ROM 53 has three inputs, aircraft velocity v, pulse 
repetition frequency f.sub.r and interarray spacing d. The output of the 
delay calculation ROM 53 on the line 54 is the number of PRI delays, N, 
such that .vertline.(d/2v)-(N/f.sub.r).vertline. is minimized. The 
interarray spacing, d, is a constant for all signal sets, the number of 
different pulse repetition frequencies, f.sub.r, are limited and the 
aircraft velocity, v, can be quantized to a low bit level so that the 
output, N, will generally be a small set of numbers which can be 
represented by a simple digital code when stored in the ROM. 
A particular feature of the present invention is that Fourier transforms of 
all of the data points are taken so that phase corrections for clutter 
cancellation can be applied in the frequency domain thus removing 
constraints inherently imposed by time domain cancellation. Accordingly, a 
digital FFT processor 84 is connected by the lines 80 and 82 to the memory 
70 for the left aperture and transforms the time history contained therein 
to I.sub.1 and Q.sub.1 signals on the lines 86 and 88 which provide a 
frequency domain representation of the received data. A difital FFT 
processor 94 is connected to the array storage device 72 by the lines 90 
and 92 and provides a comparable signal transform to the frequency domain 
for signals I.sub.c, and Q.sub.c, on the lines 96 and 98. A digital FFT 
processor 104 receives data stored in the array storage 74 in time 
sequence on the lines 100 and 102 and transforms this information into 
frequency domain signals I.sub.c and Q.sub.c on the lines 106 and 108. 
Likewise, in a similar fashion, a digital FFT processor 114 receives data 
from the right aperture that is stored in the array storage 76 on the 
lines 110 and 112, respectively, and provides output signals I.sub.r, and 
Q.sub.r, on the lines 116 and 118 which corresponds to this data in the 
frequency domain. The primed subscripts used (i.e., I.sub.c') indicate 
delayed data sets while the unprimed subscripts (i.e., I.sub.1) indicate 
undelayed data sets. In other words, each FFT processor converts the 
N.sub.d samples of the time history, of each of the returns in N.sub.r 
range bins into the corresponding N.sub.d samples of frequency domain 
information. At the output of each FFT processor we have the frequency 
characteristics for each range bin, this being referred to as a range 
doppler map, of a size N.sub.r range bins by N.sub.d doppler filters. 
A particular feature of the present invention is that the triple 
interferometric signal calibration unit 158 utilizes the range gated 
doppler filtered multiple component signal information and generates phase 
and amplitude calibration signals. Referring now to FIG. 2, one embodiment 
of the triple interferometric signal calibration unit will now be 
described. The triple interferometric signal calibration unit is 
essentially composed of two halves, one which operates on the left and 
center apertures and the other which operates on the right and center 
apertures. First considering the left and center half, a complex 
multiplication unit 120 is provided and is connected by two lines 86 and 
88 to receive the in-phase and quadrature components of the range doppler 
map from the FFT processor 84. The complex multiplier unit 120 is also 
connected to the lines 106 and 108 to receive the in-phase and quadrature 
components of the range doppler map from the digital FFT processor 104. 
All these input signals are derived from the undelayed data sets described 
above. The output flora the complex multiplication unit 120 is an in-phase 
and quadrature range doppler map which represents the phase angle between 
the two input signal sets and this information is presented via the lines 
122 and 124 to an angle processor 125. Angle processor 125 performs a 
linear regression on the phase data in each range bin and derives a linear 
frequency versus angle relationship for each range bin. It then averages 
all the range data to derive a best estimate of angle versus doppler 
information. The output from the angle processor 125 is provided as a 
phase calibration signal on the line 126 and this signal 
.theta..sub.1c.sbsb.i represents the measured angle versus doppler 
interferometer characteristic for the left-center interferometer. (Double 
subscripts (e.g., .theta..sub.1c.sbsb.i) indicate the apertures utilized 
(left-center) and a doppler frequency index (i)). Each half of the 
calibration unit also includes a magnitude detector, such as magnitude 
detector 128 and measures the magnitude of the complex vectors in both 
range-doppler maps of the left and center apertures. The magnitude 
detector 128 receives the output of FFT processor 84 on lines 86 and 88 
and the output of the FFT processor 104 on lines 106 and 108. The output 
from the magnitude detector 128 is presented on the lines 130 and 132 to 
the amplitude processor 133. The signals on this line represent the 
magnitude of the range doppler map from the left aperture and the range 
doppler map from the center aperture. The amplitude processor 133 provides 
an amplitude calibration signal A.sub.lc on the line 134 which is the 
ratio of the average clutter powers on the left and center apertures and 
is related to the amplitude difference in clutter spectrum between 
adjacent aperatures. 
In a similar fashion, the other half of the triple interferometric signal 
calibration unit also includes a complex multiplication unit 140 which is 
connected to the digital FFT processor 94 to receive the range doppler map 
from the center component and also to the digital FFT processor 114 to 
receive the right aperture range doppler map. An angle processor 145 
produces a phase calibration signal on the line 146 .theta..sub.cr.sbsb.i 
which is the measured angle versus doppler characteristic of the center 
right interferometer. A magnitude detector 148 receives a complex range 
doppler map from the right aperture and the center aperture and provides 
respective range doppler magnitudes to an amplitude processor 153. As 
before, the amplitude processor 153 provides an amplitude calibration 
signal on the line 154 A.sub.cr which is the ratio of the average clutter 
power in the center and right apertures. The data processed in this half 
of unit 158 is derived from the delayed data sets described above. 
A phase correction calculation is performed in a clutter phase calculator 
unit 160 in the following manner. 
A phase computation unit 169 calculates the phase difference between the 
delayed and undelayed data sets described hereabove. It accepts as an 
input the delay, N, on line 54 and generates on a doppler filter basis the 
phase 2.pi.Nf.sub.i .tau. where f.sub.i is the doppler frequency for each 
doppler filter and .tau. is the radar PRI. The output from the phase 
computation unit 169 is a phase difference signal presented on line 170 to 
phase adders 172 and 174 where it coherently added to the phase 
calibration signals on the lines 126 and 146. to form a left-center phase 
correction signal .phi..sub.1c .sbsb.i on a line 176 and a center-right 
phase correction signal .phi..sub.cr.sbsb.i. The phase correction signals 
are composite signals representing a time delay phase correction, the 
positional phase relationship between adjacent interferometers and a phase 
due to motion of the assumed moving targets. The resultant linear phase 
correction is significantly reduced by the use of the time delay feature, 
so that over a doppler filter width the phase deviation is small, allowing 
for a high degree of clutter cancellation across the whole filter. 
The phase correction signals are fed to the cancellation unit 250. It 
should be understood that the signals from the FFT processors when 
presented to the cancellation unit must be delayed for a predetermined 
period. This delay period allows the same signals to be processed by the 
triple interferometric signal calibration unit and the clutter free phase 
calculator unit prior to being combined in the cancellation unit. Lines 
176 and 178 go to a sine-cosine ROM 180 and a sinecosine ROM 182, 
respectively. The sine and cosine ROMs calculate the sine and cosine 
associated with the respective values of phase fed thereto. The output 
from the ROMs 180 and 182 on the lines 184, 186, 188 and 190 are the cos 
(.phi..sub.1c.sbsb.i) the sin (.phi..sub.lc.sbsb.i) the cos 
(.phi..sub.cr.sbsb.i) and the sin (.phi..sub.cr.sbsb.i) respectively and 
are fed to scaling units 192 and 194 where they are multiplied by the 
amplitude calibration signals for the respective interferometer. The 
outputs from the scaling unit 192 are fed via lines 200 (A.sub.1c sin 
.phi..sub.1c.sbsb.i) and 202 A.sub.1c cos .phi..sub.1c.sbsb.i) to the 
complex multiplication unit 204 and the output from the scaling unit 194 
is fed via lines 220 (A.sub.cr sin .phi..sub.cr.sbsb.i) and 222 (A.sub.cr 
cos .phi..sub.cr.sbsb.i) to the complex multiplication unit 224. The 
signals on these lines are correction factors which are applied to the 
delayed data sets which are the outputs of the digital FFT processors 94 
and 114, respectively. 
The complex multiplication unit 204 serves to scale the vector represented 
by signal lines 96 and 98 by a factor A.sub.lc and to rotate the vector by 
a phase .phi..sub.1c.sbsb.i (the subscript i indicating segmentation in 
doppler frequency). Likewise, the complex multiplication unit 224 serves 
to scale the vector represented by signal lines 116 and 118 by a factor 
A.sub.cr and to rotate the vector by a phase .phi..sub.cr.sbsb.i. 
The outputs of complex multiplication unit 204 appear on signal lines 206 
and 208 and represent compensated center aperture information. These 
signal lines go to summing units 210 and 212, respectively. The outputs of 
complex multiplication unit 224 appear on signal lines 226 and 228 and 
represent compensated right aperture information. These signal lines are 
fed to summing units 230 and 232, respectively. The subtraction of 
compensated center aperture information on lines 206 and 208 from 
uncompensated left aperture information I.sub.1 and Q.sub.1 on lines 86 
and 88 takes place in summing units 210 and 212, respectively. The 
subtraction of compensated right aperture information on lines 226 and 228 
from uncompensated center aperture information I.sub.c and Q.sub.c on 
lines 106 and 108 takes place in summing units 230 and 232, respectively. 
The outputs of the summing units 210 and 212 appear on signal lines 214 
and 216 and are the clutter cancelled signals I.sub.lc and Q.sub.lc. The 
outputs of the summing units 230 and 232 appear on signal lines 234 and 
236 and are the clutter cancelled signals I.sub.cr and Q.sub.cr. The 
clutter cancelled signals are presented to a detection and validation 
processor 350. 
The I.sub.lc signal on the line 214 and the Q.sub.lc signal on the line 216 
are stored in a bulk memory unit 300 while the I.sub.cr signal 234 and the 
Q.sub.cr signal 236 are fed into and stored in the bulk memory unit 302. 
These memory units are sized so that they contain 2N.sub.b bits of 
information for N.sub.r range bins by N.sub.d doppler filters. Magnitude 
detectors 310 and 314 are connected to the output of the bulk memory units 
300 and 302, respectively, and perform a conventional magnitude detection 
routine on the information stored in the bulk memory which involves taking 
the larger of the two quadrature components and adding to it one-half of 
the smaller component. This is a sufficiently accurate representation of 
true-magnitude. The outputs of the magnitude detectors are signal lines 
312 and 316 which go to a map addition unit 318. This unit adds magnitudes 
for each of N.sub.r range cells by N.sub.d doppler filters from its two 
input lines. The resultant range doppler map on signal line 320 goes to a 
CFAR type thresholding circuit 322. Average signal levels are computed in 
the range and/or doppler dimension and potential targets are declared if 
they exceed the local averages by a predetermined threshold factor. The 
outputs 326 are then stored temporarily while target angle processing is 
completed. 
The outputs of the bulk memory units 300 and 302 which represent the 
signals I.sub.lc, Q.sub.lc, I.sub.cr and Q.sub.cr are also presented to an 
angle calculation unit 334. This consists of a complex multiplier unit of 
conventional design and would typically contain four multipliers and two 
summing devices in which the inputs are complex multiplied to obtain 
complex signals from which the angle to the target can be derived. Also 
included in the angle calculation unit 334 is an arc tangent ROM which 
takes the inphase and quadrature information and derives an electrical 
phase angle. The electrical phase angle is then scaled by the 
interferometer scale factor to give the true angular field position for 
all signals in the total map of N.sub.r range bins by N.sub.d doppler 
filters and appears on signal line 336. 
An angle discrimination unit 338 is connected to the angle calculation unit 
334 and the target validation unit 328 and receives doppler and angle 
information from the angle calculation unit 334 and evaluates the angular 
information of all signals against the known angle doppler interferometric 
relationship. A discrete window is provided about the known linear 
relationship and signals falling within the discrete window will be 
identified as fixed targets and rejected. Candidate signals falling 
outside the discrete window will be identified as potential moving targets 
and will pass to a target validation unit 328 on signal line 340. The 
target validation unit correlates potential moving targets which have 
passed an amplitude threshold on signal line 326 with potential moving 
targets which have passed an angle discrimination test on signal line 340. 
The output of the target validation unit is the range, r.sub.t, and 
doppler, f.sub.t, locations of the true moving targets which have passed 
both amplitude and angle conditions on signal lines 330 and 332. Available 
on signal line 324 is amplitude information, a.sub.t, for these moving 
targets and available on signal line 336 is the angular location, 
.theta..sub.t, of these moving targets with respect to antenna boresite. 
Amplitude, range, doppler and angle information for the clutter-free region 
is provided in a conventional manner by noncoherently integrating the FFT 
ouputs (one per aperture) for detection and using the outer two apertures 
for angle measurement. Clutter/clutter free separation of the FFT outputs 
or, in other words, of the doppler filter outpus, is provided in a 
conventional manner based on aircraft velocity (quantized to a low bit 
level), radar waveform information (radar PRF and number of samples, ND), 
the antenna beamwidth and radar wavelength. 
Although this invention has been shown and described with respect to a 
preferred embodiment, it will be understood by those skilled in this art 
that various changes in form and detail thereof may be made without 
departing from the spirit and scope of the claimed invention.