Power control circuit for a radio frequency transmitter

A gain controller (130) for a radio frequency (RF) transmitter (102) controls a power level of a signal (123) transmitted within a predetermined range of output power levels. The gain controller (130) provides the first gain control signal (131) and the second gain control signal (133) responsive to an output power level control signal (150). The first gain control signal (131) controls a gain of a first variable gain stage (144) to vary the power level of the transmit signal (115) at an intermediate frequency causing the output power level of the transmit signal (123) to vary over a lower range of the predetermined range of output power levels. The second gain control signal (133) controls a gain of the second variable gain stage (120) to vary the power level of the transmit signal (121) at a radio frequency causing the output power level of the transmit signal (123) to vary over an upper range of the predetermined range of output power levels. The power control circuit (130) is advantageously utilized in a code division multiple access (CDMA) radiotelephone (100) to provide power control over an 85 dB range of power levels while minimizing sideband noise emissions, current drain, and complexity of the RF transmitter (102).

FIELD OF THE INVENTION 
The present invention relates generally to radio frequency transmitters, 
and more particularly to a power control circuit for a radio frequency 
(RF) transmitter which may be advantageously used in a code division 
multiple access (CDMA) radiotelephone. 
BACKGROUND OF THE INVENTION 
Performance requirements for a code division multiple access (CDMA) 
cellular subscriber mobile station are specified in Electronic Industries 
Association EIA/TIA/IS-95 "Mobile Station--Land Station Compatibility 
Standard for Dual-Mode Wideband Spread Spectrum Cellular System", 
published July 1993 (herein referred to as "IS-95 Standard"). The IS-95 
Standard specifies a minimum dynamic range for output power control of a 
transmit signal and a minimum amount of permitted transmit sideband noise 
emissions. 
The minimum dynamic range for output power control specified for a class 
III mobile station is 73 dB (-50 dBm to +23 dBm). When transmit gain 
tolerances are considered, the required dynamic range is 85 dB. 
The transmit sideband emissions specification calls out a dBc limit which 
is applicable at higher output power and an emission floor which is 
applicable at lower output power levels. For frequency offsets from the 
carrier frequency between 900 kHz and 1.98 MHz, the maximum emission must 
be less than the greater of 42 dBc/30 kHz relative to the desired transmit 
power in a 1.23 MHz bandwidth or both -60 dBm/30 kHz and -55 dBm/1 MHz. 
For frequency offsets from the carrier greater than 1.98 MHz, the maximum 
emission must be less than the greater of -54 dBc/30 kHz relative to the 
desired transmit power in a 1.23 MHz bandwidth or both -60 dBm/30 kHz and 
-55 dBm/1 Mhz. To produce high quality mobile stations, 10 dB of margin is 
added to the sideband emission specification. Therefore, the design target 
for the emissions floor (-60 dBm/30 kHz and -55 dBm/1 MHz) is -70 dBm/30 
kHz and -65 dBm/1 MHz. 
In other cellular systems (AMPS, NAMPS, NADC, GSM, PDC, etc.) the dynamic 
range for output power control required for mobile stations is typically 
much lower (i.e. 20 to 30 dB) than the dynamic range for output power 
control required (i.e. 85 dB) for CDMA mobile stations. In these other 
systems, the required dynamic range for output power control is typically 
provided by controlling a variable gain stage, such as a variable gain 
power amplifier (PA), which amplifies a radio frequency (RF) signal or by 
controlling a voltage controlled attenuator (VCA) which attentuates an 
intermediate frequency (IF) signal. Individually, these schemes do meet 
the dynamic range requirement for output power control or the sideband 
emission requirement for CDMA mobile stations. 
Good transmit sideband emission performance is obtained when the gain 
control circuitry for the RF signal is placed close to the antenna. 
Unfortunately, under this condition, it is not easy to realize 85 dB of 
gain control of the RF signal without providing very good shielding and 
grounding. 
A gain control range of 85 dB can be realized at a transmit signal in the 
IF range which is typically 100 to 200 Mhz. However, controlling an 85 dB 
dynamic range of power control in the IF range is disadvantageous because 
it does not optimize the sideband noise emissions requirement. To meet the 
sideband noise emissions requirement, the gain following the gain control 
stage must be minimized in order to minimize the sideband noise produced 
in the transmitter at low output power levels. This requires a higher 
output level out of the transmit IF gain stages. This implies high 
linearity for the transmit IF gain stages which results in higher current 
drain. For example, the SONY CXA3002N transmit gain control amplifier has 
85 dB of dynamic range at intermediate frequencies only, a +10 dBm output 
third order intercept point (OIP3), and a current drain of 35 mA. 
Another disadvantage of having the 85 dB gain control stage control the 
transmit signal in the IF range is the susceptibility to spurs and noise 
generated in other sections of the radio. For example, if the maximum 
output power out of the gain controlled stage is -5 dBm for adequate 
linearity and the worst case maximum gain following the gain controlled 
stage is 35 dB, the maximum noise and spurs picked up at this point must 
be less than both -105 dBm/30 kHz and -90 dBm/1 MHz to pass the emission 
floor with good margin. It is not impossible to achieve these levels, 
however, this would probably require the use of extra shielding and 
several board and/or IC revisions. Even if this degree of isolation is 
achieved, the current drain would still be higher than desired. 
Accordingly, there is a need for a power level control circuit for a RF 
transmitter which provides a wide dynamic range for output power control 
while minimizing the sideband noise emissions, the current drain, and the 
complexity of the RF transmitter.

DETAILED DESCRIPTION OF A PREFERRED EMBODIMENT 
FIG. 1 illustrates a block diagram of a radiotelephone 100 adapted for use 
in a code division multiple access (CDMA) radio frequency (RF) cellular 
telephone system. In the preferred embodiment of the present invention, 
radiotelephone 100 is a cellular radiotelephone. The radiotelephone 100 
may take many forms that are well known in the art, such as a vehicular 
mounted unit, a portable unit, or a transportable unit. According to the 
preferred embodiment of the present invention, the cellular radiotelephone 
is a code division multiple access (CDMA) cellular radiotelephone designed 
to be compatible with a CDMA cellular radiotelephone system as described 
in the aforementioned IS-95 Standard. 
The radiotelephone 100 generally includes a transmitter 102, a receiver 
104, a radiotelephone controller 105, and an antenna 106. The receiver 104 
generally includes a receive (Rx) bandpass filter 140, a signal receiver 
142, a decoder and demodulator 144 and an information sink 146. The 
radiotelephone controller 105 generally includes a microprocessor, read 
only memory, and random access memory. Generally, the receiver 104, the 
radiotelephone controller 105, and the antenna 106 are individually well 
known in the art, as taught in a radiotelephone having model #SUF1712, 
U.S. Pat. No. 5,321,847 and the aforementioned IS-95 Standard, each herein 
incorporated by reference. 
The transmitter 102 generally includes an information source 108, an 
encoder and modulator 110, a transmit (Tx) intermediate frequency (IF) 
local oscillator 112, a first variable gain stage 114, an upconversion 
stage 116, a transmit (Tx) radio frequency (RF) local oscillator 118, a 
second variable gain stage 120, a final stage 122 and a gain controller 
130. The upconversion stage 116 generally includes an upconversion mixer 
160 and a first RF bandpass filter 162. The final stage 122 generally 
includes an exciter amplifier 170, a second RF bandpass filter 172, a 
power amplifier 174, and a third RF bandpass filter 176. The transmit 
lineup for the upconversion stage 116 and the final stage 122 is described 
by example only. Other transmit lineups compatible with the present 
invention may be implemented, as well known to those skilled in the art of 
transmitter design. 
The encoder portion of 110 of the transmitter 102 and the decoder and 
demodulator 144 of the receiver 104 are generally embodied within an 
application specific integrated circuit (ASIC) as described in "CDMA 
Mobile Station Modem ASIC", Proceedings of the IEEE 1992 Custom Integrated 
Circuits Conference, section 10.2, pages 1-5, and as taught in a paper 
entitled "The CDMA Digital Cellular System an ASIC Overview", Proceedings 
of the IEEE 1992 Custom Integrated Circuits Conference, section 10.1, 
pages 1-7 (herein incorporated by reference). 
In operation, the radio transmitter 102 receives information from the 
information source 108, typically as voice or data. The information source 
provides an information signal 109 to be encoded and modulated by the 
encoder and modulator 110. The Tx IF local oscillator 112 generates a Tx 
IF local oscillator signal 111 having a frequency of 150 MHz, for example. 
The encoder and modulator 110 modulates the Tx IF local oscillator signal 
111 responsive to the information signal 109 to produce a modulated signal 
113. The center frequency of the modulated signal 113 is referred to as 
the Tx IF frequency and is, for example, 150 MHz. The modulated signal 113 
is amplified by a variable gain stage 114, having a gain controlled by a 
gain control signal 131 to produce a Tx IF signal 115. The Tx RF local 
oscillator 118 generates a Tx RF local oscillator signal 117 having a 
frequency 150 MHz higher than the desired Tx RF center frequency (for 
example, 824 to 894 MHz). The upconversion stage 116 frequency translates 
the Tx IF signal 115 from the Tx IF center frequency to the desired Tx RF 
center frequency and filters this signal using the first RF bandpass 
filter 162 to produce a first Tx RF signal 119. The first Tx RF signal 119 
is amplified by a second variable gain stage 120, having a gain controlled 
by a gain control signal 133 to produce a second Tx RF signal 121. The 
second Tx RF signal 121 is amplified and filtered by the final stage 122 
to produce the Tx output signal 123 to be transmitted via antenna 106. 
In the preferred embodiment, the first variable gain stage 114 and the 
second variable gain stage 120 are temperature compensated continuously 
variable voltage controlled attenuators. The gain transfer function for 
each gain stage, G(V), is largely a linear function of a control voltage 
over the range of operation where G(V) is the gain in dB, and V is the 
control voltage. Alternately, the variable gain stages could be 
implemented as digitally controlled attenuators or variable gain 
amplifiers as is well known to one skilled in the art. 
The receiver 104 provides a receive signal strength indication (RSSI) 
signal 148 and a closed loop correction signal 147 to the radiotelephone 
controller 105 in a conventional manner. In a conventional manner as 
described in the IS-95 Standard, the radiotelephone controller 105 
combines these two signals with a channel gain adjust signal indicative of 
the variation in transmitter and receiver gain versus frequency channel to 
produce a Tx output power control signal 150 indicative of the desired 
transmitter output power. A table of channel gain adjust signals versus 
frequency channel is determined during the manufacture of the 
radiotelephone 100 and is stored in the radiotelephone controller 105. The 
radiotelephone controller 105 provides the Tx output power control signal 
150 and a crossover threshold signal 151 to the gain controller 130. The 
crossover threshold signal 151 is an important feature of the present 
invention and will be described in further detail with reference to FIGS. 
2, 3, 4 and 5. The gain controller provides the first gain control signal 
131 and the second gain control signal 133 to the first variable gain 
stage 114 and the second variable gain stage 120, respectively, responsive 
to the Tx output power signal 150 and the crossover threshold signal 151 
to control the transmitter's output power while minimizing the sideband 
noise of the transmit output signal. The operation of the gain controller 
130 is described in more detail hereinbelow with reference to FIG. 2. 
The transmit output signal sideband noise can be expressed as a sum of the 
noise from independent noise sources amplified by the gain stages 
following the noise source. The sources of noise include the thermal noise 
of a gain stage referred to its input and external interference coupled to 
the input of a stage. The thermal noise of a gain stage referred to its 
input is defined as kT*B*(F-1) in terms of the noise figure (F), 
Boltzman's constant (k, where k=1.38*10-23 joule/K ), temperature in 
Kelvin (T) and the measurement bandwidth (B) in Hz, as is well known to 
one skilled in the art. The thermal noise referred to the input is 
hereinafter denoted as Nth. For example, at T=298K (25.degree. C.), the 
thermal noise referred to the input of a stage with a noise figure of 10 
measured in a 30 kHz bandwidth is 1.07 femtoWatts (fW) or--119.7 dBm. The 
external interference at the input to the stage may be produced by common 
mode coupling on the supplies and grounds of the stage and/or the pick up 
of radiated interference from the noise sources. The interference 
generally consists of clock harmonics and harmonics of high speed data 
signals generated by other circuits in the radiotelephone. In extreme 
cases the interference may also be caused by high power radio sources 
external to the radiotelephone such as television transmitters, for 
example. The total noise output of a gain stage having gain (G) is 
Nth+I!*G+No*G, where I is the interference picked up at the input and No 
is the output noise from the preceding stage. In transmitter 102, the 
total output noise (N) can be expressed by equation 1 (Eq. 1) shown 
hereinbelow. 
EQU N=(Nin1+Nmod)*G1*Gu*G2*Gf+ 
EQU Ninu*Gu*G2*Gf+Nin2*G2*Gf+Ninf*Gf Eq.1: 
where Gk is the gain of stage k, Nink=Nthk+Ik, Nthk is the thermal noise of 
stage k, Ik is the input interference at stage k, Nin is defined as the 
quantity (Nth+I), and Nmod is the output noise of the encoder/modulator 
110. The definition of the subscripts, k, are defined as follows: 
1-first variable gain stage 114 
u-upconversion stage 116 
2-second variable gain stage 120 
f-final stage 122 
Note that in equation 1, a reduction in the gain of the second variable 
gain stage 120 will reduce the contributions to total output noise from 
all sources except the final stage. Therefore, to minimize total output 
noise it is desirable to minimize the gain of the final stage 122 and 
maximize the range of the second variable gain stage 120. In the ideal 
approach, the entire output power dynamic range would be realized by 
controlling the second variable gain stage 120 only and the first variable 
gain stage would be eliminated. Practical considerations, however, 
preclude this for portable units, such as a CDMA radiotelephone, which are 
small and lightweight have low cost and low power dissipation and have 
high frequency and high dynamic range power control. 
In transmitter 102, the output power level (P) of the desired Tx output 
signal 123 can be expressed by the following equation 2: 
EQU P=Pmod*G1*Gu*G2*Gf 2: 
where Gk is the gain of stage k, and Pmod is the power level of the 
modulated signal 113. The definition of the subscripts, k, are the same as 
that described above in equation 1. 
A challenge in implementing the ideal approach is achieving the 85 dB 
output power control dynamic range at the RF frequency (for example, 
824-849 Mhz). The challenge becomes even greater at higher frequencies. At 
minimum output power, the input signal to the second variable gain stage 
120 is up to 85 dB greater than the output power. Some of the same issues 
discussed above regarding interference apply to the coupling of the second 
variable gain stage 120 input signal to the output of the stage. The 
coupling may be produced by common mode coupling on the supplies and 
grounds of the stage and/or the pick up at the output of a radiated input 
signal. Theoretically, this problem may be overcome using multiple stages 
at the radio frequency, good grounding practices, and shielding; however, 
this is typically impractical for a small, lightweight, low cost portable 
unit. 
According to the preferred embodiment of the present invention, a more 
practical solution is to divide the power control dynamic range 
requirements between a variable gain stage at the Tx RF frequency (824-849 
MHz), such as the second variable gain stage 120, and a variable gain 
stage at the Tx IF frequency (150 MHz ), such as the first variable gain 
stage 114. A power control scheme controls the second variable gain stage 
120 over as much of the power control dynamic range as possible and 
controls the first variable gain stage 114 over the remaining range. 
Therefore, the gain control range of the second variable gain stage 120 is 
maximized, limited only by practical considerations to 45 dB, for example. 
The gain control range of the first variable gain stage 114 is then 
designed to be at least 40 dB (i.e. 85 dB-45 dB). Equation 1, described 
hereinabove, shows that the output noise is highest at the highest gain 
settings. Therefore, it is desirable to adjust the second variable gain 
stage 120 over the high power end of the output power dynamic range, and 
adjust the first variable gain stage 114 over the lower power end of the 
output power dynamic range. 
According to the preferred embodiment, a practical power control scheme 
operation is further illustrated in FIGS. 3, 4 and 5. FIG. 3 illustrates a 
graph, combining the graphs shown in FIGS. 4 and 5, showing total gain 
versus total output power for a transmitter shown in the radiotelephone of 
FIG. 1. The graph in FIG. 3 shows the division of the transmitter gain 
control function between the first variable gain stage 114 and the second 
variable gain stage 120. Curve 300 is a plot of transmitter gain in dB 
versus transmitter output power in dBm. Dashed line 301 denotes the Gain 
crossover level. Dashed line 302 denotes the power crossover level. At 
point A on curve 300, both first variable gain stage 114 and second 
variable gain stage 120 are at their predetermined maximum gain settings. 
At point B on curve 300, the first variable gain stage 114 is set to its 
predetermined maximum gain setting and the second variable gain stage 120 
is set to its predetermined minimum gain setting. Point B on curve 300 
denotes a transition or crossover in the gain control between the second 
variable gain stage 120 and the first variable gain stage 114. At point C 
on curve 300, both first variable gain stage 114 and second variable gain 
stage 120 are at their predetermined minimum gain settings. Region 1 on 
the graph below dashed line 301 and to the left of dashed line 302 
corresponds to the low end of transmitter output power/gain. In this 
region the second variable gain stage 120 gain is held constant at its 
minimum value and the first variable gain stage 114 gain is varied to vary 
the transmitter output power. In Region 1 a 1 dB reduction in desired 
output power results in a 1 dB reduction in first variable gain stage 114 
gain and results in a 1 dB reduction in the noise contributions from the 
first term in equation 1, described hereinabove. Region 2 on the graph 
above dashed line 301 and to the right of dashed line 302 corresponds to 
the high end of transmitter output power/gain. In this region second 
variable gain stage 120 is varied to vary the transmitter output power, 
and the first variable gain stage 114 gain is held constant at its maximum 
setting. In Region 2 a 1 dB reduction in desired output power results in a 
1 dB reduction in second variable gain stage 120 gain and results in 
reduction of all output noise contributions except for the last term 
(final stage) in equation 1, described hereinabove. 
FIG. 4 illustrates a graph showing gain versus output power for the first 
variable gain stage 114. Curve 400 is a plot of first variable gain stage 
114 gain in dB vs. transmitter output power in dBm. Dashed line 401 
denotes the maximum gain level of the first variable gain stage. Dashed 
line 402 denotes the power crossover threshold level. At point A on curve 
400, the first variable gain stage 114 is clamped to its predetermined 
maximum gain setting. At point B on curve 400, the first variable gain 
stage 114 is clamped to its predetermined maximum gain setting. Point B on 
curve 400 denotes a transition or crossover in the gain control between 
the second variable gain stage 120 and the first variable gain stage 114. 
At point C on curve 400, the first variable gain stage 114 is at its 
minimum gain setting. Region 1 on the graph to the left of dashed line 402 
corresponds to the low end of transmitter output power/gain. In this 
region the second variable gain stage 120 gain is held constant at its 
minimum value and the first variable gain stage 114 gain is varied to vary 
the transmitter output power. Region 2 on the graph to the right of dashed 
line 402 corresponds to the high end of transmitter output power/gain. In 
this region the first variable gain stage 114 gain is held constant or 
clamped at its maximum setting. 
FIG. 5 illustrates a graph showing gain versus output power for the second 
variable gain stage 120. Curve 500 is a plot of second variable gain stage 
120 gain in dB vs. transmitter output power in dBm. Dashed line 501 
denotes the predetermined minimum gain level of the second variable gain 
stage. Dashed line 502 denotes the power crossover threshold level. At 
point A on curve 500, the second variable gain stage 120 is set to its 
maximum gain setting. At point B on curve 500, the second variable gain 
stage 120 is clamped to its predetermined minimum gain setting. Point B on 
curve 500 denotes a transition or crossover in the gain control between 
the second variable gain stage 120 and the first variable gain stage 114. 
At point C on curve 500, the second variable gain stage 120 is at its 
minimum gain setting. Region 1 on the graph to the left of dashed line 502 
corresponds to the low end of transmitter output power/gain. In this 
region the second variable gain stage 120 gain is held constant or clamped 
at its minimum value Region 2 on the graph to the right of dashed line 502 
corresponds to the high end of transmitter output power/gain. In this 
region second variable gain stage 120 is varied to vary the transmitter 
output power. 
Referring now to FIG. 2, FIG. 2. illustrates a block diagram of the gain 
controller 130 as shown in FIG. 1. The gain controller 130 is coupled to 
first variable gain stage 114 and the second variable gain stage 120 via 
gain control signal 131 and second gain control signal 133, respectively. 
The gain controller 130 is coupled to receive the transmit output power 
level control signal 150 and the gain crossover threshold signal 151. 
The gain controller 130 generally includes a first clamp 200, a first 
control signal processor 214, a first digital to analog converter (DAC) 
212, a second clamp 220, a second control signal processor 234, and a 
second digital to analog converter (DAC) 232. The first control signal 
processor 214 generally includes a first multiplier or scaler 202, a first 
summer or shift circuit 204, and a first predistortion circuit 210. The 
first predistortion circuit 210 generally includes a first gain control 
linearizing circuit 206 and a third summer 208. The second control signal 
processor 234 generally includes a second multiplier or scaler 222, a 
second summer or shift circuit 224, and a second predistortion circuit 
230. The second predistortion circuit 230 generally includes a second gain 
control linearizing circuit 226 and a third summer 228. 
In gain controller 130, the DAC 212 and the DAC 232 are preferably 
implemented in hardware. Further, in gain controller 130, the clamp 200, 
the clamp 220, the first control signal processor 214, and the second 
control signal processor 234 are preferably implemented in software. 
However, any allocation of hardware and software among the elements of the 
gain controller 130 can be used, as is well known to one skilled in the 
art. 
The desired output power level is provided to the gain controller 130 via 
an output power control signal 150 from the radiotelephone controller 105. 
A crossover threshold signal 151 is also provided to the gain controller 
130 from the radiotelephone controller 105. The crossover threshold signal 
151 is indicative of the output power level or transmitter gain level at 
which the control of the transmitter output power/gain crosses over 
between the first variable gain stage 114 and the second variable gain 
stage 120. The crossover threshold signal 151 is a function of frequency 
channel and is stored in the radiotelephone controller 105 as a table 
during the manufacture of the radiotelephone 100. The output power control 
signal 150 and the crossover threshold signal 151 are applied to the 
inputs of the first clamp 200 and the second clamp 220 circuits. 
Generally, the first clamp 200 and the second clamp 220 comprise a 
crossover circuit which provides continuous output power level control of 
the transmit signal between the lower range and the upper range of the 
predetermined range of the output power levels by controlling the first 
gain control signal 131 and the second gain control signal 133 responsive 
to the output power level control signal 150 and a crossover threshold 
signal 151. 
More particularly, the first clamp 200 generates a first clamp output 
signal 201 responsive to the output power control signal 150 and crossover 
threshold signal 151. The second clamp 220 generates a second clamp output 
signal 221 responsive to the output power control signal 150 and crossover 
threshold signal 151. When the output power control signal 150 is greater 
than the crossover threshold signal 151, the first clamp output signal 203 
is equal to the crossover threshold signal 151 and the second clamp output 
signal 223 is equal to the output power control signal 150. When the 
output power control signal 150 is less than the crossover threshold 
signal 151, the first clamp output signal 203 is equal to the output power 
control signal 150 and the second clamp output signal 223 is equal to the 
crossover threshold signal 151. 
The first clamp output signal 203 is processed by the first control signal 
processor 214 to produce a first control signal processor output signal 
209. The first control signal processor output signal 209 is converted 
from a digital signal to an analog signal by DAC 212 to produce gain 
control signal 131. In the preferred embodiment, the scaler 202 and the 
shifter 204 form a first linear transformer, coupled to receive the first 
clamp output signal 201 from the first clamp, for converting the first 
clamp output signal 201 to a first linear transformer output signal 205 
representative of the first gain control signal 131. The function of first 
control signal processor 214 is to transform the gain transfer function of 
the first variable gain stage 114 to a preferred gain transfer function. 
The gain transfer function of the first variable gain stage 114 is defined 
as the gain of the first variable gain stage 114 as a function of first 
control signal 131. The preferred gain transfer function for the first 
variable gain stage 114 is defined as the gain of the first variable gain 
stage 114 as a function of output power control signal 150. Preferably, 
the preferred gain transfer function is of the form, G(P)=P+a1, where G(P) 
is the gain of first variable gain stage 114 in dB, P is the output power 
control signal 150 value in dBm, and a1 is a constant. The constant al is 
also referred to as an offset. The slope of the desired transfer function 
is one so that an output power control signal 150 change of 1 dB results 
in a first variable gain stage 114 gain change of 1 dB. The slope of the 
preferred gain transfer function is also referred to as a sensitivity 
representing a change in gain to a change in the output power control 
signal. 
Likewise, The second clamp output signal 223 is processed by the second 
control signal processor 234 to produce a second control signal processor 
output signal 229. The second control signal processor output signal 229 
is converted from a digital signal to an analog signal by DAC 232 to 
produce second gain control signal 133. In the preferred embodiment, the 
scaler 222 and the shifter 224 form a second linear transformer, coupled 
to receive the second clamp output signal 221 from the second clamp 220, 
for converting the second clamp output signal 221 to a second linear 
transformer output signal 225 representative of the second gain control 
signal 133. The function of second control signal processor 234 is to 
transform the gain transfer function of the second variable gain stage 120 
to a preferred gain transfer function. The gain transfer function of the 
second variable gain stage 114 is defined as the gain of the second 
variable gain stage 114 as a function of second control signal 131. The 
preferred gain transfer function for the second variable gain stage 114 is 
defined as the gain of the second variable gain stage 114 as a function of 
output power control signal 150. 
Preferably, the preferred gain transfer function is of the form, G(P)=P+a2, 
where G(P) is the gain of second variable gain stage 120 in dB, P is the 
output power control signal 150 value in dBm, and a2 is a constant. The 
constant a2 is also referred to as an offset. The slope or sensitivity of 
the preferred gain transfer function is one so that a output power control 
signal 150 change of 1 dB results in a second variable gain stage 120 gain 
change of 1 dB. 
The first control signal processor 214 and second control signal processor 
234 circuits are preferably used because the gain transfer functions of 
the first variable gain stage 114 and second variable gain stage 120 are 
not perfectly represented by the preferred gain transfer function and/or 
are not perfectly represented by a linear equation over the entire range 
of operation. In the preferred embodiment, the first variable gain stage 
114 and the second variable gain stage 120 have gain transfer functions 
that are largely linear over their respective gain control ranges and are 
monotonically increasing with the control signal. In general these gain 
transfer functions are of the form G(V)=mV+b+d(V) where V is the gain 
control signal voltage, G(V) is the gain in dB, m and b are constants, and 
d(V) represents any deviation from the linear portion of the equation 
mV+b. The constant m represents a slope or sensitivity and b represents an 
offset. The first control signal processor 214 and second control signal 
processor 234 circuits are adjusted during manufacture so that the cascade 
of the control signal processor stage with the corresponding variable gain 
stage transfer function G(V) produces the preferred gain transfer 
functions G(P). In other words, G(V(P))=P+a1 for first control signal 
processor 214 or G(V(P))=P+a2 for second control signal processor 234. The 
operation of the first control signal processor 214 is further described 
hereinbelow. The operation of the second control signal processor 234 is 
identical to the operation of the first control signal processor 214, with 
the appropriate changes in nomenclature, and is omitted for the sake of 
brevity. 
In first control signal processor 214 the first clamp output signal 203 is 
multiplied by first multiplier 202 having gain k1 to produce a first 
multiplier output signal 203. The first multiplier output signal 203 is 
summed with constant, c1, in first summer 204 to produce a first summer 
output signal 205. The first summer output signal 205 is provided to a 
first predistortion circuit 210 to produce the first control signal 
processor output signal 209. The transfer function of the first control 
signal processor 214 is first described for the case in which the first 
variable gain stage 114 has a linear gain transfer function G(V)=ml*V+b1, 
i.e. d(V)=0. Again, the preferred gain transfer function G(V(P)) is of the 
form G(V(P))=P+a1. The desired first control signal processor 214 transfer 
function is then of the form V(P)=k1*P+c1 where k1=1/m1, and 
c1=(a1-b1)/m1. k1 and c1 are determined during the manufacture of the 
radiotelephone. In this equation V(P)=k1*P+c1, k1 represents a slope or 
sensitivity and c1 represents an offset. 
The gain transfer function of first variable gain stage 114 is 
monotonically increasing with the control signal voltage. Therefore, first 
predistortion circuit 210 can be implemented as described hereinbelow. The 
first summer output signal 205 (V1) is provided to the first gain control 
linearizing circuit 206 and the third summer 208. The first gain control 
linearizing circuit 206 produces one of a plurality of correction values 
e(V1) responsive to the first summer output signal 205. The correction 
value is summed with the first summer output signal 205 by the third 
summer 208 to produce the first control signal processor output signal 
209. The correction values, e(V1) are preferably predetermined based on 
the known characteristics of the first variable gain stage 114 gain 
transfer function and stored in a table in the first gain control 
linearizing circuit 206. The correction values e(V1) have the property 
that m1*e(V1)=-d(V1+e(V1)). The table of correction values e(V1) is 
indexed by V1. In an alternate embodiment, the first gain control 
linearizing circuit 206 function, e(V1) is implemented as a piecewise 
linear correction equation. Alternately, the correction values or the 
piecewise linear correction equation are determined and stored during the 
manufacture of the radiotelephone. 
The operation of the first control signal processor 214 is now described 
for the case in which the first variable gain stage 114 has a nonlinear 
gain transfer function G(V)=m1*V+b1+d(V). First, consider the cascaded 
transfer function of the first predistortion circuit 210 and the first 
variable gain stage 114 gain transfer function which is 
G(V1)=m1*(V1+e(V1))+b1+d(V1+e(V1)). Since e(V1) is such that 
m1*e(V1)=-d(V1+e(V1)), G(V1)=m1*V1+b1. The nonlinear case has now 
degenerated to the linear case described above, G(V)=m1*V+b1, where V is 
replaced by V1. Therefore, the desired transfer function from the first 
multiplier 202 input to the first summer 204 output is the same, and the 
constants k1 and c1 are the same (k1=1/m1, and c1=(a1-b1)/m1). 
In summary, a gain controller (130) for a radio frequency (RF) transmitter 
(102) controls a power level of a signal (123) transmitted within a 
predetermined range of output power levels. The gain controller (130) 
provides the first gain control signal (131) and the second gain control 
signal (133) responsive to an output power level control signal (150). The 
first gain control signal (131) controls a gain of a first variable gain 
stage (144) to vary the power level of the transmit signal (115) at an 
intermediate frequency causing the output power level of the transmit 
signal (123) to vary over a lower range of the predetermined range of 
output power levels. The second gain control signal (133) controls a gain 
of the second variable gain stage (120) to vary the power level of the 
transmit signal (121) at a radio frequency causing the output power level 
of the transmit signal (123) to vary over an upper range of the 
predetermined range of output power levels. The power control circuit 
(130) is advantageously utilized in a code division multiple access (CDMA) 
radiotelephone (100) to provide power control over an 85 dB range of power 
levels while minimizing sideband noise emissions, current drain, and 
complexity of the RF transmitter (102).