Accelerometer control

An accelerometer closed loop control system comprising: a capacitive accelerometer comprising a proof mass moveable relative to first and second fixed capacitor electrodes; a PWM generator to generate in-phase and anti-phase PWM drive signals with an adjustable mark/space ratio, wherein said drive signals are applied to the first and second electrodes such that they are charged alternately; an output signal detector to detect a pick-off signal from the accelerometer representing a displacement of the proof mass from a null position to provide an error signal, wherein the null position is the position of the proof mass relative to the fixed electrodes when no acceleration is applied; a PWM servo operating in closed loop to vary the mark/space ratio of said PWM drive signals in response to the error signal so that mechanical inertial forces are balanced by electrostatic forces.

This application claims priority to Great Britain Patent Application No. 1618930.0 filed Nov. 9, 2016, the entire contents of which is incorporated herein by reference.

TECHNICAL FIELD

The present disclosure relates to accelerometers, particularly capacitive accelerometers operated in closed loop.

BACKGROUND

Capacitive accelerometers are typically manufactured from silicon as micro-electromechanical systems (MEMS) devices. These small devices typically comprise a proof mass moveably mounted relative to a support or “substrate” using compliant support legs and sealed so that a gaseous medium trapped inside the device provides damping for the proof mass when it moves in a sensing direction in response to an acceleration being applied. In a capacitive accelerometer, there is typically provided a set of fixed electrodes and a set of moveable electrodes attached to the proof mass, with the differential capacitance between the electrodes being measured so as to detect deflection of the proof mass. The resonant frequency of the MEMS device is defined by the mass of the proof mass and the positive spring constant of the compliant support legs.

Such capacitive accelerometers can suffer from a “negative spring rate”, wherein the attractive electrostatic force between the proof mass and an electrode increases as the displacement of the proof mass increases. That is to say, the attractive electrostatic force gives rise to an effective “negative spring” that operates in the opposite direction to a conventional “positive spring” like that provided by the compliant support legs. This happens by virtue of Coulomb's law—as the proof mass is displaced from the null position, the attractive force between it and the electrode that is closer to the proof mass as a result of the displacement is increased while the attractive force between the proof mass and the other electrode decreases. Such a situation is clearly unstable, as a slight displacement of the proof mass may result in the attractive electrostatic force acting in the same direction as the force from the acceleration, rendering it more difficult to restore the proof mass to the null position under closed loop operation.

U.S. Pat. No. 5,142,921 describes the use of a “constant charge forcing” regime for operating a capacitive accelerometer under closed loop. In this case the “high tension” (HT) voltage applied to each electrode—i.e. the peak amplitude of the restorative voltage applied to the electrodes—changes differentially as the proof mass moves from the null point in such a way that the attractive electrostatic force is independent of position, nullifying the negative electrostatic spring constant. U.S. Pat. No. 5,142,921 uses a time multiplexed approach, wherein a fixed current is applied for a particular time so as to provide a fixed charge to the electrodes, alternating between an electrode on one side of the proof mass and an electrode on the opposite side of the proof mass. This gives rise to a differential peak voltage which is indicative of the proof mass being offset from the null position. This differential voltage signal is then used to drive a pulse width modulation (PWM) generator with an integral loop filter to restore the proof mass to the null position, wherein an unequal mark/space ratio (i.e. not 50:50) of the PWM drive signals offsets the inertial acceleration force.

However, such conventional approaches typically provide a suitable open loop gain within only a particular frequency range, outside of which the accelerometer suffers from bias shifts when under stress. It will be appreciated that the PWM drive signals are typically of a much greater frequency than the resonant frequency of the MEMS device. By way of non-limiting example only, 48 kHz PWM drive signals may be applied to a MEMS device with a resonant frequency of 3 kHz.

SUMMARY

According to a first aspect of this disclosure there is provided an accelerometer closed loop control system comprising: a capacitive accelerometer comprising a proof mass moveable relative to first and second fixed capacitor electrodes; a pulse width modulation (PWM) generator arranged to generate in-phase and anti-phase PWM drive signals with a drive frequency and an adjustable mark/space ratio, wherein said in-phase and anti-phase PWM drive signals are applied to the first and second fixed capacitor electrodes respectively such that they are charged alternately; an output signal detector arranged to detect a pick-off signal from the accelerometer representing a displacement of the proof mass from a null position to provide an error signal, wherein the null position is the position of the proof mass relative to the first and second fixed capacitor electrodes when no acceleration is applied; a PWM servo operating in closed loop arranged to vary the adjustable mark/space ratio of said in-phase and anti-phase PWM drive signals in response to the error signal so that mechanical inertial forces are balanced by electrostatic forces to maintain the operating point of the proof mass at the null position; and a differential voltage servo arranged to vary a difference in amplitude between the in-phase and anti-phase PWM drive signals in response to the error signal.

Thus in accordance with this disclosure, a capacitive accelerometer operated in closed loop may avoid the negative spring rate issues suffered by conventional accelerometers. By varying a differential voltage applied to the two capacitor electrodes in response to the error signal, the electrostatic force can be kept substantially constant regardless of any displacement of the proof mass due to an applied acceleration. This can substantially reduce if not completely remove the effects of the negative spring rate on the accelerometer.

The Applicant has appreciated that an accelerometer in accordance with this disclosure may enable the use of lower resonant frequency MEMS devices with higher open loop gain, reducing the impact of bias effects caused by mechanical stressing of the MEMS device. For example, a 3.5 kHz MEMS may undergo a displacement of 20 nm/g whereas a 1 kHz MEMS may undergo a displacement of 200 nm/g, such that a 1 nm stress error has ten times smaller an effect on bias offset for the latter device when compared to the former device, increasing the sensitivity of the device. In a conventional accelerometer, the “g range” (i.e. the acceleration range the device can measure) is directly related to the resonant frequency of the MEMS device. However, the g range of an accelerometer in accordance with this disclosure is decoupled from the resonant frequency of the MEMS, advantageously allowing the use of lower resonant frequencies while maintaining a high g range.

In some preferred examples, the differential voltage servo comprises a microcontroller arranged to produce first and second digital control words, wherein: said first digital control word is input to a first digital to analogue converter arranged to receive the in-phase PWM signal at a first reference input and output a scaled in-phase PWM signal; and said second digital control word is input to a second digital to analogue converter arranged to receive the anti-phase PWM signal at a second reference input and output a scaled anti-phase PWM signal. In such examples, these digital control words are converted by the appropriate digital to analogue converters to scale the analogue output voltages that are applied to the first and second capacitor electrodes.

In some examples, the output signal detector comprises a charge amplifier having an input connected to the proof mass and an output, said charge amplifier being arranged to produce at its output a voltage proportional to the capacitance between the proof mass and whichever of the first and second capacitor electrodes is charged at any given time. Such a charge amplifier acts so as to integrate the current applied at its input and produce an output voltage proportional to the integrated current (i.e. the charge accumulated over a period of time). As the charge built up will be proportional to the capacitance between the proof mass and the currently charged capacitor electrode, the voltage produced at the output of the charge amplifier is a measure of displacement of the proof mass.

In a preferred set of such examples, the output signal detector further comprises a demodulator having an input connected to the output of the charge amplifier, wherein said demodulator is arranged to: sample the output of the charge amplifier while the in-phase PWM signal is high so as to produce a first sample; sample the output of the charge amplifier while the anti-phase PWM signal is high so as to produce a second sample; and calculate a difference between said first and second samples; and produce the error signal, wherein the error signal is dependent on said difference.

Thus it will be appreciated that in accordance with such examples, the demodulator obtains a measure of the capacitance between the proof mass and each of the capacitor electrodes while the capacitor formed therebetween is “active” i.e. the capacitor electrode in question is receiving a high PWM signal at the time while the other capacitor electrode is receiving a low PWM signal (typically 0 V).

In some such examples, the PWM servo comprises an integral loop filter arranged to vary the adjustable mark/space ratio in response to the integral of the error signal. Thus it will be seen that in accordance with such examples, the error signal is used to drive a PWM generator with an integral loop filter in order to restore the proof mass to the null position, wherein the mark/space ratio of the PWM drive signals offsets any applied inertial acceleration force. The integral loop filter provides large DC gain and so the PWM servo responds to relatively steady accelerations. From the geometry of the device and a knowledge of the HT voltage, a linear relationship between the demodulator output signal and proof mass displacement may be obtained for any particular device. While in accordance with such examples integral control is used to set the PWM to null the inertial force, this may be extended to proportional-integral-differential (PID) control in order to optimize loop stability.

In a set of potentially overlapping examples, the differential voltage servo is arranged to vary the difference in amplitude between the in-phase and anti-phase PWM drive signals in proportion to the error signal. It will be appreciated that, in accordance with such examples, a proportional servo varies the HT voltages applied to each of the first and second fixed capacitor electrodes differentially. In contrast to integral control, the proportional control provided in such examples allows the differential voltage servo to respond to short-term accelerations such as shocks and vibrations. By way of non-limiting example only, this may be achieved using a digital to analogue convertor (DAC) on the output of an accelerometer such as the “Vesta” application specific integrated circuit (ASIC) product available from Silicon Sensing Systems Ltd. controlled by the output of such a proportional servo. Thus it is an approximate constant charge forcing scheme that may be compatible with existing accelerometers.

It will be appreciated that in the set of examples where the PWM servo provides integral control and the differential voltage servo provides proportional control, both will inevitably respond to accelerations that cause the proof mass to move. However under low frequency accelerations, the PWM servo dominates due to the integral term.

In some such examples, the demodulator is further arranged to receive a synchronisation signal, wherein the demodulator uses said synchronisation signal to produce the error signal at a predetermined frequency. In preferred examples, the predetermined frequency is the drive frequency. In such examples, the demodulator determines the difference between the first and second samples obtained from the proof mass at least once per cycle of the drive frequency and produces the error signal in time for the next application of the PWM drive signals to the capacitor electrodes.

While the error signal could be an analogue signal, in preferred examples the error signal is digital. This allows, for example, the error signal to be input directly into the digital to analogue converters so as to scale the voltages applied to the first and second capacitor electrodes.

According to a second aspect of this disclosure there is provided a closed loop method of controlling a capacitive accelerometer comprising a proof mass moveable relative to first and second fixed capacitor electrodes, the method comprising: applying in-phase and anti-phase pulse width modulation (PWM) drive signals to the first and second fixed capacitor electrodes with an adjustable mark/space ratio; detecting a pick-off signal from the accelerometer representing a displacement of the proof mass from a null position to provide an error signal, wherein the null position is the position of the proof mass relative to the first and second fixed capacitor electrodes when no acceleration is applied; operating in closed loop by varying the adjustable mark/space ratio of said in-phase and anti-phase PWM drive signals in response to the error signal so that mechanical inertial forces are balanced by electrostatic forces to maintain the operating point of the proof mass at the null position; and using the error signal so as to vary a differential voltage between the in-phase and anti-phase PWM drive signals.

The preferred and optional features described hereinabove in relation to the first aspect apply equally to the second aspect.

FIGS. 1A and 1Billustrate the operation of a conventional closed loop capacitive accelerometer2. The accelerometer2comprises a moveable proof mass4and a pair of fixed capacitor electrodes6a,6b. The two fixed capacitor electrodes6a,6bare arranged parallel to one another and are situated either side of the proof mass4. This accelerometer2is arranged to determine a linear acceleration in the sensing direction8and has the freedom to move along the sensing direction8in response to an applied acceleration.FIG. 1Aillustrates the case in which the accelerometer2is not experiencing any acceleration in the sensing direction8(e.g. the accelerometer2is at rest or at a constant velocity), and in this case the proof mass4remains in the “null position” i.e. it is equidistant from the two fixed capacitor electrodes6a,6b.

FIG. 1Billustrates the case in which the accelerometer2undergoes an acceleration in the sensing direction8. As can be seen from the figure, the proof mass4has moved closer to the first fixed capacitor electrode6aand further from the second fixed capacitor electrode6bsuch that the spacings are no longer equal. During regular operation pulse width modulated (PWM) signals are applied to the two fixed capacitor electrodes6a,6b. The PWM drive signals applied to one capacitor electrode6aare in anti-phase with the PWM drive signals applied to the other capacitor electrode6bsuch that for any given time one of the fixed capacitor electrodes6a,6bhas a “high” PWM voltage applied to it while the other has a “low” PWM voltage (e.g. 0 V) applied to it.

In such a conventional accelerometer2, these PWM drive signals represent voltages that are driven to the fixed capacitor electrodes6a,6b. Under a “constant charge” regime these PWM drive signals have a known width and height, a known current is applied for a specified amount of time such that a known charge is applied to the fixed capacitor electrodes6a,6b. The capacitance of a capacitor is equal to the stored charge divided by the voltage between the two plates and the charge is known from the properties of the PWM drive signals, so the capacitance can be determined by measuring the voltage at the proof mass4. Alternatively, under a “constant voltage” regime, a known voltage is applied to the fixed capacitor electrodes6a,6band the capacitance can instead be determined by measuring the charge at the proof mass4(typically by using a transimpedance or “charge” amplifier). As capacitance is directly proportional to the surface area of the plates and inversely proportional to the distance between them, and the surface area remains constant, the determined capacitance is a direct measure of the distance between the capacitor plates (i.e. the distance between the proof mass4and the fixed capacitor electrode6a,6breceiving the high PWM signal at any given moment in time).

FIG. 2shows a closed loop accelerometer control system10in accordance with the present disclosure. The closed loop accelerometer control system10is used to control the operation of a microelectromechanical systems (MEMS) accelerometer2which comprises a moveable proof mass4and fixed capacitor electrodes6a,6bas described previously with reference toFIGS. 1A and 1B. The control system10comprises: a charge amplifier12; a demodulator14; a mark space ratio servo16; a high tension servo18; and a pair of digital to analogue converters (DACs)20a,20b.

The mark space ratio servo16comprises a PWM loop filter22and a PWM generator24. The high tension servo18comprises a high tension loop filter26and an inverter28. The operation of these two servos16,18will be described in further detail below.

The charge amplifier12is arranged such that its input is connected to the proof mass4while its output is connected to the input of the demodulator14. As will be appreciated by those skilled in the art, a charge amplifier produces at its output a voltage that is proportional to the integral of a current at its input, i.e. the voltage42at the output is proportional to the charge at the input. Under acceleration, the voltage42produced at the output of the charge amplifier12will take two different values in each period of the fixed-height PWM drive signals30a,30b, wherein one value of the voltage corresponds to the first fixed capacitor electrode6abeing provided with the high PWM signal and the other value corresponds to the other fixed capacitor electrode6bbeing connected to a high PWM signal (assuming that the acceleration remains constant throughout the period). The reference voltage Vref(not shown inFIG. 2) used by the charge amplifier12may be 0 V, however for ease of illustration inFIGS. 3-6, it is shown to be a non-zero, positive voltage.

The demodulator14is arranged to sample the output voltage42of the charge amplifier12twice per period of the PWM drive signals, once when the in-phase, fixed height PWM signal30ais high, and once when the anti-phase, fixed height PWM drive signal30bis high (these timings also correspond to the scaled PWM drive signals40aand40bbeing high respectively, these signals40a,40bbeing discussed in further detail below). The demodulator14is arranged to output an error signal44proportional to the difference between these two samples and in this example is a digital signal. The demodulator14is also arranged to receive a reference signal32which is used to synchronise the demodulator14with the fixed height PWM drive signals30a,30b. This error signal44is provided to the mark space ratio servo16and the high tension servo18as described below.

The mark space ratio servo16is arranged to use integral control in order to vary the mark space ratio of the fixed height PWM drive signals30a,30bso as to restore the moveable proof mass4to the null position during an applied acceleration as is conventional for a closed loop accelerometer. The mark space ratio servo16is arranged such that the error signal44from the demodulator14is input to the PWM loop filter22which, based on the sign and magnitude of the error signal44, produces a control signal that is input to the PWM generator24. The PWM generator24is arranged to produce PWM drive signals30aand30bwith a fixed amplitude, however it uses the control signal from the PWM loop filter22to vary the mark space ratio of these signals i.e. the proportion of time that each of the signals takes its high value in each period. In other words the mark space ratio servo16varies the respective duty cycles of the fixed height PWM drive signals30a,30bin response to a displacement of the proof mass4.

By way of contrast, the high tension servo18is arranged to vary the amplitude of the PWM drive signals40a,40bthat are applied to the fixed capacitor electrodes6a,6b. The high tension servo18is arranged such that the error signal44produced by the demodulator14is input to a high tension loop filter26that employs proportional control to produce a pair of digital control words34a,34bthat are input to the pair of DACs20a,20brespectively. The first digital control word34ais taken directly from the output of the high tension loop filter26, while the second digital control word34bis first passed through the inverter28. The inverter28is arranged to “invert” the second digital control word34bin the sense that if the first digital control word34aincreases the second digital control word34bdecreases and vice versa. While the value of the two digital control words34a,34bmay vary in direct proportion to the error signal44produced by the demodulator14, typically they will vary from a standard, non-zero value produced by the high tension loop filter26when the proof mass4is in the null position.

The two DACs20a,20bare arranged to receive the in-phase, fixed height PWM drive signal30aand the anti-phase, fixed height PWM drive signal30bat their respective reference voltage inputs36a,36b. The DACs20a,20bare also arranged to receive the digital control words34a,34bat their respective digital inputs38a,38b. The outputs of the two DACs20a,20bare connected to the two fixed capacitor electrodes6a,6brespectively. It will be appreciated that due to their connection to the reference inputs of the DACs20a,20b(that are typically used to scale the range of the analogue output), the PWM drive signals30a,30bproduced by the PWM generator24selectively enable and disable the two DACs20a,20bat a duty cycle set by the mark space ratio servo16while the digital control words34a,34bare sampled by the DACs20a,20bin order to produce analogue, scaled PWM drive signals40a,40bthat are applied to the fixed capacitor electrodes6a,6brespectively.

While the mark space ratio servo16acts to vary the mark space ratio of the drive signals40a,40bapplied to the fixed capacitor electrodes6a,6bin order to maintain the proof mass4at the null position, the high tension servo18varies a differential voltage between the two drive signals40a,40bapplied to the capacitor electrodes6a,6bso as to ensure that the attractive electrostatic forces between the proof mass4and the fixed capacitor electrodes6a,6bremains substantially constant regardless of the displacement of the proof mass4.

Thus the system of the present disclosure uses a constant voltage regime and not a constant charge regime. However, whereas previous constant voltage systems applied the same voltage to each of the capacitor electrodes in their respective half-cycles, and applied the same voltage constantly from one cycle to the next, the control system10of the present disclosure varies a differential voltage applied to the capacitor electrodes in their respective half-cycles (i.e. they do not necessarily receive the same voltage) and varies this differential voltage from one PWM cycle to the next. This differential voltage is varied in accordance with displacement of the proof mass4such that the electrostatic force between the proof mass4and each of the capacitor electrodes6a,6bis substantially constant, regardless of the displacement of the proof mass4. As the electrostatic forces do not vary with proof mass displacement, there is no negative spring contribution from any attractive electrostatic forces while the PWM mark/space ratio provides a direct measure of the applied acceleration.

FIG. 3shows signals typical of the accelerometer control system10ofFIG. 2when the proof mass4is centred and no acceleration is being applied. In this case, the proof mass4is in the null position and the drive signals40a,40bhave a 50:50 mark space ratio. As the proof mass4is in the null position, the output signal42produced by the charge amplifier12stays at the reference voltage Vref.

FIG. 4shows signals typical of the accelerometer control system10ofFIG. 2when the proof mass4is offset towards one of the capacitor electrodes6aand no steady acceleration is being applied. This may occur when the accelerometer is subject to a sudden shock or vibration that momentarily displaces the proof mass4and causes the accelerometer control system10to act in a manner to null the movement as if it were an applied acceleration. In this case, the charge amplifier12produces a square wave output signal42that indicates that the proof mass4is offset towards the “upper” capacitor electrode6a. This square wave output signal42is defined by Vref+V1−V2, where V1is derived from the voltage output from the capacitor formed by the proof mass and the upper capacitor electrode6a, V2is derived from the voltage output from the capacitor formed by the proof mass and the lower capacitor electrode6b, and Vrefcorresponds to the voltage that the two capacitors would reach when the proof mass4is not offset from the null position as described previously with reference toFIG. 3(i.e. the mean value of the output signal42is Vrefand the signal fluctuates above and below it by the same amount). It will be appreciated that V1and V2may be the output voltages from the two effective capacitors themselves, or may instead be derived from them, e.g. proportional to them with a known scaling factor. The demodulator14receives this square wave output signal42and produces a positive error signal44. This positive error signal is used by the high tension servo18to set the control words34a,34bsuch that the drive signal40aapplied to the upper capacitor electrode6ais reduced by an amount ΔV and the drive signal40bapplied to the “lower” capacitor electrode6bis increased by the same amount ΔV. With the correct choice of ΔV determined by the high tension servo18, a constant spring constant can be achieved regardless of displacement of the proof mass4.

FIG. 5shows signals typical of the accelerometer control system10ofFIG. 2when the proof mass4is offset towards the other capacitor electrode6band no steady acceleration is being applied. In contrast to the case described with reference toFIG. 4, the charge amplifier12produces a square wave output signal42that indicates that the proof mass4is offset towards the “lower” capacitor electrode6b. The demodulator14receives this square wave output signal42and produces a negative error signal44. This negative error signal is used by the high tension servo18to set the control words34a,34bsuch that the drive signal40aapplied to the upper capacitor electrode6ais increased by an amount ΔV and the drive signal40bapplied to the “lower” capacitor electrode6bis reduced by the same amount ΔV (note that the value of ΔV will not necessarily be the same as described with reference toFIG. 4).

FIG. 6shows signals typical of the accelerometer control system10ofFIG. 2when an acceleration is applied. In this case, the mark space ratio servo16varies the mark space ratios of the respective duty cycles of the fixed height PWM drive signals30a,30b—and accordingly the respective duty cycles of the drive signals40a,40b—in response to a displacement of the proof mass4in order to counteract this displacement and restore the proof mass4to its null position. At first, the charge amplifier12produces a square wave output signal42and the demodulator14receives this square wave output signal42and produces a negative error signal44. This negative error signal is used by the high tension servo18to set the control words34a,34bsuch that the drive signal40aapplied to the upper capacitor electrode6ais increased by an amount ΔV and the drive signal40bapplied to the lower capacitor electrode6bis reduced by the same amount ΔV (note that the value of ΔV will not necessarily be the same as described with reference toFIGS. 4 and 5).

However, at time tnull, the proof mass4is restored to the null position even though it is still undergoing acceleration. As such, the output signal42of the charge amplifier12becomes constant and equal to the reference voltage Vrefdescribed previously and the error signal44produced by the demodulator44drops to 0 V. In turn, the control words34a,34bare set such that no significant differential voltage is applied to the capacitor electrodes6a,6b. The mark space ratio continues to be linear with the acceleration and so provides a direct means to measure the acceleration experienced by the accelerometer while the accelerometer itself is no longer in the negative spring regime, thus enabling the use of lower resonant frequency MEMS-based accelerometers with higher open loop gain and reducing the impact of bias effects caused by mechanical stressing of the MEMS-based accelerometer.

Thus it will be seen that the present disclosure provides an improved method for control of a closed loop capacitive accelerometer that does not suffer from a negative spring rate. By removing the effect of the negative spring rate, MEMS-based capacitive accelerometers may be implemented with lower resonant frequencies and higher open loop scale factors. This can help to alleviate the bias effects due to mechanical stressing of the MEMS. It will be appreciated by those skilled in the art that the examples described above are merely exemplary and are not limiting on the scope of the invention.