Recovery control for power converter

A device includes a first amplifier and a second amplifier. The first amplifier includes an inverting input configured to be coupled to a feedback node of an output of a power converter, a first non-inverting input configured to couple to a first voltage node, a second non-inverting input, and an output. The second amplifier includes an inverting input coupled to the output of the first amplifier, a non-inverting input coupled to a second voltage node, and an output. The device also includes a first transistor coupled to the output of the first amplifier and having a control terminal coupled to the output of the second amplifier, a capacitor coupled to a ground node and to the second non-inverting input of the first amplifier, and a current node coupled to the capacitor.

SUMMARY

In accordance with at least one example of the disclosure, a device includes a first amplifier and a second amplifier. The first amplifier includes an inverting input configured to be coupled to a feedback node of an output of a power converter, a first non-inverting input configured to couple to a first voltage node, a second non-inverting input, and an output. The second amplifier includes an inverting input coupled to the output of the first amplifier, a non-inverting input coupled to a second voltage node, and an output. The device also includes a first transistor coupled to the output of the first amplifier and having a control terminal coupled to the output of the second amplifier, a capacitor coupled to a ground node and to the second non-inverting input of the first amplifier, and a current node coupled to the capacitor.

In accordance with another example of the disclosure, a device includes a first amplifier and a second amplifier. The first amplifier includes an inverting input configured to couple to a feedback node of an output of a power converter, a first non-inverting input configured to couple to a first voltage node, a second non-inverting input, and an output. The second amplifier includes an inverting input coupled to a second voltage node, a non-inverting input coupled to the output of the first amplifier, and an output. The device also includes a first transistor coupled to the output of the first amplifier and a ground node and having a control terminal coupled to the output of the second amplifier, a capacitor coupled to the ground node and to the second non-inverting input of the first amplifier, and a second transistor coupled to the capacitor and to the ground node. A control terminal of the second transistor is coupled to the control terminal of the first transistor. The device also includes a current node coupled to the capacitor.

In accordance with yet another example of the disclosure, a system includes a pulse-width modulator (PWM) circuit for a power converter configured to provide an output voltage and a first amplifier configured to generate an output to control the PWM circuit based on a feedback voltage derived from the output voltage, a reference voltage, and a soft start voltage across a capacitor. The system also includes a second amplifier configured to generate an output to discharge the capacitor based on the output of the first amplifier and a clamping voltage.

DETAILED DESCRIPTION

Switching power converters convert power from a direct current (DC) or alternating current (AC) source to a DC load, such as a personal electronic device. In some cases, the power converter becomes overloaded, for example where a short circuit exists in the load and is drawing too much current (e.g., above a maximum current limit of the power converter). In other cases, the power converter experiences a brownout, where the input voltage to the power converter sags, causing the output voltage of the power converter to follow the input voltage sag. In either case (generically referred to as a fault), a feedback voltage of the power converter, which is the output voltage or a fraction of the output voltage, becomes unacceptably low. Once the fault is resolved, feedback mechanisms result in a delay in restoring the output voltage to its nominal level, or result in an overshoot to the output voltage leading to an inrush current spike, all of which are not desirable.

In one example, an offset in the feedback mechanism causes a soft start voltage, to which the feedback voltage is compared, to be higher than expected (e.g., higher than the feedback voltage following a fault), resulting in a large initial current demand by the power converter that causes an inrush current spike. In another example, the offset in the feedback mechanism causes the soft start voltage to be lower than expected (e.g., lower than the feedback voltage following a fault), resulting in a delay in beginning regulation as the soft start voltage catches up with the feedback voltage, for example as a capacitor that provides the soft start voltage is charged. These problems are exacerbated because a low output voltage (and corresponding feedback voltage), causes feedback control circuits to operate in an open-loop fashion, where the power converter is controlled to deliver a maximum current regardless of the particular output and feedback voltages. For example, since the output and feedback voltage are too low (e.g., out of range), the feedback mechanism controls the power converter to deliver a maximum current until the feedback voltage is approximately equal to the rising soft start voltage allowing proper regulation of the power converter. In either case, the presence of the offset in the feedback mechanism introduces inaccuracies regarding how the power converter will be regulated following a fault, which is also not desirable.

Examples of the present disclosure include a feedback control circuit for a power converter that regulates a capacitor that provides the soft start voltage during a fault to facilitate a soft start recovery of the output voltage once the fault is resolved. During a fault when a feedback voltage is low, an error amplifier of the feedback control circuit will attempt to drive a pulse width modulator (PWM) circuit coupled to the error amplifier output to increase a current provided by the power converter. However, a clamping amplifier also coupled to the error amplifier output limits the voltage of the error amplifier output to a clamping voltage by sinking current from the error amplifier output when it exceeds the clamping voltage. When the clamping amplifier operates to clamp the error amplifier output in this manner, one or more transistors are also operated to discharge the capacitor that provides the soft start voltage. Thus, the output of the error amplifier regulates the soft start voltage in a closed-loop manner, which avoids the need for a separate circuit to determine whether to regulate the soft start voltage, saving power and reducing complexity. Subsequently, when the fault condition ceases, since the capacitor that provides the soft start voltage is discharged, the feedback control circuit facilitates a normal soft start, which results in a smooth recovery to the output voltage without delay and avoiding an inrush current spike.

FIG. 1depicts an example circuit to address the challenges described above. InFIG. 1, a pulse width modulator circuit (PWM)116provides power conversion by modulating the output current command or duty cycle of its output based on an input received from an error amplifier102. The PWM circuit116represents the power converter for purposes of simplicity and the scope of this disclosure is not limited to a certain power converter topology, but rather addresses the problems noted above when an output voltage (or associated feedback voltage) for power converter control becomes unacceptably low, for example due to a fault. A power source101is coupled to the PWM circuit116and provides an input voltage (VIN, which is also used to refer to the node at VIN) as an input voltage for a power converter topology. An output inductor118is coupled to PWM116and a node at an output voltage (VOUT, which is also used to refer to the node at VOUT). An output capacitor120is coupled between VOUT and a ground node. A voltage divider comprising resistors122and124is also coupled between VOUT and ground. A feedback voltage (VFB, which is also used to refer to the node at NFB) is generated at the node between the resistors122and124.

A feedback control circuit100is coupled to the PWM circuit116. The feedback control circuit100includes the aforementioned error amplifier102, which in this example is a 3-input error amplifier102. The error amplifier102has two non-inverting inputs (the lower value of which controls) and one inverting input. For example, when a first of the non-inverting inputs is at a lower voltage than a second of the non-inverting inputs, the output of the error amplifier102is based on a comparison of the first of the non-inverting inputs and the inverting input. In another example, when the second of the non-inverting inputs is at a lower voltage than the first of the non-inverting inputs, the output of the error amplifier102is based on a comparison of the second of the non-inverting inputs and the inverting input. The inverting input of the error amplifier102is coupled to VFB. One of the non-inverting inputs of the error amplifier102is coupled to a node at a reference voltage (VREF, which is also used to refer to the node at VREF), while the other non-inverting input of the error amplifier102is coupled to a node at a soft start voltage (VSS, which is also used to refer to the node at VSS). In an example, VREF is generated by a separate reference circuit (not shown for simplicity) and has a value corresponding to VFB when VOUT is regulated to a particular level. A capacitor112is coupled to VSS and to a ground node, and a current node114charges the capacitor112. The error amplifier102compares VFB to the lesser of VREF and VSS, and the output of the error amplifier102is proportional to the difference between the inverting terminal and the lesser of the non-inverting terminals.

As explained above, the output of the error amplifier102controls or modulates the output current command level or duty cycle of the PWM circuit116to increase or decrease the amount of output current (IL, through the output inductor118) provided to a load coupled to VOUT. For example, an increase in the voltage of the output of the error amplifier102causes the PWM circuit116to increase its output current command level or duty cycle, resulting in an increase in output current. Similarly, a decrease in the voltage of the output of the error amplifier102causes the PWM circuit116to decrease its output current command level or duty cycle, resulting in a decrease in output current.

The feedback control circuit100also includes a clamping amplifier104, which comprises a non-inverting input and an inverting input. The inverting input of the clamping amplifier104is coupled to the output of the error amplifier102. The non-inverting input of the clamping amplifier104is coupled to a node at a clamping reference voltage (VHC, which is also used to refer to the node at VHC). In an example, VHC is also generated by a separate reference circuit (not shown for simplicity) and has a value corresponding to the maximum current that the power converter can deliver or the maximum duty cycle ratio that the power converter can tolerate. For example, when the error amplifier102output is equal to VHC, the PWM circuit116is controlled to deliver the maximum current that the power converter can deliver. The clamping amplifier104compares the output of the error amplifier102to VHC, and the output of the clamping amplifier104is proportional to the difference between VHC and the output of the error amplifier102.

The feedback control circuit100also includes transistors106,108,110, which in this example comprise metal-oxide-semiconductor field-effect transistors (MOSFETs) having a gate, a source, and a drain. In this example, the transistor106comprises a p-type MOSFET while the transistors108,110comprise n-type MOSFETs. An output of the clamping amplifier104is coupled to the gate of the p-type MOSFET106, while the output of the error amplifier102is coupled to the source of the p-type MOSFET106. The drain of the p-type MOSFET106is coupled to the drain of the n-type MOSFET108, which is also coupled to the gate of the n-type MOSFET108. The source of the n-type MOSFET108is coupled to a ground node. The gate of the n-type MOSFET108is also coupled to the gate of the n-type MOSFET110. The source of the n-type MOSFET110is coupled to a ground node, while the drain of the n-type MOSFET110is coupled to the capacitor112. In an example the n-type MOSFETs108,110together act as a current mirror.

FIG. 2depicts another example of a feedback control circuit200to address the challenges described above. The PWM circuit116, inductor118, capacitor120, and resistors122,124are similar to those described above with respect toFIG. 1. Additionally, the error amplifier102, the capacitor112, and the current node114are similar to those described above with respect toFIG. 1. The clamping amplifier204is also similar to the clamping amplifier104, although its terminals are switched as will be explained further below.

The feedback control circuit200also includes transistors206and210, which in this example comprise n-type MOSFETs having a gate, a source, and a drain. An output of the clamping amplifier204is coupled to the gate of the n-type MOSFET206, while the output of the error amplifier102is coupled to the drain of the n-type MOSFET206. The source of the n-type MOSFET206is coupled to a ground node. The gate of the n-type MOSFET206, and thus the output of the clamping amplifier204, is also coupled to the gate of the n-type MOSFET210. The source of the n-type MOSFET210is coupled to a ground node, while the drain of the n-type MOSFET210is coupled to the capacitor112. A resistor207and a capacitor209are coupled to the output of the error amplifier102and to a ground node and serve as a compensation network to provide additional stability to the output voltage of the error amplifier102.

FIG. 3shows a set of waveforms300and the function of both feedback control circuits100,200is explained with respect to these waveforms300. The VIN waveform corresponds to the voltage at VIN. The VOUT waveform corresponds to the voltage at VOUT. The VSS waveform corresponds to the voltage across the capacitor112that is provided as an input to the error amplifier102. The VFB waveform corresponds to the voltage at VFB. The VCOMP waveform corresponds to the output voltage of the error amplifier102. The VREF and VHC levels represent the reference voltages applied to the error amplifier102and the clamping amplifier104,204, respectively.

Although not depicted inFIG. 3, prior to startup of the power converter, the voltage across the capacitor112(VSS) is zero. Thus, upon startup of the power converter, the current node114begins to charge the capacitor112, causing VSS to increase linearly. VREF is provided by a reference circuit, and thus is provided as an input to the error amplifier102upon startup. Initially, VSS is lower than VREF, and thus causes the error amplifier102output to regulate VFB according to VSS. VFB ramps up according to the slew rate of VSS, which is in turn controlled by the current node114and the capacitor112. The controlled ramp up of VSS prevents an inrush current, for example to the output capacitor120. Depending on the capacitance of the capacitor112, VSS increases beyond VREF in some examples as the capacitor112is charged, and thus VREF takes control and causes the error amplifier102output to regulate VFB according to VREF.

During normal operation at time302, in an example the capacitor112is charged to produce a voltage VSS that is higher than VREF, such that VREF remains the lower of the two voltages supplied to the non-inverting terminals of the error amplifier102(e.g., the value of the capacitor112is selected to achieve such voltage levels). When a fault is not present, VFB is tracking VREF through the power stage control feedback loop (e.g., including the error amplifier102, the PWM circuit116, the output inductor118, the output capacitor120, and the voltage divider122,124), and thus the output of the error amplifier102is biased between its minimum and maximum potential. As a result, the PWM circuit116converts the error amplifier102output to influence its power converter output current or duty cycle to provide more or less current as needed, based on the output load current requested through feedback of VFB, which drops in response to an increased output load due to the capacitor120providing the instantaneous load transient, and which rises in response to a decreased output load due to a transient overcharging of VOUT by the PWM circuit116).

After time302, VIN begins to drop (e.g., a brownout) and at time304, VIN falls below VOUT, which causes VOUT to fall as well. When such a fault occurs, VFB as a function of VOUT also decreases to a low value (below VREF and VSS), causing the output of the error amplifier102to increase.

Referring to the example ofFIG. 1, when the output of the error amplifier102begins to exceed its maximum allowed clamping voltage (VHC), the clamping amplifier104generates a decreased output sufficient to turn the p-type MOSFET106on. The current through the p-type MOSFET106is mirrored through the current mirror comprising the n-type MOSFETs108,110, which discharges the capacitor112. Once the capacitor112is discharged such that VSS is approaching VFB, the output current of the error amplifier102will start to decrease. The clamping amplifier104then reduces the current through the p-type MOSFET106such that the n-type MOSFET110discharges current at approximately the same rate as provided by the current node114, and VSS approaches VFB. Thus, the output of the error amplifier102regulates the capacitor112that provides VSS in a closed-loop manner, which avoids the need for a separate circuit to determine whether to regulate VSS, saving power and reducing complexity.

Referring to the example ofFIG. 2, as explained above the terminals of the clamping amplifier204are switched relative toFIG. 1. As a result, when the output of the error amplifier102begins to exceed its maximum allowed clamping voltage (VHC), the clamping amplifier204generates an increased output sufficient to turn the n-type MOSFET206on. The n-type MOSFET210is also turned on, which discharges the capacitor112. Once the capacitor112is discharged such that VSS is approaching VFB, the output current of the error amplifier102will start to decrease. The clamping amplifier104then reduces the current through the n-type MOSFETs206,210such that the n-type MOSFET210discharges current at approximately the same rate as provided by the current node114, and VSS approaches VFB. Similar to above, the output of the error amplifier102regulates the capacitor112that provides VSS in a closed-loop manner, which avoids the need for a separate circuit to determine whether to regulate VSS, saving power and reducing complexity.

Referring to the example ofFIG. 1, when VIN begins to rise at time306, VFB recovers to a level just above the lesser of the non-inverting terminals of the error amplifier102(VSS during recovery). When VFB briefly increases above VSS, the output of the error amplifier102(VCOMP) decreases below VHC with minimal delay. The output of the error amplifier102also decreases the power stage current level to reduce VOUT overshoot, and the clamping amplifier104turns the p-type MOSFET106fully off, which in turn turns the n-type MOSFETs108,110of the current mirror off. The current node114then gradually charges the capacitor112with a controlled slew rate, and VSS gradually increases, causing a gradual increase in the output of the error amplifier102, which then increases the current of the power stage to charge VOUT in such a manner such that VFB tracks VSS. As VSS is pre-regulated close to VFB during the fault, when the fault ceases, a normal soft start takes place, resulting in a smooth recovery to the output voltage without delay.

Referring to the example ofFIG. 2, when VIN begins to rise at time306, VFB recovers to a level just above the lesser of the non-inverting terminals of the error amplifier102(VSS during recovery). When VFB briefly increases above VSS, the output of the error amplifier102(VCOMP) decreases below VHC with minimal delay. The output of the error amplifier102also decreases the power stage current level to reduce VOUT overshoot, and the clamping amplifier204turns the n-type MOSFETs206,210fully off. The current node114then gradually charges the capacitor112with a controlled slew rate, and VSS gradually increases, causing a gradual increase in the output of the error amplifier102, which then increases the current of the power stage to charge VOUT in such a manner such that VFB tracks VSS. As VSS is pre-regulated close to VFB during the fault, when the fault ceases, a normal soft start takes place, resulting in a smooth recovery to the output voltage without delay.

Although the example ofFIG. 3is described with respect to a brownout-type fault (e.g., sagging input voltage), VOUT and thus VFB could also drop in response to overloading, such as a short circuit at a load coupled to VOUT. In this example, the PWM circuit116is unable to provide sufficient output current, causing VOUT and thus VFB to decrease. The behavior of the feedback control circuits100,200is similar to as described above in such an example.

In the foregoing discussion and in the claims, the terms “including” and “comprising” are used in an open-ended fashion, and thus should be interpreted to mean “including, but not limited to . . . .” Also, the term “couple” or “couples” is intended to mean either an indirect or direct connection. Thus, if a first device couples to a second device, that connection may be through a direct connection or through an indirect connection via other devices and connections. Similarly, a device that is coupled between a first component or location and a second component or location may be through a direct connection or through an indirect connection via other devices and connections. An element or feature that is “configured to” perform a task or function may be configured (e.g., programmed or structurally designed) at a time of manufacturing by a manufacturer to perform the function and/or may be configurable (or re-configurable) by a user after manufacturing to perform the function and/or other additional or alternative functions. The configuring may be through firmware and/or software programming of the device, through a construction and/or layout of hardware components and interconnections of the device, or a combination thereof. Additionally, uses of the phrases “ground” or similar in the foregoing discussion are intended to include a chassis ground, an Earth ground, a floating ground, a virtual ground, a digital ground, a common ground, and/or any other form of ground connection applicable to, or suitable for, the teachings of the present disclosure. Unless otherwise stated, “about,” “approximately,” or “substantially” preceding a value means +/−10 percent of the stated value.