Synchronization signal generating device

A synchronization signal generation device includes a circuit that enables a phase difference between a synchronization signal and an input signal with intermittent edges to be arbitrarily and continuously varied. The synchronization signal generating device is of the second order phase locked loop and has a phase detector with the following elements: a circuit for generating pulses with widths corresponding to the phase difference between the input signal and the synchronization signal only upon occurrence of an edge of the input signal; a circuit for generating pulses with a constant width only upon occurrence of an edge of the input signal or the synchronization signal; a variation circuits which varies one or both of the amplitudes of the aforesaid pulses; and a combining circuit which adds or subtracts the pulses from the variation circuits to derive a phase comparison signal.

FIELD OF THE INVENTION 
This invention relates to a device which generates a synchronization signal 
from digital input signals and, more particularly, it concerns a 
synchronization signal generating device which generates a synchronization 
signal of any arbitrary phase from signals, the edges of which occur 
intermittently, such as digital data signals. 
BACKGROUND OF THE INVENTION 
In the process of reading digital data from a digital recording device, the 
bit error rate changes with the phase relationship of the digital data 
signal and the synchronization signal. Therefore, in digital recording 
devices, it is necessary to measure the changes in bit error rate with 
changes in the phase of the synchronization signal and to investigate the 
allowable range of phase difference and jitter of the synchronization 
signal. 
This invention is used especially in the measurement of bit errors of 
digital signals which are read from digital recording devices, etc. 
Moreover, by means of this invention, which can vary the phase of the 
synchronization signal arbitrarily, it is possible to optimize the phase 
of a synchronization signal so that bit errors which occur during read-out 
of a digital recording device are minimized. 
Hereafter, an example will be discussed of a change in the bit error rate 
due to changes in the phase of the synchronization signal, by using FIGS. 
6a-6e and 7a-7e. The signal F is the original digital signal comprising 
information which has been recorded by a recording device; the signal read 
out by the digital recording device is ordinarily in a distorted waveform, 
as shown by waveform A (FIG. 6b). B is the synchronization signal (FIG. 
6c). The small vertical lines shown in signal waveform F are not signals, 
but signs which show the synchronization positions. Moreover, the dots 
shown in Signal A are signs which show the sampling positions, as 
discussed below. 
In FIGS. 6a-6e, there is no phase difference between original digital 
signal F, i.e., read-out signal A, and synchronization signal B. 
Therefore, if signal A is sampled at the rising edge of synchronization 
signal B, the correct value of signal A is sampled and a discrete signal G 
is obtained (the dots in FIG. 6d). This signal G is converted to binary 
digits by a threshold value J, shown by the horizontal dotted line in FIG. 
6d, and digital signal H (FIG. 6e) is obtained. Signal H includes no 
errors and is similar to original digital signal F. 
However, if as shown in FIGS. 7a-7e, there is a large phase difference 
between read-out signal A and synchronization signal B, the timing of the 
sampling during data reproduction is displaced from the original timing, 
and correct values are not sampled. As a result, when sampled signal G is 
converted to binary digits by threshold value J, sample K gives rise to a 
bit error and reproduced signal H contains an error. In FIGS. 7a-7e, even 
if no errors arise, the margin between the samples with values near the 
threshold and noise, in an actual circuit, are small, and the probability 
of bit errors arising is high. 
As discussed above, the bit error rate changes with the displacement of the 
sample timing, due to the phase difference of the synchronization signal. 
In order to measure the changes in the bit error rate with respect to 
changes in the phase of the synchronization signal, it must be possible to 
generate a synchronization signal from the input digital signal and to set 
the phase of the synchronization signal arbitrarily. 
In generating a synchronization signal from an input signal, a second order 
phase locked loop is generally used, such as shown in FIG. 8. In the phase 
locked loop, the output of a voltage-controlled oscillator 8 is 
synchronization signal B. Input signal A and synchronization signal B are 
compared by a phase detector 19, and a current E, corresponding to the 
phase difference of the two signals, is output from phase detector 19. 
Current E and an offset current L (with a constant magnitude) from a 
direct current source 20 are combined by current-joining means 21 and 
input into an integrator 7. The current is integrated in integrator 7 and 
fed back to the input of voltage-controlled oscillator 8, completing the 
feed-back loop. 
Due to the negative feed-back of the loop, the oscillation frequency and 
phase of voltage-controlled oscillator 8 finally settle and become 
constant; and input signal A and synchronization signal B are phase-locked 
with a certain phase difference. Since the control voltage of 
voltage-controlled oscillator 8, i.e., the output voltage of integrator 7, 
is held constant by feed-back in the state in which the phases are locked, 
the input of integrator 7 is held at zero. That is, the phases are locked 
in a state in which the output current of phase detector 19 and the offset 
current L cancel each other. 
When offset current L from direct current source 20 is changed to a new 
value, the feed-back of the loop acts in the direction to make the input 
of integrator 7 zero; hence, the output of phase detector 19 changes in 
such a way that the new value of offset current L is canceled out, and the 
loop locks to the new phase. Therefore, by controlling the magnitude of 
direct current L, it is possible to change the phase difference of input 
signal A and synchronization signal B to any arbitrary value. The 
conventional phase locked loop controls the phase of the synchronization 
signal to any arbitrary value by this kind of method. 
This method presupposes that input signal A is a repetitive waveform, i.e., 
the edge of the input signal A comes in a regular manner. In cases of 
input signals with edges which occur intermittently, like digital signals, 
this method does not operate well. The reasons for this will be discussed 
below, by means of examples shown in FIGS. 9 and 10. Furthermore, in the 
diagrams in which a digital input signal A, discussed below, is shown, the 
waveform is not drawn in a distorted manner, as in FIG. 6, for purposes of 
simplification. 
Output signal E of phase detector 19, in FIGS. 9 and 10, consists of pulses 
with widths which extend from the rising or falling edge of digital input 
signal A to the rising edge of synchronization signal B. The feedback acts 
in such a way that a phase difference arises by which the mean value of 
signal E and the offset current L cancel each other. That is, the phase 
difference becomes such that the areas shown by the diagonal hatching in 
the figure become equal. 
FIG. 9 is an example in which the rising or falling edges of digital signal 
A alternatively come once each period of the synchronization signal. In 
FIG. 10, on the other hand, an example is shown in which the rising or 
falling edges of the digital signal alternatively come once in two periods 
of the synchronization signal. Comparing the outputs E of phase detector 
19, the widths of the pulses in FIGS. 9 and 10 are the same, but the 
numbers of pulses are in the ratio of 1 to 2. Therefore, the mean current 
of E in FIG. 10 is 1/2 that of FIG. 9. In order to control the phase 
difference so that it is the same as in FIG. 9, the direct current offset 
current L of FIG. 10 must be made 1/2. In FIG. 10, the current L is drawn 
as 1/2. Conversely, if the magnitude of the offset current L is not 
changed, the feedback acts in a direction such that the pulse width of E 
broadens so that a phase difference is caused which cancels this, and the 
phase difference doubles between the digital signal and the 
synchronization signal. 
Thus, the output E of phase detector 19 appears only when an edge of the 
input signal occurs, so that, in the case of an input signal the edges of 
which come intermittently, output E of phase detector 19 varies with the 
frequency of the edges, even though the phase difference does not vary. 
Therefore, in a case in which the offset current is constant, the phase 
difference varies in the opposite manner. 
In the case of a digital signal input, the frequency with which the edges 
of input signal A occur is completely random; therefore, even if the 
magnitude of offset current L is constant, the phase of synchronization 
signal B varies randomly and cannot be kept a constant value. 
In the case of an input signal, the edges of which come intermittently, 
such as a digital signal, the phase locked loop is settled in a stable 
manner and a synchronization signal with a constant phase is obtained only 
when the offset current L is zero. That occurs when the edges of input 
signal A and the edges of synchronization signal B coincide with each 
other. The output E of the phase detector 19 is also always zero. 
Thus, in the means of FIG. 8, the phase of the synchronization signal 
cannot be changed to an arbitrary value with respect to input signals, the 
edges of which come intermittently, such as digital signals. 
Therefore, as a means for generating a synchronization signal which has a 
phase difference with respect to an input signal, the edges of which come 
intermittently, a method which uses a phase locked loop and a delay 
circuit, as shown in FIG. 11, has been proposed. Phase locked loop 22 of 
FIG. 11 is a phase locked loop of the kind shown in FIG. 8 above; it holds 
at zero the phase difference between the input signal and the 
synchronization signal. If its output is output through the delay circuit 
23 as synchronization signal B, synchronization signal B is delayed by a 
quantity which is determined by the characteristic of delay circuit 23 
with respect to the output of phase locked loop 22. Therefore, a 
synchronization signal B can be obtained which has a constant, arbitrarily 
chosen phase difference from input signal A. However, this phase 
difference is a constant, corresponding to the characteristic of delay 
circuit 23; it is not possible to make it variable. Conversely, in order 
to make the phase variable, a delay circuit with a variable delay time is 
needed, which is hard to realize in practice. 
As a means for outputting a synchronization signal with a variable phase 
with respect to input signals with intermittent edges, such as digital 
data, there is a method which uses a phase locked loop with a frequency 
divider, as shown in FIG. 12. 
In FIG. 12, divider 24 divides the output of voltage-controlled oscillator 
8 into n parts, and n signals are output, each of which has a phase which 
differs by 360/n degrees from the previous one. If one specific signal 
among these n signals is returned to phase detector 19, the signal is 
phase-locked, with a phase difference of zero with respect to input signal 
A. Therefore, in the phase-locked state, the frequency of the signal is 
maintained so that it is equal to that of input signal A, and the phase 
difference becomes zero; hence, the output of the voltage-controlled 
oscillator 8 is a frequency n times that of input signal A, and it becomes 
a signal which is synchronized with the input signal. The output of 
divider 24, which divides by n, is equal to the frequency of the input 
signal A, and becomes n signals, such that their phase differences are 
integral multiples of 360/n degrees. 
By selecting a suitable signal from these n signals by means of selection 
switch 25, a signal can be obtained which is synchronized with the input 
signal A and the phase of which is an integral multiple of 360/n degrees, 
and is output as the synchronization signal. For example, if n is 8, 
synchronization signals with phases which can be varied by intervals of 
360/8=45 degrees can be generated. 
Since the steps by which the phase is varied are limited to 360/n degrees, 
it is necessary to make the division ratio n of the divider large when one 
wishes to vary the phase very precisely. If the division ratio n becomes 
large, divider 24 and selection switch 25 become complex. Moreover, if the 
division ratio n is made large, the frequencies handled by 
voltage-controlled oscillator 8 and divider 24 also become high, which 
leads to technical difficulties. Moreover, even if the division ratio n 
can be made large, the phase of the synchronization signal can only be 
varied in steps, and not continuously. 
The prior art thus made it possible to vary the phase of a synchronization 
signal to arbitrary values, in cases in which the input signals were 
repeated waveforms. But the phase of a synchronization signal with respect 
to an input signal, in cases of signals with edges which come 
intermittently, such as digital signals, was either constant at a specific 
value, or even if it was variable, it could only be varied in certain 
steps, and could not be varied continuously to any arbitrary phase. 
This invention has the purpose of solving the aforementioned problems by 
providing a synchronization signal generating device which has a means by 
which the phase difference of the synchronization signal can be varied 
arbitrarily and continuously, even with respect to input signals with 
intermittent edges. 
SUMMARY OF THE INVENTION 
This invention, as shown in the block diagram of FIG. 1, is one in which 
the phase detector of a second order phase locked loop of a negative 
feed-back type comprises a phase detector 1, which consists of a means 2 
which generates pulses with widths corresponding to phase differences 
between an input signal and a synchronization signal only when the edges 
of the input signal arrive; a means 3 which generates pulses of constant 
width only when the edges of the input signal or the synchronization 
signal arrive; means 4 and 5 which vary the amplitudes of one or both of 
these pulses; and a means 6 which subtracts or adds the pulses.

EXPLANATION OF SYMBOLS 
1: Phase detector of this invention; 
2: Means for generating pulses with widths corresponding to the phase 
difference; 
3: Means for generating pulses with constant width; 
4: Means for changing the amplitude of the pulse; 
5: Means for changing the amplitude of the pulse; 
6: Means for subtracting 2 pulses; 
7: Integrator; 
8: Voltage-controlled oscillator; 
9: Flip-flop; 
10: Flip-flop; 
11: AND gate; 
12: Variable-gain amplifier; 
13: Variable-gain amplifier; 
14: Subtracter amplifier circuit; 
15: Low-pass filter; 
16: Low-pass filter; 
17: Low-frequency amplifier; 
18: Low-frequency amplifier; 
19: Phase detector; 
20: Direct-current offset current source; 
21: Current junction means; 
22: Phase locked loop; 
23: Delay circuit; 
24: Divider; 
25: Selection switch; 
A: Input signal to phase locked loop (read-out digital signal); 
B: Synchronization signal; 
C: Pulse of width corresponding to the phase difference of A and B; 
C': Pulse which varies the amplitude of pulse C; 
D: Pulse with constant width; 
D': Pulse which varies the amplitude of pulse D; 
E: Output of phase detector; 
F: Original digital signal; 
G: Sampled signal; 
H: Reproduced digital signal; 
J: Threshold; 
K: Sample causing error; 
L: Direct-current offset current. 
DETAILED DESCRIPTION OF THE INVENTION 
A block diagram of a circuit embodying the invention is shown in FIG. 1, 
and a timing diagram which explains the operation of FIG. 1 is shown in 
FIG. 2. In FIG. 1, a phase detector 1 is made up of pulse generating means 
2 and 3, amplitude varying means 4 and 5, and a means 6 for subtracting 
two pulses. A phase locked loop is made up of phase detector 1, an 
integrator 7, and a voltage-controlled oscillator 8. Furthermore, the same 
reference numbers are given to the structural elements with the same 
functions as in the conventional technology. 
Pulse generating means 2 outputs a pulse C, only upon the rising edge of 
input signal A, pulse C having a width which corresponds to the phase 
difference between input signal A and a synchronization signal B. Pulse 
generating means 3 outputs a pulse D with a constant width only upon the 
rising edge of input signal A or synchronization signal B. In FIG. 2, in 
order to make the operation easier to understand, pulse D rises at the 
rising edge of synchronization signal B which comes after the rising edge 
of input signal A. However, the temporal position of pulse D may be any 
arbitrary one, as long as it is within one period of the synchronization 
signal. 
The amplitudes of the pulses produced in pulse generating means 2 and 3 
pass through the means 4 and 5 which change them, respectively, and pulses 
C' and D' are obtained (in the figure, the case of a gain of one is shown 
as an example). Pulses C' and D' are input into means 6 for subtracting 
two pulses and the difference E of pulses C' and D' is output from phase 
detector 1. Since negative feed-back is used to drive to zero the input of 
integrator 7, the phase locked loop locks in a state in which the pulse 
areas of pulses C' and D' are equal (the diagonal hatching of E in FIG. 
2). 
Now, if we assume that the amplitudes of pulses C and D are equal, and the 
changes of amplitude with respect to pulses C and D by means for changing 
amplitudes 4 and 5 are zero, the amplitudes of pulses C' and D' become 
equal. In this case, as shown in FIG. 2, the pulse width of pulse D', 
which has a constant width, yields the phase difference between the input 
signal and the synchronization signal. 
In the aforementioned phase-locked state, if the amplitude of the pulse D' 
is varied, as shown in FIG. 3, the pulse area of D' changes, and the 
equality between the areas of the pulses C' and D' is destroyed. Since the 
feedback of the phase locked loop acts in such a way as to cancel out this 
change, that is, to change the pulse area of C' so that they are equal, 
the phase difference between the input signal A and the synchronization 
signal B changes until the pulse areas of C' and D' become equal. By 
varying the amplitude of pulse D' continuously in this manner, the phase 
difference between the input digital signal A and the synchronization 
signal B can be changed to any arbitrary value. 
In this manner, a synchronization signal B which has a constant phase, 
determined by the area of the pulse D' in the phase locked state, with 
respect to a digital signal input A with an intermittent edge can be 
obtained. 
Furthermore, it is possible to vary at will the phase, not only by changing 
the amplitude of pulse of D', but also the amplitude of pulse C' or the 
amplitudes of both pulses C' and D'. 
Moreover, it is clear that, as a modification of this example, the same 
purpose can also be accomplished by a means which changes the width of 
pulse D. 
Furthermore, in the aforementioned explanation, an example is shown in 
which the operation occurs only upon the rising edge of the input signal, 
but it is also possible to make it occur with the falling edge, rather 
than the rising edge, or with both the rising and falling edges. Neither 
of these modes of operation conflict with the operation of this invention. 
In the example described above, pulses C' and D' have the same polarity; 
their difference is obtained by means 6 for subtracting the pulses C' and 
D', and this difference is taken as the output E of the phase detector. 
The same operation can also be obtained by adding pulses C' and D', with 
either one given reverse polarity, and taking this sum as the output of 
the phase detector. 
Additional examples of the constituent elements of the phase detector of 
this invention are shown in FIGS. 4 and 5. In the example of FIG. 4, means 
2 and 3 for generating pulses in FIG. 1 are realized by circuits which 
combine two flip-flops 9 and 10 and one AND gate. The output terminal Q of 
flip-flop 9 is held at a low level until occurrence of the rising edge of 
the input signal A, at which point it changes to a high level. When the 
rising edge occurs of synchronization signal B, following the rising edge 
of input signal A, the output terminal Q of flip-flop 10 changes to a high 
level, flip-flop 9 is reset, and output terminal Q changes to the low 
level. By this means, a pulse is output with a width corresponding to the 
time difference between the rising edge of input signal A and the rising 
edge of synchronization signal B, i.e., the phase difference between A and 
B. The pulse output occurs only at the time of the rising edge of input 
signal A. That is, output terminal Q of flip-flop 9 outputs the signal C 
shown in FIG. 2. 
On the other hand, output terminal Q of flip-flop 10 changes to the high 
level after the rising edge of synchronization signal B and is in the high 
level during one period of synchronization signal B. As a result, a high 
level for one period of the synchronization signal B is applied to the 
input of AND gate 11 only after occurrence of the leading edges of the 
synchronization signal B and input signal A. Thus, a pulse with a width 
equal to the high level period of the synchronization signal B is output 
from the AND gate only after the rising edge of synchronization signal B. 
That is, the output of the AND gate is signal D shown in FIG. 2. 
Pulses C and D are passed through respective variable gain amplifiers 12 
and 13 and are applied to wide-band subtraction amplifier circuit 14. That 
is, the difference between pulses C' and D' is extracted by wide-band 
subtraction amplifier circuit 14 and is taken as the output of the phase 
detector (E of FIG. 2). As a result, since the mean value of the output of 
the phase detector is zero in the phase-locked state, the areas of the 
pulses C' and D' become equal. If we assume that the amplitudes of pulses 
C and D are equal, and the gains of the variable gain amplifiers 12 and 13 
are equal, the phase difference between input signal A and synchronization 
signal B, which is the width of pulse C, is equal to the high time of the 
synchronization signal, which is the width of pulse D. If the duty ratio 
of the synchronization signal is 1:1, a synchronization signal B with a 
phase of 180 degrees with respect to digital input signal A is obtained. 
Moreover, if either or both of the amplitudes of pulses C and D are changed 
at will by wide-band variable gain amplifiers 12 or 13, it is possible to 
change the phase of synchronization signal B at will, as shown in FIG. 3. 
As discussed above, the phase difference between input signal A and 
synchronization signal B can be changed by changing the amplitudes of 
pulses C and D. However, it is not easy, in general, to change the 
amplitudes to any arbitrary value, since the pulses contain high-frequency 
components. Since the information needed for the output of the phase 
detector is the direct-current component of the pulse, there is no problem 
with removing the high-frequency component of the pulse. Therefore, in the 
example of FIG. 5, low-pass filters 15 and 16 are attached to the output 
terminals of flip-flop 9 and AND gate 11, respectively. The high-frequency 
component is dropped, and the low-frequency-component only is used as the 
signal; the gain with respect to this low-band signal is changed by 
low-frequency amplifiers 17 and 18, and the subtraction is performed by 
the subtraction amplifier circuit 14. Therefore, since the constituent 
elements need only act on the low-frequency component, the adjustment of 
the phase can be performed by simpler circuits. The low-pass filters and 
low-frequency amplifiers may also be combined in other arrangements than 
that shown in this example. 
Examples of this invention have been shown, but they do not limit the form, 
arrangement, and other aspects of the invention; changes in the make-up of 
the invention are permitted, if desired, as long as the gist of the 
invention is preserved. 
EFFECTIVENESS OF INVENTION 
According to this invention, a synchronization signal generating circuit 
which generates a synchronization signal of any arbitrary phase difference 
with respect to a signal with intermittent edges, such as digital data, 
can be realized by a simple circuit structure. By doing so, the 
measurement of the bit error rate with respect to variations in the phase 
difference of the synchronization signal, which is an important evaluation 
item for digital recording devices, is made easier. Moreover, it is 
possible to optimize the phase of the synchronization signal during 
read-out in digital recording devices, so that bit errors are minimized.