Pulse count type FM demodulator

An FM demodulator having improved demodulation sensitivity, and suitable for monolithic integration. As demonstrated in various embodiments, increased sensitivity is made possible= by narrowing the demodulation band on a low frequency side. According to the various embodiments, a monostable multivibrator or first pulse generator receives an input signal and provides an output having first and second states having a combined duration equal to one period of the input signal, or alternatively equal to one-half period of the input signal. A second pulse generator, responsive to the output of the first pulse generator, generates another output having third and fourth states whose combined duration is the same as the combined duration of the first and second states. Demodulation is accomplished finally through a low-pass filter, which integrates the output of the second pulse generator.

BACKGROUND OF THE INVENTION 
The present invention relates to a frequency modulation (FM) demodulator 
and, more particularly, to a pulse count type FM demodulator. 
A pulse count type demodulator of the prior art, as will be described in 
more detail later, usually consists of a limiter circuit, a monostable 
multivibrator and a low-pass filter (LPF). The monostable multivibrator 
generates a pulse having a fixed time width in response to a transition 
point, for instance the leading edge point, of the limiter circuit's 
output. The LPF integrates the output of the monostable multivibrator to 
supply a demodulated output. Since the width from the trailing edge of the 
monostable multivibrator's output to its next leading edge is proportional 
to the frequency of the input signal, the output voltage of the LPF is 
proportional to the frequency of the input signal, so that FM demodulation 
is achieved. 
In the above described pulse count type demodulator of the prior art, the 
lower limit of the frequency band of demodulated signals is zero Hz 
(D.C.), and linearity is maintained over a wide band ranging from zero to 
the upper limit eetermined by the output pulse width of the monostable 
multivibrator. Though having such a wide frequency band, the prior art 
demodulator is poor in demodulation sensitivity. For this reason, where an 
FM signal whose maximum frequency deviation is extremely small relative to 
the center frequency, i.e., an FM signal whose normalized bandwidth is 
narrow, is to be demodulated, a demodulated signal is vulnerable to the 
adverse effect of external noise, such as source voltage fluctuation. 
SUMMARY OF THE INVENTION 
Therefore, an object of the present invention is to provide an FM 
demodulator having a higher demodulation sensitivity. 
Another object of the invention is to provide an FM demodulator suitable 
for an FM signal having a narrow normalized bandwidth. 
Still another object of the invention is to provide an FM demodulator 
suitable for monolithic integration. 
According to the invention, there is provided a frequency modulation (FM) 
demodulator comprising: limiter means for converting an input signal into 
a rectangular signal; first pulse generator means responsive to the 
transition points of said rectangular signal for generating a first pulse 
train having first and second states for the period of said rectangular 
signal, said first state having a predetermined period of time; second 
pulse generator means for generating a second pulse train having third and 
fourth states for the period of said rectangular signal, the duration of 
said third state being reduced by a predetermined time length based on the 
duration of said second state of said first pulse train; and low-pass 
filter means for integrating said second pulse train to provide a 
demodulated signal.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
To facilitate understanding of the present invention, a pulse count type FM 
demodulator of the prior art will be described first with reference to 
FIGS. 1, 2A to 2D and 3. 
Referring to FIG. 1, the demodulator comprises a limiter 61, a monostable 
multivibrator 62 and a low-pass filter (LPF) 63. The limiter 61 
amplitude-limits an input signal 2a to provide a rectangular signal 2b as 
shown in FIG. 2B. The monostable multivibrator 62 generates a pulse 2c 
(FIG. 2C) rising from a transition point of the rectangular signal 2b and 
having a fixed time duration (.tau..sub.c), and supplies it to the LPF 63, 
which provides a demodulated output 2d (FIG. 2D) by integrating the pulse 
2c. 
Thus, the demodulated output of the demodulator of FIG. 1 is calculated by 
Equation (1) below: 
##EQU1## 
where Vo is the demodulated output; 
T, the period of the input signal (=1/f.sub.in); 
f.sub.in, the frequency of the input signal; 
##EQU2## 
E, the amplitude of the output pulse. 
Integrating Equation (1) gives Equation (2) below: 
EQU Vo=E.tau..sub.c f.sub.in (2) 
Equation (2) indicates that the demodulated output Vo is proportional to 
the input frequency f.sub.in, and its demodulation (F/V) characteristic is 
shown in FIG. 3. 
As may be apparent from the characteristic shown in FIG. 3, the lower limit 
of the demodulation band is zero Hz (D.C.), and linearity is maintained 
over a wide band ranging from zero to the upper limit determined by the 
pulse width .tau..sub.c. This demodulation characteristic, however, has 
the disadvantages of low demodulation sensitivity and, where the 
normalized bandwidth of input FM signals is narrow, of vulnerability to 
external noise. 
FIG. 4 illustrates a pulse count type FM demodulator, which is a preferred 
embodiment of the present invention. In FIG. 4, an input terminal 10 is 
supplied with a rectangular signal 5a (FIG. 5A), similar to the 
rectangular output from the limiter 61 shown in FIG. 1. A first pulse 
generator circuit 11, like the monostable multivibrator 62 of FIG. 1, 
generates a pulse 5b (FIG. 5B) having a fixed pulse width commencing at 
the leading edge of the rectangular signal 5a. The pulse interval .DELTA.t 
of the pulse 5b of FIG. 5B is a function of the input frequency f.sub.in. 
Thus holds the relationship of Equation (3) below: 
EQU .DELTA.t=T-.tau..sub.c (3) 
A second pulse generator circuit 12, receiving the pulse 5b, narrows the 
pulse width .tau..sub.c of the pulse 5b only for a period of time 
proportional to its interval .DELTA.t (the proportional constant being a, 
which is a positive real number), and thereby generates a pulse 5c having 
a pulse width of .tau..sub.c -a.DELTA.t as shown in FIG. 5C. An LPF 13 
integrates the pulse train 5c to provide a demodulated output, which is 
represented by Equation (4) below, derived from Equation (1) above: 
EQU Vo=E{(a+1).tau..sub.c f.sub.in -a} (4) 
The characteristic of Equation (4), as shown in FIG. 6, indicates a 
narrower demodulation band on the low frequency side and, correspondingly, 
an (a+1) times higher demodulation sensitivity than the F/V characteristic 
of the prior art illustrated in FIG. 3. Accordingly, even where the 
normalized bandwidth of an FM signal is narrow, the demodulator is hardly 
vulnerable to external noise. 
FIG. 7 illustrates a pulse count type FM demodulator, which is another 
preferred embodiment of the present invention. In FIG. 7, a monostable 
multivibrator 21, serving as a first pulse generator circuit, generates 
pulses 8b (FIG. 8B) having a pulse width .tau..sub.c commencing at the 
leading and trailing edges of an input rectangular wave signal 8a to 
increase the demodulation sensitivity, and is so adjusted as to keep the 
.DELTA.t smaller than .tau..sub.c. The output pulse 8b of the monostable 
multivibrator 21 is divided into two branches, of which one is directly 
fed to one of the inputs of an AND gate 28 and the other, to the other 
input of the AND gate 28 through a delay circuit 29. The delay circuit 29 
is comprised of a first integration circuit comprising a resistor 22 and a 
capacitor 23, a first inverter 24 to receive the integrated output, a 
second integration circuit which comprises a resistor 25 and a capacitor 
26 and receives the output of the first inverter 24, and a second inverter 
27 to receive the output of this second integration circuit. The delay 
time .tau..sub.1 of the delay circuit 29 is set as represented by 
Inequality (5) below: 
EQU .DELTA.t.sub.max &lt;.tau..sub.1 &lt;.tau..sub.c (5) 
where .DELTA.t.sub.max is the pulse interval of the pulse 8b when the input 
frequency is at its minimum. 
The output 8d of the AND gate 28, as shown in FIG. 8D, has a pulse width 
smaller by .DELTA.t than the output pulse 8b of the monostable 
multivibrator 21. Integrating the pulse 8d with the LPF 13 gives the F/V 
characteristic of a=1, in FIG. 6. 
FIG. 9 illustrates an FM demodulator, which is still another preferred 
embodiment of the present invention. In FIG. 9, the output of a monostable 
multivibrator 21 is the same as that of the corresponding one in the 
embodiment of FIG. 7. Delay circuits 31 to 34 are connected in tandem, and 
the respective outputs of these delay circuits and of the monostable 
multivibrator 21 are applied to an AND gate 35. For the optimal design, it 
is recommended to set the values of the delay time .tau..sub.1 of each of 
the delay circuits 31 to 34 and the number n of the delay circuits as 
represented by the following Equation-Inequality pair (6): 
##EQU3## 
The modulation sensitivity of the modulator of FIG. 9, set as described 
above, is (n+1) times that of the conventional modulator of FIG. 1. 
FIG. 10 illustrates an FM demodulator, which is yet another preferred 
embodiment of the present invention. In FIG. 10, the output of a 
monostable multivibrator 21 is the same as that of the corresponding one 
in the embodiment of FIG. 7. A switching circuit 41 is intended to turn on 
and off constant current sources 42 and 43. When an input signal 11b is at 
its "high" level, the switching circuit 41 keeps the constant current 
source 42 on while holding the constant current source 43 off and open, 
and vice versa when the input signal 11b is at its "low" level. Therefore, 
when the input 11b to the switching circuit 41 is "high", the constant 
current source 42 charges a capacitor 44 by a high-potential power source 
48. Conversely, when the input 11b to the switching circuit 41 is "low", 
the constant current source 43 discharges the capacitor 44 to a 
low-potential power source 49 (a ground potential in this particular 
instance). The higher potential between the two electrodes of capacitor 44 
is clamped at a fixed potential by a clamp circuit 45. 
The relationship of Equation (7) below is to be maintained between the 
output current I.sub.1 of the constant current source 42 and the output 
current I.sub.2 of the constant current source 43. 
EQU I.sub.2 =nI.sub.1 (7) 
Since the capacitor 44 is charged and discharged by the constant currents 
I.sub.1 and I.sub.2, respectively, the inclinations of its charge and 
discharge waveforms, shown in FIG. 11C, are constant as represented by 
Equation (8) below: 
##EQU4## 
where c is the capacity of the capacitor 44. 
If the upper limit of these charge and discharge waveforms is clamped at 
the clamp potential Vc of the clamp circuit 45, the potential variation 
.delta.V (FIG. 11C) during the period of discharge by the current I.sub.2 
will be represented by Equation (9) below: 
##EQU5## 
where, as is evident from FIG. 11C, .delta.t.sub.2 equals .DELTA.t. 
Therefore, .delta.V is given by Equation (10) below: 
##EQU6## 
Then, the time .delta.t.sub.1 required for returning the potential 
variation .delta.V to V.sub.c during the period of discharge by the 
current I.sub.1 is calculated as follows: 
##EQU7## 
According to Equations (10) and (12), Equation (13) can be developed as 
follows: 
##EQU8## 
According to Equation (7), Equation (13) can be converted into Equation 
(14) below: 
EQU .delta.t.sub.1 =n.DELTA.t (14) 
Therefore, by waveform-shaping the charge and discharge waveforms shown in 
FIG. 11C with a reference voltage obtained from a voltage source 46, which 
is slightly lower than the clamp voltage V.sub.c, there is provided a 
pulse (FIG. 11D) having a width of .tau..sub.c -n.DELTA.t. Thus, by 
varying the ratio n between the constant currents I.sub.1 and I.sub.2, the 
demodulation sensitivity can be changed. Integrating the pulse shown in 
FIG. 11D with the LPF 13 will provide the desired demodulation output. 
FIG. 12 is a more specific circuit diagram of an integration circuit 100 of 
FIG. 10 which comprises the switching circuit 41, the constant current 
sources 42 and 43, and the clamp circuit 45. An input signal 11b is 
supplied to the base of a first transistor 51 whose emitter is connected 
to the emitter of a second transistor 52 and to a constant current circuit 
53. The collector of the first transistor 51 is connected to the cathode 
of a first diode 54. The base of the second transistor 52 is grounded 
through a bias constant voltage source 56, and the collector of same is 
connected to the cathode of a second diode 55. The anodes of both the 
first and second diodes 54 and 55 are connected to a power source. 
Further, the collectors of the first and second transistors 51 and 52 are 
connected to the bases of third and fourth transistors 57 and 58, 
respectively, whose emitters are both connected to the power source and 
collectors are connected to those of fifth and sixth transistors 59 and 
510, respectively. 
The fifth transistor 59, with its collector and base being short-circuited, 
is used as a diode, and to their connecting point is further connected the 
base of the sixth transistor 510, the emitters of both the fifth and sixth 
transistors being grounded. Being so connected, the fifth and sixth 
transistors 59 and 510 serve as a current mirror circuit. Further, the 
emitter area of the sixth transistor 510 is made n times as great as that 
of the fifth transistor 59. The collector of the sixth transistor 510 
serving as the output point is connected to a first terminal of the 
capacitor 44 whose second terminal is grounded. In parallel to the 
capacitor 44 is connected a series circuit of a third diode 511 and a 
reference voltage source 512. 
The above described structure enables the capacitor 44 to be charged and 
discharged by switching the first and second transistors 51 and 52 with 
the input signal 11b and the ratio between the charging and discharging 
currents to be determined by that between the emitter areas of the fifth 
and sixth transistors 59 and 510. The third diode 511 and the reference 
voltage source 512 constitute the clamp circuit 45, whose clamp value is 
represented by V.sub.ref +V.sub.r, where V.sub.r represents the on-voltage 
of the diode 511 and V.sub.ref, the voltage of the reference voltage 
source 512. 
FIG. 13 shows the F/V characteristics, simulated by the simulation program 
of SPICE-F, of an FM demodulator circuit composed of the bipolar 
transistor circuit illustrated in FIGS. 10 and 12. The abscissa represents 
the frequency, and the ordinate, the D.C. component of the demodulated 
output. The characteristics were simulated with an intermediate frequency 
of 455 kHz, with temperature variations from -20.degree. to +70.degree. 
taken into consideration. As is evident from these simulated 
characteristics, a demodulation performance sufficiently close to 
linearity can be achieved in the temperature range of -20.degree. to 
+70.degree. and in the frequency (f) range of 390 kHz to 490 kHz. 
Incidentally, the ratio between the constant currents I.sub.1 and I.sub.2 
is set at 1.75. 
As hitherto described, an FM demodulator according to the present invention 
makes it possible to increase the demodulation sensitivity. This is 
achieved by adding a circuit which subjects the pulse width .tau..sub.c of 
a monostable multivibrator output to pulse width modulation by the pulse 
interval .DELTA.t to alter the pulse width to .tau..sub.c -a.DELTA.t. An 
FM demodulator according to the invention is especially suitable for 
monolithic integration of a demodulator circuit for use with input signals 
whose normalized bandwidth is comparatively narrow.