High frequency voltage regulating transformer based converter

A switching regulator and method of fabricating a switching regulator is disclosed. In one embodiment a switching regulator comprises a transformer having a primary winding and a secondary winding, an inductor (L) connected to an input of the primary winding, and a capacitor (C) connected across the primary winding of the transformer. The inductor and capacitor form a L-C tank circuit. The switching regulator also includes a frequency source connected to the inductor, wherein the frequency source provides a switching frequency to the inductor that is substantially equal to a resonant frequency of the L-C tank circuit.

TECHNICAL FIELD

The present invention relates to electrical circuits and more particularly to direct current (DC) to direct current (DC) power conversion and regulation.

BACKGROUND

Power converters typically serve to accept energy from an unregulated energy source, such as a voltage source, and derive therefrom a regulated voltage which is applied to a load circuit. The regulation function is performed by interposing a voltage regulator device between the source of energy and the load circuit. One such type of voltage regulator is known as a switching regulator or switching power supply. These devices employ switching devices that operate in either a fully on state or a fully off state. The switching device is periodically turned on for a time interval to permit energy transfer around the various elements of the power train for purposes of maintaining the voltage output at a predetermined level. The device is then periodically turned off to allow the energy to decay into the load. A feedback signal from the output of the regulator is fed back to a control circuit. The control circuit utilizes the feedback signal to continuously adjust the duty cycle and/or frequency of the control signal driving the power switches in responses to variations in the output load and input voltage, and as a result, regulating the output voltage.

Typically, the power switches are turned on and off at a frequency which varies with the application and given size, cooling, power output, temperature and efficiency constraints. In certain applications, it may be required that the output signal is electrically isolated from the input signal, for example, via a transformer. In these applications, it will be required that the feedback signal be electrically isolated from the control circuit. Therefore, the switching regulator will require DC feedback components that work across said isolation boundary such as opto-isolators or additional signal transformers that add cost and complexity to the switching regulator.

SUMMARY

One aspect of the present invention relates to a switching regulator. The switching regulator comprises a transformer having a primary winding and a secondary winding, an inductor connected to an input of the primary winding, and a capacitor connected across the primary winding of the transformer. The inductor and capacitor form a L-C tank circuit. The switching regulator also includes a frequency source driving switch elements connected to the inductor, wherein the frequency source provides a switching frequency to the inductor that is substantially equal to a resonant frequency of the L-C tank circuit.

Another aspect of the invention relates to a voltage regulating transformer based converter. The converter comprises a transformer having a primary winding with an overwinding and a secondary winding, an inductor connected to an input of the primary winding, and a capacitor connected across the primary winding of the transformer. The inductor and capacitor form a L-C tank circuit. The converter also comprises a phase locked loop (PLL) device connected to the switch elements which are connected to the inductor, wherein the PLL provides a switching frequency to the inductor that is substantially equal to a resonant frequency of the L-C tank circuit. Additionally, the overwinding provides voltage gain to a change in voltage of the tank circuit to enhance saturation of the transformer such that a clipped sine wave voltage is provided to the secondary winding of the transformer. By this, any input perturbation is minimized at the secondary.

Another aspect of the invention relates to a method for fabricating a switching regulator. The method comprises fabricating an integrated inductor-transformer assembly having a primary winding, a secondary winding and an inductor connected between the primary winding and an input terminal. A tank capacitor is connected across the primary winding of the transformer wherein the inductor and capacitor form a L-C tank circuit. A frequency source is connected to the inductor, wherein the frequency source provides a switching frequency to the inductor that is substantially equal to a resonant frequency of the L-C tank circuit. The methodology includes sweeping the frequency source along a range of frequencies about the resonant frequency, while measuring a response voltage of the tank circuit, and setting the frequency of the frequency source at a frequency that corresponds to a maximum peak response voltage of the tank circuit.

DETAILED DESCRIPTION

The present invention relates to a switching power supply or regulator that provides a regulated output voltage from an unregulated input voltage without the need of a feedback signal from an output voltage to compensate for variations in input voltage and output load. In one aspect of the invention, the switching power supply is a transformer-based converter that employs an inductor-capacitor (L-C) resonant tank circuit to transfer energy from an input voltage source to an electrically isolated load. A frequency source (e.g., a phase locked loop (PLL)) provides a switching frequency to the L-C tank circuit that is substantially equal to the resonant frequency of the L-C tank circuit. Therefore, regulation is controlled by the switching frequency of the converter as opposed to the DC feedback of the output signal. This eliminates the need to crossover the electrical isolation boundary with a voltage representative signal associated with the output signal.

Regulation is achieved by selecting a transformer that saturates at a volt-time product that is less than the volt-time product provided by the tank circuit. This operation appears as a clipped sine wave at the secondary. Additionally, regulation can be improved further by providing a transformer with an overwinding that provides additional gain to the tank circuit, since the Q value (Quality Factor or Figure of Merit) of the tank circuit is proportional to the voltage across the capacitor of the tank circuit. The Q value of a resonator is a measure of quality of the resonator, such that the lower the dissipation the higher the Q value of the resonator.

FIG. 1illustrates a switching regulator10in accordance with an aspect of the present invention. The switching regulator10can be, for example, a half bridge based converter. The present invention can be employed with most bi-directional, voltage fed power topologies including but not limited to, full bridge, phase shifted full bridge and push pull. The switching regulator10includes a frequency source12that is operative to provide a first control pulse to a first power switch Q1and a second control pulse to a second power switch Q2. The first control pulse and the second control pulse can oscillate at the same frequency at 180° out of phase. The first power switch Q1is connected to an unregulated input voltage (VIN) at its drain terminal and a node14at its source terminal. The second power switch Q2is connected to its drain terminal at the node14and to a first ground (GND1) at its source terminal. An input capacitor CINis connected between the unregulated input voltage (VIN) and the first ground (GND1). The frequency source12is operative to run each switch Q1and Q2at approximately 50% duty cycle with a little deadtime between transitions to avoid cross conduction.

The node14is connected to first end of an inductor L1, with the second end of the inductor L1being connected to an input terminal of a transformer T1. Switching of the first end of the inductor L1between VINand ground cause the inductor L1to provide the energy to charge a first capacitor C1through an overwinding winding22, and provides the energy to charge a second capacitor C2through a primary winding20. The overwinding22is another winding that is closely magnetically coupled to the primary winding to add additional winding turns to the primary winding20. The first inductor L1and the first capacitor C1form an L-C tank circuit18. The second capacitor C2charges to a voltage of VIN/2 at a node17due to the symmetry of the first and second control pulse. The second capacitor holds the return side of the primary winding20of the transformer T1at a voltage VIN/2. The second capacitor C2and the input capacitor CINcan be selected to be substantially large. By this, load transitions, input voltage variations and surge currents do not change the voltage of the second capacitor C2and the input capacitor CIN

The primary winding20and the overwinding22are magnetically coupled to a secondary winding24of the transformer T1. A first end of the secondary winding24is connected to an anode of a diode D1and a second end (return side) of the secondary winding24is connected to an anode of a diode D2. The cathodes of the diodes D1and D2are connected to a first end of an output inductor L2. A second end of the output inductor L2is connected to a third capacitor C3. The capacitor C3is connected between an output voltage VOUTand a second ground (GND2) that is electrically isolated from the first ground (GND1).

The L-C tank circuit18is selected to have a resonant frequency that is substantially equal to a switching frequency of the frequency source12. The switching frequency of the frequency source12should be substantially constant. The overwinding22on the transformer primary allows the energy stored in L1to ring C1to a higher voltage, thus achieving a higher Q resonant tank circuit18, than can be achieved by simply connecting C1to L1. For example, for an arbitrary L1and C1pair, energy is exchanged in a resonant fashion between L1and C1. During exchange of the LI2/2 component of L1to C1, the overwinding22amplifies the voltage and reduces the current into C2proportionally, such that the impedance ZC1of the capacitor C1as seen by the inductor L1is:
ZC1=1/jωC1* ((NOVERWIND+NPRIMARY)/NOVERWIND)2EQ. 1

where NOVERWINDis the number of winding turns of the overwinding22and NPRIMARYis the number of winding turns of the primary20. Therefore, without the overwinding22the impedance of the capacitor C1as seen by the inductor L1is simply:
ZC1=1/jωC1EQ. 2
Additionally, if we review the Quality factor or Figure of Merit Q for nonzero overwinding, it follows that:
Q=Q *((NOVERWIND+NPRIMARY)/NOVERWIND)2EQ. 3

such that the higher the Q of the resonant circuit, the more constant the flux excursions will be resulting in a more constant output voltage and thus a regulated output voltage without the use of feedback. The upper limit on Q will be governed in practice by practical voltage limits on the capacitor C1, creepage and clearance distance requirements across the windings and interwinding capacitance. Additionally, L1can be sized to allow safe operation into shorted secondary conditions, such that
ZMIN=jωL1EQ. 4

The transformer T1can be designed in such a way as to saturate at some volt-time product less than the maximum volt-time product developed across C1. Therefore, as long as there is enough energy in L1, the primary waveform will be a clipped sine wave. Additionally, the transformer T1can operate safely, while being driven in and out of saturation. L1should be selected to be substantially larger than the leakage inductance of the primary20of the transformer T1to mitigate any parasitic interactions caused by the transformer T1. Additionally, L1should have substantially constant inductance over line, load and temperature. L1can also serve to limit the current into T1under short circuit conditions at the secondary, thus limiting the maximum current drawn by the regulator10to a safe value.

As the load (not shown) on the output of the regulator10increases, more current is drawn through L1. This stores more energy in L1, which drives more energy into the transformer T1and the load (not shown) into C1. The higher energy in C1sustains the clipped sine waveform across the primary20and the secondary24, while the remaining current couples into the secondary24and flows to the load. As a result of the L-C tank circuit18having a resonant frequency that is substantially equal to a switching frequency of the frequency source12, and the constant clipped sine waveform provided to the secondary24. The clipped sine waveform is rectified and filtered to provide a regulated output voltage (VOUT).

FIG. 2illustrates voltage waveforms at an input of inductor L1and at an output of capacitor C1at initialization of the voltage regulator10. A first voltage waveform (VPH)30is provided at the node14. The first voltage waveform30is a square wave that oscillates between VINand the first ground (GND1) having a frequency substantially equal to the resonant frequency of the LC tank circuit18. A second voltage waveform (VC1)32illustrate the voltage response at the node16coupled to the output of the capacitor C1. As illustrated inFIG. 3, the capacitor voltage grows exponentially over time from an initial peak magnetic field (B) state of B=0 to a final peak state of B=BSAT, such that the peak amplitude voltage of the capacitor C1exceeds the saturation state of the transformer T1. Therefore, after initialization, the voltage waveform provided to an input of the transformer is a clipped sinewave having a frequency substantially equal to the resonant frequency of the LC tank circuit18.

FIG. 3illustrates steady state voltage waveforms after initialization of the voltage regulator10. A first voltage waveform (VPH)40is provided at the node14. The first voltage waveform40is a square wave that oscillates between VINand the first ground (GND1) having a frequency substantially equal to the resonant frequency of the LC tank circuit18. A second voltage waveform (VC1)32illustrate the voltage response at the node16connected to the output of the capacitor C1. The second voltage waveform42at steady state is a clipped sinewave having a frequency substantially equal to the resonant frequency of the LC tank circuit18. The clipped sinewave of the second voltage waveform is provided to the primary20and overwinding22of the transformer T1. A resultant voltage waveform (VSEC)44is provided at the secondary24of the transformer T1at node26. A current waveform46illustrates a current IL2through the output inductor L2in response to the resultant voltage waveform44.

FIG. 4illustrates a switching regulator50in accordance with another aspect of the present invention. The switching regulator50can be, for example, a half bridge based converter. The present invention can be employed with most bi-directional, voltage fed power topologies including but not limited to, full bridge, phase shifted full bridge and push pull. The switching regulator50includes a phase locked loop (PLL) frequency source52that is operative to provide a first control pulse to a first power switch Q3and a second control pulse to a second power switch Q4through a buffer54. The first power switch Q3is connected to an unregulated input voltage (VIN) at its drain terminal and a node57at its source terminal. The second power switch Q4is connected to its drain terminal at the node57and to a first ground at its source terminal. An input capacitor CINis connected between the unregulated input voltage (VIN) and the first ground (GND1). The first control pulse is phased at 180° with respect to the second control pulse, which is phased at 0°. Therefore, the PLL frequency source52is operative to run each switch Q3and Q4at approximately 50% duty cycle.

The node57is connected to an integrated inductor-transformer assembly40. The integrated inductor-transformer assembly60includes an inductor L3having a first end connected to a primary winding62and an overwinding66, and a second end connected at a first input terminal (1) of the integrated inductor-transformer assembly60. An end of the overwinding66is connected to a second input terminal (2) and an end of the primary winding62is connected to a third input terminal (3). A secondary winding64of the integrated inductor-transformer assembly60includes a first output terminal (4) connected to a center tap of the secondary winding64, a second output terminal (5) connected to a first end of the secondary winding64and a third output terminal (6) connected to a second end of the secondary winding64. The primary winding62, the overwinding66and the secondary winding64form a transformer T2. The second output terminal (5) is connected to an anode of a diode D3and the third output terminal (6) is connected to an anode of diode D4. The cathode of the diodes D3and D4are connected to a first end of an output inductor L4, and a second end of the output inductor L4is connected to a capacitor C6. The capacitor C6is connected between an output voltage (VOUT) and a second ground (GND2) that is electrically isolated from the first ground (GND1).

The integrated inductor-transformer assembly60provides for an integrated solution to providing a regulated output voltage by connecting an accurate frequency source52to the first input terminal (1) and a tank capacitor C4across the second input terminal (2) and the third input terminal (3) of the integrated inductor-transformer assembly60. The tank capacitor C4is selected such that the resonant frequency of the tank capacitor C4and the inductor L3is substantially equal to the switching frequency of the PLL52. Additionally, a number of rectification techniques can be employed at the first and second output terminals (5,6) to provide a regulated output without employing feedback.

For the present realization, a capacitor C5can be connected from the third input terminal to the first ground (GND1). The capacitor C5charges to a voltage of VIN/2 due to the symmetry of the first and second control pulse. The capacitor C5holds the return side of the primary winding62of the integrated inductor-transformer assembly60at a voltage VIN/2. The capacitor C5can be selected to be substantially large so that load transitions and surge currents do not change the voltage of the capacitor C5.

The inductor L3provides the energy to charge the tank capacitor C4through the overwinding66of the primary62of the integrated inductor-transformer assembly60. The overwinding66on the transformer primary62allows the energy stored in L3to ring C4to a higher voltage, thus achieving a higher Q resonant tank circuit, than can be achieved by simply connecting C4to L3. The transformer T2of the integrated inductor-transformer assembly60can be designed in such a way as to saturate at some volt-time product less than the maximum volt-time product developed across C4. Therefore, the primary waveform will be a clipped sine wave. The clipped sine waveform is rectified and filtered at the secondary to provide a regulated output voltage without feedback.

In view of the examples shown and described above, a methodology in accordance with the present invention will be better appreciated with reference toFIG. 5. While, for purposes of simplicity of explanation, a methodology is shown and described as executing serially, it is to be understood and appreciated that the methodology is not limited by the order shown, as some aspects may, in accordance with the present invention, occur in different orders and/or concurrently from that shown and described herein. Moreover, not all features shown or described may be needed to implement a methodology in accordance with the present invention. Additionally, such methodology can be implemented in hardware (e.g., one or more integrated circuits), software (e.g., running on a DSP or ASIC) or a combination of hardware and software.

FIG. 5illustrates a methodology for fabricating a switching regulator without feedback in accordance with an aspect of the present invention. The methodology begins at 100 where an integrated inductor-transformer assembly is fabricated. The integrated inductor-transformer assembly can include a tank inductor with a first end connected between a primary winding and an overwinding of the primary winding. A second end of the tank inductor can be connected to an input of the integrated inductor-transformer assembly. The methodology then proceeds to110. At110, a tank capacitor is connected across input terminals of the integrated inductor-transformer assembly. The input terminal can include a first input connected to the overwinding and a second input connected to a return side of a primary winding. The methodology then proceeds to120.

At120, a frequency source (e.g., a PLL) is connected to the second end of the inductor through the input of the integrated inductor-transformer assembly. At130, a measurement device (e.g., voltage measurement) is connected to the integrated inductor-transformer assembly. For example, the input device can be connected to the overwinding end of the integrated inductor-transformer assembly, or the secondary winding of the integrated inductor-transformer assembly. The measurement device is operative to measure the peak response of the tank circuit formed by the tank inductor and the tank capacitor. The methodology then proceeds to140.

At140, sweeping of the frequency of the frequency source is performed, while measuring the peak response of the tank circuit. The frequency range of the frequency sweep can be, for example, within +/−10% of the resonant frequency of the tank circuit. The sweeping of the frequencies is performed to accommodate for manufacturing variations of the integrated inductor-transformer assembly. At150, the frequency source is set at a frequency that corresponds to a maximum peak response of the tank circuit. At160, the remaining circuitry is connected to the switching regulator. The remaining circuitry can include coupling capacitors, rectifier circuitry and switching devices.