Communicating over coaxial cable networks

A method for communicating over a coaxial cable network is described. The method includes identifying at least one port in the coaxial cable network that provides high mutual isolation among nodes of the coaxial cable network when the port is terminated with an impedance that matches a characteristic impedance of coaxial cable in the coaxial cable network. The method also includes terminating the identified port with an impedance that is substantially mismatched with the characteristic impedance of the coaxial cable, and transmitting a signal from a first node in the network to a second node in the network.

TECHNICAL FIELD

This invention relates to communicating over coaxial cable networks.

BACKGROUND

Coaxial cable transmission lines can be used to route radio frequency (rf) signals throughout a home. The characteristics of a coaxial cable determine what maximum frequency the cable will support for high quality (e.g., high signal-to-noise ratio) transmission of analog or digital signals. Older cable existing in many homes may support high quality transmission of signals up to around 900 MHz. Other types of cable (e.g., cable used for satellite television signals) may support higher frequencies up to around 1700 MHz. The frequency limit also determines the maximum data rate limits for digital signals (e.g., digital video or internet protocol (IP) data packets).

A cable signal typically enters a home over a single source port and from there is distributed throughout the home. A distribution network of coaxial cable is typically formed by connecting cables to splitters that passively couple an incoming signal to two or more output ports. This network typically has a tree topology in which information flows downstream from the source (at the “root” of the tree) to each terminating device such as a television, set top box, or cable modem (the “leaves” of the tree). In some cases (e.g., for a cable modem or interactive television service) information also flows upstream from a terminating device to the source port.

SUMMARY

In a first aspect, the invention features a method for communicating over a coaxial cable network. The method includes identifying at least one port in the coaxial cable network that provides high mutual isolation among nodes of the coaxial cable network when the port is terminated with an impedance that matches a characteristic impedance of coaxial cable in the coaxial cable network. The method also includes terminating the identified port with an impedance that is substantially mismatched with the characteristic impedance of the coaxial cable, and transmitting a signal from a first node in the network to a second node in the network.

Preferred implementations of this aspect of the invention may incorporate one or more of the following:

The identified port includes an input port to a splitter having at least two output ports that are mutually isolated when the input port is terminated with an impedance that matches the characteristic impedance of the coaxial cable.

The splitter includes a hybrid splitter.

The identified port is positioned in the network to distribute an incoming signal from a source to terminal nodes of the coaxial cable network.

The source is a cable television feeder cable, a terrestrial antenna, or a satellite dish.

Terminating the identified port with the mismatched impedance includes coupling the incoming signal from the source to the identified port with an output impedance that is substantially mismatched with the characteristic impedance of the coaxial cable.

Terminating the identified port with the mismatched impedance includes uncoupling the source from the identified port.

Transmitting the signal from the first node to the second node includes coupling a signal from the first node with an output impedance that is substantially mismatched with the characteristic impedance of the coaxial cable.

The output impedance is substantially smaller than the characteristic impedance of the coaxial cable.

The output impedance is smaller than about 10% of the characteristic impedance of the coaxial cable.

Transmitting the signal from the first node to the second node includes coupling a signal to the second node with an input impedance that is that is substantially mismatched with the characteristic impedance of the coaxial cable.

The input impedance is substantially larger than the characteristic impedance of the coaxial cable.

The output impedance is larger than about 300% of the characteristic impedance of the coaxial cable.

The coaxial cable network has a tree topology with the identified port at the root of the tree.

In a second aspect, the invention features a coaxial cable network. The network includes a source port providing an input signal, a coaxial cable coupling the source port to a first splitter, and a plurality of coaxial cables providing an interface for nodes of the network. At least some of the coaxial cables are coupled to the source port over a path that includes at least one splitter. At least one splitter port provides high mutual isolation among nodes of the coaxial cable network when the splitter port is terminated with an impedance that matches a characteristic impedance of coaxial cable in the coaxial cable network. The splitter port is terminated with an impedance that is substantially mismatched with the characteristic impedance of the coaxial cable.

Preferred implementations of this aspect of the invention may incorporate one or more of the following:

The splitter port includes an input port to the first splitter, the first splitter having at least two output ports that are mutually isolated when the input port is terminated with an impedance that matches the characteristic impedance of the coaxial cable.

The coaxial cable network has a tree topology with the input port to the first splitter at the root of the tree.

The first splitter is positioned in the network to distribute the incoming signal from the source port to terminal nodes of the coaxial cable network.

The coaxial cable network further includes a node coupled to a coaxial cable interface with an output impedance that is substantially mismatched with the characteristic impedance of the coaxial cable.

The coaxial cable network further includes a node coupled to a coaxial cable interface with an input impedance that is that is substantially mismatched with the characteristic impedance of the coaxial cable.

Among the many advantages of the invention (some of which may be achieved only in some of its various aspects and implementations) are the following.

Mismatching the impedance at one or more splitters in a coaxial cable network reduces attenuation due to isolation between nodes in the network which increases the data rate and reliability of communication between nodes. Placing an impedance mismatched gateway device at the root node of a tree network enables communication among the leaf nodes while maintaining the ability to distribute a source signal to the leaf nodes. Coupling transmitting devices to a coaxial cable network with a low output impedance and coupling receiving devices to the coaxial cable network with a high input impedance provides low-loss communication over a wide range of network characteristics including, for example, various cable lengths and various numbers of splitters.

Other features and advantages of the invention will be found in the detailed description, drawings, and claims.

DETAILED DESCRIPTION

There are a great many possible implementations of the invention, too many to describe herein. Some possible implementations that are presently preferred are described below. It cannot be emphasized too strongly, however, that these are descriptions of implementations of the invention, and not descriptions of the invention, which is not limited to the detailed implementations described in this section but is described in broader terms in the claims.

System Overview

Referring toFIG. 1, a coaxial cable network100in a home includes a source port104for a source cable106that carries an incoming signal from a source108outside of the home. For example, the source108can be wired source that provides a signal over a distribution network that is fed from a head-end at a cable television distribution center to distribution coaxial cables (e.g., “trunk” or “feeder” cables). Alternatively, the source108can be wireless source such as a terrestrial antenna that receives a signal from a broadcast tower, or a satellite dish that receives a signal from a satellite.

The coaxial cable network100distributes a signal throughout the home from the source port104, through a gateway device102, to standard devices110(e.g., cable or satellite television set top boxes) and network devices112over coaxial cable111(e.g., RG6 type coaxial cable). The coaxial cable network100includes splitters that split input signal power among multiple output ports. In this exemplary network100, the first splitter113is a 4-port, 3-way splitter that divides the signal at the input port evenly among three output ports. Alternatively, some splitters provide more power to some ports than to others. These uneven splitters can be used to ensure certain devices (e.g., cable modems) have a large enough signal, or to provide more power to ports that will undergo further splitting to feed more downstream terminal nodes or “leaf” nodes. The coaxial cable network100also includes 3-port, 2-way splitters114that divide the signal at the input port evenly between two output ports. The coaxial cable network100includes a bridge device116that couples the network100to a secondary network120such as a power line communication network that uses existing AC wiring in a house to exchange information between nodes that interface with AC outlets.

The gateway device102enables the network devices112to communicate with each other, while continuing to distribute the incoming signal from the source port104to the standard devices110. In a typical cable distribution network in a home, to reduce interference on the network, the splitters113and114provide high isolation among the output ports such that a signal entering one output port of the splitter is coupled to the input port and effectively cancelled at the other output port(s). For example, a “hybrid splitter” (or “magic tee” splitter) is typically designed to provide high isolation among output ports for a given impedance at the input port. As explained in more detail below, the impedance at which this high isolation occurs is designed to match the characteristic impedance of a given type of coaxial cable. Isolation of 20 to 60 dB is typical in practice depending on the precision of the components. This high attenuation would reduce the signal-to-noise ratio (SNR) which would in turn reduce the channel capacity (data rate).

The gateway device102terminates the “root” port122of the coaxial cable network with an impedance that is mismatched with the characteristic impedance designed to provide high isolation. As described in more detail below, this mismatch “propagates” throughout the tree-structured network100to mismatch the input ports of the other splitters enabling any node in the network to communicate with any other node without suffering drastic reduction in SNR due to high isolation. Alternatively, the root port122can be disconnected from the source port104to mismatch the network100without the need for a gateway device102(though this configuration would no longer distribute the incoming signal to the standard devices110).

The standard devices110are configured to receive the signal from the source port104(and optionally to transmit signals to the source port104) without interfering with each other. In particular, the standard devices110terminate the coaxial cables111with the characteristic impedance Z0of the cable111(e.g., for RG6 coaxial cable Z0=75 Ohms). Even though the splitters no longer provide high isolation, this impedance matching effectively eliminates reflections of a signal from the input of one standard device110that could interfere with another standard device110.

The coaxial cable network100is coupled to network devices112that are configured to transmit signals to and receive signals from other network devices112coupled to the network100. The network devices112are half-duplex devices that switch between a transmit state and a receive state (the default state). The network devices112can use any of a variety of types of medium access control (MAC) protocols such as a carrier sense multiple access with collision avoidance (CSMA/CA) protocol to coordinate communication over the network100. The network devices112can optionally terminate the coaxial cables111with an impedance that depends on whether the device is in the transmit state or the receive state to improve signal characteristics such as signal-to-noise ratio (SNR), as described in more detail below.

The standard devices110and the network devices112communicate over different frequency bands using filters to reduce any potential interference between the standard and network devices. For example, in one scenario the standard devices receive a signal in the 50 to 800 MHz range and the network devices communicate in the 2 to 28 MHz range. Each network device112includes a 35 MHz low-pass filter (LPF) to interface with the network100, and each standard device includes a 50 MHz high-pass filter (HPF) to interface with the network100. The combination of the LPFs and HPFs reduce potential interference caused by signal energy transmitted from or reflected from unmatched network devices112.

Alternatively, all of the devices coupled to the output ports of the splitters can be network devices112, in which case, the filters are not necessarily used.

Impedance Matching and Mismatching

The characteristics of impedance matching and mismatching can be understood by examining simplified circuit models of the coaxial cable network100and the various devices coupled to the network acting as transmitters and/or receivers. Referring toFIG. 2A, when a device is transmitting a signal into a port of the coaxial cable network, that device can be modeled as a “source” circuit element200having a voltage source202that provides a source voltage signal VS(t) in series with an impedance Zoutthat represents the output impedance of the device. Referring toFIG. 2B, when a device is receiving a signal over a coaxial cable of the network100, that device can be modeled as a “load” circuit element204having an impedance Zinthat represents the input impedance of the device.

Referring toFIG. 2C, a transmitting device210, represented by source circuit element200, is connected to a receiving device212, represented by load circuit element204, over a coaxial cable modeled as a transmission line220having a length l. The voltage signal VR(t) that is received by the receiving device212is a function of the source voltage signal VS(t), but also depends on the impedances Zoutand Zinand the characteristic impedance Z0of the transmission line220. In general, to the extent that either Zoutor Zindiffers from the characteristic impedance Z0, there will be reflections that propagate between the input port222and output port224of the transmission line220causing distortions in the received voltage signal VR(t) including frequency selective distortions and time distortions such as multiple delayed versions of a signal arriving over a time period called “delay spread.” For a transmission line terminated with a “mismatched” load impedance at the output port224that differs from the characteristic impedance Z0, the effective impedance seen at the input port222is transformed by the transmission line (e.g., as given by a Smith Chart). For example, depending on the length l, a real load impedance (i.e., resistance) of RLcan be transformed to an inductive or capacitive impedance or to a real impedance of Z20/RL(when l is a quarter wavelength). However, a mismatched impedance remains mismatched for any length l or signal frequency. The expected behavior of a given network can be predicted according to standard transmission line theory where each section of coaxial cable in the network is modeled as a transmission line.

Typically, the input and output impedances of devices coupled to the network100are “matched” to the characteristic impedance of the coaxial cable (i.e., Zout=Z0and Zin=Z0). In this matched case, the reflections are eliminated (or in practice, due to the limited precision of the components, at least greatly reduced) and the received voltage signal VR(t) is related to the source voltage signal as VR(t)=0.5 VS(t−l/v), where v is the propagation velocity of the transmission line (typically around 0.6-0.8 times the speed of light for coaxial cables). In practice, for a matched transmission line the received voltage signal is a scaled and delayed version of the source voltage signal over a wide range of frequencies, and does not suffer the frequency distortions or delay spread of the mismatched transmission line.

A typical splitter is designed to terminate a coaxial cable coupled to its input port with a matched impedance when the output ports of the splitter are terminated with matched load impedances. The typical splitter is also designed to provide a matched output impedance to each load. Thus, the splitter is designed to preserve the impedance matching characteristics of a network. In addition to preserving impedance matching, a typical splitter is designed to provide high isolation among its output ports.

Referring toFIG. 2D, one example of a 3-port, 2-way splitter114having high isolation among output ports is a hybrid splitter modeled as a circuit230that has a single input port231and two output ports232and233. The input port231is coupled to a 2:1 impedance transformer234that transforms the output impedance of a device coupled to the input port231by a factor of ½ (e.g., a transformer with a turns ratio of √2:1 yields an impedance ratio of 2:1). The three ports are connected to a center-tap autotransformer236which couples signals among some of the ports under certain conditions. A shunt resistor238is connected to the autotransformer236to establish conditions such that the output ports232and233can be mutually isolated.

Due to the symmetry of the circuit230, an input signal at port231is evenly divided between ports232and233. However, when a signal is applied to the output port232, the circuit230sets a voltage at the other output port233based on the impedance at the input port231. Referring toFIG. 2E, a source240coupled to the output port232sees the equivalent circuit242due to the impedance transformation properties of the autotransformer236. In particular, the autotransformer236transforms the impedance 2Z0of the shunt resistor238by a factor of ¼ (since the turns ratio is ½) to a value of Z0/2. Similarly, the impedance transformer234transforms the impedance Z1at the input port231by a factor of ½ to a value of Z1/2. Thus, the source240sees the equivalent circuit244(FIG. 2F) and applies a source voltage VS(t) across three impedances: the output impedance Zoutan impedance Z0/2 due to the splitter circuit230, and an impedance Z1/2 due to the termination of input port231.

The properties of autotransformer236ensure that the voltage drop Vx(t) across the top half of the autotransformer236is the same as the voltage drop across the bottom half of the autotransformer. When the impedance Z1at the input port231is equal to the characteristic impedance Z0, the voltage drop Vx(t) across the top half of the autotransformer236is equal to the voltage drop from the mid-point of the autotransformer236to ground. Therefore, in this “matched input port” case, the voltage drop Vx(t) across the bottom half of the autotransformer236sets the voltage at the output port233to ground, regardless of the value of the source voltage VS(t) or source output impedance Zout. In this case, all of the power delivered into output port232is coupled to the input port231(neglecting internal splitter losses). This ideal model exhibits complete isolation, however, in practice hybrid splitters suffer from leakage current and leakage inductance such that isolation of 20 to 60 dB is possible over an operating bandwidth, depending on the precision of the splitter components.

When the impedance Z1at the input port231is not equal to the characteristic impedance Z0, the voltage drop Vx(t) across the top half of the autotransformer236is not equal to the voltage drop from the mid-point of the autotransformer236to ground. Therefore, in this “mismatched input port” case, the voltage drop Vx(t) across the bottom half of the autotransformer236sets the voltage at the output port233to some proportion of the source voltage VS(t) depending on the ratio of the impedances Z1and Z0. Thus, even in the ideal case, the isolation degrades and a signal can pass from output port232to output port233without suffering severe attenuation.

FIGS. 3A-3Dshow transfer responses for a simulation of a coaxial cable network based on an ideal hybrid splitter circuit model. The simulated network includes a voltage controlled voltage source with series output resistor connected to the input port “Port1” of the splitter over a 50 feet length of 75-Ohm coaxial cable to provide a variable impedance drive to the network. Two additional voltage controlled voltage sources with shunt input resistors are connected to the output ports “Port2” and “Port3” over 50 ft. lengths of 75-Ohm coaxial cable, respectively, to provide variable impedance output loads for the network.FIGS. 3A-3Dshow the transfer response between ports of the simulated network under a variety of terminating conditions for the source and loads.

FIGS. 3A and 3Bshow transfer responses with the cable termination impedances for all three ports “matched” to the cable characteristic impedance of 75 Ohms. In the plot ofFIG. 3A, showing an input-to-output response, the attenuation in decibels (dB) of the path from Port1to Port2is nearly flat as a function of frequency over a bandwidth of 0 to 30 MHz. Internal splitter power losses (e.g., due to resistive power dissipation) are minimal in practice and are modeled as 1 dB in this example. The nominal total attenuation of around 4 dB is due to the combination of this internal splitter loss, the dielectric loss of the coaxial cable (which increases with frequency), and loss due to a voltage divider effect where some power is dissipated in the output resistor of the source. The simulation models the coaxial cables using characteristics of an RG59 type coaxial cable.

In the plot ofFIG. 3B, showing an output-to-output transfer response as a function of frequency, the input port cable termination is set to 74 Ohms to simulate the likely conditions of imperfect impedance matching which results in output port isolation that is not infinite. The cable termination at Port2and Port3are 75 Ohms. The resulting transfer response plot shows the high attenuation of the path from Port2to Port3of over 50 dB. The oscillation in the transfer response is due to the changing impedance transformation properties of the 50 ft. coaxial cable with changing frequency (according to standard transmission line theory).

FIG. 3Cshows an output-to-output transfer response as a function of frequency with the cable termination impedance for Port1set to 250 Ohms, for Port2set to 5 Ohms, and for Port3set to 250 Ohms. This configuration corresponds to a simple two leaf tree network in which the root node is terminated with a mismatched high impedance, one leaf node is terminated with a mismatched low impedance, and the other leaf node is terminated with a mismatched high impedance. As described in more detail below, in some implementations network devices112are configured to use a low impedance for transmission and a high impedance for reception. The resulting transfer response plot shows the lowered attenuation of the path from Port2to Port3of around 0 to 10 dB.

FIG. 3Dshows an output-to-output transfer response as a function of input Port1cable termination impedance as it is varied from 5 to 250 Ohms. The frequency for the response shown inFIG. 3Dis assumed to be 15 MNHz. The cable termination impedances of Port2and Port3are the same as in the plot ofFIG. 3C. The resulting transfer response plot shows the dramatic rise in attenuation (or equivalently the fall in transfer response) of the path from Port2to Port3that occurs when the cable termination impedance at the input Port1approaches the 75-Ohm characteristic impedance of the transmission line at which the splitter is designed to have high output port isolation.

Signal Modulation

A coaxial cable network in which one or more are mismatched tends to suffer from increased passband ripple in the frequency domain and increased delay spread in the time domain. Both are artifacts caused by reflection of a signal at a mismatched end of a coaxial cable transmission line. Some high-speed digital communications signal modulation techniques do not tolerate excessive passband ripple or delay spread.

To achieve robust communication performance in the presence of passband ripple and delay spread, the network devices112use Orthogonal Frequency Division Multiplexing (OFDM), also known as Discrete Multi Tone (DMT). OFDM is a spread spectrum signal modulation technique in which the available bandwidth is subdivided into a number of narrowband, low data rate channels or “carriers.” To obtain high spectral efficiency, the spectra of the carriers are overlapping and orthogonal to each other. Data are transmitted in the form of symbols that have a predetermined duration and encompass some number of carriers. The data transmitted on these carriers can be modulated in amplitude and/or phase, using modulation schemes such as Binary Phase Shift Key (BPSK), Quadrature Phase Shift Key (QPSK), or m-bit Quadrature Amplitude Modulation (m-QAM).

In OFDM transmission, data are transmitted in the form of OFDM “symbols.” Each symbol has a predetermined time duration or symbol time Ts. Each symbol is generated from a superposition of N sinusoidal carrier waveforms that are orthogonal to each other and form the OFDM carriers. Each carrier has a peak frequency fiand a phase Φimeasured from the beginning of the symbol. For each of these mutually orthogonal carriers, a whole number of periods of the sinusoidal waveform is contained within the symbol time Ts. Equivalently, each carrier frequency is an integral multiple of a frequency interval Δf=1/Ts. The phases Φiand amplitudes Aiof the carrier waveforms can be independently selected (according to an appropriate modulation scheme) without affecting the orthogonality of the resulting modulated waveforms. The carriers occupy a frequency range between frequencies f1and fNreferred to as the OFDM bandwidth.

Referring toFIG. 4, a communication system400includes a transmitter402for transmitting a signal (e.g., a sequence of OFDM symbols) over a communication medium404to a receiver406. The transmitter402and receiver406can be incorporated into network devices coupled to the coaxial cable network (e.g., as part of a device transceiver). The communication medium404can represent a path from one device to another over the coaxial cable network, or a path through another type of network such as a power line network. Due to their being designed for much lower frequency transmissions, AC wiring exhibits varying channel characteristics at the higher frequencies used for data transmission (e.g., depending on the wiring used and the actual layout). As with mismatched coaxial cable network100, a power line network exhibits distortion due to multipath delay spread. The use of OFDM signals can improve reliability of communication in coaxial cable networks, power line networks, or bridged networks including both coaxial cable and power line sections, as described in more detail below.

At the transmitter402, modules implementing the PHY layer receive an input bit stream from a medium access control (MAC) layer. The bit stream is fed into an encoder module420to perform processing such as scrambling, error correction coding and interleaving.

The encoded bit stream is fed into a mapping module422that takes groups of data bits (e.g., 1, 2, 3, 4, 6, 8, or 10 bits), depending on the constellation used for the current symbol (e.g., a BPSK, QPSK, 8-QAM, 16-QAM constellation), and maps the data value represented by those bits onto the corresponding amplitudes of in-phase (I) and quadrature-phase (Q) components of a carrier waveform of the current symbol. This results in each data value being associated with a corresponding complex number Ci=Aiexp(jΦi) whose real part corresponds to the I component and whose imaginary part corresponds to the Q component of a carrier with peak frequency fi. Alternatively, any appropriate modulation scheme that associates data values to modulated carrier waveforms can be used.

The mapping module422also determines which of the carrier frequencies f1, . . . , fNwithin the OFDM bandwidth are used by the system400to transmit information. For example, some carriers that are experiencing fades can be avoided, and no information is transmitted on those carriers. Instead, the mapping module422uses coherent BPSK modulated with a binary value from the Pseudo Noise (PN) sequence for that carrier. For some carriers (e.g., a carrier i=10) that correspond to restricted bands (e.g., an amateur radio band) on a medium404that may radiate power no energy is transmitted on those carriers (e.g., A10=0).

An inverse discrete Fourier transform (IDFT) module424performs the modulation of the resulting set of N complex numbers (some of which may be zero for unused carriers) determined by the mapping module422onto N orthogonal carrier waveforms having peak frequencies f1, . . . , fN. The modulated carriers are combined by IDFT module424to form a discrete time symbol waveform S(n) (for a sampling rate fR), which can be written as

where the time index n goes from 1 to N, Aiis the amplitude and Φiis the phase of the carrier with peak frequency fi=(i/N)fR, and j=√−1. In some implementations, the discrete Fourier transform corresponds to a fast Fourier transform (FFT) in which N is a power of 2.

A post-processing module426combines a sequence of consecutive (potentially overlapping) symbols into a “symbol set” that can be transmitted as a continuous block over the communication medium404. The post-processing module426prepends a preamble to the symbol set that can be used for automatic gain control (AGC) and symbol timing synchronization. To mitigate intersymbol and intercarrier interference (e.g., due to imperfections in the system400and/or the communication medium404) the post-processing module426can extend each symbol with a cyclic prefix that is a copy of the last part of the symbol. The post-processing module426can also perform other functions such as applying a pulse shaping window to subsets of symbols within the symbol set (e.g., using a raised cosine window or other type of pulse shaping window) and overlapping the symbol subsets.

An Analog Front End (AFE) module428couples an analog signal containing a continuous-time (e.g., low-pass filtered) version of the symbol set to the communication medium404. The effect of the transmission of the continuous-time version of the waveform S(t) over the communication medium404can be represented by convolution with a function g(τ;t) representing an impulse response of transmission over the communication medium. The communication medium404may add noise n(t), which may be random noise and/or narrowband noise emitted by a jammer.

At the receiver406, modules implementing the PHY layer receive a signal from the communication medium404and generate a bit stream for the MAC layer. An AFE module430operates in conjunction with an Automatic Gain Control (AGC) module432and a time synchronization module434to provide data and timing information to a discrete Fourier transform (DFT) module436. After synchronizing and amplifying a received symbol set using its preamble, the receiver406demodulates and decodes the symbols in the symbol set.

After removing the cyclic prefix, the receiver406feeds the sampled discrete-time symbols into DFT module436to extract the sequence of N complex numbers representing the encoded data values (by performing an N-point DFT). Demodulator/Decoder module438maps the complex numbers onto the corresponding bit sequences and performs the appropriate decoding of the bits (including deinterleaving and descrambling).

Any of the modules of the communication system400including modules in the transmitter402or receiver406can be implemented in hardware, software, or a combination of hardware and software.

Network Interface

FIG. 5Aillustrates an exemplary bidirectional AFE module500that serves as a network interface for a network device112that incorporates the functions of both transmitter402and receiver406. The AFE module500uses coupling module502to receive a signal from the coaxial cable111to a receiver AFE module430, and to transmit a signal from a transmitter AFE module428into the coaxial cable111. This approach is a half-duplex approach in which the device112is either in a transmit mode or a receive mode at any given time.

FIG. 5Bshows circuitry for one implementation of a coupling module502. The circuitry includes a wideband toroidial transformer504, transient protection diodes506A and506B, and an F series 75-Ohm female connector508to accept standard RG59 or RG6 coaxial cable. Terminals from the transformer504form a bidirectional signal interface510that includes a differential pair of transmit terminals PL_TXP and PL_TXN from the transmitter AFE module428. These transmit terminals optionally include symmetric resistors with resistance R0to set the output impedance and resulting signal level. The signal interface510also includes a differential pair of receive terminals PL_RXP and PL_RXN to connect to the receiver AFE module430. The effective input impedance of the network device112is selected by setting a resistance in the receiver AFE module430to the appropriate value.

Improved communication performance can be achieved when the output impedance of a network device112driving a signal onto a cable is less than the characteristic impedance of the coaxial cable111.

Some wideband line drivers are operational amplifier circuits with feedback that achieve very low output impedances (a few Ohms or less). In some systems these drivers are matched to a system characteristic impedance using a series resistance equal to the system impedance. A voltage divider is formed by the series matching resistor and the system load impedance. One half of the driver output potential reaches the load resulting in 6 dB signal loss for the matched impedance case.

For communication techniques for which this impedance matching is not necessary (e.g., OFDM) the output impedance of a driver can be reduced to a few Ohms. The resulting loss due to the voltage divider is less than the previous case especially when low impedance loads are encountered. The low impedance driver achieves less loss and in some cases gain for many paths through the coaxial cable network100(relative to the 6 dB loss of a matched impedance driver). For example, an output impedance of about 5 Ohms for a 75-Ohm coaxial cable characteristic impedance provided robust performance for signals in the 2 to 28 MHz frequency range in a test coaxial cable network.

Improved performance can also be achieved when the input impedance of a network device112receiving a signal over a cable is larger than the characteristic impedance of the coaxial cable111. In some preferred implementations, the effective input impedance of the network device112is selected to be at least 1.2, 2, 3, or 10 times larger depending on the desired coupling properties. For example, an input impedance of about 250 Ohms for a 75-Ohm coaxial cable characteristic impedance provided robust performance for signals in the 2 to 28 MHz frequency range in a test coaxial cable network.

Network Bridges

A bridge device116can use any of a variety of techniques to couple signals between the coaxial cable network100and the secondary network120depending on the characteristics of the networks. For example, OFDM signal modulation is well-suited for the nonlinear channel characteristics of both the mismatched coaxial cable network100and a power line network. A bridge device116can couple signals between coaxial cable and power line media “passively” without necessarily changing the signal modulation characteristics. A passive bridge device is able to preserve modulation characteristics of a communication signal such as the shape of the waveform used to modulate data, and therefore does not need to delay a signal for demodulation, buffering, and/or re-modulation.

Alternatively a bridge device116can be an “active” device that demodulates a signal received over one of the networks and buffers the encoded information for subsequent transmission over the other network. An active bridge device can switch between the networks accessing them one at a time. Alternatively, an active bridge can represent two logical network nodes with one operating in the first network (e.g., the coaxial cable network) and the other operating in the second network (e.g., a power line network). This type of active bridge device can potentially communicate in both networks at the same time. Both logical nodes inside the device can be implemented with a single processor and separate physical interfaces. This active approach introduces a delay in the signal as it passes through the bridge device116.

The bridge device116can optionally be a simple coupling device that passes signals between two networks (passively or actively), or it can be incorporated into a fully functional network device112that serves as an origin and destination for transmitted signals as well as a bridge (passive or active).

In implementations in which the secondary network120is a power line communication network, the bridge device116includes components to filter out the low-frequency (e.g., 50 Hz or 60 Hz) power waveform, and components to protect against large transient surges in the power line. The communication signal waveform also carries power, however, the voltage level and corresponding average power of the communication signal (e.g., the amplitude of the OFDM symbols) is much smaller than that of a typical power waveform with a root-mean-square voltage in the range of 120-240 V.

FIG. 6shows a passive bridge600for bridging coaxial cable and power line networks in a house. The passive bridge600safely couples a communication signal (e.g., at 2-28 MHz) between the two networks while blocking the power signal (e.g., at 60 Hz) from crossing form the power line network to the coaxial cable network. The passive bridge600includes a wideband coupling transformer602that couples a differential mode signal in either direction between a coaxial cable interface606(e.g., an F series female coaxial cable connector) and a power line interface608(e.g., AC power plug prongs). In some implementations the transformer602has a 1:1 turns ratio. Alternatively, the transformer602can have a different turns ratio to provide an effective change in impedance. This bidirectional signal coupling enables the coaxial cable network and powerline network be part of the same broadcast domain in which the CSMA/CA MAC protocol operates. The transformer602also serves to block unintentional common mode energy (noise) while passing the desired differential mode signal energy. The transformer602can be fabricated with bifilar turns of wire on a ferrite toroid core. Triple insulated Teflon wire is used to provide safety isolation (with a 3 kV breakdown voltage) between the power line and coaxial cable networks.

The passive bridge600includes high-voltage series capacitors604A and604B (e.g., 0.01 microFrarad capacitors) which act as a high-pass filter to pass the desired high-frequency communication signal and block (or significantly attenuate, e.g., by a factor of 10, 100, or more) the low-frequency power waveform from passing through the transformer to the coaxial cable network100. Capacitors604A and604B with safe failure modes can be used to preserve coupler safety in the event of component failure. Shunt resistors612A and612B (e.g., 200 kOhm resistors) dissipate any residual charge present on the capacitors when the bridge600is unplugged. A high-voltage varistor610maintains a high resistance for voltages within the expected operating range and switches to a low resistance conducting state to clamp large transient arriving on the power line that could exceed the breakdown voltage of the capacitors604A and604B. Alternatively, any of a variety of transient-suppression circuit elements can be used to block (or significantly attenuate) voltage transients, including, for example, a transient voltage suppression diode.

FIG. 7shows an exemplary plastic housing700for the components of the passive bridge600with built-in AC power plug prongs702as the power line interface608. During use, the bridge600plugs into an available AC power outlet in a house. The AC power plug prongs702are non-polarized and may be inserted with either orientation. A length of coax cable (e.g., 3 to 12 feet) may be used to connect an F connector704on the bridge600with an F connector port of the coaxial cable network100.

FIG. 8shows a hybrid coupler800that couples a network device112to either or both of a coaxial cable network and a power line network, and optionally serves as a bridge between the coaxial cable and power line networks. The hybrid coupler800includes a wideband coupling transformer802with four isolated windings. The turns ratio is typically unity for all four windings. Triple insulated Teflon wire is used to provide safety isolation (with a 3 kV breakdown voltage) between the power line, coaxial cable, and the low voltage bidirectional signal interface804. The signal interface804includes a differential pair of transmit terminals TX_P and TX_N that connect to the output of the transmitter AFE module428, and a differential pair of receive terminals RX_P and RX_N that connect to the input of the receiver AFE module430. These four lines are low voltage safety isolated connections.

The hybrid coupler800includes switches806A and806B to select power line only operation, coaxial cable only operation, or hybrid operation on both power line and coaxial cable media. The power line media connection includes the capacitors604A and604B, resistors612A and612B, the varistor610, and the power line interface608, as described above. The coaxial cable media connection includes the coaxial cable interface606, as described above. The switches806A and806B are double pole single throw switches that make or break the differential connections between the coupling transformer802and the power line and coaxial cable media. The switches806A and806B can be set at the time of installation, or alternatively can be controllable via an external switch interface.

The power line and coaxial cable media are bridged together (in the manner of the passive bridge600) when both switches806A and806B are closed. For example, closing both switches allows the network device112to communicate simultaneously on both the power line and coaxial cable networks. Closing both switches in a hybrid coupled network device112at a first node linked to both networks couples the two networks together so that a second node on the power line network can communicate with a third node on the coaxial cable network through the first node as a bridge.

WORKING EXAMPLE

FIG. 9shows a plan view of a residential test site900showing AC power outlets (power line ports PL-1to PL-7) at which devices connect to a power line network, and coaxial cable ports (coaxial cable ports CX-8to CX-11) at which devices connect to a coaxial cable network. The coaxial cable network has the topology of a tree network with two 2-way splitters connected by RG6 type coaxial cable111. A source port CX-8is configured to interface with a source (or “root”) node of the tree network and to distribute a signal to devices connected to the coaxial cable ports CX-9to CX-11representing the leaf nodes of the tree network. The nominal insertion loss from port CX-8to port CX-10or port CX-11was 7 dB, and the nominal insertion loss from port CX-8to port CX-9was 3.5 dB. The AC wiring of the power line network (not shown) forms a shared communication medium such that each power outlet shares a bidirectional communication path with every other power outlet.

The signal attenuation representing the port-to-port transfer response was measured between all pairs of ports(PL-1to PL-7, and CX-8to CX-11). The transfer response was measured in both directions (e.g., transmitting from port CX-8to port CX-9, and transmitting from port CX-9to port CX-8). Since many paths have attenuation that varies with frequency (e.g., exhibiting peaks and nulls) the average attenuation was calculated and recorded.

FIG. 10shows the test setup used to perform the transfer response test measurements. A first test node1002was coupled to either a coaxial cable port1004(one of the 4 ports of the test site900) or a power outlet1006(one of the 7 outlets of the test site900). A second test node1008was coupled to either a coaxial cable port1010(one of the 4 ports of the test site900) or a power outlet1006(one of the 7 outlets of the test site900). One of the test nodes was placed in a transmit mode and the other was placed in a receive mode. If the transmitting node was coupled to a coaxial cable port, then the output impedance of the transmitting node was set to a low value of about 5 Ohms. If the receiving node was coupled to a coaxial cable port, then the output impedance of the receiving node was set to a high value of about 250 Ohms.

Some of the measurements were performed with the coaxial cable and power line networks coupled using a passive bridge600, and some of the measurements were taken with the coaxial cable and power line networks uncoupled (i.e., with the passive bridge600disconnected). In these test measurements, when the source port104was not participating in the measurement it remained disconnected (and therefore terminated with a mismatched open circuit impedance).

FIGS. 11A and 11Bshow grids representing the path attenuation measurements in which the row corresponds to the transmitting port (PL-1to PL-7, and CX-8to CX-11) and the column corresponds to the receiving port (PL-1to PL-7, and CX-8to CX-11). The shading at the intersection of a row and column is proportional to the path attenuation. The shaded squares represent attenuation levels according to the scale1100. Since a port does not transmit to itself the diagonal squares (1 to 1, 2 to 2, etc) do not represent attenuation measurements.

The grid inFIG. 11Ashows attenuation measurements between all pairs of ports and/or outlets with the passive bridge600disconnected such that the power line network and the coaxial cable network are uncoupled. The power line network connectivity is represented by the lower left quadrant (rows1-7, columns1-7) and the coaxial cable network connectivity is represented by the upper right quadrant (rows8-11, columns8-11). The average power line network attenuation is about 40 dB with a wide range of variation. The average coaxial cable network attenuation (with impedance mismatch) is less than 10 dB. The attenuation between networks is 60 dB or more (rows1-7, columns8-11, and rows8-11, columns1-7).

The grid inFIG. 11Bshows attenuation measurements between all pairs of ports and/or outlets with the passive bridge600connecting the power line and coaxial cable networks. The average attenuation between power line outlets remains about the same. The average attenuation between the coaxial cable ports also shows little change. However, the average attenuation between the power line and coaxial cable networks is greatly improved (i.e., reduced attenuation). The average attenuation levels for these power line to coaxial cable and coaxial cable to power line paths (rows1-7, columns8-11, and rows8-11, columns1-7) are similar to those for power line to power line paths (rows1-7, columns1-7), on the order of 40 dB. These new communication paths provide greater convenience and coverage.

Additionally, the communication data rates were measured over these same paths and the average throughput over a set of paths in various network configurations were calculated, as summarized in Table 1 below.

One set of paths for which the average throughput was measured corresponds to the coaxial-to-coaxial paths (rows8-11, columns8-11), with and without the passive bridge600present. Another set of paths for which the average throughput was measured corresponds to the power line-to-power line paths (rows1-7, columns1-7), with and without the passive bridge600present. Another set of paths for which the average throughput was measured corresponds to the power line-to-coaxial paths (rows1-7, columns8-11, and rows8-11, columns1-7), with the passive bridge600present. The average throughput was also measured for all paths (rows1-11, columns1-11) with the passive bridge600present.

The presence of the passive bridge600did not have a large effect on the average throughput of the existing coaxial-to-coaxial and power line-to-power line paths, while greatly increasing the total number of paths available for communicating in the test site900.

Many other implementations other than those described above are within the invention, which is defined by the following claims.