Modem with remote speed-change capability

A modem which can automatically switch between protocols at different signalling rates. The protocal with non-standard signalling rates has a training sequence which initially has distinctive frequency-domain characteristics. Thus, as soon as one modem begins to send a training sequence, the other modem can distinguish which training sequence has been started. Thus, speed changes between protocols at different signalling rates can be remotely initiated.

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protection under the copyright laws of the United States and of other 
countries. As of the first effective filing date of the present 
application, this material is protected as unpublished material. 
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reserves all copyright rights whatsoever. 
BACKGROUND OF THE INVENTION 
The present invention relates to modem systems, which permit digital data 
to be efficiently transmitted over an analog channel. A particularly 
important class of modems is those which transmit data over telephone 
lines. 
The existing telephone network provides such widespread and convenient 
linkage that it is highly desirable to be able to use it for data 
transmission. An important objective in such systems is to transmit data 
at the highest rate possible on the line being used. 
In general, modems transmit pulses at a constant rate (the "signalling 
rate" or "baud rate"), and each pulse carries one of several "symbols." 
For example, if pulses are sent at 2400 pulses per second, and each pulse 
corresponds to one of 64 symbols, the resulting data rate will be 14,400 
bits per second. (This is commonly written as 14400 bps, or 14.4 kbps.) 
That is, selecting one of 64 possible symbols is equivalent to specifying 
6 bits of data, since 64=2.sup.6. 
The set of available symbols is defined by whatever transmission protocol 
is being followed. Specifically, each symbol corresponds to a specific 
phase and amplitude value, with reference to a synchronous carrier. (This 
relation will be discussed in more detail below.) The set of all possible 
symbols is referred to as the "constellation." An example of a 
constellation is shown in FIG. 4. 
Thus, to change the data rate, it is necessary to change the constellation, 
change the signalling rate, or both. 
Changing the constellation is simpler than changing the signalling rate. 
When a pair of modems begin communications, one modem will transmit a 
standard sequence of patterns, which allows the other modem to "tune in" 
to the constellation which will be used. (This period is called the 
"training" period.) The last part of this sequence of patterns includes a 
code which specifies which constellation will be used. 
Thus, for example, a change from 14400 bps to 9600 bps could be 
accomplished by changing constellations, without varying the signalling 
rate. In fact, a modem can be commanded remotely to make such a change. 
This is very advantageous, because it permits modems to shift speeds 
during a communication session. 
Each modem normally includes a filter which is constantly watching for a 
strongly periodic signal. This filter will promptly detect when a training 
sequence is incoming, and initiate reception of the training operation. 
The last part of the training sequence will specify which constellation to 
use. 
Thus, to change to a new constellation, one modem would simply begin to 
transmit the training sequence, instead of data. The other modem detects 
the training sequence, and, at the completion of this training sequence, 
the two modems will be able to communicate using the new constellation. 
Using a constellation of 64 symbols rather than 16, at a signalling rate of 
2400 pulses per second, increases the speed only from 9600 bps to 14400 
bps. Similarly, to increase the data rate to 19200 at the same signalling 
rate, a constellation of 256 symbols would have to be used. Since the 
signal-to-noise ratio of a telephone line is limited, obtaining the high 
resolution required for such a constellation may be difficult. (In 
practice, the constellations actually used are enlarged to permit use of 
trellis coding. For example, 9600 bps protocols normally use a 32-point 
constellation, and 14400 bps protocols normally use a 128-point 
constellation.) 
Therefore, for communication at 19200 bps, many modem protocols have 
slightly increased the signalling rate. Where the number of symbols is 
large, a small increase in the signalling rate can increase the data rate 
as much as a doubling of the number of symbols in the constellation. For 
example, for communication at 19200 bps, increasing the pulse rate by only 
14%, from 2400 Hz (=19200/8) to 2742.86 Hz (=19200/7), means that only 
half as many symbols need be used. If pulses are sent at 2742.8 pulses per 
second, and each pulse can correspond to any one of 128 symbols, the 
resulting data rate will be 19,200 bits per second. (In practice, the 
19200 bps protocol of the presently preferred embodiment actually uses a 
160-point constellation, to permit implementation of coding schemes such 
as trellis coding.) 
However, the use of a higher signalling rate introduces an incompatibility. 
Remotely initiated speed changes may become more difficult, where some of 
the protocols used do not have the same signalling rate. For example, a 
change from 14400 bps to 19200 bps (or vice versa) normally requires a 
change in the signalling rate. 
Previously, modems have had much greater difficulty in detecting data-rate 
changes where a signalling rate change was necessary. Since most 19200 bps 
modems use an increased signalling rate, this difficulty has meant that 
most 19200 bps modems were not able to change speed reliably in response 
to a command from a remote modem. 
A key objective in many telephone-line modem applications (and in other 
modem types as well) is to maximize the net data rate. However, the 
maximum possible data rate is limited by the characteristics of the 
channel. For example, the frequency bandwidth of a telephone line (in the 
United States) is typically only about 3,000 Hz, and the signal-to-noise 
ratio is also severely limited. This channel quality is adequate for voice 
transmission, but makes it difficult to achieve a high data transmission 
rate. From the Shannon theorem, the absolute theoretical maximum data rate 
which could fit within the minimum bandwidth and worst-case 
signal-to-noise standards for dial-up telephone lines (in the United 
States) would be about 30,000 bits per second ("bps"). However, this is a 
theoretical limit, which cannot be readily achieved in practice. Moreover, 
telephone connection quality will vary; some connections will be better 
than the minimum standard, and some will be worse. 
The ability to remotely initiate a speed change, without requiring long 
training times or introducing significant error rates, is very desirable 
in telephone line modems. Since the transmission quality of telephone 
lines varies from line to line, and from minute to minute, it is highly 
desirable that the modem link should be able to adjust to these variable 
conditions. This is particularly desirable at higher maximum transmission 
rates, since a modem which is able to exploit a very good connection (at 
19.2 or even 38.4 kbps) must be able to "fall back" to a much lower data 
if conditions worsen. Similarly, if such a modem has had to operate at a 
lower rate than its maximum, it is advantageous if the modem can "fall 
forward" (change to a higher transmission rate) if conditions improve. 
It should be noted that not every idea discussed in the foregoing 
Background of the Invention section of the present application is 
necessarily prior art. For example, the discussion of technical 
alternatives may be colored by knowledge of some of the inventive concepts 
and their advantages. Moreover, some of the technical alternatives 
discussed may not be "prior art" under the patent laws of the United 
States or of other countries. 
Similarly, the following Summary of the Invention section of the present 
application may contain some discussion of prior art teachings, 
interspersed with discussion of generally applicable innovative teachings 
and/or with specific discussion of the best mode as presently 
contemplated. Statements made in the Summary section do not necessarily 
delimit any of the various inventions claimed in the present application 
or in related applications. Moreover, some statements made in the Summary 
section may apply to some inventive features but not to others. 
SUMMARY OF THE INVENTION 
The present invention provides a high-speed modem with improved flexibility 
for speed changes, including speed changes between protocols which have 
different signalling rates. 
The present invention provides an initial training sequence portion, at the 
higher signalling rate, which is readily distinguishable from the first 
part of the training sequences used at the lower signalling rate. 
In the standard 14400 bps modem protocol (CCITT protocol V.33), the first 
part of the training sequence is selected to provide a large amount of 
energy at the band edges. (This helps to provide rapid acquisition of 
clock phase.) This practice has been followed in many other modem 
protocols too. However, according to the present invention, the first part 
of the training sequence in one mode is defined so that it includes energy 
peaks which are shifted away from the band edges. This goes against the 
conventional wisdom, but has the advantage of providing rapid 
discrimination between incompatible protocols. 
In fact, the presently preferred embodiment transmits an initial pattern 
(to begin training in one possible protocol) which has a spectrum in which 
the carrier frequency is suppressed. The use of such a suppressed-carrier 
pattern provides even greater potential for simple filtering methods to 
distinguish among possible protocols at two (or more) different signalling 
rates. 
A filtering operation is used to distinguish between the two possible types 
of initial training sequence. That is, the use of a modified initial 
portion for the training sequence provides spectral characteristics which 
are very easily recognizable. 
Preferably a differential filtering operation is used, to protect against 
false alarms due to a sudden increase in the signal level (a "gain hit"). 
That is, in the case of FIG. 1A, a sudden increase in the overall signal 
level would also cause an increased power to be detected, in the frequency 
domain, at frequencies C and H. Such events should not be allowed to 
trigger a false decision that a training sequence is being transmitted.

DESCRIPTION OF THE PREFERRED EMBODIMENTS 
The innovative teachings of the present application will be described with 
particular reference to the presently preferred embodiment, wherein these 
teachings are advantageously applied to a 19200 bps modem which uses a 
160-point constellation at a signalling rate of 2743 Hz, and which can 
also fall back to V.33 protocol at 2400 Hz, at 14400 bps. This embodiment 
can also fall back further to a modified protocol, which is very similar 
to the V.33 protocol but which runs at 9600 bps. 
However, it should be understood that this embodiment is only one example 
of the many advantageous uses of the innovative teachings herein. For 
example, the various teachings disclosed herein can optionally be adapted 
to higher- or lower-rate modems, to modems (such as high-speed modems) 
other than telephone-line modems, to facsimile-machine modems, or to other 
M-ary QAM applications which require flexibility in shifting between 
protocols with different signalling rates. 
Data Constellations in M-ary QAM 
The present invention is particularly directed (but not necessarily 
limited) to systems using quadrature amplitude modulation ("QAM"). In QAM, 
two signals are modulated onto the same carrier frequency in phase 
quadrature. Demodulation of the signal (with respect to a synchronous 
carrier) yields two parameters, phase and amplitude. 
The "constellations" referred to above are implemented using M-ary QAM 
(where the number of available symbols at each pulse is referred to as M). 
The set of permissible combinations of phase and amplitude defines the 
constellation, i.e. the set of possible symbols. 
To decode the symbols, it is first necessary to demodulate each received 
pulse, with reference to a synchronous carrier frequency. This provides a 
phase and amplitude estimate for each received pulse. Decision logic then 
finds the best match of this phase and amplitude to one of the points in 
the constellation. A decoding operation then translates the received 
constellation point, to define log.sub.2 M bits of data output. 
For example, one way to encode a data rate of 19.2 kbps would be to use a 
constellation with 128 permissible data points ("symbols"), at a pulse 
rate of 2742.8 (=19,200/7) sec.sup.-1. Such a constellation would normally 
include all combinations of the 12 permissible nonzero values of the 
in-phase component and the 12 permissible nonzero values of the quadrature 
component, except for the four blocks of four points at each of the four 
corners of the constellation. Thus, the general appearance of the 
constellation would be cross-shaped. (Excluding a few of the 
phase/amplitude combinations near the corners of the constellation helps 
to reduce error rate.) 
In the sample constellation of FIG. 4, the two orthogonal signal components 
modulated onto the carrier are represented as an in-phase ("I") component 
and a quadrature ("Q") component. Each of these components can take one of 
14 levels. Since some combinations of levels are excluded, the total 
number of available I/Q combinations is 160. Note that, although only 7 
bits of data are required to be encoded at each symbol, the number of 
points in the constellation of FIG. 4 is actually 160, not 128. These 
additional points are used to implement a trellis coding protocol. 
General Architecture of Modem Receiver 
The organization and key elements of the modem system context of the 
presently preferred embodiment will now be described. Since the 
demodulation and decoding functions are generally more difficult than the 
encoding and modulating functions, the following description will 
primarily emphasize the operations in the modem receiver. The 
complementary operations in modem transmitter are much simpler, and will 
be immediately apparent to those skilled in the art. 
FIG. 5 shows the general architecture of a modem receiver. An incoming 
signal 502, received from a channel of some sort (e.g. a telephone line), 
is passed through an anti-aliasing filter stage 504 (which may include 
preamplification if needed). The filtered signal is further level-adjusted 
by an analog automatic gain control (AGC) stage 506 (which is governed by 
a gain-control signal 505). The analog signal is now supplied to a 
sampling converter 508, which samples it at intervals determined by a 
control signal 507, and provides a corresponding digital output. The 
sampled signal 509 is now provided to an AGC control stage 514, which 
provides feedback 507 to the analog amplifier 506 based on the digital 
signal values. The sampled signal 509 is also provided to a clock recovery 
stage 512. This stage monitors the timing of the separate pulses. (The 
pulse timing is referred to as the "baud clock.") Thus, clock recovery 
stage 512 tracks the phase of the baud clock, and accordingly provides a 
timing signal 507 to the sampling converter 508. 
The sampled signal 509 is now conditioned by bandpass filter 510, Hilbert 
transform stage 516, and adaptive equalizer stage 518. (Note that the 
Hilbert transform stage 516 provides a complex output, so that two signal 
streams are shown from Hilbert transform stage to decoding stage 522.) The 
signal thus conditioned is demodulated by demodulation stage 520, with 
reference to the recovered carrier signal 519 provided by carrier recovery 
stage 524. The demodulated values are then decoded (stage 522), to produce 
digital output data 526. The decoding operation 522 preferably uses coding 
logic (including multidimensional coding, trellis coding, convolutional 
encoding) as described, for example, in Ungerboeck, Trellis-coded 
Modulation Redundant Signal Sets: Part 2: State of the Art, IEEE 
Communications Magazine, February, 1987; in Wei, Trellis-coded Modulation 
with Multidimensional Coding, 33 IEEE Trans'ns in Information Theory 483 
(July 1987); and in Gersho and Lawrence, Multi-dimensional Signal 
Constellations for Voice Band Data Transmission, 2 IEEE Journal on 
Selected Areas in Communications 687 (Sept. 1984); all of which are hereby 
incorporated by reference. 
An overview of the elements of a modem may be found, for example, in 
Chevillat et al., "Rapid Training of a Voiceband Data-Modem Receiver 
Employing an Equalizer with Fractional-T Spaced Coefficients," 35 IEEE 
Transactions on Communications 869 (1987)(which is hereby incorporated by 
reference). A complete modem receiver will normally include automatic gain 
control (AGC) circuitry, an adaptive equalizer, sampling and clock 
recovery stages, carrier recovery and demodulation stages, and a decoding 
stage. The general outline of all of these is well-known to those skilled 
in the art. Additionally, a complete modem will also include a transmitter 
side, which must include encoding and modulation stages, as well as 
control logic. 
Many modems use error-detection or error-check-and-correct algorithms, in 
which check bits are encoded with the data. A wide variety of such 
algorithms are known. However, in any case, it is always desirable to 
reduce the raw pulse error rate. In an M-ary QAM system, achieving a low 
raw error rate requires an accurate estimation of the phase of each pulse. 
One known way to recover the clock phase of a QAM telephone line modem is 
described (e.g.) in D. N. Godard, "Passband Timing Recovery in an 
All-Digital Modem Receiver," IEEE Transactions on Communications, Vol. 
COM-26, No. 5, May 1978 (which is hereby incorporated by reference). 
However, in the system context of the presently preferred embodiment, this 
function is actually performed using (1) a stage which provides an initial 
estimate of clock phase; (2) a smoothing filter, which filters the initial 
estimate to reduce jitter levels; and (3) a frequency offset tracker, 
which can track the incoming clock even when a significant frequency 
offset exists. (Extensive discussion of this technique is found in U.S. 
patent application Ser. No. 141,499 of common assignee, which is hereby 
incorporated by reference.) 
Note that clock recovery is not the same thing as carrier phase recovery. 
Clock recovery provides the correct timing to sample the incoming signal 
to correctly recover the symbol being transmitted at each baud interval. 
The carrier recovery process, by contrast, provides a phase reference to 
demodulate the sampled data, and thereby decode the symbol. (Extensive 
discussion of an advanced technique for carrier recovery, which is 
actually used in the system context of the presently preferred embodiment, 
is found in U.S. patent application Ser. No. 090,483 of common assignee, 
which is hereby incorporated by reference.) 
Hardware and Software Implementation 
FIG. 6 schematically shows the hardware structure, including three signal 
processing chips and one control processor, which is preferably used to 
implement the modem of the present invention. In this layout, two digital 
signal processing chips (both TMS320C25) are used at location 202. One 
performs most of the receiver functions except decoding, and the other 
decodes the demodulated signals. Digital signal processing chip 204 is 
preferably an NEC 7720, and performs the transmit functions. 
Microprocessor 206 is a general-purpose performs control overhead tasks, 
and is preferably an Intel 8031. Custom LSI chip 208 implements a 
programmable divider, to serve as a local oscillator. Analog chip 209 
performs the major analog functions, including anti-aliasing filtering, 
sampling, and conversion. 
Training and Re-Training 
The functions performed during training will now be discussed. The way in 
which training sequences such as those of FIGS. 3A and 3B permit any one 
of two or more possible signalling rates to be acquired will then be 
discussed. 
General Function of Training Period 
The general purpose of the training period is to permit the transmitting 
and receiving modems to modulate and demodulate accurately with reference 
to a common data constellation, such as that shown in FIG. 4. That is, 
when one modem (whichever one is transmitting) sends a signal 
corresponding to one of the 160 points in the data constellation of FIG. 
4, it is necessary that the other modem (the one which happens to be 
receiving) should identify the received signal as corresponding to the 
same one of the data points as was intended to be sent. 
To attain this close correspondence, a predetermined "pseudo-random" 
sequence of points is sent during the training period. This pseudo-random 
sequence includes only a few points of the constellation. Instead, a 
reduced constellation with only four points is normally used, such as 
points A, B, C, and D in FIG. 2. 
This pseudo-random sequence is selected to provide a broad frequency 
spectrum, just as random data would. However, since this pseudo-random 
sequence is known a priori, the receiving modem can compare its estimates 
with the correct values in this sequence, to derive appropriate 
corrections. (In practice, the parameters of the equalizer 518 are 
adjusted to achieve these corrections.) 
However, before the pseudo-random sequence is sent, an initial sequence is 
sent, to advise the receiving modem that a training sequence is being 
initiated. 
Of course, many variations in training sequences can be used, as is well 
known to those skilled in the art. Thus, the innovative teachings set 
forth herein can be adapted for use in a wide variety of other contexts. 
Initializing Training Under V.33 
FIG. 3A shows the training sequence (defined by CCITT protocol V.33) used 
to initiate modem communications at 14400 bps, under the V.33 protocol. In 
the V.33 protocol, as in many modem protocols, training is initiated by a 
repetitive sequence of dots in a reduced training constellation (like that 
of FIG. 2). 
During the period marked "ALT," an alternating sequence of symbols is 
transmitted. (In the presently preferred embodiment, and in accordance 
with the V.33 protocol, this sequence is ABABABAB . . . , where A, B, C, 
and D are the four points shown in the reduced constellation of FIG. 2.) 
Normally, a high-speed telephone-line modem will contain filters which 
operate essentially continuously, while the modem is in data mode, and 
which are programmed to detect strongly periodic signals. Such a signal 
could result from the alternating sequence of symbols (ABABAB . . . ) 
which begins a training period. Thus, if such a strongly periodic signal 
is received for a sufficiently long time, the receiving modem can 
recognize that a new training sequence has been initiated. The receiving 
modem can then allow itself to be trained in accordance with the protocol 
is being transmitted. 
During the period marked "RANDOM," the pseudorandom sequence of symbols A, 
B, C, and D is transmitted. 
During the period marked "Rate Seq," a short coded sequence is transmitted 
which identifies whether the ensuing data will be transmitted using a 
32-point constellation (9600 bps) or a 128-point constellation (14400 
bps). 
In the presently preferred embodiment, a slight modification of the V.33 
training sequence is used to initiate communication at 9600 bps. The V.33 
standard provides two bits of data which are encoded into the rate 
sequence portion. One of these bits specifies that 12000 bps is permitted, 
and the other specifies that 14400 bps is permitted. However, the V.33 
standard does not specify any result when these two bits specify that 
neither 12000 nor 14400 bps is permitted. In the presently preferred 
embodiment, this condition is used to specify 9600 bps communication. 
During 9600 bps communication, the presently preferred embodiment uses the 
V.32 constellation. 
Initializing at New Signalling Rate 
FIG. 3B shows a novel training sequence, which is used to initiate modem 
communications at 19200 bps in the presently preferred embodiment. The 
periods marked "ALT" and "RANDOM" are generally similar to the 
corresponding periods in FIG. 3A, except that they occur at a higher 
signalling rate (2742.8 per second instead of 2400 per second). During the 
period marked "DDOT," a "double-dotted" sequence is transmitted. (In the 
presently preferred embodiment, this sequence is AACCAACCAACC . . . , 
where A, B, C, and D are the four points shown in the reduced 
constellation of FIG. 2.) 
An outstanding advantage of this DDOT period is that the spectral 
characteristics of the DDOT period at 2743 Hz are very different from 
those of the ALT period at 2400 Hz. FIG. 1A shows a diagram of frequency 
versus power, for the ABABAB . . . pattern at 2400 Hz. This spectrum is 
very simple: it includes a spike at the carrier center frequency, and two 
other narrow peaks at 1200 Hz (i.e. half the signalling rate) above and 
below the center frequency. FIG. 1B shows a diagram of frequency versus 
power, for the AACCAACCAACC . . . pattern at 2742.8 Hz. This spectrum too 
is very simple: it includes essentially no energy at the carrier center 
frequency (set at 1850, in this example), and only two narrow peaks, which 
are separated from the center frequency by .+-. one-quarter of the 
signalling frequency (2742.8/4). (The use of 180.degree. alternations in 
this initial pattern provides a spectrum in which the carrier is 
suppressed.) 
As shown by the dotted envelopes overlaid on the spectra of FIGS. 3A and 
3B, a very simple filtering operation can be used to distinguish between 
the two possible types of initial training sequence. That is, the use of a 
modified initial portion for the training sequence provides spectral 
characteristics which are very easily recognizable. 
In the presently preferred embodiment, four filters are operated 
essentially continuously: one tracks the lower band-edge; one tracks the 
upper band-edge; and two are used to detect frequency spikes which would 
result from training at a different signalling rate. Three of these 
filters (all but the filter which tracks the upper band-edge) are used to 
detect speed-change training. Thus, while operating under the V.33 
protocol (as shown in FIG. 1A), these four filters are operated at 
frequencies B (600 Hz), C (1164 Hz), F (2536 Hz), and G (3000 Hz). The 
onset of a training signal at 19200 bps is identified if, while data lock 
has been lost, frequencies C and F show power output above a certain 
(high) threshold while frequency B shows power output below a certain 
(low) threshold, for a specific minimum duration. 
In the presently preferred embodiment, these intensity measurements are 
made by a digital filtering operation which is upstream of the equalizer. 
(However, it is downstream of the automatic gain control (AGC) function.) 
The specific threshold levels within the digital filtering operation will 
now be specified, for completeness of description, in the manner in which 
those levels are actually specified in the presently preferred embodiment. 
The analog to digital converter provides output values on an arbitrary 
scale of 0(H) to 7FFF(H). Thus, a "full scale" analog input level is 
assigned the value of 7FFF(H), and any larger input level would be 
clipped. The AGC circuit is set to adjust overall average level of the 
signal to a level which, within this scale, corresponds to a value of 
4800(H). Within this scale, the threshold tests applied are: 
the high threshold test is positive if the output of the particular 
frequency bin being tested continuously exceeds a scaled value of 4800(H) 
for 100 bauds (i.e. for the duration of 100 incoming pulses); 
the low threshold test is positive if the output of the particular 
frequency bin being tested is continuously less than a scaled value of 
1000(H) for 100 bauds. (Numbers with an H after them are expressed in 
hexadecimal notation. For example, F(H)=15, 10(H)=16, 4800(H)=18,432, 
7FFF(H)=32,767, and 1000(H)=4096.) 
Once a changeover to reception at 19200 bps has been made, these four 
filters are reprogrammed, so that they are tracking frequencies A (478 
Hz), B (600 Hz), D (1800 Hz), and H (3221 Hz) respectively. The decision 
to change speeds is made if (while data lock has been lost) large power 
levels are detected at frequencies B and D, and much lower power level is 
detected at frequency A. 
In the presently preferred embodiment, the bandwidth of these four filters 
is also switched. When the signalling rate is 2400 Hz, each of these 
filters has a bandwidth of 50 Hz, and when the signalling rate is 2743 Hz, 
each of these filters has a bandwidth of 57 Hz. (The bandwidth is 
specified in Hertz, as is customary with digital filters.) 
Of course, the particular frequencies used are determined by the choice of 
carrier frequency and signalling rate. For example, the spectrum of FIG. 
1B assumes a carrier center frequency (frequency E) of 1850 Hz. 
Frequencies C and F are equal to frequency E plus or minus one quarter of 
the signalling rate, and frequencies A and H are equal to frequency E plus 
or minus one half of the signalling rate. As will be recognized by those 
skilled in the art, a given data rate can be achieved using a wide variety 
of carrier frequencies and signalling rates. 
Thus, a double-dot initial training sequence could be introduced for rapid 
discrimination in systems using other protocols, even if the carrier 
frequency and signalling rates were quite different from those of the 
presently preferred embodiment. 
It should also be noted, even more generally, that the teachings of the 
present invention can optionally be adapted to use still other patterns, 
which differ from the double-dot pattern and also from the normal 
alternating-dot training pattern. Such other patterns can be chosen, 
within the context of a simplified constellation subset used for training, 
to provide frequency-domain spikes which are far removed from the carrier 
frequency and band edges. 
For example, a "triple-dot" pattern of AAACCCAAACCC . . . (at the same 
carrier frequency and signalling rate) would have a spectrum somewhat 
similar to that of FIG. 1B, except that the two spikes C and F would be 
shifted to frequencies of 1393 Hz and 2307 Hz. (That is, in this example, 
the two frequency spikes of the triple-dot pattern would be separated from 
the center frequency by one-sixth of the signalling rate, rather than 
one-quarter.) 
Other repeating patterns of dots within a reduced constellation subset 
could less preferably be used, instead of the double-dot pattern of the 
presently preferred embodiment, to provide a spectrally distinctive 
initial segment. For assessing such alternative candidates, the preferred 
criteria are that a simple repeating pattern should be used, which does 
not carry substantial energy at the band edges. Preferably, this pattern 
should also not carry substantial energy at the carrier center frequency. 
In addition, it should be noted that some repeated dot patterns will differ 
only by a constant phase rotation, and therefore have the same 
frequency-domain characteristics. For example, a double-dot pattern of 
BBDDBBDDBBDD . . . would have the same spectrum as the AACCAACCAACC . . . 
double-dot pattern of the presently preferred embodiment, and could 
therefore readily be substituted. 
Note that, to avoid false initiation of speed-change, the filtering 
operation is done differentially. That is, while operating at 14400 bps, a 
speed change up to 19200 bps operation would be initiated only if the 
energy at frequencies C and F is high AND the energy at frequency B is 
low. This provides stability in the speed change operation. In particular, 
this differential filtering protects against false alarms due to an 
increase in noise or attenuation. 
However, note that the upper band-edge frequency G (3000 Hz in this 
example) is not used for this decision, even though this frequency is 
being filtered and tracked. This is because the varying characteristics of 
the channel may sometimes impose some high-frequency roll-off. The energy 
at the upper band edge is therefore not as reliable a gauge. 
From the foregoing description, it may be seen that the logical operations 
required to test the outputs from these filters are minimal. In the 
presently preferred embodiment, these tests are performed in the 
general-purpose DSP chips 202. When the DSP chips 202 detects that one of 
the foregoing decision thresholds has been met, the DSP chip sends a 
message to the microprocessor 206. 
Once the general-purpose microprocessor 206 detects that a signalling-rate 
change has been commanded, it commands the receiver chip to change the 
signalling rate and the filter parameters. However, it will still be 
necessary for the modem at the other end to send a second training 
sequence, before the receiver is able lock in to the new constellation. 
If the training operation is successful, the receiving modem will send an 
acknowledge signal to the modem which initiated the training. 
In the presently preferred embodiment, the transmitting modem will make 
only three attempts to train. If three attempts are not successful, the 
transmitting modem will fall back to a lower speed (if one is available). 
Thus, the innovative teachings set forth in the present application are 
particularly advantageous in modems which can shift between V.33 and 
higher-rate standards. However, these teachings can also be used to 
provide greater flexibility in the number of protocols to which a modem is 
able to interface to. For example, the innovative teachings set forth 
above could even be adapted to permit a modem to discriminate between two 
different 19200 bps protocols on the fly. Such applications may be 
particularly advantageous in the future, since no standard protocol for 
19200 bps communication has yet been universally accepted. 
As will be recognized by those skilled in the art, the innovative concepts 
described in the present application can be modified and varied over a 
tremendous range of applications. Accordingly, the scope of the innovative 
concepts is not limited except by the claims.