Ghost cancellation reference signal with bessel chirps and PN sequences, and TV receiver using such signal

Composite ghost cancellation reference (GCR) signals that make available both a chirp and a PN sequence during the same VBLI in each successive field facilitates the more rapid and efficient calculations of ghost cancellation and of equalization, on a continuing basis. A television receiver for use with such composite GCR signals includes means for separating the chirp and PN sequence portions of the GCR signals from the remainder of the composite video signal, a ghost cancellation filter and an equalization filter connected in cascade to respond to the composite video signal and provided each with adjustable filtering weights, means responding to the separated chirp portions of the GCR signals to calculate a discrete Fourier transform (DFT) therefrom, means responding to that DFT to determine the adjustable filtering weights of the ghost cancellation filter, and means responding to the separated PN sequences to determine the adjustable filtering weights of the equalization filter.

BACKGROUND OF THE INVENTION

Subcommittee T-3 of the Advanced Television Systems Committee has been meeting to determine a GCR signal for use in the United States. The GCR signal will be a compromise based from two GCR signals, one using Bessel pulse chirp signals as proposed by U.S. Philips Corp. and one using pseudo noise (PN) sequences as proposed by the David Sarnoff Research Center (DSRC) of Stanford Research Institute. The GCR signals are inserted into selected vertical blanking intervals (VBIs). The GCR signals are used in a television receiver for calculating the adjustable weighting coefficients of a ghost-cancellation filter through which the composite video signals from the video detector are passed to supply a response in which ghosts are suppressed. The weighting coefficients of this ghost-cancellation filter are adjusted so it has a filter characteristic complementary to that of the transmission medium giving rise to the ghosts. The GCR signals can be further used for calculating the adjustable weighting coefficients of an equalization filter connected in cascade with the ghost-cancellation filter, for providing an essentially flat frequency spectrum response over the complete transmission path through the transmitter vestigial-sideband amplitude-modulator, the transmission medium, the television receiver front-end and the cascaded ghost-cancellation and equalization filters.

In the conventional method for cancelling ghosts in a television receiver, the discrete Fourier transform (DFT) of the ghosted GCR signal is divided by the DFT of the non-ghosted GCR signal (which latter OFT is known at the receiver from prior agreement with the transmitter) to obtain as a quotient the DFT transform of the transmission medium giving rise to ghosting; and the inverse DFT of this quotient is then used to define the filter weighing coefficients of a compensating ghost-cancellation filter through which the ghosted composite video signal is passed to obtain a de-ghosted composite video signal. To implement the DFT procedure efficiently, in terms of hardware or of calculations required in software, an integral power of two equal-bandwidth frequency bins are used in the DFT. The distribution of energy in the Philips chirp signal has a frequency spectrum extending continuously across the composite video signal band, in contrast to the DSRC PN sequence in which the distribution of energy does not extend continuously across the composite video signal band, but exhibits nulls in its frequency distribution. Accordingly, when the number of equal-bandwidth frequency bins in the DFT is reduced in order to speed calculation time, more accurate ghost cancellation is obtained with the chirp than with the PN sequence as GCR signal, the inventors observe.

During official testing by the Subcommittee, the DSRC GCR signal has exhibited somewhat better performance in regard to equalization of the passband after ghosting, which some experts including the Philips engineers, attribute to better filter hardware.

Theoretically, equalization calculated over an entire active portion of the VBI, proceeding from the PN sequence, has an accuracy substantially the same as the accuracy available calculating equalization from the chirp signal. The entire length of the Philips chirp signal is needed to have the requisite information to implement equalization over the full composite video signal band, The PN sequence contains pulse transitions each of which transitions has substantially the entire frequency spectrum contained therein. The PN sequence contains many pulse transitions, each of which transitions has component frequencies extending over substantially the entire frequency spectrum. This property of the PN sequence, the inventors observe, permits the calculation of equalization taking samples at a prescribed sampling density only over a limited extent of the GCR signal. Taking samples over only a portion of the GCR signal causes some loss in the accuracy with which equalization can be calculated, particularly under poor signal-to-noise conditions. However, since the number of samples involved in the calculation of weighting coefficients for the equalization filter is reduced, there can be an appreciable increase in the speed with which equalization can be calculated, presuming the calculation is done using an iterative method such as least-mean-squares error reduction. Also, there is reduced complexity, in terms of hardware or of calculations required in software, with regard to calculating the equalization filter weighting coefficients.

The composite GCR signals comprised of chirps and PN sequence signals that have thus far been proposed do not make available both a chirp and a PN sequence during the same VBLI scan line.

SUMMARY OF THE INVENTION

The inventors observe that making both a chirp and a PN sequence available during the same VBLI scan line in each successive field, facilitates the more rapid and efficient calculations of ghost cancellation and of equalization, on a continuing basis, particularly when the transmission medium exhibits continual change—e.g., during the rapidly changing ghost conditions caused in over-the-air transmissions by overflying aircraft. A television receiver embodying the invention includes means for separating the chirp and PN sequence portions of the ghost cancellation reference (GCR) signal from the remainder of the composite video signal, a ghost cancellation filter and an equalization filter connected in cascade to respond to the composite video signal and provided each with adjustable filtering weights, means responding to the separated chirp portion of the GCR signal to calculate its discrete Fourier transform (DFT), means responding to that DFT to determine the adjustable filtering weights of the ghost cancellation filter, and means responding to the separated PN sequence to determine the adjustable filtering weights of the equalization filter.

DETAILED DESCRIPTION OF THE INVENTION

FIGS. 1A,1B,1C and1D show the ghost cancellation reference signals in selected scan lines of the vertical blanking intervals of four successive fields of video. Insertion may be into any one of the 11ththrough 20thscan lines of each field, the present preference being to replace the vertical interval reference (VIR) signal currently used in the 19thscan line of each field. To simplify the description that follows, insertion of GCR signal into the 19thscan line of each field will be assumed by way of specific illustration.

The ghost cancellation reference signals ofFIGS. 1A,1B,1C and1D begin with horizontal synchronization pulses11,21,31and41, respectively, which pulses are shown as being negative-going. The leading edges of the horizontal synchronization pulses are considered to be the beginning of VBLI scan lines that are each of 63.55 microsecond duration in NTSC standard television signals. The horizontal synchronization pulses11,21,31and41are respectively followed during ensuing back-porch intervals by chroma bursts12,22,32and42. The plus and minus signs near the chroma bursts12,22,32and42indicate their relative polarities respective to each other, per the NTSC standard.

Bessel pulse chirps13,23,33and43each of33microsecond duration begin 12 microseconds into the VBLI scan lines ofFIGS. 1A,1B,1C and1D, respectively. The arrows associated with each of these chirps is indicative of its relative polarity with respect to the other chirps; chirp polarity is shown as alternating from frame to frame. These chirps swing plus/minus40IRE from30IRE “gray” pedestals which extend from 12 to 48 microseconds into these VBLI lines. The gray level of the pedestals, the plus/minus swing of the chirps, the duration of the pedestals and the duration of the chirps have been specified to correspond as closely as possible to the Philips system that has been officially tested; and design variations may be expected to occur should the compromise GCR signals described herein be adopted by the Subcommittee as their official recommendation for a standard.

Beginning at 51 microseconds into the VBLI scan lines ofFIGS. 1A,1B,1C and1D127-sample PN sequences14,24,34and44respectively occur. Each of the PN sequences14,24,34and44is of the same 9-microsecond duration as the others. The PN sequence in the final field of each frame is of opposite polarity from the PN sequence in the initial field of that frame and is of the same polarity as the PN sequence in the initial field of the next frame, as indicated by the arrows associated with respective ones of the PN sequences14,24,34and44. These PN sequences have −1 and +1 values at −15 IRE and +95 IRE levels respectively. These PN sequences have been specified to correspond as closely as possible to the DSRC system that has been officially tested; and design variations may be expected to occur should the compromise GCR signals described herein be adopted by the Subcommittee as their official recommendation for a standard.

There has been opinion within the Subcommittee that the Bessel pulse chirp should be shortened to 17 microsecond duration so ghosts of up to 40 microsecond delay can be cancelled without the restriction that the VBI line following that containing the GCR signal having not to have information therein that changes from field to field. If the Bessel pulse chirp is shortened, the PN sequence could be made to be 255 pulse sample times, rather than 127 pulse sample times, in length. Adjustments to the compromise GCR signals described herein may be made so the swings of the Bessel pulse chirp and the PN sequence correspond, with suitable adjustment of the gray pedestal, if appropriate. The inventors favor the chirp swing being increased to extend over the range between the −15 IRE and +95 IRE levels and the gray pedestal being set at 40 IRE. The lesser range for the chirps was chosen by the Philips engineers for fear of overswing under some conditions, but the inventors believe that i-f amplifier AGC will forestall such overswing. Extending the gray pedestal to the beginning of the PN sequence will then provide a signal that when low-pass filtered and subsequently gated during the mid-portion of the scan line will provide a level that is descriptive of 40 IRE level and can be used for automatic gain control of the composite video signal.

FIG. 2shows the separated Bessel pulse chirp waveform that results when the GCR signals from two successive fields that are in two successive frames are differentially combined, assuming that the GCR signals are of the sort shown inFIGS. 1A,1B,1C and1D. A separated Bessel pulse chirp waveform perFIG. 2results when the GCR signals ofFIGS. 1B and 1Care differentially combined. A separated Bessel pulse chirp waveform perFIG. 2also results when the GCR signals ofFIGS. 1D and 1Aare differentially combined. A separated Bessel pulse chirp waveform perFIG. 2also results when the sum of the GCR signals ofFIGS. 1A and 1Bis differentially combined with the sum of the GCR signals ofFIGS. 1C and 1D.

FIG. 3shows the waveform that results when the sum of the GCR signals ofFIGS. 1A and 1Dis differentially combined with the sum of the GCR signals ofFIGS. 1B and 1C. The Bessel pulse chirp waveform, the “gray” pedestal and the chroma burst are supressed in this signal; and DC information concerning 0 IRE level is lost. The PN sequence is maintained as a separated PN sequence signal.

FIG. 4shows in block schematic form a television transmitter for NTSC color television signals into which are inserted GCR signals perFIGS. 1A,1B,1C and1D. A processing amplifier50generates composite video signals proceeding from color video signals and synchronizing signals. By way of example, the color video signals may be red (R), green (G) and blue (B) signals from a studio color camera and the synchronizing signals may be from a studio sync generator that also supplies synchronizing signals to the studio color camera. Alternatively, the color video signals may be from a remote location and the synchronizing signals furnished by a genlock connection. Or, if the local transmitter is a low-power transmitter re-broadcasting signals received over-the-air from a distant high-power transmitter, the color video signals may be generated by demodulating the received composite video signal and the synchronizing signals may be separated from the received composite video signal.

The processing amplifier50is shown as including a crystal oscillator51furnishing oscillations at eight times color carrier frequency fc, a counter52for counting the number of these oscillations per horizontal scan line, a counter53for counting scan lines per field, and a counter54for counting modulo-four successive fields of video signal. The processing amplifier50supplies its composite video output signal as a first input signal to an analog selector switch55. The output signal from the analog selector switch55is supplied to a video modulator56to control the vestigial-sideband amplitude modulation of the video carrier.

Sound signal is supplied to a frequency modulator57. The modulated video and sound carriers are amplified by radio-frequency ampliers58and59, respectively, and the output signals from the ampliers58and59are combined in a coupling network60to a broadcast antenna6D. A number of variants of the conventional television transmitter arrangements described in this and the previous paragraph are known to those familiar with television transmitter design.

The analog selector switch55corresponds to that previously known for inserting the vertical interval reference (VIR) signal. A decoder62detects those portions of the count from the counter52associated with the “active” portions of horizontal scan lines—i. a., the portions of horizontal scan lines exclusive of the horizontal blanking intervals—to generate a logic ONE. A decoder63responds to the scan line count from the counter53to decode the occurence of the 19thscan line in each field and generate a logic ONE. An AND gate64responds to these logic ONEs occurring simultaneously to condition the analog selector switch55to select a second input signal for application to the video modulator56, rather than the composite video signal furnished from the processing amplifier50to the analog selector switch55as its first input signal. This second signal is not the VIR signal, however, but is in successive fields successive ones of the GCR signals depicted inFIGS. 1A,1B,1C and1D (or, alternatively, inFIGS. 7A,7B,7C and7D).

These GCR signals are stored in digitized form in a read-only memory65. A first portion of the address for the ROM65is supplied from the counter54, the modulo-four field count selecting which of the GCR signals depicted inFIGS. 1A,1B,1C and1D is to be inserted in the current field. A second portion of the address for the ROM65is supplied from the counter52and scans the selected one of the GCR signals depicted inFIGS. 1A,1B,1C and1D. The digitized GCR signal read from the ROM65is supplied to a digita-to-analog converter66. The resulting analog GCR signal is supplied as the second input signal to the analog selector switch55for insertion into the “active” portion of the 19thline of the field.

FIG. 5depicts a television receiver arranged to receive television signals incorporating the ghost cancellation reference signals ofFIGS. 1A,1B,1C and1D. Television signals collected by an antenna70are amplified by a radio-frequency amplifier71and then down-converted to an intermediate frequency by a converter72. An intermediate-frequency amplifier73supplies to a video detector74and to a sound detector75amplified response to the intermediate-frequency signals from the converter72. The sound detector75demodulates the frequency-modulated sound carrier and supplies the resulting sound detection result to audio electronics76. The audio electronics76, which may include stereophonic sound detection circuitry, includes amplifiers for supplying amplified sound-descriptive electric signals to loudspeakers77and78.

The video detector74supplies analog composite video signal to an analog-to-digital converter79, to a burst detector80, to a horizontal sync separator81and to a vertical sync separator82. The separated horizontal synchronizing pulses from the horizontal sync separator81and the separated vertical synchronizing pulses from the vertical sync separator82are supplied to kinescope deflection circuitry83, which generates deflection signals for a kinescope84. A burst gate generator85generates a burst gate signal an appropriate interval after each horizontal sync pulse it is supplied from the horizontal sync separator81. This burst gate signal keys the burst detector80into operation during chroma burst interval. The burst detector80is included in a phase-locking loop for a phase-locked oscillator86. The phase-locked oscillator86oscillates at a frequency sufficiently high that the analog-to-digital converter79sampling the analog composite video signal from the video detector74once with each oscillation oversamples that signal. As is well-known, it is convenient from the standpoint of simpler digital hardware design that phase-locked oscillator86oscillate at a frequency that is an integral power of two greater than the 3.58 MHz color subcarrier frequency. Sampling chroma signals four or eight times per cycle is preferred.

The separated horizontal sync pulses from the horizontal sync separator81are supplied to a scan line counter87for counting, the scan line count from which counter87is reset to zero at the outset of each vertical sync interval by separated vertical sync pulses from the vertical sync separator82. Indication in the count from the counter87of the occurence of the 19thscan line in each field is detected by a decoder88. Indication in the count from the counter87of the occurence of the 20thscan line in each field is detected by a decoder89. The occurences of the 19thand 20thscan lines in each field is signaled to a GCR signal capture processor90, which captures the GCR signals in the 19thscan line of each field of digital composite video signal from the analog-to-digital converter79.

This capturing process will be described in greater detail in connection with the description of FIG.6.

The GCR signal capture processor90includes circuitry for separating the Bessel pulse chirp portion of the captured GCR signals, which portion is supplied to a ghost-cancellation filter weight computer91. The GCR signal capture processor90also includes circuitry for separating the PN sequence portion of the captured GCR signals, which portion is supplied to an equalization filter weight computer92. The digitized composite video signal from the analog-to-digital converter79is supplied via a cascade connection of a ghost-cancellation filter93and an equalization filter94to a luma/chroma separator96. The ghost-cancellation filter93has filtering weights adjustable in response to results of the computations by the ghost-cancellation filter weight computer91, and the equalization filter94has filtering weights adjustable in response to results of the computations by the equalization filter weight computer92.

The ghost-cancellation filter weight computer91is preferably of a type in which the Discrete Fourier Transform (DFT) of the ghosted GCR signal is divided by the DFT of the non-ghosted GCR signal to obtain as a quotient the DFT transform of the transmission medium giving rise to ghosting; and the inverse DFT of this quotient is then used to define the filter weighing coefficients of a compensating ghost-cancellation filter. As known by those skilled in the ghost-cancellation art, the ghost-cancellation filter93is preferably of a type with a sparse kernel where the positioning of the non-zero filter weights can be shifted responsive to results from the ghost-cancellation filter weight computer91. A ghost-cancellation filter with a dense kernel would typically require2048filter weights, which woud be difficult to construct in actual practice.

The equalization filter weight computer92could be of a type performing calculations using DFTs, the results of which are subject to inverse-DFT in order to define the filter weighing coefficients of a compensating equalization filter94. Preferably, however, the equalization filter weight computer92is of a type using a least-mean-square error method to perform an iterative adjustment of a 15-tap or so digital FIR filter used as the equalization filter94, adjustment being made so that there is a best match to the (sin x)/x function of the result of correlating of a portion of the de-ghosted PN sequence with the corresponding portion of the PN sequence known at the receiver as being a standard.

The luma/chroma separator95is preferably of a type using digital comb filtering for separating a digital luminance signal and a digital chroma signal from each other, which signals are respectively supplied to digital luminance processing circuitry96and to digital chrominance processing circuitry97. The digital luminance (Y) signal from the digital luminance processing circuitry96and the digital I and Q signals from the digital chrominance processing circuitry97are supplied to a digital color matrixing circuit98. Matrixing circuit98responds to the digit Y. I and Q signals to supply digital red (R), green (G) and blue (B) signals to digital-to-analog converters99,100and101, respectively. Analog red (R), green (G) and blue (B) signals are supplied from the digital-to-analog converters99,100and101to R, G and B kinescope driver amplifiers102,103and104, respectively. R, G and B kinescope driver amplifiers102,103and104supply red (R), green (G) and blue (B) drive signals to the kinescope84.

The filter94has thusfar been termed an “equalization filter” and considered to be a filter that would provide a flat frequency response through the band, which is the way this filter has been characterized by other workers in the ghost-cancellation art. In practice it is preferable to adjust the filter weights in the filter94not for flat frequency response through the band but with a frequency response known to provide some transient over- and under-shooting, or video peaking. This reduces the need for

providing transient overshooting or video peaking in the digital luma processing circuitry96.

FIG. 6shows a representative way of constructing the GCR signal capture processor90. Random access memories111,112,113and114are arranged to serve as line stores for the GCR reference signals supplied during fields00,01,10and11of each cycle of four successive fields of digitized composite video signal.

These GCR reference signals are supplied to the respective input ports of the RAMs111,112,113and114from the analog-to-digital converter79. The four successive fields in each cycle are counted modulo-4by a two-stage binary counter115that counts the ONEs generated by a decoder116that detects indications of the last scan line in a field furnished by the scan line count from the counter87. As a preparatory measure in the procedure of updating the filter weighting coefficients in the ghost-cancellation filter93and in the equalization filter94, the proper phasing of the modulo-4field count can usually be determined by correlating the most recently received GCR signal, as de-ghosted, with each of the four standard GCR signals stored in the receiver, looking for best match. Decoders121,122,123and124decode the100,101,110and111signals as generated by the 19thline decoder88supplying most significant bit and field count from the field counter115supplying the two less significant bits, thereby to furnish write enable signals sequentially to the RAMs111,112,113and114during the 19thscan lines of successive fields.

The RAMs111,112,113and114are addressed in parallel by an address counter125that counts the number of samples per scan line. The address counter125receives the oscillations from the phase-locked oscillator86at its count input connection, and is reset by an edge of the horizontal sync pulse. This addressing scan during the 19thscan line allocates each successive digital composite video signal sample to a successive addressable location in the one of the RAMs111,112,113and114receiving a write enable signal. During the 20thscan line the decoder89provides a read enable signal to all of the RAMs111,112,113and114. The addressing scan the counter125provides the RAMs111,112,113and114during the 20thscan line reads out the four most recently received and stored GCR signals parallely to a serial processor126that combines them to generate sequential samples of a separated Bessel pulse chirp signal and sequential samples of a separated PN sequence.

During the 20thscan line, the decoder89also provides a write enable signal to RAMs127and128that respectively serve as line stores for the separated chirp signal and separated PN sequence.

The decoder89at the same time conditions address multiplexers129and130to select addresses from the address counter125as write addressing for the RAMs127and128respectively. The counter125provides the RAM127the addressing scan needed to write thereinto the sequential samples of the separated chirp signal from the serial processor126. The counter125also provides the RAM128the addressing scan needed to write thereinto the sequential samples of the separated PN sequence from the serial processor126. At times other than the 20thscan line, the address multiplexer129selects to the RAM127read addressing supplied from the ghost-cancellation filter weight computer91during data fetching operations, in which operations the computer91also supplies the RAM127a read enable signal. At times other than the 20thscan line, the address multiplexer130selects to the RAM128read addressing supplied from the equalization filter weight computer92during data fetching operations, in which operations the computer92also supplies the RAM128a read enable signal.

FIGS. 7A,78,7C and7D are waveforms of the ghost cancellation reference signals in selected vertical blanking intervals of four successive fields of video, as embody the invention in one of its aspects, alternative to the aspect of the invention whichFIGS. 1A,1B,1C and1D concern. The GCR signals inFIGS. 7A and 7Dare the same as those ofFIGS. 1A and 1D. The GCR signals inFIGS. 7B and 7Cdiffer from those ofFIGS. 1B and 1Cin that the swings of the PN sequences are reversed in direction. InFIGS. 7B and 7Cthe swings of the PN sequences24′ and34′ are in the same direction as the swings of the PN sequences14′ and44′ inFIGS. 7A and 7D.

FIG. 8shows the separated Bessel pulse chirp waveform that results when the GCR signals from two successive fields that are in two successive frames are differentially combined, assuming that the GCR signals are of the sort shown inFIGS. 7A,7B,7C and7D. A separated Bessel pulse chirp waveform perFIG. 8results when the GCR signals ofFIGS. 7B and 7Care differentially combined. A separated Bessel pulse chirp waveform perFIG. 8also results when the GCR signals ofFIGS. 7D and 7Aare differentially combined. A separated Bessel pulse chirp waveform perFIG. 8also results when the sum of the GCR signals ofFIGS. 7A and 7Bis differentially combined with the sum of the GCR signals ofFIGS. 7C and 7D.

FIG. 9shows the waveform that results when the GCR signals from four (or any multiple of four) successive fields are additively combined or are averaged, assuming that the GCR signals are of the sort shown inFIGS. 7A,7B,7C and7D. The Bessel pulse chirp waveform and the chroma burst are suppressed in this signal. The DC level and “gray” pedestal are maintained in this signal as well as the PN sequence. The PN sequence can then be separated by high-pass digital filtering. The DC level and “gray” pedestal can be separated by low-pass digital filtering. The DC level and “gray” pedestal are useful in circuitry for controlling the gain and DC-offset of the analog composite signal applied to the analog-to-digital converter79. Circuits are known in the prior art in which the digital output signal of an analog-to-digital converter is selected as input signal to a first digital comparator during a portion of the digitized composite video signal known to be supposedly at 0 IRE level, there to be compared against digitized ideal 0 IRE level to develop a first digital error signal that is converted to analog error by a digital-to-analog converter and fed back to degenerate error in the 0 IRE level against which the input signal to the analog-to-digital converter is DC-restored. In certain of these circuits the digital output signal of the same analog-to-digital converter is selected as input signal to a second digital comparator during a portion of the digitized composite video signal known to be supposedly at a prescribed pedestal level, there to be compared against the prescribed pedestal level in digital form to develop a second digital error signal that is converted to analog error by a digital-to-analog converter and fed back as an automatic gain control (AGC) signal to a gain-controlled amplifier preceding the analog-to-digital converter and keeping the input signal to the analog-to-digital converter quite exactly within the bounds of the conversion range.

FIG. 10shows how theFIG. 6serial processor may be constructed for processing the ghost cancellation reference signals ofFIGS. 1A,1B,1C and1D to generate the FIG.2andFIG. 3signals. A serial adder131sums the RAM111output signal perFIG. 1Awith the RAM112output signal perFIG. 1B. Aserial adder132sums the RAM113output signal perFIG. 1Cwith the RAM114output signal perFIG. 1D. Aserial subtractor133subtracts the sum output of the adder132from the sum output of the adder131to generate a separated Bessel pulse chirp signal.

With a bit point shift of two places towards less significance, for carrying out wired division by four, this separated Bessel pulse chirp signal is theFIG. 2signal. A serial adder134sums the RAM111output signal perFIG. 1Awith the RAM114output signal perFIG. 1D. Aserial adder135sums the RAM112output signal perFIG. 1Bwith the RAM113output signal perFIG. 1C. Aserial subtractor136subtracts the sum output of the adder135from the sum output of the adder134to generated a separated PN sequence signal. With a bit point shift of two places towards less significance, for carrying out wired division by four, this separated PN sequence signal is theFIG. 3signal.

FIG. 11shows how theFIG. 6serial processor may be constructed for processing the ghost cancellation reference signals ofFIGS. 7A,7B,7C and7D to generate the FIG.8andFIG. 9signals. Serial adders131and132and serial subtractor133cooperate to generate a separated Bessel pulse chirp signal, as described in connection with FIG.10. With a bit point shift of two places towards less significance, for carrying out wired division by four, this separated Bessel pulse chirp signal is theFIG. 8signal. A serial adder137sums the sum outputs of the adders131and132to generate a separated PN sequence signal. With a bit point shift of two places towards less significance, for carrying out wired division by four, this separated PN sequence signal is theFIG. 9signal.

One skilled in the art of electronic circuits and systems design and acquainted with the foregoing disclosure will be enabled to design a number of variants of the signals and circuits specifically disclosed; and this should be borne in mind when considering the respective scopes of the claims which follow.