Redundant power system and power supply therefor

A redundant power system includes plural load-sharing power supplies connected to a common AC output bus. Identical circuitry is provided in each supply to control the redundant operation and is connected to each of the others via a common redundancy bus. An arbitration circuit in each supply selects a master power supply based on which supply detects the lack of a master supply first via the redundancy bus. A synchronization circuit in each slave supply synchronizes the polarity of the respective AC output with that of the master supply via the redundancy bus. A redundant bias circuit in each supply provides operating power to the redundancy circuitry in the respective supply from a common bulk voltage provided on the redundancy bus. A soft-start circuit in each supply allows all the supplies to start producing power at the AC output bus in unison. An overvoltage correction circuit in each supply detects an overvoltage on the AC output bus and shuts down only the supply which is causing the overvoltage.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates generally to the field of power supplies and specifically to a load-sharing redundant power system.

2. Relevant Art

Power supplies are used in many mission critical applications where it is desirable to continue to provide operating power to certain electrical and electronic systems even in the event of a utility power outage or power supply failure.

An uninterruptible power supply (UPS) system provides backup power-for the protected system from a battery source that is charged by the utility power. It is desirable to utilize an efficient connection from the battery to the load when the utility power is not available. Power management is often part of the UPS system operation.

UPSs are used in many different applications, some of which demand particular output power waveforms from the UPS. Therefore, it is advantageous for a UPS to have a configurable output waveform. It is also advantageous for the UPS to operate from a wide variety of input voltages and frequencies.

When a single UPS is used, there is still a possibility that the UPS could fail and leave the application without power. Thus, in mission critical applications, such as telecommunications, it is desirable to redundantly operate several UPSs or other power supplies connected together to provide power to the same load or system.

Such a redundant power scheme is often referred to as load sharing. If one of the power supplies operating redundantly fails or must be otherwise shutdown, the remaining redundant supply or supplies continue to supply power to the load.

Such redundant, load-sharing power supplies should have their outputs synchronized so that the desired waveform is supplied to the load and so that the power supplies do not damage one another or the load.

SUMMARY OF THE INVENTION

The present invention provides a redundant power system comprising a plurality of power supplies each connected to a common master-present bus and a common AC output bus. The plurality of power supplies each comprise an AC output connected to the common AC output bus.

Each power supply further comprises a redundant circuit for redundantly operating the power supply in cooperation with the respective redundant circuits of each of the other power supplies, the redundant circuit operating as determined by a local master/slave status as one of a master and a slave.

Each power supply further comprises arbitration logic in communication with the common master-present bus, wherein if the arbitration logic senses a master-not-present signal on the common master-present bus, then the arbitration logic sets the local master/slave status to master and transmits a master-present signal to the common master-present bus node, and wherein if the arbitration logic senses a master-present signal on the master bus and the local master/slave status is not set to master, then the arbitration logic sets the local master/slave status to slave.

According to a further aspect, the present invention provides a redundant power system comprising a plurality of power supplies each connected to a common polarity bus. The plurality of power supplies each comprises a local master/slave status settable to one of master and slave, and a synchronization circuit which reads a polarity value from the common polarity bus and if the local master/slave status is set to master, then the synchronization circuit transmits a master polarity signal to the common polarity bus. Each power supply further comprises a power section comprising an AC output and a polarity control connected to set the polarity of the AC output according to the polarity value of the common polarity bus.

According to a still further aspect, the present invention provides a redundant power system comprising a plurality of power supplies each connected to a common bias bus. The plurality of power supplies each comprises a redundant circuit for cooperation with a corresponding redundant circuit of each of the other power supplies and a power section for providing a local bias voltage. Each power supply further comprises a redundant bias circuit for contributing to a common bias voltage at the common bias bus, the redundant bias circuit providing operating power to the redundant circuit.

Each power supply further comprises a bias diode for performing a logical OR operation of the local bias voltage onto the common bias connection, wherein the redundant bias circuit is adapted to provide the operating power to the redundant circuit from the local bias voltage and to alternatively provide the operating power to the redundant circuit from the common bias connection when the local bias voltage is unavailable.

According to yet a further aspect, the present invention provides a power supply for operation in a redundant power system. The power supply comprises a power section comprising an AC output and a start-up cycle. The power supply further comprises a start-ready connection for connection to a corresponding start-ready connection of at least one other power supply. The power supply further comprises a soft-start circuit for transmitting a not-ready signal to the start-ready connection until the start-up cycle has completed, wherein the soft-start circuit disrupts the operation of the AC output until the soft-start circuit senses no not-ready signal at the start-ready connection.

According to yet a still further aspect, the present invention provides a power supply for operation in a redundant power system. The power supply comprises a power section comprising a pulse-width modulation signal and an AC output for connection to a corresponding AC output of at least, one other power supply.

The power supply further comprises an overvoltage detection circuit sensing a peak voltage level of the AC output, wherein the overvoltage detection circuit transmits an overvoltage signal if the peak voltage level exceeds a predetermined peak voltage level. The power supply further comprises a duty-cycle detection circuit sensing a duty cycle level of the pulse-width modulation signal, wherein the duty-cycle detection circuit transmits a maximum-duty-cycle signal when the duty-cycle level of the pulse-width modulation signal exceeds a predetermined maximum duty-cycle level.

The power supply further comprises an overvoltage correction circuit in communication with the overvoltage detection circuit and in communication with the duty-cycle detection circuit, wherein the overvoltage correction circuit disrupts the operation of the AC output when the overvoltage correction circuit detects both the overvoltage signal and the maximum-duty-cycle signal.

DETAILED DESCRIPTION OF THE INVENTION

Referring to FIGS. 1 and 2 , a power supply 10 according to the invention is provided with line voltage from any suitable source. Most commonly, line voltage will be 120 VAC at 60 Hz from a commercial utility. The line voltage can, however, range from 85 to 264 VAC at any frequency from DC to 400 Hz. The line voltage is input to an input stage 11 of the power supply. The input stage includes an EMI filter 12 . A resistor/relay active inrush control 14 at the EMI filter output prevents excessive current inrush to the power supply. The active inrush control 14 includes a resistor and relay connected in parallel. A high efficiency zero voltage switching (ZVS) power factor corrector (PFC) 16 boosts the line voltage to a desired DC bulk power voltage. The DC bulk power is a DC voltage with an AC ripple. For cable telecommunications amplifiers, the DC bulk power is nominally 400 VDC. The PFC presents a 0.99 or better power factor at full load to the power line. The PFC senses line voltage and shapes the input current to match it. The PFC supports the DC bulk voltage through limited brownouts. If the AC line voltage is lost or drops too low for too long, the PFC 16 sends a signal indicating an insufficient line voltage condition.

The DC bulk power is supported by electrolytic bulk:capacitors 18 . After the bulk capacitors 18 are charged through the inrush control 14 resistor, the inrush control relay is closed to short circuit the resistor. A flyback bias regulator 20 runs off of the DC bulk power, and provides initial bias required to start the power supply. A phase shifted full bridge (PSFB) converter 22 converts the DC bulk power to an inverter bulk power. The inverter bulk power can be maintained at any desired voltage. For cable telecommunications amplifiers, the inverter bulk power is regulated to a constant 59 VDC by the converter 22 . Under battery power, the inverter bulk power is normally maintained a range of 40-60 VDC. This converter provides full isolation to meet UL, CSA, and TUV requirements for an outdoor environment. The inverter bulk power feeds four sections: a battery charger/discharger 26 , an inverter section 27 including a phase shifted full bridge inverter 28 , a secondary bias regulator 30 , and a fan voltage generator & speed controller 32 . The bias regulators 20 , 30 and fan voltage generator provide power for various internal control and logic functions.

The battery charger/discharger 26 charges external batteries 34 when the line voltage is adequate and provides power from the batteries to the inverter 28 when the line voltage is inadequate. Current from the battery during discharge flows through a discharge switch S 1 , such as a FET and its corresponding body diode D 1 as shown in FIG. 2 . Charging and discharging are controlled by a microcontroller.

The inverter 28 generates an AC waveform having an arbitrary waveform determined by a microcontroller section 36 . Power to the inverter can be provided as DC in any of several forms including battery power, rectified AC, or pulse width modulated DC. A falling edge bleeder 37 section is connected across the AC output of the inverter 28 .

In the power supply 10 , all microcontroller controlled functions can be handled by a single microcontroller or by several microcontrollers dedicated to specific functions of the power supply. In one implementation, two microcontrollers are used. One microcontroller controls the inverter for output wave shaping and controls the battery charger/discharger. Another microcontroller includes user configurable logic for user interface and communication with a variety of transponders used by the cable telecommunications industry. Analog and digital hardware circuitry can also be used to configure system logic. An I 2 C common serial bus is used by the microcontrollers to communicate with each other and an internal EEPROM memory. An alphanumeric LCD display 38 provides diagnostic and other information about the power supply to a user.

Referring to FIGS. 3A , 3 B, and 4 , the inverter 28 includes an amplitude circuit 41 and a polarity circuit 46 . The amplitude circuit 41 generates a pulse width modulated output that determines the voltage amplitude of the inverter output. The amplitude circuit includes four FETs Q 1 -Q 4 operated by corresponding FET gate drivers 42 . The FET drivers 42 are operated by A side and B side power boosters 43 based on a signal from a pulse width modulator (PWM) 44 controlled by the microcontroller 36 . As schematically shown in FIG. 4 , the FETs Q 1 -Q 4 are operated as switches. The drains of the FETs Q 1 and Q 3 are connected to the inverter bulk power and their respective sources are connected across the primary of a transformer T 1 . The sources of the FETs Q 2 and Q 4 are connected to ground and their respective drains are connected across the primary of the transformer T 1 . In practice, as shown in FIG. 3A , two of the FETs Q 1 and Q 2 have their sources connected to the transformer T 1 through an inductor L 1 and respective current sensors T 2 , T 3 . The current sensors include respective transformers each having a diode connected in parallel therewith. The current sensors provide inverter current signals to the PWM 44 for analog control. The current signals are used to regulate the inverter and prevent saturation of the transformer T 1 . If an overcurrent condition is sensed, an alarm is activated. The current signal is also used for operation in a current control mode.

Diodes D 2 -D 5 are connected across each of the FETs Q 1 -Q 4 . A diode D 6 is connected between the inverter bulk voltage and a node between the transformer T 1 and the inductor L 1 . A diode D 7 is connected between ground and the node between the transformer T 1 and the inductor L 1 . The transformer T 1 secondary has a center tap connected to ground. The microcontroller 36 generates a reference voltage for the PWM 44 . The PWM controls operation of the FETs Q 1 -Q 4 to generate a pulse width modulated output from the transformer T 1 output. The energy in the transformer output determines the energy in the AC output of the inverter. Thus, the transformer T 1 output determines the absolute value of the power supply output voltage. The PWM 44 senses AC output voltage and compares it to the reference voltage from the microcontroller. The PWM controls the transformer T 1 output to obtain the desired voltage amplitude.

A current sensor including a transformer T 6 is connected on the secondary side of the transformer T 1 . The current sensor sends a current signal to the microcontroller for display on the LCD 38 and for internally monitoring the output current on the power supply.

The inverter 28 includes a polarity circuit 46 having polarity switches Q 5 -Q 8 connected between the transformer T 1 secondary and the power supply output. The polarity switches Q 5 -Q 8 are NPN insulated gate bipolar transistors. Each polarity switch Q 5 -Q 8 is connected in series with a diode D 8 -D 11 . The polarity switch and diode combinations are connected in opposing pairs in respective legs of the transformer T 1 output. The AC output of the polarity circuit is connected to a filter including an. inductor L 2 and a capacitor C 1 . The LC filter L 2 , C 1 filters the pulse width modulated output of the amplitude and polarity circuits to provide the desired output waveform.

Two polarity switch control transformers T 4 and T 5 are each provided with two sets of secondary windings. One leg of each transformer T 4 , T 5 secondary is connected through a diode and an RC filter to the gate of a corresponding one of the polarity switches Q 5 -Q 8 . The other leg of the corresponding transformer T 4 , T 5 is connected to the emitter of the respective polarity switch Q 5 -Q 8 . A bias voltage is applied to the primary side of the transformer T 4 , which is connected in series with two FETs Q 9 and Q 10 . The bias voltage is also applied to the primary side of the transformer T 5 , which is connected in series with two FETs Q 11 and Q 12 . Another FET Q 13 is connected between the gate of FET Q 12 and ground. The gates of FETs Q 9 and Q 11 are connected to a high frequency square wave generating oscillator. The gates of FETS Q 10 and Q 12 are connected to an output of the microcontroller that can provide an output shutdown signal. The output shutdown signal is normally high during operation. The gates of FETs Q 10 and Q 13 are connected to a polarity switch output of the microcontroller 36 .

When the microcontroller 36 sets the AC output to zero, the microcontroller also changes the digital state of its polarity switch output. If it was high, it becomes low. If it was low, it becomes high. This results in either FET Q 10 or FET Q 12 being on, but not both. The FETs Q 9 and Q 11 are continuously driven by the high frequency oscillator. Thus, either transformer T 4 or transformer T 5 is active, but not both. The square waves generated by the transformers T 4 and T 5 are averaged to a DC level by the corresponding diode and RC filter on each output winding. These DC voltages will turn on either the combination of polarity switches Q 5 and Q 8 or the combination of polarity switches Q 6 and Q 7 . The corresponding diodes D 8 -D 11 rectify the output of the phase shifted full bridge as either a positive or a negative voltage.

When the output is to be zero volts, FETs Q 12 and Q 10 are disabled by pulling the output shutdown signal to low, which prevents operation of the transformers T 4 and T 5 . This turns off all of the polarity switches Q 5 -Q 8 .

A transorb D 12 connected across the AC output dissipates residual energy left in the inductor L 2 when transistors Q 5 -Q 8 are all off. The dissipation period can be extended as necessary according to the inductor L 2 and filter capacitor C 1 values. A disconnect switch S 2 is located downstream of the falling edge bleeder 37 . A surge circuit 48 absorbs fast electrical transients that could damage downstream circuitry.

Referring to FIGS. 3B and 5 , the falling edge bleeder 37 includes a bleed resistor R 1 connected across the AC output of the full bridge converter 28 . A pair of FETs Q 13 , Q 14 are connected in series with the bleed resistor R 1 . A sensing voltage signal 50 representing the desired AC output voltage of the full bridge converter 28 is fed through a buffer U 1 A to the inverting input of an op amp U 1 B. The output of the buffer U 1 A is attenuated by a voltage divider R 2 , R 3 and fed to the non-inverting input of the op amp U 1 B. A capacitor C 2 connected in parallel with the resistor R 3 slows the response of the voltage divider R 2 , R 3 . The output of the op amp U 1 B is fed to the gate of the FET Q 14 and the gate of another FET Q 15 . The FET Q 15 is connected to the diode of an optocoupler U 2 . The optocoupler U 2 is connected to operate a transistor Q 16 . The transistor Q 16 emitter is connected to the gate of FET Q 13 through a diode D 13 . A 14 volt bipolar transorb D 14 is connected between the AC output and the gate of the FET Q 13 .

If the AC output voltage is controlled to move toward zero at a rate exceeding a maximum rate determined by the voltage divider R 2 , R 3 and the capacitor C 2 , the output of the operational amplifier U 1 B output will go high. This turns on the FET Q 14 . It also turns on the FET Q 15 , which pulls current through the light emitting diode of the optocoupler U 2 . If the AC output voltage is positive, current flows through the parasitic body diode of Q 13 . This places the bleed resistor R 1 in parallel with the AC output. The diode D 13 prevents current from flowing through the transistor of optocoupler U 2 and prevents harmful voltages from being impressed upon the transistor Q 16 and the optocoupler U 2 . If the AC output voltage is negative, then current flows through the FET Q 14 , a current limiting resistor R 4 , and the transistor of the optocoupler U 2 . This turns on the transistor Q 16 , which feeds a current through the diode D 13 and the parallel combination of resistor R 5 and the transorb D 14 . This places a voltage on the gate of the FET Q 13 from its gate to its source, which is limited to the breakdown voltage of the bipolar transorb D 14 . This turns on the FET Q 13 and places the bleed resistor R 1 in parallel with the AC output. Directing current through the bleed resistor R 1 accelerates the movement of the output voltage toward zero to discharge residual energy. This permits the actual output voltage to more closely follow the desired output voltage set by the microcontroller. In many cases, the desired output voltage will ramp toward zero more quickly than the actual voltage during every half cycle. Thus, the falling edge bleeder might be activated every half cycle.

When the AC output voltage begins to rise again, the voltage at the inverting input of the op amp U 1 B will exceed the time delayed voltage at the non-inverting input. The output of the op amp U 1 B will go low, which turns off the FETs Q 13 and Q 14 . If the AC output voltage is positive, the full AC output voltage appears across the FET Q 14 . If the AC output voltage is negative, the full output voltage appears across FET Q 13 . No current will flow between the AC output and return.

Thus, the amplitude circuit controls output voltage amplitude and the polarity circuit controls output voltage polarity. The falling edge bleeder ensures that a quickly declining output voltage magnitude follows the desired waveform. Frequency is determined by the rate at which the polarity circuit changes the output polarity. Normally, the amplitude circuit ramps the voltage down before polarity is changed. Smoothly synchronized microcontroller operation of the gate drivers and polarity switches generates a precise AC waveform of arbitrary shape at the output of the power supply. For example, to obtain an output waveform having a trapezoidal shape, full voltage is maintained by a relatively high duty-cycle on the amplitude circuit output. The LC filter L 2 , C 1 provides an instantaneous output voltage magnitude corresponding with the pulse width modulated voltage of the amplitude circuit. For 60 Hz operation, the voltage is set to zero and the polarity changed every {fraction (1/120)}th of a second. Before the voltage is set to zero, the duty-cycle of the inverter amplitude circuit is reduced in steps to obtain a downward sloping ramp on the output of the power supply. The falling edge bleeder dissipates energy to ensure that the output voltage decays at the correct rate. After reaching zero, the polarity circuit changes the output polarity. Then, the output voltage is ramped down to full negative voltage by increasing the inverter duty-cycle. Operation continues in this manner to obtain the desired output waveform.

Referring to FIGS. 3A and 6 , each FET driver 42 includes a transformer T 7 having its primary connected between the pulse width modulator 44 output (via the corresponding power booster 43 ) and ground. The input from the PWM 44 is a 50% duty-cycle square wave. The secondary of the transformer T 7 is connected through a resistor R 6 to the respective bases of two transistors Q 17 , Q 18 . One of the transistors Q 17 is an NPN type and the other transistor Q 18 is a PNP type. The emitter of the NPN transistor Q 17 is connected to the gate of the corresponding FET Q 1 , Q 2 , Q 3 , or Q 4 through a diode D 15 and a resistor R 7 . The collector of this transistor Q 17 is connected to the FET source through a capacitor C 3 . The emitter of the PNP transistor Q 18 is connected to the gate of the corresponding FET through the resistor R 7 . The collector of this transistor Q 18 is connected to the FET source through a capacitor C 4 and diode D 16 connected in parallel.

The FET driver 42 has two modes of operation: steady state and initial state. During steady state operation, when the voltage across the secondary of transformer T 7 goes from negative to positive, the NPN transistor Q 17 is forward biased from base to emitter. This pulls current from the capacitor C 3 through the transistor Q 17 , the diode D 15 , and the resistor R 7 to the gate of the FET. Simultaneously, the PNP transistor Q 18 is reverse biased from base to emitter and-turns off. The FET gate voltage will then rise to the level to which the capacitor C 3 is charged. After reaching the voltage of the capacitor C 3 , the energy removed from the capacitor C 3 to charge the FET is restored to the capacitor C 3 through the base-collector PN junction in the transistor Q 17 . Eventually, the transformer T 7 reverses polarity again, going from positive to negative. Then, the NPN transistor Q 17 is reverse biased from base to emitter and turns off. The PNP transistor Q 18 is forward biased from emitter to base. The capacitor C 4 is charged to a negative voltage with respect to the FET source. Current flows from the gate of the FET, through the resistor R 7 and transistor Q 18 , and into the capacitor C 4 . The capacitor C 4 becomes slightly less negative in this process. After the FET gate is at its maximum negative voltage, current flows though the collector-base PN junction in the transistor Q 18 , through the resistor R 6 and transformer T 7 secondary, and back to the capacitor C 4 , restoring the capacitor C 4 to its previous level of negative voltage. This process then repeats as the polarity of the transformer T 7 continues to alternate.

During the initial state mode when the FET driver 42 is started, the capacitors C 3 and C 4 are completely discharged. When the voltage on the transformer T 7 secondary first goes positive, all the charge flowing though the resistor R 7 flows from the transformer T 7 through the diode D 15 and the base-emitter junction of the transistor Q 17 . This will result in a low rate of rise on the waveform and a slow turn on of the FET as compared to the steady state operation. As the corresponding FET gate slowly charges, the capacitor C 3 also charges. All the energy being pulled out of the FET gate flows through the emitter-base junction of Q 18 , the resistor R 6 , and the transformer T 7 secondary. This results in a slow turn off of the FET. As the FET turns off, the capacitor C 4 gradually charges to a negative voltage. Once the capacitors C 3 and C 4 are both charged to their full normal potential, the circuit works in the steady state mode described above.

Referring to FIGS. 2 and 7 , the battery charger/discharger 26 includes a charging circuit and a discharging circuit. The charging circuit is a forward converter battery charger including two charging FETs Q 19 , Q 20 operated by a microcontroller controlled gate driver 52 . The charging FETs Q 19 , Q 20 are connected between the inverter bulk power and ground via the primary of a charging transformer T 8 . Voltage clamping diodes D 17 , D 18 are connected to the drains of the charging FETs Q 19 , Q 20 . A current sensor 54 is connected to the secondary of the charging transformer. The transformer output is rectified by a diode D 19 and smoothed by an inductor L 3 and capacitor C 5 . A freewheeling diode D 20 and ORing diode D 21 are also provided in the charging circuit. The ORing diode D 21 prevents current flow into the battery charging circuit when an external battery charger is connected to the battery. A battery disconnect switch S 3 is provided to isolate the battery 34 from the charging and discharging circuits. A voltage sensing line is connected from the battery to the microcontroller. This battery charger is capable of charging 36 V or 48 V nominal battery strings at a rate of 10 A. The battery output is normally maintained at about 41 to 54 volts. The microcontroller is connected to receive line voltage information from the power factor corrector. When the power supply loses AC line voltage on the main line, the charger is disabled, and the battery discharge circuit is activated.

The discharging circuit includes discharge FET Q 21 connecting the battery to the inverter bulk power through the disconnect switch S 3 . In FIG. 2 , the discharge FET Q 21 is schematically shown as the parallel combination of the switch S 1 and diode D 1 . If the inverter bulk drops below the battery voltage, the body diode of discharge FET Q 21 will forward bias, thus current will flow from the battery to sustain the inverter bulk voltage.

When the DC bulk voltage is lost or falls too low, the PFC sends a low voltage signal to the microcontroller, which then starts an oscillator 56 . The oscillator 56 drives control FET Q 22 . When the control FET Q 22 is on, capacitor C 6 is charged to the bias voltage (nominally 15 volts) though diode D 22 . When the control FET Q 22 is off, its drain to source capacitance is charged to the battery voltage via resistor R 8 . Thus, the voltage at node A is held higher than the battery voltage. Current then flows from the capacitor C 6 , through resistor R 9 , and diode D 23 , into the gate capacitance of the discharge FET Q 21 . When the control FET Q 22 is turned on again, diode D 23 prevents the gate of discharge FET Q 21 from losing the stored charge immediately. Thus, the discharge FET Q 21 is turned on if the failure of the AC line voltage to maintain the inverter bulk voltage is more than a transient condition. The repeated charging and discharging of the capacitor C 1 by the oscillator 56 maintains sufficient voltage at the gate of the discharge FET Q 21 to hold the discharge FET on.

When the AC line voltage is restored, but before the inverter bulk voltage is supported by the full bridge inverter 22 again, the oscillator 56 is stopped by the microcontroller. The capacitor C 6 no longer delivers charge to discharge FET Q 21 . The voltage on the gate of discharge FET Q 21 decays slowly through resistor R 10 . Once the gate voltage drops below the threshold voltage of discharge FET Q 21 , the FET turns off and current flow through the discharge FET shifts from the switch S 1 to its intrinsic body diode D 1 . This state continues until the power supply is fully restored to normal operation so the inverter bulk voltage exceeds the battery voltage. Thus, the body diode is reverse biased and there is no current flow through the discharge FET Q 21 . The battery charger is then restarted, and normal operation commences.

The battery discharger is provided with a protection circuit 58 . If the output of the battery charger, node B, for example, is shorted out, the voltage at the battery output will drop below a protection circuit bias voltage. When this occurs, current flows through transistor Q 23 from its emitter to its base. The base current flows through diode D 24 and resistor R 11 to the battery output. This will feed current into transistor Q 25 , which pulls the anode of diode D 23 to ground. This holds the discharge FET Q 21 off in a short circuit condition and prevents current from the inverter bulk voltage from flowing into the short circuit. Diode D 24 prevents damage to transistor Q 23 under normal operating conditions.

Referring to FIG. 8 , the fan controller 32 operates a cooling fan 60 for the power supply. The fan controller 32 includes a buck regulator 62 used to set a fan voltage in a range of 14 to 27 VDC. Voltage at the output O of the regulator 62 is fed back to the control input C through a resistor R 12 . A negative temperature coefficient, temperature dependent resistor, such as a thermistor TH 1 , is connected between the control input C and ground. The thermistor TH 1 monitors power supply temperature. As the power supply temperature increases, the resistance of the thermistor TH 1 decreases. This pulls down the voltage at the regulator control input C, which causes the regulator to increase the voltage to the fan. Thus, as the power supply temperature increases, the fan speed increases. Similarly, as the power supply temperature decreases, the fan speed decreases.

High voltage and low voltage clamps 63 , 65 clamp the regulator output voltage within upper and lower limits, respectively. If the output voltage exceeds a threshold determined by resistors R 13 , R 14 and R 15 , then the high voltage clamp 63 feeds current into the thermistor TH 1 , which maintains the regulator control input voltage at a level clamping the regulator output voltage to the high rail (27 VDC). If the output voltage drops below a threshold determined by the resistors R 13 , R 14 , and R 15 , then the low voltage clamp 65 pulls current away from the thermistor TH 1 , which holds the regulator control voltage at a level maintaining a minimum output voltage.

A current sensor 64 is connected to sense output current of the power supply. A low pass filter U 3 C buffers an analog representation of the output current from the current sensor. The LP filter U 3 C feeds the current signal into transistor Q 28 . If the power supply current increases, representing increased load, the transistor Q 28 reduces the voltage at the regulator control. This increases the regulator output voltage and increases the fan speed. If the power supply current decreases, representing decreased load, the transistor Q 28 raises the voltage at the regulator control. This decreases the regulator output voltage and decreases the fan speed.

A zener diode D 28 is an overvoltage protection diode. If the regulator output voltage exceeds the breakdown voltage of diode D 28 plus the voltage across the thermistor TH 1 , then the diode D 28 will conduct, which clamps the output voltage.

Referring to FIG. 9 a redundant power system 66 is shown comprising redundant power supplies 68 . Each redundant supply 68 has a redundancy section 70 and a power section 72 . As used herein, except where otherwise specified, the terms local and unit will generally refer to an element of a redundant supply 68 which operates independently of the other redundant supplies 68 in the power system 66 , and the term common will generally refer to an element of the redundant power system 66 which is shared or interconnected between two or more redundant supplies 68 .

The power sections 72 of each of the redundant power supplies 68 are connected together to share a common load 74 by way of a common AC output bus 76 . Likewise, the redundancy sections 70 of each of the redundant supplies 68 are connected together for intercommunication by a redundancy bus 78 . The redundancy bus 78 comprises a plurality of individual buses connected respectively between various matching circuits of the redundancy sections 70 of each of the redundant supplies 68 . These elements of the redundancy sections 70 will be enumerated and described in detail below.

While three redundant supplies 68 are shown in FIG. 9 by way of example. The present invention anticipates that more or less power supplies can be used as required for a given application. Additional redundant supplies 68 are added by-connecting their redundancy sections 70 and power sections 72 to the redundancy bus 78 and common AC output bus 76 , respectively. Further, as will become apparent, a single redundant supply 68 according to the present invention is capable operating as a stand-alone power supply without the connection of additional redundant supplies 68 .

In the present embodiment, the power section 72 of each redundant supply 68 includes a PWM and a polarity circuit and produces an AC output, such as the power supply 10 disclosed herein. However, it will become apparent from the disclosure that other power sections of different design may be substituted for the power section 72 in the redundant power system 66 without departing from the scope of the present invention.

Referring to FIGS. 10A , 10 B and 10 c , the basic elements of operation of the redundancy section 70 (see FIG. 9 ) are shown. Each redundancy section 70 comprises five redundant circuits: a redundant bias circuit 80 , an arbitration circuit 82 , a synchronization circuit 84 , a soft-start circuit 86 , and an overvoltage correction circuit 88 . Since each redundant supply 68 in the redundant power system 66 (see FIG. 9 ) contains each of the redundant circuits 80 - 88 in its respective redundancy section 70 , the redundant circuits 80 - 88 in one redundant supply 68 interact with the redundant circuits 80 - 88 in each of the other redundant supplies 68 that are running redundantly.

Referring to FIGS. 9 and 10A , a common bulk voltage is provided to redundantly supply operating power to the redundancy section 70 of each 68 by way of its redundant bias circuit 80 and a common bias bus 90 of the redundancy bus 78 . The common bias bus 90 is referenced to a common bias return bus or common ground 91 and supplies the common bulk voltage to each of the redundancy sections 70 .

Each redundant bias circuit 80 is connected to receive a local bias voltage 92 from its respective power section 72 . For example, the power supply 10 described herein could provide a bias voltage to the respective redundant bias circuit 80 from its secondary bias regulator 30 (see FIG. 1 ).

Further, when the respective power section 72 is operating correctly, each redundant bias circuit can provide the common bulk voltage to the common bias bus 90 . In the event that one or more of the redundant supplies 68 fails, the remaining functioning redundant supplies 68 , if any, will continue to maintain the common bulk voltage on the common bias bus 90 to provide operating power to the redundancy sections 70 of all of the connected redundant supplies 68 . The redundant bias circuit 80 of each redundant supply 68 generates it's own regulated bias voltage from the common bulk voltage of the common bias bus 90 , thus providing a regulated bias voltage to other redundancy circuits of the redundancy section 70 , including the arbitration circuit 82 and the synchronization circuit 84 , even when the respective power section 72 is not functioning. Further, it is contemplated within the scope of the present invention that the redundant bias circuit 80 can be used to power additional circuits of the redundant power supply 68 , as needed.

Specifically, in the redundant power system 66 , a bias diode D 25 in each of the bias circuits 80 performs a logical OR of their respective local bias voltage 92 onto the common bias bus 90 . Thus, if at least one of the bias circuits 80 is delivering its local bias voltage 92 to the common bias bus 90 , then the common bias bus 90 can provide the regulated bias voltage to the other redundancy circuits 82 , 84 of every redundant supply in the power system 66 .

Each of the local bias voltages 92 is referenced to a respective local bias ground or return 93 . A bias return diode D 26 separates the local bias return (ground) 93 of each redundant bias circuit 80 and the common bias return 91 . The redundant bias circuit 80 further comprises a redundant bias regulator 80 a and a bias clamp circuit 80 b to provide a regulated bias voltage 80 c to each of the other redundancy circuits 82 , 84 . The redundant bias regulator 80 a is referenced to the common bias return bus (ground) 91 and is connected to the common bias bus 90 through the clamp circuit 80 b . The clamp circuit 80 b protects the bias regulator 80 a from excessive voltages.

FIG. 11 shows a circuit which includes an example of an embodiment of the redundant bias circuit 80 (see FIG. 10A ) of the redundant power supply 68 according to the present invention. Specifically, a diode D 27 ORs in a local bias voltage V 1 that is referenced to the local bias return or local power supply ground G 1 . An isolation diode D 28 separates the local power supply ground G 1 from the common bias return or common ground G 2 . A zener diode D 29 clamps the ORed voltage. In the event that a connection opens up, the zener diode D 29 prevents voltages from rising too high for the input of U 4 . Capacitor C 7 provides local capacitance for U 4 . The voltage across D 29 is to the common bias and common bias return buses.

Protection diode D 30 prevents reverse voltages from damaging U 4 . Regulator U 4 , resistor R 17 and resistor R 18 define the regulated bias voltage that powers the redundancy circuitry. Thus, each power supply generates its own regulated bias voltage from the common bulk voltage on the common bias bus.

Referring to FIGS. 9 and 10B , a redundancy controller 94 , such as a microcontroller, is provided in the redundancy section 70 of each redundant supply 68 . The redundancy controller 94 implements logic to control the operation of various aspects of the redundancy section 70 and the power section 72 according the redundant power system 66 disclosed herein. Further, operating power can be provided to the redundancy controller 94 by the redundant bias circuit 80 .

Further regarding the redundancy controller 94 , it should be appreciated that, as mentioned above with regard to the microcontroller 36 of the power supply 10 , all controller functions of a power supply can be implemented using a single microcontroller or alternatively separate microcontrollers may be provided. Further, as previously mentioned with regard to the power supply 10 , other types of controller circuitry may be substituted for the microcontroller(s). Thus, the redundancy controller 94 may be incorporated with other controllers, or provided for separately.

The arbitration circuit 82 in each redundant supply 68 allows two or more redundant supplies 68 to collaboratively decide which supply 68 will operate as the master supply and provide a master synchronization signal to all the slave supplies. Arbitration logic for implementing the functions of the arbitration circuit 82 is embodied in both the arbitration circuit 82 itself and the redundancy controller 94 .

Each arbitration circuit 82 comprises a master-present transmit circuit 96 and a master present sense circuit 98 . A common master-present bus 100 of the redundancy bus 78 connects each arbitration circuit 82 to the arbitration circuits 82 of the other redundant supplies 68 for communicating a common master-present signal.

Specifically, regarding the arbitration circuit 82 of each redundant supply 68 , the common master-present bus 100 is connected to an input of the master-present sense circuit 98 . An output of the master-present sense circuit 98 is connected through a resistor to a master-present input/output 94 a of the redundancy controller 94 . When the master-present input/output (I/O) 94 a is functioning as a high impedance input, the master-present sense circuit 98 is capable of relaying a signal from the common master-present bus 100 to the redundancy controller 94 for the purpose of sensing whether any redundant supply 68 in the power system 66 has declared itself as master.

Further, the master-present I/O 94 a of the redundancy controller 94 is also connected to an input 96 a of the master-present transmit circuit 96 . An output 96 b of the master-present transmit circuit 96 is connected to the common master-present bus 100 . When the master-present input/output (I/O) 94 a is functioning as a low impedance output, the master-present transmit circuit 96 is capable of relaying a signal from the redundancy controller 94 to the common master-present bus 100 for controlling the signal of the common master-present bus 100 .

Further, a local master/slave status output 94 b of the redundancy controller 94 is connected to a control input 96 c of the master-present transmit circuit 96 to communicate a local master/slave status of the respective redundant supply 68 from the redundancy controller 94 for control of the operation of the master-present transmit circuit 96 .

The operation of the arbitration circuits 82 connected to the common master-present bus 100 in the redundant power system 66 of the present invention is as follows.

At initialization of each redundant supply, the master-present I/O 94 a of each controller 94 is set as a high impedance output and the local master/slave status output 94 b is set low as a slave. The redundancy section 70 provides a weak pull-up to bias voltage to the respective master-present transmit circuit 96 , causing the master-present transmit circuit 96 to set the common master-present bus 100 to a logic high voltage. This logic high voltage on the common master-present bus 100 acts as a master-not-present signal, indicating that none of the redundant power supplies 68 have declared themselves as master.

When each master-present sense circuit 98 senses the master-not-present (high) signal on the master-present bus 100 it relays this master-not-present (high) signal to the local master-present I/O 94 a of its respective redundancy controller 94 . The first redundancy controller 94 in the power system 66 to detect the master-not-present (high) signal at its respective local master-present I/O 94 a , declares the respective redundant supply 68 to be a master supply by changing its respective local master/slave status output 94 b to transmit a logic high signal. Then the redundancy controller 94 , having declared its respective supply 68 a master supply, changes its respective master-present I/O 94 a from being a high impedance input to a low impedance output, and sets the I/O 94 a to a logic high, maintaining the master-not-present signal. The respective master-present transmit circuit 96 detects on its control input 96 c that the local master/slave status output 94 b is set to master (high) and on its input 96 a that the master-present I/O 94 a is being driven high as a master-not-present signal, and responds by causing its output 96 b to pull the common master-present bus 100 low, as a master-present signal.

Once a master supply is declared and the common master-present bus is being pulled low, each of the other redundant supplies 68 , which have not yet declared themselves master become slave supplies. Specifically, the respective master-present sense circuit 98 transmits the low, master-present signal of the common master-present bus 100 to the master-present input/output 94 a of the respective redundancy controller 94 . All of redundant supplies 68 with controllers 94 that have master-present input/output 94 a set as an high impedance input detect the master-present (low) signal and do not try to become masters supplies.

The redundancy controller 94 of the master supply continues to hold the common master-present bus 100 low as a master-present signal, since the controller 94 is actively overriding the signal from the master-present sense circuit 98 . If the power section 72 of this master redundant supply 68 fails, the respective redundancy controller 94 and arbitration circuit 82 will release the common master-present bus 100 , allowing the common master-present bus 100 to be pulled high as a master-not-present signal. The remaining slave supplies in the power system 66 will go through an arbitration process to determine which redundant supply 68 will become the new master supply.

FIG. 11 shows a circuit which includes an example of an embodiment of the arbitration circuit 82 (see FIG. 10B ) of the redundant power supply 68 according to the present invention. Specifically, the master/slave status output of the redundancy microcontroller is connected to FET Q 25 and the local master-present I/O of the redundancy controller is connected to FET Q 26 .

At initialization, controlled switch U 5 starts in the off state. The regulated bias of each power supply is pulling the common master-present bus high via its respective resistor R 19 . This master-not-present signal voltage is fed through voltage divider resistors R 20 and R 21 to turn on FET Q 27 . When FET Q 27 is on, controlled switch U 6 turns off, and the local master-present signal goes high, turning FET Q 26 on.

The first redundancy controller to detect that the common master-present bus signal is high, i.e. a master-not-present signal, declares itself to be a master, and sets its respective local master/slave status to master. The master redundancy controller also changes its local master-present signal from being a high impedance input to a low impedance output, and sets it high, latching FET Q 26 on. This turns FET Q 25 on, and because FET Q 26 is on, controlled switch U 5 is turned on. Controlled switch U 5 pulls the common master-present bus low, turning FET Q 27 off, and allowing controlled switch U 6 to turn on, pulling the local master-present low as a master-present signal on all of the arbitration circuits connected to the common master-present bus.

All of the redundancy controllers still set as slaves see that their respective local master-present is low and do not try to become masters. The master supply, since it is actively overriding the signal from controlled switch U 6 , continues to hold the common master-present bus low. If the master supply fails or is otherwise shut down, it will release the common master-present bus, and the remaining slaves, if any, will go through an arbitration to determine which power supply will be the new master.

Referring again to FIGS. 9 and 10B , the synchronization circuit 84 in each redundant supply 68 provides two functions. First, the synchronization circuit 84 in a master supply will define when the slave supplies will begin a new AC output cycle by providing a common master-polarity signal to a common polarity bus 102 of the redundancy bus 78 . Second, the synchronization circuit 84 provides supplemental arbitration in the event that multiple master supplies exist simultaneously due to the arbitration circuits 82 of two or more redundant supplies 68 in the power system 66 each declaring their respective-redundant supply 68 as a master supply. Synchronization logic for implementing the functions of the synchronization circuit 84 is embodied in both the synchronization circuit 84 itself and in the redundancy controller 94 .

Each synchronization circuit 84 comprises a synchronization transmit circuit 104 and a synchronization read circuit 106 . The common polarity bus 102 connects the synchronization circuit 84 to the synchronization circuits 84 of the other redundant supplies 68 for communicating the common master-polarity signal.

Specifically, regarding the synchronization circuit 84 of each redundant supply 68 , a local master-polarity output 94 c of the redundancy controller 94 is connected to an input 104 a of the synchronization transmit circuit 104 . An output 104 b of the synchronization transmit circuit 104 is connected to the polarity bus 102 through a resistance. Further, the master/slave status output 94 b is connected to a control input 104 c of the synchronization transmit circuit 104 .

Further, the common polarity bus 102 is connected to an input 106 a of the synchronization read circuit 106 . An out put 106 b of the synchronization read circuit 106 is connected to a polarity-average input 94 d of the redundancy controller 94 .

The operation of the synchronization circuits 82 connected to the common polarity bus 102 in the redundant power system 66 of the present invention is as follows.

If a redundant supply 68 has been declared to be a master supply, then the synchronization transmit circuit 104 circuitry is activated by the local master/slave status (high) signal of the master supply that is transmitted to the control input 104 c . This logic high at the control input 104 c causes the synchronization transmit circuit 104 to transmit a local master-polarity signal from the local master-polarity output 94 c of the redundancy controller 94 to the common polarity bus 102 . A common master-polarity signal thus provided on the common polarity bus is an active logic high or low. The redundancy controller 94 generates the local master-polarity signal at the local master-polarity output 94 c based on a connection of the controller 94 to a polarity control or switch of the power section 72 in the master supply.

In each slave supply, the synchronization transmit circuit 104 circuitry is disabled by the local master/slave status (low) signal of the master supply that is transmitted to the control input 104 c . This logic low at the control input 104 c causes the output 104 a of the synchronization transmit circuit 104 to become high impedance.

The common master-polarity signal of a master supplies transmitted to the common polarity bus 102 defines the polarity of the AC waveform generated by the power sections 72 of all the redundant supplies 68 in the power system 66 . That is, when the common master-polarity signal is a logic high, the voltage at the AC output is positive, and when the common master-polarity signal is a logic low, the voltage at the AC output is negative.

The respective redundancy controller 94 of each of the redundant supplies 68 in the power system 66 , whether master or slave, reads the value of the common polarity bus from its respective synchronization read circuit 106 , and sets the voltage of its AC output accordingly by controlling the polarity switch of the respective power section 72 . Thus, voltages having matching polarities are transmitted by the AC outputs of each of the redundant supplies 68 to common AC output bus 76 .

As an example, if the power supply 10 described herein is used in the power section 72 , the microcontroller polarity switch of the polarity circuit 46 of FIG. 3B would be connected to the redundancy controller 94 . In this example, if the power supply 10 was in a master supply, the polarity switch would be monitored by the redundancy controller 94 to generate the local master-polarity signal at the local master-polarity output 94 c . Further, if the power supply 10 was in a slave supply, the polarity switch would be controlled by the redundancy controller 94 according to the common master-polarity signal at the polarity-average input 94 d.

Referring again to FIGS. 9 and 10C , regarding the above-mentioned supplemental arbitration function of the synchronization circuits 84 , if there are two or more master supplies in the power system 66 , the common master-polarity signal on the common polarity bus 102 can have three possible values: a logic high, a logic low, and an ambiguous value representing neither high nor low. Specifically, at any given moment, the value of the common master-polarity signal read by each synchronization read circuit 106 in the power system 66 is an average of the values of all of synchronization transmit outputs 104 b.

If only one master supply is present in the power system 66 , the high impedance synchronization transmit circuits 104 of each the slave supplies do not substantially affect the average and the one master supply controls the common master-polarity signal.

However, if two or more master supplies are present in the power system 66 , each of the master supplies' respective synchronization transmit circuit 104 has an averaging effect on the common master-polarity signal. When all of the master supplies agree on the polarity such that the logic level of their respective local master-polarity signals match at a given moment, then the common master-polarity signal on the common polarity bus 102 will be a logic high or logic low as the consensus determines. When at least one of the master supplies disagrees with the others such that the logic levels of all of the master supplies' respective local. master-polarity signals do not match at a given moment, then the common master-polarity signal on the common polarity bus 102 will be an ambiguous logic level derived from the average voltage of the logic levels of all of the master supplies. This will provide an ambiguous polarity state to the remaining slaves.

Specifically, the synchronization read circuit 106 of each redundant supply 68 comprises a differential amplifier which transmits the common master-polarity signal as a polarity-average signal to the polarity-average input 94 d of the redundancy controller 94 . An analog-to-digital converter in the redundancy controller 94 samples the polarity-average input 94 d and determines the correct polarity state based upon the polarity-average signal.

When the respective redundant supply 68 is a master supply, then the redundancy controller compares the sampled voltage from the polarity-average input 94 d to the voltage the controller 94 is placing on the local master-polarity output 94 c for the local master-polarity signal. If the logic levels of the polarity-average input 94 d and the local master-polarity output 94 c do not match or are dissimilar, then the redundancy controller 94 of the respective master supply determines that more the one master supply is present in the power system 66 and changes the respective redundant supply 68 to a slave supply. Each of the master supplies in the power system 66 that detects an ambiguous polarity-average signal will become a slave supply. If no master supply remains, the arbitration circuits 82 will operate to declare a new master supply.

FIG. 11 shows a circuit which includes an example of an embodiment of the synchronization circuit 84 (see FIG. 10B ) of the redundant power supply 68 according to the present invention. If the respective power supply is declared to be a master, then master-slave status will turn FET Q 28 is turned on, which turns controlled switch U 7 on. Controlled switch U 7 provides power to optocoupler U 8 . If the respective power supply is a slave, optocoupler U 8 is unpowered, and is a high impedance output.

If optocoupler U 8 is powered, it sends an active high or low to the common polarity bus via resistor R 22 . The common polarity bus defines the polarity of the AC waveform generated by all the redundant power supplies. If the output of optocoupler U 8 is high, the AC output voltage is positive, and if the output of optocoupler U 8 is low, the AC output voltage is negative.

However, if there are two or more masters in the system, the common polarity bus can have three states. If the masters agree on the common polarity bus voltage, then the common polarity bus will be high or low as determined by the consensus. However, if the masters disagree, then the common polarity bus voltage will be the average of the voltages the masters are putting out.

Disagreement between multiple master will provide an ambiguous polarity state to any slaves. A differential amplifier U 9 provides the common polarity bus signal to the polarity-average input of the redundancy controller. The A/D converter of each slave supply's redundancy controller samples the polarity-average signal and decides the correct polarity state based on this information.

The redundancy controller of a master power supply compares the polarity-average signal voltage to the voltage it is putting onto the its local master-polarity output as a local master-polarity signal. If the two signals do not agree, the redundancy controller of the master supply knows that an arbitration situation is occurring, and acts to resolve it. Generally, this is achieved by having all masters putting out either a 1 or a 0 become slaves. This resolves the ambiguity. This process continues, eliminating ambiguous polarity signals, until there is only one master controlling the common polarity bus, and all other redundant power supplies are slaves.

Referring now to FIGS. 9 and 10C , the soft-start circuit in each redundant power supply 68 is connected to a start-ready bus 108 of the redundancy bus 78 , so that two or more redundant power supplies 68 in the redundancy circuit 66 can start producing a voltage at their AC outputs in synchronization. The soft-start circuit delays the AC output of each of the power sections 72 until all of the operating power supplies 68 have completed initialization or a start-up cycle and are ready to begin outputting to the common AC output bus 76 .

Specifically, each soft-start circuit 86 comprises a start relay 110 and a soft-start relay latch circuit 112 . The start relay 110 has normally-closed relay contacts connected between the start-ready bus 108 and the local bias return (ground) 93 . The start relay 110 has its coil connected to an output shutdown signal of the power section 72 . The output shutdown signal stays low until the power section 72 has completed initialization. An example of an output shutdown signal is described herein with reference to the power supply 10 as shown in FIG. 3 B.

Further, still referring to FIGS. 9 and 10C , the relay latch circuit 112 of the soft-start circuit 86 has a trigger input 112 a connected to the start-ready bus 108 and a soft-start output 112 b of the relay latch communicates with the power section 72 to indicate to the power section 72 when to begin outputting to the common AC output bus 76 .

Before the initialization of each redundant supply 68 , the respective start relay 110 begins with its coil de-energized, thereby shorting the start-ready bus 108 to the local bias return (ground) 93 through the relay contacts of the start relay 110 . The relay latch circuit 112 reads the status of the start-ready bus 108 , being shorted to ground, and stays low. Further, the respective start relays 110 of each of the other redundant supplies 68 in the power system 66 are shorted to ground through the start-ready bus 108 , preventing the units from starting their respective AC outputs.

Further, during initialization of the power section 72 of each redundant supply 68 , the output shutdown signal is low, keeping the coil of the start relay 110 off and keeping the start-ready bus 108 shorted to ground. When the power section 72 has completed initializing and is ready to start its respective AC output, it sets output shutdown signal high. This energizes the coil of the start relay 110 causing the contacts to open. Thus, once all of the power sections 72 in the power system 66 have completed initializing, none of the start relays remain shorting the start-ready bus 108 to ground. The relay latch trigger input 112 a of each redundant supply 68 reads the open state of the start-ready bus 108 and responds by latching open the contacts of the start relay 110 and releasing the soft-start output 112 b which allows the respective power sections to start their AC outputs in synchronization.

Once the redundant supplies 68 have begun delivering power to the common AC output bus 76 , if one or more of the power sections 72 fail or are otherwise shut down, the respective output shutdown signal goes low which shuts the power section 72 off, resets respective the soft-start output 112 b , and de-energizes the respective start relay 110 . Since the start relays 110 of each of the other redundant supplies 68 have been latched open by their respective relay latches 112 , the other redundant units do not shut down, despite the start-ready bus 108 being shorted to ground. In other words, the relay latch 112 does not read the start-ready bus 108 after startup. The relay latch 112 is only cleared after all power is removed from the respective supply 68 .

FIG. 12 shows an example of an embodiment of the soft-start circuit 86 (see FIG. 10C ) of the redundant power supply 68 according to the present invention. The soft-start circuit begins with a coil of a soft-start relay K 1 de-energized causing the contacts of the soft-start relay K 1 to short the start-ready bus to the local power supply ground G 1 . While each of the redundant power supplies are initializing, a non-inverting input of a soft-start differential amplifier U 10 is held to 0 volts, and an output of the soft-start differential amplifier U 10 is low. This low output pulls each redundant power supply's soft-start to the respective local power supply ground G 1 , preventing the power supply from starting up.

Further, while initializing, the output shutdown signal in each power supply is low, keeping coil of the soft-start relay K 1 off and also holding the soft-start output off though a pull-down diode D 31 . Once the respective power supply has completed initializing, it sets the output shutdown high, turning FET Q 29 on. FET Q 29 thus energizes the coil of the soft-start relay K 1 causing the contacts of the relay K 1 to open.

Once the respective soft-start relays K 1 of all of the redundant power supplies have been energized K 1 the start-ready bus goes open. With the start-ready bus open, a soft-start capacitor C 8 in each power supply begins charging, providing a delay until the capacitor C 8 causes the output of the respective soft-start differential amplifier U 10 to go high.

Once the output of the amplifier U 10 goes high, a soft-start latching diode D 32 causes the respective soft-start differential amplifier U 10 to latch on. The latched output of the amplifier U 10 releases the respective soft-start, and all units begin to start their outputs in synchronization.

If a power supply must subsequently be shut down, the respective output shutdown goes low, shutting the unit off, resetting the soft-start, and de-energizing the soft-start relay K 1 . The latching of the soft-start differential amplifier U 10 prevents the other power supplies from shutting down.

The soft-start circuit receives bias voltage from a local bias V 2 of the respective power supply, rather than via the common bias bus. Thus, a reset diode D 33 disengages the latching of the soft-start differential amplifier U 10 when the local bias V 2 is lost.

Referring now to FIGS. 9 and 10C , the overvoltage correction circuit 88 is provided to detect an overvoltage on the common AC output bus 76 and to identify which redundant power supply 68 in the redundant power system 68 is creating the overvoltage. The overvoltage correction circuit 88 comprises an overvoltage detect circuit 114 , a duty-cycle detect circuit 116 and a overvoltage latch circuit 118 .

The overvoltage detect circuit 114 is connected to the AC output of the respective power section 72 for sensing the voltage on the common AC output bus 76 . The duty-cycle detect circuit 116 is connected to sense the signal from the PWM, or PWM signal , used to generate the AC output of the power section 72 . An example of a PWM signal is the signal generated by the PWM 44 to control the amplitude circuit 41 , as described herein with reference to FIGS. 3A and 4 .

Still referring to FIGS. 9 and 10C , the overvoltage detect circuit 114 reads the peak output voltage on the common AC output bus 76 . If peak output voltage is higher than a predetermined value, the overvoltage detect circuit 114 sends an overvoltage signal to the overvoltage latch circuit 118 .

The duty-cycle detect circuit 116 senses the duty-cycle of the PWM signal of the respective power section 72 . If the PWM signal is at or near a predetermined maximum duty-cycle, the duty-cycle detect circuit 116 sends a maximum-duty-cycle signal to the overvoltage latch circuit 118 .

The overvoltage latch circuit 118 is connected to shutdown its respective power section 72 if the power section 72 is the cause of an overvoltage on the common AC output bus 76 . If the overvoltage latch circuit 118 simultaneously receives signals from both the overvoltage detect circuit 114 and the duty-cycle detect circuit 116 , then the latch circuit 118 will shut the power section 72 down and hold it off via the output shutdown signal.

All of the overvoltage detect circuits 114 in the power system 66 will sense an overvoltage on the common AC output bus 76 . However, only the redundant supply 68 which causes the overvoltage will be at maximum duty-cycle and thus be shut down by the overvoltage latch 118 . The other units will be at a minimum duty-cycle and thus their latches 118 will not be triggered.

FIG. 13 shows an example of an embodiment of the overvoltage correction circuit 88 (see FIG. 10C ) of the redundant power supply 68 according to the present invention. Specifically, the overvoltage detect circuit is connected to the AC output of the respective power supply. A diode bridge D 34 rectifies the signal from the AC output. A resistor R 23 , a capacitor C 9 and a zener diode D 35 provide a bias voltage to an optocoupler U 11 and a shunt regulator U 12 . When rectified AC output voltage rises higher than the breakdown voltage of a zener diode D 36 , the zener diode D 36 conducts current through a resistor R 24 to charge a capacitor C 10 . When the capacitor C 10 reaches the threshold of the shunt regulator U 12 , current conducts through the optocoupler U 11 , a resistor R 25 and the shunt regulator U 12 . This current turns on the optocoupler U 1 .

In the duty-cycle detect circuit, a comparator U 13 connected to the PWM and a reference voltage of the respective power supply is used to detect a maximum duty-cycle of power supply. When the maximum duty-cycle is reached, the comparator U 13 turns on the FET Q 30 . If an overvoltage occurs while the FET Q 30 is turned on, the FET Q 30 and the optocoupler U 11 will be simultaneously turned on, shorting the latch signal to the local ground G 1 , in order to shut down and hold off the power supply.

Two switches S 4 and S 5 are used to selectively switch two zener diodes D 37 and D 38 , respectively, into the circuit. These diodes D 37 ,D 38 have lower breakdown voltages than the other zener diode D 36 , and are used to set the overvoltage threshold or maximum peak voltage level to a lower value, when desired.

The present disclosure describes several embodiments of the invention, however, the invention is not limited to these embodiments. Other variations are contemplated to be within the spirit and scope of the invention and appended claims.