System and method for programmable switching characteristics of an analog switch in a transconductance amplifier

Provided are a method and system for controlling impedance in a transconductance amplifier. A system includes a first transconductance amplifier and a second transconductance amplifier configured to control electrical characteristics associated with the first transconductance amplifier. An operational amplifier is provided and has at least one input port connected to the second transconductance amplifier. Also included is a first digital to analog converter (DAC) connected to receive a current signal from the operational amplifier.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates generally to controlling characteristics of transconductance amplifiers.

2. Background Art

In transconductance amplifiers, switching resistance (e.g., degeneration impedance) is a characteristic circuit designers commonly seek to control through use of a number of different techniques. One traditional technique includes providing a transistor-based switch within the transconductance amplifier to control the degeneration impedance characteristics. Although this technique provides a level of control, its use produces several undesirable side-effects. For example, when changes in the degeneration impedance occur in a nonlinear fashion, or occur too quickly, glitches can occur in the output current of the amplifier.

Other traditional approaches to minimizing the effects of these glitches, as well as other instability problems, include the use of resistor capacitor (RC) delay circuits to retard the transition of the switch. The use of RC delay circuits is especially prevalent in low-noise applications, such as within programmable gain amplifiers.

The use of an RC delay circuits, however, requires large resistors and capacitors, which consumes integrated circuit (IC) board real estate. The RC circuits also require use of an extra pin on the IC.

Thus, although traditional techniques are available to minimize the occurrence of glitches and other instability factors in transconductance amplifiers, many of these techniques are costly and/or impractical.

What is needed, therefore, is a more practical approach to minimizing glitches in the output current of transconductance amplifiers caused by controlling the switching resistance. More specifically, what is needed is a method and system to programmably control the switching resistance in a transconductance amplifier. Also, what is needed is a system and method to control the speed at which the resistance switch is turned on and off, in order to further minimize glitches and other instability by-products.

BRIEF SUMMARY OF THE INVENTION

Consistent with the principles of the present invention, as embodied and broadly described herein, the present invention includes a circuit for controlling impedance in a transconductance amplifier. The circuit includes a first transconductance amplifier and a second transconductance amplifier configured to control electrical characteristics associated with the first transconductance amplifier. An operational amplifier is provided and has at least one input port connected to the second transconductance amplifier. Also included is a first digital to analog converter (DAC) connected to receive a current signal from the operational amplifier.

The present invention provides a system and method to generate a programmable switching resistance used in a transconductance amplifier in time and voltage domains. In the present invention, a switch is provided to control the degeneration impedance of the transconductance amplifier. The switch is connected to sources of a differential pair within the transconductance amplifier. The switch can be a hard switch, to turn on and turn off the degeneration path, or a soft switch to tune the total impedance of degeneration with an analog control voltage applied on the switch. In some applications, the switch must also turn on and off within certain time constraints in order to meet system requirements, such as the gain ramping rate, glitch, or stability.

DETAILED DESCRIPTION OF THE INVENTION

The following detailed description of the present invention refers to the accompanying drawings that illustrate exemplary embodiments consistent with this invention. Other embodiments are possible, and modifications may be made to the embodiments within the spirit and scope of the invention. Therefore, the following detailed description is not meant to limit the invention. Rather, the scope of the invention is defined by the appended claims.

It would be apparent to one skilled in the art that the present invention, as described below, may be implemented in many different embodiments of hardware, software, firmware, and/or the entities illustrated in the drawings. Any actual software code with the specialized controlled hardware to implement the present invention is not limiting of the present invention. Thus, the operation and behavior of the present invention will be described with the understanding that modifications and variations of the embodiments are possible, given the level of detail presented herein.

FIG. 1is a schematic illustration of a transconductance amplifier circuit100constructed in accordance with an embodiment of the present invention. The circuit100includes a Real Section102and a Calibration Section104responsible for a control signal along a path105. The real section102includes a first transconductance amplifier106, including a first differential transistor pair107having gates108. The transconductance amplifier106is configured to receive an input voltage VINas an input to the gates108, and produce a proportional output current at output port109.

The transconductance amplifier106also includes a switching circuit110including resistors111coupled to a transistor M4. A digital-to-analog converter (DAC)112is used to provide an analog voltage control signal via an output port113, based upon an input digital data word provided at an input port114of the DAC112.

The calibration section104is used to reproduce operating conditions associated with the Real Section102. More specifically, the calibration section104is provided to create operating conditions that are necessary to produce and control a voltage control signal (VC). The control signal VCis provided along the path105as an input to a gate of the transistor M4. In other words, the calibration section104provides an operating condition necessary to produce the control signal VC. The voltage control signal VCis required to slowly turn on and turn off the switch110in the transconductance amplifier106.

The calibration section104includes a replica circuit115including a replica transconductance amplifier116(replicating the amplifier106) configured to receive, for example, an input common-mode voltage (VCOM) at a gate of a transistor M1. The transconductance amplifier116does not need to input the actual input voltage (VIN), as provided to the gate108of the transistor107within the Real Section102. Instead, a common mode voltage (VCOM) of the input voltage (VIN) is provided to the gate of the transistor M1.

The replica transconductance amplifier116also includes a switch117, comprising resistors118and a transistor M2. The transistor M2is used as a switch to turn on and off an internal resistance of the switching circuit117. In some applications, the switching circuit117is desirably turned on and off slowly. For example, if the switch117is turned on at Vdd, then the transistor M2must reach Vddslowly, and must turn off slowly, to avoid a potential current glitch.

If the transistor M2turns on too quickly, its gain will suddenly change, thus producing the glitch. In order for the transistor M2to be turned on and off, its threshold voltage (Vth2) must be known. For purposes of illustration, an effective source-drain resistance of the switch117is denoted herein as RSW. The Vth2varies with process and has body effect if its substrate voltage is different from source node. The source/drain voltage is dependent on the common mode voltage of input signal.

Also within the calibration section104, a control voltage (VCX) is applied to a gate of the transistor M2. As the control voltage VCXis increased, the resistance RSWdecreases and eventually becomes negligibly small, as indicated inFIG. 2.

FIG. 2is a graphical illustration200of a relationship between the switch resistance RSWand the control voltage VC. This relationship is conveyed in the form of a voltage resistance curve202. InFIG. 2, as the control voltage VCTRLincreases and exceeds voltage VCX(equivalent to Vth2), the switch resistance RSWsuddenly changes and becomes small, and a channel is formed within the transistor M2.

The idea behind the present invention, is that if the threshold voltage Vth2can be known and created within the calibration section104(referring back toFIG. 1), the real section102can be programmed based upon the threshold voltage Vth2to slowly turn on the transistor M4within the switch110. Therefore, the calibration section104is used to create a threshold voltage (Vth4) for application to the transistor M4. In other words, the calibration section104replicates the estimated operating conditions of the real section102.

In order to create the proper operating conditions for the real section102, the calibration section104also includes an operational amplifier119, a DAC120, and a transistor M5. The operational amplifier119produces an output121in the form of a bias voltage that is provided as an input to the DAC120and the DAC112. The DAC120produces an output at an output port122which is connected along a feedback path124to a source of the transistor M5and to an input port of the operational amplifier119. The DAC120also includes an input port126for receiving a digital programming word.

The transistor M5is desirably substantially similar to the transistor M2. Once an electrical loop along the path124is closed, a threshold voltage (Vth5) associated with the transistor M5will be substantially equal to the threshold voltage Vth2associated with the transistor M2. Also, when the loop124is closed, a voltage Vfbwill be equal to a voltage (Vrefprovided as the other input to the operational amplifier119.

The loop124forces the voltage Vfbequal to the voltage Vref. When this happens, the voltage VCX, at a node128and associated with the transistor M5, will become equal to a voltage occurring at the gate130of the transistor M2. These voltage equivalences occur because of the similarities between the transistors M2and M5. In this analogy, the voltage Vth2is the voltage that barely turns on the transistor M2. Thus, through derivation, VCX=Vth2=the voltage that barely turns on M2.

The operational amplifier119and the DAC120play an integral role in producing the operating conditions for the real section102. For example, the DAC120is used to generate a programmable switch control voltage based on turn-on threshold voltage Vth2produced by the calibration circuit.

In the calibration section104, VCOMis the common-mode voltage of the input voltage (VIN) to the replica transconductance amplifier116. Vrefis the source voltage of the transistor M1. Vfbis the source voltage of the transistor M5. The operational amplifier119forces Vrefand Vfbequal in steady state. Vref=(VCOM−VthM1), where VthM1is a gate-to-source voltage of the transistor M1. On the other hand, VCX=Vth5+VFB, where Vth5is the gate-to-source voltage of the transistor M5.

Since Vfb=Vref, a body effect of the transistor M5is the same as that of the transistor M2. Since the transistor M2is close to a pseudo ground of the differential pair transistors M6, there is no direct current (DC) flowing in/out of the transistor M2. The turn-on threshold voltage Vth2of the transistor M2, between its gate and its source, is very close to the intrinsic threshold voltage of the transistor M2with body effect, which is also very close to Vth5if the transistor M5has a low over-drive voltage.

Since the source voltage of the transistor M2=VCOM−VthM1, the turn-on threshold voltage for M2=VCOM−VthN1+Vth2=Vref+Vth5=Vfb+Vth5, which is equal to VCX. Thus, VCXis equivalent to the threshold voltage Vth2that turns on the transistor switch M2. The voltage VCXis also the output of the DAC112, as illustrated inFIG. 3.

FIG. 3is a more-detailed schematic illustration of the DAC112shown inFIG. 1. The DAC112is implemented as a non-linear DAC, based on a sum of programmable current sources I0-INwith a resistance load. The amount of current in the DAC112is controlled by an output current produced by the operational amplifier119. The DAC112is configured to receive a digital input code value via the input port114. If the digital input code is at a mid-code value, an output voltage of the DAC112will produce the voltage VCXas a central voltage value.

In the DAC112, the input port114is programmable and provides external connections to input ports D0-DNof flip-flop FF0-FFN. The flip-flops FF0-FFNinclude a clock input port CLK for receiving an input clocking signal. The DAC112will be discussed in greater detail below.

FIG. 4is a graphical illustration400of an exemplary nonlinear current source DAC, such as the DAC112ofFIG. 1. In the example ofFIG. 4, the DAC current sources are symmetrical (or asymmetrical depending on the switching characteristics) at a center location. That is, for example, the DAC112has its minimum current in a middle of the curve400. Since the calibration section104is programmed at a half-on condition, characteristics of an output of the DAC112will be centered at the voltage VCX, as illustrated below inFIG. 5.

FIG. 5is a graphical illustration500of voltage distribution within the DAC112, based upon multiple current sources within the DAC112, as illustrated inFIG. 3, discussed above.

In general, DACs can be implemented as a collection of current sources with switches. If a DAC, such as the DAC112, is, for example, a 3-bit DAC, then the DAC112will include eight switches (e.g., 23). Therefore, an exemplary input code for a 3-bit DAC might be 0, 1, 2, . . . 7. When the DAC112is configured to be half-on, only the switches 0-3 are set to their on position. The switches numbered 4-7 are set to their off position. If the current source distribution in the current sources I0-INis nonlinear, as illustrated inFIG. 4, the current will change very small in the middle, but changes more quickly on both sides (i.e., at the edge of the curve). Because the curve400desirably changes very gradually in the middle, that middle section of the curve400will have the highest slope (slow region). Therefore, a middle voltage value will desirably change very slowly.

At each edge of the curve400, however, the change of current source is large. Thus, by combining the current source distribution ofFIG. 4and the code values illustrated above, a nonlinear curve will be produced.

FIG. 5is a graphical illustration of a nonlinear curve500produced by combining the current source distribution ofFIG. 4and the code values above. InFIG. 5, in other words, at or about the mid-code value, the control voltage (VCX) changes slowly. But near the outer region, the code changes very quickly.

FIG. 2is an illustration of nonlinear resistance characteristics RSWas a function of control voltage VCTRLfor the transistor M4. When VCis less than VCX, the transistor M4is in its off state. When VCis greater than VCX, the transistor M4starts to turn on, and its associated turn-on resistance starts to drop. When combiningFIG. 5andFIG. 2, the effective impedance RSWcan be illustrated as a function of a desirable digital code value. Such an illustration if shown inFIG. 6.

FIG. 6is a graphical illustration indicating that by combiningFIGS. 2 and 5, the impedance RSWbecome substantially linear, since the nonlinear effect of the transistor switch M4and the DAC112will cancel out. For example, the characteristics of the impedance RSWcan be controlled in the time domain. More specifically, by multiplying the curve200ofFIG. 2by the curve500ofFIG. 5, the nonlinear curve600ofFIG. 6is produced. By changing the code, the on resistance will be linearly changed.

The input digital code provided at the input port114of the DAC112can be programmed by a waveform having any frequency. By manipulating the frequency of the input waveform, the frequency of sweeps of the DAC112are subsequently affected. Thus, in this manner, the input digital code of the DAC112enables the DAC112to be controlled in the time domain.

FIG. 7is a graphical illustration of an exemplary waveform700representing an input code word that can be used to sweep the DAC112. The input waveform700includes an exemplary clock period702for modulating the input code value of the DAC112between, for example, input code values 2-4, which represent a middle code value.

FIG. 8is a graphical illustration of the switch impedance RSWin the time domain. That is, a clock waveform800is illustrated which can be provided as an input to the clock input of the flip-flops FF0-FFN, to sweep the DAC112. The resolution of the programmable switch impedance RSWis a function of the resolution of the DAC112. Changing the number of current sources activated within the DAC112, changes the voltage. And by changing the clock speeds using exemplary waveforms such as the waveforms700and800ofFIGS. 7 and 8, respectively, the resistance RSWcan be programmed in two dimensions of programmability, voltage and time.

The circuit100, in an alternative embodiment, can also be used as a programmable gain amplifier (PGA). In this alternative embodiment, the resistors111and118of the switches110and117, respectively, can be removed. By removing the resistors, the switches110and117can be used not as a switch, but as a voltage-controllable resistance source. That is, if the transconductance has an impedance load, it is a amplifier. Since the gain of amplifier is dependent on the on-resistance of the switch, the programmable on-resistance of the switch can be used to control the gain of an amplifier.

FIG. 9is a flowchart of an exemplary method900of practicing an embodiment of the present invention. InFIG. 9, operating conditions of the transconductance amplifier are substantially replicated, as illustrated in step902. Once the operating conditions are replicated, a current signal is produced to control the degeneration impedance of the real transconductance amplifier based upon the operating conditions, as noted in step904. In step906, a digital word is provided to cooperate with the produced current signal and to produce a switch control signal accordingly. Finally, in step908, the switch control signal is provided to the transistor switch within the transconductance amplifier106.

CONCLUSION

The present invention provides a low cost system and method for an analog switch to control the output current in a transconductance amplifier. The output current is controlled through use of a programmable switching resistance, programmable in both time and voltage domains. Additionally, within the system of the invention, the ramp-up and ramp-down rates of the DAC can be precisely controlled using a specific input clocking waveforms.

Any such alternate boundaries are thus within the scope and spirit of the claimed invention. One skilled in the art will recognize that these functional building blocks can be implemented by analog and/or digital circuits, discrete components, application-specific integrated circuits, firmware, processor executing appropriate software, and the like, or any combination thereof. Thus, the breadth and scope of the present invention should not be limited by any of the above-described exemplary embodiments, but should be defined only in accordance with the following claims and their equivalents.