Regulated deflection system

The trace switch of a horizontal deflection circuit is coupled to a secondary winding of a flyback transformer. The primary winding is coupled to a source of energy and a regulator switch. A control circuit varies the phase angle of the regulator switch in accordance with an energy level of the deflection circuit. A regulator commutating inductance in combination with a commutating and tuning capacitance controls the duration of conduction of the regulator switch. The capacitance independently tunes with a flyback transformer winding to transfer energy from the source in a resonant manner.

BACKGROUND OF THE INVENTION 
This invention relates to voltage regulators such as used with television 
deflection circuits. 
Circuit arrangements frequently used in television receivers combine 
switched mode power supplies (SMPS) with transistor horizontal deflection. 
Various types of SMPS circuits have been used; many have a common feature 
of providing a regulated DC supply to the horizontal deflection circuit. 
The horizontal deflection circuit, however, draws an AC current from the 
power supply. By avoiding the necessity of providing a regulated DC input 
voltage, a substantial saving in circuit costs and a substantial increase 
in circuit efficiency may be obtained. 
Conventional switched mode transformers for television receiver application 
are of the flyback or backwards converter type, require a relatively close 
coupling, have critical tolerances, and are relatively expensive to 
manufacture. In a commonly used switched mode system using a backwards 
converter with transistor regulator switch, the AC voltage at the 
secondary side of the switched mode transformer is rectified and filtered 
by a capacitor. The DC voltage across the filter capacitor provides the 
input supply voltage for the horizontal output stage. It would be 
desirable to omit such a separate rectifying step. 
Other regulator circuits include a flyback transformer primary winding 
coupled to a regulator switch, the horizontal deflection winding, retrace 
capacitor, and trace switch being coupled to a flyback secondary winding. 
A capacitor tunes with the flyback transformer for energy transfer to the 
deflection circuit. In such circuits, however, the conduction time of the 
regulator switch cannot be selected independent of the tuning requirements 
for the flyback transformer. 
SUMMARY OF THE INVENTION 
A transformer includes first and second windings. A trace switch of a 
deflection circuit is coupled to the second winding and to a deflection 
winding. The first winding is coupled to a source of energy and a 
regulator switch. The regulator switch's phase angle is controlled by a 
control circuit that is responsive to an energy level of the deflection 
circuit. A tuning capacitance is coupled to the transformer for 
transferring energy from the source in a resonant manner. The capacitance 
in combination with a commutating inductance controls the duration of 
conduction of the regulator switch.

DESCRIPTION OF THE INVENTION 
In the regulated horizontal deflection circuit 20, illustrated in FIG. 1, 
AC line mains voltage, not shown, of a value 220 VAC, for example, is 
full-wave rectified and coupled to an unregulated B+ supply voltage input 
terminal 21, and is filtered by a capacitor 22. Input terminal 21 is 
coupled to a primary winding 23a of a horizontal output or flyback 
transformer 23. A bidirectionally conductive regulator switch 24 
comprising for example, an ITR, or for example, a silicon controlled 
rectifier SCR 25 and a parallel oppositely poled diode 26 is coupled to 
primary winding 23a. A regulator switch commutating circuit 27 for 
commutating off regulator switch 24 is coupled across regulator switch 24 
and comprises a series coupled inductor 28 and capacitor 29. A damping 
network comprising a resistor 30 and a capacitor 31 is also coupled across 
regulator switch 24. Other switching arrangements such as transistor 
switches may be substituted for the ITR of switch 24. 
Primary winding 23a is wound on a leg 123a of a rectangular core 123 of 
horizontal output transformer 23. Wound on an opposite leg 123b is a 
secondary winding 23b. Air gaps 223a and 223b are formed in respective 
legs 123a and 123b. 
One terminal of secondary winding 23b is coupled to a capacitor 32. Another 
terminal of winding 23b is coupled to a horizontal trace switch 33 of a 
horizontal output stage 34. Horizontal output stage 34 comprises a 
series-coupled horizontal deflection winding 35 and a trace capacitor 36, 
a retrace capacitor 37 and a trace switch 33, which itself is comprised of 
a horizontal output transistor 38 and a damper diode 39. A conventional 
horizontal oscillator and driver circuit 40 couples scan synchronized 
horizontal rate switching signals to the base or control electrode of 
horizontal output transistor 38 to turn on the transistor during the 
horizontal trace interval and to turn off the transistor to initiate the 
horizontal retrace interval. 
A high voltage winding 23c of horizontal output transformer 23 is coupled 
to a conventional high voltage circuit 41 for developing a beam current 
ultor voltage. Although high voltage winding 23c and winding 23b are 
illustrated in FIG. 1 as being adjacent each other on core 123b, in order 
to provide tight magnetic coupling between the two windings, high voltage 
winding 23c is wound over winding 23b. Other horizontal output transformer 
windings, not shown, may provide utility pulses for such functions as 
horizontal blanking and may also provide secondary supply voltages for use 
by such circuits as the vertical, audio, and video processing circuits. 
Isolation of horizontal deflection circuit 20 and the other load circuits 
of transformer 23 from the AC line mains supply is provided by transformer 
23. 
To provide for regulation of horizontal deflection circuit 20, a regulator 
control circuit 42 couples horizontal rate turn-on gating signals 45 to 
the gate of SCR 25 of regulator switch 24 through a coupling transformer 
43 and a capacitor 44. Horizontal rate pulse-width modulated signals are 
obtained from a conventional pulse-width modulator 46 such as a Texas 
Instrument SN74121, Texas Instruments, Dallas, Tex., or a Philips TDA2640, 
Philips Gloeilampenfabrieken, Eindhoven, Netherlands. The width of the 
pulses are modulated in accordance with an energy level of horizontal 
deflection circuit 20. The energy level selected is the horizontal retrace 
pulse amplitude obtained from a winding 23d of horizontal output 
transformer 23. Horizontal rate scan synchronizing signals are coupled to 
modulator 46 from horizontal oscillator and driver 40. 
The pulse width modulated signals from modulator 46 are differentiated by a 
capacitor 47 and resistors 48 and 49 and are coupled to the base of a 
pulse squaring transistor 50, the base being coupled to the junction of 
resistors 48 and 49. The collector of transistor 50 is coupled to one 
terminal of the primary winding of coupling transformer 43 through a 
resistor 51. Another terminal of transformer 43 is coupled to a +V supply. 
Transistor 50 converts the differentiated pulse width modulated signals 
from modulator 46 into the pulse position modulated gating signals 45. A 
diode 54 removes the negative portions of the differentiated pulse width 
modulated signals and a resistor 52 and a diode 53 damp transients 
developed across the primary winding of coupling transformer 43. 
The voltage V.sub.33 across trace switch 33 is illustrated in FIG. 2a and 
equals approximately zero during the trace interval between times t.sub.1 
-t.sub.4 and a retrace pulse between times t.sub.4 -t.sub.5. At a 
controlled instant t.sub.2 within the first portion of the horizontal 
trace interval, regulator control circuit 42 provides a gating signal 45 
to SCR 25 and turns on regulator switch 24. The input current i.sub.23a 
flowing in primary winding 23a of horizontal output transformer 23 begins 
to linearly increase from time t.sub.2, as illustrated in FIG. 2b. At time 
t.sub.2, a sinusoidal commutating current i.sub.24, obtained from 
regulator commutating circuit 27, begins to flow in regulator switch 24, 
as illustrated in FIG. 2d by the current i.sub.24 and by FIG. 2e, the 
voltage V.sub.24 across switch 24. After approximately one complete cycle 
of oscillation of current i.sub.24, regulator switch 24 is commutated off 
at time t.sub.3, still within the trace interval, at which time primary 
winding current i.sub.23a begins to decrease. 
With primary winding 23a and secondary winding 23b wound on opposite legs 
of core 123, a substantial leakage inductance 54 exists between the two 
windings, on the order of 2.3 millihenries, for example. The current 
i.sub.23b flowing in deflection-coupled secondary winding 23b and in 
capacitor 32 is illustrated in FIG. 2c. The voltage across secondary 
winding 23b is rectified by trace switch 33 during the start-up interval 
and charges capacitor 32 to an average DC voltage which is the DC value of 
retrace pulse voltage V.sub.33. Capacitor 32 blocks the DC short-circuit 
path from winding 23b. During steady-state operation, the average voltage 
across capacitor 32 equals the average value of retrace pulse voltage 
V.sub.33. 
With regulator switch 24 and trace switch 33 conducting during the middle 
portion of trace between times t.sub.2 -t.sub.3 of FIG. 2, a simplified 
equivalent circuit for the circuit of FIG. 1 is illustrated in FIG. 3, 
assuming, for example, a one-to-one transformation ratio between primary 
winding 23a and secondary winding 23b of flyback transformer 23. L.sub.a 
represents the inductance of winding 23a and L.sub.e represents the 
leakage inductance 54. The B+ supply voltage is coupled across La. Because 
capacitor 32 is relatively large valued, and because the interval when 
both switches 24 and 33 are conducting is relatively short, capacitor 32 
has been replaced in the equivalent circuit by a DC voltage source E equal 
in magnitude to the average voltage across capacitor 32. 
The current i.sub.a through La and the current i.sub.E through L.sub.e are 
each linearly increasing with slopes respectively depending on the B+ 
voltage and the voltage difference between B+ and E. The algebraic sum of 
these two currents equals the input current i.sub.23a. The current i.sub.E 
through L.sub.e equals the secondary winding current i.sub.23b. 
During the beginning and ending portion of the trace interval between times 
t.sub.1 -t.sub.2 and t.sub.3 -t.sub.4, regulator switch 24 is 
nonconducting whereas trace switch 33 is still conducting. The simplified 
equivalent circuit for these conditions is illustrated in FIG. 4, where 
C.sub.29 equals the capacitance of capacitor 29 of regulator switch 
commutating circuit 27 and L.sub.28 equals the inductance of inductor 28. 
A sinusoidal loop current i.sub.s flows in the circuit of FIG. 4, with a 
frequency defined by the series coupling of C.sub.29, L.sub.28, and the 
parallel arrangement of L.sub.a and L.sub.e. Also flowing is the sawtooth 
loop current i.sub.E '. The input current i.sub.23a is the algebraic sum 
of the currents through L.sub.a and L.sub.e and thus equals only the 
sinusoidal current i.sub.s. The current i.sub.23b through flyback 
secondary winding 23b is the algebraic sum of the input current i.sub.23a 
multiplied by L.sub.e /L.sub.a and the sawtooth current i.sub.E '. 
During retrace, the simplified equivalent circuit for FIG. 1 is illustrated 
in FIG. 5, where L.sub.35 equals the inductance of deflection winding 35 
and C.sub.37 equals the capacitance of retrace capacitor 37. Because the 
B+ voltage source and storage capacitor 32 are effectively in series with 
C.sub.29 and C.sub.37 respectively, they have been omitted. Similarly, 
because of its relatively large value, capacitor 36 has also been omitted. 
The current through L.sub.e equals i.sub.23b and functions to replenish 
load-derived losses occurring in the resonant retrace circuit 60 
comprising L.sub.35 and C.sub.37. This current comprises the 
superpositions of several sinewave frequencies, with the highest and most 
significant frequency typically being the resonant retrace frequency. 
Another component to i.sub.23b comprises a DC load current component. 
The inductances L.sub.a and L.sub.e are typically substantially larger than 
the inductance L.sub.35 of horizontal deflection winding 35. The input 
current i.sub.23a will therefore be proportional to i.sub.23b during 
retrace and will ideally be a portion of a sinewave 61 between times 
T.sub.1 -T.sub.2, as illustrated in the idealized waveforms of FIG. 6, 
with a peak magnitude of I.sub.1 at the beginning of retrace at time 
T.sub.1 and a peak magnitude of I.sub.2 at the end of retrace at time 
T.sub.2. Although shown to be equal, magnitudes I.sub.1 and I.sub.2 will 
differ as a function of retrace loading. 
From time T.sub.2 of FIG. 6, the beginning of the trace interval, until 
time T.sub.3, the beginning of the regulator switch 24 commutating 
interval, the input current decreases in a sinusoidal manner to a 
magnitude I.sub.3, as illustrated by the heavy solid line portion 62a of 
the sinusoidal waveform 62. The frequency of sinewave 62 is determined by 
the equivalent circuit illustrated in FIG. 4 when regulator switch 24 is 
nonconductive and trace switch 33 is conductive. Switch 24 becomes 
conductive at time T.sub.3 in response to a gating signal 45 coupled to 
SCR 25 from control circuit 42, the instant T.sub.3 of FIG. 6 being 
illustratively the turn-on instant for low AC mains voltage. Regulator 
switch 24 is conductive for the interval T.sub.3 -T.sub.4 and input 
current i.sub.23a equals a positive going sawtooth current 63, reaching a 
peak magnitude I.sub.4 at time T.sub.4. At time T.sub.4, regulator switch 
commutating circuit 27 commutates off regulator switch 24. 
The equivalent circuit between time T.sub.4 and time T.sub.5 the beginning 
of the next retrace interval is again that illustrated in FIG. 4, because, 
between times T.sub.4 -T.sub.5, regulator switch 24 is nonconductive 
whereas trace switch 33 is still conductive. Input current i.sub.23a is 
thusly a sinewave portion 62a' of a sinusoidal waveform 62'. Sinusoidal 
waveforms 62 and 62' are of the same frequency because they are both 
reepresented by the same equivalent circuit of FIG. 4. Input current 
i.sub.23a, however, differs in value at times T.sub.2 and T.sub.4, the 
beginning instants for which the equivalent circuit of FIG. 4 is a valid 
representation. Because the initial current conditions differ, the phases 
and amplitudes of the two waveforms 62 and 62' also differ. 
At time T.sub.5, the beginning of retrace, input current i.sub.23a has 
returned to the value of -I.sub.1, thereby beginning a new cycle of 
operation. Assuming constant load conditions, to provide both a relatively 
constant high voltage and a constant peak-to-peak scan current in 
horizontal deflection winding 35, input current i.sub.23a is maintained at 
a constant magnitude I.sub.1 at the beginning of retrace, at times T.sub.1 
and T.sub.5, With I.sub.1 maintained constant, the input current at the 
end of retrace reaches the amplitude I.sub.2, regardless of the AC mains 
variations. 
For high AC mains voltage, during the first portion of trace, beginning at 
time T.sub.2, when the equivalent circuit of FIG. 4 is operative, input 
current i.sub.23a follows the sinusoidal portion 162a of a sinusoidal 
waveform 162, as illustrated by the heavy dotted waveform of FIG. 6 
between times T.sub.2 -T.sub.3 '. Waveform 162, illustrating high AC mains 
conditions is of the same frequency as waveform 62, illustrating low AC 
mains conditions. The slope of waveform portion 162a, however, is steeper 
than the slope of portion 62 because sinewave 162 has a higher amplitude 
than sinewave 62 due to the total energy in the circuit being greater at 
high AC mains voltage than at low AC mains voltage. 
Thus, at the later time T.sub.3 ', the instant when regulator switch 24 is 
made conductive for high AC mains conditions, input current i.sub.23a has 
decreased to a negative value -I.sub.3 ' when compared to the positive 
value +I.sub.3 for low AC mains conditions. 
Between times T.sub.3 '-T.sub.4 ', the regulator switch 24 commutating 
interval, input current i.sub.23a equals a sawtooth current 163. Because 
the B+ voltage is greater for high AC mains conditions, the slope of 
sawtooth current 163 is greater than the slope of sawtooth current 63. The 
magnitude of input current at the end of the regulator switch commutating 
interval for high AC mains voltage at time T.sub.4 ' is I.sub.4 ' and is 
greater than the magnitude I.sub.4 at time T.sub.4 for low AC mains 
voltage. 
Between time T.sub.4 ' and time T.sub.5, the beginning of the next retrace, 
the equivalent circuit is again that of FIG. 4. Input current I.sub.23a 
equals a sinusoidal portion 162a' of a sinusoidal waveform 162', as 
illustrated by the heavy dotted waveform between times T.sub.4 ' and 
T.sub.5. 
The frequencies of sinusoidal waveforms 62' and 162' are the same since 
they are both represented by the equivalent circuit of FIG. 4. Because, 
however, for high AC mains voltage, the initial input current magnitude of 
I.sub.4 ' at the later time T.sub.4 ' is greater than the initial 
magnitude of I.sub.4 at the earlier time T.sub.4, for low AC mains 
voltage, the slope of waveform 162a' is greater than the slope of waveform 
62a'. Therefore, regardless of the AC mains voltage variations, the input 
current magnitude at the beginning of retrace is a constant I.sub.1 for 
constant load conditions, as is required to achieve high voltage 
regulation. 
With the regulator switch 24 commutating interval T.sub.3 -T.sub.4 or 
T.sub.3 '-T.sub.4 ' substantially of fixed duration, as determined by the 
fixed resonant frequency of regulator switch commutating circuit 27, 
regulation for AC mains voltage variations is achieved by varying the 
turn-on instant of regulator switch 24. The turn-on instant of regulator 
switch 24 is similarly varied with load current variations. 
At a constant B+ voltage, the magnitude I.sub.1 of the input current 
i.sub.23a, at the beginning of retrace, would decrease with increased 
loading by high voltage circuit 41 if the turn-on instant were to remain 
unchanged. This decrease in I.sub.1 with increased load current would 
cause both the high voltage and horizontal scanning or deflection current 
amplitude to decrease thereby providing a measure of picture width 
stability. However, to minimize the high voltage circuit impedance, it may 
be desirable to maintain a relatively constant magnitude I.sub.1 with load 
current variations. Thus, by advancing the turn-on instant within trace of 
regulator switch 24, the magnitude I.sub.1 is maintained relatively 
constant despite load current increases. 
FIG. 7 illustrates a portion of the circuit of FIG. 1 that includes a 
different arrangement for a regulator switch commutating circuit 127 than 
that of commutating circuit 27 of FIG. 1. An inductor 128 of commutating 
circuit 127 is coupled between flyback winding 23a and regulator switch 
24. A capacitor 129 is coupled between ground and the junction of inductor 
128 and winding 23a. The function and operation of regulator switch 
commutating circuit 127 is similar to that described previously for 
circuit 27. 
An advantage of the arrangement of FIG. 7 is that inductor 128 is only 
coupled in the transformer circuit during the regulator commutating 
interval. Using the regulator commutating circuit 27 of FIG. 1, a change 
in inductance value changes both the regulator commutating interval 
duration and also changes the tuning of the transformer during the 
remainder of the deflection cycle. With the arrangement of FIG. 7, the 
value of inductor 128 may be changed without affecting circuit operation 
during the regulator switch off-time. 
Another advantage of the arrangement of FIG. 7 is that input current 
i.sub.23a during the regulator commutating interval includes a sinewave 
component thereby reducing RFI radiation. Furthermore, with inductor 128 
in series with regulator switch 24, the di/dt of the switch current during 
switch turn-on is reduced, thereby further reducing RFI radiation. 
In either arrangement, the regulator commutating circuit capacitor performs 
a dual function. The capacitor combined with the regulator commutating 
inductor establishes the regulator commutating interval or the duration of 
conduction of regulator switch 24. The regulator capacitor also 
independently functions to tune with the flyback transformer inductances 
La and Le to transfer energy from the B+ voltage source in a resonant 
manner. Regulation as well as circuit efficiency is improved. The 
effective high voltage impedance is minimized. 
By varying the on-time of regulator switch 24 within trace and keeping the 
regulator switch nonconductive during retrace, the high voltage and 
deflection current amplitudes are relatively easily regulated. Because a 
separate commutating inductance, other than one of the flyback transformer 
associated inductances, is used in conjunction with the regulator 
capacitor, the duration of the commutating interval of the regulator 
switch may be selected substantially independently of the tuning 
requirements of the flyback transformer. Improved regulation and 
efficiency results. Typically, the commutating interval duration is 
selected at approximately one-half the trace interval duration. 
Selected FIG. 1 circuit values and component descriptions are given below. 
B+ voltage: 
285 volts, nominal 
Capacitor: 
22: 400 microfarad 
29: 68 nanofarad 
31: 1 nanofarad 
32: 3.3 microfarad 
36: 1.2 microfarad 
37: 11.5 nanofarad 
Resistor 30: 
1.2 kilohm 
Inductor 28: 
350 microhenry 
Deflection Winding 35: 
1.1 millihenry 
1.2 ohms 
L.sub.a : 4.9 millihenry 
L.sub.e : 2.3 millihenry 
Flyback Transformer 23: 
Core: UU59 3c8 material from Philips Gloeilampenfabrieken 
Air gaps: 0.3 millimeter, each leg 
Winding 23a: 100 turns 10.times.0.15 m.m. Litz wire 
Winding 23b: 119 turns 0.5 m.m. enameled copper wire 
Winding 23c: 818 turns 0.1 m.m. enameled copper wire 
Winding 23d: 6 turns 0.5 m.m. enameled copper wire