Soft-switched power converters

A soft-switched boost converter includes an active snubber to provide soft switching of all semiconductor components. Specifically, the current (“turn-off current”) in the rectifier is switched off at a controlled rate, the main switch is closed under zero-voltage switching (ZVS) condition, and the auxiliary switch in the active snubber is opened under zero-current switching (ZCS) condition. As a result, switching losses are reduced with beneficial effects on conversion efficiency and EMC performance.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates to DC/DC and AC/DC power converters. More particularly, this invention relates to DC/DC and AC/DC power converters with soft switching in all of their semiconductor components.

2. Discussion of the Related Art

The boost converter topology has been extensively used in various AC/DC and DC/DC converter applications. In fact, the boost technology is used today in most front ends of DC/DC power supplies having power-factor correction (PFC). The boost topology is also used in numerous applications in which a battery-powered low input voltage is used to generate a high output voltage. At higher power levels, the continuous conduction mode (CCM) boost converter is the preferred topology for a front end with PFC. Thus, in recent years, significant efforts have been made to improve the performance of high-power boost converters. These development efforts have focused on reducing adverse reverse-recovery characteristics that affect the conversion efficiency and the electromagnetic compatibility (EMC) of the boost rectifier.

Generally, reverse-recovery-related losses and EMC problems are minimized by “softly” switching off the boost converter at a controlled turn-off current rate. Many soft-switched boost converters have been proposed that use additional components to form a snubber circuit (passive or active) to control the rate of change of the turn-off current in the boost rectifier. In a passive snubber circuit, only passive components such as resistors, capacitors, inductors, and rectifiers are used. In an active snubber circuit, in addition to the passive elements, one or more active switches are used. Although a passive lossless snubber can improve efficiency, its performance is inadequate to make it useful in high-performance PFC circuit applications. Generally, a passive lossless snubber circuit suffers from increased component stresses and is difficult to operate with soft-switching of the boost switch, which is detrimental in high-density applications that require increased switching frequencies.

Some active snubbers can provide simultaneous reverse-recovery loss reduction and boost switch soft-switching. However, most of these active snubbers offer soft turn-off in the boost rectifier, zero-voltage switching (ZVS) in the boost switch, and “hard” switching in the active-snubber switch. Active-snubbers that implement soft-switching of all semiconductor components (e.g., soft turn-off in the boost rectifier, ZVS in the boost switch, and zero-current switching (ZCS) in the active-snubber switch) are desired.

SUMMARY OF THE INVENTION

According to the present invention, a new soft-switched boost converter includes an active snubber to provide soft switching of all semiconductor components. Specifically, the current (“turn-off current”) in the rectifier is switched off at a controlled rate, the switch is closed under zero-voltage switching (ZVS) condition, and the auxiliary switch in the active snubber is opened under zero-current switching (ZCS) condition. As a result, switching losses are reduced with beneficial effects on conversion efficiency and EMC performance.

In one embodiment, a circuit according to this invention may include an active snubber having a two-winding transformer, an auxiliary switch, a blocking diode, and a voltage-clamp circuit that is used to reset the magnetizing energy of the transformer. According to another embodiment of the present invention, the active snubber circuit includes a three-winding transformer that generates an isolated auxiliary power supply, in addition to providing soft-switching of all semiconductor devices.

In one embodiment, the power converter includes a storage inductor coupled to an input terminal of the power converter, a first switch operating to charge and discharge the storage inductor, an active snubber including a second switch, a rectifier operatively coupled to the storage inductor, the active snubber and the output terminal to transfer energy from the storage inductor to the output terminal; and a control circuit. The control circuit operates the first and second switches over a switching cycle, such that the second switch closes to cause a current in the rectifier to be diverted from the rectifier to the active snubber, so as to allow the first switch to close at a substantially zero voltage condition and, in turn, to allow the second switch to open at substantially zero current condition. The active snubber can be referenced to any stable DC voltage in the power converter, such as the voltage at the output terminal, the voltage at the input terminal, the voltage at a terminal of the storage inductor, or the negative rail of the power source.

In one implementation, the active snubber includes a voltage clamp circuit, which includes a transformer, a capacitor, and a resistor. The transformer may have a turns ratio between a primary winding and a secondary winding of less than 0.5. Where the leakage inductance of the transformer is not large enough to provide a desirable turn-off current in the rectifier, an external snubber inductor of a predetermimed inductance can be coupled between the storage inductor and the voltage clamp circuit. The active snubber can also be provide one or more isolated power supplies, that can regulated independently of the voltage at the output terminal.

The present invention is applicable to numerous converter configurations, such as the boost converter, the forward converter, the buck converter or the buck/boost converter configuration. Further, the present invention is applicable to both DC and AC power sources. In one embodiment, where the invention is applied to power conversion of an AC power source, the first switch and the rectifier are provided as parts of a full-wave rectifier. The present invention is also applicable to both single-phase and three-phase power sources.

To facilitate cross referencing among the figures, like objects in the figures are assigned like reference numerals.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 1shows soft-switched boost circuit100, in accordance with one embodiment of the present invention. Soft-switched boost circuit100includes voltage source101representing input voltage VIN, boost inductor102(inductance value LB), boost switch103, boost rectifier104, energy-storage capacitor105(capacitance value CB), load106(resistance value RL), and active snubber circuit107formed by auxiliary switch108, transformer109, blocking diode110, and clamp circuit115formed by resistor111(resistance value RC), clamp capacitor112(capacitance value CC) and diode113. To facilitate explanation of circuit operation in soft-switched boost circuit100,FIG. 2provides simplified circuit model200for soft-switch boost circuit100, with voltage and current reference directions indicated.

InFIG. 2, voltage sources201and202model energy-storage capacitor105and clamp capacitor112, respectively, by assuming that the capacitance value CBof energy-storage capacitor105and the capacitance value CCof clamp capacitor112are large enough so that the voltage ripple across each capacitor is small compared to its DC voltage. In addition, constant current source IINmodels boost inductor102by assuming that inductance value LBis sufficiently large so that the current through boost inductor102over a switching cycle does not change significantly. Also, transformer109ofFIG. 1is modeled by the combination of leakage inductor203(inductance value LLK), magnetizing inductor204(inductance value LM), and ideal transformer205(turns⁢⁢ratio⁢⁢n=N1N2).
. In the “on” state, semiconductor switch components are assumed to exhibit zero resistance (i.e., they are short circuits). However, the output and junction capacitances of the switches, and the reverse-recovery charge values of the associated rectifiers are modeled with non-zero values.

FIGS. 3(a)–(k) are topological stages of circuit model200ofFIG. 2during a switching cycle. The key waveforms of circuit model200are shown inFIGS. 4(a)–4(k). The reference directions of currents and voltages plotted inFIGS. 4(a)–4(k) are annotated inFIG. 2.FIGS. 4(a) and4(b) show waveforms401and402of drive signals S1and S provided to boost switch103and auxiliary switch108, respectively. According to the present invention, soft-switched boost circuit100operates with overlapping drive signals S and S1. For example, as shown inFIGS. 4(a) and4(b), drive signal S1of auxiliary switch108is turned “on” (i.e., driving signal S1to a voltage that closes auxiliary switch108) at time T0, prior to signal S of switch103being turned “on” between times T3and T4. However, signal S1of switch108is turned “off” (i.e., driving signal S1to a voltage that opens auxiliary switch108) before signal S of boost switch103is turned “off.”

Prior to signal S1of switch108turning “on” at time T0, both boost switch103and auxiliary switch108are open and input current iINflows entirely through boost rectifier104into load106. As shown inFIG. 3(a), after switch108closes at time T0, current i1(waveform405,FIG. 4(e)) flows in primary winding N1of transformer109, thereby inducing current i2in secondary winding N2. InFIG. 3(a), representing the circuit condition between times T0and T1, output voltage VO(i.e., voltage across voltage source201) is impressed across winding N2of ideal transformer205. Consequently, transformer winding voltages v1and v2across the primary and secondary windings of ideal transformer205, respectively, are given by the equations:

σ2=VO,  (1)v1=N1N2⁢VO=nVO(2)
wheren=N1N2<1.
Since voltage v1across the primary winding of ideal transformer205is assumed to be substantially constant, the voltage applied across leakage inductance LLKof transformer109is also accordingly substantially constant, so that current i1(waveform405,FIG. 4(e)) increases linearly with a slope ofⅆi1ⅆt=VO-v1LLK=VO-nVOLLK=(1-n)⁢VOLLK.(3)
At the same time, magnetizing inductor current iM(waveform408,FIG. 4(h)) of transformer109also increases, with a slope given byⅆiMⅆt=VOLM,(4)
so that current iS1(waveform406,FIG. 4(f)) in auxiliary switch108is given by:iS1=i1-i2+iM=i1-N1N2⁢i1+iM=(1-n)·i1+iM(5)
applying the relationship between the primary and secondary currents in ideal transformer205(i.e., N1i1=N2i2), and recognizing that blocking diode113is reversed biased (i.e., open circuit).

As current i1in the primary winding of ideal transformer205linearly increases, current iD(waveform410,FIG. 4(j)) in boost rectifier104decreases at the same rate, as the sum of currents i1and iDequals constant input current IIN, (i.e., i1+iD=IIN), when boost switch103is open. Therefore, current iDin boost rectifier104of circuit100has a turn-off rate given by:ⅆiDⅆt=-(1-n)⁢VOLLK(6)
According to equation (6), the turn-off current rate in boost rectifier104can be controlled in a proper design of transformer109. Specifically, the turn-off current rate of change is determined by leakage inductance LLKand turns ratio n. For today's fast-recovery rectifiers, the turn-off boost rectifier current rate of changeⅆiDⅆt
can be kept around 100 A/μs.

At time T1, boost rectifier current iDfalls to zero. Due to a stored charge in boost rectifier104, boost rectifier current iDcontinues to flow between times T1 and T2in the negative direction (“reverse-recovery current”), as shown inFIGS. 3(b) and4(j). Generally, for a properly selected leakage inductance value LLKfor transformer109and turns ratio n, this reverse-recovery current is substantially reduced, as compared to the reverse-recovery current in a circuit without boost rectifier turn-off current rate control. After the stored charge in boost rectifier104falls to zero at time T2, boost rectifier104regains its voltage blocking capability and the condition of circuit100can be represented by the topological stage ofFIG. 3(c). During this topological stage (i.e., between times T2and T3), junction capacitor302of boost rectifier104(capacitance value CD) is charged and output capacitor301of boost switch103(capacitance COSS) is discharged through a resonance between parallel connection of capacitors301and302and leakage inductor203(inductance LLK). Between times T2and T3, current i1in leakage inductor203and voltage vS(waveform404,FIG. 4(d)) across boost switch103are given, respectively, by:i1=IIN+IRR⁡(PK)+(1-n)⁢VOZC⁢sin⁡(ωR⁢t)(7)
and
σS=VO−(1−n)VO(1−cos(ωRt)),  (8)
where characteristic impedance ZCand resonant angular frequency ωRare defined asZC=LLKCOSS+CD(9)ωR=1LLK⁡(COSS+CD),(10)
and IRR(PK)is the residual reverse-recovery current in boost rectifier104.

Equation (8) shows that the condition for completely discharging output capacitor301of boost switch103at time T3(therefore, allowing zero-voltage closing of boost switch103at time T3) is given by:
νS(t=T3)=VO−(1−n)VO(1−cos π)=0,  (11)
Accordingly, the maximum turns ratio nMAXof transformer109is provided by:
nMAX=0.5  (12)

If turns ratio is less than 0.5, output capacitor301of boost switch103can always discharge to zero regardless of the load and line conditions. Once capacitor301fully discharges at time T3, current i1(waveform405,FIG. 4(e)) continues to flow through antiparallel diode303of boost switch103, as shown inFIG. 3(d). (FIG. 3(d) represents the circuit condition between times T3and T4.) During this time, voltage v1is impressed in the negative direction across leakage inductor203, so that current i1in leakage inductor203linearly decreases at the rate given byⅆi1ⅆt=-nVOLLK,(13)
as illustrated inFIG. 4(e). As a result, current is, (waveform406,FIG. 4(f)) in auxiliary switch108also decreases linearly, while current is of boost switch103(waveform407,FIG. 4(g)) increases linearly from a negative peak. To achieve ZVS of boost switch103, boost switch103closes before its current (i.e., current is) becomes positive at time T4(i.e., when current is begins to flow through the antiparallel diode303of boost switch103).

Boost-switch current is continues to flow through closed boost switch103after current is becomes positive at time T4, as shown inFIGS. 3(e) and4(g). Between times T4and T5, current i1in leakage inductor continues to decrease linearly toward zero, while current is in boost switch103continues to linearly increase at the same rate. When current i1becomes zero at time T5, boost-switch current is reaches IINso that the entire input current IINflows through boost switch103, as shown in the topological stage ofFIG. 3(f), between times T5and T6. At same time, auxiliary switch108, controlled by signal S1, carries only the magnetizing current in magnetizing inductor204. If the magnetizing inductance of transformer109is made high, magnetizing current iM(waveform408,FIG. 4(h)) in magnetizing inductor204can be minimized (i.e. current iMcan be made much smaller than input current IIN), so that auxiliary switch108can open with virtually zero current, at time T6.

When auxiliary switch108opens near ZCS (zero current switching) at time T6, magnetizing current iMstarts charging output capacitor305(capacitance COSS1) of auxiliary switch108, as shown inFIG. 3(g). At time T7, when voltage vS1(waveform403,FIG. 4(c)) across auxiliary switch108reaches clamp voltage VO+VC, where VCis voltage across clamp capacitor112(capacitance CC, represented by voltage source202), magnetizing current iMis commutated into the voltage source202. As shown inFIG. 3(h), between times T7and T8the negative voltage VCacross voltage source202resets the magnetizing current iMin magnetizing inductor204at a rate given by:ⅆiMⅆt=-VCLM,(14)
until magnetizing current iMbecomes zero at time T8.

FIG. 3(i) shows the circuit condition of circuit100after transformer109is reset at time T8until boost switch103opens at time T9and input current IINis commutated from boost switch103to charge boost switch103's output capacitor301. As shown inFIG. 3(j), between times T9and T10, capacitor301charges with constant input current IIN, voltage vS(waveform404,FIG. 4(d)) increases linearly, reaching voltage VOat time T10. At time T10, input current IINis instantaneously commutated to boost rectifier104, as shown inFIG. 3(k). As shown inFIG. 3(k), current IINflows as current iD(waveform410,FIG. 4(j)) boost rectifier104until time T11, when auxiliary switch108is closed again, as in time T0.

In the above description, the junction capacitance of blocking diode110is assumed to have no significant effect on the operation of converter circuit100. In fact, this capacitance plays a role only during a brief interval after current i1reaches zero at time T5. Specifically, after time T5, the junction capacitance of blocking diode110and leakage inductor203resonate, thus creating a small negative current i1in leakage inductor203. If current i1is greater than magnetizing current iMin magnetizing inductor204, current iS1of auxiliary switch108flows in the negative direction through an antiparallel diode of auxiliary switch108. Because of this conduction in the antiparallel diode, voltage vS1of auxiliary switch108(i.e., voltage waveform403,FIG. 4(c)) does not immediately increase after auxiliary switch108opens at time T6(i.e., shortly after current iS1in switch108reaches zero). As a result, the rise of voltage vS1across auxiliary switch108occurs after a brief delay—i.e., after the current is, through the antiparallel diode of auxiliary switch108resonates back to zero. This delay has no significant effect on the operation or the performance of circuit100. However, if current i1in leakage inductor203is smaller than magnetizing current iM, the rise of voltage vS1(waveform403,FIG. 4(c)) across auxiliary switch108occurs immediately after time T6.

In summary, circuit100of the present invention allows soft-switching of all semiconductor devices. Specifically, boost switch103is closes under ZVS condition, auxiliary switch108opens under ZCS condition, and current ID of boost diode104is turned off at a controlled rate. As a result, the turn-on switching loss of boost switch103, the turn-off switching loss of auxiliary switch108, and reverse-recovery-related losses of boost rectifier104are eliminated, thereby minimizing overall switching losses and maximizing conversion efficiency. In addition, soft-switching provides beneficial effect on electromagnetic interference (EMI) that may result in a reduced size requirement for an input filter.

Because of ZVS in boost switch103, a MOSFET (Metal Oxide Semiconductor Field Effect Transistor) device, or a parallel combination of MOSFET devices, can implement boost switch103of circuit100. Similarly, due to the ZCS of auxiliary switch108, either an IGBT (Insulated Gate Bipolar Transistor) or a MOSFET can implement auxiliary switch108without a performance penalty. In fact, an IGBT boost switch can also implement boost switch103, provided that a turn-off snubber capacitor is connected across the IGBT to reduce the turn-off loss due to IGBT's current-tail effect. In such an implementation, boost switch103should close with ZVS, so that the snubber capacitor does not contribute to the turn-on switching loss. Also, in such an implementation, the IGBT is preferably provided a co-packaged antiparallel diode, or an external diode.

In circuit100, the voltage and current stresses on boost switch103and boost rectifier104are substantially the same as the corresponding stresses in a conventional boost converter without a snubber. The voltage stress on auxiliary switch108is given by:
VS1(MAX)=VO+VC,  (15)
while the current stress on auxiliary switch108, neglecting residual reverse-recovery current IRR(PK)(waveform410,FIG. 4(j)) isiS1⁡(MAX)≅(1-n)⁡[IIN+(1-n)⁢VOZL].(16)
Voltage vS1(max)and current iS1(max)are shown in waveforms403and406ofFIGS. 4(c) and (f).

According to Equation (15), the voltage stress of auxiliary switch108is controlled by the selection of clamp voltage VC, which is generally determined by the energy stored in magnetizing inductor204, while auxiliary switch108is conducting, and the resistive value RCof clamp resistor111. If the capacitance value CCof capacitor112is selected to be large enough, so that the voltage ripple across output capacitor112is much smaller than the average voltage value of capacitor112, voltage VCcan be calculated from12⁢LM⁡(VOLM⁢DS1⁢TS)2⁢fS=VC2RC,(17)
where DS1is duty cycle of auxiliary switch108, TSis the switching period, and FS=1/TSis the switching frequency.

From Equation (17),VC=RC2⁢fS⁢LM·(DS1⁢VO)(18)
the voltage value VCcan be minimized by maximizing inductance value LMin magnetizing inductor204, so that the power loss in the clamp circuit (i.e., the power dissipation in clamp resistor111) is also minimized. Typically, for a properly designed transformer, the clamp-circuit loss is negligible compared to the output power, so that conversion efficiency is practically unaffected.

The inductance of leakage inductor203of transformer109is determined from the desired turn-off rate of the boost rectifier current defined in Equation (6), i.e.,LLK=(1-n)⁢VOdiD/dt.(19)

According to Equation (19), to minimize the inductance value LLKof leakage inductor203, one can increase the turns ratio n of transformer109. Since nMAXis 0.5, the turns ratio of transformer109should not be much less than 0.5. A value of n in the 0.3–0.5 range is desirable. If VO is400 V, n is 0.5, and diD/dt is 100 A/μs, leakage inductance value LLKwould be 2 μH. If inductance value LLKis too large to be achieved by the leakage inductance of a transformer, an external snubber inductor501can be used to adjust the desired circuit inductance, such as shown in circuit500ofFIG. 5. The operation of circuit500inFIG. 5is substantially the same as that of circuit100ofFIG. 1.

According to the present invention, a voltage clamp circuit (e.g., voltage clamp circuit107) in an active snubber that is used to reset the magnetizing inductor (e.g., magnetizing inductor204) of the transformer can be implemented in numerous ways. For example, the voltage clamp circuit can be connected to any DC potential in the circuit. InFIGS. 1 and 5, voltage-clamp circuits107and502are each connected to the output terminals of the converter.FIGS. 6 and 7show circuits600and700having voltage clamp circuits602and702that are connected to the negative rail and the input source, respectively. Furthermore, the voltage-clamp circuit can also be fitted across the primary winding of transformer109, as illustrated inFIG. 8.FIG. 9shows circuit900with voltage clamp circuit902, according to another embodiment of the present invention. Many other variations of the voltage-clamp circuit are also possible.

A soft-switched boost converter of this invention can also be implemented using an integrated isolated auxiliary power supply, such as shown in circuit1000ofFIG. 10. In circuit1000, an isolated auxiliary output voltage VAUXis provided by three-winding transformer1009, active-snubber switch108, windings N2and N3, rectifier1013(DAUX), and filter capacitor1012(having capacitance value CAUX), forming flyback converter1002with input terminals connected across the output terminals of boost converter1000. Assuming a discontinuous conduction mode (DCM) of operation of transformer1009, auxiliary output voltage VAUXis given by:VAUX=RAUX2⁢fS⁢LM·(DS1⁢VO),(20)
where RAUXis the load on the auxiliary output.

Since output voltage VOof a regulated boost converter is constant, with the duty cycle DS1of auxiliary switch108being constant, auxiliary voltage VAUXin snubber1002changes only if the load is variable (i.e., if load resistance RAUXchanges). For a variable auxiliary load RAUX, auxiliary voltage VAUXcan be maintained constant by appropriate modulation of duty cycle DS1.

For example,FIG. 11shows, conceptually, circuit1100providing a close-loop control of auxiliary output voltage VAUXfor circuit1000ofFIG. 10. Of course, many other ways of closed-loop modulating duty cycle DS1are possible. InFIG. 11, two independent feedback-control loops are provided. Specifically, output voltage VOis regulated by modulating duty cycle D of boost switch103, while auxiliary-output voltage VAUXis regulated by modulating the duty cycle DS1of auxiliary switch108. To maintain proper timing of drive signals S and S1for boost switch103and auxiliary switch108(i.e., to ensure drive signal S1is asserted before drive signal S for a predetermined time interval), the rising edge of drive signal S1is generated by controller1117from the control loop that regulates output voltage VO. The turning-off of drive signal S1is controlled by the loop that controls auxiliary output voltage VAUXwhich generates the falling edge signal.

Soft-switched boot converter with integrated isolated power supply can also provide multiple outputs. In addition, the active snubber according to the present invention can be applied to boost converters used in single-phase and three-phase AC/DC applications such as, for example, single-phase and three-phase power-factor correction circuits.FIG. 12shows single-phase AC/DC boost converter1200that is integrated with the full-wave rectifier, in accordance with one embodiment of present invention. In the circuit inFIG. 12, during positive half cycles, boost switch103aand boost rectifier104aoperate in the manner described above for boost switch103and boost rectifier104of circuit100ofFIG. 1, respectively, in conjunction withFIGS. 3(a)–3(k) and4(a)–4(k). Similarly, during negative half cycles, boost switch103band boost rectifier104boperate in the manner described above for boost switch103and rectifier104of circuit100ofFIG. 1. Because of its rectifier configuration, which has one rectifier less than the conventional configuration of a full-wave bridge rectifier followed by a boost power stage, AC/DC boost converter1200has reduced conduction loss relative to such a conventional configuration. InFIG. 12, active snubber1207includes rectifiers110aand10bon the primary side of transformer109, as AC/DC boost converter1200has boost switches103aand103band boost rectifiers104aand104bconfigured to operate as two boost switch-boost rectifier pairs that do not operate simultaneously. In AC/DC boost converter1200, rectifiers110aand110bare connected to the same primary winding of transformer109, as illustrated inFIG. 12.

Many variations of AC/DC boost converter1200are possible, such as AC/DC boost converters1300and1400ofFIGS. 13 and 14. In AC/DC boost converters1300and1400ofFIGS. 13 and 14, rectifiers104aand104bare replaced by switches103cand103d, respectively, to further reduce the conduction loss. Furthermore, AC/DC boost converter1400ofFIG. 14integrates an auxiliary power supply with a main converter.

Three-phase AC/DC boost converters1500,1600and1700according to the present invention are shown inFIGS. 15–17.FIG. 15shows three-phase AC/DC boost converter1500with DC-rail rectifier104.FIGS. 16 and 17show three-phase AC/DC boost converter1600and1700, without a DC-rail diode and with an integrated auxiliary power output (voltage VAUX), respectively. In3-phaseAC/DC boost converters1600and1700, diodes110a,110band10cconnect with the primary side of an active snubber transformer (i.e., transformer109or1009) to achieve ZVS and to minimize reverse-recovery charges of three pairs of boost switches and boost rectifiers, respectively.

The active snubber of the present invention can be applied to any isolated or non-isolated DC/DC converter, or any single- or three-phase AC/DC converter. For example,FIGS. 18 and 19show, respectively, buck converters1800and buck-boost converter1900, each having an active snubber.FIGS. 20,21and22show, respectively, forward converter2000, flyback converter2100, and two-inductor boost converter2200each having an active snubber, in accordance with the present invention.

FIG. 23shows soft-switched boost converter2300, which differs from soft-switched boost converter1000ofFIG. 10by providing an integrated auxiliary isolated power supply with auxiliary switch2308, which is controlled by control signal SAUXprovided by two-loop control circuit2301. Control circuit2301provides two independent feedback control loops to regulate both output voltage VOand voltage VAUXin the auxiliary power supply. InFIG. 23, output voltage VOis regulated by modulated duty cycle D of boost switch103, and auxiliary output voltage VAUXis regulated by modulating the duty cycle of auxiliary switch2308.

The above detailed description is provided to illustrate specific embodiments of the present invention and is not intended to be limiting the scope of the present invention. Numerous variations and modifications of the present invention are possible. The present invention is set forth in the following claims.