Ionospheric sounding

An ionospheric sounding system for frequency management of a HF communications system operates between a transmitter and a remote receiver and makes use of code-modulated narrow band sounding pulses transmitted at frequencies throughout the HF band. The code has an impulsive auto-correlation function and the frequency selection is pseudo-random. The code may be a two-part complementary code. Alternatively the code may be a selected one of a family of codes possessing high auto-correlation and low cross-correlation properties, thereby enabling communications management information to be conveyed by the choice of code. Measurements of signal and noise are made at the receiver for each transmitted frequency to assist establishing a HF communications link. Synchronous detection of the received signal is used, employing correlators (103, 104) in phase quadrature.

The invention relates to an ionospheric sounding unit for providing 
information to improve HF communications utilising long range sky-wave 
paths. 
Communicating over HF radio circuits particularly long range sky-wave paths 
has always been recognized as a sporadic and a generally unrealiable 
method for sending messages. 
This is due, to a large extent, to the naturally sporadic and transitory 
nature of the propagating medium being used which is in the main caused by 
the variability in the free electron density of the ionosphere. Also to be 
considered is the intensity and density of interference from other HF band 
users. 
These factors have alone given rise to a situation where on-demand 
communications cannot normally be expected even for relatively short-haul 
circuits. Neither can high quality, low error rate circuits be acquired or 
maintained without regular changes in operating frequency and/or without 
using very high power radio transmitters in conjunction with high gain 
directional antennas. 
For the Royal Naval warship this situation is even worse. Here, because of 
the limited size of the platform, high transmitter powers are 
impracticable and large HF directional antenna arrays are impossible. 
Coupled with this are the intractable problems of working in a hostile 
environment where it is normally operationally necessary to simultaneously 
transmit and/or receive over a very broad spectrum of frequencies. These 
may include ELF, VLF, LF, MF, HF, V/UHF and also much higher X band radar 
frequencies. 
It is therefore very easy to appreciate why it is that HF communications 
have been historically losing popularity to the more reliable and robust 
satellite communications networks for long distance circuits. 
Nevertheless from the military point of view the vulnerability of these 
satellite systems, including physical attack, will always dictate the 
necessity for having an operational requirement for HF. 
An efficient and effective high quality HF back-up system is therefore an 
important capability for the RN during any hostilities and it has the 
added advantage that it can always be entirely under our national control. 
This is currently not the case for satellite systems. 
It is unfortunate that although major advances have recently been made 
towards increasing the sophistication, complexity and reliability of most 
HF communication systems and equipments (principally through the 
exploitation of large scale electronic circuits and in particular 
microprocessors) little has been done to overcome or reduce the three 
principal HF propagation characteristics which are responsible for the 
degradation of transmitted signals. 
These are multipath or dispersion, interference and fading. 
For a mobile platform on the land, sea or in the air, the situation is 
complicated by the constraints imposed in fitting HF systems within the 
restricted space available, particularly with regard to antennas. 
Transmitter powers for these mobile systems will also be constrained 
because of the limited space available and the inordinate cost and 
technical complexity of fitting radio transmitters with rf powers much 
higher than a few kilowatts. The typical power for an HF circuit on an RN 
warship is usually limited to 1 kW peak and mean per radio circuit. The 
problems that arise when trying to simultaneously provide up to ten or 
perhaps more transmitting and receiving channels would be insurmountable 
if greater transmitter powers were to be used. Common Antenna Working 
(CAW), intermodulation products, reciprocal mixing (in the receivers) and 
wide-band transmitter noise floors would all seriously degrade the 
operational performance of HF radio circuits, particularly at the lower 
received signal levels which are necessary for long range circuits. 
Intermodulation frequency products caused by the non-linearities in the 
transmitters and by the ships superstructure (rusty bolt effect) can 
however be avoided by judicious frequency planning although this will 
inevitably reduce the available spectrum that can be used. Moreover this 
can only be accomplished if narrow-band systems are used and provided the 
higher order products can be ignored. If transmitter powers were to be 
increased these higher order products will become progressively more 
significant and will have to be considered in any subsequent frequency 
planning arrangements. 
When wide-band systems are to be used the constraints that have to be 
placed on the frequencies for transmission and reception will become even 
more restrictive. This factor alone will seriously curtail any future 
policies for implementing frequency agile or frequency hopping and 
wide-band transmission systems. Furthermore, antennas used for reception 
will unavoidably have to be placed in close proximity to transmitting 
antennas and the ships superstructure. Variations in the radiation pattern 
in the azimuthal and elevation planes will be found as a direct result of 
this, producing nulls of up to 30 dB or more in any of these antennas. 
These radiation distortions will alter with changes in frequency and with 
the attitude of the ship, i.e. pitch, roll and yaw. 
All these factors, as well as the general unpredictability of HF sky-wave 
propagation, pose a problem of numerous variables having a multiplicity of 
possible permutations. 
Significant improvements in the realistic use of sky-wave propagation in a 
HF communications system can only be achieved if the user is given 
Real-Time data regarding all the relevant path characteristics. 
The alternative to this is to rely upon long term predictions based on 
computer programmed models, Bluedeck etc, which can at any instant be 
grossly in error particularly for circuits operating into or through the 
higher latitudes. 
Over the last decade or so many attempts have been made to use the HF band 
in a more systematic manner by employing various back-scatter and oblique 
incidence ionospheric sounding techniques. Unfortunately these systems 
invariably used transmitters having output powers of 30 or 100 kW and 
sometimes even more. Furthermore these transmitters were nearly always 
connected to antennas having broad-band but more importantly directional 
gain properties. With antenna gains of between 10 and 15 dBs the radiated 
powers (ERP) of these early sounders was often 1 MW or more. Using this 
amount of pulsed rf power inevitably caused considerable interference to 
other HF band users so to avoid jamming a major part of the spectrum these 
sounders were necessarily restricted in the number of frequencies which 
could be used at any one time and also in the period for which they could 
actually be turned on. The pulsed signals were received by a time 
synchronised receiver and an ionogram plot was produced for each frequency 
scan. Because the pulse widths of these signals were perhaps a millisecond 
or so wide the resolution obtained was unavoidably small but later Barker 
sequence coding was used to improve this to 100 .mu.s without any loss in 
sensitivity. Code lengths of 11 bits or sometimes more were used to phase 
modulate the transmitter carrier signal at 10 kb/s. This produced a signal 
processing gain, on reception, of about 12 dB or more but because of 
certain implementation losses this was often reduced in practice to 
perhaps 6 dB or so. 
The object of the present invention is to improve the performance of HF 
communications by means of judicious exploitation of the HF spectrum by 
real-time channel sounding. Thus the object is to provide a low power 
ionospheric sounder which will permit a real-time assessment of the 
quality of different frequency bands in the HF spectrum thereby enabling 
improved communications links to be established using an optomised choise 
of frequencies. 
The invention consists of an ionospheric sounding system for providing 
frequency management information for high frequency (HF) communications 
comprising: 
a. a HF radio transmitter having: 
(i) frequency selection means operating such that pulses of energy can be 
transmitted at respective frequencies pseudo-randomly selected from the HF 
range of radio frequencies; and 
(ii) modulation means to modulate each transmitted pulse with a code having 
an impulsive autocorrelation function; and 
b. a remote HF radio receiver having: 
(i) frequency selection means capable of being programmed to sensitise the 
receiver to the transmitted sequence of pseudo-random frequencies; and 
(ii) correlation means for correlating the received signal during each 
pulse interval with a replica of the transmitted code and producing an 
output signal indicating detection of a transmitted signal. 
The term "impulsive autocorrelation function" applied herein to the code 
modulating the transmitted signal is a noise-like function with no 
recognizable pattern. When a signal, modulated by such a code, is 
correlated against itself, the correlation function is zero or near zero 
everywhere except when the modulated signal is in exact register with its 
replica at which time the correlation is high. Preferably the code is a 
complementary code having two parts whose separate autocorrelation 
functions add to produce an impulsive function. Preferably also double 
side band amplitude modulation (dsb AM) is used. Advantageously the 
transmitter is provided with a time code modem and a modulation control, 
the modulation control selectively connecting the sounding code or the 
time code to the modulation means. The receiver is provided with a similar 
time code modem to provide time of day information and facilitate 
synchronising of the receiver with the received signal. In an advantageous 
arrangement the two parts of the complementary code are separated by 
greater than about 10 ms to prevent corruption of the second received part 
by echo signals of the first received part. 
As an alternative to using a single sounding code the code may be a 
selected one of a pleurality of codes with the receiver being provided 
with a means to receive and distinguish each code thereby enabling 
information to be transmitted. 
Advantageously synchronous signal detection is employed including means to 
compensate for frequency and phase variations in the received carrier 
frequency. This is achieved by connecting the received signal to sin and 
cos product detectors, and also connecting a local frequency signal to the 
product detectors, the local frequency being derived from the transmitted 
carrier signal by filtering out the dsb modulation from the received 
signal. The outputs from the product detectors are connected to respective 
real and imaginary cross-correlators and the outputs from the 
cross-correlators are then combined to produce the phase-insensitive 
modulus impulse response. 
Preferably the receiver has an automatic gain control (AGC) and there is 
provided means to measure the AGC level during each received pulse. 
Advantageously there may be provided means to modify the measured receiver 
AGC level such that the measured peak impulse response level can be 
calibrated. Also it is advantageous to include means to measure the mean 
noise level in the receiver prior to a measurement interval or window when 
the peak impulse response is measured. A comparator may be included such 
that an output signal is generated whenever the measured peak exceeds the 
mean noise level by a predetermined amount. The receiver may be provided 
with means to record the received signal strength measured in dB, the 
measured noise level and the mode structure (impulse response) for each 
transmitted frequency.

Choosing the best frequency for transmission is important for two well 
known basic reasons: 
The first would be that a frequency chosen more or less at random will not 
necessarily be able to propagate via the ionosphere to a distant receiver 
and certainly not for 100 percent of the frequencies tried. Frequency 
predictions and planned optimum working frequencies used in conjunction 
with short-term predictions derived from limited ionospheric soundings do 
help but the variability of the ionosphere is generally so great and so 
rapid that this method of frequency selection can still only be used as a 
guide. Real-Time channel sounding on the other hand should provide a 
relevant solution to this particular problem. 
The second reason for wanting to operate on an optimal working frequency is 
that even if a randomly chosen frequency can propagate through the 
ionosphere it will more than likely be totally corrupted by radio 
interference from other HF users when it arrives at the remote receiver. 
Choosing the best or optimal working frequency therefore means finding a 
frequency or frequencies that will not only propagate over long-distant 
sky-wave paths but will also have relatively low levels of in-band 
interference at the remote radio receiver. 
The basic model of a HF sounding system is shown in FIG. 1. This shows two 
sky-waves S.sub.1 and S.sub.2 propagating between a sounder transmitter 
101 and a remote sounder receiver 102. Sky-wave S.sub.1 reaches the 
receiver after one reflection in the ionosphere while the multipath 
sky-wave S.sub.2 is reflected twice in the ionosphere and once at the 
ground. The actual number and type of signals received will depend 
primarily upon: 
a. the height and density of the reflecting ionospheric layers; 
b. the frequency of the transmitted signal; and 
c. the distance between the transmitter and the receiver. 
In practice the difference in propagation times .DELTA.T for these two 
sky-waves will be shorter than the duration of the two signals S.sub.1 and 
S.sub.2 so that there will be substantial overlap, and not as shown in the 
illustrative receiver response 103. 
In order that the sounder may be operated alongside other equipment the 
transmitter sounder 101 has a low power amplifier output connected to a 
transmitter aerial 104. The transmitted signals are received by a 10 m 
whip aerial 105 connected to the remote sounder receiver 102. 
FIG. 2 is a block diagram of the sounder transmitter. A transmit sounder 
modem 201 produces random frequency and code modulation drives 202 and 203 
respectively for a RF synthesiser 204. After power amplification in a 1 
kW(peak) broad-band amplifier 205 the synthesiser signal is fed to a 
log-periodic directional antenna 206. The transmit frequencies are 
selected randomly by the modem 201 from the whole HF band from 3 to 30 
MHz. Code modulation of each transmitted frequency is then used such that 
high receiver processing gain can be achieved to enable signals to be 
received without need for high transmitted power. The modulation of the 
output of the synthesiser 204 is controlled by a modulation control 
interface 207. Connected to respective inputs 208-210 of the modulation 
control interface 207 is the code modulation drive 203, a time code modem 
211 and a framing 1 kHz tone generator 212. The modulation control 
interface 207 selectably connects the sounder, the time signal or the 
framing 1 kHz tone to the synthesiser 204 where the frequency hopping 
output is 90% double side band amplitude modulated (dsb AM). As can be 
seen from FIGS. 3A and 3B the sounder transmitter signal hops between 
pseudo-random frequencies, 301-306 chosen from 550 discrete frequencies 
distributed between 3.8 MHz and 30 MHz at ten frequency hops per second. 
At any one selected frequency when transmitting the sounder code the 100 
ms transit frame 307 includes a first settling interval 308 of 20 ms to 
allow sufficient time after frequency changing for the synthesisers at 
both ends of the sounder link to stabilise and also to allow the receiver 
automatic gain control (AGC) to settle. To avoid spectral splatter the 
synthesiser output signal is attenuated using raised cosine shaping 309 
just prior to and also after changing frequency. The modulating code 
consists of two parts, Code A and Code B, separated by a 12.8 ms interval 
310 and transmitted at 10 kb/s. The two halves of the modulating code form 
a special complementary coded sequence 512 bits long (256+256 bits). 
Coded sounding pulses are pulse compression signals formed from a unique 
sequence of binary data. The binary sequence is designed to have an 
impulsive auto-correlation function. Barker codes are one particular type 
of digital sequence but these can only be produced for relatively short 
sequences. An example of a complementary code sequence is shown by the 
signal 311 in FIG. 3C, the first half of the signal being code A and the 
second half being code B. The trace 312 shows the auto-correlation 
functions of code A (.phi.AA) together with code B (.phi.BB). This 
demonstrated that the auto-correlations .phi.AA and .phi.BB produce a poor 
peak (313) to side-lobe (314) ratio, however the side lobes e.g. 314 and 
315 produced by these two codes are the exact inverse or complement of 
each other. The peaks 313 and 316 are of the same amplitude and polarity. 
Thus when these two code auto-correlation functions are added together the 
side lobes will cancel but the peaks will add. The resulting function 
.phi.XX (317) for the complete sequence will have an infinite peak to side 
lobe ratio. It can be shown that the impulse response of a system under 
test(the mode structure of the radio path between the transmitter and the 
receiver) can be completely determined using a pseudo-random digital code 
sequence providing the auto-correlation function of the sequence is 
impulsive and its spectral response is greater than the bandwidth of the 
system under test. For a complementary code digital signal the bit rate 
must be made high enough to produce a frequency response which is 
reasonably flat over the bandwidth concerned. The length of the code is 
particularly important because a long sequence will produce a greater 
processing gain on reception. This improves the detectability of the 
signal in poor signal to noise conditions and also the selectivity in 
rejecting any other input signals. Further information on complementary 
codes is published in "Complementary Series" by M J E Golay, in IRE 
Transactions on Information Theory, Vol 17-7 pp 82-87, April 1961. 
A 512 bit complementary code was adopted giving a processing gain of 27 dB. 
This gives a maximum resolution of about 100 .mu.s at a detection 
bandwidth of 10 KHz or less. 
Referring again to FIG. 2, the time clock modem 211 is described in UK 
patent application No 8127713. It is controlled by a 1 MHz reference 
timing clock and produces a pseudo-random binary coded sequence which 
uniquely encodes the time of day for transmission and enables a similar 
modem in the remote receiver to achieve timing synchronism within a very 
small timing error. Accurate timing and stability is essential to achieve 
and then maintain Synchronism between transmitter and receiver. 
Although it is desirable to know the time of day to within .+-.5 ms or 
better, in order to guarantee instant synchronism, a significantly larger 
timing error could be easily accommodated because the receiver timing 
could be manually advanced or retarded to obtain perfect synchronism after 
starting up. Initial timing errors of up to plus or minus one or two 
seconds could in practice be coped with without too much difficulty 
provided there were a reasonable number of propagating frequencies. When 
however the number of operational frequencies becomes too small, through 
poor propagation and interference, this method of achieving 
synchronisation becomes progressively more difficult and time consuming to 
reasonably undertake. 
As shown in FIG. 4 a standard radio receiver 401 operates with an external 
local oscillator drive from a synthesiser unit 402. A 10 m whip antenna 
403 connects the signal to the receiver 401. The output from the receiver 
401 carries a 100 KHz intermediate frequency signal to a system control 
interface unit 404 where pseudo-synchronous complex amplitude demodulation 
occurs. This produces the appropriate in-phase (real) and quadrature phase 
(imaginary) detection components for the two cross-correlation units 405, 
406. The circuit producing the separate detection components is described 
later with reference to FIG. 5. The digitised modulus (see FIG. 6) of the 
outputs from the cross-correlators 405 and 406 is fed to a central 
analysis and recording computer 407. After some preliminary analysis this 
data along with time and frequency markers are recorded by a digital tape 
recorder 408. The data can be simultaneously displayed on a printer 407 
and can provide output signals for a video display 410 of ionograms and 
impulse responses for recording by a 16 mm camera 411. The receiver 
includes a time clock modem 412 operated by a 1 MHz reference source to 
achieve synchronism with the transmitter and for accurate time control by 
the system control interface 404 of the synthesisor 402 and the 
cross-correlators 405 and 406. A tape reader 413 is provided for program 
control of the computer 407. 
The demodulation of the signal is done using the synchronous signal 
detection circuit shown in FIG. 5. The 100 kHz IF signal from the output 
of the receiver 401 is connected to the input 501 of the detection 
circuit. The input 501 is connected via an amplifier 502 to two product 
detectors 503 and 504 where the 100 kHz IF signal is mixed in phase and in 
phase quadrature with a 100 kHz signal 505 derived from the If carrier 
signal. The insertion signal 505 is obtained by connecting the 100 kHz If 
signal via a second amplifier 506 to a band pass (BP) filter 507 which 
filters out the dsb AM received sounding signal as well as most of the 
interference. The BP filter 507 has a centre frequency of 100 kHz and a 
band-width of .+-.50 Hz. The filtered carrier signal is limited (508) and 
after further filtering (not shown) is connected to a phase splitter 509. 
The signals at the phase splitter outputs, in phase quadrature, are 
connected to the respective mixers 503 and 504. 
The product terms in the outputs from the mixers 503 and 504 are then low 
pass filtered (510, 511) to give two phase-quadratured audio outputs 512 
and 513. 
The sounding signal uses conventional double-sideband amplitude modulation. 
These signals are usually demodulated in a radio receiver using envelope 
or diode type detectors. Unfortunately although these methods are very 
simple they only work properly when the signal-to-noise ratio (S/N) is 
good. To obtain proper demodulation in lower S/N it is important to use 
synchronous detection. If the frequency stability of the system and the 
sky-wave radio path was good enough then synchronous detection could be 
easily done by re-inserting the known carrier frequency in a product 
detector. True synchronous detection cannot be used in this application 
however because the end to end frequency stability of the complete system 
cannot be guaranteed to be less than 4 Hz at all times. This degree of 
stability is necessary to ensure the accurate operation of the 
cross-correlators. The synchronous signal detection circuit shown in FIG. 
5 has a fast response time, normally taking less than 20 ms to correct the 
phase and frequency of the mixer signals. It can also cope with fading 
signals provided these are not too deep or the S/N is not too poor. 
A cross correlation circuit shown in FIG. 6 is provided to receive the 
baseband audio signal from each of the outputs 512, 513 of the detection 
circuit (FIG. 5). The signal at input 601 is sampled by an 8-bit A/D 
converter 602 controlled by a signal from a 40 kHz clock 603. The 
digitised signal is clocked into a special shift register 604 comprising 
three series-connected shift register stores 605-607. The central shift 
register 606 acts as a time delay of about 12.8 ms equal to the time 
separation of the two parts A and B of the complementary coded transmitted 
signal (FIG. 3). The stores 605 and 607 are of such capacity that when 
code A is completely stored in the register 607, code B is stored in the 
first register 605. The stores 605 and 607 are recirculating stores such 
that the separate parts A and B can be entirely recirculated, in 
synchronism, in the respective stores 607 and 605 between successive 40 
kHz Clock pulses. Parts A and B of the correlation process are stored in 
the receiver memory stores 608 and 609. The output from the shift register 
code B store 605 and the code B memory store 609 are connected to inputs 
of a first 8-bit multiplier 610 and similarly the stores 607 and 608 for 
the transmitted code A and the stored code A are connected to a second 
multiplier 611. The contents of both A and B shift registers are 
multiplied by the appropriate code sequence after every shift register 
clock pulse. These two product streams are then summed in respective 
summers 612, 613 over the code length and then added together in adder 614 
before digital to analogue conversion in a D/A converter 615 and low pass 
filtering in a LPF 616. By this means the two halves of the code are 
cross-correlated separately and then added to produce the desired output. 
The correlation peak is obtained when the first bit of the first half of 
the code is in the last stage of the code A shift register 607. The first 
bit of the second half of the code will then be in the last stage of the 
code B shift register 605. 
Although the delay between the two parts of the code has been set at 12.8 
ms it could be varied. It is important however that it should exceed the 
maximum path dispersion. This has been found to be typically 10 ms or 
less. If the delay were less than this maximum path dispersion then the 
output signal 617 from the cross-correlator would be corrupted by echo 
signals of code A arriving at the receiver at the same time as Code B. 
This would produce "ghost" peaks in the output impulse response. 
FIG. 7 illustrates the timing requirements of the receiver. The transmitted 
signal pulse 701 occupies a 100 ms time frame 702 and the receiver 
frequency changes 703, 704 take place every 100 ms in synchronism with the 
frequency changes in the received pulses. The correlator output signal is 
sampled for the presence of a transmitted signal during a 10 ms window 705 
centred on the time 706 when the last stage of the code B register should 
be filled by the first bit of the second part of the coded transmission. 
The window 705 is selected to be less than the guard interval between the 
two halves A and B of the code. The peak signal is measured in this window 
705 as described below. Immediately prior to the signal window 705 the 
mean noise level is measured over the time interval indicated by the line 
707. From the ratio of these two quantities the signal-to-noise can be 
computed for that particular frequency. The gain of the receiver is 
continually monitored by measuring the radio receiver automatic gain 
control (AGC) level in each 100 ms pulse frame. The measurement is made a 
short time after the frequency change 704 to allow settling of the 
receiver AGC. 
The processing of the signal output from the cross-correlators 405, 406 is 
shown in more detail in FIG. 8. The baseband in-phase signal at the output 
512 and the quadrature signal at the output 513 from the synchronous 
signal detector (FIG. 5) are connected to the imputs 801, 802 of the real 
and imaginary cross-correlators 803, 804. The output signals from the 
correlators representing the complex impulse response are connected to a 
unit 805 which derives the modulus of the output signals. The output 
signals are connected to respective squaring circuits 806, 807 and the 
outputs from these circuits are combined in an adder 808 and the square 
root taken from the sum in the circuit 809. The impulse response signal at 
the output 810 is sampled during the window period 705 by a peak detector 
811 and this peak is then converted into a binary number level signal 
measured in dBs in a level circuit 812. The AGC level of the receiver 401 
(FIG. 4) is modified by adjustment of the origin and slope of its response 
curve. The receiver AGC signal, measured as shown in FIG. 7, is connected 
to a first input of a difference amplifier 813, the second input voltage 
being adjustable so as to adjust the zero level for the amplifier. The 
output from the origin adjusting amplifier 813 is connected to a first 
input of a second difference amplifier 814 with adjustable feedback to the 
second input so as to adjust the slope of the AGC signal. The adjusted AGC 
signal output from the amplifier 814 is digitised by a 7-bit A/D converter 
815. The adjustments are made such that the digital output from the A/D 
converter 815 is a dB ratio referred to 1 .mu.V. This digital output is 
then added in a binary adder 816 to the measured peak impulse response 
level (also converted to dBs) to produce the desired output. 
It is necessary to measure the sounding signal level using this particular 
method because readings will sometimes be required when the input 
signal-to-noise to the system is low. 
The processing gain of the receiving system will produce an output S/N 
which is considerably better than the input to it. But the actual output 
signal level (the peak level of the impulse response) will depend upon the 
input S/N to the radio receiver. Meanwhile, the receiver AGC level 
represents the total input signal to the radio which will normally include 
the sounding signal and/or any noise. This AGC however is used to ensure 
that the audio output level from the receiver will be constant, 
irrespective of the input signal-to-noise ratio. It therefore follows that 
the signal (sounding) level to and out of the cross-correlators will fall 
as the input signal-to-noise ratio to the radio receiver falls. 
For input S/N ratios greater than 10 dB, the signal to the correlators will 
be constant because the AGC is `captured` by the sounding signal. Below 10 
dB the AGC is affected by signal and noise. At 0 dB the receiver AGC is 
operating on equal levels of noise and signal so the signal output level 
will be 3 dB lower. At lower input S/N ratios the signal output level will 
be proportionally lower. 
Correct recognition of the sounding signals can be accomplished using the 
circuit shown in FIG. 9. The modulus of the impulse response signal at the 
output 810 from the received signal level detector (FIG. 8) is connected 
to the input 901 to a multiplexer 902a at the imput of a recirculating 
shift register store 902 clocked at 10 kHz. The impulse response at the 
input 901 is also connected to a peak detector 903 which measures the peak 
signal in the time window 705 as did the peak detector 811 in the FIG. 8 
arrangement. The output from the recirculating store 902 is connected to a 
calculation unit 904 which computes the mean value of noise in the period 
707 (FIG. 7) prior to the peak measurement window 75. The outputs from the 
noise calculation unit and the peak signal detector 903 are 7 bit binary 
numbers which are compared in a comparator 905 arranged to produce an 
output binary "1" if S/N.gtoreq.8 dB and a "0" otherwise. 
Thus for each pulse there will be produced a status "1" or a status "0" 
depending upon whether that frequency is providing a usable sounding 
channel. At a transmission rate of 10 pseudo-random frequency hops per 
second there will be produced a 10 bps stream of status bits together with 
signal level and impulse response characteristics relayed back to a 
communications receiver/transmitter co-located with the sounder 
transmitter to enable the communications transmitter to optimise the 
frequencies selected for communicating over the sounding path. Scattering 
functions for each channel can be derived from the received sounding 
pulses and characteristics for perhaps several hundred frequencies can be 
accumulated in a minute or so. These characteristics can then be updated 
at the scanning rate. The received signal strength (dB .mu.V) and mode 
structure for every tested frequency can be stored to await subsequent 
interrogation and processing to perhaps decide which frequency or 
frequencies have the highest probability of propagating a transmitted 
signal with minimum distortion and with the highest received 
signal-to-noise ratio. 
Trials have shown that it would be necessary on occasions to change 
frequency up to 200 times in a day in order to maintain a communications 
circuit having the highest possible received signal strength. In practice 
this may not be required however because a good signal to noise ratio is 
far more important than having the strongest signal. This may considerably 
reduce the number of frequency changes required. 
The early experimental work has been sufficiently encouraging to conceive 
of the adoption of facilities to enable a more exhaustive evaluation of an 
operational HF communications management system. It is possible to use a 
more sophisticated coding structure for the sounding pulses so as to allow 
for the simultaneous transmission of data over the broadcast sounding net. 
These sounding pulses will therefore not only provide the mobile operator 
with the characteristics of the radio path but also provide information 
about interference levels and channel availability at the base station. In 
the simplest example, two semi-orthogonal binary codes may be used to 
represent replectively a data "1" or a data "0". In an extention of this 
basic idea the inventor has discovered a 512 set of semi-orthogonal 
100-bit binary codes. Thus by replacing the single 512-bit complementary 
code by a code-selected 100-bit binary sequence, additional data or coded 
information can be transmitted as an aid to improving the overall 
frequency management of a communications system. This extra information is 
of particular benefit if the sounder is to be incorporated into an HF 
Management System. The coded information or data could be used to convey: 
a) Interference levels for the given frequency at the sounder transmitter 
location (or colocation); 
b) Selective call--each receiver of the sounding broadcast would be 
identified by a unique code/number; 
c) Frequency availability broadcast--frequencies already in use would be 
identified; 
d) Radiated power control--instructions could be sent over the sounding 
broadcast to increase or reduce power; 
e) Message error control--requests for retransmissions of messages could be 
made; and 
f) Link engineering instructions--other instructions such as crypto input, 
data speed, message urgency, service being used (e.g. RATT or facsimile) 
etc. 
An HF channel management system linked to a sounding broadcast station 
would provide a very powerful ability for automatic control of 
communications systems from numerous remote stations (mobile air, land and 
sea) to work into a single control station with high reliability, 
availability and performance. 
Given an ability to select an operational frequency having specific 
propagating characteristics the communicator may choose to select a 
channel having one or more of the following parameters: 
(1) minimum radio path loss 
(2) minimum interference level 
(3) single mode path structure 
(4) highest usable frequency (when any other receiver closer than the 
transmitter receiver separation will be unable to receive the transmitter 
signal via a sky-wave) or 
(5) maximum received S/N ratio. 
In practice it is found that most naturally occurring propagation paths 
tend to be perturbed by significant amounts of interference, mainly from 
other users. This is because a path that can propagate a signal for one 
particular user is similarly likely to do so for other users, particularly 
if the transmitters and receivers are more or less geographically 
co-located. Fortunately however most HF signals are narrow band and so 
interference from these other users can nearly always be avoided by simply 
changing the operating frequency by a few kHz to a relatively more quiet 
channel in which to operate. Thus, the invention provides the means for 
real time frequency sounding which can be arranged to automatically tune 
to the optimum frequency or frequencies.