Quadrature frequency changer, tuner and modulator

A quadrature frequency changer comprising first and second mixers and a local oscillator and a tuner and modulator including the quadrature frequency changer are provided. The local oscillator provides first and second commutating signals which are nominally in phase-quadrature and includes an arrangement for changing the phase of the first commutating signal by 180° to provide a third commutating signal. The first mixer comprises first and second mixing stages for mixing an input signal with the first and second commutating signals and a summer for summing the mixer stage output signals. The second mixer comprises third and fourth mixing stages for mixing the same or a different input signal with the second and third commuting signals and a summer for summing the third and fourth mixer stage output signals.

CROSS-REFERENCE TO RELATED APPLICATION

This application claims priority to British Patent Application Serial Number GB 0516768.9, filed Aug. 16, 2005, which is herein incorporated by reference.

FIELD OF THE INVENTION

Embodiments of the present invention relate to a quadrature frequency changer and to a tuner and a modulator including such a frequency changer. Such a tuner may be used, for example, for receiving analog or digital television, audio, data or telephony broadcasts via, for example terrestrial or satellite broadcast or cable distribution. Such a modulator may be used, for example, to encode signals for transmission or distribution at radio frequency.

BACKGROUND

FIG. 1of the accompanying drawings illustrates a known type of quadrature frequency converter. The frequency converter comprises two signal paths containing an in-phase (I) mixer1and a quadrature (Q) mixer2. The mixer receives commutating signals, which are intended to be in phase-quadrature with respect to each other, from a local oscillator3via a quadrature splitter4and mixes the signal in each signal path with the respective commutating signal to generate in-phase and quadrature output signals IOand QO. The “quality” of the quadrature output signals is determined at least in part by amplitude imbalances between the signal paths and quadrature phase imbalances between the quadrature commutating signals.

Gain imbalances may be reduced on the order of 0.1 dB by good design and layout techniques. However, quadrature phase imbalances are more difficult to reduce to levels which provide acceptable quadrature frequency changing performance. For example, in arrangements where the frequency of the local oscillator3is required to vary by an octave or more, it is difficult to reduce phase imbalances to less than 3° across the frequency band of operation.

A known technique for reducing quadrature phase errors is to supply the quadrature commutating signals to a double-balanced or a double-double-balanced mixer. Such a mixer produces an error signal proportional to the phase error from 90° from the commutating signals and this error signal may be used to control a feedback loop including the quadrature splitter4so as to reduce phase imbalance. However, such an arrangement is relatively complex and inconvenient.

InFIG. 1, the phase error from 90° between the commutating signals is represented by a phase error or shift of φ° added to the in-phase or 0° commutating signal whereas the quadrature or 90° signal is uncontaminated. The commutating signals are illustrated as vectors in the graph ofFIG. 2in the accompanying drawings and, when applied to the mixers1and2, result in the same phase imbalance or error in the converted output signals IOand QO. The phase error φ° has to be reduced to a value which is sufficiently low to provide acceptable operation of the quadrature frequency converter.

SUMMARY OF THE INVENTION

One embodiment of the invention provides a quadrature frequency changer. The quadrature frequency changer generally includes first and second mixers and a local oscillator, said local oscillator being arranged to provide first and second commutating signals which are nominally in phase-quadrature and including an arrangement for changing a phase of said first commutating signal by 180° to provide a third commutating signal, said first mixer comprising first and second mixing stages for mixing a first input signal with said first and second commutating signals, respectively, and a first linear combiner for forming a linear combination of output signals of said first and second mixing stages, said second mixer comprising third and fourth mixing stages for mixing a second input signal with said second and third commutating signals, respectively, and a second linear combiner for forming a linear combination of output signals of said third and fourth mixing stages.

Another embodiment of the invention provides a tuner. The tuner generally includes a quadrature frequency changer comprising first and second mixers and a local oscillator, said local oscillator being arranged to provide first and second commutating signals which are nominally in phase-quadrature and including an arrangement for changing a phase of said first commutating signal by 180° to provide a third commutating signal, said first mixer comprising first and second mixing stages for mixing a first input signal with said first and second commutating signals, respectively, and a first linear combiner for forming a linear combination of output signals of said first and second mixing stages, said second mixer comprising third and fourth mixing stages for mixing a second input signal with said second and third commutating signals, respectively, and a second linear combiner for forming a linear combination of output signals of said third and fourth mixing stages.

Yet another embodiment provides a modulator. The modulator generally includes a quadrature frequency changer comprising first and second mixers and a local oscillator, said local oscillator being arranged to provide first and second commutating signals which are nominally in phase-quadrature and including an arrangement for changing a phase of said first commutating signal by 180° to provide a third commutating signal, said first mixer comprising first and second mixing stages for mixing a first input signal with said first and second commutating signals, respectively, and a first linear combiner for forming a linear combination of output signals of said first and second mixing stages, said second mixer comprising third and fourth mixing stages for mixing a second input signal with said second and third commutating signals, respectively, and a second linear combiner for forming a linear combination of output signals of said third and fourth mixing stages.

Like reference numerals refer to like parts throughout the drawings.

DETAILED DESCRIPTION

FIG. 3illustrates a tuner of the zero intermediate frequency (ZIF) type, in which any desired channel in a broadband radio frequency input signal can be selected and converted to quadrature baseband in-phase (I) and quadrature (Q) output signals. The tuner may have an input1, for example, for connection to a terrestrial aerial, a satellite aerial system or a cable distribution network. The input1may be connected to a tracking radio frequency bandpass filter2, whose passband tracks the frequency of the selected channel and whose bandwidth is sufficient to pass the selected channel with minimal attenuation together with several adjacent channels. Alternatively, the filter2may be fixed, for example, to provide a band limit function or may be omitted. The output of the filter2may be supplied to an automatic gain control (AGC) stage3, which provides amplification and control of gain so as to supply a consistent signal level to the frequency changer.

The frequency changer may comprise an I mixer, comprising mixing stages4and5and a summer6, and a Q mixer, comprising mixing stages7and8and a summer9. The frequency changer may further comprise a local oscillator10, a quadrature splitter11and a phase shifter12for shifting the phase of the incoming signals by π radians or 180°.

The mixing stages4,5,7and8may have signal inputs, which are connected together and to the output of the stage3. The outputs of the mixing stages4and5may be vectorially summed by the summer6, whose output may comprise the output of the I mixer. Similarly, the outputs of the mixing stages7and8may be summed by the summer9, whose output may provide the Q output of the Q mixer. The mixing stages5and8may have commutating inputs connected to receive a commutating signal from the quadrature splitter11having a relative phase of 90°. The mixing stage4may have a commutating input connected to the quadrature splitter11to receive a commutating signal having a nominally 0° relative phase, but which may include a phase imbalance of φ°. This signal may also be supplied to the phase shifter12, which may provide accurate phase shifting by 180°, for example by inverting the input waveform. The output signal of the shifter12should thus have a relative phase of 180°+φ°, and this may be supplied as the commutating signal to the commutating input of the mixing stage7.

The output of the I mixer from the summer6may be supplied to an I filter13, which may have a passband sufficient for passing the desired I component at baseband and for attenuating adjacent channels. The output of the filter13may be supplied to another AGC stage14, whose output may be connected to the I output17of the tuner. Similarly, the output of the Q mixer from the summer9may be supplied to a Q filter15, which may be substantially identical to the filter13. The filtered output may be supplied to another AGC stage16, whose output may be connected to the Q output18of the tuner.

FIG. 4illustrates the actual phases of the nominally quadrature-phase local oscillator signals ILOand QLO. There may be a quadrature imbalance of φ° between the local oscillator signals, and this is illustrated (without any loss of generality) as being superimposed on the in-phase commutating signal. The relative phase of the local oscillator quadrature signals substantially determines the relative phase of the I and Q mixer output signals in a conventional quadrature frequency changer of the type shown inFIG. 1and the resulting quadrature imbalance results in contamination or crosstalk between the quadrature output signals.

FIG. 5is a vector diagram illustrating the operation of the I mixer. The mixing stage4may receive the commutating signal ILOand convert the input signal to a frequency-converted signal having the same relative phase. This signal is illustrated at IAinFIG. 5. Similarly, the mixing stage5may convert the input signal to the frequency-converted signal with the phase illustrated at IBinFIG. 5. The summer6may form the vector sum of the signals IAand IBto form the output signal IOwith the vector IObisecting the angle 2θ1between the vectors IAand IB. The angle θ1from the horizontal axis is given by

FIG. 6illustrates the operation of the Q mixer in the same format. The mixing stage8may frequency-convert the incoming signal to the output signal QBwith a relative phase of 90°. The mixing stage7may produce the frequency-converted output signal QAwith a relative phase of 180°+φ, and the summer9may form the vector sum in an effort to provide the Q mixer output signal QOwith a phase angle of θQbelow the horizontal axis, which phase angle is given by:

θQ=90+Φ2
The relative phase between the output signals IOand QOis given by:

The effect of this should be that the phase error φ is effectively cancelled by the mixing processes in the frequency changer. Thus, quadrature phase errors may be theoretically removed so that the quadrature output signals are theoretically in perfect phase-quadrature. In practice, perfect cancellation may not be achieved, but the reduction in quadrature phase imbalance provided by the frequency changer represents a substantial improvement in quadrature balance and does not require any complicated control loops.

FIG. 7illustrates an example of the circuit structure and interconnection of the mixing stages4and7and the phase shifter12. The mixing stages may be of the Gilbert cell type comprising transconductance stages4aand7aand current switching cells4band7b. As shown in the inset at20, each transconductance stage may comprise a long tail pair of transistors21and22provided with emitter degeneration resistors23and24and a constant tail current source25. The inset26illustrates each switching cell, which may comprise transistors27to30. The emitters of the transistors27and28may be connected to the collector of the transistor21, whereas the emitters of the transistors29and30may be connected to the collector of the transistor22. The collectors of the transistors27and29may be connected together to form a first differential output line whereas the collectors of the transistors28and30may be connected together to form a second differential output line. The bases of the transistors27and30may be connected together to form a first differential commutating input line, whereas the bases of the transistors28and29may be connected together to form a second differential commutating input line.

The differential commutating inputs of the switching cell4bmay be connected to receive the local oscillator commutating signal LO1having a nominal phase of 0°, but an actual phase of (0+φ)°. The commutating signal inputs of the switching cell7bmay be connected with reverse polarity or in “anti-phase” to the same commutating signal so that the switching cell7bmay receive the inverted commutating signal, which is equivalent to the non-inverted signal shifted in phase by 180°. This interconnection, therefore, may provide a relatively simple phase shifter12.

FIG. 8illustrates another application of the frequency changer4to12shown inFIG. 3. This arrangement may be used as a single sideband (SSB) modulator or as an image cancelling mixer in an effort to provide a classical or near zero intermediate frequency. The I and Q mixers may supply their output signals to phase shifters31and32, respectively, which may provide phase shifts of −45° and +45°, respectively. The outputs of the phase shifters31and32may be supplied to a summer33, which may form the vector sum of the phase-shifted I and Q signals. In the case of an SSB modulator, one of the sidebands may be effectively suppressed, whereas the other may be “constructed” and supplied to the output34of the modulator. In the case of an image cancelling mixer, the image channel may be effectively suppressed, whereas the selected desired channel may be constructed and supplied to the output34. The phase shifters31and32may also perform a filtering function to attenuate adjacent channels.

In the case of an image cancelling mixer, the I and Q mixer outputs may be in the intermediate frequency band so that the phase shifters31and32may be required to perform accurate phase shifting over a relatively small frequency range. However, in the case of a SSB demodulator, the frequency of the mixer output signals may vary substantially so that broadband phase shifters would be required. However, in order to provide the sideband suppression, it may be sufficient for there to be a relative phase shift of 90° between the signal paths including the I and Q mixers. Thus, as illustrated inFIG. 9, the phase shifters31and32may be disposed ahead of the I and Q mixers, respectively. Because the input signal to such a modulator should be in the same frequency band irrespective of the required output frequency, this arrangement may allow narrow band phase shifters to be used.

FIG. 10illustrates a more general form of modulator using the same quadrature frequency changer4to12as inFIG. 3. In this case, the I and Q mixers may receive separately generated modulating signals from a data encoder and a digital signal processor (DSP)40. Such an arrangement may be used to generate a wide variety of types of modulation, for example, including vestigial sideband (VSB) typically used in analog television transmission and quadrature amplitude modulation (QAM) as typically used in cable distribution systems.