Radio frequency transmitter having translational loop phase equalization

A Radio Frequency RF transmitter includes a translational loop architecture that supports non-constant envelope modulation types and includes by adjusting the envelope of the translational loop at the translational loop output. The RF transmitter includes a phase equalizer, an Intermediate Frequency (IF) modulator, a translational loop, an envelope time delay adjust block, an envelope adjust block, and a time delay calibration block. The phase equalizer receives a modulated baseband signal and phase equalizes the modulated baseband signal to produce a phase equalized modulated baseband signal. The IF modulator receives the phase equalized modulated baseband signal and produces a modulated IF signal having a non-constant envelope. The translational loop receives the modulated IF signal and produces a modulated RF signal having a constant envelope. The envelope time delay adjust block receives an envelope signal corresponding to the original modulated signal and produces a time delayed envelope signal based upon a time delay control signal. The envelope adjust block adjusts the modulated RF signal based upon the time delayed envelope signal to produce an envelope adjusted modulated RF signal. Finally, the time delay calibration block receives the envelope adjusted modulated RF signal and produces the time delay control signal.

FIELD OF THE INVENTION

This invention relates generally to wireless communications; and more particularly to the operation of a Radio Frequency (RF) transmitter within a component of a wireless communication system.

BACKGROUND OF THE INVENTION

The structure and operation of wireless communication systems is generally known. Examples of such wireless communication systems include cellular systems and wireless local area networks, among others. Equipment that is deployed in these communication systems is typically built to support standardized operations, i.e. operating standards. These operating standards prescribe particular carrier frequencies, channels, modulation types, baud rates, physical layer frame structures, MAC layer operations, link layer operations, etc. By complying with to these operating standards, equipment interoperability is achieved.

In a cellular system, a governmental body licenses a frequency spectrum for a corresponding geographic area (service area) that is used by a licensed system operator to provide wireless service within the service area. Based upon the licensed spectrum and the operating standards employed for the service area, the system operator deploys a plurality of carrier frequencies (channels) within the frequency spectrum that support the subscribers' subscriber units within the service area. These channels are typically equally spaced across the licensed spectrum. The separation between adjacent carriers is defined by the operating standards and is selected to maximize the capacity supported within the licensed spectrum without excessive interference. In most cases, severe limitations are placed upon the amount of adjacent channel interference that may be caused by transmissions on a particular channel.

In cellular systems, a plurality of base stations is distributed across the service area. Each base station services wireless communications within a respective cell. Each cell may be further subdivided into a plurality of sectors. In many cellular systems, e.g., GSM cellular systems, each base station supports forward link communications (from the base station to subscriber units) on a first set of carrier frequencies and reverse link communications (from subscriber units to the base station) on a second set carrier frequencies. The first set and second set of carrier frequencies supported by the base station are a subset of all of the carrier frequencies within the licensed frequency spectrum. In most, if not all cellular systems, carrier frequencies are reused so that interference between base stations using the same carrier frequencies is minimized but so that system capacity is increased. Typically, base stations using the same carrier frequencies are geographically separated so that minimal interference results.

Both base stations and subscriber units include Radio Frequency (RF) transmitters and RF receivers, together “RF transceivers.” RF transceivers service the wireless links between the base stations and subscriber units. The RF transmitter receives a baseband signal from a baseband processor, converts the baseband signal to an RF signal, and couples the RF signal to an antenna for transmission. In most RF transmitters, because of well-known limitations, the baseband signal is first converted to an Intermediate Frequency (IF) signal and then the IF signal is converted to the RF signal. The RF receiver receives an RF signal, converts the RF signal to an IF signal, and then converts the IF signal to a baseband signal, which it then provides to the baseband processor.

The fast growth of the mobile communications market demands multi-band RF transceivers that are small in size, low in cost, and have low power consumption. These market demands require that the architecture of the RF transceiver to be suitable for a high level of system integration on a single chip for reduced cost and miniaturized mobile device size. Low power consumption is very critical for increasing mobile device battery life and is very important for small mobile devices that include small batteries. To meet these design challenges, some RF transmitters now use translational loop architecture to convert the IF signal to an RF signal. Translational loop architectures are useful for constant envelope modulated wireless systems, such as the new generation Global Standards for Mobile Communications (GSM) and General Packet Radio System (GPRS) phones that employ Gaussian Minimum Shift Keying (GMSK) modulation. However, so far, the translational loop architecture has not been successfully applied in systems that employ a non-constant envelope modulation format, such as QPSK for CDMA (IS-95) and US-TDMA (IS-136) standardized systems, for 8-PSK for EDGE standard based mobile devices, and for mobile devices that support other non-constant envelope modulation formats, such as 16 QAM, 32 QAM, 64 QAM, 128 QAM, etc.

Thus, there is a need in the art for a lower power consumption RF transmitter that supports both constant envelope modulation formats and non-constant envelope formats, among other shortcomings of the prior devices.

SUMMARY OF THE INVENTION

Thus, in order to overcome the above-described shortcomings as well as other shortcomings of the present devices and methodologies, an RF transmitter constructed according to the present invention includes a translational loop architecture that supports non-constant envelope modulation types, e.g., QPSK, 8-PSK, 16 QAM, 32 QAM, 64 QAM, 128 QAM, etc. The translational loop architecture of the present invention adjusts the envelope of the translational loop so that it supports non-constant envelope modulation types. The RF transmitter may be contained in a mobile device or a stationary device.

In particular, the RF transmitter includes an Intermediate Frequency (IF) modulator, a translational loop, an envelope time delay adjust block, an envelope adjust block, and a time delay calibration block. The IF modulator receives a modulated baseband signal and produces a modulated IF signal having a non-constant envelope. The translational loop receives the modulated IF signal and produces a modulated RF signal having a constant envelope. The envelope time delay adjust block receives an envelope signal corresponding to the modulated signal and produces a time delayed envelope signal based upon a time delay control signal. The envelope adjust block adjusts the modulated RF signal based upon the time delayed envelope signal to produce an envelope adjusted modulated RF signal. Finally, the time delay calibration block receives the envelope adjusted modulated RF signal and produces the time delay control signal.

In one embodiment, the time delay calibration block includes a down converter, an Analog to Digital Converter (ADC), a Low Pass Filter (LPF), a Band Pass Filter (BPF), and a level detector and control block. The down converter converts the envelope adjusted modulated RF signal to a complex baseband signal. The ADC samples the complex baseband signal. The LPF couples to the ADC and filters the complex baseband signal to produce a LPF output. The BPF also couples to the ADC and filters the complex baseband signal to produce a BPF output. With this structure, the level detector and control block receives the LPF output and the BPF output and produces the time delay control signal based upon the LPF output and the BPF output. These components of the RF transmitter may be embodied within resources resident in a coupled baseband processor.

The time delay calibration block of the RF transmitter determines a desired channel power corresponding to the RF signal. The time delay calibration block also determines an alternate channel power (or adjacent channel power) that is emitted by the translational loop that corresponds to an alternate channel (or adjacent channel). The time delay calibration block then determines the time delay control signal based upon a ratio of the channel power and the alternate channel power. With the time delay set properly, the envelope of the RF signal produced by the translational loop is substantially or fully matched with the phase of the RF signal produced by the translational loop. Resultantly, the alternate channel power (and the adjacent channel power) produced by the translational loop is reduced/minimized to meet systematic requirements.

The RF transmitter may also include an envelope detection block that produces the envelope signal. In one embodiment, the envelope detection block determines the envelope signal based upon the complex baseband signal received from the baseband processor (not the complex baseband signal generated within the time delay calibration block). In another embodiment, the envelope detection block determines the envelope signal based upon the modulated IF signal. In still another embodiment, the envelope detection block receives the envelope signal from a coupled baseband processor.

In at least some embodiments, the envelope signal produced by the envelope detection block is a digital signal while the time delayed envelope signal is an analog signal. With this signal format, the time delay block delays a digital envelope signal by a delay that is based upon the time delay control signal. Then, a digital to analog converter receives the output of the time delay block and produces the time delayed envelope signal.

The translational loop may have a phase response that varies with frequency such that the phase response of the translational loop is constant within the desired channel but not constant within an adjacent channel and/or in an alternate channel, i.e., the phase response is a function of frequency within the adjacent channel and/or the alternate channel. Because of this non-constant phase response of the translational loop, even if the envelope and phase at the output of the translational loop are perfectly matched for the desired channel, they will be mismatched in the adjacent channel and/or alternate channel. Resultantly, the adjacent channel power and the alternate channel power output by the translational loop may violate systematic requirements.

Thus, with another RF transmitter constructed according to the present invention, an incoming baseband signal (or IF signal) is phase equalized by a phase equalizer to compensate for the non-constant phase response of the translational loop. With this phase equalization performed prior to the translational loop, the non-constant phase response of the translational loop is overcome so that the adjacent channel power and alternate channel power output by the translational loop is further reduced.

Other features and advantages of the present invention will become apparent from the following detailed description of the invention made with reference to the accompanying drawings.

DETAILED DESCRIPTION OF THE DRAWINGS

FIG. 1Ais a system diagram illustrating a cellular system within which the present invention is deployed. The cellular system includes a plurality of base stations102,104,106,108,110, and112that service wireless communications within respective cells/sectors. The cellular system services wireless communications for a plurality of wireless subscriber units. These wireless subscriber units include wireless handsets114,118,120, and126, mobile computers124and128, and desktop computers116and122. When wirelessly communicating, each of these subscriber units communicates with one (or more during handoff) of the base stations102through112. Each of the subscriber units114–128and each of the base station102–112include radio frequency (RF) circuitry constructed according to the present invention.

The RF circuitry of the present invention may be contained in any of the subscriber units114–128, any of the base stations102–112or in another wireless device, whether operating in a cellular system or not. Thus, for example, the teachings of the present invention may be applied to wireless local area networks, two-way radios, satellite communication devices, or other devices that support wireless communications.

The RF circuitry of the present invention supports both constant envelope and non-constant envelope modulation types. The RF transmitter section of this RF circuitry supports non-constant envelope modulation formats such as QPSK for CDMA (IS-95) and US-TDMA (IS-136) standardized systems and 8-PSK for EDGE standardized systems. The RF transmitter of the present invention also supports other non-constant envelope modulation types, e.g., QPSK, 8-PSK, 16 QAM, 32 QAM, 64 QAM, 128 QAM, etc, whether standardized or not. The structure and operation of the RF transmitter is described further with reference toFIGS. 2–10B

The structure and operation of the RF transmitter of the present invention may also be implemented to service other wireless communications as well. For example, the RF transmitter may be used to service premises based Wireless Local Area Network (WLAN) communications, e.g., IEEE 802.11a and IEEE 802.11b communications, and ad-hoc peer-to-peer communications, e.g., Bluetooth. In a WLAN system, the structure would be similar to the structure shown inFIG. 1Abut, instead of the base stations102–112, the WLAN system would include a plurality of Wireless Access Points (WAPs). Each of these WAPs would service a corresponding area within the serviced premises and would wirelessly communicate with serviced wireless devices. For peer-to-peer communications, such as those serviced in Bluetooth applications, the RF transmitter of the present invention would support communications between peer devices, e.g., lap top computer124and hand-held wireless device126.

FIG. 1Bis a block diagram generally illustrating the structure of a wireless device150constructed according to the present invention. The general structure of the wireless device150will be present in any of the wireless devices114–128illustrated inFIG. 1A. The wireless device150includes a plurality of host device components152that service all requirements of the wireless device150except for the RF requirements of the wireless device150. Of course, operations relating to the RF communications of the wireless device150will be partially performed by the host device components152.

Coupled to the host device components152is the RF interface154. The RF interface154services the RF communications of the host device150and includes a baseband processor160, an RF transmitter156, and an RF receiver158. The RF transmitter156and the RF receiver158both couple to an antenna160. One particular structure of a host device is described with reference toFIG. 8. Further, the teachings of the present invention are embodied within the RF transmitter156of the RF interface154.

FIG. 2is a block diagram illustrating an RF transmitter200constructed according to the present invention. The RF transmitter200includes an I-Q Intermediate Frequency (IF) modulator202that receives an I-Q baseband signal from a baseband processor and converts the I-Q represented baseband signal into a modulated IF signal. The modulated IF signal produced by the I-Q IF modulator serves as the input to a translational loop204. The translational loop204produces an RF signal at the Transmit (TX) frequency that is received by an envelope adjust block206. Note that the modulated IF signal received by the translational loop204is a non-constant envelope signal (corresponding to a non-constant envelope modulation type) while the RF signal output by the translational loop204is a constant envelope signal. The envelope adjust block206adjusts the envelope of the RF signal produced by the translational loop204based upon a time delayed envelope signal. The envelope adjust block206produces an envelope adjusted RF signal that is amplified by Power Amplifier (PA)208. The output of the PA208is applied to an RF switch, a RF filter or a duplexer that couples the amplified envelope adjusted RF signal to an antenna.

An envelope detection block210, an envelope time delay adjust block212, and a time delay calibration block214work in cooperation to produce the time delayed envelope signal. Envelope detection block210detects the envelope of the IF signal using one of three (or more) techniques/embodiments. In a first embodiment, the envelope detection block210receives an envelope indication from a coupled baseband processor. In a second embodiment, the envelope detection block210measures I and Q components of the baseband signal at the input of the I-Q IF modulator to calculate the envelope. In a third embodiment, the envelope detection block210measures the envelope of the IF signal at the output of the I-Q IF modulator202.

The envelope signal, which represents a magnitude of the modulated signal, is output by the envelope detection block210and received by the envelope time delay adjust block212. The envelope time delay adjust block212outputs a time delayed envelope signal. The time delayed envelope signal, with respect to the envelope signal, has been delayed so that it correctly corresponds to the RF signal output of the translational loop204. Thus, the delay introduced by the envelope adjust time delay212corresponds to the delay introduced by the translational loop204, and in addition, the delay introduced by the I-Q IF modulator202when the envelope is detected from the baseband signal at the input to the I-Q IF modulator202. Thus, the RF signal output of the envelope adjust block206is a phase and magnitude matched RF signal having a non-constant envelope.

A time delay calibration block214adjusts the delay introduced into the detected envelope by the envelope time delay adjust block212. The operation of the time delay calibration block214will be described in detail with reference toFIGS. 3–6and9–10B. Generally speaking, the time delay calibration block214adjusts the time delay introduced by the envelope time delay adjust block212until the ratio of the signal level within a subject channel to the signal level of an adjacent channel (or an alternate channel) satisfies the operating requirements of the RF transmitter200.

FIG. 3is a block diagram illustrating in more detail the RF transmitter200ofFIG. 2. As shown, the I-Q IF modulator202receives an IF reference signal. In the described embodiment, the IF reference signal is equal to ⅙ of the transmit frequency (TX) reference signal. A phase shifter304receives the IF reference signal (I component) and produces a corresponding Q component by shifting in phase the I component by 90 degrees. An I-Q mixer302separately mixes the I and Q components of the baseband signal with the I and Q components of the IF reference signal. The output of the I-Q mixer302is summed and band pass filtered by block306, which produces the modulated IF signal.

The translational loop204includes a Phase and Frequency Detector (PFD)308that receives the modulated IF signal and produces an output that is received by a charge pump310. Low Pass Filter (LPF)312filters the output of the charge pump310and the output of the LPF312drives a TX Voltage Controlled Oscillator (VCO)314. The output of the TX VCO314serves as one input to the envelope adjust block206, which is a mixer in the illustrated embodiment. The translational loop204also includes a mixer316that receives the output of the TX VCO314and a 7/6 TX reference signal from a local oscillator. The output of mixer316with ⅙ of the TX frequency serves as a second input to the PFD308.

The envelope detection block210implements the second embodiment described above by calculating the I and Q components in a digital format. The envelope detection block210also includes processing operations to determine the magnitude of the envelope by calculating the square root of (I2+Q2). When the envelope detection block210implements the third embodiment described above, it digitizes the modulated IF signal produced at the output of the I-Q IF modulator and determines the magnitude thereof.

The envelope time delay adjust block212includes a time delay block322and a Digital to Analog converter (DAC)324. The time delay calibration block214includes a mixer326that mixes the RF signal produced by mixer206(envelope adjust block) with a local oscillator signal at a frequency equal to the TX carrier frequency to down convert the modulated TX signal to a complex baseband signal at the output of the mixer326. An Analog to Digital Converter (ADC)328receives the output of the mixer326and converts the signal to a digital equivalent of the complex baseband signal. Both a Low Pass Filter (LPF)330and a Band Pass Filter (BPF)332filter the complex baseband signal. As an example of an operation that would be performed in an EDGE standard based cellular system, the LPF330would have a cutoff frequency (Fc) of 200 kHz, and the BPF332would have an Foof 500 kHz and a Bandwidth of 200 kHz.

The outputs of the LPF330and the BPF332are received by a level determination and control block334that produces the time delay control signal that is an input to the time delay block322. In the described embodiment, the LPF330produces a signal that corresponds to a subject channel and the BPF332produces a signal that corresponds to an alternate channel (or adjacent channel). The level determination and control block334uses these signals to calculate an Alternate Channel Power Ratio (ACPR) that is the ratio of the Channel Power (CP) to the Alternate Channel Power (ACP). The ACPR is then compared to an allowable limit. If the ACPR meets this limit, the input to the time delay block322is properly set. However, if the ACPR does not meet this limit, the level determination and control block334adjusts the time delay control signal to the time delay block322until the limit is met.

FIG. 4is a block diagram illustrating another embodiment of the time delay calibration block214ofFIG. 2that performs the ACPR measurement operations. As shown inFIG. 4, for hardware efficient filtering of the alternate channel signal, the BPF332includes a DDFS Look Up Table (LUT)402, a complex mixer404, and a LPF406. Further, the level detection and control block334includes a power level detector410and a time delay control signal block412. The power level detector410outputs detected levels of the CP and the ACP. Based upon a ratio of these measurements, the ACPR, the time delay control signal block412produces the time delay control signal.

FIG. 5Ais a block diagram illustrating a LPF employed in an ACPR measurement block constructed according to the present invention. The LPF includes filter operation blocks502,506,510, and512and gain operator504. The low pass filter ofFIG. 5Amay be employed for both/either of LPFs330and406ofFIG. 4.

FIG. 5Bis a block diagram illustrating a clock generation circuit constructed according to the present invention. As was described with reference toFIG. 3, various reference signals must be produced for the RF transmitter200. These frequencies include 1/6 TX, TX, and 7/6 TX. In order to produce these frequencies, the clock generation circuit ofFIG. 5Bincludes a Local Oscillator552that produces a clock reference signal. A Phase Locked Loop (PLL)554receives the clock reference signal and produces a frequency-multiplied output based thereupon. A first divider556receives the frequency-multiplied output and divides the signal to produce the 7/6 TX reference signal. A second divider558receives the frequency-multiplied output and divides the signal to produce the 1/6 TX (IF) reference signal. Further, in order to produce the TX reference signal (for use by the time delay calibration block214), the 7/6 TX reference signal and the 1/6 TX reference signal are mixed via mixer560to produce the TX reference signal.

FIG. 6is a block diagram illustrating a power level detector constructed according to the present invention. As is shown, the power level detector includes a first mixer602that is employed to produce the Power Spectral Density (PSD) of the signal output by the LPF330. Further, a second mixer604is employed to produce the PSD of the signal output by the BPF332. Filtering blocks606and610filter the outputs of the mixers602and604, respectively. Subsequently, the outputs of filtering blocks606and610are down sampled by down sampling blocks608and612, respectively, to produce the CP and ACP signals, respectively. Note that the input to the power level detector410is at a sample rate of 26 MHz and that the output of the power level detector410is at a sample rate of 10 kHz.

FIG. 7is a block diagram illustrating a time delay block322of the envelope time delay adjust block212constructed according to the present invention. The time delay block322receives as its input the envelope signal produced by the envelope detection block210and the time delay control signal produced by the time delay calibration block214. The time delay block322includes a plurality of serially coupled delay elements702–712. Multiplexer714receives as its input the envelope signal and the output of each of the serially coupled delay elements702–712. The multiplexer714receives as its control input the time delay control signal from the time delay calibration block214. As its output, the multiplexer714produces the time delayed envelope signal. The delay introduced by the delay elements702–712is determined based upon expected minimum and maximum delays that are required to compensate for the delay introduced by the translational loop204(and the I-Q IF modulator202).

FIG. 8is a block diagram illustrating a subscriber unit802constructed according to the present invention. The subscriber unit802operates within a cellular system, such as the cellular system described with reference toFIG. 1Aand according to the operations previously described with reference toFIGS. 2–7and as will subsequently be described with reference toFIGS. 9–10B. The subscriber unit802includes an RF unit804, a processor806that performs baseband processing and other processing operations, and a memory808. The RF unit804couples to an antenna805that may be located internal or external to the case of the subscriber unit802. The processor806may be an Application Specific Integrated Circuit (ASIC) or another type of processor that is capable of operating the subscriber unit802according to the present invention. The memory808includes both static and dynamic components, e.g., DRAM, SRAM, ROM, EEPROM, etc. In some embodiments, the memory808may be partially or fully contained upon an ASIC that also includes the processor806. A user interface810includes a display, a keyboard, a speaker, a microphone, and a data interface, and may include other user interface components. The RF unit804, the processor806, the memory808, and the user interface810couple via one or more communication buses/links. A battery812also couples to and powers the RF unit804, the processor806, the memory808, and the user interface810.

The RF unit804includes the RF transmitter components described with reference toFIG. 2and operates according to the present invention to adjust the envelope of an RF signal produced by a translational loop contained therein. The structure of the subscriber unit802illustrated is only one particular example of a subscriber unit structure. Many other varied subscriber unit structures could be operated according to the teachings of the present invention. Further, the principles of the present invention may be applied to base stations, as are generally described with reference toFIG. 1A.

FIG. 9is a logic diagram illustrating a method of operation according to the present invention. Operation according to the present invention is initiated upon power up or reset. In such case, the operations ofFIG. 9will be performed along with a number of other operations required to set/reset the operation of a corresponding RF transmitter200. The operations ofFIG. 9will be described with additional reference toFIGS. 10A and 10B.FIG. 10Ais a graph illustrating the power spectral density of an RF signal generated by the RF transmitter200of the present invention with the time delay control signal improperly set.FIG. 10Bis a graph illustrating the power spectral density of an RF signal generated by the RF transmitter200of the present invention with the time delay control signal properly set.

Referring again toFIG. 9, upon power up or reset, the calibration operations of the present invention are initiated (step902). In such case, the components of the RF transmitter200, e.g., RF transmitter200ofFIG. 2, operate to produce the RF signal. The RF signal may have a power spectral density such as that shown inFIG. 10A. With the operations of the present invention, the CP (central lobe of the PSD) and the ACP (side lobe of the PSD) are measured (step904). Then, the ratio of the CP to the ACP (or the ACP to the CP) is determined, i.e., ACPR. The ACPR is then compared to an ACPR threshold (step906). This ACPR threshold relates directly to the permissible interference that operation on the channel may produce in the alternate channel (or adjacent channel).

With the PSD shown inFIG. 10A, the ACPR does not meet the threshold and the time delay control signal provided by the time delay calibration block214to the envelope time delay adjust block212is adjusted (step908). Operation returns from step908to step904where the CP and ACP are again measured and the ACPR is determined. After one or more iterations of step908, the PSD ofFIG. 10Bis produced such that the ACPR threshold is met at step906. From step906, with the ACPR met, operation proceeds to step910wherein the currently set time delay control signal produced by the time delay calibration block214is used. Operation remains at step912until a calibration event occurs. A calibration event may occur periodically or when a threshold is met.

FIG. 11is a graph illustrating the phase response of a translational loop employed by an RF transmitter constructed according to the present invention. As shown inFIG. 11, the translational loop, e.g., translational loop204ofFIG. 2, has a phase response that varies with frequency. Within the desired channel, the phase response of the translational loop is substantially constant. Further, within an adjacent channel, the phase response of the translational loop is constant within a portion of the adjacent channel but varying with frequency in another portion of the adjacent channel. Within an alternate channel, the phase response of the translational loop varies substantially with frequency. Other translational loops may exhibit different phase response that varies greater with frequency or lesser with frequency. The graph ofFIG. 11is shown only to describe the teachings of the present invention.

In order to reduce/minimize the adjacent channel power and alternate channel power produced by the translational loop, the phase and envelope of the translational loop must be closely matched for all frequencies, including the desired channel, the adjacent channel, and the alternate channel. The teachings of the present invention, as described with reference toFIGS. 2–10show how this goal may be fully accomplished for translational loops having a constant phase response and mostly accomplished for translational loops having a phase response that varies with frequency. However, for translational loops exhibiting the behavior ofFIG. 11, the phase and envelope of the translational loop cannot be fully matched over the full frequency spectrum of the desired channel, the adjacent channel, the alternate channel. Because envelope and phase matching can only be fully performed within the desired channel (having a constant phase response), adjacent channel power and alternate channel power cannot be fully minimized for translational loops exhibiting the phase response ofFIG. 11.

Thus, according to the present invention, adjacent channel and alternate channel components of the baseband signal are phase equalized prior to their application to the translational loop. In one particular operation according to the present invention, a baseband signal is phase equalized according to the phase equalization curve shown inFIG. 11prior to its conversion to IF. Phase equalization of the baseband signal pre-conditions the baseband signal to overcome the operational shortcomings introduced by the translational loop due to its phase response. In effect, phase equalizing the baseband signal counteracts the phase response of the translational loop. Thus, when the envelope and phase of the output of the translational loop are matched, adjacent channel power and alternate channel power are significantly reduced.

FIG. 12is a block diagram illustrating another embodiment of an RF transmitter constructed according to the present invention. The structure of the RF transmitter1200is similar to the structure of the RF transmitter200described with reference toFIG. 2and elements common toFIG. 2retain common numbering. However, the RF transmitter1200ofFIG. 12further includes a phase equalizer1202and a Digital to Analog Converter (DAC)1204. The phase equalizer1202receives a modulated baseband signal from the baseband processor and produces a phase equalized modulated baseband signal. Phase equalization performed by the phase equalizer1202is performed so as to produce a phase equalization curve illustrated inFIG. 11, for example. The actual phase equalization that the phase equalizer1202produces is based upon calibration operations performed on the translational loop204. These calibration operations will be described further with reference toFIG. 15.

With this structure of the RF transmitter1200, the phase equalizer1202receives digital I and Q components of the modulated baseband signal from a coupled baseband processor. The phase equalizer1202phase equalizes the digital modulated baseband signal and produces a digital phase equalized modulated baseband signal (including both I and Q components). The DAC1204receives the digital phase equalized modulated baseband signal and converts the digital phase equalized modulated baseband signal to an analog phase equalized modulated baseband signal (including both I and Q components) that it provides to the IF modulator202.

In this digital embodiment of the phase equalizer1202, a digital all pass filter that operates in the time domain may be employed. Such a structure is described further with reference toFIG. 14. Alternately, the operations of the phase equalizer1202may be performed in the frequency domain. A structure that performs phase equalization in the frequency domain is described with reference toFIG. 13.

In an alternate construction of the RF transmitter, the phase equalizer1202is an analog component that performs analog phase equalization operations. In such case, the phase equalizer1202would receive an analog modulated baseband signal from the coupled baseband processor and produce an analog phase equalized modulated baseband signal. In one construction in this embodiment, the phase equalizer1202is an analog all pass filter having a phase response that corresponds to the phase equalization curve ofFIG. 11.

With the RF transmitter1200ofFIG. 12, a DAC1206is shown to be external to the envelope time delay adjust block212. With this structure, the envelope time delay adjust block212produces a digital time delayed envelope signal that is converted to an analog equivalent thereof by the DAC1206. The output of the DAC1206is then low pass filtered by low pass filter1208prior to its application to envelope adjust block206.

As was the case with the RF transmitter200ofFIG. 2, with the RF transmitter1200ofFIG. 12, the envelope detection block202may receive input from three separate sources. In a first operation, the envelope detection block202receives its input from the baseband processor. In a second operation, the envelope detection block202receives its input from the output of the phase equalizer1202. In a third operation, the envelope detection block202receives its input from the output of the IF modulator202.

FIG. 13is a block diagram illustrating one embodiment of a phase equalizer of the RF transmitter ofFIG. 12constructed according to the present invention. With the phase equalizer1300ofFIG. 13, phase equalization is done in the frequency domain upon digital signals. The phase equalizer1300includes Fast Fourier Transform (FFT) operational block1302, phase adjust block1304, and Inverse Fast Fourier Transform (IFFT) operational block1306that operate upon the I component of the digital modulated baseband signal. Likewise, the phase equalizer1300includes Fast Fourier Transform (FFT) operational block1308, phase adjust block1310, and Inverse Fast Fourier Transform (IFFT) operational block1312that operate upon the Q component of the digital modulated baseband signal.

The FFT operational blocks1302and1304receive the I and Q components of the digital modulated baseband signal in the time domain, respectively, and convert the I and Q components of the digital modulated baseband signal to the frequency domain. The phase adjust block1304and the phase adjust block1310receive phase equalizer calibration settings1314and, based upon the phase equalizer calibration settings1314, adjust the I and Q components of the modulated baseband signal in the frequency domain to produce I and Q components of the phase equalized modulated baseband signal in the frequency domain. Then, IFFT operational blocks1306and1312receive the I and Q components of the phase equalized modulated baseband signal in the frequency domain and produce the phase equalized modulated baseband signal in the time domain. The phase equalizer calibration settings1314cause the phase adjust blocks1304and1310to implement the phase equalization curve illustrated inFIG. 11.

FIG. 14is a block diagram illustrating another embodiment of a phase equalizer of the RF transmitter ofFIG. 12constructed according to the present invention. The phase equalizer includes all pass filter1402, all pass filter1404, and phase equalizer calibration settings1406. All pass filter1402receives the I component of the modulated baseband signal and produces the I component of the phase equalized modulated signal. All pass filter1404receives the Q component of the modulated baseband signal and produces the Q component of the phase equalized modulated signal. All pass filter1402and all pass filter1404receive respective phase equalizer calibration settings and perform phase equalization operations based thereon. All pass filters are well known in the art. For one description of all pass filters, see Adel S. Sedra and Kenneth C. Smith,Microelectronic Circuits, Fourth Edition, Oxford University Press, 1998.

FIG. 15is a logic diagram illustrating a method for calibrating a phase equalizer according to the present invention. Calibration may be initiated at power up, reset, periodically, or otherwise (step1502). Because the operational characteristics of the components of the RF transmitter including the translational loop may change with temperature, periodic calibration may be preferred. With calibration operations commenced, a first selected desired channel is chosen (step1504). Then, with the RF transmitter transmitting in the desired channel with a test signal, as provided by the baseband processor, the phase response of the translational loop within the adjacent and alternate channels are measured (step1504).

If the chosen desired channel is the last desired channel to be considered, operation proceeds to step1510. If not, operation returns to step1504wherein the next desired channel is selected. Typically, the phase response of the translational loop varies not only with the difference in frequency between the desired channel and adjacent/alternate channels but also with the frequency of the desired channel. As such, calibration operations are performed for each desired channel that is serviced by the translational loop. However, with some translational loops, the phase response may be only a function of the difference between the desired channel and the adjacent/alternate channels. In such case, only a single desired channel is considered in the operations ofFIG. 15.

With measurements obtained for each desired channel, operation proceeds to step1510in which phase equalizer calibration settings are determined. The phase equalizer calibration settings determined will be based upon the structure of the phase equalizer employed. When the phase equalizer is an all pass filter, the phase equalizer calibration settings will include filter coefficients. When the phase equalizer employs frequency domain operations, the phase equalizer calibration settings may include complex frequency domain coefficients.

FIG. 16is a logic diagram illustrating operation of a phase equalizer according to the present invention. RF transmissions at any time will be within a desired channel. Thus, as a first operation, the desired channel is selected and the local oscillator is set accordingly (step1602). Then, the phase equalizer settings for the desired channel are selected and retrieved (step1604). The phase equalizer settings are then loaded into the phase equalizer (step1606). Operation continues with these phase equalizer settings until the desired channel changes (step1608) at which point, a new desired channel is selected (step1610) and operation proceeds to step1602. In another embodiment, a single set of equalizer calibration settings are employed for all desired channels.

The invention disclosed herein is susceptible to various modifications and alternative forms. Specific embodiments therefore have been shown by way of example in the drawings and detailed description. It should be understood, however, that the drawings and detailed description thereto are not intended to limit the invention to the particular form disclosed, but on the contrary, the invention is to cover all modifications, equivalents and alternatives falling within the spirit and scope of the present invention as defined by the claims.