Programmable current source for filter or oscillator

In order to tune the frequency characteristics of a filter or oscillator as an exponential function of the position of a note on a musical scale, a pulse stream with an exponentially scaled duty cycle is used to gate an on-off switch which controls an electronic tuning signal for the first or second stage of the filter or oscillator. The pulse stream source can be a pulse code generator responsive to a priority note generator, or it can be an oscillator-driven one-shot with an exponentially scaled power supply governing the charging rate of a reactive component which determines the one-shot cycle time. There are also pitch offset and pitch modulation features in the circuit, the latter being usable in a constant deviation or a constant interval mode.

BACKGROUND OF THE INVENTION 
This invention relates to electronic musical instruments, particularly 
those employing tunable circuits such as voltage-controlled oscillators 
and voltage-controlled filters. 
Electronic musical instruments typically employ electronically tunable 
frequency-dependent circuits such as voltage-controlled oscillators 
(VCO's) and voltage-controlled filters (VCF's). The oscillators are used 
as musical tone signal sources, and the filters are used to process tone 
signals to achieve desired musical effects. The tuning is conventionally 
accomplished by applying an electronic control signal to select the 
operating frequency of the oscillator or the frequency characteristics of 
the filter. The tunable oscillator and filter circuits are commonly 
denominated "voltage-controlled". In most instances it is actually the 
control signal current which governs the tunable frequency 
characteristics, but ordinarily the current input to the tunable circuit 
is a function of the signal voltage, so the name is not entirely a 
misnomer even when current is the controlling parameter. 
It is recognized in the electronic musical instrument art that pitch varies 
exponentially with position on the musical scale. For example, in terms of 
musical scale distance, middle C is halfway between the two notes an 
octave below and an octave above it; i.e. the scale is linear. But the 
frequency of middle C is not halfway between the frequencies of those 
other two notes; on the contrary, the frequency of middle C is twice that 
of the octave below it, while the upper octave frequency (twice the 
frequency of middle C) is four times the frequency of the octave below 
middle C. Thus, if the oscillator or filter tuning signal is linearly 
proportional to the musical scale position of a selected note, some means 
must be used to convert this linear relation into an exponential one. 
The prior art has typically accomplished this (and incidentally 
accomplished the necessary voltage to current conversion as well) by using 
a conventional bipolar junction transistor as a signal conversion device. 
The transistor is first biased into the exponential region of its 
characteristic response curve, and then an input signal voltage linearly 
proportional to the musical scale position of the selected note is applied 
across its base and emitter electrodes. The resulting collector output 
current is an exponential function of the input voltage, and hence of the 
musical scale position of the selected note. Such an exponentially related 
current output is exactly what is needed to provide a tuning control 
signal for the so-called VCO's and VCF's discussed above. VCO and VCF 
circuits employing such control circuits are described in considerable 
detail in "Musical Engineer's Handbook", 1975 edition, Chapters 5 b and 5 
d respectively. The foregoing work is available from the author and 
publisher, Bernie Hutchins, at 1 Pheasant Lane, Ithaca, New York 14850. 
The problem with using bipolar transistors as exponential conversion 
devices in this fashion is that they drift with temperature and age. While 
various adjustment and compensation techniques are available, it is deemed 
preferable to attack the problem by providing an entirely different type 
of exponential conversion circuit, as the present invention does. 
SUMMARY OF THE INVENTION 
The present invention provides a tuning signal, the controlling parameter 
of which is linearly proportional to the musical scale position of a 
particular note, for example, a note selected by operating a conventional 
musical keyboard. Before the signal is used, however, to control a tunable 
circuit such as the first or second stage of VCO or VCF, it is chopped by 
an on-off switch under the gate control of a stream of pulses having a 
duty cycle which is related exponentially to the musical scale position of 
the selected note. As a result, the controlling parameter of the tuning 
signal is exponentially modulated. The exponential pulse stream can be 
generated by digital techniques, or by the use of appropriately scaled 
resistors to govern the charging rate of a capacitor or other reactive 
component in a time-base circuit. In either case, the circuit exhibits 
superior stability over temperature and with age.

DESCRIPTION OF THE PREFERRED EMBODIMENTS 
FIG. 1 illustrates in general terms one form of the invention. The circuit 
there shown is a fragment of an electronic musical instrument, the omitted 
portions of which are either conventional or are disclosed in detail in 
the above-cited earlier patent application. The musical instrument is 
preferably one of those which has a keyboard, such as a synthesizer or 
organ. Each of the keys on the keyboard, when operated by the musician, 
closes a switch which energizes an individual one of the note selection 
signal lines 10. In response to this signal, the circuit of FIG. 1 
generates and processes an electronic tone signal, the fundamental 
frequency of which is equal to the pitch of a selected musical note. That 
note may be either the note represented by the particular key which the 
musician has depressed, or it may be another note having a predetermined 
harmonic relationship to the key note. 
The circuit which generates the electronic tone signal is a priority note 
generator 12, a specialized digital circuit which is preferably realized 
in the form of an integrated circuit. In brief terms, the priority note 
generator is a circuit in which a high frequency clock pulse train is 
divided down to any desired musical frequency by means of a digitally 
controlled programmable divider to produce a musical tone signal. Further 
details of the priority note generator and its operation are disclosed in 
the earlier patent application cited above, and also in the co-pending 
U.S. patent application of Richard S. Swain and Douglas Moore entitled 
"Tone Generating System for Electronic Musical Instrument", Ser. No. 
835,832, now U.S. Pat. No. 4,186,637 filed Sept. 22, 1977, which is 
assigned in common with this application. The priority note generator has 
a number of output lines. Line 14 is the one on which the tone signal 
appears. At the same time that it produces the tone signal on line 14, the 
priority note generator 12 produces a drive signal on line 16 to activate 
one or more envelope generator circuits 18. The latter respond to the 
input on line 16 by producing an envelope signal which mimics the onset 
and decay of an acoustical tone over time. The envelope generator 18a 
produces its envelope output on line 20, which is connected to activate a 
keyer circuit 22. The keyer circuit also receives the tone signal on line 
14 from the priority note generator 12. The keyer uses the envelope signal 
to modulate, at sub-audio frequencies, the audio frequency tone signal. 
The resulting keyer output, on line 24, is an electrical representation of 
a musical note having the same audio frequency as the tone signal on line 
14, and the onset and decay characteristics of the envelope signal on line 
20. 
The envelope generator circuits 18 and the keyer 22 may be conventional, 
and in any event are described in detail in the above-mentioned Gross 
patent application. 
The envelope-modulated audio signal on line 24 normally requires further 
processing by a filter circuit, preferably a conventional 
voltage-controlled filter (VCF) 26 which changes its frequency response 
characteristics linearly in response to a control signal applied over line 
29. The linear VCF, which makes the filtered audio output signal available 
at terminal 28, is a conventional circuit described in the above mentioned 
Hutchins text. 
As so far described, the circuit is in part conventional and in part the 
subject of the two co-pending patent applications cited above. The present 
application, however, focuses on ways of assuring that the filter control 
signal on line 29 is an exponential function of the musical scale position 
of the note being played (i.e. the note represented by the energized one 
of the input lines 10). Since the VCF responds linearly to the control 
signal on line 29, if that signal is exponentially related to the musical 
scale position of the note, then the VCF response will also be 
exponentially related to the musical scale position of the note. Since 
musical frequency (or pitch) is exponentially related to musical scale 
position, then the VCF will accurately track the pitch of the notes played 
on the keyboard. 
In accordance with this invention, the filter control signal on line 29 is 
governed by an electronic switch 30, for example an MOS integrated 
circuit, catalog number CD 4016 or equivalent, which turns fully on in 
response to the presence of an enabling input on its gate input line 32, 
and turns fully off when that gate input is absent. The source of the 
filter control signal which is applied to the gate 30 over line 34 is a 
drive circuit 36. The drive output is not a function of note selection; 
i.e. it does not depend on which of the n keyboard lines 10 is energized. 
Instead, the drive signal depends only on the shape of an envelope input 
arriving on line 38 and an offset instruction signal which may arrive on 
line 40. For a given condition of these inputs 38 and 40, the drive output 
on line 34 is the same for whatever musical note selection is mandated by 
the keyboard lines 10. Thus the responsibility for imparting the desired 
exponential characteristic to the filter control signal on line 29, as a 
function of musical note selection, rests entirely with the gate signal on 
line 32. 
The latter is derived from a circuit 12a, referred to as a pulse code 
generator, which responds to musical note selection information contained 
in a multi-bit digital code instruction word arriving over a cable 42 from 
the priority note generator 12. The pulse code generator 12A is preferably 
realized as part of the same I.C. chip as the priority note generator 12. 
Various different embodiments of pulse code generators, all suitable for 
the present application, are disclosed in connection with FIGS. 11 through 
17 of the above-cited Gross application. Each of these circuits is 
basically a digitally controlled set of gates which selectively passes or 
blocks a pulse stream in such a way that the output pulse train has 
certain desired time patterns dictated by the input instructions to the 
gates. The input instructions vary according to the musical note selection 
at the keyboard. Each of the possible pulse code generator embodiments 
produces as its output a train of pulses; in some of them these output 
pulses have a repetition rate which is constant for all musical note 
selection conditions, but the individual pulse duration varies as a 
function of the musical scale position of the selected note. In other 
circuits the output pulse duration is constant for all musical note 
selection conditions, but the output pulse repetition rate varies as a 
function of the musical scale position of the selected note. Whether the 
pulse train is duration-modulated or repetition-rate modulated, in either 
case the average duty cycle of the pulse train is a function of note 
selection. Furthermore, in the circuits described in the above-cited 
patent applications, the pulse duty cycle varies exponentially with 
musical scale position. The pulse train output on gate line 32 necessarily 
turns the switch 30 on each time there is a pulse, and off each time the 
pulse terminates. Since the pulse train is duty-cycle-modulated as an 
exponential function of musical scale position, the on-off duty cycle of 
switch 30 is similarly modulated, which is exactly what is required to 
provide an exponential control signal on line 29 to the VCF. 
To complete the discussion of FIG. 1, the inputs to the filter control 
drive circuit 36, although conventional, will be described. The modulation 
input on line 38 can be taken from either of two sources or shut off 
entirely by means of a panel switch 44. If the switch selects the off 
position (terminal 46), then there is no input to the drive circuit 36 on 
line 38. As a result, the effect of drive circuit 36 on the frequency 
response of the filter 26 will be unvarying throughout the rise and fall 
of the musical tone envelope signal generated by circuit 18A. On the other 
hand, in some types of electronic musical instruments, such as 
synthesizers, it may be desirable for the frequency response of the filter 
26 to vary over time as a function of a rising and falling envelope 
voltage. If this is desired, switch 44 can connect the drive circuit 
modulation input line 38 via line 20 to the same envelope generator 18A 
which shapes the musical tone output, or it can connect it over line 50 to 
a separate envelope generator 18B which has a different rise and fall 
characteristic from circuit 18A. The latter alternative permits the 
amplitude of the musical tone and its frequency characteristics to vary 
over time in different ways. The alternate envelope generator 18B is 
similar to the primary envelope generator 18A, except for component 
values, and responds in the same way to the signal derived over line 16 
from the priority note generator 12. 
Finally, there is a constant offset or reference voltage input on drive 
circuit input line 40. Internal circuitry within box 36 (to be described 
below) uses this offset voltage to determine whether the center frequency 
to which the filter 26 is tuned is the pitch of the selected musical note, 
or another pitch harmonically related to it. In the former case, the 
filter 26 is tuned to a center frequency equal to the pitch of the 
keyboard selected note, or, if one of the modulation inputs on line 20 or 
50 is selected by the switch 44, the initial value of the modulation 
voltage tunes the filter initially to that center frequency as the 
starting point for its frequency sweep. Alternatively, the internal 
circuitry of box 36 can use the offset reference voltage in such a way 
that the center frequency (or initial center frequency) of the filter is a 
selected musical interval above or below the frequency of the 
keyboard-selected note, thus affording further flexibility in the choice 
of harmonic effects. 
FIG. 2 shows a variant of the FIG. 1 circuit in which the priority note 
generator is not employed as a musical tone source. Thus its tone signal 
output (line 14 of FIG. 1) is not used at all in FIG. 2. Instead there is 
a voltage-controlled oscillator (VCO) 60, which supplies the required 
musical tone signal on an output line 14.1. In accordance with this 
invention, the VCO 60 is tunable to choose its operating frequency and 
thereby select the musical pitch called for by the keyboard. For tuning 
purposes, the VCO responds linearly to a current input on line 29.1. The 
latter input, however, using the technique described above in connection 
with FIG. 1, is made to vary exponentially with musical position of the 
selected note by means of an on-off switch 30.1 gated by a 
duty-cycle-modulated pulse train derived from the pulse code generator 
12A. The latter in turn derives the required note selection information in 
the form of a multi-bit digital code word arriving over cable 42 from 
priority note generator 12. The tuning signal chopped by the switch 30.1 
is derived over line 34.1 from a drive circuit 36.1. The latter in turn 
has available to it alternative modulation inputs connectable to line 38.1 
by switch 44.1. These are derived over lines 20.1 and 50.1 from envelope 
generators 18.1 and 18.3 respectively. The envelope generators in turn 
respond to the input on line 16 from the priority note generator 12. The 
third position of switch 44.1 (terminal 46.1) is the off position, which 
shuts off the modulation input otherwise applied to line 38.1. Thus it 
will be appreciated that exponential control of the VCO in FIG. 2 can be 
achieved in precisely the same way as exponential control of the VCF in 
FIG. 1. 
In order to correlate FIG. 2 with the explanation given above in connection 
with FIG. 1, note that identical circuit elements have been given the same 
reference numerals in both drawings, while those circuit elements which 
are not necessarily identical, but do perform corresponding functions, are 
given reference numerals in both drawings which are numerically similar 
except for the presence of a decimal suffix. 
Thus it will be appreciated that, while the priority note generator 12 is 
not employed as a tone source in this embodiment, it is still employed as 
a source of controlling inputs to the pulse code generator 12A and the 
envelope generators 18.1 and 18.3. As in FIG. 1, the pulse code generator 
12A provides the exponential control gate input to switch 30.1 for tuning 
the VCO to the proper pitch by chopping the input from drive circuit 36.1. 
If the latter is connected to one of the envelope generators 18.1 or 18.3 
by switch 44.1, then the VCO-generated musical pitch will sweep over time 
as a function of envelope voltage. Or if the modulation input is turned 
off by connecting switch 44.1 to terminal 46.1, the VCO frequency or pitch 
will remain constant. The constant offset input on line 40.1 is used by 
the internal circuitry of block 36.1 to determine whether the VCO pitch 
(or its initial pitch, if a frequency sweep is to be performed) is equal 
to, or offset harmonically from, the keyboard-selected pitch. 
Although the musical tone signal in FIG. 2 is derived from VCO 60 instead 
of from priority note generator 12, it is processed in the same way. Thus 
the musical tone signal on line 14.1 goes to a keyer 22.1. The latter is 
driven by a signal derived over line 20.1 from envelope generator 18.1, 
which is one of the envelope generators selectable by switch 44.1 for VCO 
frequency sweeping purposes. The musical tone output of the keyer 22.1 
goes over a line 24.1 to a filter 26.1 which processes the musical tone 
before passing it on to the audio output terminal 28. 
The filter 26.1 may be fixed, or it may be tunable by means of some 
technique other than that described in connection with FIG. 1, in which 
case the balance of the elements in FIG. 2 are unnecessary. But if filter 
26.1 is a linear VCF tunable by the exponential technique described above, 
then, as described in connection with FIG. 1, its tuning is governed by an 
exponential current control input on line 29.2. The latter is obtained by 
chopping a drive input arriving over line 34.2 from drive circuit 36.2. 
The chopping is performed by switch 30.2 under control of the 
exponentially duty-cycle-modulated pulse train on line 32 coming from the 
same pulse code generator which controls the VCO switch 30.1. The drive 
circuit 36.2 has a constant pitch offset input on line 40.2, and a choice 
of constant or varying pitch characteristics by means of switch 44.2. The 
latter either connects or disconnects one of two frequency-sweeping 
varying voltages available on lines 20.1 and 50.2 from envelope generators 
18.1 and 18.2 respectively. Envelope generator 18.1 provides VCF 26.1 with 
the same frequency control profile as VCO 60, whereas envelope generator 
18.2 provides separate control of the VCF and VCO. 
As seen in FIG. 3, each of the tuning control drive circuits 36 of FIGS. 1 
and 2 includes a summing circuit 100 which sums the modulating voltage (if 
any) on line 38 with the offset voltage on line 40, and provides the 
resulting sum output to a voltage multiplier 102. The multiplication ratio 
of the latter circuit is adjustable (by means subsequently described) to 
determine the pitch tracking interval between the VCO 60 or VCF 26 of 
FIGS. 1 and 2 and the musical note selected by the keyboard. The voltage 
output of multiplier 102 is then processed by a voltage-to-current 
converter circuit 104 to provide the VCO or VCF control drive signal on 
lines 34 of FIGS. 1 and 2. In this signal, as a result of the 
voltage-to-current conversion, the parameter which is linearly 
proportional to the sum of the inputs on lines 38 and 40 is current rather 
than voltage. 
A fuller schematic representation of the drive circuit 36 and filter 26 
appears in FIG. 4. The summer 100 comprises amplifier A1 and its resistor 
network R.sub.o, R.sub.I and R.sub.f. The voltage multiplier 102 comprises 
amplifier A2, resistor R.sub.A and potentiometer R.sub.B. The gain of A2 
is determined by the ratio of R.sub.A to R.sub.B. Adjusting the gain of 
amplifier A2 (by means of potentiometer R.sub.B) selects the 
multiplication ratio of the multiplier 102, which in turn determines the 
initial tracking interval or offset between the pitch to which the 
frequency-responsive circuit (VCO 60 or VCF 26) is tuned and the pitch of 
the keyboard-selected note. This initial interval is altered over time 
when one of the modulation voltages appears on line 38. The 
voltage-to-current converter 104 comprises amplifier A3, resistor RE and 
transistor Q1. The current output of the converter 104 on line 34 is the 
collector current of Q1, which equals the current through RE, and that in 
turn is proportional to the voltage output of amplifier A2. The switch 30 
chops the current output on line 34, under control of the exponentially 
duty-cycle-modulated pulse code input on gate line 32. 
The VCF 26 comprises two stages: a transconductance amplifier A4 and a 
unity gain FET buffer Q2; as is conventional (see Hutchins, cited above). 
The transconductance amplifier A4 is preferably an RCA integrated circuit 
type CA3080, and has a resistor network R6, R2 and R3. The unity gain 
buffer FET Q2 has a capacitor C2 and resistor R4. A feedback resistor R5 
couples the output of FET Q2 to the input of amplifier A4. The 
pulse-modulated output of switch 30 appears on line 29 and is directed to 
the gain control port of amplifier A4. The tone signal input on line 24 is 
coupled to the transconductance stage A4 through resistor R2. If desired, 
conventional fan-out techniques (such as those described by Hutchins, 
supra) may be employed to permit the signal on line 29 to drive more than 
one VCF and/or VCO. 
A simplified, but less accurate, version of the drive circuit 36 and switch 
30 is illustrated in FIG. 5. This circuit has the economic advantage of 
combining some functions in the same circuit components, but its accuracy 
is impaired by emitter-base offsets. Transistor Q3 and emitter resistor R8 
form a voltage-to-current converter (corresponding to circuit 104 of FIG. 
4). The offset voltage (corresponding to the input on line 40 of FIG. 4) 
here is determined by voltage divider R9, R7. The exponentially modulated 
pulse input on line 32 is applied through a diode D1 to the base of Q3, 
which also performs the function of switch 30 in FIG. 4. The modulation 
signal on line 38 is applied to the base of Q4, causing its collector to 
add the modulation signal to the offset voltage biasing the base of switch 
Q3. The control pulses on line 32 cut off Q3 when they go high, but enable 
it when they go low. In this circuit, the initial frequency tracking 
interval is set by potentiometer R8 in the emitter circuit of switch Q3. 
The output current for controlling the VCO 60 or VCF 26 appears on Q3's 
collector output line 29. 
Another variation of the circuit in FIGS. 3 and 4 is presented in FIGS. 6 
and 7. As first seen in simplified form in the block diagram of FIG. 6, 
this circuit places the voltage-to-current converter 104 downstream of the 
exponential control switch, here designated 30A. The switch 30A itself, 
moreover, as shown here schematically, has a single pole, double throw 
operating characteristic. As a result, cycling of the switch 30A 
alternately charges and discharges capacitor C1 through resistor R10. 
Since the source of the capacitor charging voltage is the output of the 
multiplier 102, the average DC voltage level on the capacitor is the 
product of the multiplier output voltage and the duty cycle of the switch 
30A. This capacitor voltage on line 29c, which now constitutes the VCO or 
VCF control signal voltage, is applied through a high impedance buffer 
stage 108 to the converter circuit 104 for conversion to a control signal 
current. The advantage of this FIG. 6 embodiment, compared to the 
embodiments of FIGS. 4 and 5, is that the FIG. 6 version does not require 
the internal circuitry of the controlled VCO 60 or VCF 26 to switch at 
high speed, because the chopped output of the switch 30A is now smoothed 
or integrated by capacitor C1 before reaching the VCO or VCF. A 
disadvantage, however, is that the ability of the VCO or VCF to track the 
control signal accurately now depends on the linearity of its internal 
circuitry. 
FIG. 7 shows the embodiment of FIG. 6 in greater circuit detail. The summer 
100, multiplier 102 and VCF 26 are the same as they are in FIG. 4. The 
exponential control switch 30A, preferably an integrated circuit of the 
CO4016 type, comprises two solid state switches SW1 and SW2 connected in 
parallel to the capacitor charging and discharging resistor R1, and 
controlled in common by the exponentially modulated pulse stream on line 
32. Switches SW1 and SW2 are operated inversely to each other, however, by 
virtue of the inverter I. Thus, when SW1 is on, causing the multiplier 
output voltage on line 34A to charge C1, SW2 is off to prevent discharge. 
Conversely, when SW2 is on, discharging C1 to ground, SW1 is off, to 
prevent charging. The capacitor voltage is coupled through the high 
impedance buffer 108, comprising operational amplifier A5, to 
voltage-to-current converter 104, which provides the tuning signal drive 
over line 29B to VCF 26. 
Circuit 104 here is also the same as it is in FIG. 4, with the optional 
exception of a modulation voltage input M' applied through a resistor 
R.sub.E ' to the emitter of Q1. A modulation input (e.g. one of the 
envelope signals available on line 38) can be applied to terminal M' 
instead of, or in addition to, the modulation input on line 38 applied to 
summer 100. When applied to terminal M', the modulation input produces 
constant deviation modulation; i.e. a given increment in the modulation 
voltage applied to terminal M' produces a given numerical deviation in the 
frequency (or musical pitch) to which the VCF 26 or VCO 60 is tuned by the 
signal on line 29. Such constant deviation modulation is to be 
distinguished from constant musical interval modulation; the latter is the 
effect produced by the modulation input on line 38 applied to the summer 
100. To appreciate this distinction, consider that, if a given increment 
delta V in the modulation voltage on terminal M' produces a change from F 
to 2F Hertz (a deviation of F Hertz) in the frequency to which the VCF 26 
or VCO 60 is tuned, then an increment of twice delta V in the modulation 
voltage on terminal M' will produce a pitch deviation at the VCF 26 or VCO 
60 which is twice as large in numerical frequency terms (i.e. a deviation 
of 2F, from F to 3F Hertz). The first pitch deviation (of F Hertz) will 
have a given musical interval value (in this case the change from F to 2F 
is one octave), but the second pitch deviation (of 2F Hertz), although 
twice as great in frequency terms, is not twice as great in musical 
interval terms (i.e. a change from F to 3F is not two octaves). Thus, an 
arithmetic increment in modulation voltage on terminal M' produces a 
merely arithmetic deviation in pitch, whereas to preserve musical interval 
values there must be a geometric change in pitch. A change in VCO or VCF 
pitch from F to 4F, not merely from F to 3F, is required to increase 
musical pitch by two octaves, which would be twice the musical interval 
change of one octave achieved by the initial increment in modulation 
voltage. The latter sort of change, which produces constancy of musical 
interval value instead of constancy of pitch deviation, is what happens 
when the modulation voltage is injected upstream of the multiplier 102 
(e.g. when it is applied over line 38 to summer 100), because then the 
change in modulation voltage (delta V) is multiplied by circuit 102, and 
not merely added. The result then is a geometric, rather than arithmetic, 
change in frequency at the VCF 26 or VCO 60, producing constant interval 
modulation. But when the modulation voltage is injected downstream of the 
summer 102, as at terminal M', then the effect is merely additive, and 
constant deviation modulation results. 
In the light of the circuit details portrayed in FIG. 7, the advantages of 
this embodiment may now be more fully appreciated. In the preceding 
embodiments, the input to the gain control port of the filter amplifier A4 
on line 29 is pulsed because it is derived directly from the output of the 
on-off switch 30 without any integration. In following the output of 
switch 30, the amplifier A4 must itself act as a switch, because every 
time the output of switch 30 goes to zero, the gain of amplifier A4 goes 
to zero also, i.e. the amplifier shuts off. The switching of the amplifier 
places difficult design constraints upon it. Of course, the switching on 
and off of the amplifier A4 does not appear in the audio output of filter 
26 appearing on line 28, because of the effect of capacitor C2 which 
integrates the output of amplifier A4 as it is applied to the input of FET 
Q2. Nevertheless, in the preceding embodiments such switching does occur, 
and it is something of a design problem. The problem is avoided by using 
capacitor C1 to integrate the output of switch 30A before that output 
reaches amplifier A4. As a result, the amplifier now merely follows the 
integrated average capacitor voltage, and need now switch completely off. 
The high impedance buffer 108 effectively couples the capacitor C1 to the 
amplifier A4 without providing a leakage path for capacitor discharge. 
This circuit, however, places greater reliance on the linearity of filter 
26 in order to track accurately the control signal on line 34B, because 
now all of the information in that signal is presented in analog form, 
i.e. a current level. Previously, at least part of the information in that 
signal was presented in digital form because of the exponential chopping. 
One more variation of the basic circuit is shown in FIG. 8, where the 
general arrangement is the same as in FIG. 4 except that switch 30, 
instead of coupling the V-to-I converter 104 to the filter 26, couples the 
first, or transconductance amplifier stage 26A of filter 26 to the second, 
or buffer stage 26B thereof. The output of the V-to-I converter 104 on 
line 34B goes directly to the filter tuning control (the gain control port 
of amplifier A4) in this embodiment, while the output of amplifier A4 on 
line 34C provides the input of switch 30. The switch output on line 29B 
then goes to the input to buffer stage FET Q2. 
Here again, the advantage of this embodiment is that it avoids the 
necessity for amplifier A4 to follow the on-off operation of the switch 
30. Instead, the input to the amplifier gain control port is a smoothly 
varying signal from the drive circuit 36 on line 34B. It is only the 
output of the amplifier A4 which is coupled to the switch 30. The pulsed 
switch output, instead of controlling the amplifier gain directly, is 
applied to the input of the second (or buffer) stage 26B of filter 26, 
where it is integrated by the capacitor C2. The only effect of switch 30 
on amplifier A4 is the effect produced by the integrated output of 
capacitor C2 via FET Q2 and feedback resistor R5. Thus, once again, 
amplifier A4 is spared the necessity of switching on and off, because its 
gain control voltage is never entirely cut off. 
The circuit of FIG. 2 differs from FIG. 1 in that it does not rely on the 
priority note generator as a tone source. But it does not entirely 
eliminate the priority note generator, since the latter circuit (and its 
component part, the pulse code generator) are used in FIG. 2 to develop 
the exponentially modulated pulse train. If it is desired to eliminate the 
priority note generator and pulse code generator entirely, then the 
arrangement of FIG. 9 may be used. In FIG. 9 (where the same numbering 
scheme as in FIG. 2 is used) once again VCO 60 is used (instead of a 
priority note generator) to generate the musical tone signal on line 14.1. 
Once again that signal is processed by filter 26.1 before appearing at 
audio output terminal 28. And once again, one or both of the circuits 60 
and 26.1 is of the so-called voltage controlled type (VCO and/or VCF) 
which responds linearly to tuning control current inputs on lines 29.1 
and/or 29.2 respectively. Once again these inputs are derived from drive 
circuits 36.1 and/or 36.2 respectively, and chopped exponentially by 
switches 30.1 and/or 30.2 respectively. But here the exponentially 
duty-cycle-modulated pulse train on line 32, which gates the switches, is 
derived from a one-shot (monostable multivibrator) circuit 200. Each pulse 
in the train represents one cycle of the one-shot 200. The latter is 
repeatedly cycled because it is driven by a free-running oscillator 202. 
Thus the pulse train on line 32 consists of a stream of one-shot output 
pulses occurring at a fixed repetition rate equal to the frequency of 
oscillator 202. In order to modulate the duty cycle of this pulse train, 
the pulse duration must be controlled. This is accomplished, in the 
required exponential fashion, by an exponential scaling circuit 204. The 
latter applies a voltage over line 206 to the one-shot 200, and the 
magnitude of this voltage determines the one-shot cycle time (in a manner 
to be described). The magnitude of the voltage, in turn, varies 
exponentially as a function of which one of the musical note selection 
lines 10 is energized by the keyboard (here designated 10a) of the musical 
instrument. Thus, as the musician indicates his note selections by 
pressing successive keys on the keyboard 10A, the note selection 
informaion represented by lines 10 causes scaling circuit 204 to modulate 
the voltage on line 206. This in turn varies the cycle time of one-shot 
200, resulting in modulation of the duty cycle of the pulse train on line 
32 which gates the switches 30.1 and 30.2. 
The switches in turn chop the VCO and VCF control signals derived from the 
drive circuits 36.1 and 36.2. Each of these drive circuits has a choice of 
modulation inputs on lines 38.1 and 38.2, determined by the positions of 
switches 44.1 and 44.2 as described above, in connection with FIGS. 1 and 
2. One of these inputs, the special effects input on line 20.1 or 20.2, is 
derived from envelope generator 18.1 or 18.2 respectively, just as in 
FIGS. 1 and 2. Here, however, since there is no priority note generator, 
the drive line 16 for the envelope generators 18 is energized directly 
from the keyboard 10A by means of a switch which closes when any of the 
musical keys is operated. 
FIG. 10 shows in detail how the exponential scaler 204 and one-shot 200 
cooperate to produce exponentially scaled pulse duration modulation. The 
pulse duration timing network of the one-shot 200 is an RC circuit 
including capacitor 210 in the one-shot circuit and a selected one of n 
different-valued resistors 212 in the scaling circuit 204. When one of the 
n keyboard keys is operated, its corresponding line 10 is energized. The 
keyboard lines 10 are connected to the respective gate electrodes of n 
switches S (preferably the same solid state type of switch as the switch 
30 described above). The selected one of the switches S is turned on by 
the gate voltage on its associated line 10. As a result, the supply 
voltage is connected through the one enabled switch S to its associated 
resistor 212. However, the process of charging the one-shot timing 
capacitor 210 does not begin at this time, because the capacitor is 
connected through a diode 214 to a ground (not shown) internal to the 
flip-flop circuit 216, which is initially reset. 
When the next cycle of the oscillator 202 activates the one-shot circuit 
200, however, a spike is coupled to the Set input of the flip-flop through 
a differentiating network comprising series capacitor 218 and shunt 
resistor 220. This spike sets the flip-flop 216, causing its Q output to 
go high, back-biasing diode 214 and thereby blocking the discharge path of 
timing capacitor 210. At this point, charging of the timing capacitor 210 
begins. The charging rate of course depends on the value of the particular 
one of the resistors 212 which is in use, which in turn depends on which 
switch S is enabled, and that is a function of which line is selected by 
the keyboard. Thus, the capacitor charging rate is a function of musical 
note selection. 
After the passage of a period of time, the duration of which depends on the 
capacitor charging rate, the capacitor voltage rises to the level of a 
reference voltage on supply terminal 222. This condition is then detected 
by a comparator 224, which thereupon produces a positive output to reset 
the flip-flop 216, terminating the one-shot cycle, and allowing the timing 
capacitor 210 once again to discharge rapidly to ground through the Q 
terminal of the flip-flop. During the time that the flip-flop is set, a 
high Q output appears on the one-shot output line 32; that is the pulse 
which serves to gate the switches 30 in FIG. 9. When the flip-flop 216 is 
reset, the Q output on line 32 goes low, terminating the pulsed gate drive 
to switches 30. The duration of each Q output pulse on line 32 is equal to 
the flip-flop set time, which is longer when the timing capacitor 210 
takes a longer time to charge (i.e. when a higher value resistor 212 is 
selected). 
Thus the duration of the switch gate pulses on line 32 is a function of the 
value of the selected resistor 212. All that remains, then, to make the 
pulse duration vary exponentially with musical scale position, is to 
select the values of resistor 212 so that they vary exponentially from 
step to step along the musical scale. Thus, when a low note is selected by 
the keyboard, a low-valued resistor 212 is used to charge the timing 
capacitor 210 quickly, and the duration of the one-shot output pulses (see 
waveform 226) is brief. Conversely, when a high note key is operated, a 
high-valued resistor 212 is used to charge the timing capacitor 210 
slowly, and the duration of the one-shot output pulses (see waveform 228) 
is longer. In waveforms 226 and 228 the corresponding pulses start at the 
same time, i.e. times t.sub.1, t.sub.2, t.sub.3, so that their repetition 
rate (governed by the frequency of oscillator 202) is the same. But their 
durations are different, and scaled in proportion to the values of 
resistors 212, that is, exponentially. The resulting exponentially 
modulated pulse train is equivalent to that produced in FIGS. 1 and 2 for 
VCO and VCF control purposes, but does not require a digital priority note 
generator or pulse code generator. 
It will be appreciated that the various embodiments of this invention 
merely exemplify a fundamental technique of VCO or VCF control, by 
providing an exponentially modulated switching function to chop a control 
signal which tailors the frequency characteristics of the circuit 
according to the inherent nonlinear nature of musical pitch relationships, 
and that it is this technique which is broadly stated in the appended 
claims.