Long range police radar warning receiver with multiple array averaging

A police radar warning receiver includes a DSP circuit having a correlator, multiple array averager and a peak detector to provide after each sweep of a swept local oscillator respective dynamic thresholds associated with each averager array and against which information from that sweep is evaluated to determine whether to generate an alarm enable indicative of receipt of a police radar signal. The multiple averager arrays allow for different averaging or weighting ratios to maximize sensitivity of the receiver on the one hand and enhance response time on the other rather than trading off one for the other.

BACKGROUND OF THE INVENTION 
I. Field of the Invention 
The present invention is directed to an improvement in vehicle-mounted 
police radar warning receivers, and more particularly to such receivers 
which detect police radar signals before the vehicle is within the 
detection range of the police radar unit. 
II. Description of the Prior Art 
Police radar generally operates in the X-band and K-band of the frequency 
spectrum as discussed in U.S. Pat. No. 4,313,216, assigned to Cincinnati 
Microwave, Inc., the assignee herein. There are, generally, two types of 
police radar. One emits a continuous radar signal so long as the radar 
unit is turned on. The other emits a brief burst of radar signal when the 
police officer triggers the unit. This latter type is referred to as 
pulsed or instant-on radar. While transmitting, both continuous and pulsed 
radar transmit a signal which is at a fixed frequency within the selected 
band. 
An electronic assembly referred to as a police radar warning receiver has 
been devised to detect the presence of police radar signals. The receiver 
is mountable in a vehicle, such as a passenger car or truck, motorcycle, 
boat or the like, which travels on land or water in areas subject to 
speed-monitoring radar surveillance by police, and functions to detect the 
presence of the police radar and provide the driver or user with an 
audible and/or visual indication that his speed is being checked by radar. 
The receiver is contained in a box-like housing which is set on the dash 
or clipped to the visor in the vehicle. Extending from the rear of the 
housing may be a power cord which terminates in a plug adapted to be 
received in the cigarette lighter socket of the vehicle. The front panel 
of the receiver faces the driver and has various indicators and control 
knobs. 
When police radar is operating within range of the radar warning receiver, 
the circuitry of the receiver is able to detect the presence thereof. The 
ESCORT and PASSPORT radar warning receivers, manufactured by the assignee 
herein, Cincinnati Microwave, Inc. of Cincinnati, Ohio utilize a 
superheterodyne circuit for this purpose. As explained in aforementioned 
U.S. Pat. No. 4,313,216, and in U.S. Pat. Nos. 4,581,769 and 4,750,215, 
which are also assigned to the assignee herein, a superheterodyne circuit 
employs two local oscillators, one of which sweeps in frequency over a 
range of frequencies related to one or both radar bands. To this end, a 
first local oscillator signal is mixed with the incoming police radar or 
other signal to produce a first IF signal. The first IF signal is then 
mixed with the second local oscillator signal to produce a second IF 
signal. Due to the sweep of one of the local oscillators, the second IF 
signal presents a sweep pattern which extends over a band corresponding to 
the X and K bands and including noise and signals, the latter at locations 
in the sweep pattern corresponding to the frequency of received signals in 
the X and/or K bands. 
The sweep pattern of the second IF signal is then passed through an FM 
discriminator circuit. The FM discriminator provides a second sweep 
pattern output including S-curves having positive- and negative-going 
portions to define time-related positions relative the start of the sweep 
corresponding to the frequency at which an incoming signal is received. As 
is well understood, such a heterodyning process will result in generation 
of a "duplicate" or image of the police radar signal within the receiver. 
Hence, the discriminator generates one S-curve related to the actual 
frequency signal received as well as a second S-curve related to the image 
frequency of the received signal. One or both of these S-curves may then 
be utilized to indicate reception of a police radar signal so as to alert 
an operator to the presence of police radar surveillance as described in 
aforementioned U.S. Pat. Nos. 4,581,769 and 4,750,215, and U.S. Pat. No. 
4,862,175, also assigned to the assignee hereof. 
The ability of radar warning receivers to detect police radar signals, 
however, is limited primarily by the sensitivity of the receiver 
electronic circuitry which defines a capture range. That is, signals 
emitted by police radar units may travel a substantial distance from the 
unit. As is well understood, the police radar signal must travel to the 
vehicle under surveillance and then return altered by a Doppler shift 
representing speed of the vehicle. However, as the police radar signal 
travels to and from the vehicle under surveillance, the weaker the signal 
becomes. Thus, the further the vehicle under surveillance is from the 
police radar unit, the weaker the return radar signal is such that at some 
distance and beyond, the police radar signal is too weak to return to the 
police radar unit and be evaluated for speed of the vehicle (detection 
range). 
It is desirable that the radar warning receiver detect the police radar 
signal while it is still so weak as to be beyond the detection range of 
the police radar unit. However, as with the police radar unit, the further 
the police radar warning receiver is from the source of the police radar 
signal, the weaker the signal. At some distance from the police radar unit 
corresponding to the capture range of the police radar warning receiver, 
the police radar signal may be so weak that the police radar warning 
receiver is unable to distinguish signal from noise, meaning that a police 
radar signal will not be detected until the vehicle moves closer to the 
police radar unit. The difference in distance between the detection range 
of the police radar unit and the capture range of the police radar warning 
receiver defines a reaction zone during which the vehicle operator must 
react to the presence of a police radar signal. It is desirable that the 
reaction zone be as large as possible so that the operator of the vehicle 
under surveillance will have sufficient time to react before the vehicle 
comes within the detection range of the police radar unit. 
Additionally, some police radar units are of the "instant-on" type meaning 
that they may be used in a manner to intermittently emit only short bursts 
of police radar signals. Where the bursts are given only infrequently, the 
first burst may be given when the police radar warning receiver is too far 
away to detect that burst, i.e., at that distance, the signal from the 
police radar unit is below the threshold of the receiver. The second burst 
may come after the vehicle is within the detection range of the police 
radar unit. Under such circumstances, the operator will have had no 
advance warning that the vehicle is under surveillance. Accordingly, it is 
desirable to extend the capture range of the police radar warning by 
providing the receiver with as much sensitivity, i.e., as low a threshold, 
as possible so that police radar signals may be received as far from the 
police radar unit's detection range as possible. 
Typical of many radar warning receivers is that their sensitivity is 
generally low enough to be able to detect most police radar signals 
somewhat beyond the detection range of the associated police radar unit 
and, thus, provide a reaction zone. However, greater improvement is 
desired. One approach could be simply to select a lower threshold above 
which all signals are accepted as valid, thus extending the capture range. 
However, this approach may allow too much random noise to pass through the 
receiver circuitry and appear as police radar signals resulting in 
irritating and misleading false alarms. Another approach has been to 
continuously vary the threshold based on the level of random noise as a 
sweep of the local oscillator progresses. However, in such an approach, 
the threshold may be varied part way through the sweep but before a weak 
police radar signal is about to be received. Thus, if the threshold were 
caused to increase as the sweep progressed, the vehicle may be too far 
away to detect a weak police radar signal. As any signal not above the 
threshold is discarded, a police radar signal may be missed. Thus, rather 
than increase the range of the receiver, such an approach may actually 
reduce the capture range capabilities thereof, because the sensitivity of 
a receiver with such a varying threshold may be effectively, but 
undesirably, reduced. Alternately, the threshold may be caused to decrease 
to such a level that noise may lead to undue false alarms. 
III. Description of Prior Applications 
In my prior applications, Ser. Nos. 07/558,668; 07/481,509; and 07/421,525, 
the disclosures of which are incorporated herein by reference as if set 
out fully herein, I explained that sensitivity could be enhanced by 
processing the sweep pattern provided by digital correlation of the FM 
discriminator output and then peak detecting to evaluate the processed 
sweep pattern based upon a cumulative history of processed sweep patterns. 
The result was to average data over a plurality of sweeps and after each 
sweep generate a threshold level unique to the information content level 
of signals received over a plurality of sweeps whereby to adjust the 
sensitivity of the police radar warning receiver for maximum capture range 
under the circumstances. In that way, the capture range was improved over 
prior art police radar warning receivers while reducing the risk of 
missing a weak signal or causing undue false alarms which might result 
from continuously varying the threshold during a sweep. 
As more fully described in my prior applications, during each sweep of the 
local oscillator, the FM discriminator output is digitally sampled at 
successive sample intervals to generate a series of digital sample words 
representative of the sweep pattern produced by the FM discriminator. 
Thus, the magnitude of each digital sample word corresponds to the 
magnitude of signals and noise received at the X- and/or K-band 
frequencies to which the receiver is tuned at the time the sample is 
obtained. As each sample word is generated, it is manipulated in the 
digital correlator by correlating each digital sample word and several of 
its predecessors in that sweep with a complex correlation function 
representative of the FM discriminator response to produce a series of 
complex digital correlator words having improved signal-to-noise ratio as 
compared to the sample words. The complex digital correlator words so 
produced are coupled to an averager which separately accumulates and 
averages for each sample interval or group of intervals the complex 
digital correlator word(s) generated in the same sample interval(s) over a 
plurality of sweeps of the local oscillator. The series or array of 
complex averager words thus obtained are converted to magnitude and 
represent the RF signal energy received in each of the sample intervals of 
the sweep. After each sweep, a peak detector calculates the mean square 
value of the series of the complex digital averager words whereby to 
provide a dynamic threshold for that sweep as affected by all of the 
previous sweeps. Also, after each sweep, the digital averager words in the 
array are examined by the peak detector against the current dynamic 
threshold whereby any averager word larger than the dynamic threshold is 
indicative of receipt of a police radar signal in that sample interval. 
While the foregoing is believed generally to provide improved sensitivity 
over prior radar warning receivers, there was found to be an associated 
trade-off between response time and sensitivity based upon the ratio of 
averaging used. While it was desired to retain as much of the information 
from the prior sweep as possible for averaging, the optimum trade-off was 
believed to be a 0.9/0.1 weighting factor or ratio, where 10% of the value 
of the complex digital correlated word from a sample interval was to be 
added to 90% of the prior averaged value for that sample interval time 
from prior sweeps. As the ratio increased, sensitivity would likewise 
increase but response time would also be degraded. The substitution 
technique for large signed values described in my application Ser. No. 
07/481,509 helped improve response time but further improvements were 
desired. 
SUMMARY OF THE INVENTION 
The present invention provides improved circuitry for and operation of a 
radar warning receiver which provides both enhanced sensitivity and 
improved response time so as to increase the available reaction zone 
without some of the drawbacks of threshold adjustment approaches of the 
prior art and without the tradeoff associated with my prior single 
averager array process. More specifically, I have discovered that 
providing a plurality of averager arrays, utilizing the same digital 
correlation words but different weighting ratios, allows for optimization 
of sensitivity and response time. To this end, and in accordance with the 
principles of the present invention, two or more sets of arrays of 
averager words are obtained from the same complex correlation words, each 
set based upon different weighting factors to optimize particular aspects 
of radar warning receiver performance. With two such sets, for example, 
one array may be based upon a very large weighting of old to new such as 
0.95/0.05 or 0.98/0.02 for extremely good sensitivity, while the other 
array may have a lower weighting of old to new such as 0.50/0.50 to 
provide good response time in those areas where the first array is not as 
efficient. After each sweep, a dynamic threshold level may be calculated 
and peak detection accomplished for the first (sensitivity) array just as 
previously described in my prior applications. If an averager word in that 
array exceeds the associated dynamic threshold value, a police radar 
signal may have been received. However, if the sensitivity array does not 
yet indicate presence of a police radar signal, the reason may be due to a 
tradeoff in response time resulting from setting the first array weighting 
ratio for such good sensitivity. In accordance with my present invention, 
the second (response time) array is provided to enhance response time 
rather than sensitivity to eliminate the drawback associated with the 
single array tradeoff of my prior applications. To this end, the second 
array may similarly be evaluated for a dynamic threshold and then peak 
detected to determine whether there has been receipt of a police radar 
signal notwithstanding that no such signal was as yet indicated from the 
sensitivity array. Consequently, the highly averaged (larger weighting 
factor) sensitivity array will provide the desired improved sensitivity 
but the drawback of a tradeoff in response time is reduced or eliminated 
by the lightly averaged (lower ratio weighting) response time array. More 
than two averager arrays, each with a different weighting factor, may be 
utilized to provide the best response time for a segment of the total 
response time curve of the radar warning receiver. 
By virtue of the foregoing, there is thus provided a radar warning receiver 
which has increased sensitivity along with enhanced response time relative 
to prior art receivers, and in which the capture range is maximized in 
view of the signal and noise conditions existing at the time a signal is 
received. Consequently, the radar warning receiver of the present 
invention is adapted to alert operators of a vehicle to the presence of 
police radar surveillance with a desirably large reaction zone, while 
reducing the risk that a weak police radar signal will be overlooked or 
false alarms will occur too often and be annoying or irritating. 
These and other objects and advantages of the present invention shall be 
made apparent from the accompanying drawings and the description thereof.

DETAILED DESCRIPTION OF THE DRAWING 
With reference to the FIGURE, there is shown a block diagram of a radar 
warning receiver 5 according to the principles of the present invention. 
Pulsed or continuous signals occurring in the X-band or K-band are 
received by an antenna system 6 which includes horn antenna 8 coupled to 
first mixer 10 to which the signals received by horn antenna 8 are 
directed. Horn antenna 8 may include filters for preventing signals at 
unwanted frequencies from being passed to first mixer 10, and it may 
further include other filters for preventing outward radiation therefrom 
of internally generated radio frequency signals. Coupled to first mixer 10 
is a variable frequency first local oscillator 12 adapted to be swept in 
frequency over a range of frequencies as described in aforementioned U.S. 
Pat. No. 4,313,216, the disclosure of which is incorporated herein by 
reference. First mixer 10 and first local oscillator 12 may be built into 
and made a part of antenna system 6 as provided in the ESCORT or PASSPORT 
radar warning receivers available from Cincinnati Microwave, Inc., the 
assignee herein, the latter as described in U.S. Pat. No. 4,613,989. The 
disclosure of said U.S. Pat. No. 4,613,989 is incorporated herein by 
reference. The oscillator disclosed in U.S. Pat. No. 4,583,057, also 
assigned to the assignee herein, may be utilized for local oscillator 12. 
The disclosure of said U.S. Pat. No. 4,583,057 is incorporated herein by 
reference. 
The frequency of the oscillator signal generated by local oscillator 12 
varies, i.e., it is swept, across a predetermined band of frequencies by a 
sweep generator 13 which causes the local oscillator signal to vary from a 
first frequency to a second frequency in a controlled manner in response 
to receipt of a sync pulse on sync out line 14. The varying frequency 
signal generated by local oscillator 12 is mixed with received signals 
from horn antenna 8 in first mixer 10 to generate first IF signals which 
are coupled to first IF amplifier 15. The output of amplifier 15 is 
coupled to a second mixer 16 such as a SAW mixer whereat the first IF 
signals are mixed with a fixed frequency oscillator signal from second 
local oscillator 17 to produce second IF signals which are amplified by 
second IF amplifier 18. The frequency of oscillator 17 is selected so that 
during the sweep of oscillator 12, any received signal in the desired 
band(s) mixed with the signal from oscillator 12 will generate two IF 
signals, one being a primary and one being an image as is well understood. 
The output of amplifier 18 is coupled through bandpass filter 20 and 
limiting second IF amplifier 21 to frequency discriminator circuit 22. The 
output of discriminator 22 is coupled to low pass filter 23 which 
preferably has a corner frequency of less than 8 KHz and more preferably 
about 6 KHz. Operation of the foregoing is generally as described in 
aforesaid U.S. Pat. No. 4,313,216, although the sweep rate preferably is 
doubled to 16 Hz (and the sweep period reduced to 62.5 ms) to further 
improve response time. 
FM discriminator 22 is preferably part of an FM receiver IF 24 (such as an 
LM3089 available from National Semiconductor Corporation, Santa Clara, 
Calif.; or an NE604 available from Signetics Corporation, Sunnyvale, 
Calif.) and includes an AM detector 25 driven by the same IF output from 
amplifier 21 to produce another analog signal on output 26 corresponding 
to the strength of the signal received at the frequency to which receiver 
5 is tuned at that time. 
The output of low pass filter 23 is coupled to a first input of analog 
multiplexer 27. Similarly, the AM output 26 of IF 24 is coupled to a 
second input of analog multiplexer 27. In a first state of multiplexer 27, 
as determined by the state of select signal 28, filter 23 output is 
coupled to analog-to-digital (A/D) converter 29 whereas, in a second state 
of multiplexer 27, output 26 is coupled to A/D converter 29 for purposes 
to be described hereinafter. Low pass filter 23 also has gain in the pass 
band which is set such that the noise level from frequency discriminator 
22 is nearly full-scale into A/D converter 29. A/D converter 29 includes a 
sample hold storage buffer which stores the input signal in response to a 
clock A/D signal 30. 
In the first state of multiplexer 27, A/D converter 29 produces a digital 
word corresponding to the level of the output from FM discriminator 22 at 
any given instant of time. As each sweep of oscillator 12 progresses, A/D 
converter 29 samples the FM discriminator output at sample intervals as 
dictated by clock A/D pulse 30 to produce a series of digital sample 
words, each corresponding to the magnitude of RF energy received by 
antenna 8 at the frequency or frequencies to which receiver 5 is tuned by 
oscillator 12 at that interval. 
Preferably, a sample rate of about 119 KHz is used to produce 7,486 digital 
sample words in each sweep, each sample word preferably comprised of six 
bits. As they are generated, sample words from the output of A/D converter 
29 are sequentially coupled to DSP circuit 32, such as a DSP56000 or 
DSP96002 integrated circuit available from Motorola Semiconductors, Inc. 
DSP circuit 32 includes a control section 34 which causes the DSP circuit 
32 to perform a series of programmed instructions stored in program memory 
36 in sequence with pulses from system clock 38 by which to control timing 
and communication within DSP circuit 32, as is well understood. Control 
section 34 also causes generation of the pulse 14 to sweep generator 13 
which initiates a sweep of local oscillator 12 such that operation of DSP 
circuit 32 is synchronized to the sweep. Control section 34 further causes 
generation of the appropriate select 28 followed by a conversion request 
(clock A/D 30) to converter 29 by which to determine the nature and timing 
of signals input to DSP circuit 32. DSP circuit 32 is programmed in 
accordance with the principles of the present invention to advantageously 
maximize the sensitivity of receiving and minimize response time under the 
signal and noise conditions prevailing at the time, as will now be 
described. 
Each digital sample word generated at each sample interval is coupled to a 
software recursive infinite impulse response (IIR) low pass filter 40 to 
produce a filtered sample word for each sample word generated. Low pass 
IIR filter 40 has at least a second order s-domain low pass transfer 
function as follows: 
##EQU1## 
where d is the filter damping function (preferably equal to about 
1.414214), and f.sub.c is the filter cutoff frequency (point at which the 
filter gain is equal to l/d--preferably equal to about 350 Hz). 
Alternatively, filter 40 could be a band pass IIR filter having the 
transfer function: 
##EQU2## 
The output of IIR filter 40 is coupled to complex correlator 42 which 
utilizes a Discrete Fourier Transform to correlate the current filtered 
sample word and the previous 127 filtered sample words with a series of 
complex or correlation function words representative of the response of FM 
discriminator 22 to provide improved signal-to-noise ratio, as is well 
understood. The preferred correlation function is 
##EQU3## 
although other correlation functions may be used. 
Correlator 42 preferably correlates the filtered sample words according to 
the formula (3): 
##EQU4## 
where N=128 points, L=point index, T=1/sample rate (e.g., =1/119791), so 
as to produce on its output a complex correlator word as each new filtered 
sample word is input thereto. It will be appreciated that as each sweep 
first progresses, there will be fewer than 127 filtered sample words 
available for purposes of the correlation, i.e., there will be several 
missing words during the earliest portion of the sweep. As each sample 
interval occurs in the sweep, more of the missing words will be supplied, 
but until they are, they are assumed equal to zero. 
To reduce the number of digital words to be dealt with by the remainder of 
DSP circuit 32, it is preferred to decimate the complex correlator words 
generated by correlator 42. To this end, the complex correlator words 
produced by correlator 42 are coupled to a complex IIR low pass filter 44 
including a pair of IIR filters identical to IIR filter 40 (one for the 
real components and one for the imaginary components of the complex 
correlator words). The outputs of complex IIR low pass filter 44 are 
decimated by sixteen in decimator 46 whereby to produce approximately 467 
complex digital decimator words through the course of the sweep containing 
the pertinent contents of the data in the original 7,400 plus complex 
words. Preferably, the function of decimator 46 is provided merely by 
providing only every sixteenth complex word output of filter 44 to 
averager 48, whereby each decimator 46 output represents primarily the 
magnitude of RF signal energy within the group of 16 undecimated sample 
intervals (i.e., the frequency segment). The contribution of RF noise in 
each frequency segment is minimized as a result of correlator 42. The 
foregoing is generally as described in my prior applications referred to 
hereinabove although the number of sample points and decimation rate is 
altered for the doubled sweep rate. 
In order to process the data as desired, each sweep is broken into about 
400 segments representing continuous segments of frequency within the RF 
band(s) of interest, with each segment assigned a set of index numbers 
related to the number of averager arrays to be utilized. As each decimator 
word is generated for each index, a plurality of averager words are 
calculated based upon a plurality of weighting ratios, resulting in a 
plurality of averager word arrays for each sweep. In the embodiment 
described herein, two such arrays are provided. Hence, two averager words 
are computed for each decimator word to provide a first array which 
enhances sensitivity of the receiver 5 and a second array which minimizes 
response time of receiver 5, respectively. To this end, as each decimator 
word is generated, an averager word of the first array is computed by 
sensitivity averager 48 utilizing a high weighting ratio of old to new to 
maximize sensitivity and an averager word of the secondary array is then 
computed by response time averager 49 utilizing a lower weighting ratio of 
old to new to minimize response time. Consequently, two arrays of averager 
words will be produced for each sweep of Osciallator 12. For purposes of 
the two array embodiment, each frequency segment is assigned an associated 
pair of index numbers I.sub.s and I.sub.R for the sensitivity averager 
array and the response time averager array, respectively. More arrays may 
be used and the index numbers Il, I.sub.2...I.sub.n may be assigned where 
I is the index or frequency segment and n indicates the array number. 
With respect to sensitivity averager 48, as each decimator word is 
generated, averager 48 causes a small percentage of the decimator word 
value (such as not more than about 5% or, preferably, as little as 2%) to 
be stored in a memory location associated with its respective sensitivity 
index number (I.sub.S) in a page of a supplemental memory 50 exterior of 
DSP circuit 32, along with a very large percentage (such as 95% or, 
preferably, as much as 98%) of the value previously stored in that same 
location. Similarly, with respect to response time averager 49, as each 
decimator word is generated, averager 49 causes a relatively large 
percentage of the decimator word value (preferably at least about 50%) to 
be stored in a memory location associated with its respective response 
time index number (I.sub.R) in the page of supplemental memory 50, along 
with a relatively large percentage (preferably about 50%) of the value 
previously stored in that same location. In each case, of averagers 48 and 
49, the percentages of old and new preferably equal 100%. To facilitate 
the averager processes and for other purposes to be described, memory 50 
communicates over bidirectional bus 52 with averagers 48 and 49, A/D 
converter 29, and peak detector 54, as needed in response to control 34 
and program 36 of DSP circuit 32. 
To conserve memory capacity, I have also discovered that only a portion of 
the second array averager words needs to be stored in memory 50. To this 
end, in the first array, the averager words are 24 bits long (stored, for 
example, in three 8 bit words in memory 50) to provide the desired 
arithmetic precision. However, such precision is not necessary for the 
lower weighting situation as there is less signal "build-up" with each 
sweep. Consequently, I have found that only the highest 16 order bits of 
the 24 bit averager words calculated for the second array need to be 
stored in memory locations I.sub.R (such as in two 8 bit words) thereby 
reducing the amount of extra memory required for the second array by about 
1/3 as compared to the first array. 
The resulting accumulated or average value in each location I.sub.R and 
I.sub.S of memory 50 is a complex digital word referred to herein as an 
averager word, there being two averager word locations for each frequency 
segment. Thus, as the very first sweep progresses, the two memory 
locations for each index number will start with zero and thereafter a 
digital word having a value preferably equal to 2% of the associated 
decimator word will be added to location I.sub.S and a digital word having 
a value preferably equal to 50% of the associated decimator word will be 
added to location I.sub.R. Thereafter, in each sweep, the averager word in 
each memory location I.sub.S will be updated or accumulated by averager 48 
as described above (2% new, 98% old) and the averager word in each memory 
location I.sub.R will be updated or accumulated by averager 49 as 
described above (50% new, 50% old). Also as each complex averager word is 
determined, a value corresponding to the averager word is determined 
(e.g., by the sum of squares) to produce associated arrays of real 
averager words. The averager magnitude words for each array during each 
sample interval are also stored in memory locations associated with their 
respective index numbers I.sub.R and I.sub.S, such as on a second page of 
memory 50. 
The magnitude of the words stored in locations I.sub.S will tend to 
increase in response to received RF signals in that segment of frequency 
while tending to further minimize and even cancel contribution from RF 
noise in that segment whereby to improve signal-to-noise ratio and, hence, 
the sensitivity of receiver 5. Due to the averaging process with a large 
weighting ratio, however, it is possible that relatively strong police 
radar signals may not generate sufficient contribution to the averaging 
process to drive the averager magnitude in the first array above the 
associated dynamic threshold as quickly as would be desired, i.e., 
response time to relatively strong police radar signals may be too long. 
Transient response of receiver 5 is thus delayed by the sensitivity array 
averaging process suggesting that a different ratio of old to new should 
be used in the averager. That is, more of the new should be added to less 
of the old. However, the signal-to-noise ratio of the averaging process is 
adversely affected by increasing the contribution of new sweep values. 
Thus, rather than increase the contribution of new correlator words, it is 
preferred to reduce same. The first array averaging process does so but at 
the expense of response time. 
A solution to the transient response problem brought about by the averaging 
process was described in my prior application, Ser. No. 07/481,509, 
involving substitution of selected averager magnitude words with the 
magnitude of the decimator words under certain circumstances related to 
strong signals. However, I have found that the expediency of a second (or 
multiple) averager array with a weighting factor selected to enhance 
response time rather than sensitivity will provide better results than the 
substitution technique of that prior application. 
After each sweep of oscillator 12 is concluded, the contents of 
supplemental memory 50 related to the arrays are evaluated by peak 
detector 54 to determine for that sweep a dynamic threshold associated 
with each averager array and to locate received signals in the sweep which 
may be indicative of a police radar signal at antenna 8. In this regard, 
peak detector 54 calculates a digital threshold word representing the mean 
square magnitude of the first array of averager words stored in 
supplemental memory 50 by summing the real components (magnitude squared) 
in all of the associated averager magnitude word memory locations I.sub.S, 
dividing the sum by the number of memory locations involved, and 
multiplying by a constant K.sub.S. The constant K.sub.S is determined 
empirically based upon a tradeoff between sensitivity and false alarm rate 
and is equal to 14 in a preferred embodiment. Similarly, peak detector 54 
calculates a digital threshold word representing the mean square magnitude 
of the second array of averager words stored in supplemental memory 50 by 
summing the real components (magnitude squared) in all of the associated 
averager magnitude word memory locations I.sub.R, dividing the sum by the 
number of memory locations involved, and multiplying by a constant 
K.sub.R. K.sub.R could be higher than K.sub.S to account for the lower 
signal-to-noise ratio of the second array as compared to the first array, 
however, in a preferred embodiment, K.sub.R is also equal to 14. 
Peak detector 54 then examines the magnitude squared in each of the 
above-described second page of memory locations I.sub.S and I.sub.R of 
supplemental memory 50 to determine whether the magnitude of the averager 
word in an array for any segment exceeds the dynamic threshold associated 
with that array. Theoretically, whenever the magnitude squared (real 
component) of any word in an averager array exceeds the dynamic threshold 
associated with that array, a police radar signal may have been received. 
Accordingly, if any such segment meets the associated criterion (its 
averager word value exceeds the associated dynamic threshold), an alarm 
enable signal may be output from DSP circuit as at 56 to remaining 
circuitry 58 to generate audible and/or visually perceptible alarms as 
desired to alert an operator of the vehicle (not shown) to the presence of 
police radar surveillance. 
For very weak police radar signals, the first array averager process will 
provide receiver 5 with enhanced sensitivity to allow detection of such 
signals with a very large capture range. Similarly, for very strong police 
radar signals, where the first array averager process would be too slow to 
respond, the second array averager process is provided for enhanced 
capture range. Further averager array processes may be provided to improve 
capture range under various signal conditions. As a consequence, both weak 
police radar signals and strong police radar signals will be detected with 
as much capture range as is believed possible and without trading off 
performance for one type of condition (e.g. weak signal) for another (e.g. 
strong signal). 
Selection of the dynamic threshold for each array as described above by 
which signals may be accepted as indicating receipt of a police radar 
signal greatly reduces falsing problems possible with prior art approaches 
and generally allows the lowest possible threshold setting (and 
consequently, the best possible sensitivity) under the signal and noise 
conditions then-existing. Under conditions in which a multitude of signals 
appear, the dynamic threshold associated with an array might actually 
exceed the value of each of the averager words in that array. To minimize 
the risk of missing a police radar signal under those conditions, peak 
detector 54 is further programmed to constrain the dynamic threshold for 
each array within limits such that if the calculated dynamic threshold for 
that array exceeds a predetermined maximum, it is set equal to that 
maximum. For the sensitivity array, the maximum is preferably 0.003052 or 
-25.15 dB relative to full scale. A maximum dynamic threshold for the 
response time array is not always required but, if used, would be 0.183105 
or -7.37 dB. Similarly, in situations where there are very few signals, 
the threshold may go so low as to allow noise alone to cause an averager 
word to exceed the associated threshold. To this end, if the threshold is 
below a predetermined minimum, it is set equal to that minimum to minimize 
false alarms under such conditions. The predetermined minimum for the 
sensitivity array is preferably 0.0003052 (-35.15 dB relative to full 
scale). Similarly, the minimum for the response time array is in the range 
of 0.0146 to 0.0256 (-18.36 dB to -15.92 dB) and is preferably selected as 
low as possible (e.g. 0.0183105 or -17.37 dB) in that range without 
creating a problem with false alarms. 
Preferably, and to avoid the possibility of false alarms due, for example, 
to interference from variable frequency signal sources such as 
superhomodyne radar warning receivers operating in the vicinity of 
receiver 5, DSP circuit 32 includes an index memory 60 which operates in 
conjunction with peak detector 54 to provide sweep-to-sweep comparison for 
false alarm elimination in a fashion similar to that provided by the 
circuit shown in U.S. Pat. No. 4,581,769 assigned to the assignee hereof. 
Aforesaid U.S. Pat. No. 4,581,769 is incorporated herein by reference. 
Rather than provide the alarm enable signal merely because one or more 
averager words in an array exceed the associated dynamic threshold, it is 
preferred to not generate the alarm enable signal until a match is found 
in index numbers exceeding the associated dynamic threshold in two 
consecutive sweeps, for example. To this end, after a given sweep, the 
index numbers of those segments whose averager word exceeded the 
associated dynamic threshold may be stored in memory 60. Thereafter, on 
the next sweep, the index number of each segment exceeding the 
newly-calculated dynamic threshold may be compared to the index numbers 
stored in memory 60. A match between any such index number from the 
present sweep and a stored index number from the prior sweep provides 
confirmation that an actual police radar signal was likely received and 
the alarm enable signal may be placed on output 56 as previously 
described. 
To allow for variations due to instabilities in receiver 5 and the like, 
DSP circuit 32 is programmed to allow some margin in comparing segments 
from sweep to sweep. For example, if segment number 6.sub.S (or, for 
example, 30.sub.R) in sweep A had an averager word value which exceeded 
the dynamic threshold for the sensitivity array in sweep A, and in sweep B 
it was the averager word segment 7.sub.S (or, for example, 31.sub.R) which 
exceeded the dynamic threshold calculated from the sensitivity array in 
sweep B, an alarm would still be given. Thus, the sweep-to-sweep segment 
comparisons allow for some mismatch to still be deemed a valid signal. The 
amount of mismatch is preferably plus or minus 5 segments. After the 
sweep-to-sweep comparison, the index numbers of the averager words 
exceeding their associated dynamic threshold in the present sweep may be 
stored in memory 60 in place of the previously stored index numbers. 
Due to limitations in the amount of memory available in memory 60, as well 
as the amount of time available to do all of the calculations, it may be 
possible to maintain no more than a few of the index numbers of segments 
from a given sweep for comparison to the results of the next sweep. It is 
believed that storing no more than six index numbers/array of those 
segments having averager word magnitudes exceeding the associated dynamic 
threshold is sufficient to provide satisfactory operation of receiver 5. 
Thus, if seven such segments from an array satisfy the criteria, the index 
numbers for only the six largest averager words in that array (those which 
exceed the associated dynamic threshold by the most) will be stored for 
use in the subsequent sweep, whereas if fewer than six test positive, only 
the index numbers for those fewer number of segments will be stored. 
Alternatively, it has been found that the validity of the comparison is 
not hampered if the index numbers of the six segments having the largest 
averager word magnitude, irrespective of whether they exceed the 
associated dynamic threshold, are stored from a first sweep for comparison 
to the results of a second sweep. In this regard, if the index numbers of 
any segment in the present sweep whose averager word values exceed the 
associated dynamic threshold match a stored index number (within the 
above-described tolerance), even though the averager word value for that 
same segment did not exceed the associated dynamic threshold in the prior 
sweep, a match is found and an alarm enable provided. 
Because police radar operates in more than one band, it is desirable to 
know the band in which the signal is received. Thus, for example, it is 
useful to know whether the police radar unit emitting the received signal 
is operating in the X-band or the K-band. To this end, assuming that the 
sweep-to-sweep comparison is positive, peak detector 54 evaluates the 
spacing between those segments (up to six) which have an averager word 
magnitude exceeding the associated dynamic threshold in the current sweep 
to determine whether the alarm enable should indicate X- or K-band. As a 
result of generation of an image signal when the local oscillator is 
swept, at least one pair of averager words in an array having a 
predetermined difference in index numbers should be produced having 
magnitudes exceeding the associated dynamic threshold for each signal 
received. The index numbers for all of the six or fewer segments which are 
retained for an array after the current sweep are investigated for their 
spacing. The index numbers are paired and, if it is possible to pair them 
up such that the index numbers in each pair are spaced apart approximately 
35 (.+-.7) segments, then the alarm enable signal will indicate that a 
K-band alarm is to be given. If, however, two of the segments in any pair 
are separated by approximately 70 (.+-.13) segments, the above-pairing for 
K-band is not possible, or only one averager word in the array exceeds the 
associated threshold, then the alarm enable signal will indicate that an 
X-band alarm is to be given. Of course, the comparison is between index 
numbers associated with the same array; the relationship or spacing 
between an I.sub.S index number and an I.sub.R index number stored in 
memory 60 is not evaluated. 
Additionally, certain rules may be applied where there is a conflict in 
alarm enable between the two arrays. For example, if the alarm enable 
results from second averager data, and thereafter, but during the alarm 
condition, an alarm enable results from the first averager, the alarm 
enable (and thus peak detection and band determination) will be made from 
the second averager for two seconds and thereafter from the data provided 
by the first averager. Conversely, if the alarm enable is first provided 
by the first averager array, then throughout that alarm condition, the 
alarm enable is to be made from that first array and the second array 
essentially ignored. Finally, if both arrays first provide an alarm enable 
after the same sweep, they are to be treated as if the second array gave 
the first enable for two seconds as described in the former situation. 
Additionally, an alarm enable (preferably for an X-band alarm) may be 
provided for police radar signals that are too weak to cause an averager 
word for either array to exceed the associated dynamic threshold. This 
weak signal alarm enable is provided in the event that one or two of the 
same averager words in the sensitivity array otherwise present the largest 
magnitude for an extended period of time, such as two seconds. For this 
purpose, in any sweep where no index numbers are selected which match any 
index number stored in memory 60 from the prior sweep as described above, 
a counter 62 and an integrator 63 are reset by peak detector 54. Also, the 
index number of the two averager words in the sensitivity array having the 
largest magnitude (irrespective of their relationship to the associated 
threshold) are stored in low signal store or memory 64. For each sweep 
thereafter, if no matches occur between the index numbers selected in that 
sweep and the index numbers stored in memory 60 from the prior sweep as 
described earlier, counter 62 is either incremented or reset and 
integrator 63 is either updated or reset. The counter is incremented if, 
in that sweep, the index number of either of the two averager words with 
the largest magnitude in that sweep match one of the two index numbers 
stored in memory 64 when counter 62 was last reset. Similarly, the value 
in the integrator 63 is summed with the largest averager word from that 
sweep. If, as a result of such operation over a plurality of sweeps, 
integrator 63 sums up to a predetermined value (e.g., exceeding ten times 
the minimum dynamic threshold limit for the sensitivity array) and counter 
62 counts up to a predetermined number (corresponding to about two seconds 
worth of sweeps) without resetting, peak detector 54 will provide an alarm 
enable irrespective of whether the two averager words involved are above 
or below the associated dynamic threshold. If, in any sweep, no match 
occurs from either of memory 60 or memory 64, counter 62 and integrator 63 
are again reset and two new index numbers will be stored in memory 64 
based on the results of that sweep. 
As is well understood, remaining circuitry 56 of receiver 5 may provide an 
audible alarm (not shown) which beeps at a rate proportional to the 
strength of a received signal when alarm enable 56 is provided. Similarly, 
an LED bar graph (not shown) functioning as a signal strength meter may be 
employed to show visually that same signal strength. To this end, at each 
sample interval, analog multiplexer 27 is caused to switch between the 
first and second state by select signal 28 so as to provide at each sample 
interval (1) a digital sample word from FM discriminator 22 to be 
processed as previously described, and (2) a digital AM word which is 
routed to and stored in a secondary memory location in supplemental memory 
50 corresponding to the sample interval. In conjunction with provision of 
an alarm enable 56 (resulting from a positive sweep-to-sweep match, for 
example), the digital AM word stored in supplemental memory 50 at a 
location corresponding to the index number of the largest averager word in 
that sweep is routed through peak detector 54 to remaining circuitry 58 
over AM line 66. That AM word corresponds to the amplitude of the signal 
received in that sample interval and, thus, may be utilized to control the 
beep rate of the audible alarm and/or the LED bar graph display. Although 
not shown, output 26 of AM detector 26 may be amplified such that the LED 
bar graph just barely begins to display intensity at full signal when 
receiver 5 is exposed to a calibrated police radar signal reference. 
While the present invention has been illustrated by the description of an 
embodiment thereof, and while the embodiment has been described in 
considerable detail, it is not the intention of applicant to restrict or 
in any way limit the scope of the appended claims to such detail. 
Additional advantages and modifications will readily appear to those 
skilled in the art. For example, multiple as opposed to dual averager 
arrays may be used. Also, the second (or subsequent) array may be ignored 
when the peak detection sequence for the sensitivity array indicates 
possible reception of a police radar signal. Furthermore, if the memory 
and calculation capacity of the DSP circuit utilized is sufficient, it may 
be possible to dispense with complex IIR filter 44 and decimator 46 and, 
thus, provide sufficient memory locations in supplemental memory 50 to 
accumulate all the correlator output words produced by correlator 42 for 
purposes of operation of peak detector 54 as described above. Indeed, IIR 
filters 40 and 44 could alternatively be finite impulse response filters. 
The invention in its broader aspects is therefore not limited to the 
specific details, representative apparatus and method, in the illustrative 
example shown and described. Accordingly, departures may be made from such 
details without departing from the spirit or scope of applicant's general 
inventive concept.