PWM control circuit and motor equipped with the same

The PWM control circuit is provided. The PWM control circuit includes: a PWM control signal generator that generates a PWM period signal defining a period of a PWM signal and a PWM resolution signal specifying a resolution in one period of the PWM period signal; and a PWM unit that generates the PWM signal based on the PWM period signal and the PWM resolution signal, wherein the PWM control signal generator changes a frequency of the PWM resolution signal while keeping a frequency of the PWM period signal unchanged.

CROSS-REFERENCE TO RELATED APPLICATIONS

The present application claims the priority based on Japanese Patent Application No. 2007-289222 filed on Nov. 7, 2007, the disclosure of which is hereby incorporated by reference in its entirety.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to PWM control.

2. Description of the Related Art

One proposed PWM control technique is disclosed in Japanese Patent Laid-Open No. 2004-364366.

This related art technique forms a PWM fundamental wave from a basic frequency signal of a preset frequency and divides the frequency of the PWM fundamental wave to generate a PWM period signal. The PWM fundamental wave specifies a resolution to set a duty cycle in one period of the PWM period signal.

In a system of this related art technique, a change in frequency of the PWM fundamental wave for varying the accuracy of PWM control leads to a change in frequency of the PWM period signal. The changed frequency of the PWM period signal may coincide with a resonance frequency of a load structure (for example, a motor main body) under PWM control to cause undesirable vibration and noise.

SUMMARY

An object of the present invention is to provide technology that is able to allow a change of a resolution in one period of a PWM period signal constructed to define a period of a PWM signal, while keeping a frequency of the PWM period signal unchanged.

According to an aspect of the present invention, a PWM control circuit is provided. The PWM control circuit comprises: a PWM control signal generator that generates a PWM period signal defining a period of a PWM signal and a PWM resolution signal specifying a resolution in one period of the PWM period signal; and a PWM unit that generates the PWM signal based on the PWM period signal and the PWM resolution signal, wherein the PWM control signal generator changes a frequency of the PWM resolution signal while keeping a frequency of the PWM period signal unchanged.

The PWM control circuit according to this aspect of the invention allows a change of the resolution in one period of the PWM period signal, while keeping the frequency of the PWM period signal unchanged.

The present invention may be actualized by diversity of other applications, for example, a PWM control method, a PWM control device, a PWM control system, integrated circuits configured to attain the functions of PWM control, computer programs configured to attain the functions of PWM control, and recording media where such computer programs are recorded.

DESCRIPTION OF THE PREFERRED EMBODIMENT

Next, aspects of the present invention will be described in the following order on the basis of embodiments:A. First Embodiment:A1. Overview of Motor Configuration and Operation:A2. Configuration of Drive Circuit Unit:B. Second Embodiment:C. Modified Examples:

A. First Embodiment

A1. Overview of Motor Configuration and Operation

FIG. 1Adepicts in sectional view the configuration of the motor unit of a brushless motor pertaining to a first embodiment of the present invention. This motor unit has a stator portion10, an upper rotor portion30U, and a lower rotor portion30L. Each of these components10,30U,30L has generally disk-shaped contours.FIG. 1Bis a horizontal sectional view of the lower rotor portion30L. The lower rotor portion30L has four permanent magnets32L each having generally fan-shaped contours. The upper rotor portion30U is identical in design to the lower rotor portion30L and has been omitted from the illustration. The upper rotor portion30U and the lower rotor portion30L are fastened to a center shaft64and rotate simultaneously. The direction of magnetization of the magnets32U,32L is parallel to the rotating shaft64.

FIG. 1Cis a horizontal sectional view of the stator portion10. As shown inFIG. 1A, the stator portion10has a plurality of phase A coils12A, a plurality of phase B coils12B, and a support member14supporting these coils12A,12B.FIG. 1Cdepicts the phase B coils12B. In this example, there are provided four phase B coils12B each of which is wound in a fan-shaped configuration. The phase A coils12A have this same design. A drive circuit unit500is installed in the stator portion10as well. As shown inFIG. 1A, the stator portion10is fixed in a casing62.

FIG. 1Dis a conceptual diagram depicting the relationship of the stator portion10and the two rotor portions30U,30L. On the support member14of the stator portion10are provided a magnetic sensor40A for phase A use and a magnetic sensor40B for phase B use. The magnetic sensors40A,40B are used to detect the position of the rotor portions30U,30L (i.e. the phase of the motor). These sensors will hereinafter be referred to as the “phase A sensor” and the “phase B sensor.” The phase A sensor40A is positioned at a center location between two of the phase A coils12A. Similarly, the phase B sensor40B is positioned at a center location between two of the phase B coils12B. In this example, the phase A sensor40A is positioned together with the phase B coils12B at the lower face of the support member14, but it could instead be positioned at the upper face of the support member14. This applies to the phase B sensor40B as well. As will be understood fromFIG. 1C, in this embodiment, the phase A sensor40A is positioned inside one of the phase B coils12B, which has the advantage of ensuring space for placement of the sensor40A.

As shown inFIG. 1D, the magnets32U,32L are each positioned at a constant magnetic pole pitch Pm, with adjacent magnets being magnetized in opposite directions. The phase A coils12A are arranged at constant pitch Pc, with adjacent coils being excited in opposite directions. This applies to the phase B coils12B as well. In the present embodiment, the magnetic pole pitch Pm is equal to the coil pitch Pc, and in terms of electrical angle is equivalent to π. An electrical angle of 2π is associated with the mechanical angle or distance of displacement when the phase of the drive signal changes by 2π. In the present embodiment, when the phase of the drive signal changes by 2π, the rotor portions30U,30D undergo displacement by the equivalent of twice the magnetic pole pitch Pm. The phase A coils12A and the phase B coils12B are positioned at locations phase-shifted by π/2 from each other.

The magnets32U of the upper rotor portion30U and the magnets32L of the lower rotor portion30L are positioned with their magnetic poles which face towards the stator portion10having mutually different polarity (N pole and S pole). In other words, the magnets32U of the upper rotor portion30U and the magnets32L of the lower rotor portion30L are positioned with their opposite poles facing one another. As a result, as shown at the right end inFIG. 1D, the magnetic field between these magnets32U,32L will be represented by substantially straight magnetic field lines and will be closed between these magnets32U,32L. It will be appreciated that this closed magnetic field is stronger than the open magnetic field shown inFIG. 26discussed previously. As a result, magnetic field utilization efficiency will be higher, and it will be possible to improve motor efficiency. In preferred practice, magnetic yokes34U,34L made of a ferromagnetic body will be disposed respectively on the outside faces of the magnets32U,32L. The magnetic yokes34U,34L make it possible to further strengthen the magnetic field in the coils. However, the magnetic yokes34U,34L may be omitted.

FIGS. 2A-2Dillustrate the relationship of sensor output and back electromotive force waveform.FIG. 2Ais identical toFIG. 1D.FIG. 2Bshows an exemplary waveform of back electromotive force generated by the phase A coils12A.FIGS. 2C and 2Dshow exemplary waveforms of sensor outputs SSA, SSB of the phase A sensor40A and the phase B sensor40B. These sensors40A,40B can generate sensor outputs SSA, SSB of shape substantially similar to the back electromotive force of the coils during motor operation. The back electromotive force of the coils12A shown inFIG. 2Btends to rise together with motor speed but its waveform shape (sine wave) maintains substantially similar shape. Hall ICs that utilize the Hall effect, for example, could be employed as the sensors40A,40B. In this example, the sensor output SSA and the back electromotive force Ec are each a sine wave or waveform approximating a sine wave. As will be discussed later, the drive control circuit of this motor, utilizing the sensor outputs SSA, SSB, applies voltage of shape substantially similar to the back electromotive force Ec to the respective coils12A,12B.

An electric motor functions as an energy conversion device that converts between mechanical energy and electrical energy. The back electromagnetic force of the coils represents mechanical energy of the electric motor converted to electrical energy. Consequently, where electrical energy applied to the coils is converted to mechanical energy (that is, where the motor is driven), it is possible to drive the motor with maximum efficiency by applying voltage of similar waveform to the back electromagnetic force. As will be discussed below, “voltage of similar waveform to the back electromagnetic force” means voltage that generates current flowing in the opposite direction from the back electromagnetic force.

FIG. 3Ais a model diagram illustrating the relationship of applied voltage and electromotive force of a coil. Here, the coil is simulated in terms of AC back electromotive force Ec and resistance Rc. In this circuit, a voltmeter V is parallel-connected to the AC application voltage Ei and the coil. The back electromotive force Ec is also termed “induced voltage Ec” and the application voltage Ei is also termed “exciting voltage Ei.” When AC voltage Ei is applied to the coil to drive the motor, back electromotive force Ec will be generated a direction of current flow opposite that of the application voltage Ei. When a switch SW is opened while the motor is rotating, the back electromotive force Ec can be measured with the voltmeter V. The polarity of the back electromotive force Ec measured with the switch SW open will be the same as the polarity of the application voltage Ei measured with the switch SW closed. The phrase “application of voltage of substantially similar waveform to the back electromagnetic force” herein refers to application of voltage having the same polarity as, and waveform of substantially similar shape to, the back electromotive force Ec measured by the voltmeter V.

FIG. 3Billustrates an overview of the driving method employed in the present embodiment. Here, the motor is simulated by the phase A coils12A, the permanent magnets32U, and the phase A sensor40A. When the rotor having the permanent magnets32U turns, AC voltage Es (also termed “sensor voltage Es”) is generated in the sensor40A. This sensor voltage Es has a waveform shape substantially similar to that of the induced voltage Ec of the coil12A. Thus, by generating PWM signal which simulates the sensor voltage Es for on/off control of the switch SW it will be possible to apply to the coils12A exciting voltage Ei of substantially similar waveform to the induced voltage Ec. The exciting current Ii at this time will be given by Ii=(Ei−Ec)/Rc.

As noted previously, when driving a motor, it is possible to drive the motor with maximum efficiency through application of voltage of waveform similar to that of the back electromagnetic force. It can be appreciated that energy conversion efficiency will be relatively low in proximity to the midpoint (in proximity to 0 voltage) of the sine wave waveform of back electromotive force, while conversely energy conversion efficiency will be relatively high in proximity to the peak of the back electromotive force waveform. Where a motor is driven by applying voltage of waveform similar to that of the back electromotive force, relatively high voltage can be applied during periods of high energy conversion efficiency, thereby improving efficiency of the motor. On the other hand, if the motor is driven with a simple rectangular waveform for example, considerable voltage will be applied in proximity to the position where back electromotive force is substantially 0 (midpoint) so motor efficiency will drop. Also, when voltage is applied during such periods of low energy conversion efficiency, due to eddy current vibration will be produced in directions other than the direction of rotation, thereby creating a noise problem.

As will be understood from the preceding discussion, the advantages of driving a motor through application of voltage of similar waveform to the back electromotive force are improved motor efficiency and reduced vibration and noise.

FIG. 4A-4Dare illustrations depicting forward rotation operation of the brushless motor of the embodiment.FIG. 4Adepicts the state just before the phase reaches 0. The letters “N” and “S” shown at locations of the phase A coils12A and the phase B coils12B indicate the excitation direction of these coils12A,12B. When the coils12A,12B are excited, forces of attraction and repulsion are generated between the coils12A,12B and the magnets32U,32L. As a result, the rotor portions30U,30L turn in the forward rotation direction (rightward in the drawing). At the timing of the phase going to 0, the excitation direction of the phase A coils12A reverses (seeFIGS. 2A-2D). FIG.4B depicts a state where the phase has advanced to just before π/2. At the timing of the phase going to π/2, the excitation direction of the phase B coils12B reverses.FIG. 4Cdepicts a state where the phase has advanced to just before π. At the timing of the phase going to π, the excitation direction of the phase A coils12B again reverses.FIG. 4Ddepicts a state where the phase has advanced to just before 3π/2. At the timing of the phase going to 3π/2, the excitation direction of the phase B coils12B again reverses.

As will be apparent fromFIGS. 2C and 2Das well, at times at which the phase equals an integral multiple of π/2 the sensor outputs SSA, SSB will go to zero, and thus driving force will be generated from only one of the two sets of coils12A,12B. However, during all periods except for times at which the phase equals integral multiples of π/2, it will be possible for the sets of coils12A,12B of both phases to generate driving force. Consequently, high torque can be generated using the sets of coils12A,12B of both phases.

As will be apparent fromFIG. 4A, the phase A sensor40A is positioned such that the location at which the polarity of its sensor output switches will be situated at a location where the center of a phase A coil12A faces the center of a permanent magnet32U. Similarly, the phase B sensor40B is positioned such that the location at which the polarity of the sensor output switches will be situated at a location where the center of a phase B coil12A faces the center of a permanent magnet32L. By positioning the sensors40A,40B at these locations, it will be possible to generate from the sensors40A,40B the sensor outputs SSA, SSB (FIGS. 2C and 2D) which have substantially similar waveform to the back electromotive force of the coils.

FIG. 5A-5Dare illustrations depicting reverse rotation operation of the brushless motor of the embodiment.FIG. 5A-5Drespectively depicts states where the phase has reached just before 0, π/2, π, and 3/π2. Reverse rotation operation can be accomplished, for example, by reversing the polarity of the drive voltages of the coils12A,12B to from that of the respective drive voltages during forward rotation operation.

A2. Configuration of Drive Circuit Unit

FIG. 6is a block diagram depicting an internal configuration of a drive circuit unit in the present embodiment. The drive circuit unit500has a CPU110, a drive controller100, a regeneration controller200, a driver circuit150, a rectifier circuit250, and a power supply unit300. The two controllers100,200are connected to the CPU110via a bus102. The drive controller100and the driver circuit150are circuits for carrying out control in instances where driving force is to be generated in the electric motor. The regeneration controller200and the rectifier circuit250are circuits for carrying out control in instances where power from the electric motor is to be regenerated. The regeneration controller200and the rectifier circuit250will be referred to collectively as a “regeneration circuit.” The drive controller100will also be referred to as a “drive signal generating circuit.” The power supply unit300is a circuit for supplying various power supply voltages to other circuits in the drive circuit unit500. InFIG. 6, for convenience, only the power lines going from the power supply unit300to the drive controller100and the driver circuit150are shown; power lines leading to other circuits have been omitted.

FIG. 7shows a configuration of a phase A driver circuit120A and a phase B driver circuit120B included in the driver circuit150(FIG. 6). The phase A driver circuit120A is an H bridge circuit for delivering AC drive signals DRVA1, DRVA2to the phase A coils12A. The white circles next to terminal portions of blocks which indicate drive signals denote negative logic and indicate that the signal is inverted. The arrows labeled IA1, IA2respectively indicate the direction of current flow with the A1drive signal DRVA1and the A2drive signal DRVA2. The configuration of the phase B driver circuit120B is the same as the configuration of the phase A driver circuit120A.

FIG. 8A-8Eare explanatory views showing the internal configuration and the operations of the drive controller100(FIG. 6). The drive controller100includes a PWM control signal generator600, PWM units530, a moving direction register540, multipliers550, encoders560, AD converters570, voltage control value registers580, and excitation interval setters590. The drive controller100is a circuit configured to generate both a driving signal for the phase A and a driving signal for the phase B. The PWM control signal generator600and the moving direction register540are commonly used for the phase A and the phase B. The other components of the drive controller100are provided individually for the phase A and the phase B. While only the components for the phase A are shown inFIG. 8Aas a matter of convenience, another set of the same components are provided for the phase B in the drive controller100.

The PWM control signal generator600generates a clock signal SDC having a preset frequency and a clock signal PCL having a frequency of N times as much as the frequency of the clock signal SDC. The value N is set in advance by the CPU110. The internal structure of the PWM control signal generator600will be explained later. The PWM unit530generates AC drive signals DRVA1and DRVA2(FIG. 7), based on the clock signals PCL and SDC, a multiplication result Ma output from the multiplier550, a forward/reverse directional value RI output from the moving direction register540, a positive/negative sign signal Pa output from the encoder560, and an excitation interval signal Ea output from the excitation interval setter590. The operations of these components will be described later.

A value RI indicating the direction for motor rotation is established in the moving direction register540, by the CPU110. In the present embodiment, the motor will rotate forward when the forward/reverse direction value RI is L level, and rotate in reverse rotation when H level. The other signals Ma, Pa, Ea supplied to the PWM unit530are determined as follows.

The output SSA of the magnetic sensor40is supplied to the AD converter570. This sensor output SSA has a range, for example, of from GND (ground potential) to VDD (power supply voltage), with the middle point thereof (=VDD/2) being the π phase point of the output waveform, or the point at which the sine wave passes through the origin. The AD converter570performs AD conversion of this sensor output SSA to generate a digital value of sensor output. The output of the AD converter570has a range, for example, of FFh to 0h (the “h” suffix denotes hexadecimal), with the median value of 80h corresponding to the middle point of the sensor waveform.

The encoder560converts the range of the sensor output value subsequent to the AD conversion, and sets the value of the middle point of the sensor output value to 0. As a result, the sensor output value Xa generated by the encoder560assumes a prescribed range on the positive side (e.g. between +127 and 0) and a prescribed range on the negative side (e.g. between 0 and −127). However, the value supplied to the multiplier560by the encoder560is the absolute value of the sensor output value Xa; the positive/negative sign thereof is supplied to the PWM unit530as the positive/negative sign signal Pa.

The voltage control value register580stores a voltage control value Ya established by the CPU110. This voltage control value Ya, together with the excitation interval signal Ea discussed later, functions as a value for setting the application voltage to the motor. The value Ya can assume a value between 0 and 1.0, for example. Assuming an instance where the excitation interval signal Ea has been set with no non-excitation intervals provided so that all of the intervals are excitation intervals, Ya=0 will mean that the application voltage is zero, and Ya=1.0 will mean that the application voltage is at maximum value. The multiplier550performs multiplication of the voltage control value Ya and the sensor output value Xa output from the encoder560and conversion to an integer; the multiplication value Ma thereof is supplied to the PWM unit530.

FIGS. 8B-8Edepict operation of the PWM unit530in instances where the multiplication value Ma takes various different values. Here, it is assumed that there are no non-excitation intervals, so that all intervals are excitation intervals. The PWM unit530is a circuit that, during one period of the clock signal SDC, generates one pulse with a duty factor of Ma/N. Specifically, as shown inFIGS. 8B-8E, the pulse duty factor of the single-phase drive signals DRVA1, DRVA2increases in association with increase of the multiplication value Ma. The first drive signal DRVA1is a signal that generates a pulse only when the sensor output SSA is positive and the second drive signal DRVA2is a signal that generates a pulse only when the sensor output SSA is negative; inFIGS. 8B-8E, both are shown together. For convenience, the second drive signal DRVA2is shown in the form of pulses on the negative side.

FIG. 9is a block diagram showing the internal structure of the PWM control signal generator600. The PWM control signal generator600includes a fixed frequency oscillator602, a frequency divider604, a PLL circuit606, a frequency division value R storage element608, and a frequency division value N storage element610. The fixed frequency oscillator602is a circuit generating a fixed clock signal FCLK of a fixed frequency and may be constructed by, for example, a crystal oscillator or a ceramic oscillator. The frequency divider604divides the frequency of the fixed clock signal FLCK to 1/R and outputs a frequency-divided clock signal RCLK. The PLL circuit606generates the clock signal SDC in synchronism with the frequency-divided clock signal RCLK and the clock signal PCL having the frequency of N times as much as the frequency of the clock signal SDC. The value ‘N times’ represents a frequency division value N of a frequency divider provided in the PLL circuit606as explained later. The frequency division value N is stored in the frequency division value N storage element610and is arbitrarily rewritable by the CPU110. Similarly a frequency division value R is stored in the frequency division value R storage element608and is arbitrarily rewritable by the CPU110.

FIG. 10is a block diagram showing the internal structure of the PLL circuit606. The PLL circuit606includes a phase comparator620, a loop filter622, a voltage control oscillator624, and a frequency divider626. The frequency-divided clock signal RCLK output from the frequency divider604(FIG. 9) is input into the phase comparator620as a reference signal. The clock signal SDC output after frequency division by the frequency divider626is input into the phase comparator620as a return signal. The phase comparator620generates an error signal CPS representing a phase difference between the two input signals RCLK and SDC. The error signal CPS is sent to the loop filter622including a charge pump circuit. The charge pump circuit included in the loop filter622generates and outputs a voltage control signal LPS having a voltage level corresponding to a pulse level and a pulse number of the error signal CPS.

The voltage control oscillator624outputs the clock signal PCL having an oscillation frequency corresponding to the voltage level of the voltage control signal LPS. The clock signal PCL is subjected to frequency division to 1/N by the frequency divider626, based on the frequency division value N stored in the frequency division value N storage element610. The clock signal SDC output from the frequency divider626is input into the phase comparator620to be subjected to phase comparison with the frequency-divided clock signal RCLK as explained previously. The frequency of the clock signal PCL is converged to decrease the phase difference between the two input signals RCLK and SDC to zero. A frequency fPCL of the converged clock signal PCL is equal to the product of a frequency fRCLK of the frequency-divided clock signal RCLK and the frequency division value N. The frequency fPCL of the converged clock signal PCL is also equal to the product of a frequency fSDC of the clock signal SDC and the frequency division value N.

There are the following relations between a frequency fFCLK of the fixed clock signal FCLK, the frequency fRCLK of the frequency-divided clock signal, the frequency fSDC of the clock signal SDC, and the frequency fPCL of the clock signal PCL.
fFCLK/R=FRCLK  (1)
fRCLK=FSDC  (2)
fSDC×N=FPCL(3)

In the above structure, rewriting the frequency division value N changes only the frequency of the clock signal PCL, while keeping the frequency of the clock signal SDC unchanged. Increasing the frequency of the clock signal PCL with the unchanged frequency of the clock signal SDC allows the duty cycle to be set more finely. The frequency of the clock signal SDC should be set in advance not to coincide with resonance frequency of a load structure, such as a motor main body. Such setting effectively prevents the occurrence of vibration or noise from the load structure like the motor main body in the state of changing the frequency of the clock signal PCL. The frequency of the clock signal SDC is set preferably out of an audio frequency range.

Rewriting the frequency division value R stored in the frequency division value R storage element608(FIG. 9) changes the frequency of the frequency-divided clock signal RCLK and the frequency of the clock signal SDC. Increasing the frequency of the clock signal SDC ensures PWM control at cycles of narrower time intervals and thereby allows control with high precision (for example, attitude control). In this state, the relation of Equation (3) given above is held as the relation between the frequency fSDC of the clock signal SDC and the frequency fPCL of the clock signal PCL. As mentioned above, it is preferable to change the frequency of the clock signal SDC in such a manner that the frequency of the clock signal SDC does not coincide with the resonance frequency of the load structure.

FIGS. 11A and 11Bare timing charts showing the operations of the fixed clock signal FLCK, the frequency-divided clock signal RCLK, the clock signal SDC, and the clock signal PCL.FIG. 11Ashows the operations of these signals at the frequency division value N equal to 7. In this case, seven pulses of the clock signal PCL are generated in one period of the clock signal SDC. At the frequency division value N equal to 14, fourteen pulses of the clock signal PCL are generated in one period of the clock signal SDC as shown inFIG. 11B.

FIGS. 12A-12Cdepict correspondence between sensor output waveform and waveform of the drive signals generated by the PWM unit530. In the drawing, “Hiz” denotes a state of high impedance where the magnetic coils are not excited. As described inFIGS. 8B-9E, the single-phase drive signals DRVA1, DRVA2are generated by PWM control using the analog waveform of the sensor output SSA. Consequently, using these single-phase drive signals DRVA1, DRVA2it is possible to supply to the coils effective voltage that exhibits changes in level corresponding to change in the sensor outputs SSA.

The PWM unit530is constructed such that drive signals are output only during the excitation intervals indicated by the excitation interval signal Ea supplied by the excitation interval setting unit590, with no drive signals being output at intervals except for the excitation intervals (non-excitation intervals).FIG. 12Cdepicts drive signal waveforms produced in the case where excitation intervals EP and non-excitation intervals NEP have been established by the excitation interval signal Ea. During the excitation intervals EP, the drive signal pulses ofFIG. 12Bare generated as is; during the non-excitation intervals NEP, no pulses are generated. By establishing excitation intervals EP and non-excitation intervals NEP in this way, voltage will not be applied to the coils in proximity to the middle point of the back electromotive force waveform (i.e. in proximity to the middle point of the sensor output), thus making possible further improvement of motor efficiency. Preferably the excitation intervals EP will be established at intervals symmetric about the peak point of the back electromotive force waveform; and preferably the non-excitation intervals NEP will be established in intervals symmetric about the middle point (center) of the back electromotive force waveform.

As discussed previously, if the voltage control value Ya is set to a value less than 1, the multiplication value Ma will be decreased in proportion to the voltage control value Ya. Consequently, effective adjustment of application voltage is possible by the voltage control value Ya as well.

As will be understood from the preceding description, with the motor of the present embodiment, it is possible to adjust the application voltage using both the voltage control value Ya and the excitation interval signal Ea. In preferred practice, relationships between desired application voltage on the one hand, and the voltage control value Ya and excitation interval signal Ea on the other, will be stored in advance in table format in memory in the drive circuit unit500(FIG. 6). By so doing, when the drive circuit unit500has received a target value for the desired application voltage from the outside, it will be possible for the CPU110, in response to the target value, to set the voltage control value Ya and the excitation interval signal Ea in the drive controller100. Adjustment of application voltage does not require the use of both the voltage control value Ya and the excitation interval signal Ea, and it would be acceptable to use either one of them instead.

FIG. 13is a block diagram depicting the internal configuration of the PWM unit530(FIG. 8A). The PWM unit530has a counter531, an EXOR circuit533, and a drive waveform shaping circuit535. Their operation will be described below.

FIG. 14is a timing chart depicting operation of the PWM unit530during forward rotation of the motor. The drawing show the two clock signals PCL and SDC, the forward/reverse direction value RI, the excitation interval signal Ea, the multiplication value Ma, the positive/negative sign signal Pa, the counter value CM1in the counter531, the output SI of the counter531, the output S2of the EXOR circuit533, and the output signals DRVA1, DRVA2of the drive waveform shaping circuit535. For each one cycle of the clock signal SDC, the counter531repeats an operation of decrementing the count value CM1to 0, in sync with the clock signal PCL. The initial value of the count value CM1is set to the multiplication value Ma. InFIG. 14, for convenience in illustration, negative multiplication values Ma are shown as well; however, the counter531uses the absolute values |Ma| thereof. The output S1of the counter531is set to H level when the count value CM1is not 0, and drops to L level when the count value CM1is 0.

The EXOR circuit533outputs a signal S2that represents the exclusive OR of the positive/negative sign signal Pa and the forward/reverse direction value RI. Where the motor is rotating forward, the forward/reverse direction value RI will be at L level. Consequently, the output S2of the EXOR circuit533will be a signal identical to the positive/negative sign signal Pa. The drive waveform shaping circuit535generates the drive signals DRVA1, DRVA2from the output S1of the counter531and the output S2of the EXOR circuit533. Specifically, in the output S1of the counter531, the signal during intervals in which the output S2of the EXOR circuit533is at L level will be output as the drive signal DRVA1, and the signal during intervals in which the output S2of the EXOR circuit533is at H level will be output as the drive signal DRVA2. In proximity to the right edge inFIG. 14, the excitation interval signal Ea falls to L level thereby establishing a non-excitation interval NEP. Consequently, neither of the drive signals DRVA1, DRVA2will be output during this non-excitation interval NEP, and a state of high impedance will be maintained.

FIG. 15is a timing chart depicting operation of the PWM unit530during reverse rotation of the motor. Where the motor is rotating in reverse, the forward/reverse direction value RI will be at H level. As a result, the two drive signals DRVA1, DRVA2switch relative toFIG. 12, and it will be appreciated that the motor runs in reverse as a result.

FIGS. 16A and 16Billustrate the internal configuration and operation of an excitation interval setting unit590. The excitation interval setting unit590has an electronic variable resistor592, a voltage comparators594,596, and an OR circuit598. The resistance Rv of the electronic variable resistor592is set by the CPU110. The voltages V1, V2at either terminal of the electronic variable resistor592are supplied to one of the input terminals of the voltage comparators594,596. The sensor output SSA is supplied to the other input terminal of the voltage comparators594,596. The output signals Sp, Sn of the voltage comparators594,596are input to the OR circuit598. The output of the OR circuit598is the excitation interval signal Ea, which is used to differentiate excitation intervals and non-excitation intervals.

FIG. 16Bdepicts operation of the excitation interval setting unit590. The voltages V1, V2at the terminals of the electronic variable resistor592are modified by adjusting the resistance Rv. Specifically, the terminal voltages V1, V2are set to values of equal difference from the median value of the voltage range (=VDD/2). In the event that the sensor output SSA is higher than the first voltage V1, the output Sp of the first voltage comparator594goes to H level, whereas in the event that the sensor output SSA is lower than the second voltage V2, the output Sn of the second voltage comparator596goes to H level. The excitation interval signal Ea is a signal derived by taking the logical sum of the these output signals Sp, Sn. Consequently, as shown at bottom inFIG. 16B, the excitation interval signal Ea can be used as a signal indicating excitation intervals EP and non-excitation intervals NEP. The excitation intervals EP and non-excitation intervals NEP are established by the CPU110, by adjusting the variable resistance Rv.

FIGS. 17A and 17Bare illustrations comparing various signal waveforms in the case where the motor of the embodiment discussed above is driven by a rectangular wave, and where driven by a sine wave. Where a rectangular wave is employed for driving, a drive voltage of rectangular wave shape is applied to the coils. While the drive current is close to a rectangular wave at startup, it decreases as rotation speed increases. This is because the back electromotive force increases in response to the increased rotation speed (FIG. 2B). With a rectangular wave, however, despite increased rotation speed the current value will not decline appreciably in proximity to the timing of switching of the drive voltage at phase=nπ, so a fairly large current will tend to flow.

On the other hand, where a sine wave is employed for driving, PWM control is employed for the drive voltage so that the effective values of the drive voltage have sine wave shape. While the drive current is close to a sine wave at startup, as rotation speed increases the drive current will decrease due to the effects of back electromotive force. With sine wave driving, the current value declines appreciably in proximity to the timing of switching of the drive voltage polarity at phase=nπ. As discussed in the context ofFIGS. 2A-2C, generally speaking the energy conversion efficiency of a motor is low in proximity to the timing of switching of the drive voltage polarity. With sine wave driving, the current value during intervals of low efficiency is lower than with rectangular wave, making it possible to drive the motor more efficiently.

FIG. 18depicts another configuration example of the phase A driver circuit120A and the phase B driver circuit120B included in the driver circuit150(FIG. 6). These driver circuits120A,120B are furnished with amplifier circuits122situated in front of the gate electrodes of the transistors which make up the driver circuits120A,120B shown inFIG. 8. While the type of transistor also differs from that inFIG. 8, transistors of any type can be used as the transistors. In order to be able to drive the motor of the present invention over a wider operating range with regard to torque and speed, it will be preferable to establish variable power supply voltage VDD of the driver circuits120A,120B. Where the power supply voltage VDD has been changed, the level of the drive signals DRVA1, DRVA2, DRVB1, DRVB2applied to the gate voltages of the transistors will change proportionally therewith. By so doing the motor can be driven using a wider power supply voltage VDD range. The amplifier circuits122are circuits for changing the level of the drive signals DRVA1, DRVA2, DRVB1, DRVB2. In preferred practice the power supply unit300of the drive circuit unit500shown inFIG. 6will supply variable power supply voltage VDD to the driver circuit150.

FIG. 19shows the speed of the motor of the embodiment in the absence of load. As will be apparent from the graph, in the absence of load the motor of the embodiment will rotate at stable speed down to very low speed. The reason is that since there is no magnetic core, cogging does not occur.

FIG. 20illustrates the internal configuration of the regeneration controller200and rectifier circuit250shown inFIG. 6. The regeneration controller200comprises an phase A charge switching unit202and a phase B charge switching unit204, both connected to the bus102, and an electronically variable resistor206. The output signals of the two charge switching units202,204are applied to the input terminals of the two AND circuits211,212.

The phase A charge switching unit202outputs a signal of a “1” level when the regenerative power from the phase A coils12A is recovered, and outputs a signal of a “0” level when the power is not recovered. The same is true for the phase B charge switching unit204. The switching of those signal levels is conducted with the CPU110. The presence or absence of regeneration from the phase A coils12A and the presence or absence of regeneration from the phase B coil12B can be set independently. Therefore, for example, electric power can be regenerated from the phase B coils12B, while generating a drive force in the motor by using the phase A coils12A.

The drive controller100, similarly, may have a configuration such that whether or not the drive force is generated by using the phase A coils12A and whether or not the drive force is generated by using the phase B coils12B can be set independently. In such a case, the motor can be operated in an operation mode such that a drive force is generated in any one of the two sets of coils12A,12B, while electric power is regenerated in the other coils.

The voltage across the electronically variable resistor206is applied to one of the two input terminals of the four voltage comparators221-224. The phase A sensor signal SSA and phase B sensor signal SSB are applied to the other input terminal of the voltage comparators221-224. The output signals TPA, BTA, TPB, BTB of the four voltage comparators221-224can be called “mask signals” or “permission signals”.

The mask signals TPA, BTA for the phase A coils are inputted into the OR circuit231, and the mask signals TPB, BTB for the phase B are inputted into the other OR circuit232. The outputs of those OR circuits231,232are supplied to the input terminals of the above-mentioned two AND circuits211,212. The output signals MSKA, MSKB of those AND circuits211,212are called “mask signals” or “permission signals”.

The configurations of the four voltage comparators221-224and the two OR circuits231,232are identical to two sets of the voltage comparators594,596, and the OR circuit598of the excitation interval setting unit590shown inFIG. 14A. Therefore, the output signal of the OR circuit231for the phase A coils is similar to the excitation interval signal Ea shown inFIG. 14B. Further, when the output signal of the phase A charge switching unit202is at a “1” level, the mask signal MSKA outputted from the AND circuit211for the phase A coils is identical to the output signal of the OR circuit231. Those operations are identical to those relating to the phase B.

The rectifier circuit250has the circuitry for the phase A coils which includes a full-wave rectifier circuit252comprising a plurality of diodes, two gate transistors261,262, a buffer circuit271, and an inverter circuit272(NOT circuit). The identical circuitry is also provided for the phase B. The gate transistors261,262are connected to the power wiring280for regeneration. It is preferable to use Schottky diodes which have excellent characteristics of low Vf as the plurality of diodes.

During power regeneration, the AC power generated in the phase A coils12A is rectified with the full-wave rectifier circuit252. The mask signal MSKA for the phase A coils and the inverted signal thereof are supplied to the gates of the gate transistors261,262, and the gate transistors261,262are ON/OFF controlled accordingly. Therefore, within a period in which at least one of the mask signals TPA, BTA outputted from the voltage comparators221,222is at an H level, the regenerated power is outputted to the power source wiring280. On the other hand, within an interval in which both mask signals TPA, BTA are at an L level, power regeneration is inhibited.

As clearly follows from the explanation provided hereinabove, the regenerated power can be recovered by using the regeneration controller200and rectifier circuit250. Furthermore, the regeneration controller200and rectifier circuit250can restrict the interval in which the regenerated power from the phase A coils12A and phase B coils12B is recovered, according to the mask signal MSKA for the phase A coils and the mask signal MSKB for the phase B coils, thereby making it possible to adjust the quantity of the regenerated power.

As described above, the PWM control signal generator600of the first embodiment readily changes only the frequency of the clock signal PCL while keeping the frequency of the clock signal SDC unchanged by simply rewriting the frequency division value N. The PWM control signal generator600and the PWM unit530(FIG. 8A) correspond to the ‘PWM control circuit’ of the invention. The clock signal SDC and the clock signal PCL are respectively equivalent to the ‘PWM period signal’ and the ‘PWM resolution signal’ of the invention. The output S1of the counter531and the drive signals DRVA1and DRVA2correspond to the ‘PWM signal’ of the invention.

B. Second Embodiment

FIG. 21is an explanatory view showing the structure of a PWM control circuit generator600bin a second embodiment of the present invention. The PWM control signal generator600bof the second embodiment directly uses the frequency-divided clock signal RCLK as the clock signal SDC but otherwise has the similar structure to that of the PWM control signal generator600of the first embodiment shown inFIG. 9.

The frequency-divided clock signal RCLK and the clock signal SDC have the same frequencies. The frequency-divided clock signal RCLK is thus directly usable as the clock signal SDC on the assumption that an interval between two rising edges of the frequency-divided clock signal RCLK is one cycle of PWM control. Like the arrangement of the first embodiment, the arrangement of the second embodiment allows a change of only the frequency of the clock signal PCL while keeping the frequency of the clock signal SDC unchanged. The frequency-divided clock signal RCLK is equivalent to the ‘reference signal’ of the invention.

C. Modified Examples

The present invention is not limited to the embodiments described hereinabove, and may be reduced to practice in various other ways without departing from the spirit thereof. Modifications such as the following are possible, for example.

The present invention is applicable to various kinds of devices. For example, the present invention is implemented in a motor in any of various devices such as fan motors, clocks (for driving the hands), drum type washing machines (single rotation), jet coasters, vibrating motors, and the like. Where the present invention is implemented in a fan motor, the various advantages mentioned previously (low power consumption, low vibration, low noise, minimal rotation irregularity, low heat emission, and long life) is particularly notable. Such fan motors can be employed, for example, as fan motors for various devices such as digital display devices, vehicle on-board devices, fuel cell type PCs, fuel cell type digital cameras, fuel cell type video cameras, fuel cell type mobile phones, various other fuel cell-powered devices, and projectors. The motor of the present invention may also be utilized as a motor for various types of household electric appliances and electronic devices. For example, a motor in accordance with the present invention may be employed as a spindle motor in an optical storage device, magnetic storage device, polygon mirror drive, or the like. The motor of the present invention may also be utilized as a motor for a movable body or a robot.

FIG. 22is an illustration depicting a projector which utilizes a motor according to the present invention. This projector800has three light sources810R,810G,810B for emitting light of the three colors red, green, and blue; liquid crystal light valves840R,840G,840B for modulating light of the three colors; a cross dichroic prism850for synthesizing modulated light of the three colors; a projection lens system860for projecting light synthesized from the three colors onto a screen SC; a cooling fan870for cooling the interior of the projector; and a controller880for controlling the entire projector800. Any of the various brushless motors described above may be used as the motor for driving the cooling fan870.

FIGS. 23A to 23Cillustrate a fuel cell type mobile phone which utilizes a motor according to the present invention.FIG. 23Ashows an exterior view of a mobile phone900, andFIG. 23Bshows an example of internal configuration. The mobile phone900includes an MPU910for controlling operation of the mobile phone900; a fan920; and a fuel cell930. The fuel cell930supplies power to the MPU910and to the fan920. The fan920blows air into the mobile phone900from the outside in order to supply air to the fuel cell930, or in order to expel moisture evolved in the fuel cell930from the inside of the mobile phone900to the outside. The fan920may also be positioned on the MPU910as shown inFIG. 23C, to cool the MPU910. Any of the various brushless motors described above can be used as the motor for driving the fan920.

FIG. 24is an illustration depicting an electrically powered bicycle (power assisted bicycle) as one example of a moving body that utilizes a motor/generator according to the embodiments of the present invention. This bicycle1000is provided with a motor1010on its front wheel; and with a control circuit1020and a rechargeable battery1030disposed on the frame below the saddle. The motor1010uses power from the rechargeable battery1030to drive the front wheel, thereby assisting travel. During braking, regenerative power from the motor1010is used to charge the rechargeable battery1030. The control circuit1020is a circuit for controlling driving and regeneration of the motor. Any of the various brushless motors described above can be used as the motor1010.

FIG. 25is an illustration showing an example of a robot which utilizes a motor according to the embodiments of the present invention. This robot1100has first and second arms1110,1120, and a motor1130. This motor1130is used during horizontal rotation of the second arm1120as the driven member. Any of the various brushless motors described above can be used as the motor1130.

The PWM control circuit of the invention is not restrictively incorporated in the brushless motor as described in the above embodiment but may be mounted on any of various devices under PWM control.

The structure of the embodiment uses the analog PLL circuit606(FIG. 10) to implement the technique of the invention. The analog PLL circuit606is, however, neither essential nor restrictive but may be replaced by a digital PLL circuit or a combination of multiple digital counters arranged to have the same functions as those of the digital PLL circuit.