Method and system for pre-equalization in a single weight spatial multiplexing MIMO system

Certain aspects of the method may comprise receiving a plurality of spatially multiplexed communication signals from a plurality of transmit antennas at a base station. A plurality of vectors of baseband combined channel estimates may be generated based on phase rotation of the received plurality of spatially multiplexed communication signals. A plurality of pre-equalization weights may be generated based on the generated plurality of vectors of baseband combined channel estimates. The received plurality of spatially multiplexed communication signals may be modified based on the generated plurality of pre-equalization weights. At least a portion of the generated plurality of pre-equalization weights may be fed back to the base station for modifying subsequently transmitted spatially multiplexed communication signals which are transmitted from at least a portion of the plurality of transmit antennas at the base station.

FIELD OF THE INVENTION

Certain embodiments of the invention relate to pre-equalization. More specifically, certain embodiments of the invention relate to a method and system for pre-equalization in a single weight spatial multiplexing multi-input multi-output (MIMO) system.

BACKGROUND OF THE INVENTION

In most current wireless communication systems, nodes in a network may be configured to operate based on a single transmit and a single receive antenna. However, for many of current wireless systems, the use of multiple transmit and/or receive antennas may result in an improved overall system performance. These multi-antenna configurations, also known as smart antenna techniques, may be utilized to reduce the negative effects of multipath and/or signal interference may have on signal reception. Existing systems and/or systems which are being currently deployed, for example, CDMA-based systems, TDMA-based systems, WLAN systems, and OFDM-based systems such as IEEE 802.11 a/g/n, may benefit from configurations based on multiple transmit and/or receive antennas. It is anticipated that smart antenna techniques may be increasingly utilized both in connection with the deployment of base station infrastructure and mobile subscriber units in cellular systems to address the increasing capacity demands being placed on those systems. These demands arise, in part, from a shift underway from current voice-based services to next-generation wireless multimedia services that provide voice, video, and data communication.

The utilization of multiple transmit and/or receive antennas is designed to introduce a diversity gain and array gain and to suppress interference generated within the signal reception process. Such diversity gains improve system performance by increasing received signal-to-noise ratio, by providing more robustness against signal interference, and/or by permitting greater frequency reuse for higher capacity. In communication systems that incorporate multi-antenna receivers, a set of M receive antennas may be utilized to null the effect of (M-1) interferers. Accordingly, N signals may be simultaneously transmitted in the same bandwidth using N transmit antennas, with the transmitted signal then being separated into N respective signals by way of a set of N antennas deployed at the receiver. Systems that utilize multiple transmit and multiple receive antenna may be referred to as multiple-input multiple-output (MIMO) systems. One attractive aspect of multi-antenna systems, in particular MIMO systems, is the significant increase in system capacity that may be achieved by utilizing these transmission configurations. For a fixed overall transmitted power, the capacity offered by a MIMO configuration may scale with the increased signal-to-noise ratio (SNR).

However, the widespread deployment of multi-antenna systems in wireless communications, particularly in wireless handset devices, has been limited by the increased cost that results from increased size, complexity, and power consumption. The necessity of providing a separate RF chain for each transmit and receive antenna is a direct factor in the increased the cost of multi-antenna systems. Each RF chain generally comprises a low noise amplifier (LNA), a filter, a downconverter, and an analog-to-digital converter (A/D). In certain existing single-antenna wireless receivers, the single required RF chain may account for over 30% of the receiver's total cost. It is therefore apparent that as the number of transmit and receive antennas increases, the system complexity, power consumption, and overall cost may increase.

In the case of a single RF chain with multiple antennas, there is a need to determine or estimate separate propagation channels. A simple method may comprise switching to a first receive antenna utilizing, for example, an RF switch, and estimate a first propagation channel. After estimating the first propagation channel, another receive antenna may be selected and its corresponding propagation channel may be estimated. In this regard, this process may be repeated until all the channels have been estimated. However, switching between receive antennas may disrupt the receiver's modem and may lower throughput. Moreover, this approach may require additional hardware and may also result in propagation channel estimates at different time intervals.

Further limitations and disadvantages of conventional and traditional approaches will become apparent to one of ordinary skill in the art through comparison of such systems with the present invention as set forth in the remainder of the present application with reference to the drawings.

BRIEF SUMMARY OF THE INVENTION

A method and/or system for pre-equalization in a single weight spatial multiplexing multi-input multi-output (MIMO) system, substantially as shown in and/or described in connection with at least one of the figures, as set forth more completely in the claims.

DETAILED DESCRIPTION OF THE INVENTION

Certain aspects of the method may comprise receiving a plurality of spatially multiplexed communication signals from a plurality of transmit antennas at a base station. A plurality of vectors of baseband combined channel estimates may be generated based on phase rotation of the received plurality of spatially multiplexed communication signals. A plurality of pre-equalization weights may be generated based on the generated plurality of vectors of baseband combined channel estimates. The received plurality of spatially multiplexed communication signals may be modified based on the generated plurality of pre-equalization weights. At least a portion of the generated plurality of pre-equalization weights may be fed back to the base station for modifying subsequently transmitted spatially multiplexed communication signals which are transmitted from at least a portion of the plurality of transmit antennas at the base station.

In another aspect of the method, the pre-equalization parameters may be generated based on least mean squares (LMS) algorithm, recursive least squares (RLS) algorithm, direct matrix inversion, and/or a cost function. In this regard, the parameters of the cost function may be modified in accordance with the application. The pre-equalization weights may be determined periodically or continuously. The pre-equalization weights may be fed back to a transmitter via an uplink channel. The received plurality of spatially multiplexed communication signals may be spatially demultiplexed. The various embodiments of the invention may provide a good compromise between implementation complexity and performance gains to reduce the effects of, for example, inter-symbol interference (ISI) and/or inter-carrier interference (ICI) in MIMO systems.

Spatial multiplexing (SM) may provide a mode of signal transmission predicated upon the use of multiple antennas at both a transmitter and a receiver, for example, in such a way that the capacity of a wireless radio link may be increased without correspondingly increasing power or bandwidth consumption. In a case in which N antennas are used at both a transmitter and a receiver, an input stream of information symbols provided to the transmitter is divided into N independent substreams. Spatial multiplexing contemplates that each of these N independent substreams may occupy the same “space-time channel”, for example, time slot, frequency, or code/key sequence, of the applicable multiple-access protocol. Within the transmitter, each substream may be separately applied to the N transmit antennas and propagated over an intervening multipath communication channel to a receiver. Error correction coding may be applied to each of the N streams separately or in a combined space-time methodology.

The composite multipath signals may then be received by an array of N or more receive antennas deployed at the receiver. At the receiver, a “spatial signature” defined by the N phases and N amplitudes arising at the receive antenna array for a given substream may be then estimated. Signal processing techniques may be then applied in order to spatially separate the received signals, which may allow the original substreams to be recovered and synthesized into the original input symbol stream. An overall system capacity of the order of the minimum of M and N, min(M,N), for example, may be achieved, where M may be the number of receive antennas and N may be the number of transmit antennas for flat fading channel conditions. The principles of spatially multiplexed communication and exemplary system implementations are further described in, for example, “Optimum combining for indoor radio systems with multiple users”, by J. H. Winters, IEEE Transactions on Communications, Vol. COM-35, No. 11, November 1987, which is hereby incorporated by reference in its entirety.

FIG. 1is a block diagram of exemplary 2 Tx antenna and M Rx antenna spatially multiplexed wireless communication system with receiver channel estimation, in accordance with an embodiment of the invention. Referring toFIG. 1, the wireless system100may comprise a dedicated physical channel (DPCH) block127, a plurality of mixers128,130and132, a plurality of combiners134and136, a first transmit antenna (Tx1)138and an additional transmit antenna (Tx2)140on the transmit side. On the receive side, the wireless system100may comprise a plurality of receive antennas1061 . . . M, a single weight generator (SWG)110, a plurality of RF blocks1141 . . . P, a plurality of chip matched filters (CMF)1161 . . . P, a spatially multiplexed baseband (SMBB) processor126and a single weight generator baseband processor (SWGBB)121. The SWGBB121may comprise a channel estimator122and a single weight generator (SWG) algorithm block124.

The DPCH127may be adapted to receive a plurality of input channels, for example, a dedicated physical control channel (DPCCH) and a dedicated physical data channel (DPDCH). The DPCH127may simultaneously control the power of DPCCH and DPDCH. The mixer128may be adapted to mix the output of DPCH127with a spread and/or scrambled signal to generate a spread complex valued signal that may be input to mixers130and132. The mixers130and132may weight the complex valued input signals with weight factors W1and W2, respectively, and may generate outputs to a plurality of combiners134and136respectively. The combiners134and136may combine the outputs generated by mixers130and132with common pilot channel1(CPICH1) and common pilot channel2(CPICH2) respectively. The common pilot channels1and2may have a fixed channelization code allocation that may be utilized to measure the phase amplitude signal strength of the channels. The weights W1and W2may be utilized, for example, phase and or amplitude adjustments and may be generated by the single weight generator (SWG) algorithm block124. The antennas138and140may receive the generated outputs from the combiners134and136and may transmit wireless signals.

The plurality of receive antennas1061 . . . Mmay each receive at least a portion of the transmitted signal. The SWG110may comprise suitable logic, circuitry, and/or code that may be adapted to determine a plurality of weights to be applied to each of the input signals R1 . . . M. The SWG110may be adapted to modify the phase and amplitude of a portion of the transmitted signals received by the plurality of receive antennas1061 . . . Mand generate a plurality of output signals RF1 . . . P.

The plurality of RF blocks1141 . . . Pmay comprise suitable logic, circuitry, and/or code that may be adapted to process an RF signal. The RF blocks1141 . . . Pmay perform, for example, filtering, amplification, and analog-to-digital (A/D) conversion operations. The plurality of transmit antennas138and140may transmit the processed RF signals to a plurality of receive antennas1061 . . . M. The single weight generator SWG110may comprise suitable logic, circuitry, and/or code that may be adapted to determine a plurality of weights, which may be applied to each of the input signals. The single weight generator SWG110may be adapted to modify the phase and amplitude of at least a portion of the signals received by the plurality of receive antennas1061 . . . Mand generate a plurality of output signals RF1 . . . P. The plurality of RF receive blocks1141 . . . Pmay comprise suitable logic, circuitry and/or code that may be adapted to amplify and convert the received analog RF signals RF1 . . . Pdown to baseband. The plurality of RF receive blocks1141 . . . Pmay each comprise an analog-to-digital (A/D) converter that may be utilized to digitize the received analog baseband signal.

The plurality of chip matched filters (CMF)1161 . . . Pmay comprise suitable logic, circuitry and/or code that may be adapted to filter the output of the plurality of RF receive blocks1141 . . . Pso as to produce in-phase (I) and quadrature (Q) components (I, Q). In this regard, in an embodiment of the invention, the plurality of chip matched filters (CMF)1161 . . . Pmay comprise a pair of digital filters that are adapted to filter the I and Q components to within the bandwidth of WCDMA baseband (3.84 MHz). The outputs of the plurality of chip matched filters (CMF)1161 . . . Pmay be transferred to the SMBB processor126.

The SMBB126may be adapted to receive a plurality of in-phase and quadrature components (I, Q) from a plurality of chip matched filters (CMF)1161 . . . Pand generate a plurality of baseband combined channel estimates {circumflex over (h)}1to {circumflex over (h)}P. The SMBB126may be adapted to generate a plurality of estimates {circumflex over (X)}1to {circumflex over (X)}Pof the original input spatial multiplexing sub-stream signals or symbols X1to XP. The SMBB126may be adapted to separate the different space-time channels utilizing a Bell Labs Layered Space-Time (BLAST) algorithm, for example, by performing sub-stream detection and sub-stream cancellation. The capacity of transmission may be increased almost linearly by utilizing the BLAST algorithm.

The channel estimator122may comprise suitable logic, circuitry, and/or code that may be adapted to process the received estimates {circumflex over (h)}1to {circumflex over (h)}Pfrom the SMBB processor126and may generate a matrix Ĥ of processed estimated channels that may be utilized by the single weight generator (SWG) algorithm block124.

The SWG algorithm block124may determine a plurality of amplitude and phase values Aiand φI, respectively, which may be utilized by SWG110to modify the phase and amplitude of a portion of the transmitted signals received by the plurality of receive antennas1061 . . . Mand generate a plurality of output signals RF1 . . . P.

FIG. 2ais a flow diagram illustrating exemplary steps for channel estimation in a 2-Tx and M-Rx antennas wireless communication system, in accordance with an embodiment of the invention. Referring toFIG. 2a, after start step202, in step204, the single channel communication signals, sT, may be transmitted from the transmit antennas Tx_1138and Tx_2140inFIG. 1. In step206, the first and additional receive antennas,1061 . . . M, may receive a portion of the transmitted single channel communication signals. In step208, the signals received by the additional receive antennas1061 . . . Mmay be multiplied by, for example, rotation waveforms, such as sine, square, or triangular waveforms for example, in the SWG110. In this regard, the rotation waveforms may have a given set of amplitude and phase component values. In step210, the SWG110may combine output of the receive antennas1061 . . . Mmultiplied by the rotation waveforms and generate the single channel communication signal, sRC.

In step212, the SMBB126may determine a plurality of baseband combined channel estimates {circumflex over (h)}1to {circumflex over (h)}P. In step214, the SWG channel estimator122in the SWBBG121may determine the matrix Ĥ of propagation channel estimates. In this regard, the propagation channel estimates {circumflex over (h)}1to {circumflex over (h)}Pmay be determined concurrently. In step216, the pre-equalizer125may calculate or determine the pre-equalization weight parameters or weight factors W1and W2that may be applied to the mixers130and132inFIG. 1respectively. The pre-equalization weights W1and W2may be transferred to a transmitter, such as a base station, to pre-equalize the signals being transmitted from the transmit antennas Tx_1138and Tx_2140.

In step218, the wireless communication system150may determine whether a closed loop operating mode that supports transmit diversity modes CL1and CL2is active. When the closed loop operating mode is active, the process may proceed to step224. In step224, the (M−1) maximum SINR channel weights that comprise amplitude and phase components, A1to AM-1and φ1to φM-1, may be generated concurrently with the diversity pre-equalization weight parameters supported by CL1or CL2. The channel weights may be based on the propagation channel estimates determined after the application of pre-equalization weight parameters W1and W2to the transmitter. The diversity pre-equalization weight parameters supported by CL1or CL2may be transferred to a transmitter, such as a base station, to pre-equalize the signals being transmitted from the transmit antennas Tx_1138and Tx_2140. After step224, the process may proceed to step222.

Returning to step218, when the closed loop operating mode is not active, the process may proceed to step220. In step220, the SWG algorithm block124may generate the (M−1) maximum SINR channel weights that comprise amplitude and phase components, A1to AM-1and φ1to φM-1. In step222, the (M−1) maximum SINR channel weights may be applied to the SWG110.

After steps222or224, the process may proceed to end step226where additional single channel communication signals received may be phase and amplitude adjusted based on the maximum SINR channel weights applied to the mixers SWG110. The channel estimation phase rotation and the maximum SINR phase/amplitude adjustment described in flow chart200may be performed continuously or may be performed periodically.

FIG. 2billustrates an exemplary periodic phase rotation for an I signal component, in accordance with an embodiment of the invention. Referring toFIG. 2b, for the wireless system100inFIG. 1, by rotating the phase at the receive antennas1061 . . . Mfrom 0 to 360 degrees, it may be possible to estimate propagation channels, h1 . . . M, at the same time utilizing complex multiplication and integration. This operation is equivalent to orthogonalizing all the channels at the Rx antennas.FIG. 2billustrates the periodic rotation of the I component in an RF signal.

FIG. 3Ais a block diagram of an exemplary system for providing phase rotation, channel estimation and for determining optimal phase and amplitude parameters or settings for an additional receive antenna, in accordance with an embodiment of the invention. Referring toFIG. 3A, a receiver system300may comprise a first receive antenna Rx1302, an additional antenna Rx2304, a combiner306, a complex multiplier308, and a single weight generator baseband (SWGBB) processor310. The SWGBB processor310may comprise a phase rotation start controller block314, a delay block316, a SWG channel estimator318, a single weight generator (SWG) algorithm block320, a RF phase and amplitude controller312and a pre-equalizer322. The SWGBB processor310provides similar functionality as the SMBB processor126inFIG. 1.

The receive antennas Rx1302and Rx2304may each receive a portion of the transmitted signal. The combiner306may be adapted to combine the received signals into a single RF signal RF1, for example. The complex multiplier308may be adapted to receive a plurality of input signals from the additional receive antenna Rx2304and the RF phase and amplitude controller312and may generate an output signal to the combiner306.

The phase rotation start controller block314may comprise suitable logic, circuitry and/or that may be adapted to start after receiving a reset signal and may generate a plurality of output signals to the delay block316and the RF phase and amplitude controller312. The delay block316may be adapted to receive an input signal from the phase rotation start controller block314and generate a delayed output signal to the SWG channel estimator318. The SWG channel estimator318may comprise suitable logic, circuitry, and/or code that may be adapted to process the received baseband combined channel estimates per transmit antenna ĥ1. . . ĥNfrom the SMBB processor126and may generate a matrix Ĥ2×Nof processed estimated channels. The SWG channel estimator318may be adapted to generate an algorithm start signal indicating the end of integration that may be utilized by the single weight generator (SWG) algorithm block320.

The SWG algorithm block320may be adapted to receive a plurality of signals from the SWG channel estimator318, for example, a matrix Ĥ2×Nof processed baseband combined channel estimates, an algorithm start signal from the SWG channel estimator318and a noise power estimation signal. The SWG algorithm block320may generate phase and amplitude correction signals and an algorithm end signal to the RF phase and amplitude controller312. The RF phase and amplitude controller312may be adapted to receive the phase and amplitude values and the algorithm end signal to modify the phase and amplitude of a portion of the transmitted signals received by the receive antenna Rx2302and generate an output signal RF1. The pre-equalizer322may comprise suitable logic, circuitry, and/or code that may be adapted to determine a plurality of pre-equalization parameters based on the matrixH2×Nof propagation channel estimates ĥ11. . . ĥ1N, ĥ21. . . ĥ2N. In this regard, the pre-equalizer may be adapted to generate pre-equalization weight parameters or weight factors W1and W2and/or closed loop diversity pre-equalization weight parameters.

The SWG channel estimator318may receive baseband combined channel estimates ĥ1. . . ĥN, which may include all transmission channels from N Tx antennas and each Tx antenna may have a different channel estimation sequence, so that the different combined channels ĥ1. . . ĥNmay be separated and estimated. The SWG channel estimator318may generate a matrix of channel estimates Ĥ2×Nto the SWG algorithm block320. A reset signal may be utilized to start the phase rotation block314. The combined channel estimates from the SMBB126inFIG. 1may be transferred to the channel estimator318for processing. When processing is complete, the SWG channel estimator318may indicate to the SWG algorithm block320that the determination of the appropriate phase and amplitude correction for the portion of the received signal in the additional antenna Rx2304may start. The SWG algorithm block320may utilize an estimation of the noise power and interference in determining the phase and amplitude values in addition to the matrix of channel estimates Ĥ2×N. The SWG algorithm block320may indicate to the RF phase and amplitude controller312the end of the weight determination operation and may then transfer to the RF phase and amplitude controller312, the determined phase and amplitude values. The RF phase and amplitude controller312may then modify the portion of the received signal in the additional antenna Rx2304via the complex multiplier308.

In operation, the RF phase and amplitude controller312may apply the signal ejwrtto the mixer308inFIG. 3Abased on control information provided by the phase rotator start controller314. The switch340may select the rotation waveform source342based on the control information provided by the phase rotator start controller314. Once the channel weights are determined by the SWG algorithm block320and the phase and amplitude components have been transferred to the RF phase and amplitude controller312, the algorithm end signal may be utilized to change the selection of the switch340. In this regard, the switch340may be utilized to select and apply the signal Aejφto the mixer308inFIG. 3A.

FIG. 3Bis a block diagram of an exemplary system for providing phase rotation, channel estimation and for determining optimal phase and amplitude parameters or setting for additional K-1 receive antennas, in accordance with an embodiment of the invention. Referring toFIG. 3B, a receiver system330may correspond to a portion of the wireless communication system100inFIG. 1and may differ from the receiver system300inFIG. 3Ain that (K-1) additional receive antennas, Rx_2304to Rx_K305, and (K-1) mixers308to309may be utilized. The combiner306may combine the received signals into a single RF signal RF1, for example. In this regard, the SWG channel estimator318may be adapted to process the combined channel estimates, ĥ1. . . ĥN, and determine the propagation channel matrix estimate ĤK×N.

Referring to theFIG. 1, multiple receive antennas may be connected to each of the RF chains RF1. . . RFNas shown inFIG. 3Bfor the single RF chain RF1. In this regard, the combined channel estimates ĥ1. . . ĥNand consequently the channel estimate matrix ĤK×Nmay be determined per each RF chain RF1. . . RFN. Consequently, following this example, N matrices ĤK×Nmay form a channel estimate matrix ĤMxNinFIG. 1(M=NK).

The SWG algorithm block320may also be adapted to determine (K-1) channel weights per RF chain, that may be utilized to maximize receiver SINR, for example, to be applied to the mixers308to309to modify the portions of the transmitted single channel communication signals received by the additional receive antennas Rx_2304to Rx_K305. The (K-1) channel weights per RF chain may comprise amplitude and phase components, A1to AK-1and φ1to φK-1. The RF phase and amplitude controller312may also be adapted to apply rotation waveforms ejwr1tto ejwr(K-1)tor phase and amplitude components, A1to AK-1and φ1to φK-1, to the mixers308to309. In this regard, the RF phase and amplitude controller312may apply the rotation waveforms or the amplitude and phase components in accordance with the control signals provided by the phase rotator start controller314and/or the algorithm end signal generated by the SWG algorithm block320. The pre-equalizer322inFIG. 3Bmay also be adapted to determine a plurality of pre-equalization parameters based on the matrix ĤK×Nof propagation channel estimates ĥ11. . . ĥ1N, ĥ21. . . ĥ2N, . . . ,ĥK1. . . ĥKN.

FIG. 3Cis a block diagram of an exemplary RF phase and amplitude controller, in accordance with an embodiment of the invention. Referring toFIG. 3C, the RF phase and amplitude controller312may comprise a switch340, rotation waveform sources342, and a plurality of SWG algorithm determined weights344. The switch340may comprise suitable hardware, logic, and/or circuitry that may be adapted to select between the rotation waveforms ejwr1tto ejwr(K-1)tand the SWG algorithm determined weights A1ejφ1to AK-1ejφK-1. The rotation waveform source342may comprise suitable hardware, logic and/or circuitry that may be adapted to generate the signal ejwrkr, where wrk=2πfrkand frkis the rotation frequency that preserves orthogonality of the received signals at the multiple receiving antennas. The rotation frequency that preserves the signal orthogonality at the receiving antennas may be selected as wrk=kwrwhere k=1, 2, 3 . . . K-1. Other rotation waveforms such as triangular or square may be utilized with the same frequency relationships. In addition, waveforms representing different orthogonal codes of the same frequency may be utilized, similar to the CDMA orthogonal codes with the same spreading. In this embodiment ejwrktis used as an exemplary waveform. The weights344may comprise suitable hardware, logic, and/or circuitry that may be adapted to generate the signals A1ejφ1to AK-1ejφK-1from the amplitude and phase components, A1to AK-1and φ1to φK-1, respectively.

In operation, the RF phase and amplitude controller312may apply the signals ejwr1tto ejwr(K-1)tto the mixers308to309inFIG. 3Bbased on control information provided by the phase rotator start controller314. The switch340may select the rotation waveform source342based on the control information provided by the phase rotator start controller314. Once the channel weights are determined by the SWG algorithm block320and the phase and amplitude components have been transferred to the RF phase and amplitude controller312, the algorithm end signal may be utilized to change the selection of the switch340. In this regard, the switch340may be utilized to select and apply the signals A1ejφ1to AK-1ejφM-1to the mixers308to309inFIG. 3B.

FIG. 4ais a flow diagram illustrating exemplary steps in the operation of the single weight baseband generator (SWBBG) that may be utilized in a 2-Tx and M-Rx antennas system, in accordance with an embodiment of the invention. Referring toFIG. 4a, after start step402, in step404, the phase rotator start controller314inFIG. 3Bmay receive the reset signal to initiate operations for determining propagation channel estimates and channel weights in the SWBBG310. The phase rotator start controller314may generate control signals to the delay block316and to the RF phase and amplitude controller312. The control signals to the delay block316may be utilized to determine a delay time to be applied by the delay block316. The control signals to the RF phase and amplitude controller312may be utilized to determine when to apply the rotation waveforms or the channel weights determined by the SWG algorithm block124to the mixers308to309inFIG. 3B, for example.

In step406, the RF phase and amplitude controller312may apply the signals ejwr1tto ejwr(K-1)tto the mixers308to309inFIG. 3B. In step408, the delay block316may apply a time delay signal to the SWG channel estimator318to reflect the interval of time that may occur between receiving the combined channel estimates, {circumflex over (h)}1and {circumflex over (h)}2, modified by the rotation waveform and the actual rotating waveform at the mixer308. For example, the time delay signal may be utilized as an enable signal to the SWG channel estimator318, where the assertion of the time delay signal initiates operations for determining propagation channel estimates. In step410, the SWG channel estimator318may process the first and second baseband combined channel estimates, {circumflex over (h)}1and {circumflex over (h)}2, and may determine the matrix Ĥ2×Nof propagation channel estimates ĥ11to ĥ1Nand ĥ21to ĥ2N. The SWG channel estimator318may transfer the propagation channel estimates ĥ11to ĥ1Nand ĥ21to ĥ2Nto the SWG algorithm block320. In step412, the pre-equalizer322may calculate or generate the pre-equalization weight parameters or weight factors W1and W2. The pre-equalization weight parameters may be transferred to a wireless transmitter, such as a base station.

In step414, the receiver system330inFIG. 3Bmay determine whether a closed loop operating mode that supports transmit diversity modes CL1and CL2is active. When the closed loop operating mode is active, the process may proceed to step418. In step418, the (M-1) maximum SNIR channel weights that comprise amplitude and phase components, A1to AK-1and φ1to φK-1, may be generated concurrently with the diversity pre-equalization weight parameters supported by CL1or CL2. The channel weights may be based on the propagation channel estimates determined after the application of pre-equalization weight parameters W1and W2to the transmitter. The diversity pre-equalization weight parameters supported by CL1or CL2may be transferred to a transmitter, such as a base station, to pre-equalize the signals being transmitted. After step418, the process may proceed to step420.

Returning to step414, when the closed loop operating mode is not active, the process may proceed to step416. In step416, the SWG algorithm block320may generate the (M-1) maximum SNIR channel weights that comprise amplitude and phase components, A1to AK-1and φ1to φK-1, based on the propagation channel estimates ĥ11to ĥ1Nand ĥ21to ĥ2Nand/or noise power estimates and interference channel estimates, for example. The SWG algorithm block320may transfer the channel weights to the RF phase and amplitude controller312. The SWG algorithm block320may generate the algorithm end signal to indicate to the RF phase and amplitude controller312that the channel weights are available to be applied to the mixers308to309. In step420, RF phase and amplitude controller312may apply the maximum SNIR weights with phase and amplitude components, A1to AK-1and φ1to φK-1, to the mixers308to309inFIG. 3B, in accordance with the control signals provided by the phase rotator start controller314and/or the SWG algorithm block320.

In step422, the receiver system330inFIG. 3Bmay determine whether the phase rotation operation on the received single channel communication signals is periodic. When the phase rotation operation is not periodic but continuous, the process may proceed to step408where a new delay may be applied to the SWG channel estimator318. In instances when the phase rotation operation is periodic, the process may proceed to step424where the receiver system330may wait until the next phase rotation operation is initiated by the reset signal. In this regard, the process may return to step404upon assertion of the reset signal on the phase rotator start controller314.

FIG. 4bis a flow diagram illustrating exemplary steps for determining channel weights in additional receive antennas utilizing signal-to-noise ratio (SNR) or signal-to-interference-and-noise ratio (SINR), in accordance with an embodiment of the invention. Referring toFIG. 4b, after start step452, in step454, the SWG algorithm block320may determine whether the signals received in the receive antennas are noise limited. The SWG algorithm block320may receive noise statistics and/or other noise information from either the CPP5161 . . . P(FIG. 5) and/or from the spatial multiplexing processor518. When the received signals are noise limited, the flow diagram control may proceed to step458. In step458, the SWG algorithm block320may generate models for the received signals. For example, the models for a 1-Tx and 2-Rx antennas system may be represented by the following expressions:
r1=h1s+n1,
r2=Aejθh2s+Aejθn2, and
y=r1+r2=s(h1+Aejθh2)+n1+Aejθn2,
where r1may represent a model of the signal received in a first receive antenna, r2may represent a model of the signal received in the second receive antenna, s may represent the transmitted signal, and n1may represent a noise component at the first receive antenna, whose time varying impulse response is represented by h1. The parameter n2may represent a noise component at the second receive antenna, whose time varying impulse response is represented by h2, θ may represent the phase factor between the signal received in the first and second receive antennas, and A may represent an amplitude factor. The parameter y may represent the sum of the received signal models and may comprise a combined signal component s(h1+Aejθh2) and a combined noise component n1+Aejθn2.

For the case of a MIMO system with N-transmit and M-receive antennas, the models may be represented by the expressions:

rk=∑i=1N⁢⁢(Ak⁢ⅇj⁢⁢ϑ⁢khik⁢s+Ak⁢ⅇj⁢⁢ϑ⁢knk),⁢y=∑k=1M⁢(rk),
where rkmay represent the model of the signal received from the N transmit antennas by the kthreceive antenna, hikmay represent the time varying impulse response of the propagation channel between the ithtransmit antenna and the kthreceive antenna, and s may represent the transmitted signal, nkmay represent a noise component at the kthreceive antenna. The parameter Akmay correspond to the amplitude factor associated with the kthreceive antenna, θkmay correspond to the phase factor associated with the kthreceive antenna, and y may represent the sum of the M received signal models. In this regard, Ak(k=1)=1 and θk(k=1)=0.

In step460, the received signal models may be utilized to determine a signal strength parameter. In this regard, the signal-to-noise ratio (SNR) may correspond to the signal strength parameter to be determined. For example, for a 1-Tx and 2-Rx antennas system, the SNR may be determined by maximizing the following expression for various phase, θ, and amplitude, A, factors:

SNR=h1+A⁢⁢ⅇj⁢⁢ϑ⁢h22E⁢n12+E⁢A⁢⁢ⅇj⁢⁢ϑ⁢n22=h1+A⁢⁢ⅇj⁢⁢ϑ⁢h22σ2⁡(1+A2).
The SNR numerator may correspond to the y parameter's combined signal component while the SNR denominator may correspond to the y parameter's combined noise component. The phase factor, θ, may be selected, for example, from a 360-degrees phase rotation while the amplitude factor, A, may be selected, for example, from an set amplitude range. In one embodiment of the invention, the phase factor may be varied in a plurality of phase factor steps over the 360-degrees phase rotation to find the maximum SNR value. In another embodiment of the invention, the phase factor may be varied in a plurality of phase factors steps over the 360-degrees phase rotation and the amplitude factor may be varied in a plurality of amplitude factor values over the amplitude range to find the maximum SNR value.

In step470, after determining the maximum SNR in step460, the SWG algorithm block320may utilize the amplitude factor and phase factor that corresponds to the maximum SNR to determine the amplitude and phase to be provided to the RF amplitude and phase controller312in step470. For example, in one embodiment of the invention, the amplitude and/or phase factors that correspond to the maximum SNR may be utilized as the amplitude and phase to be transferred to the RF amplitude and phase controller312. After application of the appropriate amplitude and phase by the RF amplitude and phase controller312to the receive antennas, the flow diagram control may proceed to end step472until a next phase and amplitude determination is necessary.

Returning to step454, when received signals are not noise limited, the flow control may proceed to step456where a determination may be made as to whether multiple interfering signals may be present and may need to be considered during channel weight determination. When a single interferer is considered, the flow diagram control may proceed to step462. In step462the SWG algorithm block320may generate models for the received signals. For example, the models for a 1-Tx and 2-Rx antennas system may be represented by the following expressions:
r1=h1s+hI1sI+n1,
r2=Aejθ(h2s+hI2sI+n2), and
y=r1+r2=s(h1+Aejθh2)+n1+sI(hI1+AejθhI2)+Aejθn2,
where r1may represent a model of the signal received in a first receive antenna, r2may represent a model of the signal received in the second receive antenna, s may represent the transmitted signal, sImay represent the interference signal, and n1may represent a noise component at the first receive antenna whose time varying impulse response is h1. The parameter n2may represent a noise component at the second receive antenna whose time varying impulse response is h2, θ may represent the phase factor between the signal received in the first and second receive antennas, and A may represent an amplitude factor. Moreover, the time varying impulse response hI1may correspond to the propagation channel between the interference signal source and the first receive antenna and the time varying impulse response hI2may correspond to the propagation channel between the interference signal source and the second receive antenna. The parameter y may represent the sum of the received signal models and may comprise a combined signal component s(h1+Aejθh2) and a combined noise plus interference component n1+sI(hI1+AejθhI2)+Aejθn2.

For the case of a MIMO system with N-transmit and M-receive antennas, the models may be represented by the expressions:

rk=∑i=1N⁢(Ak⁢ⅇj⁢⁢ϑ⁢khik⁢s+Ak⁢ⅇj⁢⁢ϑ⁢khlk⁢sl+Ak⁢ⅇj⁢⁢ϑ⁢knk),⁢y=∑k=1M⁢(rk),
where rkmay represent the model of the signal received from the N transmit antennas by the kthreceive antenna, hikmay represent the propagation channel between the ithtransmit antenna and the kthreceive antenna, s may represent the transmitted signal, sImay represent the interference signal, nkmay represent a noise component at the kthreceive antenna, hIkmay represent the time varying impulse response of propagation channel between the interference source and the kthreceive antenna. The parameter Akmay correspond to the amplitude factor associated with the kthreceive antenna, θkmay correspond to the phase factor associated with the kthreceive antenna, and y may represent the sum of the M received signal models. In this regard, Ak(k=1)=1 and θk(k=1)=0.

In step464, the received signal models may be utilized to determine a signal strength parameter. In this regard, the signal-to-interference-and-noise ratio (SINR) may correspond to the signal strength parameter to be determined. For example, for a 1-Tx and 2-Rx antennas system, the SINR may be determined by maximizing the following expression for various phase, θ, and amplitude, A, factors:

SINR=h1+A⁢⁢ⅇj⁢⁢ϑ⁢h22E⁢n12+E⁢A⁢⁢ⅇj⁢⁢ϑ⁢n22+hl1+A⁢⁢ⅇj⁢⁢ϑ⁢hl22=h1+A⁢⁢ⅇj⁢⁢ϑ⁢h22σ2⁡(1+A2)+hl1+A⁢⁢ⅇj⁢⁢ϑ⁢hl22.
where σ2is the noise power. The above SINR equations may be easily extended, by one skilled in art, to the single channel MIMO case.

The transmit antennas may include CL1or CL2transmit diversity weights. The joint transmit-received solution may be formed in that case that may include the transmit CL weights and the additional transmit antenna channel components in the SINR numerator. The SINR numerator may correspond to the y parameter's combined signal component while the SINR denominator may correspond to the y parameter's combined noise plus interference component. The phase factor, □, may be selected, for example, from a 360-degrees phase rotation while the amplitude factor, A, may be selected, for example, from an set amplitude range. In one embodiment of the invention, the phase factor may be varied in a plurality of phase factor steps over the 360-degrees phase rotation to find the maximum SNR value. In another embodiment of the invention, the phase factor may be varied in a plurality of phase factors steps over the 360-degrees phase rotation and the amplitude factor may be varied in a plurality of amplitude factor values over a range of amplitudes to find the maximum SINR value.

After determining the SINR in step464, the SWG algorithm block320may determine the amplitude and phase to be provided to the RF amplitude and phase controller312in step470. After application of the appropriate amplitude and phase by the RF amplitude and phase controller312, the flow diagram control may proceed to end step472until a next phase and amplitude determination is necessary.

Returning to step456, when multiple taps or multiple paths in the channel impulse response may need to be considered, the flow diagram control may proceed to step466. In step466, the SWG algorithm block320may generate the received signal models for cases in which multiple taps or interference sources are considered. In step468, the SWG algorithm block320may utilize the received signal models to determine the SINR for multiple interferers. When the desired signal has i=1, . . . , P taps or multiple paths with different delays and the interfering signal has k=1, . . . , R taps or multiple paths with different delays, then the maximum SINR solution for the 1-Tx and 2-Rx antenna system in that case may be as follows:

SINRmax=∑i=1P⁢h1+A⁢⁢ⅇj⁢⁢ϑ⁢h22σ2⁡(1+A2)+∑k=1R⁢hl1+A⁢⁢ⅇj⁢⁢ϑ⁢hl22.
The SINRmaxexpression may be extended to the single channel MIMO case with or without CL transmit antenna weights.

After determining the SINR in step468, the SWG algorithm block320may determine the amplitude and phase to be provided to the RF amplitude and phase controller312in step470. After application of the appropriate amplitude and phase by the RF amplitude and phase controller312, the flow diagram control may proceed to end step472until a next phase and amplitude determination is necessary.

The operations to maximize the signal strength described for steps460,464, and468may be based on a search algorithm. In an exemplary embodiment of the invention, a search algorithm may be utilized to search over 360-degrees phase rotation in 45-degree or 90-degree phase factor steps and over a 0-5 amplitude range in 0.25 amplitude values or steps, for example. For a 1-Tx and 2-Rx antenna system, with 90-degree phase factor steps, a phase only search algorithm may calculate 4 SNR or SINR values, for example. For a 2-Tx and 2-Rx antenna system with STTD transmit mode, with 90-degree phase factor steps, a phase only search algorithm may calculate 4 SNR or SINR values. For a 2-Tx and 2-Rx antenna system with the CL1diversity mode, with 90-degree phase factor steps at both receiver and transmitter, a phase only search algorithm may calculate 4×4=16 SNR or SINR values. For a 2-Tx and 2-Rx antenna system with the CL2diversity mode, with 90-degree phase factor steps at the receiver and 45-degree phase factor steps and two power scaling weight levels at the transmitter, a phase only search algorithm may calculate 4×8×2=64 SNR or SINR values, for example. The maximum value generated by the algorithm may be the output of the search algorithm.

In another embodiment of the invention, a closed-form mathematical expression may also be utilized to maximize the SNR and/or the SINR. Utilizing an algorithm or closed-form expression that maximizes the SINR or SNR may provide a good compromise between implementation complexity and performance gains. Notwithstanding, the invention is not limited in this regard, and other channel weight algorithms may also be utilized.

FIG. 5is a block diagram of exemplary N Tx antenna and M Rx antenna with pre-equalization in a single weight single channel spatial multiplexing wireless communication system, in accordance with an embodiment of the invention. Referring toFIG. 5, there is shown a transceiver system500that comprises a baseband pre-equalizer and pre-coding block502, a plurality of RF transmit blocks5041 . . . N, a plurality of transmit antennas5061 . . . N, a plurality of receive antennas5081 . . . M, a single weight generator (SWG)510, a plurality of RF receive blocks5121 . . . P, a plurality of chip matched filters (CMF)5141 . . . P, a plurality of cluster path processors CPP5161 . . . P, a spatial multiplexing baseband processor (SMBB)518and a single weight generator baseband processor (SWGBB)521. The SWGBB521may comprise a single weight generator (SWG) channel estimation block520, a pre-equalization weight calculation block522and a single weight generator (SWG) algorithm block524.

The baseband pre-equalizer and pre-coding block502at the transmitter may contain suitable logic, code and/or circuitry to process a plurality of received weights generated by the pre-equalization weight calculation block522to convolve with the transmitted signal. At least a portion of the generated plurality of pre-equalization weights may be fed back to the base station for modifying subsequently transmitted spatially multiplexed communication signals which are transmitted from at least a portion of the plurality of transmit antennas at the base station. The pre-equalization weights may be based on the propagation channel estimates and may be determined by utilizing least-mean squares (LMS), recursive least squares (RLS), or a cost function analysis. The pre-equalization weights may be fed back to a transmitter via an uplink channel. The various embodiments of the invention may provide a good compromise between implementation complexity and performance gains to reduce the effects of, for example, inter-symbol interference (ISI) and/or inter-carrier interference (ICI) in MIMO systems.

The baseband pre-equalizer and pre-coding block502may generate a frequency selective signal by utilizing a 2D filtering process that may comprise, for example, matrix multiplication of the calculated weights and the transmitted data sequences and effectively transform the channel from a frequency selective channel to a flat fading channel. In this regard, the baseband pre-equalizer and pre-coding block502may be adapted to utilize, for example, an adaptive algorithm to adaptively calculate weights and iteratively search for an optimal weight solution. In accordance with an embodiment of the invention, the baseband pre-equalizer and pre-coding block502may be adapted to utilize, for example, a least mean square (LMS) algorithm for the weight calculation. Notwithstanding, the invention is not limited in this regard, and other weight calculation algorithms may be utilized.

The RF transmit blocks5041 . . . Nmay comprise suitable logic, circuitry, and/or code that may be adapted to process an RF signal. The RF transmit blocks5041 . . . Nmay perform, for example, filtering, amplification, and analog-to-digital (A/D) conversion operations. The plurality of transmit antennas5061 . . . Nmay transmit the processed RF signals from the plurality of RF transmit blocks5041 . . . Nto a plurality of receive antennas5081 . . . M. The single weight generator (SWG)510may comprise suitable logic, circuitry, and/or code that may be adapted to determine a plurality of weights to be applied to each of the input signals R1 . . . Mto modify the phase and/or amplitude of at least a portion of the signals transmitted from a base station and received by the plurality of receive antennas5081 . . . Mand generate a plurality of output signals RF1 . . . P. The plurality of RF receive blocks5121 . . . Pmay comprise suitable logic, circuitry and/or code that may be adapted to amplify and convert the received analog RF signals RF1 . . . Pdown to baseband. The plurality of RF receive blocks5121 . . . Pmay each comprise an analog-to-digital (A/D) converter that may be utilized to digitize the received analog baseband signal.

The plurality of chip matched filters (CMF)5141 . . . Pmay comprise suitable logic, circuitry and/or code that may be adapted to filter the output of the plurality of RF receive blocks5121 . . . Pso as to produce in-phase (I) and quadrature (Q) components (I, Q). In this regard, in an embodiment of the invention, the plurality of chip matched filters (CMF)5141 . . . Pmay comprise a pair of digital filters that are adapted to filter the I and Q components to within the bandwidth of WCDMA baseband (3.84 MHz), for example.

The plurality of cluster pair processors CPP5161 . . . Pmay generate a plurality of channel estimates {circumflex over (h)}1Nto {circumflex over (h)}PNthat may correspond to the plurality of receive antennas5081 . . . M. U.S. application Ser. No. 11/173,854 provides a detailed description of signal clusters and is hereby incorporated herein by reference in its entirety. The SWG channel estimation block520may process these estimates {circumflex over (h)}1Nto {circumflex over (h)}PNand may generate a matrix Ĥ1×Mto ĤN×Mof processed baseband combined channel estimates that may be utilized by the pre-equalization weight calculation block522and the single weight generator (SWG) algorithm block524.

The SMBB518may be adapted to receive a plurality of in-phase and quadrature components (I, Q) from a plurality of chip matched filters (CMF)5141 . . . Pand a plurality of baseband combined channel estimates {circumflex over (h)}1Nto {circumflex over (h)}PNfrom a plurality of cluster path processors CPP5161 . . . Pto generate a plurality of channel estimates {circumflex over (X)}1to {circumflex over (X)}Pof the original input signals X1to XP. The SMBB518may be adapted to separate the different space-time channels utilizing a Bell Labs Layered Space-Time (BLAST) algorithm, for example, by performing sub-stream detection and sub-stream cancellation. The capacity of transmission may be increased almost linearly by utilizing the BLAST algorithm. The pre-equalization technique may improve the performance of the receiver by transforming the frequency selective channel to a flat fading channel.

The pre-equalization weight calculation block522may comprise suitable logic, circuitry and/or code that may be adapted to calculate the effective weights to be sent to the baseband pre-equalizer and pre-coding block502at the transmitter. The weight calculation may be based on a cost function or a second order statistical technique based on the pre-equalization method used. Certain pre-coding techniques may require less complicated processing on the receiver side. The pre-equalizer weight calculation block522may be adapted to determine the pre-equalization parameters based on, for example, a least-mean squares (LMS) algorithm, a recursive least squares (RLS) algorithm, direct matrix inversion, a cost function analysis, or a second order statistical technique.

When utilizing a cost function analysis, for example, coefficients utilized by the pre-equalizer to determine the pre-equalization parameters may be obtained based on the minimization of a cost function, J, of the form J=f(SINR) or J=f(SNR), where f(x) denotes a function of variable x and SINR and SNR are the signal-to-interference-and-noise ratio and signal-to-noise ratio of the received signals, respectively. For example, a cost function J=(SINR)−1may be minimized to obtain pre-equalizer coefficients that may be utilized to determine the pre-equalization parameters. The pre-equalizer may apply and/or modify cost function parameters associated with variables utilized with the cost function. In certain instances, pre-coding techniques may be utilized in order to require less complicated processing of the pre-equalization parameters on the receiver side.

The SWG algorithm block524may determine a plurality of phase and amplitude values Aiand φiwhich may be utilized by SWG510to modify the phase and amplitude of a portion of the transmitted signals received by the plurality of receive antennas5081 . . . Mand generate a plurality of output signals RF1 . . . P. The SWG algorithm block524may also be adapted to calculate the effective weights W1and W2to be transmitted to the baseband pre-equalizer and pre-coding block502at the transmitter. The weight calculation may be based on a cost function or a second order statistical technique based on the pre-equalization method used.

FIG. 6is a block diagram of an exemplary receiver illustrating spatial multiplexing in a MIMO communication system that may be utilized in connection with an embodiment of the invention. Referring toFIG. 6, there is shown a receiver600that comprises a plurality of receive antennas6101,2, . . . ,M, a plurality of amplifiers6121,2, . . . ,M, a SWG block614, a plurality of filters6201,2, . . . ,N, a local oscillator622, a plurality of mixers6241,2, . . . ,N, a plurality of analog to digital (A/D) converters6261,2, . . . ,Nand a spatial multiplexing baseband processor SMBB630.

The antennas6101,2, . . . ,Mmay be adapted to receive the transmitted signals. The amplifiers6121,2, . . . ,Mmay be adapted to amplify the M received input signals. The SWG block614may comprise a plurality of amplitude and phase shifters to compensate for the phase difference between various received input signals. Weights may be applied to each of the input signals A1 . . . Mto modify the phase and amplitude of a portion of the transmitted signals received by the plurality of receive antennas6121 . . . Mand generate a plurality of output signals RF1 . . . N. The plurality of filters6201,2, . . . ,Nmay be adapted to filter frequency components of the RF substreams. The mixers6241,2, . . . ,Nmay be adapted to downconvert the analog RF substreams to baseband. The local oscillator622may be adapted to provide a signal to the mixers6241,2, . . . ,N, which is utilized to downconvert the analog RF substreams to baseband. The analog to digital (A/D) converters6261,2, . . . ,Nmay be adapted to convert the analog baseband substreams into their corresponding digital substreams. The spatial multiplexing baseband processor SMBB630may be adapted to process the digital baseband substreams and multiplex the plurality of digital signals to generate output signals {circumflex over (X)}1. . . {circumflex over (X)}N, which may be estimates of the original signals X1. . . XN.

In operation, the MT RF signals transmitted by a plurality of transmitters may be received by a plurality of M receive antennas6101,2, . . . ,Mdeployed at the receiver600. Each of the M received signals may be amplified by a respective low noise amplifier6121,2, . . . ,M. A plurality of weights may be applied to each of the input signals A1 . . . Mto modify the phase and amplitude of a portion of the transmitted signals received by the plurality of receive antennas6121 . . . M. A plurality of output signals RF1 . . . Nmay be generated, which may be filtered by a plurality of filters6201,2, . . . ,N. The resulting N filtered signals may then be downconverted to baseband utilizing a plurality of N mixers6241,2, . . . ,N, each of which may be provided with a carrier signal that may be generated by a local oscillator622. The N baseband signals generated by the mixers6241,2, . . . ,Nmay then be converted to digital signals by a plurality of analog to digital (A/D) converters6261,2, . . . ,N. The N digital signals may further be processed by a spatial multiplexing baseband processor SMBB530to generate output signals or symbols {circumflex over (X)}1. . . {circumflex over (X)}Nwhich may be estimates of the original spatial multiplexing sub-stream signals or symbols X1. . . XN.

FIG. 7is a flowchart illustrating exemplary steps that may be utilized for pre-equalization in a spatially multiplexed wireless communication system, in accordance with an embodiment of the invention. Referring toFIG. 7, the exemplary steps may start at step700. In step702, a plurality of spatially multiplexed communication signals may be received from a plurality of transmit antennas. In step704, a plurality of vectors of baseband combined channel estimates may be generated based on phase rotation of the received plurality of spatially multiplexed communication signals. In step706, a plurality of pre-equalization weights may be generated based on the generated plurality of vectors of baseband combined channel estimates. In step708, the received plurality of spatially multiplexed communication signals may be modified based on a generated plurality of weights. In step710, the subsequent received plurality of spatially multiplexed communication signals may be modified before transmission from the plurality of transmit antennas by utilizing at least a portion of the generated plurality of pre-equalization weights. Control then passes to end step712.

Another embodiment of the invention may provide a machine-readable storage, having stored thereon, a computer program having at least one code section executable by a machine, thereby causing the machine to perform the steps as described above for pre-equalization in a single weight spatial multiplexing multi-input multi-output (MIMO) system.

In another embodiment of the invention, a plurality of receive antennas5081 . . . M(FIG. 5) may be adapted to receive a plurality of spatially multiplexed communication signals from a plurality of transmit antennas5061 . . . Nat a base station. A channel estimator, for example, the SWG channel estimation block520may generate a plurality of vectors of baseband combined channel estimates based on phase rotation of the received plurality of spatially multiplexed communication signals. For example, the channel estimator122may comprise suitable logic, circuitry, and/or code that may be adapted to process the received estimates {circumflex over (h)}1to {circumflex over (h)}Pfrom the SMBB processor126and may generate a matrix Ĥ of processed estimated channels that may be utilized by the single weight generator (SWG) algorithm block124.

At least one processor may generate a plurality of pre-equalization weights W1and W2based on the generated plurality of vectors of baseband combined channel estimates ĤN×M. At least one of the processors may be adapted to modify the received plurality of spatially multiplexed communication signals based on a generated plurality of weights Aiand φi. At least one of the processors may be adapted to feed back at least a portion of the generated plurality of pre-equalization weights to the base station for modifying subsequently transmitted spatially multiplexed communication signals which are transmitted from at least a portion of the plurality of transmit antennas at the base station.

In another embodiment of the invention, at least one of the processors may generate the pre-equalization parameters based on least mean squares (LMS) algorithm, recursive least squares (RLS) algorithm, direct matrix inversion, and/or a cost function. In this regard, the parameters of the cost function may be modified in accordance with the application. At least one of the processors may be adapted to generate the pre-equalization parameters periodically or continuously. The pre-equalization weights W1and W2may be fed back to a transmitter via an uplink channel. At least one of the processors may be adapted to spatially demultiplex the received plurality of spatially multiplexed communication signals.