Apparatus for defeating radar speed detection signals

Acquisition and interpretation of reflections from a target in response to a Doppler radar probe signal from a seeker are inhibited by providing at the target a receiver, preferably a high-speed sweeping transceiver, which is operative to injection lock quickly to the probe signal and thereafter to repeat a low-power replica of the probe signal with frequency modulation of the carrier from the repeater with a deviation greater than the locking bandwidth of the local oscillator to generate a random aperiodic signal. The vehicle operator is notified of the presence of seeker signals to prompt the operator to verify compliance with vehicle operating regulations. The carrier frequency modulation is selected to be of a frequency and deviation sufficient to confuse phase locking and limiting circuitry in a seeker receiver and thereby to inhibit acquisition of an echo from the target.

BACKGROUND OF THE INVENTION 
This invention relates to traffic radar detection and more specifically to 
down-the-road Doppler radar detection. 
In Doppler radar systems and particularly in down-the-road Doppler radar 
systems, where the axis of the antenna is directed along the line of 
travel of a target, there are two basic shortcomings with both enforcement 
and compliance with vehicle speed regulations. First, down-the-road 
Doppler radar systems are highly susceptible to improper target 
identification. Furthermore, Doppler radars are also highly susceptible to 
spurious target-speed readings. A Doppler radar's range typically exceeds 
800 meters with a half-power beamwidth of 0.21 to 0.31 radian, 
substantially more than the cross-sectional area of a vehicle. As a 
consequence, the radar operator must make a manual determination with a 
high degree of uncertainty as to the identity of the target vehicle. 
The problem of spurious speed readings is a phenomenon of electromagnetic 
and electromechanical interference effects from common sources, generally 
from AM or FM transmitters operating in or near the seeker source, 
including a seeker vehicle's own ignition system, ventilation equipment or 
the like. While vehicle radar detectors are in widespread use, 
conventional radar detectors are themselves subject to false responses due 
to spurious signals. 
One way to safeguard a targeted object against the potential shortcomings 
of Doppler radar is to attempt to defeat echoes or reflections of a seeker 
radar signal directed at the target. In connection with general 
countermeasures, such as used in military applications, two active 
approaches have been suggested: barrage jamming and spot jamming. Barrage 
jamming has been used in surface vehicle applications, albeit 
unsuccessfully. It is not known whether spot jamming has ever been used in 
surface vehicle applications. In barrage jamming, a transmit-only barrage 
signal spreads countermeasure energy substantially continuously over a 
bandwidth deemed sufficiently wide to include all possible seeker 
frequencies in order to mask target echoes. Such an approach presupposes 
the availability of substantial power, with that power spread over a 
substantial bandwidth. The barrage method is necessarily wasteful of 
energy and inhibits or even prevents detection of the presence of a seeker 
signal. 
Alternatively, a spot signal has been used where the operating frequency of 
the seeker signal was known or could be tracked. If the seeker signal is 
diverse, or if the exact frequency is not precisely known, there is a need 
to resort to a look-through scheme wherein a targeted receiver scans to 
locate an intruding seeker signal and thereafter causes energy to be 
transmitted on the specific frequency. Such a scheme is limited by the 
required time to acquire the seeker signal versus the seeker's dwell time 
at a given frequency and to generate a countermeasure signal at the same 
frequency. The sophistication of systems to perform such a task is often 
so costly that cost outweighs the value of the countermeasure. What is 
needed is a low-cost, effective and efficient countermeasure suitable for 
general use and which minimizes false positive responses to a seeker 
system while encouraging compliance of the target with vehicle operating 
regulations. 
SUMMARY OF THE INVENTION 
According to the present invention, acquisition and interpretation of 
signal echoes or reflections from a target in response to a Doppler radar 
probe signal from a seeker are inhibited by providing at the target a 
sweeping receiver which is prone to injection lock to the probe signal and 
thereafter to repeat a low-power replica of the probe signal with 
frequency modulation of the carrier from the repeater with a deviation 
greater than the locking bandwidth of the local oscillator. The vehicle 
operator is notified of the presence of seeker signals to prompt the 
operator to verify compliance with vehicle operating regulations. The 
characteristics of the frequency modulation is selected to be of a 
frequency and deviation sufficient to confuse phase locking and limiting 
circuitry in a seeker receiver and thereby to prevent acquisition of an 
echo from the target. The frequency modulation induces random aperiodic 
energy redistribution among FM side pairs lying within the passband of 
interest of the seeker receiver so that the repeated signal which is to 
mask the echoes appears to be frequency incoherent and variable in 
amplitude. Since seeker radar equipment typically inhibits the display of 
target information when interference is present, all positive indications 
are eliminated and locking is prevented to provide the target operator 
adequate time to verify compliance with vehicle operating regulations. 
In a specific embodiment, the X-band at 10,525 MHz transceiver employs a 
Gunn diode oscillator with a voltage tunable Varactor and a Gallium 
Arsenide mixer diode in a heterodyne arrangement to sweep over a 50 MHz 
bandwidth and to produce an injection locked repeated signal within an 8 
kHz bandwidth typical for a seeker receiver. 
In a further specific embodiment, a K-band transceiver at 24,150 MHz 
employs a voltage tunable Gunn-diode oscillator and Gallium Arsenide mixer 
diode in a heterodyne arrangement to sweep over a 200 MHz bandwidth and to 
produce an injection locked repeated signal within a 15 kHz receiver 
bandwidth. A loaded Q of about 25 is typical for a non-iris-coupled cavity 
oscillator. Carrier frequency modulation is initiated after injection lock 
to assure that lock is broken at the seeking receiver. In a specific 
embodiment, an audio frequency (4 kHz) tone is frequency modulated by a 
low-frequency (25 Hz) sawtooth which modulates the carrier of the target 
transmitter to produce apparent noise of 0 Hz to 15 kHz in the passband of 
an intermediate frequency stage of the seeker receiver. The apparent noise 
is similar to noise in the absence of a target echo. 
The invention will be explained with respect to K-band operation. However, 
observations, theory and conclusions apply with equal validity to X-band 
and Ka-band operation.

DESCRIPTION OF SPECIFIC EMBODIMENTS 
In order to understand the operation of the invention, it is helpful to 
review certain first principles of radar operation. 
Equation 1 is the so-called radar equation. This equation describes the 
returned or echo power P.sub.e as a function of a radar's transmitted 
power, wavelength, antenna gain, radar cross-section and target range. A 
target's radar cross-section is the equivalent flat reflective surface 
which is orthogonal to a radar beam which would return the same echo 
strength to the radar as does the target. 
##EQU1## 
where P.sub.e =returned or echo power 
P.sub.R =radar's transmitted power 
.lambda.=wavelength 
G.sub.R =radar antenna gain 
.sigma.=radar cross-section; and 
r=target range 
Equation 2 below expresses the countermeasure power received by the seeker 
as a function of the transmitted power, antenna gain and polarization. The 
power returned to the seeker of an echoed signal of Equation 1 is 
sensitive to the fourth power of the range, whereas the power received by 
the radar from the countermeasure is sensitive only to the second power of 
range. As will be seen, this provides an advantage for a countermeasure 
signal. 
##EQU2## 
where P.sub.c =countermeasure power received by radar 
P.sub.T =countermeasure's transmitted power 
G.sub.T =countermeasure's antenna gain 
p=polarization mismatch factor 
Equations 1 and 2 may be combined to express an advantage ratio AR which 
describes relative countermeasure power versus the echo power received by 
the seeker radar receiver. A large advantage can be realized if the 
countermeasure induces a signal in the receiver of the seeker radar which 
lies within the passband of the seeker radar, owing to the term r.sup.2 
/.sigma., which is quantified as: 
##EQU3## 
This advantage ratio holds if the countermeasure is able to maintain 
stability within the passband of the seeker receiver, an effect which is 
almost impossible to maintain for an autonomous transmitter given 
temperature effects and other effects on the oscillator in the 
countermeasure apparatus. 
One way to keep the transmitted countermeasure signal within the passband 
of the seeker receiver intermediate frequency is to injection lock the 
countermeasure signal to the transmit frequency of the seeker signal. A 
countermeasure signal will be injection locked to a seeker signal whenever 
the frequency offset between the two is less than the injection lock 
bandwidth BW.sub.L, which is given by: 
##EQU4## 
where f.sub.o =operating frequency 
P.sub.i =power injected by radar into countermeasure's cavity oscillator 
Q.sub.L =countermeasure cavity's loaded Q 
P.sub.T =countermeasure's transmitted power 
The injection power P.sub.i can be expressed in a form following Equation 2 
to account for mismatch, range, antenna gains and radar transmitted power 
as follows: 
##EQU5## 
This value for injected power can be substituted into Equation 4 to yield 
an injection lock bandwidth expression as follows: 
##EQU6## 
For specific values of loaded Q (Q.sub.L) equal to about 25, G.sub.T equal 
to 80, and P.sub.T equal to 10 mW, for a circularly polarized seeker 
signal of 20 mW into an antenna with a gain of 200, Equation 6 reduces to 
the following expression: 
##EQU7## 
where r is expressed in meters. 
Equation 7 defines the locked bandwidth in the specific case of a K-band 
system of the type herein explained. Equation 7 is illustrated by FIG. 1. 
Curve 100 holds for a constant K equal to 857, which in turn is based on 
the foregoing specific values selected for this calculation. If frequency 
modulation is used to cause random aperiodic energy redistribution among 
FM side pairs lying within the radar passband, the apparent echo to the 
seeker will be frequency incoherent and highly variable in amplitude, 
thereby precluding intelligibility of a radar's seeker receiver. For a 
strong field, a broad lock 110 is effected, as illustrated in FIG. 2A. In 
a weak field, only a narrow lock 112 is possible, as shown in FIG. 2B. 
To create the FM side pairs in the far field with attendant minimal locking 
bandwidths, the target transmitter must place a signal on or near the 
frequency of the seeker signal. At 760 meters distance, which is the 
approximate maximum effective range of a Doppler radar system, the 
passband or lock bandwidth 102 is 1.12 MHz, as illustrated in FIG. 1. The 
target receiver must therefore sweep between 24,050 MHz and 24,250 MHz in 
the K-band to locate the seeker frequency. 
According to the invention, when the frequency of the sweeping oscillator 
in the target approaches the frequency of the seeker radar signal, an 
oscillation locking effect is used at the target to lock onto a seeking 
radar signal in order to permit a spot countermeasure signal to be 
generated to defeat the seeking radar. The locking effect initially tends 
to force the frequency of a sweeping oscillator in a transceiver element 
of the target to the frequency of the seeker radar signal. The 
countermeasure signal may be a frequency modulated carrier signal which 
forces the sweeping oscillator out of lock randomly and aperiodically so 
as to generate a signal which appears as noise to a seeker receiver. 
In a specific implementation where the sweeping oscillator is subject to 
chirp effects which could break lock, a primary and secondary sweep are 
employed to assure that lock is maintained. As the frequency of the output 
of the mixed or heterodyned signal of the sweeping oscillator and the 
seeking radar passes (quickly) through the range of the passband of the 
intermediate frequency amplifier in the countermeasure apparatus, a burst 
of energy is passed through a filter to a sensor (comparator) which causes 
the sweeping oscillator to terminate sweep at a short fixed interval after 
the sensed burst. Initially sweep is terminated at a time after initial 
capture and at an intended frequency greater than the frequency of the 
characteristic chirp effect caused by rapid voltage changes on Gunn 
oscillators, so that if the oscillator is pulled out of lock by the chirp 
effect (as in the weak field case) it can be reliably returned to be 
within the locking bandwidth. This offset is always reliably greater than 
a 1-3 MHz chirp frequency uncertainty. Thereafter, using the known 
separation and known sign of the separation, a secondary sweep of 
substantially lower rate is used to pull the oscillator frequency back 
relatively slowly toward the seeker frequency to be within the locking 
bandwidth of oscillation. The chirp effect at the lower rate sweep is 
incapable of breaking lock. Where a primary sweep rate of about 5 ms is 
used, the secondary sweep rate is only about 1/20th as fast. The primary 
sawtooth sweep is fast enough to inhibit radar seeker acquisition of a 
target before a countermeasure signal is generated. The minimum allowable 
sweep rate assures capture and countermeasure generation at any frequency 
of interest within 100 milliseconds, which is the typical acquisition time 
for a seeking Doppler radar system. 
The secondary sweep produces only a minute amount of chirp, due to low 
power level and smaller sweep range (preferably between 6 MHz and 8 MHz). 
This secondary sweep allows the oscillator circuit to be precisely tuned 
within the radar passband as far away as the maximum range of the seeker 
radar system. A secondary sweep requires less than about 2 milliseconds. 
After the secondary sweep has been terminated thus tuning the oscillator 
circuit to a range within the radar's passband, the oscillator in the 
target device is held at a frequency within the passband. 
Chirp effect problems are thermally related and are typical of Gunn diode 
type oscillators. There may be other sweeping oscillator types which 
suffer from frequency drift problems and which may benefit from 
primary/secondary sweep techniques. However, the scope of this disclosure 
is intended to include sweeping oscillators capable of injection locking 
with a radio/radar/optical beam, as in a transceiver cavity. 
FIG. 3A and FIG. 3B depict a typical acquisition and hold cycle as 
explained above using primary and secondary sweep. Where chirp effect is 
not a concern, the secondary sweep is unneeded. FIG. 3A illustrates 
oscillator tuning voltage as a function of time. FIG. 3B illustrates a 
detail of oscillator frequency as a function of time. In operation, the 
primary sweep 150 repeats each 5 ms in the absence of a radar signal. A 
first radar intercept point or firing point 1 (F.P. 1) occurs at some 
point during a sweep. The sweeping oscillator locks to the seeking radar 
according to the invention, as soon as the frequency of the sweeping radar 
falls within the oscillation locking bandwidth (FIG. 3B F.P. 1). A burst 
of energy from the mixed signal extracted from the oscillator is used as a 
trigger for a countermeasure signal. If chirp effects are not a concern, 
the trigger starts a signal which frequency modulates the oscillator 
signal, causing it to be transmitted. The deviation of the frequency 
modulated signal is sufficient to cause the oscillator to break lock 
randomly and aperiodically during period 157. Otherwise, at a point 151 
after a fixed interval following intercept, and assuming that the chirp 
effect could cause a break in lock due to a relatively narrow locking 
bandwidth 160, a secondary sweep 152 is initiated to reverse the sweep 
direction. Such a break in lock is illustrated in FIG. 3B as segment 163 
following lock segment 161. The secondary sweep 152 brings the carrier 
frequency of the oscillator within the locking bandwidth 160 at a 
termination point 154. After the time point 151, or thereafter, at time 
point 156, modulation 157 is applied. The burst of energy emitted by the 
mixer is sensed in an intermediate frequency amplifier 28 through a low Q 
bandpass filter 30. The bandwidth of the bandpass filter is selected to be 
about 100 kHz so as to pass energy bursts caused by the injection locking 
transitions. This 100 kHz passband permits use of a tuned intermediate 
frequency amplifier which enhances noise rejection in the circuitry of the 
receiver. 
Referring to FIG. 4, there is shown a block diagram of a circuit in 
accordance with the invention, namely, a countermeasure apparatus 10 at a 
potential radar target. A horn antenna 12 is electrically and mechanically 
coupled to an oscillator and mixer stage 14 through a rectangular aperture 
21 (a standard TE10 waveguide). The structure of such an oscillator and 
mixer stage 14 is conventional art, and it is therefore unnecessary to 
describe in detail. The oscillator and mixer stage 14 is electrically 
controlled by a sawtooth voltage ramp generator 16 whose output is 
provided through a first sample and hold circuit 18, as hereinafter 
explained. A second sample and hold circuit 20 is also provided to 
maintain a carrier frequency of the oscillator and mixer 14 stage, as 
hereinafter explained, by providing an alternative input to the first 
sample and hold circuit 18. 
K-band and X-band circuit embodiments are substantially identical except 
that in the K-band embodiment, the sweep signal from the sweep generator 
16 is applied directly to a Gunn diode 23 within the oscillator and mixer 
stage 14 so as the voltage tune the oscillator. In the X-band, however, 
the oscillator diode preferably operates at a fixed voltage, and sweep is 
applied to a Varactor diode within the cavity in a manner known to the 
art. Sweep is provided via a Varactor diode since a Gunn diode does not 
have an adequate tuning range in the X-band. 
Direct Gunn-type modulation produces carrier amplitude modulation, since 
the output power is a function of the Gunn bias voltage. This is a 
secondary effect which is tolerable and inconsequential to the operation 
according to the invention. 
The antenna 12 is preferably a low-cost pyramidal horn affixed to the 
cavity of the oscillator and mixer stage 14 at a rectangular aperture 21 
(shown diagrammatically). The configuration results in a rectangularly 
polarized radio beam when emitted from the cavity through the antenna. 
Alternatively, a circularly polarized scheme may be used, subject to a 
trade-offs of cost since seeker signals are expected to be circularly 
polarized. For a rectangularly polarized signal, a polarization mismatch 
factor P is 3 dB, and it is easily tolerated due to the inherent large 
power advantage of the transmitted signal 22 over a seeker signal 24. 
The oscillator/mixer stage 14 is coupled to an intermediate frequency 
amplifier 28, the output of which is provided through a filter 30 having a 
passband of approximately 100 kHz to a comparator 32. The comparator 
threshold level is set by a reference voltage element 36. The output of 
the comparator 32 is provided as a clock signal to a first dual D-type 
flipflop 34. A first data input D1 is held high so that a first output 
Q1bar of the first flipflop 34 is also normally high. This signal is 
coupled to drive sweep generator 16 in a normal active high, increasing 
voltage ramp ("ramp up") through a first delay 40. Any negative-going 
change on this signal line is propagated through the delay 40 after 400 
.mu.s. (An increase in the voltage through the sweep generator 16 causes 
the frequency of a Gunn oscillator 23 to decrease but the frequency of a 
Varactor-controlled oscillator to increase). The second output Q1 
complementing Q1bar is coupled to a second delay 41. The second delay 41 
is coupled to the second data input D2 of the first dual D flipflop 34. 
Any positive-going change in Q1 is propagated to input D2, which is 
otherwise normally low, after 1 ms. Thus, the output of third output Q2 is 
not triggered until a second clock is observed from the comparator 32. The 
second clock is caused by the second intercept at point 154 (FIG. 3A or 
FIG. 3B). Through third output Q2 "ramp down" signal is provided to the 
sweep generator 16. In this embodiment, the "ramp down" signal is only 
effective in the absence of a "ramp up" signal. The second output Q1 also 
provides a start signal to a hold timer 38 (which is for timing the 
duration of the countermeasure signal), and third output Q2 provides an 
alert signal to an alert element 49, such as an audible alarm and/or 
visual indicator, to alert a vehicle operator to the presence of a seeking 
signal and prompt the operator to verify compliance with applicable 
regulations affecting the vehicle. 
The third output Q2 also provides a reset signal to a second dual D 
flipflop 42 and an enable signal to a modulator 56. The second flipflop 42 
is clocked by the flyback of sweep generator 16 which in turn enables the 
second sample and hold element 20. The second sample and hold element 20 
provides an alternative voltage input control to the oscillator/mixer 
stage 14. The output of the hold timer 38 is mixed with the flyback output 
of the sweep generator 16 at a mixer 44 to provide a reset signal to the 
first flipflop 34. 
Modulator/oscillator 56, which operates at 4 kHz, is enabled by the third 
output Q2 from the first flipflop 34. Modulator/oscillator is coupled to 
an oscillator 58 which frequency modulates modulator/oscillator 56 at 25 
Hz. The output is coupled to a postamplifier 46 for modulating the voltage 
at the output of the sample and hold circuit 18 during the period of the 
countermeasure signal. 
Operation is as follows: The sweep generator 16 induces a varying sweep 
voltage across the oscillator portion (Gunn diode or Varactor diode) of 
the oscillator and mixer stage 14, exciting a mixer diode (not shown) 
within the cavity which, upon encountering a seeker radar signal, develops 
a difference frequency signal. The difference frequency signal is passed 
through to intermediate frequency amplifier 28, which in turn drives 
filter 30. As the oscillator carrier is swept toward intercept with the 
seeker radar signal, the filter 30 will pass energy of the heterodyned 
(mixed) signal within the IF passband and thus emit a burst sensed by the 
comparator 32 as it exceeds the reference threshold of voltage reference 
36. The reference threshold determines system sensitivity and hence a 
maximum distance detection range in conjunction with the limiting level of 
the IF amplifier 28. 
The comparator 32 output clocks the first dual flipflop 34 causing first 
output Q1 bar to issue an enable signal which terminates the primary (ramp 
up) sweep after a preselected delay set by the delay element 40. This 
delay allows the frequency of the carrier to pass the intercept point and 
to terminate sweep in order to mask or swamp the expected chirp of the 
Gunn oscillator and to provide a signal with a known direction of carrier 
offset. The sweep generator 16 and the sample and hold circuit 18 then 
apply a secondary sweep to the oscillator and mixer stage 14 as shown as 
segment 152 in FIG. 3A and FIG. 3B. Segment 150 represents the primary 
sweep (voltage in FIG. 3A, frequency in FIG. 3B), an up-ramp with a fast 
fly-back. It is continued as segment 163 after the first intercept. The 
secondary sweep signal applied to the oscillator mixer stage 14 causes the 
sweep to go in the opposite direction from the primary sweep at a much 
lower sweep rate (e.g. 1/20th the primary sweep rate), thus tuning the 
oscillator mixer stage 14 back toward the intercept frequency at intercept 
point 154 (FIG. 3B). The filter 30 then passes the captured signal once 
again to the comparator 32 causing it to re-clock the flipflop 34 causing 
its second output Q2 to go high, which in turn causes the sweep generator 
16 to cease functioning and the sample and hold circuit 18 to maintain the 
current voltage at the value which caused the secondary intercept. The 
output Q1 also triggers the hold timer 38 to an active state to start a 
time-out for the mixer 44 to control the reset signal on the first 
flipflop. After the desired hold interval (of about 40 ms), transitions 
from the hold timer 38 reset the flipflop 34 through the mixer 44, which 
again initiates a new sweep operation to again acquire radar signals from 
a seeker radar. This hold period is short enough to ensure that sample and 
hold drift will not affect carrier accuracy and to ensure that there are 
periodic updates of the radar frequency within the sweep range. 
A start signal from the flipflop 42 also enables modulator 56 driven by 
oscillator source 58 so as to impress a carrier frequency modulation on 
the voltage at the output of the sample and hold circuit 18. The 25 Hz 
oscillator 58 is a very low frequency oscillator which frequency modulates 
the 4 kHz carrier modulator and thus prevents radar display of the speed 
corresponding to 4 kHz when the target is stationary. A stationary target 
develops no Doppler shifted signal to suppress a weak 4 kHz component 
which would otherwise be displayed in a seeker receiver. This modulation 
also forces random aperiodic breaking of lock of the injection locked 
signal, since deviation is selected to be generally substantially greater 
than the locking bandwidth for the sweeping oscillator. The locking 
bandwidth increases with field strength, as during approach to a seeker, 
but deviation is generally selected so that lock can always be broken. 
The system 10 is set up so that after lock has been established, the 
circuit assumes that multi-path cancellation has occurred if subsequent 
sweeps fail to detect a seeker radar signal. The circuit therefore 
triggers second dual flipflop 42 to cause its output to go high, which 
inserts a sample and hold signal from second sample and hold circuit 20 
into the input to the Gunn oscillator and mixer stage 14 in order to 
maintain oscillator tuning at the last known secondary input value. The 
condition will be held until a seeker radar is re-acquired, as would be 
evidenced by a toggling of the output Q2 of the flipflop 34 resetting the 
second flipflop 42 to take the sample and hold circuit 20 off line. 
If and when the target vehicle is in compliance with operating regulations, 
the vehicle operator may turn off the countermeasure unit 10 (at the power 
supply), thereby allowing the seeker to acquire the target and display a 
report. However, if the unit is not turned off, even a large 
crosss-section target will not generate a report so long as the 
lock-breaking countermeasure signal is effective, i.e, so long as the 
deviation of the countermeasure signal is greater than the locking 
bandwidth, which is typically over separation distances greater than 20 
feet. 
The invention has now been explained with reference to specific 
embodiments. Other embodiments will be apparent to those of ordinary skill 
in the art. It is therefore not intended that this invention be limited 
except as indicated by the appended claims.