SWITCHING POWER CONVERTERS CONFIGURED TO INJECT CURRENT INTO AN OUTPUT NODE

A switching power converter includes a plurality of power stages and a blocking capacitor. Each power stage includes a respective power transformer. The blocking capacitor and a respective secondary winding of each power transformer are electrically coupled in series between an output power node of the switching power converter and a reference node of the switching power converter. Another switching power converter includes a plurality of power stages, a boost winding, and a blocking capacitor. Each power stage includes a respective power transfer winding, and the boost winding forms at least one turn around a respective leakage magnetic flux path of each power transfer winding. The blocking capacitor and the boost winding are electrically coupled in series between an output power node of switching power converter and a reference node of the switching power converter.

BACKGROUND

Switching power converters are widely used in electronic devices, such as to provide a regulated electrical power source. A switching power converter is configured such that its solid-state power switching devices do not continuously operate in their active states; instead, the switching devices repeatedly switch between their on-states and off-states. Although switching power converters can achieve high efficiency, particularly under heavy load conditions, they typically exhibit ripple current due to switching action of their switching devices. Ripple current is generally undesirable because it causes power losses and ripple voltage. Additionally, switching power converters may not respond to a transient loading event as quickly as desired due to time required to change amount of energy stored in one or more energy storage devices, e.g., inductors and/or capacitors, of switching power converters. Speed at which a switching power converter is capable of responding to a change in load powered by the switching power converter is referred to as “transient response” of the switching power converter. As such, the faster the transient response of a switching power converter, the quicker the switching power converter can respond to a change in load.

Some switching power converters include one or more transformers. A transformer exhibits magnetizing inductance and leakage inductance. Magnetizing inductance is inductance associated with magnetic flux linking the primary and secondary windings, while leakage inductance is inductance associated with magnetic flux generated by current flowing through one of the primary and secondary windings that does not couple to any other winding of the transformer.

Additionally, some switching power converters include one or more discrete inductors, where a discrete inductor is an inductor that is not magnetically coupled to any other inductor. Furthermore, some switching power converters include one or more coupled inductors, where a coupled inductor is a device including two or more inductors that are magnetically coupled. A coupled inductor exhibits magnetizing inductance, which is inductance associated with magnetic flux linking the windings of the coupled inductor. Additionally, each winding of a coupled inductor exhibits leakage inductance, which is inductance associated with magnetic flux that flows only around the particular winding, i.e., magnetic flux that does not couple to any other winding. A coupled inductor is typically designed to have a finite leakage inductance, such as to achieve a desired ripple currently magnitude and/or transient response, while a transformer is typically designed to minimize leakage inductance.

DETAILED DESCRIPTION OF THE EMBODIMENTS

Disclosed herein are switching power converters configured to inject current into a power node of the switching power converter, such as an output power node of the switching power converter. Certain embodiments of the switching power converters are configured such that current injected into the power node at least partially cancels ripple current flowing through power transfer windings, or transformer primary windings, of the switching power converter, thereby promoting low total ripple current magnitude. Certain other embodiments are configured such that current injected into the power node adds to alternating current flowing through power transfer windings, or transformer primary windings, of the switching power converter, thereby promoting fast transient response of the switching power converter. Accordingly, the new switching power converters may achieve higher performance, e.g., lower ripple current magnitude or faster transient response, than conventional switching power converters.

FIG.1is a schematic diagram of a multi-phase switching power converter100, which is one embodiment of the new switching power converters disclosed herein that are configured to inject current into an output power node102. As discussed below, switching power converter100is optimized for minimal total ripple current magnitude at the output. Switching power converter100includes N power stages104, a controller106, a blocking capacitor108, and an optional tuning inductor110, where N is an integer greater than one. In this document, specific instances of an item may be referred to by use of a numeral in parentheses (e.g., power stage104(1)) while numerals without parentheses refer to any such item (e.g., power stages104). Each power stage104corresponds to a respective phase of switching power converter100, such that switching power converter100is an N-phase switching power converter. In particular, power stage104(1) corresponds to a first phase of switching power converter100, power stage104(2) corresponds to a second phase of switching power converter100, as so on.

Each power stage104includes a switching stage112and a power transformer114, where the switching stage112is electrically coupled to a primary winding P of the power transformer114at a switching node X. The primary winding P of each power transformer114is electrically coupled between the switching node X of its respective power stage104and output power node102. For example, primary winding P of power transformer114(1) is electrically coupled between switching node X(1) and output power node102, and primary winding P of power transformer114(2) is electrically coupled between switching node X(2) and output power node102. Accordingly, each primary winding P is electrically coupled between the switching stage112of its respective power stage104and output power node102. Output power node102has a voltage Vo, and an output current Ioflows to a load (not shown) electrically coupled to output power node102. Output current Iocould have a negative polarity without departing from the scope hereof. One or more capacitors116are optionally electrically coupled to output power node102.

Each switching stage112is configured to repeatedly switch the switching node X of its power stage104between an input power node118and a reference node120, in response to control signals U and L generated by controller106. Specifically, switching stage112(1) is configured to repeatedly switch switching node X(1) between input power node118and reference node120in response to control signals U(1) and L(1), switching stage112(2) is configured to repeatedly switch switching node X(2) between input power node118and reference node120in response to control signals U(2) and L(2), and so on. Input power node118is at a voltage Vin, and each switching stage112accordingly repeatedly switches switching node X of its power stage104between voltage Vinand zero volts relative to reference node120. An input current Iinflows from an electrical power source (not shown) to switching power converter100via input power node118. Input current Iincould have a negative polarity without departing from the scope hereof. A primary winding P of a given power transformer114(1) in switching power converter100is driven “high” when its respective switching node X is at voltage Vin, and the primary winding P is driven “low” when its respective switching node X is at zero volts relative to reference node120. For example, primary winding P of power transformer114(1) is driven high when switching node X(1) is at voltage Vin, and primary winding P of power transformer114(1) is driven low when switching node X(1) is at zero volts relative to reference node120. Reference node120is depicted as being a ground node, such as an earth ground node or a chassis ground node. It is understood, though, that reference node120need not be a ground node, and reference node120accordingly could be at a different electrical potential than an earth ground or a chassis ground.

Secondary windings S of power transformers114, blocking capacitor108, and tuning inductor110are electrically coupled in series between output power node102and reference node120. The respective topological location of each of blocking capacitor108and tuning inductor110could vary as long as it is electrically coupled in series with secondary windings S. For example, blocking capacitor108could be electrically coupled between respective secondary windings S of power transformers114(1) and114(2), instead of being electrically coupled between tuning inductor110and output power node102. Additionally, one or more additional elements (not shown) could be electrically coupled in series with blocking capacitor108, tuning inductor110, and secondary windings S of power transformers114without departing from the scope hereof. While tuning inductor110is depicted as being a discrete inductor, tuning inductor110could instead be embodied by intrinsic inductance of a circuit including secondary windings S, particularly in applications where tuning inductor110need only have a small inductance value. Furthermore, tuning inductor110could be omitted, or tuning inductor110could be replaced with a plurality of tuning inductors, without departing from the scope hereof. Similarly, blocking capacitor108could be replaced with a plurality of blocking capacitors electrically coupled in series and/or in parallel without departing from the scope hereof.

Each primary winding P forms a quantity of turns equal to N1, and each secondary winding S forms a quantity of turns equal to N2. In certain embodiments, a ratio of N1/N2 is equal to N to help minimize magnitude of ripple in output current Io. As discussed above, N is equal to the number of phases of switching power converter100.

FIG.2illustrates one possible implementation of switching stages112of switching power converter100. Specifically,FIG.2is a schematic diagram of N switching stages202, where switching stages202are an embodiment of switching stages112ofFIG.1. Each switching stage202includes an upper switching device206and a lower switching device208. Each upper switching device206is electrically coupled between input power node118and the switching node X of its respective power stage104. Each lower switching device208is electrically coupled between the switching node X of its respective power stage and reference node120. For example, upper switching device206(1) is electrically coupled between input power node118and switching node X(1), and lower switching device208(1) is electrically coupled between switching node X(1) and reference node120. Each upper switching device206switches in response to a respective control signal U from controller106, and each lower switching device208switches in response to a respective control signal L from controller106. For example, in some embodiments, each upper switching device206operates in its on (conductive) state when its respective control signal U is asserted, and the switching device operates in its off (non-conductive state) when its respective control signal U is de-asserted. Similarly, in some embodiments, each lower switching device208operates in its on (conductive) state when its respective control signal L is asserted, and the switching device operates in its off (non-conductive state) when its respective control signal L is de-asserted. Each switching device206and208includes, for example, one or more transistors.

Referring again toFIG.1, controller106is implemented, for example, by analog and/or electronic circuitry. In some embodiments, controller106is at least partially implemented by a processor (not shown) executing instructions in the form of software and/or firmware stored in a memory (not shown). Although controller106is depicted as being a discrete element for illustrative simplicity, controller106could be partially or fully integrated with one or more other elements of switching power converter100. For example, some subsystems of controller106could be incorporated in one or more of switching stages112. Additionally,FIG.1should not be construed to require that there be a separate control bus for each control signal. For example, controller106could be implemented by a combination of a central integrated circuit and local control logic integrated in each switching stage112, with a single control bus running from the central integrated circuit to each switching stage112. Furthermore, controller106may include multiple constituent elements that need not be co-packaged or even disposed at a common location.

Controller106is configured to generate control signals U and L to control duty cycle (D) of power stages104, where duty cycle is a portion of a switching cycle that a primary winding P of a power transformer114is driven high, to regulate at least one parameter of switching power converter100. Controller106is configured to vary duty cycle of power stages104, for example, using a pulse width modulation (PWM) technique and/or a pulse frequency modulation (PFM) technique. Examples of possible regulated parameters include, but are not limited, magnitude of input voltage Vin, magnitude of input current Iin, magnitude of output voltage Vo, and magnitude of output current Io. For example, in some embodiments, controller106is configured to generate control signals U and L to regulate magnitude of output voltage Vo, and controller106accordingly generates control signals U and L during continuous conduction operation of switching power converter100such that duty cycle of power stages104is equal to a ratio of output voltage magnitude Voover input voltage magnitude Vin. For example, if output voltage Vois to be regulated to two volts and input voltage Vinis eight volts, controller106would generate control signals U and L such that duty cycle of power stages104is 0.25, which would result in each primary winding P being driven high for 25 percent of each switching cycle of switching power converter100. Controller106is optionally configured to generate control signals U and L such that power stages104switch out-of-phase with each other. For example, in some embodiments, controller106is configured to generate control signals U and L such that each power stage104switches360/N degrees out of phase with an adjacent power stage104in the phase domain.

Blocking capacitor108prevents direct current (DC) shorting of output power node102to reference node120, while allowing alternating current (AC) to flow between secondary windings S and output power node102. Accordingly, in particular embodiments, blocking capacitor108has a sufficiently large capacitance value such that a corner frequency established by blocking capacitor108and tuning inductor110is significantly lower than a switching frequency of switching power converter100, to enable injection of current flowing through secondary windings S into output power node102. Additionally, in certain embodiments, tuning inductor110has an inductance of Lc, where Lc=Lco/(N1/N2)2. Lcois an inductance of tuning inductor110that would be selected if a ratio of N1/N2 were one instead of N, such as to achieve a desired performance of switching power converter100if the ratio of N1/N2 were one instead of N.

Importantly, switching power converter100is configured such that current flowing through secondary windings S of power transformers114at least partially cancels ripple current flowing through primary windings P of power transformers114, at output power node102during steady state operation of switching power converter100. In particular, a respective direct current IDCand a respective alternating current IACflow through each primary winding P, as illustrated inFIG.1. Under steady state operating conditions of switching power converter100, each alternating current IACis a ripple current generated by switching of switching stages112. The sum of direct current IDCand alternating current IACflowing through a given primary winding P may be referred to as current122flowing through the primary winding P, as illustrated inFIG.1. Additionally, a common injection current Iinj, which is an alternating current, flows through secondary windings S, tuning inductor110, and blocking capacitor108, as illustrated inFIG.1. Injection current Iinjis also a ripple current under steady state operating conditions of switching power converter100. Injection current Iinjis injected into output power node102. Secondary windings S are electrically coupled to output power node102such that injection current Iinjis at least substantially out-of-phase with respect to alternating currents IACflowing through primary windings P. WhileFIG.1illustrates one possible polarity of power transformers114, polarity of power transformers114could be different as long as secondary windings S are electrically coupled to output power node102in a manner such that injection current Iinjis at least substantially out-of-phase with respect to alternating currents IACflowing through primary windings P. Additionally, as discussed above, in particular embodiments, a ratio of N1 to N2 is equal to N, such as that magnitude of injection current Iinjis substantially equal to a magnitude of a sum of all alternating currents IACflowing through primary windings P. Accordingly, injection current Iinjmay significantly cancel alternating currents IACat output power node102, thereby advantageously helping minimize magnitude of ripple in output current Io.

For example,FIGS.3A and3Bare graphs302and304, respectively, of current versus time of simulated operation of one embodiment of switching power converter100where N=4, switching frequency is one Megahertz (MHz), magnetizing inductance (Lm) of each power transformer114is 150 nanohenrys (nH), N1/N2=4, Vin=12 volts, Vo=1 volt, and inductance (Lc) of tuning inductor130is 7.5 nH. Graphs302and304have a common time base. Graph302includes curves representing each of currents122(1),122(2),122(3) and122(4) flowing through primary winding P of power transformers114(1),114(2),114(3), and114(4), respectively. Graph304includes curves representing (a) total current124flowing through all primary windings P (seeFIG.1) under a zero load condition for switching power converter100, (b) injection current Iinjinjected into output power node102, and (c) output current Iounder a zero load condition for switching power converter100. If load is applied to the output power node102of switching power converter100, then total current124and output current Iowill have a non-zero DC current added to the waveforms inFIG.3A, while phase currents122inFIG.3Awill also have a corresponding DC level added to the waveforms. Total current124has a peak-to-peak ripple current magnitude of almost 20 amperes, as illustrated inFIG.3B. If injection current Iinjwere not injected into output power node102, output current Iowould be the same as total current124, and output current Iowould therefore also have a peak-to-peak ripple current magnitude of almost 20 amperes. However, injection current Iinjis out-of-phase with respect to total current124, and peak-to-peak magnitude of injection current Iinjis almost as large as the peak-to-peak magnitude of total current124. Therefore, injection current Iinjsubstantially cancels ripple current flowing through primary windings P, at output power node102. Consequently, output current Iouthas a relatively small ripple current magnitude of only around 2.9 amperes. As such, injecting injection current Iinjinto output power node102substantially reduces magnitude of ripple in output current Io.

As another example,FIGS.4A and4Bare graphs402and404, respectively, of current versus time of simulated operation of one embodiment of switching power converter100where N=4, switching frequency is one MHz, Lmof each power transformer114is 250 nH (instead of 150 nH as in the simulation ofFIGS.3A and3B), N1/N2=4, Vin=12 volts, Vo=1 volt, and Lcof tuning inductor130is 7.5 nH. Graphs402and404have a common time base. Graph402includes curves representing each of currents122(1),122(2),122(3) and122(4) flowing through primary winding P of power transformers114(1),114(2),114(3), and114(4), respectively. Graph404includes curves representing (a) total current124flowing through all primary windings P, (b) injection current Iinjinjected into output power node102, and (c) output current Io. Total current124has a peak-to-peak ripple current magnitude of about 18 amperes, as illustrated inFIG.4B. However, injection current Iinjsubstantially cancels ripple current flowing through primary windings P, at output power node102. Consequently, output current Iouthas a relatively small ripple current magnitude of only around 1.65 amperes. As evident when comparing the simulations ofFIGS.3A and3Bto the simulations ofFIGS.4A and4B, increasing magnetizing inductance of power transformers114may reduce magnitude of ripple in output current Io.

While injection of injection current Iinjinto output power node102reduces magnitude of ripple in output current Io, such injection also degrades transient response of switching power converter100. For example,FIG.5is a graph500of current versus time illustrating simulated operation of an embodiment of switching power converter100in response to a change in load. Prior to time t1, a load powered by switching power converter100is zero, and a DC component of output current Iois therefore zero. At time t1, though, the load powered by switching power converter100rapidly increases, and total current124therefore ramps up. However, injection current Iinjinjected into output power node102ramps up in the opposite direction of total current124, and the injected current therefore slows the rise of output current Ioin response to the load increase, thereby degrading the transient response of switching power converter100.

Switching power converter100can be modified so that injection current Iinjadds to alternating currents IACin output power node102, instead of subtracting from (canceling) alternating currents IAC, thereby promoting fast transient response of the switching power converter at the expensive of larger magnitude of ripple of output current Io. For example,FIG.6is a schematic diagram of switching power converter600, which is an alternate embodiment of switching power converter100that is optimized for fast transient response instead of for low ripple current magnitude. Switching power converter600differs from switching power converter100only in the manner that secondary windings S are electrically coupled to output power node102. In particular, in switching power converter100(FIG.1), secondary windings S are electrically coupled in series between output power node102and reference node120such that injection current Iinjis out of phase with respect to alternating currents IACflowing through primary windings P. In switching power converter600(FIG.6), in contrast, secondary windings S are electrically coupled in series between output power node102and reference node120such that injection current Iinjadds to alternating currents IAC flowing through primary windings P, thereby promoting fast transient response of switching power converter600with the potential drawback of larger magnitude of ripple in output current Io. It should be noted, though, that magnitude of ripple current decreases with increasing number of phases of switching power converter600. Therefore, in embodiments of switching power converter600where Nis large, increase in magnitude of ripple in output current Iocurrent caused by injecting injection current Iinjinto output power node102may be immaterial because ripple current magnitude is very small due to N being large. WhileFIG.6illustrates one possible polarity of power transformers114, polarity of power transformers114could be different as long as secondary windings S are electrically coupled to output power node102in a manner such that injection current Iinjadds to alternating currents IACflowing through primary windings P.

FIG.7is a graph700of current versus time illustrating simulated operation of an embodiment of switching power converter600in response to a change in load. Prior to time t1, a load powered by switching power converter600is zero, and magnitude of output current Iois therefore zero (neglecting ripple current). At time t1, though, the load powered by switching power converter600rapidly increases, and total current124therefore ramps up. Additionally, injection current Iinjinjected into output power node102ramps up in the same direction as total current124, and the injected current adds to total current124and increases the rate of rise of output current Ioin response to the load increase, thereby improving the transient response of switching power converter100. However, the fact that in steady state injection current Iinjadds to alternating currents IACflowing through primary windings P increases magnitude of ripple in output current Io.

FIG.8is a schematic diagram of a multi-phase switching power converter800, which is another embodiment of the new switching power converters disclosed herein configured to inject current into an output power node. As discussed below, switching power converter800is optimized for minimal ripple current magnitude. Switching power converter800includes N power stages804, a controller806, a blocking capacitor808, an optional tuning inductor810, and a boost winding811, where N is an integer greater than one. Each power stage804corresponds to a respective phase of switching power converter800, such that switching power converter800is an N-phase switching power converter. In particular, power stage804(1) corresponds to a first phase of switching power converter800, power stage804(2) corresponds to a second phase of switching power converter800, and so on.

Each power stage804includes a switching stage812electrically coupled to a power transfer winding814at a switching node X. Each power transfer winding814is electrically coupled between the switching node X of its respective power stage804and output power node802. For example, power transfer winding814(1) is electrically coupled between switching node X(1) and output power node802, and power transfer winding814(2) is electrically coupled between switching node X(2) and output power node802. Output power node802has a voltage Vo, and an output current Ioflows to a load (not shown) electrically coupled to output power node802. Output current Iocould have a negative polarity without departing from the scope hereof. One or more capacitors816are optionally electrically coupled to output power node802.

Each switching stage812is configured to repeatedly switch the switching node X of its power stage804between an input power node818and a reference node820, in response to control signals U and L generated by controller806. Specifically, switching stage812(1) is configured to repeatedly switch node X(1) between input power node818and reference node820in response to control signals U(1) and L(1), switching stage812(2) is configured to repeatedly switch node X(2) between input power node818and reference node820in response to control signals U(2) and L(2), and so on. Input power node818is at a voltage Vin, and each switching stage812accordingly repeatedly switches node X of its power stage804between voltage Vinand zero volts relative to reference node820. Reference node820is depicted as being a ground node, such as an earth ground node or a chassis ground node. It is understood, though, that reference node820need not be a ground node, and reference node820accordingly could be at a different electrical potential than an earth ground or a chassis ground.

An input current Iinflows from an electrical power source (not shown) to switching power converter800via input power node818. Input current Iincould have a negative polarity without departing from the scope hereof. A given power transfer winding814in switching power converter800is driven “high” when its respective switching node X is at voltage Vin, and the power transfer winding814is driven “low” when its respective switching node X is at zero volts relative to reference node820. For example, power transfer winding814(1) is driven high when switching node X(1) is at voltage Vin, and power transfer winding814(1) is driven low when switching node X(1) is at zero volts relative to reference node820. In certain embodiments, switching stages812are similar to switching stages202ofFIG.2.

Boost winding811, blocking capacitor808, and tuning inductor810are electrically coupled in series between output power node802and reference node820. The respective topological location of each of blocking capacitor808and tuning inductor810could vary as long as it is electrically coupled in series with boost winding811. For example, blocking capacitor808could be electrically coupled between boost winding811and reference node820, instead of being electrically coupled between tuning inductor810and output power node802. Additionally, one or more additional elements (not shown) could be electrically coupled in series with blocking capacitor808, tuning inductor810, and boost winding811without departing from the scope hereof. While tuning inductor810is depicted as being a discrete inductor, tuning inductor810could instead be embodied by intrinsic inductance of a circuit including boost winding811, particularly in applications where tuning inductor810need only have a small inductance value. Furthermore, tuning inductor810could be omitted, or tuning inductor810could be replaced with a plurality of tuning inductors, without departing from the scope hereof. Similarly, blocking capacitor808could be replaced with a plurality of blocking capacitors electrically coupled in series and/or in parallel without departing from the scope hereof.

Each power transfer winding814forms a quantity of turns equal to N1, and boost winding811forms a quantity of turns equal to N2. In certain embodiments, a ratio of N1/N2 is equal to N to help minimize magnitude of ripple in output current Io. As discussed above, N is equal to the number of phases of switching power converter800.

Power transfer windings814are magnetically coupled by a magnetic core813. Boost winding811is magnetically coupled to a leakage component (not illustrated inFIG.8) of each power transfer winding814by magnetic core813. In particular, boost winding811is configured such that it forms at least one turn around a respective total magnetic flux path of each power transfer winding814. Importantly, boost winding811is additionally configured such that net mutual magnetic flux within the turns of the boost winding is essentially zero during steady state operation of switching power converter800. Equivalently, boost winding811can be configured such that it forms at least one turn around only a respective leakage magnetic flux path of each power transfer winding814. Such configuration of boost winding811advantageously helps minimize risk of a magnetic saturation from mutual magnetic flux within a magnetic flux path of boost winding811while enabling injection of injection current into output power node802. In this document, a winding forming a turn around an element need not completely surround the element. For example, a winding forming a turn around a leg of a magnetic core need not completely surround the leg. As another example, a winding forming a turn around a magnetic flux path need not completely surround the magnetic flux path.

Power transfer windings814, boost winding811, and magnetic core813are part of a boosted coupled inductor815. Magnetic core813is formed, for example, of a ferrite magnetic material, a composite magnetic material, or an iron powder magnetic material. However, magnetic core813could alternately be an “air core,” or in other words, magnetic core813could be implemented by placing windings814and811, or breaking these windings in sections and placing these sections in pairs in sufficient proximity, to achieve magnetic coupling without use of a tangible magnetic coupling structure. Boost winding811is drawn with a heavier line weight than power transfer windings814to help a viewer distinguish boost winding811from power transfer windings814. This difference in line weight should not be construed to imply that boost winding811is necessarily formed of a thicker conductor material than power transfer windings814. Boost winding811could actually be implemented with a smaller amount of conductor material because it carries only alternating current ripple and does not carry any load current.

Discussed below with respect toFIGS.9-29are several example embodiments of boosted coupled inductor815. It is understood, however, that boosted coupled inductor815is not limited to these example embodiments.

FIG.9is a perspective view of a boosted coupled inductor900, which is one possible embodiment of boosted coupled inductor815where N is equal to 4.FIG.10is an elevational view of a side902of boosted coupled inductor900,FIG.11is a top plan view of boosted coupled inductor900,FIG.12is a cross-sectional view of boosted coupled inductor900taken along line12A-12A ofFIG.11, andFIG.13is a cross-sectional view of boosted coupled inductor900taken along line13A-13A ofFIG.11.FIG.14is a top plan view of boosted coupled inductor900with windings omitted, to further show a magnetic core of the boosted coupled inductor.

Boosted coupled inductor900includes a magnetic core904(seeFIG.14), a plurality of power transfer windings906, and a boost winding908. Power transfer windings906are embodiments of power transfer windings814, and boost winding908is an embodiment of boost winding811. Magnetic core904is an embodiment of magnetic core813. Magnetic core904is formed, for example, of a ferrite magnetic material or a powdered iron magnetic material. Magnetic core904includes a first rail910, a second rail912, and a plurality of legs914. Although magnetic core904is illustrated as including four legs914, the number of legs914of magnetic core904will vary with the number of phases supported by boosted coupled inductor900. For example, in embodiments of boosted coupled inductor900intended for use with three phases, i.e., with N=3, boosted coupled inductor900will have three legs914instead of four legs914. First rail910and second rail912are separated from each other in a direction916, and legs914are disposed between first rail910and second rail912in direction916. Legs914are separated from each other in a direction918, where direction918is orthogonal to direction916.FIGS.9,10,12, and13further show a third direction920which is orthogonal to each of directions916and918. In some embodiments, legs914join first and second rails910and912in direction916, and in some other embodiments, one or more of legs914is separated from first rail910and/or second rail912by a respective gap (not shown), such as to help prevent saturation of magnetic core904. Each leg914optionally also forms a respective gap (not shown) along direction918, such that the leg is broken into two or more portions separated from each other in direction916by the gap.

A respective power transfer winding906is wound at least partially around each leg914, and boost winding908is wound at least partially around all legs914, such that boost winding908forms a common turn around all legs914. Accordingly, boost winding908is strongly magnetically coupled to each power transfer winding906. Boost winding908is electrically isolated from power transfer windings906. Although each power transfer winding906and boost winding908is depicted as being a single-turn winding formed of electrically conductive foil, such as copper foil, the configurations of power transfer windings906and boost winding908may vary. For example, one or more of these windings may form a plurality of turns, and/or one or more of these windings may be formed of wire instead of electrically conductive foil. As another example, a ratio of a quantity N1 of turns formed by each power transfer winding906to a quantity N2 of turns formed by boost winding908may be a function of number of phases in a switching power converter including boosted coupled inductor900, e.g., N1/N2=N.

FIG.15is a top plan view that is similar toFIG.11and is marked-up to symbolically show several mutual magnetic flux paths in boosted coupled inductor900. Lines1502,1504, and1506represent mutual magnetic flux flowing from power transfer winding906(1) to power transfer windings906(2),906(3), and906(4), respectively. While not shown inFIG.15, there are additional mutual magnetic flux paths between other power transfer winding906instances that pass through the turn formed by boost winding908. As can be appreciated fromFIG.15, although mutual magnetic flux from each power transfer winding906flows through the turn of boost winding908, net mutual magnetic flux flowing through boost winding908may be zero in some applications, e.g., mutual magnetic flux from power transfer winding906(1) may cancel itself because all return paths of such flux are also included under the boost winding908, leading to the net zero flux. The same is true for the mutual magnetic flux from any other power transfer windings906in boost winding908, such that boost winding908“sees” zero mutual magnetic flux.

Additionally, leakage magnetic flux generally flows through the turn of boost winding908, such that boost winding908is strongly magnetically coupled to leakage magnetic flux associated with power transfer windings906. For example,FIG.16is a perspective view that is similar toFIG.9and is marked-up to symbolically show a few example leakage magnetic flux paths in boosted coupled inductor900. In particular,FIG.16illustrates three example leakage magnetic flux paths1602,1604, and1606associated with power transfer winding906(1). It should be noted that all shown leakage magnetic flux paths1602,1604, and1606pass through the turn formed by boost winding908. The thick dashed lines ofFIG.16represent leakage magnetic flux flowing internal to magnetic core904, while the thin dashed lines ofFIG.16represent leakage magnetic flux flowing external to magnetic core904. Additionally, while not shown inFIG.16, there are additional leakage magnetic flux paths for power transfer winding906(1), as well as leakage magnetic flux paths for power transfer windings906(2)-906(4), which pass through the turn formed by boost winding908. Accordingly, boost winding908forms a turn around respective leakage magnetic flux paths for each power transfer winding906. The fact that boost winding908is within mutual magnetic flux paths helps maximize leakage magnetic flux coupling to boost winding908, by reducing potential for leakage magnetic flux to escape from magnetic core904before coupling to boost winding908. Nevertheless, some leakage magnetic flux associated with power transfer windings906(1)-906(4) does not pass through the turn formed by boost winding908.

Boost winding908forms a common turn around all legs914, such as illustrated inFIG.13. However, boost winding908could be modified to form a respective turn around each leg914, with all of the turns electrically coupled in series. For example,FIG.17is a top plan view of a boosted coupled inductor1700, andFIG.18is a cross-sectional view of boosted coupled inductor1700taken along line18A-18A ofFIG.17. Boosted coupled inductor1700is an alternate embodiment of boosted coupled inductor900where boost winding908is replaced with a boost winding1708forming a respective turn1709around each leg914. Turns1709are electrically coupled in series. Accordingly, boost winding1708has similar electrical properties to boost winding908. For example, boost winding1708is strongly magnetically coupled to leakage elements of power transfer windings906, and boosted coupled inductor1700is capable of being operated such that net mutual magnetic flux flowing through boost winding1708is essentially zero.

Boosted coupled inductors900and1700are configured to minimize leakage inductance of power transfer windings906, as large leakage inductance is not required to achieve small leakage current magnitude in switching power converter800. Additionally, small leakage inductance values promote good transient response of switching power converter800. Nevertheless, should larger leakage inductance be desired or required, boosted coupled inductors900and1700could be modified to include features for increasing leakage inductance, such as one or more magnetic elements configured to provide a leakage magnetic flux path between first rail910and second rail912.

For example,FIG.19is a perspective view of a boosted coupled inductor1900, which is an alternate embodiment of boosted coupled inductor900further including leakage elements1905and1907. Leakage element1905is joined to rail910, and leakage element1907is joined to rail912. Leakage elements1905and1907are formed of a magnetic material, such as a ferrite magnetic material or a powdered iron magnetic material. Leakage elements1905and1907extend towards each other in direction916to provide a relatively low reluctance path for leakage magnetic flux to flow between rails910and912. Leakage elements1905and1907are optionally separated by a gap1909in direction916where gap1909is filled with, for example, air, paper, plastic, adhesive, and/or a magnetic material having a lower magnetic permeability that magnetic material forming leakage elements1905and1907. Gap909can be split into two or more smaller gaps to decrease the fringing flux. The dotted lines delineating leakage elements1905and1907from rails910and912, respectively, are to assist a viewer in distinguishing features of boosted coupled inductor1900, and these lines do not necessarily represent discontinuities in boosted coupled inductor1900.

Boosted coupled inductor1900could be modified so that its boost winding is wound around one or more leakage elements1905and1907, instead of being wound around legs914. For example,FIG.20is a perspective view of a boosted coupled inductor1900, andFIG.21is a cross-sectional view of boosted coupled inductor1900taken along a line21A-21A ofFIG.20. Boosted coupled inductor2000is an alternate embodiment of boosted coupled inductor1900where boost winding908is replaced with a boost winding2008, where boost winding2008is another embodiment of boost winding811. Boost winding2008forms a turn around leakage element1905, which results in boost winding2008forming a turn around leakage magnetic flux paths of each power transfer winding906. Additionally, boost winding2008is outside of mutual magnetic flux paths of power transfer windings906. Consequently, mutual magnetic flux does not flow through the turn of boost winding2008, and net mutual magnetic flux flowing though boost winding2008is accordingly zero.

Boosted coupled inductor2000could be modified to have a different leakage element configuration. For example,FIG.22is a top plan view of a boosted coupled inductor2200, andFIG.23is a cross-sectional view of boosted coupled inductor2200taken along line23A-23A ofFIG.22, where boosted coupled inductor2200is an alternate embodiment of boosted coupled inductor2000with a different leakage element configuration.FIG.24is a top plan view of boosted coupled inductor2200with windings omitted, to further show a magnetic core of the boosted coupled inductor.

Boosted coupled inductor2200includes a magnetic core2204(seeFIG.24), a plurality of power transfer windings2206, and a boost winding2208. Power transfer windings2206are embodiments of power transfer windings814, and boost winding2208is an embodiment of boost winding811. Magnetic core2204is formed, for example, of a ferrite magnetic material or a powdered iron magnetic material. Magnetic core2204includes a first rail2210, a second rail2212, and a plurality of legs2214. Although magnetic core2204is illustrated as including four legs2214, the number of legs2214of magnetic core2204will vary with the number of phases supported by boosted coupled inductor2200. First rail2210and second rail2212are separated from each other in direction916, and legs2214are disposed between first rail2210and second rail2212in direction916. Legs2214are separated from each other in direction918. In some embodiments, legs2214join first and second rails2210and2212in direction916, and in some other embodiments, legs2214are separated from first rail2210and/or second rail2212by a respective gap (not shown), such as to help prevent saturation of magnetic core2204. Each leg2214optionally also forms a respective gap (not shown) along direction918, such that the leg is broken into two portions separated from each other in direction916by the gap.

Magnetic core2204further includes a leakage element2205disposed between first rail2210and second rail2212in direction916. In some embodiments, leakage element2205is separated from first rail2210and/or second rail2212by a respective gap (not shown). Leakage element2205optionally also forms a gap (not shown) along direction916, such that the leakage element is broken into two or more portions separated from each other in direction916by the gap. WhileFIGS.22-24depict leakage element2205being disposed between legs2214(2) and2214(3) in direction918so that the leakage element is centrally located with respect to legs2214, location of leakage element2205could vary as long as it is disposed between first rail2210and second rail2212in direction916. For example, leakage element2205could alternately be disposed between legs2214(1) and2214(2) in direction918. While not required, it is anticipated that leakage element2205will typically have a larger cross-sectional area (in directions918and920) than each leg2214, such as illustrated inFIG.23, because leakage element2205sees leakage magnetic flux from each leg2214. Additionally, while not required, it is anticipated that a gap in leakage element2205, when present, will be larger than gaps in the legs2214, when present, so that the leakage inductance value will be lower than the magnetizing inductance values of power transfer windings2206, which also promotes a higher saturation current for the leakage.

A respective power transfer winding2206is wound at least partially around each leg2214. Additionally, boost winding2208is wound at least partially around leakage element2205, which results in boost winding2208forming a turn around leakage magnetic flux paths of each power transfer winding2206. Additionally, boost winding2208is outside of mutual magnetic flux paths of power transfer windings2206. Consequently, mutual magnetic flux does not flow through the turn of boost winding2208, and net mutual magnetic flux flowing though boost winding2208is accordingly zero. Boost winding2208is electrically isolated from power transfer windings2206. Although each power transfer winding2206and boost winding2208is depicted as being a single-turn winding formed of electrically conductive foil, such as copper foil, the configurations of power transfer windings2206and boost winding2208may vary. For example, one or more of these windings may form a plurality of turns, and/or one or more of these windings may be formed of wire instead of electrically conductive foil.

Leakage element2205and boost winding2208could be replaced with two or more leakage elements and boost windings, respectively. For example,FIG.25is a top plan view of a boosted coupled inductor2500, andFIG.26is a cross-sectional view of boosted coupled inductor2500taken along line26A-26A ofFIG.25. Boosted coupled inductor2500is an alternate embodiment of boosted coupled inductor2200(a) where leakage element2205is replaced with two leakage elements2505and2507and (b) boost winding2208is replaced with two boost windings2508and2509, where boost windings2508and2509are collectively an embodiment of boost winding811.FIG.27is a top plan view of boosted coupled inductor2500with windings omitted, to further show the magnetic core of the boosted coupled inductor.

Each leakage element2505and2507is disposed between first rail2210and second rail2212in direction916, and leakage elements2205and2207are separated from each other in direction918. WhileFIGS.25-27depict leakage elements2505and2507being disposed at opposing ends of boosted coupled inductor2500, the location of leakage elements2505and2507may vary, as long as each leakage element is disposed between first rail2210and second rail2212in direction916. Leakage elements2505and2507optionally also form a gap (not shown), such that the leakage element is broken into two or more portions separated from each other in direction916by the gap. Boost winding2508is wound around leakage element2505, and boost winding2509is wound around leakage element2507. As such, each boost winding2508and2509forms a respective turn around leakage magnetic flux paths of each power transfer winding2206. Additionally, each boost winding2508and2509is outside of mutual magnetic flux paths of power transfer windings2206. Consequently, mutual magnetic flux does not flow through the respective turns of boost windings2508and2509, and net mutual magnetic flux flowing through each boost winding2508and2509is accordingly zero.

Boost windings2508and2509will typically be electrically coupled in series, as symbolically shown by a dashed line2699inFIG.26. For example, boosted coupled inductor2500may include an electrical conductor (not shown) electrically coupling boost windings2508and2509in series. As another example, boost windings2508and2509may be electrically coupled in series external to boosted coupled inductor2500, such as by a printed circuit board (PCB) (not shown) supporting boosted coupled inductor2500.

Boosted coupled inductors900,1700,1900,2000,2200, and2500are scalable in that they can be configured to support any number of phases by adjusting the number of legs and power transfer windings. However, boosted coupled inductor815ofFIG.8could also be a non-scalable boosted coupled inductor in embodiments of switching power converter800where N is equal to two. For example,FIG.28is a perspective view of a boosted coupled inductor2800, which is another possible embodiment boosted coupled inductor815where N is equal to two. Boosted coupled inductor2800includes a magnetic core2802, a first power transfer winding2804, a second power transfer winding2806, and a boost winding2808. Power transfer windings2804and2806are each an embodiment of a power transfer winding814, and boost winding2808is an embodiment of boost winding811.

Magnetic core2802is formed, for example, of a ferrite magnetic material or a powdered iron magnetic material. Magnetic core2802includes a first element2810and a second element2812stacked in a direction2814.FIG.29is a perspective view of boosted coupled inductor2800with second element2812removed to show an interior of boosted coupled inductor2800. Magnetic core2802forms a passageway2816extending through magnetic core2802in a direction2818, where direction2818is orthogonal to direction2814. Passageway2816has a width2820in a direction2822, where direction2822is orthogonal to each of directions2814and2818. Magnetic core2802could be formed of a single element, or magnetic core2802could be formed of three of more elements, without departing from the scope hereof.

Each of first power transfer winding2804, second power transfer winding2806, and boost winding2808are wound through passageway2816. Second power transfer winding2806is separated from first power transfer winding2804in direction2822, and boost winding2808is disposed between first power transfer winding2804and second power transfer winding2806in direction2822. In some embodiments, each of first power transfer winding2804, second power transfer winding2806, and boost winding2808is a staple style winding. Passageway2816has a height2824in direction2814. In some embodiments, height2824varies along width2820. For example, in certain embodiments, height2824at the boost winding2808is less than height2824at each of the first and second power transfer windings2804and2806, to achieve requisite leakage inductance values. The configuration of boosted coupled inductor2800enables boost winding2808be strongly magnetically coupled to leakage elements of power transfer windings2804and2806. In particular, boost winding2808forms a turn around leakage magnetic flux paths of power transfer windings2804and2806. Additionally, boosted coupled inductor2800is capable of being operated such that net mutual magnetic flux flowing through the turn of boost winding2808is essentially zero.

Referring again toFIG.8, controller806is implemented, for example, by analog and/or electronic circuitry. In some embodiments, controller806is at least partially implemented by a processor (not shown) executing instructions in the form of software and/or firmware stored in a memory (not shown). Although controller806is depicted as a discrete element for illustrative simplicity, controller806could be partially or fully integrated with one or more other elements of switching power converter800. For example, some subsystems of controller806could be incorporated in one or more of switching stages812. Additionally,FIG.8should not be construed to require that there be a separate control bus for each control signal. For example, controller806could be implemented by a combination of a central integrated circuit and local control logic integrated in each switching stage812, with a single control bus running from the central integrated circuit to each switching stage812. Furthermore, controller806may include multiple constituent elements that need not be co-packaged over even disposed at a common location

Controller806is configured to generate control signals U and L to control duty cycle of power stages804, where duty cycle is a portion of a switching cycle that a power transfer winding814is driven high, to regulate at least one parameter of switching power converter800. In some embodiments, controller806is configured to control duty cycle of power stages804using pulse width modulation and/or pulse frequency modulation. Examples of possible regulated parameters include, but are not limited, magnitude of input voltage Vin, magnitude of input current Iin, magnitude of output voltage Vo, and magnitude of output current Io. For example, in some embodiments, controller806is configured to generate control signals U and L to regulate magnitude of output voltage Vo, and controller806accordingly generates control signals U and L during continuous conduction operation of switching power converter800such that duty cycle of power stages804is equal to a ratio of output voltage magnitude Voover input voltage magnitude Vin. For example, if output voltage Vois to be regulated to two volts and input voltage Vi, is eight volts, controller806would generate control signals U and L such that duty cycle of power stages804is 0.25. Controller806is optionally configured to generate control signals U and L such that power stages804switch out-of-phase with each other. For example, in some embodiments, controller806is configured to generate control signals U and L such that each power stage804switches360/N degrees out of phase with an adjacent power stage804in the phase domain.

Blocking capacitor808prevents direct current shorting of output power node802to reference node820, while allowing alternating current to flow between boost winding811and output power node802. Accordingly, in particular embodiments, blocking capacitor808has a sufficiently large capacitance value such that a corner frequency established by blocking capacitor808and tuning inductor810is significantly lower than a switching frequency of switching power converter800, to enable injection of current flowing through boost winding811into output power node802. Additionally, in certain embodiments, tuning inductor810has an inductance of Lc, where Lc=Lco/(N1/N2)2. Lcois an inductance of tuning inductor810that would be selected if a ratio of N1/N2 were one instead of its actual value, such as to achieve a desired performance of switching power converter800if the ratio of N1/N2 were one instead of its actual value. As discussed below, in some applications, it may be desirable for the ratio of N1/N2 to be greater than N, instead of being equal to N.

Importantly, switching power converter800is configured such that current flowing through boost winding811at least partially cancels ripple current flowing through power transfer windings814, at output power node802. In particular, a respective direct current IDCand a respective alternating current IACflow through each power transfer winding814, as illustrated inFIG.8. Under steady state operating conditions of switching power converter800, each alternating current IACis a ripple current caused by switching of switching stages812. The sum of direct current IDCand alternating current IACflowing through a given power transfer winding814may be referred to as current822flowing through the power transfer winding, as illustrated inFIG.8. Additionally, an injection current Iinj, which is an alternating current, flows through boost winding811, tuning inductor810, and blocking capacitor808, as illustrated inFIG.8. Injection current Iinjis also a ripple current under steady state operating conditions of switching power converter800. Injection current Iinjis injected into output power node802. Boost winding811is electrically coupled to output power node802such that injection current Iinjis at least substantially out-of-phase with respect to alternating currents IACflowing through power transfer windings814. Additionally, as discussed above, in particular embodiments, a ratio of N1 to N2 is equal to N, such as that magnitude of injection current Iinjis substantially equal to a magnitude of a sum of all alternating currents IACflowing through power transfer windings814. Accordingly, injection current Iinjmay significantly cancel alternating currents IACat output power node802, thereby advantageously helping minimize magnitude of ripple in output current Io.

For example,FIGS.30A and30Bare graphs3002and3004, respectively, of current versus time of simulated operation of one embodiment of switching power converter800where N=4, switching frequency is one MHz, magnetizing inductance (Lm) of boosted coupled inductor815is 400 nH, each power transfer winding814has a leakage inductance (Ls) of 150 nH, N1/N2=4, Vin=12 volts, Vo=1 volt, and inductance (Lc) of tuning inductor130is 7.5 nH. Graphs3002and3004have a common time base. Graph3002includes curves representing each of currents822(1),822(2),822(3) and822(4) flowing through power transfer windings814(1),814(2),814(3), and814(4), respectively. Graph3004includes curves representing (a) total current824flowing through all power transfer windings814(seeFIG.8) assuming zero load at the converter output, (b) injection current Iinjinjected into output power node802, and (c) output current Iofor the zero load at the converter output. Total current824has a peak-to-peak ripple current magnitude of around 18 amperes, as illustrated inFIG.30B. If injection current Iinjwere not injected into output power node802, output current Iowould be the same as total current824, and output current Iowould therefore also have a peak-to-peak ripple current magnitude of around 18 amperes. However, injection current Iinjis out-of-phase with respect to total current824, and peak-to-peak magnitude of injection current Iinjis almost as large as the peak-to-peak magnitude of total current824. Therefore, injection current Iinjsubstantially cancels ripple current flowing through power transfer windings814, at output power node802. Consequently, output current Iouthas a relatively small ripple current magnitude of only around 3.3 amperes. As such, injecting injection current Iinjinto output power node802substantially reduces magnitude of ripple in output current Io.

While injection of injection current Iinjinto output power node802reduces magnitude of ripple in output current Io, such injection also degrades transient response of switching power converter800in a manner analogous to that discussed above with respect to switching power converter100. For example,FIG.31is a graph3100of current versus time illustrating simulated operation of an embodiment of switching power converter800in response to a change in load. Prior to time t1, a load powered by switching power converter800is zero, and magnitude of output current Iois therefore zero. At time t1, though, the load powered by switching power converter800rapidly increases, and total current824therefore ramps up. However, injection current Iinjinjected into output power node802ramps up in the opposite direction of total current824, and the injected current therefore slows the rise of output current Ioin response to the load increase, thereby degrading the transient response of switching power converter800.

Switching power converter800can be modified so that injection current Iinjadds to alternating currents IACin output power node802, instead of subtracting from (canceling) alternating currents IACin output power node802, thereby promoting fast transient response to the switching power converter at the expensive of larger magnitude of ripple in output current Io. For example,FIG.32is a schematic diagram of switching power converter3200, which is an alternate embodiment of switching power converter800that is optimized for fast transient response instead of for low ripple current magnitude. Switching power converter3200differs from switching power converter800only in the manner that boost winding811is electrically coupled to output power node802. In particular, in switching power converter800(FIG.8), boost winding811is electrically coupled between output power node802and reference node820such that injection current Iinjis out of phase with respect to alternating currents IACflowing through power transfer windings814. In switching power converter3200(FIG.32), in contrast, boost winding811is electrically coupled between output power node802and reference node820such that injection current Iinjadds to alternating currents IACflowing through power transfer windings814, thereby promoting fast transient response of switching power converter3200with the potential drawback of larger magnitude of ripple in output current Io. It should be noted, though, that magnitude of ripple current decreases with increasing number of phases of switching power converter3200. Therefore, in embodiments of switching power converter3200where N is large, increase in magnitude of ripple in output current Iocaused by injecting injection current Iinjinto output power node802may be immaterial because ripple current magnitude is very small due to N being large.

FIG.33is a graph3300of current versus time illustrating simulated operation of an embodiment of switching power converter3200in response to a change in load. Prior to time t1, a load powered by switching power converter3200is zero, and magnitude of output current Iois therefore zero (neglecting ripple current). At time t1, though, the load powered by switching power converter3200rapidly increases, and total current824therefore ramps up. Additionally, injection current Iinjinjected into output power node802ramps up in the same direction as total current824, and the injected current thereby adds to total current824and increases the rate of rise of output current Ioin response to the load increase, thereby improving the transient response of switching power converter3200. However, the fact that injection current Iinjadds to alternating currents IACflowing through power transfer windings814increases magnitude of ripple in output current10.

FIG.34is a top plan view of boosted coupled inductor900(FIG.9) illustrating one example of how electrical connections to a boost winding can be varied to achieve either a switching power converter optimized for low ripple current magnitude or a switching power converter optimized for fast transient response.FIG.34assumes that (a) boosted coupled inductor815of switching power converters800and3200is embodied by boosted coupled inductor900and (b) N=4 in switching power converters800and3200. It is understood, though, that switching power converters800and3200are not limited to use with boosted coupled inductor900or to N=4.FIG.34assumes that power transfer windings906, which are embodiments of power transfer windings814, are connected such that currents822flow from left to right when boosted coupled inductor900is viewed from its top, as illustrated inFIG.34. In this scenario, injection current Iinjflowing through boost winding908, where boost winding908is an embodiment of boost winding811, flows from right to left, as illustrated inFIG.34. Accordingly, if boosted coupled inductor900is used switching power converter800, end A of boost winding908is electrically coupled to output power node802and end B of boost winding908is electrically coupled to reference node820, as illustrated inFIG.8, so that injection current Iinjat least partially cancels alternating currents IACflowing through power transfer windings814. On the other hand, if boosted coupled inductor900is used switching power converter3200, end B of boost winding908is electrically coupled to output power node802and end A of boost winding908is electrically coupled to reference node820, as illustrated inFIG.32, so that injection current Iinjadds to alternating currents IACflowing through power transfer windings814.

As discussed above, magnitude of injection current Iinjis substantially equal to a magnitude of a sum of all alternating currents IACflowing through power transfer windings814if a ratio of N1 to N2 is equal to N. However, this condition applies only when boost winding811is perfectly magnetically coupled to power transfer windings814, or on in other words, when boosted coupled inductor815has infinite magnetizing inductance Lm. A practical implementation of boosted coupled inductor815, though, will not achieve infinite magnetizing inductance Lm, and magnitude of injection current Iinjtherefore will not be identical to a magnitude of a sum of all alternating currents IACflowing through power transfer windings814if a ratio of N1 to N2 is equal to N.

As such, it may be desirable to configure boosted coupled inductor815such that a ratio of N1 to N2 is larger than N to compensate for magnetizing inductance Lmof boosted coupled inductor815being finite. However, while increasing the ratio of N1 to N2 beyond N to a degree may improve operation of multi-phase switching power converter800or3200, increasing the ratio of N1 to N2 too much degrades performance of the switching power converter800or3200. Accordingly, there will typically be an optimal ratio of N1 to N2 that is greater than N, but is not too large, to compensate for magnetizing inductance of boosted coupled inductor815being finite.

For example,FIGS.35A-37Bcollectively illustrate three examples of how ratio of N1 to N2 affects operation of switching power converter800. Specifically,FIGS.35A and35Billustrate a first example where (a) N=4, (b) there is no load on switching power converter800, and (c) a ratio of N1 to N2 is smaller than optimal, such as due to finite magnetizing inductance Lmof boosted coupled inductor815.FIG.35Ais a graph3500of magnitude versus time including curves representing (a) total current824flowing through all power transfer windings814(seeFIG.8) and (b) injection current Iinjinjected into output power node802.FIG.35Bis a graph3502of magnitude versus time including a curve representing output current Io. As evident fromFIGS.35A and35B, injection current Iinjhelps cancel ripple current flowing through power transfer windings814, but the cancelation is not ideal.

FIGS.36A and36Billustrate a second example where (a) N=4, (b) there is no load on switching power converter800, and (c) a ratio of N1 to N2 is larger than in the example ofFIGS.35A and35B, such as greater than 4.FIG.36Ais a graph3600of magnitude versus time including curves representing (a) total current824flowing through all power transfer windings814and (b) injection current Iinjinjected into output power node802.FIG.36Bis a graph3602of magnitude versus time including a curve representing output current Io. As evident when comparingFIGS.35B and36B, increasing the ratio of N1 to N2 from the value ofFIGS.35A and35Bto the value ofFIGS.36A and36Bsignificantly decreased ripple in output current Io.

FIGS.37A and37Billustrate a third example where (a) N=4, (b) there is no load on switching power converter800, and (c) a ratio of N1 to N2 is larger than in the example ofFIGS.36A and36B.FIG.37Ais a graph3700of magnitude versus time including curves representing (a) total current824flowing through all power transfer windings814and (b) injection current Iinjinjected into output power node802.FIG.37Bis a graph3702of magnitude versus time including a curve representing output current Io. As evident when comparingFIGS.36B and37B, increasing the ratio of N1 to N2 from the value ofFIGS.36A and36Bto the value ofFIGS.37A and37Bincreased ripple in output current Ioand changed polarity of output current Io. Therefore, the ratio of N1 to N2 in the example ofFIGS.37and37Bis too large, and the optimal value of the ratio of N1 to N2 is between (a) the value of the example ofFIGS.36A and36Band (b) the value of the example ofFIGS.37A and37B.

Referring again toFIG.8, in some applications, it may be desirable for N1 to be less than N. For example, in high current applications, it may be desirable for N1 to be equal to one to minimize conduction losses in power transfer windings814. However, it is not feasible for the ratio of N1 to N2 to be equal to N, or to be greater than N, if N1 is less than N, due to impracticality of implementing a fractional value of N2. Consequently, is not feasible to select a ratio of N1 to N2 where magnitude of injection current Iinjis substantially equal to a magnitude of a sum of all alternating currents IACflowing through power transfer windings814if N1 is less than N.

Applicant has found that the aforementioned limitation can be overcome by including a transformer electrically coupled in series with boost winding811to enable additional freedom in setting the relationship between magnitude of injection current Iinjand magnitude of currents822flowing through power transfer windings814. For example,FIG.38is a schematic diagram of a multi-phase switching power converter3800, which is an alternate embodiment of switching power converter800(FIG.8) further including a transformer3826. Transformer3826includes a primary winding P and a secondary winding S. Primary winding P forms a quantity of turns equal to N3, and secondary winding S forms a quantity of turns equal to N4. Primary winding P of transformer3826is electrically coupled in series with boost winding811, and each of boost winding811and primary winding P of transformer3826is also electrically coupled to reference node820. Secondary winding S of transformer3826is electrically coupled in series with blocking capacitor808and tuning inductor810, and secondary winding S of transformer3826is also electrically coupled to reference node820. Accordingly, boost winding811is electrically coupled in series with blocking capacitor808via transformer3826.

A ratio of current Ibstflowing through boost winding811to injection current Iinjis given by the following relationship: Iin=(Ibst)*(N3/N4). Consequently, magnitude of injection current Iinjis substantially equal to a magnitude of a sum of all alternating currents IACflowing through power transfer windings814if the following relationship holds true, assuming boosted coupled inductor815has infinite magnetizing inductance Lm: (N1/N2)*(N3/N4)=N. As such, switching power converter3800can be configured so that magnitude of injection current Iinjis substantially equal to a magnitude of a sum of all alternating currents IACflowing through power transfer windings814even if N1 is less than N, with proper selection of N3 and N4. For example, if N1 is equal to one and N2 and N4 are each equal to one for simplicity, N3 may be equal to N to satisfy the following relationship: (N1/N2)*(N3/N4)=N. In some applications, it may be desirable for the quantity (N1/N2)*(N3/N4) to be somewhat greater than N to compensate for boosted coupled inductor815having finite leakage inductance, for reasons analogous to those discussed above with respect to multi-phase switching power converters800and3200. In a manner analogous to that discussed above with respect to switching power converter800, the respective topological locations of blocking capacitor808, tuning inductor810, and secondary winding S of transformer3826may vary as long as these three elements are electrically coupled in series between output power node802and reference node820.

While tuning inductor810is a discrete element in switching power converter3800, tuning inductor810could alternately be implemented by leakage inductance of transformer3826. For example,FIG.39is a schematic diagram of a multi-phase switching power converter3900, which is an alternate embodiment of switching power converter3800(FIG.3) where transformer3826is replaced with a transformer3926. Transformer3926is configured to implement tuning inductor810by leakage inductance3910of transformer3926, and tuning inductor810is therefore omitted. Transformer3926is configured, for example, such that leakage inductance3910has an inductance of Ls, where Ls=Lco/(N1/N2*N3/N4)2. Lcois an inductance of leakage inductance3910that would be selected if a value of (N1/N2*N3/N4) were one instead of its actual value, such as to achieve a desired performance of switching power converter3900if the value of (N1/N2*N3/N4) were one instead of its actual value. Switching power converter3200(FIG.32) could be modified to further include a transformer in a manner similar to switching power converter3800or3900.

FIGS.40-44collectively illustrate a transformer4000, which is one possible embodiment of transformer3926ofFIG.39.FIG.40is a perspective view of transformer4000,FIG.41is a bottom plan view of transformer4000,FIG.42is a side elevational view of transformer4000,FIG.43is a cross-sectional view of transformer4000taken along line43A-43A ofFIG.42, andFIG.44is a cross-sectional view of transformer4000taken along line44A-44A ofFIG.41.FIGS.40-44collectively illustrate three directions4002,4004, and4006, where (i) direction4002is orthogonal to each of directions4004and4006, (ii) direction4004is orthogonal to each of directions4002and4008, and (ii) direction4006is orthogonal to each of directions4002and4004.

Transformer4000includes a magnetic core4008, a winding4010(seeFIGS.41,42,43, and44) and a winding4012(seeFIGS.40,41,43, and44). Magnetic core4008includes a first rail4014(seeFIGS.40-43), a second rail4016(seeFIGS.40-44), a first leg4018(see FIGS.41-44), a second leg4020(seeFIGS.41and44), and a leakage element4022(seeFIGS.40,42, and44). Magnetic core4008is formed, for example, of a ferrite magnetic material or a powdered iron magnetic material. First rail4014and second rail4016are separated from each other in direction4002, and each of first leg4018and second leg4020is disposed between first rail4014and second rail4016in direction4002. First leg4018is separated from second leg4020in direction4004. In other embodiments, first leg4018and/or second leg4020are separated from first rail4014and/or second rail4016by a respective gap (not shown), such as to help prevent saturation of magnetic core4008. Each of first leg4018and second leg4020optionally also forms a respective gap (not shown) along direction4004, such that the leg is broken into two or more portions separated from each other in direction4002by the gap.

Winding4010is wound around leg4018, and winding4012is wound around leg4020. Winding4010is an embodiment of primary winding P of transformer3926, and winding4012is an embodiment of secondary winding S of transformer3926. Winding4010is depicted as forming a plurality of turns, while winding4012is depicted as being a single-turn winding. However, the quantity of turns formed windings4010and4012may vary.

Leakage element4022is joined to first rail4014, and leakage element4022extends towards second rail4106in direction4002provide a relatively low reluctance path for leakage magnetic flux to flow between first rail4014and second rail4016. Leakage element4022is separated from second rail4106by a gap4024in direction4002where gap4024is filled with, for example, air, paper, plastic, adhesive, and/or a magnetic material having a lower magnetic permeability that magnetic material forming leakage element4022. Gap4024can be split into two or more smaller gaps to decrease the fringing flux. The dotted lines delineating leakage element4022from first rail4014are to assist a viewer in distinguishing features of transformer4000, and these lines do not necessarily represent discontinuities in transformer4000. Leakage inductance of transformer4000, which is an embodiment of leakage inductance3910ofFIG.39, can be controlled, for example, by the configuration of gap4024. For example, leakage inductance of transformer4000can be increased by decreasing thickness of gap4024in direction4002and/or by increasing cross-sectional area of leakage element4022in a plane extending in directions4004and4006.

However, it may not be possible to obtain sufficiently low leakage inductance in some applications of transformer4000, such as when a ratio of N3 (quantity of turns formed by winding4010) to N4 (quantity of turns formed by winding4012) is large.FIGS.45-48collectively illustrate a transformer4500, which is another embodiment of transformer3926and can achieve lower leakage inductance than transformer4000.FIG.45is a front elevational view of transformer4500,FIG.46is a side elevational view of transformer4500,FIG.47is a top plan view of transformer4500, andFIG.48is a bottom plan view of transformer4500.FIGS.45-48collectively illustrate three directions4502,4504, and4506, where (i) direction4502is orthogonal to each of directions4504and4506, (ii) direction4504is orthogonal to each of directions4502and4508, and (ii) direction4506is orthogonal to each of directions4502and4504.

Transformer4500includes a magnetic core4508, a winding4510, and a winding4512. Magnetic core4508is formed, for example, of a ferrite magnetic material or a powdered iron magnetic material. Magnetic core4508forms a passageway4514extending through magnetic core4508in direction4506, and magnetic core4508optionally forms one or more gaps (not shown) to help prevent saturation of magnetic core4508. Each of winding4510and winding4512is wound through passageway4514and around a leg4516of magnetic core4508. Winding4510is an embodiment of primary winding P of transformer3926, and winding4512is an embodiment of secondary winding S of transformer3926. Winding4510is depicted as forming a plurality of turns, while winding4512is depicted as being a single-turn winding. However, the quantity of turns formed windings4510and4512may vary.

Winding4510is depicted as partially overlapping winding4512, or stated differently, winding4510is depicted as being partially wound over winding4512, although winding4510is electrically isolated from winding4512.FIG.45also illustrates an offset4518of winding4510with respect to winding4510, where offset4518is proportional to an extent that winding4510does not overlap winding4512. Leakage inductance of transformer4500is minimized if winding4510completely overlaps winding4512, and leakage inductance of transformer4500is therefore a function of offset4518. For example, leakage inductance of transformer4500will decrease as offset4518is decreased, and leakage inductance of transformer4500will increase as offset4518is increased.

Transformer3926(FIG.39) could be embodied in manners other than those discussed above with respect toFIGS.40-48as long as a required turns ratio and leakage inductance can be achieved. For example, primary winding P and/or secondary winding S of transformer3926could be embodied by windings in a substrate, such as PCB windings. It may be particularly advantageous to embody windings of transformer3926by PCB windings in applications where low leakage inductance of transformer3926is required because PCB windings can typically be configured in a manner that achieves low leakage inductance. Additionally, leakage inductance of PCB windings is usually readily adjustable during PCB design, and PCB windings can achieve tightly controlled and repeatable leakage inductance values in typical manufacturing environments.

Referring again toFIGS.38and39, switching power converters3800or3900could be modified to include one or more additional boosted coupled inductors. For example,FIG.49is a schematic diagram of a multi-phase switching power converter4900, which is an alternate embodiment of multi-phase switching power converter3800further including M power stages4904and a boost winding4911, where M is an integer greater than one. In some embodiments, M is the same as N, and in some other embodiments, M is different from N. Each power stage4904corresponds to a respective phase of switching power converter4900, such that switching power converter includes N+M phases, i.e., N phases associated with power stages804and M phases associated with power stages4904. Switching power converter4900further includes a controller4906in place of controller806. Control signals generated by controller4906are not shown inFIG.49for illustrative clarity.

Each power stage4904includes a switching stage4912electrically coupled to a power transfer winding4914at a switching node Y. Each power transfer winding4914is electrically coupled between the switching node Y of its respective power stage4904and output power node802. Each switching stage4912is configured to repeatedly switch the switching node Y of its power stage4904between input power node818and reference node820, in response to control signals (not shown) generated by controller4906. For example, switching stage4912(1) is configured to repeatedly switch node Y(1) between input power node818and reference node820in response to control signals (not shown) generated controller4906. Controller4906is also configured to generate control signals to control switching stages812in a manner analogous to how controller806generates control signals to control switching stages812in switching power converter800.

Boost winding811, boost winding4911, and primary winding P of transformer3826are electrically coupled in series, and a closed circuit including these three elements is partially embodied by reference node820. In some alternate embodiments, boost winding811, boost winding4911, and primary winding P of transformer3826are electrically coupled in series without being electrically coupled to reference node820.

Power transfer windings4914are magnetically coupled by a magnetic core4913. Boost winding4911is magnetically coupled to a leakage component (not illustrated inFIG.49) of each power transfer winding4914by magnetic core4913. In particular, boost winding4911is configured such that it forms at least one turn around a respective total magnetic flux path of each power transfer winding4914. Boost winding4911is additionally configured such that net mutual magnetic flux within the turns of the boost winding is essentially zero during steady state operation of switching power converter4900. Equivalently, boost winding4911can be configured such that it forms at least one turn around only a respective leakage magnetic flux path of each power transfer winding4914. In some embodiments, boosted coupled inductor4915is embodied by one of boosted coupled inductors900,1700,1900,2000,2200,2500, or2800, discussed above. Importantly, each boosted coupled inductor of switching power converter4900has a common turns ratio. Specifically, each power transfer winding814and each power transfer winding4914forms common a quantity of turns equal to N1, and each boost winding811and each boost winding4911forms a common quantity of turns equal to N2.

In view of switching power converter4900including two boosted coupled inductors configured to cancel the output current ripple from the main phases, magnitude of injection current Iin; is substantially equal to a magnitude of a sum of all alternating currents IACflowing through power transfer windings814and power transfer windings4914if the following relationship holds true, assuming that each of boosted coupled inductor815and boosted coupled inductor4915has infinite magnetizing inductance Lm: (N1/N2)*(N3/N4)=QP, where QP is total quantity of phases in the switching power converter and is therefore equal to N+M in switching power converter4900. It should be noted that switching power converter4900could be modified to include one or more additional boosted coupled inductors with boost windings electrically coupled in series with primary winding P of transformer3826, which would result in the switching power converter4900including additional phases and QP therefore being larger than the sum of N+M. In some applications, it may be desirable for the quantity (N1/N2)*(N3/N4) to be somewhat greater than QP to compensate for boosted coupled inductor815and boosted coupled inductor4915each having finite leakage inductance, for reasons analogous to those discussed above with respect to multi-phase switching power converters800and3200.

In some embodiments of switching power4900, tuning inductor810has an inductance of Lc, where Lc=Lc0/(PH)2. Lc0is an inductance of tuning inductor810that would be selected if a ratio of (N1/N2)*(N3/N4) were one instead of its actual value, such as to achieve a desired performance of switching power converter4900if the ratio of (N1/N2)*(N3/N4) were one instead of its actual value. As discussed above, in some applications, it may be desirable for the ratio of (N1/N2)*(N3/N4) to be greater than QP, instead of being equal to QP.

In an alternate embodiment, switching power converter4900is configured such that injected current Iinjflowing through secondary winding S of transformer3826, tuning inductor810, and blocking capacitor808adds to the current flowing through power transfer windings814and9414, which increases output current Ioduring a transient loading or unloading event, thereby promoting fast transient response of switching power converter4900. In this alternate embodiment, injected current Iinjdoes not have to ideally match the total current ripple from the main phases and therefore the turns ratio of transformer3826does not have to be approximately equal (N1/N2)*(N3/N4)=PH. Increasing the turns ratio of transformer3826should generally increase the total output current Ioduring a transient loading or unloading event, but increasing the turns ratio too much can cause limitations in terms of parasitics in transformer3826, or in transient current slew rate.

As mentioned above, switching power converter4900could be modified to include one or more additional boosted coupled inductors sharing transformer3826, i.e., where the respective boost winding of each boosted coupled inductor is electrically coupled in series with primary winding P of transformer3826. Switching power converter4900could also be modified to include two or more inductor-transformer sets, where each set includes a respective transformer3826and one or more respective boosted coupled inductors, where the boost winding of each boosted coupled inductor of the set is electrically coupled in series with the primary winding P of the respective transformer3826of the set. In these alternate embodiments, the power transfer windings of the one or more respective boosted coupled inductors of each set are, for example, electrically coupled to output power node802. The respective turns ratio and inductance of tuning inductor810for each set are determined, for example, as discussed above with respect toFIG.49.

Referring again toFIGS.1,6,8,32,38,39, and49, while the switching power converters discussed above have a buck-type topology, the concept of injecting current into a power node is not limited to use in a buck-type topology. Instead, injection current could be injected into a power node in other topologies where current flowing through a plurality of windings sums at a power node. It should be noted that injection current could be injected to a power node other than an output power node, such as an input power node, in certain topologies.

Switching Power Converters With Injection

Applicant has additionally determined that use of injection in a switching power converter can also help reduce magnitude of ripple in output current. For example,FIG.50is a schematic diagram of a multi-phase switching power converter5000including transformers and an injection stage. Switching power converter5000includes N power stages5002, an injection stage5004, and a controller5006, where N is an integer greater than one. Each power stage5002corresponds to a respective phase of switching power converter5000, such that switching power converter5000is an N-phase switching power converter. In particular, power stage5002(1) corresponds to a first phase of switching power converter5000, power stage5002(2) corresponds to a second phase of switching power converter5000, as so on.

Each power stage5002includes a power switching stage5008and a power transformer5010, where the power switching stage5008is electrically coupled to a primary winding P of the power transformer5010at a switching node X. The primary winding P of each power transformer5010is electrically coupled between the switching node X of its respective power stage5002and a common output node5012. Output node5012has a voltage Vo, and an output current Ioflows to a load (not shown) electrically coupled to output node5012. Output current Iocould have a negative polarity without departing from the scope hereof. One or more capacitors5014are optionally electrically coupled to output node5012.

Each power switching stage5008is configured to repeatedly switch the switching node X of its power stage5002between an input power node5016and a reference node5013, in response to control signals U and L generated by controller5006. Specifically, power switching stage5008(1) is configured to repeatedly switch node X(1) between input power node5016and ground in response to control signals U(1) and L(1), power switching stage5008(2) is configured to repeatedly switch node X(2) between input power node5016and reference node5013in response to control signals U(2) and L(2), and so on. Input power node5016is at a voltage Vin, and each power switching stage5008accordingly repeatedly switches node X of its power stage5002between voltage Vinand zero volts relative to ground. An input current Iinflows from an electrical power source (not shown) to switching power converter5000via input power node5016. Input current Iincould have a negative polarity without departing from the scope hereof. A primary winding P of a given power transformer5010(1) in switching power converter5000is driven “high” when its respective switching node X is at voltage Vin, and the primary winding P is driven “low” when its respective switching node X is at zero volts relative to reference node5013. In certain embodiments, power switching stages5008are similar to switching stages202ofFIG.2. Reference node5013is depicted as being a ground node, such as an earth ground node or a chassis ground node. It is understood, though, that reference node5013need not be a ground node, and reference node5013accordingly could be at a different electrical potential than an earth ground or a chassis ground.

Injection stage5004includes an injection switching stage5018and an injection transformer5020, where injection switching stage5018is electrically coupled to a primary winding P of injection transformer5020at a switching node X(N+1). Primary winding P of injection transformer5020is electrically coupled between switching node X(N+1) and injection output node5022. Injection output node5022, which is separate from output power node5012, is at a voltage Vo_z, and one or more capacitors5024are electrically coupled to injection output node5022. Injection switching stage5018is configured to repeatedly switch node X(N+1) between input power node5016and reference node5013in response to control signals UI and LI. Similar to primary windings P of power transformers5010, primary winding P of injection transformer5020is driven high when switching node X(N+1) is at voltage Vin, and primary winding P of injection transformer5020is driven low when switching node X(N+1) is at zero volts relative to reference node5013. Injection stage5004does not handle a direct current component of output current Io. Instead, controller5006controls injection stage5004to reduce, or even essentially eliminate, alternating current voltage across leakage inductances of power transformers5010, as discussed below. In certain embodiments, injection switching stage5018is similar to an instance of switching stage202ofFIG.2.

Secondary windings S of power transformers5010, as well as secondary winding S of injection transformer5020, are electrically coupled in series with each other. WhileFIG.50depicts the series connections of secondary windings S being partially embodied by reference node5013, secondary windings S could alternately be isolated from reference node5013, as long as they are electrically coupled in series. An optional tuning inductor5030is electrically coupled in series with secondary windings S of power transformers5010, as well as with secondary winding S of injection transformer5020. The topological location of tuning inductor5030could vary as long as it is electrically coupled in series with secondary windings S. While tuning inductor5030is depicted as being a discrete inductor, tuning inductor5030could instead be embodied by intrinsic inductance of a circuit including secondary windings S, particularly in applications where tuning inductor5030need only have a small inductance value. Furthermore, tuning inductor5030could be omitted, or tuning inductor5030could be replaced with a plurality of tuning inductors, without departing from the scope hereof.

Controller5006is configured to generate control signals U and L to control duty cycle of power stages5002, where duty cycle is a portion of a switching cycle that a primary winding P of a power transformer5010is driven high, to regulate at least one parameter of switching power converter5000. Controller5006is configured to vary duty cycle of power stages5002, for example, using a pulse width modulation technique and/or a pulse frequency modulation technique. Controller5006is optionally configured to generate control signals U and L such that power stages5002switch out-of-phase with each other.

Controller5006is further configured to generate control signals UI and LI to control injection stage5004such that the injection stage injects magnetic flux in each power transformer5010in a manner which reduces voltage across leakage inductance of each power transformer5010. Such reduction in voltage across leakage inductances advantageously reduces, or even essentially eliminates, magnitude of ripple current associated with charging and discharging of leakage inductances. Not only does such collective operation of injection stage5004and controller5006reduce magnitude of ripple in each current ILflowing through a respective power transfer winding P, it also reduces magnitude of ripple in output current Io.

For example,FIG.51Ais a graph5102of output current Ioversus time illustrating simulated operation of an embodiment of switching power converter5000with injection stage5004disabled. Peak-to-peak magnitude of output current Iois around 20 amperes with injection stage5004disabled, as illustrated in graph5102.FIG.51Bis a graph5104of output current Ioversus time illustrating simulated operation of the same embodiment of switching power converter5000that is simulated in theFIG.51Asimulation but with injection stage5004enabled. Peak to peak magnitude of output current Iois around 3.6 amperes with injection stage5004enabled, as illustrated in graph5104. As such, operation of injection stage5004in switching power converter5000reduces magnitude of ripple in output current Ioby about 5.5 times, in the embodiment of switching power converter5000simulated inFIGS.51A and51B.

FIG.52is a schematic diagram of a multi-phase switching power converter5200including a boosted coupled inductors and an injection stage. Switching power converter5200includes N power stages5202, an injection stage5204, and a controller5206, where Nis an integer greater than one. Each power stage5202corresponds to a respective phase of switching power converter5200, such that switching power converter5200is an N-phase switching power converter.

Each power stage5202includes a power switching stage5208electrically coupled to a power transfer winding5210at a switching node X. Each power transfer winding5210is electrically coupled between the switching node X of its respective power stage5202and a common output node5212. Output node5212has a voltage Vo, and an output current Ioflows to a load (not shown) electrically coupled to output node5212. Output current Iocould have a negative polarity without departing from the scope hereof. One or more capacitors5214are optionally electrically coupled to output node5212.

Each power switching stage5208is configured to repeatedly switch the switching node X of its power stage5202between an input power node5216and a reference node5217, in response to control signals U and L generated by controller5206. Specifically, power switching stage5208(1) is configured to repeatedly switch node X(1) between input power node5216and reference node5217in response to control signals U(1) and L(1), power switching stage5208(2) is configured to repeatedly switch node X(2) between input power node5216and ground in response to control signals U(2) and L(2), and so on. Input power node5216is at a voltage Vin, and each power switching stage5208accordingly repeatedly switches node X of its power stage5202between voltage Vinand zero volts relative to reference node5217. Reference node5217is depicted as being a ground node, such as an earth ground node or a chassis ground node. It is understood, though, that reference node5217need not be a ground node, and reference node5217accordingly could be at a different electrical potential than an earth ground or a chassis ground. In certain embodiments, power switching stages5208are similar to switching stages202ofFIG.2. An input current Iinflows from an electrical power source (not shown) to switching power converter5200via input power node5216. Input current Iincould have a negative polarity without departing from the scope hereof. A given power transfer winding5210in converter5200is driven “high” when its respective switching node X is at voltage Vin, and the power transfer winding5210is driven “low” when its respective switching node X is at zero volts relative to reference node5217.

Injection stage5204includes an injection switching stage5218electrically coupled to a boost winding5220at a switching node X(N+1). Boost winding5220is electrically coupled between switching node X(N+1) and injection output node5222. Injection output node5222, which is separate from output power node5212, is at a voltage Vo_z, and one or more capacitors5224are electrically coupled to injection output node5222, such that each capacitor5224is electrically coupled in series with boost winding5220. A first tuning inductor5230is electrically coupled in series with boost winding5220. Although first tuning inductor5230is illustrated as being electrically coupled between boost winding5220and capacitor5224, first tuning inductor5230could be at a different topological location as long as it is electrically coupled in series with boost winding5220. First tuning inductor5230is omitted in some alternate embodiments of switching power converter5200, such as in embodiments where a circuit including boost winding5220has sufficient inductance such that first tuning inductor5230is not required. Switching power converter5200optionally further includes a second tuning inductor5232electrically coupled in parallel with boost winding5220. Second tuning inductor5232, when present, typically has a relatively large inductance value.

Injection switching stage5218is configured to repeatedly switch switching node X(N+1) between input power node5216and ground in response to control signals UI and LI. Similar to power transfer windings5210, boost winding5220is driven high when switching node X(N+1) is at voltage Vin, and boost winding5220is driven low when switching node X(N+1) is at zero volts relative to reference node5217. Injection stage5204does not handle a direct current component of output current Io. Instead, controller5206controls injection stage5204to reduce, or even essentially eliminate, alternating current voltage across leakage inductances of power transfer windings5210, as discussed below, thereby reducing magnitude of ripple current flowing through power transfer windings5210, as well as reducing magnitude of ripple in output current Io. In certain embodiments, injection switching stage5218is similar to an instance of switching stage202ofFIG.2.

Power transfer windings5210are magnetically coupled by a magnetic core5226. Boost winding5220is magnetically coupled to a leakage component (not illustrated inFIG.52) of each power transfer winding5210by magnetic core5226. In particular, boost winding5220is configured such that it forms at least one turn around a respective leakage magnetic flux path of each power transfer winding5210. Importantly, boost winding5220is additionally configured such that net mutual magnetic flux within the turns of the boost winding is essentially zero during steady state operation of switching power converter5200. Such configuration of boost winding5220advantageously helps minimize risk of a magnetic saturation from mutual magnetic flux within a magnetic flux path of boost winding5220while enabling injection stage5204to reduce ripple current magnitude in switching power converter5200. Power transfer windings5210, boost winding5220, and magnetic core5226are part of a boosted coupled inductor5228. In certain embodiments, boosted coupled inductor5228is configured similar to one of the boosted coupled inductors discussed above with respect toFIGS.9-29.

Controller5206is configured to generate control signals U and L to control duty cycle of power stages5202, where duty cycle is a portion of a switching cycle that a power transfer winding5210is driven high, to regulate at least one parameter of switching power converter5200. In some embodiments, controller5206is configured to control duty cycle of power stages5202using pulse width modulation and/or pulse frequency modulation. Examples of possible regulated parameters include, but are not limited, magnitude of input voltage Vin, magnitude of input current Iin, magnitude of output voltage Vo, and magnitude of output current Io. Controller5206is optionally configured to generate control signals U and L such that power stages5202switch out-of-phase with each other.

Controller5206is further configured to generate control signals UI and LI to control injection stage5204such that the injection stage injects magnetic flux in magnetic core5226in a manner which reduces voltage across leakage inductances of power transfer windings5210. Such reduction in voltage across leakage inductance of power transfer windings5210advantageously reduces, or even essentially eliminates, magnitude of ripple current associated with charging and discharging of the leakage inductance, thereby reducing magnitude of ripple current flowing through power transfer windings5210, as well as reducing magnitude of ripple in output current Io.

For example,FIG.53Ais a graph5302of output current Ioversus time illustrating simulated operation of an embodiment of switching power converter5200with injection stage5204disabled. Peak-to-peak magnitude of output current Iois around 26 amperes with injection stage5204disabled, as illustrated in graph5302.FIG.53Bis a graph5304of output current Ioversus time illustrating simulated operation of the same embodiment of switching power converter5200that is simulated in theFIG.53Asimulation but with injection stage5204enabled. Peak to peak magnitude of output current Iois around 3.6 amperes with injection stage5204enabled, as illustrated in graph5304. As such, operation of injection stage5204in switching power converter5200reduces magnitude of ripple in output current Ioby about 8.1 times, in the embodiment of switching power converter5200simulated inFIGS.53A and53B.

FIG.54is a block diagram of an electrical system5400, which is one possible application of the new switching power converters disclosed herein. System5400includes a switching power converter5402configured to power a load5404. Switching power converter5402may be any one of the new switching power converters disclosed herein. For example, switching power converter5402may be any one of switching power converters100,600,800,3200,3800,3900,4900,5000, or5200. Load5404includes, for example, one or more integrated circuits, including but not limited to, a processing unit (e.g. a central processing unit (CPU) or a graphics processing unit (GPU)), a field programmable gate array (FPGA), an application specific integrated circuit (ASIC) (e.g. for artificial intelligence and/or machine learning), and/or a memory unit.

Combinations of Features

Features described above may be combined in various ways without departing from the scope hereof. The following examples illustrate some possible combinations.

(A1) A switching power converter includes (i) a plurality of power stages, each power stage including a respective power transformer, and (ii) a blocking capacitor. The blocking capacitor and a respective secondary winding of each power transformer are electrically coupled in series between an output power node of the switching power converter and a reference node of the switching power converter.

(A2) The switching power converter denoted as (A1) may further include a tuning inductor electrically coupled in series with the injection capacitor and the respective secondary winding of each power transformer.

(A3) Either one of the switching power converters denoted as (A1) or (A2) may further include a controller configured to control duty cycle of the plurality of power stages to regulate at least one parameter of the switching power converter.

(A4) In any one of the switching power converters denoted as (A1) through (A3), each power transformer may include a respective primary winding electrically coupled to the output power node of the switching power converter.

(A5) The switching power converter denoted as (A4) may be configured such that current flowing through the secondary windings of the power transformers at least partially cancels ripple current flowing through the primary windings of the power transformers, at the output power node of the switching power converter.

(A6) The switching power converter denoted as (A4) may be configured such that current flowing through the secondary windings of the power transformers adds to alternating current flowing through the primary windings of the power transformers, at the output power node of the switching power converter.

(A7) In any one of the switching power converters denoted as (A4) through (A6), each power stage may further include a respective switching stage electrically coupled electrically coupled to the primary winding of the respective power transformer of the power stage.

(A8) Any one the switching power converters denoted as (A1) through (A7) may have a buck-type topology.

(B1) A switching power converter includes a first power stage, a second power stage, and a blocking capacitor. The first power stage includes (i) a first switching stage and (ii) a first power transformer including a first primary winding and a first secondary winding, where the first primary winding is electrically coupled between the first switching stage and an output power node of the switching power converter. The second power stage includes (i) a second switching stage and (ii) a second power transformer including a second primary winding and a second secondary winding, where the second primary winding is electrically coupled between the second switching stage and the output power node of the switching power converter. The blocking capacitor, the first secondary winding, and the second secondary winding are electrically coupled in series between the output power node of the switching power converter and a reference node of the switching power converter.

(B2) The switching power converter denoted as (B1) may further include the third power stage, where the third power stage includes a third switching stage and a third power transformer. The third power transformer includes a third primary winding and a third secondary winding, where the third primary winding is electrically coupled between the third switching stage and the output power node of the switching power converter. The blocking capacitor, the first secondary winding, the second secondary winding, and the third secondary winding may be electrically coupled in series between the output power node of the switching power converter and the reference node of the switching power converter.

(B3) Either one of the switching power converters denoted as (B1) or (B2) may further include a controller, where the controller being configured to control at least the first switching stage and the second switching stage to regulate at least one parameter of the switching power converter.

(C1) A switching power converter includes (i) a plurality of power stages, each power stage including a respective power transfer winding, (ii) a boost winding forming at least one turn around a respective leakage magnetic flux path of each power transfer winding, and (iii) a blocking capacitor. The blocking capacitor and the boost winding may be electrically coupled in series between an output power node of the switching power converter and a reference node of the switching power converter.

(C2) The switching power converter denoted as (C1) may further include a controller configured to control duty cycle of the plurality of power stages to regulate at least one parameter of the switching power converter.

(C3) In the switching power converter denoted as (C1), each power stage may further include a respective switching stage electrically coupled to the respective power transfer winding of the power stage.

(C4) The switching power converter denoted as (C3) may further include a controller configured to control the respective switching stage of each power stage to regulate at least one parameter of the switching power converter.

(C5) Any one of the switching power converters denoted as (C1) through (C4) may further include a tuning inductor electrically coupled in series with the blocking capacitor and the boost winding.

(C6) Any one of the switching power converters denoted as (C1) through (C5) may be configured such that current flowing through the boost winding at least partially cancels ripple current flowing through the power transfer windings of the power stages, at the output power node of the switching power converter.

(C7) Any one of the switching power converters denoted as (C1) through (C5) may be configured such that current flowing through the boost winding adds to alternating current flowing through the power transfer windings of the power stages, at the output power node of the switching power converter.

(C8) Any one of the switching power converters denoted as (C1) through (C7) may have a buck-type topology.

(C9) Any one of the switching power converters denoted as (C1) through (C8) may further include a transformer, where the boost winding is electrically coupled in series with the blocking capacitor via the transformer.

(D1) A switching power converter includes a first power stage, a second power stage, a boost winding, and a blocking capacitor. The first power stage includes (i) a first switching stage and (ii) a first power transfer winding electrically coupled between the first switching stage and an output power node of the switching power converter. The second power stage includes (i) a second switching stage and (ii) a second power transfer winding electrically coupled between the second switching stage and the output power node of the switching power converter. The boost winding forms at least one turn around a respective leakage magnetic flux path of at least the first power transfer winding and the second power transfer winding. The blocking capacitor and the boost winding are electrically coupled in series between the output power node of the switching power converter and a reference node of the switching power converter.

(D2) The switching power converter denoted as (D1) may further include a third power stage, where the third power stage includes (i) a third switching stage and (ii) a third power transfer winding electrically coupled between the third switching stage and the output power node of the switching power converter. The boost winding may further form at least one turn around a leakage magnetic flux path of the third power transfer winding.

(D3) Either of the switching power converters denoted (D1) or (D2) may further include a controller, where the controller is configured to control at least the first switching stage and the second switching stage to regulate at least one parameter of the switching power converter.