Transistor switch device

A transistor switch device includes a main power transistor, an auxiliary solid-state switch, and an electricity replenishing means, wherein the electricity replenishing means is connected to the auxiliary solid-state switch which is connected between the collector and the base of the main power transistor so that the voltage drop at the collector of the main power transistor is reduced.

BACKGROUND OF THE INVENTION 
1. Field of the Invention 
The present invention relates to a transistor switch device capable of 
operation with a reduced voltage drop at the collector of the main power 
transistor. 
2. Description of the Prior Art 
High-power transistors, in comparison with thyristors, have a self turn-off 
function but cause a high collector voltage drop in the high-voltage 
element during conduction, resulting in a large loss. In a Darlington 
transistor circuit, the loss is especially large, amounting to two to 
three times that in a thyristor circuit. In applications where large power 
is controlled or converted, the Darlington transistor circuit has been 
considered impracticable in view of its need for considerable cooling and 
its low current utilizing efficiency. 
With respect to FIGS. 1(a) and 1(b), there are shown circuit diagrams of 
prior art power transistor switch devices. FIG. 1(a) shows a conventional 
Darlington circuit, comprising power sources 100a and 100b, a load 200, 
main power transistors 1a and 1b, and auxiliary transistors 2a and 2b 
connected between the collectors and the bases, respectively, of the main 
power transistors 1a and 1b. In this Darlington circuit, the necessary 
collector-emitter voltage V.sub.CE.sbsb.2 (ON) of the auxiliary transistor 
2 in the ON state is supplied from the collector-base voltage 
V.sub.CB.sbsb.1 (ON) of the main transistor 1 in the ON state and hence 
the collector-emitter voltage V.sub.CE.sbsb.1 (ON) of the main transistor 
1 in the ON state becomes large causing the collector loss to be increased 
during power supply. This has lowered the current utilizing efficiency and 
made cooling difficult. For these reasons, the Darlington transistor 
circuit as in FIG. 1(a) is not practical for use as a large-capacity power 
switch device. 
FIG. 1(b) shows another power transistor switch device in which the voltage 
drop at the collector of the main transistor 1 in the ON state as well as 
loss can be reduced. However, a rather large amount of power is required 
from power sources B.sub.1 and B.sub.2 to drive the bases thereof. The 
power can be reduced but at the sacrifice of greater collector voltage 
drop and loss in the main power transistor 1. 
SUMMARY OF THE INVENTION 
An object of the invention is to provide a transistor switch device 
comprising an electricity replenishing means connected to the auxiliary 
solid-state switch in order to reduce the voltage drop in the main 
transistor and to save the necessary capacity of the base drive power 
source. 
Another object of the invention is to provide a transistor switch device in 
which the base drive power for the main transistor is self-sufficiently 
replenished. 
The foregoing and other objects are attained in accordance with one aspect 
of the present invention through the provision of a transistor switch 
device comprising a main power transistor having a collector, a base and 
an emitter, an auxiliary solid-state switch, electricity replenishing 
means, means connecting the auxiliary solid-state switch between the 
collector and the base of the main power transistor, means connecting the 
electricity replenishing means to the solid-state switch whereby the 
voltage drop at the collector of the main power transistor is reduced.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
Referring now to the drawings, wherein like reference numerals designate 
identical or corresponding parts throughout the several views, and more 
particularly to FIG. 2(a) thereof, there is shown a transistor switch 
circuit comprising main power transistors 1a and 1b, auxiliary solid-state 
switches 2a and 2b connected between the collectors and the bases, 
respectively, of the main power transistors 1a and 1b, and a replenishing 
power source 40 connected in series to the auxiliary solid state switches 
2a and 2b. FIG. 2(a) shows an example wherein the replenishing power 
source 40 is used in common for the two power transistors. The auxiliary 
solid-state switches 2a and 2b may be of transistor, gate-turn-off 
thyristor as in FIG. 2(b), thyristor (with an extinction circuit provided 
separately), or the like. The replenishing power source 40 may only be 
capable of offering a voltage as low as 0.5 to 2 V. According to the 
invention, even if the collector-base voltages of the main transistors 1a 
and b are low, that is, the collector-emitter voltages thereof are low, 
the voltage drops in the auxiliary solid-state switches 2a and 2b 
(hereinafter briefly, auxiliary transistors) while they are in the ON 
state are replenished with the voltage from the power source 40. As a 
result, the necessary collector potential of the auxiliary transistor 2 is 
duly maintained, and the current amplification factor hfe.sub.2 remains 
sufficiently high. This serves to suppress the voltage drop in the main 
transistor 1. 
This operation will be described more specifically by referring to FIG. 3. 
FIG. 3(a) is an equivalent circuit diagram of the prior art circuit as in 
FIG. 1(b). In FIG. 3(a), the reference Vd.sub.3 denotes the minimum 
voltage required of the base drive power sources 3a and 3b. The voltage 
Vd.sub.3 is the sum of the base-emitter voltage V.sub.BE.sbsb.1 of the 
main transistor 1 and the collector-emitter voltage drop V.sub.2 of the 
auxiliary transistor 2, or Vd.sub.3 =(V.sub.BE.sbsb.1 +V.sub.2). 
In the circuit of the invention as in FIG. 2(a), the voltage Vd.sub.4 of 
the replenishing power source 40 is lower than the collector potential 
(i.e., Vd.sub.3) of the auxiliary transistor 2 by V.sub.CE.sbsb.1 which is 
the collector-emitter voltage of the main transistor 1. That is, Vd.sub.4 
=(Vd.sub.3 -V.sub.CE.sbsb.1)=(V.sub.BE.sbsb.1 +V.sub.2 -V.sub.CE.sbsb.1). 
In other words, under the application of the same base-drive voltage, the 
voltage V.sub.2 of the auxiliary transistor 2 can be maintained high and 
its current amplification factor can be increased, with the result that 
the collector voltage drop V.sub.CE.sbsb.1 of the main transistor is 
reduced and the loss is reduced accordingly. Under the application of the 
same base-drive current Ib.sub.1 (the same base-drive power source 
current) and with the same collector voltage drop V.sub.CE.sbsb.1 of the 
main transistor, the necessary base-drive power source voltage Vd.sub.4 is 
lowered by V.sub.CE.sbsb.1, permitting the necessary capacity of the 
base-drive power source to be reduced. It is apparent that the collector 
voltage drop V.sub.CE.sbsb.1 is lowered for the same base-drive power 
source capacity. Thus, according to the invention, the voltage drop and 
the collector loss in the main transistor are reduced to make cooling 
easy. For the same loss, greater current is allowed to flow in the 
circuit. Furthermore, because the current amplification factor hfe.sub.2 
of the auxiliary transistor can be maintained high, the external turn-on 
base current Ib.sub.2 may be made small. 
Furthermore, as in FIG. 3, the main current I.sub.L =Ic.sub.1 +Ic.sub.2. 
That is, the collector-emitter current Ic.sub.2 is branched to the main 
transistors. In other words, the current Ic.sub.2 passes through the main 
power path and sustains the power path current, enabling the current 
capacity of the auxiliary transistor to be effectively utilized in sharing 
the load current. The replenishing power source 40 of FIG. 2(a) may be of 
a DC source such as a battery or other DC source provided by the use of a 
rectifier or chopper. The replenishing power source used in the circuit as 
in FIG. 2(c) is the combination of a current source 41 and a diode 42. 
(The diode 42 is used also as a branch means when the auxiliary transistor 
switch is cut off.) The replenishing power source 40 and its current 
source 41 will be described below in more detail. 
FIGS. 4(a) to 4(c) show examples of replenishing power source 40. 
In FIG. 4(a), the DC replenishing power source 40 having a drooping 
characteristic (current limiting characteristic) is obtained from an AC 
power source 43 by way of an AC impedance 44 and a rectifier 45. Having a 
current limiting characteristic, the DC replenishing power source 40 
operates half as the current source 41. When the impedance 44 is of 
reactance X, the loss will be small. 
In FIG. 4(b), the DC replenishing power sources 40a to 40n are obtained 
from a high frequency AC power source 46 by way of a transformer 47 and 
rectifiers 45a to 45n. In this example, a current source 41 is obtained by 
the use of a current limiting transformer such as a leakage transformer in 
place of the transformer 47. 
FIG. 4(c) shows an example wherein the replenishing power source (current 
source 41) 40 is obtained from an m-phase AC power source 43 through 
rectifiers 45a to 45n by way of current transformers 48a to 48n and an 
impedance Z. The impedance Z may be of a power network from which power is 
supplied to the main power transistor circuits such as, for example, the 
input circuit of the commercial AC power source used as the AC source 43 
or may be provided separately. 
Thus, according to the invention, the replenishing power is distributed by 
way of a transformer whereby the necessary capacity of the drive power 
souce is reduced and the size of the transformer which occupies a 
considerable area of the base drive circuit is reduced. 
The current source systems shown in FIGS. 2(b), 2(c) and 4 are advantageous 
for the following reasons. When the replenishing source voltage Vd.sub.4 
is too high and the impedance is low, the shunt current to the side of the 
auxiliary solid-state switch is likely to become too large. However, a 
current source system having a drooping and current limiting 
characteristic does not give rise to such a problem. At the same time, 
influence from variations in the voltage drop V.sub.2 in the auxiliary 
solid-state switch is largely diminished. Furthermore, the circuit of the 
invention operates at a minimum of drive voltage Vd.sub.4 at all times for 
individual operating states, permitting the necessary power to remain at a 
minimum. This makes it possible to reduce the size of the transformer to a 
minimum. Still further, the circuit as shown in FIG. 2(a) operates like a 
known Darlington circuit even without the replenishing voltage Vd.sub.4 if 
the period for which the replenishing voltage is absent (ripple or dip) is 
short enough or the ratio of the period for which the replenishing voltage 
is absent or too low to the entire operating period is small enough. This 
means that the need for smoothing the replenishing power source (base 
drive power source) is obviated or the smoother used is simplified. Also, 
for a short overcurrent load, the collector voltage V.sub.CE.sbsb.1 of the 
main power transistor rises to initiate the function of the 
self-sufficient power supply. Hence, a shortage of the replenishing power 
does not immediately cause an abrupt rise in the collector voltage of the 
main power transistor and possible resultant breakdown. 
FIG. 5 shows another embodiment of the invention wherein a plurality of 
pairs of bidirectional conductive power control converter units (a 
plurality of pairs of half-bridge circuits) are used to realize a method 
for replenishing a plurality of power transistors with power. This 
embodiment can be applied to inverters, DC power control, time-ratio 
modulation power amplifiers and the like. 
In FIG. 5, the reference numeral 46 denotes a high frequency AC current 
source, and 48 a current transformer. Because the necessary power supply 
voltage may be low at a high frequency, the number of the secondary 
windings of the transformer can be made as small as one turn 
(through-winding) to several turns. On the side of primary winding, it is 
desirable that the voltage be made high and the current be made low 
because this approach facilitates forming the high frequency current 
source. (This principle does not apply to the instance in which a high 
frequency current source is set up from a power source whose voltage is as 
low as ten or several volts.) For this reason, FIG. 5 shows an example 
wherein the current transformer has more turns of winding on the primary 
side than on the secondary side. In large power devices, power can be 
supplied from a group of simple current transformers whose secondary 
winding is of one to several turns, and hence power distribution through 
current transformers of a primary series connection type is desirable. In 
the embodiments as in FIGS. 4(c) and 5, the current or voltage present due 
to turn-on at the main transistor or a group of main transistors can be 
used as the input to the transformers 47 or 48. 
According to the invention, as described above, an auxiliary solid-state 
switch is installed between the collector and the base of the main power 
transistor, and an electricity replenishing means is connected to the 
auxiliary solid-state switch whereby the collector voltage drop in the 
main power transistor is reduced. For the same collector voltage drop, a 
low power source voltage suffices for base drive, permitting the necessary 
drive power to be reduced. 
FIG. 6 is a circuit diagram showing another embodiment of the invention, 
which comprises a power circuit connecting a power source 100 (an AC power 
source when one of the directions of an AC circuit is controlled) to a 
load 200, and a main power transistor 1 connected in series to at least 
one power path X-Y of the power circuit which is controlled in the on-off 
mode. An auxiliary solid-state switch 2 is provided between the collector 
and the base of the power transistor in order to supply base current 
Ib.sub.1. An electric transformer 3 is provided for deriving electricity 
(such as voltage, current or power) produced in the power circuit due to 
the function of the power transistor. The output .nu..sub.3 and I.sub.2 of 
the transformer 3 is supplied to a branch circuit C.sub.2 --2--B.sub.1 of 
the auxiliary solid-state switch 2. This embodiment exemplifies a single 
closed-loop power circuit. 
In this embodiment, the transformer 3 is a current transformer with its 
primary winding or primary through-conductor 31 inserted in the power 
path, and its secondary winding 32 (which includes the primary winding in 
the case of an autotransformer connection) inserted in series with the 
solid-state switch branch circuit. The auxiliary solid-state switch 2 is a 
transistor or thyristor. When the solid-state switch 2 is a transistor, 
this transistor switch can be used as an electricity replenishing means 
(as will be described later), which makes it possible to form a Darlington 
transistor circuit. In such an application, increase in the voltage drop 
in the main power transistor is obviated. In prior art techniques, the 
voltage drop in the main power transistor is great even in a two-stage 
(two subordinate collector common) Darlington circuit, and hence is not 
suited for large power capacity switch devices. In a three-stage 
Darlington circuit, the voltage drop in the main power transistor (with 
capacity greater than 50 A, 200 V) is too high and the resultant loss is 
too large for practical use. The auxiliary solid-state switch 2 is 
preferably a gate-turn-off type thyristor. When the ordinary thyristor is 
used for the auxiliary solid-state switch 2, it is necessary to also 
provide an extinction means. 
In the embodiment using a current transformer as in FIG. 6, the auxiliary 
solid-state switch 2 is cut off. The shunt means for the secondary current 
I.sub.2 which flows when the solid-state switch 2 is cut off is 
constituted of rectifier elements and resistors. The shunt means 4 may be 
connected into the circuit as indicated by 4'. When the shunt means 4 is a 
rectifier element in the position indicated by the solid line, it is 
necessary to provide one or a plurality of such rectifier elements 
connected in series, depending upon the voltage drop in the auxiliary 
solid-state switch 2 in the ON state. Instead of this rectifier element, a 
rectifier element whose forward voltage drop is large, or a series of 
resistors (or nonlinear resistance elements) and a rectifier element may 
be used. When the rectifier element is in position 4' indicated by the 
dotted line, it is not necessary to provide a plurality of rectifier 
elements connected in series to increase the voltage drop in the shunt 
means. 
The rectifier elements 5a and 5b serve as base reverse biasing current 
paths for turn-off operation, through which the carrier stored in the main 
transistor 1 or auxiliary solid-state switch 2 is released. The Darlington 
transistor switch 2 within the frame may be formed of a one-package 
transistor or a monolithic IC. The numeral 6 denotes an external turn-on 
control means including the reverse biasing path. 
The core 33 of the current transformer 3 may be a ferrite system core or 
permalloy system core for repetitive high frequency operation or a silicon 
steel system core for low frequency operation. In a current power network, 
an ordinary AC current transformer can be used as the transformer 3. The 
core of the transformer 3 may be in the shape of cut, noncut, lamination, 
etc. with a multi-leg construction. In the circuit of FIG. 1, the 
transformer is inserted, together with the power transistor 1, in series 
with the DC power path X-Y, to make this transformer operate as a 
repetitive pulse current transformer as will be described below. 
In FIG. 7, assume that the initial state 1 where the main power 
transistor 1 and the auxilairy solid-state switch 2 are ON is at the point 
1 on the core B-H curve (represented by the exciting current I and the 
number of interlinking fluxes .phi.) as shown in (a). This initial point 
corresponds to the timing 1 on the operating waveform shown in FIG. 8. 
When a turn-on signal Ib.sub.2 as in FIG. 8(a) is applied to the auxiliary 
solid-state switch 2, the switch 2 turns on after a delay td.sub.2. The 
turn-on of the main power transistor 1, i.e., the presence of the full 
load current I.sub.L at the main power transistor 1, is delayed beyond the 
delay td.sub.2, to allow the voltage on the power path X-Y to be 
sustained. The turn-on delay time td.sub.1 of the main power transistor 1 
consists of the flux reset time t.sub.R of the current transformer (or 
saturable reactor in a generic sense) and the delay td.sub.10 after the 
increase of base current Ib.sub.1 as in FIGS. 8(b) and 8(c). 
A negative voltage VR is applied to the secondary winding 32 in FIG. 6 
(where the dot mark indicates the positive polarity) during the delay 
period td.sub.1 (especially t.sub.R), and the core flux falls along the 
B-H loop as in FIGS. 7(a) (i) and 8(b) (i). 
When the main transistor starts turning on before it reaches the negative 
saturation level, the base current Ib.sub.1 (which is equal to the current 
I.sub.2 of the auxiliary solid-state switch 2) of the main transistor 1 
increases along the solid line Q of FIG. 8(c) and thereafter steady 
current transformer function starts. In this case, the minimum flux level 
stands at point 2 of FIG. 7(a) or point 2 of FIG. 8(b). 
When the main transistor turns on after it reaches the negative saturation 
level, the base current Ib.sub.1 of the main transistor 1 exhibits a peak 
as indicated by the dotted line P in FIG. 8(c). 
During the turn-on start delay period td.sub.1 of the main transistor 1, 
the flux of the core 33 is reset and thereafter steady current transformer 
function starts. 
In the event of a heavy load, i.e., a large I.sub.L, the voltage drop 
(substantial turn-on) of the main transistor 1 is delayed and the product 
of the negative voltage of the current transformer and the time for which 
the negative voltage is sustained, that is, the flux reset value 
.phi..sub.r as in FIG. 7(b), increases. The greater the load I.sub.L, the 
longer the delay td.sub.1, with the result that the core reaches the 
negative saturation level -.phi..sub.s, causing a peak as indicated by the 
dotted line P in FIG. 8(c). 
The voltage drop delay time td.sub.1 of the main transistor depends upon 
the ratio of the excitation current I.sub..epsilon. (indicated by the 
dotted encircled line in FIG. 8(c) to the secondary winding of the current 
transformer 3 (FIG. 6), to the power path current I.sub.L. The larger the 
ratio I.sub.L /I.sub..epsilon., the longer the delay td.sub.1. Thus, the 
core flux is reset to the negative saturation level -.phi..sub.s. 
When the main transistor 1 turns on, the potential difference between the 
power paths X-Y and C.sub.1 -Y decreases whereby the winding 31 functions 
as the primary winding, and the winding 32 as the secondary winding with a 
main current I.sub.L. 
The potential V.sub.C2 at the terminal C.sub.2 of the solid-state switch is 
the sum of the potential V.sub.B1 at the base B.sub.1 of the main 
transistor 1 and the voltage drop V.sub.(C.sbsb.2.sub.-E.sbsb.2.sub.) in 
the solid-state switch, and is larger than the collector voltage 
V.sub.(C.sbsb.1.sub.-E.sbsb.1.sub.) of the main transistor; that is, 
V.sub.(C.sbsb.1.sub.-E.sbsb.1.sub.) &lt;V.sub.C2. Therefore, the current 
transformer voltage .nu..sub.3 is expressed as 
EQU .nu..sub.3 =V.sub.C2 -V.sub.(C.sbsb.1.sub.-E.sbsb.1.sub.) =V.sub.B1 
+V.sub.(C.sbsb.2.sub.-E.sbsb.2.sub.) -V.sub.(C.sbsb.1.sub.-E.sbsb.1.sub.) 
(1) 
In the prior art Darlington circuit, the transformer 3 is not used and 
.nu..sub.3 =0. As a result, the condition 
V.sub.(C.sbsb.1.sub.-E.sbsb.1.sub.) =V.sub.B1 
+V.sub.(C.sbsb.2.sub.-E.sbsb.2.sub.) serves as a limiting factor, causing 
the voltage drop V.sub.(C.sbsb.1.sub.-E.sbsb.1.sub.) in the main 
transistor to be increased. 
However, according to the invention, the transformer voltage .nu..sub.3 is 
replenished by virtue of the current transformer 3. During this 
replenishing period ii in FIG. 7, the core flux rises along ii as in FIG. 
7(a) and (b), where .nu..sub.3 =(.nu..sub.32 +.nu..sub.31). For this 
period, the flux .phi. assumes a curve for the positive low voltage period 
ii as in FIG. 8(b). Then the solid-state switch current I.sub.2 is: 
##EQU1## 
where N.sub.31 and N.sub.32 denote the numbers of turns of the current 
transformer windings. In FIG. 6, the number of turns N.sub.32 of the 
secondary winding includes those of the windings 31 and 32 since this 
transformer is an autotransformer. 
As in FIG. 6, 
EQU I.sub.L =I.sub.1 +I.sub.2 (3) 
The base current Ib.sub.1 of the main transistor 1 is supplied proportional 
to the collector current I.sub.1 of the main transistor. The ratio 
N.sub.32 /N.sub.31 is determined as: 
##EQU2## 
wherein hfe(min) denotes the minimum current amplification factor applied 
for the maximum current in the main transistor. 
The time toneff of the steady period ii can be determined to be 
sufficiently long relative to the flux reset time t.sub.R. 
The flux level reaches the point 3 in FIGS. 2(a), (b) and 8(b) 
immediately before the main transistor 1 is turned off. When the reverse 
bias -Ib.sub.1 is supplied, the auxiliary solid-state switch 2 turns off 
after its delay ts.sub.2, causing the base current Ib.sub.1 to be shunted 
to the shunt means 4. Then the main transistor 1 turns off also after its 
delay ts.sub.1. At this turn-off, the reverse bias pulse -Ib.sub.1 is 
supplied from the turn-on control means 6 when necessary. In FIG. 8(a), 
the reference -Ib.sub.2 denotes a steady reverse bias current. 
The core flux .phi. reaches the point 4 of FIG. 2(a) and (b) when the 
turn-off of the main transistor is completed. In this operation, the flux 
curve is nearly as 3 as in FIG. 7 when the shunt means 4 serves as the 
rectifier element indicated by the solid line in FIG. 6. When the shunt 
means 4 is a resistor or rectifier element indicated by the dotted line, a 
high positive voltage is applied to the current transformer 3 during the 
period ts.sub.1 for which the auxiliary solid-state switch 2 is in the OFF 
state and the main transistor 1 is in the ON state. As a result, the flux 
becomes more positive as 4 in FIG. 7(i c). If the flux level of 3 is 
near the positive flux saturation level +.phi..sub.s, the period ts.sub.1 
is reduced. 
Then, for the period the main transistor 1 is OFF, the core flux changes 
toward the excitation current zero line (Y axis), at a velocity depending 
upon the shunt means 4,4' or 4R. A series of the above operations is 
repeated. 
In the above operation, the condition .nu..sub.3 .ltoreq.0.5 to 1 V (where 
.nu..sub.3 is the replenishing voltage) holds as long as the auxiliary 
solid-state switch 2 is one transistor (not a Darlington connection) or 
one thyristor. This value of .nu..sub.3 is below the base-emitter voltage 
drop in the main transistor. When the auxiliary solid-state switch is a 
Darlington transistor switch, .nu..sub.3 1 to 2 V. As the voltage 
.nu..sub.3 becomes higher, the voltage drop in the auxiliary solid-state 
switch is allowed to be higher and the voltage drop in the main transistor 
1 can be lower. This means that the switch device of the invention is 
operable with a minimum of power loss, ease of cooling, and is applicable 
to large power capacity devices. 
As shown in FIG. 9(a), ON-OFF of the auxiliary solid-state switch is 
assumed to occur in the operation for the minimum OFF time toffmin. The 
turn-off delay time ts of the main transistor 1 is included in the 
substantial ON time toneff. The flux reset time t.sub.R of the current 
transformer and the turn-on delay time td.sub.10 of the main transistor 
are included in the substantial OFF time toffeff. These time relations are 
shown in FIG. 9(b). 
The current transformer operating condition dependent on the worst flux 
reset is expressed as: 
##EQU3## 
where E stands for the voltage immediately before turn-on of the power 
path X-Y, and the flux reset voltage is assumed to be VR=E. 
Here the number of turns of the secondary winding is equal to that between 
terminals C.sub.1 and C.sub.2, as in Eq. (2). 
Also the following equation is established. 
##EQU4## 
That is, the turn-on duration ton is E/.nu..sub.3 times the flux reset time 
t.sub.R. The turn-on duration multiplying factor K can range from 100 to 
600 at a power path voltage of 200 to 300 V. Accordingly, when td.sub.10 
.ltoreq.t.sub.R, the maximum turn-on time ratio .alpha. max is: 
##EQU5## 
This maximum value is readily feasible. 
In the embodiment shown in FIG. 6, as described above, the switch device 
functions substantially as a DC current transformer in spite of its use of 
a magnetic current transformer whereby a high turn-on ratio .alpha. is 
maintained. In other words, the device makes DC power control possible. 
Even for low-frequency AC control, the use of a high-frequency (time-ratio 
control frequency) current transformer suffices. Hence the size of the 
transformer can be reduced. For example, the use of a one-turn through 
type current transformer is sufficient, the size and construction of which 
may be about the same as a saturable reactor which has hitherto been 
utilized in suppressing the turn-on di/dt of the thyristor. 
Furthermore, a considerable amount of base-drive power for the main power 
transistor is supplied self-sufficiently from the main power path. At the 
same time, the device of the invention offers the effect of reducing the 
loss in the main power transistor. These advantages enhance the usefulness 
of the device of the invention. 
FIGS. 10(a) and 10(b) are circuit diagrams showing other embodiments of the 
invention, wherein a shunt means 7 is provided for shunting the excitation 
current I.sub..epsilon. to the base of the main transistor 1 for the flux 
reset period t.sub.R at the beginning of turn on. This shunt means 
(transistor) is self-controlled through a tertiary winding 34 which 
detects the core flux reset voltage. Operating waveforms of the circuits 
are shown in FIG. 11. 
In FIGS. 10(a) and 10(b), when the auxiliary solid-state switch 2 starts 
turning on, the excitation shunt transistor 7 turns on by the emf of the 
tertiary winding 34 through an impedance 8, thereby maintaining the main 
transistor 1 in the OFF state. (The shunt transistor 7 turns on before the 
main transistor turns on with the turn-on delay td.sub.1.) As a result, 
the excitation current I.sub..epsilon. of the current transformer 3 flows 
in the shunt transistor 7. The waveforms thereof are indicated by I.sub.7 
and I.sub..epsilon. in FIG. 11(c). 
When the core flux approaches near or reaches the negative saturation 
level, the tertiary winding voltage decreases to cause the shunt 
transistor 7 to turn off. Consequently, the current I.sub.2 of the 
auxiliary solid-state switch 2 is supplied to the base of the main 
transistor 1 whereby the main transistor is turned on, as in FIG. 11(d) 
and (e). When the main transistor turns on, the tertiary winding is at a 
slightly negative voltage or a very low voltage and hence it is unlikely 
for the shunt means 7 to be turned on. The circuit will thereafter operate 
in the same manner as the one shown in FIG. 6. 
FIG. 10(b) shows an example wherein a voltage transformer 3 is installed in 
a turn-on power path beside the main transistor power path (main 
transistor branch) X-Y. 
In the embodiments shown in FIG. 10, the core flux can securely be reset to 
about the negative saturation level irrespective of the turn-on delay of 
the main transistor and irrespective of the ratio of the main current 
I.sub.L to the core flux current I.sub..epsilon.. 
FIG. 12 is a diagram showing operation for maximum continuous turn-on in 
the circuit as in FIG. 10. In FIG. 12(b), the auxiliary solid-state switch 
2 is kept turned on continuously. At the same time, the excitation shunt 
means 7 is turned on for a given time t.sub.7 at the maximum ON time 
tonmax. The period t.sub.7 is approximately equal to the turn-on delay 
td.sub.10 of the main transistor subtracted from the sum of the turn-off 
delay ts.sub.1 of the main transistor and the flux reset time t.sub.R, or 
t.sub.7 =ts.sub.1 +t.sub.R -td.sub.10. Thus, as in FIG. 12(c), the main 
transistor repeats ON-OFF at an ON-OFF ratio at which the main transistor 
can be considered to be substantially in a continued ON state. 
In the circuit of FIG. 10, the excitation shunt means 7 can be turned 
on-off by an external pulse of given width which synchronizes with the 
turn-on signal Ib.sub.2. In this case, the need for the tertiary winding 
34 is obviated. 
FIG. 13 shows another embodiment of the invention in connection with 
improvements in the turn-on initial flux reset method. This circuit 
comprises a diode or breakover switch 13, which may be a thyristor with 
anode igniting means, 5-layer semiconductor switch such as SSS, dynistor, 
PNPN switch, or the like. When the breakover voltage of the breakover 
switch is higher than the base-emitter reverse peak voltage of the main 
transistor 1 and hence the voltage (during flux resetting) of the tertiary 
winding 34 is to be made high, a protective means such as reverse peak 
protection rectifier element 14, resistor 15, etc. should be provided. 
In FIG. 13(a), when the auxiliary solid-state switch 2 is turned on, a 
negative voltage is applied to the current transformer 3 before the main 
transistor 1 starts turning on, causing a negative voltage to be induced 
in the tertiary winding 34 whereby the breakover switch 13 turns on and 
the auxiliary solid-state switch current I.sub.2, i.e., the excitation 
current I.sub..epsilon., flows in the excitation shunt means 7 which 
comprises the tertiary winding 34 and the breakover switch 13. During this 
operation, the base of the main transistor 1 is negatively biased. 
When the voltage (negative) of the tertiary winding becomes small at the 
end of core flux reset, the base potential of the main transistor 1 rises 
to cause the main transistor to turn on. Thereafter the circuit will 
operate in the same manner as in FIG. 6. When the breakover switch 13 is 
an element whose forward voltage drop and holding current IH are large and 
is capable of being readily turned off (such as having a V-I 
characteristic as indicated by (v) in FIG. 13(b)), the breakover switch 
turns off while the main transistor 1 is being turned on. In this manner, 
this circuit operates as in FIG. 11. 
FIG. 14 shows another embodiment of the invention wherein the main 
transistor 1 is kept turned on continuously and the function of the 
current transformer 3 is maintained continuously. 
When the auxiliary solid-state switch 2 and the main transistor 1 are 
continuously in the ON state as in FIG. 14(b), an intermittent flux reset 
switch 17 is turned on for the period t.sub.R at intervals of time T. In 
this operation, the period t.sub.R can be controlled automatically under 
self-control by the tertiary winding 34. Instead of the tertiary winding 
34, an external signal may be used to turn on the switch 17 for a given 
period t.sub.R. With self-control, it is desirable that the impedance 18 
be a differential element comprising, for example, a parallel 
resistor-capacitor circuit. During the period t.sub.R for which the flux 
reset switch 17 is in the ON state, a negative voltage VR is applied to 
the secondary winding voltage .nu..sub.32. During this operation, the 
diode 16 blocks the voltage VR, thereby preventing it from 
short-circuiting over the side of the main transistor. In this state, the 
secondary winding current I.sub.2 is larger than the magneto-motive force 
(N.sub.31.I.sub.1) of the primary winding current I.sub.1 by the 
excitation current, that is, 
##EQU6## 
(where I.sub..epsilon. is the excitation current changing from negative to 
positive). In other words, the function of the current transformer is 
maintained in the above state. When the flux reset switch 17 is turned 
off, the circuit operation shifts to the foregoing steady operation. Thus 
the secondary winding current I.sub.2, i.e., the base current of the main 
transistor 1, assumes the waveform I.sub.2 shown in FIG. 14(b), and the 
current transformer voltage .nu..sub.32 assumes the waveform .nu..sub.32. 
As described above, the function of the current transformer 3 is steadily 
maintained while the main transistor is kept turned on continuously. In 
other words, the invention achieves its aim regarding "perfect turn-on by 
DC" or makes it possible to realize a 100% turn-on ratio. 
It is apparent that the switch circuit of the invention can be effectively 
utilized within a finite turn-on time. The embodiments shown in FIGS. 1, 
10, 13 and 14 are uniquely advantageous in that a means for applying the 
pulse reset voltage VR is provided and the current transformer function is 
maintained by a single power transistor circuit by the use of a single 
core having substantially a full time range. Prior art techniques have 
failed in offering power transistor switch devices capable of DC operation 
by the use of a transistor or the like; it has been thought theoretically 
impracticable. One example known regarding such operation depends upon the 
use of plural-core AC (alternating) type construction. 
FIGS. 15(a) and 15(b) are diagrams showing other embodiments of the 
invention wherein the reference 3.nu. denotes a voltage transformer, 
31.nu. and 32.nu. the primary and secondary windings respectively, 36.nu. 
a reset winding, and 201 a main current commutating diode (such as a 
flywheel diode or a feedback rectifier) used while the main transistor is 
in the OFF state. 
In FIG. 15, when the auxiliary solid-state switch 2 is turned on, the main 
transistor 1 turns on, a voltage is applied to the transformer 3.nu., a 
voltage .nu..sub.3 (whose positive polarity is indicated by the dot (.)) 
is induced across the secondary winding 32.nu., and this voltage is 
supplied to the auxiliary solid-state switch branch. As a result, the 
necessary voltage drop in the auxiliary solid-state switch and the 
base-emitter voltage drop in the main transistor are duly maintained even 
if the collector voltage drop in the main transistor is insufficient, as 
in the embodiment shown in FIG. 1. In other words, the collector-emitter 
voltage drop and consequent power loss in the main transistor are reduced. 
When the auxiliary solid-state switch 2 and the main transistor 1 are 
turned off, a reverse magneto-motive force is developed through the reset 
winding 36.nu. (used in common with the winding 32.nu. in FIG. 15(a)) by 
the current of the main current path, such as by the commutating diode 
201, and thus the core flux is reset. 
In this voltage transformer system, core flux reset cannot be sufficiently 
done to make it impossible to operate the circuit over a wide range of 
turn-on time ratios if the single DC circuit as in FIG. 15 is employed. To 
solve this problem, it is necessary to use a means (such as a power 
source) for providing a potential difference between anodes x.sub.1 and 
x.sub.2. This corresponds to an AC control as will be described later. 
Also, a means is provided for allowing a given voltage to be applied to or 
induced across the primary winding 31.nu.. 
FIG. 16 is a circuit diagram showing another embodiment of the invention 
comprising a plurality of main power transistors. This circuit consists 
essentially of a power network group A and a power network group B, which 
have a main transistor group 1A comprising mA numbers of main transistors, 
and a main transistor group 1B comprising mB numbers of main transistors, 
respectively. They are connected to an XA-YA group of mA numbers of power 
paths and an XB-YB group of mB numbers of power paths respectively. The 
switch device further comprises nA numbers of transformers 3A and nB 
numbers of transformers 3B, and also a plurality of power sources 100A and 
100B, and loads 200A and 200B. FIG. 16 is a conceptual diagram showing one 
group of closed loops comprising main power transistors, power sources and 
loads. In this arrangement, the power sources and loads are commonly 
operated in the individual power networks. 
The two power networks A and B may be consolidated together into one as 
shown by simplified diagrams in FIG. 1. 
In FIG. 15, assume that the two circuits (a) and (b) are a power 
closed-loop. Then it becomes possible for the two power loops to supply 
the base drive current (drive voltage) of the main power transistor on the 
side opposite to each other. A simple way to effect this operation is to 
turn on and off the main power transistors 1A and 1B synchronously with 
each other as in FIG. 16. 
Assume that both groups A and B comprise a plurality of closed-loops in 
which at least one of the transformers is in operation and at least one of 
the transformer secondary windings connected to one power transistor is in 
operation. Then this power transistor can be arbitrarily turned on-off and 
the supply of its base-drive power is secured. In this case, it is not 
necessary to satisfy the foregoing synchronized turn-on-off condition for 
the power transistors. 
What is important is that at least one of the transformers connected to the 
main transistor be operated at least for the period when this main 
transistor is turned on. To meet this requirement, there may be a variety 
of types of power networks and transformer networks available. 
Basic principles and ideas of the invention have been described above. For 
a better understanding of the invention, several application examples will 
be described below. 
Referring to FIGS. 18(a) and 18(b), there are shown examples of application 
of the invention to an inverter of the single-way connection type; (a) is 
a current transformer system, and (b) a voltage transformer system, 
wherein an output transformer 200 and a voltage transformer 3 are used on 
common. The voltage transformer 3 has its core used in common for main 
power transistors 1a and 1b. In the voltage transformer system as in FIG. 
18(b), a base-drive current limiting resistor r (with a small resistance) 
may be used when the tap voltage is too high at the secondary windings 
32.nu.a and 32.nu.b. The circuit as in FIG. 17 may be applied to various 
single way connections and multiphase connections. 
The circuits shown in FIGS. 18(a) and 18(b) constitute a series-pair 
connection (half-bridge) used for bidirectional power conversion control 
or bidirectional conductive power conversion control and are suited for 
applications to inverters and DC power controls. 
The circuit shown in FIG. 18(a) is an example wherein an AC current 
transformer is employed with its core used in common. This circuit can be 
used in the voltage transformer system when the number of turns of the 
primary winding 31 is increased and the primary winding is connected in 
parallel to the load 200. 
The circuit in FIG. 18(b) is an example wherein a current transformer is 
disposed for each main power transistor. This example is suited for 
applications to control of a DC load (reversible-polar load 200 or 
unipolar voltage load 200'), time-ratio modulation type inverters operable 
in a wide frequency range, etc. 
In FIG. 18, the rectifier element 201 may be connected in the position 
indicated by the dotted line 201'. The circuit in FIG. 18 can be applied 
to various bridge circuitry. 
FIG. 19 is a circuit diagram showing a switch circuit embodying the concept 
of the invention illustrated in FIG. 16. In FIG. 19, the numeral 300 
denotes a one unit arm comprising a main transistor 1, a rectifier element 
201, an auxiliary solid-state switch 2, and a branch means 4. There are 
also provided a main current terminal X-Y, and base-drive current supply 
terminals C.sub.1 and C.sub.2. The reference 20 indicates a rectifier for 
a base-drive current supply, and 400A and 400B and bridge type inverter 
units having a construction similar to each other. The group A inverter 
unit 400A receives power from a primary current transformer 3B installed 
in the output current path of the group B inverter unit 400B. This power 
supply may be generated from a current transformer 3A inserted in its own 
output current path. The secondary output current of the primary current 
transformer 3B is supplied commonly through a rectifier 20a to three unit 
arms of the positive group. The secondary output current of this primary 
current transformer 3B is connected to the primary windings of the 
secondary current transformers 3a, 3b and 3c. The secondary outputs of 
these secondary current transformers are connected to rectifiers 20b, 20c 
and 20d and are supplied to three unit arms, respectively, of the negative 
group. 
In this case, it is necessary to operate the two groups 400A and 400B 
simultaneously where the turn-on-off synchronism among power transistors 
need not be established. This example is advantageous in that the load 
current is balanced between the two groups A and B. 
In this example, a DC voltage transformer or, in particular, a DC current 
transformer, may be used instead of the above voltage transformer. Also, a 
DC current transformer operable without generating ripples may be used. 
Further, a current transformer having a response to a wide frequency range 
may also be used. With the use of a DC current transformer, the limitation 
on the turn-on holding time is eliminated. 
According to the invention, as has been described with reference to its 
embodiments illustrated in FIGS. 6 to 19 in connection with the switch 
device wherein an auxiliary solid-state switch is connected between the 
collector and the base of a main power transistor whereby the base-drive 
current is supplied to the main power transistor, the voltage or current 
in the main power circuit comprising the main power transistor and the 
other main power transistors is transformed and supplied to the auxiliary 
solid-state switch branches and thus the voltage drop in the main power 
transistor in the ON state is reduced, the loss in the main power 
transistor is diminished, and the base-drive power of the main power 
transistor is self-sufficiently replenished. 
Obviously, numerous modifications and variations of the present invention 
are possible in light of the above teachings. It is therefore to be 
understood that within the scope of the appended claims the invention may 
be practiced otherwise than as specifically described herein.