System and method for avoiding DC bias in a homodyne receiver

A homodyne radar system includes a mixer that outputs a signal having a mixer output frequency that is a frequency difference between two input signals. When the frequency difference is small, low frequency noise may mask the actual signal. A variable phase shifter is added to one of the mixer inputs to change the phase at a predetermined rate of change. The phase shifter shifts the frequency of the input signal so that the mixer output frequency is offset by the predetermined rate. The low frequencies are mapped to frequencies that are above the noise frequencies. The phase shift may be achieved by adding a constant phase at predetermined time intervals. The sampling frequency for the resulting signal may need to be increased to accommodate the higher frequencies.

TECHNICAL FIELD

This application generally relates to avoiding low frequency noise in a radar system.

BACKGROUND

A radar system may be utilized for many purposes in an automotive vehicle. For example, a radar system enables safety features such as collision warning and adaptive cruise control. The effectiveness of such systems may depend upon the performance of the radar system. A radar operates by transmitting an electromagnetic wave that is reflected from an object back to a radar receiver. The reflected signal may be shifted in frequency from the transmitted signal. The radar electronics, using a mixer, may generate a signal that is the difference in frequency between the transmitted and received signals. This frequency difference may then be processed to calculate the range and relative velocity of the object.

A homodyne receiver may down-convert a radar signal directly to a baseband frequency without first converting the signal to any intermediate frequencies. Non-moving (no Doppler content) returns from zero or near-zero range may result in DC or near-DC frequency signals in the baseband. High-level leakage and noise signals and the frequency spread due to oscillator phase noise may mask the DC and low-frequency (near-range) returns.

The noise resulting from the homodyne down-conversion process may be primarily a DC bias noise, sometimes referred to as mixer bias. The bias contributes to noise referred to as 1/f, 1/f2, and 1/f3noise. This noise, in addition to inherent device noise in the case of 1/fncomponents, may be the result of equal, or nearly equal, frequency components reacting in the down-conversion mixer and the phase noise of the signals themselves. Leakage of the local oscillator (LO) and radio-frequency (RF) signals may self-mix to DC. The leakage may also reflect off the mixer ports internally due to imperfect matching.

The transmit-to-receive antenna isolation may be greater than 50 dB and radar return signals may be greater than 50 dB below transmit levels. The leakage in the mixer itself may be approximately 20-25 dB below the oscillator signal levels. In automotive radars, the oscillator signal level to the homodyne mixer may only be 3-5 dB below the transmit signal level. Near range target return levels are limited by the R4roll-off dictated by the radar range equation. In addition, the radar cross section (RCS) of the targets may be physically limited by the illumination of the antenna, which may be a very small spot at near range. Thus, the leakage signals in the mixer may dominate the returns of near range targets with low RCS that have little or no relative motion.

Pedestrians may have a small RCS and low Doppler content. At near range, a return signal from a pedestrian may be masked by the mixer bias noise. In stop-and-go situations, a radar instrumented vehicle may be following a target vehicle to a stop. Issues may arise when the target vehicle exhibits a low RCS (e.g., motorcycles, certain cars). As the vehicle approaches the target vehicle and reduces speed, the target vehicle return may become masked by mixer bias. The result may be reduced performance of functions that rely on the radar system to detect objects.

SUMMARY

A homodyne receiver includes a mixer configured to receive a first input signal based on a transmitted signal and a second input signal based on a received signal and a phase shifter configured to apply a predetermined rate of phase change to one of the first input signal and the second input signal, wherein a mixer output frequency is shifted by the predetermined rate. The phase shifter may be configured to provide the predetermined rate of phase change to the first input signal. The homodyne receiver may include a coupler configured to couple the transmitted signal to the first input signal and the phase shifter may be configured to provide the predetermined rate of phase change to the first input signal. The phase shifter may be configured to provide the predetermined rate of phase change to the second input signal. The homodyne receiver may include a low noise amplifier configured to process the received signal and output the second input signal and the phase shifter may be configured to provide the predetermined rate of phase change to the second input signal. The predetermined rate of phase change may be a predetermined constant value. The homodyne receiver may include at least one controller configured to sample a mixer output amplitude, wherein a sampling frequency for the mixer output amplitude may be greater than a highest expected mixer output frequency.

A method of avoiding DC bias noise in a homodyne receiver includes changing phase of a first input signal to a mixer at a predetermined rate, mixing the first input signal and a second input signal to the mixer, and outputting a signal with a frequency that is shifted by the predetermined rate such that a mixer output amplitude is above a corresponding noise amplitude. Changing the phase of the first input signal may include adding a constant phase to the first input signal at predetermined time intervals. The predetermined rate may be a predetermined constant value. The first input signal may be based on a transmitted signal. The first input signal may be based on a received signal.

A radar system includes an oscillator configured to generate a transmitted signal, an antenna configured to transmit the transmitted signal and receive a received signal, and a homodyne receiver including a mixer configured to mix a first input signal based on the transmitted signal and a second input signal based on the received signal and a phase shifter configured to apply a predetermined rate of phase change to one of the first input signal and the second input signal, wherein a mixer output frequency is shifted by the predetermined rate of phase change. The radar system may include at least one controller configured to sample the mixer output amplitude, wherein a sampling frequency for the mixer output amplitude is greater than a highest expected mixer output frequency. The predetermined rate of phase change may be selected such that a mixer output amplitude is above a corresponding noise amplitude at a base frequency. The predetermined rate of phase change may be a predetermined constant value. The phase shifter may be configured to apply the predetermined rate of phase change to the first input signal by adding a constant phase to the first input signal at predetermined time intervals. The homodyne receiver may be configured to apply the predetermined rate of phase change to the first input signal. The homodyne receiver may be configured to apply the predetermined rate of phase change to the second input signal. The radar system may include a coupler configured to provide the transmitted signal to the antenna and provide a coupled output based on the transmitted signal, and wherein the first input signal may be the coupled output.

DETAILED DESCRIPTION

FIG. 1shows an example block diagram of a homodyne Linear Frequency Modulated (LFM) radar system12in a vehicle10. A homodyne radar system12may be configured to directly down-convert a signal frequency to zero intermediate frequency. A radar system12may generate an electromagnetic wave30having a frequency and amplitude. A Voltage Controlled Oscillator (VCO)14may be configured to output a chirp signal22. The VCO14may include an amplifier to provide a chirp signal22at an appropriate power level for transmission. The chirp signal22may be a signal that varies in frequency over time. The VCO14may generate a chirp signal22that varies linearly with time, such as f(t)=f0+γt. The frequency of the chirp signal22may repeat over a fixed range of frequencies. The frequency of the chirp signal22may also be selected such that the frequency repeatedly increases and decreases linearly within a frequency range. The range of frequencies may be denoted as the bandwidth (BW) and may be equal to the difference between the highest and lowest frequencies swept by the chirp signal22. The VCO frequency output22may be fed to a coupler16. The coupler16may allow a main transmit signal28to pass through to a transmit antenna18. The coupler16may pass a majority of the signal energy to the transmit antenna18. The coupler16may also provide a coupled signal20that is indicative of the frequency and amplitude of the main transmit signal28. The coupled signal20may resemble the main transmit signal28in terms of frequency for use in the radar receiver circuit. The main transmit signal28may be routed to a transmit antenna18.

The transmitted chirp signal28may be converted by the transmit antenna18to an electromagnetic wave30. The transmitted wave30may be of the form
s(t)=Atcos(2π(f0t+γt2))  (1)
where f0is a frequency of the chirp signal22, γ is equal to a chirp rate that may be defined as the ratio of the pulse bandwidth (BW) over the pulse length of the transmitter (Tp).

The transmitted electromagnetic wave30may travel out from the antenna18and may be reflected when striking an object32located a distance R from the radar unit12and traveling at a velocity υ relative to the radar unit12. There may be multiple objects32located at different distances from the vehicle and traveling at different speeds relative to the vehicle. When striking the object32, the electromagnetic wave30may be reflected back to the radar unit12. The reflected wave34may be received by a receive antenna36. Although the transmit antenna18and the receive antenna36are shown as separate antennas, a single antenna may be utilized for both purposes. The reflected waveform34may have different amplitude, frequency and phase than the transmitted wave30. The reflected wave34may be of the form:
s(t+τ)=Arcos {2π[f0(t+τ)+γ(t+τ)2+2υ/cf0(t+τ)]}  (2)
where c is the speed of light, and τ is the propagation delay of the signal and is equal to 2R/c.

The reflected wave34may be converted to an electrical signal38by the receive antenna36. The received signal38may be passed through a low noise amplifier (LNA)40to increase the amplitude of the received signal. The amplifier40may provide an amplified receive signal42. The amplified received signal42may be routed through a received signal power divider44to split the received signal for use in different parts of the circuit.

The coupled signal20may be passed to an input of a variable phase shifter78. The phase shifter78may be configured to apply a periodic phase shift to the transmit-based input signal20. The output signal86of the phase shifter78may be routed through a reference power divider46to split the reference frequency for different parts of the circuit. A first output48of the reference power divider46may be routed to an input of a mixer50. One output52of the received signal power divider44may be routed to a second input of the mixer50. The receive-based signal52may have a different frequency than the reference signal48due to the speed and distance of the object32from the radar system12. The mixer50may remove the frequency of the reference signal48from the received signal52. In the frequency domain, the frequency of the reference signal48may be subtracted from the frequency of the received signal52. An output54of the mixer50may be routed through a filter56. The filter56may be a low pass filter to remove higher frequency components from the mixer output54. A final in-phase output signal58may have a frequency based on the speed and range of the object32that reflected the wave.

In some applications, a second output60of the reference power divider46may be routed through a constant phase shifter62that shifts the phase of the transmit-based signal60by a constant 90 degrees. The constant phase shifted reference signal64may be routed to an input of a second mixer66. An output68of the received signal power divider44may be routed to an input of the second mixer66. The second mixer66may remove the frequency of the phase-shifted reference signal64from the received signal68. A second mixer output70may be routed through a filter72. The filter72may be low pass filter. This configuration generates a quadrature data output74that is 90 degrees out of phase with the in-phase output signal58.

The in-phase signal58and quadrature signal74may be routed to a controller76. The controller76may include one or more analog to digital converters to sample the signals. The sampled signals may be processed to ascertain the amplitude and frequency content of the signals. The processed signals may be used to calculate the range and relative velocity of the object32. The range and relative velocity data may be used to perform collision warning (CW) functions that may warn the driver when a collision with an object is possible. Additionally, the CW function may command the brake system to apply brake pressure to slow the vehicle to avoid a collision. The processed signals may also be used for adaptive cruise control (ACC) functions that control the vehicle speed according to the distance between the object and the relative speed of the object. The ACC function may control the propulsion torque and braking system to maintain a desired distance and/or speed. For example, when ACC is active, the system may first attempt to control the vehicle speed to a desired set speed. When a slower moving object is detected in front of the vehicle the system may control the vehicle speed so as to maintain a set distance between the object and the vehicle. As the object in front slows, the vehicle may be slowed to maintain the separation distance. If the object stops, the vehicle may be stopped as well.

Prior art homodyne radar systems do not include the variable phase shifter78. Prior art homodyne systems may connect the coupled transmit signal20directly to the input of the reference power divider46.

FIG. 2depicts a simplified example of a homodyne radar system in which a phase shifter78is introduced into the coupled transmit-based signal path20in the homodyne architecture.FIG. 2depicts only the in-phase portion of the homodyne radar system but the following discussion may apply to the quadrature portion as well. The phase shifter78may also be placed in the transmit antenna path after coupling off the de-chirp signal but this placement may cause intolerable attenuation to the transmit signal. The phase shifter78may be configured to apply a periodic phase shift to the transmit-based input signal20.

The periodic phase shift may be selected to be linear with the phase shifting being imparted at a given sample rate. The sample rate may correspond to an associated A/D converter sampling frequency. Frequency is related to the rate of change or time derivative of phase (dφ/dt). A constant frequency offset may be introduced into a signal by changing the phase of the signal at a constant rate (e.g., dφ/dt=K, where K is constant). This may be accomplished by periodically applying a constant phase adjustment to the signal. The frequency offset may be calculated as Δφ*f/(2π) where Δφ is the constant phase adjustment in radians and f is the frequency at which the phase is adjusted. As an example, the constant phase addition may be selected to be nπ/2. Note that any value may be chosen for the constant phase addition and the analysis will be similar. The constant rate may be selected as a sampling rate or frequency, fs. The frequency offset may then be calculated as nfs/4.

The frequency of the input signal20may be shifted by configuring the phase shifter78to provide a predetermined rate of phase change to the input signal20. The frequency offset is related to the rate of phase change. Therefore, by adding a constant phase change at predetermined time intervals (e.g., a sampling frequency of the unshifted mixer output signal), a constant frequency offset may be achieved. The frequency offset may be selected such that the amplitude of the mixer output signal54is greater than a corresponding noise amplitude at a given frequency.

For example, by selecting a periodic phase shift to be n/2 (i.e., at each sample time, n/2 radians are added to the signal path), the frequency offset imparted may be one-fourth of the sample frequency, fs. Other values of differential phase shift may be utilized resulting in different offset frequencies. In addition, the rate at which the phase is added may be adjusted to other values. The resulting frequency shift in the baseband signal may require an increase in an A/D converter sampling frequency of the controller76to ensure that the high frequency end of the band may be adequately sampled.

In this example, after the transmit signal20is passed through the variable phase shifter78, the frequency of the output signal86may be f+fs/4, where f is the oscillator frequency. In the absence of the variable phase shifter, the frequency of signal86would be the oscillator signal22frequency, f. Using the variable phase shifter, the local oscillator signal20may be shifted in frequency by a frequency of fs/4. The frequency of the mixer output54may then be calculated as fs/4+Δf, where Δf is the frequency difference between the transmitted signal and the reflected signal. A controller76may be configured to sample the output54of the mixer50. The controller76may also be configured to adjust the phase of the phase shifter. The controller76may provide an output82that may be the phase adjustment for the phase shifter78. The output82may be updated at a particular frequency to provide a constant rate of phase change. The phase shifter78may be analog or digital. The controller76may be configured to provide an output82that is a variable voltage to control an analog phase shifter. The controller76may be configured to output82a digital signal or signals to control a digital phase shifter.

The radar return signals may be replicas of the transmitted signal in frequency space with receive-time delay and amplitude differences. The homodyne de-chirping process provides a difference between the transmit signal frequency and the return signal frequency in frequency space. The delay of the return signal encodes the range to a target into a constant frequency. The band of interest is dependent on the chirp rate, the maximum range of interest, and the maximum Doppler frequency expected. Non-moving (no Doppler content) returns from zero or near-zero range may result in DC or near-DC frequency signals in the baseband (i.e., zero intermediate frequency). High-level leakage and noise signals and frequency spread due to oscillator phase noise may mask the DC and low-frequency (near-range) returns.

FIG. 4provides an example of a near-range and a far-range frequency return signal for homodyne radar system. The frequencies of the transmitted and received signals may be plotted as a function of time. The frequency output of the transmitter may vary linearly with time. The transmitted frequency may be depicted over time as a line200starting from a base value, F0210, and rising linearly. A return signal for a near-range object202may resemble the frequency characteristic of the original signal200but may have a frequency offset of Δfn206from the transmitted signal200at a given time. A return signal for a far-range object204may resemble the frequency characteristic of the transmitted signal200but have a frequency offset of Δff208from the transmitted signal200at a given time. Notice that the near-202and far-range 204 return signal frequencies may vary linearly with time in the same manner as the transmitted signal200frequency.

The output of the variable phase shifter (78FIG. 1) may be represented by a shifted base frequency curve212. The frequency may be offset by a constant frequency fs/4214from the transmitted signal200. A near-range signal202that is offset by Δfn206from the transmitted signal200may result in a phase shifter output frequency profile216that is offset from the shifted base frequency profile212by Δfn206at a given time. When down converted, the final output frequency may be Δfn′218which may be the sum of the offset frequency fs/4214and Δfn206. A far-range signal204that is offset by Δff208from the transmitted signal200may result in a frequency profile222that is offset from the shifted base frequency profile212by Δff208at a given time. After down conversion, the final output frequency may be Δff′220which may be the sum of the offset frequency fs/4214and Δff208. Note that without the variable phase shifter, the down converted frequency is Δfn206for the near-range frequency and Δff208for the far-range frequency.

FIG. 5depicts the amplitude of the frequency components. The amplitudes of the frequency components of the converted return signals may be plotted as a function of frequency. A DC noise amplitude216may also be plotted as a function of frequency. The DC noise amplitude216may be present over a relatively low range of frequencies. An amplitude profile of a prior art homodyne receiver232is depicted. In addition, an amplitude profile of the homodyne receiver with variable phase shifters226is shown.

For the prior art amplitude profile232, it is readily observed that at a near-range frequency Δfn206, the amplitude of the received signal234may be less than an amplitude of the noise236. In the range in which the DC noise bias amplitude216is greater than the return signal amplitude232, target returns may not be distinguishable from the noise. In this low range of frequencies, it may not be possible to distinguish between actual return signals and noise. Hence, the performance of the radar system at detecting near-range objects may be limited. At higher frequencies that are above the DC noise bias amplitude, such as Δff208, the amplitude may not be impacted by the low frequency noise.

The amplitude profile of the improved homodyne receiver226resolves this issue by shifting the amplitude profile of the return signals without shifting the DC characteristics. The amplitude226of the output frequency components of the phase shifted return signals may be plotted as a function of frequency. The phase-shifted output for a near-range signal may be offset to a frequency of Δfn′218which may be the sum of the offset frequency fs/4214and Δfn206. The magnitude of the phase-shifted output signal224at the offset frequency Δfn′218may be greater than the amplitude of the DC noise228at the same frequency. The far-range frequency230may be likewise shifted and may remain above the DC noise at the frequency Δff′220. Zero range returns may now be mapped to fs/4214in the frequency space.

FIG. 3depicts an alternative configuration in which a variable phase shifter90is alternatively placed in the receive antenna path38. The input to the phase shifter90may be an output of a low noise amplifier40. In the absence of a low noise amplifier40, the received signal38from the antenna may be input into the phase shifter90. Placing the phase shifter90after the low noise amplifier (LNA)40may negatively impact the system noise figure. A similar analysis may be performed as above with similar resulting waveforms.

The addition of the variable phase shifter to the homodyne receiver system provides a benefit in that the amplitudes of the signals may now be above the amplitude of low-frequency noise. This may permit better detection of near-range objects that as the shifted return signals may be distinguishable from the noise. The addition of the variable phase shifter allows the homodyne architecture to be maintained without having to resort to more complex and expensive heterodyne receiver designs.

Since the expected frequency range has been shifted, it may be desired to modify the sampling rate of the resulting signal. As an example, assume that the original sampling frequency is 5/4 times the highest expected frequency in the original homodyne architecture. The highest expected frequency in the original architecture may be Δffand a sampling frequency 5Δff/4 may be chosen in order to adequately sample the complex returns. Assume that a sequential phase shift of nπ/2 will be added to the transmit-based signal using the phase shifter. As discussed, the frequencies will be shifted by a predetermined amount. The sampling frequency of the new signal may need to be increased in order to accommodate the increased frequencies.

A new sampling frequency, fsn, for the frequency shifted baseband signals of 1.5 times the original sampling frequency, fs, may be selected. This choice moves the frequency of the nearest range target signal up by 0.375 times (Δfn+fsn/4=Δfn+3fs/8) the original sampling frequency. Assuming that the nearest range signal is at 0 Hz, then the lowest frequency is moved to 0.375 times the original sampling frequency. The effect may be to raise the lowest frequency of interest out of the DC noise range.

The highest frequency of interest in the baseband target signal spectrum is also shifted up by 0.375 times the original sampling frequency. The highest frequency maps to 47/60 times the new sampling frequency, fsn. The new sampling frequency, fsn, is more than 1.25 times the highest frequency in the complex spectrum of interest so the spectrum may be adequately sampled.

Referring again toFIG. 2, the phase may be adjusted at the new sampling frequency, fsn. The controller76may sample the mixer output signals54at the new sampling frequency. In addition, the controller76may apply the phase adjustment82to the phase shifter78at the same frequency. Similarly, inFIG. 3, the controller76may be configured to read the mixer output94and provide a signal96to the phase shifter90to adjust the phase.

The configuration described introduces a phase shifter78configured to apply a rate of phase change to the signal. The rate of phase change creates a frequency offset to the signal. The quadrature leg ofFIG. 1also depicts an additional phase shifter92. This phase shifter62is present to provide a quadrature output that is shifted by ninety degrees of phase from the non-shifted signal. The result of adding a fixed phase to a signal is a signal of the same frequency that is delayed or advanced in phase from the original. The phase shifter62is configured to apply a fixed phase offset to the signal and does not shift the frequency of the signal. The phase shifter78is configured to apply a rate of phase change that effectively changes the frequency of the signal.

FIG. 6illustrates a flowchart for the operation of the homodyne radar system. The system may begin operation300based on a system power-up or ignition on. The radar may generate the transmit signal302. After the signal is transmitted, a reflected signal may be received. The radar system may receive and filter the reflected signal304. The filtering may include amplification of the received signal. A predetermined rate of phase change may be applied to the transmit-based signal306. The effect of applying a rate of phase change may be to shift the frequencies of the mixed signals above DC noise. The system may then down convert the signals to baseband frequencies308. A controller may then process the down converted signal310to calculate the range and velocities of objects in the radar path. The system may repeat or when shut-down conditions are satisfied, operation may terminate312. The above sequence may be implemented via a combination of hardware circuitry and a microprocessor-based controller.