Body composition analyzer

An analyzer determines the amount of body fat on a patient by using a measurement of body impedence is based upon the nature of the conduction of an applied electrical current in the human body. The analyzer has a constant current source circuit for inducing a high frequency low-voltage signal in the body, and magnitude and phase detection circuits for measuring the magnitude and phase shift of the induced signal.

BACKGROUND OF THE INVENTION 
1. Field of the Invention 
This invention relates to electronic devices for the measurement of body 
fat, and more particularly to those devices which measure body fat by 
applying a low-level constant RMS value alternating current to the body. 
2. Description of the Prior Art 
Measurement of body composition is assuming greater importance in assessing 
nutritional status in both health and disease. Direct measurement of body 
composition has been limited to research centers using hydrostatic 
weighing or isotope-dilution techniques. An indirect measurement technique 
has been developed based upon a determination of body electrical 
impedance. 
The method for determining body impedance is based upon the nature of the 
conduction of an applied electrical current in the human body. In 
biological structures, application of a constant, low-level alternating 
current produces an impedance to the spread of the current that is 
frequency dependent. The human body contains intra- and extracellular 
fluids that act as electrical conductors and cell membranes that act as 
imperfect reactive elements. At low frequencies (around 1 kHz), the 
current mainly passes through the extracellular fluids, while at higher 
frequencies (500-800 kHz) it penetrates the intra- and extracellular 
fluids. Thus, body fluids and electrolytes are responsible for electrical 
conductance (the inverse of resistance) and cell membranes are involved in 
capacitance. 
The use of bioelectrical impedance measurements to determine fat-free mass 
is based upon the principle that the impedance of a geometrical system is 
related to conductor length and configuration, the impedance to the flow 
of current can be related to the flow of current as 
##EQU1## 
where Z is impedance in ohms, p is volume resistivity in ohm-cm, L is 
conductor length in cm, and A is conductor cross-sectional area in 
cm.sup.2. Multiplying both sides of the equation by L/L gives: 
##EQU2## 
where AL equals the volume V. Rearranging this question yields, 
##EQU3## 
In the human body, electrical conduction is related to the water and 
electrolyte distribution in ,.the biological conductor. Because fat-free 
mass contains virtually all the water and conducting electrolytes in the 
body, conductivity is far greater in the fat-free mass that in the fat 
mass of the body. The electrically determined biological volume V is 
inversely related to Z, and thus it is also inversely rated to resistance 
R and reactance Xc, since 
##EQU4## 
Because the magnitude of reactance is small relative to resistance, and 
resistance is a better predictor of impedance than is reactance, volume 
can be expressed as 
##EQU5## 
where L is standing height in cm and R is resistance in ohms. Although 
there are difficulties in applying this general principle in a system with 
as complex geometry and bioelectrical characteristics as the human body, 
this relationship has been used to derive models for the prediction of 
human body composition by assuming that the body is a series of connected 
cylinders. 
Determinations of resistance and reactance have been made using four 
terminal impedance plethysmographs. Examples of such plethysmographs 
include the model 101, manufactured by RJL Systems of Detroit, Mich. The 
four terminal method has been used to minimize contact impedance or 
skin-electrode interactions. As a general procedure, measurements were 
made about two hours after eating and within 30 minutes of voiding. The 
patient, clothed but without shoes or socks, lied supine on a cot. 
Aluminum foil spot electrodes were positioned in the middle of the dorsal 
surfaces of the hands and feet proximal to the metacarpal-phalangeal and 
metatarsal-phalangeal joints, respectively, and also medially between the 
distal prominences of the radius and the ulna and between the medial and 
lateral malleoli at the ankle. A thin layer of electrolyte gel was applied 
to each electrode before application to the skin. An excitation current of 
800 .mu.A at 50 kHz was introduced into the patient at the distal 
electrodes of the hand and foot and the voltage drop was detected by the 
proximal electrodes. 
According to Ohm's Law the electrical impedance Z to alternating current of 
a circuit was measured in terms of voltage E and current I as 
##EQU6## 
EQU Z=E/I. 
By using phase sensitive electronics, one could quantify the geometrical 
components of Z. Resistance R is the sum of in-phase vectors, and 
reactance Xc is the sum of out-of-phase vectors. A phase discriminator was 
used to measure the phase angle to produce resistance and reactance 
measurements. 
This technique provided a deep homogenous electrical field in the variable 
conductor of the body. Determinations of resistance and reactance were 
made using electrodes placed on the ipsilateral and contralateral sides of 
the body. The lowest resistance value for an individual was used to 
calculate conductance (h.sup.2 /R) and to predict fat-free mass. The 
accuracy of this method has been found to be within 2%. Statistical models 
have been developed to estimate total body water and fat-free mass in 
adults. 
To provide an accurate measure of resistance and reactance, it is important 
to provide a constant current source. Existing plethysmoqraphs have been 
able to provide a constant current source only by using expensive 
circuitry. Plethysmographs using less expensive circuitry have been unable 
to supply a constant current source with the associated accuracy to 
provide accurate resistance and reactance measurements. 
In addition, it is apparent that the measurement device must be properly 
connected to the body of the patient. If any of the four electrodes is 
improperly located on the patient's body or is improperly connected to the 
measurement device, the resulting measurements will be erroneous. 
SUMMARY OF THE INVENTION 
The present invention provides a body composition analyzer which overcomes 
the problems of the prior art and provides unique advantages which have 
not been available before. The body composition analyzer of the present 
invention includes a constant current source which provides an accurate 
low-level alternating current signal. 
The present invention provides a constant current source circuit comprising 
a modified Wien bridge oscillator. The signal from the oscillator is fed 
through an instrumentation amplifier and a full wave rectifier to an error 
amplifier which provides an error signal to the oscillator to maintain the 
oscillator in phase at constant current. 
In the constant current source circuit of the present invention, the error 
of the current change is very low, varying by less than 0.5% between 0 
K.OMEGA. and 1 K.OMEGA. output resistance. This is four times better than 
the prior art, which typically had a current source error of 2%. 
The present invention also provides a magnitude detection circuit and a 
phase detection circuit to measure the magnitude and phase lag of the 
signal induced in the patient by the constant current source. The 
magnitude detection circuit includes an instrumentation amplifier and a 
full wave rectifier which are essentially identical to those used in the 
constant current source network to reduce the effect upon the phase 
detection circuit. 
In the phase detection circuit of the body composition analyzer of the 
present invention, the circuit compares a sine wave in phase with the 
constant current signal with a sine wave in phase with the induced 
voltage. Since both of these signals are offset by the phase lag caused by 
going through the non-ideal differential amplifiers (which is a function 
of the operational amplifier processing and the external gain and 
frequency compensation), the differential amplifiers of both circuits are 
matched so that the differential phase offset between the two circuits 
will be at a minimum and so that errors will be minimized. 
The body composition analyzer of the present invention is provided with 
receptacles for the electrodes that are connected to the patient. These 
receptacles are arranged within the image of a patient's hand and foot, so 
that the user can quickly and easily determine the proper receptacle for 
the corresponding electrode. This design reduces the possibility that one 
of the electrodes will be incorrectly connected to the analyzer, resulting 
in erroneous readings. 
These and other advantages are provided by the body composition analyzer of 
the present invention. The body composition analyzer comprises four 
electrodes for connection to a patient. A constant current source circuit 
is connected to one of the electrodes. The constant current source circuit 
comprises an oscillator providing an output to the electrode, an 
instrumentation amplifier connected to the output of the oscillator, and a 
full wave rectifier connected to the output of the instrumentation 
amplifier and supplying an error signal to the oscillator. A magnitude 
detection circuit is connected to two of the electrodes for measuring the 
current from the electrodes. A phase detection circuit is connected to 
receive a signal from the constant current source circuit and a signal 
from the magnitude detection circuit for measuring the phase shift between 
the two signals. Microcomputer means are connected to the phase detection 
circuit and the magnitude detection circuit for measuring resistance and 
reactance of the patient and converting measurements to an indication of 
the amount of body fat. 
The body composition analyzer is contained in a housing which has images of 
a hand and a foot at the point at which the electrode is connected to the 
housing in order to assist in properly placing each of the electrodes at 
the appropriate location on the patient. The housing may also have 
calibration bars for connection of the ends of the electrodes, and a fixed 
resistance element connecting the two calibration bars, to permit the 
analyzer to be calibrated through the electrodes.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
Referring more particularly to the drawings, and initially to FIG. 1, there 
is shown the body composition analyzer 20 of the present invention. The 
body composition analyzer functions by measuring the resistance and 
reactance of the human body. Electrodes are placed on the distal and 
proximal hand and foot. A known constant current, 500 .mu.A at 50 kHz, is 
applied by an electrode connected to the distal foot, and goes to ground 
from an electrode connected to the distal hand. The induced voltage is 
measured, using high input impedance electrodes connected to the proximal 
hand and the proximal foot. Height, weight, and age are input, using a 
keypad, and the body resistance is measured by the circuitry. These 
variables are used in accordance with known prediction techniques to 
predict percent body fat. 
As shown in FIG. 1, the body composition analyzer 20 of the present 
invention comprises a housing 21, which includes an on/off power switch 
22, a display screen 23, and an input keypad 24. The keypad 24 includes 
numeric keys for inputting numeric information, such as height, weight and 
age, and four function keys for designating the type of numeric 
information input and controlling the operation of the analyzer. The 
electrodes are connected at four receptacles 26, 27, 28, and 29. The 
receptacle 26 is for the electrode attached to the distal hand. The 
receptacle 27 is for the electrode attached to the proximal hand. The 
receptacle 28 is for the electrode attached to the distal foot. The 
receptacle 29 is for the electrode attached to the proximal foot. 
The receptacles 26-29 are surrounded by graphic indications of the hand and 
foot of the patient. Specifically, the receptacles 26 and 27 are located 
on the image 31 of a hand to show the location of connection of the 
electrodes, and the receptacles 28 and 29 are similarly located on the 
image 32 of a foot. The hand and foot images 31 and 32 enable the operator 
to quickly identify the receptacle for each electrode by matching the 
graphic indication around of the receptacle to the location that the 
electrode is connected on the patient. This assures that the body 
composition analyzer will be properly connected to the patient and that 
the measurements will be accurately received. 
The analyzer is also provided with two calibration bars 34 and 35 which are 
used to verify that the unit is in proper calibration. One calibration bar 
34 is provided for the hand electrodes, and the other calibration bar 35 
is provided for the foot electrodes. An appropriate resistance element, 
e.g., 511 ohms, is provided between the calibration bar 34 and the 
calibration bar 35. By connecting the electrodes to the calibration bars 
34 and 35, and setting the analyzer in its calibration mode, the analyzer 
provides the user with an accuracy indication. 
The internal circuitry of the body composition analyzer 20 of the present 
invention is shown in schematic form in FIG. 2. The analyzer 20 includes 
four electrodes 41, 42, 43, and 44 for connection to the body of the 
patient. The electrode 41 is used for attachment to the distal foot, and 
provides a constant current input to the patient's body. The electrode 44 
is attached to the distal hand, and is connected by means of the 
receptacle 26 to ground. The electrodes 42 and 43 are connected to the 
proximal foot and hand, respectively, and provide the inputs to the body 
composition analyzer for the measurement of resistance and reactance of 
the body. 
As shown in FIG. 2, the distal foot electrode 41 is connected by means of 
the receptacle 28 to a constant current source circuit 46. The constant 
current source circuit 46 comprises an oscillator 47 providing output to a 
differential amplifier or instrumentation amplifier 48. The output of the 
instrumentation amplifier 48 is fed back to the oscillator 47 through a 
full wave rectifier 49 and an error amplifier 50. 
The details of the constant current source circuit 46 can be seen in FIG. 
3. The oscillator 47 is a modified Wien bridge oscillator comprising an 
N-channel JFET 52 used as a voltage-controlled resistor. A 20 K.OMEGA. 
resistor 53 is connected around the JFET 52. Wien bridge oscillators are 
well known and widely used and take advantage of the fact that the phase 
of the voltage across the parallel branch of a series and a parallel RC 
network connected in series is the same as the phase of the applied 
voltage across the two networks at one particular frequency and that the 
phase lags with increasing frequency and leads with decreasing frequency. 
A 3.16 K.OMEGA. resistor 54 and a 1000 pF capacitor 55 provide the series 
RC network, and a 2.74 K.OMEGA. fixed resistor 56 and 1 K.OMEGA. 
calibration resistor 57 and a 1000 pF capacitor 58 provide the parallel RC 
network. An operational amplifier 59 is connected in parallel with the 
resistor 54 and the capacitor 55. A 15 K.OMEGA. gain resistor 60 is 
provided between the output and the inverting input of the operational 
amplifier 59. 
The JFET 52 provides the inverting input to the operational amplifier 59. 
Between the JFET 52 and the operational amplifier 59 is a 1 .mu.F 
capacitor 62 which is a low frequency roll-off capacitor preventing offset 
voltage and offset current errors from being multiplied by the amplifier 
gain. The output of the operational amplifier 59 is fed on a line 63 to 
the instrumentation amplifier 48. 
As shown in FIG. 3, the oscillating output on the line 63 causes a current 
flow across a 412 .OMEGA. resistor 65. The differential amplifier or 
instrumentation amplifier 48 actually comprises three operational 
amplifiers 66, 67, and 68. A 1.65 K.OMEGA. gain resistor 69 connects the 
inverting inputs of the amplifiers 66 and 67 while the outputs of the 
amplifiers 66 and 67 are each connected to the inverting input through 
matched resistors 70. The outputs of the amplifiers 66 and 67 are 
connected to the inputs of the amplifier 68 through matched resistors 71, 
and the inputs of the amplifier 68 are connected to the amplifier output 
(for the inverting input) or to ground (for the noninverting input) 
through matched resistors 72. The output of the operational amplifier 68 
provides the output of the instrumentation amplifier 48 and can be 
described by the relationship: 
##EQU7## 
where E.sub.OUT is the output of the amplifier 68, E.sub.IN.sup.- is the 
input to the amplifier 66, E.sub.IN .sup.+ is the input to the amplifier 
67, R.sub.G is the value of the gain resistor 69, R.sub.1 is the value of 
the resistor 70, R.sub.2 is the value of the resistor 71, and R.sub.3 is 
the value of the resistor 72. 
The gain of the instrumentation amplifier 48 is determined by the value of 
the resistor 69, and the value of the resistor 69 does not affect the 
common mode error signal. Thus, common mode rejection theoretically 
increases in direct proportion to gain. Furthermore, common-mode signals 
are only amplified by a factor of one regardless of gain because no 
common-mode voltage will appear across the resistor 69, hence no 
common-mode current will flow in it. This means that large common-mode 
signals can be handled independent of gain. Because of the symmetry of the 
instrumentation amplifier 48, first order common-mode error sources in the 
operational amplifiers 66 and 67 tend to be cancelled out by the 
operational amplifier 68. 
The output of the instrumentation amplifier 48 is provided on a line 74 
which is connected to a 0.1 .mu.F capacitor 75. The capacitor 75 
eliminates direct-current offsets. This output is then provided to the 
full wave rectifier 49. 
As shown in FIG. 3, the full wave rectifier 49 comprises the input signal 
applied through an input resistor 77 to the summing node of an inverting 
operational amplifier 78. The output of the operational amplifier 78 is 
connected back to the same input through a diode 79. The output is also 
provided through another diode 80. When the input signal is positive, the 
diode 80 is forward-biased and develops an output signal across a resistor 
81. As with any inverting amplifier, the gain is equal to the ratio of 
resistances of the resistor 81 divided by the input resistor 77. When the 
signal goes negative, the diode 80 is nonconducting and there is no 
output. However, a negative feedback path is provided by the diode 79. The 
path through the diode 79 reduces the positive output swing and prevents 
the amplifier 78 from saturating. The operational amplifier 78 and the 
diodes 79 and 80 together provide a half-wave rectifier, the output of 
which is then provided through a resistor 82 to an inverting amplifier 83. 
The amplifier 83 sums the half-wave rectified signal from the resistor 82 
and the input signal provided through a resistor 84 to provide a full wave 
output. A gain resistor 85 connects the output of the amplifier 83 to the 
inverting input. 
For negative input signals, the output of the amplifier 78 is zero, no 
current flows through the resistor 82, and the output of the amplifier 83 
is 
##EQU8## 
where E.sub.OUT is the output of the amplifier 83, E.sub.IN is the input 
to the resistor 77, R.sub.4 is the value of the resistor 85, and R.sub.5 
is the value of the resistor 84. For positive input signals, the amplifier 
83 sums the current through the resistors 82 and 84 and 
##EQU9## 
where E.sub.OUT is the output of the amplifier 83, E.sub.IN is the input 
to the resistor 77, R.sub.4 is the value of the resistor 85, R.sub.5 is 
the value of the resistor 84, and R.sub.6 is the value of the resistor 82. 
The output of the full wave rectifier 49 feeds the error amplifier 50 
through a 10 K.OMEGA. resistor 87. The error amplifier 50 comprises an 
operational amplifier 88 receiving an inverting input from the full wave 
rectifier 49 and receiving a noninverting input from a band gap reference 
voltage source extracted between a 100 K.OMEGA. resistor 89 and a Zener 
diode 90, providing a 1.235 volt band gap reference voltage. A 0.1 .mu.F 
capacitor 91 in parallel with the amplifier 88 provides filtering. The 
operational amplifier 88 compares the band gap reference voltage to that 
provided from the full wave rectifier 49. The output of the amplifier 88 
is provided through a 20 K.OMEGA. resistor 92 to the JFET 52 and is used 
to adjust the channel voltage of the JFET 52 and adjust the oscillator 47 
to eliminate the error. 
The operational amplifiers 59, 66, 67, 68, 78, 83 and 88 are preferably 
formed of OP-471 low-noise operational amplifiers manufactured by 
Precision Monolithics Inc. of Santa Clara, Calif. 
To explain the operation of the constant current source circuit 46, at 
power up, the noninverting input of the amplifier 88 quickly goes to 1.235 
volts because the diode 90 provides a 1.235 volt band gap reference. 
Meanwhile, the inverting input stays near ground. This causes the output 
of the amplifier 88 of the error amplifier 50 to go to the positive rail. 
A positive signal on the gate of the JFET 52 causes the channel resistance 
to lower, thus lowering the voltage at the inverting input of the 
operational amplifier 59. This lowering of the negative feedback causes 
the oscillator 47 to output a rail-to-rail square wave on the line 63. The 
square wave results in a current flow across the resistor 65. The voltage 
across this resistor 65 is amplified by the instrumentation amplifier 48, 
rectified by the full wave rectifier 49, and provides an input to the 
error amplifier 50. The error amplifier 50 compares the band gap reference 
voltage to that received from the full wave rectifier 49 and adjusts the 
channel voltage of the JFET 52 to eliminate the error. As the error 
decreases, the output of the oscillator 47 becomes a sine wave. To 
maintain control in all situations, the JFET 52 operates at a negative 
gate voltage of -1 volts, thus allowing swings in either direction. 
The constant current source circuit 46 provides a constant current output 
through a 3.16 K.OMEGA. resistor 94 and a 0.1 .mu.F capacitor 95 to the 
distal foot electrode 41. 
An advantage of the circuit is that it is immune to the 60 Hz pick-up. In 
some prior art circuits, where the patient is in the feedback loop of an 
operational amplifier, the patient lead would feed directly into the input 
of the operational amplifier, and this would cause the pick-up of the 60 
Hz signal. In the present invention, the patient is not in the feedback 
loop, so that a constant current source is maintained without the 60 Hz 
signal being picked up. 
Among the other advantages of the constant current source circuit of the 
present invention is that the error of the current change is very low, 
varying by less than 0.5% between 0 K.OMEGA. and 1 K.OMEGA. output 
resistance. This is four times better than the prior art, which typically 
had a current source error of 2%. The error of the constant current source 
is very low because of the high differential gain of the amplifier 88 of 
the error amplifier 50. The OP-471 component used as the amplifier 88 has 
a typical DC gain of 500,000 V/V. 
Referring again to FIG. 2, the electrodes 42 and 43 which receive input 
from the proximal foot and proximal hand, respectively, are connected to a 
magnitude detection circuit 101 by means of the receptacles 27 and 28. The 
magnitude detection circuit 101 comprises a differential amplifier or 
instrumentation amplifier 102 and a full wave rectifier 103. 
As shown in more detail in FIG. 4, the instrumentation amplifier 102 is 
essentially the same as the instrumentation amplifier 48, comprising three 
operational amplifiers 105, 106, and 107. The pickup from the electrodes 
42 and 43 is fed to the noninverting inputs of the amplifiers 105 and 106, 
respectively, after passing through 0.1 .mu.F capacitors 108 which are 
provided to filter direct current signals. A pair of 1 M.OMEGA. resistors 
109 are provided to keep the noninverting input to the amplifiers 105 and 
106 from floating due to input bias current. A 1.65 KO gain resistor 110 
connects the noninverting inputs of the amplifiers 105 and 106 and is 
essentially the same as the gain resistor 69 of the instrumentation 
amplifier 48. Similarly, matched resistors 111, 112 and 113 perform the 
same function as the matched resistors 70, 71 and 72 of the 
instrumentation amplifier 48. 
The instrumentation amplifier 102 takes the voltage induced in the patient 
by the electrode 41 from the constant current source circuit 46 and picked 
up by the electrodes 42 and 43 and amplifies it. This amplified voltage is 
fed on line 115 through a 0.1 .mu.F capacitor 116 to the full wave 
rectifier 103. 
The full wave rectifier 103 is essentially the same as the full wave 
rectifier 49 and comprises an input resistor 118, an operational amplifier 
119, diodes 120 and 121, and a resistor 122. Together, these elements form 
the half-wave rectifier which feeds an amplifier 123. The amplifier 123 
receives the half wave signal through from the amplifier 119 through a 
resistor 124 and receives the input signal through a resistor 125. A 0.1 
.mu.F capacitor 126 is placed across the resistor 127 and rolls off the 
frequency response of the amplifier 123 to give an output equal to the 
average value of the input. This provides filtering or averaging to obtain 
a pure DC output. The filter time constant is RC, using the resistor 127 
and the capacitor 126, and must be much greater than the maximum period of 
the input signal. 
The filtered output of the full wave rectifier 103 of the magnitude 
detection circuit 101 is provided to an analog-to-digital (A/D) converter 
129, which then provides a digital indication of the measured voltage to a 
microcomputer 130. The microcomputer 130 also receives input from the 
keypad and provides output through the display 23. 
As shown in FIG. 2, the output of the instrumentation amplifier 48 of the 
constant current source circuit 46 and the output of the instrumentation 
amplifier 102 of the magnitude detection circuit 101 both feed a phase 
detection circuit 132 which measures the phase angle between the input 
constant current signal and the induced signal to produce resistance and 
reactance measurements. As shown in more detail in FIG. 5, the phase 
detection circuit 132 comprises a pair of comparators 133 and 134 which 
square up the sine waves received from the instrumentation amplifiers 48 
and 102. The instrumentation amplifier 48 outputs a sine wave in phase 
with the constant current signal, while the instrumentation amplifier 102 
outputs a sine wave in phase with the induced voltage. However, both are 
offset by the phase lag caused by going through the non-ideal differential 
amplifiers (which is a function of the operational amplifier processing 
and the external gain and frequency compensation). To minimize these 
errors, the differential amplifiers of both sections are matched so that 
the differential phase offset between the two sections will be at a 
minimum. As with the operational amplifiers 59, 66, 67, 68, 78, 83 and 88 
of the constant current source circuit 46, the operational amplifiers 105, 
106, 107, 119 and 123 of the magnitude detection circuit 101 are also 
preferably formed of OP-471 low-noise operational amplifiers manufactured 
by Precision Monolithics Inc. of Santa Clara, Calif. 
The comparators 133 and 134 feed an exclusive-OR gate 136 through 10 
K.OMEGA. resistors 135. The 10 K.OMEGA. resistors 137 and diodes 138 are 
provided for filtering. The output of the exclusive-OR gate is RC-filtered 
(by means of a 100 K.OMEGA. resistor 139 and a 0.1 .mu.F capacitor 140) 
and buffered by an operational amplifier 141. The exclusive-OR gate 136 
operates on a five-volt power supply, so that a 90-degree phase shift 
causes a 2.5 volts output to the A/D converter 129. 
The microcomputer 130 receives a measurement of the induced voltage through 
the magnitude detection circuit 101 as converted by the A/D converter 129 
and receives a measurement of the phase shift from the phase detection 
circuit 132 as converted by the A/D converter 129. Height, weight, and age 
are input, using the external keypad 24. The microcomputer 130 then 
calculates the percent of body fat, using the resistance and phase shift 
information and the input data in accordance with known calculations. The 
results are displayed through the output display device 23. 
While the invention has been shown and described with respect to a 
particular embodiment thereof, this is for the purpose of illustration 
rather than limitation, and other variations and modifications of the 
specific embodiment herein shown and described will be apparent to those 
skilled in the art all within the intended spirit and scope of the 
invention. Accordingly, the patent is not to be limited in scope and 
effect to the specific embodiment herein shown and described nor in any 
other way this is inconsistent with the extent to which the progress in 
the art has been advance by the invention.