Reconfigurable fir filter

A series of digit processing units (DPUs) are connected to form a finite impulse response (FIR) filter. Each DPU includes a register, a multiplexer, and a coefficient multiplier. The register stores and delays an input digital signal to be filtered. The multiplexer has inputs connected to the input node and to an output of the register, an output of the multiplexer for connecting to a next stage DPU. The coefficient multiplier is connected to the output of the register and multiplies the input signal by a coefficient or part of a coefficient. A group of DPUs can have multiplexers set so that the register of each DPU stores the same part of the input signal for processing a single filter coefficient. An adder is provided to sum output of the DPUs and output a filtered signal. The critical path of the FIR filter is independent of coefficient number and precision.

BACKGROUND OF INVENTION

1. Field of the Invention

The present invention relates to digital signal processing, and more specifically, to a programmable digital finite impulse response (FIR) filter.

2. Description of the Prior Art

Finite impulse response (FIR) filters are important components in digital communications systems. Much effort has been made to improve filter performance, reduce hardware, and increase operating speed. In addition, software radios, such as those introduced in J. Mitola, “The Software Radio Architecture,”IEEE Communications Magazine, vol. 33, pp. 26-38, May 1995 or E. Buracchini, “The Software Radio Concept,”IEEE Communications Magazine, vol. 38, pp. 138-143, September 2000, have recently gained much attention due to the need for integrated and reconfigurable communications systems. To this end, reconfigurability has become an important issue for future filter design.

FIR filters can be used to perform a wide variety of tasks such as spectral shaping, matched filtering, noise rejection, channel equalization, etc. Hence, various architectures and implementation methods have been proposed to improve the performance of filters with respect to speed and complexity. However, due to the recent explosive proliferation in wired and wireless communication standards, traditional hardwired devices may be less suitable for future communication needs.

On the other hand, software radio has gained much attention from researchers worldwide due to a strong demand for reconfigurable communication systems capable of performing multi-standard operations. In light of this trend, programmability and reconfigurability need be taken into account in filter architecture design.

A typical N-tap FIR filter can be described by:

y[n] is a filtered digital signal, n being an index of elements of the signal;

hiis a filtering coefficient; and

x is an unfiltered digital signal.

It is well known in the art that a canonical signed digit (CSD) representation can be used to reduce the complexity of a digital FIR filter implementation as in R. M. Hewlitt and E. S. Swartzlantler Jr., “Canonical Signed Digit Representation for FIR Digital Filters,” inProc. of IEEE Workshop on Signal Processing Systems,2000, pp.416-426; M. Tamada and A. Nishihara, “High-Speed FIR Digital Filter with CSD Coefficients Implemented on FPGA,” inProc. of the ASP-DAC,2001, pp. 7-8; and Y. M. Hasan, L. J. Karem, M. Falkinburg, A. Helwig, and M. Ronning, “Canonic Signed Digit Chebyshev FIR Filter Design,”IEEE Signal Processing Letters, vol. 8, pp. 167-169, June 2001, for example. Encoding filter coefficients using a CSD representation reduces the number of partial products and thus saves silicon area and power consumption in hardware implementation. Hence, this technique has been popular for fixed-coefficient implementation of FIR filters. According to the CSD representation:

di,kis an element of the set {1, 0, −1};

pkis an element of the set {0, . . . , L}, where L+1 is the length of the coefficients;

Miis the number of nonzero digits in hi.

When applying the CSD representation to implementing programmable, rather than fixed-coefficient, FIR filters, it is only natural to implement the same number of programmable CSDs for each filter coefficient to maintain regularity. However, for most filters, only a few taps require high-precision coefficients. Valuable hardware resources will be wasted if all taps are implemented with the highest precision. To minimize hardware complexity, programmable FIR filters restricting the number of allowable nonzero CSDs in every tap have been proposed in T. Zhangwen, Z. Zhanpeng, Z. Jie, and M. Hao, “A High-Speed, Programmable, CSD Coefficient FIR Filter,” inProc. of4th International Conference on ASIC,2001, pp.397-400; and in K. T. Hong, S. D. Yi, and K. M. Chung, “A High-Speed Programmable FIR Digital Filter Using Switching Arrays,” inProc. of IEEE Asia Pacific Conference on Circuits and Systems,1996, pp. 492-495. Unfortunately, this restriction may lower the coefficient precision and degrade the frequency response of the filter, and it may also induce a large overhead by assigning more CSDs than necessary to most taps. Another hardware-efficient implementation of programmable FIR filters with CSD coefficients has been presented in K. Y. Khoo, A. Kwentus, and A. N. Willson Jr., “A Programmable FIR Digital Filter Using CSD Coefficients,”IEEE Journal of Solid-State Circuits, vol. 31, pp. 869-874, June 1996. This implementation includes a 32-tap linear-phase filter with two nonzero CSDs in each tap. Additional nonzero CSDs can be allocated to specific filter taps, making it a reconfigurable FIR filter architecture. Nevertheless, some computational resources can still be unused and the critical path can be quite longin some cases.

Another state of the art programmable FIR filter is taught by Willson, Jr. et al. in U.S. Pat. No. 5,479,363, which is included herein by reference. ConsiderFIG. 1showing taps of a filter of a kind taught in U.S. Pat. No. 5,479,363. The filter comprises a series of p-taps70a-fthat include tap coefficient multipliers74a-f, adders78a-f, unit delays (registers)77a-f, and delay bypass lines75a-ffor filtering digital data on a line72. Assuming each of the p-taps70a-fhas a two-digit signed coefficient multiplier, bypass lines77a-fcan be selectively connected to bypass specific unit delays, merging p-taps to effectively increase the precision of the coefficient multipliers. This is shown inFIG. 1, where bypass line75bis active and bypasses the corresponding register77bsuch that a four-digit coefficient is realized by multipliers74b,74cand adders78b,78c. A six-digit coefficient is realized in a similar way. A fundamental shortcoming of the filter ofFIG. 1is that the critical path depends on coefficient precision. In the four-digit coefficient, for example, the critical path includes the multiplier74band the two adders78b,78c, while the six-digit coefficient has a longer critical path including a multiplier and three adders. This dependence of critical path on precision results in slow, inefficient, and somewhat unpredictable performance.

Generally, the prior art programmable FIR filters suffer from drawbacks of program inflexibility, speed, precision range, and critical path dependence on precision.

SUMMARY OF INVENTION

It is therefore a primary objective of the present invention to provide a highly flexible, reconfigurable FIR filter in which both a tap number and a number of nonzero digits in each tap can be arbitrarily assigned, and in which critical path is independent of coefficient precision.

Briefly summarized, a digit processing unit (DPU) for providing a CSD coefficient to a FIR filter according to the present invention includes a register, a multiplexer, a coefficient multiplier, and an adder. The register is connected to an input node and stores and delays an input digital signal to be filtered. The multiplexer has inputs connected to the input node and to an output of the register, an output of the multiplexer is for connecting to a second DPU. The coefficient multiplier is connected to the output of the register and multiplies the input digital signal by a CSD coefficient and outputs a product. The adder is connected to the coefficient multiplier and adds the product to products of other DPUs, the output of the adder being a component of the filtered digital signal.

According to a preferred embodiment of the present invention, DPUs are connected in series to form a FIR filter, and a group of DPUs can have multiplexers set so that the register of each DPU stores the same part of the digital signal for processing a single filter coefficient. Additionally, the adders of the DPUs are consolidated into a single optimized adder.

A method according to the present invention for filtering an input digital signal according to a function defined by a series of coefficients is also provided. The method serially receives the input digital signal as a series of equal length elements, then, simultaneously multiplies each element of the serially received digital signal by a corresponding coefficient of the series of coefficients, and further adds the products of the multiplications, before finally outputting the sum of the products of the multiplications as the filtered digital signal.

It is an advantage of the present invention that the multiplexers allow DPUs to be combined to process coefficients having a wide range of precisions in the same FIR filter.

It is a further advantage of the present invention that the critical path is a coefficient multiplier and an optimized consolidated adder and is independent of an amount of DPUs processing a single coefficient, that is, coefficient precision or number of digits.

It is a further advantage of the present invention that the FIR filter can be easily configured as a matched filter, a pulse-shaping filter, or other filters.

It is a further advantage of the present invention that the FIR filter has scalability, modularity, and cascadability amenable to VLSI implementation.

DETAILED DESCRIPTION

A generalized digit processing unit (DPU)10according to the present invention is illustrated inFIG. 2. The DPU10can be connected in a series of stages of like DPUs to form a filter, such as a finite impulse response (FIR) filter common in digital signal processing applications. The DPU10includes a delay unit, such as a register12, for storing and delaying an input digital signal (indicated inFIG. 2by “Data in”), and a multiplexer14for selecting output of the DPU10as either the input digital signal or the delayed digital signal output by the register12. A coefficient multiplier16is connected to an output of the register and multiplies the data output by the register12by a filter coefficient or part of a filter coefficient, outputting this product to an adder18. The adder18sums the product with output of a prior stage DPU connected at adder input20. Output of the adder18is sent to a next stage DPU, or when the DPU10is the last stage, to a filter output.

The coefficient multiplier16is programmed with a unit set of canonical signed digits (CSDs) of a filter coefficient, the advantages of using CSDs having been explained previously. For instance, the unit set of CSDs can be a single CSD. When the filter coefficient comprises a single CSD, the multiplexer16is set to receive input from the register12thereby delaying output to the next stage DPU, which processes another filter coefficient. However, when the CSD representation of the coefficient comprises two CSDs, the multiplexer16is set to combine the DPU10with the next stage DPU (by forwarding the un-delayed input digital signal) such that two coefficient multipliers operate on the same input digital signal data to realize a two CSD coefficient. In this way, one or more CSDs can be realized with a single or a series of DPUs10.

FIG. 3illustrates a DPU30according to the preferred embodiment of the present invention. The DPU30is similar to the DPU10while including further components to optimize operation. In the preferred embodiment, the DPU30is set up to accept digital data in 8-bit segments to be multiplied by a single CSD, and accordingly, the DPU30has a 14-bit output and a 1-bit output. The DPU30includes a delay unit, such as a register32, for storing and delaying an input digital signal, and a multiplexer34for selecting output of the DPU30as either the input digital signal or the delayed digital signal output by the register32. The DPU30further comprises a multiplier36and a shifter37for performing a multiplication of the input data by a CSD of a filter coefficient. A serial-in-parallel-out (SIPO) shift register38is provided so that a control string can be loaded. The control string can be serially shifted through the SIPO register38to a next stage DPU and includes a configuration bit “config” for setting the multiplexer32, a “zero” and a “plus” bit for setting the multiplier36, and three “shift” bits for controlling the shifter37. Addend and sign output of the shifter37and multiplier36respectively are forwarded to an adder (not shown). The DPU30is to be used in a series of like DPUs to form a filter.

Referring to Eqn. 3 and Table 1, the multiplier36is set with the multiplicand di,k, the zero bit indicating a zero value and the plus bit indicating a positive value. The shifter37is set to evaluate the multiplicand 2−pk,

the three shift bits being a binary representation of pk. Working in conjunction, the multiplier3and shifter37evaluate a single CSD multiplication, that is, the term
di,k·2−pk·x[n−i]
of Eqn. 3.

As mentioned with reference to Table 1, the multiplier36is used to multiply the input data x[n−i] by di,k, which can have values of “1”, “0”, and “1”. If di,kis “0, the zero signal will be “1” forcing the output of the multiplier 36 to be “0” regardless of input. Otherwise, the zero signal will be 0″ and if the CSD coefficient is 1, the plus signal will be “1” and the multiplier output is the same as the input. If the CSD coefficient, di,k, is “−1”, the plus signal will be “0” and the output is equivalent to the one's complement representation of the input data. The “1 ” required to form the two's complement can be added by the multiplier36or, as in the preferred embodiment, accumulated and later added into a summed filter output when the DPU30is incorporated into a filter.

The shifter37is used to multiply the output of the multiplier36di,kx[n−i] by 2−pk,

where pkranges from “0” to “7”. In the preferred embodiment, the shifter37performs an arithmetic left shift and expands the 7-bit multiplier output (excluding the most significant bit—MSB) into a 14-bit output by shifting the input left by 7−pk

bits. Zeros are padded at the least significant bit (LSB) if di,kis “1 or 0” and ones are padded if di,kis −1″.

Please refer toFIG. 4illustrating architecture of a reconfigurable FIR filter40employing a series of DPUs30. The FIR filter40includes a series of adders42corresponding to the series of DPUs30, and a register44for storing the accumulated “1”s output required to form the two's complement by the multipliers36of the DPUs30as previously mentioned. The adders42receive corresponding sign and addend signals from the DPUs30and output a filtered digital signal, each adder42processing a partial sum. A pre-calculated control string can be serially fed into the filter40, via the SIPO registers38of the DPUs30, to link adjacent DPUs with the multiplexers34, and set the multipliers36and shifters37according to the desired CSD coefficients. Thus, the FIR filter40can reconfigurably process an input digital signal with CSD represented coefficients limited in size and precision only by the number of DPUs30incorporated.

Refer toFIG. 5, showing a reconfigurable FIR filter processing element50according to the present invention. The processing element50has a similar structure to the FIR filter40, however, the processing element50is optimized for reduced latency and efficient IC fabrication. The processing element50includes a series of DPUs30, a combined adder52, a register54, and a sign extension generator56. Because of different precision between the output of DPUs30and the accumulated sum of the adder52, the sign extension generator56is required for generation of sign extension bits based on the sign outputs of the DPUs30. The register54stores “Acc”, the number of negative CSDs in the DPUs30, for performing two's complement arithmetic required by the multipliers36in the DPUs30. The adder52sums the addend output of the DPUs30, the output of the sign extension generator56, and the accumulated sum at the register54and outputs a filtered digital signal. The processing element50is readily incorporated into a pipeline arrangement that is well known in the art, such as a plurality of processing elements50connected in series.

The sign extension generator56is required as the accumulated sum at the adder52has a longer bit length than the addend output of each DPU30. For power saving reasons, it is better to handle sign extension bits of the DPUs30individually rather than extend the addends of the DPUs30to the bit length of the adder52. The sign extension generator56evaluates the sum of the sign extension bits based on the sign signals of the DPUs30by examining relations between the number of non-negative sign signals and the sum of the corresponding sign extension bits.

Suppose, for example, that each DPU30used in the processing element50processes 8-bit data with 8-bit filter coefficients so as to produce a 15-bit output (a 14-bit addend signal and a 1-bit sign signal, referring toFIG. 3). Suppose, for example, that the filter output as well as “Acc” is 24-bit wide, then,the sign extension generator56output is accordingly ten bits. The sign extension generator56includes a multiplexer that selects the seven MSBs of the output as “1111111” when any DPU30has a sign signal of “1”, or “0000000” when no sign signal is “1”. The sign extension generator56sets the three LSBs to equal three LSBs of a binary representation of the number of non-negative sign signals. Thus, in this example, a 10-bit sign extension signal is output by the sign extension generator56to the adder52.

Continuing the example above, the adder52sums eight 14-bit addend signals from the eight DPUs30, one 24-bit accumulated sum at the register54, and the 10-bit sign extension signal. The adder52includes five 14-bit full adder arrays in a two-level arrangement that compress the fourteen LSBs of the accumulated sum at the register54and the eight addend signals into four 14-bit signals. A two-level carry save adder is provided to add the ten MSBs at the register54, the sign extension signal, and the above four 14-bit signals. The adder52further comprises an ELM adder, such as in T. P. Kelliher, R. M. Owens, M. J. Irwin, and T. T. Hwang, “ELM-A Fast Addition Algorithm Discovered by a Program,”IEEE Transactions on Computers, vol. 41, pp.1181-1184, September 1992, modified to reduce the critical path delay and compute the final sum.

It can be seen inFIG. 3andFIG. 5that the critical path of the processing element50includes a coefficient multiplier (multiplier36and shifter37) and an adder52regardless of how many CSDs, and consequently how many DPUs30, are required to express a filter coefficient. This is also the case for the FIR filter40ofFIG. 4provided that the FIR filter40is constructed by cascading several processing elements50in a pipeline fashion. To be precise, the critical path of the present invention is independent of the filter coefficients.

In practical application, the present invention can be implemented with single poly quadruple-metal 0.35-μm CMOS technology. In accordance with the above-mentioned example of eight DPUs30processing 8-bit signal data, measurement results have shown that a fabricated chip consumes 16.5 mW of power when operating at 86 MHz under 2.5V.

In contrast to the prior art, the present invention has a critical path including a coefficient multiplier and an optimized consolidated adder that is independent of coefficient precision or number of digits. Furthermore, the present invention DPUs can be combined to process coefficients having a wide range of precisions in the same configurable FIR filter or processing element, and such a FIR filter is thus scalable, modular, cascadable, and well suited to VLSI implementation.

Those skilled in the art will readily observe that numerous modifications and alterations of the device may be made while retaining the teachings of the invention.

Accordingly, the above disclosure should be construed as limited only by the metes and bounds of the appended claims.