Self-calibrating, self-correcting transceivers and methods

A transceiver has a digital signal processor which can insert calibration signals of known level and frequency into transmitters for calibration and correction of transmitter parameters. An output of the calibrated and corrected transmitter is subsequently coupled into a calibration mixer along with a mixing signal (e.g., from a local oscillator generator. The outputs of the calibration mixer have known levels and frequencies and are inserted into receivers for calibration and correction of receiver parameters.

BACKGROUND OF THE INVENTION 
1. Field of the Invention 
The present invention relates generally to transceivers and, more 
particularly, to built-in calibration structures for transceivers. 
2. Description of the Related Art 
The flexibility, power and performance of digital transceivers have caused 
them to become an essential part of mobile radio systems, cellular 
telephone systems, paging systems, automotive entertainment systems and 
military communications. 
Digital transceivers are typically complex systems and their performance is 
a function of a number of system parameters (e.g., transmitter gain, 
receiver gain, receiver noise, receiver intermodulation distortion and 
receiver dynamic range). Periodic monitoring and recalibration of these 
parameters is desirable if transceiver performance is to be maintained at 
a high level. 
Monitoring and recalibration costs can be greatly reduced if the 
transceiver includes significant built-in test and calibration structures. 
These structures can also reduce initial manufacturing cost because they 
can expose problems at early manufacturing stages and can reduce the need 
for expensive test equipment. These structures can also enhance customer 
satisfaction because they reduce the cost and time required to maintain 
transceivers in a high-performance state. 
Various built-in, self-test (BIST) systems have been proposed. For example, 
a BIST system for mixed analog-digital (MAD) structures was described by 
M. F. Toner, et al. (Toner M. F., et al., "On the Practical Implementation 
of Mixed Analog-Digital BIST", IEEE 1995 Custom Integrated Circuits 
Conference, March, 1991, pp. 525-528). In this BIST system, a precision 
analog test stimulus was generated with an over-sampling oscillator. The 
test stimulus included single and/or multiple sine-waves with 
digitally-programmable amplitudes, frequencies and phases. The analog 
output was imbedded in a pulse-density modulated digital bit stream. 
Measurement of responses to the test stimulus was facilitated with an 
analog-to-digital converter (ADC) whose output was coupled through a 
narrow-band digital filter. Analog signal and noise powers were extracted 
by applying analog waveforms to the ADC, selecting a passband frequency 
for the digital filter and computing the sum of squares of samples 
emerging from the filter. 
In a sequence of operations, the oscillator was used to calibrate the ADC 
which was then used to measure analog signals. Toner, et al. suggested 
that the ADC could be used to calibrate an analog-to-digital converter 
(DAC) and the two converters used as a precision analog stimulus and 
volt-meter. 
Although the suggestion of an ADC and a DAC as analog stimulus and 
volt-meter is helpful, it fails to enable the calibration and correction 
of complex systems such as transceivers. 
SUMMARY OF THE INVENTION 
The present invention is directed to a self-calibrating, self-correcting 
transceiver. This goal is achieved with a signal processor which can 
inject calibration signals of known levels and frequencies into 
transmitters of the transceiver for calibration and correction of 
transmitter parameters. Subsequently, transmitter signals from a 
calibrated transmitter and mixing signals from another signal source 
(e.g., a local oscillator generator) are applied to inputs of a 
calibration mixer. Outputs of this mixer have known known levels and 
frequencies and are injected into receivers of the transceiver for 
calibration and correction of receiver parameters. The signal processor 
measures responses of the transmitters and receivers and parameter 
corrections are communicated to the transmitters and receivers from a 
controller. 
In an exemplary calibration and correction method, transmitter calibration 
signals are sent through a transmitter, responses to the calibration 
signals are monitored and transmitter responses corrected if necessary. 
Subsequently, a calibrated transmitter signal is mixed with a mixing 
signal to generate receiver calibration signals which are injected into a 
receiver. Receiver responses to the calibration signals are monitored and 
receiver responses corrected if necessary. 
The novel features of the invention are set forth with particularity in the 
appended claims. The invention will be best understood from the following 
description when read in conjunction with the accompanying drawings.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
A self-calibrating, self-correcting transceiver 20 is shown in FIGS. 1, 2, 
3 and 5 and an exemplary calibrating and correcting process for the 
transceiver is shown in FIG. 4. The transceiver 20 is structured to 
facilitate insertion of calibration signals with known levels and 
frequencies into its transmitters and receivers. The transceiver is 
further structured to measure the transmitter and receiver responses to 
these calibration signals and to facilitate any required corrections of 
transceiver parameters. 
A description of the process of FIG. 4 is preceded by the following 
description of the transceiver 20 of FIGS. 1, 2 and 3. After the process 
description, another receiver embodiment is described with reference to 
FIG. 5. 
The transceiver 20 of FIG. 1 includes at least one transmitter 22 (other 
transmitters are exemplified by a second transmitter 22A) and at least one 
receiver 24 (other receivers are exemplified by a second receiver 24A and 
a receiver 190). 
In the transmitter 22, a transmitter input is formed by a serial 
combination of a low-pass filter 26 and a video amplifier 28. A 
transmitter output is taken from a high-power amplifier (HPA) 30. In an 
exemplary fabrication process which enhances transceiver miniaturization, 
a major portion of the transmitter is carried on an integrated circuit 34. 
In the receiver 24, a receiver input is formed by the serial combination of 
a band-pass filter 36 and a low-noise amplifier (LNA) 38. Similar to the 
transmitter 22, a major portion of the receiver 24 is carried on an 
integrated circuit 44. For clarity of illustration, the details of the 
integrated circuits 34 and 44 are shown respectively in FIGS. 2 and 3 and 
will be described hereinafter. 
Although the transmitters and receivers can share antennas, they are 
preferably each coupled to a respective antenna. Thus, the output of the 
transmitter 22 is coupled to an antenna 46 and the input of the receiver 
24 is coupled to an antenna 48. 
The inputs of the transmitters and the outputs of the receivers are 
connected to a signal processor 50. The flexibility of modern 
communication systems (e.g, systems which process signal modulations such 
as binary phase shift keying, quadrature phase shift keying and .pi./4 
quadrature phase shift keying) is enhanced by the use of digital 
processing. Accordingly, the signal processor 50 is preferably a digital 
signal processor and it is coupled to each transmitter input by a 
respective digital-to-analog converter (DAC) 52 (for simplicity of 
illustration, only one is shown) and coupled to the receiver outputs by an 
analog-to-digital converter (ADC) 54. In particular, the receiver outputs 
are connected to the analog-to-digital converter 54 through a serial 
combination of a summing amplifier 56 and a low-pass filter 58. The signal 
processor 50 is associated with a controller 60 and the transmitters and 
receivers are coupled to the controller 60 over a control bus 59. 
As detailed in FIG. 2, the transmitter's integrated circuit 34 has first 
and second mixers 61 and 62 for double upconversion of video signals from 
the transmitter video amplifier (28 in FIG. 1). A low-pass filter 64 
precedes the first mixer 61 and a band-pass filter 66 and an 
intermediate-frequency amplifier 67 are positioned between the first and 
second mixers 61 and 62. A radio-frequency amplifier 68 and a bandpass 
filter 69 connect the second mixer 62 to the transmitter's HPA (30 in FIG. 
1). 
The first and second mixers 61 and 62 are of a mixer type (e.g., Gilbert 
mixers) whose gain is responsive to a control signal. A feedback system 
comprising a serially-connected logarithmic detector 70 and an automatic 
gain control (AGC) circuit 72 is connected between the output of the HPA 
30 and the first and second mixers 61 and 62. The gain of the transmitter 
22 is stabilized by this feedback system. The gain can be adjusted through 
commands to the AGC circuit 72. 
The upconversion frequencies for the first and second mixers 61 and 62 are 
provided by a local oscillator (LO) generator 73. The adaptability of the 
transmitter 22 is enhanced if its output frequency is selectable from a 
band of frequencies. Accordingly, the LO generator outputs are each 
tunable over a frequency band. In an exemplary tunable LO generator, a 
plurality of frequency steps are generated with phase-lock loops which are 
locked in selectable divide-by ratios to a voltage-controlled oscillator 
which, in turn, is locked by selectable ratios to a reference. In the 
transceiver 20, this reference is provided by a clock generator 74 which 
is shown in FIG. 1. The clock signal is coupled to the LO generator 73 
over a clock line 75. 
In the integrated circuit 34, the AGC circuit 72 and the LO generator 73 
are connected to a serial interface 76 which, in turn, is connected 
through a control line 77 and the control bus to the controller (59 and 60 
in FIG. 1). Through the serial interface 76, the controller can also 
communicate with a chip enable circuit 82 which applies operating power to 
the integrated circuit. With signals to the chip enable circuits of the 
transmitters, the controller (60 in FIG. 1) can enable any selected one of 
the transmitters. 
The transmitter 22 has a monitor multiplexer (MUX) 84 which sends a 
plurality of monitor signals from the transmitter to the controller 60 via 
a MUX line 86. For example, a temperature diode 88 reports the transmitter 
temperature (e.g., the temperature of the integrated circuit chip 34). 
Other monitor signals include a signal that indicates the frequency of a 
voltage-controlled oscillator in the LO generator 73, AGC levels from the 
AGC circuit 72 and various DC voltage supply levels. These monitor signals 
are communicated to the controller 60 by the monitor MUX 84 which is 
responsive to controller commands via the serial interface 76. 
As detailed in FIG. 3, the receiver's integrated circuit 44 has first and 
second mixers 91 and 92 for double downconversion of a radio-frequency 
signal from the LNA (38 in FIG. 1). A radio-frequency amplifier 93 and and 
a bandpass filter 94 precedes the first mixer 91 and a bandpass filter 96 
and an intermediate-frequency amplifier 97 are positioned between the 
first and second mixers 91 and 92. A video amplifier 98 and a low-pass 
filter 99 connect the second mixer 92 to the receiver output. 
Similar to the transmitter mixers (61 and 62 of FIG. 2), the gains of the 
first and second mixers 91 and 92 are responsive to a control signal. In 
addition, the integrated circuit 44 has an AGC module 100 (e.g., a pin 
diode) which is positioned ahead of the radio-frequency amplifier 93 and 
which is also responsive to a control signal. A feedback system controls 
the receiver gain. It comprises a serially-connected logarithmic detector 
102 and an AGC circuit 104 which are connected between the output of the 
integrated circuit 44 and the first and second mixers 91 and 62 and the 
AGC module 104. 
The upconversion signals for the first and second mixers 91 and 92 are 
provided by a LO generator 106 which is similar to the transmitter's LO 
generator (73 in FIG. 2). A reference frequency for this LO generator is 
also provided by the clock generator 74 over a clock line 108. 
The AGC circuit 104 and the LO generator 106 are connected to a serial 
interface 110 which, in turn, is connected through a control line 112 and 
the control bus to the controller (59 and 60 in FIG. 1). Through the 
serial interface 110, the controller can communicate with a chip enable 
circuit 113 on each receiver and, thereby, enable a selected one of the 
receivers. 
The receiver 24 also has a monitor MUX 114 which sends a plurality of 
monitor signals to the controller 60 via a MUX line 116. Similar to the 
transmitter, these monitor signals include one from a temperature diode 
118, a signal that indicates the frequency of a voltage-controlled 
oscillator in the LO generator 106, AGC levels from the AGC circuit 100 
and various DC voltage supply levels. These monitor signals are 
communicated to the controller 60 by the monitor MUX 114 which is 
responsive to controller commands via the serial interface 110. 
Because the MUX signals from the transmitter 22 and the receiver 24 are 
typically analog, they are communicated to the controller 60 (which is 
typically digital) through a MUX 120 and an analog-to-digital converter 
122 (as illustrated in FIG. 1). A blanking detector 124 is coupled ahead 
of the first mixer 91 to monitor for the presence of interference signals. 
This detector is responsive to the controller 60 via the serial interface 
110 and can send a blanking signal over a blanking line 126 to the signal 
processor 60 (as also illustrated in FIG. 1). 
Having described details of the integrated circuits 34 and 44 of the 
transmitter 22 and receiver 24, attention is returned to FIG. 1. In an 
exemplary embodiment, the signal processor 50 includes a communication 
signal processor 130 and a digital tuner 132. The communication signal 
processor is configured to convert an input digital bit stream 134 into 
modulated in-phase and quadrature (I/Q) baseband samples 136 which are 
coupled to the digital tuner. 
This modulation is typically selectable from various communication 
modulations (e.g., binary phase shift keying, quadrature phase shift 
keying and .pi./4 quadrature phase shift keying). The modulated I/Q 
baseband samples 136 are processed by the digital tuner 132 into a digital 
signal 142 having a selected frequency rate and this signal is coupled to 
the transmitter 22 through the DAC 52. Preferably, the communication 
signal processor 130 and tuner 132 are configured with multiple channels 
so that respective DAC's (not shown) couple signals to the other 
transmitters (as exemplified by transmitter 22A). 
On a receive path, baseband signals from receivers, e.g., receivers 24, 24A 
and 190, are summed in amplifier 56 and coupled to the digital tuner 132 
through a common ADC 54. The tuner 132 splits the digitized baseband 
signals 144 into I/Q components 146 from which the communication signal 
processor 130 recovers an output data stream 148. Preferably, the 
communication signal processor 130 and tuner 132 have multiple processing 
channels so they simultaneously process signals from multiple receivers. 
Although many other transceiver embodiments may be used with the teachings 
of the invention, one structured with a common ADC (as shown in FIG. 1) 
reduces costs because high-performance ADC's are typically an expensive 
item. 
FIG. 1 shows that signals from the transmitter 22 are coupled over a signal 
line 149 to one input of a calibration mixer 150. Another input of the 
calibration mixer 150 is coupled over a signal line 151 from a signal 
source that is preferably responsive to the controller 60. In the 
transceiver embodiment 20, this signal source is the receiver's LO 
generator 106 as shown in FIG. 3. The output of the calibration mixer 150 
is coupled through a summer 152 to the input of the receiver's integrated 
circuit 44. 
In other transceiver embodiments, the input to the calibration mixer 150 
can be taken from the output of the transmitter, i.e., from the output of 
the HPA 30 as indicated by the broken line 154 in FIG. 1. Although the 
output of the summer 152 can be injected into the receiver input as 
indicated by the signal arrow 156 in FIG. 1, the receiver's noise figure 
is enhanced if the insertion loss of the summer 152 is positioned after 
the LNA 38. 
In operation of the transceiver 20, a modulated, baseband signal 142 from 
the digital tuner 132 is coupled through the DAC 52 to the transmitter 22 
where it is amplified, filtered and upconverted and passed to the antenna 
46 for radiation. This transmission path is exemplary of other transmitter 
paths (e.g., through the transmitter 22A) and, although not shown for 
clarity of illustration, a respective DAC and antenna may be dedicated to 
each transmitter path. 
In reception, a radio-frequency signal from the antenna 48 is amplified, 
filtered and downconverted by the receiver 24 and passed through the 
summer 56 and ADC 54 as a baseband signal 144 to the tuner 132. 
An exemplary process of calibrating and correcting the transceiver 20 is 
shown in the flow chart 160 of FIG. 4. In a first process step 162, 
calibration signals are sent through a transmitter (e.g., the transmitter 
22 of FIG. 1). These calibration signals are preferably generated by the 
digital signal processor 50 of FIG. 1 so that they have a known level and 
can be swept over a known transmitter passband. In process step 164, the 
transmitter response is monitored. For example, signals from the detector 
70 of FIG. 2 are coupled through the monitor MUX 84 of FIG. 2 and the MUX 
path 86 to the signal processor 50 of FIG. 1 for measurement. 
In decision step 166, the transmitter response is compared to a 
predetermined range. If the response is out of range, it is corrected in 
process step 168. For example, the transmitter gain can be adjusted by the 
AGC circuit 72 of FIG. 2 in response to commands from the controller 60 of 
FIG. 1. 
The transmitter responses that can be calibrated and corrected in with 
calibration signals from the signal processor 50 include transmitter gain 
over frequency, transmitter gain over temperature (using data from the 
temperature diode 88 of FIG. 2), transmitter gain over AGC range and AGC 
transfer function (using data from the AGC circuit 72 of FIG. 2) and 
frequency responses of the LO generator 73 of FIG. 2 (e.g., frequency 
response of a voltage-controlled oscillator and frequency steps of a 
phased-lock loop as function of commanded divide-by ratios). This process 
of calibration and correction can be conducted for each transmitter of the 
transceiver 20. 
In process step 170, a signal from a transmitter is used in a mixing 
process with the calibration mixer 150 of FIG. 1. Because this transmitter 
has been calibrated and corrected, the signal levels sent over the signal 
line 149 of FIG. 1 have known levels. Because the conversion loss of the 
mixer 150 and the summer 152 are known, the receiver calibration signal 
levels coupled into the receiver 24 are also known. Either of the input 
signals to the mixer 150 can be used as a high-level signal to the mixer 
so that the mixer's output level is reduced from the other input signal by 
the mixer's conversion loss. 
The frequency of the signals can be varied with commands from the 
controller 60 that cause frequency changes in either or both of the mixer 
input signals, i.e., the signal in signal line 149 from the transmitter 22 
and the signal in signal line 151 from LO generator 106. 
In process step 172, the receiver responses to the receiver calibration 
signals from the summer 152 are monitored. In decision step 174, the 
receiver response is compared to a predetermined range. If the response is 
out of range, it is corrected in process step 176. For example, the 
receiver gain can be adjusted by the AGC circuit 100 of FIG. 3 in response 
to commands from the controller 60 of FIG. 1. 
The receiver responses that can be calibrated and corrected in this manner 
include receiver gain over frequency, receiver gain as a function of AGC 
signals (using data from the AGC circuit 100 of FIG. 3), receiver gain 
over temperature (using data from the temperature diode 118 of FIG. 3), 
receiver intermodulation distortion (using two mixer signals to the mixer 
150 from the transmitter 22), receiver dynamic range, gain matching of 
different receivers, isolation of one receiver channel from another and 
gain and phase match of I/Q signals. 
By monitoring frequency changes at the receiver output, the frequency 
response of phase-lock loops in the LO generator 106 can be measured as a 
function of divide-by ratios that are commanded from the controller 60. 
Also the frequency of a voltage-controlled oscillator in the LO generator 
106 can be monitored as a function of commanded tuning voltages. In 
addition, all transmitter calibration signals can be interrupted and the 
receiver and ADC noise measured by the signal processor 50. 
This process of calibration and correction can be conducted for each 
receiver of the transceiver 20. The calibration and correction of the 
transceiver ends at terminator 180 of flow chart 160. 
The teachings of the invention can be applied to a variety of transmitters 
and receivers. For example, one of the receivers of the transceiver 20 of 
FIG. 1 is particularly structured for image rejection. This is 
accomplished by processing received signals with a pair of mixers whose 
local oscillator signals are in a quadrature relationship. 
This image-rejection receiver is the receiver 190 which is shown in FIGS. 1 
and 5. The receiver 190 is similar to the receiver 24 with like elements 
indicated by like reference numbers. Similar to the receiver 24, a major 
portion of the receiver 190 is carried on an integrated circuit 192 which 
is detailed in FIG. 5. In contrast to receiver 24, the receiver 190 has a 
pair of parallel-arranged mixers 194 and 196 which receive local 
oscillator signals from a LO generator 198. The local oscillator signals 
from this generator are in a quadrature relationship, i.e., they have a 90 
degree phase difference. The downconverted signals from the mixers 194 and 
196 are respectively filtered in filters 198 and 200 and combined in a 
summing amplifier 202 whose output is filtered by a capacitor 204. Because 
of the quadrature relationship of the local oscillator signals, the 
outputs of the mixers 194 and 196 will add for an expected radio-frequency 
signal into the receiver 190 and cancel for signals at the image 
frequency, i.e., a frequency on the other side of the local oscillator 
frequency. 
Calibration and correction of the receiver 190 is carried out in a manner 
similar to that described in flow chart 160 of FIG. 4. To facilitate this, 
the receiver 190 has a calibration mixer 150 and a summer 152 which are 
shown in FIG. 5 (for clarity of illustration, these elements are not shown 
in FIG. 1). 
Transceivers of the invention can be adapted to operate at any 
communication frequency band, e.g., UHF, VHF, X-band and Ku-band. Although 
the invention has been described with reference to double-upconversion 
transmitters, double-downconversion receivers and image-rejection 
receivers, these are exemplary and the teachings of the invention can be 
practiced with any transmitters and receivers. Transceivers of the 
invention are particularly suited for calibration and correction of their 
performance parameters. Accordingly, they are structured to facilitate 
calibration of transmitter paths and receiver paths and are further 
structured to facilitate correction of parameters to bring them into 
predetermined performance ranges. 
While several illustrative embodiments of the invention have been shown and 
described, numerous variations and alternate embodiments will occur to 
those skilled in the art. Such variations and alternate embodiments are 
contemplated, and can be made without departing from the spirit and scope 
of the invention as defined in the appended claims.