Method and device relating to supervision and control of an oscillator signal

The present invention relates to methods and devices for such control and supervision of an oscillator signal from a controllable oscillator that is done mainly to control the frequency variation of the oscillator signal. According to the invention, the controllable oscillator is controlled by a controlling voltage, which in turn is modified by a correction signal, generated in a control loop. A time discrete representation of a secondary phase is generated in the control loop, the secondary phase corresponding to a frequency being the difference between the frequency of the oscillator signal and a constant frequency. A time discrete approximation signal is generated in dependence of the time discrete representation of the secondary phase. A time discrete error signal is generated in dependence of the time discrete approximation signal, the time discrete error signal indicating the difference between the actual frequency slope of the oscillator signal and a desired frequency slope. The correction signal is generated in dependence of the time discrete error signal. The control loop can also be adaptive, meaning that data from one control sequence is being used in a later control sequence.

THE TECHNICAL FIELD OF THE INVENTION 
The invention relates to methods and devices for such control and 
supervision of an oscillator signal that is done mainly to control 
frequency variation of the oscillator signal. More specific the invention 
relates to control of an oscillator signal from a controllable oscillator 
and the supervision of the oscillator signal for determining the frequency 
slope of the oscillator signal. 
PRIOR ART 
A voltage controlled oscillator--commonly named VCO--is a well-known 
device, commonly used within varying technical fields, e.g. within the 
field of radio technology. The voltage controlled oscillator generates an 
oscillating signal the frequency of which is controlled through a control 
voltage fed to the oscillator. 
In a number of technical applications the need arises to control the 
oscillator signal frequency such that the frequency varies with time in a 
manner suitable for the relevant application. 
One such application exists in an aircraft radar equipment using a 
HPD-waveform (High Pulse Repetition Frequency Doppler Radar). In such a 
radar it is desirable to generate an oscillator signal, the frequency of 
which varies as follows. At first the frequency is to be constant for a 
set period of time. Thereafter the frequency is to increase linearly with 
time up to a maximum frequency. Finally the frequency is to decrease 
linearly with time from the maximum frequency down to the original 
frequency level. 
This frequency variation is utilised in modulating the signal emitted from 
the radar antenna. The emitted signal is reflected against the target and 
is then received by the radar antenna. Through observation of the 
variations in the differences in frequency between the emitted signal and 
the received signal the distance to the target and the speed of the target 
can be obtain. In order to obtain a good accuracy in measurements of the 
target distance and to obtain an effective suppression of ground echoes a 
very good linearity is required as to the frequency variations. 
A further such application is testing equipment for spectral analysis. In 
such an analysis the aim is to establish the spectral composition of a 
test signal. This is accomplished by mixing the test signal with an 
oscillating reference signal, the frequency of which varies linearly with 
time. Through the mixing a new signal is obtained, and this new signal is 
passed through a narrow bandpass filter. 
It can be shown that the output signal from the bandpass filter is a signal 
which instantaneously oscillates with a frequency equal to the difference 
between the reference signal frequency and the centre frequency of the 
bandpass filter. It can also be shown that the output signal from the 
bandpass filter has an instantaneously peak value which is proportional to 
the value of the fourier transform of the test signal at a frequency which 
is equal to the difference between the reference signal frequency and the 
centre frequency of the bandpass filter. 
The value of the Fourier transform of the test signal as a function of the 
frequency may easily be illustrated with the use of an oscilloscope. In 
order to accomplish this, the output signal from the bandpass filter is 
first fed to a peak value detector and thereafter the output signal from 
the peak value detector is fed to the oscilloscope. 
The relationship between the frequency and the control voltage in a voltage 
controlled oscillator is usually not totally linear. This makes it harder 
to generate a frequency sweep, the frequency of which varies linearly with 
time. If the relationship between the control voltage and the frequency 
had been linear the result had been a linear frequency sweep if the 
control voltage fed to the oscillator had been a linearly, with respect to 
time varying voltage, a ramp voltage, and such a ramp voltage is 
relatively easy to generate. 
In order to generate an oscillator signal, the frequency of which varies 
with respect to time in a desired manner, using a voltage controlled 
oscillator, the control voltage thus must vary with time in a rather 
complicated way. The variation in the control voltage is determined partly 
by the desired frequency variations, and partly by the relation between 
frequency and control voltage in the voltage controlled oscillator. 
It is of course possible to measure the correlation between frequency and 
control voltage, and having this correlation construct a control voltage 
generator, which generates a control voltage giving the desired result. In 
order to obtain high performance in e.g. a radar application, a very 
accurate mapping of the correlation between frequency and control voltage 
is required, the accuracy may often be in the order of thousands of 
measurement values. 
A further difficulty is that the correlation between the frequency and the 
control voltage, i.a. is dependent of the ambient temperature. This 
implies that the measurements have to be made over again as the 
temperature changes; alternatively, measurement values have to be stored 
which describe the correlation between frequency and control voltage at a 
great number of temperatures. The first alternative is time consuming. The 
second alternative requires a great quantity of memory, and as the 
correlation between the control voltage and the frequency also changes as 
the oscillator components ages the measurements have to be repeated at 
even time intervals. 
In order to circumvent these problems some form of a closed control 
sequence of the oscillator signal is used. In such a type of control 
sequence, an error indication signal generally is generated, which in some 
way indicates the deviation of the oscillator signal actual variation as 
compared to the desired variation. The error indication signal is used in 
order to generate a correction signal, which in turn is used for modifying 
the control signal fed to the oscillator, such that the oscillator emits 
an oscillator signal, the variation of which, as time passes, corresponds 
to the desired variation. 
Several methods describing how to control the frequency of a frequency 
controlled oscillator such that the frequency vies with time in a desired 
manner are can be found in the patent literature. 
In U.S. Pat. No. 4,129,832 is described that the aim is to accomplish a 
linear frequency variation from a voltage controlled oscillator. This is 
accomplished through the calibration of a control signal. The control 
signal is obtained by successive D/A-conversion of values stored in a RAM 
memory, through integration of the results of the D/A-conversion and by 
feeding the result of the integration in the form of a control signal to 
the voltage controlled oscillator. 
In order to obtain a linear sweep the values in the RAM-memory have to be 
calibrated, which calibration proceeds as follows. The oscillator signal 
is mixed with a time delayed version of itself, whereat the result of this 
mixing is lowpass filtered. The resulting signal obtains a frequency, 
which mainly is proportional to the frequency slope--i.e. the frequency 
change per second--in the oscillator signal. This signal, the frequency of 
which depends on the frequency slope, is compared on an oscilloscope with 
a signal having constant frequency corresponding to the desired frequency 
slope of the sweep. The values in the RAM-memory are manually adjusted 
until the two curves are made to coincide on the screen of the 
oscilloscope. 
One drawback of this method is that the method is manual. In a system 
having high demands on linearity in the frequency variations the 
calibration must be made automatically. 
In U.S. Pat. No. 5,172,123 the aim is to obtain a linear frequency 
variation from a voltage controlled oscillator. the substance described in 
this patent publication may to, some extent, be said to be an automation 
of the method described in the above patent publication. 
A detector signal is generated in similar manner as in the above mentioned 
patent publication, which detector signal frequency is principally 
proportional to the frequency slope of the oscillator signal. The detector 
signal is analysed through a zero crossing detector, which generates a 
square wave, the zero crossings of which are simultaneous to the zero 
crossings of the detector signal. The square wave is used to control two 
counters, which when they are activated counts the number of pulses which 
are generated having a fixed frequency. The number of pulses counted 
becomes an indication of the frequency slope of the oscillator signal. A 
desired value corresponding to the number of pulses that the counter would 
count if the oscillator signal exhibited the desired frequency slope is 
deducted from the number of pulses, whereat a time discrete error 
indication signal is obtained. 
The correction values stored in a RAM-memory are successively D/A-converted 
and the result of the D/A-conversion is integrated. The result of the 
integration is fed, after amplification, as a control signal to the 
voltage controlled oscillator. 
Every value of the time discrete error indication signal is added to a 
corresponding correction value which is derived from the RAM-memory. The 
resultant value of the addition is stored at the location of the 
correction values in the RAM-memory, whereby a new improved correction 
value is obtain, which is used in the next control sequence of the 
frequency variations. 
A drawback of this method is that it demands quite an amount of complicated 
electronics. 
In U.S. Pat. No. 4,647,873 the described aim is to obtain a linear 
frequency variation from a voltage controlled oscillator. For this 
purpose, a circuit is used in controlling phase or frequency of the 
oscillator signal. The circuit comprises a voltage controlled oscillator, 
an error measuring system, an adaptive correction system and a control 
voltage generator. 
The error in regard to phase or in frequency measured by the error 
measurement system is used for correction of the, by the control generator 
generated control voltage, which is used in controlling the oscillator. 
This correction is done partly in a broadband loop, and partly in an 
adaptive loop. The broadband loop is used for correction of fast varying 
random errors and the adaptive loop is used for correction of slowly 
varying non-linearity in the voltage controlled oscillator, e.g. caused by 
temperature variations. 
In the broadband loop the error signal is used directly for correction of 
the control signal, simply by adding the error signal to the control 
signal from the control voltage generator. 
In the adaptive loop the error signal is used according to the following. 
The error signal is sampled at determined points of time and are 
A/D-converted. The correction values, which are retrieved from a 
RAM-memory are partly added to the A/D-converted error signals. This 
process is called "fading memory"-process in the document and is used with 
the aim to obtain a stable system. The result from the "fading 
memory"-process, at a certain sampling point of time, is stored in the 
same memory location as the correction value for the corresponding 
sampling points of time were stored, and are thus used in the process as a 
new correction value for the corresponding sampling point of time in the 
next sweep. 
The correction value which at a given sampling point of time is retrieved 
from the RAM-memory for the use in the "fading memory"-process is also 
used to correct the control signal for the oscillator. The correction 
value is D/A-converted, the result of the conversion of the D/A-conversion 
is lowpass filtered and is thereafter used as a corrections signal which 
is added to the control signal from the control voltage generator. 
This method has a draw-back in that the error signal is based on deviations 
in phase or frequency. In generating a linear frequency variation, a time 
dependent desired value must be generated, which makes higher demands on 
time precision and precision in the error measuring system. 
DESCRIPTION OF THE INVENTION 
The invention is aimed at solving the problem of an effective way of 
supervision and control of a oscillator signal from a controllable 
oscillator, for example a VCO, in such a way that the frequency of the 
oscillator may be made to vary with time in a predetermined way; it must 
especially in this respect be possible to vary the frequency linearly in 
respect of time with high precision. 
The problem may generally be solved according to the following. A 
correction signal is generated. The correction signal is generated in a 
control loop and is used to modify the control signal, which controls the 
oscillator. In the control loop a time discrete representation of a 
secondary phase is generated, the time discrete representation 
representing the secondary phase at a number of points of time. The 
secondary phase is such that the frequency corresponding to the secondary 
phase is equal to the difference between the oscillator signal frequency 
and a constant frequency. In the control loop a time discrete 
approximation signal is generated from the time discrete representation of 
the secondary phase. The signal values of the time discrete approximation 
signal, corresponding to different points of time, represent 
approximations to the second derivative with respect to time (second time 
derivative) of the secondary phase at the different points of time. The 
second time derivative of the secondary phase is related to the actual 
frequency slope of the oscillator signal. The time discrete approximation 
signal thus contains information as to the actual frequency slope of the 
oscillator signal. A time discrete error indication signal is generated in 
dependence of the time discrete approximation signal. The time discrete 
error indication signal indicates the deviation of the oscillator signal 
actual frequency slope as compared to the desired frequency slope. In the 
control loop the correction signal is generated in dependence of the time 
discrete error indication signal. The control loop may be adaptive whereby 
data generated in one control sequence are used in the next control 
sequence. The aim of the invention is thus to use the time discrete 
approximation signal for determining the oscillator signal frequency slope 
and to use a closed-loop control sequence in controlling the oscillator to 
control the oscillator signal frequency, such that this frequency varies 
with time in a predetermined manner, and that the invention comprises 
methods and devices for these purposes. 
This problem is more specifically solved according to the following. In the 
control loop the oscillator signal is quadrature demodulated. In the 
quadrature demodulation of the oscillator signal a first quadrature signal 
is generated having the secondary phase and a second quadrature signal 
having a phase .pi./2 from the secondary phase. The first and the second 
quadrature signal (in-phase and quadrature phase) are A/D-converted at a 
number of points of time, whereby a time discrete first quadrature signal 
(in-phase) and a time discrete second quadrature signal (quadrature phase) 
are generated. The time discrete first quadrature signal combined with the 
time discrete second quadrature signal constitute the time discrete 
representation of the secondary phase. In dependence of the time discrete 
quadrature signals, a time discrete differential signal is generated, 
which represents the first differences of the secondary phase between 
different points of time. In dependence of the time discrete differential 
signal the time discrete approximation signal is generated, such that it 
represents second differences of the secondary phase between different 
points of time. In an adaptive control sequence of the oscillator, a 
number of stored correction values, which are generally generated at 
preceding control sequence of the oscillator have been stored in a memory. 
In dependence of the stored correction values and the time discrete error 
indication signal new correction values are generated in the control loop. 
The generation of the new correction values may herewith be made by 
generating a first and a second time discrete lowpass signal, 
respectively, through time discrete lowpass filtering of the time discrete 
error indication signal and the stored correction values, respectively, 
whereby the new correction values are generated in dependence of the first 
and the second time discrete lowpass signal. The new correction values may 
substitute the stored old correction values. The new correction values are 
D/A-converted and the result of the D/A-conversion constitutes the 
correction signal. 
The invention exhibits, above solving the above stated problem, the 
following advantages. Pro primo, the solution offered to the problem of 
the invention is comparatively simple and provides possibilities for an 
effective and accurate signal treatment. Secondly the invention uses a 
error indication signal, which is based on a deviation in the actual 
frequency slope from a desired frequency slope, which simplifies the 
control of a linear frequency variation, since the set value information 
in such a case is independent of time.

PREFERRED EMBODIMENTS 
In FIG. 1 is shown how an open control sequence for a voltage controlled 
oscillator 1--a VCO--fundamentally may be accomplished according to prior 
art. 
The oscillator 1 in FIG. 1 is controlled by a control signal in the form of 
a voltage varying with time V(t) and generates, in dependence of the 
control signal V(t), an oscillator signal cos(.phi. (t)). Herein .phi.(t) 
represents the phase of the oscillator signal cos(.phi. (t)). The control 
signal V(t) is generated by a control signal generator 3. 
The frequency f(t) of the oscillator signal cos(.phi. (t)) and the 
frequency slope .mu.(t)--being variables, which will be used below are 
defined by the relationships below 
##EQU1## 
The dots designate, as is common usage, derivation in respect of time. 
The oscillator signal cos(.phi. (t)) frequency f(t) is dependent on the 
control signal V(t), and as is mentioned above the need often arises of 
controlling the oscillator 1 in such a manner that its frequency f(t) 
varies with time in some specific desired manner. 
In FIG. 3 is shown, schematically, one example of such a desired frequency 
variation f.sub.D (t). As shown, it is the desire that the frequency f(t) 
should vary linearly with respect to time, of which of course follows, 
that the desired frequency slope is a constant value .mu..sub.0. 
In FIG. 2 is shown, schematically, the correlation between the control 
signal V and the oscillator signal frequency f. For an ideal oscillator 
this correlation should have been totally linear. The oscillator 1 in FIG. 
1 is, however, no ideal oscillator, and the correlation which is shown in 
FIG. 2 is not linear. This poses some problems, as the oscillator 1 must 
be fed a control signal V(t), which will give the desired frequency 
variation f.sub.D (t). 
In FIG. 4 is shown, schematically, the control signal V.sub.D (t) which 
gives the desired f.sub.D (t). The control signal V.sub.D (t) becomes 
rather complicated as it, as indicated in FIG. 4, must be generated both 
with respect to the desired frequency variations f.sub.D (t) and the 
correlation between the control signal V and the frequency f of the 
oscillator 1. 
The correlation between the control signal V and the frequency f is 
influenced, as already mentioned, by the ambient temperature. This makes 
it hard to, with good precision, perform an open control sequence of a 
VCO, which usually leads to the use of some type of closed control 
sequence. In FIG. 5 is shown, in accordance with prior art how such a 
control sequence fundamentally may be performed. 
In FIG. 5 is shown how a voltage controlled oscillator 5 is fed a control 
signal in the form of a voltage V(t). The control voltage V(t) is 
generated by a control signal generator 7, which generator also recieves a 
correction signal K(t). The control signal generator 7 modifies the 
control signal V(t) in dependence of the received correction signal K(t) 
in order to make the oscillator signal frequency vary with time in the 
desired manner f.sub.D (t). 
In FIG. 5 the correction signal K(t) is generated in a control loop 
according to the following. 
The oscillator signal cos(.phi. (t)) is measured, and an error measuring 
system 9 generates a error indication signal e in dependence of the 
measured oscillator signal cos(.phi. (t)). The error indication signal e, 
in FIG. 5, indicates the deviation of in the oscillator signal cos(.phi. 
(t)) of the frequency slope .mu.(t) from the desired frequency slope 
.mu..sub.D. In FIG. 5 the information about the desired frequency slope 
.mu..sub.D may be found in a set value B, which is supplied to the error 
measuring system 9. 
There is also a possibility to use an error indication signal which 
indicates the deviation from a desired phase .phi.(t) or a desired 
frequency f.sub.D (t). 
In FIG. 5 may further be seen, that a error processing system 11 generates 
the correction signal K(t) in dependence of the error indication signal e. 
The error processing system 11 may be realised in different ways. The 
error processing system 11 may in its simplest form more or less directly 
transfer the error indication signal e as a correction signal K(t) to the 
signal generator 7. A more advanced error processing system 11 may be 
adaptive, whereby the system can store data such that the system for each 
time the generation of the desired frequency variation f.sub.D (t) is 
attempted it will succeed better by using the data from earlier attempts. 
The present invention will now be described with reference to FIGS. 6 to 9. 
In FIG. 6 a block diagram discloses a closed and adaptive control sequence 
of the frequency f(t) of a oscillator signal cos(.phi. (t)) a voltage 
controlled oscillator 13 in accordance with the present invention. The 
block diagram in FIG. 6 shall in the first place be seen as descriptive of 
the methodological build up of the invention, even though the block 
diagram naturally in some parts may be considered to indicate possible 
constructive solutions. 
In FIG. 6 the voltage controlled oscillator 13 receives a control signal in 
the form of a varying voltage V(t). The oscillator 13 generates an 
oscillator signal cos(.phi. (t)); the frequency f(t) of which is dependent 
on the received control signal V(t). 
In FIG. 6 the control signal V(t) is generated by adding a voltage U0 (17) 
to a correction signal K(t) , whereby the, by the addition received, 
cumulative signal sum is integrated 15. The result of the integration 15 
is fed to the oscillator 13 as the control signal V(t). The control signal 
V(t) is thus constantly modified at all times in dependence of the 
correction signal K(t). The voltage U0 determines the nominal slope of the 
control signal V(t), and the value of U0 is chosen, such that the control 
signal nominal slope assumes a suitable value. 
In FIG. 7 is shown the desired frequency variation f.sub.D (t). The 
frequency f(t) of the oscillator 13 is to be controlled such that it 
varies linearly with time under a time period .tau.. The desired linear 
frequency variation corresponds f.sub.D (t) to a desired frequency slope 
.mu..sub.D. 
Sometimes the desired frequency variation may be such, that it comprises 
several time periods during which the frequency shall vary linearly with 
respect to time, whereby the desired frequency slope corresponding to the 
different time periods may be different. The present invention can, under 
such circumstances, naturally be used to control the frequency variations 
for a optional number of these time periods. 
In the embodiment according to FIG. 6 the correction signal K(t) thus shall 
be so generated that the control signal V(t) is modified in such a way 
that the desired frequency variation f.sub.D (t) is accomplished. The 
generation of the correction signal is done in a control loop shown i FIG. 
6 which is described below. 
The control loop in FIG. 6 starts with a quadrature demodulation of the 
oscillator signal (cos(.phi. (t)). In the quadrature demodulation of the 
oscillator signal cos(.phi. (t)) a first quadrature signal (I(t)) (the 
in-phase signal) and a second quadrature signal Q(t) (the quadrature phase 
signal) are generated. The phase displacement of the two quadrature 
signals I(t) and Q(t) are separated by .pi./2 . 
The quadrature demodulation in FIG. 6 begins by measuring the oscillator 
signal cos(.phi. (t)). In FIG. 6 is indicated that the measurement is 
performed by means of a coupler 19, but the measurement may of course be 
performed by other means. 
The quadrature demodulation of the oscillator signal cos(.phi. (t)) in FIG. 
6 continues in that the oscillator signal cos(.phi. (t)) is separated into 
a first part and a second part, the first part and the second part being 
of equal size. 
The in-phase signal I(t) is generated by multiplying 21 the first part of 
the measured oscillator signal cos(.phi. (t)) with a first harmonically 
oscillating signal 2cos(.omega..sub.0 t) whereat the result of the 
multiplication 21 is lowpass filtered 27. The first harmonically 
oscillating signal 2cos(.omega..sub.0 t) oscillates with a constant 
frequency .omega..sub.0 /2.pi.. 
The quadrature phase signal Q(t) is generated in a similar manner by 
multiplying 23 the second part of the measured oscillator signal cos(.phi. 
(t)) with a second harmonically oscillating signal -2sin(.omega..sub.0 t) 
whereat the result of the multiplication is lowpass filtered 25. The 
second harmonically oscillating signal -2sin(.omega..sub.0 t) exhibits the 
same peak values and same frequency as the first harmonically oscillating 
signal 2cos(.omega..sub.0 t); the phase of the second harmonically 
oscillating signal -2sin(.omega..sub.0 t) lies however .pi./2 ahead of the 
phase of the first harmonically oscillating signal 2cos(.omega..sub.0 t). 
In FIG. 6 the first and the second harmonically oscillating signals are 
indicated as 2cos(.omega..sub.0 t) and -2sin(.omega..sub.0 t). The peak 
value two and the absolute phase displacement have been chosen in order to 
simplify the description, and these choices should not be understood as 
indicative of the limitations of the invention. 
In FIG. 7 the desired frequency variation f.sub.D (t) on a relatively high 
basic level f.sub.b. The changes in the frequency f(t) of the oscillator 
signal cos(.phi.(t)) which arises during the control sequence are small as 
compared to this basic level f.sub.b. The frequency .omega..sub.0 /2.pi. 
of the first and the second harmonically oscillating signal 
2cos(.omega..sub.0 t) and -2sin(.omega..sub.0 t) has been chosen such that 
it is of the same order as the basic level f.sub.b. Against this 
background it has become practical to reformulate the oscillator signal 
phase accordingly 
EQU .phi.(t)=.omega..sub.0 t+.theta.(t) (3) 
This introduces a secondary phase .theta.(t). For this secondary phase the 
frequency .theta.(t)/2.pi. corresponding to the secondary phase .theta.(t) 
equals the difference between the oscillator frequency f(t) and the 
constant frequency .omega..sub.0 /2.pi. of the first and the second 
harmonically oscillating signal 2cos(.omega..sub.0 t) and 
-2sin(.omega..sub.0 t). It is immediately evident that 
EQU .theta.(t)=.phi.(t) (4) 
Accordingly to what was said previously the following must also be valid 
EQU .vertline..theta.(t).vertline.&lt;&lt;.omega..sub.0 (5) 
When the first part of the measured oscillator signal cos(.theta. (t)) is 
multiplied 21 with the first harmonic oscillating signal 
2cos(.omega..sub.0 t) the following is given, using the equation (3), 
EQU 2cos(.omega..sub.0 t) cos(.omega..sub.0 
t+.theta.)=cos(.theta.)+cos(.theta.)cos(2.omega..sub.0 
t)-sin(.theta.)sin(2.omega..sub.0 t) (6) 
When the second part of the measured oscillator signal cos(.theta. (t)) is 
multiplied 23 with the second harmonic oscillating signal 
-2sin(.omega..sub.0 t) the following is correspondingly given 
EQU -2sin(.omega..sub.0 t) cos(.omega..sub.0 
t+.theta.)=sin(.theta.)-cos(.theta.)sin(2.omega..sub.0 
t)-sin(.theta.)cos(2.omega..sub.0 t) (7) 
When lowpass filtering 25 and 27 the signals according to the equations (6) 
and (7) the fastes varying terms are filtered off, and essentially only 
the first terms in the right side of the equations (6) and (7) will 
remain. The in-phase signal I(t) and the quadrature phase signal Q(t) can 
thus be written 
EQU I(t)=cos(.theta. (t)) (8) 
EQU Q(t)=sin(.theta. (t)) (9) 
The phase of the in-phase signal I(t) is, as can be seen, the secondary 
phase .theta.(t). The in-phase signal I(t) and the quadrature phase signal 
Q(t) exhibit the same frequency, but the phase of the quadrature phase 
signal Q(t) is separated by .pi./2 from the phase of the in-phase signal 
I(t). 
The control loop in FIG. 6 continues with A/D-conversion of the in-phase 
signal I(t) and of the quadrature phase signal Q(t) 31 and 29. 
The A/D-conversion 31 and 29 occurs successively at a number of points of 
time t.sub.k ; k=0, 1, , . . . , N during the time period .tau. during 
which the control sequence of the oscillator frequency f(t) is intended to 
be performed. In FIG. 7 some of the points of time t.sub.k are indicated. 
From FIG. 7 is evident that that the first time position t.sub.0 
essentially coincides with the start of the time period .tau. and that the 
last point of time t.sub.N essentially coincides with the end of the time 
period .tau.. From FIG. 7 also is evident that the time difference between 
two consecutive points of time, of the in FIG. 6 shown embodiment, is a 
constant time interval T. 
In the description below a number of time discrete signals are told of; 
when referring to these time discrete signals the subscript k is used and 
when referring to any signal value of these time discrete signals the 
subscript n or some other subscript is used. 
The A/D-conversion 31 of the in-phase signal I(t) gives rise to a time 
discrete in-phase signal I.sub.k ; k=0, 1, . . . , N being generated and 
in the corresponding manner the A/D-conversion of the quadrature phase 
signal Q(t) gives rise to a time discrete quadrature phase signal Q.sub.k 
; k=0, 1, . . . , N being generated. Using the equations (8) and (9) one 
will find that at a given point of time t.sub.n the signal value I.sub.n 
of the time discrete in-phase signal and the signal value Q.sub.n of the 
time discrete quadrature phase signal corresponding to the point of time 
t.sub.n may be written 
EQU I.sub.n =cos(.theta..sub.n) (10) 
EQU Q.sub.n =sin(.theta..sub.n) (11) 
where 
EQU .theta..sub.n .tbd..theta.(t.sub.n) (12) 
At a given point of time t.sub.n the signal value I.sub.n of the time 
discrete in-phase signal in combination with the signal value Q.sub.n of 
the time discrete quadrature phase signal constitute a representation of 
the secondary phase .theta..sub.n at the point of time t.sub.n. That the 
two signal values I.sub.n and Q.sub.n constitute a representation of the 
secondary phase .theta..sub.n is here taken to mean that the secondary 
phase .theta..sub.n may be calculated directly (except for an arbitrary 
multiple of 2.pi.) from the signal values I.sub.n and Q.sub.n. 
The quadrature demodulation of the measured oscillator signal cos(.phi.(t)) 
thus renders a double profit; on one hand signals I(t) and Q(t) with 
slower variations with time are obtained, which simplifies the 
A/D-conversions 31 and 29, and on the other hand a direct representation 
of the secondary phase .theta..sub.n is obtained. 
In FIG. 6 the control loop continues by generating a first time discrete 
complex signal X.sub.k ; k=0, 1, . . ., N. Corresponding to a given point 
of time t.sub.n the signal value X.sub.n is generated, such that its real 
part corresponds to the signal value I.sub.n of the first time discrete 
in-phase signal corresponding to the point of time t.sub.n, and such that 
its imaginary part corresponds to the signal value Q.sub.n of the time 
discrete quadrature phase signal corresponding to the point of time 
t.sub.n, i.e. 
EQU X.sub.n .tbd.I.sub.n +jQ.sub.n =e.sup.j.theta.n (13) 
Above j symbolises the imaginary unit. 
The control loop in FIG. 6 continues by generating a time discrete 
differential signal Y.sub.k ; k=0, 1, . . . , N. Corresponding to a given 
point of time tn the signal signal value Y.sub.n of the time discrete 
differential signal is generated, such that it corresponds to the value of 
a multiplication 35 of the signal value X.sub.n of first time discrete 
complex signal, corresponding to the point of time t.sub.n, and the signal 
value X*.sub.n-1 of the complex conjugate of the first time discrete 
complex signal at the immediately preceding point of time t.sub.n-1. The * 
signifies the complex conjugation. 
In FIG. 6 is indicated that in generation of the time discrete differential 
signal Y.sub.k a first delay block 33 is used. This should be understood 
such, that at a given point of time t.sub.n the signal value X.sub.n-1 of 
the first time discrete complex signal corresponding to the last preceding 
point of time t.sub.n-1 has been delayed, e.g. by being stored in a 
memory. The stored signal value X.sub.n-1 is complex conjugated and 
multiplied 35 with the first time discrete complex signal's recently 
generated signal value X.sub.n. After the multiplication 35 the new signal 
value X.sub.n is stored and is thus delayed such that it may be used at 
the next point of time t.sub.n+1. The first delay block 33 indicates in 
this case that the signal value X.sub.n preferably is stored in the memory 
space where the signal value X.sub.n-1 previously was stored. 
As regards the generation of the signal value Y.sub.0 of the time discrete 
differential signal corresponding to the first point of time t.sub.0 there 
is no earlier signal value X.sub.-1 to use, as such a signal value never 
has been generated. X.sub.-1 is therefore a predetermined starting value. 
In the embodiment depicted in FIG. 6 this starting value X.sub.-1 has been 
set to zero. The starting value X.sub.-1 is thus stored in the memory 
space indicated by the first delay block 33 already when the control 
sequence is begun. The starting value X.sub.-1 is of course only a fictive 
auxiliary value corresponding to a likewise fictive auxiliary point of 
time t.sub.-1. 
Accordingly the signal value Y.sub.n of the time discrete differential 
signal corresponding to a given point of time t.sub.n may be written 
EQU Y.sub.n .tbd.X.sub.n X'.sub.n-1 =e.sup.j(.theta.n-.theta.n-1) 
=e.sup.j.DELTA.-.theta.n (14) 
In the above equation a difference operator .DELTA..sup.- has been 
introduced, which here designates the back-difference according to 
EQU .DELTA..sup.- .theta..sub.n .tbd..theta..sub.n -.theta..sub.n-1(15) 
The control loop in FIG. 6 continues by generating a time discrete 
approximation signal Z.sub.k ; k=0, 1, . . . , N. Corresponding to a given 
point of time t.sub.n the signal value Z.sub.n of the time discrete 
approximation signal is generated such that it corresponds to the value of 
a multiplication 39 between the signal value Y.sub.n of the time discrete 
differential signal corresponding to the point of time t.sub.n and the 
signal value Y'.sub.n-1 , of the complex conjugate of the time discrete 
differential signal corresponding to the immediately preceding point of 
time t.sub.n-1. 
In FIG. 6 is indicated that in the generation av the time discrete 
approximation signal Z.sub.k a second delay block 37 is used. This should 
be interpreted in the same manner as for the first delay block 33. 
As regards the generation of the signal value Z.sub.0 of the time discrete 
difference signal corresponding to the first point of time t.sub.0 there 
is no earlier signal value Y.sub.-1 to use, as such a signal value never 
has been generated. Y.sub.-1 is therefore a predetermined starting value. 
In the embodiment depicted in FIG. 6 this starting value Y.sub.-1 has been 
set to zero. The starting value is thus stored in the memory space 
indicated by the first delay block 33 already when the control sequence is 
begun. The starting value Y.sub.-1 is of course only a fictive auxiliary 
value corresponding to a likewise fictive auxiliary point of time 
t.sub.-1. 
Accordingly the signal value Z.sub.n of the time discrete difference signal 
corresponding to a given point of time t.sub.n may be written 
EQU Z.sub.n .tbd.Y.sub.n Y'.sub.n-1 
=e.sup.j(.DELTA.-.theta.n-.DELTA.-.theta.n-1) 
=e.sup.j.DELTA.-(.DELTA.-.theta.n) =e.sup.j(.DELTA.-)2.theta.n(16) 
The time discrete approximation signal Z.sub.k gives information about the 
actual frequency slope .mu.(t) of the oscillator signal cos(.phi.(t)), 
exactly how this is accomplished will be thoroughly explained further 
below in the description. 
The control loop in FIG. 6 continues by generating another time discrete 
complex signal E.sub.k ; k=0,1, . . . , N. The signal value E.sub.n of the 
second time discrete complex signal corresponding to a given point of time 
t.sub.n is generated such that it correspond to the value of a 
multiplication 41 between the signal value Z.sub.n of the time discrete 
approximation signal corresponding to the point of time t.sub.n and a 
complex set value b=e.sup.-j2.pi.T2.mu.D. From this follows that the 
signal value E.sub.n of the second time discrete complex signal 
corresponding to a given point of time may be written 
EQU E.sub.n =Z.sub.n b=e.sup.j((.DELTA.-)2.theta.n-2.pi.T2.mu.D)(17) 
In FIG. 6 the control loop continues by generating a time discrete error 
indication signal e.sub.k ; k=0,1, . . . , N. The signal value e.sub.n of 
the time discrete error indication signal corresponding to a given point 
of time t.sub.n is generated such that it corresponds to the value of the 
multiplication 45 between the imaginary part Im{E.sub.n } 43 of the signal 
value E.sub.n of the second time discrete complex signal corresponding to 
the point of time t.sub.n and a real loop constant .alpha.. The signal 
value e.sub.n of the time discrete error indication signal corresponding 
to a given point of time t.sub.n can thus be written 
EQU e.sub.n .tbd..alpha.Im{E.sub.n }=.alpha.sin((.DELTA..sup.-).sup.2 
.theta..sub.n -2.pi.T.sup.2 .mu..sub.D (18) 
The loop constant .alpha. has been introduce on account of the 
non-linearity of the oscillator characteristics. The loop constant .alpha. 
is needed to stabilise the control loop. The value of the loop constant 
.alpha. is determined essentially by the oscillator 13 maximum frequency 
amplification, a fact which is well-known in the art. 
Using the approximations 
##EQU2## 
and sin(x).apprxeq.x, (when x is small) it is further found that 
EQU .alpha.sin((.DELTA..sup.-).sup.2 .theta..sub.n -2.pi.T.sup.2 
.mu..sub.D).apprxeq..alpha.(T.sup.2 .theta.(t.sub.n)-2.pi.T.sup.2 
.mu..sub.D)=2.pi..alpha.T.sup.2 (.mu.(t.sub.n)-.mu..sub.D (19) 
The signal value e.sub.n of the time discrete error signal indicates, as is 
shown, the deviation in the frequency slope .mu.(t.sub.n) in the 
oscillator signal cos(.phi. (t)) at the given time position t.sub.n from 
the desired frequency slope .mu..sub.D. 
The control sequence of the oscillator 13 in FIG. 6 is adaptive. This means 
that there are stored correction values K.sub.k ; k=0,1, . . . , N 
corresponding to each one of the points of time t.sub.k. The correction 
values K.sub.k are stored in a memory, and in FIG. 6 is indicated that 
this memory for example comprises a RAM-memory 47. 
If it is the first time the control sequence of the frequency variations 
f(t) is run, the stored correction values K.sub.k, are when the control 
sequence starts set to predetermined start values, they may e.g. be set to 
zero. 
If it is not the first time the control sequence is run of the frequency 
variations f(t), then the stored corrections values K.sub.k, when the 
control sequence starts, has values which adaptively have been improved 
during earlier control sequences of the frequency variations f(t) of the 
oscillator 13. How the adaptive improvement of the stored corrections 
values K.sub.k is done will be understood from the description below. 
In FIG. 6 the control sequence is continued by the correction signal K(t) 
being generated. The correction signal K(t) is generated in dependence of 
on the one hand the time discrete error signal e.sub.k, and on the other 
hand the stored correction values K.sub.k. 
In order to generate the correction signal K(t) a first time discrete low 
pass signal LP1(e.sub.k); k=0,1, . . . , N and a second time discrete low 
pass signal LP2(K.sub.k); k=0,1, . . . , N are generated. 
The signal value LP1(e.sub.n) of the first time discrete low pass signal 
corresponding to a given point of time t.sub.n is generated by a time 
discrete lowpass filtering 49 of the time discrete error indication signal 
e.sub.k at the given point of time t.sub.n. 
The lowpass filtering 49 of the time discrete error indication signal 
e.sub.k comprised in FIG. 6 may be a FIR-filtering (Finite Impulse 
Response filter) corresponding to a first time discrete impuls response 
function .sup.1 h.sub.k ; k=0,1, . . . m1. The signal value LP1(e.sub.n) 
of the first time discrete lowpass signal, corresponding to a given point 
of time t.sub.n, may in such a case be written 
##EQU3## 
The signal value LP2(K.sub.n) of the second time discrete lowpass signal 
corresponding to a given point of time t.sub.n is generated by a time 
discrete lowpass filtering 51 of the stored correction values K.sub.k at 
the given point of time t.sub.n --the stored correction values K.sub.k is 
naturally understood in this connection as a stored time discrete signal. 
The lowpass filtering 51 of the stored correction values K.sub.k is in FIG. 
6, e.g. a FIR-filtering corresponding to a second time discrete impulse 
response function .sup.2 h.sub.k ; k=-m2, -m2+1, . . . , m3-1, m3. The 
signal value LP2(K.sub.n) of the second time discrete lowpass signal, 
corresponding to a given point of time t.sub.n, may in such a case be 
written 
##EQU4## 
How the two time discrete impuls response functions .sup.1 h.sub.k and 
.sup.2 h.sub.k should be chosen in order to achieve the lowpass filtering 
effect is a fact known to the man skilled in the art. 
The lowpass filtering 49 which is used in generating the first time 
discrete lowpass signal LP1(e.sub.k) i. a. a causal FIR-filtering. The 
reason for this is that the filtering must be done in real time. As to the 
generation of the second lowpass signal LP(K.sub.k) there is no claim that 
the filtering 51 should be such that it works in real time, since the 
stored correction values K.sub.k corresponding to all points of time 
t.sub.k already are available. The lowpass filtering 51 of the second time 
discrete lowpass signal LP2(K.sub.k) may therefore be a non-causal 
FIR-filtering, as is indicated in the equation (21). 
When it comes to the generation of the second time discrete lowpass signal 
LP2(K.sub.k) one may, of course, use other types of lowpass filtering 51, 
which are not usable in real time, e.g. a FFT-based filtering (Fast 
Fourier Transform). 
Corresponding to a given point of time t.sub.n a new correction value 
K.sub.n is generated i FIG. 6. The new correction value K.sub.n 
corresponding the point of time t.sub.n is generated such that it equals 
the value of an addition 53 of the first time discrete lowpass signal 
value LP1(e.sub.n) and the second time discrete lowpass signal value 
LP2(K.sub.n) corresponding to the point of time t.sub.n. That is 
EQU K.sub.n =LP1(e.sub.n)+LP2(K.sub.n) (22) 
In FIG. 6 the new correction value K.sub.n, corresponding to a given point 
of time t.sub.n are stored in the memory space in the RAM-memory where the 
stored correction values K.sub.n corresponding to the point of time is 
stored. The stored correction value K.sub.n is overwritten by the new 
correction value K.sub.n, which should be understood such that the stored 
correction value K.sub.n corresponding to the time position t.sub.n is 
changed to the new correction value K.sub.n. 
The new correction values K.sub.k (k=0,1, . . . , N) are succesively 
D/A-converted 55. The result of the D/A-conversion 55 in FIG. 6 
constitutes the correction signal K(t). 
In the above description all of the control loop has been described. The 
functional mode of the invention will now be explained in more detail and 
generalised. 
In the control loop the time discrete error indication signal e.sub.k is 
generated. The time discrete error indication signal e.sub.k indicates the 
deviation of the real frequency slope .mu.(t.sub.k) of the osillator 
signal from the desired frequency slope .mu..sub.D. 
The new correction values K.sub.k are generated in dependence of the time 
discrete error indication signal e.sub.k and the stored correction values 
K.sub.k. In generating the new correction values K.sub.k, time discrete 
lowpass filterings 49 and 51 are performed of the time discrete error 
indication signal e.sub.k and the stored correction values K.sub.k. The 
reason for having to perform these lowpass filterings 49 and 51 lies in 
the stability characteristics of the control loop. As is well-known to the 
man skilled in the art, the stability characteristics of a control loop 
depends generally on the loop bandwidth and the loop amplification. In 
generating the time discrete error indication signals e.sub.k, two delays 
are used, indicated by the two delay blocks 33 and 37. This gives rise to 
a specific stability problem. 
On account of the two delays 33 and 37 and the fact that there are always 
components in the control loop which generate noise, more noise would be 
generated for each round in the control loop than was being removed 
therefrom, and this would also be true even if the time discrete error 
indication signal e.sub.k was forcefully lowpass filtered. The result is 
that the stored corrections values K.sub.k would contain more noise for 
each turn of the control loop. In order to get around this, two lowpass 
filterings 49 and 51 are performed and the noise is thereby restricted, 
and the control loop stabilised. 
The correction signal K(t) is generated by an D/A-conversion 55 of the new 
correction signal K.sub.k. The control signal V(t) is generated in 
dependence of the correction signal K(t). In the manner (shown in FIG. 6) 
in which the control signal V(t) is generated, the correction signal K(t) 
will influence the slope of the control signal V(t). Generation of the 
control signal V(t) may, however, be performed in other ways. The 
correction signal K(t) could, e.g., be added after the integration 15, the 
correction signal would then act directly on the value of the control 
signal V(t). The generation of the control signal V(t) in dependence of 
the correction signal K(t) may of course be performed according to any 
other know manner. 
The manner in which the invention generates the time discrete error 
indication signal e.sub.k deserves further attention. The signal value 
Z.sub.n of the time discrete approximation signal, corresponding to a 
given point of time t.sub.n, represents as may be seen from the equation 
(16), a second difference (.DELTA..sup.-).sup.2 .theta..sub.n and may thus 
be said to represent an approximation 
##EQU5## 
of the second derivative of the secondary phase in respect of time 
.theta.(t.sub.n) at the point of time t.sub.n. The term time discrete 
approximation signal signifies here generally a time discrete signal, the 
signal value of which corresponding to a given point of time t.sub.n 
represents an approximation of the secondary phase second time derivative 
.theta.(t.sub.n) at the given point of time t.sub.n. 
The secondary phase .theta.(t) second time derivative .theta.(t) is equal 
to the second derivative with respect to time .phi.(t) of the phase of the 
oscillator signal .phi.(t). The second derivative with respect to time 
.phi.(t) of the phase of the oscillator signal .phi.(t) is related, 
according to the equation (2), directly to the frequency slope .mu.(t) of 
the oscillator signal cos(.phi.(t)). The signal value Z.sub.n of the time 
discrete approximation signal at a given point of time t.sub.n, thus 
establishes the frequency slope .mu.(t.sub.n) of the oscillator signal 
cos(.phi.(t)) at the given point of time t.sub.n. When the information 
about the oscillator signal frequency slope .mu.(t) thus has been acquired 
via the time discrete approximation signal Z.sub.k, then having knowledge 
of the desired frequency slope .mu..sub.D, it is simple to generate the 
time discrete error indication signal e.sub.k in dependence of the time 
discrete approximation signal Z.sub.k. 
The time discrete approximation signal Z.sub.k is generated in dependence 
of the time discrete in-phase signal I.sub.k and the time discrete 
quadrature phase signal Q.sub.k In FIG. 6 this is done by first generating 
the time discrete differential signal Y.sub.k. The time discrete 
difference signal Y.sub.k is such that the signal value Y.sub.n of the 
same, corresponding to a given point of time t.sub.n, represents a 
difference .DELTA..sup.- .theta..sub.n between the secondary phase 
.theta..sub.n at the given point of time t.sub.n and the secondary phase 
.theta..sub.n-1 at the immediately preceding point of time t.sub.n-1, 
which most easily may be seen by looking at the equations (14). The time 
discrete approximation signal Z.sub.k may then be generated in dependence 
of the time discrete difference signal Y.sub.k. 
The method according to which the approximation signal Z.sub.k, in FIG. 6, 
is generated in dependence of the time discrete in-phase signal I.sub.k, 
and the time discrete quadrature phase signal Q.sub.k is a preferred 
method. The invention should, however, not be viewed as being limited to 
this method of generating the approximation signal Z.sub.k in dependence 
of the time discrete in-phase signal I.sub.k, and the time discrete 
quadrature phase signal Q.sub.k, the generation of said signal may of 
course be performed by some other method. 
One further such method is to calculate the secondary phase .theta..sub.k 
at the points of time t.sub.k from the representation of the secondary 
phase .theta., which is given by the time discrete in-phase signal I.sub.k 
and the time discrete quadrature phase signal Q.sub.k. In such a 
calculation it should be remembered that the secondary phase .theta.(t), 
as defined here is derivable and continuous with respect to time. A man 
skilled in the art, however, has no problem to construct a calculation 
algorithm for the calculation of the secondary phase .theta..sub.k at the 
points of time t.sub.k, which is consistent with the definition of the 
secondary phase0. 
When the secondary phase .theta..sub.k has been calculated for different 
points of time t.sub.k it is a simple task to, from theses calculations, 
determine an approximation of the secondary phase second time derivative 
.theta.(t.sub.n) for different points of time t.sub.n. This may e.g. be 
performed in a way similar to the one in FIG. 6, through a direct 
generation of a time discrete signal of first differences .DELTA..sup.- 
.theta..sub.k ; k=0,1, . . . , N through a direct generation of a time 
discrete signal of second differences (.DELTA..sup.-).sup.2 .theta..sub.k 
; k=0,1, . . . , N in dependence of the time discrete signal of first 
differences .DELTA..sup.- .theta..sub.k. The time discrete signal of 
second differences .DELTA..sup.-).sup.2 .theta..sub.k k are in this 
respect preferably used as time discrete approximation signal Z.sub.k. 
The approximations to the secondary phase second derivative in respect of 
time .theta.(t) may of course be calculated from the calculations of the 
secondary phase .theta..sub.k at different points of time t.sub.k, in a 
more sophisticated manner--here is referred to the mathematical 
literature, e.g. the theory of series expansion. 
The manner, according to the present invention, in which the frequency 
slope .mu.(t) is determined--i.e. the forming of a time discrete 
approximation signal Z.sub.k --may of course be used in other contexts 
where a wish exists to determine the frequency slope .mu.(t) of an 
oscillator signal, e.g. in other methods for controlling a controllable 
oscillator. 
The method which is illustrated in FIG. 6 may with certain modifications be 
used in controlling a controllable oscillator when the desired frequency 
variation f.sub.D (t) is non-linear. In such a case, the desired frequency 
slope .mu..sub.D (t) will vary in respect of time, which will have the 
result that the set value information, which is used in the control 
sequence also must vary with time. Thus, in order to use the method in 
FIG. 6 a time discrete complex set value signal b.sub.k 
=e.sup.-j2.pi.T2.mu.D(tk) corresponding to the different points of time 
t.sub.k, instead of the complex set value b--in other respects the method 
may be used unchanged. As the set value information varies with time 
higher demands are made on the time accuracy and the precision of the 
system. 
In FIG. 8 the method steps involved in the control sequence according to 
FIG. 6 are summarise from a general point of view. 
The first step 61 in FIG. 8 is the generation of a control signal V(t) for 
the control of the controllable oscillator 13. 
The second step 63 in FIG. 8 is the generation of a time discrete 
representation I.sub.k and Q.sub.k of a secondary phase .theta.(t) from an 
oscillator signal cos(.phi. (t)). For the secondary phase .theta.(t) it 
shall be valid that the frequency .theta.(t)/2.pi. corresponding to the 
secondary phase .theta.(t) is equal to the difference between the 
oscillator signal frequency f(t) and a constant frequency .omega..sub.G 
/2.pi.. 
The third step 65 in FIG. 8 is the generation of a time discrete 
approximation signal Z.sub.k in dependence of the time discrete 
representation I.sub.k and Q.sub.k of the secondary phase .theta.(t). The 
time discrete approximation signal Z.sub.k is here generated such that it 
gives information about the real frequency slope .mu.(t) of the oscillator 
signal cos(.phi.(t)). 
The fourth step 67 in FIG. 8 is the generation of a time discrete error 
indication signal e.sub.k in dependence of the time discrete approximation 
signal Z.sub.k. The time discrete error indication signal e.sub.k is here 
generated, such that it indicates the difference between the real .mu.(t) 
and the desired .mu..sub.D frequency slope of the oscillator signal 
cos(.phi. (t)). 
The fifth step 69 in FIG. 8 is the generation of the correction signal K(t) 
in dependence of the time discrete error indication signal e.sub.k. The 
correction signal K(t) is used to modify the control signal V(t) 
In FIG. 9 a device is shown for generation of an oscillator signal having a 
predetermined frequency variation f.sub.D (t). The device in FIG. 9 may be 
used for generating under a time period .tau., one in respect of time 
linear frequency variation, and the control method according to the 
invention may preferably be used in this device. 
The device shown in FIG. 9 comprises a controllable oscillator 71, and in 
FIG. 9 this oscillator 71 is a VCO. The controllable oscillator comprises 
a control signal input 75, through which the controllable oscillator 71 
receives a control signal V(t), and an oscillator signal output 73, 
through which the controllable oscillator 71 emits an oscillator signal 
cos(.phi.(t)). The oscillator signal frequency f(t) is here dependent of 
the control signal V(t). 
The control signal V(t) is generated in FIG. 9 by a control signal 
generator 77. The control signal generator 77 comprises a control signal 
output 79 and a correction signal input 81. The control signal generator 
77 is adapted to emit the control signal V(t) in dependence of a received 
correction signal K(t), received through a correction signal input 81. The 
control signal output 79 is connected to the control signal input 75. In 
the control signal generator 77 the correction signal K(t) and a voltage 
U0 are fed to the signal inputs of an adder 83. The signal output of the 
adder 83 is coupled to an integrator 85 and the control signal V(t) is in 
this case the output signal from the integrator 85. The voltage U0 
establishes the nominal slope of the control signal V(t), and the value of 
the voltage U0 is so chosen that the nominal voltage of the control signal 
receives an adequate value. This construction of the control signal 
generator 77 should only be interpreted as one possibility, and the 
control signal generator 77 may of course be designed in any other 
possible manner, e.g. in the manner described in the known technology. 
The correction signal K(t), which is fed to the control signal generator 
77, is generated in FIG. 9 in a control loop as stated below. 
The control loop in FIG. 9 comprises means for quadrature demodulation of 
the oscillator signal cos(.phi. (t)). A measurement means 87 for measuring 
of the oscillator signal cos(.phi.(t)) is coupled to the oscillator signal 
output 73. In FIG. 9 the measurement means 87 is a coupler. A signal input 
91 in a quadrature demodulator 89 is coupled to the measurement means 87. 
The quadrature demodulator 89 is adapted to quadrature demodulate the 
signal fed to the signal input 91 in respect of a constant frequency 
.omega..sub.0 /2.pi. and to emit via a first and second quadrature signal 
output 93 and 95 an in-phase signal I(t) and a quadrature phase signal 
Q(t) respectively. The in-phase signal exhibits a secondary phase and the 
quadrature phase signal Q(t) exhibits a phase that lies .pi./2 from the 
secondary phase .theta.(t). The construction of a quadrature demodulator 
is is known to the man skilled in the art. 
The control loop in FIG. 9 further comprises a control unit 97 and one to 
the control unit coupled analogue indata/outdata-unit 99. The control unit 
97 comprises processor means, memory means and clock means. The control 
unit 97 further comprises program, stored in the memory means, for control 
of the work of the control unit 97. The indata/outdata-unit 99 comprises a 
first and a second analogue indata port 103 and 101, and analogue outdata 
port 105. The indata/outdata-unit 99 further comprises A/D-converters 
connected to the indata port 101 and 103 and a D/A-converter connected to 
outdata port 105. The control unit 97 is connected to the 
indata/outdata-unit 99 in such a way that that it may control where and 
when indata and outdata is to be obtained and to be emitted, respectively. 
In FIG. 9 the first and the second indata port 101 and 103, respectively, 
are connected to the in-phase signal and the quadrature phase signal 
output 93 and 95, respectively, and the outdata port 105 is connected to 
the correction signal input 81. 
The device in FIG. 9 may be used for obtaining a oscillator signal 
cos(.phi. (t)) having a predetermined frequency variation f.sub.D (t); the 
device may specifically be used for obtaining a linear frequency 
variation, wherein any of the control sequences described according to the 
present invention may be used. 
The control unit 97 may thus in combination with the indata/outdata unit 99 
accomplish an A/D-conversion of the quadrature signals I(t) and Q(t) at a 
number of points of time t.sub.k, whereby time discrete quadrature signals 
I.sub.k and Q.sub.k are obtained. 
On the basis of the time discrete quadrature signals I.sub.k and Q.sub.k 
the control unit may further generate a time discrete approximation signal 
Z.sub.k, indicating the real frequency slope .mu.(t) of the oscillator 
signal cos(.phi. (t)). The control unit 97 further generates in dependence 
of the time discrete approximation signal Z.sub.k a time discrete error 
indication signal e.sub.k, indicating the deviation of the oscillator 
signal cos(.phi. (t)) actual frequency slope .mu.(t) from the desired 
frequency slope .mu..sub.D, corresponding to the desired linear frequency 
variation f.sub.D (t). The control unit 97 in combination with the 
indata/outdata port 99 may further in dependence of the time discrete 
error indication signal e.sub.k generate a correction signal K(t), which 
is emitted through the outdata port 105. The generation of the correction 
signal K(t) may be adaptive, whereby the stored correction values are 
stored in the control unit 97 memory means. 
The manner in which the device in FIG. 9 is adapted to determine the 
oscillator frequency slope-i.e. by forming a time discrete approximation 
signal Z.sub.k --may of course be used in other devices where it is 
desirable to determine the frequency slope of an oscillator signal, e.g. 
in other types of devices for the control of a controllable oscillator.