System and method for controlling a position sensorless permanent magnet motor

A position sensorless interior permanent magnet drive system and methods for use in electric and hybrid electric vehicles. The interior permanent magnet drive system and method use two rotor position estimation techniques for low and high speed operation and an initial rotor magnet polarity detection technique used at stand-still conditions. The use of the two different rotor position estimation techniques in combination enhances system efficiency and accuracy.

BACKGROUND 
The present invention relates generally to electric and hybrid electric 
vehicles, and more particularly, to an improved permanent magnet motor 
control system for use in electric and hybrid electric vehicles. 
The assignee of the present invention designs and develops electric and 
hybrid electric vehicles. In recent years various techniques have been 
published in the literature that address position sensorless operation of 
permanent magnet synchronous motors. Most of these techniques are based on 
either a fundamental component voltage model of the motor or on the 
spatial inductance of the salient rotor motor. 
Permanent magnet synchronous motors have been considered for electric and 
hybrid electric vehicle applications because of their high torque to 
inertia ratio, superior power density and high efficiency. For high 
performance control applications, permanent magnet drive systems require 
absolute rotor position and speed signals which can be obtained from rotor 
position sensors such as resolvers and hall effect sensors mechanically 
coupled to the rotor shaft of the motor. This coupling as well as 
electrical connectors and signal wires from the sensor to the controller 
reduce mechanical robustness of the overall system. This reduction in 
mechanical robustness and cost of the sensors and electrical interface 
make elimination of these devices very desirable. 
The conventional control techniques described above have limited 
operational range. The first technique, based on the fundamental component 
voltage model, cannot be used at or near zero speed operation because it 
requires integration of the stator voltages which are zero at stand-still 
and are very small near zero speed. This will cause oscillatory torque 
during start-up. The second technique based on the spatial information is 
not efficient in high speed operation because it requires high frequency 
injection in order to realize the absolute rotor position. This limits the 
available DC link voltage, which is not desirable in electric and hybrid 
electric vehicle applications. 
It would therefore be desirable to have an improved permanent magnet motor 
control system for use in electric and hybrid electric vehicles, and the 
like. 
SUMMARY OF THE INVENTION 
The present invention provides an improved position sensorless approach for 
implementing an interior permanent magnet drive system for use in electric 
and hybrid electric vehicle applications. The interior permanent magnet 
drive system uses a combination of two rotor position estimation 
techniques for low and high speed operation in addition to initial rotor 
magnet polarity detection at stand-still conditions. Combining the two 
different rotor position estimation techniques enhances system efficiency 
and accuracy.

DETAILED DESCRIPTION 
Referring to the drawing figures, FIG. 1 is a block diagram illustrating an 
exemplary interior permanent magnet motor drive control system 10 in 
accordance with the principles of the present invention. The present 
invention provides for a novel position sensorless approach for 
implementing the interior permanent magnet motor drive control system 10 
for electric and hybrid electric vehicle applications. The interior 
permanent magnet motor drive control system 10 uses a combination of two 
rotor position estimation techniques for low and high speed operation, 
respectively, along with initial rotor magnet polarity detection at 
stand-still conditions. The overall efficiency and accuracy of the control 
system 10 are enhanced by combining the two different rotor position 
estimation techniques. 
The exemplary interior permanent magnet motor drive control system 10 is 
used to control operation of an interior permanent magnet motor 16 of an 
electric vehicle 30. A torque command (T.sub.e *) derived from an 
accelerator pedal of the electric vehicle 30 is applied to synchronous 
current regulators 11 along with a voltage (V.sub.BATT) from electric 
vehicle propulsion batteries and d and q axis current estimates (i.sub.ds 
e, i.sub.qs e) output by the control system 10. The regulators 11 output d 
and q axis voltage signals (U.sub.d, U.sub.q) which are input to a 
synchronous-to-stationary coordinate transform circuit 12. Output signals 
generated by the synchronous-to-stationary coordinate transform circuit 12 
comprise .alpha. and .beta. axis duty cycle signals (d.sub..alpha., 
d.sub..beta.) that are added to .alpha. and .beta. axis balanced high 
frequency voltage signals derived from software switches (S/W) 13, 36, 37 
using first and second adders 14a, 14b. The signals output by the adders 
14 are applied to a voltage source inverter 15 having high frequency 
signal injection. The three phase output of the voltage source inverter 15 
is applied to the interior permanent magnet motor 16. 
The interior permanent magnet motor drive control system 10 includes a 
controller 20 that comprises a terminal voltage calculation module 21 
(equations 5 and 6 below) that processes the battery voltage signals 
(V.sub.BATT) and the .alpha. and .beta. axis duty cycle signals 
(d.sub..alpha., d.sub..beta.) and outputs machine terminal line--line 
voltages (V.sub.ab,V.sub.bc) that are input to a motor voltage model 23 
(equations 7 and 8 below). A hardware (H/W) low pass filter 22 filters two 
phases of current signals (i.sub.a, i.sub.b) sensed by current sensors at 
machine terminals and applies the filtered current signals to the motor 
voltage model 23. The motor voltage model 23 outputs stator voltages 
(V.sub.LR, V.sub.LI) subtracting voltage drop across stator resistances. 
The current signals (i.sub.a, i.sub.b) output by the low pass filter 22 are 
processed by a three phase to two phase conversion circuit 24 and are 
input to a second low pass filter 25 and to a fundamental component 
synchronous frame filter 31. The fundamental component synchronous frame 
filter 31 processes the current signals along with the rotor position 
angle signal (.theta..sub.r) to filter out the fundamental frequency of 
the rotor using a form of a notch filter, converting the signals to 
stationary frame positive and negative rotating carrier current signals 
(i.sub.qsi s.sub.pn, i.sub.dsi s.sub.pn). The positive and negative 
rotating carrier current signals (i.sub.qsi s.sub.pn) i.sub.dsi s.sub.pn) 
are input to a positive rotating carrier current synchronous frame filter 
32 along with an angular position signal (.omega..sub.i t) to filter out 
the injection frequency using a form of a notch filter, to produce 
stationary frame negative rotating carrier current signals (i.sub.qsi 
s.sub.n, i.sub.dsi s.sub.n). The negative rotating carrier current signals 
(i.sub.qsi s.sub.n, i.sub.dsi s.sub.n) are input to a known heterodyning 
process 33. The heterodyning process 33 supplies input signals to a known 
position observer 34 that outputs parameter insensitive, zero lag, rotor 
velocity and position signals (.omega..sub.r.sbsb.--.sub.low, 
.theta..sub.r.sbsb.--.sub.low) and which are input to the software 
switches 38 and 39, respectively. 
The stator voltages (V.sub.LR,V.sub.LI) output by the motor voltage model 
23 along with a rotor velocity signal (.omega..sub.r), are input to a high 
speed flux angle estimation module 26. The high speed flux angle 
estimation module 26 integrates the stator voltages (V.sub.LR,V.sub.LI). 
The line flux angle estimate (.theta..sub..psi.L) has a predetermined 
angle (30.degree.) subtracted from it in a third adder 14c to produce a 
phase flux angle signal (.theta..sub..psi.s) which is input to a fourth 
adder 14d. A load torque angle estimation module 27 processes d and q axis 
synchronous reference frame current (i.sub.ds e, i.sub.qs e) to produce 
the load torque angle (.delta.). The third adder 14c subtracts the torque 
load angle (.delta.) from the phase flux angle signal (.theta..sub..psi.s) 
to produce the high speed rotor position signal 
(.theta..sub.r.sbsb.--.sub.high). 
The high speed rotor position signal (.theta..sub.r.sbsb.--.sub.high) is 
input to the software switch 39 and to a derivative (dv/dt) module 28 
which produces a high speed rotor speed signal 
(.omega..sub.r.sbsb.--.sub.high). The high speed rotor speed signal 
(.omega..sub.r.sbsb.--.sub.high) is applied to the software switch 39. The 
software switches 38 and 39 output the rotor speed signal (.omega..sub.r) 
and the rotor position angle signal (.theta..sub.r), respectively. 
The output of the second low pass filter 25 is applied to the software 
switch 13 along with the rotor speed signal (.omega..sub.r). The software 
switch 13 selects the output of the low pass filter 25 during low speed 
operation and the unfiltered signals output from block 24 during high 
speed operation. The software switch 13 outputs fundamental stationary 
frame current signals (i.sub.qsf s, i.sub.dsf s) which are input to a 
stationary-to-synchronous coordinate transform module 35 along with the 
rotor position angle signal (.theta..sub.r). The stationary-to-synchronous 
coordinate transform module 35 outputs the d and q axis synchronous 
reference frame currents (i.sub.ds e, i.sub.qs e) which are input to the 
synchronous current regulators 11. 
For low speed operation, the rotor magnetic saliency of the interior 
permanent magnet motor 16 is used to estimate the absolute rotor position. 
The motor 16 acts as an electromagnetic resolver and the inverter 15 
(power converter) applies carrier frequency voltages to the stator of the 
motor 16 which produce high frequency currents that vary with rotor 
position. These currents are then filtered by the fundamental component 
and positive rotating carrier current synchronous frame filters 31, 32. 
Negative rotating carrier currents are then processed using the 
heterodyning process 33 to produce a signal that is approximately 
proportional to the difference between the actual and estimated rotor 
position. This signal is then used as an input to a Luenberger style 
position observer 34, for example, to produce parameter insensitive, zero 
lag, rotor position signal. 
For high speed operation, the stator flux angle estimate 
(.theta..sub..omega.L), derived by integrating the stator voltages 
(V.sub.LR, V.sub.LI), and the torque load angle (.delta.) are used to 
estimate the rotor position. This technique works very well at high speed 
since the back EMF of the motor 16 is of higher amplitude. 
Smooth transition between low and high speed rotor position estimation 
methods is obtained by using the software switch 13. The switching 
mechanism implemented by the software switch 13 is as follows. The 
switching mechanism is illustrated in FIG. 3. 
For low speed operation, the rotor magnetic saliency of the interior 
permanent magnet motor 16 is used to estimate the absolute rotor position. 
This method utilizes an injected signal at a known frequency to extract 
information from spatial saliency. The injected signals are balanced 
stationary frame voltage signals as shown in equation (1) and (2). 
EQU V.sub.ds.sup.s =V.sub.i cos .omega..sub.i t (1) 
EQU V.sub.qs.sup.s =-V.sub.i sin .omega..sub.i t (2) 
where, .omega..sub.i is injection frequency in radians/second. Software 
switches 36 and 37 are controlled to inject the injected signals only upon 
entry and during the low speed operation as described below, so as not to 
limit the power available in the stator field coils during high speed 
operation. 
The motor 16 acts as electromagnetic resolver and the power converter 
applies carrier frequency voltages to the stator which produce high 
frequency currents that vary with rotor position. 
EQU i.sub.ds.sup.s =I.sub.p sin .omega..sub.i t-I.sub.n cos(h.theta..sub.r 
-.omega..sub.i t) (3) 
EQU i.sub.qs.sup.s =-I.sub.p cos .omega..sub.i t-I.sub.n sin(h.theta..sub.r 
-.omega..sub.i t) (4) 
where, I.sub.p and I.sub.n are positive and negative sequence carrier 
signal currents, h is a number identifying the harmonic, and .theta..sub.r 
is the rotor position. 
Only the negative component of the high frequency currents contain rotor 
position information as illustrated in equations (3) and (4). These 
currents are filtered by the fundamental component and positive rotating 
carrier current synchronous frame filters 31, 32. Negative rotating 
carrier currents are then processed with the heterodyning process 33 to 
produce a signal that is approximately proportional to the difference 
between actual and estimated rotor position. This signal is then used as 
input to the Luenberger style position observer 34 to produce parameter 
insensitive, zero lag, rotor position. 
This technique has two important features. First, its steady state tracking 
ability is not dependent on the parameters of the motor 16. The term 
I.sub.n is dependent on the inductance of the motor 16. However, this term 
is only a scaling term and does not effect the accuracy of the position 
estimate since its spatial angle is being tracked not the amplitude. 
Second, the magnitude of In is also independent of speed if a linear 
observer controller is used. This technique cannot differentiate between 
north and south pole of the magnets. So initial magnet polarity detection 
is required as discussed herein. 
For high speed operation, the back EMF of the motor 16 is utilized to 
estimate stator flux angle. Stator flux angle can be calculated by 
integrating the stator voltage behind the stator resistance. FIG. 2 shows 
a space vector diagram for permanent magnet motor 16 in which V.sub.s is 
the line to neutral back EMF vector and V.sub.L is the line to line 
voltage vector. 
The line to line flux linkage vector .omega..sub.L is obtained by 
integrating V.sub.L. The line to line EMF vector V.sub.L leads V.sub.s by 
30.degree. in angular space. Thus, by calculating line to line flux vector 
.omega..sub.L, the stator flux vector .psi..sub.s can be easily derived by 
subtracting 30.degree. as shown in FIG. 1. Since the motor terminal 
voltages are very noisy due to pulse width modulated operation, command d 
and q axis voltages (i.e., .alpha. and .beta. axis duty cycle signals) are 
used to derive line to line machine voltages in the terminal voltage model 
21. 
##EQU1## 
where, d.sub.d and d.sub.q are d and q axis duty cycles respectively, and 
V.sub.d is the DC link voltage. Real and imaginary components of the 
stator line to line voltage V.sub.L can be expressed as, 
##EQU2## 
where, R.sub.s is stator resistance, I.sub.a, I.sub.b, and I.sub.c are 
machine terminal currents. These real and imaginary components of line to 
line stator voltages are then processed through an integration module 26 
comprising the high speed flux angle estimation 26, to calculate real 
.omega..sup.s.sub.LR and imaginary .psi..sup.s.sub.LI line to line flux 
vectors. The integration module 26 comprises cascaded programmable low 
pass filters, which automatically compensate for 90.degree. phase shift as 
well as amplitude attenuation introduced by hardware or software filters. 
##EQU3## 
The space angle of the line to line flux is, 
##EQU4## 
As shown in FIG. 2, the line to line flux vector .psi..sub.L leads the 
stator phase flux vector .psi..sub.s by 30.degree.. Therefore, the stator 
flux vector position can be obtained as, 
EQU .theta..sub..psi.s =.theta..sub..omega.L -30.degree. (12) 
The load torque angle .delta. can be calculated as, 
##EQU5## 
where, L.sub.d and L.sub.q are d and q axis inductance values, .psi..sub.f 
is the flux linkage due to magnets, and i.sup.e.sub.ds and i.sup.e.sub.qs 
are d and q axis currents in the synchronous reference frame. Once the 
load torque angle is available, the rotor position angle can be obtained 
as, 
EQU .theta..sub.r =.theta..sub..psi.s -.delta. (14) 
The back EMF technique works very well at high speed since back EMF of the 
motor 16 is of higher amplitude. 
Each of the two rotor position detection techniques described above has 
limitations. The low speed injection technique requires high frequency 
signal injection to realize the rotor position, which limits the available 
DC link voltage. For electric and hybrid electric vehicles 30, a 
limitation on the DC link voltage availability is not desirable since it 
degrades the drive system efficiency near and above the base speed of the 
motor 16. This technique provides robust performance at and near zero 
speed. On the other hand, the high speed back EMF technique does not work 
at and near zero speed since it requires stator voltage information to 
detect rotor position. At zero speed the stator voltages are zero and near 
zero speed amplitude of stator voltages are very small compared to the 
signal to noise ratio. Integration of the noisy signal provides poor rotor 
position information, which may result in oscillatory output torque. Also 
near zero speed, the stator resistance drop is dominant so that any 
variation in stator resistance of the motor 16 adds error in rotor 
position estimation. Thus, the back EMF technique does not perform very 
well near zero speed. At high speed this technique provides accurate rotor 
position information since the amplitude of the stator voltages are higher 
and also voltage across stator resistance is negligible. 
In the sensorless techniques used in the control system 10, advantages of 
both techniques have been combined. Smooth transition between low and high 
speed rotor position estimation methods is obtained by using the software 
switches 13, 36, 37, 38, 39. FIG. 3 shows the mechanism implemented by the 
software switches 13, 36, 37, 38, 39. FIG. 3 is a state flow diagram 
showing the transition between low and high speed rotor position 
estimation techniques. 
For the purposes of completeness, FIG. 4 is a flow diagram illustrating an 
exemplary control method 40 in accordance with the principles of the 
present invention. The exemplary control method 40 controls a permanent 
magnet motor 16 having a rotor and a stator and comprises the following 
steps. 
An initial start-up determination 41 is made regarding the control method 
40. If it is determined 41 that the method 40 has started up (Yes), then 
the control method 40 is initialized, and the magnet polarity of the rotor 
at stand-still conditions of the motor 16 are determined 42. After 
initialization, or in the event that start-up has previously been done 
(No), estimates of the rotor position is processed 43 when the motor 
operates at low speed. This is achieved in the following manner. 
Stator currents are processed 44 to extract negative rotating carrier 
currents that vary with rotor position during low speed operation. 
Negative rotating carrier currents are processed 45 to produce heterodyned 
signals that are proportional to the difference between actual and 
estimated rotor positions. The heterodyned signals are processed 46 to 
produce parameter insensitive, zero lag, rotor position signals. 
Machine (motor 16) terminal currents are sensed and line to line voltages 
are calculated 51 using duty cycles and battery voltage. Line to line 
stator flux is generated 52 by integrating the line to line voltages using 
cascaded low pass programmable filters. The phase flux and flux angle of 
the stator are then generated 53. A rotor position signal is then 
generated 54 using load torque angle information for high speed operation. 
A selected one of the two rotor position signals are processed using a 
software switch 60 to generate 61 synchronous current signals. An inverse 
vector rotation is performed 62 to calculate feedback currents. The d and 
q axis synchronous frame voltages are generated 63 using current 
regulators. Stationary reference frame duty cycle signals are generated 64 
using rotor position information. Pulse width modulated signals are then 
generated 65 to control the permanent magnet motor 16. 
The present method has been implemented and tested on a 70 KW interior 
permanent magnet motor test bench to prove out the principles of the 
present invention. In the reduced to practice embodiment of the control 
system 10, the high speed rotor position estimation technique is enabled 
all the time while the low speed injection technique is disabled beyond 
stator or fundamental frequency of 25 Hz preventing using the DC link 
voltage due to high frequency injection. The transition from high speed to 
low speed takes into account that injection is required before the 
position observer 34 starts estimating the rotor position. Therefore, 
around a stator or fundamental frequency of 20 Hz electric frequency, the 
software module enables the high frequency signal injection and then waits 
until t.sub.d time for the position observer 34 to converge to the correct 
rotor position, and then switches to low speed mode. The time t.sub.d was 
determined by experiment. Rotor polarity detection is only required at the 
initial stage. Once the magnet polarity is detected, then even when the 
vehicle 30 completely stops, rotor position observer 34 remembers the last 
rotor position. In case of a processor reset, magnet polarity detection 
operation is regenerated. 
Compared to conventional approaches, the present invention does not require 
a position sensor. Reduced mechanical and electrical interface costs are 
provided by the present invention. A reduced overall drive production cost 
is also achieved using the present invention. The present invention also 
has improved reliability. 
Thus, an improved permanent magnet motor control system for use in electric 
and hybrid electric vehicles, and the like has been disclosed. It is to be 
understood that the above-described embodiment is merely illustrative of 
one of the many specific embodiments that represent applications of the 
principles of the present invention. Clearly, numerous and other 
arrangements can be readily devised by those skilled in the art without 
departing from the scope of the invention.