High-efficiency low distortion parallel amplifier

A high-efficiency low-distortion parallel amplifier or regulator is driven by an input signal source V.sub.i. An input signal passes to a linear circuit LC, and the output current i.sub.1 from the linear circuit passes to the input of a switching circuit SC. Alternatively, the input to the switching circuit may be any other signal that is a function of the linear-circuit output current. The output of the switching circuit passes through a coupling network CN and is applied with the output current of the linear circuit to a load impedance ZL. The output of the switching circuit acts to reduce the output current of the linear circuit so that the power it supplies is low. A low linear-circuit output current is usually sufficient to drive the switching circuit and coupling network to the maximum current required by the load. The system automatically adjusts independently of the load impedence so that with high output current most of the current is supplied by the switching circuit.

This invention relates generally to amplifiers, and in particular audio 
power amplifiers, the power amplifiers employed in inverters, dc power 
converters, and voltage regulators. The invention is specifically 
concerned with obtaining the attributes of high efficiency together with 
low distortion in the case of an amplifier, or with low output ripple in 
the case of a regulator. With this invention it is possible to build 
high-performance amplifiers and regulators of low cost, small size, low 
weight and high reliability of operation. Other advantages are also 
achieved by the use of this invention. 
In the prior art of amplifier and regulator technology it has been 
difficult to achieve high-efficiency operation simultaneously with low 
distortion or low ripple. (In the context of the disclosure of this 
invention, low distortion or low ripple occurs when the amplifier or 
regulator output closely approximates the desired output.) High efficiency 
means that there is little wasted power. High efficiency is desirable 
because minimal power is consumed, the electrical requirements of the 
associated power supply are minimal, and the hardware requirements for 
dissipating the wasted power are small. This leads to the attributes noted 
above. Low distortion or low ripple is a necessary performance parameter 
of amplifiers or regulators in many practical applications. 
In the prior art high-efficiency amplifiers and regulators have been of the 
switching type; the active power devices of these systems operate in a 
switching rather than a linear mode. Switching (also known as class-D) 
amplifiers and regulators may either be of the pulse-width-modulation type 
(see, for example: H. R. Camenzind, "Modulated pulse audio power 
amplifiers for integrated circuits", IEEE Trans. Audio Electroacoust., 
vol. AU-14, pp. 136-140, September 1966; J. A. Dutra, "Digital amplifiers 
for analog power", IEEE Trans. Consumer Elec., vol. CE-24, pp. 308-316, 
August 1978; R. Mammano, "Simplifying converter design with a new 
integrated regulating pulse width modulator", SG1524 Application Note, 
Silicon General Inc.) or alternatively, of the self-oscillating type (see, 
for example: A. G. Bose, "Signal translation apparatus", British Pat. No. 
1,055,788, Jan. 18, 1967; J. J. Spijkerman and C. L. Sturgeon, "The 
Sturgeon amplifier--a new switching technique", Proceedings of the Fourth 
National Solid-State Power Conversion Conference (Powercon 4), pp. 
H3-1-H3-3, May 1977; J. L. Jensen, "Voltage and current regulation", U.S. 
Pat. No. 2,776,382, January 1, 1957). In this context the term pulse-width 
modulation is synonymous with driven or synchronous, and the term self 
oscillating is synonymous with free-running or ripple. Switching 
amplifiers have failed to become popular for high-fidelity audio 
applications for two reasons: their distortion is too high (and is not 
readily reduced by simply applying high levels of negative feedback) and 
their output contains considerable unwanted high-frequency (hf) energy. 
Similarly, switching regulators are plagued by high ripple and significant 
hf energy in their output. A further problem with switching amplifiers and 
regulators is their inferior large- and small-signal transient responses 
compared with those of their linear counterparts. It is an object of the 
present invention to ameliorate some or all of these defects of switching 
amplifiers and regulators. 
The present invention combines low-efficiency linear circuits with 
high-efficiency switching circuits, such a combination being referred to 
here as a hybrid system. In the present invention the output ports of 
linear and switching circuits are connected together via one or more 
coupling networks, both circuits supplying power in a parallel manner to a 
load and thus forming a parallel hybrid system. The efficiency of this 
system is high because the output current is predominantly supplied via 
the switching circuits; the linear circuit operates with low average 
output current. A different method of realising a hybrid system, a series 
hybrid system, is the subject of a previous invention (see: P. Garde, 
"High-efficiency low-distortion amplifier", Australian Patent Application 
No. 59694/80, June 26, 1980). In this previous method the outputs of 
switching circuits drive the supply rails of the output stage of a linear 
circuit thereby reducing the voltage across the output stage and achieving 
high efficiency. Power is supplied in a series manner from the switching 
circuits to the linear circuit and from there to the load. Both parallel 
and series hybrid systems have low distortion or ripple in their output 
signals because this depends upon their linear circuits, rather than their 
switching circuits alone; the linear circuits correct any error in the 
output of the switching circuits. As the linear circuits only operate at a 
power level sufficient to counteract the error in the output of the 
switching circuits, the power wasted by the linear circuits is very small 
compared with the power capability of the system, and high efficiency is 
obtained. 
If very high efficiency is required, the parallel and series hybrid systems 
may be combined. In this scheme the linear circuit of a parallel hybrid 
system may itself be a series hybrid system. Alternatively, the linear 
circuit of a series hybrid system may be a parallel hybrid system. 
Similarly, more than two hybrid systems may be combined; however, such 
arrangements possess the disadvantage of high complexity. 
The present invention may be employed in the realisation of a parallel 
hybrid amplifier or regulator. The circuitry of a hybrid regulator is 
usually simpler than that of a hybrid amplifier because a regulator need 
only supply output current in one direction; however the linear circuit of 
a hybrid regulator requires a reference source. 
The present invention consists in a high-efficiency low-distortion parallel 
amplifier or regulator system comprising a high-efficiency switching 
circuit, a low-distortion linear-amplification means, a coupling network, 
and a coupling node, wherein a system input signal or regulator reference 
voltage is fed to an input port of the linear-amplification means, or 
wherein a reference voltage is generated within said linear-amplification 
means, and wherein an output voltage signal produced at an output port of 
the linear-amplification means is adapted to be fed to a load via a system 
output port, and a signal which is a function of a current flowing in the 
output port of the linear-amplification means is fed to an input port of 
the switching circuit, and a switching output current flowing in the 
output port of the switching circuit is fed predominantly to the coupling 
node which is disposed between the output port of the linear-amplification 
means and the system output port, at which node the switching output 
current is combined with the current flowing in the output port of the 
linear-amplification means to produce a system output current which flows 
via the system output port to the load, the switching output current 
having a maximum time rate of change limited by the coupling network which 
is either incorporated into the switching circuit or alternatively is 
provided between the output port of the switching circuit and the coupling 
node, the amplifier or regulator system further characterised in that the 
system output current when averaged over a period of time is substantially 
supplied by the switching circuit when said averaged output current is 
large. 
In particular embodiments of the present invention, low-pass-filter means 
can be incorporated into the linear-amplification means to attenuate 
high-frequency noise such as switching noise entering an input port of the 
linear-amplification means. 
Particular embodiments of the present invention can also include 
low-pass-filter means provided between the coupling node and the system 
output port. 
In particular embodiments of the present invention a plurality of switching 
circuits can be provided in parallel connection, the output current of 
each switching circuit being fed predominantly to the coupling node and 
having a maximum time rate of change limited by a coupling network, the 
number of coupling networks provided being less than or equal to the 
number of switching circuits. 
It will be apparent to those skilled in the art that there are many 
different techniques for realising the present invention and that 
embodiments may be produced with many different features, the choice of 
those features for a particular application being dependent upon the use 
to which the system is to be put. Some of these techniques and features 
will now be discussed by way of example; however, it is not intended that 
the invention be limited to these examples. 
Embodiments of the present invention may include a low-pass filter at their 
inputs. This filter attenuates any hf noise such as that which originates 
from hf signal sources or switching of electric currents. 
The peak-current capability of the linear circuit of particular embodiments 
of the present invention may be of almost any value, high or low, in any 
given application. In all cases the average power dissipated by the linear 
circuit can be low and thus high efficiency achieved. The advantage of a 
high-current capability is that the switching circuits and coupling 
networks can have a slow response. A slow network response infers that the 
network greatly attenuates the switching transients. In this case it is 
easy to keep the switching frequency low, thus minimising switching 
losses. Under transient or other input signal conditions which are too 
fast for the switching circuits and coupling networks, the distortion is 
kept low by high currents supplied by the linear circuit. As transients 
form only a small proportion of the input signal in most practical 
applications, such as in audio systems, these high currents do not add 
significantly to the average power dissipated by the linear circuits. 
If the linear circuit is designed with a low-peak-current capability, the 
cost of the linear circuit is low; the linear circuit may be a low-power 
integrated-circuit amplifier. In this case, if the system is to respond 
linearly to fast signals, the response of the switching circuits and 
coupling networks must also be fast. A problem with conventional power 
amplifiers of the prior art is the difficulty of stabilising the bias 
current of the output devices. The problem is particularly severe with 
high-power amplifiers. The present invention substantially eases the 
difficulty because the linear circuit of a high-power amplifier or 
regulator need only have low-power capability. For the same reason the 
problem of secondary breakdown in the output transistors of a high-power 
amplifier is also greatly reduced. In addition, it is simple to fabricate 
a very fast linear circuit when its output requirement is low. The 
advantage of a fast linear circuit is that a large amount of negative 
feedback can be employed and very low distortion achieved; switching 
transients are greatly attenuated. 
It is unnecessary that the system utilise negative feedback to realise the 
present invention; the requirement for substantial attenuation of the 
noise and distortion from the switching circuits at a given frequency is 
that the linear-circuit output impedance be small compared with that of 
the coupling networks at that frequency. Nevertheless, in most 
applications this requirement is most easily met and high performance 
obtained by the judicious use of negative feedback around the linear 
circuit. The linear circuit may include more than one negative-feedback 
loop. In particular, the use of multiple feedback loops which all include 
the node to which the coupling networks from the switching circuits are 
connected may greatly reduce the noise and distortion from the switching 
circuits. At audio frequencies, for example, the loop gain around the node 
may thereby be increased to almost any desired degree and the switching 
noise and distortion correspondingly reduced to virtually any desired 
level. 
Within the switching circuit the control signals of the switching devices 
may be derived so that the circuit operates in either 
pulse-width-modulation or self-oscillating mode; several other modes of 
operation are also possible. 
Dc isolation between the power source and the terminals of an amplifier or 
regulator is a requirement in many applications of the present invention. 
The switching circuits of particular embodiments of the invention may be 
designed so that they inherently possess dc isolation. In this case the 
use of a large heavy mains-frequency power transformer or an hf switching 
supply to power the system are avoided. 
It is a requirement of the present invention to limit the time rate of 
change of the output currents of the switching circuits. This may be 
achieved by including inductance in series with the switching-circuit 
outputs within the coupling networks. The maximum time rate of change is a 
strong function of the difference between the switching-circuit supply 
voltage and the system output voltage. To avoid unduly limiting the 
large-signal (power) bandwidth of the system when driving a load it may be 
desirable for the switching circuits to be connected to higher supply 
voltages than those which power the linear circuit. 
The average switching frequency may be reduced and switching losses 
minimised if the supply voltage of the switching circuit is varied as a 
function of the load voltage and of the time rate of change of the load 
current. Continuous or discrete-level supply-voltage variation is 
possible. For example, a selection of discrete voltages may be available 
to power the switching circuits, the most appropriate voltage at any 
instant in time being fed to the switching circuits by suitable logic 
circuitry. 
The average switching frequency may also be reduced and switching losses 
minimised if two or more switching circuits and coupling networks are 
operated in parallel. The various switching circuits and their coupling 
networks may or may not be of different design and thus may switch at 
different switching-circuit input levels and have different response 
speeds. When a fast response is required more than one switching circuit 
may turn on to supply the output, or alternatively, a single switching 
circuit and coupling network with an appropriately fast response may 
supply the output. The switching circuits may be powered from the same or 
different supply voltages. 
If two or more switching circuits and coupling networks operate in 
parallel, means may be provided so that when the load current is high the 
current is shared between the circuits according to their capabilities. 
At high frequencies it may be desirable for the voltage transfer functions 
of the coupling networks to be low pass, although coupling networks should 
not exhibit undue time delay for the particular application. One purpose 
of a low-pass transfer function may be to attenuate hf switching noise 
coupled from a switching circuit to the system output. Another purpose may 
be to reduce the linear-circuit peak-current requirement. If the 
low-pass-filter cut-off frequencies are low compared with the switching 
frequency, the current requirement of the linear circuit is similarly low 
as long as it is unnecessary for the system to respond to transients or 
other fast signals. This may be the situation with a voltage regulator, 
particularly if its output is capacitively bypassed, for example. 
Embodiments of the present invention may include a low-pass filter at the 
output between the coupling network and the load. Such a filter attenuates 
the hf noise fed to the load and decouples the load from the system 
feedback; the effect of the hf impedance of the load on system performance 
is minimised. For maximum attenuation of the noise the cut-off frequency 
of the filter must be as low as possible, the limiting frequency being set 
by the required large-signal bandwidth of the system. This is usually 
considerably below the required small-signal bandwidth (cut-off frequency) 
of the system. 
If the filter cut-off frequency is to be below the system small-signal 
bandwidth then the filter characteristic must be equalised elsewhere in 
the system. Exact equalisation is often difficult to achieve, particularly 
if the load impedance is not well defined. To avoid this difficulty the 
filter may be enclosed in a negative-feedback loop. Provided that the loop 
has sufficient gain, the desired system small-signal bandwidth can be 
attained. The negative-feedback loop enclosing the filter may or may not 
be the only loop enclosing the linear circuit. When a feedback loop 
encloses the filter the load is an integral part of the negative-feedback 
loop at high frequencies. It may therefore be desirable to decouple the 
load at such frequencies with another low-pass filter. 
Embodiments of the present invention may include special circuitry to 
protect the system from failure such as is possible during signal 
overdrive, overload conditions, or high-temperature operation.

Consider the proposed design for a high-efficiency low-distortion amplifier 
shown in FIG. 1. The amplifier is driven by an input signal source 
v.sub.i. The input signal passes to the linear circuit LC and the 
switching circuit SC. The switching circuit draws negligible power from 
the input-signal source and by a very efficient switching process 
generates an output signal v.sub.s which is an approximately linear 
function of v.sub.i plus noise and distortion components. The switching 
circuit is able to provide sufficient power to drive the load impedance 
Z.sub.L as desired. 
The linear circuit amplifies the input signal and applies it to the load. 
The transfer function v.sub.o /v.sub.i of the linear circuit is also that 
of the overall amplifier. It is unnecessary for the linear circuit by 
itself to be capable of driving the load because most of the output power 
is supplied by the switching circuit. Although the linear circuit has low 
efficiency this does not appreciably degrade the efficiency of the overall 
system because the average power supplied by the linear circuit is low. On 
the other hand the linear circuit has the benefit of low distortion. As 
the linear circuit determines the system output signal, any noise or 
distortion components from the switching circuit appear across the 
coupling network CN and any resulting currents are absorbed by the linear 
circuit and therefore do not pass to the load. For these reasons the 
overall system has both high efficiency and low distortion. 
The coupling network interposes impedance between the linear- and 
switching-circuit outputs. This avoids large circulating currents when the 
circuit outputs are unequal and decouples the linear and switching 
circuits to help avoid hf oscillation when the linear circuit utilises 
negative feedback. It is desirable that a coupling network be a 
pure-reactance network otherwise system efficiency is degraded. The load 
impedance together with the transfer functions of the coupling network and 
the overall amplifier determine the required transfer function of the 
switching circuit with the assumption that ideally the linear circuit does 
not supply power to the load. 
There are two serious problems with the amplifier shown in FIG. 1. First, 
if there is a finite tolerance on the significant amplifier-component 
parameters or on the load impedance, or if the delay times of the linear 
and switching circuits (from amplifier input to output) are unequal, then 
the linear circuit may be forced to supply a large current to the load 
and/or switching circuit. As a consequence system efficiency is degraded. 
Second, the design and behaviour of the amplifier depends critically on 
the load impedance. In practice this impedance is poorly defined and may 
vary over wide limits. The application of the amplifier of FIG. 1 is 
severely limited by both of these problems. It is an object of the present 
invention to overcome them. 
An embodiment of the present invention, a high-efficiency low-distortion 
parallel amplifier or regulator, is shown in FIG. 2. In comparison with 
FIG. 1 it is apparent that the transfer function v.sub.o /v.sub.i of the 
linear-circuit is again that of the overall amplifier; however, the input 
of the switching circuit SC is the linear-circuit output current i.sub.1 
rather than the input voltage v.sub.i. Alternatively, the input to the 
switching circuit may be any other signal that is a function of the 
linear-circuit output current. The principle of operation of the present 
invention is that the output of the switching circuit generally acts to 
reduce the output current of the linear-circuit so that the power it 
supplies is low. With the invention a finite tolerance on system 
components is allowable because system operation does not depend 
critically upon matching or tracking of transfer functions within the 
system. Also, system performance is not unduly affected by the load 
impedance because a low linear-circuit output current is usually 
sufficient to drive the switching circuit and coupling network to the 
maximum current required by the load. Independent of the load impedance, 
the system automatically adjusts so that with high output current most of 
the current is supplied by the switching circuit. 
While the linear circuit used in embodiments of the present invention may 
be a linear amplifier of conventional design, it is nevertheless 
advantageous if it is specially designed to attenuate the noise and 
distortion from the switching circuits. One example of such a design, a 
multi-loop negative-feedback linear-circuit with an input filter, is shown 
in FIG. 3. The low-pass filter F1 at the input attenuates hf noise 
entering the system input. In the forward path of the circuit there are n 
amplifying stages A1 to An. The feedback networks B1 to Bn complete the n 
loops which include the coupling-network connection node. In this 
arrangement the total loop gain around the node may be very high, well in 
excess of Bode's limit for a single feedback loop, so the noise and 
distortion from the switching circuits via the coupling networks are 
reduced to a low level. The arrangement must be carefully designed to 
avoid oscillation; design procedures are well known to those skilled in 
the art. 
A parallel hybrid system may supply uni- or bi-directional current to the 
load. Usually a system which supplies only uni-directional current is 
simpler than a bi-directional current system. A dc power supply is an 
example of a uni-directional-current system and an audio amplifier is an 
example of a bi-directional-current system. 
Independent of whether a parallel hybrid system is designed to supply uni- 
or bi-directional current to the load, the output current of the 
associated linear-circuit may itself be uni- or bi-directional. The 
advantage of a system whose linear-circuit current is uni-directional may 
be simple implementation. The advantage of a bi-directional linear-circuit 
current may be very high efficiency. The nature of the linear-circuit 
current is determined by the design of the switching circuits and coupling 
networks. 
There is considerable freedom in the biasing arrangements of a 
linear-circuit output stage of a system which supplies bi-directional 
current; class-A, -B, or -AB biasing is possible, for example. When a 
linear circuit which supplies uni-directional current is used, a novel 
class-A arrangement may be employed wherein the output stage is single 
ended with the bias current supplied by the switching circuits, via the 
coupling networks. 
There are many ways of implementing the switching circuit; operation may be 
based upon pulse-width-modulation or self-oscillating principles, for 
example. The partial schematic of one arrangement for implementing a 
pulse-width-modulator switching circuit for a system which can supply 
bi-directional current is shown in FIG. 4. The comparators C1 and C2 
compare a voltage proportional to the linear-circuit output current with 
two oscillatory waveforms v.sub.a and v.sub.b which may have appreciable 
dc components. The comparator outputs v.sub.1 and v.sub.2 operate two 
switches S1 and S2, a high output voltage causing a switch to open. The 
clamp diodes D1 and D2 avoid extreme voltage swings of the 
switching-circuit output when driving an inductive coupling network. 
Examples of voltage waveforms suitable for v.sub.a and v.sub.b when the 
linear circuit supplies bi-directional current are shown as a function of 
time t in FIG. 5. Although the waveforms are triangular many other 
waveform shapes are equally suitable. However, for proper operation it is 
desirable that v.sub.a should always be more positive than v.sub.b at any 
instant. In the first example FIG. 5(a), there is no output from the 
switching circuit at low input-signal levels and the linear circuit alone 
drives the output. This feature avoids switching noise and distortion at 
low signal level. In the second example FIG. 5(b), switching occurs at all 
signal levels. In the third example FIG. 5(c), the waveforms v.sub.a and 
v.sub.b are almost identical. If the waveforms are identical as shown in 
FIG. 5(d), either S1 or S2 is closed at any given instant and therefore D1 
and D2 are redundant. The disadvantage of identical waveforms is that 
non-ideal behaviour of the switches may cause their conduction periods to 
overlap and result in current flow between the supply rails V+ and V-. By 
slightly offsetting the waveforms as shown in FIG. 5(c), this problem is 
avoided. While it is unnecessary that v.sub.a and v.sub.b be in phase, 
this is nevertheless desirable if v.sub.a is to be always more positive 
than v.sub.b. In addition, when v.sub.a and v.sub.b are in phase the least 
time that switches S1 and S2 are both off is maximised, and conduction 
overlap is most unlikely. 
If the switching circuit operates in the self-oscillating mode then it is 
unnecessary to generate two oscillatory waveforms. The partial schematic 
of FIG. 4 is also suitable for implementing a self-oscillating switching 
circuit if va and vb are dc voltages and if comparators C1 and C2 have 
appreciable hysteresis. This switching circuit is suitable for a 
self-oscillating system which can supply bi-directional current. 
Examples of the transfer characteristics of a self-oscillating switching 
circuit driving a resistive load, when the associated linear circuit 
supplies bi-directional current, are shown in FIG. 6. In the first example 
FIG. 6(a), switch S1 closes when the linear-circuit output current is 
sufficiently large. When the current subsequently reduces to a low value, 
switch S1 opens. Switch S2 closes when the current is large and negative, 
and it opens when the current becomes small. At low input-signal levels 
there is no output from the switching circuit and in this situation, as 
with the scheme associated with FIG. 5(a), switching noise and distortion 
are avoided. In the second example FIG. 6(b), switching occurs at all 
signal levels. In the third example FIG. 6(c), the magnitudes of the 
currents at which the switches open and close are almost equal. If the 
current magnitudes are equal as shown in FIG. 6(d), the transfer 
characteristic degenerates into a single hysteresis loop. In this case 
either S1 or S2 is closed at any given instant and therefore D1 and D2 are 
redundant. The disadvantage of this type of operation is that conduction 
overlap of the switches may result in current flow between the supply 
rails. By making the current magnitudes unequal as shown in FIG. 6(c), 
this problem is avoided. 
The pulse-width-modulator switching circuit shown in the partial schematic 
of FIG. 4, which may be adapted for operation in the self-oscillating 
mode, is suitable for systems supplying bi-directional current. If only 
uni-directional current is supplied, the circuit may be simplified. In 
particular, one comparator together with its associated switch is 
redundant and only one voltage v.sub.a or v.sub.b is required. The voltage 
waveforms of FIG. 5 and the transfer characteristics of FIG. 6 may be 
appropriately modified to correspond with this simplification. It is 
apparent that in general the magnitude of the current extremes of the 
linear circuit are determined independently by the switching circuit and 
in this respect there is considerable design flexibility. 
In a particular application it may be advantageous if the switching-circuit 
voltages v.sub.a and v.sub.b are a function of the system operating 
conditions. For example, if the magnitudes of v.sub.a and v.sub.b decrease 
as temperature increases, the power dissipated by the linear circuit 
reduces and stress on the linear circuit is relieved. 
Embodiments of the present invention may be designed so that they possess 
dc isolation between the power source and the system terminals by 
including an hf transformer within the switching circuit; additional 
isolation devices may also be required. Methods of embedding an hf 
transformer within a switching circuit are well known (see, for example: 
L. Rensink et al, "Design of a kilowatt off-line switcher using a Cuk 
converter", Proceedings of the Sixth National Solid-State Power Conversion 
Conference (Powercon 6), pp. H3-1-H3-26, May 1979; S. Cuk, "A new 
zero-ripple switching dc-to-dc converter and integrated magnetics", IEEE 
Power Electronics Specialists Conference, 1980 Record, pp. 12-32). The 
linear circuit may be powered from a low-power auxiliary supply. Thus dc 
isolation between the power source and the system terminals is readily 
achieved without the disadvantages of a large heavy mains-frequency 
transformer. 
Embodiments of the present invention may have multiple switching circuits 
and coupling networks operating in parallel as shown in FIG. 7. The input 
to each of the switching circuits SC1 to SCn and their associated coupling 
networks CN1 to CNn is the linear-circuit output current i.sub.1. The 
output currents from the coupling networks add together at the system 
output. 
Two realisations of coupling networks are shown in FIG. 8. A very simple 
realisation FIG. 8(a) consists of an inductor L1 connected between the 
input and output ports. The more complex realisation FIG. 8(b) includes an 
additional inductor L2 and capacitor C1. These increase the attenuation of 
hf switching noise from the switching circuit. If desired, additional LC 
sections may be included in the coupling network. 
Embodiments of the present invention may include various filters connected 
at the output of the amplifier or regulator system as shown in FIG. 9. In 
FIG. 9(a) a low-pass filter F2 is connected in series with the output of 
the parallel hybrid system PHS1. In FIG. 9(b) the low-pass filter F3 is 
enclosed in a negative-feedback loop. In FIG. 9(c) the load is decoupled 
by filter F5 from the negative-feedback loop enclosing filter F4. 
The circuit diagram of a high-efficiency negative-voltage switching 
regulator which is an embodiment of the invention is shown in FIG. 10. The 
regulator has a reference voltage determined from a negative voltage 
source V2 by the capacitor C1, the resistor R1, and the Zener diode Z1. 
The reference voltage is applied to the positive input of the operational 
amplifier A1. The power input terminals of this amplifer are connected to 
earth, and resistor R2. This resistor senses the current flowing to the 
negative voltage V2; this current is approximately equal to the output 
current of A1. When the output current is low, bipolar transistors Q1 and 
Q2 together with MOS transistor M2 are off, and MOS transistor M1 is on. 
If the current increases sufficiently, the conduction states of the 
transistors reverse and transistor M2 passes current from the amplifer 
output via the coupling network (comprising capacitor C2, and inductors L1 
and L2) to the negative voltage source V2. The coupling network limits the 
time rate of change of the current flow. As the current in M2 increases, 
the current in R2 becomes small, and hence M2 is eventually switched off. 
When M2 is off, diodes D5 and D6 limit the voltage swing at the drain of 
M2 to a safe value and maintain current flow in L2. Resistors R3 and R4 
provide hysteresis in the transfer characteristic. The collector current 
of Q1 is set by R5 and the voltage source V1 whose potential is more 
positive than V2. Diodes D3 and D4 avoid saturation of Q2, and resistor R7 
limits the peak current in Q2 to a safe value. Diodes D1 and D2 protect 
the output of A1 from any excessive voltage swing. In this embodiment the 
operational amplifier is connected for unity gain so the regulator output 
v.sub.o is approximately equal to the Zener voltage. 
The circuit diagram of an audio power amplifier which is an embodiment of 
the invention is shown in FIG. 11. The amplifier has an input signal 
v.sub.i which is ac coupled via capacitor C10 to a low-pass filter 
composed of capacitor C11, and resistors R10 and R11. The filtered signal 
passes to the positive input of operational amplifier A10. This amplifier 
is powered from a positive voltage source V12 and a negative voltage 
source V13, via resistors R14 and R15. These resistors sense the 
operational-amplifier output current. When the output current is low, 
bipolar transistors Q10 and Q11 are both off, but if the current increases 
sufficiently either Q10 or Q11 turns on. As a result, MOS transistor M12 
or M13 turns on (the operation of the switching circuits is similar to 
that for the switching regulator just described and consequently is not 
considered further) thus passing current via the coupling network 
(comprising capacitor C13, and inductors L11 and L12) to or from the 
power-amplifier output whose voltage is v.sub.o. Resistor R12 in 
conjunction with R13 sets the gain of the amplifier (the gain is also 
slightly dependent on R10 and R11). Capacitor C12 provides lead 
compensation of the feedback amplifier. A low-pass filter at the amplifier 
output (composed of capacitor C14, inductor L10, and resistor R26) 
decouples the amplifier load from the negative-feedback loop at high 
frequencies.