Samples of the amplitude of a quadrature amplitude modulated (QAM) subcarrier signal are taken at times corresponding to each axis crossing of each of the two quadrature related signal components. The polarity of alternate ones of the samples is inverted and the signal is reconstructed from the inverted and non-inverted samples whereby the phase of one of the signal components is reversed without alteration of the phase of the other. Where the QAM signal is a television chrominance signal, means are provided for periodically inhibiting inversion of the samples on alternate lines whereby type chrominance signals may be converted to or from NTSC or NTSC "like" chrominance signals.

This invention relates to transcoders and particularly to a transcoder for 
reversing the phase of one of two signal components of a quadrature 
amplitude modulated signal. 
Quadrature double sideband suppressed carrier amplitude modulation 
(QDSSC-AM, hereinafter QAM) is utilized for the transmission of color 
difference signals R-Y and B-Y in both the NTSC and the color 
television systems and has also been proposed for transmission of position 
differences signals L-R and F-B in quadraphonic stereo systems (see, 
generally, IEEE Transaction on Broadcast and Television Receivers, Vol. 
BTR-19, No. 4, November, 1973). Since the carrier is suppressed in QAM, it 
is necessary that it be regenerated in a receiver to be able to 
synchronously detect and recover the original color (or position) 
difference signals. To facilitate this regeneration, the QAM signal in 
both the NTSC and color television systems includes a burst component 
of a few cycles of subcarrier frequency transmitted during the "back 
porch" interval of the horizontal synchronizing period. In the proposed 
quadraphonic transmission systems carrier regeneration may be provided by 
means of a continuously transmitted low level pilot signal having a 
predetermined frequency and phase relationship with the subcarrier. 
It is known that one may utilize a regenerated subcarrier for processing an 
analog type of QAM signal in such a manner as to reverse the phase of one 
of the two difference signal components. One way of doing this is to 
multiply the QAM signal with a suitably phased double subcarrier frequency 
reference signal as shown for example, in U.S. Pat. No. 3,968,514 which 
issued to Narahara, et al., July 6, 1976. That approach, however, results 
in the production of triple subcarrier frequency products in the output 
signal which are relatively difficult to remove by filtering. 
An alternative to the "multiplier" approach is the 
"demodulator-remodulator" technique disclosed in the U.S. patent 
application of Carnt, et al., Ser. No. 822,659 which was filed Aug. 8, 
1977 and issued Apr. 29, 1980, as U.S. Pat. No. 4,200,881. There the 
analog QAM signal components are synchronously demodulated to base band, 
the polarity of one base band component is inverted and then (after base 
band filtering) both components are remodulated on respective quadrature 
related subcarriers. The difficulty with this approach is that the base 
band filtering requires the use of relatively high valued circuit elements 
which increases the size and cost of the transcoder and makes construction 
in integrated circuit form relatively more difficult. 
The present invention is directed in one respect to meeting the need for a 
transcoder which avoids the problems associated with base band filtering 
and in which principal undesired products of the transcoding process occur 
at a frequency greater than three times the subcarrier frequency. 
In accordance with one aspect of the invention, a transcoder for reversing 
the phase of one of two signal components of a quadrature amplitude 
modulated signal includes means for producing samples of the amplitude of 
the subcarrier at times corresponding to each axis crossing of each of the 
signal components, means for inverting alternate ones of the samples and 
means for reconstructing the subcarrier from the inverted and non-inverted 
samples. 
The invention is directed in another aspect to meeting the need for a 
transcoder for converting a chrominance signal of a first form to one of a 
second form wherein one of said forms conforms to standard chrominance 
and burst phasing, and, the other conforms to a standard such as NTSC or 
the "NTSC-like" standard proposed by Carnt, et al., in their 
aforementioned application. 
In accordance with a further aspect of the invention, a conversion between 
chrominance signals of the forms noted above may be facilitated by 
periodically inhibiting polarity inversion of alternate ones of the 
samples.

In FIG. 1 a sample and hold circuit 10, a sample inverter circuit 12 and a 
low pass filter 14 are connected in cascade in that order between a QAM 
input terminal 16 and a transcoder output terminal 18. A reference 
subcarrier input terminal 20 is connected to the input of a phase 
adjusting circuit 22 which is connected at the output thereof to a control 
or enabling input of the sample and hold circuit and also, via a divider 
24, to an enabling or control input of the sample inverter circuit. 
In transcoding applications where the QAM signal is of analog form, sample 
and hold circuit 10 may be of conventional design. A suitable 
implementation would be a transmission gate arranged to charge a holding 
capacitor with the QAM signal when the gate is enabled, the charge being 
retained when the gate is disabled. It is not necessary, however, that the 
holding time equal the entire period that the gate is disabled. In other 
words, the sample and hold circuit may, if desired, be reset to some 
reference level prior to the taking of a new sample. Other suitable sample 
and hold circuits featuring improved accuracy, reduced aperture time and 
minimized settling time are described, for example, in "Applications of 
Operational Amplifiers--Third Generation Techniques" by J. G. Graeme 
published by McGraw-Hill Book Company in 1973, pp. 132-139. 
The purpose of sample and hold circuit 10 is to produce samples of the 
amplitude of the QAM signal applied to terminal 16 at times corresponding 
to each axis crossing of each of the two quadrature related components of 
the QAM signal. The reason that samples are taken at axis crossings of the 
component signals is that it is at that instant of time that the QAM 
signal amplitude exactly equals that of one of its two individual 
components. The significance of this may be more fully appreciated by 
considering waveform B of FIG. 4. There, the signal waveform R represents 
the QAM signal at terminal 16 which is the resultant vector sum of the two 
signal components U and V. Component U is shown as a sinewave of unit 
amplitude and component V is shown as a quadrature related sinewave of 
half unit amplitude lagging U by 90.degree.. Since in quadrature 
modulation the U and V components always differ in phase by 90.degree., 
any sample taken of R when U is zero will represent the polarity and 
magnitude of only the V component. Similarly, any sample taken of R when V 
goes through a zero crossing will represent the polarity and magnitude of 
only the U component of R. 
The purpose of phase adjuster 22 is to control precisely when sample and 
hold circuit 10 is enabled so that the samples produced alternately 
represent the U and V components of R. The input to phase adjuster 22 is a 
regenerated reference subcarrier of four times the QAM subcarrier 
frequency. Where the QAM signal is a television chrominance signal the 
reference signal would be phase locked in the conventional manner to the 
color burst component of the chrominance signal. Where the QAM signal is 
used for position difference signal transmission as previously mentioned 
the reference subcarrier would be phase locked at a multiple of the pilot 
signal frequency to thereby have a fixed phase relation to the subcarrier 
and a frequency of four times the subcarrier frequency. 
Phase adjustor 22 (which may be a conventional lead or lag network) should 
be adjusted by an amount such that transitions (or peaks) of the phase 
shifted quadruple frequency reference subcarrier (4f.sub.sc) coincide with 
axis crossings of the QAM signal components. If the regenerated subcarrier 
exhibits this property after frequency multiplication then phase adjuster 
22 may be omitted. Waveforms A and B of FIG. 4 illustrate the preferred 
phase relationship between the U and V components of R and the output of 
phase adjuster 22. Note that at even numbered positive transistions 
(t.sub.o,t.sub.2,t.sub.4,etc.) of the phase adjusted reference subcarrier 
(waveform A) the U component of R is zero and the V component equals R. At 
all odd numbered positive transistions (t.sub.1,t.sub.3,t.sub.5,etc.) the 
V component is zero and the U component of R equals R. 
As shown in waveform C sample and hold circuit 10 produces samples of the 
component U of width W.sub.u and samples of the component V of width 
W.sub.v. It is not necessary that W.sub.u be equal to W.sub.v. In fact, in 
cases where one may wish to alter the relative amplitudes of U and V, one 
may do so by separately pulse width modulating the samples, as for 
example, by varying the holding time of sample and hold circuit 10. It is 
thus a feature of the invention that, in addition to reversing the phase 
of one of the QAM signal components without alteration of the other, one 
may also vary the amplitude of one of the QAM signal components 
independently of the other. 
In cases where it is desired to preserve with high accuracy the relative 
magnitudes of the U and V components it is preferred to set W.sub.u equal 
to W.sub.v rather than the alternative of scaling the amplitudes of the 
samples in proportion to the differences in W.sub.u and W.sub.v. To put it 
another way, a given relationship between U and V can be maintained by 
control of two parameters of the samples, namely, height and width. By 
maintaining the product of these parameters constant for each of U and V, 
it is possible upon reconstruction of the QAM signal to accurately 
preserve the original amplitude relationships between U and V while at the 
same time reversing the phase of one of the components. This may be 
achieved in accordance with a further aspect of the invention by 
maintaining equal gain through the cascade connection (10, 12, 14) for U 
and V samples and letting W.sub.u =W.sub.v. In cases where amplitude 
control by sample width modulation is not needed and where maximum 
conversion gain upon reconstruction of the sampled signal is desired it is 
preferred that the sample width for both U and V be substantially equal to 
one quarter of the QAM subcarrier frequency which is the same as one 
complete period of the reference subcarrier frequency 4f.sub.sc. If this 
is done, then there would be no dwell at the zero level between samples as 
in waveform C but rather a stair step type waveform would be produced with 
the amplitude of each sample remaining constant until the instant the next 
sample is taken. 
Waveform D illustrates operation of sample inverter 12 which, controlled by 
divider circuit 24, inverts the polarity of alternate ones of the samples 
produced by sample and hold circuit 10. As previously explained, all even 
numbered samples represent only the V component of R since they are taken 
when U is zero. Divider 24 divides the reference subcarrier signal of 
4f.sub.sc by two thereby supplying an enabling signal of frequency 
2f.sub.sc to inverter circuit 12. Since circuit 12 is enabled at a rate of 
exactly half of the sample rate (4 fsc) every other sample (V) will be 
inverted but intermediate samples (U) will not be inverted. 
In order to insure that the desired sample is inverted (in this case V) it 
is preferable that divider 24 be suitably initialized during the start of 
the transcoding operation or, alternatively, that means be provided for 
reversing the phase of its output relative to waveform A. It would be 
convenient where the QAM signal is a television chrominance signal to 
perform this initialization (i.e., resetting, presetting, etc.) of divider 
24 during the color burst interval since at that time precise vector 
relationships are known. Alternatively, initialization may be done once 
each period of the QAM signal as shown and described subsequently. 
Low pass filter 14 provides the means by which the QAM subcarrier may be 
reconstructed from the inverted (V) and non-inverted (U) samples produced 
at the output of inverter circuit 12. Referring now to waveform E of FIG. 
4 the components U and V are shown as smoothed separate sine waves which 
would result if the U and V components were filtered by separate low pass 
filters. If the U and V samples were so separated and separately filtered 
one could then sum the filtered outputs to provide the resultant vector R 
which, as will be recognized in comparison with waveform B, corresponds to 
the original QAM signal with the phase of one component (V) reversed. 
There are three significant problems with the separate component filtering 
approach to reconstructing the QAM signal. First, it would be necessary to 
provide means for steering alternate samples to respective filters and 
this in addition to the requirement for two filters, adds to the cost and 
complexity of the transcoder. Secondly, different gains or losses in the 
filters could result in unbalancing the initial amplitude relationship 
between U and V. Thirdly, the filter cutoff frequencies which would be 
required would be less than twice the subcarrier frequency (2f.sub.sc) 
which further complicates the filtering problem. Needless to say, 
undesired double frequency transcoding products are even more difficult to 
remove by filtering than the undesired triple subcarrier frequency 
products characteristic of the previously discussed "multiplier" or 
"modifier" type of non-baseband QAM transcoder. 
All of the above mentioned difficulties may be overcome by use of a single 
low pass filter 14 for reconstructing the QAM signal. Since only one 
filter is involved the U and V components will be treated equally in terms 
of both amplitude and phase. Moreover, the filter cutoff frequency may be 
double that of the two-filter summer approach and the cutoff frequency may 
be higher than that required to remove triple frequency products from 
modifier type transcoder. As a practical matter the lowest dominant 
undesired transcoding product occurs at the sampling rate of four times 
the subcarrier frequency and so the filter size, cutoff frequency accuracy 
and stop band slope requirements are all reduced. 
FIG. 2 illustrates one way in which the sample inverter circuit 12 of FIG. 
1 may be implemented and interfaced with divider 24, sample and hold 
circuit 10 and low pass filter 14. Specifically, the output of divider 24 
is connected to control a single pole two position switch 30. 
Illustratively, switch 30 may comprise a pair of transmission gates 
controlled by the true (Q) and complemented (Q) outputs of divider 24. 
When in position N (normal or non-invert) switch 30 connects the output of 
sample and hold circuit 10 to the input of low pass filter 14. When in 
position R (reverse polarity or invert) switch 30 couples the output of 
sample and hold circuit 10 to the input of low pass filter 14 via an 
inverting amplifier 32. 
The general operation of the transcoder of FIG. 2 is substantially the same 
as in the discussion of FIG. 1 with switch 30 and inverting amplifier 32 
performing the function of sample inverter circuit 12. One difficulty with 
the arrangement of FIG. 2 is that abrupt changes in voltage may occur at 
the output of sample and hold circuit 10 (see waveform C of FIG. 4). To 
accommodate these step function voltage changes it is preferable that 
amplifier 32 exhibit a slew rate and a settling time which are negligible 
compared with the sample width, otherwise, the V samples which pass 
through amplifier 32 may be distorted. 
The arrangement of FIG. 3 reduces the slew rate and settling time 
requirements on amplifier 32 by placing it in a portion of the transcoder 
where the signals are continuous rather than discrete. Specifically, 
inverting amplifier 32 is connected at its input to input terminal 16 and 
at its output to the input of a second sample and hold circuit 34. The 
enabling or control input of circuit 34 is connected to the output of 
phase adjuster 22 so that the two sample and hold circuits operate 
simultaneously the only difference being that one (10) takes non-inverted 
samples of the QAM signal while the other (34) takes inverted samples. 
Switch 30 then operates alternately to select the normal (N) or reversed 
(R) samples which are then reconstructed by low pass filter 14 as 
previously described to provide the transcoded QAM output signal. 
To assure substantially equal processing of the U and V components it is 
desirable that the product of the gain and sample width of circuit 10 
equal that of circuit 34. In FIG. 3, as illustrated, only the circuit gain 
needs to be balanced since both circuits receive the same sample pulse 
width. This, however, is not essential to achieve balanced treatment of 
the U and V components and, if desired, the sample pulses may be of 
different widths and the gains of the sample and hold circuits may also 
differ. Balancing may be done, as previously explained, by setting the 
product of the sample width and gain through one path equal to that of the 
other path. It is further desirable that means (e.g. delay equalizers) be 
provided for equalizing the phase shift or delay through the two signal 
paths to assure maximum accuracy in generation of the true and inverted 
samples. 
FIG. 5 illustrates how the transcoder of the invention may be modified for 
converting a chrominance signal of the type in which the phase of the U 
and V color difference subcarrier components is constant from line-to-line 
to one of the type in which the phase of the V component alternates from 
line-to-line. FIG. 5 also illustrates a desirable method of interfacing 
the transcoder with a video disc player for producing standard output 
video signals from video disc recordings of the standard proposed by Carnt 
et al., in U.S. patent Application Ser. No. 822,659, entitled "VIDEO DISC 
SYSTEMS" filed Aug. 8, 1977 and which issued Apr. 29, 1980, as U.S. Pat. 
No. 4,200,881. 
There are three salient features of the Carnt, et al. recording 
standard relevant to the video disc player of FIG. 5. The first is that 
the subcarrier is "de-switched", that is, the normal phase alternation 
by line of the V component of the chrominance subcarrier is inhibited for 
recording purposes. Secondly, the chrominance subcarrier frequency is 
shifted lower in frequency from nominally 4.43 MHz to about 1.52 MHz and, 
in effect, "buried" within the luminance band. Thirdly, chrominance burst 
is recorded at a constant phase angle of 45.degree. relative to the U and 
V subcarriers so that burst exhibits substantially equal U and V 
components. 
In FIG. 5 the transcoder is arranged to provide line-by-line reversal of 
the chrominance V component at the buried subcarrier frequency. This is 
done prior to time base correction and conversion of the subcarrier to 
standard. Since burst includes equal U and V components, the type 
"swinging" burst is automatically formed by the transcoder with the 
periodic reversal of the V component. 
It may be noted that if burst had no V component, as in the NTSC standard, 
then reversal of the V component phase would have no affect on the burst 
phase. Accordingly, if one wished to apply the principles of the present 
invention to conversion of a chrominance signal with NTSC burst phasing to 
one of format, it would be necessary to suitably change the mode of 
operation of the transcoder during the burst interval. This could be done, 
as an example, by providing means for phase shifting the chrominance 
signal by 45.degree. during the burst interval, thereby resulting in a 
Carnt, et al. type of burst phasing which, as previously explained, is 
automatically converted to the swinging burst form as the phase of the 
V component is alternated. This is discussed in more detail in connection 
with the example of FIG. 8. 
The video disc player in FIG. 5 comprises a turntable 50 for rotating video 
disc 52 and a pickup transducer 54 for recovering video information from 
the disc. Illustratively, it will be assumed that the player is intended 
for use with a record in which information is stored in the form of 
topological variations and recovered by sensing capacitance variations 
between transducer 54 and the record 52. It will be appreciated, however, 
that the transcoder in accordance with the present invention may be used 
in connection with other types of disc players, tape players, camera 
equipment, frame storage units, transmission systems, etc. For purposes of 
illustration, it will be assumed that disc 52 is recorded with video 
information in the previously mentioned Carnt, et al. format. 
The output of transducer 54 is coupled to the input of a pickup converter 
circuit 56 which comprises a capacitance-to-voltage converter responsive 
to capacitance variations between a stylus in transducer 54 and the record 
being played for producing an FM output signal voltage representative of 
the recorded information. Suitable circuits for implementing the 
capacitance-to-voltage conversion function of pickup circuit 56 are well 
known. See, for example, U.S. Pat. No. 3,783,196 entitled "HIGH-DENSITY 
CAITIVE INFORMATION RECORDS AND PLAYBACK APATUS THEREFOR" which 
issued to T. O. Stanley, Jan. 1, 1974, U.S. Pat. No. 3,972,064 entitled 
"APATUS AND METHODS FOR PLAYBACK OF COLOR PICTURES/SOUND RECORDS" which 
issued to E. O. Keizer, July 27, 1976, and U.S. Pat. No. 3,711,641 
entitled "VELOCITY ADJUSTING SYSTEM" which issued to R. C. Palmer, Jan. 
16, 1973. 
Video FM demodulator circuit 58 converts the FM signal produced by pickup 
circuit 56 to a video output signal. The video signals recorded on the 
disc are in "buried subcarrier" (BSC) format rather than the conventional 
NTSC format. As explained in the Carnt, et al. application, (see also U.S. 
Pat. No. 3,872,498 which issued to D. H. Pritchard, Mar. 18, 1975), in the 
BSC format chrominance information is represented by a color subcarrier of 
the general form employed in the well known NTSC format. However, the 
chrominance component in the BSC format is not located in the high end of 
the luminance signal video band, as in NTSC, but rather is buried in a 
lower portion of the video band. An illustrative subcarrier frequency 
choice is in the vicinity of 1.52 MHz, with the color subcarrier side 
bands extending .+-.500 KHz thereabout and with the luminance signal band 
extending well above the highest color subcarrier frequency (to 3 MHz, for 
example). 
FM demodulator 58 illustratively may be of the pulse counting type or of 
the phase lock loop (PLL) type. A suitable pulse counting type FM 
demodulator is disclosed in U.S. Pat. No. 4,038,686 entitled "Defect 
Detection And Compensation" which issued to A. L. Baker, July 26, 1977. An 
FM demodulator of the phase lock loop type is described in the U.S. Patent 
Application, Ser. No. 984,013 of T. J. Christopher, et al., entitled "FM 
SIGNAL DEMODULATOR WITH DEFECT DETECTION" which was filed Oct. 2, 1978 and 
issued May 13, 1980, as U.S. Pat. No. 4,203,134. 
The composite video signal produced by FM demodulator 58 is separated into 
luminance and chrominance components by means of a variable center 
frequency comb filter 60. Examples of filters of this type are given in 
U.S. Pat. No. 3,966,610 which issued to H. Kawamoto Dec. 7, 1976, and the 
U.S. Patent Application of T. J. Christopher and L. L. Tretter entitled 
"VIDEO PROCESSING SYSTEM INCLUDING COMB FILTERS", Ser. No. 966,512 which 
was filed Dec. 4, 1978 and issued Mar. 25, 1980, as U.S. Pat. No. 
4,195,309. 
The reason for using a variable comb filter (rather than one of fixed 
frequency) is to maximize filtering efficiency by changing the filter 
center frequency in accordance with time base errors which may be present 
in the video signal. This requires supplying a relatively high frequency 
control signal to filter 60 to operate clock drivers that control the rate 
of charge transfer in a CCD delay line in the filter. This high frequency 
signal is one of five output signals produced by timing signal generator 
62. 
Generator 62 has input terminals 64 and 66 for receiving, respectively, the 
BSC chrominance signal and the luminance signal produced by filter 60 and 
outputs 68, 70, 72, 74 and 76 for producing, respectively, a line 
frequency signal f.sub.H, a burst key signal BK, a regenerated reference 
subcarrier signal F.sub.BSC (1.52 MHz), a reference signal of four times 
subcarrier frequency 4f.sub.BSC and a reference signal of eight times 
subcarrier frequency 8f.sub.BSC. Terminal 66 is connected to the input of 
sync detector 80 which detects horizontal synchronizing pulses present in 
the luminance signal and supplies synchronizing pulses at the horizontal 
line rate f.sub.H to output terminal 68 and to the input of burst key 
generator 82. Generator 82 produces an output burst key pulse, BK, during 
the burst interval of the horizontal synchronizing period. The burst key 
pulse is supplied to the enabling input terminal of phase detector 84 and 
to output terminal 70. 
The output of detector 84 is coupled via a cascade connection of low pass 
filter 86, voltage controlled oscillator 88, divider 90 and divider 92 to 
one of its two phase comparison inputs the other of which connects to 
terminal 64 thereby forming a plural output burst keyed multiplying phase 
locked loop. The outputs of VCO 88, divider 90 and divider 92 are 
connected, respectively, to terminals 76, 74 and 72. 
Phase detector 84 compares the phase of the burst component of the 
chrominance signal supplied to terminal 64 with the divided output of VCO 
88 and produces an error signal which is filtered by filter 86 (which may 
be simply an error voltage holding capacitor) and applied as a negative 
feedback signal to VCO 88 for minimizing any phase errors. Since the 
output of VCO 88 is divided by two by divider 90 and divided by four by 
divider 92, VCO 88 thus operates (when phase locked) at a frequency of 
eight times the burst frequency and this high frequency signal is supplied 
via terminal 76 to the center frequency control input of comb filter 60. 
Since VCO 88 is always phase locked to a multiple (X8) of the buried 
subcarrier burst signal any time base errors in the burst are present in 
the VCO output. The feedback to filter 60 is of a sense to change its 
center frequency by an amount such that both its luminance and chrominance 
pass bands remain centered about the luminance and chrominance spectra 
when time base errors occur thereby maximizing filtering efficiency. 
Terminals 74 and 72 being connected to the outputs of dividers 90 and 92, 
respectively, receive regenerated buried subcarrier reference frequencies 
of 4f.sub.BSC and f.sub.BSC, respectively. 
It is a feature of the video disc player of FIG. 5 that timing signal 
generator 62 provides in one unified arrangement a source of five timing 
signals which are used for controlling three functions of the player. One 
signal is supplied to filter 60 as noted above for maximizing filtering 
efficiency. As will now be discussed in detail, three of the signals are 
supplied to transcoder 90 for controlling chrominance phase conversion and 
a further signal is supplied to a time base corrector 100 for controlling 
both time base correction and frequency conversion of the chrominance 
signal. 
Transcoder 90 is similar to the transcoder of FIG. 3 but has been modified 
in two respects. One of the modifications comprises the addition of means 
for periodically inhibiting the operation of switch 30. The other 
comprises means for initializing divider 24. Switch 30 is inhibited 
periodically by AND gate 35 which has been interposed between the output 
of divider 24 and the control or enabling input of switch 30 and is 
supplied with half line rate control signals f.sub.H /2 from the output of 
a further divider 36. The input of divider 36 is connected to a further 
transcoder input terminal 38 which is connected to output terminal 68 
where the line rate signal f.sub.H is produced. Initialization (or 
periodic presetting) of divider 24 is provided by means of an added pulse 
shaper 37 (e.g. a monostable multivibrator) which is connected at its 
input (via added terminal 40) to output terminal 72 of generator 62 and at 
its output to a preconditioning input (i.e., a jam, set or reset input) of 
divider 24. Preferably, divider 24 is of the type which may be 
preconditioned (i.e., directly set or reset) without regard to the state 
of the signal at its clock input. 
In operation of the portion of the player described thus far, comb filter 
60 and timing signal generator 62 produce output signals as previously 
described. The chrominance output signal is supplied to input 16 of 
transcoder 90 and the quadrupled regenerated reference buried subcarrier 
(4f.sub.BSC) signal is supplied to terminal 20. Inverting amplifier 32, 
the two sample and hold circuits 10 and 34 and phase adjuster 22 operate 
as previously described with circuit 10 producing non-inverted samples of 
the chrominance signal and circuit 34 producing inverted samples of the 
chrominance signal, each sample instant being taken at a time equivalent 
to the U and V axis crossings whereby every other sample represents a U 
(or-U) sample and intermediate samples represent V (or-V). 
The horizontal line rate signal f.sub.H is divided by two in divider 36 
which supplies a half line rate signal f.sub.H /2 to AND gate 35. During 
alternate television lines AND gate 35 is primed to supply the double 
subcarrier reference frequency signal 2f.sub.BSC to switch 30 and during 
intermediate lines it blocks the 2f.sub.BSC signal and holds switch 30 in 
its normal (N) position. Accordingly, during the intermediate lines only 
the samples from circuit 10 are supplied to filter 14 and since their 
polarity is unchanged, filter 14 provides a reconstructed chrominance 
signal to output 18 which is identical to that supplied to input 16. 
During the alternate lines however, when gate 35 is enabled, switch 30 
alternately selects inverted and non-inverted samples of the chrominance 
signal and so, for the reasons previously explained, the phase of one of 
the U or V chrominance components at the output of filter 14 is reversed 
relative to its phase at input terminal 16. 
Selection of which of the U and V components is phase reversed on alternate 
lines is important for the proper reproduction of a format output 
signal. As previously mentioned in the discussion of FIG. 1, one may 
insure that a desired one of the components is phase reversed by proper 
initialization of divider 24. Of the two methods of performing the 
initialization previously mentioned, it is preferred to do it 
periodically. A preferred rate of periodic preconditioning is once during 
each subcarrier cycle. In transcoder 90 this function is provided by pulse 
shaper 37 which, illustratively, may be a differentiator or a monostable 
multivibrator. 
Recall that the f.sub.BSC signal at terminal 72 is phase locked to the 
burst component of the chrominance signal and this reference is 
continuously available. Pulse shaper 37 thus continuously produces pulses 
having a fixed relation to burst. These pulses are applied as preset 
signal to divider 24 such that the initial state of divider 24 causes 
switch 30 to select the output of sample and hold circuit 10 when the V 
component is at an axis crossing. Thus the first clock signal supplied to 
divider 24 after initialization will cause switch 30 to select the V 
component when the U component is zero. If divider 24 should ever fail to 
toggle for an odd number of clock signals, pulse shaper 37 will 
immediately resynchronize it when its toggle operation returns to normal. 
(Failure to toggle for an even number of clock signals, of course, does 
not result in loss of synchronization for obvious reasons). 
The remaining elements of FIG. 5 comprise a time base corrector and 
frequency translation circuit 100 and a summing circuit 110. Circuit 100 
has an input terminal 101 for receiving the burst key signal BK from 
timing signal generator 62, a second input terminal 102 for receiving the 
format chrominance signal produced at output terminal 18 of transcoder 
90 and an output terminal for supplying a frequency translated and time 
base corrected standard output signal to one input of summing circuit 
110. The other input of circuit 110 is connected to receive the luminance 
output signal produced by filter 60 to thereby a composite video output 
signal of format at the output of circuit 110. 
Recall from the previous discussion that the player video output signal 
includes time base errors and that one of the three functions of timing 
signal generator 62 is to provide an 8Xf.sub.BSC signal containing those 
time base errors to the comb filter 60 to vary its center frequency in 
proportion to the errors thereby maximizing filtering efficiency. This 
process, of course, has no effect at all on the time base errors and they 
are present at all five outputs of generator 62 as well as in the 
chrominance signal produced by filter 60. Since the chrominance errors and 
timing signal errors vary together, however, there is no interference or 
detrimental effect on the operation of transcoder 90. The significance of 
this is that where the transcoder of the present invention is used in 
video apparatus of any type (tape, disc, studio, transmission) which 
includes time base correction means one has two options. The first is that 
where the transcoder precedes the time base corrector in the video 
processing chain, it should be supplied with timing signals having time 
base errors proportional to the color burst errors in the chrominance 
signal. The other option is that where the chrominance signal is time base 
corrected (or otherwise stabilized or resynchronized) prior to application 
to the chrominance transcoder, the timing signals supplied to the 
transcoder should be derived from the corrected burst signal. If one 
observes these two principles, then the transcoder will introduce minimal 
differential phase errors in the video processing chain thereby providing 
maximal color purity. This is particularly important with regard to NTSC 
transcoding since in the NTSC system differential chrominance subcarrier 
phase errors represent changes in hue which can not be averaged out 
optically or electronically as in the transmission system. 
Differential amplitude errors, a source of the Hanover bar or Venetian 
Blind effect in the system, may be minimized if proper care is taken 
in balancing the gain in the U and V signal paths through the transcoder 
as previously mentioned. 
Turning now to the details of circuit 100, phase detector 103 when enabled 
by the BK signal, compares the output of a standard oscillator 104 
(about 4.43 MHz) with a band pass filtered product of the burst component 
of the transcoded chrominance signal at terminal 102 multiplied by the 
output of a voltage controlled oscillator 105 having a nominal center 
frequency equal to the sum of the buried subcarrier frequency (1.52 MHz) 
and the frequency (4.43 MHz). If a time base error exists detector 103 
supplies a correction signal via low pass filter 106 to oscillator 105 in 
a sense to correct it. The multiplier (108) may be a conventional balanced 
modulator or mixer and the pass band of the band pass filter 107 should be 
centered at the subcarrier standard of 4.43 MHz. 
The transcoder of FIG. 6 is similar to that of FIG. 1 but includes three 
modifications to provide conversion of a chrominance input signal having 
NTSC burst phasing. Two of the modifications are the same as in transcoder 
90 of FIG. 5, namely, the addition of an AND gate 35 and divider 36 for 
periodically inhibiting polarity reversals of the samples every other 
horizontal line and the addition of a subcarrier frequency pulse shaper 
circuit 37 for periodically presetting or initializing divider 24 to 
assure that inverter 12 is properly synchronized so as to invert only the 
V component of the chrominance input signal. 
The third modification comprises a switch 601 and a phase shift circuit 602 
interposed between chrominance input terminal 16 and the input of sample 
and hold circuit 10 and controlled by a burst key signal supplied to a 
further input terminal 603 so as to shift the phase of the chrominance 
signal by 45.degree. during the color burst interval (but not during the 
active video span interval). Preferably, the 45.degree. shift is in a 
sense to rephase burst at an angle of 135.degree. relative to the B-Y 
color difference signal axis as prepared by Carnt, et al. In this way the 
NTSC chrominance signal, insofar as burst phasing is concerned, is 
converted to a Carnt, et al. type of signal prior to chrominance 
conversion to format. With the exception of this periodic phase shift 
during the burst interval, the circuit operation is the same as discussed 
in connection with FIG. 5. 
The phase shift circuit in this example is implemented by connecting one 
pole A of switch 601 directly to chrominance input terminal 16. The other 
pole B is coupled to terminal 16 via the 45.degree. phase shift network 
602. Switch 601 receives burst key signals supplied to terminal 603 from a 
source (not shown) and when the key signal is present couples pole B to 
the input of sample and hold circuit 10. Where the key signal is absent 
switch 601 couples pole A to circuit 10 whereby the burst component of the 
NTSC signal (which normally has no V component) is converted to one of the 
Carnt, et al. standard (which has equal U and V components) which, in 
turn, is processed by the remainder of the circuitry to form as 
previously described. 
It is not necessary that the burst key signal supplied to terminal 603 
include only the burst interval. It may, for example, extend throughout 
the entire horizontal synchronizing interval, if desired, since 
information is not visibly displayed on a television monitor during that 
period of time. Phase shift network 602 may be a conventional lead or lag 
network or it may be a delay line of a length equal to one-eighth of the 
subcarrier period. In either case, the phase shift is necessarily a 
function of the subcarrier frequency so that appropriate changes or 
adjustments should be made to accommodate different subcarrier 
frequencies. 
Subcarrier frequency conversion may be performed by conventional 
heterodyning circuits prior to application of the chrominance signal to 
terminal 16, or between the output of switch 601 and the input of circuit 
10 or it may be done after chrominance conversion (i.e., at output 
terminal 18 as in FIG. 5). For any one of these three options the 
parameters of elements 602,22,37 and 14 should be chosen to conform to the 
subcarrier frequency so that all phase shifts and pass bands conform to 
the requirements previously described. 
FIG. 7 illustrates how the transcoder of FIG. 6 may be modified for 
converting a format chrominance input signal to an NTSC type of 
chrominance output signal. As previously explained, the actual subcarrier 
frequency choice is arbitrary, that is, it may be 4.43 MHz (), 3.58 MHz 
(NTSC) or any other suitable value (e.g., 1.52 MHz as in the Carnt, et al. 
standard). The modification comprises connecting the input of pole A and 
phase shifter 602 to output 18 and applying the chrominance input 
signal to the input of sample and hold circuit 10. 
In operation, the transcoder elements between terminals 16 and 18 function 
as previously described to reverse the phase of the V chrominance 
component on alternate lines. Since the V component itself alternates from 
line-to-line in the system, the signal produced at output terminal 18 
is therefore converted to one of the Carnt, et al. type of burst phasing. 
The rearrangement of switch 601 and circuit 602 shifts the phase of the 
burst 45.degree. into alignment with the -(B-Y) axis (NTSC standard) 
without altering the "de-switched" chrominance phase during the active 
scan interval. 
The principles of the invention, while illustrated in connection with 
processing analog QAM signals, apply also to the transcoding of digitized 
QAM signals. In FIG. 1, for example, the function of sample and hold 
circit 10 may be performed with digital means such as a data latch, 
register, analog-to-digital converter or the like. Inversion of the 
sampled digital QAM signal may readily be performed, for example, by 
complementing the sign bit of the sample (where the data is in 
sign-magnitude form) so that inverter 12 could be implemented by means of 
an exclusive-OR gate. Other changes and modifications may be made, such as 
replacing sample and hold circuit 10 with an analog-to-digital converter 
arranged to sample the QAM signal at the times shown and described in 
connection with FIG. 4 and inverting the digital samples with suitable 
arithmetic means (e.g., the X-OR gate previously mentioned) and then 
applying the signal to a digital-to-analog converter (rather than filter 
14) for purposes of reconstruction of the QAM signal. 
FIG. 8 shows how the transcoder of FIG. 1 may be modified as described 
above. As may be seen, elements 10, 12 and 14 are replaced by an 
analog-to-digital converter 810, a sign bit inverter 812 and a 
digital-to-analog converter 814 which perform analogous functions to the 
replaced elements. Where the QAM signal is a television chrominance signal 
means (such as an AND gate and divider, etc.) should be provided for 
periodically inhibiting operation of the sign bit inverter as in the 
previously discussed examples of the invention.