Integrated circuit for generating a reference voltage

A circuit for generating a reference voltage including a first transistor and a second transistor of which the bases being commonly connected together. The area of the emitter of the first transistor being smaller than the area of the emitter of the second transistor, the emitter of the first transistor being connected to the ground, and the emitter of the second transistor being connected to the ground via a first resistor. The circuit also includes a current supply means which supplies an equal current to the collectors of the first and second transistors and a second resistor which is connected between an output terminal and a connection point of the commonly connected bases of the first and second transistors. The circuit additionally includes a current generator circuit which is connected between the connection point of the commonly connected bases and the ground to produce a current which is proportional to the emitter current of the first transistor or the second transistor, such that a constant voltage is generated at the output terminal.

BACKGROUND OF THE INVENTION 
The present invention relates to a circuit for generating a reference 
voltage, and more specifically to an integrated circuit for generating a 
reference voltage which is in agreement with a band gap of a semiconductor 
material that forms the transistor and which assumes a predetermined value 
irrespective of the temperature. 
The reference voltage must, usually, assume a constant value independently 
of the temperature. This requirement can be satisfied by using a band-gap 
reference circuit. As represented, for example, by an integrated circuit 
LM 117 manufactured by National Semiconductor Co., the band-gap reference 
circuit consists of a first transistor and a second transistor of which 
the bases are commonly connected and which are supplied with an equal 
current from a current mirror circuit, the area of the emitter of the 
second transistor being N times greater than that of the first transistor. 
Further, a first resistor is connected to the emitter of the second 
transistor, and a connection point between the other end of the first 
resistor and the emitter of the first transistor is grounded via a second 
resistor. The collector voltage of the first transistor, on the other 
hand, is fed back to the power supply of the current mirror circuit via a 
feedback amplifier, and the output voltage is taken out from the base 
potential of the first and second transistors. 
In such a conventional circuit for generating the reference voltage, the 
potential of the power supply for supplying a current to the current 
mirror circuit must be higher than the collector potential of the first 
transistor. When the reference voltage is 1.2 volts, the potential of the 
power supply of the current mirror circuit must be greater than 2.1 volts 
at room temperature. The potential of the power supply of the current 
mirror circuit is supplied from the power supply of the feedback 
amplifier. Therefore, the feedback amplifier requires a higher 
power-supply voltage. The requirement of such a high power-supply voltage 
is not desirable for integrated circuits. 
SUMMARY OF THE INVENTION 
The object of the present invention is to provide a reference voltage 
generator circuit which operates on a small power-supply voltage. 
Another object of the present invention is to provide a reference voltage 
generator circuit which can be suitably obtained in the form of an 
integrated circuit. 
The above objects of the present invention can be achieved by a circuit for 
generating a reference voltage, including: a first transistor and a second 
transistor of which the bases being commonly connected together. The area 
of the emitter of the first transistor being smaller than the area of the 
emitter of the second transistor, the emitter of the first transistor 
being connected to the ground, and the emitter of the second transistor 
being connected to the ground via a first resistor. The circuit also 
includes current supply means which supplies an equal current to the 
collectors of the first and second transistors and a second resister which 
is connected between an output terminal and a connection point of the 
commonly connected bases of the first and second transistors. The circuit 
additionally includes a current generator circuit which is connected 
between the connection point of the commonly connected bases and ground to 
produce a current which is proportional to the emitter current of the 
first transistor or the second transistor, so that a constant voltage is 
generated at the output terminal. 
Further features and advantages of the present invention will become 
apparent from the ensuing description with reference to the accompanying 
drawings to which, however, the scope of the invention is in no way 
limited.

DESCRIPTION OF THE PREFERRED EMBODIMENTS 
FIG. 1 shows a conventional band-gap reference circuit in which the feature 
resides in a pair of npn transistors Q.sub.1 and Q.sub.2 that produce a 
current proportional to the absolute temperature, and a resistor R.sub.1. 
The transistors Q.sub.1 and Q.sub.2 of which the bases are commonly 
connected are supplied with an equal current from a current mirror circuit 
1 comprising of pnp transistors Q.sub.3 to Q.sub.5, and the area of the 
emitter of the transistor Q.sub.2 is N times greater than that of the 
transistor Q.sub.1. One end of a first resistor R.sub.1 is connected to 
the emitter of the transistor Q.sub.2, and another end of the resistor 
R.sub.1 and the emitter of the transistor Q.sub.1 are grounded via a 
second resistor R.sub.2. Therefore, the base potential of the transistors 
Q.sub.1 and Q.sub.2, i.e., a reference voltage V.sub.B at the output 
terminal B is given by, 
EQU V.sub.B =V.sub.BE1 +I.sub.2 R.sub.2 (1) 
where V.sub.BE1 denotes a voltage across the base and emitter of the 
transistor Q.sub.1, and I.sub.2 denotes a current which flows through the 
resistor R.sub.2. 
If emitter currents of the transistors Q.sub.1 and Q.sub.2 are each denoted 
by I.sub.E, there is the relation I.sub.2 =2I.sub.E. 
Since the transistors Q.sub.1 and Q.sub.2 have different emitter areas, the 
voltage V.sub.BE2 across the base and emitter of the transistor Q.sub.2 is 
different from the voltage V.sub.BE1 across the base and emitter of the 
transistor Q.sub.1. Namely, 
##EQU1## 
where k denotes Boltzmann's constant, T denotes the absolute temperature, 
q denotes the electric charge of an electron, N denotes a ratio of emitter 
areas, and I.sub.S denotes a saturated current. 
In the connection mode of FIG. 1, 
EQU V.sub.BE1 =V.sub.BE2 +I.sub.E .multidot.R.sub.1 (4) 
If relations (2) and (3) are inserted into the above relation (4), there is 
obtained the relation, 
EQU I.sub.E .multidot.R.sub.1 =V.sub.R1 =V.sub.T l.sub.n N (5) 
By using the above relation (5), the relation (1) can be rewritten as 
follows: 
##EQU2## 
The temperature dependency, therefore, is as shown in FIG. 2. Namely, 
V.sub.BE1 which is the first term on the right side of the relation (6) 
decreases with the increase in the temperature T, and 
##EQU3## 
which is the second term increases with the rise in the temperature T. 
Therefore, if the changing ratios are equalized by adjusting R.sub.2 
/R.sub.1, the two values are cancelled by each other, and the reference 
voltage V.sub.B remains constant (compensated for the temperature). This 
constant value is nearly equal to a band-gap voltage (1.2 volts in the 
case of a silicon semiconductor) of a semiconductor material which forms 
transistors Q.sub.1 and Q.sub.2. 
Here, if a voltage across the collector and emitter which does not saturate 
the transistor is denoted by V.sub.S, the potential V.sub.A at a point A 
which supplies a current to the current mirror circuit 1 must assume a 
value which is greater than a potential V.sub.B -V.sub.BE1 +V.sub.S at the 
collector (point C) of the transistor Q.sub.1 by a quantity of two stages 
of V.sub.BE of the transistors Q.sub.3 and Q.sub.5, i.e., 
EQU V.sub.A .gtoreq.V.sub.B +V.sub.BE +V.sub.S (7) 
Practical values at room temperature are V.sub.B =1.2 V, V.sub.BE =0.7 V, 
and V.sub.S =0.2 V. Therefore, the relation V.sub.A .gtoreq.2.1 V must 
hold true. The voltage V.sub.A is supplied from the power-supply voltage 
V.sub.CC of the feedback amplifier 2a. Therefore, requirement of a high 
voltage V.sub.A means that the power-supply voltage V.sub.CC must be high. 
Symbols R.sub.3 and R.sub.4 denote resistors of the output stage, which 
feed base currents to the transistors Q.sub.1 and Q.sub.2. 
FIG. 3 is a circuit diagram illustrating a fundamental setup of the present 
invention, in which the same portions are denoted by the same symbols. 
What makes the circuit of FIG. 3 different from the circuit of FIG. 1 is 
that the second resistor R.sub.2 is connected between the output terminal 
B and a point D where bases of the transistors Q.sub.1, Q.sub.2 are 
commonly connected; this resistor is denoted by R.sub.12. Further, a 
transistor (or a diode) Q.sub.6 is connected between the point D where the 
bases are commonly connected and ground, so that the electric current 
I.sub.2 will flow through the second resistor R.sub.12 in proportion to 
the absolute temperature. The transistor Q.sub.6 forms a current mirror 
circuit together with the transistor Q.sub.1. It is therefore possible to 
flow an electric current which is proportional to the ratio of emitter 
areas of the two transistors. In other words, it is possible to adjust the 
current flowing through the resistor R.sub.12 to become equal to the 
current I.sub.2 of FIG. 1. Consequently the above-mentioned relation (1) 
holds true even with the circuit of FIG. 3. Therefore, the temperature 
characteristics of V.sub.BE1 of the transistor Q.sub.1 are compensated by 
the temperature characteristics of voltage drop I.sub.2 R.sub.12 across 
the resistor R.sub.12, and the reference voltage V.sub.B (=1.2 V) is 
maintained constant as shown in FIG. 2. Further, since the emitter of the 
transistor Q.sub.1 can be grounded, the potential at the point C can be 
lowered to V.sub.S, and the potential V.sub.A at the point A can be 
lowered to, 
EQU V.sub.A .gtoreq.2V.sub.BE +V.sub.S (8) 
If the aforementioned numerical figures are inserted V.sub.A .gtoreq.1.6 V; 
i.e., the power-supply voltage V.sub.CC can be lowered by 0.5 V as 
compared with the case of the relation (7). As is well known, the power 
supply of the integrated circuits has a small voltage, and is often 
established by storage cells. Therefore, the decrease of the power-supply 
voltage by 0.5 volt gives such a great effect that the number of storage 
cells can be reduced, for example, from three to two. 
The resistor R.sub.4 works to reduce the potential difference (1.6-1.2) V 
between V.sub.A and V.sub.B. The resistor R.sub.4, however, may be 
replaced by a diode or a transistor. FIG. 4 illustrates an embodiment of a 
circuit based upon the fundamental setup of FIG. 3, in which symbols 
Q.sub.8 and Q.sub.9 denote transistors which comprise an amplifier 2a, and 
C.sub.1 denotes a capacitor for compensating the phase. Further, a 
resistor R.sub.S connected between the power supply V.sub.CC and the point 
A has a high resistance and works to start the operation. The emitter area 
of the transistor Q.sub.2 is set to be, for example, 5 times (.times.5) 
that of the transistor Q.sub.1. In the embodiment of FIG. 4, a potential 
difference of about 0.7 V is maintained between V.sub.A and V.sub.B by a 
diode D.sub.1. 
FIG. 5 illustrates a modified embodiment of the fundamental setup of FIG. 
3. What makes the circuit of FIG. 5 different from the circuit of FIG. 3 
is that a series circuit comprising the transistor Q.sub.2 and the 
resistor R.sub.1 is connected in series with the collector of the 
transistor Q.sub.3, the collector of the transistor Q.sub.1 is connected 
in series with the base of the transistor Q.sub.3, and the feedback 
amplifier 2b is fed back to the potential V.sub.A from the collector of 
the transistor Q.sub.2. In this case, the input phase and the output phase 
of the amplifier are reversed relative to each other. The principle of 
operation, functions and effects are quite the same as those in the case 
of FIG. 3. FIG. 6 illustrates an embodiment of the setup of FIG. 5, 
wherein a transistor Q.sub.10 works as a feedback amplifier, and its 
output phase and the input phase are reversed relative to each other. 
FIG. 7 illustrates a modified embodiment of FIG. 4, in which a transistor 
Q.sub.7 is used in place of the resistor R.sub.4 that is employed in FIG. 
3, and transistors Q.sub.8 and Q.sub.9 form an amplifier. This circuit 
features a large output current since the transistor Q.sub.7 is connected 
in a manner of emitter follower. FIG. 8 illustrates a further modified 
embodiment of FIG. 4. Namely, the circuit of FIG. 8 does not have the 
transistor Q.sub.3 and the diode D.sub.1 that are used in the circuit of 
FIG. 4, and requires a further decreased power-supply voltage V.sub.CC. 
FIGS. 9A and 9B illustrate important portions of the embodiment of FIG. 3 
when the offset compensation is effected. The reference voltage generator 
circuit of this type is constructed in the form of a semiconductor 
integrated circuit, and an offset voltage (usually on the order of several 
millivolts) is generated in the voltages V.sub.BE of the transistors 
Q.sub.1 and Q.sub.6. Symbols R.sub.E1 and R.sub.E2 are small resistances 
which are inserted in the side of the emitter to cancel the offset 
voltage. These resistances generate voltages which are sufficient to 
cancel the offset voltages. 
According to the present invention as mentioned in the foregoing, the 
power-supply voltage of a band-gap reference circuit can be lowered, and 
the number of storage cells can be reduced from, for example, three to 
two. Or, even when the same number of storage cells are used, for example, 
even when two storage cells are used, the circuit can be operated 
maintaining sufficient margin.