Semiconductor switching device with leakage current detecting junction

First main electrodes of second and third semiconductor elements are connected to the first main electrode of the first semiconductor element, control electrodes of the second and third semiconductor elements are connected to the control electrode of the first semiconductor element, a second main electrode of the second semiconductor element is connected to a first resistor, and a second main electrode of the third semiconductor element is connected to a second resistor. Second main-electrode voltages of the first and second semiconductor elements are compared with each other by a first comparator, a control voltage is supplied a control voltage to the control electrodes of the first and second semiconductor elements according to an output of the first comparator by control means. Second main-electrode voltages of the first and third semiconductor elements are compared with each other by a second comparator. The transistor widths of the second and third semiconductor elements are each smaller than the transistor width of the first semiconductor element. The first semiconductor element is connected to power supply elements in parallel with one another between a power source and a load, and a weak current is detected when the load and power supply elements are OFF. There is no shunt resistor when detecting an overcurrent or a weak current. Accordingly, a heat loss is minimized and detective sensitivity is improved.

CROSS REFERENCE TO RELATED APPLICATIONS
 The subject application claims benefit of the earlier filing dates of
 Japanese Patent Application Nos.Hei 11-74259 and 2000-32364 filed on Feb.
 14, 1999 and Feb. 9, 2000 under the Paris Convention, the entire contents
 of which are incorporated by reference herein.
 BACKGROUND OF THE INVENTION
 1. Field of the Invention
 The present invention relates to a semiconductor switching device provided
 with a weak-current detecting function.
 2. Description of the Related Art
 It may be shown as a conventional semiconductor switching device with a
 current detection function in FIG. 1. This switching device is installed
 between a power 101 and a load 102. This device has the structure which
 connected a shunt resistor RS with the switching device which consists of
 power FETs, etc. in series. Usual the shunt resistor and the switching
 device are included in the identical substrate. The current detection is
 carried out using the potential difference which arises in shunt resistor
 double end. The potential difference arises in the double end of shunt
 resistor RS, when load current flows. This potential difference is
 amplified in a differential amplifier 911 and a direct current amplifier
 913. The amplified potential difference by going through an A/D converter
 902, it is read in the microcomputer, and the microcomputer judges
 overcurrents and less currents, etc., Negligible currents such as the
 leakage current can be also detected in theory by this method. Since the
 value of the detection current is small, it becomes a problem that the
 detection sensitivity is raised. In the reason, the countermeasure of
 raising the amplification factor of the direct current amplifier 913 and
 the differential amplifier 911 and of raising the value of the shunt
 resistor is required. It becomes a problem that the exothermic reaction of
 the shunt resistor as a load current flowed increases, when the shunt
 resistor is increased. In addition, there is a problem that the voltage
 supplied for the load lowers, when the voltage drop which arises in the
 shunt resistor increases. In the meantime, there is a problem that the
 detection accuracy becomes bad, since an S/N ratio deteriorates, when an
 amplification factor of the differential amplifier 911 and direct current
 amplifier 913 is raised. In method using this shunt resistor, when the
 value of detecting negligible current decreases, this problem becomes
 difficult and the accomplishing goal becomes difficult.
 An original function of the semiconductor switching device is to supply the
 electric power from the power source 101 in the load 102. Except for it,
 it has the function, namely the overcurrent protection function, when the
 trouble in which the wiring between the load 102 and the shunt resistor RS
 touched ground arose, which prevents that a large current flows for the
 wiring with the shunt resistor. It has the function which prevents that
 large current flows for the wiring with the shunt resistor. This
 overcurrent protection function becomes a essential, when the leakage
 current from the power line is measured using the shunt resistor. There is
 a case in which the semiconductor switching device contains heating and
 cutting off function. The semiconductor switching device contains a power
 element (main FET) QM, a resistor RG, a temperature sensor 121, a latch
 122 and an overheat breaking element FETQS, as it is shown in FIG. 2. It
 has heating and cutting off function which compulsorily turns off
 temperature sensor built-in FETQF by containing gate interception circuit,
 when the junction temperature of FETQF rises until exceeding the
 regulation value. Namely, if the temperature sensor 121 detects that the
 temperature of the power element QM increases above the predetermined
 value, the latch 122 holds the increased temperature information to turn
 on the breaking element QS, which forcibly turns off the power element QM.
 The temperature sensor 121 consists of four diodes that are connected in
 series, are made of, for example, polysilicon, and are integrated in the
 vicinity of the power element QM. As the temperature of the power element
 QM increases, a forward voltage-drop of the four diodes of the temperature
 sensor 121 decreases. The gate potential of an FET Q51 drops to low, the
 FET Q51 changes from ON to OFF. This pulls up the gate potential of an FET
 Q54 to the potential of a gate control terminal G of the element QF, to
 change the FET Q54 from OFF to ON. As a result, the latch 122 latches "1"
 to provide an output of high. This output changes the breaking elements QS
 from OFF to ON to short-circuit the true gate TG and source S of the power
 element QM. Consequently, the power element QM changes from ON to OFF.
 Namely, the power element QM is turned off.
 In FIG. 1, a zener diode ZD1 keeps a voltage of 12 V between the gate
 terminal G and source terminal S of the element QF and serves as a bypass
 for an overvoltage so that the overvoltage may avoid the true gate TG of
 the power element QM. The driver 901 has differential amplifiers 911 and
 913 serving as a current monitor circuit, a differential amplifier 912
 serving as a current limit circuit, a charge pump 915, and a driver 914.
 According to an ON/OFF control signal from the microcomputer 903 and an
 overcurrent signal from the current limit circuit, the driver 914 drives
 the true gate TG of the element QF. If an overcurrent exceeding an upper
 limit is detected through the differential amplifier 912 due to a voltage
 drop at the shunt resistor RS, the driver 914 makes the element QF
 nonconductive. If the current again decreases below the upper limit, the
 driver 914 makes the element QF conductive. On the other hand, the
 microcomputer 903 always monitors a current through the current monitor
 circuit made of the differential amplifiers 911 and 913, and if detects an
 abnormal current exceeding a normal value, makes the driver 914 turn off
 the element QF. If the temperature of the element QF exceeds a
 predetermined value before the microcomputer 903 issues an OFF instruction
 to the driver 914, the overheat breaking function mentioned above turns
 off the element QF.
 The prior art must employ the shunt resistor RS connected to a power line
 in series, to detect a current in the power line. When a current supplied
 from a power source to a load through the power line is large, the shunt
 resistor RS causes a large heat loss that is unignorable.
 To detect a weak current in the power line, the resistance value of the
 shunt resistor RS must be large. This, however, produces a large amount of
 heat when a large current is passed through the power line. To avoid this,
 the resistance value of the shunt resistor RS must be decreased, and then,
 it becomes difficult to detect a weak current in the power line.
 The prior art detects a current in the power line according to a voltage
 drop at the shunt resistor RS. To achieve this, the prior art must have a
 current monitor circuit involving the shunt resistor RS, A/D converter
 902, microcomputer 903, etc. These parts need a large space and are
 expensive, thereby increasing the size and cost of the switching device.
 Even if the load controlled by the switching device is OFF, a leak current
 will flow if the power line between the load and the power source causes a
 grounding fault or a short circuit due to, for example, abrasion,
 moisture, or corrosion. If no measures are taken, the leak current will
 increase to cause a fire. It is strongly required, therefore, to monitor a
 weak leak current while the load is OFF.
 SUMMARY OF THE INVENTION
 An object of the present invention is to provide a semiconductor switching
 device having a weak-current detecting function, capable of detecting a
 weak leak current in a power line without a shunt resistor connected in
 series to the power line. This switching device minimizes a heat loss, is
 easy to integrate, and is manufacturable at low cost.
 In order to accomplish the object, the present invention provides an
 apparatus for detecting a weak current, comprising a switching device that
 consists of a first semiconductor element having a first main electrode, a
 second main electrode and a control electrode, a second semiconductor
 element having a first main electrode connected to the first main
 electrode of the first semiconductor element, a control electrode
 connected to the control electrode of the first semiconductor element and
 a second main electrode connected to a first resistor, a first comparator
 for comparing second main-electrode voltages of the first and second
 semiconductor elements with each other, control means for supplying a
 control voltage to the control electrodes of the first and second
 semiconductor elements according to an output of the first comparator, a
 third semiconductor element having a first main electrode connected to the
 first main electrode of the first semiconductor element, a control
 electrode connected to the control electrode of the first semiconductor
 element and a second main electrode connected to a second resistor, and a
 second comparator for comparing second main-electrode voltages of the
 first and third semiconductor elements with each other. The transistor
 widths of the second and third semiconductor elements are each smaller
 than the transistor width of the first semiconductor element. The
 switching device is connected to power supply elements in parallel with
 one another between a power source and a load. The switching device is
 operated alone to detect a weak current when the load and power supply
 elements are OFF.
 The first to third semiconductor elements may be FETs (field-effect
 transistors), SITs (static induction transistors), or BJTs (bipolar
 junction transistors). They may also be ESTs (emitter switched
 thyristors), MOS composite elements such as MCTs (MOS-controlled
 thyristors), or insulated-gate power elements such as IGBTs (insulated
 gate bipolar transistors). The semiconductor elements may be either of n-
 and p-channel types. The first main electrode is any one of emitter and
 collector electrodes in the case of BJTs and IGBTs, and any one of source
 and drain electrodes in the case of IGFETs such as MOSFETs or MOSSITs. The
 second main electrode is the other of the emitter and collector electrodes
 in the case of BJTs and IGBTs, and the other of the source and drain
 electrodes in the case of IGFETs. If the first main electrode is an
 emitter electrode, the second main electrode is a collector electrode, and
 if the first main electrode is a source electrode, the second main
 electrode is a drain electrode. The control electrode is a base electrode
 in the case of BJTs, or a gate electrode in the cage of IGBTs and IGFETs.
 The power supply controller is a current oscillation switching circuit. The
 first semiconductor element of the power supply controller may be a power
 MOSFET. The present invention detects the difference between a terminal
 voltage of the first semiconductor element and a terminal voltage
 (reference voltage) of the second semiconductor element, and according to
 the difference, determines a deviation of the terminal voltage of the
 first semiconductor element, which is in the power supply path, from a
 normal level. This is equal to determining a deviation of a current
 passing through the power supply path from a normal current. By connecting
 the current oscillation switching circuits, i.e., the power supply
 controllers in parallel with one another, a large current will be handled.
 To detect a weak current, a reference voltage corresponding to the weak
 current may optionally be set.
 The present invention needs no current-detecting shunt resistor of the
 prior art in a power supply path, thereby suppressing a heat loss. The
 present invention is capable of responding to not only an overcurrent due
 to a dead short but also an abnormal current due to an incomplete short
 circuit such as layer short that involves a certain extent of
 short-circuit resistance. The present invention detects an overcurrent
 without a shunt resistor and carries out ON/OFF control of the
 semiconductor elements with a hardware circuit without a microcomputer,
 thereby reducing the space and cost of the switching device.
 The weak-current detector is also a current oscillation switching circuit
 provided with a weak-current detecting function. In the following
 explanation, the power supply controller, weak-current detector, and
 current oscillation switching circuit are equally handled.
 The effect of the present invention improves in case that an ON resistance
 value provided when the switching device is operated alone is set to be
 greater than an ON resistance value provided when the power supply
 elements are operated alone. This increases a voltage generated in the
 switching device, thereby improving the detecting accuracy of a weak
 current.
 Other and further objects and features of the present invention will become
 obvious upon an understanding of the illustrative embodiments about to be
 described in connection with the accompanying drawings or will be
 indicated in the appended claims, and various advantages not referred to
 herein will occur to one skilled in the art upon employing of the
 invention in practice.

DESCRIPTION OF THE PREFERRED EMBODIMENTS
 Various embodiments of the present invention will be described with
 reference to the accompanying drawings. It is to be noted that the same or
 similar reference numerals are applied to the same or similar parts and
 elements throughout the drawings, and the description of the same or
 similar parts and elements will be omitted or simplified.
 FIG. 3 shows a switching device according to an embodiment of the present
 invention. The switching device has current oscillation switching circuits
 or power supply controllers 110a to 110c for passing a current to drive a
 load 102 and a current oscillation switching circuit or a weak-current
 detector 114 for detecting a weak current such as a leak current. The
 switching circuits 110a to 110c and 114 are connected in parallel with one
 another and are arranged in a path between the load 102 are a power source
 VB. The switching circuits 110a to 110c have terminals T2 connected to a
 switch SW1 and a resistor R10. The other end of the switch SW1 is
 grounded, and the other end of the resistor R10 is connected to the power
 source VB. A terminal T12 of the switching circuit 114 is connected to a
 switch SW11 and a resistor R101. The other end of the switch SW11 is
 grounded, and the other end of the resistor R101 is connected to the power
 source VB. When the switch SW1 is ON, the switching circuits 110a to 110c
 are conductive. Each load 102 incorporates a switch, and when this switch
 is ON, the load 102 passes a current. If the load 102 is entirely OFF, no
 normal load current flows even if the switch SW1 is ON. Under this state,
 the switch SW1 is turned off and the switch SW11 on, to turn off the
 switching circuits 110a to 110c and on the switching circuit 114. The
 switching circuit 114 will pass no current if there is no leak current in
 the line between the switching circuit 114 and the load 102 or in the load
 102 itself. If there is a leak current in the line between the switching
 circuit 114 and the load 102 or in the load 102 itself, the switching
 circuit 114 will pass the sum of such leak currents, which is detectable.
 The number of parallel-connected switching circuits 110a to 110c is
 dependent on a current to be passed to the load 102, etc. When a large
 current is passed to the load, the number of switching circuits is
 increased to reduce a voltage drop in the switching device. A fuse FL of
 FIG. 3 is not always required.
 If the switch SW1 is ON and the switch SW11 OFF, the terminals T1 and T3 of
 the switching circuits 110a to 110c are conductive to apply the power
 source voltage to the load 102. Each load 102 has its own switch, which is
 turned on to drive the load 102 and is turned off to stop the load 102
 irrespective of the power source voltage. Namely, the switching circuits
 110a to 110c serve to apply the power source voltage to the load 102 and
 are not intended to switch on and off the load 102. If the switch SW1 is
 OFF and SW11 ON, the terminals T1 and T3 of the switching circuits 110a to
 110c become nonconductive, to stop the power source voltage to the load
 102. At this time, if the switch of the load 102 itself is OFF, a current
 flowing between the terminals T11 and T13 of the switching circuit 114 is
 only a leak current from the load 102. The switching circuit 114 compares
 the leak current with a reference value and provides a resultant signal.
 If the switch of the load 102 itself is ON, a load driving current
 overlaps the leak current, and the overlapping currents flow to the
 switching circuit 114. Then, the switching circuit 114 is unable to detect
 the leak current.
 The switching circuits 110a to 110c serve to minimize a voltage drop with
 respect to a load current. If the switching circuits 110a to 110c are ON
 when detecting a leak current, the leak current is distributed to the
 switching circuits 110a to 110c and becomes very weak. Then, it is very
 difficult for the switching circuit 114 to detect the leak current. To
 avoid this and to fully pass a leak current to the switching circuit 114,
 the present invention employs the circuit configuration mentioned above.
 FIG. 4 is a circuit diagram showing a current oscillation switching circuit
 110 employed by the switching device of the present invention. The
 switching circuit 110 of FIG. 4 corresponds to any one of the switching
 circuits 110a to 110c of FIG. 3. The switching circuit 110 has a first
 semiconductor element QA serving as a main power element, and a
 controller. The controller detects an abnormal current flowing to the
 first semiconductor element QA, turns on and off the first semiconductor
 element QA in response to the abnormal current, to cause current
 oscillations to break a conductive state of the first semiconductor
 element QA. The first semiconductor element QA and controller are
 integrated on a single substrate. The first semiconductor element QA and
 controller may form a hybrid IC employing an insulating substrate made of
 ceramics or glass epoxy, or an insulating metal substrate. Preferably, the
 first semiconductor element QA and controller are integrated into a
 monolithic power IC formed on a single semiconductor substrate (a single
 chip).
 The power IC is connected between the power source 101 for supplying an
 output voltage VB and the load 102 such as a lamp. In FIG. 4, the first
 semiconductor element QA serving as a main element of the switching
 circuit 110 has a function of sensing heat and breaking the switching
 circuit 110 in response to the sensed heat. The first semiconductor
 element QA of FIG. 4 may be that of FIG. 2 involving a temperature sensor.
 In the following explanation, the first semiconductor element QA is that
 of FIG. 2 associated with a temperature sensor. The sensing and breaking
 function is not needed. The first semiconductor element QA has a first
 main electrode, second main electrode, and control electrode. The first
 semiconductor element QA may be a power MOSFET of, for example, DMOS
 structure, VMOS structure, or UMOS structure, or a MOSSIT having a similar
 structure. The first semiconductor element QA may be a MOS composite
 element such as an EST or an MCT, or an insulated gate power element such
 as an IGBT. If the gate of the first semiconductor element QA is always
 with a reverse bias, the first semiconductor element QA may be a junction
 FET, a junction STI, or a SI thyristor. The first semiconductor element QA
 may be of any one of n-and p-channel types. Namely, the current
 oscillation switching circuit 110 may be of an n-channel type or a
 p-channel type.
 In FIG. 4, the switching circuit 110 is of an n-channel and is
 monolithically formed on a single semiconductor substrate. The switching
 circuit 110 consists of the first semiconductor element QA and a second
 semiconductor element QB that is connected in parallel with the first
 semiconductor element QA. The second semiconductor element QB serves as a
 reference element and may be an FET. The switching circuit 110 also has a
 comparator CMP1 for comparing a main electrode voltage of the first
 semiconductor element QA with a main electrode voltage of the second
 semiconductor element QB, and a driver 111 for supplying a control voltage
 to the control electrodes of the first and second semiconductor elements
 QA and QB. Each of the first and second semiconductor elements QA and QB
 has first and second main electrodes. The main electrode is any one of the
 emitter and collector regions of an IGBT, or any one of the source and
 drain regions of a power IGFET such as a power MOSFET or a power MOSSIT.
 The second main electrode is the other of the emitter and collector
 regions, or the other of the source and drain regions. If the first main
 electrode is an emitter region, the second main electrode is a collector
 regions, and if the first main electrode is a source region, the second
 main electrode is a drain region. The control electrode is a gate
 electrode in IGBT or power IGFET. The first and second semiconductor
 elements QA and QB have similar current-voltage characteristic curves and
 the same main and control electrode configurations.
 In this embodiment, the first semiconductor element QA is that of FIG. 2
 associated with a temperature sensor. In FIG. 2, the first semiconductor
 element QA has a power element QM which is a power FET, a resistor RG
 connected to a true gate TG of the power element QM, a temperature sensor
 121, an FET Q51 whose gate is connected to the temperature sensor 121, a
 latch 122 connected to the output side of the FET Q51, and an overheat
 breaking element QS which is an FET whose gate is connected to the output
 side of the latch 122. The output side of the breaking element QS is
 connected to the true gate of the power element QM. The power element QM
 may be a multi-channel power element consisting of a plurality of unit
 cells. The power element QM is connected in parallel with the second
 semiconductor element QB. Namely, the second semiconductor element QB is
 arranged in the vicinity of the first semiconductor element QA. The second
 semiconductor element QB may have no temperature sensor, latch, or
 overheat breaking element. Since the second semiconductor element QB is
 formed adjacent to the power element QM in the same process, no variation
 will be involved in the electric characteristics of the second
 semiconductor element QB and power element QM due to temperature drift or
 lot irregularities. The current capacity of the second semiconductor
 element QB is set to be smaller than that of the power element QM by
 adjusting the number of parallel-connected unit cells in the second
 semiconductor element QB. For example, the second semiconductor element QB
 has one unit cell, while the power element QM is composed of 1000 unit
 cells. In this case, the ratio of the channel width of the second
 semiconductor element QB to that of the power element QM is 1:1000. The
 temperature sensor 121 consists of diodes connected in series. These
 diodes are formed from, for example, a polysilicon thin film deposited on
 an interlayer insulation film formed on the second semiconductor element
 QB and power element QM. The temperature sensor 121 is integrated around a
 channel region of the power element QM. As the temperature of the power
 element QM increases, a forward voltage-drop of the four diodes of the
 temperature sensor 121 decreases. This drops the gate potential of an FET
 Q51 to low to change the FET Q51 from ON to OFF. This pulls up the gate
 potential of an FET 54 to the potential of a gate control terminal G of
 the first semiconductor element QA, to change the FET Q54 from OFF to ON.
 Then, the latch 122 latches "1" and provides an output of high level to
 change the overheat breaking element QS from OFF to ON. As a result, the
 change makes to short-circuit the true gate TG and source S of the power
 element QM. Namely, the power element QM changes from ON to OFF. In this
 way, the power element QM is interrupted due to overheat.
 Returning to FIG. 4, the current oscillation switching circuit 110 has a
 second semiconductor element QB, resistors R1, R2, R5, R8, RG, Rr, a zener
 diode ZD1, a diode D1, the comparator CMP1, and the driver 111. These
 parts and the first semiconductor element QA are monolithically formed on
 the same semiconductor substrate. The zener diode ZD1 keeps a voltage of
 12 V between the gate terminal G and source terminal S of the first
 semiconductor element QA to bypass an overvoltage so that the overvoltage
 may not be applied to the true gate TG of the power element QA. A resistor
 R10 and a switch SW1 are arranged outside the switching circuit 110. It is
 desirable that the resistor Rr is arranged outside the switching circuit
 110.
 The driver 111 has a source transistor Q5 whose collector is connected to a
 potential VP and a sink transistor Q6 whose emitter is connected to a
 ground potential GND. The transistors Q5 and Q6 are connected in series.
 In response to an ON/OFF signal from the switch SW1, the transistors Q5
 and Q6 are turned on and off to supply a drive control signal to the
 control electrodes of the first and second semiconductor elements QA and
 QB. The driver 111 in FIG. 4 consists of BJTs. Instead, the driver 111 may
 be made of MOSFETs. Forming the driver 111 with MOSFETs involves simple
 MOSFET manufacturing processes, thereby simplifying the manufacturing of
 the current oscillation switching circuit 110. Forming the driver 111 with
 BJTs involves BIMOS manufacturing processes. The output voltage VB of the
 power source 101 is, for example 12 V, and the charge pump output voltage
 VP is, for example, VB+10 V.
 The first main electrode (drain) of the first semiconductor element QA and
 the first main electrode (drain) of the second semiconductor element QB
 are connected to each other and are maintained at a common potential. The
 second main electrode (source) of the second semiconductor element QB is
 connected to a first resistor or a reference resistor Rr. The resistance
 of the reference resistor Rr is determined according to the ratio of the
 channel width of the second semiconductor element QB to that of the power
 element QM. For example, the ratio of the channel width of the second
 semiconductor element QB to that of the first semiconductor element QA is
 1:1000. In this case, the resistance of the reference resistor Rr is set
 to be 1000 times larger than the resistance of the load 102 under an
 overload state. Then, the second semiconductor element QB will generate a
 drain-source voltage V.sub.DS corresponding to an overload current that is
 caused by an abnormal operation and flows to the first semiconductor
 element QA.
 Between the first main (drain) and second main (source) electrodes of the
 first semiconductor element QA, there are resistors R1 and R2 connected in
 series. A main electrode voltage (source-drain voltage) V.sub.DS of the
 first semiconductor element QA is divided by the resistors R1 and R2, and
 the divided voltage is passed through the resistor R5 to a positive input
 terminal of the comparator CMP1. A negative input terminal of the
 comparator CMP1 receives a source voltage VS of the second semiconductor
 element QB. If potential at the positive input terminal of the comparator
 CMP1 is greater than potential at the negative input terminal thereof, the
 comparator CMP1 provides an output of high level, and therefore, the
 driver 111 provides a voltage to the gates controlled by the driver 111.
 If potential at the positive input terminal of the comparator CMP1 is
 smaller than potential at the negative input terminal thereof, the
 comparator CMP1 provides an output of low level, and therefore, the driver
 111 turns off the gates controlled by the driver 111. The comparator CMP1
 has predetermined hysteresis.
 FIG. 5 shows a minimum essential area of the current oscillation switching
 circuit of the switching device according to the present invention. The
 minimum area 113 includes the first semiconductor element QA and second
 semiconductor element QB which are made of identical transistor cells 119.
 Gate electrodes (not shown) of the transistor cells 119 are collectively
 connected to a terminal T9 of FIG. 4. Drain electrodes (not shown) of the
 transistor cells 119 are collectively connected to a terminal T6 of FIG.
 4. The first semiconductor element QA consists of 80 transistor cells 119
 whose source electrodes 118 are collectively connected to a terminal T7.
 The second semiconductor element QB consists of one transistor cell 119
 whose source electrode 118 is connected to terminals T8 and T10.
 In FIG. 5, the ratio of the number of cells in the second semiconductor
 element QB to that in the first semiconductor element QA is 1:80. Instead,
 the ratio may be 1:1000. In this case, an integer multiple of 1001, for
 example, 1001 or 4004 transistor cells are arranged in a matrix. In FIG.
 5, the 81 cells 119 are arranged adjacent to one another on a single chip
 so that they may simultaneously be processed. Then, even if manufacturing
 conditions fluctuate process by process, the cells may be processed under
 the same fluctuating conditions, to have an identical structure. Arranging
 the cells adjacent to one another on a single chip may provide the cells
 with an equal structure even if a manufacturing machine involves process
 unevenness chip by chip. The cell 119 serving as the second semiconductor
 element QB is positioned at the center of the 81 cells, so that the second
 semiconductor element QB may have a mean electric characteristic among the
 81 cells 119. If the electric characteristics of the cells 119 involve
 positions dependency over the chip, the central cell may have a mean
 electric characteristic.
 When the first semiconductor element QA is turned on to pass a current
 therethrough, the 80 cells 119 of the element QA generate heat. Even if
 the cells equally generate heat, there will be a temperature difference
 between a peripheral cell and the central cell. This temperature
 difference causes an electric characteristic difference. It is preferable
 that the cell 119 for the second semiconductor element QB has the same
 temperature and electric characteristic as the cells for the first
 semiconductor element QA. Accordingly, the cell 119 for the second
 semiconductor element QB must not be arranged at the periphery of the area
 113 where the temperature is low. A current passing through the cell 119
 for the second semiconductor element QB is substantially equal to a
 current passing through the cells 119 for the first semiconductor element
 QA, and therefore, heat generated by the cell 119 for the second
 semiconductor element QB is substantially equal to that generated by any
 one of the cells 119 for the first semiconductor element QA. In the
 operation as a weak-current detecting detecting element, there is no
 problem of the exothermic reaction, because the current is small.
 FIG. 6 shows the details of the current oscillation switching circuit 114
 having a function for detecting a weak current such as a leak current, of
 the switching device of the present invention. Compared with the structure
 of FIG. 4, the structure of FIG. 6 additionally has a semiconductor
 element QC, a resistor Rr2, and a comparator CMP2, to form a weak-current
 detector. The resistance value of the resistor Rr2 is determined by
 multiplying a load resistance value, which corresponds to a leak-current
 reference value, by a channel width ratio. For example, if the
 leak-current reference value is 60 mA under a power source voltage of 12
 V, the corresponding load resistance value is 200 .OMEGA.. If the channel
 width ratio is 1:1000, the resistance value of the resistor Rr2 is
 determined as 200 K.OMEGA.. The larger the resistance value of the
 resistor Rr2, the greater the sensitivity of detecting a leak current. The
 drain and gate of the element QC are connected to the drain and gate of a
 first semiconductor element QA, respectively. The source of the element QC
 is connected to a terminal of the resistor Rr2. The other end of the
 resistor Rr2 is grounded. A positive input terminal of the comparator CMP2
 receives a source voltage from the first semiconductor element QA, and a
 negative terminal thereof receives a source voltage from the element QC.
 If the voltage to the negative terminal is smaller than that to the
 positive terminal, the comparator CMP2 provides an output of high level to
 indicate that a voltage drop at the first semiconductor element QA is
 small and that a weak current such as a leak current to be detected is
 below a reference value. If the voltage to the negative terminal is higher
 than that to the positive terminal, the comparator CMP2 provides an output
 of low level to indicate that a voltage drop at the first semiconductor
 element QA is large and that a weak current to be detected is above the
 reference value. Increasing the resistance of the resistor Rr2 decreases a
 voltage drop at the element QC to make the reference value for a weak
 current smaller, thereby improving the detecting sensitivity of a leak
 current. An area 114 surrounded with a dotted line in FIG. 6 is an
 analog-integrated chip area.
 The elements QA and QC of FIG. 6 are essential parts for the current
 oscillation switching circuit 114 having the function of detecting a weak
 current, of the switching device of the present invention. The elements QA
 and QC of FIG. 6 are formed in one chip. The structure of the elements QA
 and QC of FIG. 6 is the same as that of the elements QA and QB of FIG. 5
 with the element QB being replaced with the element QC. Namely, the
 elements QA and QC are formed of identical transistors 119 that are
 connected as shown in FIG. 5. According to a difference .DELTA.ID between
 a weak current (leak current) and the reference value, as well as an ON
 resistance value Ron of the element QA, the comparator CMP2 detects a
 value of .DELTA.ID.times.Ron. If the difference .DELTA.ID is unchanged, a
 produced voltage increases as the ON resistance value Ron increases,
 thereby improving a weak current detecting sensitivity. To increase the ON
 resistance value Ron, the number of cells 119 that form the element QA may
 be reduced, the channel length of each cell 119 may be elongated to
 increase the ON resistance of the cell 119, or an impurity concentration
 may be decreased to increase the resistance of a semiconductor substrate.
 The switching circuit 114 is arranged as shown in FIG. 3. Since the
 switching circuit 114 needs no power supply, the ON resistance value Ron
 may be increased to improve the weak current detecting sensitivity.
 In FIG. 6, a zener diode ZD1 keeps a voltage of 12 V between the gate
 terminal G and source terminal S of the element QA to bypass an
 overvoltage so that the overvoltage may not be applied to the true gate TG
 of a power element QM integrated in the element QA (FIG. 2).
 FIG. 7 is an equivalent circuit diagram showing the first semiconductor
 element QA of the current oscillation switching circuit of the present
 invention. The element QA is represented with an equivalent current source
 g.sub.m V.sub.1, drain resistance rd, gate-source capacitance C.sub.GS,
 gate-drain resistance C.sub.GD, and drain-source resistance C.sub.DS. The
 equivalent circuit of FIG. 7 shows a power supply path from the power
 source 101 to the load 102. The load 102 includes wiring inductance LU and
 wiring resistance RU.
 FIG. 8 shows characteristic curves of drain-source voltage V.sub.DS of the
 first semiconductor element QA when the element QA changes from OFF to ON
 with the load 102 being short-circuited, being under a normal operating
 condition, and being in a state having a resistance value of 1 K.OMEGA..
 The characteristic curves are dependent on the impedance of the power
 supply path including the wiring inductance and resistance of the power
 supply path that includes the element QA.
 The drain-source voltage V.sub.DS curve with the load 102 having a
 resistance value of 1 K.OMEGA. will be explained. In this example, the
 first semiconductor element QA is HAF2001 of HITACHI company. Due to the
 characteristics of this element, a true gate-source voltage V.sub.TGS is
 substantially maintained at a threshold voltage of 1.6 V when a drain
 current I.sub.D is 12 mA. The driver 111 continuously charges the true
 gate TG of the first semiconductor element QA. Accordingly the true
 gate-source voltage V.sub.TGS will increase, if nothing happens. However,
 the drain-source voltage V.sub.DS decreases to increase the true
 gate-drain capacitance C.sub.GD, to absorb charge, thereby preventing the
 true gate-source voltage V.sub.TGS from increasing. As a result, the true
 gate-source voltage V.sub.TGS is maintained at about 1.6 V. Namely, after
 the first semiconductor element QA is changed to ON, the drain-source
 voltage V.sub.DS absorb charge supplied by the driver 111 to the gate G
 and maintains the voltage V.sub.TGS at the true gate TG at the constant
 value.
 If the load 102 has a resistance value of R that is lower than 1
 K.OMEGA.the difference between the drain-source voltage V.sub.DS with the
 load resistance value of 1 K.OMEGA. and that with the load resistance
 value of R is .DELTA.V.sub.DS GAP. When the load has the resistance value
 R, there is a true gate-source voltage of V.sub.TGS R. By detecting charge
 for Q.sub.GD of the following expression (7) from the true gate-source
 voltage V.sub.TGS R, the true gate-source voltage V.sub.TGS R becomes 1.6
 V.
EQU Q.sub.GD =.DELTA.V.sub.DS GAP.times.C.sub.GD +(V.sub.TGS R-1.6
 V).times.C.sub.GS (7)
 This means that the true gate-source voltage V.sub.TGS is increased from
 1.6 V by the charge Q.sub.GD as expressed as follows:
EQU (V.sub.TGS R-1.6V).times.C.sub.GS +((V.sub.TSG R-1.6 V)-.DELTA.V.sub.DS
 GAP).times.C.sub.GD
EQU =(.DELTA.V.sub.DS GAP-(V.sub.TGS R-1.6 V)).times.C.sub.GD
EQU (V.sub.TGS R-1.6 V)(C.sub.CS +2C.sub.GD)=.DELTA.V.sub.DS
 GAP.times.2C.sub.OD
EQU .DELTA.V.sub.DS GAP=(V.sub.TGS R-1.6 V)(C.sub.GS/ 2C.sub.GD +1) (1)
 Namely, .DELTA.V.sub.DS GAP is proportional to "V.sub.TGS R-1.6 V." If the
 drain current I.sub.D is zero, the drain-source voltage V.sub.DS curve is
 determined only by the circuit for charging the true gate and mirror
 capacitance. When the drain current I.sub.D flows, it generates counter
 electromotive force due to the whole inductance L.sub.C of the circuit.
 This provides the same effect as an increase in the resistance of the
 load. When the whole inductance L.sub.c changes, resistance that is
 equivalent to the inductance is produced, and therefore, the equivalent
 resistance value of the load never decreases below a constant value, which
 is determined by the circuit inductance, even if the pure resistance value
 of the load becomes very small due to, for example, a dead short. As a
 result, the gradient of a rise of the drain current I.sub.D converges to a
 constant value, and therefore, the true gate-source voltage V.sub.TGS
 curve also converges.
 If the ratio of the channel width of the second semiconductor element QB to
 that of the first semiconductor element QA is 1:1000 and if a drain
 current I.sub.DQA of the element QA is 5A and a drain current I.sub.DQB of
 the element QB is 5 mA, the drain-source voltage V.sub.DS is equal to the
 true gate-source voltage V.sub.TGS in each of the elements QA and QB.
 Namely, V.sub.DSA =V.sub.DSB and V.sub.TGSA =V.sub.TGSB, where V.sub.DSA
 and V.sub.DSB are the drain-source voltages of the elements QA and QB,
 respectively, and V.sub.TGSA and V.sub.TGSB are the true gate-source
 voltages of the elements QA and QB, respectively.
 When the second semiconductor element QB is completely ON, the ends of the
 first reference resistor Rr substantially receive the source voltage VB.
 Accordingly, the resistance of the first reference resistor Rr is
 determined as Rr=12 V/5 mA=2.4 K.OMEGA.. Here, the resistance of the first
 reference resistor Rr serves as load on the second semiconductor element
 QB and corresponds to the load of 5 A connected to the first semiconductor
 element QA.
 An operation of the switching device of the present invention in a triode
 region will be explained. When the first semiconductor element QA changes
 to ON, the drain current I.sub.DQA increases to a final load current value
 determined by circuit resistance. At this time, the true gate-source
 voltage V.sub.TGSA of the element QA is determined by the drain current
 I.sub.DQA. The true gate-source voltage V.sub.TGSA increases although it
 is braked by a mirror effect of the capacitance C.sub.GD due to a decrease
 in the drain-source voltage V.sub.DSA. The second semiconductor element QB
 operates as a source follower according to a gate voltage determined by
 the first semiconductor element QA with the resistor Rr serving as a load
 resistor.
 The true gate-source voltage V.sub.TGSA of the first semiconductor element
 QA increases as the drain current I.sub.DQA increases. V.sub.DSA
 =V.sub.TGSA +V.sub.TGD and V.sub.DSB =V.sub.TGSB +V.sub.TGD, and
 therefore, V.sub.DSA
 -V.sub.DSB=V.sub.TGSA-V.sub.TGSB=(I.sub.DQA-n.times.I.sub.DQB)/ Gm, where
 Gm is the transfer conductance of the first semiconductor element QA and n
 is a channel width ratio between QA and QB.
 Accordingly, by detection the difference "V.sub.DSA -V.sub.DSB " between
 the drain-source voltages, the drain current difference "I.sub.DOA
 -n.times.I.sub.DOB " is obtainable.
 The drain-source voltage V.sub.DSB of the second semiconductor element QB
 is directly supplied to the comparator CMP1. The drain-source voltage
 V.sub.DSA of the first semiconductor element QA is divided by the
 resistors R1 and R2, and the divided voltage is supplied to the comparator
 CMP1. The voltage V.sub.+ applied to the positive input terminal of the
 comparator CMP1 is expressed as follows without considering the variable
 resistor RV:
EQU V.sub.30 =V.sub.DSA.times.R1/(R1+R2) (2)
 If the load is normal, Rr/n&lt;R, where R is the load resistance.
 Accordingly, V.sub.+ &lt;V.sub.DSB to keep the first semiconductor element
 QA ON. If the load is in an overload state, then Rr/n&gt;R. Accordingly,
 the first semiconductor element QA is turned off in an ohmic region. Once
 the first semiconductor element QA is turned off, source potentials
 V.sub.SA and V.sub.SB of the first and second semiconductor elements QA
 and QB decrease toward the ground potential GND to increase V.sub.DSA and
 V.sub.DSB. Before V.sub.SA and V.sub.SB reach the ground potential GND,
 V.sub.30 &lt;V.sub.DSB is established to again turn on the first
 semiconductor element QA. Just after the first semiconductor element QA is
 turned on, the element QA is in a pinch-off region. The first
 semiconductor element QA keeps the ON state toward the ohmic region and is
 turned off when V.sub.30 &gt;V.sub.DSB is established. These operations
 form an ON/OFF cycle. Keeping an OFF state once turned off and keeping an
 ON state once tuned on are due to the inductance of the load. This
 inductance works like a resistor with respect to a current change. With
 respect to a decreasing current, the equivalent resistor has a negative
 sign to decrease the resistance of the load, and with respect to an
 increasing current, the equivalent resistor has a positive sign to
 increase the resistance of the load. This is the reason why the first
 semiconductor element QA keeps and ON or OFF state once it is turned on or
 off. The resistance Rr of the second semiconductor element QB of the
 reference circuit is n times greater than the load resistance R, and
 therefore, the inductance effect on the second semiconductor element QB is
 very small and ignorable. Accordingly, the circuit involving the second
 semiconductor element QB is considered to be a pure resistance circuit.
 For the comparator CMP1, the diode D1 and resistor R5 form hysteresis. When
 the first semiconductor element QA changes to OFF, the sink transistor Q6
 of the driver 111 grounds the gate potential, and the cathode potential of
 the diode becomes V.sub.SA -0.7 V (a forward voltage of the zener diode
 ZD1), to make the diode D1 conductive. As a result, current passes from
 the resistor R1 to the resistor R5 to the diode D1. The signal V.sub.+ to
 the positive input terminal of the comparator CMP1 becomes greater than a
 value determined by the expression (2) with the drive 111 carrying out ON
 control. The first semiconductor element QA keeps the OFF state until the
 drain-source voltage difference V.sub.DSA -V.sub.DSB reaches a
 predetermined level. Thereafter, V.sub.DSA increases further so that the
 signal V.sub.+ may become smaller than V.sub.DSB. Then, the output of the
 comparator CMP1 changes from low to high to again turn on the first
 semiconductor element QA. The comparator CMP1 may have various hysteresis
 characteristics. The hysteresis mentioned above is only an example.
 By setting the drain-source voltage V.sub.DSA of the first semiconductor
 element QA that changes to OFF as a threshold value V.sub.DSth, the
 following is established:
EQU V.sub.DSAth -V.sub.DSB =R2/R1.times.V.sub.DSB (3)
 An overcurrent determining value in the triode region is determined by the
 expression (3).
 An operation of the switching device of the present invention in the
 pentode region will be explained. If there is no trouble in wiring, the
 first semiconductor element QA keeps ON once it is turned on. Accordingly,
 after the true gate-source voltages V.sub.TGSA and V.sub.TGSB exceed each
 a pinch-off voltage, the elements QA, QB, and QC operate in the pentode
 region. In the case of HAF2001 of HITACHI company, the ON resistance
 R.sub.DS(ON) is 30 m.OMEGA. when the gate-source voltage V.sub.TGSA is 10
 V. Therefore, the following is obtained:
EQU V.sub.DSB =5A.times.30[m.OMEGA.]=0.15[V] (4)
EQU V.sub.DSA =I.sub.DQA.times.30[m.OMEGA.] (5)
EQU V.sub.DSA -V.sub.DSB =30[m.OMEGA.].times.(I.sub.DQA -5[A]) (6)
 If a wiring short circuit occurs to increase the drain current I.sub.DQA,
 the value of the expression (6) becomes larger, and if the drain current
 exceeds the overcurrent determination value, the first semiconductor
 element QA is changed to OFF. In this case, the first semiconductor
 element QA passes through the pinch-off point and the operation in the
 triode region and changes to OFF. Due to the hysteresis defined by the
 diode D1 and resistor R5 of FIG. 4, The signal V.sub.+ to the positive
 input terminal of the comparator CMP1 becomes smaller than V.sub.DSB after
 a predetermined period of time. As a result, the output of the comparator
 CMP1 changes from low to high, to again change the first semiconductor
 element QA to ON. In this way, the first semiconductor element QA is
 repeatedly turned on and off, and finally, is turned off due to overheat.
 When the wiring restores a normal state (in the case of an intermittent
 short circuit) before the first semiconductor element QA is turned off due
 to overheat, the element QA is made to keep ON.
 FIG. 9A shows a drain current I.sub.D of the current oscillation switching
 circuit of the present invention, and FIG. 9B shows a drain-source voltage
 V.sub.DS corresponding to the drain current I.sub.D. In FIGS. 9A and 9B,
 curves (3) represent an overload state and (2) a normal state. Under the
 overload state (3), the first semiconductor element QA is repeatedly
 turned on and off as mentioned above to largely fluctuate the drain
 current 1.sub.D as indicated with the curves (3). This results in
 cyclically heat the first semiconductor element QA, so that the element QA
 is turned off due to overheat.
 In summary, the switching device of the present invention needs no shunt
 resistor for detecting a current in a power supply path, thereby detecting
 negligible currents such as the leakage current at the good accuracy,
 reducing a heat loss and quickly responding to an abnormal current caused
 by an incomplete short circuit such as a layer short having a certain
 extent of short-circuit resistance. This switching device is easy to
 integrate and is manufacturable at low cost.
 The switching device of the present invention is capable of detecting a
 leak current that may flow even when the load is an OFF state and when
 other electric power supply controller is made to be an OFF state. Namely,
 the switching device of the present invention easily monitors an increase
 in the leak current.
 ON/OFF control parts of the switching device of the present invention may
 monolithically integrated to eliminate a microcomputer. This arrangement
 greatly reduces a chip area as well as cost.
 Compared with the prior art that monitors a drain-source voltage V.sub.DS
 and compares it with a threshold at predetermined timing to detect an
 overcurrent, the present invention needs reduced numbers of capacitors and
 resistors, thereby minimizing detection errors due to parts variations. In
 addition, the current oscillation switching circuit 110 of the present
 invention formed on a chip needs no external capacitor, thereby reducing
 the space and cost of the switching device.
 The present invention sets the current capacity of the second semiconductor
 element QB to be smaller than that of the first semiconductor element QA,
 so that the ratio of the resistance of load to that of the second resistor
 may substantially be inverse proportion to the ratio of the current
 capacity of the first semiconductor element QA to that of the second
 semiconductor element QB. This arrangement reduces the size of the second
 semiconductor element QB and second resistor, thereby reducing the size
 and cost of the switching device.
 The overheat protective function of the present invention turns off the
 switching device if the switching device overheats. When an incomplete
 short circuit having a certain extent of short-circuit resistance occurs,
 the first semiconductor element QA is repeatedly turned on and off to
 greatly vary a current. This cyclically heats the element QA, so that the
 overheat protective function may quickly turn off the element QA.
 Consequently, the switching device of the present invention quickly
 responds to an abnormal current caused by an incomplete short circuit.