Adjustable impedance matching network

An impedance matching network includes a first terminal, a second terminal, and a reference potential terminal. The impedance matching network further includes a first shunt branch between the first terminal and the reference potential terminal, the first shunt branch including a capacitive element. The impedance matching network also includes a second shunt branch between the second terminal and the reference potential terminal, the second shunt branch including an inductive element. Furthermore, the impedance matching network includes a transmission line transformer with a first inductor path and a second inductor path, wherein the first inductor path connects the first terminal and the second terminal. An alternative impedance matching network includes a transformer and an adaptive matching network. The transformer is configured to transform an impedance connected to a first port so that a corresponding transformed impedance lies within a confined impedance region in a complex impedance plane.

TECHNICAL FIELD

Embodiments of the present invention relate to an impedance matching network. Further embodiments of the present invention relate to a method for matching an impedance. Further embodiments of the present invention relate to an antenna circuit. Further embodiments of the present invention relates to a monolithically, fully integratable matching network with broad impedance range. Further embodiments of the present invention relate to an adaptive matching circuit with pre-transformation.

BACKGROUND

In electrical or electronic systems, it is often desirable to design the input impedance of an electrical load (or the output impedance of a source of electrical energy) to maximize the power transfer and/or minimize reflections from the load. Maximum power transfer is typically obtained when the load impedance is equal to the complex conjugate of the source impedance. In contrast, minimum reflection can typically be achieved when the load impedance is equal to the source impedance.

Current radio frequency (RF) or high-frequency (HF) front end systems typically still comprise, at the transmitter end, a power amplifier (PA) for amplifying the signal to the required level, a filter (typically a harmonic filter), a power detector, and an antenna switch that performs a switching between the transmission bands, the reception bands, as well as between transmitter operation and receiver operation. Thereafter the signal is typically forwarded via an antenna impedance matching network to the antenna.

This antenna impedance matching is designed that, averaged over all use cases, frequencies and operating modes, as well as over their respective probabilities, an optimum is achieved. As can readily be seen, the optimum is only reached very seldom, because the frequency spectrum of the mobile communications frequencies constantly becomes broader and also the antenna itself provides very different matching for all frequencies and environmental conditions that may occur.

SUMMARY OF THE INVENTION

Embodiments of the present invention provide an impedance matching network comprising a first terminal, a second terminal, a reference potential terminal, a first shunt branch, a second shunt branch, and a transmission line transformer. The first shunt branch extends between the first terminal and the reference potential terminal. The first shunt branch comprises a capacitive element. The second shunt branch extends between the second terminal and the reference potential terminal. The second shunt branch comprises an inductive element. The transmission line transformer comprises a first inductor path and a second inductor path, where in the first inductor path connects the first terminal and the second terminal.

Further embodiments of the present invention provide an impedance matching network comprising a first port, a second port, a transformer, and an adaptive matching network. The transformer is configured to transform an impedance connected to the first port so that a corresponding transformed impedance lies within a confined impedance region in a complex impedance plane. The adaptive matching network is at adjustable to match the transformed impedance located anywhere within the confines in ketones region to a second impedance connected to the second port.

According to further embodiments, an antenna circuit comprises an antenna, a signal terminal configured to relay a signal to a receiver or from a transmitter, and an impedance matching network. The impedance matching network interconnect the antenna and the signal terminal and comprises a Pi-network having a first inductor path of a transmission line transformer in a series branch of the Pi-network.

According to further embodiments, an antenna circuit comprises an antenna, a signal terminal configured to relay a signal to a receiver or from a transmitter, and an impedance matching network interconnecting the antenna and the signal terminal. The impedance matching network comprises a transformer configured to transform an impedance connected to the first port so that a corresponding transformed impedance lies within a confined impedance region in a complex impedance plane. The impedance matching network further comprises an adaptive matching network that is adjustable to match the transformed impedance located anywhere within the confined impedance region to a second impedance connected to the second port.

Further embodiments of the present invention provide a method for matching an impedance using an impedance matching network that comprises a first shunt branch with a capacitive element, a second shunt branch with an inductive element, and a transmission line transformer with a first inductor path and a second inductor path. The first inductor path (inter-) connects the first terminal and the second terminal. The method comprises at adjusting a real part of the impedance by adjusting a transmission ratio of the transmission line transformer. The method further comprises at adjusting an imaginary part of the impedance by adjusting at least one of the capacitance element and the inductive element.

According to further embodiments of the present invention a method for matching an impedance connected to a first port of an impedance matching network is provided. The method comprises transforming the impedance to a corresponding transformed impedance that lies within a confined impedance region in a complex impedance plane. This transforming is performed by (or using) a transformer of the impedance matching network. The method also comprises matching the transformed impedance located anywhere within the confined impedance region to a second impedance connected to a second port off the impedance matching network. This matching is performed by an adaptive matching network or using an adaptive matching network.

Equal or equivalent elements or elements with equal or equivalent functionality are denoted in the following description by equal or similar reference numerals.

DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS

In the following description, a plurality of details is set forth to provide a more thorough explanation of the embodiments of the present invention. However, it will be apparent to one skilled in the art that embodiments of the present invention may be practiced without these specific details. In other instances, well-known structures and devices are shown in block diagram form rather than in detail in order to avoid obscuring embodiments of the present invention. In addition, features of the different embodiments described hereinafter may be combined with each other, unless specifically noted otherwise.

FIG. 1shows a schematic block diagram of a simplified radio frequency (RF) or high-frequency (HF) frontend system as it may be used, for example, in a mobile station or a base station of a mobile communications network, such as a cell phone, a smart phone, a tablet computer, an USB wireless modem, a wireless router, or a base transceiver station. The front end system comprises a transceiver2, a power amplifier (PA)3, a harmonic filter4, an antenna switch, an antenna matching network6, and an antenna7. When functioning in a transmitter operating mode, the transceiver2provides a transmit signal at its output TX to the power amplifier3. An amplified transmit signal provided by the power amplifier3is fed to the harmonic filter4, which reduces frequency components of the amplified transmit signal outside an intended transmit frequency range. An output of the harmonic filter4is connected to one of the plurality of inputs of the antenna switch5. In the example ofFIG. 1, the antenna switch5is currently configured to connect said input to an antenna switch output. The other inputs of the antenna switch5may be connected to respective outputs of further harmonic filters (not illustrated inFIG. 1) having a different frequency response than the harmonic filter4so that the HF frontend system illustrated inFIG. 1may be configured to support several transmit frequencies and/or several mobile communications standards. The antenna switch5is further configured to connect the antenna matching network6with an RX input (i.e., a receiver input) of the transceiver2via a connection8, when the HF frontend system operates in a receiver mode.

The antenna switch output is connected to an input of the antenna matching network6. The antenna matching network6is, in the illustrated example, implemented as a basic LC network comprising a series inductance and a capacitance connected in parallel to an output of the antenna matching network6(i.e., a “shunt capacitance”). The output of the antenna matching network6is connected to the antenna7.

As the HF frontend system can be reconfigured by means of the antenna switch5to support several frequencies, mobile communications standards, and/or further parameters related to the transmission or reception of radio signals, the antenna matching network6has to be selected taking into account the various possible use cases, frequencies and operating modes, as well as their respective probabilities in order to provide a weighted optimum. This task becomes more and more difficult, because the frequency spectrum of the mobile communications frequencies is getting wider and wider and also the antenna itself requires different impedance matching settings for different environmental conditions. In addition, the problem of antenna mismatch due to the different environment of the antenna has to be considered. For example, the impedance of an antenna may vary very strongly, when the antenna is being touched by, e.g., a finger of a mobile phone user, as could be observed with some mobile phone models sold in the past. Moreover, a mismatch leads to additional non-linearities at the power amplifier3and to a modification of the filter behavior of the harmonic filter4, because the mismatch is passed through the antenna switch5. In consequence, the overall system is negatively affected at several points by the mismatch. These problems are more exactly considered only recently, because mobile communications systems used to be specified for 50 Ohm measuring systems, only.

For some time past, additional requirements with respect to the total radiated power (TRP) exist, which have to be fulfilled by mobile communications devices for specific network providers. This means that for a wide range of mismatches the system has to reach the required radiated power. A mismatch occurs if the antenna impedance differs from the source impedance, e.g., the output impedance of the antenna switch5inFIG. 1.FIG. 2schematically illustrates a selection of different antenna impedances that have may occur during the operation of a wireless communications device in a Smith chart and hence may lead to a mismatch. Note that the possible impedances are located in a confined region centered to the origin of the Smith chart. An outer ring of the Smith chart plane does not contain any impedance values to be considered, i.e., the corresponding impedance region is not considered as very likely to occur during normal operation. Indeed, the outer ring of the Smith chart corresponds to impedances that differ significantly from a reference impedance Z0(e.g., 50 ohm) which is located at the center of the Smith chart, and are therefore rather unlikely to occur.

In other words,FIG. 2illustrates a plurality of test cases in a Smith chart representation that a HF or RF frontend system has to pass in order to be admitted for operation in certain mobile communications networks. In particular,FIG. 2shows the possible impedance inside a circle of a given VSWR (Voltage Standing Wave Ratio), here for VSWR=10. Thus, every mismatch below VSWR=10 may be a valid point, and as this can be infinite much, the amount of points is limited to 121 (11 lines with each line having 11 points) for the purpose of illustration inFIG. 2.

It can be expected that simple arrangements might not be sufficient in the future for meeting current and future requirements. For this reason, switchable matching networks are currently used in some first devices, where in accordance with the frequency being used a modification of the matching is performed.

FIG. 3illustrates a schematic block diagram of an HF front end system comprising an adjustable antenna matching network9. The adjustable antenna matching network9is in the example illustrated inFIG. 3configured as a Pi-network with tunable capacitances connected in parallel to the input and the output, respectively.

With respect to the impedance matching network, it has to be considered that with a given impedance matching network topology not every impedance may be realized, i.e., there are so called “forbidden regions.”FIG. 4illustrates some basic LC networks and below each basic LC network a corresponding schematic Smith diagram in which the forbidden region is represented as a hatched region. A load impedance ZLis connected to the various LC basic networks.

In order to cover a wide range of possible impedances, adjustable impedance matching networks typically have a Pi-topology (Π-topology) or a T-topology. A Pi-topology having a series inductance, a parallel input capacitance, and a parallel output capacitance forms a low pass filter which attenuates harmonics generation. Furthermore, variable capacitances are available (rotary capacitor, BSR capacitor (i.e., (Ba,Sr)RuO3capacitor) . . . ), whereas a variable inductance typically requires a variable tap.

FIG. 5schematically illustrates a Pi-topology of an adjustable impedance matching network.FIG. 6illustrates a corresponding diagram of the forward transmission S(2,1) over frequency, and a Smith chart of the input reflection coefficient S(1,1) and the output reflection coefficient S(2,2) as a function of the frequency. An impedance of 12.5 ohms shall be matched to an impedance of 50 ohms. The target frequency is 900 MHz. The Pi-structure is illustrated inFIG. 5. The series inductance of the Pi-structure is 4 nH with a series resistance of 2 ohms. The upper diagram illustrates the insertion loss S(2,1), and the Smith chart in the lower part ofFIG. 6illustrates the input reflection coefficient S(1,1) as a full stroke line, and the output reflection coefficient S(2,2) as a dashed line.

FIG. 7shows a schematic circuit diagram of an adjustable Pi-network comprising three adjustable capacitive elements and three constant inductive elements (of fixed inductance value). In comparison to the network structure ofFIG. 5, the network structure ofFIG. 7makes it possible to cover a larger range of the voltage standing wave ratio (VSWR) and also to cover a larger frequency range.

With a network having a Pi-structure, typically only the capacitances are adjusted, whereas the inductance is fixed and should have a high quality factor.

The problem of the Pi-structure is that typically only the capacitances are varied, whereas the inductance is constant and is required to have a high figure of merit or quality factor. In the case in which no matching has to be performed, that is 50 ohms are matched to 50 ohms, the circuit functions as a pure phase shifter. As can be seen when examining the circuit illustrated inFIG. 5, losses occur in this situation, i.e., it would be desirable to bridge the inductance. In cases in which it is not possible to bridge the inductance, an adjustable capacitance may be series-connected in order to reduce the effective series inductance. Unfortunately, this very action inevitably leads to a loss off quality factor, i.e., the imaginary part becomes smaller but the series resistance remains constant, at best (typically it even increases). Hence, the topologies illustrated inFIGS. 5 and 7form narrowband systems of first order, i.e., a good matching can be achieved in a very small frequency range only, or, alternatively, in a broader frequency range by accepting inferior figures of merit/quality factors, resulting in higher losses. For this reason, broadband solutions requiring as few components as possible, and/or possibly monolithically integratable, would be desirable.

This basically means that a multistage matching structure is chosen, as shown inFIG. 8, which obviously leads to several matching elements, or that a transformer is used. Particularly for impedance transformations in the real plane so-called Guanella or Ruthroff transformers are known in the literature. While these historically were implemented as macroscopic transformers with bifilar windings, they may also be realized as planar transformers on or within a printed circuit board, a silicon substrate, or a laminated structure.

As an example,FIG. 9shows a circuit schematic of a Ruthroff transformer functioning as a 1:4 impedance transformer which is assumed to have two identical inductances. The basic idea of these transformers is the combination of signal portions in an additive manner. In the case illustrated inFIG. 9, the output voltage is a sum of the voltage V2across the series inductance and of the voltage V1across the ground path. At the same time, the electric current is split up over the two inductances so that, due to a doubling of voltage and a halving of the electric current, an impedance transformation of 1:4 is achieved (RL, appears to be more high-ohmic to the voltage source Vg).

FIG. 10shows a 4:1 Ruthroff transformer structure within a Pi-network.FIG. 11shows a diagram of the forward transmission coefficient as a function of frequency, and a Smith chart of the input reflection coefficient S(1,1) and the output reflection coefficient S(2,2). The series inductance between the nodes3and4is LS=4 nH and has a series resistance of 2 ohms. The inductance in the path to ground is substantially equal, i.e., LP=4 nH with a series resistance of 2 ohms. The parallel input capacitance is Cs1=3.4 pF and the parallel output capacitance is Cs2=2 pF. The magnetic coupling factor between the series inductance and of the ground path inductance is k=0.8. Again, a small signal simulation (S parameters) has been performed and shows the behavior illustrated in the diagrams ofFIG. 8. The solution using a transformer typically has the following characteristics in comparison to, for example, a solution using a (classical) Pi-structure as shown inFIG. 5. For the sake of comparison, the same degree of matching as inFIG. 6is desired, namely matching of 12.5 ohms to 50 ohms. The target frequency is 900 MHz. As can be seen in the upper diagram ofFIG. 11illustrating the forward transmission factor as a function of the frequency, a smaller insertion loss can be achieved using an identical series inductance having an identical figure of merit/quality factor. In particular, the Pi-structure shown inFIG. 5produced an instruction loss of approximately 0.66 dB, whereas the impedance matching network using a Ruthroff transformer illustrated inFIG. 10has an insertion loss of only 0.49 dB. This is a difference of 0.15 dB, even though an additional second inductance having the same figure of merit/quality factor is involved.

The comparison of the input reflection coefficients S(1,1) and of the S(2,1) curve of the Pi-structure shown inFIG. 6with the same curves of the Ruthroff transformer structure shown inFIG. 11further reveals that the transformer solution also offers an improved bandwidth behavior, in particular a broader bandwidth. While the Pi-network ofFIG. 5has an attenuation of already 18 dB at 3 GHz (see forward transmission factor diagram inFIG. 6), an attenuation of only 2 dB can be achieved using the transformer-implemented impedance matching network as illustrated inFIG. 10, as can be seen in the forward transmission factor diagram ofFIG. 11. This is of particular interest with respect to production tolerances, because narrowband systems might lead to yield problems.

The transformer structure shown inFIG. 10further provides an additional electrostatic discharge (ESD) protection, as illustrated inFIG. 12. While sensible capacitances of the Pi-structure inFIG. 5have to sustain an ESD pulse primarily on their own (which typically is not possible, or at least difficult to implement, using silicon-based integrated circuit), the capacitance within the transformer-implemented structure is protected by the inductance LP. The discharge current path is indicated inFIG. 12as a thick line. Another feature of the transformer-implemented impedance matching network that may be of interest for adaptive matching, will be described below in more detail: by nature, the voltages split up over the two inductances of the transmission line transformer (Ruthroff transformer).

By choosing different inductances for the series inductance LSand the ground path inductance LP, transformation factors other than 1:4 may be implemented. Furthermore, it is possible to invert one of the inductances in order to convert the 1:4 transformer into a 4:1 transformer. Conveniently, it is not necessary to change or reconnect the series inductance Lsfor such reconfiguration of the Ruthroff transformer, but it is sufficient to “turn around” the ground path Lp. In the circuit schematic shown inFIG. 13the ground path inductance LPis inverted in comparison to the configuration of the Ruthroff transformer shown inFIG. 9, because node1of the ground path inductance LPis now connected to node4of the series inductance LS, while node2of the ground path inductance LPis connected to the ground potential. The connection of the series inductance LSis the same inFIG. 9and inFIG. 13.

In order to obtain an adjustable impedance transformation, it is possible to either switch the series inductance LSor the ground path inductance LP. Reactive components, such as inductive or capacitive components, may then be adjusted as in the original approach using variable capacitances C1and C2, as illustrated inFIG. 10.FIG. 10shows a circuit schematic of an adjustable impedance matching network according to at least one embodiment.

One further challenge that may need to be overcome is the need to adjust the imaginary part/reactive component. While the capacitive portion can be adjusted in a relatively simple manner using (adjustable) capacitors against ground, the classical approach for impedance matching using a Pi-network is a series inductivity Lsas schematically illustrated in the circuit schematic ofFIG. 14. Actually, the series inductivity Lsis fixed and an adjustable capacitor is connected in series with the series inductivity Lsin order to change the inductance of the series branch of the Pi-network.

FIG. 15shows in a Smith chart how the impedance of a series inductivity changes with increasing inductance. A very small inductance (L≈0) behaves much like a short circuit so that the input impedance (or overall impedance) of a network comprising the series inductivity (i.e., substantially a short circuit for a small inductance) and a load impedance is equal to the load impedance. The corresponding point in the Smith chart is at or very close to the center of the Smith chart at (1,0). With increasing inductance of the series inductivity, the input impedance becomes increasingly inductive. InFIG. 15, the inductance is varied between 1 nH and 10 nH.

Furthermore, the quality factor of an adjustable series inductivity typically has to be taken into account. In case the adjustability of the adjustable inductivity is actually performed by a series connected adjustable capacitance, the imaginary part is automatically reduced, while the series resistance remains substantially constant, which leads to a reduction of the quality factor. In addition, losses caused by the capacitance itself and/or by a switch (in case such a switch is used, e.g., a CMOS switch) may further degrade the quality factor. A slightly better way would then be to either switch or commutate several coils (inductivities) or to design one coil with one or more taps, with which the inductivity can be adjusted. The provision of several coils leads to higher costs and space requirements. A coil with tap(s) typically requires a corresponding buildup technology, such as LTCC (low temperature co-fired ceramic) or monolithic integration. For this reason, this approach is typically not found with laminate modules, inter alia because taps are simply too big due to the required vias and due to other parameters.

No matter which approach is chosen, one has to accept significant additional losses. Moreover, ESD-related issues may arise that are caused by the capacitance or capacitances, in particular if the capacitance is a shunt capacitance. Therefore, another way appears to be more appropriate, namely the use of an inductivity against ground, i.e., a “shunt L.”

FIG. 16shows a circuit schematic of an inductivity against ground (“shunt L”).FIG. 17shows the impedance behavior of the network inFIG. 16for varying inductance values between 1 nH and 10 nH. For a very small inductance (L≈0), the shunt inductivity substantially produces a short circuit so that the resulting input impedance (or overall impedance) of the network is at left most point (0,0) of the Smith chart, i.e., real part and imaginary part are both zero. It can be seen inFIG. 17that a substantially different behavior is obtained, which does not appear to be of use for a pure Pi-structure, in particular when the impedance to be matched is a very small impedance. The reason is that a high-pass behavior is obtained. Accordingly, a T-structure with series capacitors and a shunt coil would be the result, as illustrated in circuit schematic form inFIG. 18. However, due to ESD problems related to the capacities, this approach is typically not further pursued.

As mentioned above, it may be difficult to match very low impedance values of the impedance to be matched. However, a transformer makes it possible to transform impedances from low to high, or vice versa, depending on the transformation ratio. When using a transformer as mentioned above for creating real-valued start values, it is possible to place the real-valued start value in such a manner that one shunt inductivity is sufficient.

FIG. 19shows a schematic circuit diagram of a corresponding impedance matching network100having a transmission line transformer120between a capacitive shunt branch171and an inductive shunt branch172. In the example ofFIG. 19, it is assumed that a relatively high impedance is connected to the left side of the impedance matching network100(at a first terminal101of the impedance matching network100), and a smaller impedance is connected to the right side of the impedance matching network100(at a second terminal102of the impedance matching network100).

The adjustable impedance matching network100shown inFIG. 19comprises a first terminal101, a second terminal102, and a reference potential terminal103. In the embodiment shown inFIG. 19the reference potential corresponds to a ground potential for the circuit100. The circuit100may be connected to the reference potential at a plurality of locations via a plurality of reference potential terminals103. An impedance ZLor RLthat is to be matched to the input impedance or the output impedance of another circuit (e.g., the output impedance of a power amplifier3as illustrated inFIG. 3) may be connected to the first terminal101or the second terminal102. The other circuit is then connected to the second terminal102or the first terminal101, respectively.

The adjustable impedance matching network100also comprises a transmission line transformer120. The transmission line transformer120comprises a series inductance (first inductor path)121and a ground path inductance (second inductor path)122which are magnetically coupled (coupling coefficient k). The series inductance121forms (or is part of) a first inductor path of the transmission line transformer120. The ground path inductance122forms (or is part of) a second inductor path of the transmission line transformer120. The transmission line transformer120is connected as a Ruthroff transformer in the embodiment shown inFIG. 11. In alternative embodiments the transmission line transformer could be connected as a Guanella transformer. The first inductor path121is connected to the first terminal101at one of its ends and to the second terminal102at its other end. The ground path inductance or second inductor path122is conductively couplable between the first input terminal101and the reference potential terminal103, as will be described in more detail below. In an alternative embodiment, the first inductor path121could be conductively couplable between the first terminal101and the second terminal102in a reconfigurable manner, as well. In particular, the first inductor path121could be connected with reversed polarity. Note that the polarity of the first inductor path101and the second inductor path102has to be considered, due to the magnetic coupling of the first and second inductor paths101,102. This means that the transmission line transformer120exhibits different behaviors when one of the first and second inductor paths101,102is connected with reversed polarity.

FIGS. 20 to 22schematically illustrate a procedure for adjusting (matching) an impedance.FIG. 20illustrates a step of adjusting the real part of the impedance using the transmission line transformer. The impedance to be matched has a relatively small real part. The arrow and the start inFIG. 20represent the result of the transformation performed by the transmission line transformer.

InFIG. 21, the result of an activation of the shunt-L172inFIG. 19is illustrated in the Smith chart. In an inverse manner,FIG. 22shows the influence of a capacitive impedance based on a capacitor bank in the capacitive shunt branch171. It can be seen that using this methodology, only shunt element are needed in addition to the (switchable) transformer. Conveniently, the inductive shunt branch171also provides an ESD protection in case the secondary winding122of the transformer120is not used for some reason.

It appears reasonable to design such a shunt inductivity as a coil with taps in order to coarsely obtain the necessary inductivity, and then to perform a fine tuning with the capacitor bank in the capacitive shunt branch171, so that no significant loss of quality factor occurs.FIG. 23shows a schematic top view of an inductive element372that may be used within the inductive shunt branch172. The inductive element372comprises a first terminal372aand a second terminal372f. Furthermore, the inductive element372comprises a plurality of taps372b,372c,372dand372e.

FIG. 24shows a schematic circuit diagram of the inductive element372and a plurality of switch elements472b,472c, and472dthat are connected to the taps372b,372c, and372d(tap372eis not shown inFIG. 24). Each of the switch elements472b,472c,472dis configured to selectively connect the corresponding tap372b,372c,372dwith the ground potential. In other words, the inductive element172comprises a coil372with a tap372b, and a switching element472bconnected to the tap372band configured to selectively bypass a second inductive portion that extends (in an electrical sense) along the inductive element372between the tap372band the reference potential terminal103.

The inductive element372may comprise a first inductive portion, a second inductive portion connected in series with the first inductive portion, and a switching element472bconnected to a circuit node (e.g., tap372b) between the first inductive portion and the second inductive portion and configured to selectively connect the circuit node372bwith the reference potential.

FIG. 25shows a schematic circuit diagram of an impedance matching network100according to at least some embodiments. The impedance matching network100ofFIG. 25can be regarded as a combination of the elements described above.

The series inductance121of the transmission line transformer120comprises a plurality of sub-sections121a,121b, . . .121k. Each sub-section of the plurality of sub-sections121a. . .121kextends between two inductor nodes of a plurality of inductor nodes21a,21b, . . .21k,21k+1. The second inductor path122also comprises a plurality of sub-sections122a,122b, . . .122jand corresponding inductor nodes22ato22j+1. The number of sub-sections of the first inductor path121may be equal to the number of sub-sections of the second inductor path121, i.e., j=k. According to some embodiments however, the first and second inductor paths121,122may have different numbers of sub-sections, i.e., j≠k.

The adjustable impedance matching network100also comprises a plurality of switching elements131a,131b, . . .131k+1. The switching elements131a. . .131k+1 are connected between a circuit node to the left of a corresponding sub-section121ato121kand to the second terminal102(i.e., for example, the switching element131ais connected between the circuit node to the left (with respect toFIG. 25) of sub-section121aand the second terminal102). Each switching element131a. . .131k+1 may be a semiconductor switching element. Each of the plurality of switching elements131a. . .131kmay bridge one or more sub-sections121a. . .121kof the first inductor path121when the switching element is controlled to be in a conducting state by means of a suitable control signal. In particular, the entire first inductor path121may be bridged when the switching elements131ais conducting so that the first terminal101and the second terminal102are connected via the semiconductor switching element131a.

The adjustable impedance matching network100shown inFIG. 25further comprises a reconfiguration unit150that may be used to configure the adjustable impedance matching network100according to at least two different configurations. In a first possible configuration a first end (coinciding with node22j+1 in the embodiment shown inFIG. 25) of the second inductor path122is connected to the first terminal101, and a second end (coinciding with inductor node22a) of the second inductor path122is connected to the reference potential terminal103. In a second possible configuration the first end (i.e., inductor node22j+1) of the second inductor path122is connected to the reference potential terminal103and the second end (i.e., inductor node22a) of the second inductor path122is connected to the first terminal101. Thus, in the first configuration the transmission line transformer120is connected as a Ruthroff transformer according toFIG. 9. In the second configuration the transmission line transformer120is connected as a Ruthroff transformer according toFIG. 13. As explained in connection withFIG. 13, the impedance transformation ratio can be changed, for example from a 1:4 impedance transformation ratio to a 4:1 impedance transformation ratio, by changing from the first configuration to the second configuration.

The reconfiguration unit150comprises switching elements451,452,453, and454. The switching elements may be transistors, e.g., MOSFETs, NMOS transistors, etc. The transistors451and453provide the functionality of a changeover switch. The transistors452and454provide the functionality of another changeover switch152. The transistors451and452are connected to the inductor node22aof the second inductor122, and, at their respective opposite sides, to the reference potential and the second terminal102, respectively. The transistors453and454are connected to the inductor node22j+1 of the second inductor122, and, at their respective opposite sides, to the reference potential and the first terminal101, respectively.

The reconfiguration unit150may also be regarded or used as a pole reversal element that is configured to reverse a polarity of the first inductor path121or the second inductor path122. When used as a pole reversal element for the first inductor path121, the common terminals of the two changeover switches151,152would be connected to the first inductor node21aand the second inductor node21k+1 of the first inductor path121. The other terminals of the two changeover switches151,152would then be connected to the first terminal101and the second terminal102.

The adjustable impedance matching network100ofFIG. 25also comprises a capacitive shunt branch171and an inductive shunt branch172. The capacitive shunt branch171comprises a bank of parallel, individually switchable capacitors371a,371b,371cand a plurality of further (semiconductor) switching elements471a,471b,471c. In this manner, the capacitive shunt branch171is variable or adjustable. The capacitive shunt branch171is connected in parallel to the remainder of the adjustable impedance matching network100between the first terminal101and the reference potential terminal103. The inductive shunt branch comprises the adjustable inductive element372and the switching elements472b,472c,472dillustrated inFIG. 24.

The adjustable impedance matching network100forms a Pi-network with the capacitive shunt branch171and the inductive shunt branch172being the parallel impedances. The first inductor path121of the Ruthroff transformer120forms a series impedance or series element of the Pi-network.

The transmission line transformer120may be a classical transformer, a bifilar transformer or a planar transformer. In the case of a planar transformer it may be implemented as a printed circuit board integrated transformer, semiconductor transformer, redistribution layer technologies (EWLB—Embedded Wafer Level Ball Grid Array), a LTCC structure, a HTCC structure, or a combination thereof.

As a further option for adjustment, the series inductance Lscould be designed to be unequal to the parallel inductance Lp, and/or the parallel inductivity Lpcould be made switchable, as well (not shown inFIG. 25). For example, the parallel inductivity Lpcould be made switchable with respect to its inductance value by commutating between a series connection and a parallel connection of the sub-sections122ato122jof the second inductor path. This would make the range of possible transformation ratios larger so that additional real values in the complex plane may be reached.

Following the above, it is proposed to not employ known T-networks or Pi-networks, but to use a transformer instead and to implement its transformation ratio in at least one sub-section of its winding. Thus, a number of benefits can be achieved:

Lower losses/insertion loss when using equivalent components.

Larger bandwidth and thus less sensibility to production tolerances.

Usable for several frequency bands; a transformer can handle the 900 MHz band as well as the 1.8 GHz band, whereas a Pi-structure according toFIG. 5would require two different inductivities (which may be achieved using a switch, but this would require an additional switch and thus leads to additional losses).

A ground connection is always provided via the second inductor path so that the capacitances (if present) are better protected against electrostatic discharge (ESD).

As the switching elements (e.g., switching transistors) only see a small portion of the voltage swing, devices having a lower breakdown voltage may be used: devices with smaller on-resistance Ronand/or off-capacitance Coffmay be used.

The inductive part is adjusted using a shunt inductivity which contributes less losses, because “only” it influences the reactive component.

The switching elements131ato131k+1,471ato471c,472bto472d, and451to454may be implemented as CMOS transistors, typically NMOS transistors.

FIG. 26shows in a schematic manner a top view of a planar transmission line transformer820. In general, the transmission line transformer may be realized in a passive integration or, alternatively, accommodated in a laminate (e.g., a printed circuit board), as a LTCC or HTCC (low/high temperature co-fired ceramic) structure, etc. Other implementations of the transmission line transformer are also possible, such as a transformer integrated in a silicon substrate or EWLB (Embedded Wafer Level Ball Grid Array). The transmission line transformer820comprises the first inductor path121and the second inductor path122. The first inductor path121extends between the terminals numbered1and2inFIG. 18. The second inductor path122extends between the terminals numbered2and4inFIG. 18. In other configurations, the transmission line transformer may comprise a third inductor path and possibly even further inductor paths. When using a planar transformer820as shown inFIG. 18, the transformer820may be tapped at suitable locations in order to obtain the desired impedance switching ratio.

The capacitive element or shunt branch171and the inductive element or shunt branch172may both be adjustable. The capacitive element171may have a finer adjustment resolution than the inductive element172.

The capacitive element171, the inductive element172, and the transmission line transformer120may be integrated monolithically within an integrated circuit. Alternatively, the transmission line transformer may be formed within a laminate module.

As already mentioned above in the description relative toFIGS. 5 and 7, covering a large frequency concurrently with a covering a large impedance region presents a challenge for the design of impedance matching networks. For example, frequencies between 700 MHz and 2.69 GHz have to be taken into account in mobile phones.

When designing the series inductivity for the lowest frequency (typically resulting in the highest inductance value), depending on the specification of the adjustment region to be reached, one ends up with an inductance of 22 nH. In particular high impedance values (≈500 ohm) require such inductance values. When the same impedance has to be reached (i.e., matched) at 2.69 GHz, the series inductivity may be substantially smaller, namely approximately 3 nH. Hence, the inductance has to be reduced significantly in an impedance matching network having the structure shown inFIG. 7. Even when we assume an ideally adjustable capacitance, the series resistance of the series inductivity remains constant, which results in a notably reduced quality factor. Accordingly, the mobile communications device will present large losses, or it may not be possible to even be able to cover this particular region of impedances.

A similar behavior can be observed with adjustable capacities: at 700 MHz relatively large capacitance values are necessary, whereas at 2.69 GHz only small capacitance values are needed. Once more, given a substantially constant real part of the impedance, the reduction of the imaginary part of the impedance inevitably causes a reduction of the quality factor. For this reason, only technologies are used for these matching circuits that provide the highest possible quality factor. Furthermore, the allowable frequency range may also be limited.

FIG. 27relates to a concept for an adaptive matching circuit with pre-transformation.FIG. 27shows a schematic block diagram of a frontend system of a wireless communications device comprising an impedance matching network500that uses a pre-adjustment or pre-transformation700in order to exclude certain regions of the possible impedance values before the impedance is passed on to an adaptive matching network600, for example, a Pi-network. More precisely, the proposed impedance network500comprises a first port501and a second port502. The impedance matching network further comprises a transformer700configured to transform an impedance (e.g., the impedance of the antenna7) connected to the first port501so that a corresponding transformed impedance lies within a confined impedance region799in a complex impedance plane797. The impedance matching network500also comprises an adaptive matching network that is adjustable to match the transformed impedance located anywhere within the confined impedance region to a second impedance507connected to the second port502.

FIG. 28shows the result of a simulated example. The simulation was performed for a frequency of 850 MHz as a field simulation of a planar transformer in CMOS technology. The upper part ofFIG. 28shows a Smith chart representing a complex impedance plane797. Referring toFIG. 27, the antenna7may offer a plurality of different impedances, depending on environmental influences, operating conditions, etc. The plurality of impedances offered by the antenna7are located within a confined original impedance region798. The confined original impedance region798has the shape of a circle, but this is not required. Without loss of generality, it is assumed that the impedance at the second port502corresponds to the point (1,0) in the complex impedance plane797, i.e., the center of the circular confined original impedance region798. Impedances that are represented by points close to the center (1,0) of the circle798are matched already in a relatively good manner. Points that are far away from the center of the circle798correspond to relatively badly matched impedances. The confined original impedance region798is confined, or bounded, which means that theoretically there may be impedance values that are outside the confined original region. However, these outlying impedance values are not expected to occur in practice, or at least only very seldom.

It can be seen inFIG. 28that the pre-transformation masterfully shrinks the impedance region. Due to the large bandwidth of the Ruthroff transformer, the result substantially looks the same at high frequencies (2 GHz). In order to match this reduced region, substantially smaller inductance values are necessary (approximately 3 nH at 800 MHz). In addition, these smaller inductivites can be implemented in a monolithic manner much easier.

As a result, one obtains a smaller insertion loss and also benefits from the large bandwidth of the transformer. This means that globally the bandwidth is improved.

The underlying principle is the use of a six interposed pre-transformation in order to limit the possible VSWR region/impedance region. In this manner, a simplified adaptive matching network can be obtained which requires less extreme boundary conditions, or becomes suitable for monolithic integration in the first place.

With respect to the pre-transformation, a number of options exist. For example, a pi-network is imaginable, at higher frequencies a λ/4 transformer, a transducer, and the transmission line transformers already mentioned for this purpose. The Guanella transformer and the Ruthroff transformer are transmission line transformers. Although the Guanella transformer and the Ruthroff transformer historically come from macroscopic transformers with bifilar windings, they may also be implemented in principle as planar transformers on laminate or silicon. As an example,FIG. 9shows a Ruthroff transformer as 1:4 impedance transformer, for which two identical inductivities are assumed.

The basic idea of these transformers is the combination of signal portions in an additive manner. In the case illustrated inFIG. 9, the output voltage is a sum of the voltage V2across the series inductance and of the voltage V1across the ground path. At the same time, the electric current is split up over the two inductances so that, due to a doubling of voltage and a halving of the electric current, an impedance transformation of 1:4 is achieved (RLappears to be more high-ohmic to the voltage source Vg).

In general, the transformer-based solution has some advantages over a Pi-Network-based solution:

Insertion loss

Bandwidth

ESD robustness

Reference is made toFIGS. 5,6,10and11for a comparison of a classical pi-network and a pi-network with a transmission line transformer.

When applying an impedance located within the confined original impedance region798to the transformer700, the impedance is transformed according to the transformation ratio. When the transformation ratio is appropriately chosen, this leads to the confined original impedance region798to be compacted to the smaller confined impedance region799, which is illustrated in the lower part ofFIG. 28. In other words, the impedance connected to the first port501may be located within the confined original impedance region798in the complex impedance plane797that is larger than the confined impedance region799. Note that the confined original impedance region798leaves out an outer ring of the complex impedance plane797, because this outer ring contains extreme impedance values that are either very small or very large.

The confined impedance region799may be a function of the capabilities of the adaptive matching network600. In other words, when the set of impedances is known that can be matched using the adaptive matching network600, this defines the confined impedance region799. The confined original impedance region798may be determined on the basis of the confined impedance region799using the transformation ratio of the pre-transformer700.

The transformer700may have a fixed transformation ratio. The transformer700may be a transmission line transformer.

After the pre-transformation performed by the transformer700, the new impedance region has to be matched to the target impedance in a next step. Basically, a structure as schematically illustrated inFIG. 5or18may be employed to this end. However, in order to not limit the bandwidth by performing this matching (which is needed for broadband LTE signals, for example), it may be advisable to perform a multistage matching, as schematically illustrated by a multistage pi-network as shown inFIG. 8.

Instead of using two coils as shown inFIG. 8, it may be even more advisable to use a single coil having a tap because in this manner the coupling between the inductivities can be exploited, as well.FIG. 29shows a schematic circuit diagram of a corresponding adaptive matching network600according to some embodiments. The adaptive matching network600is a two-stage Pi-network and comprises a first terminal601and a second terminal602. The first terminal six of one and the second terminal six and two are connected by a series connection of a first inductive element634and a second inductive element644. In the embodiment shown inFIG. 29the first and second inductive elements634,644are inductively coupled as indicated by the coupling factor k. It may be beneficial to exploit the coupling between the first and second inductive elements. This may be achieved by using a single coil that has a tap654. In other embodiments it is also possible that the first inductive element634and the second inductive element644are implemented as two different coils having no or only negligible inductive coupling. The adaptive matching network600shown inFIG. 29further comprises three adjustable capacitance elements632,642and652.

The adaptive matching network600may comprise a Pi-network having a series inductance634and shunt capacitances632,642. According to some embodiments, the adaptive matching network600may comprise at least two stages. The adaptive matching network600may comprise a series inductance634,644with an intermediate tap654and a capacitive element642connected to the intermediate tap. The series inductance may comprise a first inductive portion associated to a first stage of the at least two stages, and a second inductive portion associated to a second stage, the second inductive portion being inductively coupled to the first inductive portion.

The adaptive matching network600may comprise a first adjustable capacitive shunt branch and a second adjustable capacitive shunt branch.

The adaptive matching network600may comprise a fixed inductance and an adjustable capacitive element.

FIG. 30shows as an example of a schematic chip layout of the two coupled inductive elements634and644, as well as to capacitive elements642aand642bconnected to taps654aand654b, respectively. The coil shown inFIG. 30differs from the schematic circuit diagram inFIG. 29in that the coil comprises two taps654a,654b. The capacitive elements642aand642bare implemented as metal-insulator-metal capacitors.

FIG. 31shows a schematic circuit diagram of the impedance matching network500comprising the transformer700and the adaptive impedance matching network600.

An antenna circuit according to further embodiments may comprise an antenna7(see for exampleFIG. 3), a signal terminal configured to relay a signal to a receiver or from a transmitter (inFIG. 3for example the output of the antenna switch5) and an impedance matching network interconnecting the antenna7and the signal terminal. Departing fromFIG. 3, the impedance matching network comprises a Pi-network having a first inductor path of a transmission line transformer in a series branch of the Pi-network. As an alternative, an antenna circuit according to further embodiments may comprise the antenna7, the signal terminal, and an impedance matching network as schematically illustrated inFIG. 31.

FIG. 32shows a schematic flow diagram of a method for matching an impedance using an impedance matching network comprising a first shunt branch with a capacitive element, a second shunt branch with an inductive element, and a transmission line transformer with a first inductor path and a second inductor path, wherein the first inductor path connects the first terminal and the second terminal. The method comprises a step902of adjusting a real part of the impedance by adjusting a transmission ratio of the transmission line transformer. The method further comprises a step904of adjusting an imaginary part of the impedance by adjusting at least one of the capacitive element and the inductive element.

The step902of adjusting the transmission ratio of the transmission line transformer may comprise controlling a switching element that is connected in parallel to a sub-section of the first inductor path or of the second inductor path, in order to selectively bridge the sub-section and to adjust the inductance of the first inductor path or the second inductor path.

The step904of adjusting the imaginary part of the impedance may comprise performing a coarse adjustment by adjusting the inductive element. Subsequently, a fine adjustment by adjusting the capacitive element may be performed.

The step904of adjusting the inductive element may comprise controlling a further switching element connected to a tap of the inductive element, the tap being connected to a node between a first inductive portion and a second inductive portion of the inductive element, in order to selectively bypass the second inductive portion of the inductive element.

FIG. 33shows a schematic flow diagram of a method for matching an impedance. The impedance to be matched is connected to a first port of an impedance matching network. The method comprises a step952of transforming the impedance to a corresponding transformed impedance that lies within a confined impedance region in a complex impedance plane. Said transforming is performed by a transformer of the impedance matching network. The method further comprises a step954of matching the transformed impedance located anywhere within the confined impedance region to a second impedance connected to a second port of the impedance matching network, wherein said matching is performed by an adaptive matching network.

The impedance connected to the first port may typically be located within a confined original impedance region in the complex impedance plane that is larger than the confined impedance region.

The adaptive matching network may comprise at least one of an adjustable capacitive element and an adjustable inductive element. Matching the transformed impedance to the second impedance may comprise adjusting the adjustable capacitive element or the adjustable inductive element.

The step954of adjusting the adjustable capacitive element or the adjustable inductive element may comprise controlling a switching element to selectively engage or disengage a particular inductive portion of the adjustable inductive element, or a particular capacitive portion of the adjustable capacitive element.

In the foregoing Detailed Description, it can be seen that various features are grouped together in embodiments for the purpose of streamlining the disclosure. This method of disclosure is not to be interpreted as reflecting an intention that the claimed embodiments require more features than are expressly recited in each claim. Rather, as the following claims reflect, inventive subject matter may lie in less than all features of a single disclosed embodiment. Thus the following claims are hereby incorporated into the Detailed Description, where each claim may stand on its own as a separate embodiment. While each claim may stand on its own as a separate embodiment, it is to be noted that—although a dependent claim may refer in the claims to a specific combination with one or more other claims—other embodiments may also include a combination of the dependent claim with the subject matter of each other dependent claim or a combination of each feature with other dependent or independent claims. Such combinations are proposed herein unless it is stated that a specific combination is not intended. Furthermore, it is intended to include also features of a claim to any other independent claim even if this claim is not directly made dependent to the independent claim.

It is further to be noted that methods disclosed in the specification or in the claims may be implemented by a device having means for performing each of the respective steps of these methods.

Furthermore, in some embodiments a single step may include or may be broken into multiple sub steps. Such sub steps may be included and part of the disclosure of this single step unless explicitly excluded.