Subscriber's line circuit for telecommunications networks

A subscriber's line loop including a terminal station and symmetrically arranged in the loop relative to the station two constant current sources. Each current sources has a power unit defining an output connected to the terminal station, a feedback output, and a control input, a differential amplifier having a non-inverting input connected to a source of reference voltage. In order to suppress interference signals induced into the power unit, a bandpass filter is connected between the output of the differential amplifier and an adder connected to the inverting input of the amplifier. The other input of the adder is connected to the feedback output of the power unit. In a modification, a band rejection filter is interconnected between the control input of the power unit and the output of the differential amplifier.

BACKGROUND OF THE INVENTION 
The present invention relates in general to a subscriber's line circuit for 
a telecommunications network, particularly for a telephone network, and is 
of the type which includes at least one user's line loop connected to a 
user's terminal station, two sources of constant current arranged 
symmetrically to the terminal station in the user's line loop for keeping 
a constant current irrespective of the length of the line loop. 
A line circuit of this kind is known for example from German published 
patent application No. 28 50 905. In this prior-art circuit, a source of 
constant current is applied both to the a-trunk line and to the b-trunk 
line of the user line circuit, in order to obtain symmetrical damping 
conditions of the voice-modulated alternating voltage, and thus to avoid 
crosstalk phenomena. Due to an inductive or capacitive coupling, 
relatively large hum voltages in the subscriber's line will occur, caused 
for example when overland lines or railroad lines extend parallel to the 
subscriber's line. As a consequence, the symmetrical arrangement of 
current sources for damping the voice-carrying voltages is no longer 
sufficient, especially when high standards for suppression of crosstalk 
are to be met, because the interference voltages cause an additional 
current to flow in unison with the frequency of the hum voltages, thus 
disturbing the symmetry of the alternating voice voltage. Since this known 
line circuit arrangement enables a simple monitoring of the operational 
conditions of the line loop, such as loop open, loop closed, ground key 
activated, it is desirable that the susceptibility of this monitoring 
function to interfering voltages be reduced, especially due to the fact 
that a disturbance signal is coupled into the line, the monitoring 
function can be impaired, for the damping of the crosstalk becomes 
ineffective. 
SUMMARY OF THE INVENTION 
It is therefore a general object of the present invention to ovcrcome the 
aforementioned disadvantages. 
More particularly, it is an object of the invention to provide an improved 
subscriber's line circuit of the aforedescribed kind which improves the 
resistance of the condition-detecting function of the line against 
interfering signals. 
A further object of the invention is to provide such an improved line 
circuit in which the symmetry of the alternating voltage relative to the 
subscriber's station is increased. 
In keeping with these objects and others which will become apparent 
hereafter, one feature of the invention resides, in a line circuit of the 
aforedescribed type, in the provision of two constant current sources 
arranged symmetrically to the terminal station in the user line loop, each 
of the current sources comprising a power unit having a power output 
connected to the terminal station, a regulating or feedback output which 
may contain the interference signal, and a control input, a differential 
amplifier having its output connected to the control input of the power 
unit, a non-inverting input and an inverting input, a source of reference 
voltage connected to the non-inverting input of the amplifier, and, in one 
embodiment, a bandpass filter connected to the control input of the power 
unit and via an input of an adder connected to an inverting input of a 
differential amplifier; the output of the differential amplifier is 
connected to the control input of the power unit, and the non-inverting 
input of the amplifier is connected to a source of reference voltage; the 
other input of the adder is connected to the feedback or regulating output 
of the power unit. 
In another embodiment of this invention, the feedback output of the power 
unit is directly connected to the inverting input of the differential 
amplifier without the use of the adder, and a band rejection filter for 
the interference signals is connected between the control input of the 
power unit and the output of the differential amplifier. 
The novel features which are considered characteristic for the invention 
are set forth in particular in the appended claims. The invention itself, 
however, both as to its construction and its method of operation, together 
with additional objects and advantages thereof, will be best understood 
from the following description of specific embodiments when read in 
connection with the accompanying drawing.

DESCRIPTION OF THE PREFERRED EMBODIMENTS 
Referring firstly to FIGS. 1 and 2, a subscriber's terminal station Tln is 
connected through symmetrically arranged a-trunk line and b-trunk line to 
the user line circuit of a telephone central station. Resistance of 
individual trunk lines which may differ according to the length of 
individual branches of the line loop, is indicated by resistors R. 
Connection terminals A and B of the user line circuit are each connected 
to a source of constant current. Each of the current sources includes a 
power unit L1 and L2 and a controlling differential amplifier V1 and V2. 
The power unit L2 is connected to the negative pole -U.sub.B of a DC power 
source, and the other power unit L1 is connected to the grounded positive 
pole of the DC power source. The non-inverting input (+) of differential 
amplifier V1 or V2 is connected to a source of reference voltage U.sub.r. 
The reference voltage applied to the amplifier input is more negative or 
positive than the potential on corresponding poles of the DC battery 
voltage U.sub.B. 
In the embodiment according to FIG. 1, there is employed an adder A1 or A2 
connected at its output to the non-inverting input of the amplifier V1 or 
V2. A regulating or feedback signal derived from respective power units L1 
or L2, which may also contain a superimposed interference signal caused by 
coupled voltages from long-distance lines for example, is applied to the 
first input of the adder A1 or A2. The output of the differential 
amplifier V1 or V2 is connected via a bandpass filter BP1 or BP2 to the 
second input of adder A1 or A2 and is also connected to the control input 
of power unit L1 or L2. The bandwidth of each bandpass filter BP1 or BP2 
is designed in dependence on particular applications, that is in 
dependence on frequency of predominant interference signals in a 
particular location. For example, in the case of overland power lines 
extending parallel to the subscriber's circuit loop, the frequency of 
coupled interference signals is for instance 50 cycles per second, whereas 
in the case of railroad lines the frequency is 162/3 cycles per second. 
Accordingly, the bandpass filters BP1 and BP2 are dimensioned so as to 
pass through the frequency band between 16 and 50 cycles per second. Under 
other circumstances, it may also be of advantage to raise the upper limit 
frequency to pass through interference voltages of higher frequency, which 
may be generated during phase clipping control of trains. By virtue of the 
line circuit of this invention, the interference signals are neutralized 
at the output of differential amplifiers V1 and V2, and consequently a 
line circuit monitoring device D connected between the outputs of 
respective amplifiers V1 and V2 is practically undisturbed by the 
inteference signals. In addition, the symmetrical distribution of the 
alternating voltage of the entire line circuit is substantially improved. 
A second embodiment of the line circuit of this invention is illustrated in 
FIG. 2. In this embodiment, the regulating or feedback signal derived from 
the power units L1 and L2 is applied directly to the inverting input of 
the differential amplifier V1 or V2. A band rejection filter BS1 or BS2 is 
interconnected between the control input of power units L1 and L2 and the 
output of respective amplifiers V1 and V2. Both band rejection filters are 
designed such that the frequency range of interference signals which may 
be present in the regulating or feedback signal from power units L1 and L2 
is effectively stopped. The determination of the required frequency band 
to be rejected is made under the same considerations for designing the 
bandpass filters in the preceding example according to FIG. 1. The 
monitoring device D is connected between the control inputs of power units 
L1 and L2, where the interference signals no longer appear. 
A detailed circuit diagram of the embodiment according to FIG. 1 is shown 
in FIG. 3, whereas FIG. 4 shows a detailed diagram of the embodiment of 
FIG. 2. In both cases, power units L1 and L2 of the constant current 
sources have the same construction, consisting of feeding transistors T1 
or T3, the collectors of which are connected to respective connection 
points B or A of the user line loop b or a. The emitter of feed transistor 
T1 in power unit L1 is connected via resistor R1 to grounded positive pole 
of a power supply battery U.sub.B, whereas the emitter of feed transistor 
T3 in power unit L2 is connected via resistor R3 to the negative pole of 
the battery U.sub.B. Current delivered by the two feed transistors T1 or 
T3 depends on control signals applied to the base of each feed transistor 
via the collector-emitter path of transistors T2 and T4 having emitters 
interconnected through resistor R7 and collectors connected to the 
assigned poles of the power supply battery U.sub.B. The emitter of 
transistor T2 is connected to the base of feed transistor T1 through 
resistor R5, and to the collector of transistor T1 through resistor R2. In 
the same fashion, the emitter of transistor T4 is connected to the base of 
transistor T3 through resistor R6, and to the collector of transistors T3 
through resistor R4. The resistors R5 and R6 establish a required bias for 
the base of the corresponding feed transistor. The coupling resistor R7 
has a relatively high ohmic value. The base of control transistor T2 is 
connected via resistor R8 to the positive pole of supply battery U.sub.B 
and further is connected through diode D1 to the output of an operational 
amplifier V1 connected as a differential amplifier. Similarly, the base of 
the other control transistor T4 in power unit L2 is connected through 
resistor R9 to the negative pole of the battery, and through an oppositely 
connected diode D2 to the output of the second operational amplifier V2, 
which is also connected as a differential amplifier. The interconnection 
of diodes D1 or D2 between the output of the amplifier and the control 
input of the assigned power unit L1 or L2 serves for limiting the 
collector-base current of the control transistors T2 or T4 when the user 
line loop is in open condition. A secondary winding of a coupling 
transformer U is connected between the connection points A and B of the 
subscriber's line loop a and b to apply voice-modulated carrier voltage 
into the latter. The secondary winding is constituted by two separate 
coils coupled by 2 series-connected capacitor for blocking direct current 
in the line loop. The primary winding of transformer U is connected to 
coupling means of the network of the central telephone station. 
Bandpass filters BP1 and BP2 in the embodiment of FIG. 3 comprise 
capacitors C1, C3, resistor R15 and operational amplifier V3 or capacitors 
C2, C4, resistor R25 and operational amplifier V4. The group of resistors 
R10-R14 in connection with differential amplifier V1, or the group of 
resistors R20-R24 in connection with the other differential amplifier V2, 
act respectively as an analog adder. Resistors R10 and R20 in respective 
adders connect the inverting input of operational amplifiers V1 and V2 to 
feedback outputs of power units L1 and L2 at the emitters of feed 
transistors T1 and T3 and are connected via low-ohmic reference resistors 
R1 to the positive pole and through the resistor R3 to the negative pole 
of the supply battery U.sub.B. 
The output of operational amplifiers V3 and V4 in respective bandpass 
filters BP1 and BP2 are also connected via resistors R11 or R21 to the 
inverting input of the differential amplifiers V1 and V2. As a result, 
interference signals from the feedback output of the power units are added 
in opposite phase to the filtered part of the control signal, and 
consequently the interfering component of the control signal at the output 
of differential amplifiers V1 and V2 is eliminated. The resistance ratio 
R14/R10 or R24/R20 determines the amplification of the control signal 
together with the interference signal component and the resistance ratio 
R14/R11 or R24/R21 determines the amplification of about 180.degree. 
phase-inverted signal component. If the amplification of the active 
elements V3 and V4 in the bandpass filters BP1 and BP2 is adjusted to one, 
and if the values of resistors R10-R13 and R20-R23 are the same, no 
interference signal occurs at the output of differential amplifier V1 or 
V2, and the monitoring or condition-detecting device D which is connected 
between the outputs of the differential amplifiers is not influenced by 
the interference signal. 
Referring now to the function of the condition-detecting device D, it is 
necessary to refer briefly to the function of the feeding circuit of the 
latter. 
The non-inverting input of the differential amplifiers V1 and V2 is 
connected to a source of reference voltage U.sub.r. This reference voltage 
at the differential amplifier V1 is more negative by a certain value than 
the grounded positive pole of the supply battery, whereas the reference 
voltage U.sub.r at the non-inverting input of the other differential 
amplifier V2 is more positive by a certain value than the negative pole of 
the supply battery U.sub.B. This value of the voltage difference U.sub.r 
depends on the magnitude of current flowing through the user line loop and 
on the magnitude of reference resistors R1 and R3. Differential amplifiers 
V1 and V2 are power-supplied by voltages +U.sub.V1 and -U.sub.V1 or 
+U.sub.V2 and -U.sub.V2. The reference potential for the supply voltage of 
the differential amplifier V1 is the grounded positive pole and for the 
supply voltage of amplifier V2 the negative pole of the supply battery 
U.sub.B. 
When the user line loop is open, no current flows therethrough, but the 
feed transistors T1 and T3 and the control transistors T2 and T4 are 
switched on. Through resistors R8, R7 and R9, the two control transistors 
T2 and T4 are biased in such a manner that the feed transistors T1 and T3 
are in operative condition and are ready, upon closing of the user line 
loop, to immediately resume their function as the source of a constant 
current. Since voltage drop across resistors Rl and R3 is smaller than the 
corresponding reference voltage U.sub.r, the differential amplifiers V1 
and V2 are adjusted at their inputs to saturation, that is their outputs 
are at a potential which approaches their power-supply voltage. If the 
power-supply voltage of the differential transistors is set for example to 
U.sub.V1 =15 V relative to ground, and if U.sub.V2 =15 V relative to 
-U.sub.B, the potential at the output of amplifier V1 approaches -15 V 
with respect to ground, and the potential at the output of amplifier V2 
approaches +15 V relative to -U.sub.B. In order to limit the 
collector-base current of control transistor T2 or T4, diodes D1 and D2 
are connected between the outputs of differential amplifiers V1 and V2 and 
the bases of control transistors T2 and T4. 
If the user line loop is closed, that is if after the lifting of the 
telephone receiver by the user, a current starts flowing through the user 
line loop. If for instance the feed transistor T1 delivers lower constant 
current than the other feed transistor T3, which case may occur due to 
structural tolerances in the circuit, the feed transistor T3 has a 
tendency to become saturated, that is the resistance of the 
collector-emitter path decreases and the potential at the collector 
electrode of T3 grows more negative. Consequently, through resistor R4 and 
the high-ohmic coupling resistor R7, as well as through user line loop 
branches a and b and the resistor R2, the potential on the emitter of 
control transistor T2 grows more negative and so does, via resistor R5, 
the base of control transistor T1. The control transistor T2 is thus 
activated to increase current in feed transistor T1. Due to the increased 
current through transistor T1, the voltage drop across resistor R1 is also 
increased so that the excitation of control transistor T2 via the 
differential amplifier V1 is still further increased until the feed 
transistor T1 achieves a stable working condition corresponding to the 
designed working point of its characteristic and the current flowing 
through the control transistor T2 and resistor R2 is adjusted to its full 
regulating magnitude at which the difference between the constant current 
of transistors T3 and T1 is neutralized. In opposite case, when feed 
transistor T1 delivers a higher constant current than the feed resistor 
T3, the above described equalizing process takes place in the control 
transistor T4 and in the resistor R4 in an analogous manner. When the 
subscriber's loop is closed, that is when both sources of constant current 
are operative, the two differential amplifiers V1 and V2 operate in the 
linear range, that is their output voltage varies in the range of the 
battery supply voltage U.sub.B. 
If upon activation of the grounding key the a-trunk line is applied to 
ground, the part of the power supply circuit in the b-trunk line is 
without current, and feeding current from transistor T3 flows through the 
a-trunk line. This means that the differential amplifier V1 operates in 
its linear range, and the differential amplifier V2 is at its saturated or 
fully excited condition. 
In this manner, as the outputs of the two differential amplifiers V1 and V2 
resume a quasi-digital state depending on the condition of their user line 
loop, it is possible by means of the simple condition-detecting device D 
having the form of a logic circuit for example to monitor the condition of 
the user line loop, such as loop is open, loop is closed, ground key is 
activated, and to transmit directly the detected condition to a central 
control station St. The detecting or monitoring logic circuit assigns for 
example a logic "o" to the outputs of differential amplifiers V1 and V2 
when operating in linear range and a logic "l" when operating in saturated 
region. Since the aforementioned interfering and/or long-line voltages are 
coupled into the a- or b-trunk lines of the user line loop at the same 
phase, failures of the loop condition detecting function may result unless 
the arrangement of this invention is employed. 
FIG. 4 illustrates an exemplary embodiment of a detailed circuit of the 
concept of FIG. 2. Power units L1 and L2 and the differential amplifiers 
V1 and V2, as well as the condition-detecting device D, are the same in 
structure and operation as in FIG. 3. In contrast, the suppression of 
interfering signals at the inputs of V1 and V2 differential amplifiers, 
operates as follows: The output signal from differential amplifier V1 or 
V2 is applied via resistors R31, R32 or R41 and R42 to both the inverting 
and non-inverting inputs of operational amplifiers V5 or V6. The output of 
the latter operational amplifiers V5 or V6 is fed back through resistors 
R30 or R40 to their inverting input. The non-inverting input of amplifier 
V5 or V6 is connected via active resonance circuits Rk1 or Rk2 both the 
ground and to the minus pole of supply battery U.sub.B. These series 
resonance circuits are adjusted to pass through the frequency or frequency 
bands of the interference signals so that the latter are connected to 
ground potential. If the feedback resistors R30 or R40 are equal to 
resistors R31, R32 and R41, R42, the amplification of amplifiers V5 and V6 
is equal to one, and no interference signals occur at the output of the 
latter amplifiers V5 and V6. Accordingly, the condition-detecting device D 
interconnected between these two outputs no longer is influenced by the 
interference signals and also the symmetry of the carrier alternating 
voltage in the subscriber's line circuit is increased. 
In the embodiment shown, the resonance circuits Rk1 or Rk2 include 
respectively a capacitor C5 or C6 and a gyrator consisting of an 
operational amplifier V7 or V8, capacitors C7 or C8 and resistors R33, R34 
or R43 and R44. If desired, there is always the possibility to connect 
more of such resonance circuits to the non-inverting inputs of amplifier 
V5 or V6, if for instance the frequency range of the band rejecting 
filters is to be increased. It will be understood that each of the 
elements described above, or two or more together, may also find a useful 
application in other types of constructions differing from the types 
described above. 
While the invention has been illustrated and described as embodied in line 
circuits for use in telephone networks, it is not intended to be limited 
to the details shown, since various modifications and structural changes 
may be made without departing in any way from the spirit of the present 
invention. 
Without further analysis, the foregoing will so fully reveal the gist of 
the present invention that others can, by applying current knowledge, 
readily adapt it for various applications without omitting features that, 
from the standpoint of prior art, fairly constitute essential 
characteristics of the generic or specific aspects of this invention.