DC/RF blood cell detector using isolated bridge circuit having automatic amplitude and phase balance components

A differential DC/RF bridge-configured flowcell particle detector includes a flowcell and an adjustable circuit model of the flowcell, which are differentially coupled through output amplifier circuits and galvanically isolated from sources of signal degradation. The output of the difference amplifier is coupled to a DC/RF discriminator and associated downstream processing circuitry. Respective amplitude and phase outputs of the DC/RF discriminator are used to control amplitude and phase adjustment circuits of the adjustable circuit model, so as to automatically track amplitude and phase variations in the non-linear behavior of the flowcell, to mirror the characteristic impedance of the flowcell, thereby making the bridge insensitive to variations in the flowcell.

FIELD OF THE INVENTION
 The present invention relates in general to detectors of the type used for
 conducting electrical measurements of parameters of objects, such as but
 not limited to the detection of particles (e.g., blood cells) contained in
 a carrier fluid supplied to a hematology analyzer. The invention is
 particularly directed to a new and improved DC/RF bridge-configured object
 parameter detector having an automatic amplitude and phase balance circuit
 that models the behavior of the object, particularly an object having
 non-linear characteristics, and compensates for (non-linear) variations in
 conditions other than the parameter being measured.
 BACKGROUND OF THE INVENTION
 As an adjunct to the diagnosis and treatment of disease, the medical
 industry commonly employs various types of particle flow systems, such as
 that diagrammatically illustrated in FIG. 1, to analyze particles or cells
 in a patient's body fluid (e.g., blood cells). To this end, a carrier
 fluid (e.g., saline) stream 1, containing particles/cells 2 of a
 centrifuged blood sample stored in a blood sample holding chamber 3, is
 directed along a flow channel 4 through a restricted flowcell
 `measurement` aperture 5 of a flowcell 6 into a receiving chamber 7. The
 flowcell measurement aperture 5 is sized and configured to allow the
 particles to be counted one at the time as they pass through the flowcell,
 and includes a pair of electrodes 8 and 9, to which a DC electrical field
 for measuring the size or volume of each particle and an RF field for
 measuring the density of each particle passing through the flowcell
 aperture 5 are applied.
 In particular, the dimensions of the flowcell measurement aperture 5 define
 a "steady state" flowcell characteristic impedance R.sub.a, which may be
 represented by a single capacitance and resistance value at the frequency
 of interest. As particles pass through the flowcell measurement aperture
 5, they introduce changes in the resistance of the flowcell in proportion
 to their size or volume. These changes in aperture resistance are
 reflected as DC voltage pulses at the electrodes 8 and 9, and can be
 measured directly.
 In addition, the density or opacity of a blood cell or particle is
 reflected as a change in the reactance of the flowcell aperture, and has
 been conventionally measured by coupling the electrodes 8 and 9 in
 parallel with the resonance (LC tank) circuit of an associated RF
 oscillator-detector circuit 10. This change in reactance of the flowcell
 causes a corresponding change in the operation of the RF oscillator, which
 can be measured by means of an RF pulse detector/demodulator. For an
 illustration of non-limiting examples of U.S. patent literature detailing
 such conventional oscillator-based flowcell RF detector circuits attention
 may be directed to the U.S. patents to Coulter et al, U.S. Pat. No.
 3,502,974; Groves et al, U.S. Pat. No. 4,298,836; Groves et al, U.S. Pat.
 No. 4,525,666; and Coulter et al, U.S. Pat. No. 4,791,355.
 Now although an RF oscillator-based flowcell measurement circuit of the
 type generally shown in FIG. 1 is effective to provide an indication of
 both size and density of each blood cell, it suffers from a number of
 problems which are both costly and time-consuming to remedy. One
 fundamental shortcoming is the fact that the particle detection mechanism
 was originally designed as and continues to be configured as a tube-based
 RF Hartley oscillator circuit. This potentially impacts circuit
 availability, as the number of manufacturers of vacuum (as well as gas
 filled) electronic tubes continues to decline.
 In addition, the effective lifetime of a newly purchased and installed tube
 in the Hartley oscillator is not only unpredictable, but experience has
 shown that the effective functionality of most tubes within the Hartley
 oscillator--detector circuit is very limited, (even though a tube tester
 measurement shows a tube to be good). At best a tube can expect to last
 somewhere in a range of three to nine months--and typically involves on
 the order of two repair/maintenance service calls per year per flowcell.
 SUMMARY OF THE INVENTION
 In accordance with the present invention, rather than use a
 change-in-reactance based, RF Hartley oscillator-configured detector to
 measure particle/cell density, both cell volume and internal cellular
 conductivity are measured by a DC/RF-stimulated bridge detector. The
 bridge detector of the invention has a circuit configuration generally of
 the type employed in a Wheatstone bridge, and uses opto-isolator
 components for galvanic isolation from sources of signal degradation that
 might otherwise substantially impair the ability of the bridge to conduct
 accurate particle detection measurements.
 Like a conventional Wheatstone bridge, the DC/RF-driven bridge of the
 invention includes a first voltage divider branch, in which the object
 being monitored (e.g., a flowcell) is installed. The first branch of the
 bridge also includes a linear impedance element connected in a series
 circuit path between bridge stimulation terminals, across which a high
 frequency voltage (on the order of several tens of MHz), and a DC
 excitation voltage are applied. Also coupled in circuit with the flowcell
 and one of the stimulation terminals is an automatic amplitude and phase
 balancing, non-linear network (such as a resistor-capacitor network).
 The DC/RF stimulated bridge detector of the invention also has a second
 voltage divider branch containing a flowcell circuit model, which mirrors
 the impedance of the actual flowcell, and another linear impedance element
 connected in a series circuit path between the bridge stimulation
 terminals. The flowcell circuit model functions as an automatic amplitude
 and phase balance circuit, and comprises an adjustable non-linear network,
 such as, but not limited to a variable capacitor and a linear resistor
 coupled in circuit between the high frequency voltage terminal and a
 bridge output node. The linear resistance elements of the DC/RF bridge of
 the invention virtually eliminate second order Laplacian effects
 associated with coupling amplifier circuits. The input capacitance of each
 coupling amplifier circuit--together with the linear resistor--forms a
 first order filter having a cut-off frequency defined by the values of the
 resistor and the input capacitance of an associated coupling amplifier.
 A first bridge output node is coupled to a first current gain amplifier,
 whose output is coupled to a difference amplifier. The difference
 amplifier is also coupled to the output of a second current gain
 amplifier, the input of which is coupled to a second bridge output node.
 This differential amplifier connection effectively cancels inherent
 common-mode noise, as well as residual noise caused by the imbalance in
 the two branches of the bridge. The output of the difference amplifier is
 coupled to a DC/RF discriminator and associated downstream processing
 circuitry.
 By virtue of opto-isolator coupling and its isolated self-powered
 architecture, the modified Wheatstone bridge detector of the invention is
 effectively a "floating" bridge, that galvanically isolates the front-end
 signal detection circuits from very high frequency noise components
 sourced from the RF oscillator. As a result, filter bandwidths in the
 downstream signal processing circuits can be made much wider to
 accommodate all of the signal energy density, with virtually no
 interference from RF noise in the detected signal path.
 A fundamental drawback of a standard Wheatstone bridge network is the
 degradation of signal quality, and a complete loss of signal in cases
 involving very high frequency detection schemes, such as RF pulse
 detection in a flowcell. This signal degradation is mainly due to
 resistances and reactances parasitic in the interconnect components. For a
 properly functioning RF pre-amplifier, the parasitics inherent in the
 bridge must be virtually eliminated. This is effectively accomplished in
 the invention by using commercially available high noise rejection
 components, that allow parasitic-minimizing, flip-chip technology using a
 bare IC die that virtually eliminates capacitive reactance with bridge
 components, thereby making the bridge virtually immune to signal loading
 by parasitic capacitance of the coupling circuits.
 In addition to `floating` the network and using flip-chip mounting, the
 bridge's automatic amplitude and phase balancing circuits serve to track
 variations in the behavior of the flowcell, which has non-linear (both
 resistive and capacitive) characteristics, and is susceptible to
 continuous impedance changes due to temperature, and conductivity of a
 fluid through its measurement aperture. These amplitude and phase
 balancing circuits automatically `tune` their resistive and capacitive
 elements to mirror the behavior of the flowcell in response to
 environmental conditions, so as to minimize common mode noise generated by
 the network, and optimize the signal-to-noise ratio. This automatic phase
 and amplitude adjustment thus makes the bridge virtually immune to
 flowcell load tolerances and varying impedances due to the environment.

DETAILED DESCRIPTION
 As described briefly above, in accordance with the present invention, both
 cell volume and internal cellular conductivity are readily measurable by
 means of a galvanically isolated, Wheatstone bridge-configured DC/RF
 detector. In order to fully appreciate the manner in which the
 bridge-based particle detector of the invention is able to perform very
 sensitive cell measurements in the presence of substantial noise inputs
 from both the environment and the components of the circuit itself, it is
 initially useful to review the fundamental circuit configuration and
 operation of a conventional Wheatstone bridge.
 The basic circuit configuration of a Wheatstone bridge is diagrammatically
 shown in FIG. 2 as comprising a first voltage divider branch 19 containing
 a pair of circuit (impedance) elements 11 and 12, that are connected in
 series between bridge stimulation terminals 13 and 14, and a second
 voltage divider branch 20 containing a pair of impedance elements 21 and
 22 connected between terminals 13 and 14. For stimulating the bridge a
 current or voltage source, either AC or DC, is applied to terminals 13 and
 14.
 The two voltage divider branches 19 and 20 form a dual voltage divider
 network, in which three of the elements, such as impedance elements 11, 12
 and 21, are typically implemented as linear circuit components such as
 resistors, having fixed characteristic values (resistance). The remaining
 (fourth) element 22, which may also be a linear element, has a parameter
 that varies as a function of the environment being measured by the
 network.
 In a static condition, the Wheatstone bridge operates so as maintain
 electrical equilibrium of its two branches; i.e. with all four impedance
 elements being equal, the differential between network output terminals 25
 and 26 will be zero. However, a change in either voltage or current
 proportional to the change in the value of the variable impedance element
 22 will cause the bridge to fall out of its electrical equilibrium, which
 is detected as a non-zero voltage level at the outputs 25 and 26 of the
 bridge. The magnitude of this change may then be calculated and processed
 using conventional downstream-coupled amplification circuitry.
 FIG. 3 diagrammatically illustrates a potential bridge architecture based
 upon the standard Wheatstone bridge circuit of FIG. 2, described above,
 but which is stimulated by means of an RF oscillator, as an approach to
 provide for the detection of particles/cells that may be present in a
 flowcell measurement aperture of a particle flow analysis system of the
 type shown in FIG. 1, described above. Like the Wheatstone bridge of FIG.
 2, the DC/RF-driven bridge of FIG. 3 includes a first voltage dividing
 circuit branch 30 containing a pair of linear impedance circuit elements
 31 and 32, such as resistors of equal value, that are connected in series
 between bridge stimulation terminals 33 and 34.
 The bridge stimulation terminals 33 and 34 are coupled to an RF oscillator
 50, such as a 20 MHz sinusoidal oscillator producing an output voltage at
 40 Vpp. The output of RF oscillator 50 may also be inductively
 (transformer-) coupled to a rectification and scaling circuit 52, which is
 operative to appropriately scale down the RF voltage to provide isolated
 DC power to the bridge proper, and to downstream signal processing
 circuits.
 The DC/RF bridge further includes a second voltage divider branch 40
 connected between the bridge stimulation terminals 33 and 34. One of the
 impedance elements of the second branch 40 is a flowcell 41, whose
 characteristic value (impedance) is expected to vary due to the presence
 of a particle, in particular, a blood particle in the flowcell s
 measurement aperture. The other impedance element 42 is comprised of a
 resistor-capacitor circuit network that is configured to model or very
 closely approximate the behavior of the flowcell 41, tracking (non-linear)
 variations in impedance the flowcell in response to changes in the
 environment, such as temperature and the conductivity of a fluid through
 its measurement aperture, and keep the bridge in balance in a
 self-adjusting manner.
 For this purpose, a DC component as well as a phase component in the RF/DC
 output signal obtained from output nodes 45 and 46 of the bridge are used
 to maintain closed loop balancing. As will be described, these DC and RF
 components are fed back to respective threshold comparators 84 and 85, the
 outputs of which are monitored by the flowcell's control processor 90 to
 detect and correct for bridge imbalances. In particular, the DC/RF output
 pulses from the output node 45 between the circuits 41 and 42 of the
 flowcell-containing branch, and the output node 46 between reference
 branch resistors 31 and 32 are coupled through respective high speed
 buffer amplifiers 60 and 61 and passed therefrom to a DC/RF discriminator
 62, which is operative to separate the DC and RF components from the
 composite signal produced at output nodes 45,46.
 The RF component is coupled to an RF conditioning circuit path 70 comprised
 of an RF pre-amp detector 71 coupled in cascade with first and second RF
 pre-amp stages 72 and 73. The output of the second RF pre-amplifier stage
 73 is coupled through a linear opto-isolator 74 to downstream RF interface
 circuitry (DC restoration, and Peak Detection circuits) for appropriate
 pulse sizing and sorting. In like manner, the DC component is coupled to a
 DC conditioning circuit path 80 comprised of cascaded first and second DC
 pre-amp stages 81 and 82. The output of the second DC pre-amp stage 82 is
 coupled through a linear opto-isolator 83 to downstream DC interface
 circuitry. The outputs of the first DC pre-amp stage 81 and the RF
 preamplifier stage 72 are coupled to respective threshold comparators 84
 and 85, the outputs of which are monitored by the flowcell control
 processor 90 to detect and correct bridge imbalances, such as those that
 may be attributable to small impedance drifts due to the flowcell proper,
 ISOTON.RTM. conductivity, etc.
 Now although the potential RF/DC bridge circuit architecture of FIG. 3
 ostensibly provides an alternative to a conventional change-in-reactance
 Hartley oscillator referenced above, our investigation of its performance
 and circuit properties has revealed that it is very `noisy` and needs to
 be modified in order to realize a commercially practical embodiment for
 conducting accurate flowcell measurements.
 More particularly, a first aspect that requires adjustment involves the
 fact that the input impedance of the output coupling buffer (typically on
 the order of 3 pF-5 pF) undesirably loads down the output signal due to
 the change in the impedance of the flowcell as a blood particle passes
 through it. As will be discussed in detail below, the impedance of the
 flowcell is both resistive and capacitive. This reactance couples with the
 input impedance of the buffer and acts as a second order filter that
 substantially suppresses the desired signal.
 Secondly, the large resistor values in the bridge network, which are
 typically used in a Wheatstone bridge network to optimize signal
 detection, together with the parasitic reactances of their leads act as
 lumped low pass filters. It is common knowledge that a Wheatstone bridge
 network typically produces erroneous output measurements due to parasitic
 resistances inherent in the network. These parasitics can also vary due to
 temperature, conductivity or other conditions present in the environment
 of the object parameter being measured. In most standard applications,
 measurements are conducted at DC or at very low frequencies close to DC
 (e.g., 60 Hz). The measurement error in the bridge under these conditions
 is negligible.
 However, at very high frequencies (e.g. on the order of 20 MHz or above,
 commonly used in medical instrumentation), large resistor values (e.g., on
 the order of several tens of kilohms) coupled with parasitic capacitance
 inherent in the bridge network (typically on the order of 1-3 pF) further
 degrade the performance of the bridge and make measurements unpredictable
 and inaccurate. As a non-limiting example, a resistance 38.5 K.OMEGA. and
 a parasitic capacitance on the order of 1 pF yields a roll-off frequency
 of 1/2.pi.*38.5 K.OMEGA.*1 pF=4.1 MHz, which significantly reduces the
 energy in a 20 MHz excitation source used to drive the bridge of FIG. 3,
 and makes RF measurements effectively impossible.
 In order to gain an appreciation of the performance of a DC/RF driven
 bridge that led to a modification to realize a working embodiment of the
 invention, the present inventors conducted an analysis of a typical
 flowcell and its impedance characteristics. Using a spectrum analyzer test
 set-up as diagrammatically illustrated in FIG. 4, the impedance of a
 flowcell 100 of the type used in a system of the type diagrammatically
 illustrated in FIG. 1 was measured as a function of frequency. The output
 of the sweep oscillator 101 of the test circuit set-up of FIG. 4 was swept
 over a prescribed range (e.g., 0-40 MHz), using the tracking generator of
 the spectrum analyzer 103. A voltage divider resistor 105 coupled in
 series with the flowcell 100 has negligible loading effects, due to the
 flowcell and parasitic reactances of the test set-up.
 FIG. 5 is an impedance plot obtained from the flowcell measurement test set
 up of FIG. 4, while FIG. 6 is an impedance plot of a mathematical
 (electrical) flowcell model.
 Using field theory, the flowcell was further analyzed to determine its
 approximate capacitance and reactance. In this analysis, the following
 conditions were assumed: 1--ISOTON is a conductor; 2--the wavelength of
 the test frequency (20 MHz) is considerably greater than length (Lap) and
 width or diameter (Dap) dimensions of the flowcell aperture, which are of
 an order approximating the size of the particle passing through it; and
 3--a lumped parasitic capacitance of the flowcell.
 FIG. 7 is a plot of the variation in the operating point of an empty
 flowcell (i.e., a flowcell with no particle) in the complex plane, as its
 capacitance is varied between 0.05 pF and 0.3 pF (values obtained from
 field theory). The flowcell impedance values approximate both actually
 measured and modeled values. Using these measured and calculated values,
 the approximate `change` in impedance of the flowcell due to the presence
 of a particle in the measurement aperture was determined, and plotted in
 FIG. 8.
 As can be seen from the impedance variation diagram of FIG. 8, the change
 in the impedance of the flowcell due to the presence of a particle (such
 as a blood cell) is extremely small (on the order of only about 10 in
 10,000). It is readily apparent, therefore, that a conventional Wheatstone
 bridge circuit design, such as that of FIG. 3, will not successfully
 detect very minute complex electrical changes in its network and, more
 specifically, in a very high frequency environment, such as a flowcell
 employing a high frequency RF signal to measure cell density.
 This shortcoming of a conventional Wheatstone bridge approach is remedied
 by the modified bridge architecture of FIG. 9, which shows a differential
 amplifier-coupled, DC/RF-stimulated, bridge-configured detector having
 automatic amplitude and phase balance circuits in accordance with the
 present invention, the output of which is coupled to the downstream
 amplifier and opto-isolator components of FIG. 3, detailed above.
 As will be described, the automatically balanced bridge detector
 architecture of the invention is preferably implemented using flip-chip
 technology, using a bare IC die that virtually eliminates capacitive
 reactance with bridge components, thereby making the bridge virtually
 immune to signal loading by parasitic capacitance of the coupling
 circuits. The DC/RF-driven bridge detector architecture of the invention
 is operative to measure, in the presence of very large common mode noise
 commonly encountered in a bridge environment, and especially in a very
 high frequency environment, extremely small and complex electrical
 signals, reliably and virtually unaffected by parasitic resistances that
 commonly exist in the network.
 As noted above with reference to the flowcell impedance change plot of FIG.
 8, the change in impedance of the flowcell due to the presence of a
 particle in the cell's measurement aperture is extremely small--on the
 order of only 10 .mu..OMEGA.. Since it is well established that the
 characteristic impedance of the flowcell is determined by its aperture
 length L.sub.ap and diameter D.sub.ap, the diameter being the most
 dominant factor, it is clear that variations in the dimensions of the
 flowcell (which are attributable to tolerances in the flowcell
 manufacturing process) will significantly change the operating point
 (impedance) of the flowcell. Therefore, in order to maintain adequate
 performance of the RF pre-amplifier in the RF conditioning circuit path,
 it would appear that the dimensions of the flowcell should be tightly
 controlled. Unfortunately, this tolerance requirement places a significant
 strain on the manufacturing process, and makes manufacturing of flowcells
 difficult and expensive.
 On the other hand, the automatic amplitude and phase balanced bridge
 circuit architecture of FIG. 9 makes RF detection and measurement
 virtually insensitive to such flowcell dimensional variations as commonly
 occur in manufacturing processes. More particularly, the DC/RF bridge
 design of FIG. 9 comprises a first voltage dividing circuit branch 120
 containing a flowcell 122 and a first linear impedance element (resistor)
 124 connected in a series circuit path between first and second bridge
 stimulation terminals 126 and 128, across which +/-20 MHz voltages are
 applied. The peak-to-peak excitation voltage applied across terminals 126
 and 128 should be sufficiently large to optimize the signal-to-noise ratio
 between output nodes 135 and 155.
 Also coupled in circuit with the flowcell 122 and the +20 MHz terminal 126
 is an automatic amplitude balance circuit 130 comprising a variable
 resistor 131, having its variable tap coupled through a capacitor 133 to a
 +20 MHz terminal 127. The value of variable resistor 131 is controlled by
 an amplitude control signal coupled to terminal 134. The first bridge
 output node 135 is coupled to the common connection between flowcell 122
 and the first linear impedance element 124.
 The DC/RF bridge design of FIG. 9 further comprises a second voltage
 dividing circuit branch 140 containing a flowcell circuit model 142 and a
 second linear impedance element (resistor) 144 connected in a series
 circuit path between the first and second bridge stimulation terminals 126
 and 128. The flowcell circuit model 142, which functions as an automatic
 phase balance circuit, comprises a variable capacitor 151 and a linear
 resistor 153, coupled in circuit between the +20 MHz terminal 126 and the
 second bridge output node 155. The value of variable capacitor 151 is
 controlled by a phase control signal coupled to terminal 154.
 Installing linear elements 124 and 144 in the bottom legs of the bridge
 architecture of FIG. 9 virtually eliminates second order Laplacian effects
 associated with coupling circuits. The input capacitance of each coupling
 circuit together with the linear resistor (124/144) forms a first order
 filter having a cut off frequency determined by the value of the resistor
 and the input capacitance of an associated coupling amplifier.
 The first bridge output node 135 is coupled to a first front-end signal
 gain, current amplifier 160, the output of which is coupled to a first
 input 171 of a difference amplifier 170. A second input 172 of difference
 amplifier 170 is coupled to the output of a second front-end amplifier
 180, the input of which is coupled to the second bridge output node 155.
 The difference amplifier 170 is operative to subtract common-mode noise
 inherent in the network, as well as residual noise caused by the imbalance
 in the two branches 120 and 140 of the bridge. The output of the
 difference amplifier 170 is coupled to a DC/RF discriminator and
 associated downstream processing circuitry corresponding to that employed
 to process the output of the bridge design of FIG. 3, described above.
 By virtue of its differential coupling configuration and the linear
 opto-isolation described above, the modified bridge implementation of FIG.
 9 provides what is effectively a "floating" bridge and associated pulse
 detection circuits. Rather than being referenced to common `metallic`
 ground, as are conventional bridge networks, the bridge circuit of FIG. 9
 galvanically isolates the front-end signal detection circuits from
 downstream signal processing and filtering circuits (so that there is no
 metallic connection path therebetween).
 A fundamental drawback to using a common ground reference is the need to
 employ extensive filtering in the signal detection and processing
 circuits, in order to remove unwanted high frequency noise components that
 are buried in the signal. The bandwidths of these filters have to be made
 narrow enough to minimize the effects of noise and to optimize the
 signal-to-noise ratio. However, the narrow bandwidth of the filter, in
 addition to filtering noise, also causes signal attenuation. As a result,
 compensation for the loss of signal amplitude due to attenuation requires
 a gain adjustment circuit; this, in turn, undesirably amplifies the
 inherent noise of the processing circuits.
 The "floating" scheme of FIG. 9, on the other hand, prevents high frequency
 components, inherent in the oscillator, from interfering with the actually
 detected pulse. This means that the filter bandwidths in the downstream
 signal processing circuits can be made much wider to accommodate all of
 the signal energy density, with virtually no interference from noise in
 the signal path.
 As noted earlier, one of the fundamental drawbacks of a standard Wheatstone
 bridge network is the degradation of signal quality and, in cases
 involving very high frequency detection schemes, such as RF pulse
 detection, a complete loss of signal. This signal degradation is mainly
 due to resistances and reactances parasitic in its connecting components.
 For a properly functioning RF pre-amp, the parasitics inherent in the
 bridge must be virtually eliminated, i.e. the coupling amplifiers 160, 170
 and 180 must have very low input capacitance, which real life components
 do not typically exhibit.
 In accordance with a preferred implementation, the front end coupling
 amplifiers employed in the bridge architecture of FIG. 9 may be
 commercially available components, such as AD9631 model components
 manufactured by Analog Devices. These devices have been found to exhibit
 an excellent gain-bandwidth product, slew rate and offset specifications
 required for flowcell particle measurements. However, because the input
 capacitance C of this coupling amplifier is not insignificant (on the
 order of 3 pF.ltoreq.C.ltoreq.7 pF, which is typical of high performance
 amplifiers), it introduces substantial loading at high frequencies, and
 must be compensated.
 Pursuant to a further aspect of the invention, this is accomplished by
 mounting the output signal coupling amplifier circuits using flip-chip
 technology, and thereby make the bridge virtually immune to signal loading
 by the coupling circuits. In particular, virtually all signal loading due
 to the input impedance of the coupling amplifiers is eliminated by
 flip-chip mounting the bare die of the amplifier integrated circuit. It
 has been found that when the Analog Devices part No. AD9631, referenced
 above, is flip-chip mounted to the printed circuit board containing the
 other components of the bridge, the input capacitance of the coupling
 amplifier is effectively reduced to less than 0.3 pF, and thereby
 significantly reduces loading at the high operational frequency of
 interest (e.g., 20 MHz). Such mounting also affords high circuit
 integration at lower manufacturing costs and improved circuit performance.
 In addition to `floating` the network and using flip-chip mounting, as
 described above, the modified bridge implementation of FIG. 9 incorporates
 automatic amplitude and phase balancing circuits 130 and 142,
 respectively. As pointed out earlier with reference to FIGS. 1 and 2, a
 typical Wheatstone bridge network contains three linear elements. When
 measuring a change in the characteristic of an environment, it is common
 practice to physically place in the environment a fourth element, whose
 value varies as a function of the environment being measured. It is also a
 common practice to maintain a "good" balance between the two voltage
 divider branches of the bridge, so as to minimize the effects on common
 mode noise.
 In the modified Wheatstone bridge architecture of the present invention,
 the fourth element is a flowcell 122. Unlike typical bridge elements, the
 flowcell has non-linear (both resistive and capacitive) characteristics,
 as described above with reference to FIGS. 4-8B, and is prone to
 continuous impedance changes due to temperature, and conductivity of a
 fluid (such as ISOTON.RTM.) through its measurement aperture. To
 compensate for this non-linear behavior, the bridge circuit of FIG. 9 also
 employs the balancing circuits 130 and 142, whose characteristics closely
 resemble those of the flowcell proper.
 The values of the variable components of automatic amplitude and phase
 balancing circuits are controllably adjusted by the flowcell control
 processor 90 during a calibrate mode of operation to correct bridge
 imbalances, such as those that may be attributable to small impedance
 drifts due to the flowcell proper, ISOTON conductivity, etc. Bridge
 calibration may be performed prior to a blood sample analysis, by flowing
 a blood sample-free saline solution through the flowcell, and monitoring
 the bridge's DC voltage output as extracted by the DC/RF discriminator 62
 of FIG. 3, referenced above. If there has been a drift or offset in the
 flowcell impedance, for example due to temperature, the bridge's control
 processor will adjust the parameters of the flowcell balancing circuitry
 so as to drive the bridge's differential voltage output to zero.
 Namely, during bridge calibration, the control processor is operative to
 `tune` the resistive and capacitive elements of the balance circuit so as
 to mirror the characteristics of the flowcell, and thereby automatically
 balancing the amplitude and phase in the bridge, minimizing common mode
 noise generated by the network, and optimizing the signal-to-noise ratio.
 This automatic adjustment makes the network virtually immune to flowcell
 load tolerances and varying impedances due to the environment.
 As will be appreciated from the foregoing description, shortcomings of
 conventional flowcell detectors, such as change-in-reactance based, RF
 Hartley oscillator-configured circuits for measuring both cell volume and
 internal cellular conductivity are effectively obviated by the
 differential DC/RF bridge-configured detector of the present invention.
 The respective legs of the bridge, which contain both the flowcell proper,
 and an adjustable flowcell circuit model, are differentially coupled
 through output amplifier circuits and galvanically isolated from sources
 of signal degradation that would otherwise substantially impair the
 ability of the bridge to conduct accurate particle detection measurements.
 Effectively floating the bridge serves to galvanically isolate the
 front-end signal detection circuits. As a result, filter bandwidths in the
 downstream signal processing circuits can be made much wider to
 accommodate all of the signal energy density, with virtually no
 interference from noise in the signal path.
 While we have shown and described an embodiment in accordance with the
 present invention, it is to be understood that the same is not limited
 thereto but is susceptible to numerous changes and modifications as known
 to a person skilled in the art, and we therefore do not wish to be limited
 to the details shown and described herein, but intend to cover all such
 changes and modifications as are obvious to one of ordinary skill in the
 art.