Precision reversed bandgap voltage reference circuits and method

A circuit producing a reversed bandgap reference voltage circuit VRBG includes first and second resistors coupled as a voltage divider between ground and a first conductor, a base of a first transistor being coupled to the voltage divider to produce a first voltage VBE1(1+1/M) between the first conductor and ground, M being a ratio of the resistances of the first and second resistors. A third resistor is coupled between a base of the second transistor and ground to produce a second voltage VBE2+VRBGP between the second conductor and ground. First circuitry forces the collector current of the first transistor to be equal to the collector current of the second transistor, and second circuitry forces the first voltage VBE1(1+1/M) to be equal the second voltage VBE2+VRBGP. One of the first circuitry and second circuitry includes an operational amplifier coupled to effectuate the forcing.

BACKGROUND OF THE INVENTION

The present invention relates generally to reversed bandgap voltage reference circuits, and more particularly to reversed bandgap voltage reference circuits which are capable of operating from power supply voltages of less than 1 volt and also are more accurate than those of the prior art.

Traditional bandgap voltage reference circuits (see D. A. Johns and K. Martin, “Analog Integrated Circuit Design”, Wiley, 1997, page 357) provides Vout approximately equal to 1.2 volts with less than 50 ppm/degrees Centigrade temperature coefficient (TC) and better than 2% scattering of the output voltage in production volumes without the need for trimming of resistor values. Since the output bandgap voltage is approximately 1.2 volts, the disclosed bandgap reference voltage circuit can not operate from supply voltages below approximately 1.3V.

FIG. 1illustrates a conventional bandgap reference voltage circuit which includes a PNP transistor Q1having its base and collector connected to ground, with its emitter receiving a current from a P-channel transistor6of a current mirror circuit also including P-channel transistors5and7. PNP transistor Q2has an emitter area that is N times greater than that of transistor Q1. The base and emitter of transistor Q2are connected to ground, and its emitter is coupled by a resistor4to receive a current from P-channel current mirror transistor7equal to the current supplied by transistor6to transistor Q1. The emitter of transistor Q1is connected to the (−) input of operational amplifier1, and the upper terminal of resistor4is connected to the (+) input of operational amplifier1. The difference in the base-emitter voltages VBE1and VBE2of transistors Q1and Q2, respectively, due to the scaling ratio N of their emitter areas is equal to the voltage across resistor4and is used to generate the output bandgap reference voltage Vref. The circuit ofFIG. 1is capable of operation from a 1 volt power supply. A relatively large amount of 1/f noise is generated in the current mirror transistors is in the feedback loop and therefore causes large corresponding noise errors in the generated output voltage Vref. Consequently, a large external filter capacitance is needed to limit the noise bandwidth.

To satisfy the growing need for reference voltage circuits capable of functioning at lower supply voltages, various other attempts have been made to create circuits based on temperature properties of the threshold VTHand carrier mobility μ of the MOS transistor. (See I. M. Filanovsky, A. Allam, “Mutual Compensation of Mobility and Threshold Voltage Temperature Effects with Applications in CMOS Circuits”, IEEE TCAS-I, vol. 48, no. 7, pp. 876-884, 2001). However, relatively poor manufacturing repeatability and poor control of the process-defined VGS threshold voltage VTHprevents the circuits disclosed in these references from being widely adopted by the industry. Accuracy and production “scattering” of such bandgap reference voltage circuits are significantly worse than for the traditional bandgap reference voltage circuit shown inFIG. 1.

FIG. 2shows a known current mode bandgap reference voltage circuit which includes an NPN transistor Q1having its emitter connected to ground, its base connected to one terminal of a resistor R1having its other terminal connected to ground. The collector of transistor Q1is connected to Vref and to the emitter of a diode-connected NPN transistor Q2having an emitter area that is N times that of transistor Q1. The base and collector of transistor Q2are connected to one terminal of a current source and to one terminal of a resistor R2, the other terminal of which is connected to the base of transistor Q1. This circuit has the shortcomings that transistor Q1operates close to saturation and therefore the circuit is subject to errors caused by large base currents. This circuit also requires the current source Ibias to be a complicated circuit capable of providing a complicated temperature coefficient.

Various other attempts also have been made to create bandgap reference voltage circuits based on current-mode operation, by combining positive-TC and negative-TC current sources to create a temperature independent current. This current is transferred to a resistor by a current mirror to generate the reference voltage. (See P. Malcovati, F. Maloberti, C. Fiocci, and M. Pruzzi, “Curvature-compensated BiCMOS bandgap with 1-V supply voltage”, IEEE JSSC, vol. 36, no. 7, pp. 1076-1081, 2001). However, the main drawback of the current-mode reference voltage circuits disclosed in these references is the presence of the current mirror. Regardless of the circuit techniques and components used to create the positive-TC and negative-TC currents, the accuracy of such reference can not be better than accuracy of the current mirror and the resistor (considering matching and noise). In general, an improvement in current mirror accuracy can be achieved with sampling techniques. The noise can be reduced by using a large filtering capacitor at the output. However, this leads to a more complicated circuit, larger die area and increased current consumption, with somewhat compromised accuracy.

The voltage reference in FIG. 2 of the Banba reference using the “reversed bandgap principle” has been implemented with NPN transistors. One of the core NPN transistors operates with ˜190 millivolts collector-to-emitter voltage (VCE). Being that close to saturation, the parasitic substrate PNP structure, which is present in vertical NPN transistors on all but SOI (silicon on insulator) processes, becomes activated. This in turn increases the value of the base current and decreases its predictability. This circuit also requires a separate bias with a complicated TC. As a result, the accuracy is poor and this reference voltage circuit cannot compete with traditional bandgap reference voltage circuits at higher supply voltages.

Thus, there is an unmet need for a reversed bandgap voltage reference circuit which provides a more precise reference voltage than has been previously obtainable from reference voltage circuits capable of operating from power supply voltages of less than 1 volt.

There also is an unmet need for a reversed bandgap voltage reference circuit which provides a more precise reference voltage having substantially lower noise than has been previously obtainable from reference voltage circuits capable of operating from power supply voltages of less than 1 volt.

There also is an unmet need for a reversed bandgap voltage reference circuit which provides a more precise reference voltage having substantially lower noise than has been previously obtainable from reference voltage circuits capable of operating from power supply voltages of less than 1 volt and which avoids the need for providing complex current source circuitry to compensate for temperature coefficient errors.

SUMMARY OF THE INVENTION

It is an object of the invention to provide a reversed bandgap voltage reference circuit which provides a more precise reference voltage than has been previously obtainable from reference voltage circuits capable of operating from power supply voltages of less than 1 volt.

It is another object of the invention to provide a reversed bandgap voltage reference circuit which provides a more precise reference voltage having substantially lower noise than has been previously obtainable from reference voltage circuits capable of operating from power supply voltages of less than 1 volt.

It is another object of the invention to provide a reversed bandgap voltage reference circuit which provides a more precise reference voltage having substantially lower noise than has been previously obtainable from reference voltage circuits capable of operating from power supply voltages of less than 1 volt and which avoids the need for providing complex current source circuitry to compensate for temperature coefficient errors.

Briefly described, and in accordance with one embodiment, the present invention provides a circuit producing a reversed bandgap reference voltage circuit VRBGincluding first (R1) and second (R2) resistors coupled as a voltage divider between ground and a first conductor (17), a base of a first transistor (Q1) being coupled the voltage divider to produce a first voltage VBE1(1+1/M) between the first conductor and ground, M being a ratio of the resistances of the first and second resistors. A third resistor (R4) is coupled between a base of the second transistor and ground to produce a second voltage VBE2+VRBGPbetween the second conductor and ground. First circuitry forces the collector current (IC1) of the first transistor (Q1) to be equal to the collector current (IC2) of the second transistor (Q2), and second circuitry forces the first voltage VBE1(1+1/M) to be equal the second voltage VBE2+VRBGP. One of the first circuitry and second circuitry includes an operational amplifier coupled to effectuate the forcing.

In one embodiment, a circuit for producing a reversed bandgap reference voltage VRBGPincludes a first transistor (Q1) and a second transistor (Q2) having an emitter area substantially greater than that of the first transistor (Q1). First (R1) and second (R2) resistors are coupled in series between a reference voltage conductor (GND) and a first conductor (17in FIGS.4A,B or17A in FIGS.5A,B) and a base of the first transistor (Q1) is coupled to a junction (32A) between the first and second resistors, to produce a first voltage VBE1(1+1/M) between the first conductor (17or17A) and the reference voltage conductor (GND), wherein VBE1is the base-emitter voltage of the first transistor (Q1) and M is a ratio of the resistances of the first (R1) and second (R2) resistors. A third resistor (R4in FIGS.4B,5A,B or R5inFIG. 4A) across which the reversed bandgap reference voltage VRBGPis produced is coupled between a base of the second transistor (Q2) and the reference voltage conductor (GND) and a second conductor (17in FIGS.4A,B or17B in FIGS.5A,B) to produce a second voltage VBE2+VRBGPbetween the second conductor (17or17B) and the reference voltage conductor (GND), wherein VBE2is the base-emitter voltage of the second transistor (Q1). First circuitry is coupled to effectuate forcing the collector current (IC1) of the first transistor (Q1) to be equal to the collector current (IC2) of the second transistor (Q2), and second circuitry is coupled to effectuate forcing the first voltage VBE1(1+1/M) to be equal the second voltage VBE2+VRBGP. In all but one of the disclosed embodiments, at least one of the first circuitry and second circuitry includes an operational amplifier coupled to effectuate the forcing.

In one embodiment, the first (Q1) and second (Q2) transistors are NPN transistors, an emitter of the second transistor (Q2) is coupled to a first terminal of the third resistor (R5) by means of a conductor conducting the reversed bandgap reference voltage VRBGP, a second terminal of the third resistor (R5) is coupled to the reference voltage conductor (GND), and an emitter of the first transistor (Q1) is coupled to the reference voltage conductor (GND). The first circuitry includes a first operational amplifier (12), a P-channel transistor (M1) having a gate coupled to an output of the first operational amplifier (12) and a drain coupled by the first conductor (17) to a first terminal of a fourth resistor (R3) and a first terminal of a fifth resistor (R4). The fourth resistor (R3) has a second terminal coupled to a collector of the first transistor (Q1) and a first input of the first operational amplifier (12). The fifth resistor (R4) has a second terminal coupled to a collector of the second (Q2) transistor and a second input of the first operational amplifier (12). The second circuitry includes the first conductor (17) coupled to a base of the second transistor (Q2).

In another embodiment, the first (Q1) and second (Q2) transistors are PNP transistors, a collector of the first transistor (Q1) is coupled to a first terminal of a fourth resistor (R3), a collector and a base of the second transistor (Q2) are coupled to the first terminal of the third resistor (R4) by means of a conductor conducting the reversed bandgap reference voltage VRBGP, emitters of the first (Q1) and second (Q2) transistors are coupled to the first conductor (17). The first circuitry includes a first operational amplifier (12), a P-channel transistor (M1) having a gate coupled to an output of the first operational amplifier (12) and a drain coupled to the first conductor (17). The third (R4) and fourth (R3) resistors each have a second terminal coupled to the reference voltage conductor (GND). The first terminal of the third resistor (R4) is coupled to a first input of the first operational amplifier (12), and the first terminal of the fourth resistor (R3) is coupled to a second input of the first operational amplifier.(12). The second circuitry includes the first conductor (17) connected to the emitters of the first (Q1) and second (Q2) transistors.

In another embodiment, the first (Q1) and second (Q2) transistors are PNP transistors, emitters of the first (Q1) and second (Q2) transistors are coupled to the first (17A) and second (17B) conductors, respectively. A collector of the first transistor (Q1) is coupled to the reference voltage conductor (GND), a base and collector of the second transistor (Q2) are coupled by means of a conductor conducting the reversed bandgap reference voltage VRBGPto a first terminal of a third resistor (R4), and a second terminal of the third resistor (R4) is coupled to the reference voltage conductor (GND). The first circuitry includes matched first (M1) and second (M2) P-channel transistors. The second circuitry includes a first operational amplifier (12), the first (M1) and second (M2) P-channel transistors having gates coupled to an output of the first operational amplifier (12). The first P-channel transistor (M1) has a drain coupled by the first conductor (17A) to a first input of the first operational amplifier (12), and the second P-channel transistor (M2) has a drain coupled by the second conductor (17B) to a second input of the first operational amplifier (12). In that described embodiment, a matching resistance (R2A,R1A) equal to a series resistance of the first (R1) and second (R2) resistors may be coupled between the second conductor (17B) and the reference voltage conductor (GND).

In another embodiment, the first (Q1) and second (Q2) transistors are NPN transistors, collectors of the first (Q1) and second (Q2) transistors are coupled to the first (17A) and second (17B) conductors, respectively, an emitter of the first transistor (Q1) being coupled to the reference voltage conductor (GND), an emitter of the second transistor (Q2) being coupled by means of a conductor conducting the reversed bandgap reference voltage VRBGPto a first terminal of a third resistor (R4). A second terminal of the third resistor (R4) is coupled to the reference voltage conductor (GND), and a base and collector of the second transistor (Q2) are coupled to the second conductor (17B). The first circuitry includes matched first (M1) and second (M2) P-channel transistors. The second circuitry includes a first operational amplifier (12), the first (M1) and second (M2) P-channel transistors having gates coupled to an output of the first operational amplifier (12). The first P-channel transistor (M1) has a drain coupled by the first conductor (17A) to a first input of the first operational amplifier (12), and the second P-channel transistor (M2) has a drain coupled by the second conductor (17B) to a second input of the first operational amplifier (12). The first circuitry may include a matching resistance (R2A,R1A) equal to a series resistance of the first (R1) and second (R2) resistors coupled between the second conductor (17B) and the reference voltage conductor (GND).

In another embodiment, the first (Q1) and second (Q2) transistors are PNP transistors, emitters of the first (Q1) and second (Q2) transistors are coupled to the first (17A) and second (17B) conductors, respectively, collectors of the first (Q1) and second (Q2) transistors being coupled to the reference voltage conductor (GND). A base of the second transistor (Q2) is coupled by means of a third conductor (34B) conducting the reversed bandgap reference voltage VRBGPto a first terminal of a third resistor (R4) and a first terminal of a fourth resistor (R5). A second terminal of the third resistor (R4) is coupled to the reference voltage conductor (GND). The first circuitry includes matched first (M1) and second (M2) P-channel transistors and a third P-channel transistor (M3) and a first operational amplifier (12). The first (M1), second (M2) and third (M3) P-channel transistors have gates coupled to an output of the first operational amplifier (12). A drain of the third P-channel transistor (M3) is coupled by a fourth conductor (34A) to a first input of the first operational amplifier (15) and to a first terminal of a fifth resistor (R6) having a second terminal coupled to the ground reference voltage (GND). A second input of the first operational amplifier (15) is coupled to the third conductor (34B). The second circuitry includes a second operational amplifier (15), the first P-channel transistor (M1) having a drain coupled by the first conductor (17A) to a first input of the second operational amplifier (15). The second P-channel transistor (M2) has a drain coupled by the second conductor (17B) to a second input of the second operational amplifier (15). In output of the second operational amplifier (15) is coupled to a second terminal of the fourth resistor (R5) by means of an output conductor conducting a scaled-up voltage (Vref) representative of the reversed bandgap voltage VRGBP.

In another embodiment, the first (Q1) and second (Q2) transistors are PNP transistors, emitters of the first (Q1) and second (Q2) transistors being coupled to the first conductor (17). A collector of the first transistor (Q1) is coupled to a first terminal of a fourth resistor (R6). A collector of the second transistor (Q2) is coupled to a first terminal of a fifth resistor (R7). The first circuitry includes a first operational amplifier (12), a P-channel transistor (M1) having a gate coupled to an output (19) of the first operational amplifier (12) and a drain coupled to the first conductor (17). The third (R4), fourth (R3), and fifth (R7) resistors each have a second terminal coupled to the reference voltage conductor (GND). The first terminal of the fourth resistor (R6) is coupled to a first input of the first operational amplifier (12). The first terminal of the first resistor (R1) is coupled to a base of the first transistor (Q1) and a second input of the first operational amplifier (12). The circuitry also includes a second operational amplifier (15) having a first input coupled to the first terminal of the fourth resistor (R6) and a second input coupled to the first terminal of the fifth resistor (R7). In output of the second operational amplifier (15) is coupled to a first terminal of a sixth resistor (R5) by means of a conductor conducting a scaled-up voltage (Vref) representative of the reversed bandgap voltage VRBGP.The sixth resistor (R5) has a second terminal coupled to the first terminal of the third resistor (R4). The second circuitry includes the first conductor (17) connected to the emitters of the first (Q1) and second (Q2) transistors.

In all embodiments of the invention, the value of the third resistor is sufficiently low to prevent saturation of the second transistor (Q1).

In one embodiment, the invention provides a method for producing a reversed bandgap reference voltage VRBGP, including providing a first transistor (Q1) and a second transistor (Q2) having an emitter area substantially greater than that of the first transistor (Q1), producing a first voltage VBE1(1+1/M) between a first conductor (17in FIGS.4A,B or17A in FIGS.5A,B) and a reference voltage conductor (GND), wherein VBE1is the base-emitter voltage of the first transistor (Q1) and M is a ratio of the resistances of first (R1) and second (R2) resistors coupled in series between the reference voltage conductor (GND) and the first conductor (17in FIGS.4A,B or17A in FIGS.5A,B). The method includes a second voltage VBE2+VRBGPbetween a second conductor (17in FIGS.4A,B or17B in FIGS.5A,B) and the reference voltage conductor (GND), wherein VBE2is the base-emitter voltage of the second transistor (Q1), a third resistor (R4inFIG. 4Bor R5inFIG. 4A) across which the reversed bandgap reference voltage VRBGPis produced being coupled between a base of the second transistor (Q2) and the reference voltage conductor (GND). The collector current (IC1) of the first transistor (Q1) is forced to be equal to the collector current (IC2) of the second transistor (Q2), and the first voltage VBE1(1+1/M) is forced to be equal to the second voltage VBE2+VRBGP. Each of the first (Q1) and second (Q2) transistors is prevented from operating in its saturated region.

In one embodiment, the invention provides circuit for producing a reversed bandgap reference voltage VRBGP, including a first transistor (Q1) and a second transistor (Q2) having an emitter area substantially greater than that of the first transistor (Q1), means for producing a first voltage VBE1(1+1/M) between a first conductor (17in FIGS.4A,B or17A in FIGS.5A,B) and a reference voltage conductor (GND), wherein VBE1is the base-emitter voltage of the first transistor (Q1) and M is a ratio of the resistances of first (R1) and second (R2) resistors coupled in series between the reference voltage conductor (GND) and the first conductor (17in FIGS.4A,B or17A in FIGS.5A,B), means for producing a second voltage VBE2+VRBGPbetween a second conductor (17in FIGS.4A,B or17B in FIGS.5A,B) and the reference voltage conductor (GND), wherein VBE2is the base-emitter voltage of the second transistor (Q1), a third resistor (R4inFIG. 4Bor R5inFIG. 4A) across which the reversed bandgap reference voltage VRBGPis produced being coupled between a base of the second transistor (Q2) and the reference voltage conductor (GND), means for forcing the collector current (IC1) of the first transistor (Q1) to be equal to the collector current (IC2) of the second transistor (Q2), and means for forcing the first voltage VBE1(1+1/M) to be equal to the second voltage VBE2+VRBGP.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

The invention provides a number of reversed bandgap reference voltage circuits which provide low, stable reference voltages that are proportional to absolute temperature and have lower noise than the prior art, are more accurate than the prior art, and are capable of operation from a supply voltage less than 1 volt.

The traditional bandgap reference output voltage is equal to
VBGP=VBE+M*VPTAT=1.2 volts,  Eq. (1)
where M is the gain coefficient of the proportional-to-absolute-temperature voltage VPTAT. This voltage is generated from the difference between the base-emitter voltages (VBEvoltages) of two bipolar transistors having different current densities.

FIG. 3shows a reversed bandgap reference voltage circuit10-1, which is a substantial modification of a circuit shown inFIG. 4in the above-mentioned Widlar reference. The circuit inFIG. 3includes an NPN transistor Q1having its emitter connected to ground. A resistor R1has one terminal connected to the collector of transistor Q1and one terminal of a current source I0and has another terminal connected to the base of transistor Q1and one terminal of a resistor R2. The other terminal of resistor R2is connected to ground. A second NPN transistor Q2has its base and collector connected to the collector of transistor Q1. The emitter of transistor Q2is connected to one terminal of a resistor R3, the other terminal of which is connected to ground. The reversed bandgap output voltage VRBGPis provided on the conductor connecting the emitter of transistor Q2to resistor R3.

A problem with the circuit10-1inFIG. 3is that transistor Q1operates in saturation, and therefore causes significant inaccuracy in VRBGP. The reversed bandgap voltage reference circuit ofFIG. 3as illustrated is impractical because the current source I0operates in saturation and therefore is not very accurate. Reversed bandgap reference voltage circuit10-1requires a supply voltage of at least 1 volt for operation. Also, the gain characteristic of reversed bandgap circuit10-1is so complicated that it is unstable with almost any load.

The reversed bandgap output voltage VRBGPgenerated by the reversed bandgap reference voltage circuit ofFIG. 3is obtained by dividing both sides of Equation (1) by the VPTATgain coefficient M. This results in the expression
VRBGP=VBGP/M=VPTAT+VBE/M˜(1.2 volts)÷M˜200 millivolts.  Eq. (2)
Neglecting base currents, the reversed bandgap reference voltage VRBGPis given by the expression

VRBGP=⁢VBE⁢⁢1⁢{(R⁢⁢1+R⁢⁢2)/R⁢⁢2}-VBE⁢⁢2=⁢VBE⁢⁢1⁡(R⁢⁢1/R⁢⁢2)+VT⁢ln⁡(N)~200⁢⁢mV,Eq.⁢(3)
wherein N is the ratio of current density in transistor Q2to the current density in transistor Q1, and wherein VTis the thermal voltage of silicon. The reversed bandgap reference voltage circuit10-1ofFIG. 3requires careful selection of the resistor ratios and bias current I0. Its minimum supply voltage VDDis
VDD(min)=VRBGP+VBE2+Vsat  Eq. (4)
where Vsat is the voltage across the current source I0and can be as small as 10-50 millivolts for a PMOS implementation. The minimum supply voltage VDD(min) is 0.85 volts at room temperature, and increases to approximately 1 volt at −40 degrees Centigrade.

The present invention provides various reversed bandgap voltage reference circuit structures wherein all of the important operational circuit parameters are controlled by dedicated feedback loops. In accordance with the reversed bandgap reference voltage circuits of the present invention, a pair of important circuit conditions to be fulfilled are given by the expressions
IC1=IC2, and  Eq. (5A)
VBE1(1+1/M)=VBE2+VRBGP.  Eq. (5B)

In the reversed bandgap reference voltage circuit10-1ofFIG. 3, the condition of Equation (5B) is satisfied simply by connecting the collector of transistor Q1to the base of transistor Q2. Therefore, the current source I0must be designed in such a way that it causes the collector voltage of transistor Q1to be equal to the base voltage of transistor Q2. However, the ratio IC1/IC2of the collector currents of transistors Q1and Q2depends on the values of the current produced by current source I0and the absolute values of resistors R1, R2and R3.

In accordance with the present invention, this reliance on absolute component values for the circuit ofFIG. 3is eliminated by means of the feedback circuitry disclosed in the various reversed bandgap voltage reference circuits shown in the following drawings.

Reversed bandgap reference voltage circuits10-2and10-3shown inFIGS. 4A and 4B, respectively, each use a single feedback loop to control the ratio IC1/IC2. InFIG. 4A, reversed bandgap reference voltage circuit10-2includes NPN transistor Q1having its emitter connected to ground and its base connected by conductor32A to one terminal of each of resistors R1and R2. (Note that the resistance values of the various resistors in the drawings are also used as their reference characters.) The other terminal of resistor R2is connected to ground, and the other terminal of resistor R1is connected by conductor17to one terminal of resistor R3and one terminal of resistor R4. The other terminal of resistor R3is connected by conductor16A to the collector of transistor Q1and to the non-inverting input of operational amplifier12. The other terminal of resistor R4is connected by conductor16B to the inverting input of operational amplifier12and to the collector of transistor Q2. The base of transistor Q2is also connected to conductor17. The emitter area of transistor Q2is N times that of transistor Q1. The output of operational amplifier12is connected to the gate of P-channel transistor M1, the source of which is connected to VDDand the drain of which is connected to conductor17.

InFIG. 4A, the current through transistor Q1is equal to VRBGP/R4. The absolute value of R4should be chosen to avoid deep saturation of transistor Q2. The absolute value of the current through transistor Q2is independent of temperature and is equal to IC2=VRBGP/R5.

The configuration of reversed bandgap voltage reference circuit10-3ofFIG. 4Bis somewhat similar to that ofFIG. 4A. InFIG. 4B, PNP transistor Q1has its emitter connected to conductor17and its base connected by conductor32A to one terminal of each of resistors R1and R2. The other terminal of resistor R2is connected to conductor17, and the other terminal of resistor R1is connected to ground. The collector of transistor Q1is connected by conductor16A to one terminal of resistor R3and to the inverting input of operational amplifier12. The other terminal of resistor R3is connected to ground. One terminal of resistor R4is connected by conductor16B to the non-inverting input of operational amplifier12and also to the base and collector of PNP transistor Q2. The emitter of transistor Q2is connected by conductor17to the emitter of transistor Q1and to the drain of P-channel transistor M1, the source of which is connected to VDD. The emitter area of transistor Q2is N times that of transistor Q1. The output of operational amplifier12is connected to the gate of transistor M1.

In the reversed bandgap reference voltage circuits ofFIGS. 4A and 4Bthe condition Equation (5B), that is, VBE1(1+1/M)=VBE2+VRBGP, is met by the direct connections of conductor17as shown. Specifically, Equation (5B) is met by the direct connection of conductor17to the upper terminals of resistors R1and R3and to the base of transistor Q2inFIG. 4A, and by the direct connection by conductor17to the upper terminal of resistor R2and to the emitters of transistors Q1and Q2inFIG. 4B. InFIG. 4A, the condition of Equation (5A), i.e., IC1=IC2, is met because operational amplifier12forces the collector currents to be equal by causing the voltages across, and hence the currents through, matched resistors R3and R4to be equal. Similarly, inFIG. 4Bthe condition of Equation (5A), i.e., IC1=IC2, is met because feedback amplifier12forces the two collector currents to be equal because the resistances R3and R4are equal. (The PTAT gain coefficient M is equal to R2/R1forFIG. 4Aand is equal to R1/R2forFIG. 4B.)

InFIGS. 4A and 4B, and similarly for the subsequently described bandgap reference voltage circuits of the present invention, the feedback loops create a second stable operating point when all currents in the circuit are equal to zero and therefore require a conventional start-up circuit.FIGS. 8 and 10show such start-up circuits.

Two other reversed bandgap reference voltage circuits are shown inFIGS. 5A and 5B, wherein the requirement of Equation (5A) is met by using PMOS transistors M1and M2as matched current sources to establish the collector currents IC1=IC2, and feedback loops including an operational amplifier12maintains the condition VBE1(1+1/M)=VBE2+VRBGPof Equation (5B).

InFIG. 5A, reversed bandgap voltage reference circuit10-4includes NPN transistor Q1having its collector connected to ground, its base connected by conductor32A to one terminal of resistor R1and resistor R2, and its collector connected by conductor17A to the other terminal of resistor R2, the drain of P-channel transistor M1, and the inverting input of operational amplifier12. The emitter of NPN transistor Q2is connected by conductor17B to the non-inverting input of operational amplifier12, the drain of P-channel transistor M2, and to one terminal of resistor R2A. The other terminal of resistor R2A is coupled to ground by resistor R1A. The emitter area of transistor Q2is N times that of transistor Q1. The output of operational amplifier12is connected by conductor19to the gates of transistors M1and M2, the sources of which are connected to VDD.

The configuration of reversed bandgap voltage reference circuit10-5inFIG. 5Bis somewhat similar. The emitter of NPN transistor Q1is connected to ground, its base is connected by conductor32A to a first terminal of each of resistors R1and R2, and its collector is connected by conductor17A to the other terminal of resistor R1, the drain of P-channel transistor M1, and the inverting input of operational amplifier12. The other terminal of resistor R2is connected to ground. The emitter and base of NPN transistor Q2are connected by conductor17B to the inverting input of operational amplifier12, the drain of P-channel transistor M2, and to one terminal of resistor R1A, the other terminal of which is coupled by resistor R2A to ground. The emitter of transistor Q2is connected by conductor32B is to one terminal of resistor R4, the other terminal of which is connected to ground. The emitter area of transistor Q2is N times that of transistor Q1. The reversed bandgap voltage VRBGPis produced on conductor32B. The output of operational amplifier19is connected by conductor19to the gates of transistors M1and M2, the sources of which are connected to VDD.

In the reversed bandgap reference voltage circuits of bothFIGS. 5A and 5B, the requirement of Equation (5B) that VBE1(1+1/M)=VBE2+VRBGPis met because feedback amplifier12forces the voltages on conductors17A and17B to be equal.

InFIGS. 5A and 5B, if R1=R1A, R2=R2A, and M1=M2, then operational amplifier12forces the current produced by transistors M1and M2to deliver equal currents into the conductors17A and17B. Equal portions of that current therefore flow in the circuit paths R2,R1and R2A,R1A and therefore IC1=IC2, and VRBGPthen is defined by Equation (3).

Note that reversed bandgap reference voltage circuits inFIGS. 4A and 4BandFIGS. 5A and 5Beach have only one feedback loop to control either collector currents or voltages to satisfy Equations (5A) and (5B). However, by adding a second feedback loop, both of the above-mentioned circuit operation conditions in Equations (5A) and (5B) can be controlled by feedback. This allows, for example, scaling of the output voltage to a desired value, as shown inFIG. 6. Also, the second feedback loop can establish different TCs for the transistor collector currents to allow VBEcurvature compensation.

InFIG. 6, reversed bandgap reference voltage circuit10-6includes PNP transistor Q1, which has its collector connected to ground, its base connected by conductor32A to one terminal of each of resistors R1and R2, and its collector connected by conductor17A to the other terminal of resistor R2, the source of P-channel transistor M1, and the non-inverting input of operational amplifier15. The other terminal of resistor R1is connected to ground. NPN transistor Q2has its collector connected to ground, its base connected by conductor34B to one terminal of each of resistors R4and R5, and its emitter connected by conductor17B to the inverting input of operational amplifier15, to the source of P-channel transistor M2, and to one terminal of resistor R2A. The emitter area of transistor Q2is N times that of transistor Q1. The other terminal of resistor R4is connected to ground. The other terminal of resistor R2A is coupled by resistor R1A to ground. The other terminal of resistor R5is connected to the output of operational amplifier15. The reversed bandgap voltage VRBGPis produced on conductor34B.

A scaled-up reference voltage Vref is produced from VRBGPby the output of operational amplifier15. Conductor34B also is connected to the inverting input of operational amplifier12, the output of which is connected by conductor19to the gates of P-channel transistors M1, M2, and M3, the sources of which are connected to VDD. The drain of transistor M3is connected by conductor34A to the non-inverting input of operational amplifier12and to one terminal of resistor R6, the other terminal of which is connected to ground.

In the reversed bandgap reference voltage circuit ofFIG. 6, amplifier15controls the base voltage of transistor Q2in order to satisfy Equation (5). Bias currents are regulated by amplifier12using the temperature-independent voltage VRBGPas a reference. The output voltage Vref can be scaled to a required value given by the expression
Vref=VRBGP(R4+R5)/R4.  Eq. (6)

The resistive voltage divider including resistors R5and R4controls the base voltage of transistor Q2, and operational amplifier15forces the voltages on conductors17A and17B to be equal, thereby meeting the requirement VBE1(1+1/M)=VBE2+VRBGPof Equation (5B). The requirement IC1=IC2of Equation (5A) is met by designing transistors M1, M2and M3to be perfectly matched so as to deliver equal collector currents, and also by designing the resistances of resistors R4and R6to be precisely equal. Operational amplifier12establishes the value of the equal currents in matched transistors M1, M2and M3to be equal to the currents forced to flow through matched resistors R6and R4. This occurs as a result of feedback amplifier12forcing the voltages on conductors34and34B to be equal.

If both of the previously described circuit operation conditions, rather than just one of them, are controlled using feedback amplifiers, that provides a way of scaling up the reversed bandgap voltage VRBGPto obtain the output reference voltage Vref. This is advantageous, because the reversed bandgap voltage reference circuits shown inFIGS. 4A and 4BandFIGS. 5Aand B produce an output voltage VRBGPwhich is in the range of only 90 to 100 millivolts, whereas reversed bandgap voltage reference circuit10-6ofFIG. 6produces a value of Vref of up to 600-900 millivolts. Another advantage of reversed bandgap voltage reference circuit10-6is that transistors Q1and Q2can be substrate transistors having their collectors formed in the semiconductor wafer substrate, and therefore can be readily implemented in any CMOS process.

Another variant of the dual-feedback reversed bandgap reference is shown inFIG. 7, wherein reversed bandgap reference voltage circuit10-7includes PNP transistor Q1which has its base connected by conductor35to one terminal of each of resistors R1and R2and to the non-inverting input of operational amplifier12, its emitter connected by conductor17to the emitter of NPN transistor Q2, the source of P-channel transistor M1, and the other terminal of resistor R2. The collector of transistor Q1is connected by conductor38to the inverting input of operational amplifier12, the input of operational amplifier15, and one terminal of resistor R6. The other terminal of resistor R1is connected to ground, and the other terminal of resistor R6also is connected to ground. The base of transistor Q2is connected by conductor36to one terminal of each of resistors R4and R5. The collector of transistor Q2is connected by conductor39to one terminal of resistor R7and to the non-inverting input of operational amplifier15. The other terminal of resistor R5is connected to the output of operational amplifier15, which produces a scaled-up reference voltage Vref. The other terminal of resistor R4is connected to ground, as is the other terminal of resistor R7. The output of operational amplifier12is connected by conductor19to the gate of transistor M1, the collector of which is connected to VDD. The emitter area of transistor Q2is N times larger than the emitter area of transistor Q1. The output voltage of the reference inFIG. 7is defined by Equation (6).

InFIG. 7, the requirement VBE1(1+1/M)=VBE2+VRBGPof Equation (5B) is met by the direct connection of conductor17to the emitter of transistor Q2and to the collector of transistor Q1. The requirement IC1=IC2of Equation (5A) is met by the operation of feedback amplifier12to control the current through transistor Q1, thereby causing the voltages on conductors35and38across equal resistors R1and R6, respectively, to be equal, and also by the operation of feedback amplifier15to cause the collector currents of transistors Q1and Q2to be equal by causing the voltages on conductors38and39across matched resistors R6and R7, respectively, to be equal.

FIG. 8shows a practical implementation of the same basic structure as the one shown inFIG. 4B, with amplifier12being implemented as shown inFIG. 9by an input stage including transistors Q3and Q4and a folded cascode stage including transistors M5, M6, M7, and M8. P-channel transistor M3inFIG. 8is used to implement the tail current source I1inFIG. 9.

FIG. 10shows a practical implementation of the basic circuit shown inFIG. 7, further including a conventional start-up circuit, which is formed by P-channel junction field effect transistor J3and transistors M11and M12. The start-up circuit is off during normal circuit operation. The capacitors C1, C2, and C3are frequency compensation capacitors. Amplifiers A1and A2can be the same as amplifier12shown inFIG. 9.

The main advantage of the above described invention, especially the embodiments ofFIG. 4AthroughFIG. 10, over the prior art is better accuracy than the prior art bandgap circuits. For example, in Prior ArtFIG. 1, the current mirror circuitry generates a great deal of noise, which is avoided by the embodiments ofFIG. 4AthroughFIG. 10because the current mirror circuitry is outside of the feedback loops. In Prior artFIG. 2, transistor Q1is in saturation, resulting in low current gain and consequently in low accuracy which is avoided by the embodiments ofFIG. 4AthroughFIG. 10because none of the bipolar transistors are in or near saturation.

Thus, the described invention provides a set of circuit implementations of reversed bandgap reference voltage circuits which produce low output voltage (˜200 millivolts) and which are capable of operating from a supply voltage of less than 1 volt, and which have accuracy and noise parameters comparable with conventional 1.2 volt bandgap voltage references. One of the low voltage reversed bandgap voltage reference circuits, disclosed inFIG. 7, can be implemented using substrate transistors.

While the invention has been described with reference to several particular embodiments thereof, those skilled in the art will be able to make various modifications to the described embodiments of the invention without departing from its true spirit and scope. It is intended that all elements or steps which are insubstantially different from those recited in the claims but perform substantially the same functions, respectively, in substantially the same way to achieve the same result as what is claimed are within the scope of the invention. For example, the reversed bandgap reference voltage circuits inFIGS. 4A,4B andFIG. 7use resistors for current sensing. This also can be accomplished by other means, such as using MOS current mirrors with low-threshold transistors.