Beam forming with double-null-steering for in-band on-channel reception

Various exemplary embodiments relate to a method for improving reception of transmissions with first adjacent interference signals, the method including selecting one or more time samples from each of two or more antennas; generating a lower first adjacent interference (LFAI) signal, a desired signal, and an upper first adjacent interference (UFAI) signal for each of the time samples; calculating a lower weighting co-efficient based on the LFAI signal; calculating a middle weighting co-efficient based on the desired signal; calculating a upper weighting co-efficient based on the UFAI signal; combining the lower weighting co-efficient with a filtered LFAI signal into a weighted lower signal; combining the middle weighting co-efficient with a filtered desired signal into a weighted middle signal; combining the upper weighting co-efficient with a filtered UFAI signal into a weighted upper signal; and combining the weighted lower signal, the weighted middle signal, and the weighted upper signal.

TECHNICAL FIELD

The “In-band on-channel” (IBOC) radio transmission system is generally used to transmit digital radio and analog radio broadcast signals simultaneously on the same frequency. There is also an all-digital version in which two digital signals are combined.

BACKGROUND

The IBOC system multiplexes digital information by utilizing additional digital subcarriers or sidebands, thus avoiding re-allocation of broadcast bands. However, by putting RF energy outside of the normally-defined channel, interference is increased when using digital sidebands.

SUMMARY

Various exemplary embodiments relate to a method for improving reception of transmissions with first adjacent interference signals, the method including selecting one or more time samples from each of two or more antennas; generating a lower first adjacent interference (LFAI) signal, a desired signal, and an upper first adjacent interference (UFAI) signal for each of the time samples; calculating a lower weighting co-efficient based on the LFAI signal; calculating a middle weighting co-efficient based on the desired signal; calculating a upper weighting co-efficient based on the UFAI signal; combining the lower weighting co-efficient with a filtered LFAI signal into a weighted lower signal; combining the middle weighting co-efficient with a filtered desired signal into a weighted middle signal; combining the upper weighting co-efficient with a filtered UFAI signal into a weighted lower signal; and combining the weighted lower signal, the weighted middle signal, and the weighted upper signal.

In some embodiments of the method, calculating the lower weighting co-efficient based on the LFAI signal further includes shifting the LFAI signal to zero; and calculating the upper weighting co-efficient based on the UFAI signal further includes shifting the UFAI signal to zero. In some embodiments of the method, calculating the lower weighting co-efficient based on the LFAI signal further includes filtering the LFAI signal to include the lowest half of the LFAI signal; calculating a middle weighting co-efficient based on the desired signal further includes filtering the desired signal to include the middle portion of the desired signal which includes half the desired signal; and calculating the upper weighting co-efficient based on the UFAI signal further includes filtering the UFAI signal to include the upper-most half of the LFAI signal.

In some embodiments of the method, calculating the lower weighting co-efficient based on the LFAI signal further includes generating an inverse co-variance matrix based on the LFAI signal; calculating a middle weighting co-efficient based on the desired signal further includes generating an inverse co-variance matrix based on the desired signal; and calculating the upper weighting co-efficient based on the UFAI signal further includes generating an inverse co-variance matrix based on the UFAI signal. In some embodiments of the method, calculating the lower weighting co-efficient based on the LFAI signal further includes calculating a lower weighting co-efficient that maximizes the Signal-to-Interference-plus-Noise-Ratio (SINR) of the LFAI signal; calculating a middle weighting co-efficient based on the desired signal further includes calculating a middle weighting co-efficient that maximizes the SINR of the desired signal; and calculating the upper weighting co-efficient based on the UFAI signal further includes calculating an upper weighting co-efficient that maximizes the SINR of the UFAI signal.

Alternative embodiments of the method further include generating a filtered LFAI signal; generating a filtered desired signal; and generating a filtered UFAI signal. In some embodiments of the method, generating the filtered LFAI signal further includes shifting the LFAI signal to zero; and generating the filtered UFAI signal further includes shifting the UFAI signal to zero. In some embodiments of the method, generating the filtered LFAI signal further includes filtering the LFAI signal to include a lower digital sideband; generating the filtered desired signal further includes filtering the desired signal to include an analog band; and generating the filtered UFAI signal further includes filtering the UFAI signal to include an upper digital sideband.

Various exemplary embodiments relate to a device for improving reception of transmissions with first adjacent interference signals, the device including an antenna array including two or more antennas; a radio front-end block including one or more radio front ends connected to each of the two or more antennas; one or more analog-to-digital converters connected to the one or more radio front-ends; one or more baseband blocks connected to the one or more analog-to-digital converters; and a digital adaptive beam-former block connected to each of the one or more baseband blocks. In some embodiments, the digital adaptive beam-former block further includes a training block connected to each of the one or more baseband blocks; a coefficient-update block connected to the training block; one or more finite impulse response (FIR) filter blocks connected to the coefficient-update block and each of the one or more baseband blocks; and a combiner block connected to each of the one or more FIR filter blocks.

In some embodiments, the one or more baseband blocks is configured to select one or more time samples from each of the two or more antennas; the training block is configured to generate a lower first adjacent interference (LFAI) signal for each of the time samples, generate a desired signal for each of the time samples, and generate an upper first adjacent interference (UFAI) signal for each of the time samples; and the coefficient-update block is configured to calculate a lower weighting co-efficient based on the LFAI signal, calculate a middle weighting co-efficient based on the desired signal, and calculate a upper weighting co-efficient based on the UFAI signal. In some embodiments, the coefficient-update block is further configured to generate an inverse co-variance matrix based on the LFAI signal, generate an inverse co-variance matrix based on the desired signal, and generate an inverse co-variance matrix based on the UFAI signal. In some embodiments, the coefficient-update block is further configured to calculate a lower weighting co-efficient that maximizes the Signal-to-Interference-plus-Noise-Ratio (SINR) of the LFAI signal, calculate a middle weighting co-efficient that maximizes the SINR of the desired signal, and calculate an upper weighting co-efficient that maximizes the SINR of the UFAI signal.

In some embodiments, the one or more finite impulse response (FIR) filter blocks is configured to generate a filtered lower first adjacent interference (LFAI) signal, generate a filtered desired signal, and generate a filtered upper first adjacent interference (UFAI) signal. In some embodiments, the one or more finite impulse response (FIR) filter blocks is further configured to shift the LFAI signal to zero, and shift the UFAI signal to zero. In some embodiments, the one or more finite impulse response (FIR) filter blocks is further configured to filter the LFAI signal to include a lower digital sideband, filter the desired signal to include an analog band, and filter the UFAI signal to include an upper digital sideband. In some embodiments, the one or more finite impulse response (FIR) filter blocks is further configured to combine a lower weighting co-efficient with the filtered LFAI signal into a weighted lower signal, combine a middle weighting co-efficient with the filtered desired signal into a weighted middle signal, and combine the upper weighting co-efficient with the filtered UFAI signal into a weighted lower signal.

In some embodiments, the one or more baseband blocks is configured to select one or more time samples from each of the two or more antennas; the training block is configured to generate a lower first adjacent interference (LFAI) signal for each of the time samples, generate a desired signal for each of the time samples, and generate an upper first adjacent interference (UFAI) signal for each of the time samples; the coefficient-update block is configured to calculate a lower weighting co-efficient based on the LFAI signal, calculate a middle weighting co-efficient based on the desired signal, and calculate a upper weighting co-efficient based on the UFAI signal; the one or more finite impulse response (FIR) filter blocks is configured to generate a filtered LFAI signal, generate a filtered desired signal, generate a filtered UFAI signal, combine the lower weighting co-efficient with the filtered LFAI signal into a weighted lower signal, combine the middle weighting co-efficient with the filtered desired signal into a weighted middle signal, combine the upper weighting co-efficient with the filtered UFAI signal into a weighted lower signal; and the combiner block is configured to combine the weighted lower signal, the weighted middle signal, and the weighted upper signal.

In some embodiments, the one or more baseband blocks configured to select one or more time samples from each of the two or more antennas; the digital adaptive beam-former block configured to generate a lower first adjacent interference (LFAI) signal, a desired signal, and an upper first adjacent interference (UFAI) signal for each of the time samples; calculate a lower weighting co-efficient based on the LFAI signal; calculate a middle weighting co-efficient based on the desired signal; calculate a upper weighting co-efficient based on the UFAI signal; combine the lower weighting co-efficient with a filtered LFAI signal into a weighted lower signal; combine the middle weighting co-efficient with a filtered desired signal into a weighted middle signal; combine the upper weighting co-efficient with a filtered UFAI signal into a weighted lower signal; and combine the weighted lower signal, weighted middle signal, and weighted lower signal.

It should be apparent that, in this manner, various exemplary embodiments enable improved reception of in-band on-channel transmissions. In particular, by training a signal processor using the lowest and highest sub-bands.

DETAILED DESCRIPTION

The description and drawings presented herein illustrate various principles. It will be appreciated that those skilled in the art will be able to devise various arrangements that, although not explicitly described or shown herein, embody these principles and are included within the scope of this disclosure. As used herein, the term, “or,” as used herein, refers to a non-exclusive or (i.e., and/or), unless otherwise indicated (e.g., “or else” or “or in the alternative”). Additionally, the various embodiments described herein are not necessarily mutually exclusive and may be combined to produce additional embodiments that incorporate the principles described herein. Further, while various exemplary embodiments are described with regard to IBOC signal processing, it will be understood that the techniques and arrangements described herein may be implemented to facilitate reception of near-band transmissions in other types of systems that implement multiple types of data processing or data structures.

The “In-band on-channel” (IBOC) radio transmission system is used to transmit digital radio and analog radio broadcast signals simultaneously on the same frequency; in an all-digital version, two digital signals are combined—thus an IBOC signal may be in hybrid or all-digital form. The in-band on-channel digital radio broadcasting standard for the FM-band is defined by the FM-part of the NRSC-5 standard “National Radio Systems Committee (NRSC) NRSC-5-C, “In-band/on-channel Digital Radio Broadcasting Standard”, September, 2011”. This standard is the basis for the transmitted IBOC-signals of the digital radio corporation iBiquity™ that can be received by an HDradio™ certified receiver.

The well-known Maximal Ratio Combining (MRC) may be used to improve reception of IBOC signals within a multiple-antennas scenario, where the received-power of each antenna is controlled in such a way that it is constructively (coherently) added. However, due to the fact that MRC is, by definition, only capable of combining the power that is collected by each antenna it: i) has limited performance due to the fact that a receiving-antenna receives stronger electromagnetic waves in some directions than in others, i.e., it is a non-ideal omni-directional antenna, ii) cannot control suppression of undesired-signals, and iii) is costly since for each antenna requires a complete detection path before signal combinations can take place.

An IBOC signal includes an OFDM-signal including a central-part centered on a carrier frequency, a lower sideband below the lowest frequency of the central-part and an upper-sideband above the highest frequency of the central-part. Thus, the desired signals and the interference will be in three different frequency bands—a lower frequency interference band, a desired signal, and a higher interference frequency band.

Focusing on the location of the desired signal while suppressing the locations of origination of interference signals may allow filtering of undesired interference. As described below, an electronically steered complex base-band beam forming approach may be able to remove or significantly reduce adjacent interference signals by double-null-steering with a Uniform-Linear-Antenna-array (ULA) consisting of two isotropic antennas that are spaced by half a wave-length (λ/2) of a carrier frequency, e.g., fc=100 MHz→λ≈3 m for transmissions within the FM-band. Double-null-steering exploits the fact that both the first adjacent interference signals as well as the desired IBOC-signal are separated in three different frequency-bins; the desired and interference signals are orthogonal.

For complex base-band beam forming with double-null-steering, a portion of a frequency-bin may be used to obtain representative training-signals or surrogate-signals of the desired-IBOC-signal, the lower first adjacent interference signal, and the upper first adjacent interference signal. These surrogate-signals may be used to compute optimal steering-weights based on maximization of the Signal-to-Interference-plus-Noise-Ratio (SINR). As such, where there is an interference signal the beam former may distinguish the desired signal and the interference signal using the known properties of the signals. FM signals are symmetric, so the lowest part of the lower interference signal and the highest part of the higher interference signal may be used as a training signal for the beam former/canceller; at the closest first adjacent frequency portion the interference signal may be derived and subtracted from the desired signal. Furthermore, the strongest portion of the desired signal may be used as a training signal to derive the direction (point of origin) of the desired signal to prevent self-nulling. As a result, the reception of IBOC signals may be improved by using the properties of the signals received by two antennas by reducing the effects of interference signals detected by the antennas.

The multiple antennas may form a Uniform-Linear-Antenna-array (ULA) including two isotropic antennas, with electronically steered antenna radiation-patterns. Instead of MRC, the reception of transmitted IBOC signals may be improved by steering using complex base-band signals. The electronic steering may be based on estimated phase-shifts and estimated amplitudes of multiple received signals on the ULA. Steering the radiation-pattern using complex base-band signals, phase-estimations, and amplitude-estimations may be called “electronically-steered complex base-band beam forming.” Such beam forming may allow “null-steering” to suppress or remove interference signals, for example, to suppress possible 1st-adjacent interference signals for an IBOC-transmission in either the hybrid-mode or the all-digital mode. In order to improve users' experience, for example, to minimize interruption of reception or delay when seeking a desired signal, beam-forming may be conducted as quickly as possible, with the lowest latency and least complexity. Thus two antennas may be used to minimize the expense and added complexity of adding additional antennas.

Either or both of the upper adjacent and the lower adjacent signals may interfere with the digital side band of the IBOC signal, causing co-channel interference for the desired signal. Further, the neighboring signals may have more power than the desired signal, for example an interference signal may have 100 times more power than a desired signal. However, beam forming enables co-channel-interference cancellation (CIC) since the 1st adjacent interference signals which are co-channel interference signals for the digitally-modulated side-bands of an IBOC-transmission may originate from different directions. The lowest-frequency part of the IBOC signal, the central-frequency part of the IBOC signal and the upper-most frequency part of the IBOC-signal may all be used to perform low-complexity, fast beam forming with double-null-suppression. In hybrid-mode the analog central-part of the HIBOC-signal may be used and in the all-digital mode the digital central-part of the IBOC-signal may be used. Note, that phase and amplitude-estimations may be used to apply main-lobe steering and null-steering for improved reception of the digitally-modulated side-bands without actually demodulating the orthogonal frequency division multiplexing (OFDM)-signals, as shown below. Complex base-band beam forming may use specific frequency-bins to obtain training-signals or surrogate-signals in order to calculate beam forming weights for each frequency-bin. Furthermore, the computation of each weight may be based on maximizing the SINR of the desired signal. Frequency-bin based complex base-band beam-forming with maximization of the SINR results in improved reception of the desired signal.

FIG. 1illustrates an exemplary hybrid signal100in simplified form and down-converted from a carrier frequency. A beam former may perform constructive or destructive addition of the signal power for 600 kHz around a baseband (0 kHz), in other words, from −300 kHz to +300 kHz. A hybrid signal100may be a combination of an analog signal102and a digitally modulated signal104and106. The analog FM-signal102may occupy a bandwidth of 200 kHz, i.e., between −100 kHz and +100 kHz spanning the carrier frequency. The digitally-modulated signal occupies a bandwidth of approximately 200 kHz. However, the digitally-modulated signal is separated into a lower-sideband104and an upper-sideband106each with a bandwidth of approximately 100 kHz. The lower-sideband104may be spectrally positioned at a distance of 100 kHz below the carrier-frequency. The upper side-band106may be spectrally positioned at a distance of 100 kHz above the carrier-frequency.

The total power of the digitally-modulated signal104,106may be approximately a factor of hundred smaller than the power of the analog host-FM-signal102. The digitally-modulated signal may use OFDM, where the number of subcarriers may vary depending on the selected service/transmission-mode. The “channel-grid” channel-bandwidth reserved by regulation for an analog FM-signal is 200 kHz. As a consequence, the lower and upper digital OFDM-sidebands104,106transmit in the 1st-adjacent lower and upper neighboring FM-channels.

An IBOC signal may also be implemented as an all-digital signal, where the analog FM-signal is replaced by a secondary digitally-modulated signal. In the all-digital mode the bandwidth of the primary digital sidebands may be fully expanded with lower-power secondary sidebands. The below description applies to both implementations, with a few adjustments. An all-digital IBOC signal has a bandwidth of roughly 400 kHz, where also in the all-digital mode approximately 100 kHz of the lower and upper adjacent channels is occupied (outside the 200 kHz “channel-grid”).

As discussed above, a 1st-adjacent signal of an IBOC transmission introduces co-channel interference for the digitally modulated lower-sideband104and upper-sideband106of a signal100. By regulation the co-channel interference can be up to a factor of hundred stronger in power than the digitally-modulated lower-sideband104and upper-sideband106. Either or both of the 1st-adjacent interference signals may be present at a time, in which case the lower-sideband104and upper-sideband106are distorted by a neighbor transmission.

FIG. 2Aillustrates an exemplary power spectral density (PSD) of a desired signal202, with analog band204, lower digital sideband206, and upper digital sideband208;FIG. 2Billustrates an exemplary PSD of a lower-sideband 1st-adjacent interference signal210;FIG. 2Cillustrates an exemplary PSD of an upper-sideband 1st-adjacent interference signal212; andFIG. 2Dillustrates a stylized exemplary PSD of an original exemplary PSD as it may be recorded, including the desired signal202(including analog band204, lower digital sideband206, and upper digital sideband208), lower-sideband 1st-adjacent interference signal210, and upper-sideband 1st-adjacent interference signal212. It would be understood by one of skill in the art thatFIGS. 2A, 2B, and 2Cmay represent the component parts of a signal that may be recorded, such that the average of the signal is represented byFIG. 2D. Thus, the hybrid IBOC signal202may be plotted as a noisy FM-signal. As illustrated inFIG. 3, the lower-sideband204and upper-sideband206of the IBOC-transmission202are heavily distorted, respectively, by the lower 1st-adjacent neighbor transmission210and the upper 1st-adjacent neighbor transmission212. However, the IBOC-signal and both of the 1st-adjacent interference signals may originate from different geographic locations, thus the space-domain may be exploited by a beam forming approach to remove or reduce the 1st-adjacent interference signals.

FIG. 3illustrates an exemplary input signal300for a beam forming process. Signal300may be the summation of a desired IBOC transmission and two 1st-adjacent transmissions after down-sampling. Electronically-steered complex base-band beam forming may be used to separate the three different signals by their different spatial information, which may be referred to as each signal's “spatial-signature”. The base-band300of the received radio signal may occupy a bandwidth of approximately 600 kHz, i.e., between −300 kHz and +300 kHz. This received-signal may include the desired IBOC signal, occupying approximately 400 kHz between −200 kHz and +200 kHz, which may correspond to desired signal202. Although two 1st adjacent interference signals are shown, the received-signal or input-signal300may include zero, one or two 1st adjacent interference signals. The 1st adjacent interference signals of signal300each occupy approximately 200 kHz, where the lower 1st adjacent interference-signal may be between −300 kHz and −100 kHz, which may correspond to lower-sideband 1st-adjacent interference signal210, and the upper 1st adjacent interference-signal may be between +100 kHz and +300 kHz, which may correspond to upper-sideband 1st-adjacent interference signal212. A 1st adjacent interference signal occupies the same frequencies as the lower-sideband and upper-sideband and thus is for the digitally-modulated lower-sideband and upper-sideband of the IBOC-signal a co-channel interference-signal.

A frequency bin that is a portion of input frequency300of approximately 100 kHz may be used to obtain representative training-signals for the desired IBOC signal and the lower- and upper-1stadjacent interference signals.

FIG. 4illustrates an exemplary method400for beam forming with double-null-steering. The method begins402where a sample frequency base band, for example, 650 kHz, may be selected and down-converted such that all of the information from a frequency domain/time domain of −325 kHz to +325 kHz may be captured. At step404A number N of data samples may be collected. Note that although signals202,210,212, and signal300may be illustrative of a sample frequency, the N-data samples are time samples, rather than frequency samples. N may be a number sufficient that the training signals are effectively representative of the interference and desired signals.

For example, 2000 samples may be used to make a training signal for a frequency of 650 kHz. A person of skill in the art will understand that the length of each sample will depend on the sample frequency—for instance, in the case of 650 KHz, 1.65 micro-seconds (μs), such that 2000 samples for 650 kHz may result in, at the most, latency on the order of 3 milli-seconds (ms). Note that analog-to-digital conversion of the received signals will occur for the IBOC-rendering process; thus, no additional processing other than for writing the samples will be required to render samples as the samples are merely the digital representation of the received signal repeatedly processed (divided) into N time-samples. Thus, the sampling may be constant, so that new samples are continually available for the beam forming process. Thus, step404may repeat many times before moving to steps in paths406to414as discussed below. As such, the most recent N-data samples may be maintained, such that older samples may be discarded as new samples are obtained. In some embodiments, rather than constantly being updated, samples may be updated with a frequency dependent on the speed of movement of the antenna array or the interference level of the desired signal.

Further, in part due to the added complexity and processing power required, a new range for the beam former may not be calculated for every updated sample that is collected—thus, the samples may be collected more frequently than the range is updated so that the samples are available as needed when it is determined that it is time that a new range be calculated. As noted above, if no samples have been collected when a range is to be calculated (e.g. when first tuning to the frequency), the latency will be on the order of the time required to collect N samples, which may be expressed as, if recalculating at time t, the samples will be taken from t−(N*lengthsample), and the beam former will not continue until N samples have been collected.

Also, in one embodiment note that the intervals between recalculation of the beam former may depend on factors such as the speed of the receiver (e.g. assuming the receiver is located on a motorized vehicle)—in theory, if the receiver is moving quickly, in an extreme case the beam former may be re-calculated for every N+1 samples (e.g. the first calculation might be run on 0-2000 samples, the second calculation might be run on 1-2001 samples). Thus, the update rate would be the same as the sample rate, but such a scenario would require significant processing capability.

As illustrated by the left-most path406, a representative of the lower 1st adjacent interference signal, for example, signal210, calculated from the N samples, may be obtained in the frequency-bin around −250 kHz, between −200 kHz and −300 kHz416, where there may be little interference, for example, from the lower band206of the desired signal202. In the second path408, a representative of the desired-signal, for example, signal202(including 204, 206, and 208), calculated from the N samples, may be obtained in the frequency-bin around zero Hertz (0 kHz) between −50 kHz and +50 kHz418, where there may be relatively little interference, for example, from the lower 1st adjacent interference signal210and the upper 1st adjacent interference signal212. In the third path410, a representative of the upper 1st adjacent interference signal, for example, signal212, calculated from the N samples, may be obtained in the frequency-bin around +250 kHz, between +200 kHz and +300 kHz420, where there may be little interference, for example, from the upper band208of the desired signal202. Thus, the representative of the lower 1st adjacent interference signal, for example, signal210, and the representative of the upper 1st adjacent interference signal, for example, signal212, may be centered at steps416and420such that the signal is shifted to the Direct Current (DC) bias. Note that the desired frequency is shifted at step402, so does not need to be shifted to the DC bias in subsequent steps. Thus, path408may proceed to step418from step404, or in an alternate embodiment step418may wait for steps422and424to run in parallel.

At steps422,418and424, each of the representatives of the frequency-shifted lower 1st adjacent interference signal, desired signal, and frequency-shifted upper 1st adjacent interference signal may be filtered using a low-pass filter (LPF) of 50 kHz around 0 kHz (e.g. from −50 kHz to 50 kHz) with a finite impulse response (FIR) of 25-taps. In one embodiment, a low-pass filter is combined with a band-pass filter to shift each signal to zero (steps416and420), and to filter out half of the signal to ensure the cleanest signal possible (steps422,418, and424). Thus, three different frequency bands of 100 kHz may be processed by the beam-former as training signals as follows.

In steps426,428, and430the inverse of the N co-variance matrices of the 1stadjacent interference representative sample signals, (Rii−1, Rjj−1), and the N co-variance matrix of the representative sample desired signal, Rssmay be generated. Note that the matrices will be 2×2 matrices as there are two antennas in the array; the main diagonal will correspond to information about the interference or desired signal, and the secondary diagonal will correspond to information about the spatial correlation of the interference or desired signal (depending on which matrix is considered). Once the matrices are generated426,428, and430, beam-former step432,434,438may nearly instantaneously calculate weights that maximize the Signal-to-Interference+Noise-Ratio (SINR) based on the inverse of the N co-variance matrices of the 1st adjacent interference representative sample signals and the N co-variance matrix of the representative sample desired signal, respectively. The co-variance matrix may be used as a measure for the signal power of the desired signal and of the interference signal. The beam former maximizes the signal for SINR, the maximum power of the desired signal and minimum contribution of the interference plus noise, because this will result in a determination of the directionality; this condition will correspond to the direction of origination of the signal. The weights may be computed nearly instantaneously using “estimation-and-plug” techniques by solving an eigenvalue problem with a “principal-component-analysis” (PCA) method as described below. Consequently, the complex base-band beam forming algorithm is fast, i.e., it has a low-latency.

As the weights are generated as illustrated by steps416through434in paths406,408, and410, as illustrated in path412the lower 1st adjacent interference signal (note the frequency-bin of path412is 200 kHz wide, i.e., the channel-grid of the lower 1st adjacent), for example, signal210, filtered from the N samples, may be obtained in the frequency-bin around −200 kHz, between −300 kHz and −100 kHz436, where there may be great interference, for example, from the lower band206of the desired signal202; and in the third path414, the upper 1st adjacent interference signal, for example (as above, the frequency-bin of path414is 200 kHz wide, i.e., the channel-grid of the upper 1st adjacent), signal212, filtered from the N samples, may be obtained in the frequency-bin around +200 kHz, between +100 kHz and +300 kHz440, where there may be great interference, for example, from the upper band208of the desired signal202. Note the portion of the desired signal, such as signal204, in the middle path including step458, the lower 1st adjacent interference signal, such as signal210in the path412, and the upper 1st adjacent interference signal, such as signal212in the path414, are 200 kHz wide. Note that the lower 1st adjacent interference signal, for example, signal210, and the upper 1st adjacent interference signal, for example, signal212, may be centered at steps436and440such that the signal is shifted to DC bias. As noted above, the desired frequency is shifted at step402, so does not need to be shifted to the DC bias in subsequent steps. Thus, the method may proceed directly to step458from step404, and in alternate embodiments may wait for steps442and444to run in parallel.

At steps442,458and444, each of the representatives of the frequency-shifted lower 1st adjacent interference signal, desired signal, and frequency-shifted upper 1st adjacent interference signal may be filtered using a low-pass filter (LPF) of 100 kHz around 0 kHz (e.g. from −100 kHz to 100 kHz) with a finite impulse response (FIR) of 21-taps; as a result, the filtered signal will include, for example, a desired signal such as signal202, inclusive of analog band204, lower digital sideband206, and upper digital sideband208. Note that fewer taps may be used in steps442,458and444than in steps418,422, and424in order to generate a stronger/steeper training signal as a representative for the beam-former. Although in some embodiments the correction frequency-bins, i.e., lower-bin from −300 kHz to −100 kHz (path412), the middle-bin from −100 kHz to +100 kHz (in the path including step458), and the upper-bin from +100 kHz to 300 kHz (path414) may not have as high selectivity as the training-signals in paths406,408, and410, the number of taps may be a parameter that may be optimized to increase the accuracy of the training signals or correction frequencies.

At steps446,448, and450the weighting coefficients derived from the training signal at steps432,434,438may be combined with the lower, middle, and upper bands of the desired signal202, respectively (as well as the lower interference210and the upper interference212), derived at steps442,458, and444. Note that there are two antennas, and the beam-forming for each antenna will result in a different weighting coefficient; thus, the low446, middle448, and high 450 coefficients will be applied 3×2 times with different weighting coefficients, which as noted are based on the Signal to Interference-plus-Noise ratios for each of the three frequency bands of 200 kHz wide each, i.e., the channel-grid, as discussed further below.

However, 3×2 weights will be calculated for the training signals at steps432,434, and438; if there is no upper interference208the weights generated in step434and the weights calculated by step438will both be based on the desired signal and background noise, and thus they will look similar due to the lack of interference. For instance, in an example with only one lower-side 1st adjacent interference signal such as signal210and two results from the antennas' beam-former434,438, there may be six weights, i.e., six complex-numbers computed or estimated to be applied at steps446,448, and450. Note the beam former432,434,438will perform the same steps regardless of whether interference is present or not. The SINR (if there is interference) or SNR (in cases without interference) optimization criterion will be applied in all cases so that the correct 3×2 weights will result regardless if there are none, one or two interference signals present. Thus, there may be three sets of complex numbers applied to signal202. One set will be applied to enhance lower portion206446(i.e., by reducing210), another set will be applied to enhance middle portion204448, and yet another set will be applied to enhance upper portion208450(i.e., by reducing212). As an example, firstly, the two estimated complex-numbers (representing amplitude and phase estimates) may adjust the beam-pattern for the removal of the 1st adjacent lower-sided interference signal446. Secondly, the beam-former may process the representative-information of the desired signal202to prevent null-steering to the desired signal, which are the second pair of weights, shown by the (mid) weighing-coefficients448. More specifically, as an example, the beam-former may compute the weights of the second 1st adjacent interference signal, e.g., an upper FM-neighbor, to steer a null to the upper 1st adjacent interference-signal, shown by step434.

Note that after combination of the upper, lower, and middle signals a complete signal of 600 kHz wide, i.e., from −300 kHz to +300 kHz, will result with interference substantially removed. Thus, for the lower frequency band two weighting coefficients based on the maximum Signal to Interference-plus-Noise ratio between the lower interference and the desired bands will be applied446, and then the same for the higher interference and the desired bands450, and then for the middle band no interference will be calculated to prevent self-nulling448as shown below. These signals may be combined to determine steering so that the received power of the desired signal may be improved452, after which point the method will end454.

The generation of weighting coefficients by complex digital base-band beam-forming with double-null steering by maximizing the SINR may be as follows. Note the estimation-criterion is the maximization of the SINR, which computes the optimal weights by taking the first derivative of the Signal-to-Interference+Noise-Ratio (SINR); the result of this derivative may be set to zero which may solve the equation. The SINR may be expressed as

SINR⁢=def⁢wH⁢Rss⁢wwH⁢Ri⁢⁢n⁢w(equation⁢⁢1)
where w are the weights, (•)His the Hermitian transpose, i.e., both the complex-conjugate and the transpose operation, and

RSS⁢=Δ⁢1N⁢∑n=1N⁢⁢(s⁡[n]·sH⁡[n]),⁢Ri⁢⁢n⁢=Δ⁢1N⁢∑n=1N⁢⁢({i⁡[n]+n⁡[n]}.
{i[n]+n[n]}H) (eq. 2) are the sample co-variance matrices (an approximation of the co-variance matrices over a finite number of samples N), and n[n] represents complex-Gaussian noise with zero-mean and variance σ2=N0. In an example case the sample co-variance matrix Rinmay not be available, however, for an IBOC transmission the representative signals may be used to obtain an approximation of the sample co-variance matrix Rin(and also for the sample co-variance matrix Rjn), yielding:

Ri⁢⁢n≈Rii⁢=Δ⁢1N⁢∑n=1N⁢⁢(i^⁡[n]·i^H⁡[n])lower⁢⁢interferenceRss≈Rs^⁢s^⁢=Δ⁢1N⁢∑n=1N⁢⁢(s^⁡[n]·s^H⁡[n])desired⁢⁢signalRjn≈Rjj⁢=Δ⁢1N⁢∑n=1N⁢⁢(j^⁡[n]·j^H⁡[n])upper⁢⁢interference(eq.⁢3)
where î[n] is the stream of samples for the surrogate of the lower 1stadjacent interference signal, ŝ[n] is the stream of samples for the surrogate of the desired signal, and ĵ[n] is the stream of samples for the surrogate of the upper 1stadjacent interference signal.

An IBOC transmission with two 1stadjacent interference signals (and, in some embodiments, complex-Gaussian noise) may be expressed as a summation of three spatial different and independent signals, i.e., the lower 1stadjacent interference signal, the desired signal, and the upper 1stadjacent interference signal with independent complex-Gaussian noise. Thus, the SINR may be expressed as

For a received IBOC transmission, such as transmission200, the interference-signals210,212and the desired signal202may be separated into different frequency-bins, so that for each frequency-bin the SINR will be optimized to obtain the weights, such that

∇wiH⁢{wiH⁢Rss⁢wiwiH⁢Rii⁢wi}=∇wiH⁢{(wiH⁢Rss⁢wi)⁢(wiH⁢Rii⁢wi)-1}=0(eq.⁢7)
where ∇{•} is the expression for taking the complex gradient. Applying partial-differentiation may result in Rsswi(wiHRiiwi)−1−(wiHRiiwi)−2Riiwi(wiHRsswi)=0 (eq. 8), which may be rewritten as the expression

λi⁢=def⁢si+n=wiH⁢Rss⁢wiwiH⁢Rii⁢wi(eq.⁢10)
may be defined as the SINR of the frequency-bin for the lower 1st adjacent interference signal. This may be rewritten as Rsswi=Riiwiλi(Rii−1Rss)wi=λiwi(eq. 11) which is an Eigen-value problem whose solution provides the optimal weights for maximizing the SINR: wopt,i=P{Rii−1Rss} (eq. 12), where P{•} is the operator that returns the principal Eigen-vector of a matrix based on the PCA.

With a two-antenna ULA, the sample co-variance matrix may be a 2-by-2 matrix and the characteristic-function to compute the Eigen-vectors is a quadratic-function expressed as λ2−

tr⁢{A}⁢λ+det⁢{A}=0⇒λ1,2=tr⁢{A}±tr⁢{A}2-4·det⁢{A}2(eq.⁢13)
where ARii−1Rss, tr{A} is the trace of matrix A, and det{A} is the determinant of matrix A.

Similar results may be derived in a similar-way for the desired signal and the upper 1stadjacent interference signal. However, the sample co-variance matrix Rnnof independent zero-mean complex-Gaussian noise variables may be assumed to be a diagonal-matrix with noise variances σ2on the main-diagonal. Note the desired signal may not have an interference signal (one reason why it is used for the training signal). Because there is no interference self-nulling may be prevented—where there is no co-variance matrix from training there is no training signal—only the desired signal itself, absent of interference (other than un-correlated and/or white noise). By taking the inverse of the noise-matrix there will be values only on the main diagonal, such that the desired signal will be scaled, but will not change the Eigen-vectors. Therefore, the inverse sample covariance matrix Rnn−1may also be expressed as a diagonal matrix and Rnn−1Rss∝Rsswopt,s=P{Rnn−1Rss}∝P{Rss} (eq. 14). Finally, the optimal weights for the lower and upper 1st adjacent interference signal and the desired signal may be expressed as
wopt,i=P{Rii−1Rss} lower interference
wopt,s=P{Rss} desired signal
wopt,j=P{Rjj−1Rss} upper interference  (eq. 15)
where the Eigen-vectors are computed by solving “straightforward” quadratic characteristic-functions for each of the three frequency-bins.

Thus, for a hybrid or all-digital IBOC transmission, representative information such as Rii, Rss, and Rjjmay be made available by taking the sample-average and appropriately filtering the received baseband IBOC-signal including the 1stadjacent signals, i.e., a baseband signal with a bandwidth of roughly 600 kHz between −300 kHz and +300 kHz such as signal300. The appropriate surrogate-signals may be obtained through filtering as shown in steps416-424. In addition, from the training-signals/surrogate-signals the required training information for the desired signals as well as the training information for the interference signals may be computed. Due to the availability of this required training information for the lower 1stadjacent interference signal in the form of a first sample co-variance matrix (Rii), the desired signal with a second sample co-variance matrix (Rss), and the upper 1stadjacent interference signal with a third sample co-variance matrix (Rjj) a fast, i.e., low-latency (“estimation-and-plug”), low-complexity (inverse of a two-by-two matrix, solving a quadratic characteristic function), and accurate (three frequency-bin filtering) complex base-band beam forming with double-null-steering may be applied to remove, or significantly reduce, 1stadjacent interference signals on a transmitted IBOC-signal. As such, the power of the received signal-power of the desired IBOC-signal may be improved with minimum latency (delay) and, since existing receiving equipment may be utilized to implement the filtering, at low cost.

FIG. 5illustrates an exemplary hardware diagram for a device500for performing electronically-steered complex base-band beam forming with double-null-steering such as exemplary method400. Device500may be connected to an antenna array502,504, including two or more antennas spaced by half a wave-length506. The antennas502,504may be connected to a first so-called RF-block508that may include one or multiple radio-frontends510,512, which may be equal in number to the number of antennas used in the array502,504(e.g., two antennas). Each radio-frontend510,512may down-convert received IBOC signals to base-band In-phase (I) and quadrature (Q) signals,514,516, such that the signal received at antenna array502,504are on the carrier-frequency (e.g., the USA FM-broadcast band from 87.9 to 107.9 MHz with channel-grid sizes of 200 kHz) to complex I/Q base-band signals around 0 MHz514,516.

The I/Q base-band signals514,516may be sent to an A/D-block518that may include one or multiple analog-to digital converters520,522(which may be equal in number to the number of antennas used in the array502,504) to digitize the analog base-band I/Q-signals514,516into samples in discrete time524,526, for example, every clock-cycle two samples (I and Q)524,526, per antenna-branch502,504, may be generated by each A/D-converter block518.

Digital I/Q-signals524,526may be sent to a base-band block528, that may filter, per antenna-branch530,532, to obtain the required base-band input-signals534,536, for example, such as an exemplary signal300, for the next block538, which may be referred to as the digital adaptive beam-former block538, or beam-former538. Base-band block528may perform operations corresponding to step402of method400such that the inputs to beam former block538are534and536, i.e., a signal such as signal300may be the input534,536of beam former538.

As may be seen below, within digital-adaptive beam-former block538the training-generation block540and coefficient-update block550roughly correspond to steps406-410,416-434, and438of method400, and the complex-FIR filters542,544roughly correspond to steps412-414,436,440-450, and458of method400. Signal x1[n]534may be an input to the beam-former538originating from the 1st-antenna branch502,514,524, and xM[n]536may be an input to the beam-former538coming from the M-th antenna branch504,516,526. Thus, with a two-antenna approach xM[n] will be x2[n], and so the inputs534,536to block538will include x1[n] and x2[n]534,536. Note that the equivalent time-index n in equations 1-15 may be equivalent to index n in the beam-former block538. Signals x1[n] and x2[n]534,536represent the inputs to the beam-former538, and, in the case where the array includes two antennas, are related to equations 1-15 above by x1[n]=s1[n]+i1[n]+j1[n] for the 1stantenna-branch and x2[n]=s2[n]+i2[n]+j2[n] for the 2ndantenna-branch. In other words, for the two-antenna case, the relation with equations 1-15 above is x[n]=s[n]+i[n]+j[n], which may be derived from the input pair of signals (x1[n], x2[n])=(s1[n]+i1[n]+j1[n], s2[n]+i2[n]+j2[n]).

The down-converted signal300may illustrate a power-spectral-density of input-signal x1[n], and the down-converted signal power-spectral-density of x2[n] would appear the same, but actually would be shifted by half a wavelength. Signals x1[n]534through xM[n]536(which in a two-antenna array will be x2[n]) may enter the beam-former538to generate training signals540as described in paths406,408, and410above, and to be filtered and combined (542,544)—filtered to isolate the portions of the frequency including a desired signal such as signal202, inclusive of an analog band204, a lower digital sideband206, and an upper digital sideband208, as described above with respect to steps436,440,442,444, and458, and combined with weighting coefficients generated from the training signals as in steps446,448, and450of method400.

The computed/estimated coefficients w1[n]548through wM[n]562(which may reflect a computation-rate on sample basis n) may correspond to the computed/estimated optimal weighing-coefficients wopt,i, wopt,s, and wopt,jas shown in equation 15 above, and may reflect the outputs of steps432,434, and438of method400. Similarly, after signals x1[n]534through xM[n]536(which in a two-antenna array will be x2[n]) enter the beam-former538and are filtered and combined542,544, the resulting output signals y1[n]552through yM[n]554may represent the input signals of x1[n]534through xM[n]536, respectively, corrected/compensated by the computed weights wopt,i, wopt,sand wopt,j, corresponding to steps446,448, and450of method400. Signals y1[n] through yM[n] may be input to the complex combiner556to determine steering so that the received power of the desired signal may be improved as may be shown in step452of method400.

The complex combiner556may output a beamformer I/Q signal558, which may correspond to the output of step452of method400. The beamformer I/Q signal558may be input to the so-called (H)IBOC-receiver block (without interference compensation)560which may be a (H)IBOC receiver that performs OFDM demodulation and decoding without a 1st-adjacent interference compensation, since the beam-forming has instead performed the task of compensating for interference.

FIG. 6illustrates an exemplary hardware diagram for a device600including hardware for performing electronically-steered complex base-band beam forming with double-null-steering. The exemplary device600may include elements of device500. As shown, the device600includes a processor620, memory630, user interface640, antenna array650, and storage660interconnected via one or more system buses610. It will be understood thatFIG. 6constitutes, in some respects, an abstraction and that the actual organization of the components of the device600may be more complex than illustrated.

The processor620may be any hardware device capable of executing instructions stored in memory630or storage660. As such, the processor may include a microprocessor, field programmable gate array (FPGA), application-specific integrated circuit (ASIC), or other similar devices.

The memory630may include various memories such as, for example L1, L2, or L3 cache or system memory. As such, the memory630may include static random access memory (SRAM), dynamic RAM (DRAM), flash memory, read only memory (ROM), or other similar memory devices.

The user interface640may include one or more devices for enabling communication with a user such as an administrator. For example, the user interface640may include a display and an input for receiving user commands.

The antenna array650may include one or more devices for enabling reception of transmissions. For example, the antenna array650may include a radio front end and a receiver block for demodulation and decoding as explained above. Additionally, the antenna array650may include an RF/IF receiver and an ADC/baseband processor. Various alternative or additional hardware or configurations for the antenna array650will be apparent.

The storage660may include one or more machine-readable storage media such as read-only memory (ROM), random-access memory (RAM), magnetic disk storage media, optical storage media, flash-memory devices, or similar storage media. In various embodiments, the storage660may store instructions for execution by the processor620or data upon with the processor620may operate. For example, the storage660may store Training Instructions662for performing generation of a training signal, or Beamformer Instructions664for complex combinations of weighted coefficients and received signals according to the concepts described herein. The storage660may also store Signal Data666for use by the processor executing the Training or Beamformer Instructions662,664.

According to the foregoing, various exemplary embodiments provide for low-complexity, low-cost, and low-latency (fast acquisition) improved reception of in-band on-channel transmissions through suppression or removal of the 1st adjacent interference-signals and improvement of the received signal-power of the desired signal, whether transmitted in hybrid-mode or all-digital mode. In particular, by performing electronically-steered complex base-band beam forming with double-null-steering to remove or significantly reduce the first adjacent FM-interference signals by double-null-steering with an Uniform-Linear-Antenna-array (ULA) consisting of two isotropic antennas that are spaced by half a wave-length, exploiting the fact that both the first adjacent FM-interference signals as well as the desired IBOC-signal are separated in three different frequency bins, each of a known width, e.g. 200 kHz wide. More specifically, using a portion of roughly 100 kHz wide (a frequency-bin of 100 kHz) to obtain training-signal or surrogate-signal representatives of the desired IBOC-signal, the lower first adjacent FM-interference signal, and the upper first adjacent FM-interference-signal, which training-signals or surrogate-signals may be used to compute optimal steering-weights, i.e., the optimal steering-weight vector, based on maximization of the SINR, which is an average of the SINR of each of the three frequency-bins corresponding to the desired IBOC signal and the two 1st adjacent FM-interference signals.

It should be apparent from the foregoing description that various exemplary embodiments of the invention may be implemented in hardware and/or firmware. Furthermore, various exemplary embodiments may be implemented as instructions stored on a machine-readable storage medium, which may be read and executed by at least one processor to perform the operations described in detail herein. A machine-readable storage medium may include any mechanism for storing information in a form readable by a machine, such as a personal or laptop computer, a server, or other computing device. Thus, a machine-readable storage medium may include read-only memory (ROM), random-access memory (RAM), magnetic disk storage media, optical storage media, flash-memory devices, and similar storage media.

It should be appreciated by those skilled in the art that any block diagrams herein represent conceptual views of illustrative circuitry embodying the principals of the invention. Similarly, it will be appreciated that any flow charts, flow diagrams, state transition diagrams, pseudo code, and the like represent various processes which may be substantially represented in machine readable media and so executed by a computer or a processor, whether or not such computer or processor is explicitly shown.