Field coil type rotating electric machine

A field coil type rotating electric machine includes a rotor where both a series resonant circuit including a first winding and a capacitor and a parallel resonant circuit including a second winding and the capacitor are formed. The first winding is radially located closer than the second winding to a stator. The capacitance of the capacitor and the ratio of the number of turns of the second winding to the number of turns of the first winding are set to have the amplitude of a total resultant magnetic flux lower than the amplitude of a field resultant magnetic flux. The total resultant magnetic flux is the resultant of the field resultant magnetic flux and magnetic flux generated by harmonic currents flowing in phase windings of a stator coil. The field resultant magnetic flux is the resultant of magnetic fluxes generated by harmonic currents flowing in the first and second windings.

CROSS-REFERENCE TO RELATED APPLICATION

This application is based on and claims priority from Japanese Patent Application No. 2019-096348 filed on May 22, 2019, the contents of which are hereby incorporated by reference in their entirety into this application.

BACKGROUND

1 Technical Field

The present disclosure relates to field coil type rotating electric machines.

2 Description of Related Art

There is known a field coil type rotating electric machine. This machine includes a stator having a stator coil, a field coil including a serially-connected winding pair consisting of first and second windings, a rotor having a rotor core and a plurality of main pole portions, and a diode. The main pole portions are formed, at predetermined intervals in a circumferential direction, to radially protrude from the rotor core. The diode has its cathode connected to a first-winding-side end of the serially-connected winding pair and its anode connected to a second-winding-side end of the serially-connected winding pair. Each of the first and second windings is wound on each of the main pole portions. The stator coil is comprised of a plurality of phase windings. In operation, each of the phase windings of the stator coil is supplied with both fundamental current mainly for generating torque and harmonic current mainly for exciting the field coil.

Upon supply of the harmonic currents to the phase windings of the stator coil, main magnetic flux flows through a magnetic circuit which includes the main pole portions circumferentially adjacent to one another and the rotor core. Consequently, with the main magnetic flux flowing through the magnetic circuit, voltages are induced respectively in the first and second windings that are serially connected with each other, thereby inducing electric currents having harmonic components respectively in the first and second windings. The electric currents induced in the first and second windings are then rectified by the diode to flow in one direction, namely the rectification direction. As a result, field current flows in the field coil in the rectification direction, thereby exciting the field coil.

On the other hand, upon supply of the harmonic currents to the phase windings of the stator coil, leakage magnetic flux is also generated in addition to the main magnetic flux. The leakage magnetic flux flows between each circumferentially-adjacent pair of the main pole portions without flowing through the rotor core, crossing the field coil. Upon the leakage magnetic flux crossing the field coil, the voltages induced respectively in the first and second windings may become opposite in polarity to each other, thereby reducing the sum of the electric currents induced respectively in the first and second windings and thus the field current flowing in the field coil.

To solve the above problem, the known field coil type rotating electric machine further includes a capacitor that is connected in parallel with the second winding. Consequently, both a series resonant circuit including the first winding and the capacitor and a parallel resonant circuit including the second winding and the capacitor are formed, thereby increasing the field current.

SUMMARY

According to the present disclosure, there is provided a field coil type rotating electric machine which includes a stator, a field coil and a rotor. The stator includes a stator coil that is comprised of a plurality of phase windings. The field coil includes a serially-connected winding pair consisting of a first winding and a second winding that are connected in series with each other. The rotor includes a rotor core and a plurality of main pole portions that are formed at predetermined intervals in a circumferential direction and each radially protrude from the rotor core. Each of the first and second windings of the field coil is wound on each of the main pole portions of the rotor. Each of the phase windings of the stator coil is configured to be supplied with harmonic current to induce field current in the field coil. The field coil type rotating electric machine further includes a diode and a capacitor. The diode has its cathode connected to a first-winding-side end of the serially-connected winding pair and its anode connected to a second-winding-side end of the serially-connected winding pair. The capacitor is connected in parallel with the second winding. In the field coil type rotating electric machine, there are formed both a series resonant circuit including the first winding and the capacitor and a parallel resonant circuit including the second winding and the capacitor. The first winding is radially located closer than the second winding to the stator. The capacitance of the capacitor and the turn number ratio, which is the ratio of the number of turns of the second winding to the number of turns of the first winding, are set to have the amplitude of a total resultant magnetic flux lower than the amplitude of a field resultant magnetic flux when the harmonic currents are supplied to the phase windings of the stator coil. The total resultant magnetic flux is the resultant of the field resultant magnetic flux and magnetic flux generated by the harmonic currents flowing in the phase windings of the stator coil. The field resultant magnetic flux is the resultant of magnetic flux generated by harmonic current flowing in the first winding and magnetic flux generated by harmonic current flowing in the second winding.

DESCRIPTION OF EMBODIMENTS

In the field coil type rotating electric machine known in the art (see, for example, Japanese Patent Application Publication No. JP 2018-042401 A), the harmonic currents supplied to the phase windings of the stator coil cause harmonic currents to flow in the first and second windings. Consequently, magnetic flux is generated by the harmonic currents flowing in the first and second windings. On the other hand, magnetic flux is also generated by the harmonic currents flowing in the phase windings of the stator coil. In this case, the amplitude of resultant magnetic flux, which is the resultant of the magnetic flux generated by the harmonic currents flowing in the first and second windings and the magnetic flux generated by the harmonic currents flowing in the phase windings of the stator coil, may be increased, thereby increasing the torque ripple ratio of the rotating electric machine. Here, the torque ripple ratio denotes the ratio of an amount of variation in the torque of the rotating electric machine to a DC (Direct Current) component of the torque.

In contrast, in the field coil type rotating electric machine according to the present disclosure, upon the harmonic currents flowing in the phase windings of the stator coil, harmonic currents are induced to flow respectively in the first and second windings of the field coil. The difference in phase between the harmonic current flowing in the series resonant circuit including the first winding and the harmonic current flowing in the parallel resonant circuit including the second winding is, for example, larger than 120° and smaller than 240°. Moreover, with change in the turn number ratio, both the amplitude of the magnetic flux generated by the harmonic current flowing in the first winding and the amplitude of the magnetic flux generated by the harmonic current flowing in the second winding change. Consequently, the amplitude of the field resultant magnetic flux also changes; the field resultant magnetic flux is the resultant of the magnetic flux generated by the harmonic current flowing in the first winding and the magnetic flux generated by the harmonic current flowing in the second winding. Furthermore, with change in the amplitude of the field resultant magnetic flux, the amplitude of the total resultant magnetic flux also changes; the total resultant magnetic flux is the resultant of the field resultant magnetic flux and the magnetic flux generated by the harmonic currents flowing in the phase windings of the stator coil. That is, the amplitude of the total resultant magnetic flux can be made lower than the amplitude of the field resultant magnetic flux by changing the turn number ratio. In view of the above, in the field coil type rotating electric machine according to the present disclosure, the capacitance of the capacitor and the turn number ratio are set so that when the harmonic currents are supplied to the phase windings of the stator coil, the amplitude of the total resultant magnetic flux becomes lower than the amplitude of the field resultant magnetic flux. Consequently, it becomes possible to lower the torque ripple ratio of the rotating electric machine.

In a further implementation, the turn number ratio is set to be higher than or equal to 0.5 and lower than or equal to 5.2 with the capacitance of the capacitor set to be in a predetermined range under the constraint of keeping the resonance frequency of the series resonant circuit at a predetermined frequency and the turn number sum, which is the sum of the number of turns of the first winding and the number of turns of the second winding, at a predetermined turn number. In other words, the resonance frequency of the series resonant circuit is set to the predetermined frequency by setting the capacitance of the capacitor to be in the predetermined range and the turn number ratio to be higher than or equal to 0.5 and lower than or equal to 5.2. In addition, upon the setting of the resonance frequency of the series resonant circuit, the resonance frequency of the parallel resonant circuit is also set accordingly.

More specifically, to keep the resonance frequency of the series resonant circuit including the first winding at the predetermined frequency under the constraint of keeping the turn number sum at the predetermined turn number, it is necessary to set the capacitance of the capacitor and the turn number ratio such that the higher the capacitance of the capacitor, the higher the turn number ratio. The inventor of the present application has confirmed, through experimental investigation, that when the capacitance of the capacitor is varied in the predetermined range under the constraint of keeping the resonance frequency of the series resonant circuit at the predetermined frequency and the turn number sum at the predetermined turn number (hereinafter, to be referred to as the first constraint), the turn number ratio falls within the range of 0.5 to 5.2. Moreover, the inventor has also confirmed, through experimental investigation, that the torque of the rotating electric machine can be kept high by setting the turn number ratio to be in the range of 0.5 to 5.2 under the first constraint.

On the other hand, to keep the resonance frequency of the parallel resonant circuit including the second winding at the predetermined frequency under the constraint of keeping the turn number sum at the predetermined turn number, it is necessary to set the capacitance of the capacitor and the turn number ratio such that the higher the capacitance of the capacitor, the lower the turn number ratio. The inventor has confirmed, through experimental investigation, that: in the case of setting the turn number ratio to be in the range of 0.7 to 5.2 under the constraint of keeping the resonance frequency of the parallel resonant circuit at the predetermined frequency and the turn number sum at the predetermined turn number (hereinafter, to be referred to as the second constraint), the torque of the rotating electric machine becomes highest when the turn number ratio is equal to an intermediate value in the range; with the turn number ratio deviating from the intermediate value, the torque of the rotating electric machine considerably drops. Moreover, the inventor has also confirmed, through experimental investigation, that when the turn number ratio is set to be in the range of 0.5 to 5.2, the torque of the rotating electric machine tends to be lower under the second constraint than under the first constraint. This is because the first winding is radially located closer than the second winding to the stator and thus more susceptible than the second winding to the magnetic flux generated by the harmonic currents flowing in the phase windings of the stator coil.

Accordingly, the torque of the rotating electric machine can be kept higher by setting the capacitance of the capacitor and the turn number ratio under the first constraint pertaining to the series resonant circuit than under the second constraint pertaining to the parallel resonant circuit. Moreover, under the first constraint, the capacitance of the capacitor can be set within the predetermined range and the turn number ratio can be set within the range of 0.5 to 5.2; therefore, the degree of freedom of setting the capacitance of the capacitor and the turn number ratio is improved.

In view of the above, in the further implementation, the resonance frequency of the series resonant circuit is set to the predetermined frequency by setting the capacitance of the capacitor to be in the predetermined range and the turn number ratio to be in the range of 0.5 to 5.2. Consequently, it becomes possible to improve the degree of freedom of setting the capacitance of the capacitor and the turn number ratio to keep the torque of the rotating electric machine high. For example, the turn number ratio may be set to be in the range of 0.7 to 5.2. In this case, it is possible to effectively lower the torque ripple ratio while keeping the torque of the rotating electric machine high.

Exemplary embodiments will be described hereinafter with reference to the drawings. It should be noted that for the sake of clarity and understanding, identical components having identical functions throughout the whole description have been marked, where possible, with the same reference numerals in each of the figures and that for the sake of avoiding redundancy, descriptions of identical components will not be repeated.

First Embodiment

FIG. 1shows the overall configuration of a rotating electric machine system which includes a field coil type rotating electric machine30according to the first embodiment.

As shown inFIG. 1, the rotating electric machine system further includes a DC power supply10, an inverter20and a controller40in addition to the rotating electric machine30.

The rotating electric machine30is a field coil type synchronous rotating electric machine. More particularly, in the present embodiment, the controller40controls the rotating electric machine30to function as an ISG (Integrated Starter Generator) or an MG (Motor Generator). In addition, the rotating electric machine30, the inverter20and the controller40may be either integrated into a single drive apparatus or be configured as individual components.

As shown inFIG. 3, the rotating electric machine30includes a rotor60having a field coil70. In the present embodiment, as shown inFIGS. 2 and 3, the field coil70is constituted of a serially-connected winding pair consisting of a first winding71aand a second winding71bthat are connected in series with each other. The field coil70is formed by, for example, compression shaping to improve the space factor and the assemblability thereof. Moreover, the field coil70is formed, for example, of aluminum wires. The specific gravity of aluminum wires is relatively low. Therefore, forming the field coil70with aluminum wires, it is possible to lower the centrifugal force during rotation of the rotor60. Moreover, aluminum wires are lower in both strength and hardness than copper wires. Therefore, aluminum wires are suitable for being compression-shaped.

The rotating electric machine30also includes a stator50having a stator coil31. The stator coil31is formed, for example, of copper wires. As shown inFIGS. 1 and 3, the stator coil31includes a U-phase winding31U, a V-phase winding31V and a W-phase winding31W, which are arranged to be offset from each other by 120° in electrical angle.

As shown inFIG. 1, the inverter20includes a serially-connected U-phase switch pair consisting of a U-phase upper-arm switch SUp and a U-phase lower-arm switch SUn, a serially-connected V-phase switch pair consisting of a V-phase upper-arm switch SVp and a V-phase lower-arm switch SVn, and a serially-connected W-phase switch pair consisting of a W-phase upper-arm switch SWp and a W-phase lower-arm switch SWn.

To a junction point between the U-phase upper-arm and lower-arm switches SUp and SUn, there is connected a first end of the U-phase winding31U of the stator coil31. To a junction point between the V-phase upper-arm and lower-arm switches SVp and SVn, there is connected a first end of the V-phase winding31V of the stator coil31. To a junction point between the W-phase upper-arm and lower-arm switches SWp and SWn, there is connected a first end of the W-phase winding31W of the stator coil31. A second end of the U-phase winding31U, a second end of the V-phase winding31V and a second end of the W-phase winding31W are connected together to defined a neutral point therebetween. That is, in the present embodiment, the U-phase, V-phase and W-phase windings31U,31V and31W of the stator coil31are star-connected.

In addition, in the present embodiment, each of the switches SUp, SVp, SWp, SUn, SVn and SWn is implemented by an IGBT (Insulated-Gate Bipolar Transistor). Moreover, each of the switches SUp, SVp, SWp, SUn, SVn and SWn has a freewheeling diode connected in antiparallel thereto.

Each of the U-phase, V-phase and W-phase upper-arm switches SUp, SVp and SWp has its collector connected to a positive terminal of the DC power supply10. Each of the U-phase, V-phase and W-phase lower-arm switches SUn, SVn and SWn has its emitter connected to a negative terminal of the DC power supply10. In addition, a smoothing capacitor11is connected in parallel with the DC power supply10.

The rotating electric machine system further includes an angle detection unit41. The angle detection unit41is configured to output an angle signal indicative of a rotation angle of the rotor60of the rotating electric machine30. The angle signal outputted from the angle detection unit41is inputted to the controller40.

Next, the configuration of the stator50and the rotor60of the rotating electric machine30will be described in detail with reference toFIGS. 2 and 3. As shown inFIG. 3, both the stator50and the rotor60are arranged coaxially with a rotating shaft32. Hereinafter, the direction in which a central axis of the rotating shaft32extends will be referred to as the axial direction; the directions of extending radially from the central axis of the rotating shaft32will be referred to as radial directions; and the direction of extending along a circle whose center is on the central axis of the rotating shaft32will be referred to as the circumferential direction.

The stator50is formed by laminating a plurality of soft-magnetic steel sheets in the axial direction. The stator50has an annular stator core51and a plurality of teeth52which each protrude radially inward from the stator core51and are spaced at equal intervals in the circumferential direction. Between each circumferentially-adjacent pair of the teeth52, there is formed one slot. More particularly, in the present embodiment, the number of teeth52is set to 48; accordingly, the number of the slots is also equal to 48. In addition, each of the U-phase, V-phase and W-phase windings31U,31V and31W of the stator coil31is wound on the teeth52in a distributed winding manner or a concentrated winding manner.

The rotor60is also formed by laminating a plurality of soft-magnetic steel sheets in the axial direction. The rotor60has a cylindrical rotor core61and a plurality of main pole portions62which each protrude radially outward from the rotor core61and are spaced at equal intervals in the circumferential direction. Distal end surfaces (or radially outer end surfaces) of the main pole portions62radially face distal end surfaces (or radially inner end surfaces) of the teeth52of the stator50. More particularly, in the present embodiment, the number of the main pole portions62is set to 8.

On each of the main pole portions62of the rotor60, the first winding71aof the field coil70is wound on the radially outer side (i.e., the stator side) while the second winding71bof the field coil70is wound on the radially inner side (i.e., the non-stator side). That is, the first winding71ais located closer to the stator50(i.e., more radially outward) than the second winding71bis. Moreover, on each of the main pole portions62, the first and second windings71aand71bare wound in the same direction. Furthermore, for each circumferentially-adjacent pair of the main pole portions62, the winding direction of the first and second windings71aand71bon one of the main pole portions62of the circumferentially-adjacent pair is opposite to the winding direction of the first and second windings71aand71bon the other of the main pole portions62of the circumferentially-adjacent pair. Consequently, the magnetization directions of the main pole portions62of the circumferentially-adjacent pair are opposite to each other.

FIG. 2shows an electric circuit formed in the rotor60that has the first and second windings71aand71bof the field coil70wound on the same main pole portions62. In the rotor60, there are provided a diode80as a rectifying element and a capacitor90. A first end of the first winding71a(or the first-winding-side end of the serially-connected winding pair) is connected with the cathode of the diode80. A second end of the first winding71ais connected with a first end of the second winding71b. A second end of the second winding71b(i.e., the second-winding-side end of the serially-connected winding pair) is connected with the anode of the diode80. The capacitor90is connected in parallel with the second winding71b. In addition, inFIG. 2, L1represents the inductance of the first winding71a; L2represents the inductance of the second winding71b; and C represents the capacitance of the capacitor90.

Next, the controller40will be described in detail. It should be noted that part or the whole of each function of the controller40may be realized either by hardware such as one or more integrated circuits or by software recorded on a non-transitory tangible recording medium and a computer executing the software.

The controller40acquires the angle signal outputted from the angle detection unit41. Then, based on the acquired angle signal, the controller40generates drive signals for turning on/off the switches SUp, SVp, SWp, SUn, SVn and SWn of the inverter20.

Specifically, when driving the rotating electric machine30to function as an electric motor, to convert DC power outputted from the DC power supply10into AC power and supply the resultant AC power to the U-phase, V-phase and W-phase windings31U,31V and31W, the controller40generates drive signals for turning on/off the switches SUp, SVp, SWp, SUn, SVn and SWn and outputs the generated drive signals to the gates of the switches SUp, SVp, SWp, SUn, SVn and SWn. Moreover, when driving the rotating electric machine30to function as an electric generator, to convert AC power outputted from the U-phase, V-phase and W-phase windings31U,31V and31W into DC power and supply the resultant DC power to the DC power supply10, the controller40generates drive signals for turning on/off the switches SUp, SVp, SWp, SUn, SVn and SWn and outputs the generated drive signals to the gates of the switches SUp, SVp, SWp, SUn, SVn and SWn.

In the present embodiment, the controller40turns on/off the switches SUp, SVp, SWp, SUn, SVn and SWn of the inverter20to supply each of the U-phase, V-phase and W-phase windings31U,31V and31W with resultant current which is the resultant of fundamental current and harmonic current. As shown inFIG. 4(a), the fundamental current is electric current mainly for causing the rotating electric machine30to generate torque. As shown inFIG. 4(b), the harmonic current is electric current mainly for exciting the field coil70. As shown inFIG. 4(c), the resultant current is the resultant of the fundamental current and the harmonic current and supplied as phase current to each of the U-phase, V-phase and W-phase windings31U,31V and31W. In addition, the vertical axis inFIG. 4is graduated to indicate the relationship in magnitude between the fundamental current, the harmonic current and the resultant current. It should be noted that the harmonic current may alternatively have a triangular waveform.

As shown inFIG. 5, U-phase, V-phase and W-phase currents IU, IV and IW, which are supplied respectively to the U-phase, V-phase and W-phase windings31U,31V and31W, are offset in phase from each other by 120° in electrical angle.

In the present embodiment, as shown inFIGS. 4(a) and (b), the period of the envelope of the harmonic current is set to be ½ of the period of the fundamental current. The envelope of the harmonic current is designated by a one-dot chain line inFIG. 4(b). Moreover, the timings at which the envelope of the harmonic current reaches its peak values are offset from the timings at which the fundamental current reaches its peak values. More specifically, the timings at which the envelope of the harmonic current reaches its peak values coincide with the timings at which the fundamental current reaches its center of variation (i.e., 0). The controller40controls the amplitude and period of each of the fundamental current and the harmonic current severally.

By superimposing the harmonic current shown inFIG. 4(b)on the fundamental current shown inFIG. 4(a), it is possible to suppress increase in the maximum values of the phase currents flowing respectively in the U-phase, V-phase and W-phase windings31U,31V and31W and thus possible to bring the torque of the rotating electric machine30into agreement with a command torque without increasing the capacity of the inverter20.

As an alternative, harmonic current shown inFIG. 6(b)may be applied instead of the harmonic current shown inFIG. 4(b). The fundamental current shown inFIG. 6(a)is identical to the fundamental current shown inFIG. 4(a). The harmonic current shown inFIG. 6(b)is offset in phase from the harmonic current shown inFIG. 4(b)by ¼ of the period of the fundamental current. The resultant current shown inFIG. 6(c)is the resultant of the fundamental current shown inFIG. 6(a)and the harmonic current shown inFIG. 6(b). In this case, as shown inFIG. 6(a) (b), the timings at which the envelope of the harmonic current reaches its peak values coincide with the timings at which the fundamental current reaches its peak values. Moreover, in this case, the U-phase, V-phase and W-phase currents IU, IV and IW, which are supplied respectively to the U-phase, V-phase and W-phase windings31U,31V and31W, are as shown inFIG. 7. In addition, it is also possible to apply harmonic current whose phase is between the phase of the harmonic current shown inFIG. 4(b)and the phase of the harmonic current shown inFIG. 6(b).

In the present embodiment, the first winding71aof the field coil70, the capacitor90and the diode80together form a series resonant circuit. The series resonant circuit has a resonance frequency which will be referred to as the first resonance frequency f1hereinafter. The first resonance frequency f1can be calculated based on the inductance L1of the first winding71aand the capacitance C of the capacitor90by the following equation (eq1). Moreover, the second winding71bof the field coil70and the capacitor90together form a parallel resonant circuit. The parallel resonant circuit has a resonance frequency which will be referred to as the second resonance frequency f2hereinafter. The second resonance frequency f2can be calculated based on the inductance L2of the second winding71band the capacitance C of the capacitor90by the following equation (eq2).

Upon the harmonic current flowing in each of the U-phase, V-phase and W-phase windings31U,31V and31W, the main magnetic flux varies due to harmonics; the main magnetic flux flows through a magnetic circuit that includes the main pole portions62circumferentially adjacent to one another, the rotor core61, the teeth52and the stator core51. With the variation in the main magnetic flux, voltages are induced respectively in the first and second windings71aand71b, thereby inducing electric currents respectively in the first and second windings71aand71b. Moreover, when the voltages induced respectively in the first and second windings71aand71bare of the same polarity as in the patterns1and4shown inFIG. 8, the electric currents induced respectively in the first and second windings71aand71bare not cancelled by each other, thus increasing the total electric current induced in the field coil70. Furthermore, the electric currents induced respectively in the first and second windings71aand71bare rectified by the diode80to flow in one direction, namely the rectification direction. Consequently, field current flows in the field coil70in the rectification direction, thereby exciting the field coil70. In addition, inFIG. 8, e1represents the voltage induced in the first winding71a; and e2represents the voltage induced in the second winding71b.

On the other hand, upon the harmonic current flowing in each of the U-phase, V-phase and W-phase windings31U,31V and31W, leakage magnetic flux is also generated in addition to the main magnetic flux. The leakage magnetic flux flows between each circumferentially-adjacent pair of the main pole portions62without flowing through the rotor core61, crossing the field coil70. Upon the leakage magnetic flux crossing the field coil70, the voltages induced respectively in the first and second windings71aand71bmay become opposite in polarity to each other, thereby reducing the sum of the electric currents induced respectively in the first and second windings71aand71band thus the field current flowing in the field coil70.

To solve the above problem, in the present embodiment, the capacitor90is connected in parallel with the second winding71b. Consequently, when the voltages induced respectively in the first and second windings71aand71bare opposite in polarity to each other as in the patterns2and3shown inFIG. 8, the electric currents induced in the first and second windings71aand71bflow via the capacitor90, without being canceled by each other. More specifically, as shown inFIG. 9A, both the electric current induced in the first winding71aand the electric current induced in the second winding71bmay flow to the anode of the diode80via the capacitor90. Otherwise, as shown inFIG. 9B, electric current may flow from the capacitor90to the anode of the diode80via the second winding71b. As a result, it becomes possible to increase the field current flowing in the field coil70.

Moreover, a further investigation of the patterns2and3shown inFIG. 8has been made by the inventor of the present application. The results of the investigation will be described hereinafter.

The electric circuit shown inFIG. 2basically includes three sub-circuits as shown inFIGS. 10-12. The sub-circuit shown inFIG. 10is the series resonant circuit that is formed of the first winding71a, the capacitor90and the diode80. The sub-circuit shown inFIG. 11is the parallel resonant circuit that is formed of the second winding71band the capacitor90. The sub-circuit shown inFIG. 12is a rectification circuit that is formed of the first winding71a, the second winding71band the diode80.

In the series resonant circuit shown inFIG. 10, at the first resonance frequency f1, the impedance becomes lowest and the alternating current becomes highest. Moreover, due to the diode80included in the series resonant circuit, half-wave current flows in the series resonant circuit. On the other hand, in the parallel resonant circuit shown inFIG. 11, at the second resonance frequency f2, the impedance becomes lowest and the alternating current becomes highest.

In the series resonant circuit, the electric current, which varies at the first resonance frequency f1, is supplied to the capacitor90. The electric current supplied to the capacitor90is then half-wave rectified by the diode80into the half-wave current. In addition, in the series resonant circuit, the electric current, which is blocked by the diode80, returns to the anode of the diode80via the second winding71bincluded in the parallel resonant circuit.

The inventor of the present application has found that the torque ripple ratio of the rotating electric machine30can be lowered by suitably setting the capacitance C of the capacitor90and the ratio of the number of turns N2of the second winding71bto the number of turns N1of the first winding71a(hereinafter, to be referred to as the turn number ratio N2/N1). In the present embodiment, the torque ripple ratio denotes the ratio (ΔTr/Tdc) of an amount of variation ΔTr/2 in the torque of the rotating electric machine30to a DC component Tdc of the torque. The findings by the inventor will be described in detail hereinafter.

Upon the harmonic currents flowing in the phase windings31U-31W of the stator coil31, harmonic currents, which vary at the same frequency as the harmonic currents flowing in the phase windings31U-31W of the stator coil31, are induced to flow respectively in the first and second windings71aand71bof the field coil70. The difference in phase between the harmonic current flowing in the series resonant circuit including the first winding71aand the harmonic current flowing in the parallel resonant circuit including the second winding71bis, for example, larger than 120° and smaller than 240°.

With change in the turn number ratio N2/N1, both the amplitude of first magnetic flux ϕ1and the amplitude of second magnetic flux ϕ2change. Here, the first magnetic flux ϕ1denotes magnetic flux generated by the harmonic current flowing in the first winding71a; and the second magnetic flux ϕ2denotes magnetic flux generated by the harmonic current flowing in the second winding71b. Consequently, the amplitude of field resultant magnetic flux ϕrt, which is the resultant of the first magnetic flux ϕ1and the second magnetic flux ϕ2, also changes.

FIGS. 13A-13Cshow the waveforms of the first magnetic flux ϕ1, the second magnetic flux ϕ2and the field resultant magnetic flux ϕrt at different ratios of amplitude between ϕ1and ϕ2. The ratio of amplitude between ϕ1and ϕ2can be changed by changing the turn number ratio N2/N1. In the examples shown inFIGS. 13A-13C, the difference in phase between the first magnetic flux ϕ1and the second magnetic flux ϕ2is equal to 180°. In addition, inFIGS. 13A-13C, the vertical axes are graduated to indicate the relationship in magnitude between the magnetic fluxes ϕ1, ϕ2and ϕrt.

FIG. 13Ashows the waveforms of the first magnetic flux ϕ1, the second magnetic flux ϕ2and the field resultant magnetic flux ϕrt when the ratio of amplitude between ϕ1and ϕ2is equal to 1:1. In the example shown inFIG. 13A, since the amplitude of the first magnetic flux ϕ1is equal to the amplitude of the second magnetic flux ϕ2, the field resultant magnetic flux ϕrt has only a DC component.

FIG. 13Bshows the waveforms of the first magnetic flux ϕ1, the second magnetic flux ϕ2and the field resultant magnetic flux ϕrt when the ratio of amplitude between ϕ1and ϕ2is equal to 1.4:1. In the example shown inFIG. 13B, since the amplitude of the first magnetic flux ϕ1is higher than the amplitude of the second magnetic flux ϕ2, the field resultant magnetic flux ϕrt varies in phase with the second magnetic flux ϕ2.

FIG. 13Cshows the waveforms of the first magnetic flux ϕ1, the second magnetic flux ϕ2and the field resultant magnetic flux ϕrt when the ratio of amplitude between ϕ1and ϕ2is equal to 1:1.4. In the example shown inFIG. 13C, since the amplitude of the second magnetic flux ϕ2is higher than the amplitude of the first magnetic flux ϕ1, the field resultant magnetic flux ϕrt varies in phase with the first magnetic flux ϕ1.

With change in the amplitude of the field resultant magnetic flux ϕrt, the amplitude of total resultant magnetic flux ϕtotal, which is the resultant of stator-side magnetic flux ϕs and the field resultant magnetic flux ϕrt, also changes. Here, the stator-side magnetic flux ϕs denotes magnetic flux generated by the harmonic currents flowing in the phase windings31U-31W of the stator coil31. That is, the amplitude of the total resultant magnetic flux ϕtotal can be made lower than the amplitude of the field resultant magnetic flux ϕrt by changing the turn number ratio N2/N1. The difference in phase between the field resultant magnetic flux ϕrt and the stator-side magnetic flux ϕs is, for example, larger than 120° and smaller than 240°.

FIGS. 14A-14Cshow the waveforms of the first magnetic flux ϕ1, the second magnetic flux ϕ2, the field resultant magnetic flux ϕrt, the stator-side magnetic flux ϕs and the total resultant magnetic flux ϕtotal at different ratios of amplitude between ϕ1, ϕ2and ϕs. In the examples shown inFIGS. 14A-14C, the difference in phase between the first magnetic flux ϕ1and the second magnetic flux ϕ2is equal to 180°. In addition, inFIGS. 14A-14C, the vertical axes are graduated to indicate the relationship in magnitude between the magnetic fluxes ϕ1, ϕ2, ϕrt, ϕs and ϕtotal.

FIG. 14Ashows the waveforms of the first magnetic flux ϕ1, the second magnetic flux ϕ2, the field resultant magnetic flux ϕrt, the stator-side magnetic flux ϕs and the total resultant magnetic flux ϕtotal when the ratio of amplitude between ϕ1, ϕ2and ϕs is equal to 1:0.9:1.1 and the difference in phase between the field resultant magnetic flux ϕrt and the stator-side magnetic flux ϕs is equal to 180°. As seen fromFIG. 14A, even though it is aimed to smooth the field resultant magnetic flux ϕrt, it is not always possible to lower the amplitude of the total resultant magnetic flux ϕtotal due to the influence of the stator-side magnetic flux ϕs.

FIG. 14Bshows the waveforms of the first magnetic flux ϕ1, the second magnetic flux ϕ2, the field resultant magnetic flux ϕrt, the stator-side magnetic flux ϕs and the total resultant magnetic flux ϕtotal when the ratio of amplitude between ϕ1, ϕ2and ϕs is equal to 1.5:0.7:1.1 and the difference in phase between the field resultant magnetic flux ϕrt and the stator-side magnetic flux ϕs is equal to 180°. In the example shown inFIG. 14B, it becomes possible to more effectively lower the amplitude of the total resultant magnetic flux ϕtotal than in the example shown inFIG. 14A. Consequently, the amplitude of the total resultant magnetic flux ϕtotal becomes lower than the amplitude of the field resultant magnetic flux ϕrt.

FIG. 14Cshows the waveforms of the first magnetic flux ϕ1, the second magnetic flux ϕ2, the field resultant magnetic flux ϕrt, the stator-side magnetic flux ϕs and the total resultant magnetic flux ϕtotal when the ratio of amplitude between ϕ1, ϕ2and ϕs is equal to 0.7:1.5:1.1 and the difference in phase between the field resultant magnetic flux ϕrt and the stator-side magnetic flux ϕs is equal to 0°. In the example shown inFIG. 14C, since the difference in phase between the field resultant magnetic flux ϕrt and the stator-side magnetic flux ϕs is equal to 0°, the amplitude of the total resultant magnetic flux ϕtotal becomes higher than the amplitude of the field resultant magnetic flux ϕrt.

In view of the above, in the present embodiment, the capacitance C of the capacitor90and the turn number ratio N2/N1are set so that when the harmonic currents are supplied to the phase windings31U-31W of the stator coil31, the amplitude of the total resultant magnetic flux ϕtotal becomes lower than the amplitude of the field resultant magnetic flux ϕrt. Consequently, it becomes possible to lower the torque ripple ratio of the rotating electric machine30. It is well known in the art that harmonic current is superimposed on fundamental current in ordinary PWM controls. The inventor of the present application has first found that the torque ripple ratio of the rotating electric machine30can be made lower than that of an ordinary motor by making torque ripple due to the field resultant magnetic flux ϕrt and torque ripple due to the stator-side magnetic flux ϕs canceled by each other.

Moreover, the inventor of the present application has also found that it is more effective, in terms of securing high torque of the rotating electric machine30, to focus on resonance of the series resonant circuit than on resonance of the parallel resonant circuit. Furthermore, the inventor has also found that it is preferable to set the frequency fs of the harmonic currents supplied to the phase windings31U-31W of the stator coil31to be equal or close to the first resonance frequency f1of the series resonant circuit. The above findings by the inventor will be described in detail hereinafter.

To keep the first resonance frequency f1of the series resonant circuit at a predetermined frequency fcst under the constraint of keeping the turn number sum (N1+N2) (i.e., the sum of the number of turns N1of the first winding71aand the number of turns N2of the second winding71b) at a predetermined turn number Ncst, it is necessary to set the capacitance C of the capacitor90and the turn number ratio N2/N1such that the higher the capacitance C of the capacitor90, the higher the turn number ratio N2/N1. In addition, the predetermined frequency fcst is set to, for example, a value of fs which can be realized by the inverter20and satisfies the inequality (eq3) to be described later. Here, fs is the frequency of the harmonic currents supplied to the phase windings31U-31W of the stator coil31.

When the capacitance C of the capacitor90is varied in the range of 15 μF to 55 μF under the constraint of keeping the turn number sum (N1+N2) at the predetermined turn number Ncst and the first resonance frequency f1at the predetermined frequency fcst, the turn number ratio N2/N1falls within the range of 0.5 to 10.2 as shown inFIG. 15. In addition, inFIG. 15, the capacitance C, the turn number ratio N2/N1, the field current If, the torque and the torque ripple ratio under the above constraint are shown as the case of A-LINE.

The inventor of the present application has confirmed, through experimental investigation, that the torque of the rotating electric machine30can be kept high by setting the turn number ratio N2/N1to be in the range of 0.5 to 5.2 under the above A-line constraint.

FIG. 16Ashows the relationship of the field current If and the torque of the rotating electric machine30to the capacitance C of the capacitor90.FIG. 16Bshows the relationship of the torque ripple ratio of the rotating electric machine30to the capacitance C of the capacitor90.FIG. 17Ashows the relationship of the field current If and the torque of the rotating electric machine30to the turn number ratio N2/N1.FIG. 17Bshows the relationship of the torque ripple ratio of the rotating electric machine30to the turn number ratio N2/N1.

As seen fromFIGS. 15, 16A-16B and 17A-17B, with the first resonance frequency f1of the series resonant circuit kept at the predetermined frequency fcst, the torque of the rotating electric machine30can be kept high by setting the capacitance C of the capacitor90to be in the range of 20 μF to 50 μF and the turn number ratio N2/N1to be in the range of 0.5 to 5.2 (i.e., 20 μF≤C≤50 μF and 0.5≤N2/N1≤5.2).

On the other hand, to keep the second resonance frequency f2of the parallel resonant circuit at the predetermined frequency fcst under the constraint of keeping the turn number sum (N1+N2) at the predetermined turn number Ncst, it is necessary to set the capacitance C of the capacitor90and the turn number ratio N2/N1such that the higher the capacitance C of the capacitor90, the lower the turn number ratio N2/N1. InFIG. 15, the capacitance C, the turn number ratio N2/N1, the field current If and the torque under the above constraint are shown as the case of B-LINE.

As seen fromFIGS. 15, 16A and 17A, in the case of setting the turn number ratio N2/N1to be in the range of 0.7 to 5.2 (i.e., 0.7≤N2/N1≤5.2) under the constraint of keeping the turn number sum (N1+N2) at the predetermined turn number Ncst and the second resonance frequency f2at the predetermined frequency fcst, the torque of the rotating electric machine30becomes highest when the turn number ratio N2/N1is equal to 1.7. Moreover, with the turn number ratio N2/N1deviating from 1.7, the torque of the rotating electric machine30considerably drops.

When the turn number ratio N2/N1is set to be in the range of 0.5 to 5.2, the torque of the rotating electric machine30tends to be lower under the B-line constraint (i.e., the constraint of keeping the second resonance frequency f2at the predetermined frequency fcst) than under the A-line constraint (i.e., the constraint of keeping the first resonance frequency f1at the predetermined frequency fcst). This is because the first winding71ais radially located closer than the second winding71bto the stator50and thus more susceptible than the second winding71bto the magnetic flux generated by the harmonic currents flowing in the phase windings31U-31W of the stator coil31.

Accordingly, the torque of the rotating electric machine30can be kept higher by setting the capacitance C and the turn number ratio N2/N1under the A-line constraint pertaining to the series resonant circuit than under the B-line constraint pertaining to the parallel resonant circuit. Moreover, under the A-line constraint, the capacitance C can be set within the range of 20 μF to 50 μF and the turn number ratio N2/N1can be set within the range of 0.5 to 5.2; therefore, the degree of freedom of setting the capacitance C and the turn number ratio N2/N1is improved.

As seen fromFIGS. 16B and 17B, with the capacitance C set to be in the range of 20 μF to 50 μF and the turn number ratio N2/N1set to be in the range of 0.7 to 5.2, it is possible to effectively lower the torque ripple ratio while keeping the torque of the rotating electric machine30high.

In the present embodiment, the frequency fs of the harmonic currents supplied to the phase windings31U-31W of the stator coil31is set to satisfy the following inequality (eq3). Moreover, the controller40controls the inverter20to supply the phase windings31U-31W of the stator coil31with the harmonic currents having the set frequency fs.

The above inequality (eq3) is derived as follows. The inequality (0.5≤N2/N1≤5.2) can be transformed into (0.19≤N1/N2≤2). Moreover, for the sake of simplicity, (0.20≤N1/N2≤2) is used instead of (0.19≤N1/N2≤2). Further, (0.20×N2≤N1≤2×N2) is derived from (0.20≤N1/N2≤2). Moreover, the inductance L1of the first winding71aand the inductance L2of the second winding71bcan be expressed by the following equations:
L1=μ×N12×(S1/l1); and
L2=μ×N22×(S2/l2),
where μ is the magnetic permeability, N1is the number of turns of the first winding71a, S1and l1are respectively the cross-sectional area and length of a magnetic path of the first winding71a, N2is the number of turns of the second winding71b, S2and l2are respectively the cross-sectional area and length of a magnetic path of the second winding71b.

Substituting the equations of the inductances L1and L2respectively into the above-described equations (eq1) and (eq2), the following equations can be derived:
f1=1/(2×π×√{square root over (μ×N12×(S1/l1)×C))}=A/(2×π×√{square root over (N12×C))}; and
f2=1/(2×π×√{square root over (μ×N23×(S2/l2)×C))}=A/(2×π×√{square root over (N22×C))},
where A is a predetermined coefficient.

In addition, the predetermined coefficient A can be expressed by the following equation:
A=1/√{square root over (μ×(S1/l1))}=1/√{square root over (μ×(S2/l2))}

Furthermore, in the present embodiment, the frequency fs of the harmonic currents supplied to the phase windings31U-31W of the stator coil31is set to be equal to the first resonance frequency f1of the series resonant circuit, i.e., fs=f1=A/(2×π×√{square root over (N12×C)}).

Then, the above inequality (eq3) is derived from the above equation of fs and the inequality of (0.20>N2≤N1≤2×N2).

Moreover, the following inequality (eq5) is derived by substituting the following equation (eq4) into the above inequality (eq3).

According to the present embodiment, it is possible to achieve the following advantageous effects.

In the present embodiment, the field coil type rotating electric machine30includes the stator50, the field winding70and the rotor60. The stator50includes the stator coil31that is comprised of the U-phase winding31U, the V-phase winding31V and the W-phase winding31W. The field coil70includes the serially-connected winding pair consisting of the first and second windings71aand71bthat are connected in series with each other. The rotor60includes the rotor core61and the main pole portions62that are formed at predetermined intervals in the circumferential direction and each radially protrude from the rotor core61. Each of the first and second windings71aand71bof the field coil70is wound on each of the main pole portions62of the rotor60. Each of the phase windings31U-31W of the stator coil31is configured to be supplied with harmonic current to induce the field current If in the field coil70. The field coil type rotating electric machine30further includes the diode80and the capacitor90. The diode80has its cathode connected to the first-winding-side end of the serially-connected winding pair and its anode connected to the second-winding-side end of the serially-connected winding pair. The capacitor90is connected in parallel with the second winding71b. In the field coil type rotating electric machine30, there are formed both the series resonant circuit including the first winding71aand the capacitor90and the parallel resonant circuit including the second winding71band the capacitor90. The first winding71ais radially located closer than the second winding71bto the stator50. The capacitance C of the capacitor90and the turn number ratio N2/N1, which is the ratio of the number of turns N2of the second winding71bto the number of turns N1of the first winding71a, are set to have the amplitude of the total resultant magnetic flux ϕtotal lower than the amplitude of the field resultant magnetic flux ϕrt when the harmonic currents are supplied to the phase windings31U-31W of the stator coil31. The total resultant magnetic flux ϕtotal is the resultant of the field resultant magnetic flux ϕrt and the stator-side magnetic flux ϕs generated by the harmonic currents flowing in the phase windings31U-31W of the stator coil31. The field resultant magnetic flux ϕrt is the resultant of the first magnetic flux ϕ1generated by the harmonic current flowing in the first winding71aand the second magnetic flux ϕ2generated by the harmonic current flowing in the second winding71b.

With the above configuration, upon the harmonic currents flowing in the phase windings31U-31W of the stator coil31, harmonic currents are induced to flow respectively in the first and second windings71aand71bof the field coil70. The difference in phase between the harmonic current flowing in the series resonant circuit including the first winding71aand the harmonic current flowing in the parallel resonant circuit including the second winding71bis, for example, larger than 120° and smaller than 240°. Moreover, with change in the turn number ratio N2/N1, both the amplitude of the first magnetic flux ϕ1generated by the harmonic current flowing in the first winding71aand the amplitude of the second magnetic flux ϕ2generated by the harmonic current flowing in the second winding71bchange. Consequently, the amplitude of the field resultant magnetic flux ϕrt, which is the resultant of the first magnetic flux ϕ1and the second magnetic flux ϕ2, also changes. Furthermore, with change in the amplitude of the field resultant magnetic flux ϕrt, the amplitude of the total resultant magnetic flux ϕtotal, which is the resultant of the field resultant magnetic flux ϕrt and the stator-side magnetic flux ϕs generated by the harmonic currents flowing in the phase windings31U-31W of the stator coil31, also changes. That is, the amplitude of the total resultant magnetic flux ϕtotal can be made lower than the amplitude of the field resultant magnetic flux ϕrt by changing the turn number ratio N2/N1. In view of the above, in the present embodiment, the capacitance C of the capacitor90and the turn number ratio N2/N1are set so that when the harmonic currents are supplied to the phase windings31U-31W of the stator coil31, the amplitude of the total resultant magnetic flux ϕtotal becomes lower than the amplitude of the field resultant magnetic flux ϕrt. Consequently, it becomes possible to lower the torque ripple ratio of the rotating electric machine30.

Moreover, in the present embodiment, the turn number ratio N2/N1is set to be in the range of 0.5 to 5.2 with the capacitance C of the capacitor90set to be in the range of 20 μF to 50 μF under the constraint of keeping the first resonance frequency f1of the series resonant circuit at the predetermined frequency fcst and the turn number sum (N1+N2) at the predetermined turn number Ncst. In other words, the first resonance frequency f1of the series resonant circuit is set to the predetermined frequency fcst by setting the capacitance C of the capacitor90to be in the range of 20 μF to 50 μF and the turn number ratio N2/N1to be in the range of 0.5 to 5.2. In addition, upon the setting of the first resonance frequency f1, the second resonance frequency f2of the parallel resonant circuit is also set accordingly.

More specifically, to keep the first resonance frequency f1of the series resonant circuit including the first winding71aat the predetermined frequency fcst under the constraint of keeping the turn number sum (N1+N2) at the predetermined turn number Ncst, it is necessary to set the capacitance C of the capacitor90and the turn number ratio N2/N1such that the higher the capacitance C of the capacitor90, the higher the turn number ratio N2/N1. The inventor of the present application has confirmed, through experimental investigation, that when the capacitance C of the capacitor90is varied in the range of 20 μF to 50 μF under the A-line constraint (i.e., the constraint of keeping the first resonance frequency f1of the series resonant circuit at the predetermined frequency fcst and the turn number sum (N1+N2) at the predetermined turn number Ncs), the turn number ratio N2/N1falls within the range of 0.5 to 5.2. Moreover, the inventor has also confirmed, through experimental investigation, that the torque of the rotating electric machine30can be kept high by setting the turn number ratio N2/N1to be in the range of 0.5 to 5.2 under the A-line constraint.

On the other hand, to keep the second resonance frequency f2of the parallel resonant circuit including the second winding71bat the predetermined frequency fcst under the constraint of keeping the turn number sum (N1+N2) at the predetermined turn number Ncst, it is necessary to set the capacitance C of the capacitor90and the turn number ratio N2/N1such that the higher the capacitance C of the capacitor90, the lower the turn number ratio N2/N1. The inventor has confirmed, through experimental investigation, that: in the case of setting the turn number ratio N2/N1to be in the range of 0.7 to 5.2 under the B-line constraint (i.e., the constraint of keeping the second resonance frequency f2of the parallel resonant circuit at the predetermined frequency fcst and the turn number sum (N1+N2) at the predetermined turn number Ncst), the torque of the rotating electric machine30becomes highest when the turn number ratio N2/N1is equal to 1.7; with the turn number ratio N2/N1deviating from 1.7, the torque of the rotating electric machine30considerably drops. Moreover, the inventor has also confirmed, through experimental investigation, that when the turn number ratio N2/N1is set to be in the range of 0.5 to 5.2, the torque of the rotating electric machine30tends to be lower under the B-line constraint than under the A-line constraint. This is because the first winding71ais radially located closer than the second winding71bto the stator50and thus more susceptible than the second winding71bto the stator-side magnetic flux ϕs generated by the harmonic currents flowing in the phase windings31U-31W of the stator coil31.

Accordingly, the torque of the rotating electric machine30can be kept higher by setting the capacitance C of the capacitor90and the turn number ratio N2/N1under the A-line constraint pertaining to the series resonant circuit than under the B-line constraint pertaining to the parallel resonant circuit. Moreover, under the A-line constraint, the capacitance C can be set within the range of 20 μF to 50 μF and the turn number ratio N2/N1can be set within the range of 0.5 to 5.2; therefore, the degree of freedom of setting the capacitance C and the turn number ratio N2/N1is improved.

In view of the above, in the present embodiment, the first resonance frequency f1of the series resonant circuit is set to the predetermined frequency fcst by setting the capacitance C of the capacitor90to be in the range of 20 μF to 50 μF and the turn number ratio N2/N1to be in the range of 0.5 to 5.2. Consequently, it becomes possible to improve the degree of freedom of setting the capacitance C and the turn number ratio N2/N1to keep the torque of the rotating electric machine30high. For example, the turn number ratio N2/N1may be set to be in the range of 0.7 to 5.2. In this case, it is possible to effectively lower the torque ripple ratio while keeping the torque of the rotating electric machine30high.

Modification of First Embodiment

Each of the first and second windings71aand71bof the field coil70may be formed of a rectangular conductor wire (i.e., an electrical conductor wire having a rectangular cross-sectional shape). In this case, it is possible to improve the space factor of the field coil70, thereby improving the efficiency of the field coil type rotating electric machine30. Moreover, in this case, adjacent portions of the first and second windings71aand71bof the field coil70are in surface contact with each other; consequently, when the centrifugal force is applied to the windings71aand71b, it is possible to lower the load acting between adjacent portions of the windings71aand71b, thereby preventing damage to insulating coats of the windings71aand71b. Furthermore, in this case, it is possible to improve the ampere-turn (AT) of the field coil70, thereby broadening the excitation range of the field coil70. As a result, it is possible to improve the torque controllability of the field coil type rotating electric machine30.

In addition, each of the first and second windings71aand71bof the field coil70may be constituted of an α winding of a rectangular conductor wire, such as one shown inFIG. 5(A)of Japanese Patent Application Publication No. JP 2008-178211 A.

Second Embodiment

As shown inFIG. 18, in a field coil type rotating electric machine30according to the second embodiment, in the rotor60, there are provided partitioning members100between the first and second windings71aand71bof the field coil70; the partitioning members100are formed of a soft-magnetic material. Each of the partitioning members100is, for example, ring-shaped and has one of the main pole portions62of the rotor60inserted in a center hole thereof. Moreover, when viewed along the axial direction, each of the partitioning members100has an elongate shape extending in the circumferential direction. With the partitioning members100interposed between the first and second windings71aand71bof the field coil70, the two windings71aand71bare radially separated from each other. In addition, the partitioning members100have a smaller radial thickness than each of the first and second windings71aand71b.

Moreover, as shown inFIG. 19, each of the partitioning members100may be formed of a plurality of sheets that are made of a soft-magnetic material (e.g., magnetic steel) and laminated in a radial direction. With the above configuration, it is possible to lower eddy current loss in the partitioning members100. In addition, with the sheets being laminated in the radial direction, it is possible to set the radial thickness of the partitioning members100to a small value according to the thickness of the sheets while securing the circumferential length of the partitioning members100.

In the present embodiment, with the partitioning members100interposed between the first and second windings71aand71bof the field coil70, most of the leakage magnetic flux flows through the partitioning members100, not through the field coil70. Consequently, it becomes difficult for voltages of opposite polarities to be induced respectively in the first and second windings71aand71b; it also becomes difficult for voltages of opposite polarities to be induced respectively in different parts of each of the first and second windings71aand71b. As a result, it becomes possible to increase electric current induced in each of the first and second windings71aand71bin each of the four patterns shown inFIG. 8.

While the above particular embodiments have been shown and described, it will be understood by those skilled in the art that various modifications, changes, and improvements may be made without departing from the spirit of the present disclosure.

(1) In the above-described embodiments, the field coil70is constituted of the serially-connected winding pair consisting of the first and second windings71aand71bthat are connected in series with each other.

As an alternative, as shown inFIG. 20, the field coil70may be constituted of a serially-connected winding set consisting of a first winding71a, a second winding71band a third winding71cthat are connected in series with each other. Each of the first to the third windings71a-71cis wound on each of the main pole portions62of the rotor60. More specifically, on each of the main pole portions62, the first to the third windings71a-71care wound so that: the first winding71ais located radially outermost (i.e., closest to the stator50); the third winding71cis located radially innermost (i.e., furthest from the stator50); and the second winding71bis radially interposed between the first winding71aand the third winding71c. Moreover, on each of the main pole portions62, the first to the third windings71a-71care wound in the same direction. Furthermore, for each circumferentially-adjacent pair of the main pole portions62, the winding direction of the first to the third windings71a-71con one of the main pole portions62of the circumferentially-adjacent pair is opposite to the winding direction of the first to the third windings71a-71con the other of the main pole portions62of the circumferentially-adjacent pair. Consequently, the magnetization directions of the main pole portions62of the circumferentially-adjacent pair are opposite to each other.

FIG. 21shows an electric circuit formed in the rotor60that has the first to the third windings71a-71cof the field coil70wound on the same main pole portions62.

In the rotor60, there is further provided a second capacitor91in addition to the capacitor90(hereinafter, to be referred to as the first capacitor90). A first end of the third winding71cis connected with the second end of the second winding71b. A second end of the third winding71cis connected with the anode of the diode80. The second capacitor91is connected in parallel with the third winding71c. In addition, inFIG. 21, L3represents the inductance of the third winding71cand C1and C2respectively represent the capacitances of the first and second capacitors90and91.

The first winding71aof the field coil70, the first capacitor90and the diode80together form a first series resonant circuit. The first series resonant circuit has a resonance frequency which will be referred to as the first resonance frequency f1hereinafter; the first resonance frequency f1can be calculated by the equation (eq1) described in the first embodiment. The second winding71bof the field coil70and the first capacitor90together form a first parallel resonant circuit. The first parallel resonant circuit has a resonance frequency which will be referred to as the second resonance frequency f2hereinafter; the second resonance frequency f2can be calculated by the equation (eq2) described in the first embodiment. The first and second windings71aand71bof the field coil70, the second capacitor91and the diode80together form a second series resonant circuit. The second series resonant circuit has a resonance frequency which will be referred to as the third resonance frequency f3hereinafter; the third resonance frequency f3can be calculated by the following equation (eq6). The third winding71cof the field coil70and the second capacitor91together form a second parallel resonant circuit. The second parallel resonant circuit has a resonance frequency which will be referred to as the fourth resonance frequency f4hereinafter; the fourth resonance frequency f4can be calculated by the following equation (eq7).

The second series resonant circuit and the second parallel resonant circuit function similarly to the first series resonant circuit and the first parallel resonant circuit.

With the above configuration, when the frequency fs of the harmonic currents supplied to the phase windings31U-31W of the stator coil31deviates from a given frequency (e.g., the first resonance frequency f1), if the frequency fs is equal or close to the third resonance frequency f3or the fourth resonance frequency f4, it is still possible to increase the field current flowing in the field coil70.

In addition, the frequency fs of the harmonic currents supplied to the phase windings31U-31W of the stator coil31may deviate from a given frequency when the electrical angular frequency of the rotating electric machine30is high. This is because the higher the electrical angular frequency, the smaller the number M of cycles of the harmonic currents allowed to be superimposed per period of the fundamental currents (here, M is a natural number) and thus the larger the variation in the frequency fs when the number of cycles of the harmonic currents superimposed per period of the fundamental currents is changed from M to (M−1). For example, when the number M is changed between 4 and 3, the variation in the frequency fs is about 30%. Here, “M=3” represents that for each of the phase currents of the stator coil31, the number of cycles of the harmonic current superimposed the fundamental current of the phase current per period of the fundamental current is equal to 3; and 3 is considered to be the minimum value of M which can be used to induce the field current in the field coil70.

(2) As shown inFIG. 22, the field coil70may also be constituted of a serially-connected winding set consisting of (n+1) windings that are connected in series with each other, where n is a natural number greater than or equal to 3. In this case, the number of the capacitors included in the electric circuit formed in the rotor60is equal to n.

(3) As shown inFIGS. 23A and 23B, a capacitor100may be connected in parallel with the first winding71a. In this case, the second winding71b, the capacitor100and the diode80together form a series resonant circuit; the first winding71aand the capacitor100together form a parallel resonant circuit. In addition, in this case, the second winding71bis radially located closer than the first winding71ato the stator50. In other words, on each of the main pole portions62of the rotor60, the first winding71ais wound on the radially inner side (i.e., the non-stator side) while the second winding71bis wound on the radially outer side (i.e., the stator side).

In the electric current shown inFIGS. 23A and 23B, the capacitance C of the capacitor100and the turn number ratio N2/N1(i.e., the ratio of the number of turns N2of the second winding71bto the number of turns N1of the first winding71a) may be set so that when the harmonic currents are supplied to the phase windings31U-31W of the stator coil31, the amplitude of the total resultant magnetic flux ϕtotal becomes lower than the amplitude of the field resultant magnetic flux ϕrt. Here, as described in the first embodiment, the total resultant magnetic flux ϕtotal is the resultant of the field resultant magnetic flux ϕft and the stator-side magnetic flux ϕs generated by the harmonic currents flowing in the phase windings31U-31W of the stator coil31. The field resultant magnetic flux ϕrt is the resultant of the first magnetic flux ϕ1generated by the harmonic current flowing in the first winding71aand the second magnetic flux ϕ2generated by the harmonic current flowing in the second winding71b.

(4) In the above-described embodiments, the rotating electric machine30is of an inner rotor type where the rotor60is arranged radially inside the stator50.

As an alternative, the rotating electric machine30may be of an outer rotor type where a rotor is arranged radially outside a stator. In this case, the rotor may include a rotor core and main pole portions which each protrude radially inward from the rotor core and are spaced at predetermined intervals in the circumferential direction.

(5) In the above-described embodiments, the field coil70is formed of aluminum wires. Alternatively, the field coil70may be formed of other materials, such as copper wires or CNTs (Carbon Nanotubes).

Moreover, in the above-described embodiments, the field coil70is formed by compression shaping. Alternatively, the field coil70may be formed without compression shaping.