Integrated magnetic resonant power converter

A resonant power converter is disclosed which is constructed on an integrated magnetic core, with primary, secondary and tertiary windings occupying separate legs of the core. The tertiary winding is connected in a resonant circuit and induces a flux in the primary leg that causes the primary winding current to assume the shape of a series of generally sinusoidal pulses. Primary winding switching can thus occur at zero primary current between pulses, thereby eliminating prior interference problems. Furthermore, the converter can be operated in a pulse width modulated mode to accommodate for varying output load levels without the problems of low frequency operation encountered by prior frequency modulated resonant converters.

BACKGROUND OF THE INVENTION 
1. Field of the Invention 
This invention relates to DC-to-DC power converters, and more particularly 
to resonant power converters. 
2. Description of the Prior Art 
Various DC-to-DC power converters are available for transforming an input 
DC voltage of one magnitude to an output DC voltage with a different 
magnitude. Two conventional converter topologies are referred to as the 
flyback and the forward converters. They are discussed, for example, in a 
text by George Chryssis, "High-Frequency Switching Power Supplies: Theory 
and Design", McGraw-Hill Book Co., 1984, pages 11-13. 
With a flyback converter, a switch is connected in series with the input 
winding of a transformer. The switch is alternately turned on and off, 
producing a pulsing in the secondary winding which is fed through a diode 
to charge an output capacitor. When the primary current is switched off, 
the current in the secondary tends to surge. The rate of change of both 
the primary and secondary currents are very high, leading to 
electromagnetic interference and radio frequency interference. Complex 
filters are required to suppress the interference, thereby increasing the 
complexity and cost of the system and reducing its efficiency. 
In the forward converter design an inductor is added to the secondary 
circuit to reduce the absolute current magnitude in the secondary, while a 
second diode in the secondary circuit closes a circuit between the output 
capacitor and inductor when the input switch is off. This design uses a 
high input current, which is stressful for the switching transistor in the 
primary circuit. The output diode is stressed by large voltage and current 
swings, requiring a snubber circuit which adds to the cost and complexity 
of the system and is an interference source. The large rates of current 
change in the transformer windings and in the inductor produce 
electromagnetic and radio frequency interference, which again require 
complex filters to remove. 
Many of the problems associated with flyback and forward converter designs 
are resolved by the more recent "resonant" converter, which is exemplified 
in U.S. Pat. No. 4,415,959 to Vinciarelli. In this type of device, the 
most relevant embodiment of which is shown in FIG. 4 of the patent, a 
relatively large inductor acts as a current sink in the secondary circuit. 
A capacitor in the secondary circuit cooperates with the leakage 
inductance of the transformer to establish an effective LC circuit; this 
defines a characteristic time scale for the rise and fall of current from 
the DC voltage source. A switch device in the primary circuit can thus be 
switched on and off at essentially zero current, thereby overcoming the 
problems in both the flyback and forward converters associated with 
switching under high current levels. Following each cycle the energy 
stored in the capacitor is released by the current sink. After the 
capacitor has been discharged, the sink current is carried by a diode 
connected in parallel with the capacitor. 
While this type of resonant converter solves the interference problems 
associated with the flyback and forward converters, it itself has certain 
limitations. Its output power is modulated by operating the converter over 
a very wide frequency range, thus forcing the converter to operate at very 
high frequencies at full power if a wide dynamic range of minimum to 
maximum output power is required. Attempts to operate the power supply at 
low frequency for full output power preclude operation at low power 
levels. Additionally, peak currents in the switching elements increase 
with increasing input voltage. When the input voltage is at its maximum 
design rank, peak current stresses on the switching elements become 
severe. Moreover, the design does not lend itself well to multiple 
outputs. 
SUMMARY OF THE INVENTION 
In view of the above problems with the prior art, the object of the present 
invention is to provide a novel type of voltage converter in which primary 
side switching takes place at very low or zero currents, thus eliminating 
the interference problems encountered with the flyback and forward 
converter designs, and yet operates over a more narrow frequency range 
under varying power outputs. 
Another object of the invention is to provide a novel resonant converter in 
which the primary current is given a substantially sinusoidal shape 
through the operation of flux linkages in the transformer despite abrupt 
on-off switching of the primary circuit, and in which the primary current 
flow is terminated as a result of the flux linkages rather than the 
primary switch. 
Other objects include decreased cost, increased efficiency and reliability, 
relative simplicity in design and the capability of multiple outputs. 
These and other objects of the invention are realized in a novel resonant 
power converter construction which employs an integrated magnetic 
transformer core. The core includes at least three legs which are 
magnetically coupled with each other. Primary, secondary and tertiary 
windings are provided on the first, second and third legs of the core, 
respectively. The primary winding is connected to receive a DC supply 
voltage, the secondary winding is connected to an output terminal, and the 
tertiary winding is connected with a capacitor to form a resonant circuit. 
Current flow through the primary winding is established in a pulsed 
fashion by means of a switch and associated switch control, and the flux 
coupling between the windings. The rise and fall of each primary current 
pulse is gradual rather than abrupt, causing each pulse to assume a 
generally sinusoidal shape. Each primary current pulse rises initially 
because of the primary winding's leakage inductance, while the tertiary 
winding induces a flux in the primary winding which gradually diminishes 
and eventually terminates the current pulses. 
The switch control responds to the output load to control the timing of 
current initiation through the primary winding at the beginning of each 
cycle. This permits the power output of the converter to be controlled by 
means of frequency modulation, phase modulation, or pulse width 
modulation. The timing of the beginning of each pulse is controlled by the 
switch in response to the load, while the termination of each pulse is 
controlled by the resonant winding through a coupling flux in the core. 
The prior problems of wide frequency range operation are thus completely 
avoided. 
Various embodiments are discussed, including designs in which the tertiary 
winding and resonant capacitor are connected together either by themselves 
or with the secondary winding in a resonant circuit, in which the resonant 
circuit is connected to the primary rather than the secondary side of the 
converter, and in which dual switches and windings are provided for 
full-wave rectification. The tertiary winding is preferably connected to 
an output capacitor out-of-phase with the secondary winding, so as to 
reduce the peak current applied to the output capacitor to a level below 
the peak secondary winding current. 
These and other features and advantages of the invention will be apparent 
to those skilled in the art from the following detailed description of 
preferred embodiments, taken together with the accompanying drawings, in 
which:

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS 
FIGS. 1 and 2 show two embodiments of DC-to-DC power converters constructed 
in accordance with the invention, with FIG. 1 representing a basic version 
and FIG. 2 the more preferred version. In both cases the converter is 
formed on an integrated magnetic core 2 having at least three legs. 
Integrated magnetic cores per se are known in other applications, but they 
have not previously been used in the resonant DC to DC converter field. 
The novel employment of integrated magnetic cores in this context has been 
coupled with a new converter design that permits a dramatically different 
mode of operation with considerably enhanced results. 
The integrated magnetic core 2 consists of a magnetic core with at least 
three different legs which are magnetically coupled with each of the other 
legs to provide flux paths through each different pair of legs. As shown 
in the figures, the core consists of three legs labeled P, S and T for 
primary, secondary and tertiary, respectively. As illustrated, the primary 
leg is in the middle with the secondary and tertiary legs on the outside, 
but the positions of the legs are not critical; the primary leg could be 
located on the outside and either the secondary or tertiary legs in the 
middle. The integrated magnetic core is preferably formed by joining two 
"E" cores together, and is designed so that the coupling between each pair 
of legs is relatively loose. A specific exemplary design is discussed in 
detail hereinafter. 
In both embodiments primary, secondary and tertiary windings 4, 6 and 8 are 
wound about the corresponding core legs P, S and T, respectively. Primary 
winding 4 is connected on one side to an input DC voltage supply terminal 
10, and on the other side to a switch 12 which completes the circuit when 
closed. The secondary winding is connected on one side through a diode D1 
to an output terminal 14, which provides a regulated DC output. Although 
illustrated as a diode, D1 could be implemented as any switch having a 
suitable control to provide rectification. Operation as a multi-quadrant 
or bi-directional converter is also possible with the addition of suitable 
additional switching elements, rectifiers, and controls. An output 
capacitor C.sub.0 is connected between the output terminal 14 and the 
other side of the secondary winding 6, which is also referenced to ground. 
The output capacitor C.sub.0 smooths output pulses received from the 
secondary winding, as discussed further below, to provide a DC output 
voltage at terminal 14. 
The tertiary winding 8 is connected across a resonant capacitor C.sub.R to 
form an LC resonant circuit. The tertiary core leg T has an air gap 16 of 
sufficient size to store enough energy in the resonant circuit to generate 
an appropriate induced flux in the primary winding 4, as discussed below, 
without magnetically saturating the core. 
The resonant circuit serves two important functions. First, it builds up a 
flux in the tertiary winding 8 which is reflected back through the 
integrated magnetic core to the primary winding 4 to terminate primary 
current pulses in a gradual manner. This permits the primary circuit to be 
switched at essentially zero current, and thereby avoid the interference 
problems associated with switching under load. Second, it establishes a 
substantially constant frequency of operation for the circuit, where the 
frequency range required for regulation is much less than that required 
for previous designs, thus avoiding the problems of wide frequency range. 
To briefly summarize the operation of the FIG. 1 design, switch 12 is 
initially switched closed to initiate a cycle. With the integrated 
magnetic configuration shown, the primary current is initially determined 
principally by the leakage inductance as referred to the primary winding 
4. Since the leakage inductance is through low permeability, high 
reluctance air, the primary winding current initially increases at a 
relatively rapid rate. As the cycle proceeds, flux from the resonant 
circuit is coupled through the integrated magnetic core back to the 
primary leg P. The coupled flux presents less and less opposition to the 
flux induced in the primary leg by the primary winding current, and 
eventually reinforces the latter flux. This causes the primary current to 
first reduce its rate of increase, then peak, and finally to diminish back 
to zero in a generally sinusoidal pulse. 
The current in secondary winding 6 tracks the primary winding current in 
the normal transformer fashion, and produces a similarly sinusoidal shaped 
current pulse which is transmitted through diode D1 and charges the output 
capacitor C.sub.0. After the end of the primary current pulse the switch 
12 is opened, and remains open until the beginning of the next cycle, when 
the operation is repeated. In this manner a series of pulses are applied 
to the output capacitor C.sub.0 to hold it at the desired voltage level 
for the load at output terminal 14. As described hereinafter, the duration 
of the pulse is automatically adjusted in response to the load to regulate 
the energy transfer in a pulse width modulation scheme. 
The preferred circuit of FIG. 2 is similar to that of FIG. 1, but in this 
embodiment the resonant capacitor C.sub.R is connected between tertiary 
winding 8 and output terminal 14, rather than in a closed resonant circuit 
solely with tertiary winding 8. In this configuration, as with FIG. 1, the 
tertiary winding 8 is oriented so that the current is out of phase with 
the secondary winding 6. In this manner the resonant circuit draws off 
some of the output current from the secondary winding when that winding is 
conductive, and delivers a similar amount of current back to the output 
capacitor C.sub.0 when the secondary winding is not conductive; diode D1 
prevents the resonant circuit current from flowing to the secondary 
winding. 
The secondary winding when conductive thus contributes current to the 
resonant circuit, and thereby in effect forms a part of that circuit. The 
result is to reduce the peak current applied to the output capacitor 
C.sub.0 to a level less than the peak secondary winding current, without 
diminishing the energy transfer to C.sub.0. This helps to avoid stress on 
C.sub.0. 
The amount of load current that is allowed to flow in the primary and 
secondary windings, and hence the power processing capability of the 
converter, depends upon the amount of steady-state or continuous energy 
stored in the resonant circuit. The greater the steady-state energy stored 
in the resonant circuit, the more current will flow during each switch 
cycle, and the more energy will be processed by the converter. 
It is not critical to the invention which winding is on which leg of the 
transformer core. The invention will function as intended with any winding 
on any of the legs, so long as the primary, secondary and tertiary 
windings are on separate legs. The integrated magnetic core must have a 
minimum of three legs, but may theoretically have any number more than 
three. Multiple secondary windings may be wound on the secondary leg to 
provide multiple outputs. 
FIG. 3 provides a more detailed schematic diagram of the preferred 
embodiment illustrated in FIG. 2. Common elements in the two figures are 
indicated by the same reference numerals. It can be seen from the dotted 
winding convention that the primary and secondary windings 4 and 6 are 
wound with the same polarity, while tertiary winding 8 in the resonant 
circuit is wound 180.degree. out of phase. 
The primary winding switch is implemented as a unidirectional bipolar 
transistor Q1 with its collector-emitter circuit in series with the 
primary winding 4. A switch control circuit 24 has a control output 
connected to the base of Q1 to alternately enable and disable the 
transistor from conducting primary current. 
The input voltage supply is schematically indicated as a battery 26. In 
actual practice the input DC voltage is typically provided by a full-wave 
rectifier supplied from the house voltage. Energy storage capacitors C1 
and C2 are connected in series across the input voltage supply 26. Two 
capacitors C1 and C2 would be used only where a standard voltage doubler 
is present. Otherwise, a single capacitor of any suitable value to smooth 
transients could be employed. The primary leg includes a reset winding 28 
which is connected on one side through a diode D2 to the junction of input 
supply 26 and primary winding 4, and on its other side to ground. The 
reset winding is a conventional technique which is useful in resetting the 
primary leg of the transformer due to any remaining magnetizing flux which 
is present at the end of each primary conductive cycle. 
The tertiary leg includes a resonant circuit clamp winding 30 which is 
connected on one side through diode D3 to the same common junction as D2 
with voltage supply 26 and primary winding 4, and on its other side to 
ground. The clamp winding allows excess energy in the resonant circuit to 
be removed, and also allows the establishment of a defined operating point 
for the resonant circuit. Such a defined operating point simplifies the 
control requirements when the system is operated closed loop. 
In closed loop operation the timing cycle for turning switch transistor Q1 
on and off is controlled in accordance with the output load at output 
terminal 14. The period within each cycle during which the primary winding 
switch is closed is regulated by the switch control circuit 24. The switch 
is closed for longer periods of time under heavy output loads, producing a 
greater primary current flow and energy transfer to the output, and for 
shorter periods of time during light output loads. 
Since the secondary and tertiary windings 6 and 8 are oriented out of 
phase, the currents through these windings will likewise be out of phase. 
This situation is deliberately arranged to reduce the current loading on 
the output capacitor C.sub.0, and yet achieve the same energy transfer. 
The secondary winding current I.sub.S, tertiary winding current I.sub.T, 
and the net current delivered to the output capacitor I.sub.O are 
illustrated in FIG. 4. During each I.sub.S pulse delivered to the output 
capacitor, I.sub.T is drawing current away from the capacitor. The result 
is a smaller net current I.sub.O delivered to the output capacitor. During 
the next phase of operation I.sub.S is 0 (Q1 is open), while I.sub.T goes 
positive to deliver current to the output capacitor. During the entire 
cycle the same energy is thus delivered to C.sub.0, but the maximum net 
current I.sub.O to the capacitor is reduced from the peak value of 
I.sub.S. The peak I.sub.T is preferably about one-third the magnitude of 
the peak I.sub.S. 
In one specific implementation of the FIG. 3 arrangement, a 320 volt DC 
input supply was obtained from a full-wave rectified 115 volt AC line, 
using a voltage doubler and providing energy storage capacitors C1 and C2 
as 470 microfarad, 200 volt electrolytic devices. An EE55 ferrite core was 
used with its center leg machined down to the same cross-section as the 
outside legs. Each leg of the core had an effective area of about 1.7 
cm.sup.2. The primary winding consisted of 120 turns of 19 Ga single 
strand; its reset winding consisted of 60 turns of 26 Ga single strand 
wound bifilar with the first 60 turns of the primary winding. Two turns of 
14 Ga litz, 2 strand was used for the secondary winding. A 0.3 inch gap 
was formed in the tertiary leg for the resonant circuit. The tertiary 
winding consisted of 9 turns of 14 Ga litz, 2 strands; its clamp winding 
was 50 turns of 24 Ga single strand wound alongside the tertiary winding. 
The tertiary winding had a measured inductance of 7 microhenries. A 4 
microfarad, polypropylene film, low ESR pulse capacitor was used to 
implement C.sub.R. Together with the 7 microhenry tertiary winding, this 
established a resonant frequency of about 30 KHz. Output capacitor C.sub.0 
was implemented as four 1,000 microfarad/10 volt rating electrolytic 
capacitors in series with a 2 microhenry air inductor. The output 
rectifier D1 was a single Schottky diode rated at 55 amps. A nominal 100 
watt, 5 volt, 20 amp output was produced. 
Although the input and output sections of the converter of FIG. 3 are 
illustrated as having a common ground connection, this is not required. 
The input and output can have separate ground connections to achieve the 
well-known isolated off-line type of converter. In this case, the tank 
clamp winding 30 would have sufficient isolation dielectric from the tank 
winding itself to fulfill the isolation requirements. 
A more detailed schematic diagram of the embodiment illustrated in FIG. 1 
is provided in FIG. 5. This embodiment is similar to that of FIG. 3, 
except the resonant circuit is restricted to a closed loop consisting of 
tertiary winding 8 and capacitor C.sub.R. 
A different embodiment in which the resonant circuit is tied into the 
primary side ground is shown in FIG. 6. This approach offers an operation 
equivalent to that of FIG. 5, in which the resonant circuit is tied into 
the ground on the secondary side. 
The various embodiments discussed thus far are single-ended, meaning they 
produce a single rectified current pulse in the primary and secondary 
windings for each complete cycle of operation. A circuit which enables 
double-ended operation equivalent to full-wave rectification, with a pair 
of primary and secondary current pulses for each cycle, is shown in FIG. 
7. Two primary windings 4A and 4B are provided on the primary leg, while 
two secondary windings 6A and 6B are provided on the secondary leg. The 
primary windings are connected in series with respective switch 
transistors Q1A and Q1B. These transistors are controlled in an 
alternating switching pattern by switch control 24' such that Q1A is off 
when Q1B is on, and vice versa. A center tap in the secondary winding is 
grounded, while secondary windings 6A and 6B are connected to the output 
capacitor C.sub.0 through diodes D1A and D1B, respectively. With Q1A gated 
and Q1B off, current can flow through primary winding 4A to induce a 
corresponding current pulse in secondary winding 6A which is delivered to 
the output capacitor C.sub.0 through diode D1A. Similarly, when Q1B is 
gated and Q1A is off, a current pulse flows through primary winding 4B to 
induce a corresponding current pulse in secondary winding 6B, which is 
delivered to C.sub.0 through diode D1B. In this manner the output 
capacitor is charged with a pair of pulses during each operating cycle. 
The converter can be operated either "closed loop" or "open loop". In open 
loop operation the period of time that the primary winding switch is on is 
not phase modulated, but rather is fixed with respect to the resonant 
circuit phase, typically at the full power position. In this mode of 
operation the output voltage decreases with increasing load, but the 
no-load output voltage is always the input voltage multiplied by the turns 
ratio between the primary and secondary windings. Open loop operation 
might be desirable in uninterruptible power systems, when stability is not 
a problem, or when a separate voltage regulator is provided at the output 
of the converter. 
In closed loop operation the turn-on time for switch Q1 within each cycle 
is phase modulated with respect to the resonant circuit. The regulation 
loop can be closed by comparing the output voltage with a reference 
voltage, and phase modulating the switch times in response to the 
difference between the two voltages. A circuit for accomplishing this 
function is illustrated in FIG. 8. Switch control 24, indicated in a 
dashed line, includes an operational amplifier 32 having one input 
connected to the output voltage at output terminal 14, and its other input 
connected to a voltage reference. The operational amplifier output is 
connected to a pulse position modulator (VCO) 34, the output of which in 
turn is connected to a one-shot circuit 36. The op amp 32 produces an 
output which varies with the difference between its two sensed voltages, 
while pulse position modulator 34 triggers one-shot 36 at a time 
corresponding to the level of the op amp output. The pulse position 
modulator 34 triggers the one-shot earlier in the resonant cycle as the 
differential between the input voltages to the op amp 32 increases. 
One-shot 36 establishes an on time for the switching transistor Q1, 
automatically turning the switch off after a time interval that varies 
with the voltage differential between the op amp inputs. For steady-state 
operation, one-shot 36 allows the primary current pulse to fall to 0 under 
the influence of the resonant circuit before opening Q1. 
The operating principles of the converter will now be explained with 
respect to the fluxes generated by the windings on the transformer. The 
flux, voltage, and current conditions are illustrated in FIGS. 9(a), 9(b) 
and 9(c) for the tank, primary and secondary windings, respectively. FIGS. 
9(a)-(c) illustrate these waveforms for one complete conduction cycle of 
operation at nominal load conditions. FIGS. 9(d) and 9(e) illustrate the 
current that flows in the tank clamp and primary reset windings, 
respectively. The converter exemplified for the waveforms of FIGS. 
9(a)-(e) and for this explanation is shown in FIG. 10. It has equal 
numbers of turns for the primary, secondary and tank windings; a primary 
reset winding with half the number of turns of the primary winding; a tank 
leg gap with a suitable dimension so that the inductance of the tank 
winding in cooperation with the capacitance of the resonating capacitor 
causes the tank to resonate at the desired frequency, and such that a tank 
flux is generated as shown in FIG. 9(a), and a tank reset winding with 
half the number of turns of the tank winding. The converter is operating 
in a steady state condition with a regulated output voltage that is about 
10% less than the input voltage. The fluxes of FIGS. 9(a)-(c) are shown on 
the same vertical flux scale as one another. The currents for FIGS. 
9(a)-(c) are shown on the same vertical scale as one another. The voltages 
for FIGS. 9(a)-(c) are also shown on the same vertical scale as one 
another. The current of FIG. 9(c) is not shown on the same current scale 
as the other currents, as it would be too small to illustrate in the 
figure. FIG. 10 is a semi-schematic illustration of the circuit for the 
converter configuration pertaining to the description. In FIG. 10 the 
phase dots are located with respect to the polarities of the fluxes and 
voltages in FIGS. 9(a)-(c). The fluxes in the figures are the actual 
fluxes associated with the windings; these differ from the core flux by 
leakage flux. 
The operating principles of the converter will be given with reference to 
the three major phases of one operating cycle, where a cycle comprises the 
three successive phases of: (1) ON phase, where the switch in the primary 
winding is ON and power flows from the primary to the output; (2) DEMAG 
phase, where the switch is OFF, and the primary leg of the core is being 
magnetically reset; and (3) OFF phase, before the next ON phase and where 
only the tank winding is conducting current. It should be understood that 
for different operating conditions other than the one shown for this 
detailed description, the fraction of one cycle occupied by each phase 
changes; for example, at maximum power the OFF phase might be reduced to 
nearly zero, while the ON phase is much increased. For a better 
understanding, reference is had to FIGS. 9(a)-(e) and 10 while describing 
the three operating phases. 
In order to describe the operation of the converter with respect to the 
fluxes, a flux situation is defined where there is a flux balance 
.phi..sub.L defined as .phi.L=.phi..sub.P -.phi..sub.S -.phi..sub.T where 
.phi..sub.P is the total winding flux of the primary winding, .phi..sub.S 
is the total winding flux of the secondary winding, and .phi..sub.T is the 
total winding flux of the tank winding. .phi..sub.L represents the 
existence of a leakage flux such that whenever .phi..sub.L is some other 
value than zero, a leakage flux component exists which must be accounted 
for by current flow in the primary, secondary and tank windings. 
For the OFF phase, shown as t.sub.0 to t.sub.3 in FIGS. 9(a)-(e), the 
switch is open and no current is flowing in the primary or secondary 
windings. The transformer flux is driven by only the tank winding; thus 
the tank current, all voltages and all fluxes follow the natural 
sinusoidal oscillation of the tank. 
The ON phase is defined as the time during the cycle when the switch is 
closed, as illustrated in the figures from t.sub.3 to t.sub.5. Immediately 
before the switch is closed, .phi..sub.L =0 so there is no current flow in 
the primary or secondary windings. At a time t.sub.3 the switch is closed. 
At this time one winding on each leg is connected to a voltage source or 
sink: the primary winding is connected across the input capacitor voltage, 
the secondary winding is connected across the output capacitor voltage, 
and the tank winding is connected across the tank capacitor. Now, each 
winding is imposing its own d.phi./dt (rate of change of flux). Thus, 
.phi..sub.L becomes a non-zero sinusoidally changing value, in response to 
which a sinusoidally changing current, starting at zero, begins to flow in 
the primary, secondary and tank windings. As the ON cycle progresses, the 
sinusoidal change of the tank flux causes the magnitude of .phi..sub.L to 
increase sinusoidally to some maximum, and then become progressively and 
sinusoidally smaller, to the point at t.sub.5 where .phi..sub.L once again 
becomes zero. Primary current now becomes nearly zero (with the exception 
of a small magnetizing component, which is reset during the DEMAG phase, 
as described below). Thus there is a sinusoidal current waveform 
simultaneously flowing in the primary, secondary and tank or inductor 
windings due to the non-zero sinusoidally changing value of .phi..sub.L. 
The value and rate of change of .phi..sub.L depends on the leakage 
inductance of the transformer, which can be determined for any 
transformer, including integrated magnetic transformers, according to 
well-known principles. 
The above-mentioned sinusoidal primary winding current provides input 
energy for the power conversion process. The secondary winding current 
provides the output energy from the conversion process. 
A sinusoidal current component also flows in the tank winding; this adds 
some energy to the tank. The amount of energy added to the tank is 
proportional to the voltage difference between the input voltage and the 
output voltage. This tank current component causes the slope of the tank 
voltage to become somewhat steeper than just the natural tank voltage 
waveform alone, which increases the tank frequency. This can be seen from 
the tank current and voltage waveforms in FIG. 9(a) for the interval when 
the switch is closed. Excess tank energy is returned to the input voltage 
source through the tank clamp diode CR.sub.2, as shown by the current 
waveform in FIG. 9(d), from t.sub.2 to t.sub.4. 
The load current flowing in the secondary is essentially the same as the 
current in the primary winding, the only difference being that a small 
additional current is flowing in the primary to account for the 
magnetizing flux associated with the primary and secondary. As mentioned 
above, this secondary current flow is the output power of the converter. 
It should be understood that for primary to secondary turns ratios of other 
than one-to-one, the secondary current would be proportioned to the 
primary current by the turns ratio, according to conventional transformer 
theory. If multiple output windings were wound on the secondary leg of the 
transformer, the total output ampere-turns flowing in all of the 
secondaries would be essentially equal to the ampereturns of the primary 
winding, in accordance with transformer theory. 
The DEMAG phase occurs from t.sub.5 to t.sub.6. At t.sub.5, when the load 
current in the secondary has decreased to zero, the switch is opened. Some 
magnetizing flux is present in the core when the switch opens; this flux 
causes a reset current to flow in the reset winding as illustrated in FIG. 
9(e), from t.sub.5 to t.sub.6. This current returns magnetizing energy to 
the input voltage source. At t.sub.6 DEMAG is completed, the reset diode 
current has decreased to zero, and the cycle of operation begins over 
again with the OFF phase, as illustrated beginning with t.sub.0. 
It should be noted from observing FIG. 9(c) that from t.sub.1 to t.sub.3 
there is a voltage on the secondary winding induced by the tank flux, and 
that this voltage may cause the output diode to become forward biased some 
time before the switch is closed for the ON cycle. As soon as this 
tank-induced voltage component causes the output diode to become forward 
biased, the secondary leg behaves as if a voltage source is connected 
across the secondary winding, so the remainder of the tank flux is carried 
in the primary leg. Even so, only a small amount of current flow is 
generated in the secondary winding. 
The operation of the preferred embodiment of FIG. 3 during startup, light 
load, medium load and heavy load conditions is illustrated in FIGS. 11, 
12, 13 and 14, respectively. FIG. 11 illustrates waveforms when the power 
supply is first starting. Line A illustrates the tank voltage, which is 
initially at zero. Line B is timing of the switch, where the full input 
voltage is applied to the primary when the switch is ON, and zero volts 
applied when the switch is OFF. When the switch is initially closed, 
primary winding current begins to flow (line C). This primary winding 
current will reach a positive peak and then begin to diminish generally 
sinusoidally. Ideally, the primary winding current will have diminished to 
0 by the time the switch is turned off. However, during the initial cycle 
the tertiary winding flux will typically not build up sufficiently to 
reduce the primary current all the way to zero by the time the switch 
turns off. Accordingly, during the initial cycle of operation the primary 
winding current will typically be abruptly shut off by the switch opening, 
as indicated at 42. Thereafter the tertiary winding voltage will have 
built up sufficiently to reduce the primary current to zero by the time 
the switch opens. This switching at essentially zero current avoids prior 
problems with interference. 
The situation under a light load is illustrated in FIG. 12. The switch 
control circuit senses the light load, and accordingly sets the switch to 
turn on relatively late in the cycle at 44, after the tertiary voltage has 
entered its negative phase. Since the tertiary flux resists the buildup of 
primary current from the start of the cycle, that current will terminate 
after only a short pulse 46. 
For medium loads, illustrated in FIG. 13, the switch control turns on 
earlier in the cycle, while the tertiary voltage is still positive, and 
does not turn off again until later in the cycle than with light loads. 
More time is therefore required to terminate the primary current. For 
heavy loads, illustrated in FIG. 14, the switch is turned on even earlier 
during each cycle. Although the switch is illustrated as being turned on 
in coincidence with the positive tertiary voltage peak, it could be turned 
on even slightly before this time. This results in a large primary current 
pulse to charge the output capacitor. It will be noted that the current 
pulses are generally sinusoidal but not perfect sine waveforms in shape, 
the latter half of each pulse being slightly concave in this case. 
The converter thus employs a type of pulse width modulation that is 
achieved by frequency or phase modulation over a relatively narrow 
frequency range to adjust to different load levels. This is a considerable 
improvement over prior frequency modulated designs, since it eliminates 
the problems previously encountered with very wide frequency range 
required for regulation. The desirability of pulse width modulation is 
discussed, for example, in a text by Abraham I. Pressman, "Switching and 
Linear Power Supply, Power Converter Design," Hayden Book Company, 1977, 
pages 5-8 and 12-13. 
Various embodiments of a novel integrated magnetic resolnant converter have 
thus been shown and described. In addition to solving the prior 
interference and frequency range problems, the converter is relatively low 
in cost, increases efficiency and reliability, is relatively simple and 
inexpensive to build, is capable of a theoretically unlimited number of 
outputs, and exhibits a high power density. As numerous variations and 
alternate embodiments will occur to those skilled in the art, it is 
intended that the invention be limited only in terms of the appended 
claims.