System and a method for optically detecting an edge of a tape

A method and apparatus performs the detection of an edge of a tape and further controls the read/write head to position itself relative to the detected edge. An array of photodetectors in an integrated circuit chip senses the intensity of light illuminating the chip with the tape running between the light source and the photodetectors. A light-to-dark transition is formed which is indicative of the edge of the tape.

BACKGROUND OF THE INVENTION 
The invention relates to a system for tracking the edge of a magnetic 
medium or an optical medium as well as tracking the data thereon. 
In high speed magnetic tape reading and writing units ("tape streamers"), 
the data is read from, or written in, a plurality of data tracks which run 
parallel to the edges of a magnetic tape. The write/read head of the 
system must, therefore, be accurately positioned over a selected track to 
read the data from the selected track or to write new data in the track. 
It is known, for example, from U.S. Pat. No. 4,476,503 to position the 
write/read head by first locating an edge of the tape, and then moving the 
head a specified distance from the edge to the desired track, the tracks 
being disposed on the tape at respective known distances from the edge. 
It is a problem in this technology, however, that in practice the tape will 
not travel in a constant, perfectly straight path. The tape in moving will 
meander slightly and will follow an irregular serpentine path. The 
position of the edge must be constantly monitored with high accuracy and 
the position of the head is constantly adjusted through a feedback system 
because data tracks are usually packed so closely together on the tape 
that a small change in the edge position, in the absence of a 
corresponding change in the position of the head, can cause the head to be 
adjacent a track other than the desired track. 
Various tracking systems are known in the art for many purposes, such as a 
transducer which may be positioned relative to a recording tape edge in an 
optical system. A light transmitter, such as a point source illuminating 
both the tape and a photo sensor, may be partly covered by the tape and 
placed in a fixed position relative to the light source and the 
transducer. The diameter of the photo sensor must be greater than the 
expected transversal range of movement of the tape. A control system 
allows the transducer to follow the transversal displacements of the tape. 
Another known alternative embodiment implements a fixed position light 
emitting bar as the light source, and a photo sensor which is rectangular 
and of the same length as the light emitting bar and fixed to the 
transducer itself. This embodiment allows for the positioning of a 
recording head or a transducer relative to the tape edge for a multi-track 
recording system. Each position requires one reference input to the 
position controller of the transducer. This allows a signal proportional 
to the position of the transducer relative to the edge of the tape to be 
used as an input for the controller, which thereafter sends an error 
signal proportional to the difference between the reference and the output 
of the photo sensor to a motor which controls the position of the 
transducer. There are, however, several drawbacks to such a system. In a 
first version of the system, the transducer is normally placed between two 
tape guides and the light beam must be placed between one guide and the 
transducer itself. The problem is that the drift direction at the position 
of the light beam and at the position of the transducer can be different. 
A second version of this system corrects this drift problem; however, 
since the output from the photosensor is an analog signal, the system is 
subject to additional problems. The most severe problem is its sensitivity 
to small dust particles. With a magnetic medium, such particles settle on 
the illuminated part of the photodiode, and it is difficult to detect the 
occurrence of and to compensate for such noise. Updating of the DC output 
from the photodiode each time the transducer is placed in a given position 
is not possible. In addition, the system is sensitive to stray light 
pickup unless synchronous detection is used. Similarly, stray light pickup 
is difficult to compensate for unless the tape drive is completely 
shielded from external light sources. The use of an infrared source may 
help, but the stray light pickup is still a problem since infrared light 
may as well be present as background noise. In a dynamic tracking system, 
stray light pickup normally contains 100 Hz or 120 Hz components which 
will disturb a tape edge tracking servo unless synchronous detection is 
used. If an infrared light source is used, the photodiode may need a 
filter which is translucent for the wavelength used. This causes the 
distance from the tape to the photodiode to be increased which in turn 
reduces the sharpness of the transition zone between the light and dark 
area of the detector. 
Another known method has a magnetic tape passing over a fixed 
recording/reading head which is automatically balanced in a vertical 
direction. The nominal vertical position of the tape is determined by at 
least one set of photo sensors and light emitting diodes and arranged such 
that the tape edge(s) partially covers the photo sensor(s). A typical 
arrangement embodies two sets of sensors, one for the lower edge and one 
for the upper edge of the tape. In this embodiment, the head is adjusted 
and fixed in a position which corresponds to equal outputs from the two 
sensors when the tape is placed in its nominal position. A control signal 
is obtained by simply taking the difference between the outputs from the 
sensors. The error signal is fed to a motor in a mechanical arrangement 
capable of adjusting the position of the tape. Such a system is 
susceptible to the same type of errors as discussed in the above system. 
Another embodiment contains two sets of light emitters and receivers very 
similar to the one described in the preceding paragraph. Problems typical 
with an analog proportional system, such as difficulties with adjusting 
and maintaining equal light levels in the two emitters and a circuit for 
manually balancing or trimming the AC light levels and automatically the 
DC levels, are still present. The system is inherently susceptible to 
differences which may occur after the factory adjustments of the light in 
the two channels; such manual adjustment increases both the production and 
the component cost of the product. 
An automatic track following system is also known which uses at least two 
separate detecting heads with read gaps wider than the written tracks and 
where the gaps have azimuth angles of equal values but of opposite 
rotational sign. During tracking, the centers of the azimuth head follows 
the centers of the corresponding two signal tracks. When the tracks drift 
away from the center positions of the azimuth heads, a lead/lag error 
signal can be extracted from the two heads if the information signal 
tracks contain some type of known synchronization, e.g. if video sync 
pulses have been recorded in parallel on both tracks. The polarity of the 
lead/lag signal determines the direction to move the head, and its value 
is proportional to the error if the tracks are located within the range of 
the azimuth gaps. Since auxiliary read gaps are used, responding only to 
the video sync pulses of long wavelengths, the azimuth angle can be 
tolerated. The extra tape noise from the unrecorded data can be tolerated 
in the timing channels due to the lower bandwidth requirement. However, 
the primary disadvantage of such a system is the inherent weakness of 
using azimuth heads for tracking, since such allows for a very limited 
linear tracking range. If tracking is disturbed, the control system has no 
information available about the direction to move the head. Noise pulses 
may cause head movement in the wrong direction as well. The head must be 
moved to the nominal position before the tracking system can be activated 
after a loss of the lead/lag signal, or a track seeking algorithm must be 
activated to start recovery. Another disadvantage is the added cost of the 
extra read heads Such tracking systems are best suited for helical scan 
tape formats where the tracking can immediately lock on neighbor tracks if 
disturbed. Such a disturbance can be tolerated in some consumer analog 
video tape recorders (single frame loss or disturbance) or in helical scan 
data storage systems where interleaved data frames and error correction 
permits the loss of a track. 
Another known embodiment proposes a two channel system for data recording 
where two azimuth read heads are used to derive the tracking error from 
the time skew between the read data clocks of the channels. The timing 
pulses are not so easily available as the sync pulses used in other prior 
art devices. However, this device does not require separate azimuth read 
heads, since the two write heads also have azimuth angles of opposite 
sign. The signals are either read back while writing by two aligned read 
heads with the same azimuth angles, or by the same write heads in simpler 
tape drives. An advantage of this system is greater information packing 
density, since no guard bands between tracks in the information area of 
the tape are needed. This device, however, is limited in the linear 
tracking range since it requires a very accurate open-loop mechanical 
positioning mechanism in addition to the servo mechanism. An additional 
problem is the compensation or calibration of the time skew between the 
channels especially when reading tapes written in other drives. Yet 
another disadvantage is that if backward compatibility with older tape 
formats written without azimuth is to be maintained, at least one set of 
zero-azimuth read and write gaps must be provided. If one of the two write 
gaps is without an azimuth angle, half of the timing error is present as 
compared to a double azimuth scheme. The crosstalk from neighbor tracks 
will also increase. 
SUMMARY OF THE INVENTION 
It is, therefore, an object of the invention to provide a method for a 
tracking system with built-in redundance which allows for the continuous 
monitoring of the tape edges during read signal tracking even if the read 
gaps are off-track. 
It is a further object of the invention to provide a system which stores in 
memory a history of the positional trace of the tape edges so that tape 
edge qualification can be performed based on the stored data for the edge. 
It is a further object of the invention to provide an integrated CMOS chip 
to be mounted on the recording head itself or its carriage to obtain 
increased mechanical accuracy at a lower cost. 
It is a further object of the invention to also provide on-chip 
photosensors for automatic azimuth adjustment of the recording head. 
The system processes the analog information about the tape edge locally 
before digitizing the analog data. The analog data is numerically coded 
before it is sent off-chip to the digital controller and servo. 
The tape edge seeking method solves a problem in the art of indicating a 
false edge position when magnetic particles are torn off from the tape 
edge due to wear. The wear on the tape edge arises when the number of tape 
passes exceeds the specified number of passes which are allowed in an 
environment where the same cartridge is used until it is worn out. The 
present edge detecting method avoids this problem by detecting an optical 
edge, not the magnetic edge.

DESCRIPTION OF THE PREFERRED EMBODIMENTS 
FIG. 1 shows an integrated CMOS chip 10 mounted on a typical magnetic 
recording head 20. The active surface of the chip 10 is facing a magnetic 
tape 18 and is mounted on a magnetic recording head 20 using tape 
automated bonding technology well known in the art. The active surface of 
the chip 10 is protected by metal layers and an array of windows is 
created during the metallization. The windows are numbered from 1 to 42 as 
shown in FIG. 2. Behind the windows are integrated phototransistors which 
can be moved perpendicular to the transport direction of the magnetic tape 
18. A light source illuminates the tape 18 and the surface of the chip 10 
which is not covered by the tape 18. The tape 18 runs in close proximity 
to the surface of the chip 10 thereby creating a sharply defined boundary 
between the shadow area and the illuminated area on the active surface of 
the chip 10. The chip 10 may be mounted in contact with the tape 18 if 
care is taken to reduce wear by, for example, covering the chip 10 with 
thick silicon dioxide. 
The accurate determination of the position of the tape edge is created by 
the inherently geometrical accuracy found in the patterns created during 
the wafer fabrication process for integrated circuits. Dimensions may be 
controlled to a few parts per million. 
One or two index marks, corresponding to known pixel numbers on the 
detector chip 10, may be aligned with an imaginary center line running 
between each write and read gap 14, 16 in one or more channels as shown in 
FIG. 1. The index mark corresponding to known pixel members on the 
detector chip 10 avoids time-consuming edge-seeking of the edge of the 
tape 18 since once the pixel numbers of the tape edge have been found, the 
write/read gaps 14, 16 may also be located. 
FIG. 2 shows an example of a pattern of phototransistors laid down in 
silicon. Other semiconductor materials, such as gallium arsenide, may, as 
well, be used as is well known in the art. The pattern shown in FIG. 2 can 
be used for high resolution, one dimensional location of a contrast edge 
created when the shadow from a recording tape falls on the pattern. The 
actual, detailed geometrical shape used will depend on several factors 
including the resolution, light sensitivity, signal processing elements to 
be used and the actual integration and wiring of these elements together 
to form macro cells which make up the complete pattern shown. The 
phototransistors in FIG. 2 partly overlap in the Y-direction of 
measurement. To enhance resolution, it is possible to increase the overlap 
between the pixels. This can be done since the raw output signals are 
analog. Also, photo sensors may be contained within the middle of the 
processing cells. Other known methods for realizing photo detection may be 
implemented, such as a charge coupled device (CCD) which is capable of 
obtaining a greater servo bandwidth. 
The raw outputs from the individual pixels are processed in such a way that 
a smooth intensity profile is created. Based on this profile, the location 
of the tape edge is estimated. The phototransistors in FIG. 2 are four 
units in the "Y" direction and the length of the local signal processing 
part is eight units. Two units are used for wiring space between the 
cells. The scale of the drawing in the "X"-direction may be varied. 
Therefore, FIG. 2 does not give a realistic picture of the actual areas 
required for the processing electronics. The step size between cells, i.e. 
in the X direction, is one unit in this array. If an ideal contrast edge 
falls in the center of a phototransistor, the output current from this 
transistor will be halfway between the "dark" level and the "light" level 
of its next nearest neighbors. Its nearest neighbors will have output 
currents corresponding to 25% and 75% of the light difference. With 
"overlapping" phototransistors, the task of the analog signal processing 
will be to select the position corresponding to the cell with the 50 % 
level as the best estimate for the edge's position. 
The block diagram of the cell shown in FIGS. 3A and 3B performs the basic 
functions of detection, logarithmic compression, spatial averaging, 
spatial differentiation, spatial selection of the physical line 
corresponding to the estimated position of the tape edge and switching in 
or out of the connections between nodes WTAC N-2 and WTAC N-1. WTAC N 
refers to the communication line between multiple cells. Each WTAC has 
active sub-ranges (one range for each tape edge region) where all of the 
switches are closed. In this way, the time response of the 
"winner-take-all" can be improved by minimizing the capacitive loading of 
the communication line. In addition, a transconductance amplifier 42 and a 
current source 52 which lay outside the active sub-range can be disabled. 
The use of sub-ranging is important since the information media/magnetic 
tape typically has index holes to mark the beginning of the tape, the end 
of the tape, data load points, data termination area and an early warning. 
Data is automatically re-written to the tape when it has been lost due to 
the early warning hole. Therefore, the part of the chip where the holes 
pass by must be disabled. The WTAC switches, the current sources 52 and 
its bias switch need only be inserted at regular intervals, e.g. at every 
fortieth cell. The other type of analog processing cell is as shown in 
FIG. 3B. 
The following description applies to FIGS. 3A and 3B wherein like reference 
numerals designate like parts. The primary signal current representing the 
light flux goes from the pixel block 30 to the log block 32 where the 
logarithm of the current is computed. The numerical value of the 
computation will depend on the actual semiconductor process used in the 
area of the transistors. For the calculations presented in this 
application, a formula of 3.47*ln(I/IO) was used for the voltage 
normalized to units of kT/q. IO is dependent on the process and the area 
of the transistors and is normally computed for a unit-area transistor. 
The factor of 3.47 depends on the body-effect. The intermediate signal 
going to the transconductance amplifier 34 is represented as a voltage. 
The transconductance, Gm, of the amplifier 34 is controlled by a voltage 
on the node GM1 which is common to all active cells. The output current 
from the amplifier 34 drives node "n" to which a nonlinear resistor 36 is 
connected. The neighbor cell also connects a nonlinear resistor 38 to node 
"n". The resistances of each nonlinear resistor 36, 38 are controlled by 
the voltage on the NRB (nonlinear resistor bias) input node. Together with 
the transconductance of the amplifier 34, the small signal resistances of 
each nonlinear resistor 36, 38 set the space constant of the network 
created when multiple cells are connected in series. The space constant 
is, therefore, controlled by the external voltages applied to the node GM1 
and control inputs NRB. Voltage compliance of the non-linear resistors 36, 
38 (actually MOS "pass" transistors) is controlled by an automatic bias 
circuit 40, a nonlinear resistor bias network, which tracks both the 
voltage and bias developed at node "n" of the cells of the nonlinear 
resistors 36, 38 so as to compensate for the body-effect of the pass 
transistors 36, 38. To gain full "neural" advantages regarding the spatial 
resolution of such a resistor network, the density of nodes should be high 
and the spatial constant adjusted or should even be adaptable to the 
desired resolution. The transconductance amplifier 42 is of the 
"wide-range" type, i.e. its output voltage may be close to the supply 
rails. Amplifier 42 takes the voltage difference between nodes "n" and 
"n-1" . When the light intensity increases with increasing "n", the 
incremental current flows out of node "n" if the phototransistors are 
realized in N-wells. Then the voltage output from the log circuit 32 will 
decrease logarithmically with an increase in light input. To obtain a 
positive incremental current out of the amplifier 42 when there is a light 
intensity gradient in the positive direction, the difference must be taken 
as shown with the "+" and "-" at the inputs of the amplifier 42. The 
transconductance of the amplifier 42 is controlled by the voltage on node 
GM2 which is common for all active cells. This voltage may be switched off 
when the cell is located outside of the active range. 
The final local processing element of the cell is the winner take all 
circuit 44. It consists of two transistors 46, 48. The input signals to 
these cells are represented by currents injected by amplifier 42. In this 
application, the output signals are also represented by currents, commonly 
named OUTPUT-N, in the cells shown. During the current injection process, 
the output voltage of the amplifier 42 will rise to an appropriate level 
given by the DC conductance of transistor 48. If this conductance is too 
low, the voltage rises at the gate of transistor 48. Transistor 48 will 
then drive current into the WTAC N2-node. Its voltage rises until the 
current through transistor 46 is equal to the injected current from the 
amplifier 42. The injected current is now becoming "the winner". All of 
the winner take all blocks 44 and the WTAC N-1/WTAC N-2-wires simulate 
neurons with inhibitory responses, and they share one common signal path 
to communicate inhibition for al cells. Since the tape edge is always 
located within a narrow range, the communication distance can be broken up 
into sub-ranges by controlling the inputs to the switches 50 as shown in 
FIG. 3A. Depending on the length of the active range monitored, a 
reference current source 52 is connected to the WTAC N-2-line, but only 
one or a few occur per sub-range. The sub-ranges can be placed freely and 
symmetrically around the location of the tape where the inputs to the 
switches come from a long shift register and decoder, and a location of 
the sub-range depends on the actual pattern loaded into the register. By 
using sub-ranging, the response time of the system is greatly improved 
since the capacitive loading by the WTAC-line is minimized. Each cell is 
able to contribute a current into the node WTAC N-2, but only if its input 
current is larger than, or in extreme cases equal to, other currents. The 
largest input current will always "win". The signal currents which "lose" 
are shunted to ground by transistor 46 because the winning current 
determines the common gate voltage which is communicated to all neurons 
within the active sub-range. The conductances of the corresponding 
transistors 46, 48 are so high that very small voltages develop on these 
nodes. Thus, good suppression of the "losing" signals is due to the 
exponential relationship between gate voltage and transistor current. 
The circuitry of the cell shown here is only able to detect positive 
intensity gradients. This is an advantage for the application described 
here since during dynamic tracking of the tape edge, the transition zone 
from dark to light is known to be within the sub-range. Only a limited 
number of the switch nodes 50 need be sampled. If a small dust particle is 
present on the illuminated part of the chip, the first negative transition 
will be rejected. If the next positive transition is outside the 
sub-range, it will be suppressed. The same is true for the opposite 
polarity of transitions in a dual-edge detector described below. 
Both tape edges can be monitored to provide redundancy in the system. There 
is a need to detect both positive and negative intensity gradients. With a 
serpentine tape format written with a recording head 20 of the type shown 
in FIG. 1, the center chip can be divided into two halves: one lower half 
for detection of negative transitions where the polarity of the inputs of 
each amplifier 42 are as shown in FIG. 3, and one upper half for positive 
transitions where the inputs of each amplifier 42 are interchanged. Since 
sub-ranging is used for the WTAC-line, it can be shared for detection of 
both tape edges having two active ranges. Alternatively, to obtain maximum 
redundancy in a system, two completely independent systems may be used. 
FIG. 4 shows a processing scheme of the samples logarithmically compressed 
corresponding to the outputs from the cells in FIG. 2. The raw outputs 
from the cells are first averaged spatially. To implement this function, a 
nonlinear resistor network is used. Thereafter, differentiation of the 
intensity profile I(n) can be defined in this way: 
EQU dI(n)/dn=I(n+1)-I(n-1) 
This algorithm gives a maximum of dI(n)/dn for n=13 when the cell n=16 is 
illuminated by 50% of the "light" minus "dark" level of intensity. Due to 
weighted spatial averaging prior to differentiation, the signal-to-noise 
ratio is greatly improved. The final position is selected by comparing in 
a pure analog manner each output, ...n-2, n-1, n, n+1, n+2..., of the 
spatially differentiated pattern resulting in the estimated position shown 
in FIG. 4. The edge is still estimated at n=13. The signal levels are 
higher, but the three strongest levels are still close together especially 
since the signal levels will be inputs to the nonlinear transconductance 
amplifiers. The transconductance amplifier exhibits a hyperbolic tangent 
transfer function with a limiting current as the bias current of the input 
differential stage. This current can be set for sub-threshold operation 
with an external bias voltage to the gate of the current generator in the 
differential amplifier. The transconductance amplifier is often used as a 
voltage follower with a limited and externally controlled output current 
capability used to duplicate or buffer signals, to drive (nonlinear) 
resistive networks, or, when loaded with a capacitor, as a temporal 
integrator. 
Another embodiment of the invention uses another differentiation method. 
The step size between cells is the same as in FIG. 2, that is, one unit. 
FIG. 5 shows the signal processing of the four unit phototransistor cell 
wherein one unit of overlap exists. Differentiation of the intensity I(n) 
is performed in the following manner: 
EQU dI(n)/dn=I(n)-I(n-1) 
The highest output occurs at n=13 with a nearby point n=14. The tape edge 
is then located at n=16. Hence, an offset exists between the actual 
physical location and the estimated location. 
In FIG. 10, the analog processing cells have been marked p.sub.0, p.sub.1, 
p.sub.2, ... p.sub.n, p.sub.n+1, ... p.sub.4095. The number of cells is 
equal to the maximum number which can be represented by twelve binary 
digits. The numbering of the I/O pins of each analog processing cell 
corresponds to that shown in FIGS. 3A and B. Three global control voltage 
V.sub.GM2, V.sub.NRB and V.sub.GM1 are applied to all cells via lines 3, 5 
and 6, respectively These voltages are set up by the bias circuit shown in 
FIG. 11. They can also be programmed with external resistors indicated by 
the input pens R EXT1, R EXT2, ... as shown in FIG. 11 as inputs to the 
bias circuit. External analog control voltages may also be applied to the 
bias circuit, for example to set V.sub.NRB which determines the spacial 
resolution of the pixel averaging network. During power-up and initial 
start-up procedures to locate the edges of the tape, V.sub.GM2 can 
optionally take on a low value to reduce the total power dissipation of 
the chip when all processing cells are enabled. For this purpose, R EXT6 
together with the control signal HIGH GM2 have been added as inputs to the 
bias circuit as shown in FIG. 11. 
A row of switches, S.sub.0, S.sub.1, S.sub.2, ... S.sub.N ... shown next to 
the row of processing cells in FIG. 10 constitute the analog multiplexer. 
The current through pin 7 of cell P.sub.N has a double-arrow indicating 
this is the current from the winning cell. The switch adjacent P.sub.N 
routes this current to the upper line which can be accessed by all the 
switches and to the current sense amplifier, CSA. The current signal is 
then converted to a voltage signal waveform W1. All other switches dump 
the losing currents onto a dummy line with a fixed voltage of V.sub.REF1. 
Although the winning current is a digitized signal of a known level, the 
voltage output of the amplifier CSA is compared with a reference voltage 
V.sub.REF2 before the signal EDGEPULSE is sent to the counter's D-type 
register. This is further illustrated in FIG. 12. 
The current sense amplifier CSA contains one feature specific to this 
application, namely an auxiliary weak I.sub.REF current which must satisfy 
the following condition: 
EQU n*I-lose&lt;I.sub.REF &lt;&lt;I-winner, 
where n is the number of all losing currents. These currents are not 
necessarily equal, however, but a sum of currents may be more accurate. 
Since the losing currents are extremely small and since sub-threshold or 
weak inversion operation is used, the equation can be easily satisfied. 
The I.sub.REF current provides a defined load-level output of the CSA when 
the losing currents are sampled. Therefore, the low-to-high voltage level 
transition can be kept under control and minimized which improves the 
switching speed. Further, the exponential feedback is used for the CSA 
meaning logarithmic compression also takes place when the signal is 
converted from a current to a voltage. 
A dual current switching scheme provides a great advantage in that all 
lines are kept at a constant voltage: the CSA provides a virtual 
V.sub.REF1 (plus or minus an input offset voltage) at its negative 
feedback input node. Charging of the capacitance to ground of the two long 
lines in the multiplexer is, therefore, avoided, and the analog 
multiplexer can operate at high speeds. Furthermore, the switching 
elements themselves are made of complementary pass transistors which 
minimize the charge injection problem during switching. 
The analog switch control shift register operates by means of a single zero 
which propagates from left to right. At the end of each cycle, the zero 
disappears to the right, and the global OR-gate generates automatically a 
new zero which will be clocked into the first flip-flop called SC.sub.0 at 
the next edge of the CLK1 pulse. The output of the OR-gate is shown as 
SCAN*. This signal also resets the counter 80 as shown in FIG. 10. This is 
done to synchronize the counter 80 with the shift register 84 during the 
start-up of the system, since the initial state of the shift register 84 
is unknown at start-up. An asynchronous stop-pulse waveform W2 is 
generated at the time when pixel N is sampled. This pulse is synchronized 
by CLK1 and called EDGEPULSE. The 12-bit number for N is converted to 
serial form by the shift register 84. The SCAN* signal is also provided as 
an output to an external controller which can synchronize the read-out 
clock of the shift register 84, SCK, with the operation of the scan 
register. By doing this, read-out at times when data changes in the D-type 
register 82 is avoided. When two active sub-ranges are used (duo-edge 
detection), there must be two samplings per cycle. 
Taking the signals SCAN* and EDGEPULSE off-chip is also very useful for 
testing, monitoring and analyzing purposes. The scan-pulse can be used to 
trigger an oscilloscope, and the position in time of the edge pulse is a 
map of the location of the tape edges on the chip. When the tape is 
running and the chip is kept in a fixed position, a time interval 
analyzer, such as a Hewlett Packard type HP5371A, can be used for 
statistically analyzing the positions of the pulses. Extracted frequency 
domain data can then be used for design input to the head servo itself. 
The mask layout may require the placing of the counter 80, the D-type 
register 82 and the shift register 84 off-chip to reduce coupling of 
digital noise into the low-level analog processing circuits. The noise 
from the counter 80, the D-type register 82 and the shift register 84 will 
typically be greater than the noise from the fully differential 
implementations of the scan register in the analog multiplexer and range 
select register. 
The shift registers, R.sub.0, R.sub.1, ... R.sub.N ... are range select 
registers similar in design to the feedback shift register, but they are 
not clocked continuously. A sequence consisting of 4096 states is set up 
on the RANGE-line and clocked in by CLK2. Thereafter, the pattern is kept 
statically in the registers by stopping CLK2. The content of the registers 
will then consist of ones, followed by a series of zeroes which determines 
the active range where the tape edge must be present. The rest of the 
states are filled up by ones if a single sub-range is used or followed by 
a new pattern of zeroes and ones for dual-edge detection. 
Prior to the final sub-range selection, a static tape edge detection 
(single or dual edge) is performed when all cells take part in a global 
selection process. In some implementations, the voltage on the GM2 line is 
reduced during the static mode, i.e. the control signal HIGH GM2* is 
inactive. 
Sub-ranging is used for high-speed dynamic operation when a tape is moving. 
Before the final sub-range is selected, the position of the edge must be 
known to fall within the sub-range or sub-ranges with certainty. 
Sub-ranging is an option, and its use will depend on the response speed 
required during dynamic tracking, i.e. on the sampling rate of the 
tracking servo. However, most often sub-ranging must be used when tape 
holes disturb the edge detection process. 
Another method of improving response speed is digitizing the winning 
current to a "fixed" level. The level is actually within a narrow range 
given by offset effects or mismatches between transistors. Therefore, the 
voltage variations and stray capacitance charging on the communication 
line, as shown by line 1 in FIGS. 3A and B, between cells will be small 
when the winning current shifts from cell to cell with time. 
FIG. 3A shows the two analog switches with common control signals SWITCH N 
and SWITCH N* can optionally be used. The first is inserted in the 
winner-take-all communication line to shorten it, enhance, speed up 
response time by reducing stray capacitance The second switch is placed in 
series with the bias input for the amplifier 42 and the gate voltage of 
the reference current source 52 for the winner-take-all network 44. FIG. 
10 illustrates that switch control lines are provided to all of the 
processing cells although only a sub-set of switches actually needs to be 
used for the speed-up of the winner-take-all network. The other elements 
of the range select shift register are used to switch off the inactive 
amplifiers 42 (shown in FIGS. 3A and B) to save power. Current may be 
needed in some cases in the upper part of the weak inversion range. 
FIG. 12 shows in more detail the current sense amplifier CSA and COMP 
blocks of FIG. 10. FIG. 12 also illustrates how voltage V.sub.REF2 is 
generated. V.sub.REF2 comes from the bias circuit as shown in FIG. 11, but 
its generation has been shown within the dashed lines of FIG. 12 to ease 
understanding. FIG. 13 shows the voltage levels for the waveforms W1 and 
W2 of FIGS. 10 and 12. 
V.sub.REF1 may have a value of V.sub.dd /2. When the losing currents are 
scanned out, I.sub.ref ensures that the output of amplifier CSA, W1, is 
kept at a well defined level of about 0.5 volts above V.sub.REF1 as shown 
in FIG. 13. Since the signal voltage deviation of W1 is small due to the 
logarithmic compression, the threshold level V.sub.REF2 of the simplified 
comparator COMP in FIG. 12 must track V.sub.REF1. This is accomplished by 
connecting transistor Q35 to V.sub.REF1. The voltage which develops over 
this diode-connected transistor Q35 is also made to be dependent on 
V.sub.GM2 which controls the actual magnitude of the winning circuit. Q35 
is of the same type as the feedback transistor Q1b. Transistors Q31 and 
possibly Q32 and Q34 scale the current through Q35 to a certain fraction 
of the level of the winning current, indicated by I.sub.winner /K. The 
threshold level of the comparators may therefore be placed slightly below 
the high level for signal W1, giving a better overall noise margin. 
The comparator COMP is shown as a simple wide-range differential amplifier, 
i.e. its output waveform W2 goes rail-to-rail, as shown in FIG. 13. 
Normally, the comparator COMP needs a more complex design than is shown in 
FIG. 13. 
FIG. 14 illustrates the control environment for the tape edge detector chip 
10. The servo control starts by setting up a reference number in a 
control, filter and digital servo processor 100. A controller module 102 
receives the measured position of the tape edge from a light source 108 
projecting light onto the tape which casts a shadow on the tape edge 
detector chip 10. A SDATA signal indicative of the tape position is sent 
to the digital servo processor 102 where filtering is performed, and the 
error is calculated using the program stored in memory 110. The error 
signal goes back to the controller module 102 which contains the necessary 
hardware drivers to transmit the error signal to head drive motor 
electronics 104. In some implementations, the error signal can be 
transmitted directly from the control filtering and digital servo 
processor 100 to the motor electronics 104. In this case, the head drive 
motor or motors 106 symbolize the transversal movement of the head to 
correct for the error and the tape edge detector chip 10 measures the 
actual position of the tape edge which again is read out on the SDATA line 
by the controller module 102. All other processing functions required for 
operation of the tape drive are performed by a main processor 112. 
Auxiliary resistors 114 correspond to the inputs to the bias circuit of 
the tape edge detector chip 10 shown in FIG. 11. 
The physical integrated circuit technology and the methods used to realize 
the analog signal processing of the present invention are described in 
"Analog VLSI and Neural Systems" by Carver Mead which is herein 
incorporated by reference Conventional digital CMOS VLSI-circuits are 
based on complementary N- and P-channel MOS transistors which operate 
above or below a conduction threshold level. The threshold voltage is 
defined as the gate-to-source voltage where the mobile charges in the 
channels begin to limit the flow of channel current For MOS transistors, 
there exists a certain sub-threshold gate voltage range, i.e. a range of 
gate-to-source voltages where there is an exponential relationship between 
the gate voltage and the drain-to-source current. The current in this 
range is caused by a pure natural diffusion process. For gate voltages 
approaching the threshold voltage, the exponential increase in the current 
ceases. The mobile charges in the channel disturb the diffusion process, 
i.e. they begin to degrade the exponential law. For gate voltages above 
the threshold voltage, the current increases as the square of the gate 
voltage. Therefore, the threshold voltage is better thought of as a 
transition zone between the exponential sub-threshold region and the 
square-law region. 
For a pure exponential relationship between the drain current and the gate 
voltage, the gate voltage range may be measured between 300 mV to 700 mV. 
This range may vary somewhat depending upon the actual CMOS process used. 
In this range, the current increases exponentially over 5 decades from 30 
pA to 3 .mu.A. The transistors will be useful for analog processing in the 
transition zone and above when the exponential law is not used for 
computation, when a non-exponential or non-logarithmic limiting effect is 
desired, or when voltage followers are used for interfacing off-chip 
signals. 
Except for the Early-voltage effect, which can be partly controlled by the 
length-to-width ratio of the transistors, the drain-to-source current is 
independent of the drain-to-source voltage if this voltage is greater than 
a few thermal voltages, kT/q. At this point, the MOS transistor is said to 
be saturated 
In the sub-threshold range, the MOS transistors behave in a remarkably 
predictable way. Extremely low power consumption exists with a typical 
dissipation per transistor in the nW-range Complete detector chips are 
possible with the inclusion of integrated photo-detectors among the analog 
processing elements 
In a CMOS process, the photodetectors can be realized as vertical, bipolar 
photo-transistors. The bases of the transistors are isolated, diffused 
wells upon which light is permitted to fall through openings or windows in 
a metal mask. The emitters are diffused areas in the well, and the common 
substrate make up the collectors. If the incident photons have energies 
greater than the bandgap of silicon, electron-hole pairs will be created. 
For a N-well process, the bases of the transistors are of N-type material. 
The created base electrons will lower the energy barrier from the emitter 
to the base and cause an increase in the flow of holes from emitter to 
collector. For a conventional transistor, there is a large gain associated 
with this process. The output current from the photo-transistor will be 
proportional to the intensity of the light. 
Logarithmic compression is one of the most powerful analog processing 
functions which can be realized using the sub-threshold elements. By using 
logarithmic compression, the log detectors can operate over more than four 
decades of light intensity, and sensitivity to stray light will be greatly 
reduced. By taking the difference between the logarithmic output voltage 
from two pixels, a measure of the contrast ratio which is independent of 
the actual light intensity levels is obtained. 
In the present invention, logarithmic compression is very useful due to 
variations in light intensities and the simplicity of the implementation. 
The inherent contrast ratio of the tape medium can be found simply by 
taking voltage differences. In addition, logarithmic compression may be 
used with overlapping cells, i.e. the tape edge can be estimated within a 
fixed offset from the center of the edge. 
FIG. 3A, as previously described, shows how logarithmic compression is 
used. The difference between the logarithms of the "light" and "dark" 
levels represents the logarithm of the contrast ratio of the "light" and 
"dark" levels This number is simply defined as the contrast itself. The 
contrast can also be defined as the difference between light intensities; 
however, the relative contrast ratio number used here is independent of 
the illumination level. Instead, it depends on the translucency of the 
tape, the reflectivity of the semiconductor surface, its coating, etc. A 
recording tape is normally manufactured with a specification for a maximum 
light transmittance of approximately 2% of the incident light. 
FIG. 6 shows two curves, "VINPUT" and "VOUTPUT" computed on the basis of a 
typical minimum contrast. The axis labelled "n" represents the pixel 
numbers in the Y-direction in FIG. 2. The Y-axis in FIG. 6 is in units of 
thermal voltages, i.e. kT/q. "VINPUT" is the signal input directly after 
logarithmic compression. For n-numbers up to six, the signal level has 
been chosen as zero. For n greater or equal to 10, the signal level is 
maximum. The step size between pixels is one-fourth of the width of the 
pixel as shown in FIG. 2. The "VINPUT" signal will deviate from the ideal 
one shown here. Such deviations can be considered as noise in the system. 
The "VOUTPUT" signal, in FIG. 6 shown for a space constant of 2, will be 
less sensitive to these variations. Therefore, "VOUTPUT" is a spatially 
filtered version of "VINPUT", and the spatial noise can be partially 
removed from the input signal. 
FIG. 7 shows "VOUTPUT" from FIG. 6 after being spatially differentiated by 
taking the voltage differences from n-(n-1). The tape edge location has 
been estimated at n=7, within a distance of nearly one kT/q from the 
output at n=8. If the differentiation had been performed directly on the 
"VINPUT" signal in FIG. 6, the distance would have been 8.97 units of 
kT/q. By reducing the space constant slightly, a distance of, for example, 
2 kT/q can easily be obtained. The sample points in the signal in FIG. 7 
will be the inputs to a transconductance differential amplifier 42, as 
shown in FIGS. 3A and 3B. These amplifiers 42 are significantly non-linear 
if the differential input is greater than a few kT/q. 
FIG. 8 is an example of such nonlinearity. The actual current output 
corresponding to the samples in FIG. 7 are strongly compressed if the 
input exceeds approximately 2 kT/q. The absolute levels of the sample 
points in FIG. 7 should not be outside the +/-5 kT/q input range shown in 
FIG. 8. The higher the absolute levels of the individual points, the 
greater the distances must be to separate the points in FIG. 7. 
With a sharp transition from "black" to "white", it is possible to obtain a 
system resolution less than the width of the individual pixels. This is 
important because in a typical semiconductor process with a minimum drawn 
gate width of about 1 .mu.m, the design rule for the minimum distance 
between metal-2 traces specify a window width of 2 .mu.m. In the examples 
of FIG. 2 and FIG. 6, a resolution of one-fourth the width of the pixel 
window has been attained. In principle, a resolution of 0.5 .mu.m could be 
obtained with a 1 .mu.m process, but this will be limited by optical 
diffraction effects. 
FIG. 9 shows a more practical system in which a transition zone from 
"black" to "white" levels will always be present. The "VINPUT" and 
"VOUTPUT" signals when the transition zone goes from n=11 to n=21 is shown 
in the FIG.. The zone was modelled as a linear or "graded" transition from 
"black" to "white". The width of the zone is 1.5 times the width of one 
individual pixel sensor, and each pixel is stepped by one-fourth of the 
cell width. The "VINPUT" graph is therefore the result of integrating the 
light intensity over all nine pixels located within the transition zone 
and setting the light input to the other pixels to "black" and "white" 
levels, respectively. The space constant in FIG. 9 is 2, and the contrast 
ratio is 50 as was used in FIG. 6. 
Spatial averaging, discussed previously, may also be performed using this 
integrated circuit technology. Inputs from an array of photosensor 
circuits (pixels) each contribute a current to a node of a resistive 
network. The actual voltage which develops on the node will be a weighted 
sum of the inputs to all nodes in the network. The weights decrease 
geometrically as the distance from the node increases. Such a network has 
an associated "space constant" and the current drive to each node is 
limited by the control current set for the transconductance amplifiers. 
Therefore, a smooth average, resistant to spuriously bad inputs, is 
computed. 
The realization of a network of very high resistance in a standard CMOS 
process can be done with a so-called "horizontal resistor" which is not a 
resistor in the usual sense, but two MOS pass-transistors where the 
channel resistance can be controlled electronically. These "resistors" 
operate in the sub-threshold range. A very useful property of the 
horizontal resistor is that the current versus voltage graph is non-linear 
and follows a hyperbolic tangent function. The current saturates for input 
voltages greater than about 150 mV. For this reason, it is possible to 
compute smooth, error resistant averages and also to obtain segmentation. 
Both the transconductance amplifiers driving the network and the 
horizontal resistors themselves will saturate when the input voltages 
exceed a few kT/q thermal voltages. When a contrast boundary is present in 
the one-dimensional "picture" input to the photoreceptors, the computed 
node voltages on the horizontal resistive network will reproduce this 
contrast and segments the image into smooth areas. 
FIG. 1, in addition, shows a conventional control apparatus to position the 
edges of the magnetic medium or the data track itself using the dynamic 
position information for the magnetic recording head that is processed 
locally in analog for without the need for digitizing the input. The 
control of the position of the write/read heads 14, 16 is performed by a 
system as disclosed in U.S. Pat. No. 4,679,104 which is herein 
incorporated by reference. During the write operation, the write/read 
heads 14, 16 track either one tape edge or an average position determined 
by both edges. For some tape formats, the lower edge can be used for 
one-half of the tracks and the upper edge for the other half of the 
tracks. This is useful in reducing temperature dependent variations in 
track positions. Based on the actual tape width found and the actual tape 
format in use, the track positions are calculated so that guard bands of 
equal width are created at both tape edges. The actual positions for each 
track are stored in a write table, shown as memory 26. If a 
read-while-write head is used, the read gap lies in-line with the write 
gap, and it will automatically follow the movements of the head which is 
under servo control. 
During the read mode, the initial procedure is slightly different from the 
write procedure in that the exact or optimum position for the head may be 
determined by reference bursts placed prior to the beginning of the data 
tracks for each recording direction on the tape 18. Therefore, the 
tolerant offset between the write gap and the read gap may be eliminated. 
Using the reference burst procedure modified for servo control, the head 
is positioned below the nominal positions of the reference burst, and a 
read operation is started under tape edge servo control. The read signal 
is passed through a band pass filter with a center frequency corresponding 
to the expected frequency from the reference burst. The head is then moved 
upwards under servo control until a threshold detector signals that the 
read gap is over the lower part of the reference burst. The positional 
number for the head is stored in memory 26. Then, the read gap is moved 
well above the reference burst, and the head is moved downwards until the 
threshold detector signals that the upper part of the reference burst has 
been found. A positional number for the reference burst center line is 
then calculated. This center line will coincide with a center line through 
the corresponding data track. A typical system will then correct all of 
the numbers in the track table used for write operation and create a new 
read table. In the current serpentine track formats in use, many reference 
bursts are provided. The number of reference track alignments to use will 
depend on the actual accuracy which was used during writing, i.e. the tape 
edge detector chip 10 and tracking servo system must also read tapes 
written by other competing systems. In a quality tape drive, it is only 
necessary to read two reference bursts, one for each recording channel, 
i.e recording direction, forward or backward. 
When the detector chip 10 has been set up to work in the static mode and 
the positions of the edges are known at the beginning of the tape, one can 
either start to write data in the dynamic tracking mode or can perform a 
new tape qualification procedure. This is required since the tape edges 
may have been damaged during extensive use of the cartridge or there may 
exist production defects on the tape. Tape edge defects indicate that the 
cartridge not be used for high reliability storage of data. Two different 
operations can be performed to determine the qualification of the tape 
before writing data: a "track repeatability test" and a "tape defect 
test." 
The track repeatability test checks the specified dynamic transverse tape 
track movement variation movement. For a 0.250-inch tape cartridge, this 
is typically specified as +/-0.013 mm for the first write pass after tape 
conditioning. The variation in the opposite direction should not exceed 
+/-0.025 mm. The tape edge servo is disabled, and the tape is conditioned 
by running it from the beginning of the tape to the end of the tape, and 
back again to the beginning of the tape. Then, the tape is run in the 
forward direction. The position of both edges are monitored and stored in 
memory as a variable length data array. A spatial low pass filtering 
operation is performed by the servo processor 100, shown in FIG. 14. The 
result is again stored in memory and the raw data is discarded. The low 
pass filtering on the data may be performed when the tape is running by 
including a digital hardware filter (not shown) in the servo processor or 
by implementing the filter in the firm ware. The mean value of the two 
one-dimensional arrays are computed and stored in memory. This mean-value 
array represents the wander of the center line of the tape and the data 
tracks to be written. The deviation is computed between the measured 
center line and a "best fit" line through the same center line. If this 
deviation corresponding to a distance greater than the maximum allowed 
value, the qualification process may be stopped, and a signal indicative 
of faulty transverse tape movement in the forward direction may be 
indicated. The tape is then run in the reverse direction, repeating the 
same procedure and using the actual specification limit. 
The tape defect test provides duo-edge detection in which a first defect 
test pass follows the upper edge of the tape only, and the program module 
examines the lower edge for defects. On a succeeding pass, the servo 
follows the lower edge, and the program module examines the upper edge for 
defects. The firmwear module keeps track of the tape position by counting 
the number of pulses from the capstan motor. 
The method of the tape defect test is performed as follows: the tape speed 
is reduced well below the normal operating speed to allow for a proper 
spatial sampling of the possible edge defects. Raw data points are passed 
through a band-pass filter in the control, filtering and digital servo 
processor 100, as shown in FIG. 14. A threshold detector function follows 
the band-pass filter. The band-pass filter may be a combined low-pass one 
with a high cut-off frequency and a high-pass one with a low cut-off 
frequency. The low-pass function is needed to remove spurious noise in 
single data points, and the high pass function is used to remove the 
low-frequency content, i.e. the running average value of the tape edge 
position. If the tracks repeatability test is run first, the running 
average value for the tape edge position is computed and is stored in 
memory and that can then be used to remove the average value when 
searching for defects. This is done with high accuracy since measurements 
have shown that transversal movements of the tape tend to be reproducible 
if the tape has been properly reconditioned before the test starts. 
When a filtered signal exceeds a certain threshold, the signal is stored in 
memory together with the tape position. The defect storing routine also 
has both "pre-trig" and "post-trig" functions, so that a certain number of 
data points are stored both prior to the trig-in and after the trig-out 
points from the threshold detector. The defect data can then later be 
transmitted off-drive for external analyses or human inspection. 
The method may also be performed by stopping the tape drive and rewinding 
the tape slowly to the actual position where the defect occurred so that 
the defect is positioned just in front of the magnetic recording head. The 
user can then remove the tape cartridge from the drive and inspect the 
tape manually and decide if the defect is so severe that the tape must be 
discarded. 
In addition to the tape edge tracking servo discussed above, automatic 
adjustment of the recording head may be performed by monitoring the 
azimuth angle. The azimuth angle, as is known in the art, is the deviation 
formed from an imaginary center line through and parallel to the write and 
read gaps and normal to the edge of the tape. Ideally, this angle is 
0.degree., or it can be set at a different angle if azimuth recording is 
intentionally used to reduce cross-coupling between tracks. This automatic 
adjustment is performed by placing two additional, stand alone 
photosensors 11, 13 as shown in FIG. 1 at the upper edge of the tape 18 
and inside the outer edges of the tape edge sensor chip 10. An imaginary 
line running between the two sensors 11, 13 corresponds to one edge of the 
tape 18, the upper edge as shown in FIG. 1. The sensitive areas of sensors 
11, 13 is very small such that uniform light intensity is present within 
two local regions. Each region contains two light sensitive elements, 11A, 
11B and 13A, 13B. The distance between the two sensors 11, 13 is 
designated by "1" in FIG. 16. The outputs from 11A, 11B and 13A, 13B are 
compared independently by local threshold differential amplifiers included 
on the tape edge detector chip 10. 
The outputs of sensors 11A, 11B and 13A, 13B are provided to a circuit as 
shown in FIG. 3C. Floating gate techniques may be used to correct for the 
total, effective offset between the "REF" and "SENSE" elements in sensors 
13 when they are illuminated with constant light density during the 
testing process of the detector chip itself. 
The azimuth adjustment is performed by inserting a tape parallel to the 
reference plane of the cartridge (within small tolerances). The magnetic 
head 20 is mounted within a tolerance which is later fine tuned. 
To avoid the tape 18 sticking to the head 20 when the head 20 is moved, the 
tape 18 may be moved back and forth during an adjustment procedure. The 
magnetic head 20 and the detector chip 10 are moved upwards (or downwards 
if sensors 11, 13 are positioned for the lower edge) so that both sensors 
11, 13 are illuminated causing logical "high" outputs from both sensors 
11, 13. The offset voltage built into the differential amplifier or 
threshold comparator in FIG. 3C is of a polarity which ensures that the 
output is "high" when the input voltages from the "REF" and "SENSE 
phototransistors are equal or within an offset voltage. Therefore, the 
output from the detector circuit in FIG. 3C is "high" when both the "REF" 
11A, 13A and "SENSE" 11B, 13B elements are illuminated, either with 
"light" or "black" intensities. Then the head 20 is moved downwards until 
one (or both, if the azimuth angle is incidentally near 0) output goes 
"low". Such an embodiment is shown in FIG. 15 where the output from 
elements 11A, 11B are at "black" levels because the sensor was moved too 
great of a distance downwards. The head 20 is now moved upwards again 
until 11A is illuminated and 11B is in the shadow. The output from sensor 
11 will then be "low". Hence, a "low" signal is provided by the circuit in 
FIG. 3C only when the tape edge lies between the "REF" 11A, 13A and 
"SENSE" 11B, 13B elements. When the head 20 and the detector chip 10 are 
rotated an angle, the sensors 11, 13 are at positions 11' and 13', as 
shown in FIG. 15, both having "low" output signals. If the initial azimuth 
error is large or the geometrical distance "1" between the sensors 11, 13 
is large, the outputs from the sensors 11, 13 may also both be "high" 
during the rotation as shown in FIG. 16. If the center of rotation "0" is 
placed as shown in FIG. 16, i.e. if the nominal positions of sensors 11, 
13 are placed symmetrical with respect to the line through "0" 
perpendicular to the reference plane, then the head 20 may be rotated to 
an azimuth angle of opposite sign and stop when the sensor 13 goes "low". 
The total angle is measured during this procedure, and the head 20 is 
rotated back again to the first position or beyond taking care of any 
hysteresis, if necessary. Then, a new rotation is started with the known 
half-angle calculated by the azimuth control system. 
Another embodiment allows for a stepper motor 24 to move the head 20 upward 
so that the tape edge follows the sensor 11 and causes its output to go 
"low". This tracking is continued until the sensor 13 goes "low". 
In FIG. 16, the following distances were used in calculation. The distance 
between sensors 11, 13 is l=2.5 mm. An azimuth angle of 2.9 mrad is used, 
and a delta vertical, dv, of 7.25 .mu.m results. The azimuth detector 
therefore is required to have a resolution better than dv. A standard, 
digital CMOS process of current technology defining openings of 3 .mu.m by 
3 .mu.m is possible. 
Although various minor changes and modifications might be proposed by those 
skilled in the art, it will be understood that I wish to include within 
the claims of the patent warranted hereon all such changes and 
modifications as reasonably come within my contribution to the art.