Method of stepwise voltage control for supplying an induction motor

A method for stepwise voltage control using thyristors for supplying an induction motor at a fixed frequency and a variable voltage. A time sequence of electromotive force amplitudes corresponding to a predetermined law for torque variation during transient operating conditions is stored. Voltages applied to the motor and the current flow are determined to approximate the electromotive force generated by the motor. Static switch conduction intervals are used to adapt the electromotive force to the stored amplitude during the transient operating conditions.

This application is a 371 of PCT/FR95/00817 filed on Jun. 20, 1995. 
BACKGROUND OF THE INVENTION 
1. Field of the Invention 
The present invention concerns a method of stepwise voltage control using 
thyristors for supplying an induction motor at a fixed frequency and at a 
variable voltage, more particularly in the situation in which the load 
that it drives has a quadratic torque-speed law, noteworthy in that it 
combines great simplicity of the equipment with a level of performance in 
terms of progressive stopping and starting that has hitherto been 
unattainable using more complex equipments such as frequency converters. 
2. Discussion of the Background 
Electronic starters for induction motors have achieved significant 
penetration of the industrial market in the last ten years, replacing 
older generation electrotechnical hardware such as star-delta switches, 
autotransformers and liquid rheostats. Their success is related to 
advances in power semiconductors (increased reliability and economic 
competitivity), on the one hand, and to advances in digital control 
circuits having an increasing processing capacity, on the other hand. 
Among the various electronic speed variation systems, the electronic 
starter represents the simplest means of controlling the speed of 
induction motors: in a system of this kind, the power circuit or 
three-phase stepwise voltage controller typically comprises a set of three 
alternating current switches each in series with one phase between the AC 
line voltage and the motor, each switch typically comprising two 
thyristors connected in anti-parallel. The fact that these switches change 
state at a low frequency, namely that of the AC line voltage, and do not 
require any turn-off control, as turn-off occurs naturally when the 
current passes through zero, also simplifies the control circuits. 
However, these starters can only control the amplitude of the motor 
voltage, at the fixed frequency of the AC line voltage. By comparison, 
frequency converters have a more complex structure, as much from the point 
of view of the power circuit, typically comprising a three-phase power 
transistor bridge with the rupture capacity needed to switch the motor 
current at a frequency of at least a few kilohertz, as from that of the 
control circuit, which has to generate variable voltage and frequency 
waves by pulse width modulation. To compensate this, they have a second 
control quantity, namely the frequency, which can be varied independently 
of the magnetic flux and the torque to optimize the operation of the motor 
at each point, in particular with regard to slip and losses. As a result, 
applications are divided between frequency converters and electronic 
starters, the former offering high performance and the latter moderate 
cost. 
Given the trend for increasingly higher performance of digital control 
circuits, it has now become possible to expand the field of applications 
of electronic starters into that of frequency converters. For example, 
starting and stopping pumps represents a particular problem due to the 
existence of mechanical resonance that is manifested on the occasion of a 
rapid variation in flowrate by oscillation of the fluid in the pipe, known 
as "water hammer". This phenomenon is harmful as much through its 
reduction of the service life of the installation as through the 
accompanying acoustic noise. This noise is particularly unacceptable in 
water distribution installations in urban areas. Previous means of solving 
this problem tend to eliminate all sudden variation in the flow, and 
therefore in the speed, during stopping or starting. They use two 
techniques: 
modulation of the flowrate by a progressive action solenoid valve; 
variation of the speed of the pump by a speed regulator, generally 
consisting of a frequency converter and an alternating current motor. 
Both methods have the same drawback, namely high cost, increasingly so with 
increasing power levels, the latter being routinely between 10 kW and 500 
kW. Efforts to date to eliminate "water hammer" when stopping a pump, by 
regular deceleration commanded by a basic electronic starter, have failed. 
It is well known that varying the voltage at constant frequency introduces 
a discontinuity into the voltage-speed characteristic of the induction 
motor: below a certain speed which additionally depends on the 
speed-torque characteristic of the load but which, in the case of a pump, 
can be as high as two-thirds the nominal speed, operation becomes unstable 
and during deceleration the motor "stalls" and suddenly drops to a low 
speed, while during acceleration the motor "runs away" and is out of 
control between a low speed and the stable operating speed. This behavior 
causes "water hammer" and the electronic starter has therefore proved to 
be unsuitable for solving this problem. 
SUMMARY OF THE INVENTION 
The aim of the present invention is to propose an electronic starter 
control method that assures stable behavior of the starter over all of the 
range of speeds between zero and the nominal speed and which, in the 
presence of a load with a quadratic torque-speed characteristic, is 
capable of starting and slowing performance comparable with that of a 
frequency converter and eliminates the phenomenon of "water hammer" during 
slowing of a pump, under more advantageous economic conditions. 
The method of the invention is intended for a stepwise voltage control type 
power circuit such as that made up of a three-phase system of three static 
switches each of which comprises two thyristors connected in 
anti-parallel, placed between the AC line voltage and the induction motor 
supplied with power, each in series with one phase of the AC line voltage, 
or possibly of the motor alone in the case of a delta connected motor. It 
uses a phase variation thyristor control system, known in itself, for 
example from "Induction machine SCR voltage reduction; optimized control 
and dynamic modelling", by A. P. Van den Bossche and J. A. Melkebeeke, IEE 
Conference Publication Number 234, London 1984. Note however, that the 
method of varying the motor voltage described therein, by adjusting the 
angle of non-conduction of the switches, is to be understood as 
constituting only one example and can be replaced by any other method, 
such as the more conventional method in which the control magnitude is the 
phase at which the switches are turned on relative to the phase of their 
supply voltage. 
The invention therefore proposes a method for time control of transient 
operating conditions of a multiphase induction motor driving a load the 
resisting torque of which varies with the speed in accordance with a known 
law and supplied at variable voltage via static switches having periods of 
conduction of variable duration, characterized in that, having memorized a 
time succession of rotor electromotive force amplitudes corresponding to a 
predetermined law of torque variation under transient operating 
conditions, the voltages applied to the motor and the current passing 
through it are determined, the amplitude of the rotor electromotive force 
developed by the motor is at least approximately deduced and the 
conduction intervals of the static switches are varied to adjust the 
amplitude of the rotor electromotive force developed to the corresponding 
amplitude memorized under the transient operating conditions. 
In accordance with an advantageous development of the invention, the 
voltages applied to the motor and the current flowing through it are 
sampled in a substantially synchronous manner and the sampled values are 
converted into digital signals, the rotor electromotive force amplitude 
being deduced from the aforementioned digital signals by computation in 
the digital domain, the rotor electromotive force being treated as a 
vector. 
This approach exploits the advantageous performance of commercially 
available digital control circuits. 
In accordance with another aspect of the invention, the set point value of 
the rotor electromotive force or its approximate expression is determined 
by a time law adapted to produce the required acceleration or 
deceleration. This law is based on the torque-speed characteristic of the 
driven load, and possibly on its inertia, by application of the following 
formula: 
EQU e=(c/g).sup.1/2 
where 
e is the rotor electromotive force, 
g is the slip, 
c is the motor torque, 
e, g and c being expressed as a proportion of the nominal rotor 
electromotive force, slip and torque of the motor, respectively. 
The features and advantages of the invention will emerge from the following 
description given by way of example with reference to the accompanying 
drawings in which FIGS. 1 and 2 show two complementary parts of a block 
diagram showing the application of the invention to controlling a 
three-phase induction motor. 
In the selected embodiment shown, a short-circuit rotor three-phase 
induction motor M has three stator windings connected in a star 
configuration and supplied with power via terminals S1, S2 and S3. The 
latter are connected to the conductors of a three-phase AC line voltage 
V1, V2, V3 through respective switches I1, I2, I3 each made up of a pair 
of thyristors connected in anti-parallel. The triggers of the thyristors 
are driven by the output of respective pulse transformers T1, T2 and T3. 
Current transformers TI1, TI2 and TI3 are connected in series with the 
three phases.

DISCUSSION OF THE PREFERRED EMBODIMENTS 
The three subassemblies are described in turn hereinafter, noting that they 
include digital processing elements although for clarity these are 
represented by means of hardwired logic symbols. 
A) Measuring the Electromotive Force 
Three differential amplifiers 11, 12, 13 each associated with a respective 
attenuator network A1, A2, A3 with an attenuation approximately equal to 
100 and each made up of four resistors, deliver three voltages W1, W2, W3 
imaging the three voltages between phases of the motor M1. Likewise the 
amplifier 10 supplies a voltage I imaging the rectified value of the 
three-phase current from M provided by the secondary currents of TI1, TI2 
and TI3 flowing through the three-phase rectifier bridge 14 and a load 
resistor 15. The four signals I, W1, W2 and W3 are sampled by four sample 
and hold circuits 16, 17, 18 and 19, respectively, and passed to an 
analog-digital converter 20. The sampling time for each of the three 
switches I1, I2 or I3, near the middle of a non-conduction interval, is 
defined by the OR logic function 30, as explained further below. Dividing 
them by the nominal value of the voltage between phases Wn at 21, 22 and 
23 reduces the three voltages W1, W2, W3 to the values w1, w2, w3. 
Dividing it by the nominal value In of the current from M (24) reduces I 
to the value i. The rotor electromotive force e is calculated (25) from 
w1, w2, w3 and i using the equation: 
EQU e={1/31/3(wj+1-wj-1).sup.2 +(wj-0.02 i).sup.2 !}.sup.1/2 
B) Production of the Electromotive Force Set Point e ref 
Consider first deceleration to a stop within a time Ta. The value of e ref 
programmed in a memory 31 at the address n is written: 
EQU e ref(n)=0.2(1-n/m)(n/m).sup.-1/2 
with: 
EQU 0&lt;n.ltoreq.m 
and: 
EQU e.ltoreq.1 
m being the total numbers of registers used in the memory 31 to define the 
law in accordance with which e ref varies during stopping. The stop time 
Ta is varied by means of a binary switch C1, the set value c1 of which 
initializes the counter 32 on each passage through zero. This counter 
therefore divides the frequency f of an internal clock CLK to which it is 
connected by a switch 33 by c1. In each period of the frequency obtained 
in this way, the address of the memory 31 is incremented by one unit via 
the OR logic function 34. The theoretical stopping time Ta is given by: 
EQU Ta=m(c1/f) 
At the start of the stopping process, the value of n is initialized during 
a phase having a duration of several tens of milliseconds, defined by a 
monostable function 35 activated by a pushbutton 35a. During this phase, 
the switch 33 routes the frequency f to a switch 36 which increments via 
the OR function 34 or decrements the address n of the memory, depending on 
whether the sign of the error 
EQU e.sub.0 =e ref(n)-e 
is positive or negative at the output of the summing device 37 which 
controls the switch 36 using a comparator 38. The error signal e.sub.0 is 
additionally applied to a proportional-integral regulator 40 shunted by a 
switch 39 which prevents it acting during the time period defined by the 
monostable 35. From the initial value n0 obtained, n is incremented during 
stopping by one unit, at a period of c1/f, up to the maximal value m. 
The equation (2) given above for deceleration is justified as follows in 
the case considered here of a quadradic load. 
Starting from the equation e=(c/g).sup.1/2 with e, c and g having the 
relative values already mentioned above, consider the speed w of the 
motor, also relative to the nominal value of that speed. This relative 
speed value is therefore between 0 and 1, like the other relative values 
e, c and g already considered. 
The quadratic nature of the load in question (pump) implies: 
EQU c=w.sup.2 
whence 
EQU e=w/g.sup.1/2 
Consider now the nominal slip a of the motor, the value of which is 
routinely around 0.04. 
The relation between w and g is written 
EQU w(1-a)=1-ag 
where: 
EQU g=1-w(1-a)!/a 
whence: 
EQU e=a.sup.1/2 w1-w(1-a)!.sup.-1/2 
Taking a =0.04 and taking a to be negligible compared to 1: 
EQU e=0.2 w(1-w).sup.-1/2 
For starting at constant acceleration it is possible to write w=t, the time 
t being expressed relative to the start time. 
The following then applies: 
EQU e=0.2 t(1-t).sup.-1/2 
For stopping at constant deceleration it is possible to write: 
EQU w=1-t 
t being relative to the stopping time, whence: 
EQU e=0.2(1-t)t.sup.-1/2 
which justifies the equation (2) with: 
EQU n/m=t 
C) Thyristor Control (FIG. 2) 
The proportional-integral action regulator 40 that receives as input the 
error e.sub.0 delivers a control signal d that defines the angle of 
non-conduction of the switches Ij for each half-period of the AC line 
voltage. The extinction of Ij is observed with the aid of one of three 
circuits made up of an attenuator network A11, A12, A13, a double 
comparator B11, B12, B13 and an OR circuit C11, C12, C13. During 
conduction the two outputs of the double comparator are low because of the 
effect of the -P bias at its input, which introduces a threshold 
corresponding to a few tens of volts at the terminals of the switches. 
Extinction is manifested in the appearance of a voltage at the terminals 
of the switch which, as soon as it crosses the threshold imposed by the 
bias -P, causes one or other output of the double comparator and that of 
the OR circuit to change to the high state, depending on its sign. The 
monostable D1, D2 or D3 generates a short pulse at extinction that 
initializes two groups of programmable monostables F1, F2, F3 and G1, G2, 
G3. The time-delay of the first group is equal to the value defined by the 
angle d/2 supplied by the regulator 40 and a divider 41. At the end of 
this time-delay the monostable F1, F2 or F3 goes high and the monostable 
H1, H2 or H3 generates a short pulse which triggers the sample and hold 
circuits 16 through 19 via the OR function 30. The time-delay of the 
second group is equal to the value defined by the angle d defined by the 
regulator 40. The time-delay signals delivered in the form of short pulses 
by monostables K1, K2 or K3 when G1, G2 or G3 goes high are applied to the 
primary of T1, T2 or T3 and cause the corresponding switches to conduct. 
As each switch is commanded in this way on each half-period of the AC line 
voltage, the stepwise voltage controller delivers the required voltage to 
the motor. 
Of course, the invention is not limited to the example described, but 
encompasses all variant executions thereof within the scope of the claims.