Vibration wave motor

At least two vibration detection electro-mechanical energy conversion element areas substantially centered on the antinodes of both the standing waves are arranged between the group of driving electro-mechanical energy conversion element areas. Amplitudes of two standing waves generated upon application of AC voltages to these groups and a time-phase difference between the standing waves can be accurately detected. The amplitudes of the standing waves can be made equal to each other, and the time-phase difference can be set to be .pi./2, thereby obtaining an ideal vibration state of a travelling vibration wave generated in a vibration plate.

BACKGROUND OF THE INVENTION: 
1. Field of the Invention 
This invention relates to a vibration wave motor which is frictionally 
driven by a travelling vibration wave generated in an elastic vibration 
member. 
2. Related Background Art 
Vibration wave motors for frictionally driving a moving member by utilizing 
a travelling vibration wave are proposed in, e.g., U.S. Pat. Nos. 
4,580,073 and 4,484,099. These motors have been commercially available in 
recent years. The principle of operation of the vibration wave motor will 
be described below. 
Two groups each consisting of a plurality of piezoelectric elements are 
fixed on one surface of a ring-like elastic vibration plate having a 
circumferential length which is an integer multiple of a given length 
.lambda. to constitute a stator. These piezoelectric elements are normally 
arranged at .lambda./2 pitches in each group and alternately have opposite 
polarities. The piezoelectric elements in the groups are offset by an odd 
number multiple of .lambda./4. Electrode films are formed on the 
piezoelectric elements of the groups, respectively. When an AC voltage is 
applied to any one of the groups, a standing wave (wavelength: .lambda.) 
of flexural vibrations is generated throughout the entire circumference of 
the vibration plate such that antinode positions are located at the 
central positions of the piezoelectric elements of the group and positions 
away from the central positions every .lambda./2 intervals, and nodes are 
located at the central positions between the antinodes. When an AC voltage 
is applied to the remaining group, a similar standing wave is generated. 
However, in this case, the positions of the antinodes and the nodes are 
offset from the first standing wave by .lambda./4. When AC voltages having 
a positional difference of .pi./2 as a function of time and having the 
same frequencies are simultaneously applied to both the groups, two 
standing waves are combined to generate a travelling wave 
(wavelength:.lambda.) of flexural vibrations in the circumferential 
direction of the vibration plate. In this case, the respective points on 
the other surface of the vibration plate having a predetermined thickness 
are subjected to a kind of elliptical motion. If a ring-like moving member 
serving as a rotor is brought into tight contact with the other surface of 
the vibration plate, the moving member receives a circumferential 
frictional force from the vibration plate and is rotated. The direction of 
rotation can be reversed by changing a positive phase difference between 
the AC voltages applied to both the groups into a negative difference, and 
vice versa. The above description is concerned with the principle of 
operation of a vibration wave motor of this type. 
A driver circuit in a conventional vibration wave motor of this type is 
proposed in, e.g., Japanese Pat. Laid-Open (Kokai) No. 61-157276, U.S. 
Pat. No. 4,501,411, and Japanese Pat. Laid-Open (Kokai) No. 59-156169. One 
vibration detection piezoelectric element is fixed on the other one of the 
groups of piezoelectric elements (these elements are referred to as 
driving piezoelectric elements), and a frequency of the AC voltage applied 
to the driving piezoelectric elements is automatically changed into a 
resonance frequency in accordance with a detection output from the 
detection piezoelectric element, thereby improving efficiency of the 
vibration wave motor. 
In the vibration wave motor described above, however, the vibration 
detection piezoelectric element is fixed at the same spatial phase 
position as that of one of the groups of driving piezoelectric elements. 
More specifically, since the central point of the vibration detection 
piezoelectric element is located at a position offset from the central 
point of one area of the group of driving piezoelectric elements by a 
integer multiple of .lambda./2, thus posing the following problems. 
First, since the frequency characteristics of standing waves generated upon 
application of an AC voltage to the groups differ from each other due to 
the vibration detection piezoelectric element located at the same spatial 
phase position as that of one of the groups of driving piezoelectric 
elements, the vibration detection piezoelectric element can detect only a 
vibration state of the standing wave generated by applying the AC voltage 
to one group of driving piezoelectric elements. 
Second, a time-phase difference between both the standing waves generated 
upon application of the AC voltages to both the groups and an amplitude of 
one of the standing waves must be controlled by an open loop due to the 
first reason. The time-phases of the standing waves are greatly shifted 
from .pi./2, and a difference between the amplitudes of the standing waves 
is increased. As a result, the amplitude of the travelling wave greatly 
varies, thus causing degradation of efficiency and unstable rotation of 
the motor. 
SUMMARY OF THE INVENTION 
It is an object of the present invention to solve the conventional problems 
described above. 
It is another object of the present invention to provide a vibration wave 
motor or an actuator wherein a vibration state of a travelling vibration 
wave generated by a vibration plate becomes ideal, and driving efficiency 
and rotational stability of the motor can be improved. 
In order to achieve the above objects of the present invention, there is 
provided a vibration wave motor for frictionally driving with a travelling 
vibration wave a moving member brought into tight contact with the 
vibration plate, wherein at least two vibration detection 
electro-mechanical energy conversion element areas substantially centered 
on the antinodes of both the standing waves are arranged between the 
groups of driving electro-mechanical energy conversion element areas in 
said vibration plate.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
FIG. 1 shows part of a stator according to the first embodiment of the 
present invention. The stator in FIG. 1 includes a ring-like vibration 
plate 1 and a piezoelectric element plate 2 consisting of a piezoelectric 
ceramic material or the like and fixed to the vibration plate 1. The 
piezoelectric element plate 2 serves as an electro-mechanical energy 
conversion element. Electrodes A.sub.1 and A.sub.2 are arranged for a 
first group (to be referred to as an A phase) of piezoelectric elements, 
and electrodes B.sub.1 and B.sub.2 are arranged for a second group (to be 
referred to as a B phase) of piezoelectric elements An electrode S.sub.A 
is arranged for a piezoelectric element for detecting a vibration (to be 
referred to as an S.sub.A -phase piezoelectric element hereinafter) of a 
standing wave generated upon application of an AC voltage to the A-phase 
electrodes (this standing wave is referred to as an A-phase standing wave 
hereinafter). Similarly, an electrode S.sub.B is arranged for a 
piezoelectric element for detecting a vibration (to be referred to as an 
S.sub.B -phase piezoelectric element hereinafter) of a standing wave 
generated upon application of an AC voltage to the B-phase electrodes 
(this standing wave is referred to as a B-phase standing wave 
hereinafter). Electrodes G.sub.1 and G.sub.2 are short-circuited with the 
lower surface of the piezoelectric element plate, and therefore serve as 
ground electrodes. The piezoelectric element plate portions corresponding 
to the above electrodes are polarized in advance to constitute divided 
piezoelectric element groups. The A-phase electrodes A.sub.1 and A.sub.2 
are arranged at pitches of 1/2 wavelength, and the polarization directions 
of the corresponding piezoelectric element areas are alternately opposite 
to each other. The A-phase piezoelectric element group is phase-shifted 
from the B-phase piezoelectric element group by 90.degree., i.e., a 1/4 
wavelength. The lengths of the S.sub.A - and S.sub.B -phase piezoelectric 
elements and the electrodes S.sub.A and S.sub.B are equal to each other, 
i.e., a 1/4 wavelength. Identical AC voltages are applied to the A- and 
B-phase electrodes to simultaneously generate A- and B-phase standing 
waves, but these waves have a time-phase difference of 90.degree.. As a 
result, a composite wave becomes a travelling vibration wave. 
A central position a-a' of the electrode A.sub.1 is an antinode of the 
A-phase standing wave and a node of the B-phase standing node. A central 
position b-b' (shifted by a 5/4 wavelength from the position a-a') of the 
electrode B.sub.1 is an antinode of the B-phase standing wave and a node 
of the A-phase standing wave. A central position c-c' of the vibration 
detection electrode S.sub.A (S.sub.A -phase electrode) is a position 
shifted from the position a-a' by a 1/2 wavelength A central position d-d' 
of the vibration detection electrode S.sub.B (S.sub.B -phase electrode) is 
a position shifted from the position b-b' by a 1/2 wavelength. Therefore, 
the position c-c' is an antinode of the A-phase standing wave and the 
position d-d' is an antinode of the B-phase standing wave. 
If the position a-a' is plotted as the origin of the circumferential 
coordinates x, a waveform V.sub.A of the A-phase standing wave and a 
waveform V.sub.B of the B-phase standing wave are given as follows: 
##EQU1## 
where .lambda. is the wavelength, .omega. is an angular frequency, t is 
arbitrary time, and .theta. is a time-phase difference between both the 
standing waves. Therefore, the 
Outputs V.sub.SA and V.sub.SB from the A- and B-phase electrodes S.sub.A 
and S.sub.B are given as follows: 
EQU V.sub.SA =KV.sub.A cos.omega.t 
EQU V.sub.SB =KV.sub.B cos(.omega.t+.theta.) 
where K is a proportional constant 
The AC voltages are respectively applied to the A-and B-phase electrodes 
S.sub.A and S.sub.B to satisfy the following conditions: 
EQU .vertline.V.sub.SA .vertline.=.vertline.V.sub.SB .vertline. 
EQU .angle.V.sub.SA -.angle.V.sub.SB =.theta.=.pi./2 
therefore, the waveform of the travelling wave can be ideal as follows: 
EQU V.sub.A +V.sub.B =V.sub.A cos(2.pi.x/.lambda.- .omega.t) 
A driver circuit for equalizing the amplitude of the S.sub.A -phase 
detection voltage with that of the S.sub.B -phase detection voltage and 
providing a time-phase difference of .pi./2 between the detection voltages 
is shown in FIG. 2. 
Referring to FIG. 2, the driver circuit includes comparators 3, 4, 5, and 
6, exclusive OR gates 7, 8, 9, and 10, known AC-DC converters (AC/DC) 11 
and 12, a 2-input analog adder 13, amplifiers 14, 15, low-pass filters 
(LPFs) 16, 17, 18 and 19, voltage-controlled ocsillators (VCOs) 20 and 21, 
and voltage-controlled amplifiers (VCAs) 22 and 23. 
The comparator 3 converts an S.sub.A -phase signal from the vibration 
detection electrode S.sub.A into a logical level (voltages corresponding 
to binary values are +V/2 and -V/2). The comparator 4 converts an S.sub.B 
-phase signal from the vibration detection electrode S.sub.B into a logic 
level. The comparator 5 converts an A-phase driving voltage applied to the 
A-phase electrode into a logic level, and the comparator 6 converts a 
B-phase driving voltage applied to the B-phase electrode into a logic 
level. 
A truth table of the exclusive OR gates 7, 8, 9, and 10 is shown in FIG. 3. 
Each of the exclusive OR gates 7, 8, and 9 detects a phase difference 
between two corresponding inputs. FIG. 4 shows an output corresponding to 
the detected phase differences. As shown in FIG. 4, an average value 
V.sub.XOR of the outputs generated by the exclusive OR gates and 
representing the phase differences is changed linearly as a function of 
the phase difference. 
The exclusive OR gate 7 detects a phase difference between the S.sub.A - 
and S.sub.B -phase signals. The exclusive OR gate 8 detects a phase 
difference between the S.sub.A -phase signal and the A-phase driving 
voltage. The exclusive OR gate 9 detects a phase difference between the 
S.sub.B -phase signal and the B-phase driving voltage. The exclusive OR 
gate 10 switches inversion/noninversion of an output from the exclusive OR 
gate 7 in accordance with a rotational direction switching signal A signal 
of the phase difference between the S.sub.A - and S.sub.B -phase signals 
is input from the exclusive OR gate 7 to the voltage-controlled oscillator 
(VCO) 20 through the exclusive OR gate 10 and the low-pass filter (LPF) 
16. The phase of the output signal from the voltage-controlled oscillator 
(VCO) 20 is controlled such that an average value of the outputs from the 
exclusive OR gate 7 becomes zero. In other words, the phase shift of the 
S.sub.A -phase signal from the S.sub.B -phase signal is controlled to be 
set as +.pi./2 or -.pi./2. Switching of the sign of the phase difference 
is performed by the rotational direction switching signal The phase 
difference between the A-phase driving voltage and the S.sub.A -phase 
signal, which is detected by the exclusive OR gate 8, and the phase 
difference between the B-phase driving voltage and the S.sub.B -phase 
signal, detected by the exclusive OR gate 9, are given as shown in FIG. 5. 
Since the phase differences detected by the exclusive OR gates 8 and 9 are 
detected with reference to -.pi./2, an output from the 2-input analog 
adder 13 is set to be zero between an A-phase (B-phase) resonance 
frequency F.sub.1 and a B-phase (A-phase) resonance frequency F.sub.2, as 
shown in FIG. 6. 
A sum signal of the phase difference between the A-phase driving voltage 
and the S.sub.A -phase signal and the phase difference between the B-phase 
driving voltage and the S.sub.B -phase signal is supplied from the analog 
adder 13 to the voltage-controlled oscillator (VCO) 21 through the 
low-pass filter (LPF) 19. Therefore, the frequency is controlled such that 
the sum of the phase differences becomes zero. AC-DC converters (AC/DC) 11 
and 12 convert the amplitudes (e.g., root-mean-square values, average 
values, or peak values) of the S.sub.a - and S.sub.B -phase signals into 
DC amplitude signals, respectively. The DC amplitude signals of the 
S.sub.A - and S.sub.B -phase signals are input to the amplifiers 14 and 
15, respectively. Differences between the S-phase amplitude setting signal 
and the respective amplitude signals are amplified with high gains. The 
amplified signals are supplied to the voltage-controlled amplifiers (VCAs) 
22 and 23 through the low-pass filters (LPFs) 17 and 18, respectively The 
gains of the voltage-controlled amplifiers (VCAs) 22 and 23 are controlled 
such that the amplitudes of the S.sub.A - and S.sub.B -phase signals are 
set to be amplitudes determined by the S-phase amplitude setting signals. 
FIG. 7 shows the electrode layout according to the first embodiment of the 
present invention. Referring to FIG. 7, the electrode structure includes 
A-phase driving electrodes A.sub.1, A.sub.2, A.sub.3, A.sub.4, and 
A.sub.5, B-phase driving electrodes B.sub.1, B.sub.2, B.sub.3, B.sub.4, 
and B.sub.5, electrodes G.sub.1, G.sub.2, and G.sub.3 which are rendered 
conductive with the lower surface, i.e., ground electrodes, an A-phase 
standing wave vibration detection electrode S.sub.A, and B-phase standing 
wave vibration detection electrode S.sub.B. The areas of the A-phase 
driving electrodes A.sub.1 to A.sub.5 and the B-phase driving electrodes 
B.sub.1 to B.sub.5 are as large as possible in view of vibration 
efficiency. 
An operation of the first embodiment will be described in detail mainly 
with reference to FIG. 2. FIG. 2 shows a driver circuit consisting of 
control circuit blocks. The first block comprises the comparators 3 and 4, 
the exclusive OR gates 7 and 10, the low-pass filter (LPF) 16, and the 
voltage-controlled oscillator (VCO) 20. The first block controls the phase 
shift of the A-phase driving voltage from the B-phase driving voltage such 
that the phase shift of the S.sub.A -phase signal from the S.sub.B -phase 
signal is set to be .pi./2 or -.pi./2. The second block comprises the 
AC-DC converters (AC/DC) 11 and 12, the amplifiers 14, and 15, the 
low-pass filters (LPFs) 17 and 18, and the voltage-controlled amplifiers 
(VCAs) 22 and 23. The second block controls the amplitudes of the A- and 
B-phase driving voltages such that the amplitudes of the S.sub.A - and 
S.sub.B -phase signals are set to be the amplitudes designated by 
rotational speed control signals The third block comprises the comparators 
3, 4, 5, and 6, the exclusive OR gates 8 and 9, the analog adder 13, the 
low-pass filter (LPF) 19, and the voltage-controlled oscillator (VCO) 21. 
The third block controls the frequency of the B-phase driving voltage such 
that the frequency of the B-phase driving voltage is set to be an 
intermediate frequency between the A- and B-phase resonance frequencies. 
The three blocks will be sequentially described below from the first 
block. 
The first block will be described below. 
When the phase shift of the S.sub.A -phase signal from the S.sub.B -phase 
signal (see FIG. 2) is changed from the ideal state (.pi./2) to a state 
delayed by .pi./4 (+.pi./4), as shown in FIG. 10(b), an average value of 
the outputs from the exclusive OR gate 7 becomes larger than zero, as 
shown in FIG. 10(c). When the output from the gate 7 becomes positive in 
this manner, an average value of the outputs from the exclusive OR gate 10 
becomes positive because the rotational direction switching signal as one 
input of the exclusive OR gate 10 is set to be V/2. The signal from the 
exclusive OR gate 10 is input to the voltage-controlled oscillator (VCO) 
20 through the low-pass filter (LPF) 16. An output as an A-phase driving 
voltage from the oscillator 20 is applied to the A-phase driving 
electrodes A.sub.1 to A.sub.5 through the voltage-controlled amplifier 22. 
Therefore, the phase of the S.sub.A -phase signal is advanced, and the 
average value (see FIG. 10(c)) of the outputs from the exclusive OR gate 7 
is decreased toward zero. When this average value becomes zero (FIG. 
10(d)), one input to the exclusive OR gate 10 becomes a signal shown in 
FIG. 10(d). Therefore, the phase difference between the S.sub.A -phase 
signal (FIG. 10(a)) and the S.sub.B -phase signal (FIG. 10(c)) reaches 
.pi./2, thus restoring the ideal state. 
An operation will be described in which the rotational direction is 
reversed and the rotational direction switching signal represents -V/2. 
Assume that the phase of the S.sub.A -phase signal delayed from the S.sub.B 
-phase signal by .pi./4 and is thus changed from the ideal phase of 
-.pi./2 to -3.pi./4 (FIG. 10(f)). In this case, an average value of the 
outputs from the exclusive OR gate 7 becomes negative (FIG. 10(g)). When 
the output from the exclusive OR gate 7 is input to the exclusive OR gate 
10, an output from the exclusive OR gate 10 becomes the one shown in FIG. 
10(c) because one input signal of the exclusive OR gate 10 represents 
-V/2. Therefore, the phase of the S.sub.A -phase signal is advanced in the 
same manner as described above, and finally reaches -.pi./2. 
The operation of the second block will be described in detail below In this 
case, the frequencies of the output signals from the voltage-controlled 
oscillators 20 and 21 have been set to be appropriate values. 
The second block comprises two subblocks. The first subblock comprises the 
AC-DC converter 11, the amplifier 14, the low-pass filter 17, and the 
voltage-controlled amplifier 22 and controls the amplitude of the S.sub.A 
-phase signal. The second subblock comprises the AC-DC converter 12, the 
amplifier 15, the low-pass filter 18, and the voltage-controlled amplifier 
23 and controls the amplitude of the S.sub.B -phase signal. The subblock 
for the S.sub.A -phase signal is operated in the same manner as in the 
subblock for the S.sub.B -phase signal. Only the subblock for the S.sub.A 
-phase signal will be described, and a description of the other subblock 
will be omitted. Gains of the voltage-controlled amplifiers 22 and 23 for 
controlling the gain-controlled voltage of -V.sub.A to +V.sub.A (V.sub.A 
is a value representing the range of output voltages from the low-pass 
filters 17 and 18) fall within the range between 0 and Amax. 
The S.sub.A -phase signal is converted into a DC voltage corresponding to 
the amplitude of the S.sub.A -phase signal by the AC-DC converter 11. This 
DC voltage is input to the negative input terminal (-) of the amplifier 
14. A difference between the amplitude of the rotational speed control 
signal and the amplitude of the S.sub.A -phase signal is amplified by the 
amplifier 14. The amplified difference is smoothed and integrated by the 
low-pass filter 17. The integrated signal is input to the gain control 
input terminal of the voltage-controlled amplifier 22. Therefore, the gain 
of the voltage-controlled amplifier 22 is changed to control the amplitude 
of the A-phase driving voltage. If the amplitude of the S.sub.A -phase 
signal is smaller than the amplitude designated by the rotational speed 
control signal, an output voltage of the amplifier 14 is higher than zero, 
and an output voltage of the low-pass filter 17 is increased. The gain of 
the voltage-controlled amplifier 22 is increased, the amplitude of the 
A-phase driving voltage is increased, and the amplitude of the S.sub.A 
-phase signal can be controlled to the amplitude designated by the 
rotational speed control signal. 
The S.sub.B -phase signal can also be controlled to have the amplitude 
designated by the rotational speed control signal. Therefore, relation 
.vertline.V.sub.SA .vertline.=.vertline.V.sub.SB .vertline. can be 
established. 
An operation of the third block will be described under the condition that 
the first and second blocks are normally operated A subblock consisting of 
the comparator 3 and 5 and the exclusive OR gate 8 and a subblock 
consisting of comparators 4 and 6 and the exclusive OR gate 9 are the same 
circuit arrangement as that of the phase difference detector (3, 4, and 7) 
of the first block. A frequency of an output signal in response to an 
input voltage range of -V.sub.C to +V.sub.C (where V.sub.C is a value 
showing the range of the output voltage of the low-pass filter (LPF) 19) 
of the voltage-controlled oscillator 21 falls within the range of Fmin to 
Fmax and Fmin &lt;F.sub.1 &lt;F.sub.2 &lt;Fmax (where F.sub.1 and F.sub.2 are A- 
and B-phase resonance frequencies). The relationship between the phase 
difference and the frequency of the A-phase driving voltage and S.sub.A 
-phase signal is given in FIG. 5. FIG. 6 shows the relationship between 
the frequencies of the A- and B-phase driving voltages and the average 
value of the outputs from the analog adder 13 when the relationship 
between the frequency and the phase difference shown in FIG. 5 is 
established. If the frequency of the driving voltage is given as F.sub.2, 
inputs to the exclusive OR gate 8 are given, as shown in FIGS. 10(k) and 
10(m), respectively. Inputs to the exclusive OR gate 9 are given, as shown 
in FIGS. 10(h) and 10(i), respectively Outputs from the exclusive OR gates 
8 and 9 are shown in FIGS. 10(l) and 10(j), respectively. An average value 
of the outputs from the analog adder 13 is smaller than zero. A voltage 
smoothed and integrated through the low-pass filter 19 is decreased, and 
the frequency of the output voltage from the voltage-controlled oscillator 
21 is decreased. When the frequency of the voltage-controlled oscillator 
21 is decreased, the average value of the outputs from the exclusive OR 
gates 8 and 9 increase and the average value of the output from the analog 
adder 13 finally reaches zero. Therefore, the frequency of the driving 
voltage is controlled to be an intermediate frequency between F.sub.1 and 
F.sub.2. 
Referring to FIG. 7, the vibration detection electrodes S.sub.A and S.sub.B 
are arranged at two antinode positions of the A- and B-phase standing 
waves. In other words, the vibration detection electrode is arranged 
between the ground electrodes G.sub.1 and G.sub.2. However, if an 
vibration detection electrode is arranged to be centered on the 
corresponding antinode of the standing wave, any of the phase electrodes 
A.sub.1 and B.sub.1 to the phase electrodes A.sub.5 and B.sub.5 may be 
used in place of the vibration detection electrodes S.sub.A and S.sub.B. 
Alternatively, even if the vibration detection electrodes S.sub.A and 
S.sub.B are not accurately centered on the antinodes of the corresponding 
standing waves but substantially centered on the antinode positions, that 
is, if an output having the same phase as that of one standing wave can be 
obtained and is not adversely affected by the other standing wave, an 
electrode shape shown in FIG. 8, a split electrode arrangement for 
obtaining a composite output, as shown in FIG. 9 may be employed. 
According to the present invention, the vibration detection 
electro-mechanical energy convertion element areas are located to be 
centered on substantially the antinodes of the standing waves of the 
electrode groups. Therefore, the standing waves can be detected in a real 
time manner. In addition, the conversion elements can be driven at desired 
phases and an amplitude. Therefore, the amplitudes of the standing waves 
of both the groups can be made equal to each other, and the time-phase of 
each standing wave can be set to be .pi./2, thereby generating an ideal 
travelling wave and improving driving efficiency and rotational stability 
of the motor.