Diversity for direct-sequence spread spectrum systems

An invention for providing diversity for direct sequence spread spectrum wireless communication systems is presented. The invention provides a transmitting technique for communicating a first signal comprising one or more segments to a receiver with use of a plurality of M antennas. The first signal includes one or more signal segments. The technique comprises forming M copies of the first signal; for a segment of a signal copy, weighting each of two or more sub-segments of the segment with a distinct signal, wherein a sequence of the distinct weighting signals for the segment is distinct from sequences of signals weighting the same segment of one or more other signal copies; and for each of M weighted signal copies, transmitting a signal to the receiver using a distinct antenna, the transmitted signal based on the weighted signal copy. Illustratively, the first signal is a spread spectrum signal. Moreover, the process of weighting comprises applying a phase shift to a sub-segment. The invention further provides a receiving technique which comprises, for a copy of a received signal, despreading a segment of the received signal, demodulating a plurality of sub-segments of the despread signal segment, wherein each sub-segment is demodulated with use of one or more estimated communication channel characteristics corresponding to the sub-segment, and forming a summation signal reflecting a summation of a plurality of demodulated sub-segments. The receiving technique further comprises forming a signal reflecting a signal segment value, the formed signal based on one or more summation signals. The step of demodulating may precede the step of despreading in some embodiments.

FIELD OF THE INVENTION 
The present invention relates generally to Direct Sequence Spread Spectrum 
wireless communication systems, such as Direct-Sequence Code Division 
Multiple Access systems. 
BACKGROUND OF THE INVENTION 
In cellular radio systems, each cell is a local geographic region 
containing a base station and a plurality of mobile users. Each mobile 
user communicates directly with a base station only; there is no direct 
mobile-to-mobile communication. The base station performs, among other 
things, a relay function allowing a mobile user to communicate with a user 
in another location. So, for example, the base station provides coupling 
of a mobile user's transmission to another mobile user in the same cell, 
to another base station for coupling to a mobile user in another cell, or 
to an ordinary public switched telephone network. In this way, a mobile 
user can send and receive information to and from any other addressable 
user. 
Direct Sequence Spread Spectrum (DSSS) systems, such as Direct Sequence 
Code Division Multiple Access (DS-CDMA) systems, are attracting widespread 
attention in the personal communication fields, such as, for example, 
digital cellular radio. In a DS-CDMA communication system, both the time 
and frequency domains may be shared by all users simultaneously (this 
simultaneous sharing of time and frequency domains is to be distinguished 
from time-division and frequency-division multiple access systems, TDMA 
and FDMA, where multiple user communication is facilitated with use of 
unique time slots or frequency bands, respectively, for each user). As 
such, a base station may simultaneously transmit distinct information 
signals to separate users using a single band of frequencies. Individual 
information signals simultaneously transmitted may be isolated by each 
receiving user because of the base station's utilization of unique 
signature sequences in the transmission of the information signals. Prior 
to transmission, the base station multiplies each information signal by a 
signature sequence signal assigned to the user intended to receive the 
signal. To recover a transmitted signal from among those signals 
transmitted simultaneously in a frequency band, a receiving mobile user 
multiplies a received signal (containing all transmitted signals) by its 
own unique signature sequence signal and integrates the result. By so 
doing, the user identifies that signal intended for it, as distinct from 
other signals intended for other users. 
In wireless communication systems (such as DS-CDMA systems), an information 
signal is communicated from a transmitter to a receiver via a channel 
comprising several independent paths. These paths are referred to as 
multipaths. Each multipath represents a distinct route an information 
signal may take in traveling between transmitter and receiver. An 
information signal communicated via such routs or multipaths appears at a 
receiver as a plurality of multipath signals, one signal for each 
multipath. 
The amplitudes and phases of signals received from a transmitter through 
different multipaths of a communication channel are generally independent 
of each other. Because of complex addition of multipath signals, the 
strength of received signals may vary between very small and moderately 
large values. The phenomenon of received signal strength variation due to 
complex addition of multipath signals is known as fading. In a fading 
environment, points of very low signal strength, or deep fades, are 
separated by approximately one-half wavelength from each other. 
Multipaths encountered in wireless communication systems can be described 
by certain characteristics, such as amplitude attenuation and phase 
shifting. For example, the multipaths of a DS-CDMA channel may provide 
different amplitude attenuations and phase shifts to an information signal 
communicated from a transmitter to a receiver. These different amplitude 
and phase characteristics may vary due to, e.g., relative movement between 
transmitter and receiver, or changes in local geography of the transmitter 
or receiver due to movement. Because of the variation of multipath 
characteristics, a receiver can experience a signal which fades with time. 
This fading is a manifestation of the complex addition of multipath 
signals having time varying amplitudes and phases. 
If the characteristics of a DS-CDMA multipath vary slowly, a receiver 
experiencing a deep fade may observe a weak signal for a long period of 
time. Long fades are not uncommon in, e.g., indoor radio systems, where 
relative movement between receivers and transmitters is slow or 
nonexistent (often, one of these two is an immobile base station; the 
other is a mobile device carried by a person). Since the duration of a 
deep fade may be large in comparison to the duration of information 
symbols being communicated, long bursts of symbol errors may occur (due to 
the weakness of received signal strength for an extended period of time). 
To avoid or mitigate the detrimental effects of fading, a technique 
providing diversity may be employed. Diversity refers generally to the 
ability of a communication system to receive information via several 
independently fading channels. As a general matter, diversity techniques 
enhance a system receiver's ability to combine or select (or both) signals 
arriving from these independently fading channels, thus enabling (or 
facilitating) the extraction of communicated information. 
SUMMARY OF THE INVENTION 
The present invention provides a technique for mitigating the detrimental 
effects of fading in DSSS systems. An illustrative transmitter embodiment 
of the invention provides diversity by introducing a sequence of distinct 
weights to segments of a signal to be transmitted. Specifically, given a 
signal to be transmitted which comprises signal segments reflecting binary 
digits, the illustrative embodiment forms M copies of the signal, where M 
is the number of antennas used in transmitting the signal. For each 
segment of each signal copy, the embodiment of the invention applies a 
distinct phase shift to each of M sub-segments of the segment. As a 
result, M phase-shifted signal copies are produced, one copy for each 
antenna. The sequence of distinct phase shifts applied to a given segment 
copy is itself distinct from the sequences of phase shifts applied to any 
other copy of the given segment. Each of the M phase-shifted signal copies 
forms the basis of a signal transmitted to a receiver with use of a 
distinct antenna. 
An illustrative receiver embodiment comprises a plurality of receiver 
branches, each branch corresponding to a multipath of the communication 
channel through which transmitted signals have been sent. Each receiver 
branch performs despreading and demodulation processes. The despreading 
process comprises forming a product of a segment of the received signal 
and a signature sequence signal. Values of the despread received signal 
corresponding to a sub-segment are summed. The resulting sum is provided 
to a demodulation process which operates to remove the effects of the 
multipath on received signal amplitude and phase. Because of the distinct 
phase shifts applied by the transmitter to sub-segments of each signal 
segment reflecting a binary digit, the demodulation process operates on a 
sub-segment by sub-segment basis. The demodulated sub-segment values for 
each segment are summed. Summed sub-segment values from each receiver 
branch form the basis of a determination of the value of the binary digit 
corresponding the segment in question. 
Illustrative embodiments of the invention provide diversity of ML'th order, 
where M is the number of antennas employed by the transmitter and L' is 
the number of receiver branches corresponding to L' multipaths. While an 
illustrative receiver embodiment of the invention may incorporate a 
multi-branch RAKE receiver (as explained below), it will be apparent to 
those of ordinary skill in the art that a receiver with but one branch 
(i.e., L'=1) may also be used. Though illustrative embodiments of the 
present invention concern DS-CDMA systems, the present invention is 
applicable to indoor and outdoor DSSS systems generally, such as 
DS-Carrier Sense Multiple Access systems, etc. Therefore, the invention 
has applicability to cellular telephony, wireless PBXs, wireless LANs, 
etc., and may be used in combination with other DSSS systems to enhance 
diversity.

DETAILED DESCRIPTION 
A. Introduction 
The illustrative embodiment of the present invention concerns a wireless 
DS-CDMA communication system such as, e.g., an indoor radio communication 
system, a wireless local area network, a cellular telephone system, or 
personal communications system. In such systems, a base station commonly 
uses a plurality of antennas (e.g., two) for receiving signals transmitted 
by one or more mobile units. This plurality of antennas provides the base 
station with a form of diversity known as space diversity. In accordance 
with the present invention, a plurality of antennas at the base station 
should be used for the transmission of signals to mobile units. 
Advantageously, the same plurality of antennas used for base station 
reception may be used for transmission to the mobile units. These mobile 
units need employ but one antenna. 
1. DS-CDMA Signals 
FIG. 1 presents a basic set of signals illustrative of DS-CDMA 
transmission. Signal a(n) of FIG. 1(b) is a signature sequence signal 
associated with a particular receiver, as discussed above. Signal a(n) 
comprises a series of rectangular pulses (or chips) of duration T.sub.c 
and of magnitude .+-.1. Discrete time variable n indexes T.sub.c intervals 
(i.e., n is a sampling time at the chip rate). 
Signal b(n) of FIG. 1(a) is a signal (e.g., an information signal) to be 
communicated to a receiver. Each bit of signal b(n) is of a duration T and 
is indexed by i. As shown in FIG. 1(b), there are N chip intervals of 
duration T.sub.c in interval T (i.e., N=T/T.sub.c). 
The product of these two signals, a(n)b(n), is a spread spectrum signal 
presented in FIG. 1(c). As shown in FIG. 1(c), the first N chips of the 
spread spectrum signal are the same as the first N chips of signal a(n). 
This is because signal b(n)=1, 0.ltoreq.n.ltoreq.N-1. Moreover, the second 
N chips of the spread spectrum signal have polarity opposite to that of 
the second N chips of signal a(n), since signal b(n)=-1, 
N.ltoreq.n.ltoreq.2N-1. Thus, signal b(n) modulates signal a(n) in the 
classic sense. 
2. Fading in DS-CDMA Systems 
FIG. 2 presents an indoor radio system comprising a base station 1 having 
two antennas, T.sub.1 and T.sub.2, for transmitting a signal through, for 
example, a Rayleigh fading channel to a mobile receiver (a Rayleigh fading 
channel is a channel without a line-of-sight path between transmitter and 
receiver). Each of the antennas T.sub.1 and T.sub.2 transmits a spread 
spectrum signal, u(n), which reflects a scaled product of signals a(n) and 
b(n) shown in FIG. 1(c). Each copy of signal u(n) experiences an 
independent change in amplitude and phase due to the multipath in which it 
travels. This change in amplitude and phase to the transmitted signal is 
expressed as a complex fading coefficient, .beta..sub.l (n), where l, 
1.ltoreq.l.ltoreq.L identifies the multipath (in FIG. 2, L=2). 
The signal received by receiver R.sub.1, s(n), reflects a summation of the 
transmitted signals 
##EQU1## 
where A is a transmitter gain factor, .tau..sub.l is a transmission delay 
associated with a particular multipath, and v(n) is Gaussian noise added 
by the channel. Signal s(n) therefore comprises a summation of received 
signal phasors S.sub.l, where S.sub.l =A.beta..sub.l 
a(n-.tau..sub.l)b(n-.tau..sub.l). 
In the example of FIG. 2, signals S.sub.1 and S.sub.2, are received at 
specific points in space where a deep fade occurs. The deep fade is due to 
a destructive interference of S.sub.1 and S.sub.2. Signals S.sub.1 and 
S.sub.2 are independently and identically distributed with, e.g., Rayleigh 
amplitude and uniform phase. The complex fading characteristics of the 
channel through which phasors S.sub.1 and S.sub.2 are communicated 
(.beta..sub.1 (n) and .beta..sub.2 (n)) change slowly, so that the deep 
fade experienced by receiver R.sub.1 of FIG. 2 is essentially static. 
The deep fade shown location (b) of FIG. 2 occurs because of the weakness 
of received signal energy from each individual antenna T.sub.1 and 
T.sub.2. Thus, despite the fact that received signal phasors are not 
destructively aligned, receiver R.sub.2 experiences a fade. 
3. Path Diversity in Conventional DS-CDMA Systems 
Among the techniques used to mitigate the effects of fading in DS-CDMA 
communication systems is the path diversity technique. Path diversity in 
DS-CDMA systems entails estimation of the delay introduced by each of one 
or more multipaths (in comparison with some reference, such as 
line-of-sight delay), and using this delay in a receiver structure to 
separate (or resolve) the received multipath signals. Once separated, 
conventional techniques may be used to select the best multipath signal 
(or to combine multipath signals) so as to extract the communicated 
information. 
A receiver structure often employed to provide path diversity is the 
so-called RAKE receiver, well known in the art. See. e.g., R. Price and P. 
E. Green, Jr., A Communication Technique for Multipath Channels, 46 Proc. 
Inst. Rad. Eng. 555-70 (March 1958). 
While the path diversity afforded by conventional RAKE receivers is 
beneficial in many instances, it may not provide a significant diversity 
benefit in certain circumstances, such as some indoor radio environments. 
This is because the range of multipath delay values in these environments 
is small (on the order of 200 to 300 nanoseconds) compared with the 
duration of a DS-CDMA chip interval (which may be, for example, 1 .mu.s). 
Because of this, knowledge of delay values is insufficient to allow 
resolution of multipath signals. Thus path diversity is not generally 
available in such conventional DS-CDMA systems. 
4. Introduction to the Illustrative Embodiments 
The illustrative embodiments of the present invention provide diversity for 
a DS-CDMA systems, even in indoor radio environments. 
The illustrative transmitter embodiment of the present invention introduces 
phase shifts .theta..sub.1 (n) and .theta..sub.2 (n) to signals u.sub.1 
(n) and u.sub.2 (n) transmitted from antennas T.sub.1 and T.sub.2, 
respectively. These phase shifts are introduced for a portion of the 
interval T corresponding to each information signal bit. These phase 
shifts have the effect of repositioning signal phasors S.sub.1 and S.sub.2 
with respect to each other. Should the signal phasors be disposed such 
that they add destructively, the phase shifts work to alter signal phasor 
angle so that the signal phasors add constructively. This constructive 
addition mitigates the effects of fading. 
It will be understood by those of ordinary skill in the art that the 
relative angular position of signals S.sub.1 and S.sub.2 in FIG. 2 is 
merely illustrative of the possible relative angular positions such 
signals may take. However, signals S.sub.1 and S.sub.2, being out of phase 
by nearly .pi. radians, represent a near worst case scenario. Since the 
operation of the embodiment of the present invention works to mitigate 
worst case scenarios, less severe cases are naturally accounted for by the 
embodiment. 
The operation of the illustrative transmitter embodiment may be further 
understood with reference to FIGS. 3(a) and (b). FIG. 3(a) presents 
signals S.sub.1 and S.sub.2 as they appear in FIG. 2. As a consequence of 
the static angular orientation of these signals, the resultant sum of 
these phasors, s(n), has a magnitude, .vertline.s(n).vertline.=G.sub.1, 
which is small compared with the magnitude of the individual signals. 
Magnitude G.sub.1 is indicative of a deep fade. Assuming no changes in 
these signals, a given information bit, such as bit b(n), 
0.ltoreq.n.ltoreq.N-1 (and likely many more), would not likely be 
received. 
In FIG. 3(b), a phase shift of .pi. radians has been applied by the 
transmitter 1 to signal u.sub.2 during the first half of the bit interval 
(i.e., 0.ltoreq.n.ltoreq.N/2-1) in accordance with the invention. This 
phase shift has the effect of changing the relative angular disposition of 
S.sub.1 and S.sub.2 such that the destructive interference experienced by 
receiver R.sub.1 becomes constructive. As shown in the Figure, the 
magnitude of the sum of the phasors, .vertline.s(n).vertline. is G.sub.2 
for the first half of the interval and G.sub.1 for the second half of the 
interval. The large magnitude G.sub.2 for a portion (or time segment) of 
the bit interval enables the bit to be received by receiver R.sub.1. 
The illustrative transmitter embodiment may be extended to deal with the 
deep fades shown at location (b) of FIG. 2. All that is required is the 
use of additional transmitting antennas to help contribute to received 
signal strength. A discussion of the embodiment below is generic to the 
number of transmitting antennas, M. 
5. Embodiment Hardware 
For clarity of explanation, the illustrative embodiment of the present 
invention is presented as comprising individual functional blocks 
(including functional blocks labeled as "processors"). The functions these 
blocks represent may be provided through the use of either shared or 
dedicated hardware, including, but not limited to, hardware capable of 
executing software. For example, the functions of processors presented in 
FIG. 5 may be provided by a single shared processor. (Use of the term 
"processor" should not be construed to refer exclusively to hardware 
capable of executing software.) 
Illustrative embodiments may comprise digital signal processor (DSP) 
hardware, such as the AT&T DSP16 or DSP32C, read-only memory (ROM) for 
storing software performing the operations discussed below, and random 
access memory (RAM) for storing DSP results. Very large scale integration 
(VLSI) hardware embodiments, as well as custom VLSI circuitry in 
combination with a general purpose DSP circuit, may also be provided. 
B. An Illustrative Transmitter Embodiment 
FIG. 4 presents an illustrative transmitter embodiment in accordance with 
the present invention. The transmitter receives a signal, b(n), for 
transmission to a receiver. Signal b(n) is "spread" in the conventional 
sense of DS-CDMA systems by multiplying the signal by a signature 
sequence, a(n), provided by signal generator 12. This multiplication is 
performed by multiplier circuit 10. The result of this multiplication is a 
spread spectrum signal reflecting the product a(n)b(n). This spread 
spectrum signal is provided in parallel to a plurality of M transmitter 
circuit antenna branches. Each such antenna branch comprises a multiplier 
circuit 15, a signal generator 17, conventional transmission circuitry 20, 
and an antenna 25. 
The multiplier circuit 15 of each antenna branch applies to the spread 
spectrum signal (or weighs the spread spectrum signal by) a distinct 
time-varying signal p.sub.m (n) of the form 
EQU p.sub.m (n)=A.sub.m (n)e.sup.j.theta..sbsp.m.sup.(n), (2) 
where m indexes the antenna branch, A.sub.m (n) is signal amplitude, and 
.theta..sub.m (n) is signal phase. Signal p.sub.m (n) is generated by 
signal generator 17. Amplitude A.sub.m (n) of signal p.sub.m (n) takes the 
form 
##EQU2## 
Phase .theta..sub.m (n) of signal p.sub.m (n) takes the form 
##EQU3## 
where m indexes the antenna branch; and m'=1, 2, . . . , M indexes equal 
temporal portions (or sub-segments) of a segment of the spread spectrum 
signal. Each such segment is an interval of length T and is associated 
with a bit of b(n). The equal sub-segments are given by 
##EQU4## 
where i indexes the bits represented by signal b(n). The illustrative 
embodiment therefore applies a distinct phase shift, .theta..sub.m (n) and 
a common gain A.sub.m (n) to each sub-segment of the spread spectrum 
signal associated with a bit of b(n). If N is not an integer multiple of 
M, the length of the sub-segments should be made as equal as possible. 
The application of phase shift .theta..sub.m (n) by the operation of 
generator 17 and multiplier circuit 15 is illustrated with reference to 
FIG. 3(b). As discussed above, when M=2 a phase shift of .pi. radians is 
applied to one of two transmitted phasors during the first half (m'=1) of 
a bit interval. Given two transmitting antennas (i.e., M=2), the phase 
shift of .pi. radians applied to the spread spectrum signal in the second 
antenna branch is provided by generator 17 in accordance with expression 
(4). So, for example, the phase of p.sub.m (n), .theta..sub.m (n), is 
equal to .theta. when M=2, indicating the two antenna branches; m=2, 
indicating the second of the two branches; and m'=1, indicating the first 
of M=2 equal sub-segments. 
Generator 17 applies phase shift .theta..sub.m (n) for sub-segments defined 
in terms of n by expression (5). So, for example, assuming i=0 (i.e., 
assuming the first bit of b(n)), and substituting M=2, m'=1, and m=2, 
expression (5) simplifies to 
##EQU5## 
the first half (or sub-segment) of the interval corresponding to the first 
bit of b(n). Thus, generator 17 provides p.sub.m (n) with phase shift 
.theta..sub.m (n)=.pi. for the sub-segment defined by 
##EQU6## 
Generator 17 operates in accordance with expressions (4) and (5) to apply a 
phase shift of zero to the spread spectrum signal in the second antenna 
branch during the second half (m'=2) of the interval corresponding to the 
first bit of b(n). This zero phase shift is shown in FIG. 3(b) by the 
phasor S.sub.2 in its original position (shown in FIG. 3(a)). Moreover, 
generator 17 applies a phase shift of zero to the spread spectrum signal 
in the first antenna branch during both the first and second halfs (i.e., 
both sub-segments) of the interval corresponding to the first bit of b(n). 
Again, this is done in accordance with expressions (4) and (5). This zero 
phase shift is shown in FIG. 3(b) by phasor S.sub.1 remaining in its 
original position (shown in FIG. 3(a)) for both halves of the bit 
interval. 
The distinct weighting signals, p.sub.m (n), applied to each sub-segment of 
a bit interval (or segment) constitute a sequence of weighting signals. 
The sequence of weighting signals applied by one antenna branch of the 
embodiment for a given bit interval is distinct from the sequence of 
weighting signals applied in any other branch of the embodiment during the 
same bit interval. So, for example, the sequence of phase shifts applied 
by the first antenna branch of FIG. 4 for the segments of the bit interval 
discussed above is (0 rad., 0 rad.). This sequence is distinct from the 
sequence (.pi. rad., 0 rad.) applied by the second antenna branch for the 
sub-segments of the same bit interval, since the first phase shift of each 
sequence is not the same. 
The product of spread spectrum signal a(n)b(n) and signal A.sub.m 
(n)e.sup.j.theta..sbsp.m.sup.(n) produced by multiplier circuit 15 of each 
antenna branch 1.ltoreq.m.ltoreq.M is a signal u.sub.m (n). Each signal 
u.sub.m (n) is provided to conventional transmission circuitry 20. 
Circuitry 20 provides, inter alia, pulse- shaping, RF-modulation, and 
power amplification in preparation for signal transmission via antenna 25. 
As a result of the operation of the illustrative transmitter embodiment, 
each of M antennas 25 transmits a signal to a receiver. Each such signal 
is based on a distinctly phase shifted version of a spread spectrum 
signal. 
It will be understood by those of ordinary skill in the art that a 
transmitter embodiment in accordance with the invention may be realized 
with any number of antenna branches. Expressions (2)-(5) above are 
presented generally to allow for such extended realizations. Furthermore, 
it will be understood that the sequence of operations which constitute 
despreading, as well as the sequence of despreading and demodulation 
operations, is illustrative. Other sequences of such operations may be 
realized in accordance with the present invention. 
C. An Illustrative Receiver Embodiment 
FIG. 5 presents an illustrative DS-CDMA RAKE receiver embodiment of the 
invention. The embodiment comprises an antenna 50; conventional receiver 
circuitry 55; L' RAKE receiver branches, where L' is less than or equal to 
the number of multipaths, L; summing circuit 80; and decision processor 
85. The RAKE receiver branches are indexed by l, such that 
1.ltoreq.l.ltoreq.L'.ltoreq.L. As is conventional for RAKE-type receivers, 
each receiver branch is "tuned" to receive signals from a particular 
multipath of a communication channel. 
The illustrative receiver embodiment of FIG. 5 may be used to receive 
signals transmitted by the illustrative transmitter embodiment of FIG. 4. 
Assuming M=2 transmit antennas and L'=2 RAKE receiver branches, use of the 
illustrative transmitter and receiver in combination provides ML'th (or in 
this case fourth) order diversity. 
Each RAKE receiver branch comprises a DS-CDMA despreader 60, a demodulator 
70, and a summation memory 75. Receiver branch tuning is accomplished 
conventionally, by estimation of multipath transmission delay .tau..sub.l 
(for use by despreader 60) and the complex conjugate of the multipath 
complex fading coefficient, .beta..sub.l *(n) (for use by demodulator 70). 
Each despreader 60 comprises multiplier circuit 62, signal generator 63, 
and summation processor 64. Demodulator 70 comprises demodulation 
processor 72. 
Antenna 50 receives transmitted multipath signals from a transmitter 
embodiment of the invention. The received signals, r(t), are processed by 
conventional receiver circuitry 55 (comprising, e.g., low noise 
amplifiers, RF/IF band-pass filters, and a match filter) to produce signal 
s(n) as discussed above with reference to expression (1). Signal s(n) is 
provided to each of the L' receiver branches. 
Multiplier circuit 62 receives signal s(n) from circuitry 55 and a delayed 
version of the signature sequence from signal generator 63. The signal 
generators 63 of the embodiment are identical but for the delay they apply 
to the signature sequence. Each delay, .tau..sub.l, is an estimate of the 
transmission delay associated with the lth multipath. This delay is 
determined by generator 63 in the conventional fashion for DS-CDMA 
systems. See, e.g., Pickholtz et al., Theory of Spread Spectrum 
Communications--A Tutorial, Vol. COM-30, No. 5, IEEE Transactions on Comm. 
855,870-75 (May 1982). 
The output of multiplier 62 is provided to summation processor 64. For each 
bit of signal b(n) to be received, processor 64 forms M summations of the 
signal s(n)a(n-.tau..sub.l) provided by multiplier 62. Each summation is 
of the form 
##EQU7## 
where i refers to the ith bit of b(n), m' indexes equal length 
sub-segments of the ith bit interval, and .tau..sub.l is the 
conventionally determined multipath transmission delay. For each bit of 
b(n), processor 64 provides a despread signal segment which comprises M 
output signals, y.sub.lm'.sup.i, 1.ltoreq.m'.ltoreq.M. 
So, for example, if M=2, processor 64 will form two summations, each of 
which formed over one of the two (i.e., M) equal length sub-segments of 
the ith bit interval indexed by m'. These summations have a form given by 
expression (6): 
##EQU8## 
Therefore, summation processor 64 treats the sub-segments of the ith bit 
interval separately, since such sub-segments are subject to distinct phase 
shifts applied by the transmitter. 
The M output signals provided by summation processor 64, y.sub.lm'.sup.i, 
for the ith bit and the lth multipath, are provided as input to 
demodulation processor 72. Demodulation processor 72 multiplies each 
signal, y.sub.lm'.sup.i, by an estimate of the conjugate of the complex 
fading coefficient for the lth multipath. In conventional RAKE receivers, 
the estimate of the conjugate of the complex fading coefficient for the 
lth multipath is determined on an incremental bit by bit basis. That is, 
the estimate of the conjugate of the fading coefficient for the ith bit is 
dependent on an estimate of the conjugate of the coefficient for the i-1th 
bit. However, because of the application of different phase shifts in 
different segments of the ith bit interval (by the transmitter), this 
incremental determination of the conjugate of the complex fading 
coefficient must be modified. This modification may be understood with 
reference to FIG. 6. 
As shown in FIG. 6, for the case where M=2, each bit, e.g., i=1, has 
associated with it two complex fading coefficients, .beta..sup.m'* (iN), 
N.ltoreq.m'.ltoreq.M(=2). The second of these two coefficients, 
.beta..sup.2* (N), is not dependent on the coefficient which immediately 
precedes it, .beta..sup.1* (N), but rather on the second of the two fading 
coefficients associated with the preceding bit, .beta..sup.2* (0). This is 
because both coefficients, .beta..sup.2* (N) and .beta..sup.2* (0), 
correspond to a bit interval sub-segment specified by m'=2. Therefore, 
such coefficients reflect the same phase shift applied by the transmitter. 
The dependence of coefficients is indicated in the Figure by an arrow 
connecting a later coefficient with an earlier coefficient. As may be seen 
from the Figure, a coefficient associated with a given sub-segment m' of a 
given bit is dependent on the coefficient of the same sub-segment of the 
preceding bit. Therefore, processor 72 may be realized with M conventional 
coefficient estimation phase-locked loops, each such loop concerned with 
the same sub-segment m' in successive bit intervals. See, e.g., Gitlin, et 
al., Data Communications Principles, 403-32 (1992). It should be 
understood that processor 72 of the illustrative receiver need estimate 
coefficient phase only. This is because the illustrative transmitter 
embodiment uses only a phase shift to differentiate the signals 
transmitted by the different antennas. 
Referring again to FIG. 5, the output of processor 72 of the l multipath 
receiver branch for the ith bit comprises over time M signals of the form 
EQU Z.sub.lm'.sup.i =.beta..sub.l.sup.m'* y.sub.lm'.sup.i, (9) 
where the M signals are indexed by m'. These M signals are stored by 
summation memory 75 and added together as received. Memory 75 forms a sum 
as follows: 
##EQU9## 
Signals z.sub.l.sup.i from the memory 75 of each receiver branch are 
summed by summing circuit 80. The result is a signal z.sup.i which 
reflects each received bit i. Signal z.sup.i is provided to a conventional 
decision processor 85, which assigns a binary value for each bit, b.sup.i, 
based on z.sup.i. Processor 85 illustratively provides a threshold 
detection such that b.sup.i =1 when z.sup.i .gtoreq.0, and b.sup.i =0 when 
z.sup.i &lt;0. Binary signal b.sup.i is thus the received bit stream. 
The embodiments of the transmitter and receiver presented above concern 
binary phase shift keying (BPSK) modulation formats. However, other 
modulation formats such as binary differential phase shift keying (DPSK) 
may be used. The transmitter embodiment presented above may be augmented 
to provide DPSK modulation by use of the conventional differential encoder 
100 presented in FIG. 7. For DPSK modulation of a binary signal d(n), 
signal d(n) is presented to the conventional mod-summing circuit 110 of 
differential encoder 100. Modulo-2-summing circuit 110 also receives input 
from delay 120. The output of mod-2-summing circuit 110 is provided to the 
transmitter embodiment as signal b(n). Signal b(n) is also fed back to the 
mod-summing circuit via delay 120. 
The illustrative receiver can be modified to receive DPSK modulated signals 
from the transmitter by replacing demodulation processors 72 discussed 
above with the demodulation processors 73 presented in FIG. 8. Each 
processor 73 is shown comprising a loop comprising delay 130, conjugate 
processor 135 and summing circuit 140. 
The segments of signals discussed above in the context of the illustrative 
embodiments of the present invention concern individual binary digits (or 
bits) of a digital signal. It will be understood by those skilled in the 
art that these signal segments may reflect values of other types of 
signals in other embodiments of the present invention. For example, in 
such embodiments these segments may reflect complex-valued signals, analog 
signals, discrete-valued signals, etc.