Device for production of a tachometry signal of infinite resolution and without ripple from an inductive position sensor

A device for the production of a tachometry signal of infinite resolution and without ripple from an inductive position sensor. The sensor is fed by a sinusoidal reference signal and supplies two phase signals. A resistance-capacitance circuit tuned to the frequency of the reference signal is fed by the two phase signals of the sensor and supplies a sinusoidal signal whose phase angle is proportional to the rotation of the sensor. This signal is then digitized and processed by digital phase comparators and low-pass filters before differentiation. This includes the use of particular digital phase comparators, each associated with a particular return to zero detector, to produce by a storage flip-flop and an electronic switch, the outgoing tachometry signal with the aid of a single differentiator with a device for elimination of discontinuities.

BACKGROUND OF THE INVENTION 
1. Field of the Invention 
The invention relates generally to high-precision servomechanisms, and more 
particularly to selfsynchronizing motors in servomechanisms. 
2. Discussion of Background 
These servomechanisms often use a selfsynchronizing motor without brushes 
which requires the use of an inductive sensor for the determination of the 
position of the rotor. Moreover, the use of the servomechanism requires a 
tachometer to determine at any moment the exact rotation speed of the 
rotor. Therefore, it is tempting to try to avoid the duplication of 
equipment and to determine the speed directly from the resolver. Generally 
the problem therefore occurs of generating a tachometry signal from an 
inductive sensor such as a resolver (biphase), a synchro (triphase) or an 
Inductosyn (linear displacement). 
To solve this problem a device is already known, described in particular in 
European application No. 127 890. This prior device uses an RC circuit 
driven by the two signals of phases of the inductive sensor, itself fed by 
a sinusoidal reference signal. This RC circuit generates a sinusoidal 
analog signal whose phase angle is rigorously equal to the angular shift 
of the rotor of the sensor in relation to the stator. This analog signal 
then is digitized by a trigger circuit or the like to supply a square-wave 
input signal. Further, the oscillator which supplies said reference signal 
is made also to supply a second reference signal out of phase by .pi./2, 
these two reference signals also being digitized. 
Moreover, this device uses two digital phase comparators consisting 
essentially of two exclusive OR gates respectively comparing said input 
signal and one of the two reference signals to generate two square-wave 
width modulation signals with alternately increasing and decreasing 
widths. These signals are separately filtered and shunted to generate a 
trapezoidal signal as well as this same signal out of phase by .pi./2, 
these two latter signals themselves being separately processed by 
inverters to obtain finally four trapezoidal signals out of phase by 
.pi./2 whose amplitudes are proportional to the speed of rotation. Finally 
the device comprises a complex logic which, starting from the two filtered 
signals, samples from each of the four preceding trapezoidal signals the 
two central quarters of each half period. 
However, if a correct tachometry signal is desired, it is necessary to pair 
the components of the two ways. In adjustment to zero of the two shunting 
devices and also an adjustment to zero of the two inverters is also 
necessary, as otherwise there would be obtained, for example, for a 
constant rotation speed of nonconstant tachometry, a signal over a 
rotation. 
SUMMARY OF THE INVENTION 
Accordingly, one object of this invention is to achieve a novel device that 
also generates a tachometry signal from the two phase signals of an 
inductive sensor. 
Another object of this invention is to provide a novel device which is 
simpler and especially does not require any adjustment or component 
pairing. 
This invention is obtained like the previous device in a standard way using 
an RC circuit and a trigger circuit for producing the digitized main input 
signal, but it is distinguished from it by the fact that: 
each of the two digital phase comparators generating a square-wave width 
modulation signal consists of a circuit of simple flip-flops to establish 
the corresponding width modulation signal for each transition from a 
determined direction of an input signal of the comparator, and to 
interrupt it for each transition from a determined direction of the other 
input signal of the comparator, the two input signals of the comparator 
considered being the reference signal and the main signal for one of the 
comparators, and for the other comparator these same signals including a 
transit through an inverter to replace it by its complement; 
with each digital phase comparator is associated a return to zero detector 
consisting of another flip-flop circuit made to establish a corresponding 
return to zero signal when there are achieved conditions of passage of the 
square-wave width modulation signal from one state to another and this 
square-wave width modulation signal is already in the other state; 
the two return to zero signals thus generated drive a storage flip-flop 
which produces a selection signal; 
a switch is controlled by said selection signal to sample, in each of the 
square-wave width modulation signals after passage in a low-pass filter, 
the second half of each cycle alternately of one or the other signal to 
obtain a single sawtooth signal; and 
a single differentiator generates the tachometry signal from this single 
sawtooth signal, with devices for elimination of discontinuities. 
A more complete appreciation of the invention and many of the attendant 
advantages thereof will be readily obtained as the same becomes better 
understood by reference to the following detailed description when 
considered in connection with the accompanying drawings, wherein: 
FIG. 1 is a block diagram of the device; 
FIG. 2 is a detailed block diagram of one of the digital phase comparators 
and return to zero detectors of FIG. 1; 
FIG. 3 is a diagram of the differentiator with its initiation and storage 
devices of FIG. 1; 
FIGS. 4 and 5 are timing diagrams illustrating the functioning of the 
device of FIG. 2, respectively for a positive speed and a negative speed 
of the rotor of the inductive sensor; and 
FIG. 6 is a timing diagram of the functioning of the whole device.

DESCRIPTION OF THE PREFERRED EMBODIMENTS 
Various other objects, features and attendant advantages of the present 
invention will be more fully appreciated as the same becomes better 
understood from the following detailed description when considered in 
connection with the accompanying drawings in which like reference 
characters designate like or corresponding parts throughout the several 
views and wherein the inductive position sensor, or synchroresolver, not 
shown, is fed by a sinusoidal oscillator, not shown, of perfectly stable 
clock rate or pulse frequency w. This sinusoidal signal is also fed to 
input Po of the device, while the two phase signals of the two-phase 
sensor are fed to inputs P1 and P2 of this same device. In a standard way 
an RC circuit is mounted between P1 and P2 and supplies a sinusoidal 
signal of clock rate or pulse frequency w and phase angle .theta. in 
relation to signal Po. This sinusoidal signal is digitized in a trigger 
circuit T1 and supplies a main signal S in the form of relatively regular 
square waves, because .theta. varies only very slowly in relation to pulse 
frequency w. 
In the same way the initial sinusoidal signal Po is digitized by a second 
trigger circuit T2 which gives a reference signal Ref, also in regular 
square waves having a similar width. Finally, FIG. 1 shows the presence of 
an inverter I which transforms signal S into its complement S 
According to one of the essential features of the invention, the signals 
Ref and S, as well as Ref and S0 are compared separately in two digital 
phase comparators CPN, each associated with a return to zero detector DRZ. 
The details of these units are shown in FIG. 2. Each consists essentially 
of four type D flip-flops, referenced B1 to B4, which are coupled as 
indicated in the diagram. 
Flip-flop B1 is set by the positive transitions of S arriving at clock 
input CP, producing a signal at Q, having a square-wave width modulation 
output ML1. Flip-flop B3, on the other hand, is set by the positive 
transitions of Ref arriving by its clock input CP producing a signal at 
its direct output. Q is connected to reset R of B1, causing the 
interruption of signal ML1. Flip-flop B3 in turn is reset by the 
complementary output Q0 of B1 which drives its reset R. 
Since, as was seen above, the widths of the square waves of S and Ref are 
very close in size, the level transitions correspond approximately one for 
one, i.e., in general when a positive pulse of S arrives, flip-flop B1 has 
been previously set to the zero state, so that data input D of B2 is at 
zero at the moment this pulse arrives, which has the effect of maintaining 
output Q0 of B2 (referenced A) in state 1. Also when a pulse Ref arrives 
on clock input CP of flip-flop B4, this latter generally has its data 
input D at zero, so that its output Q0 (referenced B) remains in state 1. 
The NOT AND gate, referenced G, therefore normally keeps output signal RZ1 
(reset to zero) at zero. 
However, when S and Ref are close to coinciding, it happens, as shown in 
FIG. 4, that a positive transition of S arrives on B2 before a positive 
pulse of Ref occurs causing the resetting of B1 from the preceding 
positive pulse of S. In this case, data input D of flip-flop B2 will be in 
state 1 when pulse S arrives at its control input CP, which has the effect 
of setting A to the zero state, therefore putting RZ1 in state 1 as 
appears on the timing diagrams of FIG. 4, which corresponds to one 
direction of rotation of the sensor. The same explanation applies to the 
other direction of the sensor with flip-flop B4 and input B of G, which 
corresponds to the timing diagram of FIG. 5. 
All the above relates to digital phase comparator CPN and return to zero 
detector DRZ of the upper part of FIG. 1, and applies identically to the 
similar unit of the lower part of FIG. 1, aside from replacement of signal 
S by signal S This signal, taking into account the nature of the signals, 
is the same as a signal out of phase by .pi., with the outgoing signals, 
moreover, having subscripts 2 instead of the subscript 1. 
The square-wave width modulation signals ML1 and ML2 are each filtered 
through a low-pass filter F, and thus give rise to filtered signals SF1 
and SF2 respectively, whose timing diagram is seen in FIG. 6. For their 
part, return to zero signals RZ1 and RZ2 drive the inputs S (set) and R 
(reset) of an RS type flip-flop, referenced BS, which stores the last 
return to zero and produces at its direct output Q a selection signal Sel. 
Thus controls a switch COM which selects, as appears on the timing 
diagrams of FIG. 6, alternately signals SF1 and SF2 so as to sample the 
second half of these signals, i.e., the strictly linear part of the 
signal, thus eliminating the first half of each sawtooth whose linearity 
is affected by the enable time of the filters. 
Thus there is obtained at output DS of this switch a single sawtooth signal 
represented in the timing diagram of FIG. 6. At the same time, a 
transition detector DT sensitive to the positive or negative transitions 
of Sel, or again to the positive transitions of Sel and of Sel0 produces 
transition signals TR, also represented on the timing diagram of FIG. 6 
and whose position corresponds to the discontinuities of signal DS. 
It should be noted that single sawtooth signal DS is of rising or falling 
sawteeth depending on the direction of rotation of the sensor, and 
therefore it suffices to differentiate this single signal to determine the 
desired tachometry signal, in magnitude and sign, while taking care, 
however, to eliminate the discontinuities due to the transitions of DS. 
For this it is possible, for example, to use the differentiation circuit 
DIM with initialization and memory represented in FIG. 3, where is seen an 
operational amplifier AD mounted as a differentiator with a resistor R1, a 
differentiation capacitor CD and a current limitation resistor RL which is 
charged by signal DS. The direct input of amplifier AD is grounded, so 
that its reversing input is also set to the zero potential by the circuit 
indicated, while the output of amplifier AD is at a potential proportional 
to the desired tachometry signal. 
An electronic switch CO, controlled directly by signal TR, normally turns 
on this circuit with a memory capacitor CM and its load resistor R2, this 
capacitor therefore remaining constantly charged at a voltage 
representative of the tachometry signal. This signal, sampled between CM 
and CO, is sent to a tracking amplifier AS which supplies the outgoing 
tachometry signal ST while introducing a high impedance at the output of 
capacitor CM. When a pulse is present on transition signal TR, switch CO 
therefore switches off the circuit of the memory capacitor, this latter 
continuing to supply the voltage stored in memory to the tracking 
amplifier. At the same time this switch turns on, as indicated by its 
position represented in broken lines in FIG. 3, a short-circuit connection 
CC forces at initialization the differentiation circuit to discharge or 
charge the differentiation capacitor and allow another cycle to begin. The 
device thus assures a perfect continuity of tachometry signal ST. 
Thanks to the use of a single differentiator, it is seen that no pairing of 
components nor any zero or other balancing adjustment is necessary and the 
circuit, which is moreover relatively simple, does not need any 
adjustment. 
The precision of the tachometry signal obtained depends, as mentioned 
above, on the fact that pulse frequency w of the reference signal is 
sufficiently close to the frequency of rotation of the sensor. If this is 
not the case, or if it is desired to improve the precision still more, it 
is possible to add, as shown in FIG. 1, a correction circuit COR which 
takes into account the derivative of the position generated by circuit RC, 
by selecting values R and C so that 
EQU R.C=(1/w).multidot.(1-W/w). 
This correction of R or C can be achieved either by a digital means, such 
as a network of resistors or capacitors switched for different rotation 
speeds, or by an analog means such as a variable gain amplifier or a diode 
with variation of capacity. 
Obviously, numerous modifications and variations of the present invention 
are possible in light of the above teachings. It is therefore to be 
understood that within the scope of the appended claims, the invention may 
be practiced otherwise than as specifically described herein.