Transistor switching circuit having diode-resistor in base of transistor for fast discharge

In a switching circuit formed as an integrated circuit, a series circuit comprising a diode (D.sub.1) and a resistor (R.sub.1) is connected between a base and an emitter of an npn transistor (TR.sub.1) requiring a high speed switching operation. Therefore, a high speed operation is made possible. Furthermore, in a circuit constructed such that the above npn transistor is driven by a pnp transistor (TR.sub.3, TR.sub.4), a leakage current produced in the above pnp transistor at high temperature is allowed to flow in the above series circuit. Accordingly, a malfunction of the above npn transistor is prevented. Consequently, an integrated circuit operable even under high temperatures is achieved.

BACKGROUND OF THE INVENTION 
1. Field of the Invention 
The present invention relates generally to switching circuits, and more 
particularly, to switching circuits suitable for implementation as an 
integrated circuit. 
2. Description of the Related Art 
When a switching circuit comprising an npn switching transistor is formed 
as an integrated circuit, junction capacitance occurs between a base and 
an emitter of this transistor in view of the device configuration. Charges 
stored in this junction capacitance are discharged after passing between 
the base and the emitter of the above switching transistor, thereby to 
prevent fast switching response. 
When the switching circuit is so constructed that the npn switching 
transistor is driven by a pnp transistor, a malfunction of the switching 
transistor occurs due to the increase in leakage current in the pnp 
transistor under high temperatures. 
SUMMARY OF THE INVENTION 
An object of the present invention is to provide a switching circuit 
capable of increasing the speed of switching as well as eliminating the 
possibility of causing the above described malfunction. 
The switching circuit according to the present invention is characterized 
by comprising a first transistor subjected to on-off control by a 
switching control signal applied to its base, a second transistor in the 
output stage controlled by this first transistor, and a series circuit 
comprising a diode functional device and a resistor connected in series 
and in that this series circuit is connected between the base of the above 
first transistor and either one of an emitter and an collector thereof. 
According to the present invention, when the switching control signal for 
turning the first transistor off is applied, charges accumulated in a 
base-emitter junction region of the first transistor are rapidly 
discharged through the above described series circuit comprising the diode 
functional device and the resistor. Accordingly, an output of the first 
transistor is rapidly turned off, thereby allowing the speed of switching 
to be increased. 
Furthermore, a collector leakage current, which is several hundred 
nanoamperes in the environment at high temperatures, in the first 
transistor flows through the above described diode functional device and 
resistor. Accordingly, the collector leakage current does not flow in the 
first transistor by suitably determining the value of this resistor. 
Consequently, a malfunction of the first transistor can be previously 
prevented. 
The foregoing and other objects, features, aspects and advantages of the 
present invention will become more apparent from the following detailed 
description of the present invention when taken in conjunction with the 
accompanying drawings.

DESCRIPTION OF THE PREFERRED EMBODIMENTS 
FIG. 1 shows a switching circuit according to a first embodiment of the 
present invention. 
A first npn transistor TR.sub.1 and a current source CS.sub.1 for 
outputting a current I.sub.1 are connected in series between a power 
supply V.sub.S and ground G. A second npn transistor TR.sub.2 in the 
output stage is controlled by a potential at a node of the transistor 
TR.sub.1 and the current source CS.sub.1. A collector of the transistor 
TR.sub.2 becomes an output terminal OT. The switching transistor TR.sub.1 
is controlled by a current I.sub.2 outputted from a current source 
CS.sub.2. A switching device SW.sub.1 is connected to the current source 
CS.sub.2. 
In the above described circuit construction, when the switching device 
SW.sub.1 is turned on, the output current I.sub.2 of the current source 
CS.sub.2 is applied to a base of the transistor TR.sub.1. Accordingly, the 
transistor TR.sub.1 is turned on. As a result, the output current I.sub.1 
of the current source CS.sub.1 which has been applied to a base of the 
transistor TR.sub.2 flows in the transistor TR.sub.1, so that the 
transistor TR.sub.2 is turned off. On the other hand, when the switching 
device SW.sub.1 is turned off, the output current I.sub.2 of the current 
source CS.sub.2 is cut off. Accordingly, the transistor TR.sub.1 is turned 
off. Consequently, the output current I.sub.1 of the current source 
CS.sub.1 is inputted to the base of the transistor TR.sub.2, so that the 
transistor TR.sub.2 is turned on. 
When such a switching circuit is implemented as an integrated circuit, the 
following problems are encountered. In an integrated circuit, it has been 
known that junction capacitance C.sub.BE occurs between a base and an 
emitter of an npn transistor in view of the device configuration. The 
switching transistor TR.sub.1 is an npn transistor, its junction 
capacitance C.sub.BE being represented by a broken line. 
A part of the current I.sub.2 flowing out of the current source CS.sub.2 
when the switching device SW.sub.1 is on is accumulated in the junction 
capacitance C.sub.BE. When the switching device SW.sub.1 is changed from 
the on state to the off state, charges accumulated in the junction 
capacitance C.sub.BE are discharged passing between the base and the 
emitter of the transistor TR.sub.1. Accordingly, turn-off of the 
transistor TR.sub.1 is delayed. Consequently, the transistor TR.sub.1 can 
not respond to a high speed switching operation. 
The foregoing will be quantitatively described through specific examples. 
The current sources CS.sub.1 and CS.sub.2 shall be very small current 
sources, their output currents I.sub.1 and I.sub.2 being respectively 
taken as 4 .mu.A. In addition, let the direct current amplification factor 
.beta. of the transistor TR.sub.1 be 200. 
When the switching device SW.sub.1 is on, a base potential V.sub.A of the 
transistor TR.sub.1 is approximately 0.6 V. More specifically, the 
junction capacitance C.sub.BE is charged to approximately 0.6 V. 
The base potential V.sub.A is calculated from the following equation: 
EQU V.sub.A =(kT/q) ln (I.sub.E1 /I.sub.S1) 
k; Boltzmann's constant 
T; absolute temperature 
q; charge of an electron 
I.sub.E1 ; emitter current in the transistor TR.sub.1 
I.sub.S1 ; reverse saturation current in the transistor TR.sub.1 
Furthermore, a collector current I.sub.C1 in the transistor TR.sub.1 is 
represented by the following equation: 
##EQU1## 
The transistor TR.sub.1 is turned on in the saturation region. 
Immediately after the switching device SW.sub.1 is switched from the on 
state to the off state, the following equations hold: 
EQU I.sub.C1 =4 .mu.A, and 
EQU V.sub.A =0.6 V 
A base current I.sub.B1 in the transistor TR.sub.1 is represented by the 
following equation: 
##EQU2## 
Charges accumulated in the junction capacitance C.sub.BE begins to be 
discharged with an initial current of 0.02 .mu.A. Accordingly, the 
collector current I.sub.C1 is gradually decreased. 
In order to overcome such problems, according to the present invention, a 
series circuit comprising a series connection of a diode-connected 
transistor (diode functional device) D.sub.1 and a resistor R.sub.1 is 
connected between the base of the switching transistor T.sub.R1 and the 
ground G. 
The function of this series circuit will be quantitatively described 
through specific examples. 
The current sources CS.sub.1 and CS.sub.2 shall be very small current 
sources, their output currents I.sub.1 and I.sub.2 being taken as 4 .mu.A. 
In addition, let the direct current amplification factor .beta. of the 
transistor TR.sub.1 be 200, and let the resistance value of the resistor 
R.sub.1 be 50 K.omega.. 
The base potential V.sub.A of the transistor TR.sub.1 is represented by the 
following equation: 
##EQU3## 
where V.sub.BE1 is a voltage between the base and the emitter of the 
transistor TR.sub.1, V.sub.BED is a voltage between a base and an emitter 
of the diode-connected transistor D.sub.1, and I.sub.4 is a current 
flowing in the resistor R.sub.1. 
Accordingly, taking V.sub.T =kT/q=0.0259 V (T=300K), if I.sub.S is 
eliminated from the equation (2), the following equation (3) is obtained: 
EQU V.sub.T ln(I.sub.C1 /I.sub.S)=V.sub.T ln(I.sub.4 /I.sub.S)+R.sub.1 
.multidot.I.sub.4 (2) 
EQU R.sub.1 .multidot.I.sub.4 =V.sub.T ln(I.sub.C1 /I.sub.4) (3) 
In a case where the switching device SW.sub.1 is on, when R.sub.1 =50 
K.omega., V.sub.T =0.0259 and I.sub.C1 =I.sub.1 =4 .mu.A are substituted 
in the equation (3), the following relations hold: 
EQU I.sub.4 =0.82 .mu.A (4) 
EQU I.sub.B1 =3.18 .mu.A (5) 
Immediately after the switching device SW.sub.1 is switched from the on 
state to the off state, from the following relations, 
EQU V.sub.A =0.6 V 
EQU I.sub.C1 =4 .mu.A 
the following equation is obtained: 
##EQU4## 
Furthermore, from the equation (4), the following relation hold: 
EQU I.sub.4 =0.82 .mu.A (7) 
Charges accumulated in the junction capacitance C.sub.BE begin to be 
discharged with an initial current of (I.sub.B1 +I.sub.4). This initial 
current is approximately 40 times the above described initial current 
(0.02 .mu.A) in a case where no series circuit is provided. The current 
I.sub.C1 flowing in the transistor TR.sub.1 is rapidly decreased, so that 
the transistor TR.sub.1 is turned off at high speed. 
In the above described manner, fast response of switching is achieved. 
FIG. 2 shows a switching circuit according to a second embodiment in which 
a switching transistor TR.sub.1 is controlled through a current mirror 
circuit CM.sub.1. The current mirror circuit CM.sub.1 comprises two pnp 
transistors TR.sub.3 and TR.sub.4. If a switching device SW.sub.2 is 
turned on, an output current I.sub.3 of a current source CS.sub.3 flows 
into a base of the transistor TR.sub.1 through the current mirror circuit 
CM.sub.1 as a current I.sub.2 (I.sub.2 =I.sub.3). Accordingly, this 
transistor TR.sub.1 is turned on and a transistor TR.sub.2 in the output 
stage is turned off. On the other hand, if the switching device SW.sub.2 
is turned off, the current flowing into the base of the transistor 
TR.sub.1 from the current mirror circuit CM.sub.1 is theoretically cut 
off. Accordingly, the transistor TR.sub.1 is turned off and the transistor 
TR.sub.2 is turned on. 
A collector leakage current I.sub.CEO in the pnp transistor is several 
hundred picoamperes at ordinary temperatures, while being rapidly 
increased to approximately several hundred nanoamperes at higher 
temperatures around 100.degree. C., as shown in FIG. 5. The switching 
circuit constructed as described above has the following disadvantage. 
More specifically, if the switching circuit is driven by very small 
currents I.sub.1 and I.sub.3 of approximately 1 to 4 .mu.A, a collector 
current I.sub.C1 in the transistor TR.sub.1 is on the order of .mu.A at 
temperatures around 100.degree. C. from the equation I.sub.C1 =h.sub.fe 
.times.I.sub.CEO. Accordingly, the transistor TR.sub.1 is turned on and 
the transistor TR.sub.2 is turned off only by the collector leakage 
current in the pnp transistor TR.sub.3 constituting the current mirror 
circuit CM.sub.1. 
The foregoing will be quantitatively described as follows. 
Let the collector leakage current I.sub.CEO in the pnp transistor be 100 pA 
at a temperature of 25.degree. C., 100 nA at a temperature of 100.degree. 
C., and 1 .mu.A at a temperature of 125.degree. C. In addition, the 
current sources CS.sub.1 and CS.sub.3 shall be very small current sources, 
their output currents I.sub.1 and I.sub.3 being taken as 4 .mu.A. 
(1) When the switching device SW.sub.2 is in the off state, compare a case 
where temperature is 25.degree. C. with a case where it is 100.degree. C. 
Case Where Temperature is 25.degree. C. 
Let the direct current amplification factor .beta. in the transistor 
TR.sub.1 be 200. The collector current I.sub.C1 in the transistor TR.sub.1 
is represented by the following equation: 
##EQU5## 
Therefore, the transistor TR.sub.1 is off, so that the output transistor 
TR.sub.2 is fully driven by the current I.sub.1. Accordingly, the 
transistor TR.sub.2 is turned on. More specifically, a normal operation is 
performed. 
Case Where Temperature is 100.degree. C. 
The collector current I.sub.C1 in the transistor TR.sub.1 is represented by 
the following equation: 
##EQU6## 
Consequently, the transistor TR.sub.1 is turned on by the collector leakage 
current I.sub.CEO, so that no driving current is applied to a base of the 
output transistor TR.sub.2. Accordingly, the transistor TR.sub.2 is turned 
off. More specifically, a malfunction occurs. 
(2) Description is now made of a case where the switching device SW.sub.2 
is on. The following relation holds: 
##EQU7## 
Accordingly, the transistor TR.sub.1 is turned on, so that the driving 
current I.sub.1 to the output transistor TR.sub.2 can be fully cut off. 
Accordingly, the transistor TR.sub.2 is turned off. More specifically, a 
normal operation is performed. 
In order to the above described problems, according to the present 
invention, a series connecting circuit comprising a diode-connected 
transistor D.sub.1 and a resistor R.sub.1 is connected between the base of 
the switching transistor TR.sub.1 and ground G. 
The function of this series connecting circuit will be quantitatively 
described as follows. 
A case where the switching device SW.sub.2 is on is described by the 
expressions (4) and (5). 
Then, when the switching device SW.sub.2 is off, compare a case where the 
ambient temperature is 25.degree. C. with a case where it is 100.degree. 
C. 
When temperature is 25.degree. C., I.sub.CEO =100 pA. Accordingly, even if 
I.sub.4 =100 pA, the following relation holds from the equation (3): 
EQU I.sub.C1 .apprxeq.100 pA&lt;&lt;I.sub.1 (8) 
This collector current is substantially smaller than the driving current 
I.sub.1 in the transistor TR.sub.2 in the output stage. Accordingly, the 
transistor TR.sub.2 is turned on. More specifically, a normal operation is 
performed. 
When temperature is 100.degree. C., I.sub.CEO =100 nA. Accordingly, even if 
I.sub.4 =100 nA, the following relation holds from the equation (3): 
EQU I.sub.C1 .apprxeq.0.12 .mu.A&lt;&lt;I.sub.1 (9) 
This collector current is substantially smaller than the driving current 
I.sub.1 in the transistor TR.sub.2. Accordingly, the transistor TR.sub.2 
is turned on. More specifically, a normal operation is achieved also in 
this case. 
As described in the foregoing, the collector leakage current I.sub.CEO, 
which becomes several hundred nanoamperes in the environment at high 
temperatures, in the pnp transistor TR.sub.3 flows through the 
diode-connected transistor D.sub.1 and the resistor R.sub.1. Accordingly, 
this leakage current does not flow in the base of the switching transistor 
TR.sub.1 by setting the resistor R.sub.1 to a suitable value. 
Consequently, occurrence of a malfunction can be previously prevented. 
Meanwhile, the same effect can be theoretically obtained even if a series 
circuit comprising a diode functional device and a resistor is replaced 
with only a resistor. When a resistor is formed in an integrated circuit, 
however, the area of the resistor is significantly increased. Accordingly, 
the area which is approximately 10 to 15 times larger than that in the 
above described embodiment is required to obtain the same effect using 
only the resistor. Consequently, construction in the embodiments shown in 
FIGS. 1 and 2 (a series connecting circuit comprising a diode functional 
device and a resistor) is preferable so as to increase integration 
density. 
FIG. 3 shows an application of the above described switching circuit. A 
circuit shown in FIG. 3 is a power reset circuit provided in a proximity 
switch or the like. More specifically, immediately after the power supply 
is turned on, a detection signal of the proximity switch or the like may, 
in some cases, present an erroneous detected state due to the rise of each 
circuit. In order to prevent such a malfunction, the power rest circuit 
performs such a function as to inhibit the detection signal from being 
outputted during a constant time period immediately after the power supply 
is turned on. A signal for inhibiting output of the detection signal is 
outputted from the transistor TR.sub.2. FIG. 4 is a waveform diagram 
showing an operation of this power reset circuit. 
Referring now to FIGS. 3 and 4, the power reset circuit includes a constant 
voltage circuit 11. When the power supply is turned on, an output voltage 
of this constant voltage circuit 11 rises, to be settled at a constant 
voltage V.sub.S at the time point when a given time period has elapsed. An 
output inhibiting signal is outputted in the time T.sub.X elapsed from the 
time of turn-on of the power supply until the output voltage of the 
constant voltage circuit 11 is settled at the constant voltage V.sub.S. 
The switching circuit is incorporated in a flip-flop 10 and an output 
buffer circuit 20 driven by said flip-flop. The flip-flop 10 is initially 
reset immediately after the power supply is turned on. More specifically, 
the flip-flop 10 is reset such that a potential at the point A attains an 
H level and a potential at the point B attains an L level. As a result, 
current flows into an output buffer circuit 20 through a current mirror 
circuit comprising a transistor TR.sub.19. Accordingly, an output 
transistor TR.sub.2 is turned on (output is inhibited). The constant time 
T.sub.X is measured by charging time of a capacitor C.sub.O. When the 
capacitor C.sub.O is charged, until its output voltage (a potential at the 
point C) reaches a certain value, a transistor TR.sub.9 is turned on. In 
addition, a transistor TR.sub.17 is turned on. As a result, the potential 
at the point A is pulled down to the L level. 
This power reset circuit will be described in more detail. A current source 
CS is connected to the output side of the constant voltage circuit 11. 
Transistors TR.sub.21, TR.sub.22, TR.sub.23, TR.sub.24 and TR.sub.25 
respectively serving as current sources are driven by this current source 
CS. 
The flip-flop 10 comprises transistors TR.sub.18 and TR.sub.19 respectively 
constituting current mirror circuits and transistors TR.sub.15 and 
TR.sub.16. The transistor TR.sub.16 corresponds to the switching 
transistor TR.sub.1 shown in FIGS. 1 and 2. A series circuit comprising a 
diode D.sub.1 and a resistor R.sub.1 is connected between a base of this 
transistor TR.sub.16 and ground G. Transistors TR.sub.14 and TR.sub.17 for 
inversion are respectively connected in parallel to the transistors 
TR.sub.15 and TR.sub.16. The transistor TR.sub.14 corresponds to the above 
described switching device SW.sub.2. 
Transistors TR.sub.12, TR.sub.13 and TR.sub.14 are turned on when an output 
voltage of the constant voltage circuit 11 becomes approximately 2V.sub.BE 
(approximately 1.2 V), while a transistor TR.sub.11 is turned on when it 
becomes approximately 3V.sub.BE (approximately 1.8 V). 
When the output voltage of the constant voltage circuit 11 begins to rise, 
the capacitor C.sub.O begins to be charged through the transistor 
TR.sub.22. When the output voltage becomes approximately 2V.sub.BE, the 
transistors TR.sub.12, TR.sub.13 and TR.sub.14 are turned on. The 
transistor TR.sub.12 is turned on, thereby causing charges accumulated in 
the capacitor C.sub.O to be discharged through this transistor TR.sub.12. 
In addition, the transistor TR.sub.14 is turned on, thereby causing the 
potential at the point B to attain the L level. Accordingly, a collector 
current in the transistor TR.sub.14 is not applied to the base of the 
transistor TR.sub.16, so that the transistor TR.sub.16 is turned off. As a 
result, the potential at the point A attains the H level. This is initial 
reset of the flip-flop. A transistor TR.sub.20 is also turned off. 
Thereafter, when the output voltage of the constant voltage circuit 11 
becomes approximately 3V.sub.BE, the transistor TR.sub.11 is turned on. 
Accordingly, the transistors TR.sub.12, TR.sub.13 and TR.sub.14 are turned 
off. Consequently, charging of the capacitor C.sub.O is resumed. The 
initial reset state of the flip-flop 10 continues until a charging voltage 
of the capacitor C.sub.O reaches a predetermined value as described above. 
In the foregoing, if the transistor TR.sub.16 is not turned off in the time 
(time T.sub.1) elapsed from the time when the transistors TR.sub.12, 
TR.sub.13 and TR.sub.14 are turned on until they are turned off, the 
flip-flop is not initially reset. More specifically, let T.sub.2 be the 
time elapsed from the time of turn-on of the transistor TR.sub.12 or the 
like until the transistor TR.sub.16 is turned off. In this case, the 
relation T.sub.1 &gt;T.sub.2 must be satisfied. 
However, when the rise of the output voltage of the constant voltage 
circuit 11 becomes abrupt, the time T.sub.1 becomes short. Accordingly, 
the transistor TR.sub.16 must be quickly switched from the on state to the 
off state. This is a reason why the series circuit comprising the diode 
D.sub.1 and the resistor R.sub.1 is connected to the base of the 
transistor TR.sub.16. 
Although the present invention has been described and illustrated in 
detail, it is clearly understood that the same is by way of illustration 
and example only and is not to be taken by way of limitation, the spirit 
and scope of the present invention being limited only by the terms of the 
appended claims.