Control apparatus for thyristor motor

A control apparatus for thyristor motor comprises a voltage detector (27), a speed detecting circuit (600) and a correction circuit (800). The voltage detector (27) detects DC voltage (E.sub.d) supplied to an inverter circuit (200). The speed detecting circuit (600) detects the rotational speed of a synchronous motor (3) and provides speed voltage (E.omega.) proportional thereto. The correction circuit (800) detects a difference between the detected DC voltage (E.sub.d) and the speed voltage (E.omega.) and provides a correction signal to an excitation circuit (700) when a difference is detected. The excitation circuit (700) corrects field current in response to the correction signal so that the DC voltage (E.sub.d) and the speed voltage may be equal. Thus, a proportional relation can be maintained between the instructed torque and the generated torque.

BACKGROUND OF THE INVENTION 
1. Field of the Invention 
The present invention relates to a control apparatus for a thyristor motor 
and more particularly to a control apparatus for a thyristor motor having 
a compensation field coil for generating compensation field flux 
orthogonally intersecting with main field flux so as to compensate 
armature reaction. 
2. Description of the Prior Art 
A thyristor motor is a synchronous motor driven by semiconductor commutator 
means. FIG. 1 is a block diagram showing a conventional control apparatus 
for a thyristor motor. Although the below described direct current 
I.sub.d, DC voltage E.sub.d, main field current I.sub.f, compensation 
field current I.sub.c etc. in a real state and signals for detecting or 
instructing them do not have the same values, the real values and the 
signals will be sometimes regarded as the same in the present 
specification for the purpose of facilitating the explanation. The 
synchronous motor 3 has armature coils U, V and W and field coils 310 
comprised of a main field coil F and a compensation field coil C. The main 
field coil F generates main field flux and the compensation filed coil C 
generates compensation field flux orthogonally intersecting with the main 
field flux. To the rotating axis of the synchronous motor 3, a position 
sensor 4 and a tachometer generator 6 are connected. The position sensor 4 
provides a position signal of a phase according to the rotational angle of 
the rotating axis of the synchronous motor 3. The tachometer generator 6 
generates voltage proportional to the rotational speed of the rotating 
axis of the synchronous motor 3. A control apparatus for a thyristor motor 
comprises in rough a power supply circuit 100, an inverter circuit 200, an 
excitation circuit 70 and a speed instructing circuit 900. The power 
supply circuit 100 comprises a converter 1, a current detector 9, a 
current controller 10, a gate pulse phase shifter 11 and a coefficient 
multiplier 20. The converter 1 converts AC power of a commercial AC power 
source into DC power. The current detector 9 rectifies AC input current of 
the converter 1 and provides a signal proportional to the DC current 
I.sub.d outputted from the converter 1. The coefficient multiplier 20 
multiplies, by a predetermined coefficient, an instructed value of torque 
outputted from a speed controller 8 to be described below so as to provide 
an instructed value of current for the converter 1. The current controller 
10 amplifies a deviation between an output signal of the coefficient 
multiplier 20 and an output signal of the current detector 9. The gate 
pulse phase shifter 11 controls an ignition phase of the converter 1 
according to the output signal of the current controller 10. The inverter 
circuit 200 comprises an inverter 2 and a gate amplifier 5. The gate 
amplifier 5 provides a gate signal to the inverter 2 based on a position 
signal from the position sensor 4. The inverter 2 commutates the DC power 
from the power supply circuit 100 in response to the gate signal so as to 
supply the power to the armature coils U, V and W. The speed instructing 
circuit 900 comprises a speed instructing device 7 and a speed controller 
8. The speed instructing device 7 provides a speed instructing signal for 
making the rotational speed of the synchronous motor 3 be a predetermined 
rotational speed. The speed controller 8 examines and amplifies a 
difference between the speed instructing signal from the speed instructing 
device 7 and a speed feedback signal from the tachometer generator 6. The 
excitation circuit 70 comprises a main excitation circuit 71 and a 
compensation excitation circuit 72. The main excitation circuit 71 
comprises a field instructing device 12, a coefficient multiplier 21, an 
adder 22, a current detector 13, a current controller 14, a gate pulse 
phase shifter 15 and a converter 16. The field instructing device 12 
instructs a no-load value I.sub.fo of the main field current I.sub.f. The 
coefficient multiplier 21 multiplies by a predetermined coefficient, an 
instructed value of torque outputted from the speed controller 8 so as to 
apply a correction amount .DELTA.I.sub.f for a demagnetized amount of the 
main field current I.sub.f in the loaded condition. The adder 22 performs 
addition of an instructed value of field current outputted from the field 
instructing device 12 and a correction amount outputted from the 
coefficient multiplier 21 so as to obtain an instructed value I.sub.fp of 
field current represented by the equation I.sub.fp =I.sub.fo 
+.DELTA.I.sub.f. The current detector 13 rectifies the AC input of the 
converter 16 for control of the main field so as to detect the amount of 
main field current I.sub.f. The current controller 14 amplifies a 
deviation between the signal I.sub.fp and a detected value of current 
outputted from the current detector 13. The gate pulse phase shifter 15 
controls an ignition phase of the thyristor in the converter 16 according 
to the output of the current controller 14. The converter 16 supplies a 
main field current I.sub.f in response to a signal from the gate pulse 
phase shifter 15. The compensation excitation circuit 72 comprises a 
coefficient multiplier 23, a current detector 17, a current controller 18, 
a gate pulse phase shifter 19 and a converter 24. The coefficient 
multiplier 23 multiplies an instructed value of torque outputted from the 
speed controller by a predetermined coefficient so as to provide 
instruction of current for compensation field. The current detector 17 
rectifies the AC input to the converter 24 for control of compensation 
field so as to detect the amount of compensation field current I.sub.c. 
The current controller 18 examines and amplifies a difference between an 
instructed value of compensation filed current outputted from the 
coefficient multiplier 23 and a detected value of current outputted from 
the current detector 17. The gate pulse phase shifter 19 supplies ignition 
pulses to the thyristors in the converter 24 according to the output of 
the current controller 18. The converter 24 supplies a compensation field 
current I.sub.c in response to a signal from the gate pulse phase shifter 
19. 
Now, description will be made of a total operation. The position sensor 4, 
the gate amplifier 5 and the inverter 2 operate so that the phase of the 
armature current I.sub.a of the synchronous motor 3 may be a predetermined 
phase with respect to the rotating phase of the field flux. The tachometer 
generator 6, the speed instructing device 7 and the speed controller 8 
provide instruction of torque so that the rotational speed of the 
synchronous motor 3 may be equal to the instructed speed. The coefficient 
multiplier 20 multiplies the instruction of torque by a coefficient 
determined by various constants of the synchronous motor 3 so as to 
instruct armature current I.sub.a necessary for generating torque equal to 
the instructed value. The process in which the direct current I.sub.d is 
controlled to a predetermine value by the current detector 9, the current 
controller 10, the gate pulse phase shifter 11 and the converter 1 is well 
known. The field instructing device 12 supplies a reference value I.sub.fo 
of the main field current in the no-load condition, and this reference 
value becomes an instructe-d value of current I.sub.fp after a field 
current increment .DELTA.I.sub.f for correction of a demagnetized amount 
in the loaded condition is added to the reference value. The process in 
which the main field current I.sub.f is controlled to be a predetermined 
value by means of the current detector 13, the current controller 14, the 
gate pulse phase shifter 15 and the converter 16 is well known. The 
coefficient multiplier 23 instructs compensation field current I.sub.c 
necessary for compensation of the armature reaction determined by the 
constants of the motor. The instructed torque and the armature current 
I.sub.a are maintained in a proportional relation and the compensation 
field current I.sub.c and the instructed torque are also maintained in a 
proportional relation. Accordingly, the armature current I.sub.a and the 
compensation field current I.sub.c are controlled in proportion to each 
other. The current detector 17, the current controller 18, the gate pulse 
phase shifter 19 and the converter 24 control the compensation field 
current I.sub.c according to the instructed value. 
FIG. 2A is a vector diagram showing a relation between the voltage and the 
current of the FIG. 1 motor in the no-load condition. FIG. 2B is a vector 
diagram showing a relation between the voltage and the current of the FIG. 
1 motor in the loaded condition. The inverter 2 is an external commutated 
inverter and, therefore, it is necessary to provide current in a leading 
power factor for commutation. For this reason, the position sensor 4 is 
disposed in the synchronous motor 3 so that the armature current I.sub.a 
may flow in the direction advancing by an angle .gamma. with respect to 
the no-load induced voltage E.sub.o. In the loaded condition, the armature 
current I.sub.a causes voltage X.sub.s I.sub.a in the direction shown in 
FIG. 2B due to the armature reaction. The voltage X.sub.s I.sub.a includes 
a direct-axis component and a quadrature-axis component. Voltage X.sub.s 
I.sub.c caused by the compensation field coil is generated in the 
direction shown in FIG. 2B. This voltage X.sub.s I.sub.c compensates the 
quadrature component of the armature reaction. I.sub.f this state 
continues, the induced voltage V in the loaded condition becomes smaller 
than the induced voltage E.sub.o in the no-load condition and as a result, 
a sufficient output of the motor cannot be obtained. For this reason, 
voltage X.sub.s .DELTA.I.sub.f is generated by increasing the main field 
current by an amount .DELTA.I.sub.f, whereby the induced voltage having 
the same amount as that in the no-load condition can be obtained. 
As is understood from the foregoing description, in a conventional control 
apparatus, a compensation field current I.sub.c and a correction value 
.DELTA.I.sub.f of the main field current are made to flow in proportion to 
the armature current I.sub.a so that the armature reaction may be 
compensated. In such a method, a precise compensation can be made as far 
as a vector relation as shown in FIG. 2B is maintained, and since the 
torque generated in the motor is proportional to the armature current 
I.sub.a, torque control can also be made with precision. However, in 
reality, it is well known that commutation of the inverter 2 is not 
provided instantaneously and that an overlapping angle of commutation is 
caused. As a result, a delay from the determined angle .gamma. is caused 
in the phase of the armature current I.sub.a and this phase delay becomes 
a significant amount as the frequency (the rotational speed of the motor) 
becomes high. Furthermore, the larger is the armature current I.sub.a, the 
greater is the phase delay and accordingly, deviation in the vector 
relation changes according to the change in the instruction of torque. 
Thus, the phase .gamma. of the armature current changes according to the 
changes in the rotational speed of the motor or in the armature current 
and, therefore, the direction of the armature reaction changes. 
Accordingly, the armature reaction cannot be compensated with precision by 
applying the compensation field current I.sub.c and the correction value 
.DELTA.I.sub.f of the main field current. As a result, deviation is caused 
both in the amount and in the phase of the induced voltage V in the loaded 
condition with respect to the no-load induced voltage E.sub.o. As 
described above, a conventional control apparatus has a disadvantage in 
that a phase relation between the armature current I.sub.a and the induced 
voltage V cannot be maintained in a loaded condition and accordingly 
torque cannot be obtained in accordance with the instruction of torque. 
SUMMARY OF THE INVENTION 
Therefore, a primary object of the present invention is to provide a 
control apparatus for a thyristor motor in which torque control can be 
made precisely even if the phase of the armature current is changed due to 
an overlapping angle of commutation. 
Briefly stated, the present invention is a control apparatus for a 
synchronous motor including armature coils and field coils having a main 
field coil for generating main field flux and a compensation field coil 
for generating compensation field flux orthogonally intersecting with the 
main field flux, and the above stated control apparatus comprises: power 
supply means for supplying DC power having a certain amount of DC voltage; 
inverter means for commutating DC power and for supplying it to the 
armature coils; excitation circuit means for supplying field current to 
the field coils and for correcting the field current in response to a 
correction signal applied thereto; voltage detecting means for detecting 
DC voltage; speed detecting means for detecting the rotational speed of 
the synchronous motor and for providing speed voltage of an amount having 
a functional relation with the rotational speed; and a correction means 
for establishing a correction signal based on a difference between the 
detected DC voltage and the speed voltage and for applying the signal to 
the excitation circuit means. 
According to the present invention, if a difference is caused between the 
detected DC voltage and the speed voltage, a correction signal is applied 
from the correction means to the excitation circuit means. The excitation 
circuit means corrects the field current in response to a correction 
signal so that the DC voltage and the speed voltage may be equal. Thus, 
the DC voltage is controlled to be always equal to the speed voltage. 
Therefore, according to the present invention, it was ascertained that a 
proportional relation can be maintained between the instructed torque and 
the generated torque. 
A principal advantage of the present invention is that a proportional 
relation can be maintained between the instructed torque and the generated 
torque even if the phase of the armature current changes by the influence 
of an overlapping angle of commutation, whereby torque control can be made 
precisely. 
Another advantage of the present invention is that since the DC voltage can 
be maintained to a predetermined value, the rated voltage of the power 
supply means and the inverter means can be made small. 
These objects and other objects, features, aspects and advantages of the 
present invention will become more apparent from the following detailed 
description of the present invention when taken in conjunction with the 
accompanying drawings.

DESCRIPTION OF THE PREFERRED EMBODIMENTS 
FIG. 3 is a block diagram showing an embodiment of the present invention. A 
synchronous motor 3 comprises armature coils U, V and W and field coils 
310 having a main field coil F for generating main field flux and a 
compensation coil C for generating compensation field flux orthogonally 
intersecting with the main field flux. The synchronous motor 3 is 
connected with a position sensor 4 for providing a position signal of a 
phase according to the rotational angle of the rotating axis of the motor. 
The control apparatus comprises a power supply circuit 100, an inverter 
circuit 200, an excitation circuit 700, a voltage detector 27, a speed 
detecting circuit 600 and a correction circuit 800. The power supply 
circuit 100 supplies DC power having a certain amount of DC voltage 
E.sub.d. The inverter circuit 200 is connected to the power supply circuit 
100 and the armature coils U, V and W and serves to supply the DC power 
from the power supply circuit 100 to the armature coils U, V and W after 
commutation based on the position signal from the position sensor 4. The 
excitation circuit 700 is connected to the field coils 310 and serves to 
supply field current to the field coils 310 and to correct the field 
current in response to a correction signal applied thereto. The voltage 
detector 27 is connected to the power supply circuit 100 and serves to 
detect the DC voltage E.sub.d. The speed detecting circuit 600 comprises a 
tachometer generator 6 coupled to the synchronous motor 3 and a 
coefficient multiplier 25 connected to the tachometer generator 6 and 
serves to detect rotational speed of the synchronous motor 3 and to 
generate speed voltage E.sub..omega. of an amount having a certain 
functional relation with the rotational speed. The correction circuit 800 
is connected to the excitation circuit 700, the voltage detector 27 and 
the speed detecting circuit 600 and serves to establish the above 
described correction signal based on a difference between the detected DC 
voltage E.sub.d and the speed voltage E.sub..omega. and to apply the 
signal to the excitation circuit 700. In the following, the block diagram 
in FIG. 3 will be described in detail. 
FIG. 4 is a block diagram showing the first embodiment of the present 
invention. The power supply circuit 100, the inverter circuit 200, the 
tachometer generator 6, the position sensor 4 and the speed instructing 
circuit 900 shown in this block diagram are respectively the same as those 
in FIG. 1 and therefore, description thereof is omitted. The excitation 
circuit 700 comprises a main excitation circuit 710 and a compensation 
excitation circuit 720. The main excitation circuit 710 is the same as the 
main excitation circuit 71 in FIG. 1, while the compensation excitation 
circuit 720 is different from the compensation excitation circuit 72 in 
FIG. 1 in that the circuit 720 has an adder 30. The coefficient multiplier 
25 multiplies, by a predetermined coefficient, the voltage proportional to 
the rotational speed of the synchronous motor 3 outputted from the 
tachometer generator 6 and provides an output as speed voltage 
E.sub..omega., whereby a reference DC voltage at no-load is provided. The 
correction circuit 800 comprises a coefficient multiplier 26, a subtractor 
28 and a DC voltage controller 29. The coefficient multiplier 26 
multiplies the instruction of torque outputted from the speed controller 8 
by a predetermined coefficient and provides voltage corresponding to the 
voltage drop in the armature coils. The subtractor 28 subtracts the output 
voltage of the coefficient multiplier 26 from the DC voltage detected by 
the voltage detector 27. The DC voltage controller 29 amplifies a 
deviation obtained by subtracting the speed voltage outputted from the 
coefficient multiplier 25 from the DC voltage outputted from the 
subtractor 28 and provides a correction signal .DELTA.I.sub.c of the 
compensation field. The adder 30 makes addition of the instructed value 
I.sub. c of compensation field current outputted from the coefficient 
multiplier 23 and the correction signal .DELTA.I.sub.c outputted from the 
DC voltage controller 29 so as to obtain a new instructed value I.sub.c '. 
The torque generated by the synchronous motor 3 is proportional to the 
value obtained by dividing, by the rotational speed, the electric power 
after subtraction of copper loss from the input power. On the other hand, 
the input of the motor is equal to the DC input of the inverter 2. 
Assuming that the DC voltage is E.sub.d, the DC current is I.sub.d, the 
rotational angular speed of the motor is .omega., and the resistance for 
one phase of the armature coil is R.sub.a, the generated torque T is 
represented by the following equation. 
##EQU1## 
By transposing the above described equation, the following equation is 
obtained. 
##EQU2## 
From the equation (2), it can be understood that the rotational speed 
.omega. and the voltage (E.sub.d -2R.sub.a I.sub.d) are controlled to be 
in a proportional relation in order that torque may be generated in 
proportion to the DC current I.sub.d. 
The tachometer generator 6, the coefficient multiplier 25 and the 
correction circuit 800 in FIG. 4 constitutes a feedback control system for 
correcting the compensation field current so as to maintain the above 
described proportional relation. More specifically, the coefficient 
multiplier 25 generates speed voltage E.sub..omega. proportional to the 
rotational speed .omega.; the voltage detector 27 detects DC voltage 
E.sub.d ; coefficient multiplier 26 calculates voltage 2R.sub.a I.sub.d ; 
and the subtractor 28 calculates voltage (E.sub.d -2R.sub.a I.sub.d). The 
DC voltage controller 29 compares the voltage (E.sub.d -2R.sub.a I.sub.d) 
with the speed voltage E.sub..omega. and provides a correction signal 
.DELTA.I.sub.c in a direction for increasing the compensation field 
current I.sub.c ' if the voltage (E.sub.d -2R.sub.a I.sub.d) becomes 
larger than the speed voltage E.sub..omega., and provides a correction 
signal .DELTA.I.sub.c in a direction for decreasing the compensation field 
current I.sub.c ' if the voltage (E.sub.d -2R.sub.a I.sub.d) becomes 
smaller than E.sub..omega. reversely. As a result, the compensation field 
current I.sub.c ' changes so that the DC voltage E.sub.d may change to 
cause the output of the subtractor 28 to be always equal to the output of 
the coefficient multiplier 25. Such operation will be described in further 
detail with reference to the vector diagrams. 
FIG. 5A is a vector diagram for explaining the operation in the FIG. 4 
embodiment. FIG. 5B is a vector diagram showing a process where the 
compensation field current is regulated. Referring to FIG. 5A, a line OP 
shows a reference phase .gamma. of an armature current vector I.sub.a 
determined by a position signal from the position sensor 4. In reality, 
the armature current flows in a phase .gamma.' with a little delay from 
the reference phase due to an overlapping angle of commutation. As a 
result, a phase difference between the no-load induced voltage E.sub.o and 
the armature current I.sub.a becomes small and the power factor becomes 
good and accordingly, the generated torque becomes larger than the 
instructed torque. However, if it is assumed that the compensation field 
current I.sub.c having a little larger amount than a predetermined value 
flows as shown in FIG. 5A, voltage X.sub.s I.sub.c generated by the action 
of the compensation field coil causes the phase of the induced voltage V 
to be delayed by .theta. and as a result the power factor can be returned 
to the original state. Therefore, it is understood that by changing the 
compensation field current the power factor can be regulated and 
accordingly the torque generated by the motor can also be regulated. 
Referring to FIG. 5B, if the armature resistance R.sub.a is disregarded for 
the purpose of facilitating the explanation and assuming that an angle 
formed by the induced voltage V and the armature current I.sub.a is 
.theta., the DC voltage E.sub.d is represented by the following equation. 
##EQU3## 
where cos .theta. indicates a power factor. This equation (3) indicates 
that the DC voltage E.sub.d is proportional to a component of the I.sub.a 
direction in the vector V of the induced voltage. In FIG. 5B, it is 
assumed that the DC voltage for generating a desired torque is E.sub.d, 
the induced voltage at this time is V and the compensation field current 
is I.sub.c. Now let us assume a case where the compensation field current 
instructed from the coefficient multiplier 23 is I.sub.c1 which is smaller 
than I.sub.c. In this case, the voltage X.sub.s I.sub.c1 becomes smaller 
than the voltage X.sub.s I.sub.c and accordingly, the induced voltage 
becomes V.sub.1. As a result, the DC voltage is increased to the voltage 
E.sub.d1 which is an I.sub.a direction component of the induced voltage 
V.sub.1. Although, in reality, the vector direction of the armature 
current I.sub.a slightly changes since the overlapping angle of 
commutation is slightly changed, such slight change is disregarded in this 
case for the purpose of facilitating the explanation. In consequence, the 
output of the subtractor 28 becomes larger than the output of the 
coefficient multiplier 25 and the DC voltage controller 29 provides a 
positive output +.DELTA.I.sub.c, which is added to the current I.sub.c1 by 
the adder 30, so that the compensation field current is regulated to the 
correct value I.sub.c. On the contrary, if the compensation field current 
instructed from the coefficient multiplier 23 is I.sub.c2 which is larger 
than I.sub.c, the induced voltage becomes V.sub.2. Consequently, the DC 
voltage is decreased to E.sub.d2 and the output of the subtractor 28 
becomes smaller than the output of the coefficient multiplier 25 and the 
DC voltage controller 29 provides a negative output -.DELTA.I.sub.c, which 
is added to the current I.sub.c2 by the adder 30, whereby the compensation 
field current is brought back to the correct value I.sub.c also in this 
case. 
Thus, the voltage (E.sub.d -2R.sub.a I.sub.d) outputted from the subtractor 
28 is controlled to be always equal to the speed voltage E.sub..omega. 
outputted from the coefficient multiplier 25, and accordingly, torque T is 
proportional to the DC current I.sub.d, as is clear from the equation (2). 
More specifically, even if the vector of the armature current is changed 
due to an overlapping angle of commutation, the generated torque becomes 
proportional to the instructed torque and accordingly, the precision of 
torque control is improved. In addition, since a surplus amount of 
compensation field current is caused to flow when the overlapping angle of 
commutation becomes large, the margin time of commutation can be increased 
effectively. 
Although in the FIG. 4 embodiment, a value obtained by subtraction of the 
voltage 2R.sub.a I.sub.d corresponding to a resistance drop from a 
detected value E.sub.d of DC voltage is compared with speed voltage 
E.sub..omega., it goes without saying that the same result as in the case 
of FIG. 4 can be obtained if comparison is made between a value obtained 
by adding the voltage 2R.sub.a I.sub.d corresponding to a resistance drop 
to the speed voltage E.sub..omega. and the detected value E.sub.d of DC 
voltage, as shown in FIG. 6. 
Instead of the above described embodiment where the compensation field 
current is corrected, an embodiment where the main field current is 
corrected will be described in the following. FIG. 7 is a block diagram 
showing the second embodiment of the present invention. In the following, 
differences from the block diagram in FIG. 4 will be mainly described. The 
DC voltage controller 29' in the correction circuit 800 amplifies a 
deviation obtained by subtracting the voltage (E.sub.d -2R.sub.a I.sub.d) 
outputted from the subtractor 28 from the speed voltage E.sub..omega. 
outputted from the coefficient multiplier 25 and provides a correction 
signal .DELTA.I.sub.f of the main field. The excitation circuit 700' 
comprises a main excitation circuit 710' and a compensation excitation 
circuit 720'. The adder 22 in the main excitation circuit 710' performs 
addition of the instructed value I.sub.fo of main field current outputted 
from the field instructing device 12 and the correction signal 
.DELTA.I.sub.f outputted from the DC voltage controller 29' so as to 
provide a new instructed value I.sub.fp. The DC voltage controller 29' 
compares the voltage (E.sub.d -2R.sub.a I.sub.d) from the subtractor 28 
with the speed voltage E.sub..omega. from the coefficient multiplier 25 
and if the voltage (E.sub.d -2R.sub.a I.sub.d) becomes smaller than 
E.sub..omega., the controller 29' provides a correction signal 
.DELTA.I.sub.f in a direction for increasing the main field current and if 
the voltage (E.sub.d -2R.sub.a I.sub.d) becomes larger than E.sub..omega., 
reversely, it provides a correction signal .DELTA.I.sub.f in a direction 
for decreasing the main field current. As a result, the DC voltage E.sub.d 
changes and control is made so that the output of the subtractor 28 may be 
always equal to the output of the coefficient multiplier 25. This 
operation will be further described with reference to the vector diagrams. 
FIG. 8A is a vector diagram for explaining the operation in the FIG. 7 
embodiment. FIG. 8B is a vector diagram showing a process where the main 
field current is regulated. These figures correspond respectively to FIGS. 
4A and 4B. Referring to FIG. 8A, a line OP indicates a reference phase 
.gamma. of an armature current vector I.sub.a determined by a position 
signal of the position sensor 4. In reality, due to an overlapping angle 
of commutation, current I.sub.a flows in a phase .gamma.' having a small 
delay from the reference phase .gamma.. As a result, a phase difference 
between the induced voltage V and the armature current I.sub.a becomes 
small and the power factor is improved. Accordingly, the generated torque 
becomes larger than the instructed torque. Also in this case, the armature 
resistance R.sub.a is disregarded for the purpose of facilitating the 
explanation, and if an angle formed by the induced voltage V and the 
armature current I.sub.a is assumed to be 8, the above described equation 
(3) is established. Accordingly, based on the above described equation 
(2), the amount of generated torque changes in proportion to the I.sub.a 
direction component of the induced voltage V. 
Referring to FIG. 8B, it is assumed that the DC voltage for generating a 
desired torque is E.sub.d, the induced voltage at this time is V and the 
main field current is I.sub.f. Assuming that the main field current 
instructed from the field instructing device 12 is I.sub.f1, which is 
smaller than I.sub.f, the induced voltage becomes V.sub.1 since the 
voltage X.sub.s I.sub.f1 becomes smaller than the voltage X.sub.s I.sub.f. 
As a result, the DC voltage is decreased to E.sub.d1 which is an I.sub.a 
direction component of the voltage V.sub.1. Although in reality, the 
vector direction of the armature current I.sub.a changes slightly since an 
overlapping angle of commutation changes slightly, such change will be 
disregarded in the explanation for the purpose of facilitating it. In 
consequence, the output of the subtractor 28 becomes smaller than the 
output of the coefficient multiplier 25 and the DC voltage controller 29' 
provides a positive output +.DELTA.I.sub.f. This positive output is added 
to the instructed value I.sub.fo by the adder 22, so that the main field 
current is regulated to the correct value I.sub.f. Reversely, if the main 
field current instructed from the field instructing device 12 is I.sub.f2 
which is larger than I.sub.f, the induced voltage becomes V.sub.2. As a 
result, the DC voltage is increased to E.sub.d2. Accordingly, the output 
of the subtractor 28 becomes larger than the output of the coefficient 
multiplier 25 and the DC voltage controller 29' provides a negative output 
-.DELTA.I.sub.f. This negative output is added to the instructed value 
I.sub.fo by the adder 22 and as a result, the main field current I.sub.f 
is regulated to the correct value also in this case. 
Also in this embodiment, since the voltage (E.sub.d -2R.sub.a I.sub.d) 
which is an output of the subtractor 28 is controlled so as to be always 
equal to the speed voltage E.sub..omega. which is an output of the 
coefficient multiplier 25, the torque T is proportional to the direct 
current I.sub.d, as is clearly understood from the above described 
equation (2). More specifically, even if the vector of the armature 
current changes under the influence of an overlapping angle of 
commutation, the generated torque becomes proportional to the instructed 
torque and thus, the precision of torque control is improved. In addition, 
since the DC voltage E.sub.d is maintained to a predetermined value, there 
is an advantage that the rated voltage of the converter 1 and the inverter 
2 can be made small. 
Although in the FIG. 7 embodiment, a value obtained by subtracting, from 
the detected value E.sub.d of the DC voltage, the voltage 2R.sub.a I.sub.d 
corresponding to a resistance drop is compared with the speed voltage 
E.sub..omega., it is the same with the comparison between a value obtained 
by adding the voltage 2R.sub.a I.sub.d to the speed voltage E.sub..omega. 
and the DC voltage E.sub.d, as shown in FIG. 9. 
Although in the foregoing description, an instructed value of torque 
outputted from the speed controller 8 was used for the purpose of 
obtaining the voltage corresponding to a resistance drop, the same control 
can be made if an instructed value of current outputted from the 
coefficient multiplier 20 or a detected value of current outputted from 
the current detector 9 is used instead of the instructed value of torque. 
In case where a high precision of control is not required, it is not 
needed to apply the voltage 2R.sub.a I.sub.d corresponding to a resistance 
drop. In addition, in order to make field-weakening control in a high 
speed region, the output from the coefficient multiplier 25 is made to be 
proportional to the rotational speed of the motor 3 in a region not 
attaining the field-weakening control region and is made constant in the 
field-weakening region. 
Although the present invention has been described and illustrated in 
detail, it is clearly understood that the same is by way of illustration 
and example only and is not to be taken by way of limitation, the spirit 
and scope of the present invention being limited only by the terms of the 
appended claims.