Dual transformer device for power converters

A dual transformer device (34) for use in a power converter (30) is provided. The dual transformer device is preferably implemented as an integrated magnetics device suitable for use in high density power converters. Two transformers, T1 and T2, have series connected primary windings (300 and 302), and secondary windings (304 and 306) connected at a common node (312). T1 and T2 have magnetizing inductances, L1 and L2, that independently store energy responsive to an AC voltage, V.sub.p. Diodes, D1 and D2, operate to rectify substantially out-of-phase secondary voltages, V.sub.S1 and V.sub.S2 and produce a rectified voltage V.sub.R. D1 and D2 cause L1 and L2 to store discharge energy such that the magnitude of an output current, I5, is always equal to the sum of magnetizing currents, I.sub.L1 and I.sub.L2. An output filter (38) reduces current ripple in I5 and voltage ripple in V.sub.R. Another inductance, L3, supplies energy to an AC voltage supply (32) to reduce switching losses.

FIELD OF THE INVENTION 
This invention relates to power converters, and more particularly, to high 
density power converters. 
BACKGROUND OF THE INVENTION 
Along with the ongoing push to increase the density of electronic and 
integrated circuits, there has developed a need for higher density power 
converters that supply power to these circuits. That is, there has 
developed a need to either reduce the physical size of a power converter 
and keep its power output the same or keep the physical size of the power 
converter the same and increase its power output or both (i.e., reduce the 
physical size and increase the power output). By accomplishing one of the 
above, the power density of the power converter, i.e., electrical power 
output (watts) per cubic inch of volume taken up by the power converter, 
increases. Most power converters presently have power densities in the 
range of 1-3 watts/cubic inch. Next generation power converters, which are 
presently under development, may have densities as much as an order of 
magnitude larger than present power converter densities (e.g., 30 
watts/cubic inch). Long term projections, however, indicate that future 
power converters will be required to have much higher power densities, 
perhaps as high as several hundred watts/cubic inch. 
In order to increase the power density of present power converters to high 
enough levels to satisfy future needs, present power converters must be 
made smaller and more efficient. One way to reduce the size of a power 
converter is to reduce the size and/or number of components in the power 
converter. Similarly, the efficiency of a power converter can be increased 
by increasing the efficiency of the components and/or by reducing the 
number of components in the power converter. 
As is readily apparent to a person skilled in the power conversion art, 
power converters typically include one or more magnetic components. An 
output series inductor is typically used as an energy storage device to 
filter the power converter output. In addition, the power converter may be 
coupled to a voltage supply by a transformer. The transformer typically 
performs two functions: transforming the magnitude of the supply voltage 
to a level suitable for the circuits fed by the power converter; and, 
isolation of those circuits from the voltage supply. 
An example of a prior art power converter having these magnetic components 
is illustrated in FIG. 1. FIG. 1 is a simplified schematic diagram of a 
transformer-coupled push-pull converter 10. Push-pull converters of the 
sort illustrated in FIG. 1 are well known in the art and, hence, are not 
discussed in detail herein. However, to better understand the present 
invention, a few key components of the push-pull converter 10 illustrated 
in FIG. 1 and their function are briefly discussed. The push-pull 
converter 10 is coupled to an AC voltage supply 22 by a transformer 12 and 
comprises a rectifier 14 and an output filter 16. The rectifier 14 
consists of two diodes 17 and 18 connected in a conventional manner to 
provide a rectified output voltage. The output filter 16 consists of an 
output inductor 19 and a capacitor 20. A rectified and filtered output 
voltage is produced by the push-pull converter 10 and applied to a load 
24. Thus, as can be seen from FIG. 1, and from the above discussion, a 
conventional, prior art push-pull converter 10 comprises at least two 
magnetic devices (viz., the transformer 12 and the output inductor 19). 
One problem associated with power converters in the prior art is that the 
transformers and inductors usually have a conventional, wound-type of 
construction, which makes them relatively bulky. As a result, their size 
limits efforts to reduce the size of the associated power converters. Yet 
another problem associated with conventional transformers and inductors 
used in prior art power converters is that their construction is very 
labor intensive. As a result, they are expensive to construct and not well 
suited to high volume production methods, which increases the cost of 
prior art power converters. 
An important characteristic of power converters, such as the prior art 
pushpull converter 10, illustrated in FIG. 1, is that they be 
controllable. For example, in many instances the output of the power 
converter must be a regulated, i.e., controlled, output. Typically, the 
output voltage of a power converter is controlled by controlling the 
average value of the input voltage applied to the power converter (i.e., 
the voltage produced by the AC voltage supply 22 in FIG. 1). One method of 
controlling the average value of the input voltage is through pulse width 
modulation of the input voltage. Pulse width modulation of the input 
voltages is a relatively straightforward method of control that has been 
widely accepted in the prior art. 
Another important characteristic of power converters concerns the nature of 
their output inductor current. The output inductor current of a power 
converter may be of a continuous or discontinuous nature. For example, if 
the magnitude of the output inductor current of the power converter is 
always positive or always negative but never becomes zero, it is a 
continuous current. If the magnitude of the output inductor current does 
become zero between successive positive or negative values, than it is a 
discontinuous current. A continuous output inductor current is preferable 
over a discontinuous inductor current in many power converter 
applications. One advantage of a continuous current is that it is easier 
to filter than a discontinuous current. Accordingly, a smaller and simpler 
filter may be used, thus, reducing the size of at least one component in a 
power converter. The prior art push-pull converter 10, depicted in FIG. 1, 
produces a continuous output inductor current. As is well known in the 
prior art, the energy stored in the inductor 19 is discharged, such that a 
continuous output inductor current is produced. 
As can be readily appreciated from the foregoing discussion, there is a 
need for power converters that have very high power densities. The high 
density power converters should be capable of being manufactured using 
high volume production techniques and, thus, inexpensive to produce. 
Furthermore, the high density power converter should be easily 
controllable and should, preferably, provide a continuous current output. 
This invention is directed to a dual transformer device that may be used 
in power converters to achieve these results. 
SUMMARY OF THE INVENTION 
In accordance with the present invention, a transformer device for use in a 
power converter is provided. The power converter comprises: the 
transformer device; and, a switching device. The transformer device is 
responsive to an AC voltage and is configured to function as both a 
transformer and as an energy storage device, the transformer device when 
functioning as a transformer acting to produce a plurality of 
substantially out-of-phase transformed voltages from the AC voltage and 
the transformer device when functioning as an energy device acting to 
store energy responsive to the AC voltage. The switching device rectifies 
the plurality of substantially out-of-phase transformed voltages and 
produces a rectified voltage. The switching device also controls the 
transformer device so that the transformer device discharges the energy 
stored therein through the switching device. 
In accordance with further aspects of the present invention, the 
transformer device comprises a first transformer having a first 
magnetizing inductance and a second transformer having a second 
magnetizing inductance. When the transformer device functions as a 
transformer, the first transformer produces a first one of the plurality 
of substantially out-of-phase transformed voltages and the second 
transformer produces a second one of the plurality of substantially 
out-of-phase transformed voltages. When the transformer device functions 
as an energy storage device the first and second magnetizing inductances 
store the energy responsive to the AC voltage. Energy stored in the first 
magnetizing inductance is independent of energy stored in the second 
magnetizing inductance. 
In accordance with further aspects of the present invention, the primary 
windings of the first and second transformers are connected in series and 
the secondary windings of the first and second transformers are connected 
at a common point. 
In accordance with still further aspects of the present invention, the 
transformer device comprises at least one flat primary winding formed by 
folding a flat sheet along perpendicular fold lines, at least one flat 
secondary winding located adjacent to the at least one primary winding, 
and a flat core that substantially encloses the at least one primary 
winding and the at least one secondary winding. 
In accordance with alternative aspects of the present invention, the 
transformer device is configured to function as a second energy storage 
device that stores energy responsive to the AC voltage and discharges the 
energy stored therein to compensate for switching losses in an AC voltage 
supply providing the AC voltage.

DESCRIPTION OF THE PREFERRED EMBODIMENT 
FIG. 2 is a block diagram of a power converter 30 formed in accordance with 
the present invention. The power converter 30 comprises a dual transformer 
device 34, a rectifier 36, and an output filter 38. An AC voltage supply 
32 produces an AC supply voltage, V.sub.p. The V.sub.p voltage is applied 
to the dual transformer device 34, via line 402. The dual transformer 
device 34 includes two transformers that transform the V.sub.p voltage by 
conventional transformer action and produce voltages, designated V.sub.S1 
and V.sub.S2, on lines 404A and 404B, respectively. The rectifier 36 
rectifies the V.sub.S1 and V.sub.S2 voltages and produces a rectified 
voltage, designated V.sub.R, on line 406. A continuous current, designated 
I.sub.5, is also produced on line 406. The output filter 38 is optional, 
and if used, substantially filters out any ripple in the rectified 
voltage, V.sub.R, and the continuous current, I.sub.5. A substantially 
ripple free DC output voltage, designated V.sub.OUT, and DC output 
current, designated I.sub.OUT, are produced on line 408. 
The power converter 30 is easily controlled by conventional methods that 
are well known in the prior art. More specifically, the V.sub.OUT voltage 
may be controlled by controlling the average value of V.sub.p. As will 
become better understood from the following discussion, in one actual 
embodiment of the present invention, a conventional, pulse-width 
modulation technique is used to control the average value of V.sub.p and, 
hence, control the output voltage, V.sub.OUT. 
In addition to transforming the V.sub.p voltage (i.e., to V.sub.S1 and 
V.sub.S2), the two transformers in the dual transformer device 34 act as 
energy storage devices. At appropriate times, the rectifier 36 permits the 
energy stored in the two transformers to be discharged through the 
rectifier 36. The energy is discharged in such a way that a continuous 
output current (i.e., I.sub.5) is produced on line 406. As will become 
better understood from the following discussion, the energy if stored in 
the magnetizing inductance of the transformers and, further, that the 
energy stored in one inductor is independent of the energy stored in the 
other inductor. As a result, the independent energies may be discharged at 
different times. Thus, the use of the independently stored energies in the 
magnetizing inductances of the transformers to produce a current, such as 
a continuous current, is a significant aspect of the present invention. 
Furthermore, it is that aspect of the present invention (i.e., the use of 
stored energy in the magnetizing inductances) that permits the power 
converter 30 to function without an output inductor, such as the output 
inductor 19 in the prior art push-pull converter 10 discussed above and 
illustrated in FIG. 1. As will become better understood from the following 
discussion, the elimination of the output inductor, and the use of two 
transformers permits the dual transformer device 34, in one preferred 
embodiment, to be formed as an integrated magnetics device using 
integrated magnetics technology. Preferably, two transformers configured 
as an integrated magnetics device share a common core. The use of a common 
core, as opposed to two separate cores, results in a smaller structure. 
In addition to implementing the dual transformer device 34 in an integrated 
magnetics configuration, the use of compact winding and core designs 
permits a further reduction in size of the dual transformer device 34. As 
will become better understood from the following discussion, a dual 
transformer device 34 constructed in this manner may form part of a very 
high density power converter. 
Turning next to a discussion of two transformers configured in accordance 
with the present invention, FIG. 3 illustrates a simplified schematic 
diagram of the dual transformer device 34. The dual transformer device 34 
depicted in FIG. 3 comprises two transformers, designated T1 and T2. T1 is 
a two winding transformer having a primary winding 300, a secondary 
winding 304, and a magnetic core 308. The transformer T1 has a magnetizing 
inductance, designated L1, which is shown in FIG. 3 as a discrete inductor 
connected in parallel with the secondary winding 304. Similarly, T2 is a 
two winding transformer having a primary winding 302, a secondary winding 
306, and a magnetic core 310. The T2 transformer has a magnetizing 
inductance, designated L2, which is represented as a discrete inductor 
connected in parallel with the secondary winding 306. 
In accordance with the preferred embodiment of the present invention, and 
as will be described next, the transformers, T1 and T2, have their primary 
windings 300 and 302 connected in series and their secondary windings 304 
and 306 connected at a common point. Further, the turns ratio of the 
transformers, T1 and T2, are equal. The source voltage, V.sub.p, is 
applied to one end of the primary winding 300 via line 402. The other end 
of the primary winding 300 is series connected to one end of the primary 
winding 302. The other end of the primary winding 302 is connected to a 
return line 403. One side of the secondary winding 304 is connected to 
line 404A. The other side of the secondary winding 304 is connected to one 
end of the secondary winding 306 at a node 312. The other end of the 
secondary winding 306 is connected to line 404B. As noted above, the 
secondary voltages, V.sub.S1 and V.sub.S2, are formed on lines 404A and 
404B with respect to node 312. Accordingly, V.sub.S1 is the secondary 
voltage of T1 and V.sub.S2 is the secondary voltage of T2. Further, the 
V.sub.S1 and V.sub.S2 voltages are substantially out-of-phase with one 
another. The energy storage function of T1 and T2, noted briefly above, is 
discussed more fully below. More specifically, the charging and 
discharging of energy in the magnetizing inductances, L1 and L2, is 
discussed in conjunction with a description of the operation of the power 
converter 30, which is presented next. 
FIG. 4 is a schematic diagram of the power converter 30 illustrated in FIG. 
2 and discussed briefly above. The dual transformer device 34 is 
configured as discussed above and illustrated in FIG. 3. The rectifier 36 
comprises two voltage controlled switches, such as two diodes, designated 
D1 and D2. The output filter 38 comprises a capacitor, designated C1. As 
will be discussed more fully below, and in accordance with an actual 
embodiment of the present invention, the output filter 38 comprises a low 
profile capacitor assembly consisting of a plurality of capacitors. As 
noted above, the output filter 38 (i.e. C1) is optional, and, if used, 
reduces the amount of ripple in the rectified voltage, V.sub.R, and the 
continuous current, I.sub.5. Accordingly, the output filter 38 may be 
omitted when ripple in the output voltage, V.sub.OUT, and the output 
current, I.sub.OUT, is of little or no concern. Hence, the output filter 
38 is not required for the operation of the power converter 30, depicted 
in FIG. 4. 
An AC voltage supply 32 is connected to the primary windings 300 and 302 
vial lines 402 and 403. The AC voltage supply 32 produces the source 
voltage, V.sub.p, on line 402. The anode of D1 is connected to the 
secondary winding 304 of transformer T1, via line 404A. The anode of D2 is 
connected to the secondary winding 306 of transformer T2, via line 404B. 
The cathode of D1 is connected to the cathode of D2 and to one side of C1, 
via line 406. The other side of C1 is connected to the common node 312 
between the secondary windings 304 and 306, via line 404C. A load 40 is 
connected across C1, via lines 408 and 404C. The output voltage, V.sub.OUT 
is produced on line 408. 
In accordance with the preferred embodiment of the invention, the AC 
voltage source 32 is a switching circuit (not shown) that converts a DC 
input voltage (also not shown) into the V.sub.p voltage on line 402. In 
one actual embodiment of the present invention, the AC voltage source 32 
is a full bridge circuit that produces a pulse width modulated V.sub.p 
voltage having a shape depicted by the waveform illustrated in FIG. 5A. As 
can be seen from FIG. 5A, the V.sub.p voltage consists of alternating 
positive V.sub.p pulses 80 and negative V.sub.p pulses 82. The leading 
edge of a positive V.sub.p pulse 80 occurs at time t.sub.1 and the 
trailing edge of a V.sub.p pulse 80 occurs at time t.sub.2. Similarly, the 
leading edge of a negative V.sub.p pulse 82 occurs at time t.sub.3 and the 
trailing edge of a negative V.sub.p pulse 82 occurs at time t.sub.4. As 
can be seen from FIG. 5A, the magnitude of the V.sub.p voltage is zero 
between adjacent positive and negative pulses 80 and 82. 
The power converter 30, represented by the circuit in FIG. 4 operates in 
the following manner. For purposes of clarity, the operation of the 
circuit in FIG. 4 is discussed in its steady state. That is, after an 
initial period, C1 has charged-up to the V.sub.OUT voltage, currents in 
the circuit have reached their steady state, and energy is stored in the 
magnetizing inductance, L1 and L2. During the time period between times 
t.sub.1 and t.sub.2 a positive V.sub.p pulse 80 is produced on line 402, 
thereby causing V.sub.S1, on line 404A, to be positive and V.sub.S2, on 
line 404B, to be negative. The positive V.sub.S1 voltage forward biases 
D1, making D1 conductive, such that the V.sub.S1 voltage is clamped to the 
V.sub.OUT voltage (i.e., V.sub.S1 =V.sub.OUT). Contrariwise, the negative 
V.sub.S2 voltage causes D2 to be reverse biased, such that D2 is 
nonconductive and, hence, V.sub.S2 is not clamped to V.sub.OUT. 
As noted above, the circuit in FIG. 4 is in a steady state condition, such 
that energy is already stored in the magnetizing inductors, L1 and L2. As 
a result, currents are flowing in L1 and L2. More specifically, a 
magnetizing current, designated I.sub.L1, is flowing in L1 and a 
magnetizing current, designated I.sub.L2, is flowing in L2. As a result, 
energy is being stored in L1 and L2. As noted above, the energy stored in 
one inductance (e.g., L1) is independent of the energy stored in the other 
inductance (e.g., L2) and, thus, the L1 and L2 energies may be charged and 
discharged independently (i.e., at different times). The independent 
nature of the L1 and L2 inductances is a result of substantially 
independent magnetic paths for each of the magnetizing currents, I.sub.L1 
and I.sub.L2. As will become better understood from the following 
discussion, during the time between t.sub.1 and t.sub.2 (i.e., when D1 is 
conductive and D2 is nonconductive), the energy in L1 is decreasing and 
the energy in L2 is increasing. In other words, the magnitude of I.sub.L1 
is decreasing and the magnitude of I.sub.L2 is increasing. This behaviour 
of I.sub.L1 and I.sub.L2 is illustrated in FIGS. 5D and 5E, between times 
t.sub.1 and t.sub.2. 
As noted above, during the period of time between t.sub.1 and t.sub.2, D2 
is nonconductive. Accordingly, I.sub.L2 causes a current, designated 
I.sub.2, to circulate in the secondary winding 306 of transformer T2. The 
magnitudes of I.sub.L2 and I.sub.2 are equal (i.e., I.sub.L2 =I.sub.2). 
During this time, I.sub.L2 and I.sub.L1 are flowing in a direction 
indicated by the arrows in FIG. 4. I.sub.2 is reflected into the primary 
winding 302 of transformer T2 and causes a current, designated I.sub.2 ' 
to flow in the primary winding 302. As is well known in the art, the 
magnitude of I.sub.2 ' is related to the magnitude of I.sub.2 by the turns 
ratio of the secondary winding 306 and the primary winding 302 of 
transformer T2. 
Since the primary windings 300 and 302 are series connected, I.sub.2 ' 
flows through the primary winding 300 of transformer T1. The flow of 
I.sub.2 ' through primary winding 300 causes a current, designated 
I.sub.1, to flow in the secondary winding 304 of transformer T1. As is 
well known in the art, the magnitude of I.sub.1 is related to the 
magnitude of I.sub.2 ' by the turns ratio of the primary winding 300 and 
the secondary winding 304 of transformer T1. Since, as noted above, the 
turns ratio of T1 and T2 are equal, the transformation of current from the 
secondary to the primary of T2 and the subsequent transformation of the 
current from the primary to the secondary of T1 results in I.sub.1 having 
a magnitude equal to the magnitude of I.sub.2. Further, as noted above, 
the magnitudes of I.sub.2 and I.sub.L2 are equal and, thus, the magnitudes 
of I.sub.1 and I.sub.L2 are equal (i.e., I.sub.1 =I.sub.L2). 
The currents, I.sub.1 and I.sub.L1, sum at node 314, such that a current, 
designated I.sub.3, is equal to the sum of I.sub.1 and I.sub.L1 (i.e., 
I.sub.3 =I.sub.1 +I.sub.L1). Since I.sub.1 =I.sub.L2, I.sub.3 is equal to 
the sum of the magnetizing currents in L1 and L2 (i.e., I.sub.3 =I.sub.L1 
+I.sub.L2). Because D1 is conductive and D2 is nonconductive, the current 
flowing into node 318 is equal to I3. An output current, designated 
I.sub.5, flows out of node 318 and is equal to I3 and, hence, equal to the 
sum of the magnetizing currents (i.e., I.sub.5 =I.sub.L1 +I.sub.L2). 
During this time, I.sub.5 is increasing, as can be seen in the waveform 
depicted in FIG. 5F. 
At time t.sub.2 the V.sub.p pulse goes to zero. As a result, no current is 
induced in the secondary windings 304 and 306 (i.e., I.sub.1 =I.sub.2 =0). 
During the time period between times t.sub.2 and t.sub.3 (i.e., when 
V.sub.p =0), the energy stored in the magnetizing inductance, L1, holds 
the V.sub.S1 voltage positive. More specifically, when V.sub.p goes to 
zero, magnetizing current flowing through L1 (i.e., I.sub.L1) continues to 
flow since, as is well known, the current flowing in an inductor cannot 
change instantaneously. As a result, I.sub.L1 flows through D1 and forward 
biases D1, such that the V.sub.S1 voltage is clamped to the V.sub.OUT 
voltage, and D1 remains conductive. During this time, the magnitude of 
I.sub.3 is equal to the magnitude of I.sub.L1 (i.e., I.sub.3 =I.sub.1 
+I.sub.L1, but as noted above, I.sub.1 =0). 
At the same time, the energy stored in the magnetizing inductance, L2, 
causes the V.sub.S2 voltage on line 404B to become positive. More 
specifically, the current flowing through L2 (i.e., I.sub.L2) continues to 
flow and, further, flows through D2. I.sub.L2 forward biases D2, such that 
V.sub.S2 is clamped to V.sub.OUT. Current flowing through D2, designated 
I.sub.4, is equal to the magnitude of I.sub.2 and I.sub.L2, which sum at 
node 316. During this time the magnitude of I.sub.4 equal the magnitude of 
I.sub.L2 (since, as noted above, I.sub.2 =0). Thus, during the period of 
time between t.sub.2 and t.sub.3, I.sub.5 is equal to the sum of I.sub.3 
and I.sub.4 and, hence, equal to the sum of the magnetizing currents in L1 
and L2 (i.e., I.sub.5 =I.sub.L1 +I.sub.L2). During the time between t2 and 
t3, stored energy in L1 and L2 is being discharged, such that I.sub.L1 and 
I.sub.L2 are decreasing. As a result, I.sub.5 is decreasing. This 
behaviour of I.sub.L1 and I.sub.L2 is depicted in the waveforms 
illustrated in FIGS. 5D and 5E. The resulting I.sub.5 current is depicted 
in FIG. 5F. 
During the time period between times t.sub.3 and t.sub.4, the circuit 
depicted in FIG. 4 operates in a manner that is the reverse of the 
operation described above for the time period between times t.sub.1 and 
t.sub.2. More specifically, during the time period between t.sub.3 and 
t.sub.4, a negative V.sub.p pulse 82 is produced on line 402, causing 
V.sub.S1 voltage on line 404A to be negative and the V.sub.S2 voltage on 
line 404B to be positive. The positive V.sub.S2 voltage forward biases D2, 
making D2 conductive, such that V.sub.S1 is clamped to V.sub.OUT. 
Contrariwise, the negative V.sub.S1 voltage causes D2 to be reverse 
biased, such that D1 is nonconductive and, hence, V.sub.S1 is not clamped 
to V.sub.OUT. 
As will become better understood from the following discussion, during the 
time period between t.sub.3 and t.sub.4,the stored energy in the 
magnetizing inductance, L1, is increasing and the energy in the 
magnetizing inductance, L2, is decreasing. Accordingly, during this time 
I.sub.L1 is increasing and I.sub.L2 is decreasing. This behaviour of 
I.sub.L1, and I.sub.L2 is illustrated in the waveform depicted in FIG. 5D 
and 5E, between times t.sub.3 and t.sub.4. 
As noted above, during this time D1 is nonconductive. Accordingly, I.sub.L1 
causes a current, I.sub.1, to circulate in the secondary winding 304 of 
transformer T1. The magnitude of I.sub.1 is equal to the magnitude of 
I.sub.L1. I.sub.1 is reflected into the primary winding 300 of transformer 
T1 and causes a current, designated I.sub.1 ', to flow in the primary 
winding 300. The magnitude of I.sub.1 ', is related to the magnitude of 
I.sub.1 by the turns ratio of the secondary and primary windings 304 and 
300 of T1. Since the primary windings 300 and 302 are series connected, 
I.sub.L1 ' flows through the primary winding 302 of T2 and causes a 
current, I.sub.2, to flow in the secondary winding 306 of T2. The 
magnitude of I.sub.2 is related to the magnitude of I.sub.1 ' by the turns 
ratio of the primary and secondary windings 302 and 306 of T2. Thus, as 
noted above, since the turns ratios of T1 and T2 are the same, the 
magnitude of I.sub.2 is equal to the magnitude of I.sub.L1. 
The two currents, I.sub.2 and I.sub.L2 , are summed at node 316 to form the 
current, I.sub.4 (i.e., I.sub.4 =I.sub.2 +I.sub.L2). Since, as noted 
above, I.sub.2 =I.sub.L1 , I.sub.4 is equal to the magnetizing currents in 
L1 and L2 (i.e., I.sub.4 - I.sub.L1 +I.sub.L2). The output current, 
I.sub.5, is formed by the sum of I.sub.3 and I.sub.4 at note 318. During 
this time, I.sub.3 is zero, since, as noted above, D1 is nonconductive. 
Thus, I.sub.5 is equal to I.sub.4 and , hence, equal to the magnetizing 
currents in L1 and L2 (i.e., I.sub.5 =I.sub.L1 +I.sub.L2). During this 
time I.sub.5 is increasing, as can be seen in the waveform depicted in 
FIG. 5F. 
At time t.sub.4, the V.sub.p voltage goes to zero (FIG. 5A). As a result, 
no current is induced in the secondary windings 304 and 306 (i.e., I.sub.1 
=I.sub.2 =0). During the time period between t.sub.4 and the next positive 
V.sub.p pulse 80 (i.e., at time t.sub.1 ) the energy stored in the 
magnetizing inductance, L2, holds the V.sub.S2 voltage positive. More 
specifically, when V.sub.p goes to zero, the current flowing in L2 (i.e., 
I.sub.L2) continues to flow. Further, I.sub.L2 flows through D2, thus, 
forward biasing D2 and making D2 conductive such that V.sub.S2 is clamped 
to V.sub.OUT. The current flowing through D2 (i.e., I.sub.4) is equal to 
the magnetizing current in L2 (i.e., I.sub.4 =I.sub.2 +I.sub.L2), where 
I.sub.2 =0). At the same time, the energy stored in the magnetizing 
inductance, L1, causes the V.sub.S1 voltage on line 404A to become 
positive. More specifically, the current flowing through L1 (i.e., 
I.sub.L1) continues to flow. Further, I.sub.L1 flows through D1, thus, 
forward biasing D1 and making D1 conductive, such that V.sub.S1 is clamped 
to V.sub.OUT. The current flowing through D1 (i.e., I.sub.3) is equal to 
the magnetizing current in L1 (i.e., I.sub.3 =I.sub.1 +I.sub.L1, where 
I.sub.1 =0). During this time (i.e., between t.sub.4 and t.sub.1) I.sub.5 
is equal to the sum of I.sub.3 and I.sub.4 and, thus, is equal to the 
magnetizing currents in L1 and L2 (I.sub.5 =I.sub.L1 +I.sub.L2). During 
this time, the energy in L1 and L2 is being discharged, such that the 
magnitude of I.sub.L1 and I.sub.L2 are decreasing. Hence, the magnitude of 
I5 is also decreasing. This behavior of I.sub.L1, I.sub.L2 and I.sub.5 is 
depicted in the waveforms illustrated in FIGS. 5D, 5E, and 5F, 
respectively. 
Thus, as is readily apparent from the above discussion and from FIG. 5F, 
the power converter 30, formed in accordance with the present invention, 
provides a continuous current output (i.e., I.sub.5) on line 406. 
Accordingly, the presently preferred embodiment of the power converter 30 
operates in a continuous inductor current (i.e., I.sub.5) mode. However, 
it is to be understood that a power converter formed in accordance with 
the present invention also functions in a discontinuous inductor current 
mode and, further, functions in a discontinuous inductor current mode in 
the same manner as the prior art. 
In an alternative embodiment of the present invention, a third inductance 
designated L3, is added to the power converter 30, illustrated in FIG. 4, 
to reduce switching losses in the AC voltage supply. As is well known in 
the art, an AC voltage supply that comprises a switching circuit, such as 
the full bridge circuit discussed above has inherent switching losses. The 
switching losses are a result, in part, of inherent capacitances 
associated with the switch in the switching circuit. For example, in the 
full bridge circuit, each of the four switches has an inherent value of 
capacitance. This capacitance is charged and discharged each time the 
associated switch opens and closes. Normally, the energy is discharged in 
the form of heat and, hence, becomes lost energy. For high speed switching 
circuits, such losses become significant and, thus, reduce the efficiency 
of the switching circuit. 
The inductance, L3, depicted in FIG. 4, is located in parallel with the 
primary windings 300 and 302. That is, one end of L3 is connected to line 
402 and the other end of L3 is connected to line 403. After the circuit 
illustrated in FIG. 4 has reached a steady state, energy is stored in L3 
such that inductance current, I.sub.L3, flows in the inductance L3. The 
waveform of I.sub.L3 is depicted in FIG. 5G, and represents the behavior 
of I.sub.L3 for a V.sub.p waveform of the sort discussed above and 
illustrated in FIG. 5A. As can be seen from FIG. 5G, the magnitude of 
I.sub.L3 increases during the time period between times t.sub.1 and 
t.sub.2. Contrariwise, I.sub.L3 decreases during the time period between 
times t.sub.3 and t.sub.4. Thus, as FIG. 5A illustrates, energy is stored 
in L3. The energy stored in L3 is available (i.e., in the form of 
inductance current, I.sub.L3) to charge the capacitance associated with 
the switches in the switching circuit forming the AC voltage supply 32. 
The value of L3 can be optimized, such that an appropriate amount of 
energy is stored in L3 to reduce the switching losses of the AC voltage 
supply 32. 
FIG. 6 is an exploded view of one preferred embodiment of the power 
converter 30 illustrated in FIG. 4. More specifically, FIG. 6 is an 
exploded view of an assembly comprising the dual transformer device 34, 
the rectifier 36, and the output filter 38. As will become better 
understood from the following discussion, the presently preferred 
embodiment of the dual transformer device 34 is implemented as an 
integrated magnetics device in which the two transformers, T1 and T2 share 
a common core. As will also become better understood from the following 
discussion, the series connected primary windings 300 and 302 (FIG. 4) are 
formed by a single primary winding on a common (i.e., single) primary core 
leg. 
Beginning at the top of FIG. 6 and proceeding downward, the power converter 
30 comprises: a core top 42; gap paper 44; a top flex strip 46; a 
secondary conductive sheet 50; a bottom flex strip 54; two diodes 58A and 
58b; a capacitor assembly 60; a core bottom 62; and, a bottom conductive 
sheet 65. As will become better understood from the following discussion, 
the integrated magnetics device forming the dual transformer 34 comprises: 
the core top and bottom 42 and 62; the gap paper 44; the top and bottom 
flex strips 46 and 54; and, the secondary and bottom conductive sheets 50 
and 65. Likewise the rectifier 36 is formed by the diodes 58A and 58B and 
the output filter 38 is formed by the capacitor assembly 60. 
The various components depicted in FIG. 6 and introduced above are 
discussed below in greater detail, with a discussion of the components of 
the integrated magnetics device forming the dual transformer device 34 
discussed first. The core top 42 and core bottom 62 are made of a suitable 
magnetic material. The core top 42 is planar. The core bottom 62 has a 
center leg 86 and two parallel outer legs 84A and 84B. As will become 
better understood from the following discussion, the center leg 86 forms a 
single primary core leg shared by the series connected primary windings of 
two transformers (e.g., windings 300 and 302 in FIG. 4). The outer legs 
84A and 84B form secondary core legs for the secondary windings of two 
transformers (e.g, windings 304 and 306 in FIG. 4). Thus, the two 
transformers have different secondary core legs that provide independent 
magnetic paths for the magnetizing currents (I.sub.L1 and I.sub.L2). As 
discussed above, these independent magnetic paths permit the independent 
storage of energy in the two magnetizing inductances (L1 and L2). The core 
top 42 and the core bottom 62 combine to form a substantially flat core 
assembly 40, which is best illustrated in FIG. 7. 
The top flex strip 46 is formed from flexible, nonconductive material and 
has a central aperture 49. A flat conductor 48 is formed on the surface of 
the flex strip 46 and forms a spiral winding about the aperture 49. The 
flex strip 46 and conductor 48 are covered by an insulating layer (not 
shown in FIG. 6). Similarly, the bottom flex strip 54 is formed from 
flexible, nonconductive material and has a central aperture 57. A flat 
conductor 56 is located on the surface of the bottom flex strip 54 (as 
indicated by the dashed lines in FIG. 6), and forms a spiral winding about 
the aperture 57. The flex strip 54 and conductor 56 are coated with an 
insulating layer (not shown in FIG. 6). As will become better understood 
from the following discussion, the flat conductors 48 and 56 are 
electrically connected to form a single primary winding consisting of the 
series connected primary windings of two transformers (e.g., windings 300 
and 302 in FIG. 4). As will be discussed more fully below, in an actual 
embodiment of the present invention, the top and bottom flex strips 46 and 
54 are made by folding one flexible sheet. 
The secondary conductive sheet 60 consists of two spaced apart, parallel 
legs 51A and 51B, which are joined by a base 53. An edge of the base 53, 
opposite the legs 51A and 51B, is attached to a return conductor 52. The 
legs 51A and 51B have ends 55A and 55B opposite the base 53. As will 
become better understood from the following discussion, the legs 51A and 
52B form secondary windings of two transformers (e.g., windings 304 and 
306 in FIG. 4). The bottom conductive sheet 65 consists of a plate 66 and 
a conductor 67 joined to one end of the plate 66. As will become better 
understood from the following discussion, the bottom conductive sheet 65 
forms a common connection (node 312, FIG. 4) between the secondary 
windings of two transformers. Accordingly, in the preferred embodiment of 
the dual transformer device 34 illustrated in FIG. 6, the legs 51A and 51B 
and the bottom sheet 65 form single-turn secondary windings of two 
transformers. 
Turning next to the rectifier 36, the diodes 58A and 58B preferably have a 
substantially flat and compact shape that make them suitable for use in a 
high density power converter. In one actual embodiment of the invention, 
the diodes 58A and 58b are hermetically sealed diode packages having top 
and bottom surfaces that form the anode and cathode of the diode. A diode 
suitable for use in the present invention is manufactured by SEMETEX 
Corporation. The capacitor assembly 60 preferably, has a low profile that 
also makes it suitable for use in a high density power converter. In one 
actual embodiment of the present invention, the capacitor assembly 60 
comprises a plurality of capacitors that are arranged to provide a low 
profile structure. The diodes 58A and 58B and the capacitor assembly 60 
are represented schematically as D2, D4 and C1, respectively, in FIG. 4. 
The components of the power converter 30 illustrated in FIG. 6 may be 
assembled in the following manner. The top and bottom flex strips 46 and 
54 are placed on either side (i.e., on top and bottom) of the secondary 
conductive sheet 50, such that the central apertures 49 and 57 are aligned 
with one another and aligned with the space between the legs 51A and 51B. 
This subassembly is positioned on the core bottom 62 such that the top and 
bottom flex strips 46 and 54 and legs 51A and 51B are located between the 
outer legs 84A and 84B of the core bottom 62. Furthermore, the center leg 
86 of the core bottom 62 is aligned with, and penetrates, the central 
apertures 49 and 57 of the top and bottom flex strips 46 and 54. The gap 
paper 44 is placed on top of the outer legs 84A and 84B. The core top 42 
is positioned on top of the gap paper 44, such that air gaps are formed 
between the core top 42 and the outer legs 84A and 84B and the center leg 
86 of the core bottom 62. 
Adjusting the air gap distance between the outer legs 84A and 84B and the 
core bottom 62 changes the values of the magnetizing inductances, L1 and 
L2, of transformers, T1 and T2 (FIG. 4). By adjusting the air gap between 
the center leg 86 and the core top 42, the value of the inductance, L3, 
discussed above and depicted in FIG. 4, can be adjusted. Accordingly, the 
air gap between the center leg 86 and the bottom core 62 can be adjusted, 
such that the resulting value of L3 provides sufficient energy to minimize 
switching losses in the AC voltage supply 32 (also FIG. 4). 
The core assembly 40 encloses the top and bottom flex strips 46 and 54 and 
a portion of the secondary conductive sheet 50. The ends 55A and 55B of 
the legs 51A and 51B and the return conductor 52 protrude from the core 
assembly 40 (see FIG. 7). Top surfaces of the diodes 58A and 58B form the 
anodes of the diodes 58A and 58B. The top surfaces 59A and 59B are 
attached to the ends 55A and 55B of the legs 51A and 51B, respectively. A 
top surface 61A of the capacitor assembly 60 forms one side of the 
plurality of capacitors in the assembly 60. The top surface 61 is 
connected to the return conductor 52. The bottom conductive sheet 65 is 
attached to the foregoing subassembly. Further, an opposite surface of the 
capacitor assembly 60 (not shown) forms the other side of the capacitor in 
the assembly 60 and is connected to the conductor 67. Similarly, opposite 
surface (not shown) of the diodes 58A and 58B form the cathode of the 
diodes 58A and 58B and are attached to bottom plate 66. 
FIG. 7 is a perspective view of the completed assembly discussed above. As 
FIG. 7 illustrates, the assembled power converter 30 is a substantially 
flat and compact structure. As a result, the power converter 30 has a very 
high power density. One particular working model of a power converter 30 
formed in accordance with the present invention and having the general 
shape illustrated in FIG. 7 has a power density of approximately 400 
watts/cubic inch. Such a high power density is made feasible in part by 
the compact structure of the integrated magnetics device forming the dual 
transformer device 34. Further, the compact structure, (i.e. the close 
proximity of the primary and secondary windings) provides for low leakage 
inductance that is repeatable in manufacturing the power converter. The 
high power density is further facilitated by operating the power converter 
30 at a high frequency (e.g., 500 kHz). The high operating frequency is a 
function of the switching capabilities of the bridge circuit in the AC 
power supply 32 discussed above. 
As noted above, in an actual embodiment of the present invention, the top 
and bottom flex strips 46 and 54 are formed by folding a single flexible 
sheet. FIGS. 8A and 8B illustrate a preferred embodiment of the flex 
strips 46 and 54 formed by folding a flexible sheet 500. Turning first to 
FIG. 8A, a front surface 501A of the flexible sheet 500 is depicted. The 
sheet 500 consists of a body 502 and a tab 504 extending from one side of 
the body 502. The body 502, as depicted in FIG. 8A, is divided into four 
quadrants 508, 510, 512, and 514, which are defined by two perpendicular 
fold lines 530 and 532. Apertures 516 are centrally located in each of the 
quadrants 508, 510, 512, and 514. A flat conductor 518A is located on the 
front surface 501A and forms a spiral winding about the aperture 516 in 
the quadrant 508. One end of the conductor 518A is brought out to a 
termination point 562A located near the edge of the tab 504. As will be 
discussed more fully below, the other end of the conductor 518A is 
electrically connected to a conductor 518B on a back side 501B of the 
flexible sheet 500 (FIG. 8B). 
As illustrated in FIG. 8A, another flat conductor 520A is located on the 
front surface 501A and forms a spiral winding about the aperture 516 in 
the quadrant 512. One end of the flat conductor 512 is brought out to a 
termination point 526B located near the edge of the tab 504. As will be 
discussed more fully below, the other end of the conductor 520A is 
electrically connected to a conductor 520B located on the back side 501B 
of the flexible sheet 500 (FIG. 8B). 
An electrostatic shield 524B is located on the front surface 501A and 
substantially covers the quadrant 510 around the aperture 516. Similarly, 
an electrostatic shield 524B is located on the front surface 501A of the 
quadrant 514 about the aperture 516. The shields 524A and 524B are 
connected to a conductor 529, which is brought out to a termination point 
528 located near the edge of the tab 504. The front surface 501A, the 
shields 524A and 524B, and the conductors 518A, 520A, and 529 are covered 
by an insulating layer (now shown in FIG. 8A). 
Turning next to FIG. 8B, a back surface 501B of the flexible sheet 500 is 
depicted. A conductor 518B is located on the back surface 501B of the 
quadrant 508 and forms a spiral winding about the aperture 516. As noted 
above, one end of the conductor 518B is connected to the conductor 518A 
located on the front surface 501A of the quadrant 508. The other end of 
the conductor 581B is connected to a conductive strip 522. Similarly, a 
conductor 520B is located on the back side 501B of the quadrant 512 and 
forms a spiral winding about the aperture 516. As was also noted above, 
one end of the conductor 520B is connected to the conductor 520A located 
on the front side 501A of the quadrant 512. The other end of the conductor 
520B is connected to the conductive strip 522. Thus, the spiral winding 
formed by conductors 518A and 518B are series connected to the spiral 
winding formed by conductors 520A and 520B. The back side 502B, the 
conductors 518B and 520B, and the conductive strip 522 are covered by an 
insulating layer (not shown). 
In accordance with the presently preferred embodiment of the invention, the 
conductors 518A, 518B, 520A, 520B; the shields 524A, 524B, and the 
conductive strip 522 are formed by conventional printed circuitry 
techniques, such as by a die cutting or a photo etching process, for 
example. Electrical connections between ends of conductors 518A and 518B 
and between ends of conductor 520A and 520B are also made using 
conventional printed circuit board techniques. For example, solder may be 
applied to holes that penetrate through the ends of the conductors and the 
sheet 500, such that the solder makes electrical contact with the 
appropriate ends of the conductors. 
As will become better understood from the following discussion, the 
windings described above and illustrated in FIG. 8A and 8B form the top 
primary winding 48 and the bottom primary winding 56 of the integrated 
magnetics device illustrated in FIG. 6 and discussed above. The top and 
bottom flex strips 46 and 54 (FIG. 6) may be formed by folding the 
flexible sheet 500 in the following manner. First the flexible sheet 500 
is folded along fold line 530, such that the back surfaces 501B of 
quadrants 510 and 514 are placed adjacent to the front surfaces 501A of 
quadrants 508 and 512. Next, the flexible sheet 500 is folded along the 
fold line 532 such that the quadrants 508 and 510 are placed adjacent to 
the quadrants 512 and 514. By folding the flexible sheet 500 in the manner 
described above, the quadrants 508 and 510 form the top flex strip 46, 
illustrated in FIG. 6, and the quadrants 512 and 514 form the bottom flex 
strip 54, also illustrated in FIG. 6. Accordingly, the conductors 518A and 
518B form the top primary winding 48 and the conductors 520A and 520B form 
the bottom primary winding 56. A suitable electrical connector (not shown) 
may be connected to the termination points 526A, 526B and 528 located on 
tab 504. Thus, a bridge circuit (FIG. 4) may be connected to the windings 
on sheet 500. 
As can be readily appreciated from the foregoing description, the present 
invention provides a dual transformer device that may be implemented as an 
integrated magnetics device suitable for use in a high density power 
converter. While a preferred embodiment of the invention has been 
illustrated and described herein, it is to be understood that, within the 
scope of the appended claims, various changes can be made. For example, 
the series connected primary winding may be made from a relatively 
nonflexible material and consists of a single layer of the material 
instead of from the multiple layers of the flexible sheet discussed above. 
Likewise, the two legs of the secondary conductive sheet that form the 
secondary windings of the transformers may have a different configuration 
than as described above, such as being vertically spaced apart instead of 
horizontally spaced apart. In addition, numerous other switching devices, 
such as half-wave bridge circuits, may be used to supply an alternating 
signal to the dual transformer device. Furthermore, the dual transformer 
device may be carried out with two separation transformers interconnected 
as shown in FIG. 3 and implemented with a variety of different core and 
winding types common in the art. Hence, the invention can be practiced 
otherwise than as specifically described herein.