Control strategy for reuse system assignments and handoff

Disclosed is a method of selecting a relatively high reliability signal path between a mobile communication unit and a number of possible base sites.

FIELD OF THE INVENTION 
This invention relates to communication systems, including but not limited 
to trunked mobile communication systems. 
BACKGROUND OF THE INVENTION 
Trunked, mobile communication systems are known. Such systems typically 
allocate communication resources upon perception of a need for 
communication services. Such allocation, in some systems, is under control 
of a resource controller. In other systems, a communication unit searches 
for an unused communication resource, and upon identification of such 
resource, seizes possession of such resource. 
Upon receipt of an allocation of a communication resource communication 
units tune to allocated frequencies and begin transceiving an information 
signal over an allocated bandwidth. Transmissions between communication 
units or between a communication unit and a subscriber on a public service 
telephone network may continue until the end of a conversation or until 
one of the communication units exceeds the range of his transceiver. 
Trunked communication systems in which information signals are encoded 
using quadrature amplitude modulation (QAM) are also known. Such systems 
combine characteristics of both phase and amplitude modulation to reduce 
the bandwidth required to carry a fixed amount of information. In QAM, 
information is conveyed using changes in both the amplitude of a carrier 
wave and the relative phase angle of the carrier signal with respect to a 
reference angle. Because of the multi-dimensional aspect of QAM four or 
more bits of digital data may be transmitted per QAM signal element. 
Multi-carrier QAM is a technique in which an information-bearing signal, 
such as serial digitized voice, or digital data from a computer for 
example, is divided up into multiple, separate, frequency division 
multiplexed QAM signals. Each QAM signal occupies a discrete frequency 
band (with each of the bands being substantially frequency adjacent to the 
others) and carries a portion of the information in the complete 
information-bearing signal. 
In order to coherently detect the transmitted data, the receiver must be 
able to measure and correct phase and amplitude variations induced by the 
transmission channel. The variations may be induced by multiple path 
signal propagation and are commonly referred to as fading. 
Two types of fading can occur over the transmission channel. The types are 
differentiated by the ratio of the differential delays between multiple 
signal paths and the transmitted bit period. If the maximum differential 
delay between significant signal paths is much less than the bit period, 
then the fading process is referred to as flat fading. The term, flat 
fading, applies because the channel appears to vary uniformly across the 
transmission bandwidth as a function of time (i.e. flat across the signal 
bandwidth). If the maximum differential delay between significant signal 
paths is comparable to or greater than the bit period, then the fading 
process is referred to as frequency selective fading. In the case of 
frequency selective fading the channel variation is a function of 
frequency within the frequency bandwidth. 
The means by which phase and amplitude variations are measured and 
corrected is based upon a process of inserting known pilot symbols in the 
transmitted signal stream at fixed intervals. The process of inserting 
known symbols at predetermined intervals in the data symbol stream is 
commonly referred to as a time division multiplex (TDM) pilot reference. 
At the receiver, the location of the known pilot symbols is determined by a 
time synchronization process. The difference between the amplitude and 
phase of the transmitted and received pilot symbols is the variation 
induced by the channel. An interpolation filter is used to generate 
channel variation values for the intervening data symbols. The channel 
variation values are then used to correct for channel variation, resulting 
in relativly accurate estimates of the transmitted symbols. 
When multi-channel transmission is used, each sub-channel may experience 
its own unique variation due to frequency selective fading. To allow for 
such unique variation, and to provide a mechanism for correction of such 
variation, each sub-channel may contain pilot symbols. 
In the context of geographic reuse the limiting factor in the quality of a 
received signal is co-channel interference. Co-channel interference may be 
caused by reception of unwanted signals on the same frequency as the 
desired signal. 
Attempts to increase the quality of a transmitted signal often include 
increasing the power level of a transmitted signal. Increasing the power 
level of a transmitted signal increases the ratio of signal to noise and, 
consequently, decreases the effect of co-channel interference. Increasing 
the power level of a transmitted signal often times does result in a 
higher quality received signal. Increasing the power level, on the other 
hand, often also results in an expansion of the geographic area within 
which the resource may not be re-used by other communication systems. 
Other attempts to provide a higher quality received signal include 
implementing search algorithms to identify a signal path exhibiting the 
least amount of co-channel interference. Such algorithms in some cases 
involve measuring the quality of a signal received on different 
communication resources from different sources to identify the resource, 
and signal source, providing the highest reliability in terms of received 
signal quality. Methods used in the prior art to identify high reliability 
resources include signal strength measurements and measurement of bit 
error rates. 
Shown (FIG. 12) is a graph of a computer simulation of probability of bit 
error versus E.sub.b /N.sub.o (energy per bit divided by noise in a one 
Hertz bandwidth). The graph, as is known, gives a substantially accurate 
representation of error rates under a variety of transmitting 
environments. Transmitting environments offered include S=0 .mu.s delay 
spread (flat Rayleigh) to S=10 .mu.s delay spread (very bad hilly). 
The information shown in FIG. 12, as is known, is also commonly displayed 
using an ordinate calculated in terms of C/[I+N] where C is signal, I is 
interference and N is noise. The two methods of displaying bit error 
rates, as is known to those in the art, are used substantially 
interchangeably. 
In the context of communications between a communication unit and a 
transceiver at a base site the identification of the highest reliability 
communication resource is often a measurement of proximity of the closest 
base site transceiver. The closest base site often provides the strongest 
signal. Signal strength or bit error rate measurements may often provide 
similar results in terms of resource reliability when used to identify the 
communication resource providing access to the closest communication 
services provider. 
Bit error rate calculations, on the other hand, are time consuming. Where 
error rates are low a significant time interval must be allotted to 
accumulating and averaging errors. 
Signal strength measurements, though quick and easy to implement, do not 
directly measure resource reliability. Resource reliability in the most 
direct manner involves minimal bit error rates. Minimal bit error rates, 
on the other hand, involve resource resistance to short term interference 
factors such as frequency selective fading or short term signal strength 
variations. 
While, in the past, signal strength or bit error rate measurements have 
provided good indication of the reliability of a communication resource 
such measurements do not give indication of the effects of certain types 
of interference exemplified by frequency selective fading. Frequency 
selective fading may affect certain aspects of a received signal, thereby 
degrading signal quality, without appreciably affecting signal strength. 
By way of example the following comparison is offered wherein a 
communication unit compares the reliability of communication resources 
between the communication unit and two base sites. The maximum allowable 
bit error rate is two percent. The first base site has a detected bit 
error rate of 0.5% and 26 dB E.sub.b /N.sub.o with S=0 (flat Rayleight 
fading). The second base site has a detected error rate of 1.3% and 35 dB 
E.sub.b /N.sub.o with S=10 .mu.s (very bad hilly). The results can be seen 
plotted on FIG. 12. The first site shows a 10 dB link margin between the 
0.5% error rate (26 db) and 2% error (16 dB). The second site shows a 16 
dB link margin between the 1.3% error rate (35 dB) and 2% error (19 dB). 
In the example given the lowest bit error rate does not offer the highest 
reliability in terms of signal reception. While the first site offers a 
lower bit error rate, log-normal fluctuations in received power levels may 
disrupt signal reception in excess of allowable standards. Clearly the 
second site while offering a higher initial error rate offers the highest 
reliability signalling channel. 
While the example offered provides an indication of channel reliability, 
use of such an algorithm depends upon provisions within the receiver for 
measuring delay spread. Measurement of delay spread is well known in the 
art (see "900-MHz Multipath Propagation Measurements for U.S. Digital 
Cellular Radiotelephone", IEEE Transactions, Vol. 39, No. 2, May 1990, 
pages 132-139) and allows a receiver to enter such a graph (FIG. 12) based 
upon measured delay spread and bit error rate for each measured channel. 
Such an algorithm, as is known, may be used to gain a measurement of 
reliability of communication resources between base site transmitters and 
a mobile communication unit. 
Measurement of bit error rates, as has been mentioned, is time consuming. 
Where bit error rates are low a number of measurements may need to be 
undertaken to ensure reliable data. A communication units seeking handoff 
may not have time to measure bit error rates in an environment of a 
rapidly deteriorating received signal. 
Because of the importance of mobile communications a need exists for a 
method of measuring signal quality (reliability of a communication 
resource) that can be rapidly calculated from easily measured parameters 
and which takes into account such affects as frequency selective fading. 
Such a method should give a indication of instantaneous as well as average 
reliability of communication resources and be subject to rapid evaluation. 
Such a method would be useful both from the viewpoint of handoff of 
communication units between base sites but also in terms of selection of 
an initial base site to initiate a communication transaction. 
SUMMARY OF THE INVENTION 
Pursuant to one embodiment of the invention a method is offered of 
selecting a high reliability signal path between a communication unit and 
a number of base site transmitters where each base site transmitter is 
transmitting an information signal. The method is based upon a comparison 
of measured delay spreads and average signal quality factors calculated 
from measured signal parameters. The delay spread and signal parameters 
are measured by the communication unit from information signals received 
from at least some of the base site transmitters. 
The method includes measuring a delay spread and calculating a signal 
quality factor from measured signal parameters of a first information 
signal received from a first base site of the number of base site. A first 
link margin is then determined based upon the measured delay spread and 
the calculated signal quality factor of the first information signal. 
A delay spread is measured and a signal quality factor is calculated from 
measured signal parameters of an at least second information signal 
received from an at least second base site of the number of base sites. A 
second link margin is determined based upon the delay spread and signal 
quality factor of the at least second information signal. A high 
reliabilty signal path is then selected based, at least in part, upon the 
highest relative link margin.

DESCRIPTION OF A PREFERRED EMBODIMENT 
FIG. 1 shows a simplified block diagram of a four subchannel QAM 
transmitter (10). The QAM transmitter (10) formats information from a 
serial data source (12) into four subchannels, wherein each subchannel 
carries a fractional amount of the information in the original serial bit 
stream (12). The serial bit stream (12) may originate from virtually any 
particular source and may be digitized voice information, data from a 
computer or the like, or any other similar source of such information. 
The serial bit stream (12) is reformatted by a serial to parallel converter 
(14), which divides the serial bit stream into four different data 
streams. In the embodiment shown, the serial to parallel converter (14) 
formats 16 bits of serial data from the serial data source (12) into four, 
four bit data words. The data words from the serial to parallel converter 
(14) form a stream of discrete samples of information from the serial bit 
stream (12). In the embodiment shown, in this case, the discrete samples 
can be represented by 16 QAM symbols, which correspond to the sixteen 
points in the constellation shown in FIG. 7. 
FIG. 7 shows a constellation map of a 16 level QAM signal. Each point of 
this constellation map is assigned a binary bit pattern corresponding to 
every possible binary bit pattern representable by four bits. For example, 
a bit pattern produced by the serial to parallel converter (14) of all 
zeros might be represented by a vector with magnitude M1, at 45 degrees. 
This vector might be transmitted as a carrier wave of a certain amplitude 
M1 and with a certain phase-shift of .phi.1, identifiable by a receiver as 
a 45 degree phase shift from some other reference value. 
Synchronization and pilot symbols are inserted in a sync insertion block 
(16) as shown. (The actual symbols might be generated by a microcomputer, 
a suitable digital signal processor, or other suitable device.) Since the 
data from the serial to parallel converter (14) is a stream of samples, 
the synchronization and pilot symbols are also discrete samples of 
information. The output of the symbol insertion block (16) is coupled to a 
pulse shaping filter (18) which band limits the frequency spectrum of the 
composite signals from the symbol insertion block (18). 
The output of the pulse shaping filter (18) is coupled to a modulator (20) 
which multiplies the output of the pulse shaping filter (18) by a sine 
wave quantity equal to e.sup.(j2.pi.f.sbsp.i.sup.t) where i runs from 1 to 
4. The output of the first modulator stage (20) is a complex zero IF 
signal which is summed together in a summer (22) with the signals from 
other pulse shaping filters and modulators (18b through d and 20b through 
d, respectively) as shown. 
The complex zero IF output from the summer (22) is frequency shifted by an 
IF up-converter, or modulator (24) to some carrier frequency, f.sub.o, 
amplified by an RF Amplifier (26) for subsequent broadcasting on an 
antenna (28). Each of the QAM subchannels broadcast from the antenna 
occupies its own frequency spectrum as a result of the modulation process 
used in the transmitter (10). 
FIG. 2 shows a representation of the transmit energy spectrum output from 
the transmitter (10) shown in FIG. 1 with four subchannels (32, 34, 36, 
and 38) centered about a center frequency f.sub.o. Note that each 
subchannel (32, 34, 36, and 38) has its own center frequency f.sub.1 
+f.sub.o, f.sub.2 +f.sub.o, f.sub.3 +f.sub.o, and f.sub.4 +f.sub.o 
respectively (42, 44, 46, and 48 respectively). 
FIG. 3 shows a representative diagram of the information that might be 
present on each of the subchannels 1 through 4 shown in FIG. 2. Note that 
subchannel 1 (52) is shown with a series of synchronizing sequences 
designated S.sub.11, S.sub.12, through S.sub.1n. (These synchronizing 
sequences might be described as a sequence of symbols, represented as 
vectors that permit synchronization.) Subchannel 2 (54) has its own series 
of sychronizing symbols S.sub.21, S.sub.22, through S.sub.2n. Similarly, 
subchannels 3 and 4 (56 and 58) have sync symbols S.sub.31 through 
S.sub.3n, and S.sub.41 through S.sub.4n respectively. These synchronizing 
sequences are complex values added to the information within subchannels 1 
through 4 and are preselected values to simplify detection and decoding by 
a QAM receiver. 
FIG. 4 shows a simplified block diagram of a QAM receiver (502, FIG. 16) 
contained within a mobile communication unit receiver (60). A frequency 
preselector (62) detects the RF energy in the transmit spectrum shown in 
FIG. 2 and presents this information to an IF stage (64), the output of 
which is a zero IF signal, comprised of streams of complex quantities 
known or referred to as an in-phase and quadrature components of a zero-IF 
signal. This zero IF down converter (64) might include an automatic 
frequency control input that permits it to track shifts in frequency of 
the signal received by the receiver. A sync detection circuit (66) 
monitors these zero IF signals to find timing synchronization of the QAM 
symbols shown in the subchannels of FIG. 3. (Information in the QAM 
subchannels is transmitted as discrete packets, which are finite time 
periods of amplitude modulated and frequency shifted or phase modulated 
carrier signals. The sync detection circuit (66) includes circuitry to 
identify, from the synchronizing sequences added to the information in the 
QAM subchannels, when information in the QAM subchannels should be sampled 
for detection.) 
The zero-IF signal from the zero-IF converter 64 is coupled to four 
subchannel receivers that each include subchannel mixers (65a through 65d) 
and receiver pulse-shaping filters (67a through 67d). The subchannel 
mixers multiply the zero-IF signal by a signal, e.sup.-j2.pi.f.sbsp.t, 
where f.sub.i is the subchannel center frequency for the respective 
subchannels, one through four, and t is time. The output of a subchannel 
mixer is a signal centered about zero Hertz, which is filtered by a pulse 
shaping filter (67a through 67d) to remove noise and any undesired 
subchannel signals. 
The output of the pulse shaping filters is sampled at a rate determined by 
the sync detection circuit (66). The sampled outputs of the pulse shaping 
filters are input to symbol detector blocks (69a through 69d) that 
estimate the information symbols originally transmitted. 
Automatic frequency control is provided by an AFC block (68). The AFC block 
(68) receives the sampled output of the pulse shaping filters during the 
times when sync symbols are present, as determined by the sync detection 
block (66). Stated alternatively, the AFC block only utilizes sync symbol 
samples, x.sub.ij, (where i is the subchannel number and j is the sync 
symbol number) from the pulse shaping filters when sync symbols are 
present. The sync symbol samples, x.sub.ij, correspond to the originally 
transmitted sync symbol vectors, S.sub.ij. The output of the AFC block is 
coupled to the zero-IF down converter (64), or to possibly other frequency 
shifting stages between the preselector (62) and the symbol detectors (69a 
through 69d) to track received signal frequency shifts. 
FIG. 5 shows a simplified block diagram of a sync matched filter and other 
circuitry associated with the sync detector (66) of FIG. 4 and within the 
receiver (60). After being down-converted to zero-IF by the zero-IF down 
converter (64), a complex zero IF is input to a sync matched filter (660) 
which is a filter whose impulse response closely approximates the complex 
conjugate of the time reversed transmitted composite waveform. (Whatever 
signal appears at the antenna of the transmitter has a net composite 
waveform from the addition of the QAM subchannels, including sync 
sequences. The sync matched filter (660) tests only for the waveform from 
the transmitter due to the sequences of sync symbols S.sub.ij where i is 
the subchannel number and j is the sync symbol time index, or number, 
sent. This filter (660) does not test for information in the QAM frame.) 
The output of the synchronization matched filter (660) is coupled to a 
magnitude squaring block (670) which computes the square of the amplitude 
of the sync matched filter (660) output and allows the determination of 
the power level of the signal detected by the sync matched filter (660). 
The output of the magnitude squaring block (670) is compared against a 
threshold in a comparator (680) to determine whether or not the sync 
matched filter (660) has found a synchronization pattern from the 
transmitter (10). (As shown in FIG. 5A, the threshold can be chosen to 
discriminate against noise.) 
Still referring to FIG. 5, the output of the magnitude squaring block is 
also coupled to a peak timing detector circuit (690). The peak timing 
detector circuit (690) finds the time of the occurrence of the peak output 
value from the magnitude squaring block. The time of occurrence of the 
peak output value from the magnitude squaring block provides timing 
information of the QAM symbol times to enable accurate symbol sampling by 
the receiver (60). (In FIG. 4, the output (692) of the sync detection 
block (66) controls when symbols are to be acquired.) 
In the preferred embodiment, the receiver elements shown in FIG. 4, 
excluding the preselector (62) and zero IF (64) are performed by a digital 
signal processor, such as a Motorola DSP 56000 family device. 
It has been determined that if opposing subchannels (FIG. 3, 1 and 4, or 2 
and 3) are loaded with sync vectors that are complex conjugates of each 
other, the waveforms produced by the addition of the modulated sync 
vectors requires a simplified receiver that must only identify a signal 
representing a resultant vector that has only real quantities. For 
example, referring to FIG. 2, if subchannel 1 and subchannel 4 are 
considered first and second halves of a pair of channels, both equally 
displaced and centered about the center frequency f.sub.o (40), and if 
subchannel 2 and subchannel 3 are considered as the first and second 
halves of a second pair of subchannels, both equally displaced from and 
about f.sub.o, loading a complex vector S.sub.11 into the synchronizing 
symbol for subchannel 1, and loading its complex conjugate S.sub.41 into 
subchannel 4, (where S.sub.11 *=S.sub.41, and where the * denotes 
conjugate) produces (subsequent to pulse filtering and modulation) upon 
their addition a resultant signal that will have only a real component. 
Similarly, loading a complex vector S.sub.21 into subchannel 2 and its 
complex conjugate S.sub.31 into subchannel 3, (where S.sub.31 =S.sub.21 *, 
where * denotes the conjugate) produces upon their addition a resultant 
signal that similarly has only real components. 
Referring to FIG. 8, there is shown a map of vectors and complex conjugates 
of these vectors that are used to simplify sync and timing detection for 
the receiver. The first sync vector S.sub.11 is arbitrarily chosen at a 45 
degree angle with a particular magnitude of M1. Its complex conjugate is 
used as the sync symbol for subchannel 4 and is shown as S.sub.41. The 
sync symbols for subchannels 2 and 3 are also shown with the sync vector 
for subchannel 2 represented by S.sub.21 and sync vector for subchannel 3 
shown as S.sub.31. Similar diagrams could be shown for other symbols 
occurring at a particular time, comprising sync sequences, i.e. S.sub.12, 
S.sub.22, S.sub.32, S.sub.42, etc. 
Simplified detection of the transmitted symbol timing and synchronization 
by inclusion of complex vectors and conjugates is accomplished by the sync 
matched filter shown in FIG. 6 and contained within the receiver (60). In 
FIG. 6A, the complex zero IF, denoted as the real portion (A) and the 
imaginary portion (B) couples to a group of complex filter elements (662, 
664, 666, and 668). Each element has a real scalar input and real scalar 
output. The impulse response of this complex filter is matched to the 
quantity s(t)=s.sub.r (t)+js.sub.i (t) where s.sub.r and s.sub.i are the 
real and imaginary components, respectively of s(t). S(t) represents the 
waveform of the composite of the sync symbol elements sent by the 
transmitter. The impulse response of the matched filter in general is 
approximately equal to s*(-t)=s.sub.r (-t)-j s.sub.i (-t). Since the input 
to the matched filter is a complex signal, the actual implementation 
requires four real-valued filters as shown in FIG. 6A. 
A simplification of the receiver shown in FIG. 6A, and one which is made 
possible by the use of complex sync vectors and their complex conjugates 
in paired, or matched, QAM subchannels is shown in FIG. 6B. 
In FIG. 6B, the complex zero IF signal is coupled to a filter matched to 
the waveform produced by sync vectors that are complex conjugates of each 
other and which when modulated and added together prior to transmission 
produce a constant phase waveform. The filter shown in FIG. 6B is 
considerably simpler than that shown in FIG. 6A. 
It should be noted that in any multi-carrier QAM system described herein 
not all of the paired subchannels need to have the complex vector/complex 
conjugate vector sync sequences added to them. Of any multi-carrier QAM 
system, at least one pair of the paired subchannels must have the 
synchronizing sequence described herein added to it to permit simplified 
sync detection. (The requirement that the subchannels be centered about a 
center frequency still holds.) 
As each complex sync vector is received on a channel (101a through 101d) 
sychronization is established by the receiver to transmitted signals. 
Synchronization of the receiver to the transmitter allows the receiver to 
accurately detect data and pilot symbols between synchronization symbols. 
Inclusion of known pilot symbols allows a receiver to correct distortion 
induced into the signal during transmission. Pilot symbols, in one 
embodiment of the invention, may be inserted into the data as often as one 
pilot symbol to five data symbols as a means of measuring and correcting 
distortion within intervening data symbols. As each pilot symbol is 
received on a channel (101a through 101d) a distortion factor is 
calculated within the symbol detector (69). A complex representation of a 
received pilot is divided by a complex representation of the originally 
transmitted pilot symbol to produce a complex, pilot correction factor for 
that pilot. Two pilot correction factors, calculated from two sequential 
pilot symbols, are processed by interpolation over the data transmission 
slots between the two pilot symbols to produce a number of interpolated 
pilot correction symbols equal to the number of data symbols between pilot 
symbols. The data symbols are then processed by division by the 
interpolated pilot correction symbols to produce corrected data symbols 
having a corrected phase and amplitude for each channel. 
The corrected data symbols are then compared with the constellation chart 
to determine the closest match. The closest match is assumed to be the 
correct, transmitted symbol. 
In one embodiment of the invention an average signal quality factor (q) is 
calculated for the four channels used in the QUAD16QAM example using 
measured signal parameters comprised of the pilot correction factors and 
corrected data symbols. Signal quality factors are calculated values 
representative of delay spread and the ratio of carrier to interference 
plus noise [C/(I+N)]. A means of quickly and accurately measuring signal 
quality factors can be an important factor in efficient resource 
assignment and handoff. Given two possible resources, both of which have 
delay spreads within the design limits of the communication system, the 
resource with the best signal quality factor may provide the best relative 
quality signal transmission. 
Calculation of the signal quality factor is accomplished through the 
inclusion of the signal processing elements shown in FIGS. 9 and 10 and 
included within the receiver (60). Shown in FIG. 13 is a flow chart by 
which the functions of the signal processing elements of FIGS. 9 and 10 
and receiver 60 may be better understood. 
The pilot correction factors are calculated as described above (FIG. 13, 
200 and 201) for each channel (69, FIG. 10). Each pilot correction factor 
is squared (103) (207) to produce an absolute pilot value and to eliminate 
the imaginary components and then divided (107a through 107d) by a 
normalizing factor determined by summing (104) the absolute values, and 
amplifying (105) by an empirically determined factor. The result is four 
weighting factors (w.sub.0 through w.sub.3). 
Instantaneous signal quality factors are calculated (209) by multiplying 
(108a through 108d) (208) absolute error values by the weighting factors 
(w.sub.0 through w.sub.3). Absolute error values are determined by 
selecting the corrected data symbols and determining the closest allowable 
data symbol to the corrected data symbol within a threshold detector (109a 
through 109d) (204). The corrected data symbol is subtracted from the 
closest allowable data symbol within an adder (110a through 110d) to 
produce data error values (205). The data error values are then squared 
(111a through 111d) to eliminate imaginary components and clipped (112a 
through 112d) to eliminate the effects of spurious symbols to produce an 
absolute error value (206). 
The average signal quality factor (q) can be determined by summing (113) 
the instantaneous signal quality factors and integrating (114) (209) to 
produce the value q. Calculating an average signal quality factor q for a 
signal received on a communication resource allows for a comparison of the 
reliability of a communication path between a communication unit and a 
base site to determine the resource offering the highest reliability of 
performance (high reliability signal path). 
Shown (FIG. 11) is a computer simulation of calculated signal quality 
factor (q) for a four received signals (delay spreads of 0.0, 2.5, 10 and 
20 .mu.s) and over a range of signal to interference plus noise (C/[I+N]) 
ratios. Also shown on the same graph (FIG. 11) are calculated bit error 
rates for the same four signals and over the same range of signal to 
interference plus noise (C/[I+N]) ratios. In use a communication unit 
would calculate q and request service on a high reliability resource from 
the base site offering the lowest relative, calculated value of q. 
By way of example a communication unit (comprising a receiver (60) and 
optional delay measurement device (61)) detects signals from a first and a 
second base site transmitter (10) within small geographic areas (301 and 
302). The communication unit (60 and 61) calculates a signal quality 
factor for each. The delay spread for the first base site signal may be 
assumed to be zero (S=0 .mu.s) or the delay spread may be communicated to 
the receiver (60) on a control channel from the first base site as a 
locally measured condition. The calculated value of q is 0.18. A 
controller (500, FIG. 16) within the receiver (60) enters the graph (FIG. 
11) stored within a memory (501) of the receiver (60) based upon the 
calculated value for the first site identifies a first point (A) and a 
first value of C/[I+N] of approximately 26 db. 
The second base site may have a communicated delay spread of ten 
microseconds (S=10 .mu.s) and a calculated value of q of 0.14. Entering 
the graph (FIG. 11) based upon the calculated value identifies a second 
point (B) and a second value of C/[I+N] of 41 db. Based on the lowest 
calculated value of q the highest reliability communication resource 
selected by the controller (500) is that offered by the second base site 
(q=0.14). Based upon the lowest calculated value of q the communication 
unit (60 and 61) would request a communication resource from the second 
base site. 
According to the invention a communication unit (60 and 61) may calculate 
the delay spread using prior art methods based upon measured parameters by 
the delay spread measurement device (61) of the detected signal and would 
request service based on an algorithm (FIG. 15) calculating the highest 
decibel range between a detected signal and an unacceptable signal 
(unacceptable bit error rate) based on the calculated curve upon which the 
signal is operating. 
In the above example since delay spread is known (through measurement) the 
graph (FIG. 11) stored within a memory (not shown) of the receiver (60) 
can be used to determine projected bit error rates. The projected bit 
error rate of the first base site can be determined within the controller 
(500) by extending a line vertically down from point A. The intersection 
of the vertical line from point A and the bit error probability line for 
zero delay spread (S=0 .mu.s) occurs at substantially 0.9%. The projected 
bit error rate for the second base site (point B) can be determined in a 
substantially identical manner and in the example given for point B gives 
an indicated bit error rate of 1.1%. 
If an algorithm for assigning communication resources within a 
communication system is based upon the lowest bit error rate then the 
example indicates that the first base site may offer the most reliable 
resource. If, on the other hand, an unacceptable bit error rate is 
established (say 3%) then a channel reliability factor can be calculated 
for each base site based on a decibel range between the projected bit 
error rate and the unacceptable bit error rate. In the example the second 
base site offers a 20 db margin between the detected signal (1.1% bit 
error rate) and an unacceptable bit error rate (3%). The first base site 
offers a 7 db margin between the detected signal (0.9%) and an 
unacceptable signal (3%). If resource assignment is based upon resistance 
to signal fading, then the controller (500) would select the communication 
resource offering a high reliability signal path may be the path between 
the second base site and the communication unit. 
In another embodiment of the invention other constellation charts may be 
used in the transmission, reception, and subsequent detection of 
information signals. Examples include QPSK, 8PSK, 8AMPM, etc. 
In a further embodiment of the invention any number of subchannels may be 
used in the transmission, reception, and subsequent detection of 
information signals. The number of subchannels may be described as ranging 
from one to n.