Composite amplifier with efficient power reduction

A composite amplifier includes a plurality of signal amplifiers. The amplified signals are coupled by transmission lines to node points for combination. In order to achieve a reduction in power below full output power, some of the amplifiers are deenergized. Shorting switches at selected points along the transmission lines are selectively rendered conductive to adjust the impedances to maintain impedance match. Short- or open-circuited transmission lines may also be switched in-circuit to aid in maintaining impedance match.

This invention relates to composite power amplifiers in which the output 
powers from a plurality of separate amplifiers or amplifier modules are 
combined to produce a higher power output signal. 
Many communication systems require transponders separated by significant 
distances. Such transponders when used for radiative communication links 
between cities eliminate the need for land communication cables, which are 
very costly. The transponders cannot always be placed in the most ideal 
location, but rather must be placed at locations where towers can be 
placed, and the antennas may be required to have relatively high gain. The 
necessary gain is achievable with antennas of reasonable size and 
reasonable cost only at microwave frequencies and at frequencies higher 
than microwave. 
The transmission of signal from one transponder to another transponder 
requires a power amplifier located at the transponder which is capable of 
generating many watts of power with great reliability. In the past, the 
microwave power was generated by traveling wave tubes (TWT). Taveling wave 
tubes were, and continue to be, used for microwave transponders 
notwithstanding the reliability problem attributable to the inherent 
degradation resulting from operation over a period of time. More recently, 
solid state power amplifiers (SSPA) have been used at lower microwave 
frequencies, such as at C-band, instead of traveling wave tubes. The SSPA 
has no inherent degradation mechanism, and therefore is more reliable than 
the TWT. This reliablity is very advantageous, since microwave 
transponders are often placed at relatively inacessible locations, as for 
example the top of mountains. Generally speaking, solid state power 
amplifiers are implented by using a relatively large number of relatively 
low power solid state devices or amplifier modules. Each solid state 
device provides a small portion of the total power output, and power 
combiners are used to combine the powers from each of the individual solid 
state devices to generate the desired amount of signal power at the 
desired microwave or millimeter wave frequencies. 
For some purposes, it may be desirable to be able to vary the output power 
of the solid state power amplifier. This might be desirable to overcome 
the effects of increased or decreased path attenuation due to rainfall, 
foliage or other effects. Since some microwave transponders are located in 
areas far from a power grid, and therefore rely upon solar energy to 
generate energization voltages, it is important that control of the output 
power be performed in an efficient manner. Generally speaking, it is not 
efficient to simply reduce the amount of signal being amplified by an 
SSPA, as this reduces the amount of output power without affecting the 
energization power. 
Various types of power combiners are described in the article "Microwave 
Power Combining Techniques," by Kenneth J. Russell, published in the IEEE 
Transactions on Microwave Theory and Techniques, May 1979. The Russell 
article describes corporate or tree combiners, in which chains of 
combination are performed. Such arrangements tend to be disadvantageous 
because of the accumulation of losses in the combiners. It is extremely 
desirable that the power combination be performed with low loss. U.S. Pat. 
No. 4,641,106 issued Feb. 3, 1987, to Belohoubek et al. describes a 
low-loss radial power combiner. The radial power combiner is very 
advantageous when a large number of individual solid state amplifiers have 
their outputs combined. If one of the amplifiers fails, the net effect on 
the overall operation is small. However, if a large proportion of the 
amplifiers should fail, there might be significant degradation of the 
impedance match and combiner loss, and therefore there might also be a 
degradation in the amplifier gain. The degradation of impedance match and 
amplifier gain would also occur if some of the amplifier modules were to 
be deenergized in order to operate at reduced power levels. 
When a relatively small number of solid state power amplifier modules are 
to be have their output powers combined to produce a combined output, a 
low-loss combiner may be disadvantageous, because failure of a single 
solid state amplifier module may constitute a failure of a significant 
proportion of the total number of amplifier modules, and consequently, may 
result in degradation of the performance. Consequently, different 
considerations may govern power amplifiers which produce a sum output 
signal by combination of power when relatively few amplifier outputs are 
combined, compared with the situation which exists when relatively large 
numbers of amplifiers are combined, as in the Belohoubek et al. patent. 
U.S. Pat. No. 4,315,222, issued Feb. 9, 1982, to Saleh describes a power 
combiner arrangement in which the output power from a plurality of 
amplifier modules is combined at a single junction. Each amplifier module 
is coupled to the junction by a transmission line having an electrical 
length of one-quarter wavelength (.lambda./4) at a frequency within the 
operating frequency range. A sensing arrangement is coupled to each 
amplifier module which, in the event of failure of an amplifier, decouples 
the failed amplifier from the combining junction by way of a switch which 
effectively decouples the amplifier and its associated transmission line 
from the juncture. In one embodiment of the Saleh arrangement, a 
short-circuiting switch located at the amplifier output is closed or 
rendered conductive in order to reflect or present an open circuit to the 
combining point. The arrangement of the Saleh patent has the disadvantages 
that failure of an amplifier and the operation of the switch which 
decouples it from the combining junction results in a change in the 
impedance at the junction, and also that the source impedance of the 
combined output is low and depends upon the number of amplifiers which, at 
the moment, happen to be in operation, and therefore ordinarily requires 
an impedance transformer. Further, the Saleh arrangement is not amenable 
to efficient power reduction. 
A power amplifier of the combining type is desired in which the combining 
is performed in a low-loss manner, which is reliable, and which is capable 
of efficient power reduction. 
SUMMARY OF THE INVENTION 
A composite amplifier according to the invention includes a power divider 
or splitter adapted to be coupled to a source of signal for dividing the 
signal into equal portions. First, second, third, and fourth amplifiers 
have their inputs coupled to the power splitter for receiving one of the 
equal signal portions and for amplifying the equal signal portions for 
producing first, second, third, and fourth amplified signals at first, 
second, third, and fourth output ports. The first and second output ports 
are coupled to a first node point by first and second transmission line 
arrangements having a characteristic impedance equal to a selected 
impedance. Thus, the impedance at the first node point is half the 
selected impedance when both amplifiers are operating. The third and 
fourth output ports are coupled to a second node point by third and fourth 
transmission line arrangements also having the selected characteristic 
impedance. Thus, the second node point has an impedance equal to one half 
the selected characteristic impedance when the third and fourth amplifiers 
are operating. 
Fifth and sixth transmission line arrangements couple the first and second 
nodes, respectively, to a sum port. The fifth and sixth transmission line 
arrangements are each one-quarter wavelength long at a frequency within 
the operating frequency band, and each has the selected characteristic 
impedance. The fifth and sixth transmission line arrangements perform an 
impedance transformation between the first and second nodes, respectively, 
and the sum port, whereby the impedance at the sum port equals the 
selected impedance when all four amplifiers are operating to produce 
maximum output power. 
A controllable switch arrangement includes a first switch coupled to the 
second transmission line arrangement for short-circuiting the second 
transmission line arrangement at a location one-quarter wavelength from 
the first node point. The controllable switch arrangement includes a 
second switch coupled to the third transmission line arrangement for 
short-circuiting the third transmission line arrangement either at the 
second node point, or in some embodiments at a point one-quarter 
wavelength from the second node point in either the third (or equivalently 
at the fourth) transmission line arrangement. When the first and second 
switches are open, all four amplifiers are energized to produce maximum 
output power. When the first and second switches are conductive for 
short-circuiting their associated transmission lines, at least the second 
and third amplifiers are deenergized, whereby the signal is amplified by 
the remaining amplifier or amplifiers with high efficiency and full gain.

DESCRIPTION OF THE INVENTION 
FIG. 1a is a simplified block diagram of a composite amplifier 70 including 
an input port 72 adapted for receiving signal to be amplified and for 
coupling the signal to a power divider 74. Power divider 74 divides the 
signal into a plurality of equal portions. As illustrated in FIG. 1a, 
there are four portions, one of which appears on each of transmission 
lines 76, 78, 80, and 82 for application to input ports of amplifiers 
numbered 1, 2, 3, and 4, respectively. Amplifiers 1-4 amplify the applied 
signal portions and produce amplified signal portions at output ports 11, 
12, 13, and 14, respectively. 
The amplified signal portion produced at output ports 11-14 are applied 
through a tree network of transmission lines to a sum output port 60, from 
which the combined signals are coupled to utilization means (not 
illustrated). The amplified signal portion produced at output port 11 of 
amplifier 1 is applied over a transmission line 21 to a first node point 
31. The amplified signal portion produced at output terminal 12 of 
amplifier 2 is applied over portions 22a and 22b of a transmission line 22 
to first node point 31. Similarly, the amplified signal portion at output 
port 13 of amplifier 3 is applied over a transmission line 23 to a second 
node point 32, and the amplified signal portion produced at output port 14 
of amplifier 4 is applied over a transmission line 24 to second node point 
32. If the phases of the amplified signals at amplifier output ports 11-14 
are equal, and if the path lengths of transmission line pairs 21, 22; 23, 
24 are equal, the signal summations at node points 31 and 32 are 
constructive, and the signal power level at each node point is twice that 
at either amplifier output port. The sum signals at node 31 are applied 
over a transmission line 25 to a sum node or point 60, and the sum signals 
at node point 32 are applied over a transmission line 26 to sum node 60. 
In the embodiment of FIG. 1a, sum node 60 is the output port. With 
constructive addition, the power of the signal at sum node 60 is four 
times the power at the output port of any one of the amplifiers. All the 
transmission lines illustrated in FIGS. 1a and 1b have a characteristic 
impedance which equals a selected impedance. The characteristic impedance 
is often selected to be 50 ohms, but other impedances may be used, such as 
75 ohms. The designation Z.sub.0 may be used to denote the characteristic 
impedance of a transmission line. 
If it is desired to reduce signal power while maintaining efficiency, it 
may be desirable to deenergize certain ones of the amplifiers. For 
example, to achieve quarter-power operation (-6 dB), it may be desirable 
to maintain amplifier 1 in operation and to deenergize amplifiers 2, 3, 
and 4. Each of amplifiers 1, 2, 3 and 4 has an output impedance in the 
energized state which is nominally matched to the impedance of the 
transmission lines. However, the deenergized amplifiers present impedances 
at node points 31, 32, and 60 which perturb the operation of the amplifier 
and which, in general, will result in a power reduction other than that 
desired. In accordance with an aspect of the invention, a first switch 41 
is coupled to transmission line 22 at a node point 51 lying between 
transmission line portions 22a and 22b for short-circuiting transmission 
line 22 at node point 51. Similarly, a second switch 42 is coupled to the 
end of transmission line 23 at node point 32 for short-circuiting 
transmission line 23 at node point 32. In accordance with a further aspect 
of the invention, portion 22b of transmission line 22, and transmission 
line 26, are both selected to have a length which is an odd integer 
multiple of one-quarter wavelength. More formally, the lengths of 
transmission line portion 22b and of transmission line 26 are: 
EQU (2N+1).lambda./4 (1) 
where N is an integer and .lambda. is the wavelength. As illustrated in 
FIG. 1a, switches 41 and 42 are separated from node points 32 and 51 for 
simplicity of illustration, but those skilled in the art will understand 
that switches 41 and 42 are arranged to short-circuit their associated 
transmission lines without excess line length. As an aside, it is noted 
that a switch such as switch 41 can short-circuit its associated 
transmission line when it is conductive, but when nonconductive it does 
not open-circuit the transmission line. 
The operating mode of composite amplifier 70 of FIG. 1a is established by a 
control circuit 90 which includes conductors coupled to amplifiers 1, 2, 
3, and 4 for energization thereof, and also includes control of the 
position of switches 41 and 42, as illustrated by dashed line 92. Dashed 
line 92 may represent mechanical control or electrical control of relay 
windings associated with switch contacts 41 and 42 for remote control 
thereof, or may represent bias control of solid-state switches, all as 
known in the art. 
FIG. 1b is a redrawing of portions of composite amplifier 70 of FIG. 1a for 
convenience in comparing the structure with that of the representations of 
FIGS. 2-5. 
FIG. 1c illustrates details of a possible configuration of control circuit 
90 of FIG. 1a. Elements of FIG. 1c illustrates details of a possible 
designated by the same reference numerals. In FIG. 1c, a power supply 
illustrated as a battery 100 has its negative terminal grounded and its 
positive terminal connected through an on-off switch 102 to the common 
terminal or pole 104 of a single pole, double throw switch 103. The upper 
switched terminal 106 of switch 103 is connected by a conductor 108 to 
energize amplifiers 2, 3, and 4. Lower switched terminal 110 of switch 103 
is connected by a conductor 112 to relay windings 114 and 116 magnetically 
coupled to switches 41 and 42, respectively, for control thereof. Lower 
switched terminal 110 of switch 103 is also connected by the 
anode-to-cathode path of a diode 118 to a conductor 120 for energizing 
amplifier 1. A further diode 122 has its anode connected to conductor 108 
and its cathode connected to conductor 120. 
In operation, control circuit 90 of FIG. 1c enables the composite amplifier 
of FIG. 1a when ON-OFF switch 102 is closed, which allows energizing 
voltage to be applied to the common pole 104 of switch 103. When pole 104 
is in its upper (FULL POWER) position, supply voltage is applied over 
conductor 108 to energize amplifiers 2, 3, and 4. Supply power is also 
applied through the anode-to-cathode path of diode 122 to energize 
amplifier 1. However, diode 118 is reverse-biased by the supply voltage, 
and therefore current is not applied to relay windings 114 and 116. 
Consequently, in the full power operation condition, normally open 
switches 41 and 42 remain open. The difference in amplifier operating 
potentials due to diode 122 is assumed to be negligible, but if it is 
important, other diodes may be used to offset the supply to amplifiers 2, 
3 and 4. 
In the lower position of switch pole 104 (corresponding to QUARTER-POWER 
operation), supply voltage is applied by way of terminal 110 and conductor 
112 to relay windings 114 and 116, resulting in closure of switches 41 and 
42. Supply voltage is also applied by way of the anode-to-cathode path of 
diode 118 and conductor 120 to energize amplifier 1. Amplifiers 2, 3, and 
4 are not energized because diode 122 becomes reverse biased. Thus, in the 
quarter power position of switch 104, amplifier 1 is energized and 
switches 41 and 42 are closed. Thus, the signal power is decreased to 
one-fourth of the full output value, but the energization power is also 
reduced to one-fourth the maximum value (the power consumed by relay 
windings 114 and 116 is considered to be negligible). 
Referring now to FIG. 1b, with switch 41 closed or conductive in the 
quarter-power operating mode, transmission line portion 22a is 
short-circuited one-quarter wavelength from node point 31. As is known to 
those skilled in the art, transmission line portion 22a presents an 
impedance to node point 31 which is very high. This is sometimes known as 
"reflecting" a high impedance to node point 31. Thus, the amplified signal 
portion produced by amplifier 1 at output port 11 flows by way of 
transmission line 21 past node point 31 and onto transmission line 25 
without signal attenuation attributable to deenergized amplifier 2. In the 
quarter-power operating mode, switch 42 is also closed or conductive, 
which short-circuits the ends of transmission line portions 23 and 24 at 
node point 32, one quarter wavelength from sum node 60. Transmission line 
portion 26 therefore reflects an open circuit to sum node 60. The high 
impedance does not perturb the impedance at sum node 60, and therefore the 
amplified signal portion produced by amplifier 1 flows from transmission 
line 25 past sum node 60 to the utilization means. In the quarter-power 
operating mode, amplifier 1 is fully energized, and produces its amplified 
signal portion at output port 11 with full gain. The signal flows from 
output port 11 to sum node 60 without significant loss. 
Thus, the arrangement of FIG. 1 energizes four amplifiers and sums the 
amplified signal portions at their outputs in a full power operating mode, 
and may be switched to a quarter-operating power mode, in which only 
amplifier 1 is energized, and the signal is coupled from its output port 
to the composite output port (sum node 60) without significant loss. Full 
amplifier gain is maintained in either mode. The direct current 
energization power is reduced by a factor of four in conjunction with the 
reduction of signal output power, whereby the efficiency remains 
substantially the same in both operating modes. 
The arrangement of FIG. 2a is similar to that of FIG. 1b. Elements of FIG. 
2a corresponding to those of FIG. 1b are designated by the same reference 
numerals. The arrangement of FIG. 2a differs from that of FIG. 1b in that 
switch 42 connected to node 32 has been deleted, and a switch 242 has been 
connected to transmission line 23 at a node point 252, which divides 
transmission line 23 into portions 23a and 23b. Transmission line portion 
23b has a length of one quarter wavelength, in accordance with equation 1. 
When switches 41 and 242 are closed or conductive, only the amplified 
signal portions from output ports 11 and 14 of amplifiers 1 and 4 are 
coupled to sum node 60. Consequently, the arrangement of FIG. 2a is 
capable of switching between full power operation and half-power (-3 dB) 
operation. The half-power operation, however, results in a mismatch at sum 
node 60. This mismatch may be corrected by an arrangement including a 
further switch 243 serially coupled between sum port 60 and an inductance 
illustrated as an inductor 227, together with a length of transmission 
line 229 one-quarter wavelength long, extending from sum node 60 to a 
further node point 260, and a switch 244 serially connected between node 
point 260 and an inductance illustrated as an inductor 228. As known in 
the art, inductors 227 and 228 may be discrete wound inductors, or may be 
implemented as short-circuited transmission lines. The magnitude of the 
inductive reactance which must be coupled to sum port 60 by switch 243 in 
the half-power mode is Z.sub.0 /2, and the magnitude of the inductive 
reactance which must be coupled to node point 260 by switch 244 is 
Z.sub.0. As known to those in the art, an impedance of +JZ.sub.0 /2 may be 
implemented as a short-circuited S transmission line having a 
characteristic impedance of Z.sub.0 and a length of 0.0735 .lambda.. The 
impedance of +JZ.sub.0 necessary at node point 260 may be achieved by a 
short-circuited transmission line having a characteristic impedance of 
Z.sub.0 and a length of .lambda./8. Node 260 is the output port. 
Control circuit 290 of FIG. 2a differs from control circuit 90 in that it 
controls the arrangement to provide either full-power operation or 
half-power operation. FIG. 2b illustrates details of control circuit 290. 
In FIG. 2b, elements corresponding to those of FIG. 1c are designated by 
the same reference numerals. During full power operation, amplifiers 2 and 
3 are energized directly by way of pole 104, terminal 106, and conductor 
108. Amplifiers 1 and 4 are energized by way of forward biased diode 122 
and conductor 220 from conductor 108. However, during full power 
operation, back based diode 218 prevents any voltage from appearing on 
conductor 212, whereby no power is applied to any of relay windings 214, 
216, 222, or 224, and therefore normally open switches 41, 242, 243, and 
244 remain open. During half-power operation, pole 104 applies power by 
way of terminal 210 and diode 218 to conductor 220 and amplifiers 1 and 4. 
Amplifiers 2 and 3 remain deenergized. Also, power is applied by way of 
terminal 210 to conductor 212, and to windings 214, 216, 222, and 224, 
thereby closing or rendering switches 41, 242, 243, and 244 conductive, 
thereby establishing the conditions required for half-power operation in 
the arrangement of FIG. 2a, together with an impedance match at output 
port 260. 
FIG. 3 illustrates an arrangement similar to that of FIG. 2a, and elements 
of FIG. 3 corresponding to those of FIG. 2a are designated by the same 
reference numerals. FIG. 3 differs from FIG. 2a in that a different 
impedance matching arrangement is used for matching the output port during 
half-power operation. As in the arrangement of FIG. 2a, half-power 
operation is achieved by closing switches 41 and 242. An impedance match 
at sum port 60 is achieved by use of a series connected switch 343 
connected between node point 31 and a capacitance illustrated as a 
discrete or lumped capacitor 327, by a switch 345 serially connected 
between sum port 60 and a capacitance illustrated as a discrete capacitor 
329, and by a switch 344 serially connected between node 32 and a 
capacitance illustrated as a discrete capacitor 328. Switches 343, 344, 
and 345 are closed or conductive during half-power operation to provide an 
impedance match at sum node 60, which is the output port. The capacitive 
reactance required at each node is equal to Z.sub.0. As known, a 
capacitive reactance equal to Z.sub.0 can be implemented by the use of an 
open-circuited transmission line having a length of .lambda./8. Details of 
control circuit 390 required to achieve either full-power or half-power 
operation are believed to be self-evident in view of the preceding 
discussion. 
FIG. 4a is a simplified schematic diagram similar to FIGS. 1b and 2a of an 
arrangement which is adapted for providing full-, half-, or quarter-power 
operation. In FIG. 4b, elements corresponding to those of FIGS. 1c or 2b 
are designated by the same reference numerals. The arrangement of FIG. 4a 
includes switch 42 connected for short circuiting transmission line 23 at 
node point 32, switch 242 arranged for short circuiting transmission line 
23 at node point 252, switch 41 for short circuiting transmission line 22 
at node point 51, and switches 243 and 244 coupled to node points 60 and 
260, respectively, for coupling matching inductances thereto. FIG. 4b 
illustrates details of a control circuit adapted for controlling the 
arrangement of FIG. 4a to full-, half-, or quarter-power modes. 
Full power operation is selected in the control circuit of FIG. 4b by 
connecting pole 104 of switch 103 to full power terminal 106, which 
couples energizing power to amplifiers 2 and 3 by way of conductor 108, 
and which also couples power to amplifier 4 by way of diode 222 and 
conductor 220, and to amplifier 1 by way of diode 122 and conductor 120. 
Diodes 118 and 218 prevent application of any power to relay windings 114, 
116, 216, 224, or 226 during full power operation, therefore all switches 
(41, 42, 242, 244, and 243) remain open. 
For half-power operation, the control circuit of FIG. 4b connects pole 104 
of switch 103 to apply power to terminal 210, which forward biases diode 
218 to energize amplifier 4 by way of conductor 220, and forward biases a 
further diode 418 to apply power to amplifier 1 by way of conductor 120. 
Amplifiers 2 and 3 remain deenergized. Power is also applied during 
half-power operation from terminal 210 by way of a conductor 212 to relay 
windings 216, 224, and 226 for closing switches 242, 243, and 244. Power 
is also applied from terminal 210 of switch 103 by way of conductor 212 
and a further diode 450 to winding 114, for closing switch 41, thereby 
completing the requirements for half-power operation. 
Quarter-power operation of the arrangement of FIG. 4a is achieved in the 
control circuit of FIG. 4b by connecting common pole 104 of switch 103 to 
terminal 110 of switch 104, which energizes amplifier 1 by the way of 
diode 118 and conductor 120. Amplifiers 2, 3, and 4 remain deenergized. 
Power is also applied from terminal 110 directly to relay winding 116 by 
way of conductor 112 for closing switch 42, and power is also applied to 
relay winding 114 by way of diode 452 for closing switch 41. 
FIG. 5 is an extension of the arrangement of FIG. 4a which includes 
redundant amplifiers 505 and S06 fed from transmission lines 576 and 578, 
respectively, and which includes further switch contacts 545, 546, 547, 
548 and 549 which may be selectively closed under control of a control 
circuit 590. The signal amplified by amplifier 505 is coupled by way of a 
transmission line 525 including portions 525a and .lambda./4 portion 525b 
to first node point 31. Similarly, a signal produced at the output of 
amplifier 506 is coupled by way of a transmission line 426 including a 
portion 526a and .lambda./4 portion 526b to second node point 32. Node 
point 31 includes a further grounding switch 549 symmetrical with 
grounding switch 42 connected to node point 32. 
Any one of amplifiers 1, 2, and 505 coupled to node point 31 may serve as a 
redundant amplifier, and any two of the three may be operated for normal 
full-power operation. Similarly, any two of amplifiers 3, 4, or 506 may be 
used as a redundant amplifier, with the other two operating in a full 
power mode. The operating mode of control circuit 590 will depend upon 
which amplifiers are available for operation. The operating principles of 
the arrangement of FIG. 5 will be clear from the preceding discussions. 
FIG. 6a is a perspective view of a cross-section of a printed circuit board 
microstrip transmission line illustrating details of a relay-operated 
grounding switch suitable for operation in conjunction with FIGS. 1-5. In 
FIG. 6a, a flat dielectric plate 610 has its bottom surface coated with a 
conductive ground plane 612. Upper surface 613 of dielectric plate 610 
supports an elongated strip conductor designated generally as 614, which 
includes portions 614a and 614b separated by a through hole 618 which 
extends from the upper surface of strip conductor 614 through dielectric 
plate 610 and ground plane 612. Strip conductor 614 coats with dielectric 
plate 610 and ground plane 612 to form a microstrip transmission line, as 
well known to those skilled in the art. A metallic cap 616 is connected to 
the upper surface of transmission line 614 in conductive contact with 
portions 614a and 614b, and bridging hole 618. Cap 616 is dimensioned to 
resist the forces occasioned by switch actuation. A composite switch 
plunger 620 is slideably mounted within hole 618 and makes conductive 
contact along its sides with grounding springs 622 and 624, which in turn 
make conductive contact with ground plane 612. In addition to providing 
contact with ground, spring contacts 622 and 624 tend to retain plunger 
620 in position. The lowermost portion of composite plunger 620, as 
illustrated in FIG. 6a, is a magnetic portion 620a. Portion 620a may be 
made from soft iron, ferrite, or other magnetic material, while the 
conductive upper portion 620b is made from a nonmagnetic conductive 
material such as copper. A relay coil 626 consisting of conductive 
windings surrounds plunger 620 and has leads 640 adapted to be coupled to 
the control unit of the composite amplifier. FIG. 6b is an elevation view 
of the near face of the arrangement of FIG. 6a. More clearly visible in 
FIG. 6b is a dielectric bushing 628 which may be part of a bobbin or form 
on which windings 626 are wound, and which tends to maintain alignment of 
plunger 620. Also visible in FIG. 6b is a layer of adhesive material 630 
which holds magnetic windings 626 and bushing 628 affixed to the lower 
surface of ground plane 612. 
In operation, current applied to windings 626 tends to attract magnetic 
portion 620a of plunger 620 upward, thereby driving the entire plunger 
upward and causing the conductive tip of plunger 620 to contact cap 616. 
Since cap 616 is in conductive contact with strip conductor 614, and 
plunger 620 makes conductive contact with ground plane 612 by way of 
spring clips 622 and 624, a short, low inductance path is thereby formed 
between strip conductor 614 and ground for effectively grounding the 
microstrip transmission line at the location of hole 618. 
FIG. 6a also illustrates in phantom outline a further elongated conductive 
element 614c which intersects conductors 614b and 614c at or near through 
hole 618. When more than two strip conductors intersect at hole 618, the 
cap is dimensioned to make contact with them, as by extending cap 616 with 
a further portion 616c, as also outlined in phantom. Thus, the grounding 
provided by plunger 620 can occur at the intersection of multiple 
transmission lines. 
Other embodiments of the invention will be apparent to those skilled in the 
art. For example, transmission lines other than microstrip may be used, 
such as stripline, waveguide or coax. Any type of grounding or connecting 
switch may be used, either locally or remotely operated, mechanical or 
solid-state. The amplifiers may be single or multiple-stage, and may 
include phase trimming elements. The control circuit may be an analog, 
locally controlled system such as those illustrated in FIGS. 1c, 2b and 
4b, or it may be remotely controlled by a telemetry or data link, and/or 
implemented by digital logic circuits. While switch 242 has been 
illustrated in FIG. 2a as being connected to transmission line 23, those 
skilled in the art realize that it could alternatively be connected to 
transmission line 24, with an appropriate change of the control circuit to 
deactivate amplifier 4 rather than amplifier 3 when the switch is 
conductive. While equal power division has been described in conjunction 
with power divider 74, those skilled in the art will recognize that some 
inequality is unavoidable.