NTSC/PAL switchable video color decoder using a digital comb filter and method

Apparatus for and a method of decoding a video color signal provides a digital wide band luminance signal. An analog signal V1 representing the low band luminance component of a coded video color input signal Icv, is obtained at a first analog filter 1. An analog signal Yhc representing high band luminance and chrominance components is obtained at second means (5) which subtracts the first analog signal Y1 from a delayed input signal Icv. Signal Yhc is synchronously demodulated to provide (B-Y) and (R-Y) chrominance component signals U.sub.1, V.sub.1, which are digitized. Via digital delay means, single line delayed signals Yd1, U.sub.0, V.sub.0 and double line delayed signals U.sub.-1, V.sub.-1 are obtained. High band luminance signals Yu, Yv are then obtained as a function of at least two of the respective chrominance component signals U.sub.1, U.sub.0 and U.sub.-1 ; and V.sub.-1, V.sub.0 and V.sub.-1. These signals Yu, Yv are remodulated and combined to provide a digital high band luminance component signal Yh. Finally, by digitally summing the digital signals Yd1 and Yh a digital wide band luminance output signal Yd is obtained. Two different functions for obtaining the high band luminance components are exemplified. Wideband chrominance signals are also provided.

This invention relates to a video colour decoder using a digital comb 
filter capable of, and method for, processing signals of more than one 
television standard to provide a digital wide band luminance output 
signal. 
In an embodiment, reference is made to the use of the digital luminance and 
chrominance signals from this decoder in apparatus having a four field 
frame store to provide especially advantageous techniques for motion 
detection and adaptive line interpolation. 
BACKGROUND 
Television signals are currently broadcast and distributed mostly in 
`coded` form, that is to say that the original colour picture comprising 
red, green and blue component signals has been encoded into a single 
composite signal in accordance with the standards of the , NTSC or 
SECAM systems, or their variants. 
These systems were evolved largely on the basis of broadcast requirements, 
and their characteristics were determined by such considerations as 
compatability with existing monochrome broadcast formats. Consequently, 
there are aspects of these systems which do not ideally suit the studio 
environment, where the video signal may be processed by a long chain of 
equipment and the main requirement is that minimal cumulative degradation 
takes place. As much of this equipment is now using digital storage and 
processing techniques, it is appropriate that new video standards have 
been introduced which operate in the `component` domain, that is `RGB` 
(red, green, blue) or `YUV` (matrixed RGB, being the three signals derived 
for use in the `coded` systems), the video signal being transferred 
between items of equipment in digital P.C.M. form. There is therefore a 
need for an interface between the coded analog and a component environment 
to allow conventional analog signals to be further processed in component 
form. 
Although this interface has obviously existed for as long a time as the 
coded system itself, (for example, in a colour television receiver where 
RGB signals are utlimately required for display), there are several levels 
of refinement associated with the decoding process, the more sophisticated 
decoding techniques being devised in an attempt to remove the degradations 
introduced into the decoded component signals as a result of the 
compromises inherent in coded signal standard itself. One of the major 
compromises associated with all existing coded formats is the requirement 
that the coded colour signal is contained within a bandwidth no greater 
than the corresponding monochrome standard, this being achieved by 
modulating the colour information (the U and V components) onto a 
subcarrier situated towards the top of the video bandwidth. This shared 
bandwidth leads to crosstalk between the signal components, the exact 
nature of the crosstalk being characteristic of the coded system in use. 
These effects are generally refered to as cross-colour and cross-luminance, 
an example being the appearance of coloured fringes in areas containing 
high-frequency picture detail (cross-colour). 
The simplest form of decoder attempts to separate the luminance and 
chrominance signal components purely on the basis of their predominant 
frequency bands, the luminance information being regarded as occupying the 
lower part of the spectrum and the chrominance the upper part, and this 
technique is basically that applied in the domestic TV set, where the 
above effects may be observed. 
The more sophisticated systems apply analog comb filter techniques to 
separate the luminance and chrominance components, the difficulty of 
separating signals occupying the same parts of the spectrum being overcome 
by exploiting the redundancy of information contained in a video signal 
when several neighbouring scan lines contain very similar information. It 
must be emphasised that this assumption is fundamental to the operation of 
line-based comb filters, as information theory shows that in the general 
case, the extra chrominance information cannot be introduced as an 
independent signal, without the occurence of crosstalk. 
EXISTING ANALOG COMB FILTER DESIGNS 
There is a well-documented history of the development of analog 
comb-filter decoders which operate in the coded signal domain. These 
generally operate using a three-line `aperture` meaning that three 
consecutive lines of video are processed in such a way as to produce 
separated luminance and chrominance signals which are as close as possible 
to those which are originally coded onto the centre line signal. The 
derivation of the processing methods rests on the frequency relationship 
between the chrominance subcarrier frequency and the line frequency as 
defined in the NTSC and systems, no comb filering of SECAM signals is 
possible. 
Several analog comb-filer decoders are described in an article entitled 
Comb Filter Decoders in the March 1984 issue of "International 
Broadcast Engineer" magazine and an earlier system is described in article 
entitled Colour Picture Improvement Using Simple Analog Comb Filters 
in Vol. 87 (October 1978) of the SMPTE Journal. 
Although the systems described in these articles can give good results, 
there are several respects in which an alternative approach is desirable: 
(a) In order to gain simultaneous access to the three lines of video for 
processing, an input signal must be passed through two delays each of 
exactly one line period (or as precisely required by the particular 
design). This is generally achieved by the use of glass delay lines, or 
more latterly charge-coupled devices, neither of which possess entirely 
satisfactory characteristics. In particular, the glass delays tend to have 
a dependence of delay and other parameters on temperature, exhibt multiple 
reflections and have to be driven by a signal modulated onto a carrier 
which then must be demodulated after amplification at the receiving end. 
As the centre line of the three has been once delayed, the main signal path 
suffers the degradation associated with the passage through the filter 
performance may be difficult to maintain over a period of time. 
(b) As may be judged by the descriptions of the existing designs, there is 
considerable analog processing devoted to filtering and matching various 
signal paths to achieve comb filter operation. This demands a lengthy 
set-up procedure and great stability in the analog circuitry and would 
present a considerable challenge in trying to achieve a system capable of 
comb-filtering both NTSC and signals, where the phase relationships 
and line periods are both different. 
(c) It is generally recognised that where signals are to be delayed by a 
substantial period, the preferred method is a digital process, this being 
particularly desirable when the analog signal has to be digitalised in any 
case for subsequent storage in a digital frame store. As digital devices 
have come cheaper and more complex, digital signal processing has also 
become more attractive in relation to the equivalent analog methods, major 
advantages being ease of automatic testing and fault location, minimal 
set-up effort and long-term stability. There is an additional advantage in 
relation to the proposed application in that the delay required for the 
NTSC and comb filters can be precisely obtained by virtue of the 
number of clock periods assigned to the line periods of the two systems. 
According to the present invention there is provided a video colour decoder 
using a digital comb filter, comprising 
first means for deriving a first analog signal Y1 representing low band 
luminance component of a video colour input signal Icv; 
second means for deriving a second analog signal Yhc representing high band 
luminance and chrominance components of the video colour input signal Icv; 
third means to enable synchronous demodulation of the second analog signal 
Yhc to provide a B-Y chrominance component signal U.sub.1 and an R-Y 
chrominance component signal V.sub.1 (where B and R is the conventional 
notation for the respective RGB component signals); 
first analog to digital converter (ADC) means for digitising the first 
analog signal Y1 to produce the equivalent digital signal Yd1; 
second analog to digital converter (ADC) means for digitising the (B-Y) and 
(R-Y) chrominance component signals U.sub.1 V.sub.1 ; 
first digital delay means for providing a single line delay of the first 
digital signal Yd1; 
second digital delays means for providing a single line delay of the 
chrominance component signals U.sub.l V.sub.1 with a delay period of one 
line, the chrominance component signals so delayed by one line period 
being designated U.sub.0,V 0; third digital delay means for providing a 
further single line delay of the chrominance signals U.sub.0,V.sub.0 with 
a further delay period of one line, the chrominance component signals so 
delayed by one further line period being designated U.sub.-1, V.sub.-1 ; 
summing means arranged to reject consistent chrominance information and 
operable to provide a high band luminance signal Yu as a function of at 
least two of the respective chrominance component signals U.sub.1, U.sub.0 
and U.sub.-1 ; 
summing means arranged to reject consistent chrominance information and 
operable to provide a high band luminance signal Yv as a function of at 
least two of the respective chrominance component signals V.sub.1,V.sub.0 
and V.sub.-1 ; 
means for generating digital sinewave and cosinewave reference signals 
Uref, Vref representing the sine wave of the coded (B-Y) subcarrier 
reference phase signal and the cosinewave of the coded (R-Y) subcarrier 
reference phase signal; 
digital means for obtaining the products of the signals Yu,Yv and their 
respective sinewave and cosinewave reference signals Uref,Vref; 
means for summing these products to provide a digital high band luminance 
component signal Yh; 
and digital means for summing the digital signal Yd1 and the digital signal 
Yh to provide a digital wide band luminance output signal Yd. 
In a preferred embodiment said digital high band luminance signals (Yu,Yv) 
are obtained as a function of the respective chrominance component signals 
of three video lines represented by the undelayed chrominance components 
(U.sub.1, V.sub.1) the one line delayed chrominance components U.sub.0, 
V.sub.0 and the two line delayed chrominance components (U.sub.-1, 
V.sub.-1): said function requiring half the sum of the undelayed 
chrominance component signals (U.sub.1, V.sub.1) and the twice delayed 
chrominance component signals (U.sub.-1, V.sub.-1) to be subtracted from 
the once delayed chrominance component signals (U.sub.0, V.sub.0). This 
embodiment is employed to process signals of more than one television 
standard, namely the NTSC and standards. 
In the preferred embodiment the decoder receives an input signal selected 
from the NTSC and video signal systems, and said first and second 
means for deriving said first and second analog signals Y1 and Yhc are 
switchable for operation in the respective NTSC and mode. 
In the preferred embodiment there is provided an automatic gain control 
(AGC) loop with the third means and to provide signal scaling means 
(receiving a signal representing an inversion of gain in AGC loop) for 
scaling the digitised high band luminance frequency signal Yh. 
It is a preferred feature to provide means for detecting residual 
chrominance components in the high band luminance signal Yh and comparing 
these residual components with two threshold levels, said detecting means 
being operable to generate a control signal for controlling a data 
selector, said data selector being capable of operation on the high band 
luminance signal Yh in one of three modes, said data selector being 
operable in a first said mode to pass this luminance signal, said data 
selector being operable in a second said mode to halve the amplitude of 
this luminance signal, said data selector being operable in said third 
said mode to suppress this luminance signal; said first mode corresponding 
to those residual components being below the first threshold, said second 
mode corresponding to those residual components being between the 
thresholds, and said third mode corresponding to those residual components 
being above the second threshold. 
In a modified embodiment which is used as an alternative mode of processing 
system input signals said digital high band luminance signals (Yu,Yv) 
are obtained as a function of the respective chrominance component signals 
of two video lines represented by the undelayed chrominance components 
(U.sub.1, V.sub.1) and the two line delayed chrominance components 
(U.sub.-1, V.sub.-1); said function requiring half the difference of the 
twice delayed chrominance component signals (U.sub.-1, V.sub.-1) and the 
undelayed chrominance component signals (U.sub.1, V.sub.1). 
According to a further aspect of the present invention there is provided a 
method of decoding a video colour signal to provide a digital wide band 
luminance signal comprising the steps of: 
(a) deriving a first analog signal Y1 representing low band luminance 
component of a video colour input signal Icv, by passing the video colour 
input signal Icv through a first analog filter means; 
(b) deriving a second analog signal Yhc representing high band luminance 
and chrominance components of the video colour input signal Icv by passing 
the colour input signal Icv through second means which subtracts the first 
analog signal Y1 from a correspondingly delayed input signal Icv; 
(c) synchronously demodulating the second analog signal Yhc to provide a 
(B-Y) chrominance component signal U.sub.1 and an (R-Y) chrominance 
component signal V.sub.1 (where B and R is the conventional notation for 
the respective RGB component signals); 
(d) employing first analog to digital converter (ADC) means to digitise the 
first analog signal Y1 thereby to produce the equivalent digital signal 
Yd1; 
(e) employing second analog to digital converter (ADC) means to digitise 
the (B-Y) and (R-Y) chrominance component signals U.sub.1,V.sub.1 ; 
(f) employing first digital delay means to provide a single line delay of 
the first digital signal Yd1; 
(g) employing digital delay means to provide a single line delay of the 
chrominance component signals U.sub.1, V.sub.1 with a delay period of one 
line, the chrominance component signals so delayed by one line period 
being designated U.sub.0,V.sub.0 ; 
(h) employing third digital delay means to provide a further single line 
delay of the chrominance signals U.sub.0,V.sub.0 with a further delay 
period of one line, the chrominance component signals so delayed by one 
further line period being designated U.sub.-1, V.sub.-1 ; 
(i) employing summing means arranged to reject consistent chrominance 
information and operable to provide a high band luminance signal Yu as a 
function of at least two of the respective chrominance component signals 
U.sub.1,U.sub.0 and U.sub.-1 ; 
(j) employing summing means arranged to reject consistant chrominance 
information and operable to provide a high band luminance signal Yv as a 
function of at least two of the respective chrominance component signals 
V.sub.1,V.sub.0 and V.sub.-1 ; 
(k) generating digital sinewave and cosinewave reference signals Uref, Vref 
representing the sine wave of the coded (B-Y) subcarrier reference phase 
signal and the cosine wave of the coded (R-Y) subcarrier reference phase 
signal; 
(l) obtaining the products of the signals Yu,Yv and their respective 
sinewave and cosinewave reference signals Uref,Vref; 
(m) summing these products to provide a digital high band luminance 
component signal Yh; 
(n) and, digitally summing the digital signal Yd1 and the digital signal Yh 
to provide a digital wide band luminance output signal Yd. 
In a preferred method in steps (i) and (j) said digital high band luminance 
signals (Yu,Yv) are obtained as a function of the respective chrominance 
component signals of three video lines represented by the undelayed 
chrominance component (U.sub.1,V.sub.1) the one line delayed chrominance 
components U.sub.0, V.sub.0 and the two line delayed chrominance 
components (U.sub.-1 , V.sub.-1); said function requiring half the sum of 
the undelayed chrominance component signals (U.sub.1, V.sub.1) and the 
twice delayed chrominance component signals (U.sub.-1, V.sub.-1) to be 
subtracted from the once delayed chrominance component signals (U.sub.0, 
V.sub.0). 
Alternatively for processing signals, in steps (i) and (j) said digital 
high band luminance signals (Yu,Yv) are obtained as a function of the 
respective chrominance component signals of two video lines represented by 
the undelayed chrominance components (U.sub.1,V.sub.1) and the two line 
delayed chrominance components (U.sub.-1,V.sub.-1); said function 
requiring half the difference of the twice delayed chrominance component 
signals (U.sub.-1,V.sub.-1) and the undelayed chrominance component 
signals (U.sub.1,V.sub.1).

DESCRIPTION OF PREFERRED EMBODIMENT 
Referring to FIG. 1, there is shown schematically a circuit for a video 
colour decodeer. A or NTSC coded video colour signal Icv (which has 
already been band-limited to exclude out-of-band noise) is passed through 
a phase-compensated low-pass filter, the cut-off frequency of which is 
chosen to divide the video spectrum into the predominant regions of 
luminance and chrominance energy. As the and NTSC systems use 
differing subcarrier frequencies, it is necessary to use a cut-off 
frequency suited to each system in two switchable filters. 
The input signal Icv is also passed through a wideband analog delay line 2 
which has a delay equal to the delay of the low-pass filter 1. As the NTSC 
filter exhibits a greater delay than the filter, an extra compensating 
(analog) delay must be introduced in delay line 2 when operating the the 
NTSC mode. 
The output signal Y1, which represents the low band luminance component of 
signal Icv, of the low-pass filter 1 contains the lower frequency part of 
the luminance spectrum and very little chrominance information. This 
signal Y1 is digitised at an analog to digital converter (ADC) 3 into 
8-bit PCM form. Prior to digitising, signal Y1 has been subjected to a 
small analog delay at a compensating analog delay Trim 4. This additional 
delay at analog delay trim 4 is necessary in order that the delay of the 
luminance signal Y1 will match the exact delay in chrominance channels U, 
V (to be described), as subsequent digital delays can only introduce 
delays which are integer multiples of the 13.5 MHz sampling period. 
Signal Y1 from filter 1 and signal Icv from delay line 2 are input to a 
differential amplifier 5 where they are subtracted to produce a signal 
Yhc. This signal Yhc represents that part of the spectrum of signal Ivc 
which is complementary to signal Y1 and which contains the chrominance and 
high-frequency luminance information. It may be seen at this stage that, 
adding this signal Yhc to the output signal Y1 of the low-pass filter 1, 
reproduces the original full-band coded signal Icv. 
The output signal Vhc from the differential amplifier 5 is used to feed a 
burst-locked oscillator 6 and synchronous demodulators 7,8 forming an NTSC 
or `simple` decoder. As will be appreciated by those skilled in the 
art, oscillator 6 comprises quad oscillators at the or NTSC subcarrier 
frequencies of 4.43 MHz or 3.58 MHz. Signal Yhc from amplifier 5 is input 
to oscillator 6 and employed as a gated reference burst signal. Likewise 
oscillator 6 outputs a burst locked reference signal B. Synchronous 
demodulation is obtained by using four times subcarrier frequency 
reference oscillators (with digital division to reduce set-up effort) to 
output reference signals F and G for the (B-Y) and (R-Y) demodulators 7, 
8. A system microprocessor (not shown) is employed to generate a digital 
phase control signal P whereby the signal phase is corrected to obtain 
correct decoder reference axes F, G in all modes of operation of the 
system. The system microprocessor is employed to facilitate switching 
between the and NTSC modes of the decoder by means of signal X. 
When operating in mode, the R-Y reference axis is switched in sympathy 
with the burst phase. Signals U and V representing the R-Y and B-Y outputs 
from demodulators 7,8 are filtered at low-pass filters 9,10 to remove the 
high-order demodulation components. These B-Y and R-Y signals U, V are 
then digitised in analog to digital converters (ADC) 11, 12, each 
operating as 8-bit ADCs at the same sampling rate as the luminance channel 
(13.5 MHz) to provide signals U.sub.1, V.sub.1. 
Prior to demodulation, the `highband` signal Vhc is passed through a 
highband automatic gain-controlled (AGC) amplifier 13. The gain control 
signal for the AGC amplifier 13 is derived from ADC's 11,12. This use of 
the digitised R-Y and B-Y signals U.sub.1, V.sub.1 establishes an AGC 
loop. This system allows optimum resolution to be obtained in the 
high-band channel under typical signal conditions while allowing high 
chrominance amplitudes, such as 100% colour bars to be handled. The 
original signal amplitude can subsequently be re-established by 
multiplication by a scaling factor S, which is derived in the AGC loop by 
a measurement of the reciprocal of the gain at amplifier 13. This value is 
digitised to an unsigned 8-bit value in a slow ADC (not shown). The AGC 
system allows the gain to be increased to twice that value appropriate for 
100% colour bars. 
The digitised R-Y and B-Y signals U.sub.1 and V.sub.1 from ADC's 11, 12 are 
delayed digitally at digital delay lines 15, 16 by one line period to 
obtain a `centre` line signals U.sub.0, V.sub.0 and further delayed 
digitally at digital delay lines 17, 18 by another period of one line to 
provide third line signals U.sub.-1, V.sub.-1 for a processing aperture. 
Likewise, the digitised signal Yd1 from ADC 3 is delayed at digital delay 
line 14 by one line period. This contributes to a centre line low-band 
luminance. signal Y.sub.0. 
For the purposes of arithmetic processing: 
Let the undelayed B-Y signal be represented as U.sub.1 
Let the undelayed R-Y signal be represented as V.sub.1 
Let the once-delayed B-Y signal be represented as U.sub.0 
Let the once-delayed R-Y signal be represented as V.sub.0 
Let the twice-delayed B-Y signal be represented as U.sub.-1 
Let the twice-delayed R-Y signal be represented as V.sub.-1 
Let the once-delayed Y signal be represented as Y.sub.0 
Processing is provided at summing untis 23 and 24 to derive digital 
highband luminance signals Yu and Yv. The processing gives a signal: 
EQU Yu=U.sub.0 -(U.sub.-1 +U.sub.1)/2 
and 
EQU Yv=V.sub.0 -(V.sub.-1 +V.sub.1)/2. 
A generator 19 is provided for generating sine and cosine signals Uref, 
Vref corresponding to the B-Y and R-Y reference axes. It comprises a pair 
of 2K.times.12 bit ROMs and it receives signal L2 comprising addresses 
from a counter system generated at system sample rate. A phase comparator 
20 locks the B-Y axis frequency and phase signal Uref so derived from 
generator 19 to that of the burst-locked reference oscillator 6 from which 
it receives the burst locked reference signal B. Phase comparator 20 then 
controls the frequency of the video clock output (EBU clock) VCO 21 
generating the 13.5 MHz sample rate, signal E. This enables the line 
delays at delays 15-18 to be controlled very accurately since locking the 
sample rate to the subcarrier frequency rather than line frequency, gives 
far greater stability, particularly when locking to a noisy input signal. 
Line phase locking of all timing signals is, however, still required. 
Initially this is achieved by allowing the 13.5 MHz VCO 21 to be 
controlled by a signal L1 from a line phase comparator (not shown). 
Initially, a switch 22 is set to its first position P1 which couples the 
line phase comparator signal L1 with the VCO 21 and initiates line phase 
locking in the P1 position. When this line phase lock is achieved, the 
subcarrier frequency locking mode (previously described) is entered by 
placing the switch 22 in the P2 position. When in this mode, very fine 
line phase adjustments may be made. This is achieved by inputting a signal 
L2 representing `fine line phase adjustment` to generator 19. In practice 
this involves altering address increments, made to the look-up ROM's 
contained within generator 19, for a period in order to alter the phase of 
the sine waves relative to the line phase. Since these synthesised 
reference signals are locked to the reference subcarrier, the line phase 
must change. This is done automatically to maintain lock, by using a 
signal L2 derived from the line phase comparator (not shown) which causes 
a small phase adjustment to be made, if necessary, at the start of each 
field based on an averged line phase measurement made during the previous 
field. If the line phase error exceeds a preset limit, however, the lock 
mode reverts to simple line phase control until the error has been brought 
within this limit, whereupon fine control is re-established. 
The digital value representing the instantaneous value of the B-Y reference 
sinewave Uref. is multiplied at multiplier 25 by the quantity: 
EQU Yu=U.sub.0 -(U.sub.-1 +U.sub.1)/2 
and the corresponding digital value of the R-Y reference sinewave Vref. is 
multiplied at multiplier 26 by the quantity: 
EQU Yv=V.sub.0 -(V.sub.-1 +V.sub.1)/2. 
Multipliers 25, 26 are two 12.times.12 bit signed multipliers. The products 
of the operations in multipliers 25, 26 are summed in unit 27. It may be 
easily seen that, if the three lines in the aperture contain the same Y, 
U, V component information prior to coding, then the signals from 
multipliers 25, 26 will contain no U or V components. In the event the sum 
of the contributions from the outer two lines (U1, U-1) (V1, V-1), 
resulting from the presence of high-band luminance, will cancel leaving 
only the contribution (U,V) from the centre line itself. The overall 
process them simplifies to one of `demodulating` all high-band Y 
information in terms of two orthogonal decoding axes, Uref, Vre, and 
subsequently `remodulating`, this time digitally, upon the two same axes, 
using the multipliers 25, 26. It may be shown that the signal so 
reconstructed is the high-band luminance signal Yh, but with all 
consistent chrominance information removed. 
In the case of NTSC, the net contribution from the U and V components is 
still zero, but further analysis shows that high-band luminance Yh is 
contributed from all three lines, resulting in a doubling of the 
reconstructed high-band luminance amplitude. Compensating divide-by-2 
circuits 29,30 are inserted at the points shown when operating in NTSC 
mode. 
The resulting high-band luminance signal Yh, in either mode, is rescaled in 
a third multiplier 28 to compensate for the input AGC system: the scaling 
input level signal for multiplier 28 is obtained from the AGC amplifier 13 
(as previously mentioned). The signal is then passed through a data 
selector 32. Selector 32 can (a) pass the signal unchanged, (b) pass the 
signal shifted down one bit, i.e. at a gain of one half, or (c) pass a 
zero output. Following selector 32, the Yh signal passes to unit 33 where 
it is finally added back into the similarly delayed low-band luminance 
signal Y1 to obtain a wide-band comb-filtered luminance signal Y, with a 
controllable proportion of the high-band range present. 
By using the scaling signal S the chrominance signals U.sub.0 and V.sub.0 
(one line delayed) from digital delay means 15, 16 are digitally combined 
at multiplexer 35 to give signal Z. Signal Z is rescaled at multiplier 36 
to provide the chrominance signal for subsequent processing (e.g. motion 
detection and standards conversion by way of adaptive interpolation). 
Multiplier 36 (like multiplier 28 for the luminance signal) receives a 
level scaling signal S which is derived from the amplifier 13 in the input 
AGC loop, for rescaling the chrominance signal. 
Reference has been made to adaptation of the system as between NTSC of 
input signals, e.g. at filter 1, wide band delay 2, burst locked 
oscillator 6 and compensating divide-by-two circuits 29, 30. By these 
means, processing for NTSC or video signals can be readily selected. 
It will be appreciated that the comb filter decoder described with 
reference to FIG. 1 may form the first stage of apparatus for digitally 
processing video signals. In particular it is advantageously incorporated 
in apparatus which includes a frame store for four fields. Reference will 
be made below to the advantageous manner in which the digital luminance 
signal Y and chrominance signal U, V outputs from this decoder may be 
employed in motion detection and in adaptive interpolation (e.g. for 
standards conversion) in apparatus using a four field store. 
Analyses of the performance of the system when presented with dissimilar 
information over the three-line aperture must take into account the 
effects of differing luminance and chrominance components. 
A. Differing Luminance (Y) Components 
In the NTSC mode, the high band Y may be shown to be constructed from 
contributions of 25%, 50%, 25% across the three-line aperture. This is 
equivalent to typical analog designs and exhibits some zeroes when 
processing certain high diagonal frequencies (e.g. a frequency which 
shifts its phase by 180 degrees from one scan line to the next). 
In the first mode, the contributions from the outer lines to high-band 
Y may be shown to undergo a transformation equivalent to that produced by 
the ` modifier` system described in the previously mentioned prior art 
references. The net result is that the interfering `alias` signals reverse 
their phase every two fields, which implies that if the wideband Y signal 
could be averaged over two fields of the same type, the aliases would 
cancel leaving only the original Y signal presented to the coder (i.e. 
perfect comb filtering). 
The above-described decoder is to be incorporated in apparatus providing a 
frame store for four fields. When using four fields of storage, it is 
possible to exploit this characteristic (for the elimination of aliases). 
This is implemented by ensuring that, when processing stationary pictures, 
equal contributions to a particular picture line are received from two 
consecutive frames. 
This requires that a motion detection system be used to allow adaptive data 
manipulation. 
B. Differing Chrominance (U,V) Components 
If the coded U, V information is substantially different over the three 
lines, the cancellation of the U, V contributions to the bracketed 
quantities will fail, just as is the case with the analog design 
configurations. The occurrence of this may be detected by passing these 
Signals U, V (from units 23, 24) through unit 34. Unit 34 firstly 
comprises low-pass filters (averagers) to remove the effect of high 
luminance frequencies, (other than those very close to subcarrier 
frequency). Unit 34 further comprises comparator means for assessing the 
absolute value of the residual chrominance signals against two thresholds. 
These comparator means within unit 34 provide a control signal C2 to data 
selector 32. This control signal C2, in response to the residual 
chrominance signals (U,V) exceeding a first lower threshold, will switch 
data selector 32 to the .times.1/2 mode and will cause the recombination 
of Yh to occur at half the normal gain level. Likewise, this control 
signal in response to the residual chrominance signal (U,V) exceeding the 
second threshold will switch the data selector 32 to the X0 (times zero) 
mode and will cause the Yh to be completely supressed. This corresponds to 
the introduction of a notch or low-pass filter in the analog designs, when 
comb failure is detected. 
Motion Detection 
As already described, the characteristics of the comb filter can be 
considerably enhanced by subsequent motion adaptive processing. In the 
case, however, the detection of motion from the comb-filtered Y signal is 
made difficult in some situations by the presence of the alias components 
which tend to indicate the presence of motion in a stationary scene 
containing diagonal frequencies, because of the phase reversal of these 
components every two fields. 
For the purpose of subsequent motion detection (in the above mentioned 
apparatus providing a four field frame store), an output of the Y1 signal 
is made available. When in the mode, motion detection is carried out 
by analysis of the Y1, U and V data streams from all four stored fields as 
the data is being read out of the field store. Y1 values from two similar 
fields are compared (both pairs of fields are analysed) and the high-band 
part of the signal is analysed by comparing values of (U+V) or (U-V) from 
two similar picture lines. This quantity may be shown to be stationary in 
the general case of a stationary picture, the +/- decision being dependent 
on the state of the switch in the lines in question. 
Chrominance Comb-Filtering 
As the intial application of this filtering system is to a device which has 
to incorporate a line interpolation system having access to four 
consecutive field lines, it is relatively straightforward to modify the 
chrominance interpolation system to provide chrominance signals which are 
free from cross-colour for vertical luminance frequencies (chrominance 
comb filtering). 
FIG. 2 shows the interpolation aperture applied to the U and V signals over 
four input lines to synthesise an output line at position X. Due to the 
symmetry of the function, it can be shown that without any further 
additions, NTSC comb filtering of the U and V signals is achieved by this 
function. 
input, however, requires further processing due to the more complex 
sequence of U and V axis rotations from line to line (see FIG. 4 for a 
four-line axis sequence diagram). 
As a result of this sequence, the signals in the U and V channels produced 
by a highband luminance vertical frequency are generated with four 
different phases characteristic of the type of line during which they 
were generated. For example, the signal produced in either channel during 
a Type A line is in antiphase with that produced in the same channel 
during a Type C line. 
The net signal introduced into either channel can be seen to be dependent 
upon the difference between the interpolation coefficients assigned to the 
antiphase A and C lines and the antiphase B and D lines. It may also be 
seen that for each phase of crosstalk introduced into the U channel, there 
is another line where the same phase of crosstalk is being introduced into 
the V channel, this being either the line before or the line after 
(considering the A,B,C,D sequence to be continued). It follows that there 
is an aperture function which can easily be derived from the coefficients 
of the function in FIG. 2 which will cancel the cross-colour when applied 
to the four lines of the opposite channel, thereby introducing U-into-V 
and V-into-U crosstalk. 
The appropriate crosstalk aperture function for the arrangement of line 
coefficients shown in FIG. 2 and the four-line sequence illustrated is 
shown in FIG. 3. Note that the average value of the coefficients is zero, 
so that there is no overall D.C. crosstalk between U and V channels, while 
the cancellation of Y-into-U and Y-into-V crosstalk (cross-colour) is 
effected by the difference between the coefficients assigned to the 
antiphase line pairs. 
The crosstalk between U and V channels is easily introduced due to the fact 
that the U and V data are interleaved in a time-shared multiplex fashion 
in the same processing channel. The crosstalk is introduced only when 
there is a measurable difference between the Y1 samples and the 
corresponding comb-filtered wide-band Y samples. In the case where the two 
sets of samples are either identical or very similar, the implication is 
that either the comb fail detector 34 has caused the data selector 32 to 
cut off Yh due to dissimilar chrominance information being present across 
the aperture, or that there is virtually no Yh present in the input 
signal. If the former is true, no U-into-V or V-into-U crosstalk is 
desirable; if the latter, none is necessary. 
Alternative luminance comb filter mode 
It is possible to derive alternative expressions to be substituted in place 
of those shown for Yu and Yv, which retain the characteristic of rejecting 
consistent chrominance information, but are derived from the outer two 
lines of the three only. This substitution is made available as an 
alternative mode for input, and corresponds to an equivalent two-line 
configuration in the coded signal domain. Although, by using this 
configuration, the luminance response is considerably degraded in that 
there is cancellation of certain diagonal frequencies, and other forms of 
distortion which are not present in the three-line case, this may be 
preferable to the incomplete cancellation of consistent chrominance which 
can occur due to differential phase distortion in the input signal when 
processed accordingly to the original system. 
The expressions for Yu and Yv are modified as follows: 
EQU Yu=(U.sub.-1 -U.sub.1)/2 Yv=(V.sub.-1 -V.sub.1)/2 
The change may be implemented simply by disabling the sources of the centre 
line contributions Uo, Vo and inverting the sign of the U.sub.-1 and 
V.sub.-1 contributions. However, as the phases of the U and V axis 
components on the centre line are not consistent with the corresponding 
outer line phases, it also becomes necessary to advance the phase of the 
digital U and V reference sine wave generators by 90.degree., and invert 
the sense of the switch signal feeding them. 
This alternative process may be shown to be equivalent to a comb filter 
configuration in the coded signal domain wherein high-band luminance is 
derived from the averaged value of the two outer lines. As may easily be 
shown, this configuration produces no deviation from the original 
configuration when presented with vertical high frequencies (which are 
consistent from line to line), but as the high frequencies deviate from 
being vertical, cancellation will increasingly occur with a null present 
at 90.degree. per line, and at greater deviations, the luminance high-band 
components reappear in inverted form. The amplitude vs angle from 
verticality function is a cosine, and for this reason, the equivalent 
`coded domain` design is sometimes termed a `two-line cosine comb filter`. 
It is desirable, when using this configuration, to extend the adaptation 
function based on comb filter failure, to detect the occurrence of 
non-vertical high luminance frequencies which exceed the 90.degree. per 
line null, so that inverted luminance is suppressed. 
It is also particularly desirable that this be done in the case of to 
NTSC conversion, as the maximum occurring at 180.degree. per line 
corresponds exactly with the NTSC subcarrier to line phase relationship, 
and the presence of these luminance components, while not conveying any 
useful information, due to their inversion, does cause considerable 
cross-colour effects in the NTSC format. 
In order to extend the failure detection system to take account of this 
requirement, the inputs to the digital L.P.F.'s (34) are processed prior 
to being input, by an arrangement which, in conjunction with the existing 
processing, allows the correllation of luminance signal phase across the 
three-line aperture to be assessed, and produces the control signals 
required by the data selector (32) to produce a smooth transition between 
the various attenuation levels, as already described. 
The `comb failure` adaptation logic in the original mode of operation has 
also been enhanced in that account is now taken of the outputs of unit 
(34) over several lines at the corresponding point in the Iine, in order 
to differentiate between inconsistent chrominance and high luminance 
frequencies which are close to the chrominance subcarrier frequency. This 
involves the addition of two further single-bit one line delays.