Current sensing of switching power regulators

Apparatus and methods for current sensing in switching regulators include a current sensing circuit to sense current of a power stage of a power converter. The power converter can include first and second transistors. The current sensing circuit comprises a transistor that is a scaled version of one of the transistors of the power converter. A circuit of the current sensing circuit matches a drain-to-source voltage of the transistor of the current sensing circuit to the corresponding transistor of the power converter. A current mirror generates a current that mirrors the current flowing through the transistor of the current sensing circuit. A first resistor converts the mirrored current to a current sensed signal.

BACKGROUND

Field

Embodiments of the invention relate to electronic devices, and more particularly, to switching regulators.

Description of the Related Technology

A switching regulator can be used to generate a regulated voltage by controlling a current provided to a load through an inductor. For example, the switching regulator can include one or more switches that are turned on and off. The duty cycle and/or timing of the switches can control the voltage level of the regulated voltage. Examples of switching regulators include, for example, buck converters and boost converters.

A switching regulator can include current sensing or observation circuitry to sense the current through one or more of the regulator's switches, thereby sensing the current provided to the load through the inductor. The sensed switch current can be used to provide enhanced control over switching operations in the regulator.

There is a need for improved current sensing circuits in switching regulators.

SUMMARY

The systems, methods, and devices of the invention each have several aspects, no single one of which is solely responsible for its desirable attributes. Without limiting the scope of this invention as expressed by the claims which follow, some features will now be discussed briefly. After considering this discussion, and particularly after reading the section titled “Detailed Description,” one will understand how the features of this invention provide advantages that include improving power efficiency of power regulators by reducing power losses of current sensing circuits.

In one embodiment, an apparatus is disclosed. The apparatus comprises a current sensing circuit configured to sense current of a power stage of a power converter comprising a first transistor and a second transistor. The current sensing circuit comprises a third transistor having a gate, a drain, and a source. The gate of the third transistor is operatively coupled to a gate of the second transistor. The third transistor is a scaled version of the second transistor. The current sensing circuit further comprises a first circuit configured to match a drain-to-source voltage of the third transistor with a drain-to-source voltage of the second transistor. The current sensing circuit further comprises a current mirror configured to mirror current flowing through the third transistor. The current sensing circuit further comprises a first resistor configured to convert the mirrored current to a current sensed signal.

In another embodiment, a method of sensing a current of a power converter is disclosed. The method comprises providing a gate of a sensing transistor with a voltage corresponding to a gate voltage of a power transistor of the power converter. The sensing transistor is a scaled version of the power transistor. The method further comprises controlling a drain-to-source voltage of the sensing transistor to match a drain-to-source voltage of the power transistor. The method further comprises generating an output current based at least partly on current flowing through the sensing transistor by using a current mirror such that the output current is a scaled version of the current flowing through the sensing transistor. The method further comprises converting the output current to a current sensed signal by using a first resistor.

In another embodiment, an apparatus for sensing a current of a power converter is disclosed. The apparatus comprises means for providing a gate of a sensing transistor a voltage corresponding to a gate voltage of a power transistor of the power converter. The sensing transistor is a scaled version of the power transistor. The apparatus further comprises means for controlling a drain-to-source voltage of the sensing transistor to match a drain-to-source voltage of the power transistor. The apparatus further comprises means for generating an output current based at least partly on current flowing through the sensing transistor, wherein the output-current generating means includes a current mirror such that the output current is a scaled version of the current flowing through the sensing transistor. The apparatus further comprises means for converting the output current to a current sensed signal.

DETAILED DESCRIPTION OF EMBODIMENTS

Embodiments are described in the context of systems and methods for sensing drive or output current of buck regulators, but will be applicable to other types of switching regulators, such as boost and buck-boost converters. In one embodiment, a current sensing circuit senses current of a power stage of a power converter, such as a buck regulator. For example, the power converter can include a high-side power transistor or switch and a low-side power transistor or switch that are independently switched on and off to convert an input voltage to a regulated output voltage. The current sensing circuit includes a sensing transistor that is a (scaled) version of one of the power transistors, and includes a voltage matching circuit. The matching circuit controls a voltage of a terminal of the sensing transistor so that the voltages of the sensing transistor match the corresponding power transistor. Because the sensing transistor is a version of the power transistor, the current flowing through the sensing transistor is a version of the current flowing through the power transistor. Matching can be done with same polarity or reverse polarity to control the direction of the sensing transistor current. The sensing circuit can further include a current mirror to generate a mirrored current of the sensing transistor current. In some embodiments, a current sensed signal can be generated by passing the mirrored current through a resistor to generate a voltage that is related to the switch current.

In some embodiments, the current sensing circuit can have reduced power losses and improved efficiency. For example, one drawback of certain current sensing circuits is power loss due to, in part, using inefficient power supplies to power, supply, or bias the current sensing circuit. For example, high current buck switching regulators (such as point of load regulators) can use a power metal oxide semiconductor field effect transistor (MOSFET) having a low on-resistance RONas the low-side power switch. The sensing transistor can be sized N times smaller than the power MOSFET. But if the input voltage directly or indirectly (through a linear regulator) supplies current to the sensing transistor, power dissipation can be large even though the current of sensing transistor is only a fraction of the power MOSFET current. For example, where the input voltage is 12 volts (V), the sensing ratio N is 500, the output voltage is 1.2 V, and the output current is 10 amperes (A), the duty cycle is about 10% and the average low side current is about 9 A. Accordingly, the current of the sensing transistor is about 9/500 A (or about 18 mA). In turn, power loss due to the current sensing circuit can be calculated to be about 216 milliwatts (mW) and as having 1.8% efficiency loss with respect to the output power of the regulator.

While power dissipation can be reduced by increasing the sensing ratio N, there can be practical limitations to this approach. For example, increasing the sensing ratio N also degrades sensing accuracy due to, for example, decreased signal to noise ratio resulting from decreased current levels. Because of sensing accuracy considerations, power dissipation may only be reducible to a limit by the selection of the sensing ratio N alone. Thus, there is a need for improved current sensing circuits in switching regulators.

In certain embodiments, efficiency can be improved by powering, supplying, and/or biasing the sensing transistor with the switching regulator. In comparison with a linear regulator, the switching regulator is relatively efficient (for example, linear regulators can step down voltage by dissipating power). In one specific example, the current sensing circuit is supplied current by the switching regulator alone, without additional power supplies. For example, the currents flowing through the sensing transistor, the matching circuit, and the current mirror are supplied by the switching regulator. In another specific example, the current sensing circuit is supplied current partially by the switching regulator and partially by the input voltage and/or a linear regulator. For example, the current flowing through the sensing transistor and the matching circuit can be supplied by the switching regulator and the current flowing through the current mirror can be supplied by a separate supply, such as the input voltage and/or a linear regulator. In another example, current flowing through the sensing circuit can be partially supplied by the switching transistor and partially supplied by the input voltage and/or a linear regulator.

As previously stated, configuring the current sensing circuit to be powered, supplied, and/or biased, at least partially, by the switching regulator can aid in reducing losses and improving the efficiency of the current sensing circuit. This can be true in some situations because, for example, the output voltage of the switching regulator can be lower than the input voltage. Additionally or alternative, the output voltage of the switching regulator can be an efficient power rail in comparison to power rails powered with linear regulators.

FIG. 1is a schematic block diagram illustrating a closed-loop switching regulator system100. The closed-loop switching regulator system100includes a power stage102, a current sensing circuit104, a feedback compensator/driver106(“driver”), an optional linear regulator108, and a core circuitry load110. The power stage102includes one or more power transistors, such as a high-side transistor112and a low-side transistor114, as well as an output inductor116and an output capacitor118.

The power stage102is configured to receive a first drive control DRV1 and a second drive control DRV2 as inputs and a first supply voltage V1and a second supply voltage V2as power supplies, and is configured to generate an output voltage VOas an output power supply. The power stage102can generate the output voltage VOat a variable level, in accordance with the first and second drive controls DRV1, DRV2, by converting the received first and second supply voltages V1, V2. For example, in the illustrated embodiment the high-side transistor112of the power stage102includes a gate configured to receive the first drive control DRV1, a drain configured to receive a first supply voltage V1(for example, from a positive power rail), and a source electrically connected to a first end of the output inductor116and to a drain of the low-side transistor114(for example, at a node ND). The low-side transistor114includes a gate configured to receive the second drive control DRV2, a drain electrically connected to the source of the high-side transistor112and the first end of the output inductor116(for example, at the node ND), and a source configured to receive the second supply voltage V2(for example, from a negative supply rail or ground). The output inductor116further includes a second end electrically connected to an output of the power stage102to provide the output voltage VO. The output capacitor118is electrically disposed between the second end of the output inductor116and the second supply voltage V2. The generated output voltage VO, for example, can be effective for powering a circuit, device, or load, such as the core circuitry load110.

In operation, the first and second drive controls DRV1, DRV2 turn on and off the high-side and low-side transistors112,114to generate a variable output voltage VOby controlling an inductor current ILthrough the output inductor116. In certain implementations, the first and second drive controls DRV1, DRV2 can be used to regularly switch the state of the high-side and low-side transistors112,114between a first phase and a second phase of the power stage102. For example, during the first phase of the power stage102, the high-side transistor112can be on and the low-side transistor114can be off to increase or build up the magnetic field of the output inductor116by providing a switch current I1from the first supply voltage V1to a load (for example, the core circuitry load110) through the output inductor116. During the second phase of the power stage102, the high-side transistor112can be off and the low-side transistor114can be on such that inductor's magnetic field operates to provide a switch current I2from the second supply voltage V2to the load through the output inductor116. During switching between the first and second phases, the output capacitor118can prevent an instantaneous change in the output voltage VO. Applicable inductances of the output inductor116and applicable capacitances of the output capacitor118will be readily determined by one of ordinary skill in the art.

Although the power stage102has been described as selectively operating in two phases, the power stage102can operate in additional phases. For example, in one embodiment, the power stage102can selectively operate in the first phase, the second phase, and a third phase.

In certain implementations, the high-side and low-side transistors112,114are implemented as metal oxide semiconductor (MOS) transistors, such as vertical diffused MOS (DMOS) transistors. As used herein and as persons having ordinary skill in the art will appreciate, MOS transistors can have gates made out of materials that are not metals, such as polysilicon, and can have dielectric regions implemented not just with silicon oxide, but also with other dielectrics, such as high-k dielectrics.

The high-side and low-side transistors112,114are illustrated as n-type MOS (NMOS) transistors. However, the teachings herein are applicable to configurations using p-type MOS (PMOS) transistors and/or configurations using a combination of NMOS and PMOS transistors. For example, in certain implementations, the high-side transistor112can be implemented as a PMOS transistor and the low-side transistor114can be implemented as an NMOS transistor.

In one embodiment, the power stage102corresponds to a power stage102of buck converter. The power stage102, however, can be any applicable type of power stage, including a boost-type power stage. The choice of the type of the power stage will depend on the application, as will be appreciated by one skilled in the art.

The current sensing circuit104is configured to receive the switch voltage VDof the low-side transistor114and the second drive control DRV2 as inputs, is configured to receive the output voltage VOof the power stage102as well as the first and second supply voltages V1, V2and a third supply voltage V3as power supplies, and is configured to generate the current sensed signal VCSas an output. In the illustrated embodiments, the current sensed signal VCScorresponds to a voltage signal representative of the sensed current. For example, the current sensing circuit104ofFIG. 1is electrically connected to the power stage102and the driver106to receive the switch voltage VDand the second drive control DRV2. The current sensing circuit104can be configured to sense the switch current I2of the low-side transistor114based at least partly on the switch voltage VDand the second drive control DRV2. Thus, the current sensing circuit104can sense the approximate inductor current ILthrough the output inductor116when the low-side transistor114is on and the high-side transistor112is off. The current sensing circuit104can be configured to generate the current sensed signal VCSin relation to the sensed inductor current IL(by way of sensing the switch current I2) when the low-side transistor114is on and the high-side transistor112is off. Various embodiments of the current sensing circuit104will be described in further detail later in connection withFIGS. 2-5.

The current sensed signal VCScan aid in monitoring the performance of the power stage102. Such monitoring can be beneficial, for example, for overload protection, for output control, fault detection, and/or failure prevention of the power stage102, among other functions.

FIG. 1illustrates the current sensing circuit104in the context of a switching regulator implemented in a buck converter configuration. In addition, the teachings herein are applicable to other implementations of switching regulators, including, for example, boost converter configurations. Thus, the teachings herein are applicable to switching converters that generate not only a buck or step-down voltage, but also to switching converters that generate a boost or step-up voltage.

The power stage102and the current sensing circuit104form an output portion of the switching regulator system100for providing the output voltage VOas well as the switch voltage VD. Various embodiments of the output portion will be described in further detail with reference toFIGS. 2-5.

The driver106is configured to receive the output voltage VOof the power stage102and the current sensed signal VCSas inputs and is configured to generate the first and second drive controls DRV1, DRV2 as outputs. For example, the driver106of the illustrated embodiment can generate the first and second drive controls DRV1, DRV2 based on the output voltage VOand the current-sensed voltage VCSto regulate or control the output voltage VO. In certain embodiments, the driver106can be configured to receive, or generate internally, a reference voltage VREF(not shown). The reference voltage VREFcan indicate an intended or desired voltage of the output voltage VOof the power stage102. To track approximately the reference voltage VREF, the driver106can include one or more compensators (not shown) disposed in corresponding feedback paths, such as a voltage feedback path and/or a current feedback path, for aiding in the generation of the first and second drive controls DRV1, DRV2. The driver106can be configured to operate in a voltage mode, a current mode, and the like compensation modes. Furthermore, the current mode compensation can include peak current mode control, valley current mode control, emulated current mode control, hysteretic current mode control, and the like current mode controls.

Although not illustrated for the sake of clarity, the driver106ofFIG. 1can include switch control circuitry for generating the first and second drive controls DRV1, DRV2. For example, the driver106can include a pulse width modulator (not shown) configured to pulse width modulate the first and second drive controls DRV1, DRV2. Additionally, the driver106can be configured to generate restrictively the first and second drive controls DRV1, DRV2 in only permissible states. For example, the driver106can be configured to prevent the high-side and low-side transistors112,114from being on at the same time. In certain implementations, the driver106can include control circuitry configured to operate based on the current sensed signal VCS.

In one embodiment, the current sensing circuit104is powered from the generated output voltage VOor from the first supply voltage V1. In alternative embodiments, the optional linear regulator108is configured to receive the first supply voltage V1as a power supply and is configured to generate the third supply voltage V3as an output. For example, the optional linear regulator108can generate the third supply voltage V3(for example, VDD) for provisioning a supply voltage different from the output voltage VO. For example, the power stage102can be configured to provide a low voltage supply and the linear regulator can be configured to provide an intermediary supply voltage relative to the first supply voltage V1and the output voltage VO. Examples of types of linear regulators include passive linear regulators, active linear regulators, and the like as appreciated by one skilled in the art.

In certain embodiments, the third supply voltage V3can correspond to the first supply voltage V1. For example, the linear regulator108can correspond to an electrical path that provides the first supply voltage V1to the current sensing circuit104. In other embodiments, the third supply voltage V3can be omitted. For example, in a certain embodiment, the current sensing circuit104can be powered or biased entirely by the output voltage VOof the power stage102.

As stated, the current sensing circuit104can be supplied power or biased from various power supplies, such as the first, second, and third supply voltages V1, V2, V3and the output voltage VOof the power stage102. The output voltage VOcan provide an efficient power rail in comparison with the first and third supply voltages V1, V3. For example, linear regulators can consume the difference between the input voltage and the regulated voltage (for example, V1- V3), resulting in inefficiency. In some embodiments, the output voltage VOis used to completely supply the current sensing circuit104alone. For example, this may be possible where the output voltage VOis sufficiently large (for example, VO>2 V) for powering the current sensing circuit while providing enough headroom for powering the core circuitry load110.

As shown inFIG. 1, in some embodiments, the first and third supply voltages V1, V3can also be used as a power supply for the current sensing circuit104, for instance, for low-current bias circuits. For example, the first and third supply voltages V1, V3can be used to supply the current sensing circuit104, either separately or supplementally, with the output voltage VO. In doing so, the output voltage VOcan be a relatively low voltage supply (for example, 0.5 V or lower), but can have a voltage in a very broad range.

The core circuitry load110is configured to receive the first, second, and third voltage supplies V1, V2, V3and the output voltage VOas power supplies. The core circuitry load110can correspond to circuits configured to perform, for example, various functions related to electronic devices, electromechanical devices, electro-optical devices, electrochemical devices, and the like devices.

In one embodiment, the power stage102, the current sensing circuit104, the driver106, the linear regulator, and the core circuitry load110, as well as subcomponents thereof, are implemented on an integrated circuit. However, other configurations are possible. For example, the output inductor116, the output capacitor118, and the core circuitry load110are implemented external to the integrated circuit that includes the remaining components of the switching regulator system100.

FIG. 2is a schematic circuit diagram illustrating one embodiment of an output portion200of the switching regulator system100including a current sensing circuit204and the power stage102ofFIG. 1. Elements ofFIG. 2common toFIG. 1share common reference indicia, and only differences betweenFIGS. 1 and 2are described herein for the sake of brevity. The current sensing circuit204includes a sensing transistor206, a first amplifier208, a transistor210, first and second mirror transistors212,214, an offset current source216, and a sensing resistor218.

The sensing transistor206can include a gate configured to receive the second drive control DRV2, a drain configured to receive the switch voltage VD, and a source configured to receive a voltage corresponding the second supply voltage V2. For example, the illustrated embodiment shows the sensing transistor206as having its gate electrically connected to the second drive control DRV2, the drain electrically connected to the node NDof the power stage102, and the source electrically connected to a matching circuit formed of the first amplifier208and the transistor210. As described later in further detail, the matching circuit can be configured to provide the voltage corresponding to the second supply voltage V2.

In certain implementations, the sensing transistor206is scaled in width relative to the low-side transistor114. For example, the low-side transistor114can be sized with a width of about N times a width of the sensing transistor206such that the “on” resistance of the sensing transistor206is also about N times higher than the on resistance of the low-side transistor114. The current I2flowing through the low-side transistor114can be indirectly observed by the voltage drop generated by the current I2and the on resistance of the low-side transistor114. This voltage drop results in the switch voltage VD, which is a negative voltage during operation with the low-side transistor114on. In operation, a sensing current I3can flow through the sensing circuit204. For instance, the voltages of the gate, drain, and source of the sensing transistor206can be matched with the voltages of the gate, drain, and source of the low-side transistor114. Thus, with the on resistances scaled by a factor of N, the switch current I2can be about equal to N*I3. In one embodiment, N is selected to be about 100. In other embodiments, N is selected to be in the range of about 10 to about 10000. One benefit, among several, of selecting N to be much greater than 1 is that the sensing current I3can have a reduced impact on the inductor current ILrelative to the switch current I1. As one result, sensing the switch current I2can provide an effective indication of the inductor current IL.

The first amplifier208(for example, an operational amplifier) and the transistor210can form the matching circuit, which can be configured to receive the second supply voltage V2as an input and to generate the voltage corresponding to the second supply voltage V2as an output. The generated voltage corresponding to the second supply voltage V2can be used to set the source voltage of the sensing transistor206. For example, the illustrated embodiment shows that the first amplifier208can have a first input (for example, the non-inverting input) configured to receive the second supply voltage V2, a second input electrically connected to the source of the sensing transistor206. Further, the transistor210can have a gate electrically connected to the first amplifier208and a source electrically connected to a second input (for example, the inverting input) of the first amplifier208, forming a feedback loop with the first amplifier208and the transistor210.

In operation, the first amplifier208is in a feedback loop such that the first amplifier208can control the voltage at the gate of the transistor210such that the voltage at the source of the transistor210, which is provided as an input to the second input of the first amplifier208matches with the second supply voltage V2which is provided as an input to the first input of the first amplifier208. If the source of the sensing transistor206is set to the second supply voltage V2, then the sensing current I3can be a version of the switch current I2with a scaling based on the sizing of the low-side transistor114relative to the sensing transistor206. In addition, the sensing current I3will flow across the transistor210as a drain-to-source current I4.

The first and second mirror transistors212,214can form a current mirror configured to receive the sensing current I3(by way of the current I4flowing across the transistor210) as an input and to generate a mirrored current I6. The first mirror transistor212is diode connected. For example, the first mirror transistor212can be a scaled version or an unscaled version of the second mirror transistor214such that the current I5flowing through the first mirror transistor212can be approximately equal to a scaled version of the current I6flowing through the first mirror transistor212. In various embodiments, a width of the first mirror transistor212can be selected to be about N times a width of the second mirror transistor214. Thus, the mirrored current I6can be about equal to about I5/N. In one embodiment, N is selected to be about 1. In other embodiments, N is selected to be in the range of about 0.1 to about 100.

In operation, the transistor210can cause the sensing current I5to flow through the first mirror transistor212. In turn, the mirrored current I6flows through the second mirror transistor214of the current mirror and to a first end of the first sensing resistor218. The first sensing resistor218can convert the mirrored current I6to a current sensed signal, such as the current sensed voltage VCS. Thus, the first sensing resistor218is configured to change the current sensed voltage VCSin relation to the switch current I2.

The first sensing resistor218can be implemented using a variety of configurations. For example, in certain implementations, the first sensing resistor218is implemented using passive structures, such as polysilicon structures. However, other configurations are possible, such as implementations in which the first sensing resistor218is implemented using active devices such as transistors biased to provide a desired resistance.

The first sensing resistor218can also been configured to receive an offset current IOSfrom the offset current source216. Configuring the first sensing resistor218to receive the offset current IOScan aid in generating a voltage VCSabove a minimum offset from V2even when the inductor current ILis approximately equal to zero.

As shown, the output voltage VOsupplies the entire current sensing circuit204ofFIG. 2. In one embodiment, the output voltage VOcan be greater than about 1 V. In another embodiment, the output voltage can be in a range of about 3 to about 5 V. Other applicable values can be readily determined by one of ordinary skill in the art.

In certain implementations, the transistors206,210,212,214of the current sensing circuit204are implemented as MOS transistors. In addition, the transistors206,210of the current sensing circuit204are implemented by NMOS transistors and the transistors212,214are implemented by PMOS transistors. However, the teachings herein are applicable to configurations using other combinations of NMOS and PMOS transistors.

FIGS. 3-5are schematic circuit diagrams illustrating various embodiments of the output portion of the switching regulator system100including the current sensing circuit104and the power stage102ofFIG. 1. Elements common to the embodiments share common reference indicia, and only differences between the embodiments are described herein for the sake of brevity.

FIG. 3is a schematic circuit diagram illustrating one embodiment of an output portion300of the switching regulator system100including a current sensing circuit304and the power stage102ofFIG. 1. The current sensing circuit304further includes another transistor306and a second amplifier308.

In certain embodiments, the sensing transistor206and the matching circuit (the first amplifier208and the transistor210) can be supplied current from a supply different from the supply of the first and second mirror transistors212,214. For example, in the illustrated embodiment the drain of the transistor210is electrically connected to the output voltage VO. Thus, the current I3flowing through the sensing transistor206flows from the transistor210(as the current I4) and, in turn, from the output voltage VO.

In addition, the transistor306includes a drain and a source that forms a portion of an electrical pathway between the third and second supply voltages V3, V2. In certain implementations, the transistor306is scaled in size or drive strength relative to the transistor210. For example, the transistor306can be sized with a width about M times a width of the transistor210. In one embodiment, M is selected to be in the range of about 0.01 to about 10. However, other configurations are possible. One benefit, among several, of selecting M to be less than one is that reducing M reduces power dissipation by the portion of the circuit powered by the third supply voltage V3.

The second amplifier308has a first (non-inverting) input electrically connected to the drain of the transistor306and a second (inverting) input electrically connected to the drain of the transistor210. Furthermore, the second amplifier308has an output electrically coupled to the gates of the first and second mirror transistors212,214, thereby forming a feedback path around the output of the second amplifier308and the first input of the second amplifier308. The second amplifier308can be implementable with any applicable amplifier, such as an operational amplifier.

In operation, the second amplifier308is in a feedback loop such that the second amplifier308can control its output such that its first input approximately matches its second input, thereby matching the drain voltage of the transistor306with the drain voltage of the transistor210. Accordingly, the voltages of the gate, the drain, and the source of the transistor306can be matched with the voltages of the gate, the drain, and the source of the transistor210. Thus, the transistor carries a current I4′that can about equal to M*I4.

One benefit, among others, of the configuration of the current sensing circuit304as shown inFIG. 3is that the current sensing circuit304can be effective in some situations in which the output voltage VOis relatively low. For example, the third supply voltage V3can supply currents I4′and I6regardless of the loading on the output voltage VO.

In certain implementations, the transistor306of the current sensing circuit304is implemented as a MOS transistor. In addition, the transistor306of the current sensing circuit304is illustrated as an NMOS transistor. However, the teachings herein are applicable to configurations using a PMOS transistor.

FIG. 4is a schematic circuit diagram illustrating one embodiment of an output portion400of the switching regulator system100including a current sensing circuit404and the power stage102ofFIG. 1. In the illustrated embodiment, the source of the transistor306is electrically connected to the node with the sources of the sensing transistor206and of the transistor210. Accordingly, the current I3flowing through the sensing transistor206can be partially supplied by the output voltage VOof the power stage102and can be partially supplied the third supply voltage V3. The configuration and operation of the remaining portion of the current sensing circuit404are similar to the embodiment described in connection withFIG. 3.

Configuring the current sensing circuit404in the above-described switching manner can aid in increasing the accuracy of current sensing by attenuating an effect of an offset of the first amplifier208. For example, in operation the inverting input of the first amplifier208can be offset from the non-inverting input of the first amplifier208. As a result, the voltage of inverting input of the first amplifier208can be offset from the second supply voltage V2. If in contrast to the illustrated embodiment ofFIG. 4, the source of the transistor306is connected to the second supply V2rather than to the inverting input of the first amplifier208, then the gate-to-source voltage of the transistor210can be offset from the gate-to-source voltage of the transistor306. In turn, because the current I4can be offset from the current I4′, sensing accuracy may be affected. However, as shown in the illustrated embodiment ofFIG. 4, the sources of the transistors210,306are both connected to the inverting input of the first amplifier208. Accordingly, the gate-to-source voltage of the transistor210approximately matches the gate-to-source voltage of the transistor306, and the current I4can be approximately proportional to the current I4′. As a result, sensing accuracy can be improved.

FIG. 5is a schematic circuit diagram illustrating one embodiment of an output portion500of the switching regulator system100including a current sensing circuit504and the power stage102ofFIG. 1. The current sensing circuit504further includes a second sensing resistor506and a third sensing resistor508.

In the illustrated embodiment, the gate of the sensing transistor206can be configured to receive the second drive control DRV2, and the source of the sensing transistor206can be configured to receive the second supply voltage V2. The second sensing resistor506includes a first end configured to receive the switch voltage VDfrom the power stage102. In addition, the second sensing resistor506further includes a second end that can be electrically connected to the second input of the first amplifier208. The third sensing resistor508includes a first end electrically connected to the second input of the first amplifier208and to the second end of the second sensing resistor506. The third sensing resistor508further includes a second end electrically connected to the drain of the sensing transistor206, which is in turn electrically connected to the sources of the transistors210,306.

In operation, the first amplifier208operates in a feedback loop and controls its output such that the voltage of its second input matches the voltage of its first input (for example, V2). The voltage created across the second sensing resistor506causes a current IRto flow through the second sensing resistor506. Furthermore, the current IRflows through the third sensing resistor508and creates a voltage at the drain voltage of the sensing transistor206. In one embodiment, the resistances of the second and third sensing resistors506,508are about equal. The resistances of the second and third sensing resistors506,508are not critical as the second and third sensing resistors506,508are configured as a voltage divider. With the values of the resistances of the second and third sensing resistors506,508about equal and the second supply voltage V2at ground, the feedback loop operates to place the opposite polarity of the switch voltage VD(a negative voltage when the low-side transistor114is on) at the second end of the second sensing resistor506such that the voltage at the inverting input of the amplifier208that corresponds to an intermediate node of the voltage divider matches with ground or zero volts. The drain voltage of the sensing transistor206can then match the drain voltage of the low-side transistor114, but with opposite polarity (positive voltage). Based on the matching of the gate, source, and drain voltages and a scaling in width between the low-side transistor114and the sensing transistor206, a current I3that has a magnitude of 1/N of the switch current I2flows through the sensing transistor206and causes the currents I4and I4′to flow through the transistors210,306, respectively. The configuration and operation of the remaining portion of the current sensing circuit504are similar to the embodiment described in connection withFIGS. 3 and 4.

The second and third sensing resistors506,508can be implemented using a variety of configurations. For example, in certain implementations, the second and third sensing resistors506,508are implemented using passive structures, such as polysilicon structures. However, other configurations are possible, such as implementations in which the second and third sensing resistors506,508are implemented using active devices such as transistors biased to provide a desired resistance.

The various transistors of the foregoing description have been described as insulated gate field-effect transistors, such as MOSFETs. However, it will also be understood that the transistors can have various structural types other than MOSFETs, including, but not limited to, BJT, JFET, IGFET, MESFET, pHEMT, HBT, and the like transistor structural types. Further, the transistors described herein can also have various polarities, such as N-channel, P-channel, NPN-type, and PNP-type; and can include various semiconductor materials, such as Si, SiC, GaAs, SiGe, and the like.

Applications

Devices employing the above described schemes can be implemented into various electronic devices. Examples of the electronic devices can include, but are not limited to, consumer electronic products, parts of the consumer electronic products, electronic test equipment, medical electronic products, etc. Examples of the electronic devices can also include memory chips, memory modules, circuits of optical networks or other communication networks, and disk driver circuits. The consumer electronic products can include, but are not limited to, a mobile phone, a telephone, a television, a computer monitor, a computer, a hand-held computer, a personal digital assistant (PDA), a microwave, a refrigerator, an automobile, a stereo system, a cassette recorder or player, a DVD player, a CD player, a VCR, an MP3 player, a radio, a camcorder, a camera, a digital camera, a portable memory chip, a washer, a dryer, a washer/dryer, a copier, a facsimile machine, a scanner, a multi-functional. peripheral device, a wrist watch, a clock, etc. The medical electronic products can include, but are not limited to, a Digital-X-ray detector, a CT (Computed Tomography) scanner, an Ultrasounds system, a MRI (Magnetic Resonance Imaging) system, etc. Further, the electronic device can include unfinished products.