Heterodyne stage having precise closed-loop control of the amplitude of the injection signal thereof

A heterodyne stage of a receiver which includes a conventional mixer circuit and local oscillator circuit further includes a differential amplifier circuit which measures the amplitude of the injection signal generated by the local oscillator circuit and controls it precisely to a desired reference level by adjusting the current bias supply to the local oscillator circuit within a banded range to ensure start-up of the local oscillator upon energization and to protect against loss of injection signal under all operating conditions. The differential amplifier circuit provides for a precise amplitude reference setting and sufficient closed-loop gain for controlling the amplitude of the injection signal with a minimum of error between the measured and reference values.

BACKGROUND OF THE INVENTION 
The present invention is directed to a heterodyne stage of a radio or pager 
receiver, and more particularly, to a closed-loop control circuit for use 
therein in controlling the bias current of a local oscillator within a 
banded range to render the injection signal thereof precisely at a desired 
amplitude. 
A heterodyne stage of a radio or pager receiver includes a local oscillator 
which generates an injection signal at a predetermined frequency and at an 
amplitude that is proportional to the amount of bias current supplied to 
the oscillator, and a mixer circuit which is governed by the injection 
signal of the local oscillator to convert a mixer input signal at one 
frequency to a mixer output signal at another frequency based on the 
frequency of the injection signal. Normally, in a receiver heterodyne 
stage, the frequency of the injection signal is subtracted from the 
frequency of the input signal to the mixer to render an output signal of 
the mixer at a frequency substantially corresponding to the resulting 
frequency difference. A receiver of the aforementioned type may include 
one or more heterodyne stages for converting received radio frequency or 
RF signalling to an intermediate frequency or IF and possibly to convert 
one IF signal to another lower IF signal for utilization by further 
downstream circuitry. 
Receivers of the portable variety are battery powered and thus considerable 
emphasis is placed on conserving power consumption by such receivers to 
extend the usable operating life thereof without having to recharge or 
replace its batteries. To this end, some more recent portable receiver 
units have included power conservation measures which are designed to 
de-energize certain circuits of the receiver when not needed and then 
re-energize them according to demand. It is always of some concern in the 
transition from de-energization to re-energization that each of the 
circuits restart and operate at designed performance levels in a short 
time interval. 
For example, restarting a crystal-controlled local oscillator circuit 
requires approximately on the order of twice the bias current than what is 
needed after the oscillator circuit reaches steady state conditions. 
Accordingly, what is desired is to provide a large amount of bias current 
to the oscillator circuit initially upon re-energization and then adjust 
back to the minimum bias current needed to sustain the oscillator 
injection signal at a desired amplitude while operating under steady state 
conditions. The problem is that it is not always easy to estimate these 
two bias current level extremes under all working conditions. Selecting 
current levels that are too high may result in unpredictable oscillator 
performance, as well as excessive gain in the mixer stage and undesirable 
consumption of power from the battery which will decrease the operational 
battery life of the receiver. Selecting too low a level may cause the 
oscillator not to restart upon re-energization or to cause the amplitude 
of the injection signal to fall below operational limits, resulting in 
degraded mixer performance. 
One solution to ensuring proper restart of the oscillator circuit and 
protecting against loss of the injection signal is to provide a closed 
feedback loop at the heterodyne stage to control the bias current to the 
oscillator circuit in accordance with a desired injection signal 
amplitude. Such a solution is proposed in U.S. Pat. No 3,805,162 issued 
Apr. 16, 1974 to Clive Hoffman et al. which patent being assigned to the 
same assignee as the instant application. The proposed circuit of the 
Hoffman et al. patent is directed to detecting an oscillator injection 
signal amplitude and setting a desired amplitude thereof based on the 
difference between base-emitter voltages of a mixer transistor having high 
frequency operational characteristics and a control stage transistor 
having lower frequency operational characteristics. While the proposed 
closed loop control circuitry of Hoffman et al. is considered adequate for 
many operational conditions, it is believed not without need of 
improvement. 
Some factors which must always be considered in evaluating such types of 
feedback control loop circuits are: (1) the ability to measure the 
amplitude of the injection signal undistorted by extraneous signals such 
as broadband RF signalling provided at a first heterodyne stage 
operational at the input of such a receiver unit, (2) the ability to set a 
precise reference level for controlling the amplitude of the injection 
signal to the mixer such that it does not deviate substantially during the 
heterodyne operation of the mixer circuit, and (3) to ensure that the 
closed-loop control circuit operates within a bounded bias current range 
to sustain oscillator operation even under extreme abnormal operating 
conditions. 
In the present invention, these and other factors are provided for in a 
closed-loop control circuit for controlling the oscillator injection 
signal to a desired amplitude by adjusting the bias current to the 
oscillator circuit in a heterodyne stage. The advantages of Applicants' 
invention over the prior art will become more evident from the following 
description of preferred embodiments and accompanying drawings. 
SUMMARY OF THE INVENTION 
In accordance with the present invention, a heterodyne stage of a receiver 
comprises a local oscillator circuit which is responsive to bias current 
supplied from a bias current source for generating an injection signal 
which governs a mixer circuit to convert a receiver signal from one 
frequency to another frequency by a heterodyning process, and a 
differential amplifier circuit having one transistor stage input coupled 
to a measurement of the amplitude of the injection signal to effect a 
signal representative thereof and another transistor stage input biased at 
a reference level and including a circuit stage governed by the amplitude 
representative signal and the reference level to adjust the amount of bias 
current supplied to the oscillator circuit by the bias current source 
within a non-zero bias current range. 
In one embodiment of the present invention, the injection signal is coupled 
directly from the local oscillator circuit to the one transistor stage 
input of the differential amplifier circuit. In this embodiment, the bias 
current source includes a current mirror circuit disposed in a supply line 
between a governing current source and the local oscillator circuit, said 
governing current source setting the maximum amount of bias current to be 
supplied to the local oscillator by the bias current source. The current 
stage of the differential amplifier is coupled to the supply line for 
diverting thereto a portion of the governing current, thus decreasing the 
bias current supply to the local oscillator circuit through the current 
mirror circuit. In this manner, the bias current supplied to the local 
oscillator is controlled within a banded range to ensure performance of 
the local oscillator under all operating conditions. 
In another embodiment of the present invention, the measurement of the 
injection signal amplitude is taken from the delta current signal of the 
mixer circuit and coupled to the one transistor stage input of the 
differential amplifier circuit. The current mirror circuit of the bias 
current source is biased to supply a minimum amount of bias current to the 
local oscillator circuit. In this embodiment, the circuit stage of the 
differential amplifier circuit adjusts the amount of bias current supplied 
to the local oscillator from a minimum amount to a maximum amount which is 
set by the circuit stage. Accordingly, the bias current is adjusted within 
a banded range to control the injection signal at a desired amplitude. 
In both of the foregoing described embodiments, a precise reference level 
and sufficient closed-loop gain are provided by the differential amplifier 
circuit in each case to control the injection signal precisely at a 
desired amplitude set by the reference level.

DESCRIPTION OF THE PREFERRED EMBODIMENTS 
FIG. 1 is a functional block diagram of a heterodyne stage of a radio or 
pager receiver and includes the conventional elements of a local 
oscillator circuit shown at block 10 and a mixer circuit shown at block 
12. The oscillator circuit 10 is responsive to bias current supplied over 
line 14 from a bias current source 16 to generate an injection signal over 
line 18 at a predetermined frequency and an amplitude based on the amount 
of bias current supplied over line 14. The oscillator circuit 10 may be of 
the crystal controlled variety in which case a crystal 20 configured in a 
network of reactive tuning elements 21 and 22 governs the oscillator 10 
with a tuned frequency signal over signal line 24. The mixer circuit 12 is 
governed by the injection signal 18 to convert a receiver signal over line 
26 from one frequency to another frequency by using a well-known 
heterodyning process and the resultant receiver signal at the other 
frequency is output over signal line 28. The block 30 represents a 
closed-loop control circuit which is governed by a reference signal 32 and 
a measured signal 34 to adjust the amount of bias current supplied to the 
oscillator circuit 10 over line 14 from the bias current source 16 over 
line 36. The measured signal 34 may be the injection signal 18 itself as 
indicated by the dashed line 38 or it may be derived from a parameter 40 
of the mixer circuit 12 which parameter being representative of the 
injection signal 18. 
In operation, the controller 30 measures the amplitude of the injection 
signal as provided by the injection signal line 38 or a parameter measured 
from the mixer circuit over signal line 40 and compares that measurement 
with a reference level derived from the reference signal 32. In the event 
that the measured signal falls below the reference level, the bias current 
supplied to the oscillator circuit 10 is increased causing a proportional 
increase in the amplitude of the injection signal. Conversely, when the 
measured signal climbs above the reference level, the bias current 
supplied to the oscillator circuit 10 is decreased by the controller 30 
causing the amplitude of the injection signal to decrease commensurately 
therewith. 
The circuit details of an oscillator circuit 10, a mixer 12, a controller 
30, and current source 16 are shown in two separate embodiments depicting 
respectively in FIGS. 2 and 3. All of the transistor and resistor elements 
of the embodiments of FIGS. 2 and 3 may be constructed on the same 
substrate of an integrated circuit for each case and the reactive 
components of inductors and capacitors may be externally coupled 
respectively to their associated integrated circuit. Each integrated 
circuit may be manufactured by Motorola's Semiconductor Division using the 
well known semiconductor manufacturing process known as MOSAIC 1.5, for 
example. Reference is made to the Motorola publication "Linear and 
Interface Integrated Circuits", DL128, Rev. 2, pp. 1-8, for a more 
detailed description of this type manufacturing process. As a result of 
the integrated circuit implementation and certain trimming steps which are 
a part of the manufacturing process thereof, the characteristic parameters 
of the transistor and resistor components may be more precisely related to 
one another. In addition, better performance is expected from an 
integrated circuit implementation than that which would be expected from a 
circuit comprising discrete components. With that all said and done, it is 
now time to describe the respective circuit embodiments. 
Referring to FIG. 2, the symbol B+ shown throughout the circuit depiction 
represents the power supply potential which is provided from a battery 
source (not shown) and regulated to approximately 1 volt, for example, for 
both of the present embodiments. Continuing, the local oscillator circuit 
shown in the dot dashed lines of block 10 is a conventional Colpitts 
configuration including an NPN transistor Q3 having its collector coupled 
to the B+ supply through an inductive element L1. The base of Q3 is 
coupled to the crystal tuned frequency signal line 24, to the B+ supply 
through a resistor R1 and to ground potential through a capacitor divider 
network comprising capacitive elements C1 and C2. The node connection 
between C1 and C2 is coupled to the emitter of the transistor Q3. For 
frequency tuning purposes, a variable capacitor C3 is coupled from the 
collector of Q3 to ground potential. 
In the present embodiment, a bias current source 16 comprises in part a 
voltage source shown at 42 coupled in series with a resistor Rs. The 
voltage source 42 may be fixed at the B+ supply, for example, or variable, 
as the case may be. The governing current supplied through Rs is conducted 
over a supply line 44 to a current mirror circuit configuration comprising 
the transistors Q1 and Q2 which are also a part of the bias current source 
16. More specifically, the supply line 44 is coupled to both the collector 
and base of transistor Q1 which has its emitter coupled to ground 
potential and is also coupled to the base of transistor Q2 which has its 
collector coupled to the emitter of transistor Q3 in the oscillator 
circuit 10 and also has its emitter coupled to ground potential. In the 
present embodiment, the device geometries of the transistors Q1 and Q2 are 
designed to provide a current mirror ratio of 10-12 to 1; thus, the 
current flow through transistor Q2 is 10 or 12 times the amount of current 
flowing through transistor Q1 which is the control or bias current 
generated by the controller. A capacitor C4 may be coupled from the base 
of transistors Q1 and Q2 to ground potential to offer an enhancement to 
noise immunity and control circuit stabilization. 
The operation of a Colpitts oscillator is considered well known to anyone 
skilled in the pertinent art, the details of which contributing nothing to 
the present invention other than the fact that as more bias current is 
drawn from transistor Q3 through transistor Q2 as a result of the current 
mirror configuration of Q1 and Q2, the oscillator gain and the amplitude 
of the generated injection signal over signal line 18 will both increase 
and conversely, as less bias current is drawn, the oscillator gain and 
injection amplitude will both decrease. Accordingly, the amplitude of the 
injection signal 18 may be adjusted in accordance with the amount of bias 
current drawn through transistor Q2. 
The mixer circuit shown at 12 comprises a conventional single NPN 
transistor stage Q4 connected in a common base configuration. The mixer 
input signal which may be of a broadband, radio frequency variety may be 
AC coupled to the emitter of the mixer transistor Q4 through a capacitive 
element C5. A conventional LC tuning and impedance network comprising the 
paralleled capacitive and inductive elements C6 and L2, respectively, is 
provided at the input side of capacitor C5 and coupled conventionally to 
ground potential. The emitter of transistor Q4 is DC biased through 
resistor R2. Another conventional LC tuning and impedance matching network 
comprising the reactive capacitive and inductive elements C7, C8, and L3 
are coupled between the base of Q4 and ground potential. The node 
connection between L3 and C8 may be coupled to B+ through a resistor RB. 
The injection signal generated from the oscillator circuit 10 over line 18 
is AC coupled to the base of the mixer transistor Q4 through a capacitive 
element C9. The mixer stage 12 includes an additional LC tuning and 
matching network comprising capacitive and inductive elements C10, C10', 
and L4, respectively, which is coupled at the collector of Q4 to ground 
potential and offers selective tuning and AC coupling for the output 
signal from the collector of Q4 to the output line 28. 
In operation, the mixer input signal may be selectively filtered by the 
tuning circuit L2 and C6 to pass only frequencies within a desirable 
frequency range through capacitive element C5. The heterodyne mixing 
operations occur between the filtered mixer input signal and injection 
signal in the base emitter junction of the mixer transistor Q4 with the 
resulting frequency converted signal appearing at the collector of Q4 and 
being filtered by the tuning network C10 and L4 and AC coupled by C10' to 
the output line 28. The LC tuning circuit comprising the elements C7, C8 
and L3 at the base of Q4 substantially attenuates signals having 
frequencies other than the frequency of the injection signal and thus, 
substantially minimizes distortion to the injection signal over the signal 
line 18 caused by the heterodyne mixing process and/or the magnitude of 
the RF mixer input signal, thus providing an essential distortion free 
injection signal at the point 18. 
In accordance with the present invention, a closed loop control circuit 30 
is disposed in the heterodyne stage embodiment of FIG. 2 to control the 
injection signal at a desired amplitude by adjusting the bias current 
supplied to the oscillator circuit 10 from the bias current source 16. The 
circuit of controller 30 is configured as a differential amplifier 
including the transistor stage inputs of Q5 and Q6. The addition of 
transistors Q7 and Q8 in the differential amplifier circuit provide for 
dual differential amplification which provides adequate gain to allow for 
minimum closed-loop error between the reference and measured amplitudes 
while minimizing the current drain on the battery source, i.e. B+ supply. 
Current is supplied to the collectors of NPN transistors Q8 and Q7 through 
a dual collector PNP transistor Q11 configured as a current mirror circuit 
having one collector tied to its own base and the collector of Q7 and the 
other collector tied to the collector of Q8, the emitter of Q11 being 
coupled to the B+ supply. The emitters of transistors Q8 and Q7 are 
commonly coupled through a resistor R3 to ground potential and the bases 
thereof are respectively coupled to the collectors of transistors Q5 and 
Q6 which collectors also being respectively coupled through the resistors 
R4 and R5 to the B+ supply. The emitters of transistors Q5 and Q6 are 
coupled through respective resistors R6 and R7 to a common resistor R8 
which is coupled to ground potential. 
A signal line 46 couples the injection signal from the line 18 to the base 
of the input stage transistor Q5. The base of the other input stage 
transistor Q6 may be coupled to a reference potential or the B+supply as 
the case may be. The transistor Q6 is constructed to have an IC area which 
may be, for example, on the order of eight times that of the IC area of Q5 
in order to establish a reference level between the input transistor 
stages. It is a well-known principle that an IC area differential between 
transistors establishes a base-emitter junction voltage differential 
between the transistors which sets a reference voltage threshold level. In 
the preferred embodiment, the eight times area factor results in a 
threshold level of 54 millivolts. That is, the differential amplifier will 
be in a balanced condition when the voltage on the base of Q5 is 54 
millivolts above the voltage of the base of Q6. 
A current mirror circuit stage is coupled to the differential amplifier at 
the collector of the transistor Q8 and to the governing current supply 
line 44 to divert governing current from the supply line. More 
specifically, the current mirror circuit stage comprises NPN transistors 
Q9 and Q10 having their bases tied together and their emitters both 
coupled to ground potential with the common base connection node being 
coupled to the collector of Q8 and also to the collector of Q9. The 
collector of Q10 is coupled to the supply line 44 through a current 
limiting resistor RL In addition, the IC area of transistor Q10 may be on 
the order of eight times that of the IC area of transistor Q9 which 
permits the current diverted from the supply line 44 to be substantially 
greater than that flowing through transistor Q9. 
In operation, the input transistor stage Q5 of the control circuit 30 
measures the amplitude of the injection signal from the supply line 18 at 
the base of Q4. More specifically, the base-emitter junction of transistor 
Q5 rectifies and averages the injection signal to effect a signal which is 
representative of the amplitude thereof which is then compared to the 
reference level set for the desired amplitude of the injection signal. 
When the measured signal is below the reference level, transistors Q5 
through Q11 operate cooperatively to limit the current passing through 
transistor Q9 which in turn limits the current diverted from the supply 
line through transistor Q10, thus rendering higher bias current via 
current mirror Q1-Q2 to the oscillator 10 to cause it to increase the 
amplitude of the injection signal. Should the measured signal exceed the 
reference level, then the transistors Q5 through Q11 operate in a fashion 
to cause increased current to flow through transistor Q9 of the current 
mirror circuit which results in a higher current diversion from the 
governing current supply line 44 through transistor Q10, rendering less 
bias current to the oscillator circuit 10, thereby causing a lower 
injection signal amplitude. Of course, once brought to steady state 
conditions, the controller circuit 30 is provided with sufficient gain to 
sustain the amplitude of the injection signal substantially at the desired 
amplitude level as set by the precise reference level between the input 
transistor stages Q5 and Q6. 
The current mirror circuit comprising transistors Q9, Q10 and limiting 
resistor RL is designed to divert only a portion of the governing current 
from supply line 44, thus ensuring that the closed- loop control circuit 
30 adjusts the amount of bias current supplied to the oscillator circuit 
10 only within a non-zero bias current range. This characteristic of the 
control circuit 30 ensures against loss of oscillator signal during all 
operating conditions. 
It is recognized that during power saving operations where the oscillator 
circuit may be de-energized and then re-energized by the B+supply line, 
the bias current supply to the oscillator circuit will continue to be 
controlled by controller 30 in a closed-loop fashion using the feedback 
measurement of the injection signal as a guide to determine the required 
minimum amount of bias current needed by the oscillator circuit. During 
start-up, when the injection signal level is essentially zero, all of the 
current available from the bias current source is applied to the 
oscillator to minimize the start-up time. Accordingly, start-up of the 
oscillator circuit 10 is assured and once steady-state conditions thereof 
prevail, only the minimum necessary amount of bias current is supplied to 
the oscillator circuit to maintain the injection signal at its desired 
amplitude setting. 
Typical values for the circuit components of the embodiment of FIG. 2 are 
shown in the following table: 
TABLE 1 
______________________________________ 
Element Value 
______________________________________ 
R1 4K ohms 
R2 550 ohms 
R3 6K ohms 
R4 24K ohms 
R5 24K ohms 
R6 2K ohms 
R7 2K ohms 
R8 16K ohms 
Rs 3K ohms 
C1 19 pf 
C2 36 pf 
C3 5-35 pf (trimmer) 
C4 .05 uf 
C5 1000 pf 
C6 5-35 pf (trimmer) 
C7 5-35 pf (trimmer) 
C8 220 pf 
C9 220 pf 
C10 5-35 pf (trimmer) 
L1 70 nH 
L2 70 nH 
L3 70 nH 
L4 2.6 uH 
20 51 MHz (3rd overtone) 
21 5-35 pf (trimmer) 
22 1.2 uH 
______________________________________ 
The foregoing described heterodyne stage embodiment depicted in FIG. 2 has 
primary application to a first stage heterodyning process dealing with 
received broadband RF signalling. An alternate embodiment of a heterodyne 
stage embodying the principles of the present invention which will be 
described in connection with that shown in FIG. 3 has primary application 
to a second heterodyne stage of a receiver in which IF signalling is 
converted from one frequency to another. The receiver IF signalling input 
to a second heterodyne stage has a much narrower bandwidth and limited 
power level than that of the RF signalling input to the first heterodyne 
stage, enabling a more beneficial measuring point of the injection signal. 
Much of that depicted in FIG. 3 is substantially similar to the embodiment 
described in connection with FIG. 2. Accordingly, reference numerals for 
common or substantially equivalent components will be maintained between 
the two embodiments; however, it should be understood that while the same 
or similar components may be performing same or similar functions, their 
values in the alternate embodiment may be changed due to different design 
considerations. 
Referring to FIG. 3, the oscillator circuit 10 remains crystal controlled 
for this embodiment and is of the same Colpitts design; however, because 
the application is primarily directed to a second heterodyne stage, the 
frequency of the injection signal generated thereby will be at a much 
lower value and thus may require fewer reactive tuning components. 
Nevertheless, the functioning thereof is substantially similar to that 
described in connection with the oscillator circuit 10 of FIG. 2. The 
addition of resistor R9 and capacitor C11 provides a decoupling of the B+ 
supply from the oscillator circuit 10. 
The mixer 12 similarly comprises a single stage NPN transistor Q4, but in 
this alternate embodiment, it is configured in a common emitter 
arrangement with the emitter thereof coupled to ground potential. The 
mixer 12 of this alternate embodiment further includes a current mirror 
circuit shown at 50 comprising an NPN transistor Q12 having its collector 
tied to its base and also to the B+ supply through a resistor R10 and 
having its emitter coupled to ground through a resistor R11. To protect 
against noise and unwanted signal coupling, a capacitor C12 is coupled 
between the base of Q12 and ground potential. An inductor component L5 is 
coupled between the collector of Q12 and base of Q4 to provide a DC bias 
current path to Q4 while highly attenuating the high frequency signals at 
the base of Q4. 
Note that the mixer input signal over line 26 and the injection signal over 
line 18 are combined at the base of Q4 and that the base emitter junction 
of Q4 is still used to perform the heterodyning process. The resultant 
difference frequency signal is provided at the collector of Q4 and coupled 
to the output line 28. A resistor divider network comprising resistors R12 
and R13 is coupled between the collector of Q4 and the B+ supply and a 
capacitor C13 is coupled between the node connection of R12 and R13 and 
ground potential to provide for a stable signal at such node. The voltage 
potential appearing at the node connection of R12 and R13 is 
representative of the current drain (delta current) of the mixer 12 and 
may be used as a measure of the amplitude of the injection signal supplied 
to the mixer from the oscillator circuit 10 in that the mixer drain will 
increase as the injection level is increased. 
A narrow bandpass filter having a center frequency set at the desired 
frequency of the mixer input signal is disposed in the input line and 
shown as block 52. Variable capacitive tuning elements C14 and C15 are 
disposed respectively upstream and downstream of filter 52 and coupled to 
ground. An inductive element L6 is disposed in series with the filter 52. 
The aforementioned reactive components are provided for additional 
frequency tuning of the filter 52. In a similar manner, another narrow 
band frequency filter shown at block 54 is disposed at the output line 28 
of the mixer 12 and has a center frequency set at the frequency of the 
resultant mixer output signal. In summary, then, the primary difference up 
to this point in the description of the alternate embodiment is that the 
measurement of the injection signal representative of the amplitude 
thereof is a delta current measurement of the mixer stage 12 and is taken 
from the node between the resistor divider connection of R12 and R13 in 
the collector of the mixer transistor Q4. The measurement signal is 
connected by signal line 46 to the base of an input transistor stage Q5 of 
the differential amplifier of the controller circuit 30. 
Referring now to the controller circuit 30 of the embodiment depicted in 
FIG. 3, the input stages are similarly comprised of transistors Q5 and Q6 
having their emitters commonly coupled through a common resistor R14 to 
ground potential. However, the collectors of Q5 and Q6 are directly 
coupled respectively to the dual collectors of Q11 without the need of the 
dual differential transistors Q8 and Q7 as described in connection with 
the embodiment of FIG. 2. Another difference of the controller circuit 30 
in the present embodiment is that the reference level of the other input 
transistor stage Q6 is set by a resistor divider network comprising 
resistors R15 and R16 coupled in series coupled between the B+supply and 
the ground potential. The resistor divider network of R15 and R16 provides 
a precise voltage reference level at the node connection thereof which is 
coupled to the base of the other input transistor stage Q6 of the 
differential amplifier. Since this reference level is representative of 
the desired amplitude, the injection signal may be precisely amplitude 
controlled with a minimum of bias current supply. 
The bias current source 16 comprises a similar mirror circuit arrangement 
comprising similar transistors Q1 and Q2 with the collector of Q2 being 
coupled to the oscillator circuit 10 to control the bias current supply 
thereto. In this alternate embodiment, a resistor R17 is coupled between 
the current mirror circuit and the B+ supply for setting a minimum bias 
current supply to the oscillator circuit 10 to ensure against loss of 
injection signal under all oscillator operating conditions. A current 
adjusting resistor R18 is coupled between the emitter of Q1 and ground 
potential. As part of the control of the bias current, the differential 
amplifier of the control circuit 30 includes a circuit stage comprising a 
dual collector PNP transistor Q13 having one of its collectors tied to its 
base which is in turn coupled to the collector of the input stage 
transistor Q5. The emitter of Q13 is coupled to the B+ supply and the 
other collector thereof is coupled to the current mirror circuit of Q1 and 
Q2 for controlling the amount of bias current being supplied to the 
oscillator circuit 10. The maximum governing current through Q13 which may 
be adjusted by the value of R14 sets the maximum bias current supply via 
Q1-Q2 to the oscillator circuit 10. Accordingly, a non-zero bias current 
range is created between the minimum setting of bias resistor R17 and 
maximum setting of the governing current of Q13, which maximum setting 
ensures proper power start-up of the oscillator circuit 10. 
In operation, the control circuit 30 is governed by the measured signal 
over line 46 and the reference potential set by the resistor divider 
network R15 and R16 such that when the measured signal is below the 
reference level, governing current is conducted through Q13 to the bias 
current source circuit 16 to increase the bias current supply to the 
oscillator circuit 10 which renders an increase in amplitude of the 
injection signal generated thereby. Conversely, when the measured signal 
becomes greater than the reference level, less current is conducted 
through Q13 reducing the bias current supply to the oscillator circuit, 
thus rendering a lower amplitude of the injection signal. The control 
circuit 30 is set with tight enough control loop parameters to control the 
amplitude of the injection signal substantially at the desired level. 
Further, at re-energization, the maximum bias current supply is applied to 
the oscillator to ensure proper start up of the oscillator circuit 10. 
Accordingly, after energization and once steady state conditions prevail 
with the amplitude of the injection signal at the desired level, the 
control circuit 30 governs the bias current supplied to the oscillator 
circuit to a minimum level necessary to maintain the desired amplitude of 
its generated injection signal. Typical values for the circuit elements of 
the embodiment of FIG. 3 are shown in the following table: 
TABLE 2 
______________________________________ 
Element Value 
______________________________________ 
R1 20K ohms 
R9 100 ohms 
R10 15K ohms 
R11 3K ohms 
R12 lK Ohms 
R13 1.8K Ohms 
R14 4K ohms 
R15 20K ohms 
R16 80K ohms 
R17 30K ohms 
R18 3K ohms 
Cl 80 pf 
C2 30 pf 
C4 .01 uF 
C11 .1 uF 
C12 .1 uF 
C13 .1 uF 
C14 2-10 pf 
C15 2-10 pf 
L5 4.2 uH 
L6 2.6 uH 
52 17.9 MHz (crystal filter) 
54 455 KHz (ceramic filter) 
20 17.445 MHz (fundamental 
crystal) 
______________________________________ 
While the embodiments of FIGS. 2 and 3 were described using a crystal 
controlled oscillator circuit, this is not the case for all heterodyne 
stages of a receiver. Some stages may use another type of oscillator 
circuit, such as a voltage controlled oscillator (VCO), for example. A 
suitable circuit embodiment of a VCO for use in a heterodyne stage 
embodying the invention is shown in FIG. 4. The VCO embodiment includes a 
conventional Colpitts oscillator circuit 10 similar to the oscillator 
circuit described in connection with the depictions of FIGS. 2 and 3. 
Reference numerals are maintained for the same or similar elements. 
Instead of being driven by a tuned crystal circuit, the frequency of the 
Colpitts oscillator 10 is governed by a varactor or varicap element VCl 
which is coupled between the variable capacitor 21 and ground potential. A 
choke coil L7 is coupled between the node connection between 21 and VCl 
and a variable source of voltage potential at 60. Further, a fixed 
capacitor element may be coupled between the base of Q3 and ground 
potential to provide additional oscillator tuning. Still further, a choke 
coil L8 is disposed between the base of Q3 and B+ supply for biasing 
purposes and for decoupling the B+ supply from the varactor tuned driver 
stage. A resistor R19 coupled between the collector of Q3 and the B+supply 
provides both impedance matching and source or local impedance. 
A similar current mirror stage comprising transistors Q1 and Q2 is coupled 
to the VCO 10 to control the bias current thereof which in turn adjusts 
the amplitude of the injection signal output AC coupled through capacitor 
C9. Another choke coil L9 and capacitive C17 are disposed in the bias 
current line between the oscillator circuit 10 and current mirror 16 to 
provide AC decoupling of the bias current from the transistor Q2. Resistor 
R20 sets a minimum current level to ensure that the oscillator has 
sufficient current under all loop conditions. 
In operation, the frequency of the injection signal coupled through C9 to 
the mixer may be controlled by varying the voltage potential applied to 
the VCO at 60. For example, varying the voltage at 60 between 0.5 V and 3 
V alters the capacitance of VC1 in the present embodiment for 24 to 44 pf 
which commensurately adjusts the frequency of the oscillator about a 
center frequency of approximately 150 MHz. Typical values of the circuit 
elements for a suitable embodiment of a VCO tuned for a center frequency 
of approximately 150 MHz are shown in the table below. 
TABLE 3 
______________________________________ 
Element Value 
______________________________________ 
C1 22 pf 
C2 15 pf 
C4 .05 uf 
C9 220 pf 
C16 30 pf 
C17 125 pf 
21 5-35 pf 
VC1 24-44 pf (.5V-3V) 
L7 2.6 uH 
L8 2.6 uH 
L9 2.6 uH 
22 87 nH 
R17 3 K ohms 
R18 1 K ohms 
R19 50 ohms 
R20 1 K ohms 
______________________________________ 
In summary, both of the embodiments of the present invention described in 
connection with the depictions of FIGS. 2 and 3 provide a very ample and 
viable feedback control loop having the ability to measure the injection 
signal generated by the oscillator circuit in each case undisturbed by 
extraneous signals, the ability to set a precise reference level for 
controlling the injection to a desired amplitude signal which becomes 
particularly important when using a VCO for which precise channel 
selectivity is needed and because of the substantial increase in sideband 
noise when the amplitude of the injection signal deviates from that 
desired, and a bias current control range to ensure that the closed-loop 
controller operates within a banded range to start-up and sustain 
oscillator operation under all operating conditions with a minimum current 
drain from the battery source. 
While the present invention has been described in connection with two 
specific embodiments, it is understood that additions, deletions, and 
modifications of these embodiments may be made without deviating from the 
principles of such invention. Accordingly, the present invention should 
not be limited to any specific embodiment but rather construed in scope 
and breadth according to the claims appended hereto.