Resonator filters on dielectric substrates

A resonator filter has electrodes disposed in a confronting relationship to each other and has a dielectric base plate interposed therebetween. The filter has ground or common terminals on the electrode portions in different positions which do not include mutually confronting electrode portions. The electrodes have spiral configurations or configurations having at least one bent portion such that the electrodes have a lumped-constant inductance. Voltage signals induced in the electrodes due to mutual induction therebetween are opposite in phase with respect to each other and a parasitic distributed-constant capacitance is produced due to a potential difference between the electrodes and a dielectric constant of the dielectric base plate. As a consequence thereof, a two-terminal parallel resonance band pass filter circuit is formed on an equivalent basis which is composed of the lumped-constant inductance of one of the electrodes and the distributed-constant capacitance between the electrodes. The parallel resonance band pass filter has a resonance frequency which is lower than a frequency having its 1/4 wavelength equal to an equivalent electrical length of each electrode.

BACKGROUND OF THE INVENTION 
1. Field of the Invention 
The present invention relates to a resonator filter for use in radio and 
television transmitters and receivers and other communication equipment. 
2. Description of the Prior Art 
As more and more signals are used in radio and television broadcasting and 
communications, resonator filters for selecting the frequencies of the 
signals to be received are required to be highly stable and reliable in 
their performance. There has also been a great demand for the reduction of 
the cost of manufacture of receivers, transmitters and other communication 
equipment in which the resonator filters are installed. In particular, the 
development of a new technology is desired for tuning circuit components 
in radio-frequency stages which have been difficult to improve. 
Conventional resonator filters will be described with reference to the 
drawings. FIG. 1 of the accompanying drawings shows a basic circuit 
arrangement of a resonator filter. The resonator filter comprises a 
resonator circuit 3 composed of an inductor 1 and a capacitor 2 which are 
connected in parallel to each other. The resonator filter has 
conventionally been constructed of components as illustrated in FIGS. 2 
and 3. In one prior arrangement shown in FIG. 2, an inductor component 4 
and a capacitor component 5 are interconnected by conductors 6, 7. 
According to another prior art resonator design, a planar inductor 9 is 
placed on a plate-like dielectric 8, a capacitor 12 is composed of 
confronting electrodes 10, 11, and the inductor 9 and the capacitor 12 are 
interconnected by conductors 13, 14. 
However, the conventional arrangements have been subjected to the following 
disadvantages: 
In the resonator shown in FIG. 2, the inductor component 4 is large in size 
in comparison with the other components, and is particularly much larger 
in height, with the result that the equipment in which the resonator is 
incorporated is not rendered smaller in size and lower in profile. A 
ferrite core inserted in the inductor component is variable in position 
due to mechanical vibrations, resulting in wide drifts in tuning 
frequencies. The inductance of the inductor component is unstable due to a 
large degree of temperature-dependency of the magnetic permeability .XI. 
of the ferrite core, a feature which also causes tuning frequencies and 
also Q of the resonator filter circuit to vary widely. For keeping the 
tuning frequencies stable at their target settings, the components are 
required to be very accurately installed in predetermined positions. Where 
the resonator filters are mass-produced as RF resonator filters, it is 
difficult to maintain a desired installation accuracy and hence the tuning 
frequencies tend to differ greatly from their target settings and cannot 
be caused to converge to fixed values. Therefore, there has been 
difficulty experienced with the mass production of the resonator filters. 
The resonator filter illustrated in FIG. 3 has suffered a problem in that 
the inductor and the capacitor take up a large area to thereby prevent the 
equipment incorporating the resonator filter from being smaller in size. 
The resonator filter requires at least three electrodes, that is, the 
inductor electrode and the confronting capacitor electrodes. Since a large 
quantity of electrode material having a high conductivity and hence a high 
cost is used, the cost of manufacture of the resonator filter is high and 
efforts to reduce the amount of material are impossible to make. 
The resonator filter arrangements shown in FIGS. 2 and 3 further have 
common drawbacks. The inductor and the capacitor are constructed as 
separate components that are interconnected by long conductors. The long 
conductors tend to produce unwanted lead inductances and stray capacities 
which cause the resonator filters to operate is an unstable fashion and 
make it difficult to achieve an initial design target. Accordingly, it 
takes a long period of time to design resonator filters, including 
corrective design actions. Since the resonator filters are composed of 
discrete components which perform minimum functions, it has been 
impossible with available technological concepts to reduce the number of 
components used and cope with demands for an improved manufacturing 
process. One resultant problem is that there is a limitation on efforts to 
lower the cost of the resonator filters. 
SUMMARY OF THE INVENTION 
It is an object of the present invention to provide a resonator filter 
which is composed of an integral construction of inductor and capacitor 
components, is low in profile, small in size, mechanically stable, has 
tuning frequencies and Q less dependent on temperature, is stable at high 
frequecies without suffering from adverse influences due to connector 
leads, and has a reduced number of components which permit its 
manufacturing process to be improved. 
To achieve the above object, according to the present invention, electrodes 
are disposed in confronting relationship to each other with a dielectric 
base plate interposed therebetween and have ground or common terminals on 
electrode portions in different positions which do not include mutually 
confronting electrode portions. The electrodes have spiral configurations 
or configurations having at least one bent portion such that the 
electrodes have a lumped-constant inductance. 
With such an arrangement, voltage signals induced in the electrodes due to 
mutual induction therebetween are opposite in phase with respect to each 
other. A parasitic distributed-constant capacitance is produced due to a 
potential difference between the electrodes and a dielectric constant of 
the dielectric base plate. 
As a consequence, a two-terminal network, parallel resonance bandpass 
filter circuit is formed on an equivalent basis which is composed of the 
lumped-constant inductance of one of the electrodes and the 
distributed-constant capacitance between the electrodes. The parallel 
resonance bandpass filter has a resonance frequency lower than a frequency 
having its 1/4 wavelength equal to an equivalent electrical length of each 
electrode. 
Furthermore, the resonance frequency can be trimmed, and a resonance 
bandpass frequency can be controlled by a combination of the resonance 
filter with a variable-reactance element. 
The above and other objects, features and advantages of the present 
invention will become more apparent from the following description when 
taken in conjunction with the accompanying drawings in which preferred 
embodiments of the present invention are shown by way of illustrative 
example.

DESCRIPTION OF THE PREFERRED EMBODIMENTS 
Most resonator filters having the basic tuning circuit shown in FIG. 1 and 
used in HF, VHF, and UHF bands are constructed of individual 
lumped-constant reactance elements interconnected and put together as 
illustrated in FIG. 2. A resonator filter including individual 
distributed-constant reactance elements, as shown in FIG. 3, because of 
long wavelengths corresponding to the above frequency bands, and a 
distributed-constant resonator such as a quarter-wave resonator and a 
dielectric resonator (not shown) have their components scattered and are 
of quite a large configuration. However, distributed-constant resonator 
filters and resonators can easily be constructed of solid-state 
components. The present invention is based on the idea that where it is 
ideal and possible for resonator filters used in the above frequency bands 
to be constructed of solid-state components which are put together, it is 
best to achieve resonator filters in optimum form by allowing 
lumped-constant and distributed-constant reactors to act effectively and 
be combined together. 
Resonator filters according to embodiments of the present invention will 
hereinafter be described in detail with reference to the drawings. 
FIGS. 4(a), 4(b) and 4(c) through 7(a), 7(b) and 7(c) illustrate the 
constructions of resonator filters according to embodiments of the present 
invention. 
In FIGS. 4(a), 4(b) and 4(c), a dielectric 15 in the form of a plate is 
made of ceramic and supports an electrode 16 forming an inductor on a 
surface of the dielectric 15 and another electrode 17 disposed on an 
opposite surface of the dielectric 15 in a confronting relationship to the 
electrode 16. The electrodes 16, 17 jointly form a distributed-constant 
circuit forming a capacitor. The electrode 16 has a ground terminal 16a 
and an open terminal 16b, and the electrode 17 has a ground terminal 17a 
and an open terminal 17b which are opposite to the terminals 16b, 16a, 
respectively. Each of the electrodes 16, 17 has one bent portion having a 
desired bent angle and direction. 
In FIGS. 5(a) through 5(c), electrodes 19, 20 are mounted on opposite 
surfaces of a dielectric 18 in the from of a plate, and in FIGS. 6(a) 
through 6(c), electrodes 22, 23 are mounted on opposite surfaces of a 
dielectric 21 in the from of a plate. As illustrated in FIGS. 5(a) through 
5(c) and 6(a) through 6(c), the electrodes are mounted on the dielectrics 
and have a terminal arrangement in the same manner as described with 
reference to the embodiment of FIGS. 4(a) through 4(c). However, each of 
the electrodes has more than one bent portion having a desired bent angle 
and direction. 
In FIGS. 7(a) through 7(c), electrodes 25, 26 are mounted on opposite 
surfaces of a dielectric 24 in the from of a plate. The electrodes are 
mounted on the dielectrics and have a terminal arrangement in the same 
manner as described with reference to the embodiment of FIGS. 4(a) through 
4(c). However, each of the electrodes has a spiral configuration. 
Although in the embodiments shown in FIGS. 4(a), 4(b) and 4(c) through 
7(a), 7(b) and 7(c) each electrode has a bent portion formed as an angular 
pattern having a desired bent angle, the electrode may instead have a bent 
portion formed as an arcuate pattern having a desired curvature. 
FIGS. 8(a) and 8(b) show the construction of a resonator filter according 
to another embodiment of the present invention. A tubular dielectric 27 
supports an electrode 28 on an inner peripheral surface thereof and an 
electrode 29 on an outer peripheral surface thereof, the electrodes 28, 29 
extending in confronting relation to each other. The electrodes 28, 29 
have ground terminals located at opposite ends thereof. The dielectric 27 
may be in the form of a hollow structure having a rectangular cross 
section. 
The terminals that are designated as ground terminals in the foregoing 
embodiments may not be used as ground terminals but may serve as general 
common terminals for connection to other circuits (not shown). 
The resonator filters shown in FIGS. 4(a), 4(b) and 4(c) through 7(a), 7(b) 
and 7(c) can form relatively large distributed inductors and capacitors 
though they take up small areas. Therefore, small resonator filters having 
relatively low tuning frequencies can be achieved with a resulting 
increased space factor. 
The resonator filter shown in FIGS. 8(a) and 8(b) is smaller in size than 
the resonator filters shown in FIGS. 4(a), 4(b) and 4(c) through 7(a), 
7(b) and 7(c), but can form a sufficiently large inductor and capacitor. 
Therefore, a much smaller resonator filter having a sufficient tuning 
frequency can be accomplished. The resonator filter of FIGS. 8(a) and 8(b) 
can be mass-produced with ease by continuously forming electrodes 28, 29 
on an elongate continuous tubular dielectric 27 and then cutting the 
tubular dielectric 27 to the desired lengths. 
The electrodes, or transmission-line electrodes, in the foregoing 
embodiments may be constructed of a metal conductor, a printed metal-foil 
conductor, a printed thick-film conductor, or a thin-film conductor, or 
may comprise a combination of these conductors. The dielectric may be made 
of alumina ceramic, barium titanate, plastics, Teflon (trademark), glass, 
mica, or in the form of a printed-circuit board of resin base. 
Operation of the resonator filter in each of the foregoing embodiments will 
now be described. FIGS. 9(a) through 9(e) are circuit diagrams of 
equivalent circuits of the resonator filter of the present invention. In 
FIG. 9(a), transmission lines are composed of transmission-line electrodes 
30, 31 having an electrical length l and including ground terminals 
disposed at opposite ends. A signal source 32 for generating a voltage e 
is connected to the transmission-line electrode 30 for supplying a signal 
thereto. When the signal is applied, a traveling-wave voltage e.sub.A is 
excited at the open terminal on the distal end of the transmission-line 
electrode 30. Since the transmission-line electrode 31 is disposed in 
confronting or juxtaposed relationship to the transmission-line electrode 
30 adjacent thereto, a voltage is induced in the transmission-line 
electrode 31 by mutual induction. A traveling-wave voltage e.sub.B is then 
induced at the open terminal on the distal end of the transmission-line 
electrode 31. 
With the ground terminals of the transmission-line electrodes 30, 31 being 
disposed in an opposite relationship or remotely from each other, the 
induced traveling-wave voltage e.sub.B is opposite in phase to the excited 
traveling-wave voltage e.sub.A. Since the distal ends of the transmission 
lines are open, the traveling-wave voltages e.sub.A, e.sub.B form voltage 
standing waves in the transmission lines which are composed of the 
transmission-line electrodes 30, 31, respectively. Assuming that a voltage 
distribution coefficient indicative of the distribution of the voltage 
standing wave in the transmission-line electrode 30 is expressed by K, a 
voltage distribution coefficient of the transmission-line electrode 31 can 
be expressed by (1-K). 
A potential difference V between any confronting portions of the 
transmission-line electrodes 30, 31 is expressed by: 
EQU V=Ke.sub.A -(1-K)e.sub.B (1) 
On the condition that the transmission-line electrodes 30, 31 are of the 
same electrical length l, 
EQU e.sub.B =-e.sub.A (2) 
Thus, the potential difference V given by the above equation (1) can now be 
expressed by: 
##EQU1## 
Therefore, the potential difference V can be generated between all 
confronting portions of the transmission-line electrodes 30, 31. 
It is assumed here that the transmission-line electrodes 30, 31 have a 
width W (with their thickness being small) and are spaced a distance d 
from each other by a dielectric having a dielectric constant 
.epsilon..sub.s. A capacitance C.sub.O formed per unit length of the 
transmission lines is given as follows: 
##EQU2## 
Accordingly, the transmission lines shown in FIG. 9(a) are equivalent to 
transmission lines as shown in FIG. 9(b) having distributed capacitors 33 
of C.sub.O determined by the equation (6) per unit length. As illustrated 
in FIG. 9(c), the transmission lines can be equivalently expressed as a 
distributed-constant circuit composed of total distributed inductors 34, 
35 due to distributed inductor components of the transmission lines and 
lumped inductor components produced by the bent configuration of the 
transmission lines and a distributed capacitor 33. 
Now, the relationship between the formation of the distributed capacitor 33 
and the electrical length l of the transmission lines will be described. A 
characteristic impedance Z.sub.O per unit length of the transmission lines 
shown in FIG. 10(a) can be expressed by an equivalent circuit illustrated 
in FIG. 10(b). The characteristic impedance Z.sub.O is generally given by: 
##EQU3## 
If the transmission lines cause no loss, 
EQU Z.sub.O =L.sub.O /C.sub.O (8) 
This assumption can be applied to many of the resonator filters according 
to the embodiments of the invention. For the sake of brevity, the 
characteristic impedance Z.sub.O defined by the equation (8) will 
hereinafter be used. The capacitance C.sub.O in the equation (8) is the 
same as the capacitance C.sub.O per unit length of the transmission lines 
as determined by the equation (6). More specifically, the characteristic 
impedance Z.sub.O per unit length of the transmission lines is a function 
of the capacitance C.sub.O, and also a fuction of the dielectric constant 
.epsilon..sub.s of the dielectric, the width W of the transmission-line 
electrodes, and the distance d between the transmission-line electrodes. 
An equivalent reactance X at a terminal of a transmission line having a 
characteristic impedance Z.sub.O per unit length thereof, an electrical 
length l, and an open distal end can be expressed by: 
EQU X=-Z.sub.O cot.theta. (9) 
where 
##EQU4## 
the equivalent reactance X is given by: 
EQU X.ltoreq.0 (12) 
Therefore, the equivalent reactance at the terminal of the transmission 
line can be a capacitive reactance. Where .theta. falls in the ranges of 
the equation (11) because of the electrical length l of the transmission 
line, or by selecting the electrical length l to be .lambda./4 or less, a 
capacitor can be formed. The capacitance C of the capacitor thus formed 
is: 
##EQU5## 
Therefore, any desired capacitance C can be achieved by varying .theta. or 
selecting the electrical length l of the tranmission line. 
FIG. 11 is illustrative of the mode of operation of the transmission lines 
described with reference to the above equations (9) through (13). In FIG. 
11, the equivalent reactance X generated at the terminals is shown as 
being varied as the electrical length of the transmission lines with open 
distal ends varies. As is apparent from FIG. 11, it is possible to form a 
negative terminal reactance if the electrical length l is .lambda./4 or 
below or in the range of from .lambda./2 to 4.lambda./3, that is, a 
capacitor can equivalently be formed. Furthermore, under the condition to 
form a negative terminal reactance, the capacitance C can be of a desired 
value by selecting any desired electrical length l of the transmission 
lines. 
The capacitor C can be equivalently replaced with a lumped-constant 
capacitor 36 shown in FIG. 9 (d). The inductor formed by the total of 
distributed inductor components present in the transmission lines and 
lumped inductor components generated by the bending of the transmission 
lines can be equivalently replaced with a lumped-constant inductor 37. By 
expressing the ground terminals in FIG. 9(d) as a common ground terminal, 
the arrangement can finally be equivalent to a parallel resonator circuit 
as shown in FIG. 9(e) which is composed of a lumped-constant capacitor 36 
and a lumped-constant inductor 37, thus realizing a resonator filter. 
The resonator filter of the invention is realized on the basis of the 
arrangement and operation described above. The construction and principles 
of operation of the resonator filter according to the present invention 
are entirely different from those of the conventional resonator filters. 
To indicate that the resonator filter of the present invention differs 
entirely from a conventional resonator filter or another resonator filter 
arrangement using the same transmission lines as those in the resonator 
filter of the invention, the construction and operation of the 
conventional resonator filter or the other resonator filter arrangement 
using the same transmission lines will hereinafter be described. This will 
serve to clarify the difference between the resonator filter of the 
present invention and the conventional resonator filter and also the 
novelty of the resonator filter of the present invention. 
FIG. 12 is illustrative of the circuit arrangement of a quarter-wave 
resonator which has heretofore been used most widely. The illustrated 
prior art resonator is fully different from the resonator filter of the 
invention as to the distal ends of the transmission lines, selection of 
the length thereof, and selection of ground terminals. In FIG. 12, 
balanced-mode transmission-line electrodes 102, 103 have an electrical 
length l equal to .lambda./4 at a resonance frequency and their distal 
ends short-circuited. The transmission-line electrodes are driven in a 
balanced mode by a balanced signal source 104 which generates a voltage e. 
A ground terminal is selected to be located at a neutral point of the 
balanced signal source 104, and no ground terminal is placed at any 
terminal of the transmission-line electrodes. An equivalent terminal 
reactance X produced at the terminals of the transmission lines is given 
by: 
EQU X=Z.sub.O tan.theta. (14) 
where Z.sub.O is a characteristic impedance of the transmission lines and 
equal to the one indicated by the equation (8) and .theta. is equal to the 
one indicated by the equation (10). The electrical length l of the 
transmission lines of this resonator is: 
EQU l=.lambda./4 (15) 
and hence, 
EQU .theta.=.pi./2 (16) 
Therefore, the terminal reactance X in the equation (14) becomes: 
EQU X=Z.sub.O tan(.pi./2)=.infin. (17) 
As a consequence, parallel resonance characteristics can equivalently be 
obtained. The construction of the quarter-wave resonator above will be 
compared with that of the resonator filter of the present invention. With 
respect to the terminal condition, the terminals of the resonator filter 
of the invention are open, whereas those of the conventional quarter-wave 
resonator are short-circuited, so that the terminal conditions are 
entirely different. As to the electrical length l of the transmission 
lines, the electrical length of the resonator filter of the invention is 
selected to be .lambda./4 or shorter, and about .lambda./16 in reality. 
However, the electrical length of the prior art quarter-wave resonator is 
strictly selected to be .lambda./4 of the resonance frequency. Therefore, 
it is apparent that the electrical lengths l of the transmission lines are 
also basically different from each other. Because of the difference betwen 
the electrical lengths of the transmission lines, the resonator filter of 
the present invention can be of a smaller size when the resonator filter 
of the invention and the prior art resonator are designed for use at the 
same tuning frequency or resonance frequency. The quarter-wave resonator 
is required to have much longer transmission lines, and has to be larger 
in size. Some prior art quarter-wave resonators are rendered smaller in 
size by shortening the transmission lines with a dielectric having quite a 
large dielectric constant. The dielectric with a large dielectric constant 
is generally subjected to a high dielectric loss tan .delta., and the Q of 
the resonator tends to be greatly reduced. Furthermore, the dielectric 
constant of the dielectric with a high dielectric constant is generally 
highly temperature-dependent, and it is difficult to maintain the 
stability of a tuning frequency. 
Superior performance of the resonator filter of the present invention will 
be described with reference to experimental results indicative of 
comparison between the performance of the resonator filter of the present 
invention and that of the prior art resonator filter. FIG. 13 is a graph 
showing experimental results of the measured temperature-dependency of a 
tuning frequency, and FIG. 14 is a graph showing experimental results of 
the measrued temperature-dependency of the Q. In FIGS. 13 and 14, 
characteristic curves A are indicative of the temperature dependencies of 
the resonator filter of the present invention, with the dielectric being 
made of alumina ceramic or in the form of a printed-circuit board of resin 
base. Characteristic curves B are indicative of the temperature 
dependencies of the prior art resonator filter which has been most widely 
used in the art. It is evident from these experimental results that the 
resonator filter of the present invention, even if constructed of a 
general dielectric, has a highly stable tuning fruquency, a high Q, and is 
stable. With the prior art resonator filter, however, the magnetic 
permeability .mu. of the ferrite core of the inductor and the Q are 
basically unstable, and the inductance varies due to expansion and 
contraction of the coil, with the result that it is difficult to maintain 
desired stability of the tuning frequency and the Q. Because of these 
shortcomings, the prior resonator filter has required another temperature 
compensation component or another automatic stability compansation circuit 
for compensating for its instability. 
With the present invention, as described above, the ground terminals or 
common terminals of the electrodes disposed in a confronting relationship 
to each other with the dielectric interposed therebetween are located in 
an opposite relationship or remotely from each other. Thus, a potential 
difference is effectively produced bewteen the transmission-line 
electrodes to form a distributed capacitor, which coacts in parallel with 
a total inductor composed of a distributed-constant inductor and a 
lumped-constant inductor of the transmission lines for constituting an 
equivalent parallel resonator circuit thus realizing as a resonator 
filter. The resonator filter of the present invention has the following 
advantages: 
(1) An inductor and a capacitor can integrally be constructed through a 
simple process and of a simple arrangement composed only of two electrodes 
serving as transmission lines and a single dielectric, so that a resonator 
filter can be achieved which can be handled as one component. 
(2) The resonator filter can be of a very low profile which has not been 
possible heretofore, and is small in size and light in weight, with the 
result that the space factor of the resonator filter is greatly improved. 
Devices in which the resonator filter of the present invention is 
incorporated can also be lower in profile, smaller in size, and lighter in 
weight. 
(3) Since the resonator filter is constructed as a module having no 
mechanically moving parts, any variation in the tuning frequency and the Q 
due to changes in environmental conditions can be greatly reduced. The 
resonator filter is particularly stable against mechanical vibrations. 
(4) The resonator filter is very stable with respect to the temperature 
dependency of the tuning frequency and the Q when the dielectric is made 
of a general material such as an alumina ceramic or a printed-circuit 
board of resin base. It is therefore unnecessary to control the 
temperature dependency of each component, which has been most difficult to 
achieve in designing conventional resonator filters. As a result, it is 
quite easy to design and control the temperature dependency of the 
resonator filter. Due to the stable temperature dependency and the 
stability against mechanical vibrations, the reliabity of the resonator 
filter itself and the devices in which the resonator filter is 
incorporated can greatly be increased. 
(5) As compared with microstrip lines (a transmission line with a 
transmission-line electrode formed on one surface of a dielectric and a 
wide ground electrode on the other surface of the dielectric) which have 
widely been used heretofore, a loss due to the closely located ground 
terminal can be made smaller, so that the resonator filter has a 
sufficiently high Q. 
(6) With the inductor and the capacitor being of an integral construction, 
it is not necessary to provide the resonator filter with unwanted 
connector leads. Thus, any unstable elements can be eliminated which would 
include a lead inductance and a stray capacity generated by the connector 
leads. The tuning operation of the resonator filter can extend in a stable 
fashion into very high frequency ranges. Since no unstable element of a 
reactance component is present with respect to a target design value of 
the tuning frequency, the desired tuning frequency can easily be achieved, 
and hence the resonator filter can simply be designed. The process for 
designing resonator filters can easily be standardized so that the 
resonator filters can highly efficiently be designed. 
(7) Since the resonator filter is constructed of minimum functional 
elements including transmission-line electrodes and a dielectric only, the 
resonator filter can be fabricated of a minimum quantity of material. 
Consequently, the material used can be reduced and the resonator can be 
manufactured at a reduced cost. 
(8) Conventional resonator filters are required to include two components, 
that is, an inductor and a capacitory. The resonator filter of the present 
invention, however, is constructed of a single module component so that 
the number of required parts can be reduced, the cost of assembling the 
resonator filter can be lowered, and the time required to assemble the 
resonator filter can be shortened. Since the types and quantities of 
components in inventory can be reduced, the manufacture control can also 
be simplified. Therefore, the total cost of the resonator filter can 
greatly be reduced. 
(9) The design of resonator filters can be carried out through a simple 
artwork which determines patterns of transmission-line electrodes. No much 
skill is required for the designing of resonator filters, and any design 
changes can easily be coped with. Where resonator filters are to be 
designed in an automatic process, a design procedure relying on computer 
graphics can easily be introduced, since the artwork designing process for 
the resonator filter of the present invention is well suitable for such an 
automated design procedure, and design parameters include only the 
dimensions of transmission-line electrodes and the thickness and 
dielectric constant of the dielectric which are simple and can readily be 
converted into data. With the automatic designing process, resonator 
filters can be designed in a short period of time, with an increased 
accuracy of tuning frequencies, and with an increased degree of design 
freedom. 
FIGS. 15(a) through 15(c) are illustrative of the construction of a 
resonator filter according to still another embodiment of the present 
invention. Electrodes 39, 40 are mounted on opposite surfaces of a 
dielectric 38 in the form of a plate. The electrodes are mounted on the 
dielectrics and have a terminal arrangement in the same manner as 
described with reference to the embodiment of FIGS. 4(a) through 4(c). 
However, each of the electrodes has a spiral configuration. 
If it is desired to adjust the capacitance of the distributed capacitor and 
the inductance of the distributed inductor, the electrode 40 is cut off at 
a portion 40a. 
The severance of the electrode will be described in greater detail. 
By cutting off the electrode, a distributed inductor 105 (FIG. 16(a)) is 
cut off at an electrode portion 105a. As a result, the values of a 
distributed capacitance 106 and the distributed inductance 105 can be 
varied as desired. 
FIG. 16(b) illustrates the resonator filter in the form of a 
lumped-constant equivalent circuit which comprises a parallel-connected 
circuit of a variable inductor 107 and a variable capacitor 108. 
The inductance of the inductor of the resonator filter can be designed as 
desired by adjusting the number of turns of the spiral electrodes or the 
length of the spiral electrodes. The capacitance of the distributed 
capacitor can be designed as desired by adjusting the area in which the 
spiral electrodes confront each other, the dielectric constant .epsilon. 
and thickness of the dielectric. The formation of the distributed 
capacitance will be described further with reference to FIG. 17. The 
confronting spiral electrodes have a transmission-line equivalent length l 
which is designed to be shorter than .lambda./4 at an operating frequency 
taking into account a wavelength shortening coefficient 1.sqroot..epsilon. 
determined by the dielectric constant .epsilon. of the dielectric used. By 
selecting the ratio of the transmission-line equivalent length l to the 
.lambda./4 as desired, the value of a capacitive reactance Xc can be 
designed as desired. A capacitance C=1/2.pi.fo Xc can be determined from 
the capacitive reactance Xc and the operating frequency fo. 
If the transmission-line equivalent length l is shortened to a 
transmission-line equivalent length l', the capacitive reactance Xc is 
changed to a capacitive reactance Xc'. A capacitance C'=1/2.pi.fo Xc' can 
be determined from the capacitive reactance Xc' and the operating 
frequency fo. The capacitance is thus varied since C'&lt;C. The capacitor 
having the capacitance C is equivalent to the variable capacitor 108 shown 
in FIG. 16(b). The length of the spiral electrode (the spiral electrode 40 
in FIG. 15(c)) forming the capacitor electrode which is grounded has been 
illustrated as being the same as the length of the spiral electrode (the 
spiral electrode 39 in FIG. 15(a)) forming the inductor electrode. 
However, the capacitor electrdoe may be of any desired length shorter than 
the inductor electrode, and may be formed at any desired position 
confronting the inductor electrode. 
FIGS. 18, 19, 20 and 21 are illustrative of the manner in which the 
variable capacitor and the variable inductor of the resonator filter of 
FIGS. 15(a) through 15(c) are variably adjusted. 
FIGS. 18 and 19 are explanatory of a mode for adjusting the variable 
capacitor by cutting off the capacitor electrode. FIG. 19 shows the 
relationship between an electrode length d from the open terminal to the 
cut-off position, a distributed capacitance C over the electrode length d, 
a distributed inductance L over the electrode length d, and a 
self-resonant frequency fo for the electrode length d. As the electrode 
length d increases, the distributed capacitance C is reduced but the 
distributed inductance L remains unchanged, and the self-resonant 
frequency fo goes higher. FIGS. 20 and 21 are explanatory of a mode for 
simultaneously adjusting the variable inductor and the variable capacitor 
by cutting off the inductor electrode. FIG. 21 shows the relationship 
between an electrode length d from the open terminal to the cut-off 
position, a distributed capacitance C over the electrode length d, a 
distributed inductance L over the electrode length d, and a self-resonant 
frequency fo for the electrode length d. As the electrode length d 
increases, both the distributed inductance L and the distributed 
capacitance C are reduced, and the self-resonant frequency fo goes higher. 
The electrode may be cut off by a non-contant cutter means such as a laser 
cutter or a sand blaster which does not affect the tuning frequency during 
cutting operation. 
With the above embodiment, the inductor electrode and the capacitor 
electrode are shared by each other and the inductance of the capacitor 
electrode which is grounded is cancelled out, so that the variable 
inductor and the variable capacitor are of an integral construction. 
Each of the electrodes in the above embodiments may be composed of a metal 
conductor, a printed thin metal conductor, a thick film conductor, or a 
thin film conductor, and the confronting electrodes may comprise 
conductors of different types. The electrodes may be disposed in the 
dielectric, rather than on the surface thereof. The electrode surfaces may 
be protected by another dielectric material which hermetically seals the 
electrodes. The dielectric may be made of alumina ceramic, barium 
titanate, plastics, polyethylene fluoride, glass, mica, or in the form of 
a printed-circuit board of resin base. 
As described above, the electrodes are disposed in confronting relation to 
each other with the dielectric base plate interposed therebetween and at 
least one of the electrodes is disposed in the dielectric base plate or 
the electrodes are placed parallel to each other on one surface of the 
dielectric base plate, with the ground terminals of the electrodes being 
located in opposite relation remotely from each other. At least one of the 
electrodes is cut off at a portion thereof. The resonator filter thus 
constructed has the following advantages: 
(1) The variable inducator and the variable capacitor are of a simple 
integral construction. 
(2) The resonator filter is of an extremely low profile and an extremely 
small size. 
(3) Since the resonator filter may be of a module, its tuning frequency 
after adjustment is quite stable, and any shift in the tuning frequency 
due to mechanical vibrations is held to minimum. 
(4) Since the variable inductor and the variable capacitor are 
interconnected without leads, the circuit operation is highly stable 
without being affected by any lead inductance and stray capacity. 
(5) The number of parts used can be reduced, and the manufacturing process 
is improved, with the result that the cost of manufacture can be lowered. 
(6) The capacitor and inductor can be adjusted by a non-contact cutter 
means without affecting the tuning frequency. 
(7) The speed of adjusting the capacitor and inductor is increased. 
The initial values of the inductance and the capacitance for the setting of 
the tuning frequency of the resonator filter are dependent on a simple 
artwork for an electrode pattern, so that the resonator filter can be 
designed with increased freedom and the constants can easily be corrected. 
FIGS. 22(a) through 22(c) are illustrative of the construction of a 
resonator filter according to still another embodiment of the present 
invention. Electrodes 42, 43 are mounted on opposite surfaces of a 
dielectric 41 in the from of a plate. The electrodes are mounted on the 
dielectrics and have a terminal arrangement in the same manner as 
described with reference to the embodiment of FIGS. 4(a) through 4(c). 
However, each of the electrodes has a spiral configuration. 
If it is desired to adjust the capacitance of the distributed capacitor or 
the inductance of the distributed inductor, the electrode 43 is grounded 
at a desired portion 43a. 
Shifting the position wherein the ground terminal is connected will be 
described in greater detail. 
By selecting the position in which the ground terminal is connected as 
shown in FIG. 22(a), the ground terminal is connected to a distributed 
inductor 109 at a desired electrode portion 109a. As a result, the values 
of a distributed capacitance 110 and the distributed inductance 109 can be 
varied as desired. 
FIG. 23(b) illustrates the resonator filter in the form of a 
lumped-constant equivalent circuit which comprises a parallel-connected 
circuit of a variable inductor 111 and a variable capacitor 112. 
The inductance of the inductor of the resonator filter can be designed as 
desired by adjusting the number of turns of the spiral electrodes or the 
length of the spiral electrodes. The capacitance of the distributed 
capacitor can be designed as desired by adjusting the area in which the 
spiral electrodes confront each other, the dielectric constant and 
thickness of the dielectric. The formation of the distributed capacitance 
will be described further with reference to FIG. 24. The confronting 
spiral electrodes has a transmission-line equivalent length l which is 
designed to be shorter than .lambda./4 at an operating frequency taking 
into account a wavelength shortening coefficient 1/.sqroot..epsilon. 
determined by the dielectric constant .epsilon. of the dielectric used. By 
selecting the ratio of the transmission-line equivalent length l to the 
.lambda./4 as desired, the value of a capacitive reactance Xc can be 
designed as desired. A capacitance C=1/2.pi.fo Xc can be determined from 
the capacitive reactance Xc and the operating frequency fo. 
If the transmission-line equivalent length l is shortened to a 
transmission-line equivalent length l', the capacitive reactance Xc is 
changed to a capacitive reactance Xc'. A capacitance C'=1/2.pi.fo Xc' can 
be determined from the capacitive reactance Xc' and the operating 
frequency fo. The capacitance is thus varied since C'&lt;C. The capacitor 
having the capacitance C is equivalent to the variable capacitor 112 shown 
in FIG. 23(b). The length of the spiral electrode (the spiral electrode 43 
in FIG. 22(c)) forming the capacitor electrode which is grounded has been 
illustrated as being the same as the length of the spiral electrode (the 
spiral electrode 42 in FIG. 22(a)) forming the inductor electrode. 
However, the capacitor electrode may be of any desired length shorter than 
the inductor electrode, and may be formed at any desired position 
confronting the inductor electrode. 
FIGS. 25, 26, 27 and 28 are illustrative of the manner in which the 
variable capacitor and the variable inductor of the resonator filter of 
FIGS. 22(a) through 15(c) are variably adjusted. 
FIGS. 25 and 26 are explanatory of a mode for adjusting the variable 
capacitor by positionally adjusting the grounded position of the capacitor 
electrode. FIG. 26 shows the relationship between an effective electrode 
length d from the open terminal 44 to the ground terminal position, a 
distributed capacitance C over the effective electrode length d, a 
distributed inductance L over the effective electrode length d, and a 
self-resonant frequency fo for the effective electrode length d. As the 
effective electrode length d increases, the distributed capacitance C is 
reduced but the distributed inductance L remains unchanged, and the 
self-resonant frequency fo goes lower. FIGS. 27 and 28 are explanatory of 
a mode for simultaneously adjusting the variable inductor and the variable 
capacitor by positionally adjusting the ground terminal position of the 
inductor electrode. FIG. 28 shows the relationship between an effective 
electrode length d from the open terminal 45 to the ground terminal 
position, a distributed capacitance C over the effective electrode length 
d, a distributed inductance L over the effective electrode length d, and a 
self-resonant frequency fo for the effective electrode length d. As the 
effective electrode length d increases, both the distributed inductance L 
and the distributed capacitance C are reduced, and the self-resonant 
frequency fo goes lower. 
With the arrangement of the above embodiment, the inductor electrode and 
the capacitor electrode are shared by each other and the inductance of the 
capacitor electrode which is grounded is cancelled out, so that the 
variable inductor and the variable capacitor are of an integral 
construction. 
Each of the electrodes in the above embodiment may be composed of a metal 
conductor, a printed thin metal conductor, a thick film conductor, or a 
thin film conductor, and the confronting electrodes may comprise 
conductors of different types. The electrodes may be disposed in the 
dielectric, rather than on the surface thereof. The electrode surfaces may 
be protected by another dielectric material which hermetically seals the 
electrodes. The dielectric may be made of alumina ceramic, barium 
titanate, plastics, polyethylene fluoride, glass, mica, or in the form of 
a printed-circuit board of resin base. 
As described above, the electrodes are disposed in confronting relation to 
each other with the dielectric base plate interposed therebetween and the 
ground terminals connected to the electrodes are positioned differently 
for the respective electrodes. The resonator filter of the above 
construction has the following advantages: 
(1) The variable inducator and the variable capacitor are of a simple 
integral construction. 
(2) The resonator filter is of an extremely low profile and an extremely 
small size. 
(3) Since the resonator filter may be of a module, its tuning frequency 
after adjustment is quite stable, and any shift in the tuning frequency 
due to mechanical vibrations is held to minimum. 
(4) Since the variable inductor and the variable capacitor are 
interconnected without leads, the circuit operation is highly stable 
without being affected by any lead inductance and stray capacity. 
(5) The number of parts used can be reduced, and the manufacturing process 
is improved, with the result that the cost of manufacture can be lowered. 
(6) The capacitor and inductor can be adjusted by nondestructive means. 
(7) The capacitance or the inductance can repeatedly be increased and 
reduced. 
(8) The speed of adjusting the capacitor and inductor is increased. 
The initial values of the inductance and the capacitance for the setting of 
the tuning frequency of the resonator filter are dependent on a simple 
artwork for an electrode pattern, so that the resonator filter can be 
designed with increased freedom and the constants can easily be corrected. 
FIG. 29 is a circuit diagram of a balanced resonator filter according a 
still further embodiment of the present invention. The balanced resonator 
filter has a main electrode 46 having a neutral ground terminal 47 and 
forming a distributed inductor, the main electrode 46 also having balanced 
terminals 48, 49. The balanced resonator filter also has an auxiliary 
electrode 50 confronting the main electrdoe 46 with a dielectric (not 
shown) interposed therebetween and having opposite ends serving as ground 
terminals 51, 52, respectively. A hypothetical neutral point can be 
provided to achieve an equivalent ground terminal wihtout having to add 
the ground terminal 47 to the main electrode 46. The balanced terminals 
48, 49 may be located at any positions presenting desired impedances, 
rather than on the ends of the main electrode 46. The ground terminals 51, 
52 of the auxiliary electrode 50 may also be located at any desired 
positions. According to the embodiment of FIG. 29, the balanced resonator 
filter is of a simple construction having the individual main and 
auxiliary electrodes. 
FIG. 30 is a circuit diagram of a balanced resonator filter according 
another embodiment of the present invention. The balanced resonator filter 
has main electrodes 53, 54 having neutral ground terminals 55, 56 and 
balanced terminals 57, 58. The balanced resonator filter also has an 
auxiliary electrode 56 confronting the main electrdoes 53, 54 with a 
dielectric (not shown) interposed therebetween and having a ground 
terminal 60 disposed in substantial alignment with the balanced terminals 
57, 58 of the main electrodes 53, 54. The balanced terminals 57, 58 may be 
located at any positions presenting desired impedances, rather than on the 
ends of the main electrodes 53, 54. With this embodiment, the balanced 
terminals 57, 58 may be disposed closely to each other so that signal 
input and output leads can be of a short length. 
FIG. 31 is a circuit diagram of a balanced resonator filter according still 
another embodiment of the present invention. The balanced resonator filter 
has a main electrode 64 having a neutral ground terminal 61 and balanced 
terminals 62, 63, as with the balanced resonator filter of FIG. 29. 
However, the balanced resonator filter has auxiliary electrodes 67, 68 
having ground terminals 65, 66, respectively. The ground terminals 61, 65, 
66 and the balanced terminals 62, 63 may be arranged in the same manner as 
described with reference to FIG. 29. By cutting off the auxiliary 
electrodes 67, 68 shown in FIG. 31 and the auxiliary electrode 59 shown in 
FIG. 30 at desired positions close to the open terminals, the distributed 
capacitance can be varied to change the tuning frequency and adjust the 
balanced condition. 
FIG. 32 is a circuit diagram of a balanced resonator filter according still 
another embodiment of the present invention. The balanced resonator filter 
includes a tuning section 71 composed of a main electrode 69 and an 
auxiliary electrode 70 as with the embodiment of FIG. 29. The main 
electrode 69 has balanced terminals 74, 75 connected to voltage-variable 
capacitance diodes 74, 75 and balanced terminals of a 
balanced-to-unbalanced mode converter 76 which has an unbalanced terminal 
77. The diodes 74, 76 are connected to a control voltage terminal 78. The 
auxiliary electrode 70 is connected to an unbalanced secondary tap 
terminal 79. Although the tuning section 71 is the same as shown in FIG. 
29, it may be composed of the arrangement shown in FIG. 30 or 31. The 
voltage-variable capacitance diodes 74, 76 and the balanced-to-unbalanced 
mode converter 76 may not be connected together, but may be installed 
individually. With the embodiment of FIG. 32, there can be provided a 
balanced variable resonator filter in which the tuning frequency can be 
varied and which can be connected to an unbalanced signal circuit system. 
The voltage-variable capacitance diodes 74, 75 may be replaced with 
variable air capacitors. Where the tuning section 71 is constructed of the 
arrangement illustrated in FIG. 30 or 31, the auxiliary electrode 59 or 
the auxiliary electrodes 67, 68 may be cut off at desired positions to 
select any desired tuning frequency band and adjust the balanced 
condition. 
FIGS. 33(a), 33(b) and 33(c) and 34(a), 34(b) and 34(c) are illustrative of 
the construction of the main and auxiliary electrodes and a dielectric for 
the balanced resonator filter shown in FIG. 29. 
In FIGS. 33(a) through 33(c), the resonator filter has a dielectric base 
plate 80 and main and auxiliary electrodes 81, 82 forming a 
distributed-constant circuit providing a distributed inductor and a 
distributed capacitor. The main and auxiliary electrodes 81, 82 have 
ground terminals positioned remotely from each other between the 
confronting main and auxiliary electrodes as shown in FIGS. 29 and 31. 
(The ground terminals in FIGS. 34(a) through 34(c) are similarly 
arranged.) Sides A, B shown in FIG. 33(a) correspond respectively to sides 
A, B in FIG. 33(c). (This also holds true in FIGS. 34(a) through 34(c).) 
In FIGS. 33(a) through 33(c), the electrodes 81, 82 of a meandering 
configuration are disposed in a confronting relationship to each other 
with the dielectric base plate 80 interposed therebetween. 
In FIGS. 34(a) through 34(c), electrodes 86, 87 of a spiral configuration 
are disposed in a confronting relationship to each other with a dielectric 
base plate 85 interposed therebetween. 
The manner in which a voltage-variable capacitor is connected to the 
resonator filter shown in FIGS. 7(a) through 7(c) will be described with 
reference to FIGS. 35 and 36 which illustrate overall resonator filters 
according to different embodiments of the present invention. 
FIG. 35 shows an overall resonator filter according to an embodiment of the 
invention (only one electrode shown with the opposite electrode omitted 
from illustration, which holds true also for FIG. 36). A spiral electrode 
91 (which may form an inductor or a capacitor, this being true also for 
FIG. 36) is placed on a dielectric base plate 90 and has a ground terminal 
92. A voltage-variable capacitor 94 has an anode connected bewteen the 
ground terminal 92 and a desired portion of the spiral electrode 91, and a 
voltage control terminal 95 serving as a common cathode of the 
voltage-variable capacitor 94. 
FIG. 36 shows an overall resonator filter according to another embodiment 
of the invention. A spiral electrode 97 is placed on a dielectric base 
plate 96 and has a ground terminal 98. A voltage-variable capacitor 100 
has an anode connected bewteen the ground terminal 98 and an open terminal 
99 of the spiral electrode 97, and a voltage control terminal 101 serving 
as a common cathode of the voltage-variable capacitor 94. 
In the overall resonator filter constructions shown in FIGS. 35 and 36, the 
variable inductor and capacitor and the voltage-variable capacitor are 
connected with no lead connector conductors or by quite short paths. 
While in the embodiments of FIGS. 35 and 36 the voltage-variable capacitors 
94, 100 are of the twin type, they may be of the single type in which case 
a DC voltage blocking capacitor (not shown) is connected between the 
cathode or the voltage control terminal and the electrode forming the 
variable inductor or the variable capacitor. 
With the above embodiments, the inductor electrode and the capacitor 
electrode are shared with each other and disposed in a confronting 
relationship with the dielectric interposed therebetween, the capacitor 
electrode being grounded with its inductance cancelled. One or both of the 
electrodes are cut off or grounded at any desired portions thereof, 
thereby forming a variable inductor and a variable capacitor. The 
voltage-variable capacitor is connected directly or by short paths between 
the ground terminal of one of the confronting electrodes and any desired 
portion thereof. The resonator filter of the above construction has the 
following advantages: 
(1) The variable inductor and capacitor and the voltage-variable capacitor 
are connected with no lead connector conductors or by quite short paths. 
(2) Thus, no unwanted resistance is present in the resonator filter and the 
Q thereof can be improved. 
(3) The circuit operation of the resonator filter is highly stable without 
being affected by any lead inductance and stray capacity. 
(4) The entire resonator filter can be fabricated as a simple, low-profile, 
small-size and integral module, or can be assembled integrally into a 
dielectric circuit board. 
(5) Since the resonator filter as adjusted has no mechanical moving parts, 
any shift in the tuning frequency due to mechanical vibrations can be 
reduced to a minimum. 
(6) It is possible to form the electrodes of the resonator filter 
simultaneously with the circuit of another circuit element. The number of 
parts used can thus be reduced, and the resonator filter can be 
manufactured in an simplified process and at a lower cost. 
(7) Where the tuning frequency of the resonator filter is adjusted by 
cutting off the electrode, a non-contact adjusting means may be employed 
for frequency adjustment without influencing the tuning frequency. 
(8) Where the tuning frequency of the resonator filter is ajusted by 
adjusting the ground terminal position, a nondestructive adjusting means 
may be employed so that the tuning frequency of the resonator can 
repeatedly be adjusted up and down. 
The initial values of the variable inductance and the variable capacitance 
for the setting of the tuning frequency of the resonator filter are 
dependent on a simple artwork for an electrode pattern, so that the 
resonator filter can be designed with increased freedom and the constants 
can easily be corrected. 
While the embodiments shown in FIGS. 9(a), 9(b), 15, 18, 20, 22, 25, 27, 29 
through 36 have been described using respective typical transmission-line 
electrode patterns, all of these embodiments may be achieved, with the 
same advantages, by employing any desired transmission-line electrode 
patterns shown in FIGS. 4(a), 4(b) and 4(c) through 8(a), 8(b) and 8(c).