Two-way paging system and apparatus

Paging systems typically include a network of ground stations or orbiting satellites equipped with transmitters and antennas for transmitting paging messages to small, battery-operated paging receivers that are worn about the person, known as pagers. Conventional pagers are receive-only devices and the network receives no confirmation that paging has been successful. The present invention includes the provision of two-way pagers that contain a transmitter for transmitting an acknowledgment confirming reception of a paging message, and a network of receiving stations able to receive said acknowledgments. The network makes use of advanced phased-array antenna signal processing techniques to render the return link operable with much less transmitter power in the portable unit than employed by the base station.

BACKGROUND 
The present invention relates to systems for electronically alerting people 
using portable devices and transmitting short messages to those devices, 
for example a telephone number that should be called. This field of 
one-way radio communications of short messages is known as paging. 
Conventional paging systems are one-way communications systems in which the 
portable unit consists of only a receiver. Paging services are, however, 
facing increasing competition from two-way cellular phone systems, as the 
latest cellular phones are small, of low cost, and offer long battery life 
and voice communications. The paging services market is therefore 
responding to competition by expressing an interest in moving towards 
two-way communications services, such as short message services or voice 
mailboxes in the network, which services have to be triggered to replay 
the centrally stored message by the paging unit, thereby necessitating 
communication in the reverse direction, i.e., from the paging unit to the 
network. 
However, there are many technical problems in providing reverse 
communication from a paging unit to a base station. Paging broadcast 
transmitters are typically of high power, for example 100 to 250 watts, in 
order to compensate for the low efficiency of the paging receiver antenna 
which is usually worn close to the user's body. The provision of 
capability for pagers to communicate in the reverse direction is thus 
seriously impeded by the need for a similarly high transmitter power to 
close the link in the reverse direction. 
Conventional systems do exist which provide two-way radio communication 
from a portable unit back to a base unit with much lower portable 
transmitter power than used in the forward, i.e. base-to-portable, 
direction. A landmobile radio system such as the EDACS system manufactured 
by Ericsson Inc. in Lynchburg, Va. is an example of such a system. In 
these conventional systems, the reverse link is closed by providing 
multiple, distributed, base station receiving sites so that the range from 
a portable unit to the nearest receiving site is much less than the range 
from the base transmitter to the portable, thus allowing the portable 
transmitter power to be much lower than the base transmitter power. 
However, the provision of multiple receiver sites can be costly. 
Conventional systems also contain examples of two-way communications from a 
portable unit to a base unit using lower power than that used in the 
forward direction, and without using a much greater number of fixed 
receiver sites than fixed transmitter sites. A cellular phone system is an 
example of such a system. Each site, known as a cell-site, comprises at 
least one transmitter and at least one receiver. All sites are thus both 
transmitter sites and receiver sites. To allow the portable unit to employ 
lower power than the base unit while communicating over the same distance, 
the fixed receiving system often provides spatial-diversity reception by 
using two, spaced-apart receiving antennas. This provides a gain of 7 dB 
when signals are subject to Rayleigh fading, allowing the portable unit 
power to be, in principle, five times lower than the base power. 
Conventional cellular systems also include the use of directional antennas 
at the base station. Typically, the base station antenna includes three 
directional antennas placed around a mast at 120 degree intervals and each 
covering three 120 degree sectors or cells. A sectorized cellular system 
can be regarded as having collected together the base sites of three 
adjoining cells, the cells then being illuminated from their common edge 
instead of their separate centers, thus reducing the number of sites by 
three to save real estate costs. The extra communications distance that 
the base must serve, i.e. from cell edge to cell edge instead of 
center-to-edge, is accommodated by the extra directive gain of the 
120-degree sector antenna as compared to the omni-directional antenna that 
would be used in the case of central illumination. Sectorization in 
cellular systems is therefore a way of providing the same performance with 
reduced real estate costs. 
The antenna direction or sector to be used for serving a particular mobile 
phone is determined at call set-up time and adequate time is available to 
establish the sector to be used, due to the relatively long duration of a 
telephone call. The same antenna direction is used for transmission as 
well as for reception in cellular systems. This solution does not, 
however, function for a paging system which broadcasts a paging message 
over the full 360 degree azimuth in which a portable unit may be located 
and which system does not know in advance what antenna direction to use 
either for transmission or reception. In a paging system, messages are 
typically too short to execute a call set-up procedure similar to that 
employed in making a cellular telephone call, and which enable antenna 
directivity to be properly selected to serve the call. 
The above-incorporated by reference parent patent applications disclose 
ways and systems to enhance reception antenna directivity by the use of 
antenna arrays. The parent applications also disclose employing known 
symbol patterns transmitted by mobile transmitters at the base receiver 
site in order to determine the optimum coefficients with which signals 
from the antenna elements can be combined to enhance reception. Moreover, 
these applications also describe using signals received to determine phase 
and amplitude errors between array elements and to correct same on a 
long-term basis. These techniques are further developed below to overcome 
the deficiencies of conventional paging systems that hinder implementation 
of a two-way paging system. 
SUMMARY 
These, and other, drawbacks, limitations and problems are overcome by way 
of the present invention, which has as one objective to reduce the need 
for multiple receiver sites while still allowing a great reduction in 
portable unit transmit power. Two-way paging systems according to 
exemplary embodiments of the present invention include small, portable, 
battery-powered paging units each having a receiver section, a transmitter 
section and a control section. The control section has a low-power timer 
that controls powering up and down of the receiver such that the receiver 
is only active for a low percentage of the time to conserve battery power. 
When powered up, the receiver receives a signal and processes it to decode 
an address. Upon successfully decoding an address, a receive timing 
indication is output to the control section which in turn activates the 
transmitter to generate an acknowledgment signal having a certain timing 
relative to the receive timing. The acknowledgment signal can, for 
example, be a cyclic redunctancy check (CRC) code calculated from the 
received message bits, but can also provide an indication that further 
information follows. 
Paging networks according to the present invention include a network of 
receiver sites that may be collocated with or separated from the paging 
transmitter sites. The receiving sites can each include an array of 
elementary directive antennas whose signals can be jointly processed to 
increase probability of reception. Signals from the antenna elements are 
sampled at certain times after transmission of a paging message in order 
to capture an acknowledgment signal. If the same frequency is employed for 
reception as for transmission, the base transmitters can be inhibited 
during the receive timing window. 
The sampled signals are digitized and stored in a memory and then processed 
by a numerical processor to attempt detection of the acknowledgment. For 
example, the processor can evaluate different hypotheses of where in the 
storage medium samples corresponding to the start of the acknowledgment 
lie (i.e., time of arrival hypotheses); different hypotheses of direction 
from which the acknowledgment was received (i.e., direction of arrival 
hypotheses), and different hypotheses of the radio frequency on which the 
acknowledgment was transmitted (i.e., frequency hypotheses). As an 
alternative to frequency hypothesis, frequency inaccuracies in the 
portable pager device can be avoided by correcting for the frequency using 
a frequency error measured on the signal received by the paging receiver.

DETAILED DESCRIPTION 
A general exemplary schematic of a pager for implementing the present 
invention is shown in FIG. 1. An antenna 10 is coupled alternatively to a 
receiver section 12 or a transmitter section 13 via a transmit-receive 
antenna switch 11. The switch 11 may be constructed, for example, using 
PIN diodes and known techniques whereby the PIN diodes consume no current 
except when the unit is placed in the transmit mode. It is alternatively 
possible to arrange that the PIN diodes do not take current from the 
battery except when the transmitter or receiver are operated. 
It will be appreciated that a significant design issue in pagers is the 
minimization of battery consumption when the pagers are in standby mode. 
To this end, pagers and paging networks are designed to not require 
continuous reception by the portable unit, but cooperate to define a time 
slot in a repetitive frame period in which pages to a particular pager 
would always be transmitted. The pager can then power down its receiver or 
enter a "sleep mode" for the majority of the frame period in order to 
conserve battery energy. The only circuitry that remains active in the 
sleep mode is a low-power timer driven by a low-current oscillator that 
determines when the pager will wake up to examine the paging station's 
transmissions for a possible message containing its address. This timer 
(not shown in FIG. 1) is provided in control section 14 which issues 
power-up control signals to either the receiver or transmitter circuits 
when required. 
When the receiver 12 or transmitter 13 is to be activated, the frequency 
accuracy needed to ensure that the receiver or transmitter functions on 
the correct radio channel frequency is usually higher than the low-power 
timing oscillator in control section 14 can provide. Such low-power 
oscillators, similar to those used in electronic wristwatch chips, 
generally employ a 32 KHz crystal with an accuracy of a few parts in 
100.000. At a transmitter or receiver operating frequency of 500-1000 MHz, 
however, this translates to a frequency inaccuracy of many 10's of 
kilohertz, which is typically unacceptable. To obtain acceptable radio 
frequency accuracy, it is generally necessary to employ a higher frequency 
crystal, in the 10 MHz region, which has a particular angle of cut 
designed to ensure high temperature stability, i.e., an AT-cut crystal. 
Such an oscillator however consumes an undesirably high current from the 
battery on continuous duty and ideally should itself be powered down 
except during transmission or reception. When powered up, such a high 
stability oscillator can, moreover, take several tens of milliseconds to 
reach a stable operating condition due to the high Q factor of the 
high-stability crystal. Instead, a low-power standby timer system which 
operates without using an external 32 KHz crystal and which issues a 
pre-power-up command to the high stability oscillator prior to powering up 
the rest of the receiver, in order to give the oscillation time to 
stabilize can be provided. Moreover, the low-accuracy, non-crystal 
oscillator, which can be entirely constructed on an integrated circuit 
chip, is calibrated against the high-accuracy oscillator 16 at every 
power-up opportunity so that the number of counts the timer must count to 
the next power-up period can be recalculated. 
The control section 14 also connects to a Man-Machine Interface (MMI) 15. 
MMI is a general term used for earphones, microphones, buzzers, displays 
and keyboards that enable the human user to interact with an electronic or 
mechanical device. Typically, a pager MMI comprises a beeper, a display 
and one or two key buttons. The display can show a telephone number that 
the user is being paged to call, or even display longer text messages that 
the user can scroll through with the help of, for example, right/left or 
up/down buttons. 
According to the present invention, the control section 14 also has an 
interface with transmit section 13. When the control section detects its 
address in its receive timeslot and receives a message, the control 
section electronically assembles an acknowledgment or pre-prepared reply 
and activates the transmitter section 13 to transmit the reply. To ease 
the base receiver's task of detecting the reply, the control unit 
preferably causes transmission of the reply at a predetermined time delay 
after receipt of the paging message. The predetermined time delay need be 
no more than the time the receiver needs to verify its address after 
receiving the last message bit, plus the time needed to tune the 
transmitter onto frequency, for example 2 milliseconds to 10 milliseconds. 
It is also possible for the pager to initiate transmission of a short 
message without having been paged according to the present invention. In 
this case, control section 14 activates the transmitter section 13 at a 
predetermined time after receiving a message in the pager's own timing 
slot whether or not the paging slot contained the pager's address. The 
pager may be permitted to transmit its own message at a time when another 
pager may be responding, or alternatively such pager-initiated 
transmissions can be permitted only when the receiving slot contained an 
idle indication. Moreover, a separate receiving frequency channel can be 
provided at the base station with corresponding alternative transmit 
channels at the paging unit for sending respectively an acknowledgment to 
a paging message or a self-initiated message that was not in response to 
detection of the pager's own address. Another system option is to use the 
alternative frequency channel for transmitting self-initiated messages 
only when the receive timeslot contains the address of a different pager, 
and to use the primary reverse channel frequency when the receive timeslot 
contains an idle indication. The base network can respond with an 
acknowledgment of receipt of such a self-initiated message and the pager, 
upon non-receipt of an acknowledgment, can continue to repeat 
transmissions until successful. 
Yet another system option facilitated according to the present invention 
includes using the same frequency for forward and return links. By aging 
paging units to respond with relatively precise transmit burst timing 
relative to a timing marker received from the base station, the network 
can shut off its transmitters during the expected reply window in order to 
receive the low-power responses from pagers. All of the foregoing 
alternatives are considered to be within the scope and spirit of the 
present invention as specified by the attached claims. 
FIG. 2 shows more details of the preferred paging receiver implementation 
in the portable unit. The received signal from T/R switch 11 enters 
through an antenna filter 20 which excludes strong, out-of-band, 
interfering signals. The filtered signal enters a low-noise amplifier 21 
and quadrature downconvertors 22 and 23 where the amplified signal is 
mixed with cosine and sine signals from quadrature oscillator 24 
controlled by frequency synthesizer 35. Integrated circuit chips can be 
made which perform all of the so-called "front-end functions", i.e. those 
performed by elements 21, 22, 23 and 24, in a single chip. 
The output of the front-end chip comprises two baseband signals known as I 
and Q. The I,Q signals are low-pass filtered in active filters 25 and 26 
and amplified in baseband amplifiers 27 and 28. At this point in the 
processing the amount of amplification required is not known as signals 
can be received by the pager anywhere over a wide dynamic range. However, 
the amount of amplification should be limited to avoid a signal strong 
enough to cause saturation. Moreover, since the receiver depicted in FIG. 
2 is a homodyne receiver, it suffers from high DC offset on the outputs of 
mixers 22 and 23 and amplifiers 27 and 28 which further limits the amount 
of amplification which can be used without saturation. These offsets arise 
not only from practical circuit imbalances but also from the receiver 
receiving a signal from its own local oscillator 24, which, in a homodyne 
receiver, is typically in the center of the desired receive frequency 
channel, giving rise to coherent interference. The problem of DC offset 
can be alleviated as described in U.S. Pat. No. 5,241,702 to Paul W. Dent 
entitled "DC offset Compensation" which is incorporated here by reference. 
As discussed in more detail in this patent, the troublesome DC offsets are 
removed by differentiating the I,Q signals using capacitors 29 and 30 and 
then digitizing the differentiated I,Q signals using dual-channel AtoD 
convertor 31. An AtoD conversion technique can be used which employs 
companded delta modulation, which encompasses the differentiation of the 
I,Q signals. 
The differentiated and digitized I,Q signals are passed from AtoD convertor 
31 to digital signal processing unit 32 which can form part of the same 
CMOS integrated circuit as used to realize control section 14. The signal 
processing unit may store the digitized and differentiated I,Q signals in 
memory 33 and then recall those signals for off-line processing. Each 
sample stored would however have been received at an accurately known 
instant relative to a master clock in control unit 14 driven by reference 
crystal oscillator 16, which master clock controls the starting instant of 
AtoD convertor 31 for digitizing samples. Thus off-line signal processing 
does not entail a loss of knowledge of the signal's real-time behavior. 
This knowledge is preserved in order to be able to trigger transmitter 13 
to transmit a reply at a certain time after a specific symbol or signal 
pattern was detected in the received message. 
Signal processing 32 typically includes reintegrating the digitized I,Q 
signals to restore their original characteristics using I and Q digital 
accumulators which are initially set to zero. The signal processing may 
also include estimating an offset error in the re-integrated signals and 
compensating for the offset error according to U.S. Pat. No. 5,241,702 
incorporated above. In addition, a systematic upward or downward slope in 
the reintegrated signals may be compensated. The compensated signals are 
then digitally demodulated to extract the transmitted digital data, the 
data having been impressed by any known digital modulation technique such 
as binary FM, MSK, GMSK, PSK, QPSK, OQPSK, Pi/4-DQPSK, QAM or other 
technique. 
After processing the received signal as described above, the decoded data 
may be stored in memory with a format such as that shown in FIG. 3. There 
a time-of-arrival word 40 indicates the sample number in the memory buffer 
33 that was deemed to contain the start of a particular bit in the 
received message. This might suitably be the first bit of the address 
component 42, for example, in its coded form. The address in coded form 
might include a greater number of bits than the address in its decoded 
form due to the use of an error correction code for transmission which 
expands the number of bits by adding redundancy. Moreover, the expanded 
number of coded bits can be interleaved and transmitted in 
non-time-sequential order in order to provide protection against fading or 
non-Gaussian noise. Nevertheless the transmit format (including the 
interleaving format) is known a-priori to the receiver so that the first 
coded bit of the address can be located in the received signal sample 
stream. The time-of-arrival word 40 indicates the time, in terms of sample 
count, after the start of AtoD converting the signal as triggered by 
control section 14, at which the designated marker in the message was 
detected. Meanwhile, the real-time counter in the control section 14 
continues to increment, so that the count which will be reached at a given 
time after this marker was received can be computed by adding the 
time-of-arrival word to the counter starting state and adding the 
predetermined delay after receipt of the marker at which transmission is 
to occur. The result is compared with the rolling counter and when a match 
is obtained, the transmitter is activated to transmit the acknowledgment. 
FIG. 3 indicates that the memory also contains a frequency error word 41. 
This is a value produced by the data demodulator which is used in the 
demodulation algorithm to compensate for frequency error between the 
frequency transmitted by the base station and the portable unit's 
frequency reference. For example, this value can represent the phase drift 
per bit as computed over the received, coded message bits. This error 
value can be used to correct the output frequency of the portable unit's 
reference oscillator before the transmission of the acknowledgment by the 
transmitter section 13. In exemplary implementations, the frequency error 
word is used, at each receive opportunity, to update a correction value 
that is applied to reference oscillator 16, for example with the aid of a 
DtoA convertor controlling a varactor diode (not shown) across the 
crystal. If, however, significant frequency changes are expected after 
each receive opportunity, as may occur if sleep periods were several 
minutes long instead of just seconds or fractions of a second, then the 
correction assessed on the receive opportunity should be immediately 
applied before transmission, and this can be effected by predistorting the 
transmit signal frequency in the necessary direction, using for example a 
fractional-N transmit frequency synthesizer having five frequency steps. 
The rest of the memory format of FIG. 3 contains the received message, 
including an address 42 which is compared to the pager's own address, some 
other data 43, for example a telephone number to be called, and a CRC 
check code 44. The CRC check code 44 is a number of bits computed at the 
transmitter in dependence on the address and other data bits and is 
recomputed at the receiver in the same way. If the recomputed version 
matches the decoded, stored version 44, then the message is deemed to have 
been correctly decoded. If a correctly decoded message contains the 
pager's own address, then an acknowledgment will be transmitted with the 
transmit timing based on the time-of-arrival word 40. 
FIG. 4 shows a block diagram of an exemplary transmitter for implementing 
the present invention. A transmit power amplifier 50 is driven by a 
transmit oscillator 51 controlled by synthesizer 35, which can be, for 
economy, the same frequency synthesizer chip as used for the receiver 
section 12. The power amplifier 50 and transmit oscillator 51 can be 
packaged as a single module 52. The transmit oscillator signal is applied 
to the frequency synthesis chip 35 rather than the receive oscillator 
signal for transmitter purposes. Although this connection is illustrated 
only generally in FIG. 4, i.e., by an arrow extending from transmit 
oscillator 51 to frequency synthesizer 35, it could, for example, comprise 
two inputs on the synthesizer chip 35 (i.e., one for the receive 
oscillator signal and one for the transmit oscillator signal) of which 
only one is active at a time. Alternatively, a switch such as the T/R 
switch 11 can be used to switch signals into the synthesizer 35, or the 
transmit oscillator signal can be routed through a buffer in the receive 
chip 36 which would select the receive oscillator 24 to be routed to 
synthesizer chip 35 in reception mode and the transmit oscillator 51 to be 
routed to the synthesizer during transmit mode. Preferably, receive 
functions 36, transmit functions 52 and T/R switch 11 are all integrated 
in a combined transmit/receive chip thus avoiding the pin connections 
associated with the signal routing between them. A loop filter 34 is 
provided to reduce spurious signals on the VCO control line while 
permitting rapid retuning of the synthesizer between receive and transmit 
frequencies. Such a synthesizer is described in U.S. Pat. No. 5,180,993 
which is incorporated here by reference. 
The transmit frequency allocated to the pager can differ from the receive 
frequency. The acknowledgment frequency channel may even be in a different 
frequency band, for example UHF for receive and VHF for transmit. 
Nevertheless the synthesizer chip 35 is sufficiently flexible to be 
programmed to control a transmit frequency that is substantially different 
from the receive frequency. 
In switching from controlling the receive frequency to controlling the 
transmit frequency, however, a certain time is allowed for reprogramming 
the synthesizer and allowing the newly controlled oscillator frequency to 
settle on the desired frequency. A fast settling time may be obtained 
using, for example, the techniques described in U.S. Pat. Nos. 5,095,288 
and 5,180,933 which are incorporated here by reference and these 
techniques are embodied in the UM1005 synthesizer chip manufactured and 
sold by the Philips company. A fast settling time is useful in minimizing 
the aforesaid delay between receive and transmit modes during which time 
energy is being drained from the battery. 
Another factor in determining the delay between reception of a signal 
addressed to a pager and transmitting the acknowledgment, is the signal 
processing delay and the time needed by the receiver to check the CRC and 
verify that its own address is part of the received message. Hard-wired 
logic to perform these functions is preferable from both the power 
consumption and time delay viewpoint, but alternatively they may be 
performed by a programmable digital signal processor or microprocessor 
equipped with a suitable program. 
The transmitter activation sequence after detection by the receiver of a 
valid message may thus include, for example: 
Activation of the transmit oscillator; 
Transfer of the synthesizer control function from receive to transmit; 
Switching the antenna from the receiver to the transmitter; 
Ramping up the transmit power from zero to maximum; 
Waiting to allow the ramp-up transient and the synthesizer to settle; 
Applying data modulation to the transmit signal; 
Ramping down the transmit power from maximum to zero; 
Switching the antenna from transmit to receive; 
Switching the oscillator off; 
Switching the synthesizer, and reference oscillator off and returning all 
circuits to sleep mode. 
In the above sequence, the period of high power during which data 
modulation is applied to the transmitter should be as short as necessary 
to convey a useful number of information bits within the bandwidth 
available. The data transmission period should in any case be less than 1 
mS for a transmit frequency in the 1 GHz region to avoid significant 
changes in propagation path phase or amplitude over the transmit period 
caused by fast fading when the pager is being transported at speeds of up 
to 100 kM/Hr. Assuming a 25 KHz channel bandwidth, a bitrate of around 32 
KB/S can be supported by using a spectrally efficient binary digital 
modulation such as Offset QPSK, thus permitting a 32-bit acknowledgment to 
be sent. This can be a rate 1/2, block-coded version of a CRC check 
computed over the received message, which will verify to the base that the 
message was received correctly. Moreover, there is only a 1 in 65536 
chance that noise received at the base receiver would be interpreted 
incorrectly as receipt of an acknowledgment. It is desirable to minimize 
this probability which determines the number of bits transmitted and 
prevents further reductions in the length of the transmit burst. 
A digital data modulator that can produce any desired modulation is 
described, for example, in U.S. patent application Ser. No. 08/305,702, 
entitled "Quadrature Modulator with Integrated Distributed RC Filters" and 
filed on Sep. 14, 1994, which is incorporated here by reference. Such 
techniques may be economic candidates for use in a pager if integrated 
into a single, combined transmit/receive chip as previously referred to. 
Alternatively, a simpler modulation technique called constant-envelope 
OQPSK may be employed without use of such a modulator chip and formed 
within frequency synthesizer 35 instead, as will now be described. 
FIGS. 5(a)-5(c) show signal diagrams for constant envelope OQPSK 
modulation. The radio signal's complex vector value is constrained to move 
only around the constant radius circle of FIG. 5(a) and undergoes eight 
types of transitions associated with even and odd bit periods. 
In even bit periods, the bit information bit carried by the I bit is static 
while the information bit carried by the Q bit either remains the same or 
changes. Since the I bit can be static at 0 or 1 and the Q bit can be 
static at 0 or 1 or change from 0 to 1 or 1 to 0, this defines eight 
possible signal waveform transitions for even bit periods. The Q waveshape 
during a transition is however independent of whether the I bit is a 0 or 
a 1, so there are four possible Q waveforms over the period which are 
illustrated in FIG. 5(b). 
In odd bit periods, the Q bit is static at 0 or 1 while the I bit either 
transitions from 0 to 1 or 1 to 0 or remains unchanged. This leads to four 
possible trajectories for the I signal to take over an odd period as shown 
in FIG. 5(c). Since the sum of the squares of I and Q is at all times 
equal to the constant radius of the circle, when I changes sign it goes 
through zero and Q rises from a magnitude of 1/root(2) to unity at that 
moment. Likewise, if Q changes sign, passing through zero, the magnitude 
of I rises to unity at the instant Q passes through zero in order that the 
sum of the squares remains unity. 
The OQPSK signal can be decoded by using a matched filter which correlates 
received I and Q waveshapes received with expected waveshapes for 
different bit sequences and picks the sequence having the closest 
correlation. However, the method of sampling the I waveform in the middle 
of even bit periods and sampling the Q waveform in the middle of odd bit 
periods may be used, the signs of the samples yielding the data 
information carried by the modulation. 
Since the amplitude (represented by the radius of the circle in FIG. 5(a)) 
of the signal remains constant, only the phase angle of the vector changes 
and this can also be equated to a frequency modulation waveform through 
the relationship that an instantaneous frequency shift is equal to the 
rate of change of phase. Converting the I,Q waveforms to phase waveforms 
using an ARCTAN function and then differentiating those waveforms yields 
the frequency modulating function. The frequency modulation waveform has 
three important values of -B/4 corresponding to a clockwise phase change 
through 90 degrees over a bit period, +B/4 corresponding to an 
anti-clockwise phase change through 90 degrees over a bit period, or zero 
corresponding to two consecutive like I bits and two consecutive like Q 
bits giving rise to no phase change over a bit period. The frequency 
modulating waveform composed of these three values is then applied to a 
low-pass filter to smooth transitions and thus improve the spectral 
containment of the signal energy within an allotted radio channel. A 
suitable filter can, for example, be a raised cosine Nyquist filter with 
an additional X/sin(X) term for waveform shaping in the frequency domain. 
The provision of an additional X/sin(X) term is due to the phase change 
over a bit period being the integral of the frequency waveform over a bit 
period, which mathematical operation is equivalent to a sin(X)/X filter. 
Thus, if the frequency waveform is filtered by a Nyquist filter, the phase 
waveforms will have been filtered by a Nyquist filter plus an additional 
sin(X)/X filter waveform. The additional sin(X)/X filter is thus removed 
by applying the inverse, i.e. an X/sin(X) filter, leaving the phase 
Nyquist filtered waveform. Nyquist filtering of the phase results in the 
phase passing through the points (.+-.l .+-.j ) at bit-period intervals, 
as shown in FIG. 10 which is described in more detail below. 
The waveforms produced by this modulation are closely related to the 
waveforms produced by other constant envelope modulations such as Gaussian 
Minimum Shift Keying (GMSK), which modulation scheme has been specified 
for the Pan-European GSM digital cellular system. A GMSK waveform is shown 
for comparison in FIG. 11 which is also described in more detail below. Of 
course, any type of modulation scheme can be used to implement the present 
invention. 
After low-pass filtering, the frequency modulating function is a continuous 
waveform. Frequency modulation with a continuous waveform may be carried 
out by applying the waveform to a voltage controlled oscillator (VCO). 
When the modulation sensitivity of the oscillator is not exact, however, 
the rate of change of phase created will not be exact and thus the phase 
will rotate too much or too little over a bit period, causing the phase to 
gradually deviate from the desired information-representative values. 
Another problem is that frequency synthesizer 35 will attempt to correct 
any frequency changes that the modulation makes to the VCO's frequency 
causing another source of error. 
These problems may be solved using two-point modulation of the frequency 
synthesizer in which modulation is applied to the VCO and at the same time 
control bits are applied to the synthesizer logic to indicate whether the 
modulation is demanding a +B/4, -B/4 or zero rate of change of phase. In 
this way the synthesizer control loop is prevented from fighting the 
modulation and instead cooperates to effect the desired phase changes. For 
example, a fractional -N synthesizer of the type disclosed in U.S. Pat. 
No. 5,180,993, which disclosure is incorporated here by reference, can be 
used. Next, the base station receiving network's detection of replies from 
the low-power pagers according to exemplary embodiments of the present 
invention will be discussed. 
FIG. 6 illustrates an exemplary vertical collinear army of patch antennas 
for providing vertical beamwidth compression as well as about 5 dB of 
azimuthal directivity. The dimensions illustrated are, of course, 
exemplary. Dual-polarized patch antennas are printed on long printed 
circuit boards together with stripline phasing and coupling lines. Each 
patch 60 operates against a rear ground plane and an optional plane 
(illustrated by dotted lines in FIG. 6) of printed director patches (not 
shown) may be mounted in front of the driven patches to increase directive 
gain and narrow the beamwidth. 
Each such collinear array thus provides two outputs, for example, an output 
corresponding to a left-hand circularly polarized received wave and an 
output corresponding to a right-hand circularly polarized wave. Low-noise 
preamplifiers together with bandpass filters to reject strong out-of-band 
interfering signals may be contained on the printed circuit board close to 
the array elements, to reduce line losses. The complete assembly is 
enclosed in a radio-transparent, weatherproof tube 59. 
FIG. 7 shows more details of the internal connections between antenna 
elements to form the collinear array. A patch 60 can be fed off-center at 
two places, i.e., those connected to lines 66 and 67, spaced by 90 degrees 
relative to the center to provide orthogonal linear polarizations. 
Quadrature coupler 61 couples the two linearly polarized signal outputs to 
form circularly polarized signal outputs. Bandpass filters 62 and 63 
reject unwanted signals that may desensitize low-noise amplifiers 64 and 
65, such as a strong nearby paging transmitter operating on a different 
frequency. 
FIG. 8 illustrates part of a suitable receiver signal processing channel 
which may be connected to either of the preamplified outputs of the 
antenna of FIGS. 6 and 7. A further bandpass filter 70 can be provided, if 
necessary, to attenuate unwanted signals and precedes image-rejection 
amplifier/mixer chip 71. The chip 71 uses an externally supplied first 
local oscillator (LO) frequency to downconvert received signals to a first 
intermediate frequency (IF). The first LO is common to all receiver 
channels to preserve a fixed phase relationship between the first IF 
outputs. The first IF signal is then filtered in IF filter 72 to impose a 
channel bandwidth optimum for receiving replies from the pagers. Further 
amplification and a second downconversion using a common second LO takes 
place in IF chip 73. A suitable IF chip is, for example, the SA637 
manufactured by the Philips company, formerly known as Signetics in the 
USA. The IF chip 73 has two second IF amplifier blocks and provision to 
insert second IF filters 74 and 75 between them to improve selectivity 
against adjacent channel signals. The IF chip also provides a hardlimited 
second IF output, and a signal proportional to the logarithm of the 
instantaneous signal amplitude known as RSSI. These output signals are fed 
to logpolar digitizer 76 which functions, for example, according to the 
disclosure in U.S. Pat. No. 5,048,059 entitled "Logpolar Signal 
Processing", which is incorporated here by reference. The logpolar 
digitization method provides a sequence of numerical samples that 
represent the received signals complex vector value at sequential time 
instants. The numerical samples include bits that represent the 
instantaneous phase angle of the vector and bits that represent the 
logarithm of the amplitude. As will be seen, this format is particularly 
attractive for applying phase rotations and amplitude scalings to the 
outputs of different antennas for the purpose of signal combining to 
enhance directivity of reception. The receiving chain of FIG. 8 can also 
be included on the printed circuit board of the antenna array and housed 
within the weatherproof tube. The output from the antenna in this case 
would be a digital stream of complex numbers in logpolar format 
representing a left hand circular polarized wave and another stream 
representing fight hand circular polarization. The two streams could also 
be multiplexed into one stream to save wires, and even conveyed by optical 
fiber to a central processing point which may be located at the bottom of 
the antenna mast. FIG. 9 shows, for example, eight such collinear arrays 
80 disposed at equal angular intervals around a mast, their output 
logpolar streams being connected to a central signal processing unit 81. 
In practice, a larger number of arrays such as 16, 32 or 64 may be used. 
When a signal wavefront impinges on the array from a particular direction, 
different ones of the collinear antennas 80 will receive the signal at 
different strengths depending on the angle of arrival relative to their 
beam-centre directions, and at a different relative phase that depends on 
the angle of arrival and the position of the antenna. A feature of the 
present invention is that the individual signal processing channels, such 
as those shown in FIG. 8 do not have to be phase matched. Whatever phase 
differences exist between channels there will be a unique set of relative 
phases and amplitudes of the signals received by different antennas for 
each possible direction of arrival. By determining these characteristic 
relative phases and amplitudes adaptively, the signal processing unit 81 
learns how to combine the signals from each antenna with others in order 
to enhance reception in any and all directions. The way in which this 
adaptive learning takes place will be described later. 
Typically, if each antenna already has a restricted azimuthal beam pattern, 
for example being insignificant outside of a 120 degree sector, then only 
one third of the antennas will significantly receive a signal from a given 
direction. For example, antennas 1, 2 and 3 may significantly receive a 
signal from due North (i.e., 0 degrees); antennas 2, 3 and 4 a signal from 
North East; antennas 3, 4 and 5 a signal from East, and so on, in the case 
of an eight-antenna system. To enhance reception from, for example, the 
North, the signals from antenna signal processing chains 1, 2 and 3 should 
be combined in phase. 
Each signal comprises a stream of logpolar complex samples comprising a 
logamplitude L(i) and a phase PHI(i). The phase values are integer binary 
numbers of, for example, eight bits length, and the integer values 
wraparound when modulo 256 arithmetic is used in the same way as phases 
add or subtract in modulo 2.pi. fashion. The phase of a signal may be 
changed therefore by byte-wide addition of an adjustment value THETA(I) to 
PHI(i). 
Multiplicative amplitude scaling also simplifies to integer subtraction of 
a scaling value S(i) from the logamplitude L(i) in the logarithmic domain. 
To form a phased and weighted version of signals 1, 2 and 3 for summing, 
the signal processing unit thus forms: 
L(1)-S(1); PHI(1)-THETA(1) 
L(2)-S(2); PHI(2)-THETA(2) 
L(3)-S(3); PHI(3)-THETA(3) 
using simple integer binary subtractors. 
These modified logpolar complex numbers are then converted to Cartesian 
form with the aid of antilog and cos/sin lookup tables, discussion of 
which may be further found in the aforementioned U.S. Pat. No. 5,048,059. 
Having transformed the modified logpolar values above to Cartesian complex 
form X(i)+jY(i), the sum X1+X2+X3; Y1+Y2+Y3 is formed and represents an 
enhancement of the signal received from a particular direction determined 
principally by the choice of phasing values THETA(i). 
According to the present invention, the signal processing unit maintains a 
matrix of phasing and scaling values in an electronic memory (e.g. a RAM 
chip) corresponding to many different possible directions of arrival. 
Denoting one pair of phasing and scaling values (L(i); THETA(I) by V(i,j) 
where j indicates association direction number j, such a stored matrix is 
of the form: 
______________________________________ 
V11,V21,V31, 0 , 0 , 0 , 0 , 0 
N 
NNE 
0 ,V22,V32,V42, 0 , 0 , 0 , 0 
NE 
ENE 
0 , 0 , V33,V43,V53, 0 , 0 , 0 
E 
ESE 
0 , 0 , 0 ,V44,V54,V64, 0 , 0 
SE 
SSE 
0 , 0 , 0 , 0 ,V55,V65,V75, 0 
S 
SSW 
0 , 0 , 0 , 0 , 0 ,V66,V76,V86 
SW 
WSW 
V17, 0 , 0 , 0 , 0 , 0 ,V77,V87 
W 
WNW 
V18,V28, 0 , 0 , 0 , 0 , 0 ,V88 
NW 
NNW 
______________________________________ 
In the above matrix, the null value 0 indicates that there is no value in 
that position as opposed to the values being zero. The band matrix form 
with interspersed null elements is due to only three antennas contributing 
significantly to receiving from the given direction. Of course those 
skilled in the art will appreciate that antennas could be more or less 
densely provided or could have a more or less restricted azimuth so that 
more or fewer than three antennas could receive significant signal 
components from a source. Only rows corresponding to eight points of the 
compass are shown above for brevity, however, intermediate directions, 
such as WNW, would have corresponding rows and could have four non-null 
entries. 
One such phasing and scaling table is provided for each of the two 
orthogonal polarizations. Since the relation between the different 
polarizations can not be predicted in advance due to the arbitrary 
orientation of the pager's antenna, as well as unknown effects of the 
proximity of the user's body, it will be described later how received 
signals from each polarization are processed together non-coherently. 
The signal processing unit 81 receives a timing synchronization signal from 
the paging transmitter control unit (not shown) that triggers the signal 
processing unit to record logpolar samples in memory from each antenna and 
polarization during a predetermined window. Offline, the processor 81 then 
uses the above stored phasing and scaling matrices to combine the signals 
from antennas corresponding to left hand circular (LHC) polarization 
successively in all the different ways corresponding to different 
hypothetical directions of arrival of a reply from a pager. In parallel, 
corresponding combinations are formed for RHC polarization. The sequences 
of combined samples for both polarizations and a particular direction at a 
time are then processed to try to detect the presence of a reply from a 
pager received from that direction. The method of searching for such a 
reply involves further hypothesizing the exact time of arrival and this 
will now be elaborated with reference to FIG. 10. 
FIG. 10 shows the constant-envelope, OQPSK waveform of FIG. 5, which is 
used in this example to illustrate transmissions from the pager. FIG. 10 
indicates a number of sampling instants used by the receiver chains of 
FIG. 8 for digitizing the received complex signal vector. The AtoD 
convertor 76 is assumed to sample and digitize the signal vector at the 
rate of, for example, eight samples per bit period, and successive samples 
are numbered Z1,Z2 . . . Z10. Samples beyond Z10 have not been illustrated 
in FIG. 10. 
This complex sample sequence is then decimated into a number of bit-spaced 
sample sequences each comprising one sample per bit period, i.e. 
______________________________________ 
Sample phase 1: 
Z1 Z9 Z17 Z25 . . . 
Sample phase 2: 
Z2 Z10 Z18 Z26 . . . 
Sample phase 3: 
Z3 Z11 Z19 Z27 . . . 
Sample phase 8: 
Z8 Z16 Z24 Z32 . . . 
______________________________________ 
Since the reply expected from the pager is known, if Z1 is indeed the 
sample corresponding to the +/-1, +/-j points of FIG. 10, then it is known 
which of the four complex values should be received. Supposing this value 
is (-1+j)/root(2), corresponding to a phase angle of 135 degrees, then Z1 
is rotated through -135 degrees by complex multiplication of that sample 
with exp(-j.pi./4) to give an expected value of 1. This is done 
successively for all points Z9,Z17 . . . etc. belonging to the same sample 
phase, and using the corresponding known data bit pair for that point, the 
rotated complex values being accumulated together to form a complex 
correlation value. Since the rotated value is expected to be 1 in every 
case, the expected correlation value is simply N, where N is the number of 
samples added. 
In practice, transmission through the aether causes an unknown phase shift 
ALPHA, so that the correlation value will not be N but 
N.multidot.exp(jALPHA). The angle of the complex result yields the angle 
ALPHA of the unknown transmission phase, while the magnitude determines 
how well the received waveform of sample phase 1 matched the expected 
response. The calculation is also repeated for the same sample phase and 
corresponding samples from the opposite polarization, and the magnitudes 
of the results for each polarization are added. This process is then 
repeated for all sample phases due to the uncertainty that sample phase 1 
represents the correct timing. The number of sample phases for which 
correlation values using signals received from each polarization should be 
computed depends on the time-of-arrival uncertainty. For example, if the 
pager can be at any distance from 0 to 30 kM from a base station, the 
round trip delay can have a propagation time uncertainty of 0 to 200 
.mu.S. If bits are transmitted at 32 KB/s (30 .mu.S bit periods) then the 
time-of arrival uncertainty is approximately 7 bit periods or 56, 
1/8th-bit samples. The above calculation should thus be repeated with 56 
sample phases. Moreover, the calculation of all 56 sample phases is 
repeated for all possible directions of arrival, that is, by combining the 
signals from the antenna elements using each row of logpolar combining 
coefficients in turn. Due to the matrix combination being a linear 
operation, it can be advantageous to apply the angular rotations needed 
for the complex correlation process to the logpolar values output by 
receiver chains of FIG. 8, while the signals are still in the logpolar 
format and angular rotation is simply performed by integer addition to the 
phase values. The derotated value sequences are then combined using the 
coefficient matrix to produce correlation results, one per sample phase, 
from which the known data modulation has been removed by the derotation 
process. The magnitude squared of a result for one polarization is then 
added to the magnitude squared of a corresponding result for the other 
polarization to yield the polarization-diversity combined correlation 
result for a given direction of arrival and time-of-arrival. If the result 
for a particular direction and time of arrival exceeds a threshold, then a 
reply from the pager is deemed to have been detected. The threshold is 
determined with regard to values produced with incorrect directions or 
times of arrival or when noise alone is known to be present, such that the 
probability of a false detection is remote. 
It will sometimes be desirable to transmit more information from the pager 
than a simple acknowledgment. A variation of the present invention 
includes allowing the pager to reply to a page using either a first code 
or a second code, the codes being chosen to be maximally different, e.g. 
orthogonal codes. The base receiver then performs the above correlation 
process using both codes and whichever yields the largest correlation is 
deemed to be that transmitted. If a first coded is detected, it can 
signify for example that the response is a simple, acknowledgment; if the 
alternate code is detected however, it can be used to signify that other 
information follows. According to another aspect of the invention, the 
base receiving system may in that instance continue to process further 
samples collected in memory from the receiver chains of FIG. 8, but using 
now the single set of matrix combining coefficients corresponding to the 
detected direction of arrival, the sample phase detected to yield the time 
of arrival, and the magnitudes of the corresponding correlations produced 
for each polarization to determine a single way of combining the antenna 
element signals during the further processing, in order to extract other 
information transmitted by the pager. Third and additional codes could 
also be provided to indicate other replies from the pager. Next, an 
exemplary way in which the table of logpolar combining coefficients is 
adaptively learned will be described. 
It will be appreciated that not all pagers will lie at some maximum range 
from a base station and that over the course of even one day many replies 
from locations distributed all over the service region will be processed. 
Many of these will be from locations so close or favorable from a radio 
propagation viewpoint that replies can be detected without the extra, for 
example, 15 dB of directivity afforded by combining many collinear array 
signals together. The present invention may also therefore comprise signal 
processing of non-combined signals from each collinear array, using only 
polarization combination and time-of-arrival testing but without 
direction-of-arrival testing. Those responses detected in this way are 
then used to associate the correlation phases and amplitudes from each 
individual collinear array with that particular direction. Since the 
relative phases and amplitudes corresponding to any particular direction 
can be predicted knowing the antenna geometry, apart from any phase or 
gain mismatches between the receiver channels of FIG. 8, this allows 
re-estimation of the gain and phase mismatches, which may thus be updated 
every time a pager acknowledgment is detected. Using replies detected 
without the extra directive gain allows convergence of the learned 
coefficients even from a poor starting approximation, but once reasonable 
values have been learned, even replies detected using the array gain can 
be processed to determine if the coefficients had drifted slightly, and 
might be adjusted. The mathematics involved in such mobile assisted array 
calibration procedures may be formulated by a person of normal skill in 
the art as guided by the disclosures in the parent applications. Such 
mathematical operations may be performed offline by a low-cost 
microprocessor or other suitable computer as the array components are not 
expected to change characteristics rapidly. Since the system does not 
necessarily know whether incoming signals can be processed in an 
uncombined manner, it could, for example, first look for pager replies on 
each uncombined antenna signal and, if no reply is detected, then process 
the signals in the combined manner described above. Since the received 
signals are stored in memory, this iterative process can be readily 
accommodated. 
Using the above described invention, it is disclosed how collinear arrays 
of antennas having a reduced beamwidth in elevation and corresponding 
directive gain limited only by the vertical stacking dimension can be 
constructed. Moreover, such antennas can have a reduced beamwidth in 
azimuth (for example 120 degrees) and a corresponding additional directive 
gain of 5 dB for example. Finally it has been revealed how a circular 
disposition around a mast of a number of such collinear antennas, for 
example 32, can be used to obtain an additional directive gain of for 
example 9 dB, by combining for example the 11 antennas in any 120 degree 
sector together using a logpolar coefficient matrix. Finally, at least a 3 
dB further gain is achieved by using both polarizations, and thus avoiding 
the 3 dB or more polarization loss that normally is allowed for when the 
pager antenna is arbitrarily orientated. The total gain of about, for 
example, 17 dB compared to a paging transmitter antenna of the same 
vertical aperture, but omnidirectional in azimuth, allows at least a 50:1 
reduction in transmitter power in the reverse direction of pager to base 
transmission. By forming 32-bit correlations at the base receiver using 
the known, expected bit pattern of a pager reply, a further gain is 
obtained relative to transmitting bit-wise information in the forward 
direction. The present invention thus provides the possibility of reliably 
detecting replies from pagers that transmit very short bursts, for 
example, of only 1 watt RF power over paths that require, for example, 100 
watts or more for communication in the forward direction. 
A further capability provided by the present invention is to convey 
correlation values calculated at several different base receiving points 
to a central processing point and to combine the squared magnitudes of 
correlations from different base receivers. The correlation values from 
different bases that are combined should correspond to time-of-arrival and 
direction of arrival hypotheses consistent with the same hypothesis of 
pager location. The triangular region bounded by three base sites can, for 
example, be divided into a number of smaller regions, for example 
hexagonal cells, of dimension corresponding to one, 1/8th-bit sample 
delay, that is for example 4 .mu.S or 1.2 kM across. If the base sites 
are, for example, 60 kM apart, the triangular region bounded by them will 
comprise roughly 1600 such smaller regions. The pager can be postulated to 
be located in each of these smaller cells in turn and the direction of 
arrival and relative Time of arrival at each site predicted. This will 
determine which DOA/TOA-corresponding correlations from one site should be 
combined with those of another site in order to detect the pager's signal. 
In practice, since pagers located near one or another site would be 
expected to be strongly received using that site alone, not all possible 
locations will need the benefit of multiple-site reception to enhance 
pager signal detection. An example of this aspect of the present invention 
will now be presented with reference to FIG. 12. 
FIG. 12 shows an exemplary scenario comprising a triangular service region 
123 bounded by base station sites 1, 2 and 3 respectively; region 234 
bounded by base station sites 2, 3 and 4, and so on. These relatively 
large triangular regions, e.g., on the order of tens of kilometers on a 
side, are imagined to be divided into smaller regions or "cells" of 
perhaps 1 kilometer in diameter as typified by the cell marked "X" in 
region 123. 
If the network pages a particular unit and wishes to test the hypothesis 
that a reply was received from that unit and that the unit is currently 
located in cell X, it can be seen that the network will expect the reply 
to be received at base 1 from the direction SSE +5 degrees and since the 
distance is also known to be between 19 and 20 kM one way, the exact 
time-of-arrival can be predicted with a loop delay uncertainty of, for 
example, +4 uS or +1/8th of a symbol period, due to the inventive pager's 
feature of accurately timing transmission of an acknowledgment relative to 
receipt of a correct address. Likewise, the directions of arrival and 
relative times-of-arrival at bases 2 and 3 can be predicted, so that 
signals received respectively at bases 1, 2 and 3 by different antenna 
array elements and processed into memory, can first be subject to 
combining the signals from the antenna elements of the same site and 
polarization using the postulated direction of arrival at that site. Then, 
after correlation with the expected acknowledgment code with a time 
alignment for each site derived from the postulated time-of-arrival at 
that site, the magnitudes of the correlations obtained at different sites 
and with different polarizations are added to obtain a composite 
correlation value using the signals received at all sites. This may be 
repeated using other codes such as a code indicating further data was 
transmitted and one or more orthogonal codes or dummy codes to obtain a 
threshold value, and the composite magnitudes compared against the 
threshold value to determine if the acknowledgment was received, the 
message flag was received, or no reply was received. An alternative way of 
determining a threshold against which correlations are compared is to add 
the magnitudes of the received signal samples correlated with the 
acknowledgment code, this yielding the largest possible correlation that 
could be achieved. If the actual correlation is not too far below the 
maximum, the acknowledgment is deemed to have been detected. A person 
skilled in the art will be able to carry out computer simulations to 
predict the probabilities of correct detection, missed detection and false 
detection at various signal to noise ratios with different choices of the 
detection thresholds mentioned above in order to determine optimum values 
according to his specification criteria. 
Using the technique disclosed above of hypothesizing pager location, the 
number of direction-of-arrival hypotheses times the number of 
time-of-arrival hypotheses which have to be made is reduced while at the 
same time obtaining the benefit of multiple receiver site detection. This 
benefit is obtained by passing signals received into memory at more than 
one site to one or more common processing points. For example, all signals 
received at all sites could be forwarded to a common signal processing 
node in the network. Alternatively, in the interests of network 
homogeneity, each site could contain signal processing and could receive 
signals from all its nearest neighbor sites for the purpose of executing 
the above location hypothesis-testing algorithm. Thus according to this 
aspect of the invention, a reduction in processing, the benefits of 
multi-site detection and the approximate location of the replying unit are 
accomplished. Once the location has been established, if more information 
is to be forwarded to the unit, the network is able to choose the best 
base station transmitter for this purpose or even utilize two or more 
transmitters to effect a diversity transmission to enhance probability of 
correct reception. Diversity transmission of digital data is accomplished 
according to the best known art by employing a deliberate time offset 
between multiple transmitters such that their signals are received with a 
time offset of one or more whole digital data symbol periods at the 
receiver. The receiver then preferably employs a Viterbi equalizer to 
combine the signals from the transmitters. This scheme can be hard to 
implement in, for example, digital cellular systems, which do not normally 
compute the mobile unit's position, and therefore do not know the 
propagation delays from each transmitter to the mobile receiver. Using the 
above-described exemplary embodiment of the present invention, however, 
the network specifically derives the mobile pager unit's position or time 
delay from each transmitter and so is able to use this information to 
control the time offsets of a multi-site diversity transmission in an 
optimum manner. 
The processing described above makes use of complex signals in logpolar 
format which is particularly suited for implementing the processes 
described in simple digital logic chips adapted for fast, short 
word-length digital arithmetic. Such chips can be constructed as 
Application-Specific, Integrated Circuits (ASIC) and according to one 
aspect of the invention an ASIC chip is disclosed to be suitable for 
making use of logpolar signal processing to combine radio signals from the 
elements of an antenna array in order to enhance directivity. Such a chip 
can form these combinations very rapidly and can thus sequentially form 
combinations corresponding to many different directions of reception. Such 
a chip is called a beamformer, and the preferred implementation is called 
a logpolar beamformer, although other implementations are possible using 
digital signal processors that can perform complex multiplications between 
numbers in Cartesian representation. 
The novel two-way paging system disclosed herein is moreover not restricted 
to being able to detect a reply from a single pager at a time. The 
beamformer and correlation processes described above can be programmed to 
search for a first code expected to be received from a first pager and a 
second code expected to be received from a second pager. Whichever is 
detected at the highest correlation level is noted first and its 
contribution to the signals from each collinear array, as indicated by the 
partial correlations may then be subtracted before continuing to search 
for replies on the same frequency from other pagers, thus applying the 
principles disclosed in U.S. Pat. No. 5,151,919 entitled "CDMA Subtractive 
Demodulation" which disclosure is incorporated here by reference, and it 
will be recognized by a person skilled in the art that the inventive 
correlation with a known code expected to be transmitted from a pager is 
analogous to despreading a CDMA signal using a designated access code, to 
which the incorporated patent is applicable. Even signals transmitted from 
different pagers using the same code and the same radio frequency can be 
distinguished by direction of arrival using the invention and separately 
coded, or alternatively jointly decoded when direction of arrival 
separation is inadequate, as disclosed in the parent applications. 
The invention is suitable for use either when the portable paging unit is 
allocated a separate frequency band for transmission compared to 
reception, or shall use the same frequency band. The precise timing of 
acknowledgment or message transmission from the pager relative to signals 
received from the base network by the pager allow the network to 
accurately anticipate the timeslots in which signals will be received in 
the reverse direction, and the network can shut-off its own transmitters 
during those short timeslots to avoid interfering with its own receiving 
channels. 
While the present invention has been described in terms of the foregoing 
exemplary embodiments, the present invention is capable of many variations 
and modifications apparent to one skilled in the art. All such variations 
and modifications are deemed to fall within the spirit and scope of the 
invention as described in the following claims.