Call maintainance during position location

The present invention is a novel and improved method and apparatus for performing position location in wireless communications system. In one embodiment the invention comprises a method for performing position location in a subscriber unit in a CDMA wireless communications system having a base station, including the step of receiving a position location request during a communication, entering a position location mode, transmitting frames to the base station while performing a position location procedure, and returning to communications mode when said position location procedure has been completed.

BACKGROUND OF THE INVENTION
 I. Field of the Invention
 The present invention relates to position location. More particularly, the
 present invention relates to a novel and improved method and apparatus for
 performing position location in wireless communications system.
 II. Description of the Related Art
 Both government regulation and consumer demand have driven the demand for
 position location functionality in cellular telephones. The global
 positioning system (GPS) is currently available for performing position
 location using a GPS receiver in conjunction with a set of earth orbiting
 satellites. It is therefore desirable to introduce GPS functionality into
 a cellular telephone.
 Cellular telephones, however, are extremely sensitive to cost, weight and
 power consumption considerations. Thus, simply adding additional circuitry
 for performing GPS location is an unsatisfactory solution for providing
 position location functionality in a cellular telephone. Thus, the present
 invention is directed to providing GPS functionality in a cellular
 telephone system with a minimum of additional hardware, cost and power
 consumption.
 SUMMARY OF THE INVENTION
 The present invention is a novel and improved method and apparatus for
 performing position location in wireless communications system. In one
 embodiment the invention comprises a method for performing position
 location in a subscriber unit in a CDMA wireless communications system
 having a base station, including the step of receiving a position location
 request during a communication, entering a position location mode,
 transmitting frames to the base station while performing a position
 location procedure, and returning to communications mode when said
 position location procedure has been completed.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
 A novel and improved method and apparatus for performing position location
 in wireless communications system is described. The exemplary embodiment
 is described in the context of the digital cellular telephone system.
 While use within this context is advantageous, different embodiments of
 the invention may be incorporated in different environments or
 configurations. In general, the various systems described herein may be
 formed using software controlled processors, integrated circuits, or
 discreet logic, however, implementation in an integrated circuit is
 preferred. The data, instructions, commands, information, signals, symbols
 and chips that may be referenced throughout the application are
 advantageously represented by voltages, currents, electromagnetic waves,
 magnetic fields or particles, optical fields or particles, or a
 combination thereof. Additionally, the blocks shown in each block diagram
 may represent hardware or method steps.
 FIG. 1 is a block diagram of the Global Positioning System (GPS) waveform
 generator. The circle with a plus sign designates modulo-2 addition. In
 general, the GPS constellation consists of 24 satellites: 21 space
 vehicles (SVs) used for navigation and 3 spares. Each SV contains a clock
 that is synchronized to GPS time by monitoring ground stations. To
 determine a position and time, a GPS receiver processes the signals
 received from several satellites. At least 4 satellites must be used to
 solve for the 4 unknowns (x, y, z, time).
 Each SV transmits 2 microwave carriers: the 1575.42 MHz L1 carrier, which
 carries the signals used for Standard Positioning Service (SPS), and the
 1227.60 MHz L2 carrier, which carries signals needed for Precise
 Positioning Service (PPS). PPS is used by governmental agencies and allows
 a higher degree of accuracy in positioning.
 The L1 carrier is modulated by the Coarse Acquisition (C/A) code, a
 1023-chip pseudorandom code transmitted at 1.023 Mcps that is used for
 civil position location services. (The Coarse Acquisition code should not
 be confused with the coarse and fine acquisitions described herein, which
 both involve the use of the C/A codes.) Each satellite has its own C/A
 code that repeats every 1 ms. The P code, which is used for PPS, is a
 10.23 MHz code that is 267 days in length. The P code appears on both
 carriers but is 90 degrees out of phase with the C/A code on the L1
 carrier. The 50 Hz navigation message, which is exclusive-ORed with both
 the C/A code and P code before carrier modulation, provides system
 information such as satellite orbits and clock corrections.
 Each satellite has a different C/A code that belongs to a family of codes
 called Gold codes. Gold codes are used because the cross-correlation
 between them is small. The C/A code is generated using two 10-stage shift
 registers. A G1 generator uses the polynomial 1+X.sup.3 +X.sup.10, while a
 G2 generator uses the polynomial 1+X.sup.2 +X.sup.3 +X.sup.6 +X.sup.8
 +X.sup.9 +X.sup.10. The C/A code is generated by exclusive ORing the
 output of the G1 shift register with 2 bits of the G2 shift register.
 FIG. 2 is a highly simplified block diagram of a cellular telephone system
 configured in accordance with the use of the disclosed method and
 apparatus. Mobile telephones 10 are located among base stations 12, which
 are coupled to base station controller (BSC) 14. Mobile switching center
 MSC 16 connects BSC 14 to the public switch telephone network (PSTN).
 During operation, some mobile telephones are conducting telephone calls by
 interfacing with base stations 12 while others are in standby mode.
 As described in copending U.S. patent application Ser. No. 09/040,501 now
 U.S. Pat. No. 6,081,229 entitled "SYSTEM AND METHOD FOR DETERMINING THE
 POSITION OF A WIRELESS CDMA TRANCEIVER" assigned to the assignee of the
 present invention and incorporated herein by reference, position location
 is facilitated by the transmission of a position request message
 containing "aiding information" that allows the mobile telephone to
 quickly acquire the GPS signal. This information includes the ID number of
 the SV (SV ID), the estimated code phase, the search window size around
 the estimate code phase, and the estimated frequency Doppler. Using this
 information, the mobile unit can acquire the GPS signals and determine its
 location more quickly.
 In response to the aiding message, the mobile unit tunes to the GPS
 frequency and begins correlating the received signal with its locally
 generated C/A sequences for the SVs indicated by the base station. It uses
 the aiding information to narrow the search space and compensate for
 Doppler effects, and obtains pseudo-ranges for each satellite using time
 correlation. Note that these pseudo-ranges are based on mobile unit time
 (referenced from the CDMA receiver's combiner system time counter), which
 is a delayed version of GPS time.
 Once this information is calculated, the mobile unit sends the
 pseudo-ranges for each satellite (preferably to 1/8 chip resolution) and
 the time the measurements were taken to the base station. The mobile unit
 then retunes to CDMA to continue the call.
 Upon, receipt of the information, the BSC uses the one-way delay estimate
 to converts the pseudo-ranges from mobile unit time to base station time
 and computes the estimated position of the mobile unit by solving for the
 intersection of several spheres.
 Another parameter provided by the aiding message is the frequency Doppler
 or Doppler offset. The Doppler effect manifests as an apparent change in
 the frequency of a received signal due to a relative velocity between the
 transmitter and receiver. The effect of the Doppler on the carrier is
 referred to as frequency Doppler, while the effect on the baseband signal
 is referred to as code Doppler.
 In the GPS case, frequency Doppler changes the received carrier frequency
 so the effect is the same as demodulating with a carrier offset. Since the
 base station's GPS receiver is actively tracking the desired satellite, it
 knows the frequency Doppler due to satellite movement. Moreover, the
 satellite is so far away from the base station and the mobile unit that
 the Doppler seen by the mobile unit is effectively the same as the Doppler
 seen by the base station. In one embodiment of the invention, to correct
 for the frequency Doppler value, the mobile unit uses a rotator in the
 receiver. The frequency Doppler ranges from -4500 Hz to +4500 Hz, and the
 rate of change is on the order of 1 Hz/s.
 The effect of the code Doppler is to change the 1.023 Mhz chip rate, which
 effectively compresses or expands the width of the received C/A code
 chips. In one embodiment of the invention, the mobile unit correct for
 code Doppler by multiplying the frequency Doppler by the ratio
 1.023/1575.42. The mobile unit can then correct for code Doppler over time
 by slewing (introducing delay into) the phase of the received IQ samples
 in 1/16 chip increments as necessary.
 FIG. 3 is a block diagram of the receiver portion of a cellular telephone
 (wireless subscriber unit) configured in accordance with one embodiment of
 the invention. The received waveform 100 is modeled as the C/A signal c(n)
 modulated with a carrier at frequency w.sub.c +w.sub.d, where w.sub.c is
 the nominal carrier frequency 1575.42 MHz, and w.sub.d is the Doppler
 frequency created by satellite movement. The Doppler frequency ranges from
 0 when the satellite is directly overhead, to about 4.5 kHz in the worst
 case. The receiver analog section can be modeled as demodulation with a
 carrier at frequency w.sub.r and random phase .theta., followed by low
 pass filtering.
 The resulting baseband signal is passed through an A/D converter (not
 shown) to produce digital I and Q samples, which are stored so that they
 may be repeatedly searched. The samples are generated at two times the C/A
 code chip rate (chip.times.2) which is a lower resolution than necessary
 to perform the fine search algorithm, but which allows 18 ms of sample
 data to be stored in a reasonable amount of memory. In general, it is
 desirable to perform the searching over something greater than 10 ms in
 order to allow acquisition in most environmental conditions, with 18 ms
 being a preferred integration period. These environmental conditions
 include being inside or not having a direct view to the satellite.
 During operation, the samples are first rotated by rotator 102 to correct
 for the Doppler frequency offset. The rotated I and Q samples are
 correlated with various offsets of the satellite's C/A sequence and the
 resulting products are coherently integrated over Nc chips by integrators
 104. The coherent integration sums are squared and added together to
 remove the effect of the unknown phase offset .theta.. To augment the
 hypothesis test for a particular offset, several coherent intervals are
 non-coherently combined. This despreading is performed repeatedly at
 various time offsets to find the time offset of the satellite signal.
 Rotator 102 removes the frequency Doppler created by satellite movement.
 It uses the Doppler frequency specified by the base station (preferably
 quantized to 10 Hz intervals) and rotates the I and Q samples to remove
 the frequency offset.
 In one embodiment of the invention, the rotation is continuous only over
 the coherent integration window. That is, the rotator stops in between
 coherent integration periods of, for example, 1 ms. Any resulting phase
 difference is eliminated by the square and sum.
 FIG. 4 is another block diagram of a receiver configured in accordance with
 one embodiment of the invention, where the rotator portion of the receiver
 is depicted in greater detail.
 FIG. 5 is a receiver configured in accordance with an alternative
 embodiment of the invention. This internal embodiment of the invention
 takes advantage of the ability to stop the rotator between coherent
 integration periods by rotating the locally generated C/A sequence instead
 of the input samples.
 As shown, the C/A sequence c(n) are rotated by application to the sinusoids
 sin(W.sub.d nT.sub.c) and cos(W.sub.d nT.sub.c) and then stored. The
 rotation of the C/A sequence only needs to be done once for each
 satellite. Thus, rotating the C/A sequence reduces the amount of
 computation required. It also saves memory in the DSP used to perform this
 computation in one embodiment of the invention.
 Another significant impairment that degrades the performance of the
 position location algorithm is the frequency error in the mobile units
 internal clock. It is this frequency error which drives the use of short
 coherent integration times on the order of 1 ms. It is preferable to
 perform coherent integration over longer time periods.
 In an exemplary configuration, the mobile's free running (internal) local
 oscillator clock is a 19.68 MHz crystal that has a frequency tolerance of
 +/-5 ppm. This can cause large errors on the order of +/-7500 Hz. This
 clock is used to generate the carriers used for demodulation of the GPS
 signals, so the clock error will add to the signal acquisition time.
 Because the time available to search is very small, error of this
 magnitude due to the frequency tolerance are not tolerable and must be
 greatly reduced.
 To allow longer coherent integration times, in one embodiment of the
 invention, the CDMA receiver corrects for local oscillator error by using
 timing acquired from the CDMA pilot, or any other source of timing
 information available. This produces a control signal that is used to tune
 the local oscillator clock to 19.68 MHz as closely as possible. The
 control signal applied to the local oscillator clock is frozen when the RF
 unit switches from CDMA to GPS. Even after the correction is performed
 using the timing information from the base station (or other source),
 however, some additional clock error remains. In one embodiment of the
 invention, the resulting frequency uncertainty after correction is +/-100
 Hz. This remaining error still reduces the performance of the receiver,
 and in general prevents longer coherent integration times. In one
 embodiment of the invention, the remaining error simply avoided by
 performing non-coherent integration for duration of more than 1 ms which
 reduces performance.
 As also shown in FIG. 1, the 50 Hz NAV/system data is also modulated onto
 the L1 carrier. If a data transition (0 to 1 or 1 to 0) occurs between the
 two halves of a coherent integration window, the resulting coherent
 integration sum will be zero because the two halves will cancel each other
 out. This effectively reduces the number of non-coherent accumulations by
 one in the worst case. Although the data boundaries of all the satellites
 are synchronized, they do not arrive at the mobile unit simultaneously
 because of the differences in path delay. This path delay effectively
 randomizes the received data phase.
 In one embodiment of the invention, the solution to the problem of
 different data phases on different signals is to include the data phase in
 the aiding information sent from the base station to the mobile unit.
 Since the base station is demodulating the 50 Hz data, it knows when the
 data transitions occur for each satellite. By using knowledge of the
 one-way delay, the base station can encode the data phase in, for example,
 5 bits (per satellite) by indicating which one millisecond interval (out
 of 20) the data transition occurs on. If the coherent integration window
 straddles the 50 Hz data boundary the coherent integration is divided into
 two (2) sections. One section preceding the data boundary and one section
 following the data boundary. For example, if En1 is the coherent
 integration sum over the window preceding the data boundary the first half
 of this window and En2 is the coherent integration sum over the window
 following the data boundary, the mobile unit then selects the maximum (in
 magnitude) of (En1+En2) (in case the data stayed the same) and (En1-En2)
 (in case the data changed) to account for the phase change. The mobile
 unit also has the option of performing non-coherent combining of the two
 halves over this data window or avoiding this data window completely.
 In an alternative embodiment of the invention, the mobile unit attempts to
 find the data transitions without the aiding information from the base
 station by comparing the magnitude squared of the sum and difference in 1
 ms coherent integration.
 In one embodiment of the invention, a firmware-based DSP (Digital Signal
 Processor) approach is used to perform the GPS processing. The DSP
 receives I and Q samples at a chip.times.2 (2.046 MHz) or chip.times.8
 (8.184 MHz) rate, and stores a snapshot of 4-bit I and Q samples in its
 internal RAM.
 In the exemplary embodiment, the DSP generates the C/A sequence, performs
 rotation to eliminate frequency Doppler, and correlates over the search
 window provided by the base station for each of the satellites. The DSP
 performs coherent integration and non-coherent combining and slews an IQ
 sample decimator as necessary to compensate for code Doppler.
 To save computation and memory, the initial search is performed using 1/2
 chip resolution and a finer search to obtain 1/8 chip (higher) resolution
 is performed around the best index (or indexes). System time is maintained
 by counting hardware-generated lms interrupts (derived from local
 oscillator).
 Additionally, in one embodiment of the invention, the fine search is
 performed by accumulating the chip.times.8 samples (higher resolution)
 over the duration of one chip at various chip.times.8 offsets. The
 correlation codes are applied to the accumulated values yielding
 correlation values that vary with the particular chip.times.8 offset. This
 allows the code offset to be determined with chip.times.8 resolution.
 FIG. 6 is a flow chart illustrating the steps performed to correct for the
 local oscillator error during a position location procedure when performed
 in accordance with one embodiment of the invention. At step 500, it is
 determined whether the local oscillator has been corrected recently. If
 not, then the pilot is acquired from the base station, and error of the
 local oscillator is determined by comparing to the pilot timing at step
 502 and a correction signal generated based on that error.
 The flow then leads to step 504, where the correction signal is frozen at
 the current value. At step 506, enters GPS mode and performs the position
 location using the corrected clock. Once the position location has been
 performed, the mobile unit leaves GPS mode at step 508.
 FIG. 7 is an illustration of a DSP receiver system configured in accordance
 with one embodiment of the invention. The DSP performs the entire
 searching operation with minimal additional hardware. DSP core 308, modem
 306, interface unit 300, ROM 302 and Memory (RAM) 304 are coupled via bus
 306. Interface unit 300 receives RF samples from an RF unit (not shown)
 and provides the samples to RAM 304. The RF samples can be stored at
 coarse resolution or fine resolution. The DSP core 308 processes the
 samples stored in memory using instruction stored in ROM 302 as well as in
 memory 304. Memory 304 may have multiple "banks" some of which store
 samples and some of which store instructions. Modem 700 performs CDMA
 processing during normal mode.
 FIG. 8 is a flow chart of the steps performed during a position location
 operation. A position location operation begins when the aiding messaging
 is received, and the RF systems is switched to GPS frequencies at step
 600. When the RF is switched to receive GPS, the frequency tracking loop
 is fixed. The DSP receives aiding information from the phone
 microprocessor and sorts the satellites by Doppler magnitude.
 At step 602, the coarse search data is stored within the DSP RAM. The DSP
 receives a few hundred microseconds of input data to set an Rx AGC. The
 DSP records the system time and begins storing an 18 ms window (DSP memory
 limitation) of chip.times.2 IQ data in its internal RAM. A contiguous
 window of data is used to mitigate the effects of code Doppler.
 Once the data is stored, a coarse search is performed at step 604. The DSP
 begins the coarse (chip.times.2 resolution) search. For each satellite,
 the DSP generates the C/A code, rotates the code based on the frequency
 Doppler, and correlates over the search window specified by the base
 station, via repeated application of the C/A code to the stored coarse
 search data. Satellites are processed over the same 18 ms data window and
 the best chip.times.2 hypothesis that exceeds a threshold is obtained for
 each satellite. Although a 2 ms coherent integration time (with 9
 non-coherent integrations) is used in one embodiment of the invention,
 longer coherent integration times can be used (for example 18 ms),
 although preferably where additional adjustments are made as described
 below.
 Once the coarse search is performed, a fine search is conducted, at step
 606. Before beginning the fine search, the DSP computes the rotated C/A
 code for each of the satellites. This allows the DSP to process the fine
 search in real-time. In performing the fine (chip.times.8 resolution)
 search, the satellites are processed one at a time over different data.
 The DSP first slews the decimator to compensate for code Doppler for the
 given satellite(s). It also resets the Rx AGC value while waiting for the
 next 1 ms boundary before storing a 1 ms coherent integration window of
 chip.times.8 samples.
 The DSP processes 5 contiguous chip.times.8 resolution hypotheses on this 1
 ms coherent integration window, where the center hypothesis is the best
 hypothesis obtained in the coarse search. After processing the next 1 ms
 window, the results are combined coherently and this 2 ms sum is combined
 non-coherently for all Nn iterations.
 This step (starting from slewing the decimator) is repeated on the same
 data for the next satellite until all the satellites have been processed.
 If the code Doppler for 2 satellites is similar in magnitude, it may be
 possible to process both satellites over the same data to reduce the
 number of required data sets. In the worst case, 8 sets of 2*Nn data
 windows of Ims are used for the fine search.
 Finally, at step 608, the results are reported to the microprocessor and
 the vocoder process is restarted within the DSP so that the call can
 continue. The DSP reports pseudoranges to the microprocessor, which
 forwards them to the base station. After the microprocessor redownloads
 the vocoder program code into the DSP memory, the DSP clears its data
 memory and restarts the vocoder.
 FIG. 9 is a diagram illustrating the fine search performed after the coarse
 search. After isolating the best chip.times.2 phase in the coarse search,
 the DSP performs a fine search around this phase to gain chip.times.8
 resolution.
 The 5 phases to compare in the fine search are shown enclosed by a
 rectangle. The best chip.times.2 phase is evaluated again so that
 comparisons can be made over the same set of data. This also allows the
 coarse search and fine search to use different integration times. The fine
 search is performed separately for each satellite because each satellite
 may have a different value for code Doppler.
 FIG. 10 provides a time line of the search process when performed in
 accordance with one embodiment of the invention. The overall processing
 time (coarse+fine search) is performed in about 1.324 seconds in one
 embodiment of the invention, which does interrupt the call, but still
 allows the call to continue once the search is performed. The total search
 time of 1.324 seconds is an upper bound, because it assumes that the DSP
 needs to search all 8 satellites and each satellite has a search window of
 68 chips. The probability that the entire 1.324 seconds will be necessary
 is small, however, due to the geometry of the satellite orbits.
 During the first 18 ms 80, IQ sample data is collected at the GPS
 frequency. During the period 82, a coarse search is performed internally
 which could last up to 1.13 seconds, but which will probably terminate
 early when the satellite signals are identified. Once the coarse search is
 performed, the C/A codes are computed during time period 84, which takes
 24 ms. During time periods 86 the slew value is adjusted for code Doppler
 and the Rx AGC is further adjusted. During time periods 88, fine searches
 are performed on the IQ data samples, with continuous adjustment performed
 during time periods 86. The use of 18 ms integration times allows code
 Doppler to be neglected because the received C/A code phase will be
 shifted by less than 1/16 of a chip. Up to eight sequences of adjustments
 and fine searches are performed for the up to eight satellites, at which
 time the position location procedure is complete.
 Additionally, in some embodiments of the invention, the phone continues to
 transmit reverse link frames to the base station while the position
 location procedure is performed. These frames may contain null information
 simply to allow the base station to remain synchronized with the
 subscriber unit, or the frames may contain additional information such as
 power control commands or information request. The transmission of these
 frames is preferably performed when GPS samples are not being gathered
 when the RF circuitry is available, or while the GPS samples are gathered
 if sufficient RF circuitry is available.
 Although the use of 18 ms integration time avoids the effects of code
 Doppler, the transmission of data over the GPS signals at 50 Hz rate can
 cause problems if a data change occurs within the 18 ms processing span
 (as described above). The data change causes the phase of the signal to
 shift. The 50 Hz data boundaries occur at different places for each
 satellite. The phase of the 50 Hz transitions for each satellite have been
 effectively randomized by the varying path lengths from each satellite to
 the phone.
 In the worst case, if the data bit was inverted in the middle of a coherent
 integration interval, the coherent integration could be completely wiped
 out. For this reason, in an alternative embodiment of the invention, the
 base station must communicate the data transition boundaries for each
 satellite to the phone (also described above). Preferably, the data
 transmission boundary is also included in the aiding message transmitted
 from the base station (such as in a set of five bit messages indicating
 the millisecond interval during which the transition occurs for each
 satellite). The phone uses this boundary to split the coherent integration
 interval for each satellite into 2 pieces and decide whether to add or
 subtract the coherent integration sums in these 2 intervals. Thus, by also
 including the data boundary of each GPS signal, the reliability of the
 location procedure is increased.
 In the exemplary embodiment of the invention, any frequency uncertainty
 creates a loss in Ec/Nt that increases with the coherent integration time.
 For example, uncertainty of +/-100 Hz, the loss in Ec/Nt increases rapidly
 as the coherent integration time is increased, as shown in Table I.
 TABLE I
 Nc Loss in Ec/Nt
 1023 (1 ms) 0.14 dB
 2046 (2 ms) 0.58 dB
 4092 (4 ms) 2.42 dB
 6138 (6 ms) 5.94 dB
 8184 (8 ms) 12.6 dB
 As also noted above, there is always some unknown frequency offset of the
 local oscillator in the mobile unit. It is this unknown frequency offset
 that prevents longer coherent despreading and integration from being
 performed. Longer coherent would improve processing if the effects of the
 unknown frequency offset could be reduced.
 In one embodiment of the invention, this unknown frequency offset is
 accounted for by expanding the search space to 2 dimensions to include
 frequency searches. For each hypothesis, several frequency searches are
 performed, where each frequency search assumes the frequency offset is a
 known value. By spacing the frequency offsets, one can reduce the
 frequency uncertainty to an arbitrarily small value at the expense of
 added computation and memory. For example, if 5 frequency hypotheses are
 used, the resulting search space is shown in FIG. 10.
 For a +/-100Hz frequency uncertainty, which is the typically operating
 specification of a mobile unit, this configuration reduces the maximum
 frequency offset to 20 Hz (one hypothesis must be within 20 Hz of the
 actual frequency offset). With a 20 ms coherent integration time, the loss
 in Ec/Nt with a 20 Hz frequency offset is 2.42 dB. By doubling the number
 of frequency hypotheses to 10, the frequency uncertainty can be reduced to
 10 Hz, which causes an Ec/Nt loss of 0.58 dB. However, adding additional
 hypotheses widens the search space, which increases both the computation
 and memory requirements.
 One embodiment of the invention computes the frequency hypothesis by
 lumping the frequency offset in with the frequency Doppler, and computing
 a new rotated PN code for each frequency hypothesis. However, this makes
 the number of frequency hypotheses a multiplicative factor in the total
 computation: 5 frequency hypotheses would mean 5 times as much
 computation.
 Alternatively, since the frequency uncertainty is small compared to the
 frequency Doppler, the rotation phase can be considered to be constant
 over a 1 ms interval (8% of a period for an 80 Hz hypothesis) in another
 embodiment of the invention. Therefore, by dividing the coherent
 integration interval up into 1 ms subintervals, the integration sums of
 the subintervals are rotated to reduce the added computations needed to
 compute the frequency searches by three orders of magnitude. The result is
 that longer coherent despreading can be performed, and performance
 improved.
 FIG. 12. is a block diagram of a receiver configured in accordance with the
 use of longer coherent despreading approach. The first set of multipliers
 50 compensates for the frequency Doppler by correlating the IQ samples
 with a rotated C/A code. This is equivalent to rotating the IQ samples
 before correlation with the unmodified C/A code. Since the frequency
 Doppler can be as large as 4500 Hz, the rotation is applied to every chip.
 After coherently integrating over a 1 ms interval (1023 chips) using
 accumulators 52, the second set of multipliers 54 rotates the lms
 integration sums (.SIGMA..sub.I and .SIGMA..sub.Q) to implement the
 frequency hypothesis. The rotated sums are then added over the whole
 coherent integration interval.
 Recall that the frequency Doppler rotation was only computed on 1023 chips
 to save memory and computation. For coherent integration times longer than
 1 ms, each coherent integration sum are multiplied by a phase offset to
 make the phase of the rotation continuous over time. To show this
 mathematically, the 1 ms coherent integration sum with frequency Doppler
 rotation can be expressed as:
 ##EQU1##
 where I(n) and Q(n) are the input samples received on the I and Q channels
 respectively, c(n) is the unrotated C/A code, w.sub.d is the frequency
 Doppler, and T.sub.c is the chip interval (0.9775 us). A 2 ms coherent
 integration sum can be expressed as:
 ##EQU2##
 Here, S.sub.1 is the first Ims integration sum and S.sub.2 is the second 1
 ms integration sum computed using the same rotated C/A values that were
 used to compute S.sub.1. The term e.sup.-jwd(1023)Tc is the phase offset
 that compensates for using the same rotated values. Similarly, a 3 ms
 coherent integration sum can be expressed as
EQU S(3 ms)=S.sub.1 +e.sup.-jw.sup..sub.d .sup.(1023)T.sup..sub.c S.sub.2
 +e.sup.-jw.sup..sub.d .sup.(2046)T.sup..sub.c S.sub.3
 So to extend the integration time while using the same 1023-element rotated
 C/A sequence, the (n+1) 1 ms integration sum should be multiplied by
 e.sup.-jwdn(1 ms) before being added to the whole sum. Since this is a
 rotation of 1 ms integration sums, we can combine this operation with the
 frequency search to avoid having to perform 2 rotations. That is, since
EQU e.sup.-jw.sup..sub.d .sup.n(1 ms) e.sup.-jw.sup..sub.h .sup.n(1 ms)
 =e.sup.31 j(w.sup..sub.d .sup.+w.sup..sub.h .sup.)n(1 ms)
 we can multiply the (n+1)th 1 ms integration sum by e.sup.-j(wd+wh)n(1 ms)
 to search a frequency hypothesis and account for the frequency Doppler
 phase offset.
 Note that the frequency search can be reduced after acquiring one
 satellite, because the frequency uncertainty is not dependent on the
 satellite. A much finer frequency search can be performed if a longer
 coherent integration is desired.
 In the exemplary embodiment of the invention, the fine search is performed
 in similar manner the coarse search with 2 differences. First, the
 integration intervals are always added coherently instead of squaring and
 adding noncoherently. Second, the rotation to remove the frequency
 uncertainty (which should be known after the coarse search) is combined
 with the frequency Doppler phase offset and used to rotate the 1 ms
 coherent integration intervals before adding them together.
 In an alternative embodiment of the invention, the coherent integration
 window of chip.times.2 data is integrated for integration times longer
 than 18 ms. This embodiment is useful were additional memory is available.
 For coherent integrations longer than 18 ms, the 50 Hz data boundaries are
 treated the same as with shorter integration periods. The base station
 indicates where the boundaries are for each satellite and the DSP decides
 whether to add or subtract the sum of 20 1 ms coherent integration
 intervals to or from its running sum.
 However, because the product of the frequency uncertainty and the
 integration time constant affects the loss in Ec/Nt, the frequency
 uncertainty must be reduced to very small levels for long coherent
 integration intervals. Since a 20 ms integration with a 20 Hz frequency
 uncertainty resulted in a loss in Ec/Nt of 2.42 dB, maintaining the same
 loss with an integration time of 400 ms requires that the frequency
 uncertainty be reduced to 1 Hz. To correct for this problem, the frequency
 uncertainty is reduced down to 1 Hz in a hierarchical manner. For example,
 a first frequency search reduces the uncertainty from 100 Hz to 20 Hz, a
 second search reduces the uncertainty to 4 Hz, and a third search reduces
 the uncertainty to 1 Hz. The frequency search will also compensate for
 errors in the frequency Doppler obtained from the base station.
 Additionally, to perform longer integrations only satellites with similar
 Doppler are searched over the same data for long integration times, since
 the code Doppler is different for each satellite. The DSP computes how
 long it takes to slip 1/16 of a chip and slews the decimator as it
 collects a coherent integration data window. Additionally, multiple data
 windows are taken in this embodiment.
 Thus, a method and apparatus for performing position location in wireless
 communications system has been described. The previous description of the
 preferred embodiments is provided to enable any person skilled in the art
 to make or use the present invention. The various modifications to these
 embodiments will be readily apparent to those skilled in the art, and the
 generic principles defined herein may be applied to other embodiments
 without the use of the inventive faculty. Thus, the present invention is
 not intended to be limited to the embodiments shown herein but is to be
 accorded the widest scope consistent with the principles and novel
 features disclosed herein.