Voltage level-shifting control circuit for electronic switch

A control circuit receives complementary logic signals ranging from Vdd to 0 VDC, and outputs a drive signal Vhs1 ranging from magnitude Vhv to Vhv+Vdd to control a high-side switch. The control circuit includes an Ibias generator and a level shift circuit that preferably includes a passive current sink mechanism, coupleable between Vdd and ground. The level shift circuit includes a totem-pole configuration of a PMOS device, an NMOS device, and an NMOS device that mirrors Ibias current. An additional NMOS device is provided, whose source node is coupled between the PMOS and NMOS devices in the totem-pole, whose gate node is coupled to Vdd, and whose drain node serves as an interface to the load circuit. A capacitor coupled across the current source hastens Vhs1 transition time. The PMOS and NMOS devices in the totem pole turn on and off complementarily responsive to the logic signals, which dictate the state of Vhs1.

FIELD OF THE INVENTION

The invention relates generally to control circuits used with electronic switches, and more particularly to control circuits used with logic circuitry whose so-called high-side output signals must be level-shifted to drive an electronic switch.

BACKGROUND OF THE INVENTION

FIG. 1 depicts a prior art H-bridge circuit in which a DC voltage (Vhv) is selectively coupled via switches Sh 1 , Sh 2 , Sl 1 , Sl 2 to a load 10 . In practice, Vhv may be on the order of 100 VDC, although other magnitudes could be used. Switches Sh 1 and Sh 2 are referred to as high-side switches in that they control current flow between the high voltage potential Vhv and the load. These switches control the high side of an H-bridge (when used with such configuration), and their control signals, Vhs 1 , Vhs 2 are referenced to Vs or Vhv, rather than to a common ground that is shared with logic unit 40 . Switches Sl 1 and Sl 2 are typically referenced to system ground and are termed low-side switches.

A control unit 20 must develop high-side and low-side control signals to cause switches Sh 1 and Sl 1 turn on or off, while switches Sh 2 and Sl 2 turn off or on, e.g., switch in complementary fashion. Control unit 20 receives operating potential from a power source 30 , which also powers a logic unit 40 . Although not explicitly shown in FIG. 1 , power source 30 also provides control unit 20 with a higher potential voltage whose magnitude is at least Vhv. It will be appreciated that it is relatively easy for control unit 20 to generate control signal Vls 1 to control switch Sl 1 and control signal Vls 2 to control switch Sl 2 , as switches Sl 1 , Sl 2 are referenced to ground. However it is a more challenging task for control unit 20 to generate control signals Vhs 1 and Vhs 2 to control high-side switches Sh 1 , Sh 2 , as these control signals are referenced to the floating potential Vs. Because the present invention will be directed to generating high-side control signals Vhs 1 or Vhs 2 , these control signal paths are shown with solid lines in prior art FIG. 1 , whereas the low-side control signals are shown with phantom lines.

Regardless of how the high-side and low-side control signals are generated, it will be appreciated that if the various switch pairs Sh 1 -Sl 1 , and Sh 2 -Sl 2 can be switched correctly, current can be made to flow from power source Vhv through the load, in one direction or the other, to ground. For example, if Sh 1 and Sl 1 are closed (as shown in FIG. 1 ) current will flow from Vhv through Sh 1 through the load, through Sl 1 , to ground. If control unit 20 causes Sh 1 and Sl 1 to open, and causes Sl 2 and Sh 2 to close, current can then flow from Vhv through Sh 2 through the load through SI 2 to ground. In this fashion, DC energy from source Vhv can be effectively converted to AC current flowing through the load.

Ideally control unit 20 should provide the required high-side voltage level-shifted control signals Vhs 1 , Vhs 2 to the high-side switches Sh 1 , Sh 2 without dissipating high power. Preferably control signals Vhs 1 , Vhs 2 output from control unit 20 should exhibit high noise immunity, e.g., should maintain correct logic state in the presence of transient components, and should also exhibit short propagation delays through the control circuit. But as noted, it can be a challenging task to efficiently generate the high-side control signals Vhs 1 , Vhs 2 .

Some prior art level-shifting control circuits use continuous control signals and simply accept the resulting high power dissipation needed to achieve short propagation delays. This statement is especially true where the voltage level-shifting control circuit is implemented with discrete components, as opposed to being fabricated on a common integrated circuit (IC). Other prior art approaches use pulse circuits that include latches to conserve power dissipation while still providing short propagation delay. Unfortunately, however, these pulse circuits may be susceptible to noise resulting from transistor switching, from power supply transients, and/or from electrostatic discharge (ESD). U.S. Pat. No. 5,870,266 to Fogg (1999) entitled Bridge Control Circuit and Method discloses a control system that uses both continuous and/or continuous and pulsed control signals in an attempt to reduce power dissipation and maintain good noise immunity, while trying to achieve short propagation delay. However a Fogg type control system can be somewhat complex in its implementation.

Thus, there is a need for a control unit for an electronic switch that provides high-side level-shifted control signals Vhs 1 , Vhs 2 . while achieving short propagation delay, low power dissipation, and high noise immunity. Preferably such a control unit should generate these control signals without the complexity of using both pulsed and continuous control signals.

The present invention provides such a control unit.

SUMMARY OF THE INVENTION

The present invention provides a control unit for an electronic switch that provides level-shifted high-side control signals Vhs 1 , with short propagation delay, with low power dissipation and improved noise immunity, without using pulsed and continuous control signals. Advantageously, propagation delay improvement is obtained from passive current sinking capacitors.

The control circuit is coupled to power sources Vhv, Vdd, and to ground, and receives a logic input signal Vin (or complementary DC logic drive signals S 2 , and {overscore (S 2 )}) and outputs a high-side control signal Vhs 1 (or Vhs 2 ) that can be used to control a high-side electronic switch, e.g., an NMOS transistor Sh 1 (or Sh 2 ) in an electronic switch configuration. In one aspect, the control circuit includes a logic circuit, an current mirror Ibias generator, a level shift circuit that preferably includes a passive current sink mechanism comprising capacitors, and a load circuit from which Vhs 1 (or Vhs 2 ) is obtained. Alternatively, the present invention can function to output a high-side control signal to certain switch configurations, using the Ibias generator and a portion of the level shift circuit.

The complementary DC signals are used by the level shift circuit to determine the desired output logic state for Vhs 1 . The Ibias generator establishes and provides a holding current to maintain correct logic states after the initial transition between states, even in the presence of noise.

Within the level shift circuit, a totem-pole arrangement of solid state devices is coupled between Vdd and ground. The arrangement comprises a PMOS device, an NMOS device, and a second NMOS device that is configured as a current source that mirrors (directly or proportionally) the Ibias current. The source node of a third NMOS device is coupled between the PMOS and NMOS devices in the totem-pole, the gate node is coupled to Vdd, and the drain node serves as an interface to a load circuit coupleable to Vhv Vdd. A capacitor is coupled across the second NMOS device current source. When S 2 transitions from logical 0 to 1 is in a first DC state, current flows from the drain of the interface NMOS device. A high-side switch in an electronic switch configuration can thus be triggered with control signals that vary in amplitude from Vhv Vdd to Vdd.

Other features and advantages of the invention will appear from the following description in which the preferred embodiments have been set forth in detail, in conjunction with the accompanying drawings.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Referring to FIG. 2 , a control circuit 100 according to the present invention is coupled to power sources Vhv, Vdd, and to ground, and receives as logic input signal Vin. Control circuit 100 generates a high-side control signal Vhs 1 (or Vhs 2 ) from node Vo, which signal can be used to control a high-side electronic switch, e.g., an NMOS transistor Sh 1 (or Sh 2 ), in an electronic switch configuration that need not be an H-bridge was shown in FIG. 1 . Vhv may be about 100 VDC, Vdd may be about 10 VDC, although these values are exemplary and other values may be used.

In brief, depending upon the 0 or 1 DC state of the logic input signal Vin, circuit 100 is configured to define a first current path, path A, or a second current path, path B. In the configuration shown, when path A is continuous, the output signal from node Vo is 0 (e.g., Vhs 1 0), and when path B is continuous, the Vo node output signal is 1 (e.g., Vhs 1 1). As used herein, the notation OFF-to-ON will refer to a 0-to-1 voltage transition at node Vo, while ON-to-OFF will refer to a 1-to-0 Vo node voltage transition: As described later herein, passive current sinking capacitors (C 1 , C 2 ) within control circuit 100 advantageously shorten transition times between ON-to-OFF and OFF-to-ON without substantially increasing power dissipation.

More specifically, control circuit 100 includes a logic circuit 110 , an Ibias generator 120 , a level shift circuit 130 that includes a preferably passive current sink 135 , and a load circuit 140 whose node Vo provides the high-side control signal Vhs 1 (or Vhs 2 ). Level shift circuit 135 preferably comprises a pair of capacitors (C 1 , C 2 ) that act as current sinks when as associated transistor (M 3 or M 7 ) is turned-ON, to shorten output signal state transition time.

Logic circuit 110 receives a logic input signal Vin and outputs complementary DC signals S 2 and {overscore (S 2 )} for use by the level shift circuit 130 . The relative logical state of Vin (and thus of S 2 , and {overscore (S 2 )}) determines whether the signal at node Vo is 1 or 0 . Logic circuit 110 may be as straightforward as a single inverter or, as shown, an inverter pair U 1 and U 2 . In the configuration shown, logic output signal S 2 will be the complement of the Vin input control signal, while {overscore (S 2 )} will be a replication of Vin. In the present invention, S 2 and {overscore (S 2 )} are each complementary continuous or DC signals that are logical level 1 and 0 or 0 and 1 . FIGS. 3A and 3B depict exemplary voltage waveforms as a function of time for Vin, S 2 , and {overscore (S 2 )}. Understandably, if complementary logic signals S 2 , and {overscore (S 2 )} are available as input signals, then control circuit 100 may omit logic circuit 100 . It is noted that FIGS. 3A-3F are computer simulations in which magnitude of C 1 and C 2 were increased substantially over their nominal 0.5 pF value and in which timing sequences were slowed, to better show operation of circuit 135 .)

Ibias generator 120 is preferably configured as a current mirror and includes device M 1 , here coupled between Vdd and ground. Ibias generator 120 is coupled to device M 2 within the level shift circuit 130 , and can induce a mirrored current flow in M 2 proportional to Ibias, and in devices coupled in series with M 2 . Within the level shift circuit, devices M 2 , M 3 , M 4 and M 5 may be used to define a portion of a first current path denoted path A, and devices M 6 , M 7 , M 8 and M 9 may be used to define a portion of a second current path denoted path B. When completed, path A and path B will each include a device in load circuit 140 , and a completed current path A or path B will conductor current from Vdd Vhv to ground. An exemplary value for Ibias is about 50 nA, although other magnitudes could instead be used.

In one aspect, the present invention may be defined as including Ibias generator 120 and level shift circuit 130 , assuming that complementary logic signals S 2 and {overscore (S 2 )} are available, and that a suitable load circuit 140 is available. In a second aspect, the present invention may be defined as including Ibias generator 120 and M 2 , M 3 , M 4 , M 5 and C 1 , assuming an appropriate load circuit and switch configuration are available. However the present invention will be described with reference to what is shown in FIG. 2 .

Referring to FIG. 2 , level shift circuit 130 advantageously includes a preferably passive current sink mechanism 135 , capacitors C 1 and C 2 , each about 0.5 pF or so. C 1 sinks additional current from M 3 when M 3 turns-ON to make path A continuous, and C 2 sinks additional current from M 7 when M 7 turns-ON to make path B continuous. The additional current sunk by C 1 or by C 2 when the voltage at node Vo changes state reduces propagation delay, while reducing power consumption of circuit 100 between transitions. It will be appreciated that once C 1 or C 2 is charges, power dissipation through the charged capacitor is essentially nil, which promotes efficiency of the overall circuit. Note that device M 6 discharges capacitor C 2 in preparation for the next OFF-to-ON transition, and that current source M 2 will discharge C 1 in preparation for the next ON-to-OFF transition. FIGS. 3C and 3D depict path A and path B current flow (with amplified current resolution in FIG. 3 C), and FIG. 3E depicts voltage waveforms across capacitors C 1 and C 2 . Again it is noted that the waveforms shown are for a computer simulation in which substantially larger values of C 1 and C 2 were used, and in which time is slowed perhaps ten-fold, the better to depict operation of the invention.

Devices M 5 and M 9 interface between level shift unit 130 and load unit 140 . Unit 140 includes devices M 10 , M 11 , M 12 , M 15 on one-hand, and devices M 13 , M 14 on the other hand. As shown by FIG. 2 , device M 10 conducts current when path A is continuous, whereas device M 14 conducts current when path B is continuous. Node Vo, at the drain of device M 1 3 , provides the desired high-side control voltage Vhs 1 (or Vhs 2 ), which may be coupled to the gate of a high-side switch, e.g., an NMOS device. It will be appreciated that logic circuit 100 shown in FIG. 2 could be configured to augment or replace M 13 with a simple resistor, if desired, to facilitate 0 to 1 node Vo pull-up transitions.

When path A is continuous, M 11 , M 12 , M 15 mirror (directly or proportionally) current flow through path A into node Vo, from whence the Vhs 1 output waveform is obtained. When path B is continuous, M 13 and M 14 mirror the current in path B to the output node Vo. FIG. 3F shows the voltage waveform Vhs 1 (t) at node Vo as a function of time.

Consider operation of control circuit 100 during an ON-to-OFF transition, e.g., a condition in which voltage at the Vo node transitions from 1-to-0. In this state, path A becomes continuous while path B becomes discontinuous. Referring to FIGS. 3A , 3 B, and 3 F, at time 2.5 S, Vin goes to 0 and S 2 goes to 1 . (Again it is noted that time units in FIGS. 3A-3F are exemplary relative to real-time operation of control circuit 100 , and faster or slower times could in fact be used.) When S 2 is 1 , M 3 will turn-on, and M 4 will turn-off. At the same time, {overscore (S 2 )} will turn-on M 8 and will turn-off M 7 , thus breaking continuity for path B within level shift circuit 130 . Although path B continuity is broken, within level shift circuit 130 , at least a portion of path A continuity is established. As M 3 tries to conduct source current, capacitor C 1 advantageously acts as a current sink, permitting a larger initial surge of path A current, than would otherwise be possible. FIGS. 3C and 3D depict the spike of path A current, resulting from the i C 1 (dV/dt) action of capacitor C 1 . As depicted in phantom in FIG. 2 , if desired, an optional active dedicated device M 16 could be coupled across C 1 to promote more active discharge of C 1 before the next ON-to-OFF transition.

Consider now the various devices within load circuit 140 during this ON-to-OFF transition. As shown in FIG. 2 , M 5 is coupled to M 10 , and M 9 is coupled to M 14 . Within level shift circuit 130 , mirrored current flows (directly or proportionally) through M 3 , M 5 and thus through M 10 . The M 10 current flow is mirrored (directly or proportionally) within M 11 and M 12 , and thus through the drain of M 15 . The result is that the drain of M 13 , which is to say node Vo, is pulled low, e.g., Vhs 1 0, the desired result in an ON-to-OFF transition. During initial turn-on of M 3 , in addition to current sunk by C 1 , device M 2 allows a small amount of current to continue to flow through the source of M 3 , even after C 1 is charged and essentially no longer sinks current (e.g., dV/dt 0). This action helps maintain node Vo in a low voltage state, even in the presence of noise. M 6 serves to discharge capacitor C 2 in preparation for the next OFF-to-ON transition. In essence, C 2 can act as a current sink or path to permit current flow from the source of M 7 , as soon as M 7 turns-on.

Consider now the configuration of FIG. 2 during an OFF-to-ON transition, e.g., node Vo goes from 0 to 1, as does Vhs 1 . During an OFF-to-ON transition, path A is interrupted and path B is made continuous. Continuous control signal S 2 from logic unit 110 turns-on M 4 and turns-off M 3 , which interrupts continuity of path A within level shift unit 130 . Since current in M 10 , M 11 , M 12 , M 15 mirror (directly or proportionally) path A current, the result is an interruption of current flow from the drain of M 15 . Simultaneously the complementary continuous control signal {overscore (S 2 )} from logic unit 110 turns-on M 7 and turns-off M 8 , which make path B via level circuit 130 continuous. Current flows (mirrored directly or proportionally) within M 13 , which pulls node Vo high, which brings Vhs 1 to a logical 1 level. Within unit 135 , capacitor C 2 sinks current, which permits M 7 at turn-on to conduct more current than would otherwise be the case. The i C 2 (dV/dt) action results in a shorter OFF-to-ON propagation delay than would be the case if C 2 were omitted from the circuit. Note that current source M 2 discharges C 1 in preparation for the next ON-to-OFF transition. As shown in phantom in FIG. 2 , a device M 17 could added as a current source coupled across C 2 to enable a small amount of current to continue to flow through the source of M 9 , even after C 2 is charged.

FIG. 4 depicts a system 200 in which two high-side control units 100 , such as described with respect to FIG. 2 , are used to turn ON and turn OFF high-side switches Sh 1 , Sh 2 . It is understood that the output signals Vhs 1 , Vhs 2 from each high-side control unit 100 will be complementary in phase. A low-side controller 210 (or indeed two such controllers) provide low-side control signals Vls 1 , Vls 2 to control low-side switches Sl 1 , Sl 2 . Details of the low-side controller(s) 210 are not given is that such circuits are well known in the art. As seen in FIG. 4 , low-side switches S 1 , Sl 2 are referenced to ground, which simplifies generating and providing the low-side control signals. While FIG. 4 depicts a single logic unit 110 shared by both high-side control units 100 , and by the low side controller(s) 210 , separate logic units 110 could of course be provided.

While FIG. 4 depicts use of two units of the present invention to control high-side switches Sh 1 , Sh 2 in an H-bridge configuration, one or more units 100 may be used to control high-side switches in other configurations as well. Those skilled in the art will appreciate that, if desired, the Vhs 1 , Vhs 2 high-side control signal could be level-shifted to a potential more negative than the S 2 , {overscore (S 2 )} logic signals. If desired, the drain of M 6 could be coupled to a dedicated current source, similar to current source M 2 . Indeed, a dedicated device (e.g., M 16 ) could be coupled across C 1 while a dedicated current source is coupled to the drain of M 6 .

Modifications and variations may be made to the disclosed embodiments without departing from the subject and spirit of the invention as defined by the following claims. For example, the high-side switch Sh 1 (or Sh 2 ) controlled by the present invention may be a NMOS device or a PMOS device, depending upon configuration of the present invention. It will be appreciated that NMOS and PMOS devices may be substituted in a suitable configuration of the present invention, and that a variety of electronic switch configurations involving Sh 1 , Sl 1 , or Sh 1 , Sl 1 , Sh 2 , Sl 2 , etc. may be accommodated.