Bidirectional oscillator-based radio with integrated antenna

A packaged integrated circuit device includes a transceiver with one or more oscillators therein. The oscillators can be configured as harmonic or fundamental-frequency oscillators for signal transmitting, or they can be configured as regenerative receivers for signal receiving. An antenna is provided in the package and, preferably, on the same chip IC chip as the transceiver. The antenna includes one or more feeds, which are coupled to the one or more oscillators. A dual-function varactor/envelope detector is provided, which is electrically coupled to the one or more oscillators. A control circuit is provided, which is configured to drive nodes of the transceiver and the dual-function varactor/envelope detector with a first plurality of reference voltages during operation of the transceiver as a radio transmitter (with varactor) and a second plurality of reference voltages during operation of the transceiver as a radio receiver (with envelope detector).

FIELD OF THE INVENTION

The present invention relates to integrated circuit devices and, more particularly, to integrated circuit devices having high frequency radio transceivers therein and methods of operating same.

BACKGROUND OF THE INVENTION

There is an ongoing need for highly integrated radio systems in low-cost silicon platforms to address field-deployable and massively-producible applications in military systems and commercial markets, including wireless sensor networks, medical implantable devices, and swarm multi-robot systems. However, these applications typically place stringent requirements on the radio solutions which need to offer an ultra-compact form-factor, an ultra-low power consumption, a sufficient communication distance, and a useful data rate. Most existing integrated radio solutions, however, cannot satisfy such demanding SWaP (Size-Weight-and-Power) requirements. Accordingly, it remains a challenge to push the power consumption limit in conventional radio architectures even with various low-power design techniques. Moreover, the physical size of GHz or mm-Wave radios is fundamentally dominated by the size of the antenna at millimeter or even centimeter scales.

The continuous device scaling in silicon IC technologies (e.g., CMOS, SiGe HBT) has opened the door to radios operating at mm-Wave and THz frequencies. Such a high operating frequency allows a drastic reduction of the antenna sizes as well as the whole radio form-factor to the sub-millimeter scales. However, most nm-Wave and THz radios consume substantial DC power, often from hundreds of milli-watts to watts, incompatible with field-deployable applications.

SUMMARY OF THE INVENTION

Integrated circuit devices according to embodiments of the invention include a nano-scaled mm-Wave/terahertz (THz) radio configured as a packaged mm-Wave/THz transceiver on an integrated circuit chip with a fully integrated antenna. This antenna may be configured as an on-chip and/or co-packaged antenna (e.g., multi-slot/multi-feed antenna). According to some of these embodiments of the invention, the mm-Wave/THz radio transceiver is configured as a bidirectional circuit-sharing radio that can be selectively configured (e.g., digitally controlled) either as a harmonic-oscillator based mm-Wave/THz transmitter or as a super-harmonic regenerative mm-Wave/THz receiver (or just fundamental-frequency THz receiver), which have an ultra-compact form factor that can operate at ultra-low power. The on-chip/co-packaged multi-feed antenna structure also achieves spatial power combining/splitting and radiation for the mm-Wave/THz transmitting/receiving signals with minimal signal loss. This antenna may be patterned on-chip to achieve antenna-level power combining and obviate any need for a lossy on-chip power combining network or separate packaging. In some embodiments of the invention, the radio transceiver may be configured to include first and second matched oscillators, which are electrically coupled together, and the antenna may be configured to support oscillator synchronization with first and second feeds that are electrically coupled to the first and second oscillators, respectively.

According to additional aspects of these embodiments of the invention, corresponding first output terminals of the first and second oscillators are electrically shorted together and corresponding second output terminals of the first and second oscillators are electrically shorted together. In addition, a common-mode node of the first oscillator is electrically coupled to the first feed and a common-mode node of the second oscillator is electrically coupled to the second feed. The first and second oscillators, which may have shared differential output terminals, may operate collectively as a push-push harmonic oscillator that generates a second harmonic signal from a fundamental oscillation signal (f0). During this harmonic operation, the transceiver may operate as a radio frequency transmitter in response to an on-off-keying (OOK) modulation or amplitude-shift-keying (ASK) modulation signal. On-off-keying (OOK) or amplitude-shift-keying (ASK) is preferably chosen to, among other things, preclude on-chip coherent local oscillator (LO) signal generation.

According to still further embodiments of the invention, the transceiver can include a pair of MOS transistors having commonly-connected drain terminals and commonly-connected source terminals, which can perform a dual function as a varactor or as an envelope detector when the transceiver is disposed in a transmitting mode or a receiving mode, respectively. A gate terminal of a first of the pair of MOS transistors may be electrically coupled to a first of the shared differential output terminals and a gate terminal of a second of the pair of MOS transistors may be electrically coupled to a second of the shared differential output terminals. In addition, a first pair of load inductors is provided, which are electrically connected to the differential output terminals of the first oscillator, and a second pair of load inductors is provided, which are electrically connected to the differential output terminals of the second oscillator. These first and second pairs of load inductors may be configured as microstrip transmission lines on the integrated circuit chip. A tail current source may also be provided to support digital control of the transceiver. This tail current source may be configured as a pull-down transistor having a gate terminal responsive to the OOK modulation control signal when the transceiver is operated as an RF transmitter, or responsive to an oversampled quench signal when the transceiver is operated as an RF receiver.

According to additional embodiments of the invention, the transceiver can be controlled to operate as a fully-integrated super-harmonic (e.g., 2ndharmonic) regenerative receiver that detects incoming mm-Wave/THz signals, which are injection-locked to the receiver operating at a lower frequency (e.g., half of the incoming THz frequency). This transceiver architecture can substantially improve receiver sensitivity and reduce receiver power consumption. When operating as a transmitter, the fully integrated push-push harmonic oscillator can generate a 2ndharmonic THz output signal from a fundamental oscillation signal (f0). Two on-chip fundamental oscillators are directly coupled to each other to increase the total output power. On-off-keying (OOK) modulation can be efficiently handled by turning on/off these paired harmonic oscillators.

According to still further embodiments of the invention, an integrated circuit device includes an integrated circuit package and a transceiver in the package. The transceiver includes first and second oscillators therein, which have respective pairs of cross-coupled differential input terminals and differential output terminals that are joined in a push-push oscillator configuration. A multi-feed antenna is also provided, which has first and second feeds electrically coupled to first and second common-mode nodes of the first and second oscillators, respectively, in the package. A dual-function varactor and envelope detector is provided, which is electrically coupled to the first and second oscillators. A control circuit is provided, which is configured to drive nodes of the transceiver and the dual-function varactor and envelope detector with a first plurality of reference voltages during operation of the transceiver as a radio transmitter and a second plurality of reference voltages during operation of the transceiver as a radio receiver.

According to some of these embodiments, the transceiver includes a symmetric injection circuit. This injection circuit includes first and second serially-connected injection transistors having respective first and second gate terminals electrically coupled (directly or by transmission lines) to the first and second feeds. The first and second injection transistors may be NMOS transistors having commonly-connected emitter terminals, which are responsive to reference voltages generated by the control circuit. In addition, a drain terminal of the first injection transistor may be electrically coupled to a first of the differential output terminals of the first oscillator and a drain terminal of the second injection transistor may be electrically coupled to a first of the differential output terminals of the second oscillator. Alternatively, the transceiver may also include an asymmetric injection circuit.

The transceiver may also include a first pair of load inductors, which are electrically connected to the differential output terminals of the first oscillator, and a second pair of load inductors, which are electrically connected to the differential output terminals of the second oscillator. These load inductors may be configured as microstrip transmission lines (TLs). A tail current source may also be provided, which receives an on-off-keying (OOK) modulation control signal during a transmitting mode and a quench signal (e.g., 4× oversampled) during a receiving mode. This tail current source may be configured as a pull-down transistor having a gate terminal responsive to the OOK modulation control and quench signals.

The nano-scaled, ultra-low-power and low-cost radio transceivers described herein can be employed in a wide variety of applications, including defense and commercial applications. In the defense space, the proposed nano-scaled transceivers can be utilized to realize ubiquitous large-scaled wireless sensor networks for surveillance. If the transceivers are deployed on insects or swarm mini-robots, a distributed, yet real-time dynamic wireless sensor network, can be achieved. When operated in parallel (e.g., for large-volume commercial markets), the nano-scaled transceivers described herein can be readily adopted as the low-power and compact sensing/communication nodes for Internet-of-Things (IoT) applications, for example.

DETAILED DESCRIPTION OF EMBODIMENTS

The terminology used herein is for the purpose of describing particular embodiments only and is not intended to be limiting of the present invention. As used herein, the singular forms “a,” “an” and “the” are intended to include the plural forms as well, unless the context clearly indicates otherwise. It will be further understood that the terms “comprising”, “including”, “having” and variants thereof, when used in this specification, specify the presence of stated features, steps, operations, elements, and/or components, but do not preclude the presence or addition of one or more other features, steps, operations, elements, components, and/or groups thereof. In contrast, the term “consisting of” when used in this specification, specifies the stated features, steps, operations, elements, and/or components, and precludes additional features, steps, operations, elements and/or components.

Referring now toFIGS. 1A-1B, a bidirectional integrated circuit radio10with an on-chip multi-feed slot antenna12and a dual-function varactor/envelope detector14is illustrated. This radio10uses an oscillator core, which serves as the fundamental building block to the system and can be fully configured in either a transmitting mode (Tx) or a receiving mode (Rx) to achieve bidirectional TRx within a compact integrated circuit (IC) chip area. In particular, and as illustrated herein, the radio10can be selectively configured in a super-harmonic, sub-harmonic, or fundamental-frequency regenerative receiver architecture in the receiving mode (instead of classical heterodyne or direct-conversion architecture) to thereby improve receiver sensitivity below, around, or above transistor fmax. Also, as illustrated herein, the radio10can be selectively configured in a harmonic or fundamental-frequency transmitter, and the fundamental-frequency or harmonic oscillation signals drive the antenna for transmitter radiation below, around, or above transistor fmax. In addition, the on-chip slot antenna12, preferably with semiconductor substrate thinning, is used to achieve compact chip layout and efficient backside radiation. This multi-feed slot antenna architecture is used to achieve antenna-level power combining and avoid the need for a relatively lossy on-chip power combining network. Nonetheless, other types of antennas and possibly multiple antennas may be used.

To meet system size requirements of about 10−11m3, a CMOS THz radio10can be implemented using only 10% of the total system volume (i.e., 10−12m3) and an energy “harvester” is implemented within the other 90% of the volume. Assuming a thinned CMOS chip thickness of about 50 μm, the chip area can fit within about 2×104μm2(e.g., 250 μm×80 μm). As will be understood by those skilled in the art, a direct tradeoff exists between the size of the antenna12and the system operating frequency within such a small chip area. In some of the embodiments described herein, a 340 GHz carrier frequency is targeted. This THz level frequency can be generated as the 2nd harmonic signal from an on-chip fundamental oscillator operating at f0=170 GHz.

For deep-scaled CMOS technologies, such as 32 nm semiconductor-on-insulator (SOI), the fmaxof transistors is typically about 350 GHz. The fmaxof transistors sets the upper frequency limit for the power amplification and fundamental oscillation. However, after considering the loss of on-chip passive components, the maximum allowed oscillation frequency typically decreases substantially compared with transistor fmax. Therefore, in order to perform power generation at THz range (300 GHz-3 THz), passive frequency multipliers or active frequency multipliers should typically be used. Passive frequency multipliers can achieve THz signal generation with low DC power consumption, but often exhibit substantial conversion loss and typically rely on high power mm-wave pumping signals. Generating such high power pumping signals at mm-wave frequency is typically very energy inefficient. On the other hand, the harmonic signal can typically be extracted from a fundamental oscillator as an active frequency multiplier. For example, the push-push oscillator configuration illustrated herein can enable a 2ndharmonic extraction at its common-mode node, while the fundamental tone and all the odd-order harmonic signals are cancelled

As illustrated byFIGS. 1A-1B, a desired 340 GHz carrier frequency (2ndharmonic signal) is generated from a fundamental oscillator at 170 GHz, leading to an on-chip half-wavelength slot antenna12of 220 μm, as shown byFIGS. 1A-1B. Alternatively, as illustrated byFIG. 4, a 320 GHz carrier frequency (2ndharmonic signal) can be generated from a fundamental oscillator at 160 GHz. At higher operating frequencies, the size of the antenna12can be reduced to save chip area, but this will typically require either: (i) a fundamental oscillator at higher frequency, or (ii) generating the harmonic signal directly from the oscillator. Moreover, as the oscillation frequency approaches the transistor fmax, the output swing of the oscillator normally decreases, which makes the devices less nonlinear, generates less harmonic current, and decreases the transmitter output power.

As shown by the timing diagram inFIG. 4, on-off-keying (OOK) modulation is utilized, which eliminates on-chip coherent local oscillator (LO) signal generation to thereby save power. Other types of modulation may also be used, such as amplitude-shift-keying (ASK), frequency-shift-keying (FSK), phase-shift keying (PSK) and quadrature-amplitude modulation (QAM), as well as amplitude modulation (AM), frequency modulation (FM) and phase-modulation (PM).

The radio receiver is implemented as a power detector (with super-harmonic regenerative operation) to further simplify the receiver architecture. Based on this modulation format, the link budget can be calculated using the Friis transmission equation, as:

PR=PT+GT+GR+20×log10⁡(λ4⁢π⁢⁢R),
where PRis the received power, PTis the transmitted power, GTis the transmitter antenna gain, GRis the receiver antenna gain. For a communication range of 1 m, assuming a 200 kb/s OOK modulation data rate, a −10 dBm transmitter output power and a 3.9 dB antenna gain, the received signal power level is around −85 dBm at its input. In addition, the signal-to-noise ratio (SNR) of the receiver can be estimated with the given received power, as:
SNR=PA−174 dBm/Hz−NF+20×log(BW)=35 dB−NF,
where NF is the receiver noise figure, and BW denotes the modulation bandwidth. At a carrier frequency of 340 GHz, it is challenging to design a CMOS amplifier. Therefore, one can either choose a mixer-first topology, which normally has a high NF because of the conversion loss, or utilize the regenerative architecture described herein. Thus, in the embodiments described herein, a Tx oscillator is used as a super-harmonic regenerative receiver in the Rx mode to achieve OOK demodulation with low power consumption. Circuit block sharing also helps to substantially decrease the chip area.

Referring again toFIGS. 1A-1B, the packaged bidirectional radio10(i.e., transceiver) is illustrated as including an on-chip (or, at least, co-packaged) multi-feed slot antenna12(λ/2=220 μm at 340 GHz) and a dual-function varactor/envelope detector14. The radio10includes first and second matched oscillators having respective pairs of cross-coupled differential input terminals and differential output terminals that are joined together in a push-push oscillator configuration. As shown, the first oscillator includes NMOS transistors M3, M4(W=14.4 μm, L=40 nm) having commonly-connected emitter terminals and the second oscillator includes NMOS transistors M5, M6(W=14.4 μm, L=40 nm) having commonly-connected emitter terminals, connected as illustrated. A gate terminal of NMOS transistor M3is electrically connected to a differential output terminal of the first oscillator, which corresponds to a drain terminal of NMOS transistor M4. Similarly, a gate terminal of NMOS transistor M4is electrically connected to a differential output terminal of the first oscillator, which corresponds to a drain terminal of NMOS transistor M3. A gate terminal of NMOS transistor M5is electrically connected to a differential output terminal of the second oscillator, which corresponds to a drain terminal of NMOS transistor M6. Similarly, a gate terminal of NMOS transistor M6is electrically connected to a differential output terminal of the second oscillator, which corresponds to a drain terminal of NMOS transistor M5.

The first oscillator further includes a first pair of inductors, which are illustrated as transmission lines (TLs) having an inductance of 25 pH (at f0). The first pair of inductors are commonly-connected at a common-mode node of the first oscillator, which is electrically coupled to a first feed (1) of the multi-feed slot antenna12. The second oscillator further includes a second pair of inductors, which are illustrated as transmission lines (TLs) having an inductance of 25 pH (at f0). The second pair of inductors are commonly-connected at a common-mode node of the second oscillator, which is electrically coupled to a second feed (2) of the multi-feed slot antenna12. Moreover, according to additional embodiments of the invention, the coupling between the oscillators and the coupling between the antenna(s) and the oscillators may be electrical (as shown), magnetic or electromagnetic.

The dual-function varactor/envelope detector14includes NMOS transistors M8, M9(W=3.2 μm, L=40 nm) having commonly-connected emitter terminals and commonly-connected drain terminals. As shown, the commonly-connected emitter terminals are responsive to a variable bias voltage Vb2, which is provided by a control circuit16, and the commonly-connected drain terminals are electrically coupled by a pull-up resistor (e.g., 40K ohms) to a variable bias voltage Vb3, which equals Vb2during the transmitting mode but is greater than Vb2(e.g., Vdd) during the receiving mode. In addition, a gate terminal of NMOS transistor M8is electrically connected to a pair of differential output terminals associated with the first and second oscillators and a gate terminal of NMOS transistor M9is electrically connected to an opposite pair of differential output terminals associated with the first and second oscillators, as illustrated.

An NMOS pull-down transistor M7(W=60 μm, L=40 nm), having a drain terminal connected to emitter terminals of the oscillator transistors M3-M6, operates as a tail current source. This current source is responsive to an on-off-keying (OOK) modulation signal, which provides transmission data (Tx data) during a transmitting mode of operation, or a quench signal when the radio10is operating as a receiver during a receiving mode of operation.

Moreover, during the receiving mode of operation, a symmetric injection circuit is selectively enabled, but during the transmitting mode of operation the injection circuit is selectively disabled. This injection circuit includes NMOS transistors M1, M2(W=1.6 μm, L=40 nm) having commonly connected emitter terminals, which are responsive to a variable bias voltage Vb1generated by the control circuit16. The magnitude of Vb1determines whether the injection circuit is active (Vb1<Vdd) or inactive (e.g., Vb1=Vdd). As shown, the drain terminals of NMOS transistors M1and M2are shorted together and to corresponding differential output terminals of the first and second oscillators (and gate terminal of M9). The gate terminals of NMOS transistors M1and M2are electrically coupled to respective ones of the first and second feeds (1) and (2) of the antenna12.

As further illustrated byFIGS. 1A-1B, the power supply of the oscillator (i.e., Vdd) can be provided through the power plane of the slot antenna12. To achieve the desired operation, the two-feed slot antenna12requires two driving signals with the same phase and amplitude at its two input feeds (1) and (2). In the transmitting mode, M1and M2are turned off by setting Vb1to Vdd. Transistors M3, M4form one cross-coupled oscillator core and transistors M5, M6form another cross-coupled oscillator core. The four load inductors, which may be implemented as microstrip transmission lines, achieve 25 pH at 170 GHz. To ensure that the driving signals are in-phase with the same amplitude at the two antenna input feeds (1) and (2), the two oscillators are coupled together by shorting their positive outputs and negative outputs, respectively, as illustrated. Tail current source M7is turned on/off to provide the OOK modulation. The desired 2ndharmonic outputs are taken from the common-mode nodes of the coupled oscillators and directly radiate out through the feeds (1) and (2) of the on-chip antenna12. In addition, NMOS transistors M8and M9are configured as a varactor to provide frequency tuning. Different transceiver circuitry can be used when different numbers or types of antenna are used.

Referring now toFIG. 2A, a simulated output spectrum for the fundamental, 2nd, 3rd, 4thand 5thharmonic signals is shown. In this simulation, schematic-level transistor models in the IBM 32 nm SOI design kit and the EM simulated S-parameters of the antenna are used. The peak CW output power is −13.5 dBm at 2f0(344 GHz) and the total output power from two antenna feeds is −10.5 dBm, with a total power consumption of 15 mW at a level where Vdd=1.1 V. The harmonic suppressions of the fundamental leakage and the 4thharmonic signal are −25 dBc and −22 dBc, respectively. If the distance between Tx and Rx decreases, the transmitter output power can also be reduced to save total power consumption, as shown byFIG. 2B. This can be achieved by reducing the tail current value of the oscillators.

As previously highlighted, the bias voltage Vb2can be used for frequency tuning, which means the output power can be adjusted as shown byFIG. 2C. In particular, a total 6.5 GHz CW frequency tuning range (1.5%) is achievable from 339 GHz to 345.5 GHz. This output power variation is within 2 dB across the frequency tuning range provided by the varactor14and Vb2(where Vb3=Vb2).

As will be understood by those skilled in the art, because no amplifier can be designed above fmax, classical heterodyne receivers typically suffer from high conversion loss and noise figure at frequency near or above fmax. An alternative solution, as illustrated herein, is the regenerative receiver, which can readily demodulate OOK signals or other types of modulations, including ASK, FSK, PSK, QAM, AM, FM and PM. Thus, as shown byFIGS. 1A-1B, the radio10can be selectively configured as a super-harmonic (2ndharmonic) regenerative receiver by adjusting the bias voltages Vb1, Vb2and Vb3and periodically quenching NMOS transistor M7by driving the gate terminal of M7with a logic 1 voltage (above the threshold voltage of M7). When the quench signal is received, the coupled oscillators will gradually establish oscillation even without any incoming signal at the antenna12. When the antenna12receives an input signal at 2f0, the NMOS transistors M1and M2within the symmetric injection circuit will provide super-harmonic injection currents into the oscillator tanks. Asymmetric injection may also be utilized as described herein with respect toFIG. 4. The super-harmonic regenerative principle predicts that the oscillation start-up time is inversely proportional to the received signal power in dB scale. Therefore, with higher input signal power, the coupled oscillators will have a shorter start-up time, as shown in the simulation ofFIG. 3, where varying start-up times (x-axis) are shown as a function of antenna input power (Pin). Based on a calculated link budget, the input power strength at the receiver antenna12is about −85 dBm with 1 m communication range, while the receiver noise floor lies around −120 dBm. Moreover, the oscillation start-up time difference is about 6 ns when the input power is −80 dBm versus −120 dBm, based on which the frequency of the quench signal can be determined. Here, NMOS transistors M8, M9and the pull-up resistor load (40K ohms) form an envelope detector14for received signals, which means that signal demodulation can be performed by comparing the output voltage level of the envelope detector (at the drain terminals of M8, M9) with a reference value using a comparator (not shown).

Referring now toFIG. 4, a packaged bidirectional radio frequency transceiver10′,10″ (with modified dual-feed slot antenna12′) is illustrated under conditions of transmission (transmitting mode Tx) and reception (receiving mode Rx), with corresponding timing diagrams. As illustrated, a bias voltage applied to the emitter terminal of NMOS transistor M1can be controlled to selectively enable/disable M1, which operates as an asymmetric injection (e.g., current/voltage injection) circuit during the receiving mode Rx. The matched oscillators that make up the oscillator core (OC) are defined by NMOS transistors M3-M6and transmission lines TL3-TL6, connected as illustrated. Additional transmission lines TL1, TL2are provided to the pair of antenna feeds. The dual-function varactor/envelope detector14includes NMOS transistors M7-M8. During the transmitting mode Tx, equal bias voltages of Vctrl are applied to the varactor to support frequency tuning. However, during the receiving mode Rx, current is allowed to flow through M7-M8and an envelope detector output signal (ED_OUT) is taken from the commonly-connected drain terminals, as shown.

During the transmitting mode, the two coupled oscillators operate at f0=160 GHz to thereby generate a 2ndharmonic signal (2f0=320 GHz) as the matched THz output signals to the antenna feeds. As will be understood by those skilled in the art, the cross-coupled transistors M3-M6(within the oscillator core OC) provide differential negative transconductance (gm) for oscillation, and the drain terminal transmission lines TL3-TL6and the built-in device parasitic capacitors form the resonator tank at f0=160 GHz. The transmission lines further perform impedance matching to the on-chip slot antenna12′ at 2f0to maximize the output power (Pout), and the two in-phase 2f0currents are power-combined on the antenna12′. Transistors M7and M8are used as varactors for frequency tuning. The Tx OOK data signal directly drives the tail current source M2to thereby provide bits-to-THz transmitting. The Rx injection transistor M1is turned off in the Tx mode by setting its emitter terminal to Vdd, as shown.

The on-chip antenna12′ consists of a VDDplane and a GND plane, as shown. The VDDplane is DC-connected to the drain nodes of oscillator core transistors M3-M6for biasing purposes. Sufficient bypass capacitors are needed to decouple the VDDplane from the GND plane. However, on-chip capacitors typically exhibit poor quality factor (Q) in the THz range, which can lead to severe signal loss to the THz radiation signal and radiation efficiency degradation, especially when the antenna input impedance is low (e.g., 11Ω). To minimize the loss from the bypass capacitors, the slot antenna12′ is designed with a total length of around λ. Based on this design, the slot antenna12′ presents two current nulls and the VDDplane and the GND plane are separated and bypassed at these two current nulls, which ensures minimal signal loss through the bypass capacitors. Different supply feeding and/or bypassing circuits can be employed when different numbers or types of antenna are used.

Alternatively, in the Rx mode, the radio10″ is configured as a super-harmonic super-regenerative Rx to detect the incoming OOK-modulated THz signal at 2f0. Once a 2f0input signal is received by the on-chip antenna12′, M1injects a 2f0current into the resonator tank and creates a small asymmetry to perturb the fundamental oscillation start-up at f0. Thus, as shown by the timing diagram on the right side ofFIG. 4, the received OOK signal (“1” or “0”) leads to different Rx oscillation start-up times (“0”=long, “1”=short). As previously explained with respect toFIGS. 1A-1B, transistors M7and M8are operated as an envelope detector (ED), which generates signal ED_OUT. By periodically quenching the tail current source M2(with 4× oversampling), OOK demodulation can be realized by tracking the digitized oscillation start-up time encoded within ED_OUT. This super-harmonic super-regenerative Rx substantially improves the Rx sensitivity over non-coherent THz power-detector Rx, yet exhibits significant sensitivity.