Apparatus and method for driving and controlling brushless motor

A brushless dc motor having 3-phase stator windings and a permanent-magnet rotor is driven on a dc voltage through a 3-phase electronic switching circuit. Back emf voltages across the stator windings are individually phase-delayed and the phase-delayed voltages are individually compared in comparators with a common sawtooth-wave comparator reference voltage having a frequency proportional to a current rotor speed and an amplitude whose center voltage is one-half of the motor driving dc voltage. A drive control unit, a microcomputer, transmits to the switching circuit switching control signals in sequential driving steps according to 3-phase output signals of the comparators so that the switching circuit performs commutation of the motor drive voltage in a first motor driving mode. The time constant of the phase-delay circuits is increased when the rotational speed of the rotor is below a predetermined speed or the motor driving current is above a predetermined level. When the time constant is increased, the motor driving mode is switched from the first motor driving mode to a second motor drive mode for a predetermined time period at fixed driving steps regardless of the 3-phase comparator output signals. The motor driving mode reverts to the first motor driving mode when the predetermined time period for the second motor driving mode has lapsed.

BACKGROUND OF THE INVENTION 
1. Field of the Invention 
This invention relates to an apparatus and method for driving and 
controlling a small brushless dc motor particularly having 3-phase stator 
windings and a permanent-magnet rotor. The apparatus has a solid-state 
switching circuit that electronically commutates dc power to sequentially 
energize the stator windings. The timings for the commutation switchings 
are primarily determined by the angular positions of the rotor that are 
electronically detected from the back emit voltages across the stator 
windings induced by the revolving rotor without utilizing any physical 
angular position sensor. 
2. Description of the Prior Art 
One of such brushless motor units is described in U.S. Pat. No. 5,640,073, 
that is commonly assigned to the assignee of the present application. 
The brushless motor has Y-connected 3-phase stator windings and a 
permanent-magnet rotor. A dc voltage is provided to a solid-state 
electronic switching circuit to be converted to 3-phase voltages that are 
individuality provided to the 3-phase stator windings. The switching 
circuit consists of six solid-state switching elements (e.g. IGBT) having 
respective control terminals that are individually connected to six 
switching control outputs of a control unit. The switching elements are 
turned on and off by switching control signals transmitted from the 
control unit at specific rotor angles. The switching sequence is arranged 
to cause the stator windings to produce a rotating magnetic flux that 
interacts with the flux produced by permanent magnets on the rotor so as 
to rotate the rotor in synchronism with the rotating magnetic field. 
Back emf voltages across the 3-phase stator windings are individually 
provided to 3-phase phase-delay filter circuits so that the phase angle of 
each voltage is delayed by an electrical angle smaller than 90.degree. 
thereby. The phase-delayed output voltages of the phase-delay filter 
circuits are individually provided to the positive input terminals of 
voltage comparators. To the negative input terminals of the voltage 
comparators is commonly provided a sawtooth-wave comparator reference 
voltage, which is an output of a comparator reference voltage computation 
circuit, having a frequency proportional to a current rotor speed and an 
amplitude whose center voltage is one half of the dc power supply voltage. 
The control unit receives 3-phase output voltages of the comparators and, 
in reference thereto, transmits switching control signals to the switching 
elements so as to control the switching circuit. 
The control unit monitors the rotor speed from the frequency of the 3-phase 
comparator output voltages. And when the monitored speed becomes smaller 
than a predetermined value, the time constant of the phase-delay filter 
circuits is increased to prevent a possible rotor trip-off due to a low 
rotational speed. 
(Problems Pertaining to the Conventional Motor to be Solved by the Present 
Invention) 
In the brushless motor as described above, the signal representing the 
angular position of the rotor is obtained by comparing the phase-delayed 
output voltages of the phase-delayed filter circuits with the comparator 
reference voltage, which is an output of a comparator reference voltage 
computation circuit. However, if the time constant of the filter circuits 
is changed responsive to the variation of the rotational speed of the 
rotor, the change of the time constant causes changes of the wave forms 
and the slopes of the phase-delayed output voltages of the phase-delayed 
filter circuits. Then, the timings when the levels of the phase-delayed 
output voltages and the level of the comparator reference voltage become 
even will transiently shift. Such shiftings of the timings make it 
difficult to detect a precise current angular position of the rotor, 
thereby resulting in undesirable time shiftings of the driving steps (i.e. 
switching steps) of the electronic switching circuit. The driving steps, 
then, will deviate from properly regulatable electric angle ranges, and 
the switching circuit will be subjected to excessive currents. In order to 
cope with such excessive currents, the switching circuit will have to be 
of an undesirably large capacity, whereby the size and cost of the 
switching circuit will have to be increased. 
SUMMARY OF THE INVENTION 
It is therefore an object of the present invention to provide an apparatus 
and method for driving and controlling a brushless dc motor, which 
includes a solid-state switching circuit and phase-delay filter circuits, 
in which the switching circuit and the motor are not subjected to 
excessive driving currents when the time constant of the phase-delay 
circuits is changed. 
In order to achieve the above object, the brushless dc motor to be driven 
includes 3-phase stator windings having respective winding terminals and a 
permanent-magnet rotor. The apparatus includes a dc power supply, an 
electronic switching circuit, 3-phase phase-delay filter circuits, 3-phase 
voltage comparators, a comparator reference voltage computation circuit, a 
drive control unit, which is a microcomputer, and a current meter. The dc 
power supply provides a motor drive voltage and a midpoint voltage that is 
one half of the motor drive voltage. The electronic switching circuit is 
connected to the dc power supply for switching the motor drive voltage to 
produce 3-phase dc voltages that are applied to the 3-phase stator 
windings individually. Three-phase back emf voltages induced across the 
3-phase stator windings, while the rotor is in rotation, are individually 
provided to the 3-phase phase-delay circuits so as to delay phases of the 
3-phase back emf voltages by an electric angle of less than 90.degree. so 
that 3-phase phase-delayed voltages are obtained therefrom. 
Each of the 3-phase voltage comparators has a first input terminal, a 
second input terminal and an output terminal, and the 3-phase 
phase-delayed voltages are individually provided to the first input 
terminals. The comparator reference voltage computation circuit outputs a 
sawtooth-wave comparator reference voltage having a frequency proportional 
to a current angular speed of the rotor and an amplitude whose center 
voltage is equal to the above mentioned midpoint voltage. The comparator 
reference voltage is commonly provided to all of the second input 
terminals of the voltage comparators so as to obtain 3-phase comparator 
output voltages from the comparators. The 3-phase comparator output 
voltages are individually provided to the drive control unit. In the drive 
control unit are obtained switching control signals in sequential driving 
steps having driving step time periods according to the 3-phase comparator 
output voltages. The switching control signals in sequential driving steps 
are provided to the electronic switching circuit so that the electronic 
switching circuit performs commutation of the motor drive voltage in a 
first motor driving mode. On the other hand, in the drive control unit, a 
rotational speed of the rotor is obtained from the 3-phase comparator 
output voltages, and an amount of motor drive current supplied from the dc 
power supply and measured by the current meter is monitored. The time 
constant of all of the phase-delay filter circuits is increased when the 
rotor speed is below a predetermined speed and/or the motor drive current 
is above a predetermined amount. 
Datum of each of the driving step time periods is stored consecutively in a 
refreshing manner in a memory unit in the control drive unit. When the 
time constant of the phase-delay filter circuits is increased, the last 
step time period stored in the memory unit is timed by a driving step 
period timer in the control drive unit. The last step time period is 
multiplied by a predetermined number of steps to obtain a time period for 
a second motor driving mode. 
When the time constant is increased, motor driving mode is switched from 
the first motor driving mode, which is dependent of the 3-phase comparator 
output voltages, to a second motor drive mode, which is independent of the 
3-phase comparator output voltages, for the time period calculated for the 
second motor driving mode. The drive control unit provides switching 
control signals in sequential driving steps for the second motor driving 
mode to the electronic switching circuit so that the electronic switching 
circuit performs commutation of the motor drive voltage in the second 
motor driving mode. In the second motor driving mode, each of the 
sequential driving steps has a time period that is equal to the last step 
time period stored in the memory unit. The motor driving mode reverts from 
the second motor driving mode to the first motor driving mode when the 
time period for the second motor driving mode has lapsed. 
In an alternative embodiment according to the present invention, when the 
time constant of the phase-delay filter circuits is increased, motor 
driving mode is switched from the first motor driving mode, as described 
above, to a modified second motor drive mode, which is also independent of 
the 3-phase comparator output voltages, for a time period that equals to a 
predetermined driving step period multiplied by a predetermined number of 
driving step. Then, the driving mode reverts to the first motor driving 
mode after the time period for the modified second mode has lapsed. The 
drive control unit provides switching control signals in sequential 
driving steps, each time period of which is fixed for the modified second 
motor driving mode, to the electronic switching circuit so that the 
electronic switching circuit performs commutation of the motor drive 
voltage in the modified second motor driving mode.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
The present invention will now be described in detail with reference to the 
drawings. 
FIG. 1 is a circuit diagram of a brushless dc motor and an apparatus for 
driving and controlling the motor according to the basic embodiment of the 
present invention. A brushless dc motor 11 primarily consists of 
Y-connected 3-phase stator windings 12-1, 12-2, 12-3 and a 
permanent-magnet rotor 13. An electronic switching circuit (i.e. an 
electronic commutation circuit) 15 has three pairs of bridge-connected 
solid-state switching elements (i.e. transistors) 17-1,4 (U-phase), 17-2,5 
(V-phase) and 17-3,6 (W-phase), each switching element having a control 
terminal that is individually connected to each of six control outputs of 
a drive control unit (a microcomputer) 50. The three pairs of the 
switching elements 17-1.about.6 have respective output terminals 16-1, 
16-2 and 16-3 that are connected to the terminals of the stator windings 
12-1, 12-2, 12-3, respectively. A dc voltage VM Of a dc power supply 20, 
with a grounded negative terminal, is applied to each of the three pairs 
of the switching elements as shown. 
The terminal voltages Vu, Vv and Vw, which are back emf voltages induced by 
the rotor 13 in rotation, of the 3-phase stator windings 12-1, 12-2 and 
12-3, respectively, are provided to phase-delay filter circuits 14-1, 14-2 
and 14-3, respectively, and phase angle in each circuit is delayed thereby 
by an electrical angle of approximately 60.degree. in this particular 
embodiment. But it is also permissible if the delay angle is less than 
90.degree. but more than 30.degree. The phase-delay filter circuits 14-1, 
14-2 and 14-3 consist of main filter circuits 21-1, 21-2 and 21-3, 
respectively, and time constant increase circuits 91-1, 91-2 and 91-3, 
respectively. The main filter circuits 21-1, 21-2 and 21-3 consist of 
resistors R1, R2 and R3, respectively, connected in series to the 
terminals of the stator windings 12-1, 12-2 and 12-3, respectively, and 
capacitors C1, C2 and C3, respectively, connected in parallel between the 
outputs of the respective resistors and ground. The time constant increase 
circuits 91-1, 91-2 and 91-3 include capacitors C1', C2' and C3', 
respectively, which are connected in parallel with the capacitors C1, C2 
and C3, respectively, and on-off switches AS1', AS2' and AS3', 
respectively, that are serially connected to the capacitors C1', C2' and 
C3', respectively, and ground. The on-off switches AS1', AS2' and AS3' are 
normally in the off position so that the time constant increase circuits 
91-1, 91-2 and 91-3 are normally kept disabled. The on-off switches AS1', 
AS2' and AS3' will be turned on so that the time constant increase 
circuits 91-1, 91-2 and 91-3 are enabled upon receiving a time constant 
increase signal from the drive control unit 50, as will be discussed in 
detail later. The output terminals of the phase-delay filter circuits 
14-1, 14-2 and 14-3 are the common connecting points of R1/C1/C1', 
R2/C2/C2', and R3/C3/C3', respectively. 
The phase-delay filter circuits 14-1, 14-2 and 14-3 output phase-delayed 
output voltages Fu, Fv and Fw, respectively, that are provided to the 
positive input terminals of voltage comparators 22-1, 22-2 and 22-3, 
respectively. 
The voltage VM across the dc power supply 20 is divided in half by a 
voltage divider 23, which consists of a pair of resistors R4 and R5, so as 
to produce a midpoint voltage Vn. The midpoint voltage Vn is provided to a 
comparator reference voltage computation circuit 24. The comparator 
reference voltage computation circuit 24 performs an arithmetic-logic 
operation in reference to the midpoint voltage Vn, as will be described in 
detail later, so as to output a sawtooth-wave comparator reference voltage 
VnOUT that is commonly provided to all of the negative input terminals of 
the voltage comparators 22-1, 22-2 and 22-3. In addition to the midpoint 
voltage Vn, to the comparator reference voltage computation circuit 24 are 
individually provided a basic reference voltage Vref from a DA converter 
26 and a pair of switch on-off signals S1 and S2 from the drive control 
unit 50. Explanation will be made in detail later as to the basic 
reference voltage Vref and the switch on-off signals S1 and S2. 
FIG. 2 is a detailed schematic diagram of the comparator reference voltage 
computation circuit 24. The basic reference voltage Vref is commonly 
provided to a pair of on-off switches AS1 and AS2 connected in parallel. 
The switches AS1 and AS2 are alternately turned on and off at a frequency 
proportional to a current rotor speed by the pair of switch on-off signals 
S1 and S2, respectively, transmitted from the drive control unit 50 (FIG. 
1), so that the outputs of the switches AS1 and AS2 are alternately nil or 
the basic reference voltage Vref itself. The output of the switch AS1 is 
provided to the positive (non-inverting) input terminal of a first 
operation amplifier OP1 and the output of the switch AS2 is provided to 
the negative (inverting) input terminal of the first operation amplifier 
OP1. 
Three resistors R6, R7 and R8, each having an identical resistance value, 
are connected to the first operation amplifier OP1 as shown. Since the 
resistance value of the resistors R6, R7 and R8 are all the same, when the 
switch AS1 is "ON" (the switch AS2 is "OFF") the first operation amplifier 
OP1 will function as a non-inverting amplifier having an amplification 
factor 1, whereby an output voltage V1 thereof will be the basic reference 
voltage Vref itself. Conversely, when the switch AS2 is "ON" (the switch 
AS1 is "OFF") the first operation amplifier OP1 will function as an 
inverting amplifier having an amplification factor 1, whereby the output 
voltage V1 thereof will be an inverted basic reference voltage, i.e. 
-Vref. 
The output voltage V1 of the first operation amplifier OP1 and the midpoint 
voltage Vn are provided to the positive input terminal of a second 
operation amplifier OP2. Four resistors R9, R10, R11 and R12, each having 
an identical resistance value, are connected to the second operation 
amplifier OP2 as shown. Since the resistance values of the resistors R9, 
R10, R11 and R12 are all the same, the second operation amplifier OP2 
functions as a voltage summing amplifier with the voltages V1 and Vn being 
the input voltages to be summed up. An output voltage V2 of the second 
operation amplifier OP2 is provided to a low-pass filter LPF, which 
consists of a resistor R13 and a capacitor C5, so that the above mentioned 
sawtooth-wave comparator reference voltage VnOUT is outputted therefrom. 
FIG. 3 is a waveform diagram that will help explain the function of the 
comparator reference voltage computation circuit 24 of FIG. 2. The six 
serial steps (1).about.(6), denoted by the word "STEP", constitute one 
electrical cycle period (i.e. 360.degree. electrical angle) that 
corresponds to one energizing cycle period of the 3-phase stator windings. 
The rotor 13 keeps rotation synchronously with the revolving magnetic flux 
as the 6-step cycle is repeated. In this case, the sawtooth-wave 
comparator reference voltage VnOUT has a cycle period that equals to one 
third of the cycle period (i.e. three times in frequency) of the terminal 
voltages Vu, Vv and Vw of the stator windings 12-1, 12-2 and 12-3, 
respectively, or the cycle period for energizing the 3-phase stator 
windings. 
In FIG. 3, "Cu", "Cv" and "Cw" represent the waveforms of the output 
voltages of the voltage comparators 22-1, 22-2 and 22-3, respectively, 
shown in FIG. 1. The cycle period, or the frequency, of the output 
voltages Cu, Cv and Cw is identical to that of the winding terminal 
voltages Vu, Vv and Vw. The switchings from one step to the succeeding 
step for the steps (1).about.(6) are performed by the drive control unit 
50 in reference to the waveforms Cu, Cv and Cw with a frequency 
proportional to the rotor's angular speed. "AS1" and "AS2" represent the 
timings of the ON/OFF states of the switches AS1 and AS2, respectively, 
that alternately occur in synchronism with the steps (1).about.(6). "V1" 
represents a waveform of the output voltage V1 of the first operation 
amplifier OP1, which is a rectangular waveform having a 2-times Vref 
amplitude with a center voltage grounded and a cycle period being equal to 
a 2-step time period. "V2" represents a waveform of the output voltage V2 
of the second operation amplifier OP2, which has the same waveform, the 
same amplitude and the same cycle period as those of the output voltage 
V1, but a center voltage of the amplitude being Vn. In other words, the 
voltage V2 is a voltage of V1 shifted up by Vn. The comparator reference 
voltage VnOUT has the same center voltage (Vn) and the same cycle period 
as those of the voltage V2. 
FIGS. 4(A).about.(E) shows voltage waveforms at parts of the circuitry 
shown in FIG. 1 that determine motor driving steps (or, switching steps) 
and on-off timings of the solid-state switching elements 17-1.about.6 of 
the switching circuit 15 in a first motor driving mode. More specifically, 
FIG. 4(A) shows six motor driving steps (1), (2), (3), (4), (5) and. (6), 
each step corresponding to a 60.degree. electrical angle and the complete 
6-step period corresponds to one energizing cycle for the 3-phase stator 
windings; FIG. 4(B) shows a waveform of the terminal voltage Vu of the 
winding 12-1; FIG. 4(C) shows waveforms of the phase-delayed output 
voltage Fu of the phase-delay filter circuit 14-1 and the comparator 
reference voltage VnOUT outputted from the comparator reference voltage 
computation circuit 24; FIG. 4(D) shows a waveform of the output voltage 
Cu of the voltage comparator 22-1; and FIG. 4(E) shows "ON" state timings 
of the six solid-state switching elements 17-1(U.sup.+), 17-2(V.sup.+), 
17-3(W.sup.+), 17-4(U.sup.-), 17-5(V.sup.-) and 17-6(W.sup.-). 
The output voltages Fu, Fv and Fw of the phase-delay filter circuits 14-1, 
14-2 and 14-3, respectively, are delayed by approximately 60.degree. with 
respect to the phase angles of the winding terminal voltages Vu. Vv and 
Vw, respectively. As shown in FIG. 4(B), the winding terminal voltage Vu 
is of a trapezoidal waveform having spikes of a voltage Vsp at the ends of 
steps (2) and (5). Such spikes appear in the terminal voltages Vu, Vv and 
Vw when the corresponding switching elements are turned off, or, in other 
words, the currents to the respective stator windings 12-1, 12-2 and 12-3 
are interrupted by the switching circuit 15. The phase-delayed output 
voltage Fu, as shown in FIG. 4 (C) of the phase-delay filter circuit 14-1 
is compared with the comparator reference voltage VnouT outputted from the 
comparator reference voltage computation circuit 24 by the voltage 
comparator 22-1, and the voltage Cu as shown in FIG.4 (D) is outputted 
from the voltage comparator 22-1. The output voltage Cu has a rectangular 
waveform that rises or falls as the voltage Fu and the voltage VnOUT come 
even with each other, as shown in FIG. 4(D) along with FIG. 4(C). 
Reference is now made to FIG. 3 along with FIGS. 4(A).about.(E). 
Immediately before the voltage Cu rises at the end of step (6), as 
indicated by reference numeral 31, the voltages Cu and Cv are at level "0" 
and the voltage Cw is at level "1", and the switching elements 17-3 
(W.sup.+) and 17-5(V.sup.-) are in "ON" state. As the rotor 13 maintains 
rotation, and when the voltage Cu turns from level "0" to level "1", step 
6 is switched over to step 1. The drive control unit 50 receives this 
switching information and, simultaneously, the drive control unit 50 
transmits switching control signals individually to the switching elements 
17-3 (W.sup.+) and 17-3 (U.sup.+) so as to cause the switching element 
17-3 (W.sup.+) to be turned "OFF" and 17-3 (U.sup.+) to be turned "ON", as 
will be understood in reference to FIG. 4(E). Simultaneously, the drive 
control unit 50 causes the switch AS1 to be turned "ON" and the switch AS2 
to be turned "OFF", as shown in FIG. 3. Then, at this time, the output 
voltage VnOUT of the comparator reference voltage computation circuit 24 
starts to increase, as seen in FIG. 4(C). Likewise, when the voltage Cu 
falls at the end of step (3), as indicated by reference numeral 32, the 
drive control unit 50 transmits switching control signals individually to 
the switching elements 17-6(W.sup.-) and 17-4(U.sup.-) so as to cause the 
switching element 17-6(W.sup.-) to be turned "OFF" and 17-4(U.sup.-) to be 
turned "ON", as will be understood in reference to FIG. 4(E). 
As described above, the switching control signals provided from the drive 
control unit 50 to the phase "U" switching elements, for example, to start 
energizing the phase "U" stator winding 12-1 are obtained from the voltage 
Cu that is derived from the terminal voltage Vu of the phase "U" stator 
winding 12-1. Similarly, other switching control signals are produced in 
the drive control unit 50 responsive to the respective output voltages Cv 
(phase "V") and Cw (phase "W"), which are derived from the stator winding 
terminal voltages Vv and Vw, respectively, and transmitted to the control 
terminals of the corresponding switching elements. Thus, the stator 
windings 12-1, 12-2 and 12-3 are provided with 3-phase dc voltages, with a 
shifted phase angle of 120.degree. one another, from the dc power supply 
20 by way of the electronic switching circuit 15, and a revolving magnetic 
flux generated by the 3-phase windings being energized causes the 
permanent-magnet rotor 13 to be kept rotated. 
The drive control unit 50 detects a rotational speed of the rotor 13 from 
the frequency of the output Cu, Cv or Cw of the voltage comparators 22-1, 
22-2 or 22-3, respectively. Then, the drive control unit 50, according to 
the rotational speed of the rotor 13, transmits a basic reference voltage 
setting signal CDA, to the DA converter 26 so that the DA converter 
generates an adjusted and optimum basic reference voltage Vref that varies 
depending on the rotor speed. The basic reference voltage setting signal 
CDA transmitted to the DA converter 26 causes the basic reference voltage 
Vref to be large when the rotor speed is large, and small when the rotor 
speed is small. The increase or decrease of the basic reference voltage 
Vref causes the amplitude of the sawtooth-wave output voltage VnOUT of the 
comparator reference voltage computation circuit 24 to be increased or 
decreased, respectively. 
FIG. 5 is a graph showing phase delay angle vs. frequency characteristics 
of phase-delay filter circuits. In the graph, "fc" represents the cutoff 
frequency of the phase-delay circuits. (Namely, fc=1/2.pi.RC) As shown in 
FIG. 5, the delay angle varies as the frequency varies within a limited 
frequency range. There is no delay in the frequency range from 0 to 0.1 
fc. The delay angle increases from 0.degree. to 90.degree. as the 
frequency increases from 0.1 fc to 10 fc, but the delay angle remains 
constant at 90.degree. if the frequency exceeds 10 fc. 
Phase-delay filter circuits of a conventional brushless dc motor unit are 
intended to be used in a frequency range of the saturation region, where 
the delay angle is 90.degree. constant. The main reason for that is once 
the R and C values of the phase-delay filter circuits are determined so 
that the frequency range of the induced winding terminal voltages, which 
represents the rotor speed range, comes within the saturated region (over 
10 fc in FIG. 5), the delay angle can be maintained at 90.degree. constant 
as long as the rotor speed stays within the intended range, and this leads 
to a simple circuit structure. However, once the rotor speed (i.e. 
frequency) is out of the intended operational range and comes down into 
the non-saturation region, the delay angle becomes smaller than 
90.degree., thereby causing the switching timings to be excessively 
advanced and the motor control difficult. 
Whereas, since the phase-delay filter circuits 14-1, 14-2 and 14-3 are 
purposely operated with a delay angle smaller than 90.degree., such as 
60.degree., the operating region, indicated by letter "A" in FIG. 5 is in 
the linear region (non-saturation region). Therefore within this operating 
region, the amount of the delay varies depending on the frequency of the 
output voltages of the phase-delay filter circuits 14-1.about.3 or of the 
terminal voltages Vu, Vv and Vw of the stator windings 12-1, 12-2 and 
12-3, respectively. 
Namely, as the rotor speed increases, the delay angle of the outputs Fu, Fv 
and Fw of the phase-delay filter circuits 14-1, 14-2 and 14-3, 
respectively, also increases. This added phase angle delay causes the 
turn-on timings of the switching elements of the switching circuit 15 to 
be also delayed with respect to the current angular position of the rotor 
13. Therefore, when the rotor speed exceeds a certain speed, the amount of 
the phase-delay may become excessive and the rotor 13 may consequently 
trip off. Oppositely, when the motor speed is too slow, the amount of the 
phase-delay of the outputs Fu, Fv and Fw may become too small, causing the 
switching timings to be unwanted advanced, and the rotor may likewise trip 
off. It can be said in this case that the speed range in which the rotor 
can be run safely and reliably will have to be limited. The following 
discussion pertains to a solution to such a problem. 
The basic reference voltage Vref is made to be increased when the rotor 
speed is large, and decreased when the rotor speed is small, by the drive 
control unit 50 and the DA converter 26. The increase or decrease of the 
basic reference voltage Vref causes the amplitude of the comparator 
reference voltage VnOUT outputted from the comparator reference voltage 
computation circuit 24 to be increased or decreased, respectively. 
FIG. 6 is a graph showing a correlation between the variation of the 
amplitude of the comparator reference voltage VnOUT and the shifting of 
electrical angle where the comparator reference voltage VnOUT becomes even 
with the output voltage Fu. In reference back to FIGS. 2 an 3, when the 
basic reference voltage Vref inputted to the comparator reference voltage 
computation circuit 24 is increased, the output voltages V1 and V2 of the 
operation amplifiers OP1 and OP2, respectively, are also increased, and, 
consequently, the amplitude of the filtered sawtooth-wave output voltage, 
i.e. the comparator reference voltage VnOUT , is increased as well. In 
FIG. 6, ".DELTA..theta." represents the electrical angle difference 
between the cross point P1 of the output voltage Fu with the output 
voltage Vnout (i.e. Fu/Vnout cross point) and the cross point P2 of the 
output voltage Fu with the midpoint voltage Vn (i.e. Fu/Vn cross point). 
If the amplitude of the output voltage VnOUT increases the angle 
difference .DELTA..theta. will increase because the FU/VnOUT cross point 
P1 will shift to the left, as FIG. 6 is viewed, and, conversely, if the 
amplitude of VnOUT decreases the angle difference .DELTA..theta. will also 
decrease because the FU/VnOUT cross point P2 will shift to the right. As 
the angle difference .DELTA..theta. increases, the rise times of the 
corresponding comparator output voltage Cu will advance, and then the 
turn-on timings of the corresponding switching elements will also advance. 
The same can be said with regard to the phase-delay filter output voltages 
Fv and Fw, the comparator output voltages Cv and Cw, and the turn-on 
timings of the corresponding switching elements. Therefore, by regulating 
the amplitude of the basic reference voltage Vref according to the 
rotational speed of the rotor 13, adjusted "ON" timings of the switching 
circuit 15 can be obtained. 
A current meter 18 is installed in the power supply line on the positive 
side of the dc power supply 20. The current meter 18 measures the current 
in the line and outputs an analogue signal to an AD converter 19, where 
the amount of measured current is converted to a digital signal CAD, which 
is transmitted to the drive control unit 50. Thus, the drive control unit 
50 monitors the amount of the currents supplied from the dc power supply 
20 to the stator windings 12-1, 12-2 and 12-3 by way of the switching 
circuit 15. Then, when the amount of the currents monitored by the drive 
control unit 50 exceeds a predetermined upper value, the drive control 
unit 50 transmits a control signal CDA to the DA converter 26 to decrease 
the value of the basic reference voltage Vref. Conversely, when the amount 
of the currents is less than a predetermined lower value, the drive 
control unit 50 transmits a control signal CDA to the DA converter 26 to 
increase the value of the basic reference voltage Vref. 
Whereas, in a conventional brushless dc motor, when the motor drive 
currents increase, the turn-on timings of the solid-state switching 
elements advance for the reason that will be mentioned below. This 
phenomenon makes it difficult to maintain an efficient operation of the 
motor, and that may further lead to a rotor trip-off problem. In a 
conventional brushless dc motor, spikes appear in the terminal voltages of 
stator windings, as exemplified by spike voltage Vsp of the stator 
terminal voltage Vu shown in FIG. 4(B). The pulse widths of such spikes 
increase as the motor drive currents increase. Such increased spike pulse 
widths cause to minimize the amount of delay of the outputs of the 
phase-delay filter circuits. A solution to such a problem is to adjust the 
value of the basic reference voltage Vref according to the amount of the 
motor drive currents. In other words, adjusted turn-on timings can be 
obtained depending on the variation of the motor drive currents. 
As mentioned above, the time constant increase circuits 91-1, 91-2 and 91-3 
are provided between the output terminals of the main filter circuits 
21-1, 21-2 and 21-3, respectively, and the positive input terminals of the 
comparators 22-1, 22-2 and 22-3, respectively. The drive control unit 50 
can transmit a common switch control signal (or, a time constant increase 
signal) to all of the on-off switches AS1', AS2' and AS3' simultaneously 
so as to turn on or off the switches, thereby enabling or disabling the 
capacitors C1', C2' and C3'. 
As mentioned above, the phase delay angle of the phase-delay filter 
circuits is less than 90.degree., i.e. approximately 60.degree.. 
Therefore, as explained above in reference to FIG. 5, when the rotor speed 
decreases (i.e. the frequency decreases:) the delay angle of the main 
filter circuits 21-1, 21-2 and 21-3 becomes smaller, thereby decreasing 
the phase delay angle of the voltages having passed the phase-delay filter 
circuits 14-1, 14-2 and 14-3, provided that the time constant increase 
circuits 91-1, 91-2 and 91-3 are disabled. Such decrease of the phase 
delay angle can be compensated by decreasing the value of the basic 
reference voltage Vref as long as the rotor speed is within a normal 
range. However, when the rotor speed is low, such as less than 1,000 rpm, 
even if the basic reference voltage Vref is made zero, there is a 
possibility that no proper compensation to phase delay can be made, and 
the rotor may consequently trip off. 
In view of such a problem, the drive control unit 50 monitors the rotor 
speed from the frequency of the voltage Cu, Cv, or Cw, and when the 
monitored rotor speed becomes smaller than a predetermined value, the 
drive control unit 50 transmits a time constant increase signal to cause 
the switches AS1', AS2', and AS3' to be closed, thereby enabling the time 
constant increase circuits 91-1, 91-2 and 91-3. Then, the time constant 
and the amount of phase delay of the phase-delay circuits 14-1, 14-2 and 
14-3 will be increased, so that the possible rotor trip-off at a low speed 
can be prevented. 
FIG. 7 is a block diagram of an improved apparatus for driving and 
controlling a brushless motor according to the present invention. FIG. 1 
particularly shows details of the drive control unit 50 of the present 
invention. Between FIG. 1 and FIG. 7, the same reference numerals denote 
the same electrical devices or components having the same functions that 
have already been explained above. Therefore, no duplicate explanation 
will be made here on the devices shown in FIG. 7 if the devices are also 
shown in FIG. 1. The reference numeral "14" in FIG. 7 collectively 
represents the phase-delay filter circuits 14-1, 14-2 and 14-3 as shown in 
FIG. 1, and the reference numeral "22" in FIG. 7 collectively represents 
the comparators 22-1, 22-2 and 22-3 as shown in FIG. 1. 
The drive control unit 50 includes a first drive controller 51, which is a 
primary drive controller, a second drive controller 52, a drive signal 
output controller 53, a driving mode selection unit 54, a phase-delay 
filter time constant changing unit 55, a driving step period memory unit 
56, a driving step period timer 57, and a driving step number counter 58. 
Referring back to FIG. 1, the voltage comparators 22 individually output 
signals Cu, Cv and Cw representing current angular positions of the 
revolving rotor by comparing the phase-delayed output voltages of the 
phase-delay circuits 14 with the comparator reference voltage, i.e. the 
output voltage of the comparator reference voltage computation circuit 24. 
The output signals Cu, Cv and Cw of the voltage comparators 22 are provided 
to the first drive controller 51 according to the present invention. The 
drive current signal CAD representing the amount of total motor drive 
current measured by the current meter 18 is also provided to the first 
drive controller 51. 
The first drive controller 51, while being selected by the driving mode 
selection unit 54, provides a first drive control signal to the drive 
signal output controller 53, which in turn controls on-off switchings of 
the solid-state switching circuit 15 in a first motor driving mode 
according to the first drive control signal provided from the first drive 
controller 51. Therefore, in the first motor driving mode, the motor 
driving steps, which are the switching steps of the solid-state switching 
circuit 15, are determined according to the output signals Cu, Cv and Cw 
of the voltage comparators 22, as described above. The first drive 
controller 51 also provides data on the amount of the total drive current 
measured by the current meter 18 and a current rotational speed of the 
rotor 13 to the phase-delay filter time constant changing unit 55. The 
phase-delay filter time constant changing unit 55 transmits a time 
constant change signal to the phase-delay circuits 14 if the amount of the 
total drive current exceeds a predetermined maximum level. The phase-delay 
filter time constant changing unit 55 also transmits a time constant 
change signal to the phase-delay circuits 14 if the rotational speed of 
the rotor 13 is below a predetermined minimum speed. The time constant 
change signal, when transmitted from the phase-delay filter time constant 
changing unit 55, causes the on-off switches AS1', As2' and AS3' to be 
closed and thereby the time constant increase circuits 91-1, 91-2 and 91-3 
to be enabled so that the time constant of the phase-delay filter circuits 
14 is increased. 
The second drive controller 21, if selected by the driving mode selection 
unit 54, provides a second drive control signal to the drive signal output 
controller 53, which in turn controls on-off switchings of the solid-state 
switching circuit 15 in a second motor driving mode according to the 
second drive control signal provided from the second drive controller 52. 
Either the first drive controller 51 or the second drive controller 52 is 
selected at a time by the driving mode selection unit 54, and the selected 
controller, either 51 or 52, transmits the switching control signal to the 
drive signal output controller 53. The phase-delay filter time constant 
changing unit 55 also provides the time constant change signal to the 
driving mode selection unit 54, in addition to the phase-delay filter 
circuits 14. The time constant change signal causes the driving mode 
selection unit 54 to switch selection of the drive controller from the 
first drive controller 51 to the second drive controller 52 so that the 
drive signal output controller 53 to be switched from the first motor 
driving mode to the second motor driving mode so as to drive the 
solid-state switching circuit 15 in the second motor driving mode on a 
specified driving step and for a specified time period, which will be 
discussed in detail later. 
As mentioned before, the drive control unit 50 also includes the driving 
step period memory unit 56, the driving step period timer 57, and the 
driving step number counter 58. The control signal transmitted from the 
drive signal output controller 53 to the solid-state switching circuit 15 
is also provided to the driving step period memory unit 55 so that the 
driving step period memory unit 55 stores datum of the time period of one 
driving step, or one switching step, of the control signal transmitted 
from the drive signal output controller 53 to the solid-state switching 
circuit 15 for driving the brushless motor. The datum of the driving step 
time period stored in the driving step period memory unit 56 is always 
refreshed so that the stored datum is of the latest driving step 
transmitted from the drive signal output controller 53. As will be 
discussed later, the driving step period timer 57 reads the driving step 
time period from the driving step period memory unit 56 and performs 
timing for the driving step time period. In other words, the driving step 
period timer 57 times exactly the same driving step time period of the 
control signal transmitted from the drive signal output controller 53 to 
the switching circuit 15, which is stored in the driving step period 
memory unit 56. The driving step number counter 58 counts a predetermined 
number of driving steps having the driving step period timed and set in 
the driving step period timer 57. The second drive controller 52 performs 
a second drive control in the second motor driving mode only for a time 
period (hereinafter identified as "T2") that equals to the one-step time 
period (hereinafter identified as "Ts") timed and set in the driving step 
period timer 57 multiplied by the predetermined number (hereinafter 
identified as "Ns") of steps set in the driving step number counter 58. 
The exemplified number (Ns) of steps set in the driving step number 
counter 58 is three (3), as explained later. Namely: 
EQU T2=Ts.times.Ns 
Where: 
T2: Time period of second motor driving mode 
Ts: Time period of one driving step for the second 
motor drive mode, which is timed and set in the driving step period timer 
57 
Ns: Number of steps set in the driving step number counter 58 for the 
second motor driving mode e.g. "3") 
As soon as the time period (T2) for the second. motor driving mode has 
lapsed, the second drive controller 52 transmits a count-up signal to the 
driving mode selection unit 54 that in turn causes to select the first 
drive controller 51, thus the driving mode reverts to the first motor 
driving mode. 
FIG. 8 is a flow chart showing the function of the drive control unit 50 
shown in FIGS. 1 and 7. While the rotor 13 is running, in Step 1 (S1), the 
driving step time period (Ts) for driving the brushless motor 11 is stored 
in the driving step period memory unit 56. if, in Step 2 (S2), no time 
constant change signal from the phase-delay filter time constant changing 
unit 55 is detected by the driving mode selection unit 54, the first drive 
controller 17 is kept selected so that the first motor driving mode is 
maintained (Step 3 (S3)). If, in Step 2 (S2), the time constant change 
signal from the phase-delay filter time constant changing unit 55 is 
detected by the driving mode selection unit 54, the second drive 
controller 52 is selected, then the driving step period timer 57 reads the 
driving step time period (Ts) stored in the driving step period memory 
unit 56 and starts timing for the one-step time period (Ts) (Step 4 (S4)). 
At the same time, the drive signal output controller 53 transmits a drive 
control signal for driving the solid-state switching circuit 15 in the 
second motor driving mode according to the driving steps regulated by the 
second drive controller 21 (Step 5 (S5)). 
Next, in Step 6 (S6), a judgement is made as to whether or not the time 
counting of the one-step time period (Ts) in the driving step period timer 
57 started in Step 4 has completed. If the time period counting has not 
completed in Step 6 (S6), the sequence reverts to the start of Step 6 
(S6). Namely, the time period counting continues. If it is determined that 
the time period counting has completed in Step 6 (S6), a deduction is made 
by one count from the number (Ns) set for counting in the driving step 
number counter 58 (Step 7 (S7)). In Step 8 (S8), if the left number of the 
driving step number counter 58 is other than zero (0), the sequence 
reverts to Step 4 (S4). If the left number of the driving step number 
counter 58 is zero (0) in Step 8 (S8), the sequence reverts to Step 1, as 
marked with "A" in FIG. 8, after resetting the driving step number counter 
58 for the predetermined number (Ns), which is "three (3)" in the example 
provided (Step 9 (S9)). 
The number (Ns) (e.g. "3") to be set and counted by the driving step number 
counter 25 is initially determined in a manner that the total time period 
of the one-step period (Ts) multiplied by the number (Ns) to be counted is 
greater than the difference between the two time constants before and 
after the change of the time constant of the phase-delay circuits 14. 
Provided, for example, that the time constants before and after the change 
of the time constant are 1.14 msec and 2.45 msec, respectively, the 
difference between the two time constants, before and after, is 1.31 msec. 
This matter will be further discussed below in more detail. 
FIG. 9(a) and FIG. 9(b) are graphs showing, in the vertical axis varying 
amounts of the motor driving currents provided from the dc power supply 20 
and measured by the current meter 18 and, in the horizontal axis, elapsed 
time and the motor driving steps, or switching steps, on which the 
solid-state switching elements of the switching circuit 15 are turned on 
or off. FIG. 9(a), which is provided for a comparison purpose, represents 
an unimproved case of motor drive control in which no second motor driving 
mode is involved. FIG. 9(b) is an improved case according to the present 
invention, in which both the first motor driving mode and the second motor 
driving mode are involved. Referring to FIG. 9(a), the driving step time 
periods immediately after the time (hereinafter identified as "Tc") when 
the time constant of the phase-delay circuits is changed are substantially 
longer than those before the time constant is changed. On the other hand, 
in FIG. 9(b), little change is observed as to the driving step time 
periods before and after the time (Tc) when the time constant of the 
phase-delay circuits is changed. 
In both FIGS. 9(a) and 9(b), it is assumed, as an example, that the 
brushless motor has a 4-pole rotor and the rotational speed of the rotor 
is about 8,000 rpm. Under this rotational speed, the driving step time 
period (Ts) is obtained as follows: 
Number of Rotation of Rotor per Second: 8,000/60=133.3 
Time Period per Rotation: 1/133.3=7.50 msec 
In the case of a brushless motor, having 3-phase stator windings and a 
4-pole permanent-magnet rotor, 12 driving steps will make one rotation of 
the rotor. One six-step cycle shown in FIGS. 9(a) and 9(b) will make one 
half rotation. Therefore: 
Driving Step Time Period (Ts): 7.50 msec/12=625 .mu.sec 
As mentioned before, the difference between the exemplified two time 
constants before and after the change of the time constant is 1.31 msec. 
Whereas, 
625 .mu.sec.times.2=1,250 .mu.sec &lt;1.31 msec 
625 .mu.sec.times.3=1,875 .mu.sec &gt;1.31 msec 
Since the accumulative time period of three driving steps exceeds the 
difference (1.31 msec) of the time constants before and after the change 
of the time constant, it is desirable to adopt the number of three (3) for 
the number (Ns) to be counted by the driving step number counter 58, as 
already mentioned above. 
Referring back to FIG. 9(a), in the case of an unimproved motor drive 
control, it will be understood from the graph that excessive motor driving 
currents, which are as large as three times the normal amount of current, 
are shown after the time constant of the phase-delay circuits is changed. 
Furthermore, it is also shown in FIG. 9(a) that the driving step time 
periods become longer immediately after the time constant is changed as 
compared with the normal driving step time periods before the change of 
the time constant. The electric angles of the normal driving step time 
periods are in the range of 40.degree. to 60.degree.. However, the 
electric angles of those immediately after the change of the time constant 
increase by about 20.degree. to become about 80.degree.. As opposed to 
this example, in the case an improved control apparatus according to the 
present invention, as shown in FIG. 9(b), little change is observed in 
both the motor driving currents and the driving step time periods before 
and after the change of the time constant of the phase-delay circuits. 
FIGS. 10(a) and 10(b) are graphs showing waveforms of output voltages of 
one of the phase-delay circuits and the comparator reference voltage 
computation circuit before and after the time constant of the phase-delay 
circuits is changed in the cases of using an unimproved drive control unit 
and the improved drive control unit 50 according to the present invention, 
respectively. The waveform Fu represents an output voltage of one of the 
phase-delay circuits and the waveform VnOUT represents a sawtooth output 
voltage of the comparator reference voltage computation circuit. In both 
the graphs, the horizontal axis represents elapsed time and the motor 
driving step periods and the vertical axis represents voltage. 
In reference to both FIGS. 10(a) and 10(b), it will be understood that 
because the waveform and the slopes of the output voltage (Fu) of the 
phase-delay circuit change after the time (Tc) of the change of time 
constant, the timings when the phase-delay circuit output voltage (Fu) 
becomes even with the comparator reference voltage (VnOUT) shift. In the 
unimproved case, because of these shiftings of the timings, the time 
periods of the motor driving steps are extended immediately after the time 
(Tc) of the change of time constant, as shown in FIG. 10(a). Under such 
situation, the angular positions of the rotating rotor will not be 
precisely detected. This in turn will cause the electric angles of the 
driving step periods come off a regularable electric angle range (i.e. 
40.about.60.degree.), resulting in an excessive amount of motor driving 
current. 
On the other hand, in reference to FIG. 10(b), in the case the drive 
control unit 50 is used according to the present invention, although the 
waveform and the slopes of the output voltage of the phase-delay circuits 
14 change immediately after the time (Tc) of the change of time constant, 
the motor driving step periods are unchanged and not affected by the 
change of the waveform of the phase-delay circuit output voltage (Fu). In 
this case, the driving step periods are maintained constant because each 
of the driving step periods, immediately after the change of the time 
constant, is the same driving step time period (Ts) as of the driving step 
stored in the driving step period memory unit 56, which is nothing but the 
driving step immediately before the time (Tc) of the change of time 
constant. The driving step period is timed by the driving step period 
timer 57 and the number of steps is counted by the driving step number 
counter 58 for the accumulative period (T2), which is the time period for 
the second motor driving mode. According to the example explained above 
and shown in FIG. 10(b), the number (Ns) of driving steps counted by the 
driving step number counter 58 is three (3). At tire end of the period T2, 
the driving mode selection unit 54 selects the first drive controller 51 
so that the driving mode reverts from the second motor driving mode to the 
first motor driving mode, in which the driving steps are determined 
according to the output signals of the voltage comparators 22. 
The present invention shall not be limited to the embodiment described 
above. As an alternative embodiment, the driving step period memory unit 
56, the driving step period timer 57, and the driving step number counter 
58, as shown in FIG. 7, may be omitted. FIG. 11 is an flowchart showing 
the function of the motor control apparatus according to such an 
alternative embodiment of the present invention. 
Referring to FIG. 11, in this alternative case, a judgement is made as to 
whether or not the time constant change signal is present by the driving 
mode selection unit 54 while the rotor is running (Step 11 (S11)). If no 
time constant change signal is present, the first drive controller 51 is 
selected and the motor is controlled in the first motor driving mode, as 
described above (Step 12 (S12)). If the time constant change signal is 
detected in Step 11, the second drive controller 52 is selected and the 
motor is controlled in the second motor driving mode, as also described 
above Step 13 (S13)). In this case of the alternative embodiment the 
second drive controller 52 is preprogrammed so as to always transmit a 
drive control signal to the drive signal output controller 53 according to 
a predetermined driving step time period (e.g. 625 .mu.sec, in case of a 
4-pole rotor) and a predetermined driving step number (e.g. "3"). In Step 
14 (S14), a judgement is made as to whether or not the second motor 
driving mode for the predetermined number of driving steps has been 
completed. If completed, the sequence reverts to Step 11 (S11), as marked 
with "A" in FIG. 11. 
It should also be understood that various changes and modifications may be 
made in the above described embodiments which provide the characteristics 
of the present invention without departing from the spirit and principle 
thereof particularly as defined in the following claims.