Method and system for pre-equalization in a single weight (SW) single channel (SC) multiple-input multiple-output (MIMO) system

In wireless systems, a method and system for pre-equalization in a single weight (SW) single channel (SC) multiple-input multiple-output (MIMO) system are provided. A first receive antenna and at least one additional receive antenna may receive a plurality of SC communication signals transmitted from at least two transmit antennas. Estimates of the propagation channels between transmit and receive antennas may be performed concurrently and may be determined from baseband combined channel estimates. Channel weights may be determined to modify the signals received by the additional receive antennas. Pre-equalization weight parameters may be determined to modify subsequent signals transmitted from the transmit antennas. The pre-equalization weight parameters may be based on the propagation channel estimates and may be determined by LMS, RLS, DMI, or by minimizing a cost function. Closed loop transmit diversity may also be supported.

FIELD OF THE INVENTION

Certain embodiments of the invention relate to the processing of wireless communication signals. More specifically, certain embodiments of the invention relate to a method and system for pre-equalization in a single weight (SW) single channel (SC) multiple-input multiple-output (MIMO) system.

BACKGROUND OF THE INVENTION

In most current wireless communication systems, nodes in a network may be configured to operate based on a single transmit and a single receive antenna. However, for many of current wireless systems, the use of multiple transmit and/or receive antennas may result in an improved overall system performance. These multi-antenna configurations, also known as smart antenna techniques, may be utilized to reduce the negative effects that multipath and/or signal interference may have on signal reception. Existing systems and/or systems which are being currently deployed, for example, CDMA-based systems, TDMA-based systems, WLAN systems, and OFDM-based systems such as IEEE 802.11 a/g/n, may benefit from configurations based on multiple transmit and/or receive antennas. It is anticipated that smart antenna techniques may be increasingly utilized both in connection with the deployment of base station infrastructure and mobile subscriber units in cellular systems to address the increasing capacity demands being placed on those systems. These demands arise, in part, from a shift underway from current voice-based services to next-generation wireless multimedia services that provide voice, video, and data communication.

The utilization of multiple transmit and/or receive antennas is designed to introduce a diversity gain and/or an array gain and to suppress interference generated within the signal reception process. Such diversity gains improve system performance by increasing received signal-to-noise ratio, by providing more robustness against signal interference, and/or by permitting greater frequency reuse for higher capacity. In communication systems that incorporate multi-antenna receivers, a set of M receive antennas may be utilized to null the effect of (M−1) interferers, for example. Accordingly, N signals may be simulataneously transmitted in the same bandwidth using N transmit antennas, with the transmitted signal then being separated into N respective signals by way of a set of N antennas deployed at the receiver. Systems that utilize multiple transmit and multiple receive antenna may be referred to as multiple-input multiple-output (MIMO) systems. One attractive aspect of multi-antenna systems, in particular MIMO systems, is the significant increase in system capacity that may be achieved by utilizing these transmission configurations. For a fixed overall transmitted power, the capacity offered by a MIMO configuration may scale with the increased signal-to-noise ratio (SNR).

However, the widespread deployment of multi-antenna systems in wireless communications, particularly in wireless handset devices, has been limited by the increased cost that results from increased size, complexity, and power consumption. Providing a separate RF chain for each transmit and receive antenna is a direct factor that increases the cost of multi-antenna systems. Each RF chain generally comprises a low noise amplifier (LNA), a filter, a downconverter, and an analog-to-digital converter (A/D). In certain existing single-antenna wireless receivers, the single required RF chain may account for over 30% of the receiver's total cost. It is therefore apparent that as the number of transmit and receive antennas increases, the system complexity, power consumption, and overall cost may increase.

In the case of a single RF chain with multiple antennas, there is the need to determine or estimate separate propagation channels. A simple method may comprise switching to a first receive antenna utilizing, for example, an RF switch, and estimate a first propagation channel. After estimating the first propagation channel, another receive antenna may be selected and its corresponding propagation channel may be estimated. In this regard, this process may be repeated until all the channels have been estimated. However, switching between receive antennas may disrupt the receiver's modem and may lower throughput. This approach may require additional hardware and may also result in propagation channel estimates at different time intervals. Any mechanisms that may be utilized to compensate for the presence of multiple time-varying propagation channels may also present added complexity and cost to the design and operation of MIMO systems.

A single weight approach may work best for a single path, that is, for flat fading channels, because a single weight may not combine all paths arriving at different delays optimally. To optimally combine each multipath at receiving antennas may require multiple weights at different delays. For example, the same number of weights as multipaths arriving at different delays may be required, which may be more like a complete channel equalization approach. On the other hand, utilizing a single weight may have an average combining effect on multiple paths, with sub-optimal performance. A single weight may not be selected so that an optimized combination of multiple paths may be achieved at the receiving antennas. For example, for a Rayleigh flat fading channel, a single weight solution may result in about a 6 dB gain, while for the channels with many Rayleigh faded paths the gain may be reduced to about 2 dB.

Moreover, multi-path propagation in band-limited time dispersive channels may cause inter-symbol interference (ISI), which has been recognized as a major obstacle in achieving increased digital transmission rates with the required accuracy. ISI may occur when the transmitted pulses are smeared out so that pulses that correspond to different symbols are not discernable or separable. Meanwhile, data received from a desired user may be disturbed by other transmitters, due to imperfections in the multiple access scheme, giving rise to inter-carrier interference (ICI). For a reliable digital transmission system, it is necessary to reduce the effects of ISI and ICI.

BRIEF SUMMARY OF THE INVENTION

A method and/or system for pre-equalization in a single weight (SW) single channel (SC) multiple-input multiple-output (MIMO) system, substantially as shown in and/or described in connection with at least one of the figures, as set forth more completely in the claims.

DETAILED DESCRIPTION OF THE INVENTION

Certain embodiments of the invention may be found in a method and system for pre-equalization in a single weight (SW) single channel (SC) multiple-input multiple-output (MIMO) system. A first receive antenna and at least one additional receive antenna may receive a plurality of SC communication signals transmitted from at least two transmit antennas. Estimates of the propagation channels between transmit and receive antennas may be performed concurrently and may be determined from baseband combined channel estimates. Channel weights may be determined to modify the signals received by the additional receive antennas. Pre-equalization weight parameters may be determined to modify subsequent signals transmitted from the transmit antennas. The pre-equalization weight parameters may be based on the propagation channel estimates and may be determined by LMS, RLS, DMI, or by minimizing a cost function. Closed loop transmit diversity may also be supported. The various embodiments of the invention may provide a good compromise between implementation complexity and performance gains to reduce the effects of, for example, inter-symbol interference (ISI) and/or inter-carrier interference (ICI) in MIMO systems.

Most communication channels suffer from multipath fading. To address multipath fading, different equalizer techniques may be used. Generally, equalization algorithms may be implemented at the receiver side of a communication link. However when the equalizer weight solution is available at the transmitter, then pre-equalizer techniques may be used. The method of equalization may be the same for pre-equalization at the transmitter and post-equalization at the receiver when the optimal weights become the inverse conjugate of the channel, for example. The weights may therefore be applied at either the transmitter during pre-equalization or at the receiver during post-equalization. These weights may be optimal when there is no interference present in the system. When any interference sources are present in the system, the optimum weights for the pre-equalization and post-equalization may be different. One of the benefits of using pre-equalizer techniques lies in the simplification of the receiver architecture that results from moving the complexity of the equalization operation to the transmitter. However, pre-equalizer techniques may be related to the feedback of the channel estimates. The delay may cause some lag between the received symbols and the corresponding transmitted symbols. Pre-equalization weights may be calculated for vector and matrix channels and applied to the transmitted symbols accordingly.

An approach that supports channel pre-equalization at the transmitter may be utilized to improve upon the use of a single weight solution when multipath signals are received by multiple receive antennas. The purpose of pre-equalization is to make signals appear at the receiver as a single path, that is, a flat fading channel. In the case of flat fading channels the two receive antenna SW solution may yield maximum gain that may approach 6 dB gain in flat fading Rayleigh channels, for example. A problem with pre-equalization with multiple receive antenna systems is that the pre-equalization may not pre-equalize the channels optimally for all multiple receiving antennas. An averaging effect may occur, which pre-equalizes the multipath channel at the multiple receiving antennas partially.

FIG. 1Ais a block diagram of an exemplary two-transmit (2-Tx) and two-receive (2-Rx) antennas wireless communication, in accordance with an embodiment of the invention. Referring toFIG. 1A, there is shown a wireless communication system100that may comprise a first transmit antenna (Tx_1)138, an additional transmit antenna (Tx_2)140, a first receive antenna (Rx_1)106, and an additional receive antenna (Rx_2)108. The wireless communication system100may further comprise a mixer110, an adder112, an RF block114, a baseband (BB) processor119, a single weight baseband generator (SWBBG)121, a single weight generator (SWG) channel estimator122, and a SWG algorithm block124.

The first transmit antenna, Tx_1138, and the additional or second transmit antenna, Tx_2140, may comprise suitable hardware that may be adapted to transmit a plurality of SC communication signals, ST, from a wireless transmitter device. The first receive antenna, Rx_1106, and the additional or second receive antenna, Rx_2108, may comprise suitable hardware that may be adapted to receive at least a portion of the transmitted SC communication signals in a wireless receiver device. For example, the receive antenna Rx_1106may receive signal sR1while the receive antenna Rx_2108may receive signal sR2. The propagation channels that corresponds to the paths taken by the SC communication signals transmitted from the transmit antennas Tx_1138and Tx_2140and received by the receive antenna Rx_1106may be represented by h11and h12respectively. In this regard, h11and h12may represent the actual time varying impulse responses of the radio frequency (RF) paths taken by the SC communication signals transmitted from the transmit antennas Tx_1138and Tx_2140and received by the receive antenna Rx_1106. The actual time varying impulse responses, hxy, may contain multiple propagation paths arriving at different delays.

Similarly, the propagation channels that corresponds to the paths taken by the SC communication signals transmitted from the transmit antennas Tx_1138and Tx_2140and received by the receive antenna Rx_2108may be represented by h21and h22respectively. In this regard, h21and h22may represent the actual time varying impulse responses of the RF paths taken by the SC communication signals transmitted from the transmit antennas Tx_1138and Tx_2140and received by the receive antenna Rx_2108. In some instances, a wireless transmitter device may be adapted to periodically transmit calibration and/or pilot signal that may be utilized by a 2-Rx antennas wireless receiver device to determine estimates of h11, h12, h21, and h22. The 2-Tx and 2-Rx antennas wireless communication system100inFIG. 1Amay represent a MIMO communication system.

The mixer110may comprise suitable logic and/or circuitry that may be adapted to operate as a complex multiplier that may modify the amplitude and/or phase of the portion of the SC communication signals received by the receive antenna Rx_2108via a rotation waveform ejwrtprovided by the SWBBG121, where wr=2πfrand fris the rotation frequency. In this regard, a channel weight comprising an amplitude component and phase component may be provided by the SWBBG121for modifying the signal received by the receive antenna Rx_2108to achieve channel orthogonality between the receive antenna Rx_1106and the receive antenna Rx_2108. In some implementations, the mixer110may comprise a variable gain amplifier and a phase shifter, for example.

Through the achieved channel orthogonality, estimates of h11, h12, h21, and h22may be determined by the SWG channel estimator122in the SWBBG121. The h11, h12, h21, and h22estimates may be utilized by the SWG algorithm block124to determine an optimum amplitude A and phase φ that modify signals received by the receive antenna Rx_2108via mixer110so that the receiver signal-to-interference-and-noise ratio (SINR) is maximized, for example. In some instances, instead of utilizing the rotation waveform ejwrtto achieve the channel orthogonality between the receive antenna Rx_1106and the receive antenna Rx_2108, square or triangular waveforms may be also utilized. Moreover, waveforms representing different orthogonal codes may also be utilized, similar to the CDMA orthogonal codes with the same spreading.

The output of the mixer110may be transferred to a bandpass filter, a low noise amplifier (LNA), and/or a phase shifter for further processing of the received signals. The adder112may comprise suitable hardware, logic, and/or circuitry that may be adapted to add the output of the receive antenna Rx_1106and the output of the mixer110to generate a combined received SC communication signal, sRC. In some instances, bringing the output signals of the receive antenna Rx_1106and the mixer110together into a single electrical connection may provide the functionality of the adder112. Notwithstanding, an output of the adder112may be transferred to the RF block114for further processing of the combined received SC communication signal, sRC.

The RF block114may comprise suitable logic and/or circuitry that may be adapted to process the combined received SC communication signal, sRC. The RF block114may perform, for example, filtering, amplification, and/or analog-to-digital (A/D) conversion operations. The BB processor119may comprise suitable logic, circuitry, and/or code that may be adapted to determine a first baseband combined channel estimate, ĥ1, which may comprise information regarding propagation channels h11and h21. The BB processor119may also be adapted to process the output of the RF block114to determine a second baseband combined channel estimate, ĥ2, which may comprise information regarding propagation channels h12and h22. The BB processor119may also be adapted to determine an estimate of the transmitted SC communication signals, ŝT.

The SWBBG121may comprise suitable logic, circuitry, and/or code that may be adapted to receive the first and second baseband combined channel estimates, ĥ1and ĥ2, from the BB processor119and generate phase and amplitude components of the rotation waveform to be applied by the mixer110to modify the portion of the SC communication signals received by the receive antenna Rx_2108, sR2. The SWG channel estimator122may comprise suitable logic, circuitry, and/or code that may be adapted to process the first and second baseband combined channel estimates, ĥ1and ĥ2, generated by the BB processor119and may determine a matrix Ĥ2×2of propagation channel estimates ĥ11, ĥ12, ĥ21, and ĥ22, which correspond to estimates of a matrix Ĥ2×2of time varying impulse responses h11, h12, h21, and h22respectively. The actual time varying impulse responses, hxy, may contain multiple propagation paths arriving at different delays. In that regard, the matrix Ĥ2×2of propagation channel estimates ĥ11, ĥ12, ĥ21, and ĥ22may consist of multiple path estimates arriving at different delays. The SWG algorithm block124may comprise suitable logic, circuitry, and/or code that may be adapted to determine a channel weight to be transferred to the mixer110to modify the signal sR2so that the receiver SINR is maximized. The channel weight to be transferred to the mixer110may refer to a phase, φ, and amplitude, A, that results in a maximum SINR.

FIG. 1Bis a block diagram of an exemplary two-transmit (2-Tx) and two-receive (2-Rx) antennas wireless communication with pre-equalization, in accordance with an embodiment of the invention. Referring toFIG. 1B, there is shown a wireless communication system131that may differ from the wireless communication system100inFIG. 1Ain that the wireless communication system131further comprises mixers130and132and a pre-equalizer125.

The mixers130and132may comprise suitable logic and/or circuitry that may be adapted to multiply a signal to be transmitted, sT0, with weight factors W1and W2respectively. For example, the weight factors W1and W2may correspond to phase and/or amplitude component feedback adjustments that may be generated by the pre-equalizer125. In this regard, the pre-equalizer125may transfer the weight factors or parameters that correspond to those weight factors to the transmitter via an uplink feedback process.

The pre-equalizer125may comprise suitable logic, circuitry, and/or code that may be adapted to determine a plurality of pre-equalization parameters and/or weight factors W1and W2based on the matrix Ĥ2×2of propagation channel estimates ĥ11, ĥ12, ĥ21, and ĥ22. The pre-equalization parameters may comprise phase and amplitude information to be transferred to the transmitter portion of the wireless communication system131. The weights or weight parameters determined by the pre-equalizer125may be a single channel tap or a weight vector for a frequency selective propagation channel. Moreover, the pre-equalizer125may be adapted to determine the pre-equalization parameters based on, for example, a least-mean squares (LMS) algorithm, a recursive least squares (RLS) algorithm, direct matrix inversion, a cost function analysis, or a second order statistical technique. When utilizing a cost function analysis, for example, coefficients utilized by the pre-equalizer to determine the pre-equalization parameters may be obtained based on the minimization of a cost function, J, of the form J=f(SINR) or J=f(SNR), where f(x) denotes a function of variable x and SINR and SNR are the signal-to-interference-and-noise ratio and signal-to-noise ratio of the received signals respectively. For example, a cost function J=(SINR)−1may be minimized to obtain pre-equalizer coefficients that may be utilized to determine the pre-equalization parameters. The pre-equalizer may apply and/or modify cost function parameters associated with variables utilized with the cost function. In certain instances, pre-coding techniques may be utilized in order to require less complicated processing of the pre-equalization parameters on the receiver side.

The SWG algorithm block124inFIG. 1Bmay be adapted to support two-transmit antenna closed loop mode1(CL1) and closed loop mode2(CL2) for transmit diversity as described in the 3rdGeneration Project Partnership (3GPP), Technical Specification Group Radio Access Network, Physical Layer Procedures (FDD), Release 6 (3GPP TS 25.214 V5.5.0, 2003-06). When either CL1or CL2are active, the wireless communication system131may be said to be in an active closed loop mode of operation. The SWG algorithm block124may generate weight factors W1and W2to support two-transmit antenna CL1and CL2transmit diversity. In this regard, the SWG algorithm block124may utilize, for example, similar operations as those for determining phase and amplitude adjustments at the wireless receiver when determining the phase and amplitude adjustments to be applied at a diversity transmitter when either CL1or CL2are active.

FIG. 1Cis a block diagram of an exemplary two-transmit (2-Tx) and two-receive (2-Rx) antennas wireless communication system that supports WCDMA/HSPDA, in accordance with an embodiment of the invention. Referring toFIG. 1C, there is shown a wireless communication system135that may differ from the wireless communication system100inFIG. 1Ain that the wireless communication system135may comprise chip matching filter (CMF)116, a cluster path processor (CPP)118, and a baseband (BB) processor120.

The CMF116may comprise suitable logic, circuitry, and/or code that may be adapted to operate as a matched-filter on the digital output from the RF block114. The output of the CMF116may be transferred, for example, to the CPP118and/or to the BB processor120for further processing. The CPP118may comprise suitable logic, circuitry, and/or code that may be adapted to process the filtered output of the CMF116to determine a first baseband combined channel estimate, ĥ1, which may comprise information regarding propagation channels h11and h21. The CPP118may also be adapted to process the filtered output of the CMF116to determine a second baseband combined channel estimate, ĥ2, which may comprise information regarding propagation channels h12and h22. In this regard, the CPP118may process the received signals in clusters. U.S. application Ser. No. 11/173,854 provides a detailed description of signal clusters and is hereby incorporated herein by reference in its entirety. The CPP118may also be adapted to generate a lock indicator signal that may be utilized by, for example, the BB processor120as an indication of whether the channel estimates are valid. The BB processor120may comprise suitable logic, circuitry, and/or code that may be adapted to digitally process the filtered output of the CMF116to determine an estimate of the transmitted SC communication signals, ŝT.

FIG. 1Dis a block diagram of an exemplary two-transmit (2-Tx) and two-receive (2-Rx) antennas wireless communication system with pre-equalization that supports WCDMA/HSPDA, in accordance with an embodiment of the invention. Referring toFIG. 1D, there is shown a wireless communication system137that may differ from the wireless communication system135inFIG. 1Cin that the wireless communication system137may comprise a dedicated physical channel (DPCH) block126, a mixer128, a first combiner134, a second combiner136, and the pre-equalizer125. The pre-equalizer125and the SWG algorithm block124inFIG. 1Dmay be adapted to operate substantially as the pre-equalizer125and the SWG algorithm block124inFIG. 1B.

The DPCH block126may comprise suitable logic, circuitry, and/or code that may be adapted to receive a plurality of input channels, for example, a dedicated physical control channel (DPCCH) and a dedicated physical data channel (DPDCH). The DPCH126may be adapted to simultaneously control the power on each of the DPCCH and DPDCH channels. The mixer128may comprise suitable logic and/or circuitry that may be adapted to multiply the output of DPCH126with a spread and/or scramble signal to generate a spread complex-valued signal that may be transferred to the inputs of the mixers130and132.

The output of the mixer130may be transferred to the first combiner134and the output of the mixer132may be transferred to the second combiner236. The first and second combiners134and136may comprise suitable logic, circuitry, and/or code that may be adapted to add or combine the outputs generated by mixers130and132with a common pilot channel1(CPICH1) signal and a common pilot channel2(CPICH2) signal respectively. The CPICH1signal and CPICH2signals may comprise fixed channelization code allocation and may be utilized to measure the signal phase and amplitude and strength of the propagation channels between the transmit antennas and the receive antennas.

FIG. 1Eis a block diagram of an exemplary two-transmit (2-Tx) and multiple-receive (M-Rx) antennas wireless communication system with pre-equalization that supports WCDMA/HSPDA, in accordance with an embodiment of the invention. Referring toFIG. 1E, the wireless communication system150may differ from the wireless communication system137inFIG. 1Din that (M−1) additional receive antennas (Rx_2108to Rx_M109) and (M−1) mixers110to111may be utilized.

The first transmit antenna, Tx_1138, and the additional or second transmit antenna, Tx_2140, may comprise suitable hardware that may be adapted to transmit a plurality of SC communication signals, sT, from a wireless transmitter device. The propagation channels that correspond to the paths taken by the SC communication signals transmitted from the transmit antennas Tx_1138and Tx_2140and received by the receive antennas Rx_1106to Rx_M109may be represented by an M×2 matrix, HM×2. The matrix HM×2may comprise propagation channels h11to hM1and h12to hM2. In this regard, h11to hM1may represent the time varying impulse responses of the RF paths taken by the portion of the transmitted SC communication signals transmitted by transmit antenna Tx_1138and received by the receive antennas Rx_1106to Rx_M109respectively. Similarly, h12to hM2may represent the time varying impulse responses of the RF paths taken by the portion of the transmitted SC communication signals transmitted by transmit antenna Tx_2140and received by the receive antennas Rx_1106to Rx_M109respectively. In some instances, a wireless transmitter device comprising a first and a second transmit antenna may be adapted to periodically transmit calibration and/or pilot signals that may be utilized by an M-Rx antenna wireless receiver device to determine estimates of h11to hM1and h12to hM2. The 2-Tx and M-Rx antennas wireless communication system150inFIG. 1Bmay represent a MIMO communication system.

The CPP118inFIG. 1Emay be adapted to determine a first baseband combined channel estimate, ĥ1, which may comprise information regarding propagation channels h11to hM1. For example, a portion of ĥ1may comprise information regarding the propagation channels between the transmit antenna Tx_1138and the receive antennas Rx_1106and Rx_2108, that is, h11and h21, while another portion of ĥ1may comprise information regarding the propagation channels between the transmit antenna Tx_1138and the receive antennas Rx_1106and Rx_M109, that is, h11and hM1.

The CPP118inFIG. 1Emay also be adapted to determine a second baseband combined channel estimate, ĥ2, which may comprise information regarding propagation channels h12to hM2. For example, a portion of ĥ2may comprise information regarding the propagation channels between the transmit antenna Tx_2140and the receive antennas Rx_1106and Rx_2108, that is, h12and h22, while another portion of ĥ2may comprise information regarding the propagation channels between the transmit antenna Tx_2140and the receive antennas Rx_1106and Rx_M109, that is, h12and hM2. The combined channel estimates ĥ1and ĥ2may be determined, that is, may be separated, in the CPP118by utilizing the orthogonal relationship between the common pilot signals CPICH1and CPICH2transmitted by the antennas Tx_1138and Tx_2140, respectively.

The SWG channel estimator122inFIG. 1Emay be adapted to process the first and second baseband combined channel estimates, ĥ1and ĥ2, determined by the CPP118and may determine a matrix ĤM×2of propagation channel estimates ĥ11to ĥM1and ĥ12to ĥM2, which correspond to estimates of the matrix HM×2of time varying impulse responses h11to hM1and h12to hM2, respectively. The SWG algorithm block124may utilize the contents of the matrix ĤM×2to determine (M−1) channel weights utilized by the mixers110to111to modify the portions of the transmitted SC communication signals received by the additional receive antennas Rx_2108to Rx_M109so that the receiver SINR is maximized, for example. The (M−1) channel weights may comprise amplitude and phase components, A1to AM-1and φ1to φM-1, for example, that result in a maximum receiver SINR. The pre-equalizer125inFIG. 1Bmay be adapted to determine a plurality of pre-equalization parameters based on the matrix ĤM×2of propagation channel estimates ĥ11to ĥM1and ĥ12to ĥM2.

FIG. 2Ais a flow diagram illustrating exemplary steps for channel estimation in a 2-Tx and M-Rx antennas wireless communication system, in accordance with an embodiment of the invention. Referring toFIG. 2A, after start step202, in step204, the SC communication signals, sT, may be transmitted from the transmit antennas Tx_1138and Tx_2140inFIG. 1E. In step206, the first and additional receive antennas, Rx_1106to Rx_M109, may receive a portion of the transmitted SC communication signals. In step208, the signals received by the additional receive antennas Rx_1106to Rx_M109may be multiplied by, for example, rotation waveforms, such as sine, square, or triangular waveforms for example, in the mixers110to111. In this regard, the rotation waveforms may have a given set of phase component values. In step210, the output of the receive antenna Rx_1106and the output of the mixers110to111associated with the additional receive antennas Rx_2108to Rx_M109may be added or combined into the received SC communication signal, sRC. The combination may occur in the adder112, for example.

In step212, the CPP118may determine the first and second baseband combined channel estimates, ĥ1and ĥ2, which comprise information regarding propagation channels h11to hM1and h12to hM2. In step214, the SWG channel estimator122in the SWBBG121may determine the matrix ĤM×2of propagation channel estimates ĥ11to ĥM1and ĥ12to ĥM2. In this regard, the propagation channel estimates ĥ11to ĥM1and ĥ12to ĥM2may be determined concurrently. In step216, the pre-equalizer125may calculate or determine the pre-equalization weight parameters or weight factors W1and W2that may be applied to the mixers130and132inFIG. 1Erespectively. The pre-equalization weights W1and W2may be transferred to a transmitter, such as a base station, to pre-equalize the signals being transmitted from the transmit antennas Tx_1138and Tx_2140.

In step218, the wireless communication system150may determine whether a closed loop operating mode that supports transmit diversity modes CL1and CL2is active. When the closed loop operating mode is active, the process may proceed to step224. In step224, the (M−1) maximum SINR channel weights that comprise amplitude and phase components, A1to AM-1and φ1to φM-1, may be generated concurrently with the diversity weight parameters supported by CL1or CL2. In this regard, the SWG algorithm block124may be utilized to generate the amplitude and phase components and the diversity weight parameters W1and W2. The channel weights may be based on the propagation channel estimates determined after the application of pre-equalization weight parameters W1and W2to the transmitter. The diversity weight parameters supported by CL1or CL2may be transferred to a transmitter, such as a base station, to combine the signals being transmitted from the transmit antennas Tx_1138and Tx_2140. After step224, the process may proceed to step222.

Returning to step218, when the closed loop operating mode is not active, the process may proceed to step220. In step220, the SWG algorithm block124may generate the (M−1) maximum SINR channel weights that comprise amplitude and phase components, A1to AM-1and φ1to φM-1. In step222, the (M−1) maximum SINR channel weights may be applied to the mixers110to111inFIG. 1E.

After steps222or224, the process may proceed to end step226where additional SC communication signals received may be phase and amplitude adjusted based on the maximum SINR channel weights applied to the mixers110to111. The channel estimation phase rotation and the maximum SINR phase/amplitude adjustment described in flow chart200may be performed continuously or may be performed periodically. In this regard,FIG. 2Billustrates an exemplary periodic phase rotation for an in-phase (I) signal received in one of the additional receive antennas, in accordance with an embodiment of the invention.

FIG. 3Ais a block diagram of an exemplary single weight baseband generator (SWBBG) that may be utilized in a 2-Tx and 2-Rx antennas system, in accordance with an embodiment of the invention. Referring toFIG. 3A, a receiver system300may correspond to a portion of the wireless communication system137inFIG. 1Dand may comprise a first receive antenna (Rx_1)302, an additional receive antenna (Rx_2)304, an adder306, a mixer308, a single weight baseband generator (SWBBG)310, and a pre-equalizer322. The SWBBG310may comprise a phase rotator start controller314, a delay block316, a single weight generator (SWG) channel estimator318, an SWG algorithm block320, and an RF phase and amplitude controller312. The SWBBG310may represent an exemplary implementation of the SWBBG121inFIG. 1D.

The first receive antenna, Rx_1302, and the additional or second receive antenna, Rx_2304, may comprise suitable hardware that may be adapted to receive at least a portion of transmitted SC communication signals in the receiver system300. For example, the receive antenna Rx_1302may receive a signal sR1while the receive antenna Rx_2304may receive a signal sR2. The mixer308may correspond to, for example, the mixer110inFIG. 1D. In some instances, the output of the mixer308may be communicated to a bandpass filter and/or a low noise amplifier (LNA) for further processing of the received signals.

The adder306may comprise suitable hardware, logic, and/or circuitry that may be adapted to add the output of the receive antenna Rx_1302and the output of the mixer308to generate a combined received SC communication signal, sRC. In some instances, bringing the output signals of the receive antenna Rx_1302and the mixer308together into a single electrical connection may provide the functionality of the adder306. The output of the adder306may be transferred to additional processing blocks for RF and baseband processing of the combined received SC communication signal, sRC.

The phase rotator and start controller314may comprise suitable logic, circuitry, and/or code that may be adapted to control portions of the operation of the RF phase and amplitude controller312and to control the delay block316. The phase rotator and start controller314may receive a signal, such as a reset signal, from, for example, the BB processor120inFIG. 1D, or from firmware operating in a processor, to indicate the start of operations that determine the propagation channel estimates and/or the channel weight to apply to the mixer308. The delay block316may comprise suitable logic, circuitry, and/or code that may be adapted to provide a time delay to compensate for the RF/modem delay. The delay may be applied in order to compensate for the interval of time that may occur between receiving the combined channel estimates, ĥ1and ĥ2, modified by the rotation waveform and the actual rotating waveform at the mixer308.

The SWG channel estimator318may comprise suitable logic, circuitry, and/or code that may be adapted to process the first and second baseband combined channel estimates, ĥ1and ĥ2, and determine the matrix Ĥ2×2of propagation channel estimates ĥ11, ĥ12, ĥ21, and ĥ22. The SWG channel estimator318may also be adapted to generate an algorithm start signal to the SWG algorithm block320to indicate that the propagation channel estimates ĥ11, ĥ12, ĥ21, and ĥ22are available for processing. In this regard, the algorithm start signal may be asserted when integration operations performed by the SWG channel estimator318have completed.

The SWG algorithm block320may comprise suitable logic, circuitry, and/or code that may be adapted to determine a channel weight to be transferred to the mixer308via the RF phase and amplitude controller312to modify the signal sR2. The channel weight to be transferred to the mixer308may refer to the phase, φ, and amplitude, A. The channel weight may be based on the propagation channel estimates ĥ11, ĥ12, ĥ21, and ĥ22and on additional information such as noise power estimates and interference propagation channel estimates, for example. The SWG algorithm block320may also be adapted to generate an algorithm end signal to indicate to the RF phase and amplitude controller312that the channel weight has been determined and that it may be applied to the mixer308.

The SWG algorithm block320may also be adapted to generate a portion of the weight parameters or weight factors W1and W2related to the closed loop diversity operation. The channel weights and closed loop diversity weights may be calculated jointly to maximize the receiver SINR, for example. The pre-equalizer322may comprise suitable logic, circuitry, and/or code that may be adapted to determine a plurality of pre-equalization parameters based on the matrix Ĥ2×2of propagation channel estimates ĥ11, ĥ12, ĥ21, and ĥ22. The pre-equalizer322may also be adapted to generate a portion of the weight parameters or weight factors W1and W2. In this regard, the pre-equalizer322may generate the weight factors W1and W2when the closed loop operating mode is not active, while the SWG algorithm block320may generated the weight factors W1and W2when the closed loop operating mode is active, for example.

The RF phase and amplitude controller312may comprise suitable logic, circuitry, and/or code that may be adapted to apply the rotation waveform ejwrtto the mixer308. When phase and amplitude components, A and φ, that correspond to the channel weight determined by the SWG algorithm block320are available, the RF phase and amplitude controller312may apply amplitude A and phase φ to the mixer308. In this regard, the RF phase and amplitude controller312may apply the rotation waveform or the amplitude and phase components in accordance with the control signals provided by the phase rotator start controller314and/or the algorithm end signal generated by the SWG algorithm block320.

The phase rotation operation performed on the sR2signal in the additional receive antenna Rx_2304may be continuous or periodic. A continuous rotation of the sR2signal may be perceived by a wireless modem as a high Doppler, and for some modem implementations this may decrease the modem's performance. When a periodic rotation operation is utilized instead, the period between consecutive phase rotations may depend on the Doppler frequency perceived by the wireless modem. For example, in a higher Doppler operation, it may be necessary to perform more frequent channel estimation while in a lower Doppler operation, channel estimation may be less frequent. The signal rotation period may also depend on the desired wireless modem performance and the accuracy of the propagation channel estimation. For example, when the Doppler frequency is 5 Hz, the period between consecutive rotations may be 1/50 sec., that is, 10 rotations or channel estimations per signal fade.

FIG. 3Bis a block diagram of an exemplary single weight baseband generator (SWBBG) that may be utilized in a 2-Tx and M-Rx antennas system, in accordance with an embodiment of the invention. Referring toFIG. 3B, a receiver system330may correspond to a portion of the wireless communication system150inFIG. 1Eand may differ from the receiver system300inFIG. 3Ain that (M−1) additional receive antennas, Rx_2304to Rx_M305, and (M−1) mixers308to309may be utilized. In this regard, the SWG channel estimator318may be adapted to process the first and second baseband combined channel estimates, ĥ1and ĥ2, and determine the matrix ĤM×2of propagation channel estimates ĥ11to ĥM1and ĥ12to ĥM2.

The SWG algorithm block320may also be adapted to determine (M−1) channel weights, that may be utilized to maximize receiver SINR, for example, to be applied to the mixers308to309to modify the portions of the transmitted SC communication signals received by the additional receive antennas Rx_2304to Rx_M305. The (M−1) channel weights may comprise amplitude and phase components, A1to AM-1and φ1to φM-1. The RF phase and amplitude controller312may also be adapted to apply rotation waveforms ejwr1tto ejwr(M-1)tor phase and amplitude components, A1to AM-1and φ1to φM-1, to the mixers308to309. In this regard, the RF phase and amplitude controller312may apply the rotation waveforms or the amplitude and phase components in accordance with the control signals provided by the phase rotator start controller314and/or the algorithm end signal generated by the SWG algorithm block320. The SWG algorithm block320may also be adapted to generate a portion of the weight parameters or weight factors W1and W2related to the closed loop diversity operation. The channel weights and closed loop diversity weights may be calculated jointly to maximize the receiver SINR, for example. The pre-equalizer322inFIG. 3Bmay also be adapted to determine a plurality of pre-equalization parameters based on the matrix ĤM×2of propagation channel estimates ĥ11to ĥM1and ĥ12to ĥM2.

FIG. 3Cis a block diagram of an exemplary RF phase and amplitude controller, in accordance with an embodiment of the invention. Referring toFIG. 3C, the RF phase and amplitude controller312may comprise a switch340, a plurality of rotation waveform sources342, and a plurality of SWG algorithm weights344. The switch340may comprise suitable hardware, logic, and/or circuitry that may be adapted to select between the rotation waveforms ejwr1tto ejwr(M-1)tand the SWG algorithm determined weights A1ejφ1to AM-1ejφM-1. The rotation waveform sources342may comprise suitable hardware, logic and/or circuitry that may be adapted to generate the signal ejwrkt, where wrk=2πfrkand frkis the rotation frequency that preserves the orthogonality of the received signals at the receive antennas Rx_2302to Rx_M305inFIG. 3B, for example. The rotation frequency that preserves the signal orthogonality at the receiving antennas may be selected as wrk=kwrwhere k=1, 2, . . . , M−1. Other rotation waveforms such as triangular or square waveforms may be utilized with the same frequency relationships. Moreover, waveforms representing different orthogonal codes of the same frequency may also be utilized, similar to the CDMA orthogonal codes with the same spreading. In this embodiment, the signal ejwrktmay be utilized as an exemplary waveform. The plurality of SWG algorithm weights344may comprise suitable hardware, logic, and/or circuitry that may be adapted to generate the signals A1ejφ1to AM-1ejφM-1from the amplitude and phase components, A1to AM-1and φ1to φM-1, respectively.

In operation, the RF phase and amplitude controller312may apply the signals ejwr1tto ejwr(M-1)tto the mixers308to309inFIG. 3Bbased on control information provided by the phase rotator start controller314. The switch340may select the rotation waveform sources342based on the control information provided by the phase rotator start controller314. Once the channel weights are determined by the SWG algorithm block320and the phase and amplitude components have been transferred to the RF phase and amplitude controller312, the algorithm end signal may be utilized to change the selection of the switch340. In this regard, the switch340may be utilized to select and apply the signals A1ejφ1to AM-1ejφM-1to the mixers308to309inFIG. 3B.

FIG. 4is a flow diagram illustrating exemplary steps in the operation of the single weight baseband generator (SWBBG) that may be utilized in a 2-Tx and M-Rx antennas system, in accordance with an embodiment of the invention. Referring toFIG. 4, after start step402, in step404, the phase rotator start controller314inFIG. 3Bmay receive the reset signal to initiate operations for determining propagation channel estimates and channel weights in the SWBBG310. The phase rotator start controller314may generate control signals to the delay block316and to the RF phase and amplitude controller312. The control signals to the delay block316may be utilized to determine a delay time to be applied by the delay block316. The control signals to the RF phase and amplitude controller312may be utilized to determine when to apply the rotation waveforms or the channel weights determined by the SWG algorithm block124to the mixers308to309inFIG. 3B, for example.

In step406, the RF phase and amplitude controller312may apply the signals ejwr1tto ejwr(M-1)tto the mixers308to309inFIG. 3B. In step408, the delay block316may apply a time delay signal to the SWG channel estimator318to reflect the interval of time that may occur between receiving the combined channel estimates, {circumflex over (h)}1and {circumflex over (h)}2, modified by the rotation waveform and the actual rotating waveform at the mixer308. For example, the time delay signal may be utilized as an enable signal to the SWG channel estimator318, where the assertion of the time delay signal initiates operations for determining propagation channel estimates. In step410, the SWG channel estimator318may process the first and second baseband combined channel estimates, ĥ1and ĥ2, and may determine the matrix ĤM×2of propagation channel estimates ĥ11to ĥM1and ĥ12to ĥM2. The SWG channel estimator318may transfer the propagation channel estimates ĥ11to ĥM1and ĥ12to ĥM2to the SWG algorithm block320. In step412, the pre-equalizer322may calculate or generate the pre-equalization weight parameters or weight factors W1and W2. The pre-equalization weight parameters may be transferred to a wireless transmitter, such as a base station.

In step414, the receiver system330inFIG. 3Bmay determine whether a closed loop operating mode that supports transmit diversity modes CL1and CL2is active. When the closed loop operating mode is active, the process may proceed to step418. In step418, the (M−1) maximum SINR channel weights that comprise amplitude and phase components, A1to AM-1and φ1to φM-1, may be generated by the SWG algorithm block320concurrently with the diversity weight parameters W1and W2supported by CL1or CL2. The channel weights may be based on the propagation channel estimates determined after the application of pre-equalization weight parameters W1and W2to the transmitter. The diversity weight parameters that support CL1or CL2may be transferred to a transmitter, such as a base station, to apply the weights to the signals being transmitted. After step418, the process may proceed to step420.

Returning to step414, when the closed loop operating mode is not active, the process may proceed to step416. In step416, the SWG algorithm block320may generate the (M−1) maximum SINR channel weights that comprise amplitude and phase components, A1to AM-1and φ1to φM-1, based on the propagation channel estimates ĥ11to ĥM1and ĥ12to ĥM2and/or noise power estimates and interference channel estimates, for example. The SWG algorithm block320may transfer the channel weights to the RF phase and amplitude controller312. The SWG algorithm block320may generate the algorithm end signal to indicate to the RF phase and amplitude controller312that the channel weights are available to be applied to the mixers308to309. In step420, RF phase and amplitude controller312may apply the maximum SINR weights with phase and amplitude components, A1to AM-1and φ1to φM-1, to the mixers308to309inFIG. 3B, in accordance with the control signals provided by the phase rotator start controller314and/or the SWG algorithm block320.

In step422, the receiver system330inFIG. 3Bmay determine whether the phase rotation operation on the received SC communication signals is periodic. When the phase rotation operation is not periodic but continuous, the process may proceed to step408where a new delay may be applied to the SWG channel estimator318. In instances when the phase rotation operation is periodic, the process may proceed to step424where the receiver system330may wait until the next phase rotation operation is initiated by the reset signal. In this regard, the process may return to step404upon assertion of the reset signal on the phase rotator start controller314.

FIG. 5is a flow diagram illustrating exemplary steps for determining channel weights in additional receive antennas utilizing signal-to-noise ratio (SNR) or signal-to-interference-and-noise ratio (SINR), in accordance with an embodiment of the invention. Referring toFIG. 5, after start step502, in step504, the SWG algorithm block320may determine whether the signals received in the receive antennas are noise limited. The SWG algorithm block320may receive noise statistics and/or other noise information from either the CPP118and/or from the BB processor120. When the received signals are noise limited, the flow diagram control may proceed to step508. In step508, the SWG algorithm block320may generate models for the received signals. For example, the models for a 1-Tx and 2-Rx antennas system may be represented by the following expressions:
r1=h1s+n1,
r2=Aejθh2s+Aejθn2, and
y=r1+r2=s(h1+Aejθh2)+n1+Aejθn2,
where r1may represent a model of the signal received in a first receive antenna, r2may represent a model of the signal received in the second receive antenna, s may represent the transmitted signal, and n1may represent a noise component at the first receive antenna, whose time varying impulse response is represented by h1. The parameter n2may represent a noise component at the second receive antenna, whose time varying impulse response is represented by h2, θ may represent the phase factor between the signal received in the first and second receive antennas, and A may represent an amplitude factor. The parameter y may represent the sum of the received signal models and may comprise a combined signal component s(h1+Aejθh2) and a combined noise component n1+Aejθn2.

In step510, the received signal models may be utilized to determine a signal strength parameter. In this regard, the signal-to-noise ratio (SNR) may correspond to the signal strength parameter to be determined. For example, for a 1-Tx and 2-Rx antennas system, the SNR may be determined by maximizing the following expression for various phase, θ, and amplitude, A, factors:

SNR=h1+A⁢⁢ⅇj⁢⁢ϑ⁢h22E⁢⁢n12+E⁢⁢A⁢⁢ⅇj⁢⁢ϑ⁢n22⁢=h1+A⁢⁢ⅇj⁢⁢ϑ⁢h22σ2⁡(1+A2).
The SNR numerator may correspond to the y parameter's combined signal component while the SNR denominator may correspond to the y parameter's combined noise component. The phase factor, θ, may be selected, for example, from a 360-degrees phase rotation while the amplitude factor, A, may be selected, for example, from an set amplitude range. In one embodiment of the invention, the phase factor may be varied in a plurality of phase factor steps over the 360-degrees phase rotation to find the maximum SNR value. In another embodiment of the invention, the phase factor may be varied in a plurality of phase factors steps over the 360-degrees phase rotation and the amplitude factor may be varied in a plurality of amplitude factor values over the amplitude range to find the maximum SNR value.

In step520, after determining the maximum SNR in step510, the SWG algorithm block320may utilize the amplitude factor and phase factor that corresponds to the maximum SNR to determine the amplitude and phase to be provided to the RF amplitude and phase controller312in step520. For example, in one embodiment of the invention, the amplitude and/or phase factors that correspond to the maximum SNR may be utilized as the amplitude and phase to be transferred to the RF amplitude and phase controller312. After application of the appropriate amplitude and phase by the RF amplitude and phase controller312to the receive antennas, the flow diagram control may proceed to end step522until a next phase and amplitude determination is necessary.

Returning to step504, when received signals are not noise limited, the flow control may proceed to step506where a determination may be made as to whether multiple interfering signals may be present and may need to be considered during channel weight determination. When a single interferer is considered, the flow diagram control may proceed to step512. In step512the SWG algorithm block320may generate models for the received signals. For example, the models for a 1-Tx and 2-Rx antennas system may be represented by the following expressions:
r1=h1s+hI1sI+n1,
r2=Aejθ(h2s+hI2sI+n2), and
y=r1+r2=s(h1+Aejθh2)+n1+sI(hI1+AejθhI2)+Aejθn2,
where r1may represent a model of the signal received in a first receive antenna, r2may represent a model of the signal received in the second receive antenna, s may represent the transmitted signal, sImay represent the interference signal, and n1may represent a noise component at the first receive antenna whose time varying impulse response is h1. The parameter n2may represent a noise component at the second receive antenna whose time varying impulse response is h2, θ may represent the phase factor between the signal received in the first and second receive antennas, and A may represent an amplitude factor. Moreover, the time varying impulse response hI1may correspond to the propagation channel between the interference signal source and the first receive antenna and the time varying impulse response hI2may correspond to the propagation channel between the interference signal source and the second receive antenna. The parameter y may represent the sum of the received signal models and may comprise a combined signal component s(h1+Aejθh2) and a combined noise plus interference component n1+sI(hI1+AejθhI2)+Aejθn2.

In step514, the received signal models may be utilized to determine a signal strength parameter. In this regard, the signal-to-interference-and-noise ratio (SINR) may correspond to the signal strength parameter to be determined. For example, for a 1-Tx and 2-Rx antennas system, the SINR may be determined by maximizing the following expression for various phase, θ, and amplitude, A, factors:

SINR=h1+A⁢⁢ⅇj⁢⁢ϑ⁢h22E⁢⁢n12+E⁢⁢A⁢⁢ⅇj⁢⁢ϑ⁢n22+hl⁢⁢1+A⁢⁢ⅇj⁢⁢ϑ⁢hl⁢⁢22=h1+A⁢⁢ⅇj⁢⁢ϑ⁢h22σ2⁡(1+A2)+hl⁢⁢1+A⁢⁢ⅇj⁢⁢ϑ⁢hl⁢⁢22.
where σ2is the noise power. The above SINR equations may be easily extended to the SC MIMO case. The transmit antennas may include CL1or CL2transmit diversity weights. The joint transmit-received solution may be formed in that case that may include the transmit CL weights and the additional transmit antenna channel components in the SINR numerator. The SINR numerator may correspond to the y parameter's combined signal component while the SINR denominator may correspond to the y parameter's combined noise plus interference component. The phase factor, θ, may be selected, for example, from a 360-degrees phase rotation while the amplitude factor, A, may be selected, for example, from an set amplitude range. In one embodiment of the invention, the phase factor may be varied in a plurality of phase factor steps over the 360-degrees phase rotation to find the maximum SNR value. In another embodiment of the invention, the phase factor may be varied in a plurality of phase factors steps over the 360-degrees phase rotation and the amplitude factor may be varied in a plurality of amplitude factor values over a range of amplitudes to find the maximum SINR value.

After determining the SINR in step514, the SWG algorithm block320may determine the amplitude and phase to be provided to the RF amplitude and phase controller312in step520. After application of the appropriate amplitude and phase by the RF amplitude and phase controller312, the flow diagram control may proceed to end step522until a next phase and amplitude determination is necessary.

After determining the SINR in step518, the SWG algorithm block320may determine the amplitude and phase to be provided to the RF amplitude and phase controller312in step520. After application of the appropriate amplitude and phase by the RF amplitude and phase controller312, the flow diagram control may proceed to end step522until a next phase and amplitude determination is necessary.

The operations to maximize the signal strength described for steps510,514, and518may be based on a search algorithm. In an exemplary embodiment of the invention, a search algorithm may be utilized to search over 360-degrees phase rotation in 45 or 90 degree phase factor steps and over a 0-5 amplitude range in 0.25 amplitude values or steps, for example. For a 1-Tx and 2-Rx antenna system, with 90-degree phase factor steps, a phase only search algorithm may calculate 4 SNR or SINR values, for example. For a 2-Tx and 2-Rx antenna system with STTD transmit mode, with 90-degree phase factor steps, a phase only search algorithm may calculate 4 SNR or SINR values. For a 2-Tx and 2-Rx antenna system with the CL1diversity mode, with 90-degree phase factor steps at both receiver and transmitter, a phase only search algorithm may calculate 4×4=16 SNR or SINR values. For a 2-Tx and 2-Rx antenna system with the CL2diversity mode, with 90-degree phase factor steps at the receiver and 45-degree phase factor steps and two power scaling weight levels at the transmitter, a phase only search algorithm may calculate 4×8×2=64 SNR or SINR values, for example. The maximum value generated by the algorithm may be the output of the search algorithm.

In another embodiment of the invention, a closed-form mathematical expression may also be utilized to maximize the SNR and/or the SINR. Utilizing an algorithm or closed-form expression that maximizes the SINR or SNR may provide a good compromise between implementation complexity and performance gains. Notwithstanding, the invention is not limited in this regard, and other channel weight algorithms may also be utilized.

Determining channel weights and/or pre-equalization parameters may be performed by monitoring the baseband combined channel estimates, for example. In this regard, a SWBBG may be utilized for monitoring the baseband combined channel estimates generated by, for example, a CPP. U.S. application Ser. No. 11/174,252 provides a detailed description of monitoring baseband combined channel estimates and is hereby incorporated herein by reference in its entirety.

Another embodiment of the invention may provide a machine-readable storage, having stored thereon, a computer program having at least one code section executable by a machine, thereby causing the machine to perform the steps as described above for pre-equalization in a single weight, single channel MIMO system.

The approach described herein for determining pre-equalization parameters in a single channel MIMO system may provide a good compromise between implementation complexity and performance gains in the design and operation of MIMO systems.