High dynamic range analog front-end receiver for long range LIDAR

A system and method for operating a high dynamic range analog front-end receiver for long range LIDAR with a transimpedance amplifier (TIA) include a clipping circuit to prevent saturation of the TIA. The output of the clipping circuit is connected via a diode or transistor to the input of the TIA and regulated such that the input voltage of the TIA remains close to or is only slightly above the saturation threshold voltage of the TIA. The regulation of the input voltage of the TIA can be improved by connecting a limiting resistor in series with the diode or transistor. A second clipping circuit capable of dissipating higher input currents and thus higher voltages may be connected in parallel with the first clipping circuit. A resistive element may be placed between the first and second clipping circuits to further limit the input current to the TIA.

FIELD OF THE DISCLOSURE

The present disclosure relates to the field of remote sensing that uses light from a pulsed light source to measure ranges or distances from objects. More particularly, the present disclosure relates to shunt-feedback transimpedance amplifiers of the type used in remote sensing applications.

BACKGROUND

Remote sensing using light pulses emitted for example by lasers and retroreflected by distant objects is sometimes also referred to as LIDAR (light detection and ranging). A LIDAR receiver, hereinafter also referred to as a front-end receiver, includes an optical receiver having a photodiode (PD) or an avalanche photodiode (APD) as a receiving element, and a transimpedance amplifier (TIA), for example a shunt-feedback amplifier, which converts the photocurrent from the receiving photodiode into a voltage.

In the following, the terms photodiode (PD) and avalanche photodiode (APD) will be used interchangeably, unless otherwise stated. A photodiode is typically a p-n junction or PIN structure. When a photon of sufficient energy strikes the diode, it creates an electron-hole pair. This mechanism is also known as the inner photoelectric effect. If the absorption occurs in the junction's depletion region, or one diffusion length away from it, these carriers are swept from the junction by the built-in electric field of the depletion region. Thus holes move toward the anode, and electrons toward the cathode, and a photocurrent is produced. When used in zero bias or photovoltaic mode, the flow of photocurrent out of the device is restricted and a voltage builds up. When used in photoconductive mode, the diode is often reverse-biased (with the cathode driven positive with respect to the anode). This reduces the response time because the additional reverse bias increases the width of the depletion layer, which decreases the junction's capacitance. The reverse bias also increases the dark current without much change in the photocurrent. For a given spectral distribution, the photocurrent is linearly proportional to the illuminance (and to the irradiance).

APDs can be thought of as photodetectors that provide a built-in first stage of gain through avalanche multiplication. From a functional standpoint, they can be regarded as the semiconductor equivalent to photomultipliers. Due to their high sensitivity, a typical application for APD's is in laser rangefinders and long range fiber-optic telecommunication.

In some applications, an APD generates a current pulse proportional to the received electromagnetic power and a TIA converts the current pulse into a voltage pulse and also provides a high gain in order to detect weaker signals from distant objects. For closer objects, the magnitude of the current pulse at the input of the TIA can reach the limits of linear operation of the TIA. In such cases, the TIA becomes saturated. In shunt-feedback amplifier topology, saturation causes the output voltage pulse to widen by a certain amount, which is referred to as pulse-width distortion. Upon overloading, such transimpedance amplifiers have very long relaxation times until again the TIA can return to linear operation.

SUMMARY OF THE DISCLOSURE

A system and method for operating a high dynamic range analog front-end receiver for long range LIDAR with a transimpedance amplifier (TIA) include a clipping circuit to prevent saturation of the TIA. The output of the clipping circuit is connected via a diode or transistor to the input of the TIA and regulated such that the input voltage of the TIA remains close to or is only slightly above the saturation threshold voltage of the TIA. The regulation of the input voltage of the TIA can be improved by connecting a limiting resistor in series with the diode or transistor. A second clipping circuit capable of dissipating higher input currents and thus higher voltages may be connected in parallel with the first clipping circuit. A resistive element may be placed between the first and second clipping circuits to further limit the input current to the TIA.

It would be beneficial to provide a front-end receiver with a transimpedance amplifier (TIA), which enables detection of the photocurrent from the receiving photodiode or APD with no or only minimal pulse-width distortion caused by large photocurrents. A system and method for adaptive clipping an input voltage of a transimpedance amplifier (TIA) so as to prevent the TIA from going too deep into saturation include a first adaptive clipping circuit receiving a feedback voltage from an input of the TIA and a clip voltage. The output of the first adaptive clipping circuit is connected by way of a diode or transistor to the input of the TIA and regulated such that the input voltage of the saturated TIA is kept as close as possible to the saturation threshold voltage of the TIA and never exceeds the saturation voltage of the TIA by more than a turn-on voltage of the diode or transistor, preferably by no more than half the turn-on voltage of the diode or transistor. The regulation of the input voltage of the TIA can be improved by connecting a limiting resistor in series with the diode or transistor. A second clipping circuit capable of dissipating higher input currents and thus higher voltages than the first adaptive clipping circuit may be connected in parallel with the first adaptive clipping circuit.

In certain embodiments, a front-end receiver is provided that includes a transimpedance amplifier (TIA) configured to convert an input current applied to an input port into an output voltage, and a first adaptive clipping circuit, coupled to the input port by way of a diode path and configured to, in response to a clip voltage applied to the first adaptive clipping circuit, limiting a maximum value of an input voltage of the TIA to an externally applied clip voltage so that the TIA is not overly saturated. In some cases, the first adaptive clipping circuit can limit the maximum value of the input voltage of the TIA such that the TIA operates as close as possible to the saturation threshold voltage of the TIA.

In certain embodiments, a method is provided for operating a transimpedance amplifier (TIA) of a front-end receiver so that the TIA is not overly saturated or operating the TIA as close as possible to the saturation threshold voltage of the TIA, wherein the method includes supplying to a first adaptive clipping circuit at a first input terminal a clip voltage selected to be approximately equal to a saturation threshold voltage of the TIA, supplying to the first adaptive clipping circuit at a second input terminal a feedback voltage derived from an input voltage of the TIA, coupling an output voltage of the first adaptive clipping circuit to an input port of the TIA by way of a diode path comprising a rectifying element having a turn-on voltage; and regulating an output voltage of the first adaptive clipping circuit such that the feedback voltage is equal to the clip voltage. In this way, a maximum value of the input voltage of the TIA is limited to a value that is higher than a saturation threshold voltage of the TIA by no more than the turn-on voltage of the rectifying element, preferably half the turn-on voltage of the rectifying element, to ensure that the TIA is not overly saturated.

DETAILED DESCRIPTION

The pulsed time-of-flight (TOF) laser ranging method is based on the measurement of a transit time (ΔT) of a short laser pulse (width for example about 3 ns) to an optically visible target and back to the front-end receiver. The measured transit time can be converted to a distance (R) between the target and the receiver.

LIDAR based on pulsed TOF measurements is particularly appealing in environmental perception systems where a high measurement speed (>1000 results/s) is needed, where the dynamics of the received echo can be very wide (>1:1000) and where an accurate distance measurement (<1 cm) is needed even with a single transmitted pulse to distances of up to tens of meters to non-cooperative targets. Examples of this kind are anti-collision systems and scanners in traffic applications. For example, in automotive applications the dynamic range may exceed 1:100,000. In the case of a mirror-like reflection or a reflection from a close object, a very high input signal may appear at the input of the receiver channel. The input signal may approach 10-100 mA or even 0.5-1 A, which may saturate the receiver channel.

Of interest is also a measurement of the pulse-width of the received pulses in LIDAR applications, since the pulse-width difference between the transmitted and the received pulses in LIDAR applications also carries information about the weather (moisture, fog, etc.). The pulse-width can increase, for example, due to multiple scattering events under foggy conditions, thus distinguishing this pulse-widening from the TIA's own pulse-widening is valuable. Limiting the pulse-width also enables increasing the number of pulses in a pulse train in a given time period; which is helpful when using averaging techniques to improve SNR.

Accordingly, it is advantageous and in many situations essential to prevent deep saturation of a TIA caused by large currents that induce a voltage highly exceeding the saturation threshold voltage of the TIA, and to keep both the amount and the variation of pulse-width distortion small over a wide range of input current amplitudes.

FIG. 1Ashows an exemplary current pulse of photocurrent Iingenerated in response to a laser pulse. The photocurrent pulse Iinproduces an input voltage Vinat the input of the TIA which would mirror the shape of the input current if the TIA were to operate in its linear operating range. However, the TIA is characterized by an input saturation voltage threshold Vsat,thabove which the TIA becomes saturated. With increasing photocurrent Iin, the TIA input voltage Vinenters an “oversaturation” region, from where it decays at the end of the pulse with a time constant determined by the resistor-capacitor (RC) discharge over the feedback resistor Rf. The value of the capacitance C is determined by, for example, the capacitance to the APD and other parasitic capacitances of the system. As evident fromFIG. 1B, the width of the pulse is substantially broadened before the input voltage Vinof the TIA returns to the linear operating region (<Vsat,th). This increases the detected pulse-width of the output voltage Vout,satby an amount indicated inFIG. 1Cas “Pulse-width distortion”, which therefore does no longer resemble the shape of the received photocurrent pulse IininFIG. 1A. It would therefore be desirable to limit the input voltage Vinat the input of the TIA to a value that is close to the input saturation voltage threshold Vsat,th.

According to some embodiments of the disclosure, a schematic circuit diagram illustrated inFIG. 2shows a transimpedance amplifier (TIA)200receiving at an input an input voltage Vingenerated by a photocurrent Iinfrom an avalanche photodiode (APD) in response to a received light input signal from an unillustrated light source, for example a laser. The DC and low-frequency gain of a transimpedance amplifier can be determined by the equations (1) and (2), in cases where the gain of the amplifier is sufficiently large:

so that the gain is

The high gain of the op-amp of the TIA keeps the photodiode current equal to the feedback current through resistor Rf.

According to some embodiments of the disclosure, pulse-width distortion or pulse-widening may be reduced by limiting (clipping) the TIA input voltage Vinwith a diode, wherein the clipping path is inactive during linear operation below the saturation voltage Vsat,th, without significantly increasing noise. For example, the input voltage Vinof a TIA may be limited by bridging the input of the TIA with Schottky or Zener diodes (not shown). Disadvantageously, however, due to current leakage even when these diodes operate below their turn-on voltage (which is approximately 0.7 for Si junctions and approximately 0.3 V for Ge junctions), the clip voltage cannot be selected to be very close to the saturation threshold Vsat,th.

The forward current through a diode is given by the following equation

wherein

VD=is the applied voltage across the diode

T=is the absolute temperature in Kelvin

q=is the electron charge (1.6*10−19Coulomb)

ID=is the actual current through the diode

IS=is the diffusion current (a device dependent constant).

The so called thermal diode voltage, VT, is kT/q=26 mV at room temperature.

It is evident from (eq. 3) that IDincreases exponentially with VDand is non-zero even below the bandgap of approximately 0.7 V for a Si diode. Depending on the specific application, the forward leakage current through diode204inFIG. 2may be considered as being negligible when the forward bias across diode204is reduced from the bandgap (or theoretical turn-on voltage) of the diode by a multiple of the thermal voltage VT, for example by approximately eight times the thermal voltage VT, i.e. by approximately 8*26 mV or approximately 200 mV.

According to some embodiments of the disclosure, as illustrated inFIG. 2, the clip voltage may be adjusted by connecting an output of a buffer amplifier202which has a first input controlled by an external control voltage Vctrlto a reverse-biased diode204which is in turn connected to the input of the TIA. A second (unillustrated) input may be connected to a common mode voltage, for example to ground. The buffer amplifier may be implemented as a voltage follower. The diode may be, for example, a Si diode with a turn-on voltage of approximately 0.7 V. In order to clip the input voltage Vinat a value of, for example, ΔV=350 mV above an exemplary saturation threshold Vsat,thof 2 V, the control voltage Vctrlwould have to be 1.85 V. During linear operation of the TIA, when the input voltage Vinof the TIA is below the saturation threshold Vsat,thof 2 V, the diode204inFIG. 2is essentially non-conducting because the voltage on the diode is 0.25V and the input voltage Vinis determined solely by the feedback resistor Rf. When Vinexceeds 2.35 V, the diode becomes forward-biased and excess current Iinwill be dissipated by diode204. It should be noted that the value of ΔV=350 mV is merely chosen as a representative example and may be, for example, equal to approximately the turn-on voltage of the diode (0.7 V for a Si diode and 0.3 V for a Ge diode), preferably one-half of the turn-on voltage of the diode, or a multiple of the thermal voltage VTof the diode, for example 200 mV which is approximately eight times VT, as long as Vinis still within the normal operating range of the TIA.

Because the diode204must be able to dissipate relatively large currents above the saturation threshold Vsat,thwithout causing an excessive increase in Vin, the forward resistance of diode must be small, which would require a large diode. However, larger diodes have also a significantly larger capacitance than smaller diodes which would in turn increase the RC time constant that controls the RC discharge time inFIG. 1B, decrease the bandwidth and increase TIA noise.

The effect of the diode204in conjunction with the buffer amplifier202on the input voltage Vinand the output voltage Voutand the concomitant pulse-width is schematically shown inFIG. 3, whereinFIG. 3Ashows three different exemplary input current levels Iin_1, Iin_2and Iin_3with Iin_1>Iin_2>Iin_3. The photocurrent pulses Iin_1, Iin_2and Iin_3produce corresponding voltages Vin_1, Vin_2and Vin_3at the input of the TIA, with Vindepending on input current Iinas Vin=Vctrl+0.7 V+Rdiode,int*Iin, wherein Vctrlis the control voltage applied to buffer amplifier202, 0.7 V is the turn-on voltage of a Si diode and Rdiode,intis the internal forward resistance of the diode204. The discharge period assisted by the diode204always terminates when the current through the diode204becomes negligible, i.e. at Vin=Vctrl+Vdiode(O)=Vctrl+0.7 V. The voltage Vctrl+Vdiode(O) is in this example selected to be higher by ΔV˜450 mV than the saturation threshold voltage Vsat,thof the TIA, because the diode should be off during normal operation when Vin≤Vsat,th, i.e. the forward leakage current through the diode should be negligible for the particular application. When the input voltage Vindrops below Vctrl+Vdiode(0), the input current can from this point on only be discharged through the feedback resistor Rf. However, the discharge through Rfalways starts from Vctrl+Vdiode(0) regardless of Iinwhich implies that the pulse-widening will be the same for all illustrated currents Iin_1, Iin_2and Iin_3that cause the voltages Vin_1, Vin_2and Vin_3to exceed Vctrl+Vdiode(0).

VCC,APDindicates the (positive) supply voltage of the APD. VCCindicates the supply voltage of the ESD inFIG. 11and (not shown explicitly) of the buffer amplifiers and the TIA. VCC,APDis typically larger than VCC.

A comparison ofFIG. 3andFIG. 1shows the benefits of discharging large currents through the diode204which reduces the pulse-width distortion. The discharge time required to reduce ΔV=Vctrl+Vdiode(0)−Vsat,thto approximately zero is the main contributor of the pulse-width distortion. It is desirable to make ΔV as small as possible. Although pulse-width distortion could theoretically be reduced by making ΔV smaller, current leakage through diode204could adversely affect normal operation of the TIA, i.e. when Vctrlis selected such that Vctrl+Vdiode(0) is at most only slightly above Vsat,th(for example by no more than the turn-on voltage of the diode, preferably by no more than half the turn-on voltage of the diode, or by a multiple of the thermal voltage VTOf the Si diode, for example 8*VT=200 mV, as long as Vinis still within the normal operating range of the TIA).

Although pulse-width distortion is reduced fromFIG. 1with diode-assisted clipping illustrated by the results inFIG. 3, the time constant of the discharge of ΔV through Rfcan be on the order of tens of nanoseconds which can still be unacceptably long for some applications. The pulse-width distortion can be further reduced with adaptive clipping which will now be described.

According to some embodiments of the disclosure, illustrated inFIG. 4, the input voltage Vinmay be clipped adaptively by adjusting the voltage across a diode404commensurate with an externally applied clip voltage Vclip. The adaptive clipping circuit ofFIG. 4differs from the circuit previously described with reference toFIG. 2in that the inverting input of amplifier402is not tied to a fixed potential as inFIG. 2, such as a common mode voltage or to ground, but instead receives the voltage Vinfrom the input of the TIA.

The adaptive clipping circuit ofFIG. 4operates as follows: the output voltage of op-amp402is linearly proportional to the voltage difference between the positive input terminal (+) and the negative input terminal (−) by a gain factor. An ideal op-amp has infinite gain, infinite input resistance, and zero output resistance. A consequence of the assumption of infinite gain is that, when the output voltage is within the linear region of the op-amp, the voltage at the positive input terminal (+) is always equal to the voltage at the negative input terminal (−). Without the diode404, the circuit ofFIG. 4would be a voltage follower, in which the feedback loop formed by the amplifier402and the diode404always drives noise412such that Vin=Vclip. When the diode404having a forward voltage Vfis inserted between the output of op-amp402and the negative input terminal (−) and when a clip voltage Vclipapplied to the positive input terminal (+), the feedback loop formed by the amplifier402and the diode404is unable to equalize Vinto Vclipwhen Vin≤Vclip. For example, as described above, when Vin˜2V and Vclip˜2.2 V, the feedback loop has no effect, TIA is operating in linear region, and Vinis determined by the output common mode voltage of the amplifier200plus the input current Iintimes Rf.

During a high current pulse, Vinis charged up to Vclip, at which point, the feedback loop formed by the amplifier402and the diode404is activated and modulates the cathode of diode404such that the current flowing through diode404prevents Vinfrom increasing further to keep it from going above the desired clip voltage Vclip. With the exemplary adaptive clipping circuit ofFIG. 4, ΔV=Vclip−Vsat,thcan be set to a value smaller than the actual, temperature-dependent soft turn-on voltage of the diode404, without causing a high leakage current through diode404during normal operation, resulting in very small pulse-width distortion and a shorter time to return to normal operation. This was not possible with the diode circuit ofFIG. 2because a small value of ΔV=Vctrl+Vdiode(0)−Vsat,thwould cause a large voltage drop across the diode204and a commensurate large leakage current during normal operation.

According to some embodiments of the disclosure, as illustrated inFIG. 5, the functionality provided by the diode404, i.e. preventing the feedback loop from increasing Vin, can also be provided by a transistor504, for example an open-emitter PNP emitter follower or an open-source PMOS (p-type metal-oxide-semiconductor field-effect transistor) source follower, because both of these open-emitter/open-source stages can sink current and discharge/reduce Vin, but are unable to source current to charge/increase Vin. The feedback loop pulls the voltage down when Vinexceeds, for example, the exemplary value of 2.2 V. As mentioned above, the adaptive clipping circuits shown inFIGS. 4 and 5should be sized appropriately to handle large currents; however, the size-related capacitance limits the response time. Stated differently, the loop bandwidth of the adaptive clipping circuit has to be sufficiently high so that the loop can respond quickly to the rising edge of the input current pulse and prevent Vinfrom reaching hazardous levels. However, since the loop is off during normal operation, the loop has to be established first (wake up phase) and can only thereafter respond by lowering Vinto the desired level close to Vsat,th.

Another problem with achieving higher “maximum tolerable currents” is that higher amplitude current pulses charge Vinto Vsat,thfaster, thus requiring faster response times.

FIGS. 6A-6Cillustrate schematically the effect of different, in particularly very high input current levels Iin_1, Iin_2, Iin_3on the shape of the input voltages Vin_1, Vin_2, Vin_3at the input of the TIA and the output voltages Vout_1, Vout_2, Vout_3generated by the TIA. As discussed above, adaptive clipping limits the voltage Vinto Vclipclose to the saturation threshold Vsat,thindependent of the input current Iin. Since the voltage at which Vinis clipped is close to the saturation threshold Vsat,th, the remaining excess voltage ΔV=Vclip−Vsat,this discharged through Rfwith a time constant t=Rf*C. The pulse-width of the output pulse from the TIA can thus be reduced with adaptive clipping to close to the width of the input current for not excessively high input currents, such as exemplary input current Iin_3inFIG. 6A.

However, the dynamic response of the clipping circuits ofFIGS. 4 and 5at elevated input current levels, such as input currents Iin_1and Iin_2inFIG. 6Acan result in pulse distortion which manifests itself in the narrowing of Vin_2compared to Vin_3inFIG. 6Band the phase reversal of Vin_1inFIG. 6B, creating a negative voltage glitch at Vin_1at the falling edge of the current pulse Iin_1. If the negative voltage glitch is strong enough, Vin_1may decrease below the common mode voltage and produce at the output of the TIA a pulse with opposite-polarity, as illustrated inFIG. 6C.

According to some embodiments of the disclosure, as illustrated inFIG. 7, adaptive clipping may be combined with diode clipping, hereinafter referred to as assisted adaptive clipping. Assisted adaptive clipping adds a faster alternative current path to the adaptive clipping circuit so that the loop can respond quickly to the rising edge of the input current pulse and prevent Vinfrom reaching undesirable or hazardous levels.

With assisted adaptive clipping the overall clipping circuit is comprises a first diode clipping path701in accordance withFIG. 2that responds first and clips Vinat a relatively safe voltage with is somewhat higher than Vclipand is controlled by the control voltage Vctrl, because diode clipping has a shorter response time determined solely by the transit time of the diode in the first diode clipping path701.

Thereafter, the adaptive clipping feedback loop702illustrated inFIG. 7Aand designed in accordance withFIG. 5and the adaptive clipping feedback loop704illustrated inFIG. 7Band designed in accordance withFIG. 4, respectively, wake up and begins to dissipate the current because it clips at a lower voltage than the non-adaptive clipping circuit, as mentioned above. In other words, the non-adaptive first diode clipping path701responds first and dissipates the first surge of a high input current. Thereafter, the adaptive clipping feedback loop702or704, respectively, wakes-up, and starts dissipating also the current passing through non-adaptive clipping circuit, thereby further reducing the input voltage (because the adaptive clipping feedback loop702,704clips at a lower voltage value above Vsat,th). During this transition period, the current through the adaptive clipping feedback loop702,704increases, whereas the current through non-adaptive clipping path701decreases. Vinis thus discharged via the respective adaptive clipping feedback loops702and704down to approximately Vclip.

Since very high currents are handled initially, i.e. before the adaptive clipping feedback loop702,704is activated, by diode clipping with the first diode clipping path701, the devices of the adaptive clipping feedback loops702,704can be made smaller, thus increasing the bandwidth of the feedback loop.

According to some embodiments of the disclosure adaptive clipping may be made more robust by limiting the adaptive clipping current flowing through the transistor504in circuit802ofFIG. 8Aor similarly through the diode404in circuit804ofFIG. 8B. To this end, a current-limiting resistor Rlimmay be inserted between the input terminal of the TIA200and the open-emitter PNP emitter follower or open-source PMOS source follower504(at812) in adaptive clipping circuit802shown inFIG. 8Aor commensurately between the input terminal of the TIA200and the anode (at812) of the diode in adaptive clipping circuit804shown inFIG. 8B. Due to the voltage drop across the resistor Rlim, the current dissipated by the PNP emitter follower or open-source PMOS source in the adaptive clipping circuit802or by the diode in the adaptive clipping circuit804, respectively, is reduced.

The feedback loop in adaptive clipping circuit804between node812and the (−) input of amplifier402holds the node812at the clip voltage Vclip, allowing Vinto initially have a higher voltage Vin=Vclip+Rlim*Idiodewhen the diode is forward-biased, i.e. when Vin>Vclip. The adaptive clipping circuit804has no effect when Vin<Vclipsince diode404is then reverse-biased and blocks current flow. The adaptive clipping circuit802operates in the same manner. While Vinis at a higher voltage than Vclipby Rlim*Idiodeduring the initial discharge period of the adaptive clipping circuit804, Vinapproaches Vclipwhen Idiodegoes to zero, concluding the discharge through the adaptive clipping circuit804. The resistor Rlimoperates to create an offset between the voltage at node812and Vin. This offset provides a means to charge the node812(temporarily pulling up the voltage) and, therefore, gives the feedback loop the ability to hold node812at Vclipduring the discharge through Rlim.

The operation of the exemplary adaptive clipping circuit804is illustrated schematically inFIG. 9. The feedback loop of the adaptive clipping circuit804holds the voltage at node812at Vclip, regardless of the input current Iinwhile Vinvoltages are Rlim*Idiodeabove Vclip(FIG. 9Ashows three different input current levels Iin_1>Iin_2>Iin_3). When the amplitude of the input current pulse returns to zero, i.e. at the end of the input pulse, diode404continues to conduct current assisting Vinto be discharged. This assist continues until the current through diode404decreases to zero at which point the voltage drop on Rlimequals zero. As a result, the diode assist phase always finishes at voltage Vclip. Once the excess current has been dissipated through the adaptive clipping circuit804via the diode404and the current-limiting resistor Rlim, the remaining current is dissipated through the feedback resistor Rfof the TIA200. It should be emphasized that because the voltage at node814is pulled down below Vclipcommensurate with the forward voltage drop across the diode404, the voltage difference ΔV between Vclipand Vsat,thcan be held to values of, for example, less than 0.3-0.4 V, for example 0.1 V, which is much lower than the forward voltage of a typical Si diode404of 0.7 V.

The above discussion applies, mutatis mutandis, also to the adaptive clipping circuit802. In some embodiments, while not shown inFIG. 8, the circuits can further include assisted adaptive clipping, e.g., where the circuit includes a first diode clipping path701illustrated byFIG. 7.

In the examples illustrated inFIG. 8, Rf˜5-10 kΩ and Rlim˜20Ω, so that the time constant for the discharge through Rlimto the voltage Vclipcan be much faster than the remaining discharge through the feedback loop of the TIA.

Without a current-limiting resistor Rlimthe voltage between the nodes811and812will actually reverse polarity above a certain input current level (seeFIG. 6B), meaning that Vinwill drop below Vclipsince the feedback loop cannot act instantaneously. When Vin<Vclip, Vclipcannot pull-up the node811to Vclipbecause the diode404(or commensurately the transistor504) are reverse-biased. This situation leads to the phase reversal described above. The current-limiting resistor Rlimcreates an offset between the node812sensed by the feedback and Vin. This offset provides a means to charge the node812(a temporary pull-up path) and, therefore, gives the feedback loop the ability to hold the node812at Vclipduring the discharge.

According to some embodiments of the disclosure illustrated inFIG. 10, a resistive element Rdmay be inserted in the current path between the anode of the APD and the input of the TIA200. In the embodiment ofFIG. 10, a first clipping circuit1004may be connected to a connection point1012between the input of TIA200and a first terminal of the resistor Rd, and a second clipping circuit1002may be connected to a connection point1010between the anode of the APD and a second terminal of the resistor Rd. The second clipping circuit1002is designed to handle larger current pulses with less capacitive loading at the input of TIA200. In this structure, the purpose of the first clipping stage1004is to keep Vinat a desirable voltage level such that TIA can operate properly, i.e. as close as possible to the saturation threshold voltage of the TIA. The maximum current that is handled by the first clipping circuit1004is limited by the resistor Rd. Although the voltage drop caused by the internal resistance of the diode D1of the first clipping circuit1004is still of concern, as previously discussed in conjunction withFIG. 2, the maximum current supplied by the first clipping circuit1004is limited by the resistor Rdand by the second clipping circuit1002. With these relaxed current handling requirements, the diode D1can now have a much smaller size, since its internal resistance will be less of a concern due to the smaller currents flowing through the diode D1.

The current through diode D1will generate a voltage drop across the resistor Rd. Because of this voltage drop, assuming that for example Vclip2is selected to be equal to Vclip1, the voltage applied to diode D2will be larger than the voltage applied to diode D1. Therefore, the current through diode D2is larger than the current through diode D1even when the two diodes D1and D2are equally sized. As a result, a smaller portion of the current from APD is handled by the first clipping circuit1004while the larger remaining current from the APD is handled by the second clipping circuit1002.

Although the clipping circuits1002and1004are each shown as having a structure substantially identical to the circuit having the buffer202and the diode204shown inFIG. 2, or likewise the circuit701ofFIG. 7, it will be understood that one of the clipping circuits1002and1004may also have the structure of circuits702and704(FIG. 7) or of circuits802and804(FIG. 8), respectively.

The second clipping circuit1002deals with a substantial fraction of the high input current from the APD. However, the voltage-dependent internal resistance RD2,intof the diode D2of the second clipping circuit1002is not really of concern, because the output voltage VAPDof the APD does no longer affect the operating point of the TIA. For example, even when the output voltage VAPDis very high, for example 5V, the voltage drop across Rdis large enough so that the first clipping circuit1004need to deal with only a small portion of the current and is capable of keeping Vinclose to or only slightly above Vclip1+0.7V which is within the operating range of the TIA, i.e. as close as possible to the saturation threshold voltage of the TIA. With this architecture, D1and D2can be small while still being able to handle large input currents.

At a high current where both clipping circuits1002and1004are activated, the current Iclip1,maxof the first clipping circuit1004can be calculated as;
Iclip1,max=(Vclip2−Vclip1+Iclip2*RD2,int)/(Rd+RD1,int),assuming that the current through feedback resistor Rfis negligible. RD1,intis the voltage-dependent internal resistance of the diode D1, RD2,intis the voltage-dependent internal resistance of the diode D2. The maximum current through D1would be minimized by selecting Vclip2=Vclip1.

FIG. 12shows an example scenario of operating conditions for the circuit ofFIG. 10. As mentioned above, Vclip1and Vclip2may advantageously be chosen to be identical and, for example, equal to 1.8V. Assuming that the APD supplies a photocurrent of 500 mA, that the internal resistances of both diodes D1and D2is 10 at their respective operating points, and that the resistance of the resistive element Rdis selected to be 18Ω, the current dissipated by the first clipping stage1004will be 25 mA and the current dissipated by the second clipping stage1002will be 475 mA. The voltage at the connection point1010is then 2.975V, while the voltage at the connection point1012and thus also at the input of the TIA is 2.525V, which is well within the normal operating range of the TIA. Increasing Rdrelaxes the current handling requirement of the first clipping circuit1004, but increases the noise of the system.

While the illustrated example may not correspond to actual optimized operating conditions, it demonstrates that it operates much better than having only a single stage clipping circuit.

According to some embodiments of the disclosure illustrated inFIG. 11, the first clipping stage1002ofFIG. 10may be replaced with an ESD (electrostatic discharge) diode1102which is typically already incorporated in a photocurrent detection circuit to provide ESD protection and is connected in parallel with the APD. This circuit is equivalent to the circuit ofFIG. 10, where Vclip2=VCC<VCC,APD, and where there is no need for the buffer in1002. Using ESD diode as the second clipping stage has the advantage that no additional capacitance is introduced at the connection point1010since ESD diode is already in place. The disadvantage, however, is that Vclip2has to be equal to Vcc, which results in higher maximum current in first clipping structure. This current can be reduced by increasing Rd, however, increasing Rdincreases the noise contribution.

Detection of pulse dispersion can provide information about weather conditions in optical ranging applications since multiple scattering occurs for example on water droplets, such as fog. The weather conditions may thus be an important parameter for ranging applications because of a resulting change in the shape of the reflected laser pulse. Aside from detecting weather conditions, dispersion information is also needed to maintain the accuracy of ranging algorithms.

The components used in laser radar electronics are typically low-priced and thus this technology is potentially interesting for high-volume applications.

According to one embodiment, the polarity of the circuits including the polarity of the APD may be reversed, without affecting the performance or operation of the aforedescribed circuits.

Although the embodiments have been described with reference to LIDAR applications, it will be understood that the described and illustrated circuits can be used whenever light pulses with varying intensity need to be detected, for example in optical time domain reflectometers (OTDR) where overloading by large optical pulses can occur. Moreover, the described and illustrated circuits may be used when other type of current pulses need to be detected.

In the drawings, clipping or limitation of the input voltage Vinis shown only for one polarity. It is noted that it is possible to provide clipping for two polarities by providing clipping circuits which also clip the other polarity. This can be realized easily by, e.g., the use of Schottky diodes, which conduct current in the forward direction and block current flow in the reverse direction. This is in contrast to, e.g., Zener diodes, which are used in the reverse direction.

It should also be noted that the functions related to circuit architectures illustrate only some of the possible circuit architecture functions that may be executed by, or within, systems illustrated in the FIGURES. Some of these operations may be deleted or removed where appropriate, or these operations may be modified or changed considerably without departing from the scope of the present disclosure. In addition, the timing of these operations may be altered considerably. The preceding operational flows have been offered for purposes of example and discussion. Substantial flexibility is provided by embodiments described herein in that any suitable arrangements, chronologies, configurations, and timing mechanisms may be provided without departing from the teachings of the present disclosure.

Numerous other changes, substitutions, variations, alterations, and modifications may be ascertained to one skilled in the art and it is intended that the present disclosure encompass all such changes, substitutions, variations, alterations, and modifications as falling within the scope of the appended claims.

Note that all optional features of the device and system described above may also be implemented with respect to the method or process described herein and specifics in the examples may be used anywhere in one or more embodiments.