NTSC video signal receivers with reduced sensitivity to interference from co-channel digital television signals

A video signal receiver with reduced sensitivity to interference from co-channel digital TV signals includes circuitry for selecting a vestigial sideband amplitude-modulation signal descriptive of a video signal, converting the selected VSB AM signal to an intermediate frequency signal, and amplifying the IF signal to provide an amplified IF signal. The VSB AM signal is selected from any single one of a plurality of channels which can contain co-channel interference from a digital television signal. The amplified IF signal is synchronously detected with respect to video carrier frequency for generating an in-phase synchronous detection response and for generating a quadrature-phase synchronous detection response. All frequency components of the quadrature-phase synchronous detection response above a prescribed frequency are phase shifted substantially 90.degree. and are linearly combined with in-phase synchronous detection response for recovering lower frequency portions of the video signal, substantially free of artifacts from any co-channel interfering digital television signal. In some embodiments of the video signal receiver higher frequency portions of the video signal described in the full sideband of the VSB AM signal, but not in its vestigial sideband, are recovered by synchronously detecting the VSB AM signal after selective filtering to remove the pilot carrier signal component of any co-channel digital TV signal. The selective filtering avoids any artifact of such pilot carrier signal being generated when recovering higher frequency portions of the video signal. The lower and the higher frequency portions of the video signal are combined to obtain a fullband video signal.

The invention relates to NTSC television signal receivers and, more 
particularly, to improvements in such receivers for rendering them 
substantially less sensitive to interference from co-channel digital 
television signals. 
BACKGROUND OF THE INVENTION 
U.S. Pat. No. 5,122,879 issued 16 Jun. 1992 to Katsu Ito and entitled 
"TELEVISION SYNCHRONOUS RECEIVER WITH PHASE SHIFTER FOR REDUCING 
INTERFERENCE FROM A LOWER ADJACENT CHANNEL" describes a receiver for the 
video portion of an analog television signal that synchronously detects 
received analog television signal both in-phase and quadrature-phase. To 
improve noise figure by avoiding amplifiers with varactor diode tuning, 
the Ito receiver synchrodynes the radio-frequency (RF) amplifier response 
directly to baseband, so an adjacent lower channel may appear as an image. 
The quadrature-phase synchronous detection response is phase shifted 
90.degree. at all video frequencies above 750 kHz and linearly combined 
with the in-phase synchronous detection response to suppress image 
frequency components translated to baseband during synchronous detection 
of the video portion of the received NTSC signal. In U.S. Pat. No. 
5,122,879 Ito does not disclose the fact that this procedure also cancels 
the video components above 750 kHz. The attendant loss of luminance high 
frequencies is acceptable in small-viewing-screen television receivers, 
however, such as those used in wrist watches. 
By modifying the band-limited video signal receiver described by Ito so 
that the quadrature-phase synchronous detection response is phase shifted 
90.degree. at all video frequencies, artifacts of co-channel interfering 
digital television signals are removed from the band-limited baseband NTSC 
signal, it is here pointed out. Presuming that the time constant of the 
automatic gain control circuitry in the receiver is a few horizontal scan 
lines, the quadrature-phase synchronous detection response need be phase 
shifted 90.degree. only for frequencies above a few kilohertz to keep 
artifacts of co-channel interfering digital television signals in the 
band-limited video signal from being visible on the television viewing 
screen or interfering with horizontal synchronization. 
SUMMARY OF THE INVENTION 
A video signal receiver with reduced sensitivity to interference from 
co-channel digital television signals is constructed in accordance with a 
principal aspect of the invention to include input circuitry for selecting 
a vestigial sideband amplitude-modulation signal descriptive of a video 
signal, converting the selected vestigial sideband amplitude-modulation 
signal to an intermediate frequency signal, and amplifying the 
intermediate frequency signal to provide an amplified intermediate 
frequency signal. The vestigial sideband amplitude-modulation signal as 
originally received includes a video carrier and full sideband in addition 
to a vestigial sideband. The vestigial sideband amplitude-modulation 
signal is selected from any single one of a plurality of channels at least 
one of which is subject to containing at times co-channel interference 
from a digital television signal. Video synchrodyning circuitry 
synchronously detects the amplified intermediate frequency signal with 
respect to video carrier signal, for generating an in-phase synchronous 
detection response and for generating a quadrature-phase synchronous 
detection response. All frequency components of the quadrature-phase 
synchronous detection response above a prescribed frequency are phase 
shifted by substantially 90.degree. by an inverse Hilbert transform filter 
and are linearly combined with suitably delayed in-phase synchronous 
detection response for recovering lower frequency portions of the video 
signal described both in the full sideband and the vestigial sideband of 
the vestigial sideband amplitude-modulation signal, substantially free of 
artifacts from any co-channel interfering digital television signal. The 
term "linear combiner" as used in this specification and the claims 
appended thereto is a generic term for "additive combiner", or adder, and 
for "differential combiner", or subtractor. 
In accordance with further aspects of the invention, the video signal 
receiver includes circuitry for recovering higher frequency portions of 
the video signal described in the full sideband of the vestigial sideband 
amplitude-modulation signal, but not in its vestigial sideband. The 
vestigial sideband amplitude-modulation signal as supplied to this 
circuitry for recovering higher frequency portions of the video signal is 
selectively filtered to remove the pilot carrier signal component of any 
co-channel digital television signal. This is done to avoid any artifact 
of such pilot carrier signal being generated when recovering higher 
frequency portions of the video signal. The video signal receiver further 
includes circuitry for linearly combining those higher frequency portions 
of the video signal with the lower frequency portions of the video signal, 
which are described both in the full sideband and the vestigial sideband 
of the vestigial sideband amplitude-modulation signal, and which are 
substantially free of artifacts from any co-channel interfering digital 
television signal.

DETAILED DESCRIPTION 
FIG. 1 shows a television receiver that is capable of receiving NTSC analog 
TV signals as well as DTV signals. Over-the-air type television 
broadcasting signals as received by an antenna 1 are amplified by an 
adjustably tuned radio-frequency amplifier 2 and supplied to a first 
detector 3. The RF amplifier 2 and the first detector 3 have adjustable 
tuning and together function as a tuner for selecting said digital 
television signal from one of channels at different locations in a 
frequency band. The first detector 3 includes a first local oscillator 
supplying first local oscillations tunable over a frequency range above 
the ultra-high-frequency (UHF) TV broadcast band and a first mixer for 
mixing the first local oscillations with a TV signal selected by the 
adjustably tuned RF amplifier 2 for upconverting the selected TV signal to 
generate a UHF intermediate-frequency signal in a 6-MHz-wide UHF 
intermediate-frequency band located at frequencies above the assigned 
channels in the UHF TV broadcast band. 
The first detector 3 supplies the high-IF-band signal to a UHF-band 
intermediate-frequency amplifier 4 for NTSC audio signal, to a UHF-band 
intermediate-frequency amplifier 5 for fullband NTSC video signal, and to 
a UHF-band intermediate-frequency amplifier 6 for NTSC video highs signal. 
The responses of the UHF-band IF amplifiers 4, 5 and 6 are supplied to 
respective second detectors 7, 8 and 9 to be downconverted to respective 
VHF-band intermediate-frequency signals in a VHF band below the very high 
frequencies assigned as TV broadcast channels. The second detectors 7, 8 
and 9 share a common second local oscillator for generating second local 
oscillations and have respective second mixers for mixing those second 
local oscillations with the responses of the UHF-band IF amplifiers 4, 5 
and 6, respectively. The VHF-band IF signals from the second detector 
detectors 7, 8 and 9 are respectively supplied to a VHF-band 
intermediate-frequency amplifier 10 for NTSC audio signal, to a VHF-band 
intermediate frequency amplifier 11 for fullband NTSC video signal, and to 
a VHF-band intermediate frequency amplifier 12 for NTSC video highs 
signal. 
The UHF-band IF amplifiers 4, 5 and 6 include surface-acoustic-wave (SAW) 
filters for UHF-IF-band NTSC audio signal, for UHF-IF-band fullband NTSC 
video signal and for UHF-IF-band NTSC video highs signal, respectively. 
SAW filters with steep rejection skirts, but with pass bands having linear 
group delay and flat amplitude response, are more easily implemented at 
UHF than at VHF. This is the reason for preferring to determine overall IF 
response for UHF-IF-band NTSC audio signal, for UHF-IF-band fullband NTSC 
video signal and for UHF-IF-band NTSC video highs signal in the UHF IF 
band rather than in the VHF IF band. 
The SAW filter in the IF amplifier 5 for determining overall IF response 
for fullband NTSC video signal preferably has substantially flat amplitude 
response for those portions of the VSB AM signal ranging between 
frequencies 500 kHz to at least 3.5 MHz above the lower limit of the 
6-MHz-wide TV broadcast channel as that VSB AM signal is translated to the 
UHF IF band, that rejects in-channel and adjacent-channel NTSC audio 
signals, and that has substantially linear phase response throughout its 
passband. The SAW filter in the IF amplifier 5 may suppress or reject the 
pilot carrier of any co-channel interfering ATSC DTV signal, so long as 
linear phase response is maintained to 750 kHz from NTSC video carrier 
frequency. The self resonances of the IF filtering for fullband NTSC video 
signal, which are stimulated by impulse noise, are near the middle of the 
IF passband. So, ringing effects caused by impulse noise are less apt to 
affect baseband video response below 750 kHz if the IF filtering has at 
least 3 MHz bandwidth. 
The SAW filter in the IF amplifier 6 for determining overall IF response 
for NTSC video highs signal rejects in-channel and adjacent-channel NTSC 
audio signals, and preferably this SAW filter exhibits a roll-off for the 
lower 1.75 MHz or so of the 6-MHz-wide TV broadcast channel as translated 
to the UHF IF band and has substantially linear phase response throughout 
its passband. The roll-off for the lower 1.75 MHz or so of the 6-MHz-wide 
TV broadcast channel as translated to intermediate frequencies rejects 
adjacent-channel NTSC audio signal, the pilot carrier of any co-channel 
interfering ATSC DTV signal and the in-channel NTSC video carrier. The SAW 
filter in the IF amplifier 6 preferably exhibits a roll-off for the upper 
550 kHz or so of the 6-MHz-wide TV broadcast channel as translated to the 
UHF IF band and rejects in-channel sound signal. 
FIG. 8 shows the desired overall receiver responses, as referred to the 
lower frequency of the original transmission channel, at the output ports 
of the UHF IF amplifiers 5 and 6. 
The UHF-band IF amplifiers 4, 5 and 6 can include wideband constant-gain 
amplifiers for driving their component SAW filters from source impedances 
that minimize multiple reflections and for overcoming the insertion losses 
of their component SAW filters. The VHF-band IF amplifiers 10, 11 and 12 
include respective controlled-gain amplifiers that provide up to 60 dB or 
more amplification. The VHF-band IF amplifiers 10, 11 and 12 each include 
stages with forward automatic gain control derived in response to the 
output signal level of the IF amplifier 11, forward AGC being preferred 
for the better noise figure it affords. The RF amplifier 2 is provided 
with delayed reverse automatic gain control in response to the output 
signal level of the IF amplifier 11. 
The response of the VHF IF amplifier 10 is applied to an intercarrier sound 
detector 13, which supplies 4.5 MHz intercarrier sound 
intermediate-frequency signals to an intercarrier sound 
intermediate-frequency amplifier 14 which amplifies and in most designs 
symmetrically limits the amplified response for application to an FM 
detector 15. The FM detector 15 reproduces baseband composite audio signal 
supplied to the sound reproduction portion 16 of the NTSC receiver per 
conventional practice. The sound reproduction portion 16 of the NTSC 
receiver typically includes stereophonic decoder circuitry. If the NTSC 
audio signals are selected with narrowband filtering in the IF amplifiers 
4 and 10 that pass only the FM audio carrier as translated to intermediate 
frequencies, the intercarrier sound detector 13 can be provided by a 
multiplier that multiplies the IF amplifier 10 response by a video carrier 
supplied from a third local oscillator in circuitry 17 for synchrodyning 
the fullband NTSC video signal to baseband. 
Alternatively, if the NTSC audio signals are selected with filtering in the 
IF amplifiers 4 and 10 that passes both the NTSC video and audio carriers 
as translated to intermediate frequencies, for implementing 
"quasi-parallel" sound, the intercarrier sound detector 13 can be a simple 
rectifier or can be a square-law device. A video carrier is then no longer 
supplied from a third local oscillator in the circuitry 17 for 
synchrodyning the fullband NTSC video signal to baseband. 
Output signal from the VHF IF amplifier 11 is applied to the circuitry 17 
for synchrodyning NTSC video carrier modulation to baseband, which 
circuitry can take the form shown in FIG. 9. Both an in-phase synchronous 
detector and a quadrature-phase synchronous detector are used in the 
circuitry 17 for synchrodyning NTSC video carrier modulation to baseband. 
Synchrodyning is carried out in the analog regime in the specific 
circuitry 17 for synchrodyning NTSC video carrier modulation to baseband 
shown in FIG. 9, and the responses of the in-phase synchronous detector 
170 and the quadrature-phase synchronous detector 171 used for this 
purpose are digitized using respective analog-to-digital converters 172 
and 173. The third local oscillator 174 in the circuitry 17 supplies 
oscillations in 0.degree. phasing to the in-phase synchronous detector 170 
and supplies oscillations in +90.degree. phasing or in -90.degree. phasing 
via a phase shift network 175 to the quadrature-phase synchronous detector 
171. The third local oscillator 174 is a controlled oscillator provided 
automatic frequency and phase control (AFPC) signal responsive to the 
unwanted appearance of low frequency components in the quadrature-phase 
synchronous detector 171 response. FIG. 9 shows the AFPC signal being 
generated using the commonplace Costas loop arrangement in which the 
responses of the in-phase synchronous detector 170 and the 
quadrature-phase synchronous detector 171 are filtered by lowpass filters 
176 and 177, the responses of the lowpass filters 176 and 177 are 
multiplicatively mixed in a mixer 178, and the resulting product is 
filtered by a lowpass filter 179 to generate the AFPC signal for the third 
local oscillator 174. 
Alternatively, synchrodyning NTSC video carrier modulation to baseband can 
be done in the digital regime after converting to a final 
intermediate-frequency band just above baseband, so the final 
intermediate-frequency can be digitized. This avoids any problems with the 
two analog-to-digital converters 172 and 173 differing somewhat in 
conversion gain. 
The digital response Q of the quadrature-phase synchronous detector 171 is 
the Hilbert transform of the single sideband components of the NTSC signal 
(i. e., those components above 750 kHz in frequency) plus the artifacts of 
the DTV signal as they appear in the response I of the in-phase 
synchronous detector 170. The reader's attention is now directed back to 
FIG. 1. This Hilbert transform provided by the response Q of the 
quadrature-phase synchronous detector in the synchrodyne circuitry 17 is 
phase shifted to provide 90.degree. lag at all frequencies above a few 
kilohertz by inverse Hilbert transform circuitry 18. 
Finite-impulse-response digital filters suitable for the inverse Hilbert 
transform circuitry 18 are known in the digital television receiver art. 
The inverse Hilbert transform response of the circuitry 18 is linearly 
combined in a linear combiner 19 with the digital response I of the 
in-phase synchronous detector, to generate a luminance signal cutting off 
somewhat above 750 kHz. This luminance signal is generally free of DTV 
artifacts, owing to their single-sideband character as referred to NTSC 
video carrier frequency. Whether the linear combiner 19 is an adder or a 
subtractor depends on the whether the operation of the quadrature-phase 
synchronous detector is chosen to lead the operation of the in-phase 
synchronous detector or to lag it. 
Output signal from the VHF IF amplifier 12 is applied to a quadrature-phase 
synchronous detector 20 for synchrodyning to baseband the NTSC video 
carrier modulation that is descriptive of the higher-frequency portions of 
the composite video signal. The quadrature-phase synchronous detector 20 
supplies a digital response Q'. By way of example, if quadrature-phase 
synchronous detection is performed in the analog regime, an 
analog-to-digital converter is cascaded after the synchronous detector for 
digitizing its response. Synchronous carrier for the quadrature-phase 
synchronous detector 20 is supplied from the same source in the circuitry 
17 for synchrodyning NTSC video carrier modulation to baseband as supplies 
the quadrature-phase synchronous detector within the synchrodyne circuitry 
17 (e. g., from the phase shift network 175). The response Q' of the 
quadrature-phase synchronous detector 20 is the Hilbert transform of the 
single sideband components of the NTSC signal (i. e., those components 
above 750 kHz in frequency) plus the artifacts of the portion of the DTV 
signal passed by the SAW filter in the IF amplifier 6. This Hilbert 
transform provided by the response Q' of the quadrature-phase synchronous 
detector 20 is phase shifted to provide 90.degree. lag at least at 
frequencies above 500 kHz or so by inverse Hilbert transform circuitry 21. 
This procedure generates a response that is the same at higher frequencies 
as the response of an in-phase NTSC video detector, but which exhibits a 
low-frequency cut-off that is complementary to the high-frequency cut-off 
of the linear combiner 19. 
A linear combiner 22 combines the responses of the linear combiner 19 and 
of the quadrature-phase synchronous detector 20 to generate a fullband 
composite video signal for application to the portion 23 of the NTSC 
receiver used to reproduce pictures on a viewing screen. This portion 23 
of the NTSC receiver typically includes sync separation circuitry and 
color signal reproduction circuitry; in a combination NTSC and HDTV 
receiver circuitry will also be included for adapting the 4:3 aspect ratio 
NTSC image for presentation on a 16:9 viewscreen used for displaying DTV 
images. 
The inverse Hilbert transform circuitry 18 requires a substantial amount of 
latency (or insertion delay) in order to provide 90.degree. lag for 
frequencies as low as a few kilohertz. Providing 90.degree. lag for 
frequencies that are a fraction of horizontal scan line rate means that 
uncancelled artifacts of DTV signals will be of low enough frequency that 
receiver AGC will operate to suppress them. Shim delay is necessary in the 
I signal connection from synchrodyne circuitry 17 to linear combiner 19 
for equalizing the latencies of the I and Q signals supplied to the linear 
combiner 19. Shim delay must be cascaded with the inverse Hilbert 
transform circuitry 21 to the extent that its latency is less than that of 
the inverse Hilbert transform circuitry 18. Making the inverse Hilbert 
transform circuitry 21 the same as the inverse Hilbert transform circuitry 
18 is possible to avoid the need for such shimming. When such modification 
is made, the circuitry can be subjected to a reduction technique that 
eliminates the need for separate inverse Hilbert transform circuitry 18 
and 21. 
FIG. 2 shows a television receiver that is capable of receiving NTSC analog 
TV signals as well as DTV signals, which receiver results from such 
reduction. Elements 18-22 of the FIG. 1 television receiver are replaced 
by an adder 24 for combining the Q output signal of the synchrodyne 
circuitry 17 and the Q' output signal of the quadrature synchronous 
detector 20, inverse Hilbert transform circuitry 25 responsive to the sum 
output signal from the adder 24, and linear combining circuitry 26 for 
linearly combining the inverse Hilbert transform circuitry 25 response 
with the I output signal of the synchrodyne circuitry 17 to generate a 
luminance signal cutting off somewhat above 750 kHz. This luminance signal 
is generally free of DTV artifacts, owing to their single-sideband 
character as referred to NTSC video carrier frequency. If the quadrature 
synchronous detector 20 and the quadrature synchronous detector within the 
synchrodyne circuitry 17 are operated out-of-phase with each other, rather 
than in-phase with each other as presumed, the adder 24 is replaced by a 
subtractor to achieve equivalent operation. 
The synchronous detection of video high frequencies using quadrature-phase 
video carrier is advantageous in that cross-over between video low 
frequencies and video high frequencies is automatically correct. 
Furthermore, cross-over occurs at the highest video frequencies possible 
so that DTV artifact cancellation extends to as high frequency as 
possible. 
FIG. 3 shows a modification of the FIG. 1 television receiver, which 
modification uses an in-phase synchronous detector 27 for synchronous 
detection of video high frequencies, rather than the quadrature-phase 
synchronous detector 20. The quadrature-phase synchronous detector 20 is 
dispensed with, together with the inverse Hilbert transform circuitry 21 
and the linear combiner 22. A cross-over filter 28 lowpass filters the 
response of the linear combiner 19 and highpass filters the I' output 
signal of the in-phase synchronous detector 27 before linearly combining 
them to generate a fullband NTSC composite video signal for application to 
the portion 23 of the NTSC receiver used to reproduce pictures on a 
viewing screen. The cross-over frequency at which the lowpass filtering 
and highpass filtering cut off in the cross-over filter 28 is preferably 
at least 500 kHz. The FIG. 2 television receiver is more economical of 
hardware than the FIG. 3 receiver, since the cross-over filter 28 is not 
required in the FIG. 2 receiver. 
In the FIG. 1, FIG. 2 and FIG. 3 television receivers chroma demodulation 
circuitry is presumed to be included in the portion 23 of the NTSC 
receiver used to reproduce pictures on a viewing screen, with chroma 
signal being separated from the fullband composite video signal applied to 
that portion 23 of the NTSC receiver used to reproduce pictures on a 
viewing screen. However, it is possible alternatively to separate chroma 
signal from the high frequency component of the composite video signal 
before its combination with the low frequency component of the composite 
video signal. 
FIG. 4 shows a variant of the FIG. 3 television receiver having 
conventional chroma demodulation circuitry 29 connected to be directly 
responsive to the baseband video high frequencies as detected by the 
in-phase synchronous detector 27. The chroma demodulation circuitry 29 is 
shown as being separate from a portion 30 of the NTSC receiver used to 
reproduce pictures on a viewing screen. The chroma demodulation circuitry 
29 supplies color difference signals to that portion 30 of the NTSC 
receiver, which portion 30 receives fullband composite video signal from 
the cross-over filter 28. 
FIG. 5 shows a variant of the FIG. 2 television receiver having chroma 
demodulation circuitry 29 connected to be directly responsive to the 
baseband video high frequencies as detected by the quadrature-phase 
synchronous detector 20. Since color burst like other chroma signal 
components is phase shifted by 90.degree., the fact of the Hilbert 
transform of color signal rather than actual color signal being 
synchronously detected has no substantial effect on color difference 
signal recovery. 
The FIG. 6 television receiver differs from the FIG. 2 TV receiver in that 
the UHF-band IF amplifier 5 is replaced by a UHF-band IF amplifier 31 with 
a passband for the complete frequency spectrum of the VSB AM NTSC video 
carrier modulation, the NTSC video lows second detector 8 is replaced by a 
second detector 32 for the complete frequency spectrum of the VSB AM NTSC 
video carrier modulation, and the VHF-band IF amplifier 11 is replaced by 
a VHF-band IF amplifier 33 with a passband for the complete frequency 
spectrum of the VSB AM NTSC video carrier modulation. FIG. 10 shows the 
desired overall receiver response, as referred to the lower frequency of 
the original transmission channel, at the output port of the UHF IF 
amplifier 33. This full bandwidth response permits the UHF-band IF 
amplifier 6, the NTSC video highs second detector 9, the VHF-band IF 
amplifier 12, the NTSC video highs quadrature synchronous detector 20, and 
the inverse Hilbert transform circuitry 21 to be dispensed with entirely. 
Instead, a highpass filter 34 extracts video high frequencies from the 
response of the inverse Hilbert transform circuitry 18 for application to 
the linear combiner 22, there to be linearly combined with the video low 
frequencies supplied from the linear combiner 19. 
The FIG. 7 television receiver differs from the FIG. 3 TV receiver in that 
the UHF-band IF amplifier 5 is replaced by a UHF-band IF amplifier 31 with 
a passband for the complete frequency spectrum of the VSB AM NTSC video 
carrier modulation, the NTSC video lows second detector 8 is replaced by a 
second detector 32 for the complete frequency spectrum of the VSB AM NTSC 
video carrier modulation, and the VHF-band IF amplifier 11 is replaced by 
a VHF-band IF amplifier 33 with a passband for the complete frequency 
spectrum of the VSB AM NTSC video carrier modulation. FIG. 10 shows the 
desired overall receiver response, as referred to the lower frequency of 
the original transmission channel, at the output port of the UHF IF 
amplifier 33. This full bandwidth response permits the UHF-band IF 
amplifier 6, the NTSC video highs second detector 9, the VHF-band IF 
amplifier 12 and the NTSC video highs in-phase synchronous detector 27 to 
be dispensed with entirely. Instead, the response I of the in-phase 
synchronous detector in the synchrodyne circuitry 17 is applied to the 
cross-over filter 28 to supply it with video high frequencies. 
In variants of the FIG. 1 TV receiver, chroma demodulation circuitry can be 
arranged to directly respond to the response of the quadrature-phase 
synchronous detector 20 or of the inverse Hilbert transform circuitry 21. 
In variants of the FIG. 6 and FIG. 7 TV receivers, chroma demodulation 
circuitry can be arranged to directly respond to either the I output 
signal or the Q output signal of the synchrodyne circuitry 17 or to 
directly respond to the response of the inverse Hilbert transform 
circuitry 18; these arrangements are possible in the FIG. 1, FIG. 2 and 
FIG. 3 TV receivers as well, if their video lows IF amplifiers are 
modified to have an overall fullband response as shown in FIGURE 10. 
The effect of artifacts of co-channel interfering digital signal on chroma 
demodulation results can (like other forms of random noise) be reduced by 
transversal filtering, since digital television signals are random from 
scan line to scan line while chrominance signals tend to exhibit strong 
line-to-line anticorrelation before demodulation and strong line-to-line 
correlation after demodulation. An NTSC receiver as described above can be 
incorporated into a digital television receiver as described in patent 
application Ser. No. 08/821,944 filed by the inventor 21 Mar. 1997 and 
entitled "USING VIDEO SIGNALS FROM AUXILIARY ANALOG TV RECEIVERS FOR 
DETECTING NTSC INTERFERENCE IN DIGITAL TV RECEIVERS". Modifications of the 
television receivers thusfar disclosed to use or SECAM signals rather 
than NTSC signals are easily effected by one skilled in the art of 
television receiver design when acquainted with the foregoing disclosure. 
While the foregoing disclosure describes NTSC television receivers that 
reproduce sound and picture, the invention has application to NTSC 
television receivers that do not reproduce sound and picture, such as 
those incorporated into video tape recorders or into NTSC signal 
cancellation filters for digital television receivers. One skilled in the 
art of television receiver design when acquainted with the foregoing to 
disclosure will be enabled to design many variants of the receivers 
described as preferred embodiments, and this should be borne in mind when 
determining the scopes of the claims which follow.