Predistortion generator coupled with an RF amplifier

An in-line distortion generator is coupled to an RF amplifier on a single PC board for producing an output signal of useful amplitude but with low composite triple beat and cross modulation distortions. The backplane under the section of the PC board upon which the distortion circuit resides is removed and the portion of the heat sink under the removed portion of the backplane is also removed. This eliminates any parasitic capacitances that could degrade the performance of the RF amplifier, thereby making the distortion circuit transparent to the RF amplifier. Furthermore, the layout of the predistortion circuitry has been specifically designed to enhance the performance of the circuitry without inducing any negative operating characteristics on the associated RF amplifier.

BACKGROUND OF THE INVENTION 
1. Field of the Invention 
This invention relates generally to radio frequency (RF) amplification. 
More particularly, the invention relates to a predistortion generator 
coupled with an RF amplifier on a single printed circuit board. 
2. Description of the Related Art 
Lowering distortion in RF power amplifier circuits without compromising 
their transient response is an omnipresent problem. High frequency 
amplification is widely used in communications and broadcasting and also 
where high-speed switching is required for use in digital instrumentation. 
However, high frequency amplifier applications extend linear operation 
into areas where parasitic effects of interelectrode capacitance, wire 
inductance, stored charge and even operating frequency wavelength begin to 
adversely affect and dominate circuit behavior. 
Minimizing distortion is particularly important when a series of amplifiers 
is cascaded over a signal transmission path, such as a series of RF 
amplifiers in a CATV transmission system. Disposed throughout a CATV 
transmission system are RF amplifiers that periodically amplify the 
transmitted signals to counteract cable attenuation and attenuation caused 
by passive CATV components, such as signal splitters and equalizers. The 
RF amplifiers are also employed to maintain the desired carrier-to-noise 
ratio. Due to the number of RF amplifiers employed in a given CATV 
transmission system, each RF amplifier must provide minimum degradation to 
the transmitted signal. 
Many amplifiers are subject to a wide range of ambient operating 
temperatures. These temperature changes may affect the operating 
characteristics of certain electronic components within the amplifier, 
thereby inducing additional distortions. A temperature range of 
-40.degree. C. to +85.degree. C. is not uncommon for many amplifier 
applications in a communication environment. To ensure consistent 
performance over the operating bandwidth, and to minimize resulting 
distortions, an amplifier must be designed for a broad range of ambient 
operating temperatures. 
The distortions created by an amplifier which are of primary concern are 
second (even) and third (odd) order harmonic intermodulation and 
distortions. Prior art amplifier designs have attempted to ameliorate the 
effects of even order distortions by employing push-pull amplifier 
topologies, since the maximum even order cancellation occurs when the 
proper 180.degree. phase relationship is maintained over the entire 
bandwidth. This is achieved through equal gain in both push-pull halves by 
matching the operating characteristics of the active devices. 
However, odd-order distortion is difficult to remedy. Odd-order distortion 
characteristics of an amplifier are manifest as cross modulation (X-mod) 
and composite triple beat (CTB) distortions on the signal being amplified. 
X-mod occurs when the modulated contents of one channel being transmitted 
interferes with and becomes part of an adjacent or non-adjacent channel. 
CTB results from the combination of three frequencies of carriers 
occurring in the proximity of each carrier since the carriers are 
typically equally spaced across the frequency bandwidth. Of the two noted 
distortions, CTB becomes more problematic when increasing the number of 
channels on a given CATV system. While X-mod distortion also increases in 
proportion to the number of channels, the possibility of CTB is more 
dramatic due to the increased number of available combinations from among 
the total number of transmitted channels. As the number of channels 
transmitted by a communication system increases, or the channels reside 
close together, the odd-order distortion becomes a limiting factor of 
amplifier performance. 
There are three basic ways of correcting distortion created by a non-linear 
device (NLD): 1) reduce the signal power level; 2) use a feed forward 
technique; and 3) use a predistortion or postdistortion technique. The 
first method reduces the signal power level such that the NLD is operating 
in its linear region. However, in the case of an RF amplifier this results 
in very high power consumption for low RF output power. 
The second method is the feed forward technique. Using this technique, the 
input signal of the main amplification circuit is sampled and compared to 
the output signal to determine the difference between the signals. From 
this difference, the distortion component is extracted. This distortion 
component is then amplified by an auxiliary amplification circuit and 
combined with the output of the main amplification circuit such that the 
two distortion components cancel each other. Although this improves the 
distortion characteristics of the amplifier, the power consumed by the 
auxiliary amplification circuit is comparable to that consumed by the main 
amplification circuit. This circuitry is also complex and very temperature 
sensitive. 
The third method is the predistortion or postdistortion technique. 
Depending upon whether the compensating distortion signal is generated 
before the non-linear device or after, the respective term predistortion 
or postdistortion is used. In this technique, a distortion signal equal in 
amplitude but opposite in phase to the distortion component generated by 
the amplifier circuit is estimated and generated. This is used to cancel 
the distortion at the input (for predistortion) or output (for 
postdistortion) of the amplifier, thereby improving the operating 
characteristics of the amplifier. 
One distortion design, as disclosed in U.S. Pat. No. 5,703,530 and shown in 
FIG. 1, relies upon a traditional .pi.-attenuation network and a delay 
line for gain compensation; and a diode pair coupled with a delay line for 
distortion and phase compensation. This circuit generates a distortion 
that is equal in amplitude but opposite in phase to the distortion 
introduced by the amplifier. Plots of the distortions contributed by the 
distortion generator and the distortions manifest by the amplifier are 
shown in FIGS. 2 and 3. As shown, the distortion signal compensates for 
the distortions generated by the amplifier. However, the use of delay 
lines in such a manner is impractical since delay lines are physically 
large, are difficult to adjust and the results are inconsistent across a 
wide frequency range. Additionally, both amplitude and phase information 
are required for correct compensation. The '530 patent also states that 
the system disclosed therein is not ideal for certain applications, such 
as distortion for CATV RF amplifiers, due to the excessive losses 
introduced by the distortion circuit. 
Since a frequency response, which is flat within .+-.0.25dB over 50-1000 
MHz, is required of a CATV RF amplifier carrying over 150 channels, 
special attention must be paid not only to the high-frequency 
characteristics of the electronic components used in the RF amplifier 
design, but also to the layout and packaging techniques as well. One 
important aspect that has serious impact on high speed and high frequency 
circuits is the existence of parasitic capacitance within the circuit. The 
subtle effects of capacitance witnessed at low frequencies often dominate 
circuit behavior at high frequencies. 
Although it is paramount to eliminate distortions caused by RF amplifiers, 
most RF amplifier designs have succeeded in only reducing the distortions, 
not eliminating distortions. Accordingly, a separate circuit to compensate 
for these distortions is usually required. Coupling a distortion circuit 
to the associated RF amplifier on the same PC board is an option that is 
not typically pursued since it creates additional problems. Namely, 
parasitic capacitance of the distortion circuit components on the PC board 
causes degradation in the return loss and bandwidth performance of the RF 
amplifier. Accordingly, the performance of the RF amplifier is 
compromised. 
Accordingly, there exists a need for an integrated distortion generator 
which is coupled with an RF amplifier on a single PC board without 
degrading the performance characteristics of the RF amplifier. 
SUMMARY OF THE INVENTION 
The present invention comprises an in-line distortion generator coupled to 
an RF amplifier on a single PC board for producing an output signal of 
useful amplitude but with low composite triple beat and cross modulation 
distortions. The backplane under the section of the PC board upon which 
the distortion circuit resides is removed and the portion of the heat sink 
under the removed portion of the backplane is also removed. This 
eliminates any parasitic capacitances that could degrade the performance 
of the RF amplifier, thereby making the distortion circuit transparent to 
the RF amplifier. Furthermore, the layout of the predistortion circuitry 
has been specifically designed to enhance the performance of the circuitry 
without inducing any negative operating characteristics on the associated 
RF amplifier. 
Accordingly, it is an object of the invention to provide an RF amplifier 
coupled with a distortion generator on the same printed circuit board 
without degrading the performance of the RF amplifier. 
Other objects and advantages of the system and the method will become 
apparent to those skilled in the art after reading a detailed description 
of the preferred embodiment.

DESCRIPTION OF THE PREFERRED EMBODIMENTS 
The preferred embodiment of the present invention will be described with 
reference to the drawing figures where like numerals represent like 
elements throughout. Although the following is a description of a 
predistortion circuit coupled with an RF amplifier, those of skill in the 
art should recognize that the description is equally applicable to a 
postdistortion circuit coupled with an RF amplifier. 
The transfer function of an RF amplifier with no second order distortion is 
in the form of: 
EQU V.sub.out =k.sub.1 v.sub.in -k.sub.3 v.sub.in.sup.3 Equation (1) 
The negative sign for k.sub.3 represents the saturation phenomena. The 
output signal will be compressed at both sides when the signal swings from 
positive to negative. For an example of a typical CATV RF amplifier with 
18 dB gain and an input power level of 30 dBmv/channel for 77 channels, 
the transfer function is: 
EQU V.sub.out =7.8v.sub.in -0.056v.sub.in.sup.3 Equation (2) 
The input average peak voltage will be 0.38 volts and the output average 
peak voltage will be 3 volts if the RF amplifier is linear. Due to the 
nonlinearity of the hybrid, the final average peak amplitude is: 
EQU V.sub.abs =3-0.003 Equation (3) 
Equation 3 demonstrates that due to the nonlinearity of the RF amplifier, 
the output average voltage is compressed by 1 thousandth at the amplitude 
peak. In other words, the output signal of the RF amplifier has been 
compressed by 0.0086 dB at its amplitude peak. 
The remedy for this distortion is using the instant voltage nonlinear 
controlled attenuator in accordance with the present invention. This 
attenuator provides attenuation of 0.0086 dB at the RF signal peak. As 
will be explained in detail hereinafter, the present invention takes 
advantage of the nonlinearity of the current flowing through two diodes 
coupled together to instantly generate a correction voltage. When the 
nonlinear controlled attenuator is cascaded with an RF amplifier, the 
output signal of the combination of attenuator and RF amplifier will be 
linearized. 
The present invention will be described with reference to FIG. 4, whereby a 
.pi. attenuator network 20 is shown. The network 20 comprises a selected 
configuration of resistors Z.sub.1, R.sub.1, R.sub.2, R.sub.3, Z.sub.0, 
R.sub.p. The signal source is input at signal input 30 and the output of 
the attenuator network 20 is seen across the output 95. Z.sub.1 is the 
source of internal impedance which should be equal to the system impedance 
Z.sub.0, which is seen across the output 95. In an embodiment of the 
invention for use with a CATV system, the impedance values Z.sub.1 and 
Z.sub.0 are equal to 75 Ohms. Three of the resistors R.sub.1, R.sub.2, 
R.sub.3 form a .pi. attenuator configuration. Preferably, the values (Y) 
of resistors R.sub.2 and R.sub.3 are equal, and substantially larger than 
the value (X) of resistor R.sub.1. Resistor R.sub.p is connected in 
parallel with resistor R.sub.1. 
As one skilled in the art would clearly recognize, when the following 
condition is satisfied: 
EQU X=2Z.sub.0.sup.2 Y/(Y.sup.2 -Z.sub.0.sup.2) Equation (4) 
the attenuator network 20 is matched at input and output, from DC to very 
high frequencies. For one example of the attenuator when X=7.5 and Y=1.5 
K, the power attenuation A for this attenuator network 20 is: 
##EQU1## 
Under the condition when Z.sub.0 &lt;&lt;Y, (as is the case when X=7.5 and Y=1.5 
K): 
EQU A.congruent.(2Z.sub.0 /(2Z.sub.0 +X)).sup.2 Equation (6) 
EQU A(dB)=10 lg A Equation (7) 
When X=7.5 and Y=1.5 k, A (dB).congruent.0.42 dB. This means the attenuator 
network 20 has very low insertion losses and a good frequency response. 
When X has a small variation due to the parallel of R.sub.p, shown in FIG. 
4, from Equation (6) 
##EQU2## 
From Equation (9): 
##EQU3## 
For example, If R.sub.p =375 ohms then: 
##EQU4## 
Equation (11) shows that when R.sub.p (375 ohms) is in parallel with 
R.sub.1 (7.5 ohms), the attenuation will be reduced by 0.00868 dB. This 
amount of attenuation change is needed for non-linear compensation for an 
amplifier. This example also shows that when the value of R.sub.p 
&gt;&gt;R.sub.1, (i.e., when R.sub.p is 50 times larger than R.sub.1), adding 
R.sub.p parallel with R.sub.1 has almost no effect on the impedance match, 
and the voltage drop over the R.sub.p is mainly determined by the value of 
R.sub.1. 
However, if a linear resistor R.sub.p is used in the attenuator network 20, 
there will be no distortion signal produced. The attenuator network 20 as 
shown is a linear device. In order for a distortion circuit to operate 
effectively, diodes are used to create a non-linear resistance. 
Preferably, Schottky diodes are utilized. At small current, diode current 
is exponentially proportional to the voltage across over the diode. Thus 
diodes can be used as a non-linear resistance. For non-linear 
applications, the amount of attenuation can be calculated as: 
##EQU5## 
Where I.sub.p is the current flow through R.sub.p, (the non-linear 
resistance). I.sub.1 is the current flow through R.sub.1. Equation 12 
provides the relationship of the attenuation change due to the current 
change in I.sub.p. This equation is accurate over a broad frequency range. 
The relationship between the delta attenuation and a change in current is 
still valid when the resistance is a non-linear resistor. Accordingly, 
Equation 12 provides a good estimation of how much non-linear current is 
required for predistortion or postdistortion purposes. 
Referring to FIG. 5, when the input sinusoidal voltage wave changes from 
V.sub.1 to V.sub.2 to V.sub.3, the output current changes from I.sub.1 to 
I.sub.2 to I.sub.3 respectively. The non-linear current used for order 
correction is: 
EQU I.sub.nonlinear .congruent.I.sub.1 -2I.sub.2 +I.sub.3 Equation (13) 
From Equation 12, the non-linear current needed is: 
##EQU6## 
Only non-linear current will be useful for predistortion or postdistortion 
purposes. Equation 14 can be rewritten in the form of: 
##EQU7## 
Accordingly, I.sub.nonlinear eff in Equation 15 is the effective non-linear 
current going to the output port 114 which is shown in FIG. 6. 
I.sub.output in Equation 15 is the total current that goes to the output 
port 114. Equation 16 shows that only a small part of the non-linear diode 
current is effectively being used for correction. 
The .pi. attenuator network 20 has low insertion loss and the voltage drop 
of the input voltage on R.sub.1 (shown in FIG. 4) is proportional to the 
input voltage. This voltage may be used to drive a pair of diodes to 
produce non-linear current. The non-linear current flowing in the diodes 
will cause an attenuator to provide less attenuation at larger RF 
amplitudes, (i.e. when the input signal has a higher power). This may be 
used to compensate for the signal compression caused by amplification. 
Because of the relatively high value of the diode's non-linear resistance, 
the match of the attenuator network is almost unchanged. This match will 
not be changed even over temperature. Additionally, frequency response 
over multi-octave frequency bands is favorable. 
Referring to FIG. 6, the preferred embodiment of the attenuator 100 for 
predistortion and postdistortion is shown. The attenuator 100 of the 
present invention includes several additional components that modify a 
traditional .pi. attenuator to achieve significantly better performance 
over a wide frequency and temperature range. The attenuator 100 has an 
input port 101, an output port 114 and a bias control port 116. The 
attenuator 100 may be used in a predistortion configuration with an 
amplifier or in a postdistortion configuration. For a predistortion 
configuration, the output port 114 is connected to the input of an 
amplifier. For the postdistortion configuration as shown in FIG. 6, an 
output signal generated by an amplifier, is applied to the input port 101. 
The attenuator 100 includes resistors 105, 106, 107, 108, 112; capacitors 
102, 103, 104, 111, 113, 115; and diodes 109, 110. 
The function of the resistors 105, 106, 107, 108, 112 and the capacitors 
102, 103, 104, 111, 113, 115 is to form a modified .pi. attenuation 
network in comparison to the .pi. attenuation network 20 shown in FIG. 4. 
The capacitors 102, 103, 104, 111, 113, and 115 are also used for DC 
blocking and AC coupling. From an AC standpoint, the parallel combination 
of resistors 105 and 106 is functionally equivalent to resistor R.sub.2 of 
FIG. 4. Preferably, the values of resistors 105 and 106 should be chosen 
such that the parallel combination is equivalent to the value of 
resistance of resistor 112, (i.e. ((R.sub.105 *R.sub.106)/(R.sub.105 
+R.sub.106))=R.sub.112). Resistor 108 is functionally equivalent to 
resistor R.sub.1 of FIG. 4; and the in-series combination of resistor 112 
and capacitor 111 is functionally equivalent to resistor R.sub.3 of FIG. 
4. The value of resistor 107 has no effect on signal attenuation. 
The other function for resistors 105, 106, and 107 is to supply a DC bias 
to the diodes 109, 110. The diodes 109, 110 are first connected in series; 
and the series combination is connected to resistor 107 in parallel. 
Because resistor 107 has a low resistance value and is in parallel with 
the diodes 109, 110, the voltage drop across the diodes 109, 110 will be 
primarily determined by the resistance of resistor 107. If the current 
flow in resistor 107 is much more than the current flow in the diodes 109, 
110 the voltage drop across the diode 109, 110, will be very stable and 
will be insensitive to the presence or absence of a signal at the input 
port 101. 
The integrated functions of signal attenuation and diode bias supply avoid 
any parasitic effects due to the introduction of additional bias 
circuitry. This permits a high frequency response and a favorable 
impedance match. 
From an DC perspective, resistor 107, in parallel with capacitors 103 and 
104, provides a dissipative circuit to the capacitors 103, 104. Therefore, 
resistor 107 will discharge the accumulated electric charge of connected 
capacitors 103, 104 in every AC cycle. 
Diode 109 is connected to resistor 108 through capacitor 104 while diode 
110 is connected to resistor 108 through capacitor 103. Diode 109 is 
responsible for the RF distortion correction during the negative portion 
of the AC cycle, while the diode 110 has the same function during the 
positive half of the AC cycle. The non-linear current of diode 109 charges 
capacitor 104, and the non-linear current of diode 110 charges capacitor 
103. Due to the configuration of the circuit, the voltage produced on 
capacitors 103 and 104 have the same value but different signs. The small 
resistance from resistor 107 connected to the capacitors 103, 104 
discharges the accumulated electric charge during every AC cycle. As a 
result, there is no voltage drop across the capacitors 103, 104. This 
permits the diode 109, 110 to provide the largest non-linear current for 
the correction purpose. 
The present invention has several unique advantages over the prior art. Due 
to its symmetric structure, the attenuator 100 produces only odd order 
distortion. Consequently, the circuit does not degrade the second order 
performance of an NLD. The attenuator 100 also uses two low series 
resistances 107, 108. From a DC perspective, resistor 107 significantly 
improves the correction efficiency and reduces the susceptibility to 
ambient temperature effects. From an AC perspective, resistor 108 provides 
for distortion correction with low insertion losses. Due to the attenuator 
100 design, the voltage drop across resistor 108 fully loads the diodes 
109, 110 even under non-linear operation of the diodes 109, 110. As a 
result, maximum non-linear current is utilized for correction purposes. 
Finally, proper phasing of the distortion signals is inherent in the 
design, thereby avoiding additional phase circuitry and delay lines. This 
permits a circuit design which is much less complex and results in a 
compact and robust design. 
Table 1 provides a listing of the components shown in FIG. 6. However, one 
skilled in the art would clearly recognize that the values shown in Table 
1 are only for explanatory purposes, and should not be considered to be 
limiting to the invention. For example, the value of resistor 108 may 
range from approximately 2 .OMEGA. to 30 .OMEGA.. Likewise, the value of 
resistor 107 may range from approximately 100 .OMEGA. to 3000 .OMEGA.. 
TABLE 1 
______________________________________ 
VALUE OR 
COMPONENT IDENTIFICATION 
______________________________________ 
102 0.1 .mu.f 
103 0.1 .mu.f 
104 0.1 .mu.f 
105 6 K.OMEGA. 
106 6 K.OMEGA. 
107 330 .OMEGA. 
108 7.5 .OMEGA. 
109 HP HSMS-2822#L30 
110 HP HSMS-2822#L30 
111 0.1 .mu.f 
112 3 K.OMEGA. 
113 0.1 .mu.f 
114 75 .OMEGA. 
115 0.1 .mu.f 
______________________________________ 
As previously described, the attenuator 100 uses the non-linear current 
produced by the diodes 109, 110 to compensate for the voltage compression 
caused by an NLD. As shown, the attenuator 100 comprises capacitance, 
resistance and two diodes. The diodes are the only components that are 
sensitive to temperature change and the only components that require 
correction during operation over a wide temperature range. There are three 
factors which must be taken into consideration when operating the 
attenuator 100 over a wide temperature range: 
1) The diode operating current will change if the bias voltage remains 
constant while the ambient temperature changes. Under the same input 
voltage swing at the input port 101 and the same bias voltage, more 
non-linear diode current will be created as the ambient temperature rises. 
2) When the ambient temperature rises, the diode will produce less 
non-linear correction current for the same input signal voltage and the 
same diode bias current. 
3) NLDs typically exhibit more distortion as the ambient temperature rises. 
Accordingly, a higher diode non-linear current is required for correction 
of the greater distortion. 
All of the temperature effects experienced by the attenuator 100 are 
related to the bias voltage. Some of the effects are additive while others 
are subtractive. However, the result is that for a given temperature, 
there will be an optimum bias voltage to produce the proper correction 
output. Proper temperature correction will be achieved when there is a 
predefined change of bias voltage versus temperature. 
Referring to FIG. 7, the preferred embodiment of the temperature 
compensation circuit 200 is shown. The temperature compensation circuit 
200 controls the bias of the diodes 109, 110 (shown in FIG. 6) for optimum 
compensation of the distortion. As shown, the temperature compensation 
circuit 200 comprises two transistors 206, 213; a capacitor 216; nine 
resistors 201, 202, 203, 204, 207, 209, 210, 214, 215; two diodes 205, 
208; and a negative temperature coefficient thermistor 211. 
The negative temperature coefficient thermistor 211 is coupled in parallel 
with resistor 210 to form a linearized resistance, which is correlated to 
a change in temperature. A PNP transistor 206 provides a constant current 
source through its collector to the linearized resistor combination 210, 
211. The constant current provided by the PNP transistor 206 induces a 
linearized voltage change across the resistor combination 210, 211 as the 
temperature changes. By adjusting the value of the variable resistor 202, 
the amount of constant current through the PNP transistor 206 can be 
changed. Therefore, the voltage swing over temperature can be changed. The 
constant current also passes through the variable resistor 209, thereby 
creating a constant voltage drop that is used as a starting bias point for 
bias voltage adjustment. By selectively adjusting the resistance of 
resistors 202 and 209, any combination of voltage swing and starting bias 
voltage can be obtained. An NPN transistor 213, which is an emitter 
follower transistor, provides the control bias voltage from line 217 
through line 116 to the attenuator 100, as shown in FIG. 7. The two diodes 
205 and 208 are used to compensate for the junction voltage of the two 
transistors 206, 213 which change over temperature. 
Table 2 provides a listing of the components shown in FIG. 7. However, one 
skilled in the art would clearly recognize that the values shown in Table 
2 are only for example, and should not be considered to be limiting to the 
invention. 
TABLE 2 
______________________________________ 
VALUE OR 
COMPONENT IDENTIFICATION 
______________________________________ 
201 16 K.OMEGA. 
202 3.3 K.OMEGA. 
203 4.7 K.OMEGA. 
204 50 K.OMEGA. 
205 1N4148 
206 2N3906 
207 2 K.OMEGA. 
208 1N4148 
209 1.5 K.OMEGA. 
210 2 K.OMEGA. 
211 DKE 402N10 
212 100 .OMEGA. 
213 2N3904 
214 100 .OMEGA. 
215 3 K.OMEGA. 
216 50 .mu.f 
______________________________________ 
It should be recognized that the present invention provides an instant 
voltage controlled non-linear attenuator design combined with a bias 
supply for optimum non-linear correction efficiency and bias temperature 
stability. Even if the temperature compensation circuit 200 as disclosed 
herein is not utilized, the preferred embodiment of the present invention 
provides adequate distortion correction over a broad temperature range. 
When the temperature compensation circuit 200 is utilized, the distortion 
compensation results can be further improved. Accordingly, a trade off 
between the performance of the compensating circuit and the complexity of 
the circuit must be weighted. 
The present invention provides for correction of odd-order phase 
distortion. Third order distortion is dominant because it has the largest 
amplitude at the output of the RF amplifier; whereas higher odd-order 
distortions decrease in amplitude rapidly, making them less relevant to 
the distortion correction problem. Although the foregoing examples 
describe third order distortion due to the large difference in amplitude 
between third and higher odd-order distortion, the circuit is relevant to 
all odd-order correction and cancellation. 
Referring to FIG. 8, the preferred embodiment of the present invention 
includes a distortion circuit 27 and an amplifier circuit 29 coupled on a 
single printed-circuit (PC) board 23. It should be recognized by those of 
skill in the art that there are many types and configurations of RF 
amplifier circuits that may be utilized in accordance with the teachings 
of the present invention. This distortion circuit is device independent. 
That is, its implementation may occur in a single ended RF amplifier, a 
push-pull amplifier or a power doubler device including pre and post 
amplifiers. The topology is also irrelevant to the implementation of the 
circuit. It may be used in all silicon, all GaAs, or in a combination 
thereof. Implementation for the above selected configuration is made by 
adjusting resistor 108 and the bias voltage applied to the bias control 
point 116 to match the distortion generated by the selected configuration. 
For example, the RF amplifier circuit 29 may be silicon cascode, silicon 
darlington, GaAs cascode or a combination of GaAs and silicon in a cascode 
configuration. 
Power level is also irrelevant as this circuit may appear as a pre- or 
post-amplifier device. In addition, it may be used to correct both RF 
amplifier distortion, optical detector distortion, or a hybrid containing 
both RF and optical circuitry. One RF amplifier circuit 29, which is 
disclosed in U.S. patent application Ser. No. 09/236,175 entitled WIDEBAND 
LINEAR GAAS FET TERNATE CASCODE AMPLIFIER, is herein incorporated by 
reference. 
Coupling of the circuits 27, 29 on a single PC board 23 results in several 
advantages. First, the positioning of the distortion circuit 27 with 
respect to the RF amplifier circuit 29 can be precisely determined during 
manufacture. Once the circuits 27, 29 are deposited onto the PC board 23, 
no adjustments to the circuits 27, 29 to account for a spatial change in 
the location of the respective circuits 27, 29 relative to each other are 
necessary. Second, this eliminates the expense and performance degradation 
introduced, when for example, delay lines are utilized to couple the 
distortion circuit 27 with the RF amplifier circuit 29. 
The physical implementation of the preferred embodiment of the present 
invention is performed using surface mount devices (not shown) on a single 
side 21 of a double sided PC board 23 is shown in FIG. 9. A layer of 
copper is adhered to the second side of the PC board 23. This layer 
comprises a low inductance groundplane 25. As shown, the groundplane 25 is 
further coupled to a heat sink 31. In this manner, the groundplane 25 
provides a convenient means for soldering the PC board 23 onto the heat 
sink 31. 
Referring to FIG. 10, a side view of the PC board 23 is shown. The 
thickness of the PC board 23, the circuits 27, 29 and the groundplane 25 
have been greatly exaggerated for explanation. Due to the proximity of the 
circuits 27, 29 to the groundplane 25, an electric charge may accumulate 
between the circuits 27, 29 and the groundplane 25 as illustrated by the 
distance A. Additionally, an electric charge may accumulate between the 
circuits 27, 29 and the heat sink 31, as shown by the distance B. The 
electric charges that accumulate create parasitic capacitances which 
ultimately degrade the performance of the RF amplifier circuit 29. 
Accordingly, it is paramount to reduce or eliminate these parasitic 
capacitances. 
The PC board 23 includes foil regions for the conductive paths of the 
distortion circuit 27 and RF amplifier circuit 29. The foil template for 
the component side 21 of the PC board 23 is shown in FIG. 11. The foil 
template for the groundplane 25 side of the PC board 23 is shown in FIG. 
12. 
To protect from, and eliminate, excessive loss due to the parasitic effects 
of stray capacitance known to exist at the extended operating frequencies 
of the distortion circuit 27, a portion of the copper groundplane 25 is 
specifically removed under the distortion circuit 27. This is shown in 
FIG. 12 as the cross-hatched region 32. A small area 33 of copper is 
retained for maintaining ground continuity with the selectively configured 
heatsink 31. 
The heatsink 31 for the present invention is shown in greater detail in 
FIG. 13. The heatsink 31 is machined out of a thermally conductive 
material to complement the dimensions of the PC board 23 and the 
groundplane 25 foil pattern. The heatsink 31 can be any material which is 
thermally conductive, has a low electrical resistance and which includes 
at least one solderable surface. The top surface 35 of the heatsink 31 is 
selectively configured for matching correspondence with the distortion 
circuit 27 and the RF amplifier circuit 29 foil regions of the groundplane 
25. A small area 39 of the top surface 35 of the heatsink 31 under the 
distortion circuit 27 provides support and attachment for the PC board 23, 
and also provides an additional ground path for the groundplane 25 small 
area 33, (shown in FIG. 12). 
Two indented areas 37a, 37b on opposite sides of the top surface 35, define 
external cover 43 attachment areas of the PC board 23. This enables a 
cover 43 to protect the PC board 23 as shown in FIGS. 14 and 15 and 
protect against short circuits. Two external mounting holes 41 in the 
heatsink 31 permit attachment of the heatsink 31 to an internal thermal 
heatsinking surface of an environmental enclosure (not shown). 
Referring to FIG. 16, an alternative embodiment of the present invention is 
shown. In this embodiment, an additional non-conductive insert 34, such as 
an additional piece of PC board or a ceramic insert with a low dielectric 
constant, is inserted under the distortion circuit 27. This increases the 
distance C between the distortion circuit 27 and the groundplane 25 and 
the distance D between the distortion circuit 27 and the heatsink 31. As 
these distances are increased, the accumulation of charges, and the 
resulting parasitic capacitances, are significantly reduced. 
Referring to FIG. 17, the circuit layout 50 of the preferred embodiment of 
the distortion circuit 27 of the present invention is shown. The layout, 
or spatial relationship, between the components which comprise the 
distortion circuit 27 is critical. It should be noted that path C-C' and 
D-D' must be equal for efficient cancellation of unwanted distortion which 
may be introduced by the circuitry 27, such as second order harmonics or 
second order beats produced by the diodes 109, 110 and for efficient 
cancellation of the third order products of the RF amplifier circuit 29. 
Capacitor 111 and resistor 112 are required for efficiently matching the 
distortion circuit 27 with the RF amplifier circuit 29. 
The transmission line A-A' provides for bandwidth adjustment and interstage 
matching. It is desirable to keep the length of the transmission line from 
A to A' as short as possible to reduce the insertion losses of the 
distortion circuit 27. It should also be noted that the distortion circuit 
27 is symmetric about the transmission line A-A'. This ensures proper 
operation of the distortion circuit 27 and eliminates any undesired 
operating characteristics or unwanted distortions from being introduced by 
the distortion circuit 27. Finally, the distance between point B, which is 
the output of the diodes 109, 110, and the resistor 108 should be kept as 
short as possible to maintain cancellation at the highest frequencies. If 
this distance is too long, it could introduce a phase shift which will 
ultimately decrease the amount of distortion generated by the distortion 
circuit 27. 
Referring to FIGS. 18A and 18B, the advantages of the present invention can 
be clearly shown with an improvement in the CTB and X-mod distortion 
output by the RF amplifier circuit 29. As shown in FIG. 18A, a dramatic 
improvement in the reduction of CTB distortion can be seen in the 200-540 
MHz in an RF amplifier coupled to the distortion circuit 27 in a manner in 
accordance with the present invention. Furthermore, it should be noted 
that correction occurs across the entire bandwidth. Referring to FIG. 18B, 
an improvement in the amount of X-mod distortion is shown from the 90-640 
MHz range; and is particularly dramatic in the 300-540 MHz range when the 
predistortion circuit 27 is coupled to an RF amplifier. Again, significant 
improvement can be seen across the entire bandwidth. 
The preferred embodiment has been described using surface mount devices 
with an integral heatsink. Other construction methods may be employed 
adhering to the system and method of the claimed invention. While the 
present invention has been described in terms of the preferred embodiment, 
other variations which are within the scope of the invention as outlined 
in the claims below will be apparent to those skilled in the art.