Auto balancing duplexer for communication lines

A circuit for generating a control signal for use in a communication line duplexer or other isolation means. The transmitted signal at the output of the duplexer is phase detected to detect its real and imaginary components. These components are used to modulate the transmitted signal which is then injected into a feedback loop of the duplexer. This substantially cancels the transmitted signal at the output of the duplexer. The circuit permits the communication line to be terminated in a constant impedance.

BACKGROUND OF THE INVENTION 
1. Field of the Invention 
The invention relates to isolation means particularly those employed on 
bidirectional telephone or communication lines for isolating the 
transmitted signal from the received signal. 
2. Prior Art 
In recent years, because of judicial and regulatory decisions, many 
companies are now manufacturing equipment which connects directly to the 
public telephone network in the United States. The equipment so connected 
must meet certain specifications, for example, those involving isolation 
and impedance matching. 
One of these requirements which is particularly significant for 
modulator/demodulator units (modems) is that the telephone line must be 
terminated in a constant impedance. Typically, in the United States, this 
impedance is 600 ohms resistive with little or no reactive components. 
This impedance is, at least in theory, equal to the line impedance. 
However, in practice, the line impedance is seldom 600 ohms and often 
contains a reactive component (capacitive or inductive) which varies 
between 100 ohms to 1.2k ohms. 
A duplexer is generally used to separate or isolate the transmitted signal 
from the signal received on the bidirectional telephone lines. In FIG. 1, 
a typical duplexer is shown within the dotted line 46 and will be 
discussed along with the improvement to the duplexer provided by the 
present invention. 
In modems and other devices, transmission and reception frequently occur at 
the same time when they are used in full duplex mode. Since the 
transmitted signal is generally substantially larger than the received 
signal, circuitry must be provided to prevent the transmitted signal from 
being detected as the received signal or interfering with the proper 
detection of the received signal. By way of example, 300 baud modems in 
the answer mode transmit at 2025 Hz to 2225 Hz and receive at 1070 Hz to 
1270 Hz. In the originating mode, they transmit at 1070 Hz to 1270 Hz and 
receive at 2025 Hz to 2225 Hz. These frequency bands are relatively close, 
making it difficult to provide complete isolation. When the line impedance 
is 600 ohms, and the duplexer terminates the line in 600 ohms, the 
duplexer provides satisfactory isolation in conjunction with reasonable 
bandpass filters. However, in practice, because of the variations in line 
impedance, a duplexer may only provide 10 db or less of isolation, thus 
increasing the amount of filtering required as described below. 
Filters are typically used to provide further isolation. These filters 
limit the bandwidth of the transmitted frequency, and likewise, limit the 
bandwidth of the received signal. However, unless quite expensive filters 
are used, the "skirts" of these filters overlap, thus in some cases some 
of the transmitted signal passes through the filters (and duplexer) to the 
receiver where it may be detected as a received signal or interfere with 
the proper detection of the received signal. 
The problem of compensating for telephone line impedance fluctuations is an 
old problem, and numerous circuits have been proposed to compensate for 
such impedance variations. The prior art known to Applicant does not 
provide the terminating of the telephone line in a constant impedance. In 
some cases, a conjugate match to the line is made; obviously in these 
cases the line is not terminated in a pure resistive load (e.g., 600 
ohms). The prior art known to Applicant is U.S. Pat. Nos. 4,103,118; 
4,096,362; 3,982,080; and 3,178,521. Other related art which uses a 
passive network and manual balancing is described in U.S. Pat. Nos. 
3,496,292 and 2,186,006. 
As will be seen, the present invention provides an improved duplexer which 
includes a circuit for compensating for line impedance variations. The 
line is terminated in a pure resistive load with the described invention. 
With the use of the present invention less expensive filters may be 
employed, by way of example, in modems, since better isolation is obtained 
from the duplexer. 
SUMMARY OF THE INVENTION 
A circuit for providing a control signal for an isolation means to 
automatically reduce the magnitude of the transmitted signal at the 
receiver in the event of line impedance variations, is described. The 
circuit is particularly useful in a telephone communications system 
employing the isolation means in connection with a bidirectional telephone 
line. 
A phase detection means detects the real and imaginary components of the 
transmitted signal which appear at the received signal output of the 
isolation means. This phase detection uses the transmitted signal as a 
reference signal. The output of the phase detection means is coupled to a 
modulation means. The modulation means modulates the real and imaginary 
components derived from the transmitted signal, to provide a control 
signal for the isolation means. The control signal provides substantial 
cancellation of the transmitted signal at the output of the isolation 
means.

DETAILED DESCRIPTION OF THE INVENTION 
A circuit for providing automatic cancellation of the transmitted signal at 
the output of a duplexer is described. In the following description, 
numerous specific details, such as specific circuit components, are set 
forth to provide a thorough understanding of the present invention. It 
will be understood by one skilled in the art, however, that the invention 
may be practiced without these specific details. In other instances, 
well-known circuits are shown in block diagram form in order not to 
obscure the present invention in unnecessary detail. 
Referring to FIG. 1, a prior art duplexer (enclosed within dotted line 46), 
comprising the differential amplifier 10 and resistors 12, 13, and 14 is 
illustrated. The bi-directional telephone line 11 is connected to one 
input terminal of the amplifier 10. The output of the amplifier provides 
the received signal on line 15. This received signal is coupled to 
resistors 13 and 14 and provides a feedback signal to the amplifier from 
node 23. The transmitted signal, T.sub.x, (line 16) is coupled to the 
telephone line 11 and the non-inverting input of the amplifier 10 through 
the resistor 12. In the United States the resistor 12 is typically 600 
ohms, and it provides the termination resistance for the line 11. (The 
amplifier driving line 16 has a very low output impedance and may be 
considered a voltage source, the amplifier 10 a very high input impedance, 
thus line 11 is terminated in a pure resistive load of 600 ohms. The 
resistors 13 and 14 are of equal value, thus one-half of the output signal 
on line 15 is fed back to the amplifier 10. Moreover, one-half of T.sub.x 
is coupled to the inverting terminal of amplifier 10 through these 
resistors. However, the resistors 13 and 14 may be unequal if more or less 
gain in the receive channel is desired, without affecting the automatic 
balancing. 
Assume that the line impedance is precisely 600 ohms (resistive). For this 
case, precisely one-half of the transmitted signal is coupled to the 
non-inverting input terminal of the amplifier 10. The resistors 13 and 14 
also divide this transmitted signal in half, and this divided signal is 
coupled to the inverting input terminal of amplifier 10. For these 
conditions the difference signal between the inputs of amplifier 10 is 
zero, thus none of the transmitted signal appears on line 15. The received 
signal from line 11 is amplified by the amplifier 10 and coupled to line 
15. (One-half of this signal is fed back into the inverting input terminal 
of amplifier 10.) Therefore, when the line impedance is equal to the 
resistance of resistor 12, transmission and reception can simultaneously 
occur without the transmitted signal being sensed on line 15. 
If the line resistance is other than 600 ohms, the balanced bridge 
condition described above is not met and some of the transmitted signal 
will appear on line 15. One way of correcting this imbalance is to 
dynamically change resistor 12 to match the line impedance. This implies 
that both the resistive and reactive components of the line impedance be 
conjugatively matched by a variable impedance replacing resistor 12. This, 
however, cannot be done if the requirement to terminate line 11 in a 
constant resistive load is to be met. 
With the present invention, the component of transmitted signal due to line 
impedance mismatch which appears on line 15 is compared with the 
transmitted signal on line 16 within the automatic balance controller 20. 
The results of this comparison are used to generate an AC current whose 
magnitude and phase are of the proper values such that when it is injected 
into node 23 it effectively cancels the residue of the transmitted signal 
present on line 15. 
Before describing the presently preferred embodiment of the present 
invention, reference is made to FIG. 2 to describe the method implemented 
by the present invention. Block 25 illustrates that the component of 
transmitted signal appearing on line 15 is phase detected with reference 
to the transmitted signal (line 16). This, in effect, yields a DC value 
which represents the real component of the residue of the transmitted 
signal. This DC voltage then modulates the amplitude of the transmitted 
signal (block 26). 
The transmitted signal (line 16) is also shifted in phase by 90.degree. as 
indicated by block 24. This phase shifted signal is then used to detect 
the imaginary component of the transmitted signal present in the 
transmitted signal residue on line 15 (R) as indicated by block 27. The 
resulting DC voltage which represents the imaginary component of the 
residual transmitted signal on line 15, is used to modulate the phase 
shifted transmitted signal (block 28). 
The results of the modulation are used to generate a control signal for the 
duplexer as indicated by block 29. This control signal, in the presently 
preferred embodiment as indicated in FIG. 1, is injected into the feedback 
loop associated with the duplexer, that is, into the bridge network, i.e., 
at node 23. 
The mathematical analysis necessary to show that the phase detection of the 
real and imaginary components, along with the modulation, yields a signal 
which will cancel the effects of the transmitted signal at the output of 
the duplexer is extremely complicated. This analysis is not presented here 
since it is not necessary to practice or understand the present invention. 
Referring now to FIG. 3 and the presently preferred embodiment, the 
transmitted signal, line 30, is coupled to a duplexer (shown within dotted 
line 46) through the filter 31, capacitor 77, and buffer 32. The received 
signal from the duplexer is coupled to an automatic gain control (AGC) 
circuit shown within dotted line 48. The received signal after being 
coupled through an RC filter is connected to a filter 58 with the output 
of this filter (line 59) being coupled to a receiver 61. The filters 31 
and 58, as discussed in the Prior Art Section of this application, may be 
used to prevent the transmitted signal from being sensed on line 59. 
However, to assure that substantially none of the transmitted signal is 
sensed on line 59, these bandpass filters must have precise 
characteristics and are typically quite expensive. In the presently 
preferred embodiment, relatively inexpensive filters (capacitive-switch 
filters) are used for filters 31 and 58. The present invention provides 
substantially better isolation through the duplexer making the more 
expensive filters unnecessary. 
The duplexer again includes an amplifier 41 with the output coupled to the 
resistor 44. The inverting terminal of this amplifier receives the 
transmitted signal, after this signal is divided across the resistors 43 
and 44. The bidirectional telephone line 45 is terminated in a pure 
resistive load 42, shown as 600 ohms. 
While not necessary to practice the present invention, in the presently 
preferred embodiment, an AGC circuit is used to provide a more constant 
level of received signal. A voltage controlled amplifier 51 receives the 
output of the duplexer at its noninverting terminal. The output of the 
amplifier 51 (line 60) is connected to a detector 55. The output of this 
detector, after coupling through amplifier 52, controls the gain of the 
amplifier 51. Line 60 is also coupled through resistor 75 and amplifier 53 
to provide a DC feedback signal to the inverting terminal of the amplifier 
51. Line 60, which contains both the received signal and the residue of 
the transmitted signal, is coupled to the phase detectors 62 and 64. The 
detector 62 receives the transmitted signal from line 34. This signal is 
also coupled to the multiplier 66. 
The transmitted signal from line 34 is capacitively coupled through 
capacitor 35 to the inverting input terminal of amplifier 36. The output 
of amplifier 36 is fed back through resistor 37 to this input terminal. 
The capacitor 35, along with the amplifier 36 and its feedback, 
effectively differentiate the transmitted signal on line 34, and thus 
provide a phase shift to the transmitted signal of approximately 
90.degree. (line 39). (Note a circuit which effectively integrates T.sub.x 
may be used since it also provides a phase shift of approximately 
90.degree.). The phase shifted signal on line 39 is connected to the 
detector 64 and also to the multiplier 68. 
The output of the phase detectors 62 and 64 are connected to one input 
terminal of the multipliers 66 and 68, respectively. The DC signal from 
these detectors are smoothed by the capacitors 70 and 72. The outputs of 
the multipliers 66 and 68 are connected to the node 50. The current 
outputs of these multipliers are effectively summed at node 50. The 
current injected into node 50 is of the proper magnitude and phase to 
effectively null the transmit signal at node 60 for a wide range of 
resistive and reactive line mismatches. 
Commercial components may be used to fabricate the entire circuit of FIG. 
3. However, in the preferred embodiment, those portions of the circuit 
shown within dotted lines 48 and 82 have been realized as a custom 
integrated circuit, with the exception of the capacitors 70, 72, 80 and 
81. 
The operation of the phase detectors 60 and 64 may be best described with 
reference to specific examples. Assume that the signal applied to a 
detector from line 34 is A.sub.1 cos .omega.t. (A.sub.1 is substantially 
larger than A.sub.2.) For this case, the DC output from the detector would 
be proportional to A.sub.2. If the signal applied on line 39 is A.sub.1 
sin .omega.t and the signal on line 60 is -A.sub.2 sin .omega.t, then the 
DC output would be proportional to -A.sub.2. If the signal applied on line 
34 or 39 is A.sub.1 sin .omega.t and the signal applied on line 60 is 
A.sub.2 cos .omega.t, then the output of the detector would be zero. 
The phase detector 62, since its detection is based on an unshifted T.sub.x 
(line 34) effectively detects the real component of the residue of the 
transmitted signal on line 60. The detector 64, since its detection is 
based on T.sub.x, shifted by .+-.90.degree., detects the imaginary 
component of the residue of the transmitted signal present on line 60. 
The output of the detector 62 is used to modulate the amplitude of the 
unshifted transmitted signal by means of the multiplier 66. Similarly, the 
output of the detector 64 is used to modulate the amplitude of the 
90.degree. phase shifted transmitted signal by the multiplier 68. The 
current outputs of these multipliers are summed at node 50 with this 
output injected into the inverting terminal of the amplifier 41 in such a 
manner as to minimize the residual transmitted signal on line 60. 
The detectors 62 and 64 synchronously detect the residue of T.sub.x present 
at the output of the duplexer. Since these detectors are driven in 
quadrature, the outputs of the detector represent the real and imaginary 
components of the residue signal. (The AGC circuit has little effect on 
this residue since the residue is typically much smaller than the received 
signal. The amplitude of this residue is in the linear portion of the AGC 
circuit for practical purposes.) The summed current signals from the 
multipliers 66 and 68 when injected into the feedback path of the duplexer 
operates to null the residue of the transmitted signal from the duplexer. 
Thus, a circuit has been described which greatly enhances the performance 
of an isolation means such as a duplexer. The circuit provides 
compensation for variations in line impedance and effectively cancels the 
residue of the transmitted signal at the receiver. Unlike prior art 
circuits, the telephone line remains terminated in a fixed impedance. The 
circuit is particularly useful in modems since it eliminates the need for 
expensive bandpass filters.