Mixing stage, modulator circuit and a current control circuit

A mixing stage includes a first modulation stage that receives an input signal from a first common node of the mixing stage, a first local oscillator input that receives a local oscillator signal, and a first modulation signal output adapted to provide a first modulated signal. A second modulation stage of the mixing stage includes a second input that receives a phase inverted representation of the input signal from a second common node of the mixing stage, a second local oscillator input that receives the local oscillator signal, and a second modulation signal output adapted to provide a second modulated signal. A current generation circuit provides a supply current to the first common node and to the second common node. A current control circuit is adapted to superimpose an offset current to the current of at least one node of the first common node and the second common node.

FIELD

Embodiments relate to a mixing stage, a modulator circuit for providing a single-side band signal using a mixing stage and a current control circuit for a mixing stage.

BACKGROUND

Mixing stages or signal mixers are used in various implementations, for example in communication systems. In those applications, mixing stages may be used to mix or superimpose a baseband or intermediate frequency signal to a carrier frequency prior to the submission or sending of the generated modulated carrier frequency signal. Examples of those applications are sending or receiving stages of mobile telecommunication handsets or base stations, terrestrial radio senders and the like. Generally, mixing stages are used in multiple applications where an information signal is to be transmitted or received by means of wireless or wired transmission techniques.

One particular quality criterion for a mixing stage is the achieved signal quality, for example in terms of a spectrum of the signal provided at an output of the mixing stage. For example, it may be required that a modulator circuit for providing a single-side band modulated signal (SSB) provides a spectrum showing only the single transmitted side band without spectral components of the carrier frequency or the complementary side band. To this end, Hartley Modulators are sometimes used, inherently providing carrier suppression and suppression of one of the two side bands due to its concept. Further, mixer stages such as for example Gilbert Cells are sometimes used, which may also provide for an inherent suppression of the carrier or local oscillator (LO) frequency. Those type of mixer stages or modulators are, therefore, also called balanced devices.

Mixing stages typically comprise multiple semiconductor devices, for example bi-polar transistors or field-effect transistors, which have slightly different characteristics due to process fluctuations. Due to those differences within the participating components and also due to fluctuations within the signals used as an input to the modulation stages, also balanced devices may require some additional circuitry in order to counterbalance the imbalances present.

However, additional balancing circuitry may be costly in terms of area and production costs, in particular when the imbalances shall be counterbalanced with a high accuracy. The cost and complexity of such balancing circuitry should be reduced.

SUMMARY

Embodiments may achieve a reduction in the complexity of balancing circuits for a mixing stage comprising two modulation stages in that a current control circuit is coupled to a common node between an input of the first modulation stage and a current generation circuit and to a second common node between a second input to the second modulation stage and the current control circuit. The current control circuit is adapted to superimpose an offset current to the current of at least one node of the first common node and the second common node. That is, an offset current may be superimposed directly to a supply current or to the input signal at a node between the current generation circuit and the input of the modulation stages. This may decrease the complexity of the current control circuit and, hence, result in cost and area saving of the mixing stage as well as in an increase in the reliability of the mixing stage.

According to some embodiments, a current control circuit for superimposing an offset current to a current provided to an information signal input of a mixing stage makes use of a current mirror circuit adapted to provide the output current. To this end, the current mirror circuit is coupled to the output terminal of the current control circuit. A digital-to-analog converter is coupled to the current mirror circuit such that a variation of the output voltage of the digital-to-analog converter causes a variation of the output current of the current mirror circuit. Using a current mirror circuit to provide a current to be superimposed to the information signal input of a mixing stage may allow to directly couple the current mirror circuit to an input of a modulation stage since an output of the current control circuit has a high impedance so that a current superimposes to the input of the modulation stage without flowing into another component of a mixing stage.

A modulator circuit according to further embodiments comprises two embodiments of mixing stages to provide for the possibility of deriving or creating a single-side band modulated signal. That is, the modulator circuit comprises a first mixing stage and a second mixing stage. The outputs of the mixing stages are combined, i.e. the modulator circuit further comprises a signal combiner. The signal combiner is coupled to a first mixing stage output of the first mixing stage as well as to a second mixing stage output of the second mixing stage. The first mixing stage output is coupled to the first modulation signal output and to the second modulation signal output of the first mixing stage and the second mixing stage output is coupled to the first modulation signal output and to the second modulation signal output of the second mixing stage.

That is, the signal combiner is coupled to each of the modulation signal outputs of both mixing stages in order to be able to combine signals having a contribution of each of the modulation signal outputs of the two mixing stages. By superimposing, e.g. adding, the signals of the first mixing stage output and the second mixing stage output, a resultant signal may be provided in which one of the side bands generated by the mixing of signals within the mixing stages may be suppressed to some extent. Using embodiments of mixing stages for the modulator may also provide for the possibility of controlling the current control circuits of the mixing stages such that the contribution of a signal oscillating with the local oscillator frequency is decreased.

DETAILED DESCRIPTION

Various example embodiments will now be described more fully with reference to the accompanying drawings in which some example embodiments are illustrated. In the figures, the thicknesses of lines, layers and/or regions may be exaggerated for clarity.

Accordingly, while example embodiments are capable of various modifications and alternative forms, embodiments thereof are shown by way of example in the figures and will herein be described in detail. It should be understood, however, that there is no intent to limit example embodiments to the particular forms disclosed, but on the contrary, example embodiments are to cover all modifications, equivalents, and alternatives falling within the scope of the disclosure. Like numbers refer to like or similar elements throughout the description of the figures.

FIG. 1illustrates an embodiment of a mixing stage10. The mixing stage10comprises a first modulation stage (MOD1)20and a second modulation stage (MOD2)30. The first modulation stage20comprises an input22adapted to receive an input signal, a first local oscillator (LO) input24adapted to receive a local oscillator signal oscillating with a predetermined local oscillator frequency and a first modulation signal output26adapted to provide a modulated signal. The modulated signal depends on the local oscillator signal and on the input signal. Modulation stages or modulators of that kind are, for example, used to modulate a signal containing information onto a carrier frequency in mobile telecommunication applications. The modulated radiofrequency (RF) signal is then amplified and fed to radiators of associated antenna systems. An ideal modulator modulating an input signal having a frequency fifand a local oscillator frequency flocreates an output signal having a spectrum peaking at the frequencies flofifand flo+fif.

However, in practical implementations, frequency components corresponding to floand fifare also present within the modulated signals at a signal output of the modulators. Balanced modulators try to partly alleviate this imperfection by using redundant circuitry and differential signals such that DC-offsets of the input signal and of the local oscillator signal cancel at least partly at a differential output of the balanced mixer. For a similar purpose, the mixing stage10ofFIG. 1further comprises the second modulation stage30having a second input32, a second local oscillator input34and a second modulation signal output36. The second input32is adapted to receive a phase-inverted representation of the input signal.

The first input22of the first modulation stage20is coupled to a first common node40aof the mixing stage10and the second input32of the second modulation stage30is coupled to a second common node40bof the mixing stage10. The first common node40aand the second common node40bare coupled to an information signal input50of the mixing stage10which, therefore, provides for the possibility of operating the mixing stage10with a differential or complementary input of the information signal. I.e. a differential signal may be modulated onto the local oscillator frequency in different branches of the circuit.

The embodiment of a mixing stage10further comprises a current generation circuit60for providing a supply current, the current generation circuit60being coupled to the first common node40aand to the second common node40b.The provision of a supply current to the first common node40aand to the second common node40bmay be required in order to provide a working point for the first modulation stage20and the second modulation stage30. That is, the current generation circuit60provides a DC-current to which a current of the signal provided at the information signal input50superimposes at the first common node40aand at the second common node40bto be used at the first input22and the second input32of the current controlled modulation stages20and30.

The mixing stage10further comprises a current control circuit100coupled to the first common node40aand to the second common node40b.The current control circuit100is adapted to superimpose an offset current to the current of at least one node of the first common node40aand the second common node40b.That is, a current may be additionally superimposed to the current at the first common node40aor at the second common node40bor to both common nodes40aand40bsimultaneously. This, in turn, may provide for the possibility of balancing the signal at the first modulation signal output26with respect to the signal at the second modulation signal output36. Balancing may be used to compensate for asymmetries in the layout. Further, balancing may be used to compensate for variations within the characteristics of the semiconductor devices constituting the first modulation stage20and the second modulation stage30.

Superimposing a current to at least one of the first common node40aand the second common node40bmay provide for the possibility of suppressing signal components of the local oscillator signal in the modulated signal determined by using the first modulation signal output26and the second modulation signal output36as a differential output of the mixing stage10. As a general rule, self-biasing or asymmetry in the modulator may lead to a significant contribution of a signal with the local oscillator frequency at the output of the mixing stage10. Correspondingly, an imperfect local oscillator signal creates strong signal components with a frequency corresponding to the frequency of the input signal and its phase-inverted representation as provided to the information signal input50. In other words, self-biasing or a deviation of the duty cycle of the local oscillator signal (LO-signal) leads to or could introduce imbalances in the overall circuit. Imbalances result in a limited suppression of the LO-signal itself (LO-leakage) or induce the presence of information signal components within the modulated signals. Non-idealities within the information signal at the information signal input50lead to limited suppression of the local oscillator signal within the modulation signal outputs26and36.

The current control circuit100, however, may avoid the presence of a local oscillator signal component within the first modulated signal and the second modulated signal by injecting or superimposing an offset current to at least one of the first common node40aor the second common node40bso as to counterbalance any imbalances or so as to introduce an artificial imbalance to achieve a high suppression of the local oscillator signal LO within the output signal.

In other words, the current control circuit100may be used to achieve an effective LO-suppression of the mixing stage10, i.e. an efficient or high suppression of the presence of a component with the local oscillator frequency in the output signal of the mixing stage10.

By superimposing the current directly to the first common node40aor to the second common node40b,the complexity of the current control circuit100may be reduced as compared to conventional approaches where an injection of an additional current or the superposition of a current is performed within the current generation circuit60. If the offset current superimposed by the current circuit is modified in finite quantities, the granularity may be decreased. That is, only a comparatively small amount of different possible currents need to be generated by the current control circuit as compared to conventional approaches performing a superposition of an additional current by means of the current generation circuit60.

An example for such a conventional approach is illustrated inFIG. 3. The current generation circuit240of the conventional approach ofFIG. 3comprises two transistors242aand242bas a current source. According to the conventional approach, a current at the inputs of the first and second modulation stages220and230is modulated by a modulation of the current of the base terminals of the current sources242aand242b.That is, the intermediate frequency input250of the Gilbert cell200is coupled to the base terminals of the respective transistors242aand242b.Additional balancing of the input currents of the modulation stages220and230is achieved by an additional offset voltage applied to the base terminals of the transistors242aand242bby means of a conventional voltage control circuit260including a Digital to Analog Converter. Due to the amplification of the transistors, however, the conventional current control circuit260may require a rather high resolution.

When the current is superimposed directly to the inputs of the modulating stages20and30as according to the embodiment ofFIGS. 1 and 2, a smaller resolution may be sufficient in order to achieve a comparable result. That is, mixing stages according to some embodiments may use circuitry with a significantly lower complexity. This may also translate into cheaper devices requiring less semiconductor area and production costs as well as providing for a better reliability or long-term stability due to a decrease in complexity.

According to further embodiments, the current control circuit100according toFIG. 1is adapted to modify the offset current in finite quantities. According to some embodiments, for example,FIG. 2, the current control circuit100comprises a digital-to-analog converter to control the offset current in an efficient manner. The resolution of the digital-to-analog converter may be significantly smaller as compared to conventional approaches and, for example, be less than 10 bits or even below, for example 4, 5, 6, 7, 8 or 9 bits. Further embodiments, however, may also use another arbitrary number of bits. A lower resolution of the digital-to-analog converter may be sufficient due to direct superposition of the offset current to either one or both of the common nodes40aor40b.According to some embodiments, the current control circuit100is adapted to superimpose a first offset current to the common node40aand a complementary offset current to the second common node40bso as to superimpose a current of the same magnitude to the first common node40aand to the second common node40b.To this end, a complementary signal or current shall be understood as a current which has the same magnitude but opposite phase. Generally speaking, a complementary signal, be it current or voltage, shall be understood to be a signal of equal amplitude but with opposite phase.

According to some embodiments, the current generation circuit60is adapted to support the direct superposition of an offset current to the first common node40aand to the second common node40b.To this end, the current generation circuit60comprises two independent current sources, i.e. a first current source coupled to the first common node40aand a different second current source coupled to the second common node40b.

With respect toFIG. 1it may also be noted that a mixing stage as illustrated therein may be used for both, up-mixing of a signal or down-mixing of a signal. In either case, the signal comprising information to be processed by the mixing stage is provided to the information input signal. In the up-mixing application, as for example within a sending amplifier of a mobile telecommunication device, the signal provided to the information signal input may be the intermediate frequency signal, while the local oscillator signal may be a signal oscillating with the desired carrier frequency used for transmission of the radiofrequency signal. To this end, some of the following embodiments may denote the information signal input as an intermediate frequency (IF) input, when up-mixing scenarios are described.

To the contrary, when down-mixing is performed, the radiofrequency signal, as for example received over a wireless transmission, is provided to the information signal input. In that event, the radiofrequency signal comprises the information to be processed or to be reconstructed. Similarly, the local oscillator frequency signal would correspond to the carrier frequency and a signal component having the intermediate frequency may be derived at a modulation signal output.

To this end, further embodiments of mixing stages supporting down-mixing applications may comprise an impedance matching circuit coupled between the current control circuit100and the first common node40aand the second common node40b.The impedance matching circuit is operable to increase an input impedance of the current control circuit100so as to avoid current from leaking into the current control circuit when the radiofrequency signal is provided to the information signal inputs in the down-mixing application. This may avoid current leakage into the current control circuit itself when the same has an inherently low input impedance at high frequencies.

According to further embodiments of mixing stage10, the first modulation stage20is a balanced mixing stage and also the second modulation stage30is a balanced mixing stage. Balanced mixing stages may provide LO-suppression capabilities, i.e. suppression of a signal component oscillating with the local oscillator frequency within the signal provided at the first modulation signal output26and at the second modulation signal output36. Hence, the leakage of local oscillator signal components may be reduced further or, the requirement to superimpose offset currents at the first common node40aand at the second common node40bmay be reduced. This may result in current control circuits with lower resolution. Mixing stages according to those embodiments may also be denoted as double-balanced mixing stages since they have inherent balancing capabilities with respect to the information signal as well as with respect to the local oscillator signal.

A practical implementation of a double-balanced mixing stage according to an embodiment is illustrated inFIG. 2. The application is designed for up-mixing of the signal provided at the information signal input50, which may hence also be denoted as intermediate frequency signal input. The embodiment ofFIG. 2comprises the basic components ofFIG. 1so that the following description of the embodiment ofFIG. 2will partly rely on the description ofFIG. 1and, hence, only differences will be briefly discussed.

As already said, the mixing stage10ofFIG. 2is a double-balanced mixing stage and, hence, the first modulation stage20and the second modulation stage30are themselves balanced in order to provide inherent low-signal suppression at the first modulation signal output26of the first modulation stage20and at the modulation signal output36of the second modulation stage30. The information signal input50is operable to receive an input signal as an information signal as well as a phase-inverted representation of the input signal so as to allow a balanced or differential mode of operation. According to further embodiments, however, the information signal input may also be operable to receive a single input signal and the modulation stage10itself may be operable to derive the phase-inverted representation of the input signal by means of appropriate circuitry. Since both of the modulation stages20and30are balanced, the local oscillator input supports a differential input of the local oscillator signal and of a phase-inverted representation of a local oscillator signal. To this end, the first local oscillator input24comprises a first terminal24aadapted to receive the local oscillator signal and a second terminal24badapted to receive a phase-inverted representation of the local oscillator signal. Equivalently, the second modulation circuit comprises a second local oscillator input34having a first terminal34aand a second terminal34b.

In order to provide for a differential output allowing for balancing the signal, the first modulation signal output26comprises a first node26aand a second node26b.The first modulation stage20provides a first modulated sub-signal depending on the local oscillator signal and on the input signal at the first node26aand a second modulated sub-signal depending on the phase-inverted representation of the local oscillator signal and on the input signal at the second node26b.In the particular example embodiment ofFIG. 2, the first modulated sub-signal at the first node26ais generated by means of a first transistor28aand the second modulated sub-signal at the second node26bis generated by means of a second transistor28b.The emitters of the transistors28aand28bare coupled to the common node40a,and hence, receive the current of the input signal as provided by the information signal input50. The base terminal of the first transistor28ais controlled by the local oscillator signal24aand the base terminal of the second transistor28bis controlled by the phase-inverted representation of the local oscillator signal. The collector terminals of the first transistor28aand the second transistor28bare coupled to the first node26aand to the second node26b,respectively. Further, load-resistors29aand29bare associated with the transistors28aand28band coupled between the emitter of the transistors28aand28aand the first common node40a.The particular implementation of the modulation stages20and30is based on bi-polar NPN-transistors inFIG. 2. Further embodiments may also use other implementations, as for example PNP-transistors, field-effect transistors (FET) or the like. Since the emitter current of each of the transistors20aor20bis modulated by or corresponding to the current of the input signal while the base current is, at the same time, modulated by the local oscillator signal, the current at the first node26aand the second node26bessentially corresponds to a multiplication of the two currents. In particular, the frequencies of alternating current signals are added within the signal provided at the first node26aand the second node26b.By subtracting the signals at the first node26aand the second node26b,contributions of the local oscillator signal may principally be suppressed to some extent.

The second modulation stage30relies on the same principles so that the components within the second modulation stage30are only enumerated shortly. The second modulation stage30comprises a third transistor38aand a fourth transistor38bas well as two resistors29aand29b.The modulation signal output36furthermore comprises a third node36aand a fourth node36b.The third node36aprovides a third modulated sub-signal depending on the local oscillator signal and on the phase-inverted representation of the input signal. The fourth modulated sub-signal provided at the fourth node depends on the phase-inverted representation of the local oscillator signal and on the phase-inverted representation of the input signal.

In order to achieve suppression of DC components within the information signal input, the nodes26a,26b,36band36aare cross coupled to one another resulting in a subtraction of the respective signal components due to the fact that output nodes which depend on phase-inverted inputs are coupled to each other so that the signals sum up. In particular, node26adepending on the information signal is coupled to36bdepending on the phase inverted representation of the input signal. To this end, a mixing signal output110of the mixing stage10comprises a first terminal110aand a second terminal110b.The first terminal110ais coupled to the first node26aand to the fourth node36bwhereas the second terminal110bis coupled to the second node26band to the third node36a.

In order to allow for the direct superposition of an offset current by means of a current control circuit100, the current generation circuit60comprises two independent current sources controlled by means of a common current mirror62. That is, a first current source64ais coupled to the first common node40aand a second, different current source64bis coupled to the second common node40b.In the particular implementation ofFIG. 2, the first current source64ais formed by means of a further transistor66ahaving its base terminal controlled by a current mirror62and the second current source64bis formed by the equivalent transistor66bhaving its base current also controlled by a current mirror62. The embodiment ofFIG. 2provides for a high input impedance of the current control circuit100as well as for a high input impedance for the current generation circuit60. To this end, a current may be directly superimposed to the first common node40aand to the second common node40bby the current control circuit100and, hence, serve to additionally balance the output of the first modulation stage20with respect to the second modulation stage30if necessary.

The doubly-balanced mixing stage as illustrated inFIG. 2is sometimes also called a Gilbert cell. Hence, embodiments may also be denoted as Gilbert-cells having a current control circuit100coupled to a first common node40abetween a current generation circuit60and an input to a first modulation stage20as well as to a second common node40bbetween the current generation circuit60and an input to the second modulation stage30.

FIG. 3shortly illustrates a conventional approach as to how additional balancing of the signal at the output of a Gilbert cell can be achieved. In the conventional approach ofFIG. 3, the current at the inputs of the first and second modulation stages220and230is modulated by a modulation of the current of the base terminals of the current sources242aand242b.That is, the intermediate frequency input250of the Gilbert cell200is coupled to the base terminals of the respective transistors242aand242b.Additional balancing of the input currents of the modulation stages220and230is achieved by an additional offset voltage applied to the base terminals of the transistors242aand242bby means of a conventional voltage control circuit260, including a Digital to Analog converter. Due to the exponential characteristics of the transistors, however, the conventional current control circuit260may require a much higher resolution as compared to embodiments to achieve a fine tuning of the balanced current. Hence, its implementation complexity may be much higher than the complexity of the corresponding current control circuit100of an embodiment.

FIG. 4illustrates a possible application of a mixing stage according to an embodiment. In particular,FIG. 4illustrates a Hartley-modulator for generating a single-side band modulated signal. In particular the up-conversion of an intermediate frequency or information signal310is schematically illustrated inFIG. 4. The Hartley-modulator is also denoted as I/Q-modulator, since the individual signals to be mixed, i.e. the carrier frequency signal or LO-signal312and the intermediate frequency signal310or information input signal are used both as an in-phase and as a quadrature component. The single-side band modulator comprises a first mixing stage320ain the in-phase path330aand a second mixing stage320bin the quadrature-path330b.A first input to the first mixing stage320in the I-path is the local oscillator signal312without phase shift and the second input to the first mixing stage320is the intermediate frequency signal310without a phase shift. The second mixing stage320b,however, receives a phase-shifted representation of the local oscillator signal as well as a phase-shifted representation of the intermediate frequency signal. In particular, the local oscillator signal is phase-shifted by −90° with respect to the local oscillator signal312in the I-path330a.The intermediate frequency signal is phase-shifted by −90° at an input of the second modulator320b.Since both of the mixing stages320aand320bgenerate output signals ideally comprising frequencies at flo+fifand flo−fif, the output signals as provided by the mixing stages320aand320bhave particularly beneficial phase relations of those two signal components with respect to each other. In particular, the upper frequency component or the upper image having the frequency flo+fifof the Q-path330B is phase-shifted by −180° with respect to the same signal component in the I-path. Summing the signals of the I-path330aand of the Q-path330bat an output340of the Hartley-modulator principally cancels the upper side band signal (USB). If modulator stages providing for an inherent carrier suppression, i.e. a suppression of the local oscillator signal, are used, a signal may be derived at an output of the Hartley modulator which only has frequency components at the desired frequency of flo−fif, while the image having flo+fifas well as the leaking component of the local oscillator frequency flois suppressed.

However, due to the inherent disturbances to the signals as well as to the imperfections within the provided local oscillator intermediate frequency signals, both components are normally present within conventional Hartley-modulator implementations. That is, the output spectrum of a Hartley-modulator or a single-side band modulator as illustrated inFIG. 4generally comprises all of those components, which is schematically illustrated inFIG. 5. For the single-side band modulator, the desired signal component, i.e. the desired side band is centered around flo−fif. A portion of the not completely suppressed image, i.e. an image-side band412is centered around flo+fifwhile an undesirable component of the local oscillator signal414is situated at flo. In applications where the carrier frequency is much higher than the intermediate frequency so that the distance between the carrier frequency414and the desired side band410is small, it may be practically impossible and furthermore energy-wasting to apply filters to the output signal so as to try to filter the frequency component at floand beyond.

Use of embodiments of mixing stages within the modulator circuits may provide for the possibility of efficiently suppressing the LO-leakage of the local oscillator frequency414within the single-side band modulated signal.FIG. 6illustrates schematically an embodiment of a modulator circuit800for providing a single-side band modulated signal. The modulator circuit800comprises a first mixing stage810and a second mixing stage820according to an embodiment. The first mixing stage810has a first mixing stage output corresponding to the mixing signal output110ofFIG. 2. The first mixing stage810is used within the I-path330aof the modulator circuit800and the second mixing stage820is used within the Q-path330bof the modulator circuit. Consequently, the first local oscillator signal814of the first mixing stage has a phase relation of 90° with respect to the corresponding local oscillator signal824of the second mixing stage820. The same applies to the intermediate frequency signals or the information signals816of the first mixing stage810with respect to the intermediate frequency or information input signal826of the second mixing stage820.

The modulator circuit800further comprises a signal combiner840adapted to combine the signal of the first mixing stage output812with the signal of the second mixing stage output822to provide a representation of the single-side band modulated signal at an output850of the modulator. That is, the signal combiner may be operable to add the signals of the output of the first mixing stage810and of the second mixing stage820in order to provide a single-side band modulated signal having the signal component of the image side band strongly reduced or, ideally, completely suppressed.

Optional RF buffers832and842may also be placed between the mixing signal output of the mixing stages810and820and the mixing stage outputs812and822, respectively, in order to improve the phase balance of the signals provided to the signal combiner840.

According to the particular embodiment ofFIG. 6, the modulator circuit800furthermore comprises an optional envelope detector860which is coupled to the output850of the signal combiner840. In order to be able to control the first and/or the second mixing stage810and820properly, the single-side band signal analyzer is adapted to determine the presence of a contribution of a signal oscillating with the local oscillator frequency within the signal at the output850of the signal combiner840. The envelope detector860is coupled to the current control circuit818of the first mixing stage810and/or to the current control circuit828of the second mixing stage820. The coupling can be performed by means of a MCU.

FIGS. 7 and 8illustrate to what an extent the presence of a signal component corresponding to the local oscillator frequency or the carrier frequency may be suppressed within the single-side band modulated signal provided by the modulator circuit ofFIG. 6.

To this end, a frequency spectrum of an output of the modulator circuit800ofFIG. 6is illustrated inFIG. 7. The Y-axis illustrates the spectrum starting from 85 GHz and ending at 87 GHz. The carrier frequency, i.e. the frequency of the local oscillator signal is chosen to be 85.5 GHz and the intermediate frequency of the information signal is chosen to be 500 MHz for illustrative purposes.FIG. 7illustrates the performance of the modulator circuit ofFIG. 6without the use of the single envelope detector860. The Y-axis of the spectrum ofFIG. 7illustrates the power at the output850of the modulator circuit800in the event that a sinusoidal signal is provided as an intermediate frequency signal at the information signal inputs of the mixing stages810and820. As illustrated inFIG. 7, the desired lower side band410contains the most power within the output signal. However, also the undesired upper side band image412is clearly visible at a frequency of 86 GHz. Also, a strong LO-leakage is illustrated in image414. That is, a component oscillating with a local oscillator frequency contributes to a rather high fraction to the output power within the signal at the output850of the modulator circuit800. As illustrated inFIG. 7, the contribution of the local oscillator signal in the output spectrum amounts to roughly −18.4 dB.

FIG. 8illustrates the same spectrum in the event that the envelope detector860is operational and appropriately controls the current control circuits818and828, respectively. In particular, the envelope detector860may control digital-to-analog converters (DAC) within the current control circuits to appropriate values. As illustrated inFIG. 8, the contribution of the local oscillator signal414can be decreased by a considerable amount, down to about −56 dB and even lower than the contribution of the undesired upper side band signal, amounting to roughly −51.9 dB.

FIG. 8thus illustrates to what extent the desired carrier frequency or local oscillator frequency suppression may be achieved by using embodiments of mixing stages as disclosed before.

FIG. 9illustrates a down mixing application of an embodiment. That is, further embodiments may be used within a mixing stage used for down-conversion. This application may result in a better DC-offset and a better IP2, when the balance of the circuit is increased by means of an embodiment as illustrated inFIG. 9. When an embodiment is used as a down-converter, the application of the signal to the information signal input50is changed. In the particular embodiment used for down-conversion, the radiofrequency, i.e. the modulated carrier frequency as received by some receive antenna circuits may be applied to the information signal input50and the intermediate frequency is provided at the mixing signal output110of the circuit. This could also be denoted as swapping the radiofrequency and intermediate frequency ports of the mixing stage10ofFIG. 2. In the down-converting configuration ofFIG. 9, additional impedance matching circuits910and920may be present. For example, an impedance matching circuit910may be coupled between the current control circuit100and the first common node40aas well as the second common node40b.The impedance matching circuit may serve to increase the input impedance of the current control circuit100, when the inherent input impedance of the current control circuit100is not sufficiently high at the high carrier frequency as opposed to the lower intermediate frequency in the application ofFIG. 2. In the particular example ofFIG. 9, a λ/4thtransmission line is used as an impedance matching circuit in order to prevent current flowing into the current control circuit100rather than into the modulation stages20and30as required. Any other circuitry may also be used as an impedance matching circuit910in order to provide for the functionality, if required. In other words, any kind of radiofrequency-choke (RF-choke) may be placed in series to the output of the current control circuit in order to avoid leaking of the radio frequency signal. For the same purpose, additional impedance matching circuits920may be applied between the output of the current source60and the first common node40aand the second common node40b.

WhileFIG. 9illustrates a current generation circuit60along the lines of the current generation circuit60ofFIG. 2, further embodiments may use a simplified current generation circuit, where the two current sources of the embodiment ofFIG. 9are merged to become a single current source.

FIG. 10illustrates a particular embodiment as to how a current control circuit100may be implemented in order to serve as a current control circuit100within a mixing stage according to an embodiment.

The current control circuit100has an output terminal110for superimposing an offset current to a current provided to an information signal input of a mixing stage. The current control circuit is illustrated in schematic terms in the left illustration ofFIG. 10, while the right illustration gives an example of a practical implementation of an embodiment of a current control circuit100.

The current control circuit has an output terminal110, illustrated as a load inFIG. 10. The output terminal serves for providing an output current for superimposing an offset current to a current provided to an information signal input of a mixing stage, as for example the mixing stage illustrated inFIGS. 1 and 2. The current control circuit further comprises a current mirror circuit120which is adapted to provide the output current, wherein the current mirror circuit120is coupled to the output terminal110. In the particular embodiment ofFIG. 10, a digital-to-analog converter130is coupled to the current mirror circuit120such that a variation of the output voltage of the digital-to-analog converter130causes a variation of the output current of the current mirror circuit120. In the particular implementation ofFIG. 10, it is possible to directly couple an output of the digital-to-analog converter130to a control terminal132or a base terminal of a transistor of the current mirror120. Further embodiments are operable to supply complementary offset currents. To this end, current control circuit100may furthermore comprise a second current mirror circuit140adapted to provide a phase-inverted representation of the offset current at the output110. To this end, a differential digital-to-analog converter130may be used, having a second output coupled directly to a control terminal of a further transistor134of the current control circuit.

For the sake of completeness, an embodiment of a method for balancing a mixing stage having a first modulation stage and a second modulation stage, a current generation circuit adapted to provide a supply current to a first common node and a second common node coupled to the input of the first and second modulation stages is illustrated as a flow chart inFIG. 11.

The method comprises providing an information signal to the first common node and to the second common node at 1000.

The method further comprises superimposing an offset current to the current of at least one node of the first common node and the second common node at 1002, so that the offset current is directly superimposed to the current of the information signal.

Functional blocks denoted as “means for . . . ” (performing a certain function) shall be understood as functional blocks comprising circuitry that is configured to perform a certain function, respectively. Hence, a “means for s.th.” may as well be understood as a “means configured to or suited for s.th.”. A means configured to perform a certain function does, hence, not imply that such means necessarily is performing the function (at a given time instant).