Single-channel and multi-channel coherent demodulation devices with no pilot signal, and corresponding receiving system using a plurality of diversity paths

Coherent demodulation is effected in single-channel and multi-channel coherent demodulation devices without any knowledge of the transmitted signal (i.e. without any pilot signal). The phase shift is estimated by applying a predetermined function to produce a signed value from the argument of a summed signal, the summed signal being itself obtained from the received signal (by quadrature demodulation, complex despreading and summing over N samples). The phase shift estimate is used in a phase-locked loop so that the system converges towards a null error. The residual static phase ambiguity introduced by application of the predetermined function is resolved. A plurality of such single-channel and multi-channel coherent demodulation devices is used in a receiving system using a plurality of diversity paths.

BACKGROUND OF THE INVENTION 
1. Field of the Invention 
The field of the invention is that of digital transmission with frequency 
spreading, and in particular, although not exclusively, CDMA (Code 
Division Multiple Access) transmission. The CDMA technique, which consists 
in multiplying a source signal (included in a common frequency band) by 
means of a specific code, constitutes one application of frequency 
spreading. 
2. Description of the Prior Art 
In transmission systems of the above kind, frequency spreading modulation 
devices are generally used for transmission. They apply to input signals 
(source signals) frequency spreading followed by quadrature modulation to 
obtain the signals to be transmitted. Conventionally (and this applies in 
the remainder of the present description), the input signal or each input 
signal is deemed to have a bit rate D while the signal to be transmitted 
and the received signal each have a bit rate N*D, where N is the spreading 
factor. 
The invention is more precisely concerned with coherent demodulation 
devices of the type for regenerating, from the signals received, the input 
signals of the aforementioned frequency spreading modulation devices. 
In the context of the present invention, only complex frequency spreading 
(corresponding to the use of two spreading sequences in quadrature) is of 
interest. Real spreading (corresponding to the use of a single spreading 
sequence) has an inherent performance handicap such that it is of no 
interest. 
Two families of frequency spreading transmission are generally recognized: 
single-channel transmission: the modulation device receives a single input 
signal to which it applies frequency spreading followed by quadrature 
modulation to generate the signal to be transmitted; and 
multi-channel transmission: the modulation device receives a plurality of 
input signals and multiplies each of them by a separate orthogonal code (a 
Walsh code, for example) to obtain a plurality of channels. It combines 
this plurality of channels onto a single multi-channel signal to which it 
applies frequency spreading and then quadrature modulation to obtain the 
signal to be transmitted. 
In each of these two families, it is possible to distinguish two 
sub-families, respectively corresponding to the situations in which the 
input signal, or each input signal, is real or complex. A complex input 
signal at bit rate D generally results from passing a real signal at bit 
rate 2.D through a 1 to 2 serial/parallel converter. 
The invention has many applications, for example in digital cellular mobile 
radio systems. 
In cellular systems, single-channel transmission is typically used only for 
an uplink (mobile station to base station) channel where the mobile 
station is supposedly satisfied with the existence of a single 
communication channel to the base station. A plurality of mobile stations 
can each transmit in "single-channel mode" in the same frequency band. 
Because they use different spreading sequences or different phases of a 
common spreading sequence the base station can separate out the signals 
transmitted by the various mobile stations. 
Moreover, in cellular systems, multi-channel transmission is typically used 
in the case of a downlink (base station to mobile station) channel where 
the base station has to communicate with a plurality of mobile stations. 
The signal transmitted by the base station is then an aggregate of several 
channels broadcast to all the mobile stations. These channels are 
separated by the use of codes known as "orthogonal" codes in that they 
enable a mobile station receiver to extract the channel addressed to it 
without this being impeded by the presence of other channels. 
However, the current trend in standardizing future CDMA type cellular 
networks is to introduce multi-channel transmission in the uplink (mobile 
station to base station) direction as well. This trend is justified by the 
resulting flexibility (in particular for multimedia applications) and by 
the possibility of adopting coherent demodulation (offering higher 
performance than the non-coherent demodulation that has to be used in 
multi-channel receiving devices). 
The existing demodulation techniques (with their respective drawbacks) are 
described below for each of the two transmission families previously 
mentioned. 
First, it should be remembered that a demodulation (or receiving) device 
has the task or regenerating the input (source) signal or signals from the 
signal that it receives. The signal received corresponds to the signal 
transmitted affected by various disturbances. The type of disturbance of 
interest here is a phase shift. After the received signal is demodulated 
(using two carriers in quadrature), the resulting demodulated signal is a 
complex signal subject to a phase rotation. This rotation corresponds 
precisely to the phase shift. The phase shift is known to be due to the 
propagation medium and to the modulation and demodulation operations (and 
in particular due to the asynchronism of the local oscillators feeding the 
modulator and the demodulator). The phase shift varies in time, i.e. it is 
a dynamic phenomenon. The treatment of the phase shift varies for 
single-channel and multi-channel transmission. 
In the case of single-channel transmission, non-coherent demodulation is 
currently used, which has repercussions for the receiver as well as for 
the transmitter. The principle of non-coherent demodulation is to choose a 
transmitted sequence that can be interpreted at the receiver without 
knowing the phase shift due to the channel. 
Unfortunately, adopting non-coherent demodulation leads to a performance 
handicap. 
In the case of multi-channel transmission, coherent demodulation is 
currently used, which presupposes a knowledge of the varying phase shift 
introduced by transmission, modulation and demodulation. The current 
solution to the problem of acquiring this knowledge consists in dedicating 
one channel to the transmission of a pilot signal. In other words an all 
"one" signal is generally transmitted as one of the input signals. The 
receiver exploits the presence of the pilot signal to estimate the channel 
and in particular to determine the phase shift due to the channel. Once it 
knows this, the receiver can cancel the phase shift. 
Unfortunately, using a pilot signal also reduces system performance. The 
channel carrying the pilot signal is not available for transmitting data. 
Also, the pilot signal channel often has to have a higher power rating 
than a normal channel, in particular if the dynamic variations are fast. 
This surplus transmitted power does not convey any information and so link 
performance is degraded. 
An objective of the invention is to alleviate the various drawbacks of the 
prior art. 
To be more precise, one objective of the present invention is to provide a 
single-channel coherent demodulation device usable in the case of 
single-channel transmission and offering better performance than the 
conventional non-coherent demodulation devices referred to above. 
Another objective of the invention is to provide a multi-channel coherent 
demodulation device usable in the case of multi-channel transmission and 
offering better performance than the conventional coherent demodulation 
devices using a pilot signal referred to above. 
Another objective of the invention is to provide single-channel and 
multi-channel coherent demodulation devices of the above kind adapted to 
estimate and to correct the phase shift induced by the propagation medium 
in particular without using any hypothesis as to the transmitted signals 
and in particular without transmission of any pilot signal. 
A complementary objective of the invention is to provide a receiving system 
using a plurality of diversity paths which retains the advantages 
associated with the (single-channel or multi-channel) devices included in 
the system. 
SUMMARY OF THE INVENTION 
These various objectives, along with others that will become apparent 
hereinafter, are achieved in accordance with the invention with the aid of 
a single-channel coherent demodulation device for regenerating, from a 
received signal, an input signal of a single-channel Nth order complex 
spreading modulation device, said single-channel complex spreading 
modulation device applying to said input signal complex spreading followed 
by quadrature modulation to obtain a signal to be transmitted, said input 
signal having a bit rate D, said signal to be transmitted and said 
received signal each having a bit rate N*D, which device includes: 
quadrature demodulation means generating a demodulated signal at bit rate 
N*D from said received signal; 
complex despreading means generating a despread signal at bit rate N*D from 
said demodulated signal; 
means for summing over N samples generating a summed signal at bit rate D 
from said despread signal; 
a loop for estimating and correcting the phase shift induced in said 
demodulated signal, said loop comprising: 
means for sampling the argument of said summed signal; 
means for applying a predetermined function for moving the argument of the 
summed signal into a range of less than or equal to -.pi./(2x) but less 
than .pi./(2x), where x is equal to 1 or 2 according to whether said input 
signal is real or complex, the result of application of said predetermined 
function to said argument of the summed signal constituting an estimate of 
said phase shift; 
means for subtracting said estimate of the phase shift from the phase of 
said demodulated signal or from that of said despread signal; and 
means for regenerating said input signal from said summed signal, 
themselves comprising means for resolving a residual static phase 
ambiguity induced by application of said predetermined function. 
In the case of single-channel transmission, the general principle of the 
invention therefore consists in using coherent demodulation with no pilot 
signal, which has no effect on the transmitter and enables demodulation 
performance to be improved. It will be remembered that in the prior art 
non-coherent demodulation is used in this situation. 
It is important to note that the coherent demodulation of the invention is 
effected without any knowledge of the transmitted signal. The phase shift 
is estimated by producing a signed value from the argument of the summed 
signal. To be more precise, a predetermined function, itself based on a 
modulo function, is applied to the summed signal to move the argument of 
the summed signal into a range around zero. 
The estimated phase shift is used in a phase-locked loop (for example a 
first order loop). The system therefore converges towards a null error 
(i.e. perfect correction of the phase shift). 
Note that application of the predetermined function leads to a residual 
static phase ambiguity, which must be resolved. However, phase ambiguity 
is a well-known phenomenon in systems without frequency spreading 
(non-CDMA systems), where it is associated with carrier recovery. 
Consequently, existing solutions to the problem of resolving phase 
ambiguity can be applied here. In any event, even if the solution adopted 
to the problem of resolving the phase ambiguity leads to slightly degraded 
performance, performance is only very slightly degraded in comparison to 
the use of non-coherent demodulation. 
The invention also concerns a multi-channel coherent demodulation device 
for regenerating, from a received signal, a plurality of input signals of 
a multi-channel Nth order complex spreading modulation device, said 
multi-channel modulation device multiplying each of said input signals by 
a separate orthogonal code to obtain a plurality of channels, said 
plurality of channels being grouped on a common multi-channel signal, said 
multi-channel signal undergoing complex spreading followed by quadrature 
modulation to obtain a signal to be transmitted, each of said input 
signals having a bit rate D, said signal to transmit and said received 
signal each having a bit rate N*D, said multi-channel coherent 
demodulation device including: 
quadrature demodulation means generating a demodulated signal at bit rate 
N*D from said received signal; and 
complex despreading means generating a despread signal at bit rate N*D from 
said demodulated signal; 
a plurality of processing branches each associated with a given channel 
from said plurality and including: 
means for multiplication of said despreading signal by the orthogonal code 
specific to said given channel to obtain a despread signal specific to 
said given channel at bit rate N*D; and 
means for summing over N samples generating a summed signal specific to 
said given channel at bit rate D from said despread signal specific to 
said given channel; 
a loop for estimating and correcting the phase shift induced in said 
demodulated signal, said loop comprising: 
in each of said processing branches: 
means for sampling the argument of said summed signal specific to said 
given channel; 
means for applying a predetermined function to move the argument of the 
summed signal specific to said given channel into a range of less than or 
equal to -.pi./(2x) but less than .pi./(2x) where x is equal to 1 or 2 
according to whether said input channel of said given channel is real or 
complex, the result of application of said predetermined function to said 
argument of the summed signal specific to said given channel constituting 
an estimate of said phase shift; 
means for averaging estimates of said phase shift supplied by said 
processing branches to obtain an average estimate of said phase shift; and 
means for subtracting said average estimate of the phase-shift from the 
phase of said demodulated signal or from that of said despreading signal; 
and 
in each of said processing branches, means for regenerating said input 
signal of said given channel from said summed signal specific to said 
given channel, themselves comprising means for resolving a residual static 
phase ambiguity induced by application of said predetermined function. 
In the case of multi-channel transmission, the general principle of the 
invention is to use coherent demodulation with no pilot signal. It will be 
remembered that in the prior art coherent demodulation with a pilot signal 
is used in this case. 
The foregoing comments (referring to the single-channel device) on the 
coherent demodulation of the invention apply equally to the multi-channel 
device. 
The multi-channel device of the invention is distinguished from the 
aforementioned single-channel device essentially in that the phase shift 
is estimated on each summed signal specific to a given channel. All these 
estimates are used to calculate an average estimate which is used for 
global correction of the phase of the received signal. A residual static 
phase ambiguity has to be resolved in respect of each summed signal 
specific to a given channel. 
A number of features common to the multi-channel and single-channel 
coherent demodulation devices of the invention will now be described. 
Said predetermined function is preferably written: 
f(.PHI.)=((.PHI.+.pi./(2.x))%(.pi./x))-.pi./(2.x), where % is the "modulo" 
function. 
It will be remembered that x is equal to 1 or 2 according to whether said 
input signal is real or complex. 
Said loop advantageously further comprises at least some of the means from 
the group comprising: 
means for multiplying said estimate of the phase shift or said average 
estimate of the phase shift by a predetermined scalar quantity to adjust 
the dynamic characteristics of said loop; and 
means for integrating said estimate of the phase shift or said average 
estimate of the phase shift over a predetermined time period to obtain a 
cumulative estimate of said phase shift. 
Said input signal or each of said input signals advantageously belongs to 
the group comprising: 
the real input signals at bit rate D (x=1); and 
the complex input signals at bit rate D (x=2), each generated by a 1 to 2 
serial/parallel converter from a real source signal at bit rate 2*D. 
It is clear that in terms of bit rate it is preferable to use complex input 
signals. 
Said means for regenerating the input signal or each of said input signals 
advantageously further comprise at least some of the means belonging to 
the group comprising: 
means for sampling the real part if said input signal is a real signal; 
threshold means; and 
Viterbi decoding means if convolutional encoding means are used by the 
transmitter. 
In one particular embodiment of the invention said means for resolving a 
residual static phase ambiguity comprise differential encoding/decoding 
means. 
Note that this solution entails multiplication of the error rate (by a 
ratio of two or less) leading to slightly degraded performance. However, 
performance is degraded only very slightly in comparison to the use of 
differential (non-coherent) demodulation. 
In an advantageous variant, where convolutional encoding means are used by 
the transmitter and said means for regenerating the input signal comprise 
Viterbi decoding means, said means for resolving a residual static phase 
ambiguity comprise: 
phase shifting means for shifting the phase of the signal at the input of 
said Viterbi coding means by a value chosen from a predetermined set of 
values; and 
means for analyzing the signal at the output of said Viterbi decoding 
means, indicating to said phase shifter means the choice of one of the 
phase shift values according to the result of said analysis. 
This variant applies in particular (but not exclusively) when the 
convolutional code employed is not transparent to phase ambiguities. If 
there is no transparency, the previous solution (differential 
encoding/decoding) is inapplicable. In this case, this non-transparency 
can be used to implement the present variant which is not subject to the 
multiplication of errors phenomenon mentioned above. In the situation 
where the residual phase ambiguity is non-null, analyzing the output 
signal of the decoder (for example comparing the cumulative metric to a 
predetermined threshold) detects if the value chosen for the phase shift 
is the correct one. By trying the various possible phase shift values (by 
a trial and error process), the best value is determined, which amounts to 
resolving the phase ambiguity. 
In this particular embodiment, or in this variant, said predetermined set 
of phase shift values comprises the following values: 
0 and .pi. if said input signal or each of said input signals is a real 
signal; or 
0, .pi./2, .pi. or 3.pi./2 if said input signal or each of said input 
signals is a complex signal. 
The invention also concerns a receiving system comprising at least two 
single-channel or multi-channel coherent demodulation devices of the 
invention each corresponding to a separate diversity path, and, for said 
input signal or for each of said input signals, means for regenerating 
said input signal from summed signals supplied by the summing means of 
each of said diversity paths, 
wherein said regeneration means comprise: 
means for combining said summed signals to obtain a final combined signal 
having a maximum gain; and 
means for resolving the residual static phase ambiguity of said final 
combined signal, induced by the application, on each of said diversity 
paths, of said predetermined function. 
In accordance with the invention, the receiving system first determines a 
combined signal (resulting from the combination of the signals associated 
with the various parts) so that it has a maximum gain and then resolves 
the phase ambiguity of this combined signal. The gain is measured by 
measuring the absolute value, for example. 
It will be remembered that in systems using diversity resolving the phase 
ambiguity is slightly complicated because the ambiguity must be resolved 
on each diversity path. Unless measures are implemented to prevent it, 
there is the risk of combining paths in phase opposition rather than in 
phase. 
Said means for combining the various summed signals advantageously comprise 
n.sub.G groups of means in cascade where n.sub.G =n.sub.S -1, where 
n.sub.S is the number of summed signals to be combined (n.sub.S 
.gtoreq.2), each group of means comprising: 
phase shifter means for shifting the phase of a first of said summed 
signals or of a combined signal at the output of a preceding group of 
means by a value chosen from a predetermined set of values to generate a 
phase shifted summed signal; 
means for adding said phase shifted summed signal to another of said summed 
signals to generate a combined signal; and 
means for controlling said phase shifter means assuring a final choice of 
the phase shift value such that the combined signal has a maximal gain, a 
group of means receiving the combined signal generated by the preceding 
group of means only when said control means of said preceding group of 
means have effected said final choice, 
the combined signal generated by the final group of means constituting said 
final combined signal. 
Thus the first path is combined with the second, after which the third is 
combined with the signal resulting from combining the first two, and so 
on. 
Other features and advantages of the invention will become apparent on 
reading the following description of one preferred embodiment of the 
invention given by way of illustrative and non-limiting example, and from 
the accompanying drawings.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
The various types of frequency spreading transmission will now be briefly 
described with reference to FIGS. 1A through 1D. FIGS. 1A and 1B 
correspond to single-channel transmission, respectively when the input 
signal is real (FIG. 1A) or complex (FIG. 1B). FIGS. 1C and 1D correspond 
to the case of multi-channel type transmission, respectively when the 
input signals are real (FIG. 1C) or complex (FIG. 1D). 
The convention adopted in all the figures is that real signals are 
represented by thin lines and complex signals are represented by thick 
lines. Also, the same item is associated with the same reference number in 
all the figures. 
In the case of single-channel transmission with complex spreading of a real 
signal (FIG. 1A), the frequency spreading modulation device 1.sub.a 
receives a single input signal 2.sub.a assumed to be a digital signal made 
up of digital bit stream at bit rate D. A spread signal 5.sub.a is 
obtained by multiplication (3) of the input signal 2.sub.a by a complex 
signal 4 denoted (Pn+j Qn). Pn and Qn are two pseudo-random bit streams at 
bit rate N*D where N is the spreading factor. The spread signal 5.sub.a, 
which is a complex signal at bit rate N*D, is fed to two inputs of a 
modulator 6 using two carriers in quadrature. The output signal of the 
modulator 6, also at bit rate N*D, constitutes the signal 7.sub.a to be 
transmitted. 
In the case of single-channel transmission with complex spreading of a 
complex signal (FIG. 1B), the frequency spreading modulation device 
1.sub.b differs from that of FIG. 1A only in that the input signal 2.sub.b 
is complex and not real. The complex input signal 2.sub.b at bit rate D 
results (for example) from passing a real source signal 8.sub.b at bit 
rate 2.D through a 1 to 2 series/parallel converter 9. 
In the case of multi-channel transmission with complex spreading of a real 
signal (FIG. 1C), the frequency spreading modulation device 1.sub.c 
receives a plurality of input signals 2.sub.c,0 through 2.sub.c,k . Each 
of these input signals 2.sub.c,0 through 2.sub.c,k is multiplied (10) by a 
separate orthogonal code HO through Hk (a Walsh code, for example) to 
obtain a plurality of channels 11.sub.c,0 through 11.sub.c,k. This 
plurality of channels 11.sub.c,0 through 11.sub.c,k is grouped (12) onto a 
common multi-channel signal 13.sub.c. A spread signal 5.sub.c is obtained 
by multiplication (3) of the multi-channel signal 13.sub.c by a complex 
signal 4 denoted Pn+j Qn. The spread signal 5.sub.c, which is a complex 
signal at bit rate N*D, is fed to two inputs of a modulator 6 using two 
carriers in quadrature. The output signal of the modulator 6, also at bit 
rate N*D, constitutes the signal 7.sub.c to be transmitted. 
In the case of multi-channel transmission with complex spreading of a 
complex signal (FIG. 1D), the frequency spreading modulation device 
1.sub.d differs from that of FIG. 1C only in that each of the input 
signals 2.sub.d,0 through 2.sub.d,k is complex and not real. Each of the 
complex input signals 2.sub.d,0 through 2.sub.d,k at bit rate D results 
(for example) from passing a real source signal from a plurality 8.sub.d,0 
through 8.sub.d,k of such signals at bit rate 2.D through a 
series/parallel converter 9. 
Four embodiments of a coherent demodulation device of the invention, 
respectively corresponding to the four types of transmission of FIGS. 1A 
through 1D, will now be described with reference to FIGS. 2A through 2D. 
In the first embodiment shown in FIG. 2A the single-channel coherent 
demodulation device receives a signal 14.sub.a resulting from the 
transmission through a given propagation medium of the signal 7.sub.a 
output by the modulation device 1.sub.a from FIG. 1A. 
This first embodiment of the coherent demodulation device includes: 
a quadrature demodulator 15 generating a demodulated signal 16.sub.a (at 
bit rate N*D) from the received signal 14.sub.a ; 
complex despreading means 17 generating a despread signal 18.sub.a (at bit 
rate N*D) from the demodulated signal 16.sub.a. This complex despreading 
is effected by multiplication by the complex signal 19, denoted (Pn-j Qn), 
conjugate with the signal 4 used for the spreading; 
means 20 for summing over N samples, generating a summed signal 21.sub.a 
(at bit rate D) from the despread signal 18.sub.a ; 
a phase shift estimation and correction loop (see detailed description 
below). It will be remembered that the phase shift is induced in the 
demodulated signal 16.sub.a in particular by modulation, transmission via 
the propagation medium and demodulation; 
means 22 for regenerating the input signal 2.sub.a from the summed signal 
21.sub.a (see detailed description below). With perfect regeneration the 
regenerated signal 23.sub.a is equal to the input signal 2.sub.a (FIG. 
1A). 
In the example shown in FIG. 2A, the phase shift estimation and correction 
loop comprises: 
means 24 for sampling the argument 25.sub.a of the summed signal 21.sub.a. 
The argument 25.sub.a is in the range of less than or equal to 0 but less 
than 2.pi. 
means 26.sub.a for applying a predetermined function to move the argument 
25.sub.a of the summed signal 21.sub.a into the range of less than or 
equal to -.pi./2 but less than .pi./2. The signed value obtained in this 
way constitutes a raw estimate 27.sub.a of the phase shift. In this case, 
the predetermined function is written (for example): f.sub.a 
(.PHI.)=((.PHI.+.pi./2)%.pi.)-.pi./2, where % is the "modulo" function; 
means 28 for multiplying the raw estimate 27.sub.a of the phase shift by a 
predetermined scalar quantity (loop coefficient) c.sub.a to adjust the 
dynamic characteristics of the loop; 
means 30 for integrating the resulting signal 29.sub.a (at the output of 
the multiplication means 28) over a predetermined duration to obtain a 
cumulative phase shift estimate 31.sub.a ; 
means 32 for subtracting the phase of the demodulated signal 16.sub.a from 
the cumulative estimate 31.sub.a. This subtraction is effected by 
multiplication (33) of the cumulative estimate 31.sub.a by -1 (to obtain 
the opposite of the cumulative estimate) and then multiplication (34) of 
the demodulated signal 16.sub.a by e.sup.j.phi., where .phi. is the 
opposite of the cumulative phase shift estimate 31.sub.a. 
Note that, in one variant, the cumulative estimate 31.sub.a can be 
subtracted from the phase of the despread signal 18.sub.a (and not from 
the phase of the demodulated signal 16.sub.a) as the order of execution of 
the demodulation and despreading operations can be reversed because they 
are linear operations. 
In the example shown in FIG. 2A, the regeneration means 22 comprise: 
means 36 for sampling the real part 37.sub.a of the summed signal 21.sub.a 
; 
means 38.sub.a for resolving a residual static phase ambiguity of the real 
part 37.sub.a of the summed signal 21.sub.a caused by application of the 
function f.sub.a (.PHI.) to the argument 25.sub.a of the summed signal 
21.sub.a. In the present case, the input signal being a real signal, the 
residual static phase shift can be equal to 0 or .pi.. In other words, the 
phase ambiguity is .pi. (see detailed description of these ambiguity 
resolving means below); 
processing means 39 (for example threshold means or Viterbi decoding means) 
for obtaining the regenerated signal 23.sub.a after the phase ambiguity 
has been resolved. Clearly Viterbi decoding can be applied only if 
convolutional encoding means are used by the transmitter. 
One particular embodiment of the means 38.sub.a for resolving a residual 
static phase ambiguity will now be described with reference to FIG. 3. 
This particular embodiment applies in particular if convolutional encoding 
means (not shown) are used at the transmitting end, if the means 22 for 
regenerating the input signal comprise Viterbi decoding means 39, and in 
the situation where the encoding scheme employed is not transparent to 
phase ambiguities. In this particular embodiment, the means 38.sub.a for 
resolving a residual static phase ambiguity comprise: 
phase shifter means 38.sub.a,x for shifting by 0 or by .pi. the phase of 
the signal at the input of the Viterbi decoding means 39; and 
means 38.sub.a,y for analyzing the signal 23.sub.a at the output of the 
Viterbi decoding means 39 indicating to the phase shifter means 38.sub.a,x 
the choice of one of the phase shift values (0 or .pi.) according to the 
result of the analysis. 
The aforementioned analysis (38.sub.a,y) consists, for example, in 
comparing the cumulative metric of the signal 23 to a predetermined 
threshold. If the residual static phase shift is equal to 0, the Viterbi 
decoder operates normally. On the other hand, if the residual phase shift 
is equal to .pi., abnormally high cumulative metrics are observed at the 
output of the Viterbi decoder, higher than in the maximal case of 
disturbance. When this is observed, the phase of the signal fed to the 
input of the Viterbi decoder is modified by introducing a phase shift of 
.pi.. This resolves the residual static phase ambiguity. 
Clearly other types of residual static phase ambiguity resolving means can 
be envisaged that do not depart from the scope of the present invention. 
Accordingly, in one variant, the residual static phase ambiguity resolving 
means comprise differential encoding/decoding means. 
In the second embodiment shown in FIG. 2B, the coherent demodulation device 
is of the single-channel type and receives a signal 14.sub.b resulting 
from the transmission through a given propagation medium of the signal 
7.sub.b output by the modulation device 1.sub.b of FIG. 1B. 
This second embodiment differs from the first embodiment (see FIG. 2A) 
essentially in that: 
the regeneration means 22 do not comprise any means for sampling the real 
part of the summed signal 21.sub.b, because the regenerated signal 
23.sub.b to be obtained is a complex signal. It will be remembered that in 
the case of perfect regeneration the regenerated signal 23.sub.b is equal 
to the input signal 2.sub.b (FIG. 1B); 
in the phase estimation and correction loop, the predetermined function 
(applied by the means 26.sub.b) moves the argument 25.sub.b of the summed 
signal 21.sub.b into the range of less than or equal to -.pi./4 but less 
than .pi./4. The signed value obtained in this way constitutes a raw 
estimate 27.sub.b of the phase shift. In this case, the predetermined 
function is written (for example): f.sub.b 
(.PHI.)=((.PHI.+.pi./4)%.pi./2)-.pi./4, where % is the "modulo" function; 
the means 38.sub.b for resolving a residual static phase ambiguity are 
slightly different from those of FIG. 1A because the input signal is a 
complex signal and so the residual static phase shift can be equal to 0, 
.pi./2, .pi. or 3.pi./2. In other words, the phase ambiguity is .pi./2 (it 
is .pi. for a real input signal). 
FIG. 4 shows one particular embodiment of the means 38.sub.b for resolving 
the residual static phase ambiguity. This embodiment is evidently deduced 
directly from that described hereinabove with reference to FIG. 3. All 
that is required is to replace the set of phase shift values {0, .pi.} 
with the set {0, .pi./2, .pi., 3.pi./2}. 
In the third embodiment shown in FIG. 2C, the coherent demodulation device 
is of the multi-channel type and receives a signal 14.sub.c resulting from 
the transmission through a given propagation medium of the signal 7.sub.c 
output by the modulation device 1.sub.c from FIG. 1C. 
As in the first embodiment (FIG. 2A), a despread signal 18.sub.a is 
generated, after passing through a quadrature demodulator 15 and then 
despreading means 17. 
On the other hand, this third embodiment includes a plurality of branches 
for processing the despread signal 18.sub.a. Each branch is associated 
with a given one of k+1 channels and includes (for the ith branch, for 
example) with 0.ltoreq.i.ltoreq.k): 
means 40.sub.i for multiplying the despread signal 18.sub.c by the 
orthogonal code Hi specific to the channel concerned to obtain a despread 
signal 41.sub.c,i specific to the channel concerned at bit rate N*D; 
means 20.sub.i for summing over N samples, generating a summed signal 
21.sub.c,i specific to the channel concerned, at bit rate D, from the 
despread signal 41.sub.c,i specific to the channel concerned; 
means 22.sub.c,i for regenerating the input signal 2.sub.c,i of the channel 
concerned from the summed signal 21.sub.c,i specific to the channel 
concerned. With perfect regeneration the regenerated signal 23.sub.c,i is 
equal to the input signal 2.sub.c,i (FIG. 1C). 
The aforementioned summing means 20.sub.i and regeneration means 22.sub.c,i 
are identical to those 20 and 22, respectively, of the first embodiment 
and therefore have already been described above, with reference to FIG. 
2A. Note that it is merely for simplicity that the regeneration means 
22.sub.c,i in FIG. 2C do not include the processing means 39 from FIG. 2A. 
In this third embodiment, the phase shift estimation and correction loop 
comprises: 
in each processing branch (for example the ith branch, with 
0.ltoreq.i.ltoreq.k), means 24.sub.c,i for sampling the argument of the 
summed signal 21.sub.c,i specific to the channel concerned and means 
26.sub.c,i for moving the argument of the summed signal 21.sub.c,i 
specific to the channel concerned into the range of less than or equal to 
-.pi./2 but less than .pi./2. The signed value obtained in this way 
constitutes a raw estimate 27.sub.c,i of the phase shift. The means 
24.sub.c,i and 26.sub.c,i are identical to those 24 and 26, respectively, 
of the first embodiment and therefore have already been described above, 
with reference to FIG. 2A; 
means 42 for averaging the raw estimates 27.sub.c,i of the phase shift 
supplied by the various processing branches to obtain an average phase 
shift estimate 43.sub.c ; 
means 28 for multiplying the average estimate 43.sub.a of the phase shift 
by a predetermined scalar quantity (loop coefficient) c.sub.c to adjust 
the dynamic characteristics of the loop; 
means 30 for integrating the resulting signal 29.sub.c (at the output of 
the multiplication means 28) over a predetermined duration to obtain a 
cumulative phase shift estimate 31.sub.c ; and 
means 32 for subtracting the cumulative estimate 31.sub.c of the phase 
shift from the phase of the demodulated signal 16.sub.c (or, in a variant, 
from that of the despread signal 18.sub.c). 
The aforementioned multiplication means 28, integration means 30 and 
subtraction means 32 have already been described above, with reference to 
FIG. 2A. 
In the fourth embodiment shown in FIG. 2D the coherent demodulation device 
is of the multi-channel type and receives a signal 14.sub.d resulting from 
the transmission through a given propagation medium of the signal 7.sub.d 
output by the modulation device 1.sub.d from FIG. 1D. 
This fourth embodiment is deduced from the third embodiment in the same way 
as the second embodiment is deduced from the first embodiment. It 
therefore does not require a specific description. 
One particular embodiment of a receiving system in accordance with the 
invention using a plurality of diversity paths will now be described with 
reference to FIG. 5. It will be remembered that, with a receiving system 
of this kind, there are generally as many sources of static phase 
ambiguities as there are paths. Steps must therefore be taken to avoid the 
risk of combining the various paths in phase opposition. Means specific to 
the invention enabling in-phase combination of the various diversity paths 
are specifically described below. 
The receiving system is a "RAKE" type receiver, for example, which exploits 
the multipath phenomenon to introduce diversity gain. The multipath 
phenomenon is present when the signal received by the receiver has taken 
different paths associated with different electrical time delays. The 
"RAKE" type receiver attempts to realign the various components in time 
and then to combine them "in phase" to obtain a maximum gain. It is 
assumed here that the various diversity paths have already been realigned 
in time. 
In the particular embodiment described here, the receiving system comprises 
three single-channel devices in accordance with the second embodiment 
described above (with reference to FIG. 2B). In other words, there are 
three diversity paths on the single channel, each of the three coherent 
demodulation devices receiving a signal 14.sub.b, 14.sub.b ', 14.sub.b " 
resulting from the transmission through a given propagation medium of the 
signal 7.sub.b output by the modulation device 1.sub.b from FIG. 1B. 
The receiving system therefore comprises three phase-locked loops identical 
to that from FIG. 2B, one for each of the three coherent demodulation 
devices. 
The receiving system also comprises means 50 for regenerating the input 
signal 2.sub.b from the various summed signals S0 through S2 supplied by 
the summing means 20 of each of the three coherent demodulation devices. 
Note that in FIG. 2B the summed signal (to which each of the summed 
signals S0 through S2 corresponds) is denoted 21.sub.b . With perfect 
regeneration the regenerated signal 21 is equal to the input signal 
2.sub.b (FIG. 1A). 
The regeneration means 50 comprise: 
means 52 for combining the various summed signals S0 through S2 in phase to 
obtain a final combined signal 53 having a maximal gain (see detailed 
description below); 
means 54 for resolving the residual static phase ambiguity of the final 
combined signal 53, induced by the application of the predetermined 
function f.sub.b (.PHI.) on each of said diversity paths; and 
means 55 (for example of the threshold or Viterbi decoder type) for 
processing the signal output by the means 54 for resolving the phase 
ambiguity of the combined final signal to obtain the regenerated signal 
51. 
In the FIG. 5 embodiment, the combination means 52 comprise n.sub.G (=2) 
groups G1, G2 of means in cascade. Generally, n.sub.G =n.sub.S -1, where 
n.sub.S is the number of summed signals to be combined (n.sub.S =2 in the 
example shown). Each group G1, G2 of means comprises: 
phase shifter means 56 for shifting by 0, .pi./2, .pi. or 3.pi./2 the phase 
of the first summed signal S0 (first group G1) or of a combined signal 
S.sub.c present at the output of one of the preceding groups of means 
(second group G2, for example) to generate a phase shifted summed signal 
S.sub.d ; 
means 57 for adding the phase shifted summed signal S.sub.d with another 
one of the various summed signals to generate a combined signal S.sub.c ; 
and 
means 58 for controlling the phase shifter means 56 to assure a final 
choice of phase shift value (0, .pi./2, .pi. or 3.pi./2) such that the 
combined signal S.sub.c has a maximal gain (for example, a maximal 
absolute value after summation (57)). This can be done by trial and error. 
The combined signal S.sub.c generated by the last group G2 of means 
constitutes the combined final signal 53. 
In operation, a group G1, G2 of means receives the combined signal S.sub.c 
generated by the preceding group of means only when the control means 58 
of the preceding group of means have made their final choice. 
To summarize, with the first group G1, the first and second summed signals 
S0, S1 are combined. Then, with the second group G2, the previous 
combination (of the first and second summed signals S0, S1) is combined 
with the third summed signal S2. 
The principle can therefore be generalized to a larger number of summed 
signals (i.e. a larger number of paths). At each step, path n is added to 
the sum of paths n-1 assigned a static phase shift chosen in the range {0, 
.pi./2, .pi., 3.pi./2}. 
Additionally, it is a simple matter to go from the above example (where the 
input signal is a complex signal) to the situation where the input signal 
is a real signal. It is sufficient to replace the set of phase shift 
values {0, .pi./2, .pi., 3.pi./2} with the set {0, .pi.}. 
From the foregoing description it is clear that the invention can also be 
generalized to the situation in which the receiving system comprises any 
number (.gtoreq.2) of coherent demodulation devices with no pilot signal. 
Similarly, the coherent demodulation devices in the receiving system can be 
implemented in accordance with any of the other three embodiments 
described above (with reference to FIGS. 2A, 2C and 2D, respectively). 
Note that for generalization to the situation in which the coherent 
demodulation devices are of the multi-channel type (third and fourth 
embodiments, FIG. 2C and FIG. 2D), the explanations given with reference 
to FIG. 5 (for the situation in which the coherent demodulation devices 
are of the single-channel type) must be applied to each channel. Path 
diversity applies in the multi-channel situation to each of the channels.