Sample rate change between single-carrier and multi-carrier waveforms

A Baseband transmitter for a radio frequency (RF) communication device according to an embodiment of the present invention includes a single-carrier processor, a pulse shape filter, a multi-carrier processor and a signal combiner. The single-carrier processor generates a single-carrier signal at a first sample rate. The pulse shape filter includes multiple polyphase filters which filter the single-carrier signal according to a time shaping pulse that approximates a multi-carrier power spectrum and a sample switch that selects outputs of the polyphase filters at a second sample rate. The multi-carrier processor generates a multi-carrier signal at the second sample rate. The signal combiner combines the filtered single-carrier signal with the multi-carrier signal while maintaining phase, gain, frequency and timing alignment.

FIELD OF THE INVENTION

The present invention relates to wireless communications, and more particularly to a wireless Baseband transmitter configured to communicate using a mixed carrier signal with sample rate change between single-carrier and multi-carrier waveforms.

BACKGROUND OF THE INVENTION

The Institute of Electrical and Electronics Engineers, Inc. (IEEE) 802.11 standard is a family of standards for wireless local area networks (WLAN) in the unlicensed 2.4 and 5 Gigahertz (GHz) bands. The current IEEE 802.11b standard defines various data rates in the 2.4 GHz band, including data rates of 1, 2, 5.5 and 11 Megabits per second (Mbps). The 802.11b standard uses direct sequence spread spectrum (DSSS) with a chip rate of 11 Megahertz (MHz), which is a serial modulation technique. The IEEE 802.11a standard defines different and higher data rates of 6, 12, 18, 24, 36 and 54 Mbps in the 5 GHz band. The FCC has also approved a modified version of 802.11a to run in a licensed band near 6 GHz. It is noted that systems implemented according to the 802.11a and 802.11b standards are incompatible and were not designed to work together.

A new IEEE standard is being proposed, referred to as 802.11g (the “802.11g proposal”), which is a high data rate extension of the 802.11b standard at 2.4 GHz. It is noted that, at the present time, 802.11g is only a proposal and is not yet a completely defined standard. Several significant technical challenges are presented for the new 802.11g proposal. It is desired that the 802.11g devices be able to communicate at data rates higher than the standard 802.11b rates in the 2.4 GHz band. In some configurations, it is desired that the 802.11b and 802.11g devices be able to coexist in the same WLAN environment or wireless area without significant interference or interruption from each other, regardless of whether the 802.11b and 802.11g devices are able to communicate with each other. Thus, it is desired that 802.11g be backwards compatible with 802.11b devices. It may further be desired that the 802.11g and 802.11b devices be able to communicate with each other, such as at any of the standard 802.11b rates.

An impairment to wireless communications, including WLANs, is multi-path distortion where multiple echoes (reflections) of a signal arrive at the receiver. Other types of interferences, such as different and incompatible wireless signal types, may cause problems with WLAN communications. The Bluetooth standard, for example, defines a low-cost, short-range, frequency-hopping WLAN. Systems implemented according to the Bluetooth standard present a major source of interference for 802.11-based systems. Both the single-carrier systems and multi-carrier systems include equalizers that are designed to combat various types of distortion. The equalizers are typically designed to use the preamble to achieve good receiver acquisition. One proposal to implement 802.11g is a mixed mode configuration including a single-carrier segment with a preamble and header and a multi-carrier segment with a payload. The traditional multi-carrier system, however, was not designed to utilized the information obtained from a single-carrier preamble. Losing all information when transitioning from single-carrier to multi-carrier is not desirable in the presence of multi-path distortion or other types of interference.

There are also several potential problems with the signal transition between single- and multi-carrier signals, particularly with legacy equipment. The transmitter may experience analog transients (e.g., power, phase, filter delta), power amplifier back-off (e.g. power delta) and power amplifier power feedback change. The receiver may experience Automatic Gain Control (AGC) perturbation due to power change, spectral change, multi-path effects, loss of channel impulse response (CIR) (multi-path) estimate, loss of carrier phase, loss of carrier frequency, and loss of timing alignment.

A mixed waveform configuration for wireless communications was previously disclosed in U.S. Provisional Patent Application entitled, “Wireless Communication System Configured to Communicate Using a Mixed Waveform Configuration”, Serial No. 60/306,438 filed on Jul. 6, 2001, which is incorporated by reference in its entirety. The system described therein reused the equalizer information obtained during acquisition of the single-carrier portion of the signal. The technique provided continuity between the single-carrier and multi-carrier segments (e.g., orthogonal frequency division multiplexing or OFDM), which was achieved by specifying the transmit waveform completely for both the single-carrier and multi-carrier segments and specifying the transition. The waveform enabled continuity between the two signal segments, including AGC (power), carrier phase, carrier frequency, timing and spectrum (multi-path). It was contemplated that the signal would not have to be reacquired by the multi-carrier portion of the receiver since the information developed during the single-carrier portion (preamble/header) was valid and used to initiate capture of the multi-carrier portion. However, particular receiver architectures were not discussed.

A mixed carrier wireless architecture has been previously disclosed in U.S. Provisional Patent Application entitled, “Single-Carrier to Multi-Carrier Wireless Architecture”, Ser. No. 60/325,048 filed on Sep. 26, 2001, which is incorporated by reference in its entirety. The wireless architecture described therein is capable of communicating using the proposed mixed carrier waveform configuration. The term “mixed carrier” refers a combined signal with a single-carrier portion followed by a multi-carrier portion. The transmitter could be configured to operate in multiple operating modes including single-carrier, mixed carrier and multi-carrier modes. Furthermore, several receiver architectures were described that are configured to receive a mixed carrier signal and resolve the Baseband signals incorporated in the mixed carrier signal.

A Baseband transmitter and receiver architecture according to one embodiment of the prior disclosure achieves coherency across the single-carrier to multi-carrier transition by maintaining gain, phase, frequency, sample timing and Channel Impulse Response (CIR) from the single-carrier signal to the multi-carrier signal of a mixed carrier signal. In this manner, the signal does not have to be reacquired by the multi-carrier portion of the receiver since the information developed during the single-carrier portion is valid and used to initiate capture of the multi-carrier portion. Maintaining and accumulating information makes the signal much more robust in the face of common interferences experienced in wireless communications. A Baseband receiver architecture according to an alternative embodiment was also described that does not preserve the coherency across the transition, so that the multi-carrier portion of the receiver must completely re-acquire the signal after the transition. A multi-carrier preamble may be used for this purpose. Yet another non-coherent receiver embodiment was disclosed that utilizes selected information gained from the single-carrier portion of the waveform, such as any selected parameter associated with gain, phase, frequency or timing. Although the non-coherent architectures are less robust than the coherent configurations, the non-coherent options may be easier and cheaper to implement while remaining sufficiently robust to achieve a suitable communication system for many applications.

A technical challenge of the mixed carrier transmitters is rate changing either or both of the single-carrier and multi-carrier signals so that they may be combined in a coherent manner. Several rate changing techniques are described herein.

SUMMARY OF THE PRESENT INVENTION

A Baseband transmitter for a radio frequency (RF) communication device according to an embodiment of the present invention includes a single-carrier processor, a pulse shape filter, a multi-carrier processor and a signal combiner. The single-carrier processor generates a single-carrier signal at a first sample rate. The pulse shape filter includes multiple polyphase filters which filter the single-carrier signal according to a time shaping pulse that approximates a multi-carrier power spectrum and a sample switch that selects outputs of the polyphase filters at a second sample rate. The multi-carrier processor generates a multi-carrier signal at the second sample rate. The signal combiner combines the filtered single-carrier signal with the multi-carrier signal while maintaining phase, gain, frequency and timing alignment.

The time shaping pulse may be sampled and decomposed into the polyphase filters of the pulse shape filter in accordance with the second sample rate. In one configuration, a selected number of polyphase filters is used to up-sample the single-carrier signal to an intermediate rate, where the time shaping pulse is sampled based on the intermediate rate, where each of the polyphase filters has a selected number of taps that incorporate tap coefficients based on samples of the time shaping pulse, and where the sample switch selects outputs of the plurality of polyphase filters to down-sample to the second sample rate. Alternatively, a selected number of polyphase filters is used to up-sample the single-carrier signal to the second sample rate, where the time shaping pulse is sampled according to the second sample rate, where each of the polyphase filters has a selected number of taps that incorporate tap coefficients based on samples of the time shaping pulse, and where the sample switch selects each output of the plurality of polyphase filters at the second sample rate. In a specific example of the latter case, the first sample rate is 11 megahertz (MHz), the second sample rate is 44 MHz, the time shaping pulse is converted to discrete samples based on a 44 MHz rate, and 11 polyphase filters are used with 9 taps each.

A first of the polyphase filters may be selected to have a center tap having a coefficient that corresponds with a peak magnitude of the sampled time shaping pulse. In this case, the signal combiner of the Baseband transmitter may further include a combiner and a soft switch. The combiner combines the filtered single-carrier signal with the multi-carrier signal and provides a combined mixed carrier signal. The soft switch selects the filtered single-carrier signal until a last sample is completed and selects the combined mixed carrier signal during a transition period. Furthermore, the last sample of the single-carrier signal at the first sample rate is positioned at the center tap of the first polyphase filter at the beginning of the transition period.

The Baseband transmitter may further include a rate change filter. The multi-carrier processor generates a multi-carrier signal at a third sample rate rather than the second sample rate. In one embodiment, the rate change filter converts the multi-carrier signal from the third sample rate to the second sample rate. The rate change may include an internal rate change filter that converts a sample rate of the multi-carrier signal from the third sample rate to a fourth sample rate, and a first-in, first-out (FIFO) buffer that converts the multi-carrier signal from the fourth sample rate to the second sample rate. Alternatively, the rate change filter includes multiple polyphase filters that up-sample to an intermediate frequency and a sample switch that selects outputs of the plurality of polyphase filters at the second sample rate. In the latter case, a selected number of polyphase filters of the rate change filter may be used to up-sample the multi-carrier signal to the intermediate frequency, where each of the polyphase filters has a selected number of filter taps with coefficients to incorporate a low pass filter (LPF) based on the intermediate frequency, and where the sample switch selects outputs of the plurality of polyphase filters to down-sample the multi-carrier signal to the second sample rate. In a specific embodiment, the third sample rate is 20 MHz, the second sample rate is 44 MHz, 11 polyphase filters are used with 21 taps each, and the sample switch selects every 5thoutput of the 11 polyphase filters to achieve a 44 MHz rate. 10

A center tap of a first polyphase filter of the rate change filter may be selected to have a largest magnitude coefficient. The polyphase filters of the rate change filter may each be initialized with a cyclic extension of a first multi-carrier symbol. The signal combiner may further include a phase multiplier, a combiner and a soft switch. The phase multiplier multiplies the multi-carrier signal by a phase based on the single-carrier signal and provides a rotated multi-carrier signal. The combiner combines the filtered single-carrier signal and the rotated multi-carrier signal and provides a combined mixed carrier signal. The soft switch selects the filtered single-carrier signal until completed, selects the combined mixed carrier signal during a transition period, and selects the rotated multi-carrier signal at the end of the transition period until completed. The single-carrier signal may include consecutive chips according to a predetermined timing interval and where the transition period has a duration equivalent to the predetermined timing interval. The time shaping pulse may be sampled and decomposed into the polyphase filters of the pulse shape filter, where a center tap of a first of the polyphase filters is selected to have a coefficient that corresponds with a peak magnitude of the sampled time shaping pulse. In this case, the combiner is operated so that a last chip of the single-carrier signal is located at the center tap of the first of the polyphase filters of the pulse shape filter at the beginning of the transition period. Also, the combiner is operated so that a first full sample of the multi-carrier signal is located at the center tap of the first polyphase filter of the plurality of polyphase filters of the rate change filter at the end of the transition period.

A method of generating a mixed carrier packet for RF transmission according to an embodiment of the present invention includes generating a single-carrier segment including a preamble and header according to a single-carrier modulation scheme at a first sample rate, filtering the single-carrier samples according to a time shaping pulse that approximates a multi-carrier power spectrum and selecting filtered samples at a second sample rate, generating a multi-carrier payload using a selected multi-carrier modulation scheme that provides multi-carrier samples at the second sample rate, and combining the single-carrier segment with the multi-carrier payload to provide a mixed carrier packet in such a manner to maintain gain, phase, frequency and timing.

The combining may include selecting the single-carrier filtered samples, combining the filtered single-carrier filtered samples with the multi-carrier samples during a transition period in such a manner to maintain gain, phase, frequency and timing, and selecting the multi-carrier samples.

The filtering the single-carrier samples may include up-sampling to an intermediate rate using a plurality of finite impulse response (FIR) filters each having multiple taps with coefficients selected according to discrete samples of the time shape pulse based on the intermediate rate. In one embodiment, the intermediate rate may be equal to the second sample rate and the selecting filtered samples may include selecting each output of the plurality of FIR filters at the second sample rate. Alternatively, the intermediate rate is greater than the second sample rate. In this latter case, the selecting filtered samples may include down-sampling outputs by selecting outputs of the plurality of FIR filters to achieve the second sample rate. In another embodiment, the method may further include selecting the coefficients so that a center tap of a first FIR filter has a coefficient that corresponds with a peak value of the time shaping pulse. In this case, the combining the single-carrier segment with the multi-carrier payload may include selecting an output of the first FIR filter when a last sample of the single-carrier segment is positioned at the center tap of the first FIR filter at the beginning of a transition period between the single-carrier segment and the multi-carrier payload of the mixed carrier packet.

The generating a multi-carrier payload may include providing multi-carrier samples at a third sample rate rather than the second sample rate. In one case, the method may include rate change filtering the multi-carrier samples from the third sample rate to a fourth sample rate, and buffering the multi-carrier samples using a first-in, first-out (FIFO) buffer and selecting outputs of the FIFO buffer at the second sample rate.

If the generating a multi-carrier payload includes providing multi-carrier samples at a third sample rate rather than the second sample rate, the method may alternatively include rate change filtering the multi-carrier samples from the third sample rate to the second sample rate. Also, the rate change filtering the multi-carrier samples may include up-sampling to an intermediate frequency using a plurality of FIR filters each having a number of taps determined by the intermediate frequency and tap coefficients selected according to an LPF based on the intermediate frequency. The method in this case may further include selecting a center tap of a first FIR filter of the plurality of FIR filters to have a largest magnitude tap coefficient. The method may further include initializing filter taps of a rate change filter with a cyclical extension of a first multi-carrier symbol of the multi-carrier samples. The combining the single-carrier segment with the multi-carrier payload may include selecting an output of the first FIR filter of the rate change filter when a first full sample of the first multi-carrier symbol of the multi-carrier payload is positioned at the center tap of the first FIR filter of the rate change filter at the end of a transition period between the single-carrier segment and the multi-carrier payload of the mixed carrier packet.

Continuing in this latter embodiment, the filtering the single-carrier samples may include up-sampling using a plurality of FIR filters each having a number of taps with tap coefficients selected based on discrete samples of the time shape pulse. In this case, the method further includes selecting the tap coefficients of the plurality of FIR filters used for up-sampling so that a center tap of a first FIR filter has a coefficient that corresponds with a peak value of the time shaping pulse, and where the combining the single-carrier segment with the multi-carrier payload includes selecting an output of the first FIR filter when a last sample of the single-carrier segment is positioned at the center tap of the first FIR filter at the beginning of the transition period. The combining may further include rotating the multi-carrier payload by a phase determined from the filtered single-carrier segment. The single-carrier modulation scheme may be the Barker modulation and the multi-carrier modulation scheme may be orthogonal frequency division multiplexing (OFDM), where the rotating includes rotating an OFDM multi-carrier payload by a phase of a last Barker Word of the filtered single-carrier segment. In this case, the combining may include ramping the filtered single-carrier segment down while ramping the multi-carrier payload up during the transition period.

DETAILED DESCRIPTION OF EMBODIMENTS OF THE INVENTION

The wireless devices described herein operate in the 2.4 Gigahertz (GHz) band in 802.11b or 802.11g modes or in any of several bands (multi-band) in one or more 802.11a modes, such as 2.4 GHz, 5 GHz, 6 GHz or any other suitable band. The devices may be configured in any suitable format, such as any type of computer (desktop, portable, laptop, etc.), any type of compatible telecommunication device, any type of personal digital assistant (PDA), or any other type of network device, such as printers, fax machines, scanners, hubs, switches, routers, etc. It is noted that the present invention is not limited to the 802.11g proposal, the 802.11b standard, the 802.11a standard or the 2.4, 5 and 6 GHz frequency bands, although these standards and frequencies may be utilized in certain embodiments. The wireless devices may be configured to communicate with each other at any of the standard 802.11b rates, including 1, 2, 5.5 and 11 Mbps to maintain backwards compatibility with 802.11b devices. The wireless devices may also be configured for mixed carrier mode operation to enable communicates at different or higher data rates using a mixed signal configuration according to any one of several embodiments, such as the standard 802.11a data rates of 6, 9, 12, 18, 24, 36, 48 or 54 Mbps.

The mixed signal devices may operate and/or coexist in the same wireless operating area as 802.11b devices without significant interference with each other even while operating in mixed signal mode. The mixed signal devices are illustrated using Barker Word modulation as the single-carrier modulation scheme and orthogonal frequency division multiplexing (OFDM) as the multi-carrier modulation scheme. It is appreciated, however, that the present invention is not limited to any particular modulation schemes and that alternative single-carrier and multi-carrier modulation schemes may be used.

FIG. 1is a conceptual diagram of a mixed signal packet101implemented according to an embodiment of the present invention. The packet101includes a single-carrier section103followed by a multi-carrier section105. The single-carrier section103is intended to be modulated by a single-carrier modulation scheme and the multi-carrier section105is intended to by modulated by a multi-carrier modulation scheme. In some embodiments described herein, the single-carrier modulation is Quadrature Phase Shift Keying (QPSK) symbol rate or Binary Phase Shift Keying (BPSK), such as according to the 802.11b standard, and the multi-carrier modulation is according to OFDM, such as according to the 802.11a standard. It is understood and appreciated that other single-carrier and multi-carrier modulation schemes may be used.

In the embodiment shown, the single-carrier section103includes a Barker preamble108followed by a Barker header111. The Barker preamble108includes a Sync field107followed by a Sync Field Delimiter (SFD)109and is configured according to 802.11b for Barker Word modulation. The preamble108and the Barker header111may be modulated according to BPSK or QPSK and thus may be transmitted at 1 or 2 megabits per second (Mbps). A long version of the single-carrier section103is transmitted in 192 microseconds (μs) and a short version is transmitted in 96 μs. The multi-carrier portion105includes a preamble113, a data field115and a Short Interference Space (SIFS) Pad117. The data field115is transmitted at a selected data rate from among typical data rates of 6, 9, 12, 18, 24, 36, 48 or54Mbps using OFDM modulation. The SIFs pad is transmitted in 6 μs.

The preamble113is used for synchronization for OFDM modulation, and includes a long sync field119and a signal field121. The preamble113is transmitted in approximately 12 μs. The long sync field119includes a pair of 0.4 μs guard intervals123,125and a pair of 3.6 μs long training symbols127,129. In this manner, it is appreciated that the total duration of the long sync field119is 8 μs, which is significantly shorter than the short or long versions of the single-carrier section103consuming at least 96 μs.

FIG. 2is a simplified block diagram of a mixed carrier signal transmitter200including a Baseband transmitter201implemented according to an embodiment of the present invention. In one embodiment, the transmitter201is configured to operate in several modes, including a single-carrier mode (e.g. 802.11b), a mixed carrier mode (802.11g), and several multi-carrier modes (e.g. 802.11a). The multi-carrier modes may employ OFDM modulation in any one of several frequency bands, such as the 2.4, 5 or 6 GHz bands. A single-carrier processor or kernel203incorporates the core processing functions to configure single-carrier signals at a selected chip rate “w” MHz. The chip rate may be w=11 for 802.11b, such as Barker chips at 11 MHz. The output of the kernel203is provided to the input of a 1:2 splitter205. A first output207of the splitter205is provided to the input of an single-carrier pulse shape block209(digital filter), which outputs a single-carrier packet at a sample rate of “y” MHz. The sample rate may be any selected or suitable rate, such as 18.333, 20, 22, 40, 44 etc. MHz. The pulse shape block209changes the sample rate from w to y and need only meet the spectral mask of the selected single-carrier modulation scheme. The output of the single-carrier pulse shape block209is provided to a first input211of a 3:1 multiplexor (MUX)213, having its output coupled to the input of a digital to analog converter (DAC)215. The DAC215operates at a rate of y MHz to convert the digital Baseband signal into an analog signal for RF processing and transmission.

The analog output of the DAC215is mixed with a radio frequency (RF) signal and transmitted in a wireless medium via an antenna204as known to those skilled in the art. In the embodiment shown, the analog output of the DAC215is provided to an RF system202, which converts the Baseband signal to an RF signal that is asserted in the wireless medium via the antenna204. The transmitter200also includes control logic206coupled to the RF system202and the Baseband transmitter201to control the particular mode of operation. The control logic206controls the splitter205and the MUX213to select single-carrier packets for the single-carrier mode, to select multi-carrier packets for the multi-carrier mode, and to select mixed carrier packets for the mixed carrier mode of operation.

The RF system202and the control logic206may further be configured for multi-band operation. The RF system202may be configured to transmit packets using a selected one of several RF carrier frequencies including, but not limited to, the unlicensed 2.4, 5 or 6 GHz bands. It is contemplated that the 2.4 GHz band be used for the single-carrier mode in accordance with 802.11b. The 2.4 GHz band may also be employed for the mixed carrier mode to provide backwards compatibility with legacy 802.11b devices. It is contemplated that the 5 GHz band be used for the multi-carrier mode in accordance with 802.11a. It is further contemplated that several different bands be used for the multi-carrier mode (i.e., multi-band operation) including the 2.4, 5 and 6 GHz bands as well as any other selected or available frequency bands.

A second output217of the splitter205is provided to a mixed carrier pulse shape digital filter219. The digital filter219receives a single-carrier preamble and header signal from the kernel203and shapes or filters the signal in such as manner to have a similar power spectrum as the multi-carrier signal employed for the mixed carrier waveform. As described further below, the digital filter219includes finite impulse response (FIR) filter taps that are scaled so that the power spectrum of the single-carrier signal approximates the power spectrum of the multi-carrier signal. The mixed carrier preamble and header output from the digital filter219is provided to one input221of a combiner223, which receives a multi-carrier payload originating from a multi-carrier processor or kernel225at a second input227. As described further below, the combiner223operates to combine the mixed carrier preamble and header with the multi-carrier payload to develop a mixed carrier packet at its output228, which is coupled to a second input229of the MUX213.

The kernel225incorporates the core processing functions to configure multi-carrier packets at a selected sample rate of “x” MHz. The sample rate “x” of the kernel225may be any suitable or arbitrary rate, such as 20, 22, 40, 44 etc. MHz. A sample rate of 20 MHz is in accordance with the 802.11a standard for an OFDM kernel. The output of the kernel225is provided to a rate change filter226, which converts the sample rate of the kernel225to y MHz, which is the sample rate of the DAC215. The output of the rate change filter226is provided to the input227of the combiner223and to a third input231of the MUX213. The rate change filter226is optional and not needed if x=y or if the kernel225otherwise operates at the sample rate of y MHz. The configuration and operation of the rate change filter226is further described below.

The transmitter201operates in the single-carrier mode (e.g. 802.11b) when the control logic206controls the splitter205to select its first output207and controls the MUX213to select its first input211so that single-carrier packets generated by the kernel203and shaped by the pulse shape block209are provided to the DAC215. The transmitter201operates in the mixed signal mode (e.g. 802.11g) when the control logic206controls the splitter205to select its second output217and controls the MUX213to select its second input229so that mixed carrier packets from combiner223are provided to the DAC215. The transmitter201operates in a multi-carrier mode (e.g. 802.11a) when the control logic206controls the MUX213to select its third input231so that multi-carrier packets generated by the kernel225are provided to the DAC215.

It is noted that the kernel203may be configured to be capable of producing an entire single-carrier packet (via pulse shaper209) and that the kernel225may be configured to be capable of producing an entire multi-carrier packet. The combiner223, however, combines the first portion or preamble and header of the single-carrier signal with the payload portion of the multi-carrier signal (e.g., including the preamble113, the data field115and SIFs117) to generate the mixed carrier packet. The single-carrier kernel203is further configured to modify the header111of the single-carrier section103to include a bit or field that indicates mixed carrier mode of operation. The mixed carrier mode bit informs the receiver that the packet is a mixed carrier signal rather than a single-carrier signal.

FIG. 3is a simplified block diagram of an embodiment of the combiner223. The combiner223performs phase and time alignment between the single-carrier header and preamble received via its input221and the multi-carrier payload received via its input227. The combiner223also transitions between single-carrier header termination and multi-carrier payload onset. The combiner223includes a soft switch301that switches the output228between first, second and third terminals303,305and307, respectively. The soft switch301is not necessarily implemented as a physical or mechanical switch, but instead may be implemented in firmware or digital logic to perform smooth switching between the signals during transition. The first terminal303is coupled to the input221of the combiner223and to a first input317of a digital signal combiner309. The second terminal305is coupled to the output of the digital signal combiner309. The third terminal307is coupled to an output315of a phase rotator311, which is also provided to a second input319of the digital signal combiner309. The phase rotator311rotates or multiplies the multi-carrier signal by a phase angle “φ” relative to the single-carrier signal (or last portion thereof) to maintain phase continuity. The digital signal combiner309combines the single-carrier and multi-carrier signals during the transition between the full single-carrier portion and the full multi-carrier portion of the mixed carrier signal.

FIGS. 4A and 4Bare graph diagrams of phase relationships for an exemplary single-carrier modulation scheme using either BPSK or QPSK, respectively.FIG. 4Ais a graph diagram of a BPSK plot illustrating that BPSK incorporates both real and imaginary portions in two quadrants (1 of 2 phases). The phase angle φ is either 1 or −1.FIG. 4Bis a simplified graph diagram of a QPSK plot illustrating that QPSK incorporates both real and imaginary portions in all four quadrants (1 of 4 phases). The phase angle φ is either 1, j, −1 or −j. The particular phase of a signal is obscured during transmission so that absolute phase is indeterminate. A receiver is typically configured to determine and track the phase of the incoming signal. However, for the mixed carrier signal, the relative phase between the single-carrier and multi-carrier portions should be maintained or otherwise determinable to facilitate acquisition by the receiver. Therefore, the phase of the multi-carrier signal is based on phase information from the signal carrier signal to facilitate receiver phase acquisition, such as the phase of the last portion of the single-carrier signal.

In the CCK-OFDM configuration, the single-carrier signals employ Direct Sequence Spread Spectrum (DSSS), which are fundamentally different than the OFDM multi-carrier signal format. For CCK-OFDM, either of the BPSK or QPSK formats may be re-used for the header. The phase of the last Barker word of an 802.11b header determines the phase of the coherent OFDM signal relative to an OFDM signal generated by the kernel225. Referring back toFIG. 3, for CCK-OFDM, the phase rotator311rotates the OFDM signal by the phase angle φ of the last Barker word and asserts the rotated OFDM signal at its output315. The rotated OFDM signal is applied to the input319of the digital signal combiner309and to the third terminal307of the soft switch301. A phase angle of 1 corresponds to 0 degree rotation (no rotation), a phase angle of j corresponds to 90 degree rotation, a phase angle of −1 corresponds to 180 degree rotation and a phase angle of −j corresponds to −90 degree rotation. The multi-carrier signal, such as OFDM, is a complex number including real and imaginary parts, otherwise referred to as in-phase (I) and quadrature phase (Q) components, so that mathematically the I and Q components are multiplied by −1, j or −j.

FIG. 5is a timing diagram illustrating exemplary alignment between the single-carrier and multi-carrier portions of the signals using Barker and OFDM. The timing diagram illustrates alignment of an OFDM signal portion501with the last Barker word503of the header. The first chip of each Barker word, including the first chip of the last Barker word503, shown at507, is centered on the 1 μs alignment, and each subsequent Barker chip of each word is centered every 1/11 μs or 91 nanoseconds (ns). For onset of the OFDM signal, the first full sample of the OFDM signal, shown at509, occurs 1 μs after the zero-phase peak of first chip of the last Barker word in the header, and thus 1/11 , μs after the last chip511of the last Barker word thereby maintaining timing during a transition between the single- and multi-carrier segments. The period between the last chip511and the first full OFDM sample509forms a 1/11 μs overlap or transition period513between the last Barker word503and the first full sample of the OFDM signal. A scaled or cyclically extended OFDM sample515is shown before the first full scale OFDM sample509to demonstrate operation of the digital combiner317to smooth the transition between the waveforms. In one embodiment, the OFDM sample515is cyclically extended in that it occurs early prior to fill onset of the OFDM sample. Such transition time alignment allows the equalizer information and the timing information to carry over between the single- and multi-carrier portions of the mixed carrier signal.

Referring back toFIG. 3, for OFDM-CCK operation, the soft switch301connects the first terminal303to the output228of the combiner223until just after the last Barker chip511in order to forward the last Barker word. Then, after the last Barker chip511, the switch301switches to connect the output of the digital signal combiner309at the second terminal305to the output228. The digital signal combiner309digitally combines the single-carrier signal at input317with the rotated multi-carrier signal at input319during the transition period513. It is noted that a digital combiner is used since the signals are digitally sampled in the configuration shown, although analog combiners or the like are contemplated in alterative embodiments. In one embodiment, the digital signal combiner309ramps down the single-carrier signal while ramping up the multi-carrier signal.

In one specific configuration, the single-carrier and multi-carrier signals are both sampled at 44 MHz (y=44), and alignment is based on 11 MHz Barker chip, so there are three (3) intermediate samples between the last barker chip511and the first full OFDM sample509in the transition period513. In one embodiment, the digital signal combiner309combines 75% of the Barker signal with 25% of the OFDM signal for the first intermediate sample, combines 50% of the Barker signal with 50% of the OFDM signal for the second intermediate sample, and combines 25% of the Barker signal with 75% of the OFDM signal for the third intermediate sample during the transition, which intermediate samples are provided to the output228on consecutive 44 MHz cycles. Before the first full OFDM sample509, the soft switch301switches to connect terminal307with the rotated OFDM sample at the output315of the phase rotator311to the output228, and remains at the terminal307for the remainder of the multi-carrier section105.

FIG. 6is a graph diagram illustrating exemplary termination of the single-carrier signal, shown with a dashed curve at601and shaped consistent with 802.11b, and onset of an OFDM symbol, shown at603and shaped identical to 802.11a, during the transition period513. As illustrated in these graph diagrams, the single-carrier is terminated in a controlled fashion when transitioning from single-carrier to multi-carrier. This single-carrier termination maintains the AGC at the point of transition, minimizes the signal power gap, which in turn minimizes the corruption of one signal by the other. The single-carrier termination of the 802.11b segment is similar to that used for 802.11a OFDM shaping. The 802.11a standard specifies a windowing function for OFDM symbols, which is employed to define termination of single-carrier segment. The single-carrier signal is terminated in a predetermined window of time, such as nominally 100 nanoseconds (ns). It is not necessary to completely flush the single-carrier pulse-shaping filter. The resulting distortion to the last Barker word in the header is trivial compared to the 11 chips processing gain, thermal noise and multi-path distortion. The termination may be accomplished either explicitly in the digital signal processing or by analog filtering.

FIG. 7Ais a graph diagram of an exemplary continuous time shaping pulse p(t) that is used by the pulse shape digital filter219so that the power spectrum of the single-carrier portion of the mixed carrier signal approximates the power spectrum of a multi-carrier signal. The graph of the time shaping pulse p(t) shows normalized amplitude plotter versus time in microseconds (μs). The specific time shaping pulse p(t) shown is specified in continuous time and is derived using an infinite impulse response of a brick wall approximation. The infinite impulse response is preferably truncated using a continuous-time window that is sufficiently long to achieve desired spectral characteristics (to approximate multi-carrier modulation) but sufficiently short to reduce complexity. The resulting continuous time pulse shape p(t) may be sampled at the sample rate (y MHz) of the DAC215. For 802.11g using Barker and OFDM, the FIR taps are scaled such that the Barker preamble and header power spectrum approximates the OFDM power spectrum. It is appreciated that the time shaping pulse p(t) is exemplary only and that other filter shapes are contemplated to provide power (gain) matching between the single-carrier and multi-carrier spectrums.

FIG. 7Bis a graph diagram of discrete-time samples700of the time shaping pulse p(t) based on a sample rate of 44 MHz.FIG. 7Cis a graph diagram of the discrete-time samples700of the time shaping pulse p(t) further distributed among four polyphase filters0,1,2and3, each represented by corresponding circle, triangle, star and diamond symbols, respectively. Each of the polyphase filters0–3includes multiple taps, where each tap is programmed with a coefficient “ca,b” that corresponds to one of the discrete-time samples700. As shown, there are 35 discrete samples700, so that each polyphase filter includes 9 taps numbered0–8for a total of 36 taps. The coefficients are indexed according to tap number and filter number, where “a” is an index representing one of the tap numbers0–8and “b” is an index representing one of the polyphase filters0–3. A peak sample701is selected to correspond to coefficient c4,0, which is the center tap of the first polyphase filter0. A second consecutive sample703is selected to correspond to coefficient c4,1, which is the center tap of the second polyphase filter1. A third consecutive sample705is selected to correspond to coefficient c4,2, which is the center tap of the third polyphase filter2. A fourth sample707, being a mirrored sample of the sample703, is selected to correspond to coefficient c4,3, which is the center tap of the fourth polyphase filter3.

After the center taps are determined, the remaining discrete-time samples700are distributed among the taps of the polyphase filters0–3so that every fourth sample corresponds to the next tap of the same filter. Thus, from left to right, the samples are ordered as c0,3, c0,0, c0,1, c0,2, c1,3, c1,0, c1,1, c1,2, c2,3, c2,0, c2,1, c2,2, c3,3, c3,0, c3,1, c3,2, c4,3, c4,0, c4,1, c4,2, c5,3, c5,0, c5,1, c5,2, c6,3, c6,0, c6,1, c6,2, c7,3, c7,0, c7,1, c7,2, c8,3, c8,0, and c8,1. The final tap c8,2of the third polyphase filter2may be programmed with a zero or with the next consecutive point along the time shaping pulse p(t) on either side of center.

FIG. 8Ais a simplified block diagram of an exemplary embodiment of the pulse shape digital filter219configured to convert 11 MHz single-carrier samples to a 44 MHz sample rate and that further uses the discrete-time samples700of the time shaping pulse p(t) to shape the signal in such as manner to have a similar power spectrum as the multi-carrier signal employed for the mixed carrier waveform. The polyphase filters0–3are shown as filters801,803,805and807, each receiving the 11 MHz (w=11) samples from the single-carrier kernel203via a common input802. In this manner, each of the single-carrier samples are shifted into each tap of each of the polyphase filters0–3at a rate of 11 MHz. Each of the polyphase filters0–3has an output that is selected by a switch809operating at 44 MHz, so that the outputs of the filters are provided to an output810of the exemplary pulse shape digital filter219at a new sample rate of 44 MHz.

FIG. 8Bis a more detailed block diagram of an exemplary embodiment of the polyphase filter801(the first filter0) in the form of a FIR filter. The polyphase filter801includes series of memory locations811a–iconfigured in a similar manner as a shift register. In particular, each sample from the kernel203is loaded into the first memory location811aat a rate of 11 MHz, and then each sample is serially shifted from left to right through each of the memory locations811b–iat the same rate of 11 MHz. Each sample loaded into the each of the memory locations811a–iis provided to one input of a corresponding one of a series of multipliers813a–i, where each of the multipliers813a–ihave a second input for receiving a respective one of the coefficients c0,0–c8,0. The outputs of the multipliers813a–iare summed at a summing junction815, which asserts a summed value at an output817of the polyphase filter801. In this manner, the samples are shifted right, multiplied and added at a rate of 11 MHz to provide an 11 MHz output. Each of the polyphase filters803,805and807are configured in substantially the same way as the polyphase filter801, except that different coefficients are used in accordance with the distribution of the discrete-time samples700. Since there are four polyphase filters0–3and since each output is used by the switch809, the output rate of the pulse shape digital filter219is 44 MHz.

It is noted that the case illustrated inFIGS. 7B,7C,8A and8B are specific to w=11 and y=44 in which the rate of the mixed carrier signal is a multiple of the single-carrier rate. Different single-carrier and mixed carrier sample rates are contemplated. In an alternative example for w=11 and y=40, the new rate is not a direct multiple of the single-carrier rate. The single-carrier rate may be up-sampled by the new rate of 40, and the 440 MHz result is then filtered using a low pass filter (LPF) specified as the desired pulse shape. The filtered result is then down-sampled by the original rate of 11 to obtain the new sample rate of 40 MHz. The up-sampling may be achieved using the same number of polyphase filters as the up-sampling rate, such as 40 in this example. The taps of the polyphase filters are programmed with appropriate coefficients in a similar manner as described above for the discrete-time samples700, except that the time shaping pulse p(t) is sampled at 440 MHz rather than at 44 MHz. The number of taps of each polyphase filter is selected to at least incorporate the samples at the selected sample rate. The down-sampling is achieved by switch logic or circuitry that selects every wthoutput to achieve the desired rate, such as every 11thoutput to achieve the new rate of 40 MHz (440/11).

Referring back toFIG. 2, the rate change filter226performs a rate change between x and y and does not perform any particular pulse shaping other than low pass filtering. It is possible that x and y are equal since the sample rates are arbitrary, in which case the rate change filter226is not necessary.FIG. 9a simplified block diagram of an exemplary embodiment of the rate change filter226decomposed into two blocks including an x:z rate change filter901for converting to an intermediate and arbitrary rate z MHz and a first-in, first-out (FIFO) buffer903for converting between rates z and y MHz. Adding the FIFO buffer903to the rate change filter is not necessary but may be desired to ease implementation by allowing the rate change filter to operate asynchronously with respect to the input and output sample rates.

If x and y are not equal (or if a different intermediate rate z is desired), the rate change may be performed in a similar manner as described above for the pulse shape digital filter219using polyphase filters except that the taps are programmed according to a low pass filter (LPF). If x and y are not direct multiples, conversion may be simplified when x and y are divisible by a common multiple. For example, a conversion from 20 MHz to 44 MHz may employ up-sampling by 11 (rather than 44) and down-sampling by 5 (rather than 20) to achieve the desired conversion since each is divisible by 4In one embodiment, 11 polyphase filters may be employed, each having 21 taps to achieve the desired sample rate of 220 MHz. Any extra taps may be provided with an appropriate coefficient value, such as zero.

FIG. 10is a block diagram illustrating cyclic extension of the multi-carrier signal to initialize the rate change filter226to preclude an unnecessarily long transient. A complete multi-carrier symbol1001with seven samples x0, x1, x2, x3, x4, x5, x6is shown having form xn, denoting a sample “x” at a time “n”. In this manner, x0is first in time, x1is second and so on in which the subscript “n” indicates a time index. The last two samples x5and x6are copied and pre-pended to the multi-carrier symbol1001as shown at1003. The time index of the pre-pended samples x5and x6are then changed to x−2and x−1, respectively, to indicate that they are first in time of a cyclically extended multi-carrier symbol1005.

FIG. 11is a block diagram of a FIR filter1100similar in format and operation as the polyphase filter801. The FIR filter1100includes multiple memory locations1101a–e, and multipliers1103a–eand a summing junction1105to implement an exemplary 5-tap FIR filter with output1107. Although an exemplary 5-tap filter is shown, it is appreciated that any suitable number of taps may be employed in specific embodiments. The multipliers1103a–ereceive coefficients ct,p, where “t” is an index referencing tap number and “p” is an index referencing the phase or polyphase filter number. The memory locations1101e,1101d,1101cand1101bare pre-loaded with cyclically extended samples x−4, x−3, x−2and x−1, respectively, to initialize the history of the FIR filter1100with cyclically extended samples prior to the first actual sample x0.

Referring back toFIGS. 2 and 5, the timing of the combiner223is such that the last sample511of the single-carrier signal is multiplied by the tap in the pulse shape digital filter219with the largest magnitude at the beginning of the transition period513. Further, the timing of the combiner223is such that the first full multi-carrier sample509of the multi-carrier signal is multiplied by the tap in the rate change filter226with the largest magnitude at the end of the transition period513. If the polyphase filter801is employed as the first filter of the pulse shape digital filter219, then the last sample of the single-carrier signal from the kernel203is located at memory location811eat the beginning of the transition period513. This last sample is multiplied by the largest coefficient c4,0corresponding to the discrete sample701. Also, if the FIR filter1101is used as the first polyphase filter of the rate change filter226, then the first full sample of the multi-carrier signal is located at memory location1101cat the end of the transition period513.

FIG. 12is a block diagram illustrating the FIR filter1101used as the first polyphase filter (phase or polyphase filter number=0) with the first full sample x0of the multi-carrier signal at the middle memory location1101chaving the largest coefficient c2,0. If the FIFO buffer903is also used, then any delay through this buffer is taken into account to ensure proper timing of the symbols. It is appreciated that the last sample of the single-carrier signal is shifted through the polyphase filter801during ramp down of the single-carrier signal while the cyclically extended portion of the multi-carrier signal is propagating through the FIR filter1101during ramp up of the multi-carrier signal during the transition period513.

FIG. 13is a block diagram of an exemplary rate change filter1300that may be used to convert a 20 MHz sample rate to a 44 MHz sample rate. The rate change filter1300may be used, for example, as the rate change filter226for x=20 and y=44The signal at a 20 MHz sample rate are received at an input1301and provided to each input of 11 polyphase filters1303individually labeled0–10. Each of the polyphase filters1303operates at 20 MHz so that the collective outputs of the polyphase filters1303provides an intermediate rate or frequency of 11 times 20 or 220 MHz. Each of the polyphase filters1303is configured as a FIR filter similar to the FIR filter1100with a plurality of filter taps. The number of filter taps is determined by the intermediate frequency of 220 MHz, and in the embodiment shown each filter includes 21 taps. A sample switch1305selects every 5thfilter output to down-sample the intermediate frequency rate by 5 to achieve the new sample rate of 44 MHz. For example, in a first pass, the sample switch1305selects the outputs of polyphase filters0,5and10. In the second-sixth consecutive passes, the sample switch1305selects the outputs of polyphase filters4and9,3and8,2and7,1and6, and0,5and10, respectively. Operation repeats in this manner for the entire signal segment.

Although a system and method according to the present invention has been described in connection with the preferred embodiment, it is not intended to be limited to the specific form set forth herein, but on the contrary, it is intended to cover such alternatives, modifications, and equivalents, as can be reasonably included within the spirit and scope of the invention.