AC-DC converter and AC-DC conversion method

The present invention provides an AC-DC converter and AC-DC conversion method for converting an AC input provided by a power transfer winding. The AC-DC converter includes a rectifying means for rectifying the AC input into a rectified output, and a control means for controlling the rectifying means based on a comparison between a reference signal and a voltage feedback signal, the voltage feedback signal being based on the rectified output.

FIELD OF THE INVENTION

The present invention relates to AC-DC converters and AC-DC conversion methods. The invention will be described in the context of wireless power transfer, particularly where a power transfer coil functions as an energy-receiving coil that wirelessly receives energy in order to, for example, charge a battery included with a portable electronic device. However, it will be appreciated that the invention is not limited to this particular use.

BACKGROUND OF THE INVENTION

Traditional AC-DC power switching power supplies consist of an AC-DC power conversion stage and a DC-DC power conversion for output voltage regulation. Their output filter usually consists of an inductor and a capacitor forming an output voltage filter. The schematic of a traditional AC-DC converter with electric isolation can be shown inFIG. 1. It consists of two stages: an AC-DC power stage and a DC-DC power conversion stage.

For electrically isolated output voltage, which is required in many power supplies, electrical isolation is usually achieved with the use of an isolation transformer. The DC-DC power converter usually consists of an inverter bridge (fed by a DC voltage from a front AC-DC power stage), an isolation transformer, a rectifier and an output filter comprising an inductor and a capacitor. Generally, voltage feedback from the “secondary side” of the transformer is required and the control action for the output voltage regulation is carried out by the inverter on the “primary side” of the transformer. It is important to note that this traditional approach requires the output filter Loutand Coutto filter the switching voltage ripple.

In order to reduce the conduction loss in the diode rectifier, synchronous rectification can be used. Synchronous rectification has been utilized in switched mode power supply technology. The replacement of diodes with power MOSFETs with low on-state resistance enables synchronous rectifiers to have less conduction loss than diodes. This has been adopted in switched mode power supply for computer products which have Central Processing Units (CPUs) running at low voltage and high current conduction (e.g. 3.3 V DC at 100 A). In conventional switched mode power supply applications, closed-loop output voltage control is an essential feature because the output voltage of a power supply must be controlled within a tight tolerance.

In existing synchronous rectification technology, the output voltage regulation is primarily controlled from the primary side of the system.FIG. 2shows a typical schematic of a DC-DC power converter using a synchronous rectifier based on National Semiconductor design document titled “Synchronous Rectification in High-Performance Power Converter Design” authored by Robert Selders Jr., and available at the website: http://www.national.com/appinfo/power/files/national_power_designer112.pdf.

In this traditional DC-DC converter with an isolated diode-based rectifier, the output voltage is controlled by the driving circuit on the primary side. Secondary feedback, via isolated means, is used to control the switching action in the primary circuit in order to regulate the output DC voltage.

The diodes inFIG. 2can be replaced with power MOSFETs having low on-state resistance as shown inFIG. 3. Similar to the circuit inFIG. 2, the output voltage control is carried on the primary circuit with secondary feedback provided through isolated means. In addition, an output inductor Loutis needed.

In principle, the secondary gate controller can be eliminated if a self-driven gate drive design is adopted. A self-driven synchronous rectifier takes advantage of the polarities of the induced voltages in the secondary winding. Such a self-driven synchronous rectifier is shown inFIG. 4and a corresponding control scheme is shown inFIG. 5. Despite the fact that the secondary gate drives can be eliminated, the output voltage regulation is still controlled by the primary circuit.

In both cases, the synchronous rectifiers, regardless of using diodes or MOSFETs, provide the AC-DC rectification only. The output voltage regulation is controlled by the switching action in the primary circuit.

The problems of the traditional approach when employed in a wireless transfer system are summarized as follows:

(a) Two power stages, i.e. AC-DC and DC-DC without transformer isolation, are needed. This increases the cost and size of the circuit and is not attractive for embedding into a portable device such as a mobile phone, particularly, one with a slim design.

(b) Output voltage regulation is controlled by the inverter switching action on the “primary side” of the isolation transformer. This means an isolated feedback mechanism is required, which leads to increased cost.

(c) An output inductor Loutis needed. This increases power loss and reduces energy efficiency of the secondary circuit, leading to: thermal problems in a portable device, which typically has no ventilation; safety problems in the battery due to a high temperature rise; and a reduction in overall system efficiency.

If output voltage regulation is needed without control from the primary circuit of the transformer, one solution is to use a DC-DC converter with voltage control as shown inFIG. 6. The AC voltage induced in the secondary winding is first rectified, and then the DC-DC converter will turn the rectified voltage into a regulated DC voltage. However, this approach:

(b) requires a DC-DC converter, such as the ones described above; and

(c) requires an output inductor.

These three factors increase the cost and size of the secondary module and reduce the overall energy efficiency of the system.

SUMMARY OF THE INVENTION

The present invention provides in a first aspect an AC-DC converter for converting an AC input provided by a power transfer winding, the AC-DC converter including: a rectifying means for rectifying the AC input into a rectified output; and a control means for controlling the rectifying means based on a comparison between a reference signal and a voltage feedback signal, the voltage feedback signal being based on the rectified output.

Preferably, the rectifying means includes a synchronous rectifier. More preferably, the rectifying means includes a self-driven synchronous rectifier.

Preferably, the control means uses hysteresis control to control the rectifying means.

Preferably, the AC-DC converter includes a comparing means for providing the comparison to the control means, the comparing means comparing the voltage feedback signal with a hysteresis tolerance defined by an upper hysteresis band above the reference signal and a lower hysteresis band below the reference signal.

Preferably, the control means disables the rectified output when the voltage feedback signal exceeds the upper hysteresis band. Also preferably, the control means enables the rectified output when the voltage feedback signal falls below the lower hysteresis band.

Preferably, the AC-DC converter includes a main comparator for providing the comparison to the control means. Preferably, the main comparator has a non-inverting input and an inverting input, the voltage feedback signal being provided to the non-inverting input and the reference signal being provided to the inverting input. In one embodiment, the reference signal is a voltage across a zener diode.

Preferably, the AC-DC converter includes a voltage feedback means for sampling the rectified output and providing the voltage feedback signal. Preferably, the voltage feedback means includes a voltage feedback circuit connected to the rectified output. Preferably, the voltage feedback circuit is connected before an output capacitor. Preferably, the voltage feedback circuit includes a resistive potential divider.

Preferably, the rectifying means includes two main switches. Preferably, at least one main switch has low on-state resistance. Preferably, at least one main switch includes a power MOSFET. At least one main switch can include an N-type MOSFET or a P-type MOSFET or both. In some embodiments, at least one main switch includes a pair of component switches connected back-to-back to form a bi-directional switch.

Preferably, the control means includes a first driver for driving one main switch and a second driver for driving the other main switch, the first and second drivers disabling the rectified output when the voltage feedback signal is above the reference signal by a first predetermined value, and allowing the rectifying means to operate as a self-driven rectifier to enable the rectified output when the voltage feedback signal is below the reference signal by a second predetermined value.

Preferably, the first and second drivers disable the rectified output by maintaining both the main switches on. More preferably, the main switches form part of a closed loop circuit when both the main switches are on, the current of the AC input circulating in the closed loop circuit thereby disabling the rectified output. Preferably, the first and second drivers allow the rectifying means to operate as a self-driven rectifier by allowing the main switches to turn on and off as part of a self-driven rectifier.

In a first embodiment, each of the first and second drivers includes two driver switches.

Preferably, the control means includes a third driver for receiving the comparison between the reference signal and the voltage feedback signal, and for driving the first and second drivers based on the comparison. Preferably, the third driver includes two driver switches. Preferably, the first, second and third drivers drive the main switches through a summation point.

Preferably, the rectifying means receives the AC input via a reactive impedance formed from a leakage inductance from the power transfer winding in series with an input capacitor, the AC-DC converter including an impedance capacitor connected across the reactive impedance with an impedance capacitor switch, wherein the impedance capacitor switch is turned on when the main switches are on thereby connecting the impedance capacitor across the reactive impedance.

In a second embodiment, each of the first and second drivers includes three driver switches and a driver diode.

Preferably, each of the first and second drivers directly receives the comparison between the reference signal and the voltage feedback signal. Preferably, the control means includes an auxiliary low-power diode bridge having an auxiliary capacitor to provide DC power to the first and second drivers.

In a third embodiment, each of the first and second drivers includes two driver switches and two driver comparators. Preferably, one of the driver comparators operates as a logical OR gate.

Preferably, the control means includes a return switch in the return path of the AC input such that turning off the return switch when the rectified output is disabled reduces energy loss from continuously circulating current.

Preferably, the control means includes a third driver in the return path of the AC input, the third driver being self-biased such that the return switch is normally closed, and the third driver turning off the return switch when the rectified output is disabled thereby reducing energy loss from continuously circulating current. Preferably, the third driver includes three driver switches.

Preferably, the AC-DC converter forms part of a wireless power receiver that receives power wirelessly through the power transfer winding. Preferably, the wireless power receiver is a portable electronic device. Preferably, the power transfer winding is a transformer winding located on one side of a transformer and the control means is located on the same one side of the transformer. The control means is preferably conductively coupled to the rectifying means. Also preferably, the AC-DC converter is a single-stage AC-DC converter.

In a second aspect, the present invention provides a method of converting an AC input provided by a power transfer winding, the method including: rectifying the AC input into a rectified output; and controlling the rectification on the basis of a comparison between a reference signal and a voltage feedback signal, the voltage feedback signal being based on the rectified output.

Preferably, a synchronous rectifier is used to rectify the AC input into the rectified output. More preferably, a self-driven synchronous rectifier is used to rectify the AC input into the rectified output.

Preferably, hysteresis control is used to control the rectification.

Preferably, the method includes comparing the voltage feedback signal with a hysteresis tolerance defined by an upper hysteresis band above the reference signal and a lower hysteresis band below the reference signal.

Preferably, controlling the rectification includes disabling the rectified output when the voltage feedback signal exceeds the upper hysteresis band. Also preferably, controlling the rectification includes enabling the rectified output when the voltage feedback signal falls below the lower hysteresis band.

Preferably, a main comparator is used to compare the voltage feedback signal with the reference signal. Preferably, the main comparator is provided with a non-inverting input and an inverting input, and the method includes providing the voltage feedback signal to the non-inverting input and providing the reference signal to the inverting input. In one embodiment, a voltage across a zener diode is used to provide the reference signal.

Preferably, the method includes sampling the rectified output to provide the voltage feedback signal. Preferably, a voltage feedback circuit connected to the rectified output is used to sample the rectified output. Preferably, the voltage feedback circuit is connected before an output capacitor. Preferably, the voltage feedback circuit is provided with a resistive potential divider.

Preferably, two main switches are used to rectify the AC input into the rectified output. Preferably, at least one main switch is provided with low on-state resistance. Preferably, at least one main switch is provided with a power MOSFET. At least one main switch can be provided with an N-type MOSFET or a P-type MOSFET or both. In some embodiments, at least one main switch is provided with a pair of component switches connected back-to-back to form a bi-directional switch.

Preferably, controlling the rectification includes driving one main switch with a first driver and driving the other main switch with a second driver, the first and second drivers disabling the rectified output when the voltage feedback signal is above the reference signal by a first predetermined value, and allowing the main switches to operate as part of a self-driven rectifier to enable the rectified output when the voltage feedback signal is below the reference signal by a second predetermined value.

Preferably, the first and second drivers disable the rectified output by maintaining both the main switches on. More preferably, the main switches form part of a closed loop circuit when both the main switches are on, the current of the AC input circulating in the closed loop circuit thereby disabling the rectified output. Preferably, the first and second drivers allow the main switches to turn on and off to operate as part of a self-driven rectifier.

In a first embodiment, each of the first and second drivers is provided with two driver switches.

Preferably, controlling the rectification includes receiving the comparison between the reference signal and the voltage feedback signal with a third driver, and driving the first and second drivers with the third driver based on the comparison. Preferably, the third driver is provided with two driver switches. Preferably, the first, second and third drivers drive the main switches through a summation point.

Preferably, the AC input is provided via a reactive impedance formed from a leakage inductance from the power transfer winding in series with an input capacitor, the method including providing an impedance capacitor connected across the reactive impedance with an impedance capacitor switch, and turning on the impedance capacitor switch when the main switches are on thereby connecting the impedance capacitor across the reactive impedance.

In a second embodiment, each of the first and second drivers is provided with three driver switches and a driver diode.

Preferably, each of the first and second drivers directly receives the comparison between the reference signal and the voltage feedback signal. Preferably, the method includes providing an auxiliary low-power diode bridge having an auxiliary capacitor to provide DC power to the first and second drivers.

In a third embodiment, each of the first and second drivers is provided with two driver switches and two driver comparators. Preferably, one of the driver comparators operates as a logical OR gate.

Preferably, controlling the rectification includes providing a return switch in the return path of the AC input such that turning off the return switch when the rectified output is disabled reduces energy loss from continuously circulating current.

Preferably, controlling the rectification includes providing a third driver in the return path of the AC input, the third driver being self-biased such that the return switch is normally closed, and the third driver turning off the return switch when the rectified output is disabled thereby reducing energy loss from continuously circulating current. Preferably, the third driver is provided with three driver switches.

Preferably, the method includes receiving power wirelessly through the power transfer winding, wherein the power transfer winding is provided as part of a wireless power receiver. Preferably, the wireless power receiver is provided as a portable electronic device. Preferably, the power transfer winding is provided as a transformer winding located on one side of a transformer, and wherein the rectification is controlled with a control means located on the same one side of the transformer. Preferably, the AC input is rectified into the rectified output with a rectifying means and the rectification is controlled with a control means conductively coupled to the rectifying means. Also preferably, the AC input is converted into the rectified output in a single stage.

DETAILED DESCRIPTION OF THE BEST MODE OF THE INVENTION

Referring to the figures, there is provided an AC-DC converter1for converting an AC input2provided by a power transfer winding3. The AC-DC converter1includes a rectifying means4for rectifying the AC input2into a rectified output5, and a control means6for controlling the rectifying means4based on a comparison between a reference signal7and a voltage feedback signal8, the voltage feedback signal being based on the rectified output5.

Thus, the AC input2is converted into the rectified output5in a single stage, the AC-DC converter1thereby being a single-stage AC-DC converter. Furthermore, the power transfer winding3is a transformer winding located on one side9of a transformer10and the control means6is located on the same one side9of the transformer10. The control means6is conductively coupled to the rectifying means4.

The rectifying means4includes a synchronous rectifier11, which in the present embodiment, is a self-driven synchronous rectifier. The control means6uses hysteresis control to control the rectifying means4. In particular, there is a comparing means12for providing the comparison to the control means6, the comparing means12comparing the voltage feedback signal8with a hysteresis tolerance defined by an upper hysteresis band above the reference signal7and a lower hysteresis band below the reference signal7. The control means6disables the rectified output5when the voltage feedback signal8exceeds the upper hysteresis band, and enables the rectified output5when the voltage feedback signal8falls below the lower hysteresis band.

In the present embodiment, the comparing means12is in the form of a main comparator13. The main comparator13has a non-inverting input14and an inverting input15, the voltage feedback signal8being provided to the non-inverting input and the reference signal7being provided to the inverting input. In this particular embodiment, the reference signal7is a voltage across a zener diode16.

There is also a voltage feedback means17for sampling the rectified output5and providing the voltage feedback signal8. In particular, the voltage feedback means17includes a voltage feedback circuit18connected to the rectified output5. In the present embodiment, the voltage feedback circuit18is connected before an output capacitor19, and includes a resistive potential divider20.

The rectifying means4includes two main switches21and22. The control means6includes a first driver23for driving one main switch21and a second driver24for driving the other main switch22. The first and second drivers23and24disable the rectified output5when the voltage feedback signal8is above the reference signal7by a first predetermined value, and allow the rectifying means4to operate as a self-driven rectifier to enable the rectified output5when the voltage feedback signal8is below the reference signal7by a second predetermined value.

The first and second drivers23and24disable the rectified output5by maintaining both the main switches21and22in an on position. In particular, the main switches21and22form part of a closed loop circuit when both the main switches are on, the current of the AC input2circulating in the closed loop circuit thereby disabling the rectified output5. The first and second drivers23and24allow the rectifying means4to operate as a self-driven rectifier by allowing the main switches21and22to turn on and off as part of a self-driven rectifier.

Thus, embodiments of the present invention relate to a novel concept together with relevant circuits and control schemes for electromagnetically coupled single-stage self-driven AC-DC converters with synchronous rectifiers that have output voltage regulation functionality without using an output inductor and without using a DC-DC converter.

More particularly, embodiments of the present invention involve band-band or hysteresis control of the output DC voltage for an electromagnetically coupled secondary circuit in which a synchronous rectifier included with an AC-DC converter of the invention should provide self-driven switching and output voltage regulation without using an output filter inductor. A basic embodiment is based on the embodiment described above, and uses the voltage feedback means17, the comparing means12, the desired voltage reference7in the secondary circuit to form the control means6to control the power flow of the rectifying means4to the output capacitor19in a manner that ensures that the output voltage is regulated to a desired DC voltage level within a certain tolerance.

FIG. 7shows this basic embodiment. The power transfer winding3is in the form of a secondary winding of a magnetically coupled device, which in this embodiment, is the transformer10. The secondary winding3is fed to the rectifying means4, in the form of a synchronous rectifier circuit, through a reactive impedance X. The impedance X can consist of the stray or leakage inductance of the secondary winding3in series with an input capacitor25for impedance matching in order to achieve maximum power transfer and high efficiency.

As described above, a power flow enabling and disabling mechanism in the synchronous rectifier4, controlled by the control means6, regulates the output voltage. The synchronous rectifier4is on the one side9, that is, the secondary side, of the transformer, which is the same one side9the control means is located on so as to control the synchronous rectifier4without control from the primary side34and also without an output inductor. Thus, the synchronous rectifier4and the control means6form part of the secondary circuit.

The voltage feedback circuit18, in the form of the resistive potential divider20, is used to provide the voltage feedback signal8for comparison with the reference signal7. In this embodiment, the reference signal7is a voltage across a zener diode16, but the signal can be obtained from other means in other embodiments. This reference voltage7is usually a scaled down version of the desired output voltage level Vo.

The main comparator13has a hysteresis tolerance with an upper hysteresis band and a lower hysteresis band. The difference between the upper band and the lower band is the tolerance ΔV. If the voltage feedback signal8exceeds the reference voltage7level by Vo+ΔV/2 (i.e reaching the upper band), the main comparator13will disable the power flow from the self-driven synchronous rectifier4to the output filter capacitor19. The voltage of the output capacitor19is also the output voltage of the AC-DC converter1. When the power flow is disabled, the output capacitor19will be discharged by the load and hence the output voltage Voutwill decrease.

When Voutis decreased to the lower hysteresis band (i.e. Vo−ΔV/2), the main comparator13will change state to enable the power flow and so the voltage of the output capacitor19will increase. This bang-bang control strategy allows this secondary circuit to self-regulate the output voltage without using control from the primary circuit of the transformer10. The operation of this bang-bang or hysteresis control is illustrated with the aid of relevant waveforms of the secondary circuit inFIG. 8.

Embodiments can be applied to general AC-DC power conversion and are particularly suitable for use in the energy-receiving modules (secondary modules) of wireless energy transfer systems such as wireless battery charging systems for portable electronic device loads. If applied to wireless energy transfer through the use of loosely coupled transformers, the output voltage regulation is carried out by the rectifying means4, such as the self-driven synchronous rectifier described above, and controlled by the control means6, both on the secondary side9of the wireless energy transfer system.

This is the case in the present embodiment, in which the AC-DC converter1forms part of a wireless power receiver that receives power wirelessly through the power transfer winding3. The wireless power receiver is a portable electronic device, such as a mobile phone or a laptop computer.

Thus, both the rectifying means4and the control means6are part of the secondary circuit or the energy-receiving module in the portable electronic device, that is, the secondary side9of the wireless energy transfer system, and the control means6controls the rectifying means4on the secondary side9without control from the primary circuit or wireless charging circuit, that is, the primary side34of the wireless energy transfer system. The wireless charging circuit can be part of for example, a wireless charging pad.

In future wireless energy transfer systems, such as those that include wireless battery charging pads, the secondary modules could be designed from different manufacturers based on some common protocol agreed to by an international body such as the International Wireless Power Consortium (http://www.wirelesspowerconsortium.com). This means that the primary circuit may not be designed exactly for a particular secondary load. In this case, it is necessary that output voltage regulation be provided by the secondary circuit. While the secondary winding will receive energy via the electromagnetic coupling, the secondary circuit also needs to regulate its output DC voltage in order to protect the remaining charging circuit and battery inside the portable electronic load, such as a mobile phone. Embodiments of the present invention provide new solutions to meet the output voltage regulation requirements of self-driven synchronous rectifiers on the secondary side of the system without direct control from the primary side.

Three particular embodiments of the present invention will now be described in further detail. The same numbering is used for the same or equivalent features in different embodiments, unless otherwise indicated.

FIG. 9,FIG. 10andFIG. 11show a secondary AC-DC bridge-type rectifying circuit4. An ordinary full-bridge comprising two diodes D1and D2and two main switches21(also labeled as “Q1” inFIGS. 9,10and11) and22(also labeled as “Q2” inFIGS. 9,10and11) (Q1and Q2can be N-type MOSFETs) is fed by an AC input2having a voltage V1(which can be an induced voltage in the secondary winding of a transformer) through some input impedance (which can be stray inductance of the secondary winding or an inductive-capacitive impedance). The two main switches21(Q1) and22(Q2) are of a type with low on-state resistance (such as MOSFETs) and are intentionally utilized to replace two ordinary diodes at the lower portion of the bridge for the purpose of switching control and also for reducing conduction loss. The AC power delivered to the output load R8, with a filtering capacitor19(Cout) can be regulated to achieve a constant DC output5(Vout). It should be noted that even before this circuit becomes ready to function when the system is powered up, the diodes D1and D2and the body diodes of Q1and Q2already provide a diode rectifier. Therefore, the inherent diode rectifier function exists before the self-driven synchronous rectifier4is ready to function.

One simple control method for this self-driven synchronous rectifier4is to use a hysteresis control mechanism. A feedback circuit18formed by two resistors R6and R7is used to sense the output voltage and feed it into the non-inverting input of a main hysteresis comparator13(U1). The sampled voltage8is then compared with a predefined voltage reference7(Vref) at the inverting input of the main comparator13. This reference voltage7can be obtained from the stable voltage of a zener diode or other equivalent means. When the sampled voltage signal8is higher than the reference voltage7by a hysteresis voltage band (representing a certain small tolerance), the output of the main comparator13will go to “high”. When the sampled voltage signal8is lower than the reference voltage7by a hysteresis voltage band, the output of the main comparator13will go to “low”. That means when the output voltage is higher than desired value, the main comparator13goes to high and vice versa. The introduction of the small hysteresis band (tolerance) is to avoid unwanted chattering of the output of the main comparator13when the feedback voltage8is very close to the reference voltage7.

There are three drivers inFIG. 9,FIG. 10andFIG. 11. One driver23(also labeled as “Driver1” inFIGS. 9,10and11), formed by Q3and Q4, is used to drive the main switch21(Q1) to turn ON or OFF according to the switching voltage sensed at point B. The other driver24(also labeled as “Driver2” inFIGS. 9,10and11), formed by Q5and Q6, is used to drive the main switch22(Q2) to turn ON or OFF according to the switching voltage sensed at point A. Therefore, the drivers23and24are designed to form the self-driven gate drive system6that controls the synchronous rectifier4according to the AC voltage input2to the synchronous rectifier4so that the synchronous rectifier can replace the diode rectifier.

A third driver26(also labeled as “Driver3” inFIGS. 9,10and11) is designed for the control of the power flow from the synchronous rectifier4to the output capacitor19. The gates of Q1and Q2are intentionally connected at a summation point27by three resistors R1, R2and R3, to receive the driving signals from the three drivers23,24and26at the same time. When the third driver26, formed by Q7and Q8, does not function in the circuit, the output of the third driver26through R3will bias both Q1and Q2in the linear operation region at around 2 volts. The AC input switching signal at point A and point B will drive main switches Q1and Q2. Therefore, the full-bridge D1, D2, Q1and Q2will function as a self-driven synchronous rectifier at the lower portion of the bridge circuit. Without the use of the third driver26, the synchronous rectifier4does not have the output regulation capability.

Once the hysteresis control mechanism is in place, the bridge circuit can act as a self-regulated AC-DC converter. In particular, there are two modes: power flow disabling and power flow enabling. These modes are described in further detail below.

Power Flow Disabling: When the feedback voltage signal8is higher than the desired value, the main comparator13(U1), goes to high. The high output of U1drives Q7of the third driver26to saturation (i.e. to fully turn on). The gate voltages of both main switches Q1and Q2go high and turn on both main switches completely at the lower portion of the bridge circuit. The input2(V1), input impedance, Q1and Q2will form a closed loop circuit. Input current will circulate in this loop (without being transferred to the output to charge up the output capacitor19and thus increase the output voltage5) until the main switches Q1and Q2are turned off. When both Q1and Q2are turned on and the input current circulates within this loop, this is effectively the “disable” period inFIG. 8. Power flow is disabled from transferring to the output capacitor19in this “disable” period. During this power flow disabling period, the capacitor voltage will remain constant if there is no load discharging the output capacitor19. If the output capacitor19is loaded, the output capacitor will be discharged by the load and so the output voltage will decrease as shown inFIG. 8. This situation will continue until the output voltage decreases to the lower hysteresis voltage band where the feedback voltage8becomes less than the reference voltage7.

Power Flow Enabling: The input power can then be transferred to the output capacitor19again when the feedback voltage8is less than reference voltage7. In this case, the output of the main comparator13(U1) becomes “low”. The third driver26will not influence the normal self-driven functions of the other two drivers23and24and so the main switches Q1and Q2will turn on and off according to the normal self-driven mechanism described previously. During this power flow enabling period, the input current will charge the output capacitor19and the output voltage will increase until it reaches the upper hysteresis voltage band.

Since the hysteresis voltage band is small, the power flow disabling and enabling mechanism ensures that the output voltage5is regulated to the desired level within a tight tolerance. Since the power flow from the synchronous rectifier4is in form of a current source, the current source can be used to charge the capacitor19directly without using an extra output filter inductor. For wireless charging systems, this feature is acceptable because the voltage ripple in the output of the secondary module can be relatively large because there will be a battery charging control inside the battery pack for further control of the battery charging, such as control of the initial constant current charging and the subsequent constant voltage charging.

The power flow disabling (i.e. turning on both Q1and Q2) and enabling mechanism is a main factor in the present invention to determine or regulate the amount of input power that will go to the output load. This control signal is derived from the output of the main comparator13(U1). Hysteresis control is one application example. Other control methods that can take advantage of this disabling and enabling mechanism can also be used. In the example shown inFIG. 9andFIG. 10, the input current circulates within a loop and is prevented from transferring to the output to charge up the output capacitor19.

It should be noted that the impedance X inFIG. 7can consist of a leakage inductor and a series input capacitor25to form a series resonant circuit that is designed according to the operating frequency of the transformer. If the two main switches Q1and Q2are turned on together to form a closed current loop, there may exist a resonant situation that may lead to a high current in the loop. To avoid this high current issue, one solution is to modify the circuit to include an alternative path with a different capacitor such as an impedance capacitor28(also labeled as “A1” inFIG. 11). By inserting the impedance capacitor A1across the impedance X using an impedance capacitor switch29(S1) when both of Q1and Q2are turned on, the resonant frequency of the equivalent impedance will change and limit the loop current to a lower value.

FIG. 12andFIG. 13show a secondary AC-DC self-driven synchronous rectifier4with switches placed in the upper part of the rectifier. A full-bridge synchronous rectifier4consisting of two diodes D1and D2and two main switches21and22(also labeled as “Q1” and “Q2” respectively inFIGS. 12 and 13) is fed by the voltage of the AC input2(which could be the induced voltage in the secondary winding of a transformer) through an impedance network (which can be stray inductance of the secondary winding or an inductive-capacitive impedance). The main switches21(Q1) and22(Q2) at the upper part of the bridge are used to replace traditional diodes in a rectifier for the purpose of switching control. Each of Q1and Q2is formed by a pair of P-type MOSFETs connected in a back-to-back manner in order to form a bi-directional switch as shown inFIG. 12. When the common gate drive of each switch pair is deactivated (i.e the off-state of the bidirectional switch), one of the body diodes of the switch pair will always block the current flowing back from output load to input. The AC power flow to the output capacitor19(Cout) and the load R13can be regulated to achieve a constant DC output5(Vout) within a tight tolerance. The power flow control method can adopt the hysteresis control similar to Embodiment 1. A feedback circuit18formed by R14and R15is used to sense and feed the output voltage information to the non-inverting input of a main comparator13(U1). The sensed feedback voltage signal8is then compared with a voltage reference7(Vref,) at the inverting input of the main comparator13. The reference voltage7represents the desired output voltage level. When the feedback voltage signal8is higher than the reference voltage7, the output of the main comparator13will go to high, and vice versa.

Unlike Embodiment 1 with three drivers, there are two drivers23and24(also labeled as “Driver1” and “Driver2” respectively inFIGS. 12 and 13) in this embodiment as shown inFIG. 12andFIG. 13because the output of the main comparator13(U1) is connected to the base drives of the two drivers23and24directly through resistors R17and R18, respectively. An auxiliary low-power diode bridge30with an auxiliary capacitor31(C1) inFIG. 13is used to provide the DC voltage for the two drivers23and24. One driver23, formed by Q3to Q5, R3to R6and D9, is used to drive the main switch21(Q1) to turn ON or OFF according to the output voltage of the main comparator13(U1), and also to the switching voltage at point B. The other driver24, formed by Q6to Q8, R7to R12and D10, is used to drive the other main switch22(Q2) to turn ON or OFF according to the output voltage of the main comparator13(U1), and also to the switching voltage at point A.

The two modes of power flow disabling and power flow enabling are described further below.

Power Flow Disabling: The hysteresis control takes place when the output voltage8(Vout) is higher or lower than that of the reference voltage7(Vref). When the feedback voltage8(representing the actual output voltage) is higher than the reference voltage7(representing the desired voltage), the main comparator13(U1) generates a high signal. This high voltage will turn on Q5and Q8through R17and R18. Q5and Q8will in turn disable Q4and Q7. The bases of Q3and Q6are tied (through R1and R7respectively) to the high voltage of an auxiliary power, which is developed from an auxiliary low power diode bridge circuit30with auxiliary capacitor31(C1) inFIG. 13. The base currents of Q3and Q6saturate them and turn them on. Therefore, the gates of Q1and Q2are in the high state. The switches Q1and Q2are in the off-state as they are P-type MOSFETs. Since both the main switches Q1and Q2are turned on when the output voltage is higher than the reference voltage7, the output capacitor19(Cout) is cut off from the synchronous rectifier4and thus the power flow from the synchronous rectifier is disabled. During this power flow disabling period, the capacitor voltage will remain constant if there is no load discharging the output capacitor19. If the output capacitor19is loaded, the output capacitor will be discharged by the load and so the output voltage5will decrease as shown inFIG. 8. This situation will continue until the output voltage5decreases to the lower hysteresis voltage band where the feedback voltage8becomes less than the reference voltage7.

Power Flow Enabling: The input power can then be transferred to the output capacitor19again when the feedback voltage8is less than reference voltage7. In this case, the output of the main comparator13(U1) becomes “low”, meaning that the control of Q5and Q8is not influenced by U1under this condition. Q5is now controlled by the voltage at point B through resistor R19, while Q7is controlled by the voltage at point A through resistor R20. The main comparator13(U1) will not influence the normal self-driven functions of the drivers23and24and so the main switches Q1and Q2will turn on and off according to the normal self-driven mechanism described previously. During this power flow enabling period, the input current will charge the output capacitor19and the output voltage5will increase until it reaches the upper hysteresis voltage band.

Since the hysteresis voltage band is small, the power flow disabling and enabling mechanism ensures that the output voltage5is regulated to the desired level within a tight tolerance. Since the power flow from the synchronous rectifier4is in form of a current source, the current source can be used to charge the output capacitor19directly without using an extra output filter inductor. For wireless charging system, this feature is acceptable because the voltage ripple in the output of the secondary module can be relatively large because there will be a battery charging control inside the battery pack for further control of battery charging, such as control of the initial constant current charging and the subsequent constant voltage charging.

FIG. 14andFIG. 15show another secondary AC-DC bridge-type rectifying circuit4. Unlike Embodiments 1 and 2 in which the self-driven mechanism of the main switches21and22depend on the voltage at points A and B of the rectifier bridge, the main switches21and22in this approach use a “current-control” method. A synchronous rectifier bridge comprising two diodes D1and D2and the two main switches21and22(also labeled as “M1” and “M2” inFIGS. 14 and 15), which can be N-type MOSFETs, is fed by an AC input2having voltage V1via a reactive impedance typically consisting of the stray impedance of the secondary winding3and a series input capacitor25. M1and M2are intentionally used to replace two ordinary diodes at the lower portion of the bridge for the purpose of switching control and reduction of conduction loss. A tertiary switch32(also labeled as “M3” inFIGS. 14 and 15) is present in the return path of the AC input2. The AC power delivered to the output load R8with an output filtering capacitor19(Cout) can be regulated to achieve a constant DC output5(Vout).

The control method employs a hysteresis control mechanism. A feedback circuit18, formed by R2and R3, is used to sense the output voltage5and feed it into the non-inverting input of a main comparator13(U3) inFIG. 15. The sampled voltage8is then compared with a predefined voltage reference7(V2) at the inverting input of the main comparator13(U3). When the sampled voltage signal8is higher than the reference voltage7, the output of the main comparator13(U3) will go to high. When the sampled voltage signal8is lower than the reference voltage7, the output of the main comparator13(U3) will go to low. That means when the output voltage5is higher than desired value, the main comparator13goes to high too. When the output voltage5is lower than the desired value, the main comparator13goes to low.

There are three drivers inFIG. 14andFIG. 15. One driver23(also labeled “Driver1” inFIGS. 14 and 15), formed by U1, U4, Q3and Q4, is used to drive M1to switch ON or OFF according to the voltage sensed at the sensing resister R6. The sensing method can operate by sensing the voltage drop of the internal resistance of the switching MOSFET (M1). Another driver24(also labeled as “Driver2” inFIGS. 14 and 15), formed by U2, U5, Q5and Q6, is used to drive M2to turn ON or OFF according to the voltage sensed at the sensing resister R7. Again, the sensing method can operate by sensing the voltage drop of the internal resistance of the switching MOSFET M2. M1and M2will be turned off when their currents are reversed (simulating the turn-off mechanism of the diode-reverse-recovery behavior of a diode). This can be realized with the help of driver comparators U1and U2. When the current in either M1or M2flows in the negative sense in the sensing resistor (R6for M1and R7for M2), the inverting input of the driver comparator (U1for M1and U2for M2) is more positive than the voltage at the non-inverting input. The output of U1or U2will go low. This results in the turning off of M1or M2. Otherwise, M1and M2are normally turned on for carrying any forward current.

The output of the main hysteretic comparator13(U3) feeds a driving signal to the inputs of the logical OR gates of U4and U5, which are intentionally connected together at one of their inputs. Therefore, both sensing signals from the output feedback voltage8and current flowing through switching MOSFETs M1and M2will determine the ON time of the switching MOSFETs.

The two modes of power flow disabling and power flow enabling are described further below.

Power flow disabling: When the output feedback signal8is higher than the reference voltage7, the output of the main comparator13(U3) goes high. Through U4and U5, both M1and M2will be turned on simultaneously, resulting in points A and B being shorted together to form a closed current loop in the secondary winding3and its series impedance.

The longer the ON time, the less the power it will deliver to the output. A third driver33(also labeled as “Driver3” inFIGS. 14 and 15), formed by Q1, Q2and Q7, in the return path of the AC input2, has the function of cutting off the current flow when the output of driver comparator U1is low, as well as the output of the main comparator U3is high. A high output of the main comparator U3means that output voltage8is higher than the threshold voltage7(V2). Less power will be delivered to the load by increasing the ON times of both main MOSFETs M1and M2. However, with a continuous current flow from input2(V1), M1and M2will create unnecessary energy loss. The tertiary switch32(M3), in the form of a switching MOSFET, can help to reduce the energy loss if it is in the OFF state, but must be OFF at the instant when the current flow of switching MOSFET M1is in a reverse direction (zero current crossing)—driver comparator U1is at low state. Q7will then tie the switching MOSFET M3to the OFF state. This will avoid any current transient when input current is flowing continuously to the output load. In the normal ON state, switching MOSFET M3is driven by Q1and Q2, both of which have an auxiliary supply from the input through D3. Resistor R5biases Q1and M3connects the input and the load normally.

Power Flow Enabling: The input power can then be transferred to the output capacitor19again when the feedback voltage8is less than reference voltage7. In this case, the output of the main comparator13(U3) becomes “low”. The third driver33is self-biased in such a way that switch M3is normally in a closed condition. In this condition, it will not influence the normal self-driven functions of the other two drivers23and24and so the main switches M1and M2will turn on and off according to the normal self-driven mechanism described previously. During this power flow enabling period, the input current will charge the output capacitor19and the output voltage5will increase until it reaches the upper hysteresis voltage band.

The present invention also provides in another aspect a method of converting an AC input provided by a power transfer winding. One broad embodiment of the method includes rectifying the AC input2into the rectified output5, and controlling the rectification on the basis of a comparison between the reference signal7and the voltage feedback signal8, the voltage feedback signal being based on the rectified output5. It will be appreciated that the foregoing description also describes other embodiments of this method.

Advantageously, the present invention offers solutions to provide a “single-stage” AC-DC power converter:

(a) with output voltage regulation;

(b) with self-driven functionality for a synchronous rectifier on the secondary side of a transformer;

(c) without voltage control from the primary side of the transformer;

(d) without an extra DC-DC converter; and

(e) without an output inductor.

Based on a power flow enabling/disabling control signal, the power flow from the rectifying means of the AC-DC converter of the invention to an output capacitor is controlled in a way so that the output DC voltage is kept to a desired voltage level within a tight tolerance.

Although the invention has been described with reference to specific examples, it will be appreciated by those skilled in the art that the invention can be embodied in many other forms. It will also be appreciated by those skilled in the art that the features of the various examples described can be combined in other combinations.