A frequency modulation circuit includes a VCO, a DIV, a MIX, a single-phase differential converter, and a signal processing circuit. The signal processing circuit performs differential arithmetic processing of an intermediate frequency signal with a program of a microcomputer according to a quadrature demodulation scheme and, thereafter, measures a frequency from phase information, performs n-th order polynomial (n is an integer equal to or larger than 2) approximation on time-frequency data of an IF signal output by a chirp modulation control voltage after inverse function correction, and performs modulation correction for correcting a time error.

FIELD

The present invention relates to a frequency modulation circuit of a radar that performs frequency modulation, an FM-CW radar, and a high-speed modulation radar.

BACKGROUND

A conventional FM-CW radar adopting a Frequency Modulated-Continuous Waves (FM-CW) scheme, a circuit configuration of which is relatively simple, measures a frequency of a beat signal of a frequency-modulated transmission signal and a reception signal reflected from a target object, and calculates a relative distance and relative speed with respect to the target object. A Voltage Controlled Oscillator (VCO) is provided in the conventional FM-CW radar adopting the FM-CW scheme.

The VCO outputs an oscillation frequency signal frequency-modulated according to a modulation control voltage. The oscillation frequency signal is requested to have high modulation linearity. However, because the VCO is a semiconductor device that controls a frequency with a voltage, the VCO shows a nonlinear frequency characteristic with respect to the voltage. The frequency characteristic of the VCO deviates according to individual variability or temperature characteristics. Therefore, in a shipment inspection process, work for measuring the oscillation frequency signal of the VCO and adjusting the oscillation frequency signal concerning modulation linearity is essential. The adjusting work is a hindrance to a reduction of an inspection time for mass production.

To obtain the modulation linearity of the VCO, the conventional FM-CW radar represented by Patent Literature 1 corrects the oscillation frequency signal of the VCO according to a Look Up Table (LUT) for modulation control voltage, or is provided with a mechanism for measuring the oscillation frequency signal of the VCO on a frequency modulation circuit to perform feedback control, to thereby cope with characteristic change of the VCO due to aged deterioration after shipment.

CITATION LIST

Patent Literature

SUMMARY

Technical Problem

The conventional FM-CW radar represented by Patent Literature 1 performs frequency division of the oscillation frequency signal of the VCO with a divider (DIV), which is a frequency divider, and thereafter converts an intermediate frequency (IF) signal, which is a signal having an intermediate frequency down-converted using a local signal, into a digital signal with an Analog to Digital Converter (ADC). Thereafter, the FM-CW radar measures an instantaneous frequency with a microcomputer from instantaneous phase information of the IF signal according to a quadrature demodulation scheme.

The oscillation frequency of the VCO is calculated from a frequency and a frequency division number of the local signal. However, there is a problem in that time-frequency data measured in the quadrature demodulation scheme is poor in measurement accuracy and high linearity of frequency modulation cannot be obtained even when the feedback control is performed.

The present invention has been devised in view of the above, and an object of the present invention is to obtain a frequency modulation circuit that can obtain high linearity of frequency modulation.

Solution to Problem

A frequency modulation circuit according to an aspect of the present invention includes: a digital-analog converter to output modulation control time-dependent voltage data; a voltage control oscillator to oscillate, based on the modulation control time-dependent voltage data output from the digital-analog converter, an oscillation frequency signal; a frequency divider to perform frequency division of the oscillation frequency signal of the voltage control oscillator and output the oscillation frequency signal; a frequency converter to down-convert a frequency division signal output from the frequency divider; a single-phase differential converter to convert an intermediate frequency signal of single-phase output from the frequency converter into differential signals and output the differential signals; an analog-digital converter to convert, concerning the differential signals output from the single-phase differential converter, analog signals of the differential signals into digital signals; and a signal processing circuit to perform frequency measurement based on the differential signals of the analog-digital converter, update the modulation control time-dependent voltage data based on the measured frequency, and correct a time error of the oscillation frequency signal of the voltage control oscillator.

Advantageous Effects of Invention

According to the present invention, there is an effect that it is possible to obtain high linearity of frequency modulation.

DESCRIPTION OF EMBODIMENTS

A frequency modulation circuit, an FM-CW radar, and a high-speed modulation radar according to an embodiment of the present invention are explained in detail below with reference to the drawings. Note that the present invention is not limited by this embodiment.

Embodiment

FIG. 1is a diagram illustrating a frequency modulation circuit of an FM-CW radar according to an embodiment of the present invention. An FM-CW radar100-1illustrated inFIG. 1includes a frequency modulation circuit110-1, a transmission antenna1connected to the frequency modulation circuit110-1, and a reception antenna14connected to the frequency modulation circuit110-1.

The frequency modulation circuit110-1includes: a high-frequency circuit2connected to the transmission antenna1and the reception antenna14; and a signal processing circuit6that generates a triangular wave voltage signal, which is a modulation control voltage based on a modulated signal to be output from the high-frequency circuit2, and outputs the triangular wave voltage signal to a VCO5of the high-frequency circuit2. The frequency modulation circuit110-1includes a single-phase differential converter18, a baseband amplifier circuit11, a Low Pass Filter (LPF)24, an LPF25, a control circuit15, and an ambient temperature monitor23.

The high-frequency circuit2includes: the VCO5that generates an oscillation frequency signal, which is a modulated signal subjected to frequency modulation in accordance with a frequency control voltage transmitted from the signal processing circuit6; and a power distributor4that outputs most of an output f the VCO5to an amplifier3and outputs the remaining output to a mixer (MIX)12, which is a frequency converter, as a local signal.

The high-frequency circuit2includes: an amplifier3that amplifies an output of the power distributor4and outputs the output to the transmission antenna1; a low-noise amplifier13that amplifies a reception signal received by the reception antenna14; and the MIX12that down-converts, with a local signal, the signal amplified by the low-noise amplifier13into an IF signal and outputs the IF signal.

The high-frequency circuit2includes: a DIV19that performs frequency division of the oscillation frequency signal of the VCO5and outputs the oscillation frequency signal; a reference frequency generator21that outputs a local signal; and a MIX20that mixes the frequency division signal output from the DIV19and the local signal output from the reference frequency generator21, down-converts, with the local signal, the frequency division signal into an IF signal and outputs the IF signal. A frequency of the IF signal is equivalent to a frequency of a difference between a frequency of the frequency division signal and a frequency of the local signal.

The elements of the high-frequency circuit are configured by Microwave Monolithic ICs (MMICs).

The single-phase differential converter18converts a single-phase IF signal, that is, a single-end signal output from the MIX20into differential signals and outputs the differential signals. The LPF25reduces unnecessary waves and noise of a positive phase-side differential signal output from the single-phase differential converter18and outputs the positive phase-side differential signal. The LPF24reduces unnecessary waves and noise of an opposite phase-side differential signal output from the single-phase differential converter18and outputs the opposite phase-side differential signal. The baseband amplifier circuit11amplifies the output signal of the MIX12and outputs the amplified signal as a reception signal.

The output signal of the LPF24is input to an ADC16in the signal processing circuit6. The output signal of the LPF25is input to an ADC17in the signal processing circuit6. The output signals are used for update of data for a triangular wave voltage signal in a LUT22.

The signal processing circuit6includes: a microcomputer10, which is a main circuit unit that mainly performs transmission processing and measurement processing; and a Digital to Analog Converter (DAC)7, which is a digital-analog converter that converts a triangular wave voltage signal transmitted from the microcomputer10into an analog signal and outputs the analog signal to the VCO5of the high-frequency circuit2.

The signal processing circuit6includes: the ADC16that converts the output signal of the LPF24into a digital signal; the ADC17that converts the output signal of the LPF25into a digital signal; and an ADC9that converts the reception signal output from the baseband amplifier circuit11into a digital signal and outputs the digital signal to the microcomputer10.

The microcomputer10includes the LUT22that stores data for a triangular wave voltage signal given to the VCO5and a nonvolatile memory8. An ambient temperature monitor23that measures an ambient temperature of the microcomputer10is connected to the microcomputer10.

The control circuit15controls, with the microcomputer10, various control voltages supplied to the MMICs in the high-frequency circuit2. Specifically, the MMICs in the high-frequency circuit2have variations depending on manufacturing lots. Therefore, control voltage values adjusted and determined for each of the MMICs are stored in the nonvolatile memory in the microcomputer10. During actual operation, the microcomputer10reads out the control voltage values from the nonvolatile memory8and supplies the control voltage values to the MMICs in the high-frequency circuit2via the control circuit15.

The operation of the FM-CM radar100-1is explained below.

The VCO5generates, on the basis of the triangular wave voltage signal output from the signal processing circuit6, an FM-CW signal, which is an oscillation frequency signal of high-frequency that includes a rising modulated signal, a frequency of which rises in a fixed period, and a falling modulated signal, a frequency of which falls in the fixed period.

The FM-CM signal is input to the power distributor4. Most of the FM-CW signal is supplied to the transmission antenna1. A millimeter-wave radio wave is irradiated from the transmission antenna1toward a target object. The remaining FM-CM signal is supplied to the MIX12as a local signal.

A reflected wave reflected on the target object is captured by the reception antenna14and input to the MIX12as a reception signal. The MIX12mixes the reception signal input from the reception antenna11and the local signal supplied from the power distributor4, and outputs a beat signal having a frequency equivalent to a frequency difference between the signals. The beat signal is amplified to an appropriate level by the baseband amplifier circuit11and input to the microcomputer10via the ADC9.

The microcomputer10includes a signal processing unit10-1that calculates a distance to the target object and relative speed from a frequency in a rising modulation period and a frequency in a falling modulation period in the input beat signal, and outputs relative distance information on the distance to the target object and information on the relative speed with respect to the target object. Note that these kinds of information output from the signal processing unit10-1are transmitted to a vehicle control unit200provided in a vehicle mounted with the FM-CW radar100-1. The vehicle control unit200has a function of collectively controlling the operation of the vehicle mounted with the FM-CW radar100-1. The vehicle control unit200performs, based on these kinds of information, processing such as clutter removal and target identification.

On the other hand, the frequency of the FM-CW signal from the VCO5is reduced to a frequency of one fraction of an integer by the DIV19and input to the MIX20.

In the MIX20, the frequency division signal output from the DIV19and the local signal output from the reference frequency generator21are mixed and an IF signal is output.

The IF signal is converted into differential signals by the single-phase differential converter18. After unnecessary waves and noise are removed from the differential signals by the LPF24and the LPF25, the differential signals are input to the microcomputer10via the ADC16and the ADC17.

The microcomputer10measures, according to the quadrature demodulation scheme, a frequency from phase information of the IF signal, performs correction processing using a result of the measurement, calculates a voltage table necessary for securing modulation linearity of the frequency of the oscillation frequency signal, and updates the LUT22for a control voltage. Consequently, data for a triangular wave voltage signal output in the next cycle to the VCO5is updated. The updated data for the triangular wave voltage signal is converted into an analog signal, which is modulation control time-dependent voltage data, and input to the VCO5by the DAC7.

Note that, as for an initial value of modulation control time-dependent voltage data, predetermined default chirp data is stored in the microcomputer10and output from the microcomputer10, and the default chirp data is not output from when the LUT22is updated after a frequency is measured.

Correction processing for obtaining modulation linearity of the VCO5is explained.

FIG. 2is a flowchart illustrating a modulation correcting operation in the microcomputer illustrated inFIG. 1.FIG. 3is a timing chart for explaining the modulation correcting operation in the microcomputer illustrated inFIG. 2. A waveform of a modulation control voltage is illustrated on the upper side ofFIG. 3. A modulation frequency characteristic is illustrated on the lower side ofFIG. 3. Signs (1) to (8) illustrated inFIG. 3correspond to numbers S1to S8illustrated inFIG. 2.

The microcomputer10outputs a modulation control voltage of fault chirp, whereby the VCO5outputs a modulated signal of default chirp corresponding to the modulation control voltage (S1). The microcomputer10measures a frequency of a first-time VCO frequency division signal (S2). When the LUT22is updated, voltage-frequency data corresponding to the default chip data is subjected to n-th order polynomial approximation (n is an integer equal to or larger than 2), for example, cubic function approximation. A voltage table necessary for securing linearity is calculated from a result of the n-th order polynomial approximation. In this calculation, an inverse function thereof is used for correction and the modulation correction is performed (S3).

The microcomputer10outputs a modulation control voltage after the inverse function correction, whereby the VCO5outputs a modulated signal corresponding to the modulation control voltage (S4). The microcomputer10measures frequencies of second and subsequent-time VCO frequency division signals (S5). The microcomputer10subjects time-frequency data to the n-th order polynomial approximation, for example, the cubic function approximation. When the LUT22is updated for the second and subsequent times, the microcomputer10calculates a time error with respect to an ideal frequency straight line and performs correction of time data. In this calculation, the time error is used for correction and the modulation correction is performed (S6). The microcomputer10outputs a modulation control voltage after the time error correction, whereby the VCO5outputs a modulated signal corresponding to the modulation control voltage (S7). The modulation correction is completed (S8). The modulation frequency is corrected to a waveform illustrated in (8) inFIG. 3by the operation in S1to S8.

A calculation method for an error time in S5to S8inFIG. 2is explained in detail.FIG. 4is a diagram illustrating a configuration for measuring a frequency from phase information in accordance with a quadrature demodulation scheme using an IF signal input to the signal processing circuit illustrated inFIG. 1. The ADC16and the ADC17are equivalent to the ADC16and the ADC17illustrated inFIG. 1, respectively. After the IF signal is digitized, the microcomputer10performs differential arithmetic processing to calculate V′. The microcomputer10includes an LPF10-2, a MIX10-3, a frequency generating unit10-4, a MIX10-5, an LPF10-6, an LPF10-7, an instantaneous-phase-difference calculating unit10-8, an instantaneous-frequency calculating unit10-9, and a multiplying unit10-10. Quadrature demodulation processing is performed by the MIX10-3, the frequency generating unit10-4, the MIX10-5, and the multiplying unit10-10. Specifically, data sampled by the ADC16is separated into two signals of an I (In-phase) component and a Q (Quadrature) component according to quadrature detection. In the LPF10-2in a first stage, reduction processing for high-frequency and unnecessary wave components of the digitized IF signal is performed. After the multiplying unit10-10performs the quadrature detection to thereby separate the IF signal into two signals of an I (In-phase) signal and a Q (Quadrature) signal, the LPFs10-6and10-7in a second stage reduce a sum frequency component (fIF+fLO) generated by multiplication processing and allow only a difference frequency component (fIF-fLO) to pass. After an instantaneous phase difference Δθ=Tan−1(Q/I) of the IF signal is calculated from the I signal and the Q signal by the instantaneous-phase-difference calculating unit10-8, an instantaneous frequency f=Δθ/Δt of the IF signal is calculated by the instantaneous-frequency calculating unit10-9. Δt represents a time step.

FIG. 5is a diagram for explaining a time calculating method in the signal processing circuit illustrated inFIG. 1.FIG. 6is a diagram for explaining an ideal-frequency-curve calculating method in the signal processing circuit illustrated inFIG. 1. In each ofFIG. 5andFIG. 6, the horizontal axis indicates time and the vertical axis indicates a frequency.

(1) Frequency Measurement

As illustrated inFIG. 4, an opposite phase-side differential signal V+output from the single-phase differential converter18is input to the ADC16and a positive phase-side differential signal V−is input to the ADC17. Each of the opposite phase-side differential signal V+and the positive phase-side differential signal V−is digitized. Thereafter, differential arithmetic processing is performed by a program of the microcomputer10. Then, a frequency of the IF signal is measured from phase information according to the quadrature demodulation scheme. Note that, in S2, the same frequency measurement is performed. The time-frequency data measured in S5is represented as fDECTECT1(t).

A method of calculating a time error with respect to an ideal frequency from fDETECT1(t) is explained below.

(2) Time Error Calculation

The time-frequency data fDETECT1(t) is subjected to the n-th order polynomial approximation in a relation between time and a frequency. Frequency measurement data fDETECT1,A(t) polynomial approximation is indicated by Expression (1) described below. When Expression (1) described below is expanded to i-th discrete data, Expression (2) described below is obtained. In the expressions, an, an−1, and a0(n is a natural number) represent coefficients and Δt represents a time step.
[Math. 1]
fDETECT1,A(t)=antn+an−1tn−1Λ+a0(1)
[Math. 2]
fDETECT1,A(i)=an(i·Δt)n+an−1(i·Δt)n−1Λ+a0(2)

A time error ∂t(t) is calculated from Expression (3) described below. When Expression (3) described below is expanded to i-th discrete data, ∂t(i) indicated by Expression (4) below is obtained.

In Expression (3) described above, α′(t) represents a modulation gradient and is calculated by differentiating fDETECT1,Aonce. The modulation gradient α′(t) is indicated by Expression (5) described below. When Expression (4) described above is expanded to i-th discrete data, α′(i) indicated by Expression (6) described below is obtained.

In Expression (3) described above, fIDEAL(t) represents an ideal frequency curve and is as indicated by Expression (7) described below. When Expression described below is expanded to i-th discrete data, Expression (8) described below is obtained. In Expression (7) and Expression (8) described below, α represents a modulation gradient theoretical value.
[Math.7]
fIDEAL(t)=αt+β(7)
[Math. 8]
fIDEAL(i)=α·(i·Δt)+β  (8)

A calculation method for β in Expression (7) and Expression (8) described above is explained below. β is calculated by resetting fIDEAL(t) from a modulation center point T1+T/2 of a frequency measurement result fDETECT1,Ato t=0. Specifically, fIDEALat an A point (T1+T/2) illustrated inFIG. 6is calculated by Expression (9) described below. A relation of Expression (10) described below holds from a condition that fDETECT1,Aand fIDEALare equal at the A point (T1+T/2). From Expression (10) described below, β indicated by Expression (11) described below is calculated.

The microcomputer10calculates any i-th time error ∂t(i) from Expression (2), Expression (8), and Expression (11) described above and additionally corrects the LUT22after the inverse function correction using the calculated time error ∂t(i) to obtain high modulation linearity as illustrated in (7) inFIG. 3.

FIG. 7is a diagram illustrating a comparative example compared with the FM-CW radar according to the embodiment of the present invention. An FM-CW radar100illustrated inFIG. 7is different in the following points compared with the FM-CW radar100-1in the embodiment.

(1) In a frequency modulation circuit110of the FM-CW radar100, a single-phase IF signal output from the MIX20is input to an LPF26.

(2) An output signal of the LPF26is input to the ADC16in the signal processing circuit6and converted into a digital signal.

The microcomputer10illustrated inFIG. 7measures an instantaneous frequency from instantaneous phase information on the IF signal, in accordance with the quadrature demodulation scheme using the signal converted into the digital signal by the ADO16. The microcomputer10calculates an oscillation frequency of the VCO5from a frequency and a frequency division number of a local signal source. Time-frequency data measured in the quadrature demodulation scheme is poor in measurement accuracy. High linearity of frequency modulation cannot be obtained by performing feedback-control on the time-frequency data.

After the signal processing circuit6performs differential arithmetic processing of the IF signal with a differential arithmetic processing program in accordance with the quadrature demodulation scheme, the FM-CW radar100-1according to the embodiment of the present invention measures a frequency from phase information obtained by program execution processing of the microcomputer10, performs n-th order polynomial (n is an integer equal to or larger than 2) approximation on time-frequency data of the IF signal output based on a modulation control voltage after the correction using inverse function, and performs modulation correction for correcting a time error. Consequently, effects explained below can be obtained.

(1) Frequency Measuring Method

A DC offset, an even order harmonic, and common mode noise included in a signal can be reduced by converting an IF signal of a frequency division that is an output of a VCO into a differential output and performing the differential arithmetic processing with the differential arithmetic processing program of the microcomputer10. A measurement error during frequency measurement in the quadrature demodulation scheme can be improved. When a differential ADC is used as the ADC16illustrated inFIG. 7, not only component cost but also an occupied area of a module configuring the differential ADC increases. However, in the FM-CW radar100-1according to this embodiment, the ADCs16and17corresponding to only a single end can be used. It is possible to reduce component cost and perform highly accurate frequency measurement. Modulation correction accuracy is improved by performing the highly accurate frequency measurement. As a result, the modulation linearity is improved. The distance to the target object and the relative speed can be more accurately calculated. A modulation band of the VCO5is wide. Even after the oscillation frequency signal from the VCOS5is down-converted into an IF signal, in a low frequency range of the modulation band, in particular, a secondary harmonic component affects the modulation band. It is difficult to suppress the secondary harmonic component with the LPF26configured by hardware. However, by performing the differential arithmetic processing, the secondary harmonic component in the low frequency range of the modulation band can be reduced.

In the microcomputer10, based on a single-phase IF signal output from the MIX20, an instantaneous frequency f of the IF signal is measured. The instantaneous frequency f of the IF signal is a frequency measured by the instantaneous-frequency calculating unit10-9explained above. Concerning a frequency-time waveform of a frequency in a rising modulation period of the IF signal and a frequency in a falling modulation period of the IF signal, the microcomputer10performs the modulation correction explained with reference toFIG. 2andFIG. 3. After the FM-CW radar100-1transmits an FM-CW signal generated based on the LUT22after the modulation correction and receives a reflected wave from the target object, a reception beat signal down-converted by the MIX12is converted into a digital signal by the ADC9. The reception beat signal converted into the digital signal is subjected to Fast Fourier Transform (FFT) processing and signal processing. A distance to the target object and relative speed are calculated. As a result of the highly accurate frequency measurement of the VCO5, modulation correction accuracy is improved, the modulation linearity changes to a favorable characteristic, and calculation accuracy of a distance to the target object and relative speed by the FM-CW radar100-1is improved.

When the frequency measurement is performed by the quadrature demodulation scheme, a measurement error is caused by noise, a DC offset, and a harmonic component included in the IF signal. Therefore, when a time difference is calculated based on time-frequency data including the measurement error and time error correction is performed, it is difficult to obtain high modulation linearity with the highly accurate modulation correction. According to this embodiment, the measurement error at the frequency measurement can be absorbed by approximating the measurement error with the n-th order polynomial as in Expression (1) described above. Highly accurate modulation correction is enabled by the absorption of the measurement error. In a module of a general FM-CW radar, it can be predicted that a measurement error of a frequency is instantaneously caused by occurrence of vibration, noise, and electromagnetic noise, which are disturbances. However, the measurement error of the frequency cannot be absorbed. According to this embodiment, the instantaneously caused measurement error of the frequency can be absorbed by the n-th order polynomial approximation.

On the other hand, in the VCO5, a certain fixed error is caused in frequency measurement due to a temperature drift according to a physical property of a semiconductor. Therefore, when an intercept β in Expression (7) and Expression (8) described above of an ideal frequency curve is fixed at a theoretical value like the gradient α that is fixed, a correction amount becomes excessive. Accurate modulation correction sometimes cannot be performed in such a case. Therefore, in this embodiment, by using Expression (11) described above, it is possible to prevent the correction amount from becoming excessive. Thus, stable modulation correction is enabled.

FIG. 8is a diagram illustrating a first modification of the FM-CW radar according to the embodiment of the present invention. In a frequency modulation circuit110-2of an FM-CW radar100-2illustrated inFIG. 8, the reference frequency generator21and the MIX20illustrated inFIG. 1are omitted.

The frequency modulation circuit110-2includes the DIV19that performs frequency division of an oscillation frequency signal of the VCO5and outputs the oscillation frequency signal, and the single-phase differential converter18that converts a frequency division signal output from the DIV19into differential signals and outputs the differential signals. One of the differential signals is input to the LPF24and the other of the differential signals is input to the LPF25.

The microcomputer10of the signal processing circuit6illustrated inFIG. 8measures a frequency from phase information of differential signals according to the quadrature demodulation scheme, performs n-th order polynomial (n is an integer equal to or larger than 2) approximation on differential signals output based on a modulation control voltage of default chirp, and performs modulation correction for correcting a time error of the differential signals.

With the frequency modulation circuit110-2, the reference frequency generator21and the MIX20illustrated inFIG. 1are unnecessary. The configuration of the frequency modulation circuit110-2is simplified, manufacturing cost can be reduced, and reliability is improved.

FIG. 9is a diagram illustrating a second modification of the FM-CW radar according to the embodiment of the present invention. In a frequency modulation circuit110-3of an FM-CW radar100-3illustrated inFIG. 9, the single-phase differential converter18is omitted.

The frequency-modulation circuit110-3includes the DIV19and the MIX20that down-converts a frequency division signal output from the DIV19and converts a single-phase IF signal into differential signals and outputs the differential signals. One of the differential signals is input to the LPF24and the other of the differential signals is input to the LPF25.

The microcomputer10of the signal processing circuit6illustrated inFIG. 9measures a frequency from phase information of the IF signal in accordance with the quadrature demodulation scheme, performs n-th order polynomial (n is an integer equal to or larger than 2) approximation on the IF signal output based on a modulation control voltage of default chirp, and performs modulation correction for correcting a time difference of the IF signal.

With the frequency modulation circuit110-3, the single-phase differential converter16illustrated inFIG. 1is unnecessary. The configuration of the frequency modulation circuit110-3is simplified, manufacturing cost can be reduced, and reliability is improved.

FIG. 10is a diagram illustrating a third modification of the FM-CW radar according to the embodiment of the present invention. In a frequency modulation circuit110-4of an FM-CW radar100-4illustrated inFIG. 10, the MIX20and the reference frequency generator21illustrated inFIG. 1are omitted.

The frequency modulation circuit110-4includes a Balance unbalanced (Balun)27, which is a balance-unbalance converter, instead of the single-phase differential converter18illustrated inFIG. 1. The Balun27converts a single-end frequency division signal output from the DIV19into differential signals of a differential type and outputs the differential signals. One of the differential signals is input to the LPF24and the other of the differential signals is input to the LPF25.

The microcomputer10of the signal processing circuit6illustrated inFIG. 10measures a frequency from phase information of the differential signals according to the quadrature demodulation scheme, performs n-th order polynomial (n is an integer equal to or larger than 2) approximation on the differential signals output based on a modulation control voltage of default chirp, and performs modulation correction for correcting a time difference of the differential signals.

With the frequency modulation circuit110-4, the reference frequency generator21and the MIX20illustrated inFIG. 1are unnecessary. The configuration of the frequency modulation circuit110-2is simplified, manufacturing cost can be reduced, and reliability is improved.

FIG. 11is a diagram illustrating a fourth modulation of the FM-CW radar according to the embodiment of the present invention. In a frequency modulation circuit110-5of an FM-CW radar100-5illustrated inFIG. 11, the MIX20, the reference frequency generator21, and the single-phase differential converter18illustrated inFIG. 1are omitted.

The DIV19of the frequency modulation circuit110-5performs frequency division of an oscillation frequency signal of the VCO5and converts a frequency division signal into differential signals and outputs the differential signals. One of the differential signals is input to the LPF24and the other of the differential signals is input to the LPF25.

The microcomputer10of the signal processing circuit6illustrated inFIG. 11measures a frequency from phase information of the differential signals in accordance with the quadrature demodulation scheme, performs n-th order polynomial (n is an integer equal to or larger than 2) approximation on the differential signals output based on a modulation control voltage of default chirp, and corrects a time difference of the differential signals.

With the frequency modulation circuit110-5, the MIX20, the reference frequency generator21, and the single-phase differential converter18illustrated inFIG. 1are unnecessary. The configuration of the frequency modulation circuit110-5is simplified, manufacturing cost can be reduced, and reliability is improved.

FIG. 12is a diagram illustrating a fifth modification of the FM-CW radar according to the embodiment of the present invention. A frequency modulation circuit110-6of an FM-CW radar100-6illustrated inFIG. 12includes the Balun27instead of the single-phase differential converter18illustrated inFIG. 1.

The Balun27converts a single-phase IF signal output from the MIX20into differential signals and outputs the differential signals. One of the differential signals input to the LPF24and the other of the differential signals is input to the LPF25.

The microcomputer10of the signal processing circuit6illustrated inFIG. 12measures a frequency from phase information of the IF signal in accordance with the quadrature demodulation scheme, performs n-th order polynomial (n is an integer equal to or larger than 2) approximation on the differential signals output based on a modulation control voltage of default chirp, and performs modulation correction for correcting a time difference of the IF signal.

With the frequency modulation circuit110-6, it is unnecessary to use the differential ADC and an increase in an occupied area can be reduced as in the frequency modulation circuit110-1illustrated inFIG. 1.

The example is explained above in which the frequency modulation circuit according to this embodiment is provided in the FM-CW radar, which is an example of the radar that performs frequency modulation. However, the frequency modulation circuit according to this embodiment can be provided in a high-speed modulation radar. Both of the FM-CM radar and the high-speed modulation radar are radars that perform frequency modulation. However, the FM-CW radar is a radar that performs frequency modulation in a broad sense and the high-speed modulation radar is a radar that performs frequency modulation in a narrow sense.FIG. 13is a diagram illustrating a high-speed modulation radar according to the embodiment of the present invention.FIG. 14is a diagram representing frequency specification in the FM-CM radar according to the embodiment of the present invention.FIG. 15is a diagram representing frequency specification in the high-speed modulation radar according to the embodiment of the present invention.

A difference between the FM-CM radar100-1illustrated inFIG. 1and a high-speed modulation radar100-7illustrated inFIG. 13is that the arithmetic processing in the signal processing unit10-1is different. The high-speed modulation radar100-7illustrated inFIG. 13includes the frequency modulation circuit110-1illustrated inFIG. 1. However, the high-speed modulation radar100-7can include any one of the frequency modulation circuits110-2to110-6instead of the frequency modulation circuit110-1. By including any one of the frequency modulation circuits110-2to110-6, the same effects as the effects of the FM-CM radars100-2to100-6can be obtained. The arithmetic processing in the signal processing unit10-1provided in each of the FM-CM radars100-1to100-6and the high-speed modulation radar100-7is explained below.

The vertical axis ofFIG. 14represents a frequency and the horizontal axis ofFIG. 14represents time. The signal processing unit10-1of the frequency conversion circuit provided in the FM-CW radars100-1to100-6selects a combination of an UP frequency fupand a DOWN frequency fDNindicated by Expression (12) and Expression (13), respectively, described below and thereafter solves simultaneous equations to calculate a relative distance to the target object and relative speed. Note that, in Expression (12) and Expression (13) described below, C represents speed of light, B represents a modulation bandwidth, T represents a modulation time, λ represents a wavelength, R represents a relative distance, and v represents relative speed.

The vertical axis ofFIG. 15represents a frequency and the horizontal axis ofFIG. 15represents time. The signal processing unit10-1of the frequency modulation circuit provided in the high-speed modulation radar100-7calculates the relative distance R in accordance with Expression (14) described below. In the high-speed modulation radar100-7, because chirp speed is high compared with the FM-CW radars100-1to100-6, the item of the relative speed v can be neglected compared with the relative distance R. Therefore, 2v/λ can be regarded as 0. After collecting data for each distance bin, the signal processing unit10-1performs doppler processing to calculate the relative speed v.

In the high-speed modulation radar100-7, compared with the FM-CW radars100-1to100-6, the modulation time T is different. As the modulation time T of the high-speed modulation radar100-7, a time equal to or shorter than 1/100 of the modulation time T of the FM-CW radars100-1to100-6can be exemplary illustrated. Therefore, the FM-CW radars100-1to100-6can reduce sampling frequencies in the ADCs16and17compared with the high-speed modulation radar100-7and, therefore, can reduce power consumption. The high-speed modulation radar100-7has high modulation speed compared with the FM-CW radars100-1to100-6. The high-speed modulation radar100-7can increase processing speed of processing such as clutter removal and target identification in the vehicle control unit200.

The FM-CW radars100-1to100-6according to this embodiment have high linearity of frequency modulation. Therefore, the FM-CW radars100-1to100-6can more highly accurately calculate a relative distance to the target object and relative speed. The high-speed modulation radar100-7according to this embodiment has high linearity of frequency modulation. Therefore, the high-speed modulation radar100-7can more highly accurately calculate relative distance to the target object and relative speed. Further, the high-speed modulation radar100-7according to this embodiment has higher distinctiveness compared with the FM-CW radars100-1to100-6. The high-speed modulation radar100-7can calculate an actual distance to the target object.

The configuration explained above in the embodiment indicate an example of the content of the present invention. The configuration can be compared with other publicly-known technologies. A part of the configuration can be omitted or changed in a range not departing from the spirit of the present invention.

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