Variable attenuator

A current bias-type variable attenuator. Two PIN diodes are disposed in series to each other and in the forward direction on a bias current path. When a bias voltage is applied to a bias terminal, d.c. current flows in the order of a combination output port, an in-phase output port, a first PIN diode, an input port, a 900.degree.--delayed phase output port, and a second PIN diode, or in the order of the combination output port, the in-phase output port, the first PIN diode, the second PIN diode, the 900.degree.--delayed phase output port, and the input port. Since d.c. current flowing through the first PIN diode and d.c. current flowing through the second PIN diode have an equal value, an RF resistance of the first PIN diode and that of the second PIN diode are balanced.

BACKGROUND OF THE INVENTION 
a) Field of the Invention 
This invention relates to an attenuator for attenuating a microwave signal, 
and particularly to a variable attenuator which can vary attenuation by 
means of a control signal. 
b) Description of the Prior Art 
A microwave attenuator may be implemented as a fixed attenuator or a 
variable attenuator. A typical example of the variable attenuator is a 
reflection attenuator. The reflection attenuator can consist of a 4-port 
coupler, PIN diodes, and dummy loads. 
FIG. 9 shows the construction of a reflection attenuator according to one 
prior art. This prior art uses a 4-port coupler (3) which has four ports 
(4 to 7). The port 4 and the port 5 are in radio frequency coupling, and a 
microwave signal inputted from the port 4 appears at the port 5 in an 
in-phase half amplitude. Similarly, the port 6 and the port 7 are in radio 
frequency coupling, and a microwave signal inputted from the port 6 
appears at the port 7 in an in-phase half amplitude. The port 4 is 
connected to the port 6 by a line, and a microwave signal which is 
inputted in the port 4 appears at the port 6 in 90.degree.--delayed half 
amplitude. And, the port 5 is connected to the port 7 by a line, and a 
microwave signal which is inputted in the port 5 appears at the port 7 in 
a 90.degree.--delayed phase half amplitude. These relations constitute 
even when input and output are exchanged. Specifically, the 4-port coupler 
3 is a quadrature hybrid coupler or a 3-dB range coupler, and the ports 4 
to 7 are an input port, an in-phase output port, a 90.degree.--delayed 
phase output port and a 90.degree.--delayed phase combination output port, 
respectively. 
In FIG. 9, the port 4 is connected to a signal input terminal 1, and the 
port 7 to a signal output terminal 2, respectively. A microwave signal 
which is inputted in the signal input terminal 1 appears at the port 5 in 
an in-phase half amplitude and at the port 6 in a 90.degree.--delayed 
phase half amplitude. The port 5 is connected to a termination circuit in 
which a transmission line 11a, a PIN diode 12a, and a dummy load 13a are 
connected in series in this order. Similarly, the port 6 is connected to a 
termination circuit in which a transmission line 11b, a PIN diode 12b, and 
a dummy load 13b are connected in series. 
The transmission lines 11a and 11b have an impedance converting function. 
In this case, a characteristic impedance of the 4-port coupler 3 after the 
impedance conversion made by the transmission lines 11a and 11b is assumed 
to be Z0 as the matched impedance of the transmission line. Since the PIN 
diodes 12a and 12b function as variable radio frequency resistive 
elements, when synthesized resistance Z1a of the dummy load 13a and the 
PIN diode 12a is equal to Z0, the port 5 falls under a reflectionless 
termination state. Similarly, when synthesized resistance Z1b of the dummy 
load 13b and the PIN diode 12b is equal to Z0, the port 6 falls under a 
reflectionless termination state. 
As described above, the signal which appears at the port 5 appears at the 
port 7 in a 90.degree.--delayed phase half amplitude, and the signal which 
appears at the port 6 appears at the port 7 in an in-phase half amplitude. 
Therefore, the signal appearing at the port 7 is a combined signal of a 
reflection signal by the termination circuit connected to the port 5 among 
the signals inputted from the port 4, and a reflection signal by the 
termination circuit connected to the port 6 among the signals inputted 
from the port 4. These reflection signals are in-phase with each other 
because they are opposite relative to the signal inputted in the port 4. 
Therefore, the combination at the port 7 is an in-phase combination. When 
both Z1a and Z1b are equal to Z0, both the ports 5 and 6 are subjected to 
the reflectionless termination, and no reflection signal appears at the 
port 7 (a state of infinite attenuation). On the other hand, when both Z1a 
and Z1b are substantially infinite, both the ports 5 and 6 fall under a 
total reflection state, and the maximum reflection signal appears at the 
port 7 (a state of minimum attenuation). 
Therefore, if Z1a and Z1b could be varied successively in the range of Z0 
to infinity, the attenuation of the signal outputted from the signal 
output terminal 2 with respect to the signal inputted from the signal 
input terminal 1 can be ideally controlled successively in the range of 0 
to infinity. As a means therefor, the prior art provides the PIN diodes 
12a and 12b and a bias control circuit. 
As shown in FIG. 9, the signal input terminal 1 and the signal output 
terminal 2 have bias terminals 10a and 10b connected, respectively. Since 
the coupling of the ports 4 and 5 prevents a d.c. voltage, the d.c. 
voltage which is applied to the bias terminal 10a is applied to the anode 
of the PIN diode 12b. Similarly, since the coupling of the ports 5 and 7 
prevents a d.c. voltage, the d.c. voltage which is applied to the bias 
terminal 10b is added to the anode of the PIN diode 12a. Accordingly, the 
resistance values of the PIN diodes 12a and 12b are determined by the d.c. 
voltage (bias voltage) which is applied to the terminal 10b or 10a, 
respectively. In other words, the successive variation of the bias voltage 
which is applied to the terminals 10a and 10b can successively vary Z1a 
and Z1b in the range of Z0 to infinity. Thus, the attenuation can be 
successively controlled in the range of 0 to infinity. 
Furthermore, quarter wavelength lines 8a and 8b are respectively connected 
between the signal input terminal 1 and the bias terminal 10a and between 
the signal output terminal 2 and the bias terminal 10b. To the bias 
terminals 10a and 10b, quarter wavelength open stubs 9a and 9b which are 
quarter wavelength lines with one end open are connected, respectively. 
When the quarter wavelength open stubs 9a and 9b are observed from the 
bias terminals 10a and 10b, impedance thereof becomes 0. When this 
impedance is observed from the signal input terminal 1 or the signal 
output terminal 2, it is seen to be infinite because of the quarter 
wavelength lines 8a and 8b which are present between the signal input 
terminal 1 and the bias terminal 10a and between the signal output 
terminal 2 and between the bias terminal 10b. 
Therefore, regardless of the connection of the bias terminals 10a and 10b 
to the signal input terminal 1 and the signal output terminal 2, 
respectively, for a signal which has a relatively high frequency and 
therefore a relatively short wavelength of the same length as that of an 
electrical length of each quarter wavelength line or open stub, the bias 
terminals 10a and 10b fall in a state not visibly existing. Thus, the 
quarter wavelength lines 8a and 8b function in the same way as an 
inductance in a Radio-frequency circuit, and the quarter wavelength open 
stubs 9a and 9b function in the same way as a capacitance, so that a 
circuit consisting of the quarter wavelength line 8a and the quarter 
wavelength open stub 9a and a circuit consisting of the quarter wavelength 
line 8b and the quarter wavelength open stub 9b can be understood by the 
analogy with a resonance circuit in a Radio-frequency circuit. These 
circuits are called RF chokes. 
For accurate functioning of a variable attenuator having the circuit 
configuration as shown in FIG. 9, the resistance values of the PIN diodes 
12a and 12b must be balanced accurately. Therefore, bias voltages which 
are added to the PIN diodes 12a and 12b must be also balanced accurately. 
But, it is difficult to secure such a balance in the construction as the 
bias voltages are separately applied to the PIN diodes 12a and 12b as in 
FIG. 9. Furthermore, the PIN diodes 12a and 12b generally have a stray 
capacity or a stray inductance. 
Because the demands for securing balanced bias voltages of the PIN diodes 
12a and 12b cannot be fully met, the attenuation cannot be fully and 
accurately controlled heretofore. Therefore, the variable attenuator can 
not avoid suffering from poor I/O characteristics, reflection 
characteristics, and frequency characteristics. And, attenuation 
characteristics also deteriorate when affected by a stray capacity or a 
stray inductance. 
SUMMARY OF THE INVENTION 
The first object of this invention is to provide a variable attenuator 
which can accurately and sufficiently control attenuation. 
The second object of this invention is to satisfactorily balance control 
signals (e.g. bias voltage) which are given to variable radio frequency 
resistive elements (e.g. PIN diodes) without adversely affecting the 
characteristics of a variable attenuator, and additionally to balance 
resistance values of the variable radio frequency resistive elements. 
The third object of this invention is to realize a variable attenuator 
which is not affected by stray capacitance or stray inductance of the 
variable radio frequency resistive element. 
The variable attenuator according to this invention is provided with the 
following: 
a) a coupler having first to fourth ports: wherein when a radio frequency 
signal is inputted in the first port, a radio frequency signal which has 
an in-phase from the radio frequency signal inputted in the first port 
appears at the second port, and a radio frequency signal which has a 
90.degree.--delayed phase of the radio frequency signal inputted in the 
first port appears at the third port; and at the fourth port, a radio 
frequency signal which has a 90.degree.--delayed phase of the radio 
frequency signal appeared at the second port appears and a radio frequency 
signal which has an in-phase from the radio frequency signal appeared at 
the third port appears; 
b) a single d.c. current path through which d.c. current according to the 
required attenuation flows; and 
c) first and second variable radio frequency resistive elements which are 
disposed on the d.c. current path, and making up at least parts of the 
termination impedances of the second and third ports, respectively, by 
producing resistances of a value corresponding to a value of the d.c. 
current. 
In this invention, a radio frequency signal is inputted through the first 
port, namely an input port, to the coupler. Then, a signal which has an 
in-phase from the radio frequency signal inputted in the input port 
appears at the second port, namely an in-phase output port, and a 
90.degree.--delayed phase signal appears at the third port, namely a 
90.degree.--delayed phase output port, of the coupler. The radio frequency 
signal which has appeared at the in-phase output port without phase-delay 
and appears at the fourth port with 90.degree. delay, namely a combination 
output port of the coupler, and the radio frequency signal which has 
appeared at the 90.degree.--delayed phase output port is phase-inverted 
and appears at the combination output port. Specifically, a radio 
frequency signal which is obtained by combining the radio frequency signal 
obtained by the phase-inversion of the radio frequency signal which is 
inputted from the input port by coupling the input port and the in-phase 
output port and a radio frequency signal obtained by the phase-inversion 
of the radio frequency signal which is inputted from the input port by 
coupling the 90.degree.--delayed phase output port and the combination 
output port, is outputted from the combination output port. This 
combination is called an in-phase combination because the signals to be 
combined are in-phases. 
In this invention, the in-phase output port is terminated by a termination 
impedance containing radio frequency resistance components of a first 
variable radio frequency resistive element such as a PIN diode, and the 
90.degree.--delayed phase output port is terminated by a termination 
impedance containing radio frequency resistance components of a second 
variable radio frequency resistive element such as an PIN diode. 
Therefore, the control of the radio frequency resistance components of the 
first and second variable radio frequency resistive elements can increase 
or decrease the reflection from the in-phase output port or the 
90.degree.--delayed phase output port. An increase of reflection from the 
in-phase output port and the 90.degree.--delayed phase output port 
increases output from the combination output port, and a decrease of 
reflection from the in-phase output port and the 90.degree.--delayed phase 
output port decreases output from the combination output port. The 
variable attenuator which can variably control attenuation successively 
based on the above principle is called a reflection attenuator. 
The first and second variable radio frequency resistive elements produce 
resistance (radio frequency resistance components) of a value 
corresponding to a value of the supplied d.c. current. This invention is 
principally characterized by the fact that the first and second variable 
radio frequency resistive elements are driven by an equal current. 
Specifically, the first and second variable radio frequency resistive 
elements are disposed on a certain single d.c. current path. When a PIN 
diode is used as the first variable radio frequency resistive element, 
first and second PIN diodes are connected in the forward direction along 
the d.c. current path. 
Generally, the same current flows through two circuit elements which are 
disposed on the same d.c. current path. In this invention, the same 
current flows through the first and second variable radio frequency 
resistive elements and, as a result, the radio frequency resistance 
components of the first and second variable radio frequency resistive 
elements have the same value. It does not happen that different currents 
flow through the first and second variable radio frequency resistive 
elements, resulting in an unbalanced radio frequency resistance component 
as in the case of voltage driving. 
Accordingly, this invention does not cause unbalanced radio frequency 
resistance components of the first and second variable radio frequency 
resistive elements, and besides does not cause an unbalance of the 
termination impedance of the in-phase output port and the termination 
impedance of the 90.degree.--delayed phase output port. As a result, 
attenuation can be accurately controlled to a sufficient degree. In this 
case, no adverse effect is caused to the variable attenuator 
characteristics. 
To the above input port and the combination output port, a signal input 
terminal and a signal output terminal are connected, respectively. When a 
radio frequency signal to be attenuated is inputted to the signal input 
terminal, the attenuated radio frequency signal is outputted from the 
signal output terminal. The termination circuit of the in-phase output 
port generally consists of a line (may be provided with an impedance 
converting function) for transmitting the radio frequency signal appeared 
at the in-phase output port to the first variable radio frequency 
resistive element and a dummy load which forms at least a part of the 
termination impedance, in addition to the first variable radio frequency 
resistive element. The termination circuit of the 90.degree.--delayed 
phase output port generally consists of a line (may be provided with an 
impedance converting function) for transmitting the radio frequency signal 
appeared at the 90.degree.--delayed phase output port to the second 
variable radio frequency resistive element and a dummy load which forms at 
least a part of the termination impedance, in addition to the second 
variable radio frequency resistive element. 
First, the d.c. current path of this invention has a structure in that 
polarity of the first variable radio frequency resistive element which is 
observed from the in-phase output port, and polarity of the second 
variable radio frequency resistive element which is observed from the 
90.degree.--delayed phase output port, are opposite (reversed polarity 
connection to the coupler). For example, when observed in the direction 
that d.c. current flows, a path consisting of the terminal (bias terminal) 
for supplying the d.c. current .fwdarw.the combination output 
port.fwdarw.(coupler inside).fwdarw.the in-phase output port.fwdarw.the 
first variable radio frequency resistive element.fwdarw.the second 
variable radio frequency resistive element.fwdarw.the 90.degree.--delayed 
phase output port.fwdarw.(coupler inside).fwdarw.the input port.fwdarw.the 
ground can be adopted. When this path is adopted, there are required: a 
circuit which makes a d.c. connection of the terminal and the combination 
output port while isolating a radio frequency signal (hereafter referred 
to as the first circuit); a circuit which makes d.c. isolation of the 
first and second variable radio frequency resistive elements from the 
ground while making a radio frequency connection and also makes a d.c. 
connection of the first variable radio frequency resistive element and the 
second variable radio frequency resistive element while isolating a radio 
frequency signal (hereinafter referred to as the second circuit); and a 
circuit which makes a d.c. connection of the first port and the ground 
while isolating a radio frequency signal (hereinafter referred to as the 
third circuit). 
The first circuit can be realized by inserting a quarter wavelength line 
between the bias terminal and the combination output port and connecting 
another quarter wavelength line with one end open to the bias terminal. 
The second circuit can be realized by inserting grounding capacitors 
between the first variable radio frequency resistive element and the 
ground and between the second variable radio frequency resistive element 
and the ground, respectively, and connecting the non-grounded ends of both 
the grounding capacitors by a connection line. The third circuit can be 
realized by inserting a quarter wavelength line between the input line and 
the ground and connecting another quarter wavelength line with one end 
open to the above quarter wavelength line. 
Second, the d.c. current path of this invention has a structure in that 
polarity of the first variable radio frequency resistive element which is 
observed from the in-phase output port and the polarity of the second 
variable radio frequency resistive element which is observed from the 
90.degree.--delayed phase output port are the same (same polarity 
connection to the coupler). In particular, when the first and second PIN 
diodes which contain stray components distributed asymmetrically with 
respect to an RF resistance are used as the first and second variable 
radio frequency resistive elements, the same polarity connection to the 
coupler prevents the effects of the stray capacitance and the stray 
inductance of the variable radio frequency resistive element. Thus, it is 
effective to secure the impedance balance. 
In this type of construction, when observed in the direction that d.c. 
current flows, the d.c. current path can have a flow of the bias 
terminal.fwdarw.the first circuit.fwdarw.the combination output 
port.fwdarw.(coupler inside).fwdarw.the in-phase output port.fwdarw.the 
first variable radio frequency resistive element.fwdarw.the input 
port.fwdarw.(coupler inside).fwdarw.the 90.degree.--delayed phase output 
port.fwdarw.the second variable radio frequency resistive 
element.fwdarw.the ground. To adopt this construction, in addition to the 
first circuit, there are needed a circuit which makes a radio frequency 
connection of the first variable radio frequency resistive element and the 
ground while making a d.c. isolation and makes a d.c. connection of the 
first variable radio frequency resistive element and the input port while 
isolating a radio frequency signal (hereinafter referred to as the fourth 
circuit); and a circuit which makes a d.c. connection of the second 
variable radio frequency resistive element and the ground while making a 
d.c. isolation of the second variable radio frequency resistive element 
and the bias terminal (hereinafter referred to as the fifth circuit). 
The first circuit is sufficient in the structure described above. The 
fourth circuit can be structured by disposing a grounding capacitor 
between the first variable radio frequency resistive element and the 
ground, connecting a quarter wavelength line to the first variable radio 
frequency resistive element, connecting another quarter wavelength line 
with one end open to the above quarter wavelength line, and disposing a 
further quarter wavelength line between the quarter wavelength line with 
one end open and the input port. The fifth circuit can be structured by 
connecting a quarter wavelength line to the bias terminal, and disposing a 
coupling capacitor between the second variable radio frequency resistive 
element and the quarter wavelength line. 
For the same polarity connection to the coupler, the fourth circuit may be 
structured by disposing a grounding capacitor between the first variable 
radio frequency resistive element and the ground, connecting a quarter 
wavelength line with one end open to one end of the grounding capacitor, 
and disposing another quarter wavelength line between the above quarter 
wavelength line and the input port. 
To adopt the above same polarity connection to the coupler, it is 
preferable to additionally dispose a dumping resistance in order to 
suppress resonance sharpness of the first and second variable radio 
frequency resistive elements.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
Now, preferred embodiments of the invention will be described with 
reference to the drawings. As to the structure which is the same as or not 
the same as but similar to that of the prior art of FIG. 9, the same 
reference numerals will be used and their description will be omitted. The 
same is also applied to the structure which is common to respective 
embodiments. 
a) First embodiment 
FIG. 1 shows the structure of a variable attenuator according to the first 
embodiment of the invention. In this embodiment, the anode and the cathode 
of the PIN diode 12a are connected to the line 11a and the dummy load 13a, 
respectively, while the anode and the cathode of the PIN diode 12b are 
connected to the dummy load 13b and the line 11b, respectively. In other 
words, the PIN diode 12b has its polarity reversed from the prior art. 
Grounding capacitors 14a and 14b are disposed between the dummy loads 13a, 
13b and the grounds, respectively. The grounding capacitors 14a and 14b 
insulate a d.c. signal and short in a radio frequency range between the 
dummy loads 13a, 13b and the grounds. Furthermore, a connection line 15 
connects the connecting point of the dummy load 13a and the grounding 
capacitor 14a to the connecting point of the dummy load 13b and the 
grounding capacitor 14b. Also, a position where a bias terminal 10a is 
disposed in the prior art is now grounded. 
Accordingly, this embodiment configures only one d.c. current path 
consisting of the bias terminal 10b.fwdarw.the line 8b.fwdarw.the port 
7.fwdarw.the port 5.fwdarw.the line 11a.fwdarw.the PIN diode 
12a.fwdarw.the dummy load 13a.fwdarw.the connection line 15.fwdarw.the 
dummy load 13b.fwdarw.the PIN diode 12b.fwdarw.the line 11b.fwdarw.the 
port 6.fwdarw.the port 4.fwdarw.the line 8a.fwdarw.the ground. Since the 
PIN diodes 12a and 12b are present in the same polarity along this current 
path, the same current flows through the PIN diodes 12a and 12b, which 
have the same resistance value as a result. Thus, in the embodiment, the 
PIN diodes 12a and 12b have their resistance values preferably balanced, 
making it possible to accurately control attenuation. A termination 
circuit which is connected to the ports 5 and 6 is the same equivalent 
circuit as in the prior art in terms of a radio frequency, so that it is 
noted that the termination function does not change. 
FIG. 2 shows forward direction current-voltage characteristics of the PIN 
diodes 12a and 12b. It is assumed here that the two PIN diodes have 
different characteristics i.e. the PIN diode 12a has characteristics 
indicated by a broken line and the PIN diode 12b has characteristics 
indicated by a solid line. Then, the control by a bias voltage VF1 in the 
same manner as in the prior art results in that current IF1 flowing 
through the PIN diode 12a and current IF2 flowing through the PIN diode 
12b have different values. Since a PIN diode has the current resistance 
characteristics as shown in FIG. 3, the PIN diodes 12a and 12b have 
different resistance values R1 and R2 to each other. This results in 
inducing the unbalance between the reflections at the port 5 and the port 
6. 
On the other hand, in this embodiment, the same current flows through the 
PIN diodes 12a and 12b without exception, resulting in the same resistance 
value. Therefore, this embodiment provides the appropriately balanced 
resistance values of the PIN diodes 12a and 12b, making it possible to 
accurately control attenuation, and does not raise any problem in 
reflection properties and others. 
b) Second embodiment 
FIG. 4 shows the structure of a variable attenuator according to the second 
embodiment of the invention. In this embodiment, the PIN diode 12b has the 
same polarity as does the prior art. The cathodes of the PIN diodes 12a 
and 12b are connected each to one end of quarter wavelength lines 16a and 
16b, respectively. The other ends of the quarter wavelength lines 16a and 
16b are connected to a connection point of the quarter wavelength line 8a 
and the quarter wavelength open stub 9a and to a connection point of the 
quarter wavelength line 8b and the quarter wavelength open stub 9b. Among 
the grounding capacitors disposed in the first embodiment, 14b is not 
disposed in this embodiment. The cathode of the PIN diode 12b is connected 
to one end of the quarter wavelength line 16b through a coupling capacitor 
17. 
Therefore, this embodiment forms one d.c. current path consisting of the 
bias terminal 10b.fwdarw.the line 8b.fwdarw.the port 7.fwdarw.the port 
5.fwdarw.the line 11a.fwdarw.the PIN diode 12a.fwdarw.the line 
16a.fwdarw.the line 8a.fwdarw.the port 4.fwdarw.the port 6.fwdarw.the line 
11b.fwdarw.the PIN diode 12b.fwdarw.the dummy load 13b.fwdarw.the ground. 
Since the PIN diodes 12a and 12b are present in the same polarity along 
this current path, the same current flows through the PIN diodes 12a and 
12b, and they have the same resistance value. Thus, this embodiment 
provides the appropriately balanced resistance values of the PIN diodes 
12a and 12b, making it possible to accurately control attenuation, and 
does not raise any problem in reflection properties and others. 
As in the case of the first embodiment, the termination circuits connected 
to the ports 5 and 6 are the same equivalent circuits as in the prior art 
in terms of a radio frequency, so that it is noted that the termination 
function does not change. And, when observed from the cathode of the PIN 
diode 12a or 12b, the impedances of the quarter wavelength lines 16a and 
16b becomes a high impedance at a radio frequency because the quarter 
wavelength open stub 9a or 9b is disposed. Therefore, there is no leakage 
of an RF signal from the cathode of the PIN diode 12a or 12b to the 
quarter wavelength line 16a or 16b. An electrostatic capacity of the 
coupling capacitor 17 is set up to establish the above impedance 
relationship at a frequency of the signal subject to attenuation. 
This embodiment, as compared with the first embodiment, has an advantage of 
reducing the effect of the unbalanced stray inductances of the PIN diodes 
12a and 12b. 
As shown in FIG. 5, the PIN diodes 12a and 12b may have the structure in 
that a diode chip 18 having a surface electrode (not shown) is 
accommodated into a package 19, and the diode chip 18 is connected to an 
external circuit through terminals 20 and 21 which are disposed outside of 
the package 19. Realization of this structure needs to connect the surface 
electrode of the diode chip 18 to the terminal 21, and to dispose a wire 
22 to connect the diode chip 18 to the terminal 20. This wire 22 is 
expressed as a stray inductance LS which is connected in series to a radio 
frequency resistor R of the diode chip 18 on the equivalent circuit shown 
in FIG. 6. In the FIG., Cs represents stray capacity between the terminals 
20 and 21. 
When the PIN diode having the above structure is used for the PIN diodes 
12a and 12b in the first embodiment, for one of the PIN diodes 12a and 
12b, the stray inductance LS appears on the side of the 4-port coupler 3 
as observed from the radio frequency resistor R, and for the other, it 
appears on the side of the dummy load. This unbalance makes the reflection 
condition at the port 5 and that at the port 6 unbalance. Consequently, 
when the stray inductance LS is designed to appear on the same side as 
observed from the radio frequency resistor R for both of the PIN diodes 
12a and 12b as in the second embodiment, the reflection characteristics 
can be further improved. 
c) Third embodiment 
FIG. 7 shows the structure of a variable attenuator according to the third 
embodiment of the invention. In this embodiment, the PIN diodes 12a and 
12b are connected in the same polarity as in the second embodiment. But, 
the quarter wavelength lines 16a and 16b and the coupling capacitor 17 are 
not provided, and the connection point of the quarter wavelength line 8a 
and the quarter wavelength open stub 9a is connected to a connection point 
of the dummy load 13a and the grounding capacitor 14a. 
This embodiment forms one closed d.c. circuit consisting of the bias 
terminal 10b.fwdarw.the line 8b.fwdarw.the port 7.fwdarw.the port 
5.fwdarw.the line 11a.fwdarw.the PIN diode 12a.fwdarw.the dummy load 
13a.fwdarw.the line 8a.fwdarw.the port 4.fwdarw.the port 6.fwdarw.the line 
11b.fwdarw.the PIN diode 12b.fwdarw.the dummy load 13b.fwdarw.the ground. 
Since the PIN diodes 12a and 12b are present in the same polarity on this 
closed circuit, the same current flows through the PIN diodes 12a and 12b, 
and therefore, they have the same resistance value. Thus, this embodiment 
provides the appropriately balanced resistance values of the PIN diodes 
12a and 12b, makes it possible to accurately control attenuation, and does 
not raise any problem in reflection properties and others. And, since the 
PIN diodes 12a and 12b have the same polarity as in the second embodiment, 
the effects obtained are the same as in the second embodiment. 
As compared with the second embodiment, this embodiment has an advantage 
that the number of component parts is not high. First, this embodiment 
utilizes a fact that a radio frequency signal is grounded by the grounding 
capacitor 14a in the termination circuit of the port 5. Specifically, 
since the impedance of the grounding capacitor 14a which is observed from 
the connection point of the dummy load 13a and the grounding capacitor 14a 
is zero in terms of a radio frequency, the connection of the connection 
point of the dummy load 13a and the grounding capacitor 14a to the signal 
input terminal 1 makes it unnecessary to provide a means, such as the 
quarter wavelength line 16a, to increase the apparent impedance of the 
signal input terminal 1. Second, this embodiment utilizes the fact that a 
radio frequency signal is grounded by the termination circuit of the port 
6. Specifically, since the impedance of the ground which is observed from 
the dummy load 13b is zero, it is not necessary to provide a means, such 
as the quarter wavelength line 16b, to increase the apparent impedance of 
the signal input terminal 2. Furthermore, the coupling capacitor 17 has no 
need to be provided either. 
d) Fourth embodiment 
FIG. 8 shows the structure of a variable attenuator according to the fourth 
embodiment of the invention. This embodiment is structured by further 
adding a dumping resistance 23a to the termination circuit of the port 5 
of the third embodiment, and a dumping resistance 23b to the termination 
circuit of the port 6. The dumping resistances 23a and 23b are disposed on 
the lines 11a and 11b as observed from the PIN diodes 12a and 12b. 
The dumping resistances 23a and 23b have a function to decrease resonance 
sharpness Q of the PIN diodes 12a and 12b. When it is assumed that the 
equivalent circuits of the PIN diodes 12a and 12b are as seen in FIG. 6, 
the resonance sharpness Q of the PIN diodes 12a and 12b are expressed as 
follows: 
EQU Q=.omega.Ls/R-1/(.omega.CsR). 
Therefore, when RF resistances R of the PIN diodes 12a and 12b are 
increased in appearance by the dumping resistances 23a and 24b as in this 
embodiment, Q is decreased. When Q is decreased, the variable attenuator 
is hardly affected by resonance. In other words, the characteristics near 
the resonance frequencies: 
EQU f=1/{2.pi.(Ls-Cs)} 
of the PIN diodes 12a and 12b are improved. 
To the dumping resistances 23a and 23b, resistance segments of the dummy 
loads 13a and 13b may be utilized. Furthermore, the termination impedances 
in the termination circuits of the ports 5 and 6 are a total of the 
impedances of the dummy loads 13a and 13b, the RF resistances R of the PIN 
diodes 12a and 12b, and the dumping resistances 23a and 23b. And, it is to 
be understood that this embodiment forms the same d.c. circuit as in the 
third embodiment, and provides the same effects as in the third 
embodiment. In addition, it is easy to incorporate the characteristics of 
this embodiment into the second embodiment.