Linear motor control with triac and phase locked loop

A Triac and a phase-locked loop are used to control the amplitude of reciprocation of an AC linear motor. The phase locked loop generates a square wave that is synchronized with an AC power source but is substantially isolated from electrical noise on the power source. The phase of the square wave is varied with an amplitude control signal, and Triac firing pulses are generated at the transitions of the square wave.

TECHNICAL FIELD

The invention is in the general field of electric motor control. Specifically, the invention uses a Triac and a phase locked loop to control the amplitude of reciprocation of a linear motion AC motor, in such a way that pulses of power line noise have little effect on amplitude or phase of reciprocation.

BACKGROUND ART

Triac controllers (Ref. 2 ) offer an inexpensive means of controlling reciprocation amplitude of linear motion AC motors (Ref. 1 ), but in prior art have the disadvantage of responding to pulses of electrical noise on the AC power line if peak pulse voltage is high enough to cause the power line voltage to cross zero volts. A spurious zero crossing is treated by the prior art control as a normal zero crossing, and, since prior art Triac control generates Triac firing pulses that are timed relative to zero crossings of power line voltage, the result of a pulse of line noise can be generation of a spurious Triac firing pulse. If the linear motor drives a free piston machine such as a Stirling refrigerator, incorrectly timed Triac firing can cause damage to the machine and/or loss of amplitude control. Electrical filtering of line voltage can attenuate line noise, but suppression by filtering of pulses of line noise occurring very near normal zero crossings, to the point where a spurious zero crossing cannot occur, is impractical.

BRIEF DISCLOSURE OF INVENTION

The invention uses a Triac and a phase locked loop to control the amplitude of reciprocation of a linear motor with substantial isolation from disturbance by line noise pulses. Amplitude control in the invention is achieved, as in prior art, by connecting the series combination of the motor and a Triac across an AC power source, and varying the phase of Triac firing pulses relative to the power source voltage. The invention departs from prior art in that firing pulses are generated at transitions of the square wave output of a Voltage Controlled Oscillator (VCO) that is part of a phase locked loop (PLL). The VCO is synchronized in frequency with the AC power source by the PLL, but is practically isolated from power line noise pulses. In the invention, the phase of the VCO output and therefore of the Triac firing pulses, relative to line voltage, is varied with a control voltage that is proportional to the difference between a first voltage proportional to an amplitude set-point voltage and a second voltage that is proportional to a measured value of reciprocation amplitude. A decrease in measured amplitude causes a leading phase shift of firing pulses, which constitutes negative feedback that acts to hold the amplitude of reciprocation of the motor at a preset value determined by the amplitude set point voltage.

In one form of the invention, a voltage proportional to the amplitude of reciprocation of the linear motor is derived by analog or digital computation based on the equivalent circuit of the linear motor, thus obviating the cost and complexity of a position sensor.

The basic components of the phase locked loop used in the invention can be elements of a single integrated circuit (e.g., type 4046 , see Ref 4 ).

DETAILED DESCRIPTION

Referring to FIG. 1 , a linear motor (MOTOR) with terminals M 1 and M 2 and a Triac with main terminals MT 1 and MT 2 and a gate terminal G are connected in series across an AC power line whose terminals are L 1 and L 2 . Firing pulses applied to G cause the Triac to become practically a short circuit so that line voltage (V) is applied across the motor. After firing, the Triac remains a short circuit until current (I) falls to zero, whereupon the Triac becomes practically an open circuit and awaits the next firing pulse. Control of the reciprocation amplitude of the motor is achieved in the invention by controlling the time of occurrence of firing pulses relative to zero crossings of V. If firing pulses precede zero crossings of V by a time that is short relative to the period of V, the Triac remains a short circuit for only a small part of the AC cycle, and motor amplitude is small. As the time between a firing pulse and the next zero crossing of V increases, i.e., as firing pulses increasingly lead V in phase, the Triac remains a short circuit for a greater fraction of the AC cycle and motor amplitude increases.

In the invention, firing pulses are generated at the transitions of a square wave that is synchronized in frequency with line voltage V, and whose phase relative to V can be varied with a control voltage VCTL. Isolation from the effects of pulses of electrical noise on the power line is achieved in the invention by using a phase locked loop to generate the square wave from which firing pulses are derived, as will now be described.

Referring to FIG. 1 , a resistive voltage divider (R 1 , R 2 ) is connected across L 1 and L 2 . The attenuated replica of power line voltage that appears at the node joining R 1 and R 2 is a first input to a PHASE COMPARATOR whose second input is the output of a VOLTAGE CONTROLLED OSCILLATOR (VCO). The VCO output (designated VCOout) is a square wave whose frequency is determined by the VCO input (designated VCOin). The output of the PHASE COMPARATOR (PCout) is low pass filtered by FILTER and applied to a difference amplifier (designated DIFF) which subtracts a control voltage VCTL from the filtered PCout and applies the difference to VCOin according to the equation

thus forming a closed phase locked loop (PLL). As is known in prior art, such a PLL will synchronize the two inputs to the PHASE COMPARATOR, that is, the PLL causes VCOout to have the same frequency as the AC power line. Since there is a unique relationship between VCOin and the frequency of VCOout, it follows that VCOin remains constant regardless of VCTL, i.e.,

where K 1 constant value of VCOin.

In the invention, the PHASE COMPARATOR is of the Exclusive-OR (X-OR) type, for which

where K 2 is a constant. Combining equations 1 and 2 gives,

phase lag of VCO out relative to AC line (1/ K 2 ) ( K 1 VCTL ) equation 3

Equation 3 shows that VCTL can be used to control the phase of VCOout relative to the AC line, as illustrated by FIG. 5 , which shows the DC component of PCout for an X-OR phase comparator. In order for the PLL to be stable, it must operate on the phase lagging portion of the phase comparator characteristic, i.e., on the left in FIG. 4 . According to equation 3, the location of the point labeled VCTL 0 in FIG. 5 is determined by the constants K 1 and K 2 . By means which will be described later, K 1 and K 2 in the invention are made such that for VCTL 0, the phase lag of VCO out relative to V is (180 ) electrical degrees, where is a small angle typically less than 15 degrees. Therefore, if VCTL 0, firing pulses generated at the transitions of VCOout by the FIRING PULSE GENERATOR will precede zero crossings of V by a small fraction of the period of V and the amplitude of reciprocation of the motor will be consequently be low. Referring again to FIG. 5 and equation 3, as VCTL decreases, the phase of VCOout increasingly leads V. As a result, firing pulses precede zero crossings of V by a greater fraction of the period of V, and the amplitude of reciprocation consequently increases. The effect of VCTL is further illustrated by FIG. 4 , which shows V, VCOout, firing pulses, PCout, filtered PCout, and I for two values of VCTL. The top part of FIG. 4 applies to VCTL close to zero. Decreasing VCTL causes all of the waveforms in the lower part of FIG. 4 to lead their counterparts in the upper part of FIG. 4 . In particular, decreasing VCTL causes current (I) to increase because of increasing time during which the Triac is fired. The rounded shape of the pulses of current is a result of motor inductance, which maintains motor current at and after a zero crossing of V.

Returning to FIG. 1 , VCTL is the output of an ERROR AMPLIFIER, which amplifies the difference between a first voltage AMEAS, which is proportional to measured amplitude of motor reciprocation, and a second voltage ASET, which is proportional to required amplitude. Decreasing AMEAS decreases VCTL, which constitutes negative feedback, which maintains AMEAS nearly equal to ASET.

FIG. 2 shows a preferred embodiment of the invention, In which an X-OR phase comparator and VCO are elements of a single CMOS integrated circuit of the generic type number 4046 (Ref. 4 ). In this embodiment, K 1 is determined by resistors R 3 , R 4 , capacitor C 1 , and the positive DC supply voltage ( V). K 2 is determined by V only. R 6 and C 2 comprise the FILTER element of FIG. 1 . The time constant R 6 C 2 is typically about 0.05 seconds, which is long compared to the period of a 60 Hz. power line, so that the DC component of PCout appears at the terminal of difference amplifier U 1 (DIFF) while the alternating components of PCout are highly attenuated. Resistors R 7 and R 5 determine the gain of DIFF. Capacitors C 3 , C 4 , diodes D 1 , D 2 , D 3 , D 4 , resistor R 9 , transistor Q 1 , and a NAND gate comprise the FIRING PULSE GENERATOR. C 4 , D 1 , and D 4 form a positive pulse at the base of Q 1 when VCOout transitions from ground to V. The NAND gate, C 3 , D 2 , and D 3 form a positive pulse at the base of Q 1 when VCOout transitions from V to ground. Q 1 is an emitter follower, and R 8 controls the peak firing pulse current.

It can be shown that, with typical values for K 1 , K 2 , and FILTER time constant R 6 C 2 , a relatively severe line voltage pulse of any amplitude sufficient to produce a false zero crossing of 1 millisecond duration will typically produce only a 2 degree phase disturbance of the phase of VCOout, and that the disturbance will decay exponentially with a time constant of about 25 milliseconds. Such a disturbance has no practical consequence, so that in practice VCOout is isolated from pulses of power line noise.

A measurement of reciprocation amplitude of the linear motor, i.e., AMEAS, is necessary for amplitude feedback and can be provided by a position transducer. However, the linear motor itself is an accurate velocity transducer (Ref 3 ), and is used in one form of the invention to avoid the cost and complexity of a separate transducer. FIG. 3 shows an analog circuit for deriving AMEAS by analog computation based on the equivalent circuit of the motor, which is;

In equation 4;

a constant with units of volts/(meter/second)

v velocity of moving magnets in the linear motor. (meters/second)

L motor inductance (henrys)

VM motor voltage (volts)

I motor current (amps.)

R motor resistance (ohms)

In FIG. 3 , a voltage proportional to current (I) is generated by passing (I) through a low resistance RS, which is typically 0.2 ohms. A voltage proportional to VM appears at the output of difference amplifier U 2 . The inverting terminal of U 3 is a virtual ground and sums the following currents;

RS C 3 ( dI/dt ), the current through C 3

RS/R 13 I, the current through R 13

The sum of the three currents is, with the definition K 3 (R 11 /R 10 ) (1/R 12 )

By choosing,

it follows from equation 4 that the sum of the three currents is K 3 v , i.e., the sum of the three currents is proportional to motor velocity (v). The INTEGRATOR in FIG. 3 integrates the summed currents, and, since the integral of (v) is motor displacement, it follows that;

displacement of moving magnets of linear motor K 4 output of U 3 , equation 5

In equation 5, K 4 is a constant equal to (K 3 )/C 6 . R 14 is a high resistance that provides a DC feedback path for U 3 and has no practical effect on the validity of equation 5. AMEAS is derived from the substantially sinusoidal output of U 3 by the FULL WAVE RECTIFIER, which includes an inverter (U 4 , R 15 ), rectifier diodes D 1 , D 2 , and low pass filter R 16 , C 7 . The time constant R 16 C 7 is long compared to the period of V, so that AMEAS is substantially proportional to the peak amplitude of reciprocation.

Although FIG. 3 shows an analog computation of AMEAS, digital computation of AMEAS is also possible and is considered within the scope of the invention.