Series-resonant tuning of a downhole loop antenna

A loop antenna useful, for example, in investigation of earth formations. Embodiments of the invention provide a loop antenna circuit comprising a loop antenna disposed to generate, in response to an electromagnetic wave, a pick-up signal on an output node. Loop antenna circuit further includes a tuning network coupled to the loop antenna. The tuning network is disposed to provide simultaneous tuning at a plurality of interrogation frequencies. Further embodiments include a preamplifier circuit coupled to the loop antenna. The preamplifier circuit is disposed to receive the pick-up signal on an input node and provide a high impedance load to the loop antenna for a first of the plurality of interrogation frequencies to reduce the secondary radiation from the loop antenna to below a predetermined value at the said first interrogation frequency.

FIELD OF THE INVENTION

This invention relates to the field of well logging; in particular, it relates to electromagnetic wave propagation systems to measure an attribute which relates to at least one of the borehole and surrounding formation; and more particularly, it relates to antenna circuits that are deployable within a borehole to measure one or more attributes of an electromagnetic wave as it passes.

BACKGROUND OF THE INVENTION

In the practice of logging-while-drilling (LWD), measurement-while-drilling (MWD) and wireline logging, it is well known that by studying the propagation characteristics of an electromagnetic wave, useful clues regarding the characteristics of the earth formations can be derived. To probe sections of the earth surrounding the borehole, a pair of transmitters can be positioned within a well borehole to radiate an electromagnetic field at a particular interrogation frequency. This electromagnetic wave is influenced by the electromagnetic energy shed back from the formation. A pair of spaced differential loop antenna receivers is conventionally positioned within the borehole to measure, for example, the attenuation and/or phase shift of the electromagnetic wave as it passes between each receiver loop antenna. Various methods for analyzing the measurements to derive estimates of certain characteristics of the earth formations surrounding the well borehole are well known.

It is of increasing importance in oil and gas exploration to obtain accurate and reliable measurements of an electromagnetic wave investigating a formation. However, the accuracy of the information derived from the measurements can be degraded by the effects of magnetic field mutual cross-coupling between receiving loop antennae. Receiver cross-coupling typically results from significant circulating alternating current that is induced in a receiver loop antenna in response to an electromagnetic wave. This alternating current tends to produce a secondary electromagnetic field that can have a corrupting influence on the primary electromagnetic wave generated by a transmitter. The secondary electromagnetic field will affect the measurements obtained by any other receiving loop antenna in close proximity to the first receiver, producing an error component due to the cross-coupling. The receiving antenna closest to an active transmitter tends to receive a stronger signal and produce greater circulating currents than subsequently spaced receiving antennae. Accordingly, the magnitude of the undesirable secondary electromagnetic field radiated by an antenna tends to be greater from the receiving antenna that is closer to an active transmitter, and the magnitude of the error component due to cross-coupling tends to be higher in the next subsequently spaced receiving antenna.

Considerable effort has been expended by the industry to compensate measurements for the cross-coupling error. For example, one known method includes a calibration procedure where, under laboratory conditions, the cross-coupling error for each frequency of interest is measured and stored. Thereafter, each subsequently measured value is adjusted accordingly. Such methods tend to be cumbersome, may introduce new sources of error and may create maintenance restrictions. For example, extra components may be needed on the receiver circuits to simulate the cross-coupling effect in the lab. Parameters such as the distance between receivers, which will vary with temperature, are critical to the accuracy of the estimate of the cross-coupling error. In addition, the calibration may be invalidated by the replacement, in a receiver antenna system, of a failed component that influences the cross-coupling.

Eliminating the source of the cross-coupling error in downhole tools has proven to be problematic in the industry. For example, methods commonly employed to counter the excessive signal loss resulting from a lengthy coaxial cable, typically having a characteristic impedance less than 100 ohms, involve matching the receiver loop antenna impedance to the impedance of its load. However, the matching of receiver loop antenna impedance to that of its load typically results in significant circulating currents being induced in receiver loop that create receiver cross-coupling. Other methods employed by prior art systems include spacing receiver antennae as far apart as possible to reduce the effects of cross-coupling, such as locating each of a pair of receiver antennae on opposing sides of a pair of transmitters.

A strong pick up signal is an important consideration in obtaining accurate measurements. Prior art downhole tools that match the receiver antenna impedance to a load comprising a lengthy coaxial cable tend to employ single turn antennae, even though a multiple turn loop antenna typically provides the advantages of a strong pickup as compared to having a single turn antenna. A multiple turn loop antenna, in the range of 6 to 12 inches diameter, commonly exhibits several hundred ohms of impedance at 2 Mhz. Thus, prior art methods for matching the receiver antenna to the load impedance combined with the use of a step-up transformer, tend to limit a receiver loop antenna to no more than a single turn.

A further limitation of prior art receiver loop antenna systems is their inability to be simultaneously series tuned at multiple interrogation frequencies. It well known that it is advantageous to utilize multiple interrogation frequencies to probe earth formations with electromagnetic waves. Certain attributes of the earth formation are discoverable only when the interrogation frequency is of a specific range. Lower frequencies are able to investigate deeper regions of the earth for a given transmitter and receiver spacing. Also, lower frequencies often mitigate borehole effects. Higher frequencies yield higher phase shift and attenuation values for a given resistivity, which is advantageous for increased accuracy in highly resistive formations of commercial interest. In LWD and MWD systems where measurements are commonly obtained while the measuring tool is rotating and moving axially through the borehole, greater and more accurate information about the surrounding earth formation can be derived by obtaining simultaneous measurements of a plurality of interrogation frequencies.

There is therefore a need in the art for receiver systems, deployable within a borehole, that can utilize a loop antenna having multiple turns, as opposed to a single turn, to enable the antenna to pick-up a strong signal from which a more accurate measurement of particular attributes of an electromagnetic wave can be derived. Also, there is a need for a loop antenna that can be simultaneously series tuned at plurality of interrogation frequencies to enable it to simultaneously and accurately pick-up the plurality of interrogation frequency components from an electromagnetic wave. In addition, there is a need for a method for decreasing the design, manufacture, and maintenance cost of systems, that deploy a pair of loop antenna receivers downhole, while still diminishing the undesirable effects of mutual cross-coupling. Furthermore, there is an ever present need for downhole antenna systems that are stable over a wide range of temperatures, and that provide increasingly accurate and greater amounts of information about the earth formations surrounding a borehole.

SUMMARY OF THE INVENTION

In accordance with one aspect of the invention, a loop antenna circuit is provided for use in a borehole. This loop antenna circuit includes a loop antenna and a tuning network. The loop antenna is disposed to generate, in response to an electromagnetic wave, a pick-up signal on an output node. The tuning network is coupled to the loop antenna and is disposed to provide simultaneous tuning at a plurality of interrogation frequencies. Further embodiments may include a preamplifier circuit, which is coupled to the loop antenna. The preamplifier circuit is disposed to receive the pick-up signal on an input node. For at least one of the plurality of interrogation frequencies, the preamplifier circuit is disposed to provide a load impedance to the loop antenna that is sufficiently high to reduce the secondary radiation from the loop antenna to be below a predetermined level.

Other embodiments provide for the load presented to the loop antenna to substantially exceed the source impedance of the loop antenna at each of the plurality of interrogation frequencies. In addition, exemplary loop antennae may include at least six turns.

In accordance with another aspect of the invention, a tool is provided for measuring attributes of sections of an earth formation surrounding a borehole. This tool includes a housing that is adapted to be used within a borehole, a transmitter, a receiver loop antenna, and a tuning network. The transmitter is deployed on the housing to selectively generate an electromagnetic wave having plurality of interrogation frequency components. The receiver loop antenna is deployed on the housing, spaced apart from the transmitter. The receiver loop antenna is disposed to generate, in response to the electromagnetic wave, a first pick-up signal on a first output node that represents the relative magnitude and phase of each of the plurality of frequency components. The tuning network is coupled to the receiver loop antenna, wherein the combined source impedance of the loop antenna and the tuning network, provide the pick-up signal that is maximum for a narrow band of frequencies surrounding each of the plurality of interrogation frequency components and attenuated for frequencies that are out-of-band.

According to another aspect of the invention, a method is provided for processing an electromagnetic wave that includes: (1) receiving, by a receiver, an electromagnetic wave having plurality of interrogation frequency components; (2) selecting, simultaneously, the plurality of interrogation frequency components of the electromagnetic wave to the exclusion of the other frequencies; and (3) generating a pick-up signal that represents the relative magnitude and phase of the plurality of interrogation frequency components.

According to another aspect of the invention a method is provided for evaluating formations surrounding a borehole that includes: (1) selectively, radiating, via transmitter, an electromagnetic wave having first and second interrogation frequency components; (2) receiving the electromagnetic wave by a first loop antenna that is spaced apart from the transmitter; (3) selecting simultaneously, the first and the second of interrogation frequency components of the electromagnetic wave received by the first loop antenna to the exclusion of the other frequencies; (4) receiving the electromagnetic wave via second loop antenna that is spaced apart from the transmitter and the first loop antenna; and (5) selecting, simultaneously, the first and the second interrogation frequency components of the electromagnetic wave received by the second loop antenna to the exclusion of the other frequencies.

It is therefore a technical advantage of the invention is to provide antenna receiver systems to measure the amplitude and phase of a plurality of interrogation frequencies included in an electromagnetic wave with improved accuracy as compared with prior art systems. Additionally, the present invention provides cost effective and space efficient loop antenna receivers that work reliably in the adverse conditions commonly found while drilling in a subterranean borehole.

DETAILED DESCRIPTION

FIG. 1shows, in block diagram form, a portion of one exemplary embodiment of a measuring tool100on which the present invention may be deployed. Measuring tool100is deployable within a subterranean borehole to investigate the propagation characteristics of an electromagnetic wave passing through the surrounding earth formation. Measuring tool100may be advantageously employed to determine an attribute of either a section of the borehole or a section of the surrounding earth formation, such as, for example, its resistivity or its dielectric constant.

Measuring tool100comprises a logging collar110, which, inFIG. 1is illustrated as essentially an elongated steel shaft. Logging collar110is adapted to be positioned within a borehole140. In one embodiment, logging collar110is adapted to be in the drill string close to the drill bit to provide measurement-while-drilling or logging-while-drilling. Measuring tool100is advantageously adapted to provide accurate measurements under a wide range of ambient temperatures and adverse conditions commonly found while drilling within a subterranean borehole.

In a first exemplary embodiment, logging collar110includes two transmitters TX1122, TX2128and a pair of differential receiver antennae RX1124and RX2126. TX1122, TX2128, RX1124and RX2126each comprise a coil that is wound with one or more turns on a insulating surface within in a recess circumferential to logging collar110. In one suitable embodiment TX1122and TX2128are spaced about 4 to 8 ft apart, axially on logging collar110. TX1122and TX2128are individually controllable to selectively radiate an electromagnetic wave comprising a plurality of predetermined interrogation frequency components. In the exemplary embodiment, two interrogation frequencies 500 Khz and 2 Mhz are employed. In another exemplary embodiment, a third interrogation frequency of 1 Mhz is also employed. In the exemplary embodiment, RX1124and RX2126are spaced about 10 inches apart axially on logging collar110and centered between TX1122and TX2128. Receiver antenna RX1124and RX2126are each adapted to detect bands of frequencies centered on each of the plurality of interrogation frequencies. A pick-up signal is generated by each receiver antenna RX1124and RX2126representing the phase shift and/or attenuation of the interrogation frequency components as the electromagnetic wave passes between the differential pair of receiver loop antennae RX1124and RX2126.

One skilled in the art will recognize that the embodiments of the present invention are not limited to logging-while-drilling or measurement-while-drilling applications, and may be extended to other types of applications, such as, for example, wire line systems. Embodiments will further be appreciated to be adaptable for a wide range of logging collar geometries and axial spacing for receivers and transmitters, as well as a wide range of interrogation frequencies. In addition, embodiments of the present invention may include tools having a single transmitter.

FIG. 2Adepicts a schematic block diagram of an exemplary embodiment of an antenna receiver system200, which is suitable for the measuring tool100ofFIG. 1having receiver antennae RX1124and RX2126. Antenna receiver circuit200includes antenna202, which, in the exemplary embodiment, is a loop antenna corresponding to either of RX1124or RX2126, having about a 6 inch diameter, 6 turns, and an inductance of approximately 18 μh. Antenna202is coupled in series with the primary winding of a step-up transformer220and tuning network204, so as to generate a pick-up signal across the secondary winding of transformer220(in the form of a voltage differential across nodes235and238) in response to an electromagnetic wave passing antenna202. The secondary winding of transformer220is coupled to a preamplifier circuit206that generates an amplified version of the pick-up signal on output nodes281and282. In an exemplary embodiment, output nodes281and282couple through a 4 to 6 foot coaxial cable (having a characteristic impedance of 50 to 100 ohms) to an external amplifier circuit, which is not shown. A microprocessor-based data acquisition system (not shown) samples the preamplifier output signal to determine the relative amplitude and relative phase of the electromagnetic wave at each of the plurality of interrogation frequencies.

Still referring toFIG. 2A, step-up transformer220serves to amplify the signal received from antenna202on the primary winding to provide a pick-up signal on the secondary winding. Conductors221and224each couple to an opposite end of the primary winding of step-up transformer220. Conductor221couples to a first end of antenna202and conductor221couples to the other end of antenna202. The primary winding of transformer220is split at nodes222and223to define a first portion and a second portion of the primary winding. Tuning network204couples to nodes222and223to be in series with antenna202and the primary winding of step-up transformer220. In an exemplary embodiment, step-up transformer220is of common bobbin construction and is comprised of a standard core from TDK™ part number PC44ER11/5-Z; each of the two primary winding comprises of 5 turns of #32 wire and the secondary winding comprises 32 turns of #34 wire. One skilled in art will recognize that although it is advantageous to split the primary side of the transformer into a first and second portion to balance the effects of stray capacitance and inductance inherent in the components and conductors, other suitable embodiments are available, such as a transformer that is not split, or that has a center tap on the primary winding.

Tuning network204cooperates with antenna202to achieve “simultaneous tuning” at a plurality of interrogation frequencies. “Simultaneous tuning” results in a pick-up signal that is strong for the bands approximately centered at each of the plurality of interrogation frequencies and attenuated for the other frequencies (i.e. out-of-band frequencies).

The exemplary embodiment of antenna receiver circuit200is advantageously configured with antenna202and tuning network204coupled in series to achieve “simultaneous series tuning” at a plurality of interrogation frequencies. This is implemented by providing for the impedance of the antenna202and tuning network204combination to be negligible (advantageously close to zero) for a narrow band around each of the interrogation frequencies and, at the same time, for the impedance to rise for out-of-band frequencies so that reception by the antenna202effectively excludes (or substantially attenuates) out-of-band frequencies. The exemplary embodiment of antenna receiver circuit200provides for negligible impedance of about 10 ohms, which is sufficiently low for most applications, although the invention is not limited in this regard. One skilled in the art will understand that the bands are substantially centered on each of the interrogation frequencies and are preferably as narrow as possible to advantageously provide a high signal-to-noise ratio when in-band. However, the invention is not limited to any particular in-band bandwidth, and the width of the narrow bands may be selected to specifically attenuate particular frequencies anticipated in particular embodiments to cause interference with the electromagnetic wave measurements at the interrogation frequencies.

FIGS. 2B,2C and2D illustrate simultaneous series tuning for two exemplary embodiments.FIG. 2Bshows ZAselected to represent the impedance of the antenna202and tuning network204combination shown onFIG. 2A.FIGS. 2C and 2Dare plots of impedance ZAversus interrogation frequency approximated by an analog circuit simulator. The vertical scale represents impedance linearly. The horizontal scale is logarithmic and represents the frequency range of 100 Khz through 10 Mhz.FIG. 2Cshows the impedance ZAversus frequency response for an exemplary embodiment of tuning network204that provides simultaneous series tuning at the two interrogation frequencies of 500 KHz and 2 Mhz.FIG. 2Ddepicts the impedance ZAversus frequency response for an alternate embodiment of a tuning network204that provides simultaneous series tuning at the three interrogation frequencies of 500 KHz, 1 Mhz and 2 Mhz. As shown inFIGS. 2C and 2D, the impedance ZAis negligible for a narrow band centered on each of the interrogation frequency and rises for the out-of-band frequencies.

It will be understood that the level of source impedance of the antenna202and tuning network204combination is of interest. Consistent with the invention, a source impedance that will provide sufficient attenuation so as to effectively exclude out-of-band frequencies is dependent on the load impedance. For example, with further reference toFIG. 2A, if a preamplifier206exhibits an input impedance of 3600 ohms at a particular out-of-band frequency, then a step-up transformer220with a turns ratio of 5:5:32 (i.e. 1 to 3.2), will present a load impedance of about 350 ohms to the antenna202and tuning network204combination. Thus, when the combination of antenna202and tuning network204has a source impedance of 350 ohms, the pick-up signal across the secondary windings of transformer202is attenuated by 6 db (i.e. signal is reduced by a factor of 2). Attenuation for out-of-band frequencies can be designed to be much higher, as illustrated by another exemplary embodiment shown onFIG. 2I.

In accordance with another aspect of the invention, the impedance of the load presented to the combination of antenna202and tuning network204may be tailored to reduce the secondary electromagnetic radiation from antenna202to below a predetermined value. Generally speaking, it is understood that the magnitude of a secondary magnetic field radiated by an antenna (such as antenna202onFIGS. 2A,2B and2G) is proportional to the frequency for a given circulating current. For this reason, the load impedance may be tailored to be higher for the higher interrogation frequencies, as compared to the lower interrogation frequencies in order to reduce measurement error components due to antenna differential pair magnetic cross-coupling. The particular interrogation frequencies may be selected for each application based on the attributes of a formation being investigated. The impedance characteristics of the load presented to antenna202and tuning network204may be tailored specifically for each application based, for example, on the selected interrogation frequencies, the source impedance characteristics, and desired gain of the received pick-up signal. The invention is not limited in regard to selection of load impedance. For out-of-band frequencies, preamplifier circuit206may present a low impedance load to the antenna202and tuning network204without affecting the measurements.

FIG. 2Eshows a block diagram of an embodiment of a preamplifier206that has an input impedance that presents a load impedance to tuning network204and antenna202. InFIG. 2E, the load impedance is tailored to reduce the secondary electromagnetic radiation from antenna202to be below a predetermined level. Zp is selected to illustrate the input impedance characteristics of preamplifier206.FIG. 2Fis a plot of Zp derived by shorting the signal across nodes222and223and simulating the impedance observed across nodes221and224.FIG. 2Fshows that in a particular embodiment, at 2 Mhz the impedance Zp is about 350 ohms.

FIGS. 2G through 2Iare illustrative of the low level of circulating currents in antenna202achieved, in the exemplary embodiments, while maintaining serviceable gain at each or the plurality of interrogation frequencies.FIG. 2Gshows antenna receiver system200with the embodiment of tuning network204that has the characteristics shown inFIG. 2C, and the preamplifier that has the characteristics shown inFIG. 2F.FIG. 2His a plot of impedance ZLversus frequency response for the antenna receiver system200shown inFIG. 2G. ZLis selected, for illustration purposes, by breaking the circuit at node224and simulating the impedance ZLversus frequency response to indicate the level of circulating currents in antenna202.FIG. 2Iis a simulated plot approximating the gain Vout/Vin, as shown inFIG. 2Gwith node224coupled to antenna202(as shown by the dotted line). The gain achieved for each of the two interrogation frequencies of 500 Mhz and 2 Mhz is shown, as well as the relative attenuation of the other frequencies.

Additional considerations may be required in the tailoring of the antenna source/load impedance. The antenna load impedance primarily comprises of the “magnetizing” impedance of transformer220and the impedance presented by preamplifier circuit206. These two impedances are in parallel. Step-up transformer200may have a magnetizing impedance that varies with frequency as well as temperature. In order to maintain thermal stability of the combined impedance within the wide range of extreme temperature conditions that may be encountered within a borehole, the impedance of the preamplifier may advantageously be selected to be lower than that of the transformer at the integration frequencies.

The conductors coupling antenna202to preamplifier circuit206are preferably implemented to be short, in order to minimize the load on antenna202. In actual application, however, the tuning network204and preamplifier circuit206may effectively negate this minimized loading, as the preamplifier, in the exemplary embodiments, may need to be located within a few inches of the receiver antenna.

In accordance with another aspect of the present invention, exemplary embodiments may employ a substantial mismatch in the impedance of antenna202and its load to allow embodiments of antenna202to comprise of more than one turn. For example, a six turn antenna of about a 6 inch diameter has a characteristic inductance of about 18 μH. It is generally understood that multiple turn antenna generally provide a stronger pick-up signal than can be achieved with a single turn antenna of similar geometries. The invention is nonetheless not limited in this regard, and other embodiments may advantageously provide receiver circuits having an antenna with a single turn, two or more turns, or a different geometry.

FIG. 3is a detailed schematic of an exemplary embodiment of a tuning network300that is suitable for the tuning network204shown inFIG. 2. Tuning network300is configured, in the illustrated embodiment, to provide simultaneous series tuning at the two predetermined interrogation frequencies of 500 Khz and 2 Mhz, although it will be understood that the invention is not limited in this regard. Tuning network300onFIG. 3for loop antenna202onFIG. 2Ais coupled in series to the primary winding of step-up transformer220and antenna202. One end of tuning network300couples to node222and on the other end to node223, as shown on bothFIGS. 2A and 3. Tuning network300is comprised of inductor L1360and capacitor C1362, which are coupled together in parallel. The parallel combination is coupled in series with capacitor C2368.

As noted earlier in the discussion ofFIG. 2A, antenna202and tuning network204cooperate to achieve simultaneous series tuning by providing for a combined impedance that effectively excludes (or substantially attenuates) reception at frequencies outside each of the narrow bands of frequencies around the plurality of interrogation frequencies. For a given antenna inductance LAnt, the following simultaneous equations are satisfied for two operating frequencies (it is assumed that the real part of the various impedances are negligible):
ZAnt1+ZC2+ZC1L1=0 at FrequencyF1  (1)
ZAnt2+ZC2+ZC1L1=0 at FrequencyF2  (2)
where ZAnt>0, Zc2<0 and represents the reactance of antenna202and the reactance of tuning capacitor C2368; and ZC1L1=(ZC1*ZL1)/(ZC1+ZL1) and represents the net reactance of the parallel combination of L1360and C1362.

In an exemplary embodiment, the interrogation frequencies are selected to be about 500 Khz and 2 Mhz and antenna202has a characteristic inductance LAntof 18 μH; C1362is selected to be 620 pfd; C2368is selected to be 1250 pfd; and inductance of L1360is selected to be 47 μH. In addition, the effective inductance variation of antenna202at the different operating frequencies preferably is taken into account. Inductance variation with frequency may occur if the self-resonant frequency of the loop antenna202is not significantly higher than the highest desired operating frequency.

Typically there is also a point of maximum impedance between the two minima. Selection of inductor L1360has some influence on the frequency point of maximum impedance located between the two minima, and also on the impedances of the out-of-band frequencies; however, the values of C1362and C2368are unique for a given value of L1360and the value of LAnt.

FIG. 4shows a detailed schematic of an alternative embodiment of a tuning network400that is suitable for a tuning network204shown inFIG. 2. Tuning network400, in an exemplary embodiment, is configured to provide simultaneous series tuning at the two selected interrogation frequencies of 500 Khz and 2 Mhz. Tuning network400for loop antenna202is coupled in series to the primary winding of step-up transformer220and antenna202. One end of tuning network400couples to node222and on the other end to node223. Tuning network400comprises of the parallel configuration of two circuits: the first circuit comprises an inductor L1470and a capacitor C1472that are coupled together in series and the second circuit comprises an inductor L2474and a capacitor C2476that are coupled together in series. The inductance of the antenna202and tuning network204is preferably negligible (advantageously as close to zero) at each interrogation frequency to achieve simultaneous series tuning. Accordingly, the following two equations are solved for each of the two interrogation frequencies F1and F2to determine the particular values of inductance for L1470and L2474and particular values of capacitance for C1472and C2476:

With reference toFIG. 5, one skilled in the art will recognize that the present invention is not limited in its series tuning aspect to any particular number of interrogation frequencies. For example,FIG. 5shows a tuning network500which provides simultaneous series tuning at three interrogation frequencies of 500 Khz, 2 Mhz and 1 Mhz. One end of tuning network500couples to node222and on the other end to node223. Tuning network500comprises a parallel combination including inductor L1560and capacitor C1562, and a parallel combination including L2563and C3565. The two parallel combinations are coupled in series together and with capacitor C2566. Simultaneous series tuning is achieved by providing an equation representing the combined impedances of the tuning network500and antenna202and solving the equation to where the combined impedance is zero at each of the three interrogation frequencies.

Antenna202and tuning network204cooperate to achieve simultaneous series tuning by providing for a combined impedance that sufficiently attenuates reception by antenna200to effectively exclude reception at frequencies outside each of the predetermined narrow bands of frequencies around the interrogation frequencies. This approach leads to three equations that are solved for a 18 μH antenna202at the three interrogation frequencies of 500 Khz, 1 Mhz, and 2 Mhz. L1and L2are each selected to be 10 μH, C2is selected to equal 2400 pfd, C3is selected to equal 1100 pfd and C1is selected to equal 3300 pfd.

FIG. 6Adepicts a high-level block diagram of a circuit suitable for the preamplifier circuit206depicted in the schematic ofFIG. 2. InFIG. 6A, the impedance of the load to antenna202and tuning network204is advantageously tailored specifically for a particular application based on the selected plurality of interrogation frequencies. This objective is to reduce the effect of the secondary electromagnetic field to below a predetermined acceptable level. A pick-up signal from the secondary winding of step-up transformer220is received on input nodes235and238. An amplified version of the pick-up signal is generated on output nodes281and282. Preamplifier circuit206, in this exemplary circuit, is comprised of an amplifier circuit616, with a positive gain, and a positive-feedback path614for providing feedback that is a function of frequency to tailor the input impedance of the preamplifier circuit206.

It will be appreciated that the present invention is also able to reduce the disadvantageous effectives of cross-coupling. A beneficial result of maximizing the load impedance presented to the receiver loop antenna202is that the secondary electromagnetic field radiated by receiver loop antenna202is minimized for the interrogation frequencies, thereby diminishing the cross-coupling error in any other receiver antenna that is sufficiently close to be measurably affected by cross-coupling. In addition, the feedback mechanism may be selected specifically to maintain thermal stability of the combined load impedance of the step-up transformer202and the preamplifier circuit206.

FIG. 6Bshows a detailed schematic of an exemplary embodiment that is suitable for a preamplifier circuit206. Node238and node281are coupled to a common ground. A pick-up signal from the secondary winding of step-up transformer220is received on node235and an amplified version of the pick-up signal is generated on amplifier output node282. Amplifier circuit616is implemented with operational amplifier652in conjunction with resistors R7644, R13646, R2654and C4656to receive the pick-up signal on node235and to generate an amplified version of the pick-up signal on output node282. A positive-feedback path614is provided by the series network, linking the output node282to the positive input of operational amplifier652. Positive feedback path614comprises resistor R7644, capacitor C19642, resistor R14648, and the parallel combination of R2654and C4656.

In one embodiment of the circuit ofFIG. 6B, operational amplifier652may be deployed in the form of an integrated circuit available from Elantec, part no. EL2125. Resistor R2654and capacitor C4656are coupled together in parallel and each couple on one end to output node282and on the other end to the negative input to operational amplifier652. In this exemplary embodiment, resistor654is 100 ohms and capacitor656is 47 pf. The negative input of operational amplifier652is coupled to ground through resistor R6650, which is 10 ohms. Resistors R7644and R13646are each 750 ohms and are each coupled on one end in series to each other. One end of the resistor pair R7644and R13646is coupled to the positive input of operational amplifier652and the other end of resistor pair R7644and R13646is coupled to ground. Capacitor C19642, selected to be 1000 pf, is coupled in series to resistor R14648, selected to be 249 ohms. The series combination of capacitor C19642and resistor R14648are coupled on one end to the negative input of operational amplifier652and on the other end to the node that couples to both of the resistors R7644and R13646.

In the exemplary embodiment depicted inFIG. 6B, preamplifier circuit206is specifically tailored for interrogation frequencies 500 Khz and 2 Mhz. The addition of capacitor C19642and resistor R14648provide the positive-feedback path614shown inFIG. 6A, which increases the input impedance of the preamplifier circuit206at 2 Mhz and to a lesser degree at 500 Khz. This particular configuration of a positive-feedback path, also known as “boot-strapping,” results in the effective load impedance of the pre-amplifier206being increased by more than a factor of two at 2 MHz. In addition, the impedance at the lower operating frequency has also been increased significantly. The capacitance value of C19642and resistance value of R14648are selected to increase or decrease the amount of positive feedback at the plurality of interrogation frequencies to reduce the magnitude of the secondary electromagnetic field to below a predetermined acceptable level. Note however that, in some applications, the shunt admittance of the transformer, if dominant, may tend to create thermal instability of the input impedance presented by the step-up transformer220and preamplifier circuit206. In such situations, thermal instability considerations may set an upper boundary for efforts to increase the input impedance of the preamplifier circuit206.

With further reference to the exemplary embodiment depicted inFIG. 6B, the impedance of the preamplifier circuit206without the presence of capacitor C19642and resistor648R14would be approximately 1500 ohms, which is the sum of the two 750 ohm resistors. The addition of capacitor C19642and resistor R14648raises the input impedance of the preamplifier circuit from 1500 ohms to approximately 3500 ohms at the 2 MHz interrogation frequency. At interrogation frequency 500 Khz, the impedance of the preamplifier circuit is about 2600 ohms and the load impedance presented to the antenna is about 146 ohms due to the step-up transformer. Accordingly, the secondary electromagnetic field and error due to cross-coupling are kept below a predetermined value for each of the interrogation frequencies.

It will be appreciated that a higher value of impedance for the preamplifier circuit206could be achieved by reducing the turns ratio of the step-up transformer. However, such a reduction in the turns ratio would require additional amplification by pre-amplifier206, thereby potentially causing a decrease in the signal to noise ratio of the antenna receiver200.

FIG. 7depicts a detailed schematic diagram of an alternative exemplary circuit that would be suitable for the preamplifier206shown inFIG. 2. A series of LC circuits are provided to tailor the input impedance for a selected plurality of interrogation frequencies so as to minimize cross-coupling between receiver antennae. The positive input to operational amplifier752is coupled to the secondary winding of step-up transformer220through node235to receive the pick-up signal from antenna202. The output of the operational amplifier connects to output node282. Resistor R2754and capacitor C4756are coupled together in parallel and each couple on one end to output node282and on the other end to the negative input to operational amplifier752. In this exemplary embodiment, resistor R2754is 100 ohms and capacitor C4756is 47 pf. The negative input of operational amplifier752is coupled to ground through 10 ohm resistor R6750. The positive input to the operational amplifier752connects to ground through a DC blocking capacitor C3780and a circuit comprising two LC circuits coupled in series. The first LC circuit comprises inductor L1772coupled in parallel to capacitor C1776. The second LC circuit comprises inductor L2774coupled in parallel to capacitor C2778. The values of L1772and C1776are advantageously selected according to the following equation for the first interrogation frequency F1:

The values of L2and C2are advantageously selected according to the following equation for the second interrogation frequency F2: