Current sensor for phase inversion-modulation of AC signals

A circuit for sensing the magnitude of an AC signal applied to the circuit includes control means for reversing the polarity of the AC signal to thereby generate an output signal which is proportional in amplitude to the magnitude of the AC signal and which has a phase, with respect to the phase of the AC signal, determined by the control means. The control means also provides the capability of generating output pulses from the circuit which have an amplitude proportional to the magnitude of the AC signal and a pulse-width-spacing proportional to the duty cycle of a pulsating control signal applied to the control means.

BACKGROUND OF THE INVENTION 
1. Field of the Invention 
The present invention is related generally to electronic current sensor 
type circuits and more particularly to such type circuits for use in 
electronic metering devices wherein phase inversion or 
amplitude-mark-space modulation (time division) techniques are employed to 
make precision measurements of alternating current and/or voltage. 
2. Description of the Prior Art 
While the present invention has application in most any environment where 
the magnitude of a large AC signal has to be sensed and/or measured, it 
finds particular application in such devices as watthour meters, 
watt-transducers and the like. 
In recent years, electronic watt-hour or energy consumption meters have 
come into existence, with the expectation that they will one day replace 
the older prior art induction type meters. This electronic metering 
technology is still in the refinement stages of development. Three such 
type electronic energy consumption meters are disclosed in my U.S. Pat. 
Nos. 3,955,138; 3,875,509, and 3,875,508. The entire disclosure of these 
earlier patents is hereby incorporated by reference. The entire right, 
title and interest in and to the inventions described in the 
aforementioned patents and the entire right, title and interest in and to 
the invention herein disclosed, as well as in and to the patent 
application of which this application is a part, are assigned to the same 
assignee. 
One major problem in electronic watt-meter design is to sense or measure 
the magnitude of the current of an AC signal with high precision, while 
simultaneously inverting that signal in a circuit which utilizes a minimum 
amount of power and which presents the lowest possible load or impedance 
to the circuit being monitored. While the above mentioned precision is 
possible with the use of large and expensive electrical and electronic 
components, the cost and size of those components quickly removes the 
economic incentive for a user to purchase a meter using such components. 
As such, a need exists for a universally usable AC sensor circuit design 
having phase inversion or mark-space-modulation capabilities which can be 
fabricated from small low cost components and which design provides for 
virtually powerless sensing of the magnitude of the current of an AC 
signal. 
SUMMARY OF THE INVENTION 
The present invention fulfills the above needs by the provision of a low 
cost electronic sensor circuit suitable for fabrication in monolithic 
integrated circuit form. 
The sensor circuit of the present invention provides, in combination, a 
small low impedance current transformer terminated in an active load 
amplifier circuit through a polarity switch which can be used for 
modulation and/or phase inversion of a sensed AC signal. 
Through the combined use of a low input impedance amplifier and a 
substantially zero resistance polarity switch for connecting the current 
transformer to the input of the amplifier, the current transformer 
continuously operates virtually short circuited regardless of the position 
of the switch. By virtue of this short circuit operation, the potential or 
voltage difference across the input terminals of the amplifier is always 
negligible. Because of this low voltage difference, it is possible to use 
low voltage inexpensive transformers, analog switches and components in 
the present invention. Further, the low potential difference substantially 
reduces or eliminates switching voltage transients which normally occur at 
the switched inputs of amplifiers of the type contemplated for use in the 
present invention. 
The foregoing advantages afforded by the present invention provide a 
circuit capable of virtually powerless precision monitoring of large AC 
currents to obtain a phase controllable output voltage from the amplifier 
which is directly proportional to the magnitude of the monitored current. 
In view of the preceding, it is therefore an object of the present 
invention to provide an alternating current sensor circuit having enhanced 
operating characteristics. 
It is another object of the present invention to provide a sensor circuit 
capable of virtually powerless precision monitoring of alternating 
current. 
A still further object of the invention is to provide a sensor circuit 
suitable for applications requiring the precision sensing and/or 
measurement of alternating current signals which does not load the circuit 
being monitored. 
Yet another object of the present invention is to provide a switch operated 
phase inversion-modulation sensor circuit capable of accurately generating 
an output voltage proportional to sensed current whereby the output 
voltage is proportional to monitored current and has a phase and/or 
modulation characteristic determined by the switch position and/or the 
duty cycle of a control signal applied to the switch. 
It is still a further object of the present invention to provide a sensor 
circuit for monitoring alternating signals of large current magnitude 
capable of being fabricated from low cost, small size current and voltage 
components.

DESCRIPTION OF THE PREFERRED EMBODIMENTS 
A first illustrated embodiment of the invention is shown in the schematic 
diagram of FIG. 1, wherein a current transformer CT is connected to an 
alternating current power source (not shown) for monitoring or sensing an 
AC-current I.sub.1. As shown, the primary winding of the transformer is a 
single wire carrying the AC-current I.sub.1. The secondary winding of the 
transformer consists of many windings or turns for transforming the 
primary current I.sub.1 to a desired secondary current shown as I.sub.2. 
The secondary winding of the transformer CT is connected to inverting (-) 
and non-inverting (+) input terminals of a transresistance amplifier OA1 
via a double pole double throw switch illustrated as SW1 and SW2. There 
are any number of different types of amplifiers suitable for use with the 
present invention. One such amplifier is a type uA741 high performance 
integrated circuit operational amplifier manufactured by Fairchild 
Semiconductor, a division of Fairfield Camera and Instrument Corporation, 
313 Fairchild Drive, Mountain View, California. 
A conventional power supply is provided having a circuit ground or 
reference potential for providing two bias voltages +V1 and -V2 to 
amplifier OA1. 
Still referring to FIG. 1, the aforementioned switch (SW1 and SW2) serves 
as an analog switch for switching the polarity of the input current 
I.sub.2 applied to the first (-) and second (+) input terminals of OA1. 
While that switch may be a mechanical switch as diagrammatically shown, 
those in the art will recognize that in actual practice SW1 and SW2 may be 
realized as a plurality of active electronic switch elements controlled by 
an electronic control signal applied to a control terminal D. Preferably, 
the switch (SW1 and SW2) is realized as a low cost low voltage C-MOS 
device such as conventionally available in the art, or as C-MOS elements 
which are integrally formed in a single monolithic integrated circuit 
together with the other circuitry of FIG. 1. 
As shown in FIG. 1, SW1 is comprised of switch contacts or terminals A1, B1 
and C2 and SW2 has similarly associated terminals A2, B2, and C2. 
Terminals A1 and B2 of SW1 and SW2 respectively are secured together as 
output terminals and connected to the inverting (-) input terminal of OA1 
at a summation point S3. In a similar fashion, output terminals B1 and A2 
of SW1 and SW2 respectively are connected to the non-inverting (+) input 
terminal of OA1. As shown, the + input terminal of OA1 also serves as a 
ground return for current to flow from the amplifier OA1 through the CT 
secondary winding during circuit operation. 
As previously mentioned, the polarity of the secondary current I.sub.2 is 
switch controlled for application to the input terminals of OA1. As seen 
by observation of FIG. 1, this control is provided by the connections of 
SW1 and SW2, where the secondary winding of the CT is connected to the 
input terminals C1 and C2 of the analog switch. With the switch contacts 
C1, A1 and C2, A2 in the positions shown, an output voltage Vz is 
generated at an output terminal C of OA1 which is in phase with the line 
current I.sub.1 and which has an amplitude proportional to the magnitude 
of that current. When the switch contacts are changed to connect B1 and B2 
to C1 and C2 respectively, the polarity of the current I.sub.2 applied to 
the input terminals of OA1 is changed. This results in a change in the 
phase of the output voltage Vz with respect to the line current I.sub.1. 
With the switch contacts in this latter position, the amplitude of Vz is 
still proportional to the magnitude of I.sub.1. 
Two feedback resistors R1 and R2 are also connected from the output 
terminal C of OA1 to the input terminals of OA1 (via SW1 and SW2) for 
controlling the gain of the amplifier. As shown in FIG. 1, the resistors 
R1 and R2 are terminated at corresponding summation points S1 and S2 
respectively. With the switch contacts C1, A1 and C2, A2 in the positions 
shown, the gain of OA1 is controlled by feedback resistor R1, which is 
connected to the - input terminal of OA1 via the contacts C1, A1 of SW1. 
When the switch is changed to close contacts C1, B1 and C2, B2 of SW1 and 
SW2 respectively, R2 serves to control the gain of OA1 by being connected 
to the - input terminal of OA1 via SW2. 
The operating principles of the invention will be disclosed with reference 
to FIG. 1. However, as an aid to an understanding of some of those 
principles, reference is made to my prior U.S. Pat. No. 3,815,013. That 
patent is entitled "Current Transformer With Active Load Termination" and 
it is assigned to the same assignee of the present invention. This patent 
provides a detailed analysis and an operational description of a current 
transformer, similar to transformer CT, operating directly into a low 
impedance or transresistance amplifier similar to OA1 of FIG. 1. 
Some of the analytical details of the circuit described in my 
aforementioned U.S. Pat. No. 3,815,013 are applicable to an understanding 
of the present invention. However, those details will not be discussed 
herein, except as necessary to a complete understanding of the present 
invention. For that reason, that patent is incorporated herein as 
reference material for information purposes. 
To understand the operation of the present invention, it is considered 
advantageous to make a few observations. First, as described in my U.S. 
Pat. No. 3,815,013, the transformer CT is considered as an ideal 
AC-current source having a very low (neglible) output or source impedance 
at the operating frequency of I.sub.1 (eg. 60 Hz line voltage). Also, 
amplifier OA1 has a very low input impedance Z.sub.i which may be 
expressed by the following approximation: 
EQU Z.sub.i =R.sub.f /A.sub.0 (equation 1) 
Where R.sub.f represents a feedback impedance, such as the resistance of 
R.sub.1 or R.sub.2, connected between points C and S.sub.1 or C and 
S.sub.2 respectively, and Ao represents the open loop gain of amplifier 
OA1. This expression (equation 1) neglects any resistance in the contacts 
C1, A1 and C2, A2. The effects of this resistance will be described in the 
ensuing description. 
Because of the low source impedance of transformer CT and the low input 
impedance of amplifier OA1 (see U.S. Pat. No. 3,815,013), the secondary 
winding of the transformer is virtually short circuited. This results in 
an ideal (short circuit) current transformer operating condition. 
Also, as contemplated by the present invention, the line current I.sub.1 is 
an alternating current sign wave which goes through positive and negative 
half cycles. Thus, by reference to FIG. 1, it can be seen that I.sub.2 
will flow first in one direction through the secondary winding of CT 
during one half cycle (eg. positive), and then in the opposite direction 
during the other half cycle (eg. negative) for a complete cycle of I.sub.1 
and I.sub.2. 
With the previous observations in mind, reference is now made to FIG. 1. 
During the positive half cycle of I.sub.2 (and I.sub.1), I.sub.2 flows 
into the summing points S.sub.1 in the direction of the arrow head as 
illustrated. From S.sub.1, I.sub.2 flows through resistor R.sub.1 into the 
output terminal C of amplifier OA1. Amplifier OA1 is now functioning as a 
current sink for a load not shown. Under this condition I.sub.2 flows 
through OA1, and out of the -V2 terminal back to the power supply, where 
the current path is completed to ground at point M to return back to the 
CT secondary winding via contacts C2, A2 of SW2. 
On the next half cycle (negative) of I.sub.2 (and I.sub.1) the direction of 
current flow reverses through the secondary winding of the CT and now 
flows out of terminal C of OA1, through R1 into S1 to the CT secondary 
winding. Under this condition, OA1 is now operating as a current source 
for the load whereby current is being provided to OA1 via the +V1 terminal 
of the power supply, with the current return path to the power supply 
being through the secondary winding to poing M (ground). 
It should be noted that summing point S3 is at the same potential as 
summing point S1. As described in my U.S. Pat. No. 3,815,013, because R1 
is connected between the inverting (-) input (via SW1) and the output C of 
OA1, the potential of S3 (and S1) is virtually at zero volt, with a 
typical value being 0.5 millivolt or less. Further, the impedance between 
S1 (and S3) and M (ground) is very low. Typically, this impedance is 0.2 
ohm or less. Thus, the potential between M (ground) and S1 (and S3) is 
substantially the same (ie. zero). As a result, the summing point S3 (or 
S1) conducts no current to point M (ground). In other words, the input 
current to the inverting (-) input terminal at S3 of OA1 is so 
insignificant that it can be neglected. 
The output voltage V.sub.z, as measured between points C and M (ground), is 
proportional to I.sub.1 or I.sub.2 and can be calculated from the 
following expression: 
EQU V.sub.z =-I.sub.2 .multidot.R.sub.1 (equation 2) 
Utilizing typical values in the above equation 2 as an example, if I.sub.2 
=2 ma and R1=10K ohms, then V.sub.z =-10 volts. The negative sign appears 
in equation 2 and this example because V.sub.z is inverted 180.degree. in 
phase with respect to I.sub.1 or I.sub.2. Other values for R.sub.1 can 
also be used, depending upon the magnitude of I.sub.1 (or I.sub.2) and the 
amplitude desired for V.sub.z. 
Still referring to FIG. 1, let it now be assumed that the switch (SW1 and 
SW2) contacts are changed to close contacts C1, B1 and C2, B2 whereby, 
contacts A1 and A2 are open. The transformer secondary is now grounded at 
S1 via C1, B1 of SW1 to point M (the + input of OA1) and the other end of 
the secondary is applied to S3, the (-) input of OA1 via summation point 
S2 and C2, B2 of SW2. As can be seen, the polarity of the current I.sub.2 
applied to OA1 is now reversed 180.degree.. This in turn causes a 
180.degree. phase reversal of the output voltage V.sub.z. It should also 
be noted, that resistor R1 is removed from the circuit and replaced by 
resistor R2 to now provide the aforementioned feedback path around OA1 as 
previously described. 
With the SW1 and SW2 contacts in the last described position, the invention 
operates in the same manner as when those contacts are in the C1, A1 and 
C2, A2 closed positions. Further, the preceding expression for the 
calculation of input impedance (Zi) and the output voltage (V.sub.z) also 
apply, if R2 is substituted for R1. 
Still referring to FIG. 1, it will be noted that, as SW1 and SW2 are 
switched between positions A1, B1 and A2, B2 respectively, the resistors 
R1 and R2 are alternately grounded at one end at S1 and S2 via the 
respective switch contacts. If it is assumed that there is no resistance 
between any of the contacts of SW1 and SW2, which would be the case if a 
mechanical switch is employed, then points S1 and S2 will alternately be 
at ground (zero potential). Thus, as previously described, the secondary 
winding of the CT is always short circuited by the low input impedance of 
OA1 via SW1 and SW2. 
It is known, however, that solid state analog switches, such as the C-MOS 
type, do present a finite resistance between their respective contacts. 
This resistance, when this type of switch is utilized in the embodiment of 
FIG. 1, causes a slight voltage drop to occur between the contacts of SW1 
and SW2. 
With the switch contacts in the position shown in FIG. 1, the end of R2 at 
point S2 is normally considered to be at virtually ground potential, 
however, due to the resistance of contacts C2, A2 of SW2, point S2 is at 
some finite potential above ground. It has been found that the contact 
resistance of a typical C-MOS type switch is approximately 10 ohms. 
Neglecting any currents which flow in the circuit, except that current 
which flows through contacts A2, C2 of SW2, and assuming I.sub.2 =1 ma and 
Ron=10 ohms (Ron is the resistance of contacts C2, A2), then the voltage 
drop across A2, C2, or the potential of S2 with respect to ground, is 10 
millivolts (ie, 1 ma of current is flowing through C2, A2 contacts). 
An additional current is caused to flow through C2, A2 contacts by virtue 
of the presence of R2, which causes an additional voltage drop across 
those contacts. This additional current flow can be understood by assuming 
that I.sub.2 at S1 is in a positive half cycle. This causes I.sub.2 to now 
flow through R.sub.1 into OA1 at C as previously described. Also, note 
that S1 is positive with respect to point C, because of the negative 
output (-V.sub.z) of OA1. It is further significant to note that, even 
though the resistance of contacts C1, A1 of SW1 is also approximately 10 
ohms, that resistance has a negligible affect on the potentials of S1 and 
S3. This is because virtually no current is flowing into the inverting (-) 
input of OA1. All of I.sub.2 is being diverted through R1, thus causing S1 
to be at virtually zero volt (eg. 0.5 millivolt). Also, in the embodiment 
of FIG. 1, R1 and R2 are preferably matched resistors of the same value. 
With the preceding assumption and understanding, reference is now made to 
point S2 of FIG. 1. As shown, a second current I.sub.2 ' is shown flowing 
from S2 through R2 into point C of OA1. Using the preceding value where 
I.sub.2 =1 ma, I.sub.2 ' also equals approximately 1 ma. That is, the 
current I.sub.2 ' flowing through R2 is I.sub.2 '.apprxeq.V.sub.z /R2+Ron 
(where Ron is the C2, A2 contact resistance). This second current (I.sub.2 
') is due to the contact resistance of SW2, and it is added to I.sub.2, in 
the following manner. I.sub.2 ' now flowing into the output of OA1 exits 
the amplifier through the -V2 terminal with I.sub.2 with a combined 
current of 2 ma (I.sub.2 +I.sub.2 ') to the power supply where the added 
currents return via ground to point M through contacts C2, A2 of SW2. This 
double current (I.sub.2 +I.sub.2 ') now presents a voltage drop across C2, 
A2 or a potential at S2, of 20 millivolts instead of 10 millivolts as 
previously described. 
From the preceding, it can be seen that a very small potential exists 
between points S1 and S2. Assuming that S1 is at 0.5 millivolts and S2 is 
at 20 millivolts, the potential across the secondary winding of the CT is 
only 19.5 millivolts, which is negligible, thus still providing a 
substantially short circuit operating condition for the transformer. 
While the preceding description dealt with the direction of current flow 
through SW2 when I.sub.2 was in a positive half cycle, no further 
description is believed necessary for how the circuit operates when 
I.sub.2 is in a negative half cycle, for it is believed that those skilled 
in the art can analyze the circuit operation by the mere reversal of the 
direction of current flow through the various circuit components as 
previously described. 
Further, it should be recognized that, when contacts C1, B1 of SW1 and C2, 
B2 of SW1 are closed, contacts C1, B1 of SW1 present the Ron resistance 
previously described in connection with SW2. The circuit operates in the 
same manner as previously described except that I.sub.2 flows through R2, 
I.sub.2 ' flows through R1 and (I.sub.2 +I.sub.2 ') flows through C1, B1 
of SW1. 
In view of the foregoing, it can now be seen how the invention functions as 
a virtually powerless current sensing switch controlled inverter circuit. 
As previously mentioned, the invention also operates to perform a basic 
time-division or amplitude-mark-space modulation function. Still referring 
to FIG. 1, this function is provided when an alternating or pulsating 
control signal is applied at terminal D of the analog switch (SW1 and 
SW2). A representative input signal at D may be a pulsating signal such as 
that provided by a pulse width modulator 22 as disclosed in my 
aforementioned U.S. Pat. No. 3,955,138. That signal has a duty cycle 
proportional to sensed voltage, wherein the instantaneous pulse width is 
proportional to the corresponding instantaneous magnitude of a sensed 
input voltage variable applied to the pulse-width-modulator. When a signal 
of this type is applied to terminal D to rapidly switch SW1 and SW2 (eg. 
at a 10,000 Hz rate, as compared to the 60 Hz line frequency of I.sub.1) 
the output voltage Vz is a pulsating signal which is a product of the 
current I.sub.1 (or I.sub.2) and the control signal applied to terminal D, 
with that product being equal to power (eg. kwhr) consumption. 
While the present invention contemplates its application in a watthour 
meter or the like, it also has application in most any environment where 
large AC-current has to be sensed or measured accurately without loading 
the circuit being measured. To that end, when the invention is applied as 
a switch controlled polarity reversal circuit, the output voltage or 
signal V.sub.z is proportional to the magnitude of the current I.sub.1 (or 
I.sub.2). Also, when the invention is applied as a pulse width modulator, 
the output voltage V.sub.z is a pulsating signal with an amplitude 
proportional to I.sub.1 (or I.sub.2) and a pulse width proportional to the 
duty cycle of the signal applied to terminal D of the analog switch. 
Reference is now made to FIG. 2 which illustrates an improved embodiment of 
the invention. In this embodiment, two operational amplifiers OA2 and OA3, 
with corresponding feedback resistors R4 and R3, are added to the basic 
circuit of FIG. 1. Amplifiers OA2 and OA3 may be of the same type as OA1. 
As shown in FIG. 2, summing point S1 is connected to the inverting (-) 
input of OA3, with summing point S2 similarly connected to the inverting 
(-) input of OA2. Also, the non inverting (+) inputs of OA1, OA2 and OA3 
are connected together to a common potential source or ground at point M. 
The operation of the embodiment of FIG. 2 is basically the same as that 
previously described for FIG. 1, except for the inclusion of OA2 and OA3. 
For that reason, only that portion of the embodiment of FIG. 2 comprising 
OA2 and OA3 will be described. 
The advantage of FIG. 2 is that the use of OA2 and OA3 eliminates the need 
for low contact resistance analog switches. With the switch contacts C1, 
A1 of SW1 and C2, A2 of SW2 in the positions shown, the amplifier OA2, in 
conjunction with its resistor R4, functions as a feedback circuit with the 
"on" resistance Ron of C2, A2 in a feedback loop to summing point S2 and 
the - input of OA2. 
The input impedance of OA2 is very low, with the impedance between S2 and 
point M equal to (R4+Ron)/Ao, where Ao is the open loop gain of the 
amplifer OA2. The potential between S2 and M is thus very low. A typical 
value being much less than one millivolt. 
Reference is now made to OA3 and OA1 of FIG. 2. It will be noted that 
contacts C1, A1 of SW1 are closed to complete the circuit to point S3 to 
allow OA1 to function as previously described. Also, with the switch 
contacts as shown, OA3 has no affect on the operation of the circuit. This 
is because contacts C1, B1 are open, thus removing the amplifier as a 
feedback circuit via R3 and the "on" resistance of contacts C1, B1 of SW1. 
As previously described in connection with FIG. 1, point S1 of that 
embodiment, and of FIG. 2, is always at substantiately zero volt with 
respect to point M. As a result, as can be seen in FIG. 2, the potentials 
of S1 and S2 with respect to point M are both very small, with a typical 
measurement being less than one millivolt. The secondary of the 
transformer CT is thus terminated on both ends in an ideal substantially 
zero impedance. As a result, and in keeping with the teachings of the 
invention, the embodiment of FIG. 2 provides a virtually non-loading or 
powerless sensing circuit for monitoring the current I.sub.1. 
Still referring to FIG. 2, if SW1 and SW2 are switched to close contacts 
C1, B1 and C2, B2 respectfully, amplifier OA3, in conjunction with R3 and 
the "on" resistance of contacts C1, B1 provide the feedback closed loop 
circuit in the manner just described for the amplifier OA2. In this case, 
OA2 is now isolated from the circuit via contacts C2, A2 as previously 
described for OA3. Further, it will be noted that contacts C2, B2 are 
closed to thus reverse the polarity of V.sub.z with respect to I.sub.2. 
FIG. 3 is similar to the embodiment of FIG. 2, but including an additional 
set of switch contacts A3, B3 and C3 generally shown as SW3 to form a 
triple pole double throw switch. This embodiment operates in the same 
manner as FIG. 2, with the exception that total isolation is provided by 
the contacts of SW3 at the outputs of amplifiers OA2 and OA3. As shown in 
FIG. 3, the output of amplifier OA3 is grounded at point M via contacts 
C3, A3 to isolate OA2 from OA3, while contacts C2, A2 are closed to 
complete the feedback path around OA2. In a similar fashion, the output of 
OA2 is grounded at point M and OA3 is isolated from OA2 when contacts C3, 
B3 are closed along with contacts C1, B1 and C2, B2. In this embodiment, 
SW3 ensures that any small voltage gradient present at the output of the 
amplifier (OA2 or OA3) not being used does not affect the potential of the 
input of the amplifier connected in feedback configuration. This is 
effected by absolutely grounding the output terminal of the presently 
unused amplifier while isolating the presently used amplifier from the 
unused amplifier. Also, in the embodiment of FIG. 3, the - input of OA1 is 
connected to ground (point M). 
In the embodiments of FIGS. 2 and 3, while it is preferable that the 
resistors R1, R2, R3 and R4 be of the same value, those resistors do not 
have to be matched as preferred for the FIG. 1 embodiment. This is because 
of the use of amplifiers OA2 and OA3 which allow the use of non-low 
contact resistance switches in the aforementioned feedback loops to keep 
the potentials of S1 and S2 at virtually zero volts. 
FIG. 4 is a similar embodiment to that of FIG. 1, incorporating SW3 as a 
switchable feedback element with R1 around OA1, which eliminates R2. This 
embodiment operates in the same manner as heretofore described for FIG. 1, 
except for the control of feedback current by SW3. With the switch 
contacts in the position shown, current I.sub.2 is allowed to flow from 
point S1 through SW3 contacts A3, C3 and R1 into point C and back to S1 as 
previously described. When the switch (SW1, SW2, SW3) contacts are 
switched to the bottom positions of FIG. 4, current I.sub.2 is then 
allowed to flow from S2 through R1 via SW3 contacts B3, C3 into OA1 at 
point C. Current from OA1 is also allowed to flow back to point S2 through 
this same path with a change in the polarity of I.sub.1 or I.sub.2 in the 
same manner that it flows into point S1 from the OA1 output at point C. 
In embodiment of FIG. 4, there is some slight "on" resistance Ron across 
contacts C2, A2 and C1, B1 when they are closed causing a small voltage 
drop across those respective contacts. I.sub.2 ' is not present in this 
embodiment, thus, if I.sub.2 =1 ma and Ron=10 ohms, then the potential of 
S1 or S2 is approximately 10 millivolts. This causes a small voltage 
across the secondary winding of 9.5 millivolts (ie, the potential of S2-10 
millivolts minus the potential of S1=0.5 millivolts equals 9.5 millivolts 
and visa virsa). 
In each of the embodiments of FIGS. 1-4, two reverse parallel connected 
diodes D1 and D2 are connected across the terminals of the secondary 
winding of the transformer CT. These diodes serve to provide over voltage 
or surge current protection for the transformer and the amplifiers 
connected directly across the secondary winding. Diodes D1 and D2 are 
normally "off" (non-conducting), as the potential between S1 and S2 and 
ground is normally low enough to prevent conduction of the diodes. The 
diodes utilized in the present invention are designed to conduct when a 
potential of 500 millivolts is applied across the diodes from S1 to S2. 
When either D1 or D2 conducts (depending on the polarity of I.sub.2) the 
transformer secondary winding and the amplifier inputs are shorted out to 
thus prevent damage to the circuitry. These diodes also provide protection 
for the circuitry should some component failure cause the potential of 
either S1 or S2 to rise above the prescribed diode conduction level. 
Although this invention has been described with respect to a few particular 
exemplary embodiments, those in the art will appreciate that it is 
possible to modify many features of the exemplary embodiments without 
departing from the new and improved teachings and features of this 
invention. Accordingly, all such modifications are intended to be 
incorporated within the scope of the present invention.