Brushless motor, external AC voltage source, and electric power steering device

The present invention provides a brushless motor in which the voltage utilization ratio can be increased and the torque and output of the motor can thereby be increased, and also provides a drive method for a brushless motor. The brushless motor includes an armature constituted by an armature core having armature windings of a plurality of phases, and a field pole constituted by a field pole core having a plurality of permanent magnets. A voltage, in which at least a 5th order harmonic component is superimposed on a 1st order fundamental wave of a voltage under predetermined phase difference condition and amplitude condition in order to increase a 1st order fundamental wave peak of the applied voltage over an applied voltage peak, is applied to the armature windings.

TECHNICAL FIELD

The present invention relates to a brushless motor suitable for use in an electric power steering device or the like, and to a drive method for a brushless motor.

BACKGROUND ART

A method is known for increasing a torque while inhibiting torque ripple in a brushless motor by adjusting the induced voltage of each phase to a trapezoidal waveform in which an odd-order harmonic is superimposed on a 1st order fundamental wave.

However, when a harmonic is superimposed on the induced voltage to obtain a trapezoidal waveform shape of the induced voltage of each phase, since the phase current of each phase has a sine waveform, a peak current value is considered to be difficult to suppress. For this reason, the current that can flow in a brushless motor is restricted and the motor output is accordingly inhibited.

A brushless motor in which the inter-phase induced voltage waveform between two different phases of armature windings is provided with a trapezoidal waveform has been suggested (see, for example, Patent Document 1). In such a brushless motor, as a result of the inter-phase induced voltage waveform between two different phases being provided with a trapezoidal waveform, the phase current has a trapezoidal waveform and the current peak value of the 1st order component contributing to the motor torque can be increased over that of the sine phase current having the same phase current peak value. Therefore, a high torque and high revolution are realized while inhibiting the peak current value.

DISCLOSURE OF THE INVENTION

However, the following problems are associated with the related art.

Thus, when the phase current waveform of a brushless motor is a trapezoidal waveform such that harmonics with a phase difference θn,0(n=3, 5, 7 . . . ; n is a harmonic order) are included in the 1st order fundamental wave and the 1st order fundamental wave peak is made larger than the waveform peak, a voltage harmonic component is generated in the armature windings of the motor.

In this case, under the effect of motor resistance or inductance, the voltage harmonic component has a phase difference of θn,0+θn,1with the 1st order fundamental wave of the voltage. Therefore, the 1st order fundamental wave peak of the voltage typically does not become larger than the waveform peak.

Where a voltage applied from the outside is considered for the purpose of energizing an electric current of the above-described substantially trapezoidal waveform, it is necessary to apply a voltage with a waveform peak larger than the 1st order fundamental wave peak as a voltage equal to the above-described voltage harmonic component.

Further, in the voltage applied to the motor, the component affecting torque and output is the 1st order fundamental wave component, and a limitation is placed on the peak of the applied voltage that can be supplied from an external AC voltage source to the armature windings.

A problem caused by the generation of the aforementioned voltage harmonic component in the armature windings of the motor is that the ratio of the 1st order fundamental wave component of the voltage affecting the torque and output decreases with respect to the applied voltage peak, and the voltage utilization ratio decreases by comparison with that during the sine current energizing.

The present invention has been created to resolve the above-described problems, and it is an objective thereof to provide a brushless motor in which the voltage utilization ratio can be increased and the torque and output of the motor can thereby be increased, and to provide a drive method for a brushless motor.

A brushless motor in accordance with the invention includes an armature constituted by an armature core having armature windings of a plurality of phases, and a field pole constituted by a field pole core having a plurality of permanent magnets, wherein a voltage, in which at least a 5th order harmonic component is superimposed on a 1st order fundamental wave of a voltage under predetermined phase difference condition and amplitude condition in order to increase a 1st order fundamental wave peak of the applied voltage over an applied voltage peak, is applied to the armature windings; a harmonic voltage superimposed on the armature windings is the 5th order harmonic component; and the 5th order harmonic component has a phase difference of 150° to 210° with the 1st order fundamental wave of the voltage, where one period of the high-order harmonic component is taken as 360°, and a ratio of an amplitude of the 5th order harmonic component to an amplitude of the 1st order fundamental wave of the voltage is 2% to 12%.

Further, a brushless motor in accordance with the invention including an armature constituted by an armature core having armature windings of a plurality of phases, and a field pole includes an armature constituted by an armature core having armature windings of a plurality of phases, and a field pole constituted by a field pole core having a plurality of permanent magnets, wherein a voltage, in which at least a 5th order harmonic component is superimposed on a 1st order fundamental wave of a voltage under predetermined phase difference condition and amplitude condition in order to increase a 1st order fundamental wave peak of the applied voltage over an applied voltage peak, is applied to the armature windings; a harmonic voltage superimposed on the armature windings is the 5th order harmonic component and a 7th order harmonic component; and the 5th order harmonic component and the 7th order harmonic component have a phase difference of 120° to 240° with the 1st order fundamental wave of the voltage, where one period of the high-order harmonic component is taken as 360°, and a ratio of a sum of amplitudes of the 5th order harmonic component and the 7th order harmonic component to an amplitude of the 1st order fundamental wave of the voltage is 2% to 36%.

With the brushless motor and the drive method for a brushless motor in accordance with the present invention, a voltage, in which at least either of the 5th order harmonic component and the 7th order harmonic component is superimposed on the 1st order fundamental wave of the voltage under predetermined phase difference condition and amplitude condition such that the 1st order fundamental wave peak of the applied voltage is made larger than the applied voltage peak, is applied to the armature windings.

Therefore, by increasing the 1st order fundamental wave peak of the applied voltage over the applied voltage peak, it is possible to increase the voltage utilization ratio of the brushless motor and increase the torque and output of the motor.

BEST MODE FOR CARRYING OUT THE INVENTION

The preferred embodiments of a brushless motor and a drive method for a brushless motor according to the invention are explained below with reference to the appended drawings in which like or corresponding components are assigned with like reference numerals.

FIG. 1is a block diagram showing a brushless motor1according to Embodiment 1 of the invention together with an external AC voltage source2. InFIG. 1, the brushless motor1is driven by application of an AC voltage (motor application voltage) V from the external AC voltage source2to interphases UV, VW, WU of three phases U, V, W. The combination of the brushless motor1and the external AC voltage source2can be also considered as a brushless motor.

FIG. 2is a cross-sectional view showing the configuration of the brushless motor1shown inFIG. 1. InFIG. 2, the armature of the brushless motor1is constituted by an armature core12having three-phase armature windings11, and field poles are constituted by a field pole core14having a plurality of permanent magnets13. After the armature windings11of all phases have been wired, terminals U′, V′, W′ are connected to the phases U, V, W, respectively, of the external AC voltage source2shown inFIG. 1.

The motor shown inFIG. 2is but an example. Thus, permanent magnets13may be embedded in the field pole core14, as shown, for example, inFIG. 3, and the number of the permanent magnets13and the number of the armature windings11are not limited to those shown inFIG. 2.

FIG. 4is a circuit diagram showing equivalent circuits of the interphases UV, VW, WU of the external AC voltage source2shown inFIG. 1. InFIG. 4, the voltage (motor application voltage) V applied to the armature windings11of the brushless motor1is a sum of an induced voltage E=ωφmgenerated between the terminals by the rotation of the field poles of the motor, and the voltage drop components on an inductance L of the armature winding11between the motor terminals, the armature windings11, and the resistance R of the external AC voltage source2. For the φm, a magnetic flux generated from the permanent magnets13is taken to be interlinked with the armature windings11between the terminals U′V′, V′W′, W′U′ of the brushless motor1.

FIG. 5is a vector diagram relating to the case in which three phases of the circuit shown inFIG. 4are converted into two phases along dq axes, and an electric current is energized in the q-axis direction. InFIG. 5, Lqis a q-axis inductance, and φmdis a d-axis component of the magnetic flux φm. It follows fromFIG. 5, that the effective voltage value V applied to the motor is represented by the following Eq. (1).

It follows from Eq. (1), that the phase of the current iqand voltage V applied to the motor differs depending on the resistance component and inductance component of the motor.

Further, the motor torque is proportional to the product of the magnetic flux φmand electric current i. However, the average torque of the motor is affected by the 1st order fundamental wave component of the magnetic flux φmand electric current i, and the respective higher harmonic components cause torque ripples. The motor output is represented by a product of the motor torque and revolution speed, and where the motor torque increases at the same revolution speed, the motor output also increases.

FIG. 6is an explanatory drawing showing the waveform of the voltage V applied from the external AC voltage source2to the armature winding11of each phase shown inFIG. 4in Embodiment 1 of the invention. InFIG. 6, a substantially trapezoidal voltage is applied to the armature windings11, and the settings are such that the 1st order fundamental wave peak V1pof the voltage V is larger than the peak Vpof the voltage V.

Such settings can be explained as follows. Since the 1st order fundamental wave of the voltage applied to the circuit shown inFIG. 4is the 1st order fundamental wave of the electric current, when the voltage peak Vpis constant, the torque and output increase with the increase in the 1st order fundamental wave peak V1p. Here, V1p/Vpis a voltage utilization ratio related to a sine voltage, and as the value thereof increases, larger torque and output can be obtained for the voltage with the same peak.

InFIG. 6, the voltage V applied to the armature winding11is represented by the following Eq. (2) as a sum of the 1st order fundamental wave component and a 5th order higher harmonic component.

In Eq. (2), V1pand V5prepresent peaks of respective harmonics, and θ5stands for a phase difference between the 1st order fundamental wave component and the 5th order higher harmonic component. InFIG. 6, θ5is taken as 180° and V5p/V1pis taken as 0.06. With such settings, the V1p/Vpratio is 1.05, and the torque and output increase.

The V1p/Vpratio also increases when the ratio of V5pto V1por θ5is changed.FIG. 7shows the ratio of the 1st order fundamental wave peak V1pof the voltage to the voltage peak Vpin the case in which θ5is initially taken as 180° and V5p/V1pis changed within a range of 0 to 0.15. It follows fromFIG. 7that where V5p/V1pis 0.02 to 0.12, V1p/Vpbecomes equal to or higher than 1.02.

In a more desirable case, V1p/Vpbecomes equal to or higher than 1.03 when V5p/V1pis 0.03 to 0.11. In an even more desirable case, V1p/Vpbecomes equal to or higher than 1.04 when V5p/V1pis 0.04 to 0.09.

FIG. 8shows the maximum value of the ratio (V1p/Vp) of the 1st order fundamental wave peak V1pto the voltage peak Vpin the case in which θ5is changed within a range of 140° to 220° and V5p/V1pis changed within a range of 0 to 0.15. It follows fromFIG. 8that where θ5is 150° to 210°, V1p/Vpis equal to or higher than 1.02.

In a more desirable case, V1p/Vpis equal to or higher than 1.03 when θ5is 160° to 200°. In an even more desirable case, V1p/Vpis equal to or higher than 1.04 when θ5is 170° to 190°.

It follows from the above that in order to increase V1p/Vpand input a waveform with increased torque and output at the same voltage peak, it is desirable that V5p/V1pbe set to 0.02 to 0.12 and θ5be set to 150° to 210°.

As for the range of V5p/V1p, it is desirable that V5p/V1pbe set to 0.03 to 0.11, more desirably to 0.04 to 0.09. The desirable range of θ5is 160° to 200°, and it is more desirable that θ5be set to 170° to 190°.

FIG. 9is a block diagram illustrating a method for applying a harmonic voltage of the external AC voltage source2in Embodiment 1 of the invention. InFIG. 9, the external AC voltage source2is provided with a fundamental wave voltage application unit31that applies a 1st order fundamental wave voltage to the armature winding11on the basis of a 1st order fundamental wave voltage command value, a 5th order harmonic voltage command unit32that calculates a 5th order harmonic voltage command value on the basis of the 1st order fundamental wave voltage command value, and a 5th order harmonic voltage superposition unit33that superimposes the 5th order harmonic voltage on the armature windings11on the basis of the 5th order harmonic voltage command value.

The method for calculating the 5th order harmonic voltage command value in the 5th order harmonic voltage command unit32involves calculating the 5th order harmonic voltage command value such that the 5th order harmonic component command value with the amplitude V5p, and the phase difference θ5with the 1st order fundamental wave is superimposed on the peak V1p, of the 1st order fundamental wave command value of the voltage at predetermined values, desirably V5ps/V1psfrom 0.02 to 0.12 and θ5from 150° to 210°. The 5th order harmonic voltage is superimposed in the 5th order harmonic voltage superposition unit33on the basis of the 5th order harmonic voltage command value, and the voltage V including the 5th order harmonic voltage is superimposed on the armature winding11.

Torque ripples generated in the motor in Embodiment 1 of the invention are considered below. As mentioned hereinabove, the motor torque is proportional to the product of the magnetic flux φmand electric current i. However, the average torque of the motor is affected by the 1st order fundamental wave components of the electric current i and magnetic flux φm, and the respective higher harmonic components cause torque ripples. Thus, it is usually desirable that only the 1st order basis harmonic component, such as represented by the following Eq. (3), be provided as the magnetic flux φminterlinked with the armature windings11between the terminals of the motor.

Here, a case is considered in which a voltage V in the drive method of Embodiment 1 of the invention, which is shown inFIG. 10, is applied between the motor terminals. In the voltage V shown inFIG. 10, the 5th order harmonic component has a phase difference of 180° with the 1st order fundamental wave, and the ratio of the amplitude of the 5th order harmonic component to the amplitude of the 1st order fundamental wave is about 3.5% to 4%.

When such a voltage V is applied, the electric current i flowing in the armature windings11of the motor has a waveform including the 5th order harmonic component or the 7th order harmonic component, as shown inFIG. 11. Further, in the operation mode such that the interlinkage magnetic flux represented by Eq. (3) hereinabove is interlinked with the armature windings11of the motor by the phase relationship shown inFIG. 12, the torque waveform becomes such as shown inFIG. 13. It follows fromFIG. 13that only the 6th order harmonic component of torque ripples is generated.

Meanwhile, there is also a large number of other waveforms, such as the waveform A (120° rectangular waveform) and waveform B (120° trapezoidal waveform) shown inFIG. 14for which the 1st order basis harmonic peak V1pof the voltage V becomes larger than the peak value Vpof the voltage V. However, as shown inFIG. 14, which represents the results of frequency analysis of those voltages, this voltage V typically has a waveform including odd-order components up to higher harmonics. Therefore, when such voltage V is applied, the current i flowing in each armature winding11of the motor has a waveform including odd-order components up to higher harmonics.

For this reason, for example, when the voltage with the waveform A (120° rectangular waveform) is applied and the motor is operated such that the interlinkage magnetic flux of Eq. (3) hereinabove is interlinked with the armature windings11of the motor by the phase relationship shown inFIG. 12, the torque has a waveform in which harmonic torque ripples (6n-th order, n is an integer equal to or greater than 2) are generated in addition to the 6th order harmonic component of torque ripples generated in Embodiment 1 of the invention, as shown inFIG. 15.

Thus, in Embodiment 1 of the invention, only the 5th order harmonic component is selectively applied to the armature windings11. Therefore, it is clear that no torque ripples of the 6n-th order (n is an integer equal to or greater than 2) is generated with respect to the substantially trapezoidal voltage shown inFIG. 14.

Further, in the above-described example, the case is explained in which the sine interlinkage magnetic flux represented by Eq. (3) hereinabove is interlinked with the armature windings11of the motor, but the explanation presented hereinabove is also valid when the interlinkage magnetic flux or voltage generated in the windings by a magnetic flux interlinked with the windings includes a higher harmonic, provided that the explanation is limited to torque ripples caused by the voltage.

In the actual motor, it is difficult to superimpose only the 5th order harmonic voltage on the peak V1psof the 1st order fundamental wave command value of the voltage, and it is possible that a harmonic component of the (6n±1)-th order (n is an integer equal to greater than 2) is additionally applied. In such a case, a 6n-th order (n is an integer equal to greater than 2) of harmonic torque ripples is generated in addition to the 6th order harmonic component, in the same manner as in the case of the above-described waveform A (120° rectangular waveform) or waveform B (120° trapezoidal waveform), thereby increasing the torque ripples.

FIG. 16shows the 6th order component and 12th order component of torque ripples in the case in which the 11th and 13th order harmonic voltages are applied at a ratio of a % to the peak V1pof the 1st order fundamental wave voltage.
V=V1psin(ωt)+γV1psin(5ωt+θ5)+αV1p{sin(11ωt+θ11)+sin(13ωt+θ13)}

Here, θ11, θ13represent a phase difference between the 1st order fundamental wave component and 11th order and 13th order harmonic components. The ratio γ of the amplitude of the 5th order harmonic component to the amplitude of the 1st order fundamental wave is about 0.035 to 0.04. In this case, it is clear that the 12th order component of torque ripples does not increase till α is 0 to about 0.03. This is because, the 12th order component of torque ripples generated by harmonics included in the interlinkage magnetic flux of the motor and the 12th order component of torque ripples caused by the 11th, and 13th order harmonic voltage represented by formulas hereinabove cancel each other.

Further, till α is 0 to about 0.03, the 6th order component of torque ripples is predominant and takes about 340%, as compared with the 12th order component of torque ripples. Therefore, the effect produced by the 12th order component of torque ripples on the total torque ripples is small. It follows from the above that even when the 11th and 13th order harmonic voltages are contained in certain amounts with respect to the peak V1psof the 1st order fundamental wave command value of voltage, the effect thereof on torque ripples is small.

Further,FIG. 17shows the 6th order component and 18th order component of torque ripples in the case in which the 17th and 19th order harmonic voltages are added at β% to the peak V1pof the 1st order fundamental wave voltage. This voltage is represented by the following formula.
V=V1psin(ωt)+γV1psin(5ωt+θ5)+βV1p{sin(17ωt+θ17)+sin(19ωt+θ19)}

Here, θ17, θ19represent the phase difference between the 1st order fundamental wave component and the 17th and 19th order harmonic components, respectively. The ratio γ of the amplitude of the 5th order harmonic component to the amplitude of the 1st order fundamental wave is about 0.035 to 0.04. In this case, it is clear that the 12th order component of torque ripples does not increase till β is 0 to about 0.03. This is because, the 18th order component of torque ripples generated by harmonics included in the interlinkage magnetic flux of the motor and the 18th order component of torque ripples caused by the 17th and 19th order harmonic voltage represented by formulas hereinabove cancel each other.

Further, till α is 0 to about 0.03, the 6th order component of torque ripples is predominant and takes about 500%, as compared with the 18th order component of torque ripples. Therefore, the effect produced by the 18th order component of torque ripples on the total torque ripples is small. It follows from the above that even when the 17th and 19th order harmonic voltages are contained in certain amounts with respect to the peak V1psof the 1st order fundamental wave command value of voltage, the effect thereof on torque ripples is small.

In the above-described example, the effect produced on the 12th and 18th order components of torque ripples is described with respect to the case in which the 11th and 13th, and 17th and 19th order harmonic voltages are applied, but even when a (6n±1)-th (n is an integer equal to or greater than 2) harmonic voltage is applied in a certain amount, it is canceled by the torque generated by the interlinkage magnetic flux, the ratio to the 6th order torque ripples is small, and therefore the effect on the 6n-th order (n is an integer equal to or greater than 2) of torque ripples is small.

Further, a 3(2n−1)-th (n is an integer equal to or greater than 1) voltage is included in the waveform B (120° trapezoidal waveform) shown inFIG. 14. As shown inFIG. 18, this voltage causes a circulating current when the armature windings are in a three-phase Δ connection. Therefore, in Embodiment 1 of the invention, only the 5th order harmonic component is selectively applied to the armature winding11. As a result, the circulating current generated in the Δ connection shown inFIG. 18is not generated.

However, in the actual motor, it is difficult to superimpose only the 5th order harmonic voltage on the peak V1psof the 1st order fundamental wave command value of the voltage, and it is also possible that a 3(2n−1)-th (n is an integer equal to or greater than 1) harmonic component is additionally applied.

FIG. 19shows a 3rd order component of the circulating current generated in the Δ connection shown inFIG. 18in the case in which the 3rd order harmonic voltage is applied at δ% to the peak V1pof the 1st order base voltage. In this case, it is clear that the 3rd order component of the circulating current does not increase till β is 0 to about 0.15. This is because the 3rd order component of the circulating current generated by the harmonics contained in the interlinkage magnetic flux of the motor and the 3rd order component of the circulating current caused by the 3rd order harmonic voltage represented by the formula above cancel each other. It follows from the above that in the Δ connection, even if the 3rd order harmonic voltage is contained in a certain amount with respect to the peak V1psof the 1st order fundamental wave command value of the voltage, the effect thereof on the 3rd order component of the circulating current is small.

Further, in the above-described example, the effect on the 3rd order component of the circulating current is described with respect to the case in which the 3rd order harmonic voltage is applied, but even if the 3(2n−1)-th (n is an integer equal to or greater than 1) harmonic voltage is applied in a certain amount, it is canceled by the 3(2n−1)-th (n is an integer equal to or greater than 1) circulating current generated by the interlinkage magnetic flux. Therefore, the effect produced thereby on the 3(2n−1)-th (n is an integer equal to or greater than 1) circulating current can be said to be small.

As described hereinabove, in Embodiment 1, a voltage obtained by superimposing at least either of the 5th order harmonic component and the 7th order harmonic component on the 1st order fundamental wave under the predetermined phase difference condition and amplitude condition such that the 1st order fundamental wave peak of the applied voltage is made larger than the applied voltage peak is applied to the armature windings.

Therefore, as a result of the 1st order fundamental wave peak of the applied voltage being larger than the applied voltage peak, it is possible to increase the voltage utilization ratio of the brushless motor and increase the torque and output of the motor. Further, since at least either of the 5th order harmonic component and the 7th order harmonic component is selectively applied in order to increase the 1st order fundamental wave peak of the applied voltage over the applied voltage peak, the torque ripples of the 6n-th order component (n is an integer equal to or greater than 2) caused by the application of harmonics, and the generation of the circulation current in the Δ connection shown inFIG. 18can be inhibited.

Thus, in Embodiment 1 of the invention, a substantially trapezoidal voltage obtained by superimposing the 5th order harmonic component on the 1st order fundamental wave is applied to the armature windings, and the 5th order harmonic component has a phase difference (θ5) of 150° to 210° with the 1st order fundamental wave, where one period of the high-order harmonic component is taken as 360°, and the ratio (V5p/V1p) of the amplitude of the 5th order harmonic component to the amplitude of the 1st order fundamental wave is 2% to 12%.

As a result, the 1st order fundamental wave peak V1pof the substantially trapezoidal voltage applied to the brushless motor can be increased by 2% or more over the peak Vpof the substantially trapezoidal voltage. Therefore, the voltage utilization ratio relating to a sine voltage can be increased and the torque and output of the brushless motor can thereby be increased.

Further, when such a brushless motor is used in an electric power steering device, since the torque and output of the brushless motor are increased, the electric power steering device can be reduced in size and weight, and a vehicle where the electric power steering device is installed can be also reduced in size and weight.

FIG. 20is an explanatory drawing showing the waveform of the voltage V applied from the external AC voltage source2to the armature winding11of each phase shown inFIG. 4in Embodiment 2 of the invention. InFIG. 20, a substantially trapezoidal voltage is applied to the armature winding11, and the settings are such that the 1st order fundamental wave peak V1pis larger than the peak Vpof the voltage V.

Further, inFIG. 20, the voltage V applied to the armature windings11is represented by the following Eq. (4) as a sum of the 1st order fundamental wave component, 5th order harmonic component, and 7th order harmonic component.

In Eq. (4), V1p, V5p, and V7pstand for peaks of waveforms of respective orders, θ5is a phase difference between the 1st order fundamental wave component and the 5th order harmonic component, and θ7is a phase difference between the 1st order fundamental wave component and the 7th order harmonic component. InFIG. 20, θ5and θ7are set to 180°, V5p/V1pis set to 0.126, and V7p/V1pis set to 0.054. With such settings, V1p/Vpis about 1.077, and the torque and output are increased.

Further, V1p/Vpis also increased when the ratios of V5pand V7pto V1p, or θ5and θ7are changed. It follows fromFIG. 21that the ratio of the 1st order fundamental wave peak V1pof the voltage to the voltage peak Vpin the case in which θ5and θ7are set to 180° and (V5p+V7p)/V1pis changed within a range of 0 to 0.40.FIG. 21demonstrates that the ratio V1p/Vpis equal to or greater than 1.02 when (V5p+V7p)/V1pis 0.02 to 0.36.

More desirably, the ratio V1p/Vpis equal to or greater than 1.04 when (V5p+V7p)/V1pis 0.04 to 0.32. Even more desirably, the ratio V1p/Vpis equal to or greater than 1.05 when (V5p+V7p)/V1pis 0.06 to 0.30. Still more desirably, the ratio V1p/Vpis equal to or greater than 1.06 when (V5p+V7p)/V1pis 0.10 to 0.26. And even more desirably, the ratio V1p/Vpis equal to or greater than 1.07 when (V5p+V7p)/V1pis 0.14 to 0.22.

FIG. 22shows the maximum value of the ratio (V1p/Vp) of the 1st order fundamental wave peak V1pof the voltage to the voltage peak Vpin the case in which θ5and θ7are changed within a range of 120° to 240° and (V5p+V7p)/V1is changed within a range of 0 to 0.40. It follows fromFIG. 22that V1p/Vpis equal to or greater than 1.02 when θ5and θ7are within a range of 120° to 240°.

More desirably, the ratio V1p/Vpis equal to or greater than 1.03 when θ5and θ7are within a range of 140° to 220°. Even more desirably, the ratio V1p/Vpis equal to or greater than 1.04 when θ5and θ7are within a range of 150° to 210°. Still more desirably, the ratio V1p/Vpis equal to or greater than 1.05 when θ5and θ7are within a range of 160° to 200°. And even more desirably, the ratio V1p/Vpis equal to or greater than 1.06 when θ5and θ7are within a range of 170° to 190°.

The above-described results indicate that in order to input a waveform that increases V1p/Vpand also increase the torque and output at the same voltage peak, it is desirable that (V5p+V7p)/V1pbe set to 0.02 to 0.36, and θ5and θ7be set to 120° to 240°.

Further, the (V5p+V7p)/V1prange is more desirably set to 0.04 to 0.32, even more desirably to 0.06 to 0.30, still more desirably to 0.10 to 0.26, and even more desirably to 0.14 to 0.22, and the θ5and θ7range is more desirably set to 150° to 210°, even more desirably to 160° to 200°, and still more desirably to 170° to 190°.

FIG. 23is a block diagram showing the harmonic voltage application method of the external AC voltage source2in Embodiment 2 of the invention. InFIG. 23, the external AC voltage source2is provided with a fundamental wave voltage application unit31that applies the 1st order fundamental wave voltage to the armature windings11on the basis of the voltage command value of the 1st order fundamental wave, a 5th order harmonic voltage command unit32that computes a 5th order harmonic voltage command value on the basis of the voltage command value of the 1st order fundamental wave, a 5th order harmonic voltage superposition unit33that superimposes the 5th order harmonic voltage on the armature windings11on the basis of the 5th order harmonic voltage command value, a 7th order harmonic voltage command unit34that computes a 7th order harmonic voltage command value on the basis of the voltage command value of the 1st order fundamental wave, and a 7th order harmonic voltage superposition unit35that superimposes the 7th order harmonic voltage on the armature windings11on the basis of the 7th order harmonic voltage command value.

In the method for computing the 5th and 7th order harmonic voltage command values in the 5th and 7th order harmonic voltage command units32,34, the 5th and 7th order harmonic voltage command values are computed such that the 5th order harmonic voltage command value with an amplitude V5psand a phase difference θ5with the 1st order fundamental wave and the 7th order harmonic voltage command value with an amplitude V7p, and a phase difference θ7with the 1st order fundamental wave are superimposed on the peak V1psof the 1st order fundamental wave command value of the voltage at respective predetermined values, desirably, (V5p+V7p)/V1pin a range of 0.02 to 0.36 and θ5and 07 in a range of 120° to 240°. Harmonic voltages are superimposed in the 5th and 7th order harmonic voltage superposition units33,35on the basis of the 5th and 7th order harmonic voltage command values, and the voltage V including the harmonic voltages is superimposed on the armature windings11.

The torque ripples generated in the motor in Embodiment 2 of the invention are considered below. When the voltage V superimposed with the 5th order harmonic component and the 7th order harmonic component is applied, as in Embodiment 2 of the invention, the electric current i flowing in each armature winding11of the motor has a waveform including the 5th order harmonic component or the 7th order harmonic component, in the same manner as in Embodiment 1 described hereinabove. Therefore, in the torque ripples, only the 6th order harmonic component is generated, in the same manner as in Embodiment 1.

Thus, in Embodiment 2 of the invention, only the 5th order harmonic component and the 7th order harmonic component are selectively applied to the armature windings11. Therefore, it is clear that the torque ripples of the 6n-th order component (n is an integer equal to or greater than 2) are not generated with respect to the substantially trapezoidal voltage shown inFIG. 14.

Further, in the actual motor, it is difficult to superimpose only the 5th and 7th order harmonic voltages on the peak V1psof the 1st order fundamental wave command value of the voltage, and it is possible that a harmonic component of the (6n±1)-th order (n is an integer equal to greater than 2) is additionally applied. However, as explained in Embodiment 1, even if a harmonic component of the (6n±1)-th order (n is an integer equal to greater than 2) is applied in a certain amount, it is canceled by the torque generated by the interlinkage magnetic flux, the ratio to the 6th order torque ripples is small, and therefore the effect on the 6n-th order (n is an integer equal to or greater than 2) of torque ripples is small.

In the above-described example, the case is explained in which the interlinkage magnetic flux of a sine waveform represented by Eq. (3) hereinabove is interlinked with the armature windings11of the motor, but even when the interlinkage magnetic flux includes harmonics, the explanation above is valid, provided it relates to the torque ripples caused by the voltage.

Further, in Embodiment 2 of the invention, only the 5th order harmonic component and the 7th order harmonic component are selectively applied to the armature windings11. Therefore, the circulating current caused by the 3(2n−1)-th order (n is an integer equal to or greater than 1) in the Δ connection shown inFIG. 18can be inhibited.

In the actual motor, it is difficult to superimpose only the 5th order harmonic voltage on the peak V1psof the 1st order fundamental wave command value of the voltage, and it is possible that a harmonic component of the 3(2n−1)-th order (n is an integer equal to greater than 1) is additionally applied. However, even when a harmonic component of the 3(2n−1)-th order (n is an integer equal to greater than 1) is applied in a certain amount, it is canceled by the circulating current of the 3(2n−1)-th order (n is an integer equal to greater than 1) generated by the interlinkage flux. Therefore, the effect on the circulating current of the 3(2n−1)-th order (n is an integer equal to greater than 1) can be said to be small.

As described hereinabove, in Embodiment 2, a voltage obtained by superimposing at least either of the 5th order harmonic component and the 7th order harmonic component on the 1st order fundamental wave under the predetermined phase difference condition and amplitude condition such that the 1st order fundamental wave peak of the applied voltage is made larger than the applied voltage peak is applied to the armature windings.

Therefore, as a result of the 1st order fundamental wave peak of the applied voltage being larger than the applied voltage peak, it is possible to increase the voltage utilization ratio of the brushless motor and increase the torque and output of the motor. Further, since at least either of the 5th order harmonic component and the 7th order harmonic component is selectively applied in order to increase the 1st order fundamental wave peak of the applied voltage over the applied voltage peak, the torque ripples of the 6n-th order component (n is an integer equal to or greater than 2) caused by the application of harmonics, and the generation of the circulation current in the Δ connection shown inFIG. 18can be inhibited.

Thus, in Embodiment 2 of the invention, a substantially trapezoidal voltage obtained by superimposing the 5th order harmonic component and the 7th order harmonic component on the 1st order fundamental wave is applied to the armature windings, and the 5th order harmonic component and the 7th order harmonic component have a phase difference (θ5, θ7) of 120° to 240° with the 1st order fundamental wave, where one period of the high-order harmonic component is taken as 360°, and the ratio ((V5p+V7p)/V1p) of the sum of the amplitude of the 5th order harmonic component and the amplitude of the 7th order harmonic component to the amplitude of the 1st order fundamental wave is 2% to 36%.

As a result, the 1st order fundamental wave peak V1pof the substantially trapezoidal voltage applied to the brushless motor can be increased by 2% or more over the peak Vpof the substantially trapezoidal voltage. Therefore, the voltage utilization ratio relating to a sine voltage can be increased and the torque and output of the brushless motor can thereby be increased.

Further, when such a brushless motor is used in an electric power steering device, since the torque and output of the brushless motor are increased, the electric power steering device can be reduced in size and weight, and a vehicle where the electric power steering device is installed can be reduced in size and weight.

FIG. 18is a connection diagram showing the connection method of the armature windings11of each phase of the brushless motor1in Embodiment 3 of the invention. InFIG. 18, the Δ connection is used for the armature windings11of all phases. In addition to the connection shown inFIG. 18, the Y connection shown inFIG. 24can be also used for the armature windings11of the three-phase brushless motor1. In Embodiment 3 of the invention, the external AC voltage source2is connected between the terminals, and the brushless motor1is driven by the drive method of the above-described Embodiment 1 or 2.

Where the wire diameter and the number of turns are considered to be the same and only the connection method is changed with respect to the armature windings11of each phase shown inFIGS. 18 and 24, the resistance and inductance of the armature windings11of each phase are the same inFIGS. 18 and 24, and where the resistance and inductance between the terminals in the Y connection inFIG. 24are denoted by R and L, respectively, the resistance and inductance in the Δ connection shown inFIG. 18are ⅓R and ⅓L, respectively.

Since the resistance and inductance between the terminals shown inFIGS. 18 and 24correspond to the resistance R and inductance L between the motor terminals shown in the above-describedFIG. 4, in the Δ connection, the motor loss and voltage drop between the terminals are less than those in the Y connection, and the current i that can flow in the brushless motor1can be increased. Therefore, the motor torque and output in the drive methods of the above-described Embodiment 1 or 2 can be increased.

Where the armature windings11are wound such that the resistance values between the terminals shown inFIGS. 18 and 24are the same and the number of turns w in each phase is the same, the resistance value of the winding of each phase is inversely proportional to the second power of the wire diameter d in the motor and can be represented by the following Eq. (5).

Therefore, when the resistance values between the terminals are the same, the resistance value of the armature winding11of each phase in the Δ connection shown inFIG. 18is three times that in the Y connection shown inFIG. 24, as indicated hereinabove. Therefore, the wire diameter can be made 1/√3 that in the Y connection. As a result, even though the resistance value of the motor is the same, the operability of the winding operation of the motor realizing the drive method of the above-described Embodiment 1 or 2 can be improved.

As indicated hereinabove, according to Embodiment 3, with the Δ connection of the armature windings of all phases, in the drive method according to the above-described Embodiment 1 or 2, it is possible to increase the torque and output of the motor and improve the winding operability of the motor which is the drive object.

FIG. 25is an explanatory drawing showing the torque characteristic versus the revolution speed of the brushless motor1in Embodiment 4 of the invention. It follows fromFIG. 25that the inclination of the torque characteristic increases in the negative direction at a certain revolution speed as a boundary. This is because the peak Vpof the voltage V applied to the motor is restricted by the maximum inter-phase voltage peak Vmaxthat can be outputted by the external AC voltage source2and the current that flows in the motor is restricted by the voltage, with an inflection point as a boundary.

A combination of a DC voltage source and a frequency converter that converts the DC voltage into frequency-variable AC voltage is mostly used as the external AC voltage source2of the brushless motor1, and the maximum inter-phase voltage peak Vmaxthat can be outputted by the external AC voltage source2is mostly determined by the performance of the DC voltage source.

Therefore, the operation region of the brushless motor1can be divided into the operation region in which the inter-phase voltage V applied to the motor is less than the inter-phase voltage peak Vmaxof the external AC voltage source2and the torque is restricted by the maximum current, and the operation region in which the inter-phase voltage V applied to the motor is equal to the inter-phase voltage peak Vmaxof the external AC voltage source2(voltage saturation), and the torque is restricted by the voltage.

In Embodiment 4 of the invention, the brushless motor1is driven by the drive method of the above-described Embodiment 1, 2 or 3 in the operation region in which the peak Vpof the applied inter-phase region V is equal to the inter-phase voltage peak Vmaxof the external AC voltage source2.

By driving the brushless motor1by using the drive method of the above-described Embodiment 1, 2 or 3, it is possible to obtain the voltage utilization ratio V1p/Vprelated to the voltage peak Vpequal to or greater than 1.02, and in this operation region, the voltage peak Vpis equal to Vmax, and therefore V1max/Vmaxis equal to or greater than 1.02. In this case, V1maxhas the value of the 1st order fundamental wave peak of the Vmaxvoltage.

FIG. 26is an explanatory drawing in which the relationship between the revolution speed and torque characteristic is shown for the brushless motor1of Embodiment 4 of the invention in comparison with that obtained with the side voltage drive in the case of using the external AC voltage source2having the same inter-phase voltage peak Vmax. It follows fromFIG. 26that in the operation region in which the applied inter-phase voltage V is equal to the maximum inter-phase voltage peak Vmaxthat can be outputted by the external AC voltage source2, the torque and output can be increased by comparison with those in the side voltage drive.

Thus, with Embodiment 4, in the operation region in which the applied voltage V is equal to the maximum inter-phase voltage peak Vmaxthat can be outputted by the external AC voltage source2, V1max/Vmaxcan be made equal to or greater than 102% and the torque and output of the motor can be increased even while using the external AC voltage source2having the same inter-phase voltage peak Vmax.

FIG. 27is a block diagram showing the brushless motor of Embodiment 5 of the invention together with the external AC voltage source. InFIG. 27the external AC voltage source2connected to the armature windings11is constituted by one DC voltage source21and two frequency converters22,23that convert the DC voltage into frequency-variable AC voltages.

Concerning the phases U1, U2, V1, V2, W1, and W2of the frequency converters22,23, the phases U1, U2are connected to a terminal U′ of the armature winding11of the motor, V1, V2are connected to a terminal V′ of the armature winding11of the motor, and W1, W2are connected to a terminal W′ of the armature winding11of the motor. In Embodiment 5 of the invention, the external AC voltage source2is connected between the terminals by the connection method shown inFIG. 27, and the brushless motor1is driven by the drive method of the above-described Embodiment 1, 2, 3, or 4.

In this case a resistance component Raccreated by a switching element or a relay is present in the AC portion from each phase of the frequency converters22,23to the motor terminal. However, in the configuration shown inFIG. 27, since the two frequency converters22,23are connected in parallel, when the resistance components Racof the frequency converters22,23are the same, the resistance component is Rac/2

Since this resistance component is included in the resistance R between the motor terminals in the above-describedFIG. 4, the loss and voltage drop between the phases of the external AC voltage source2decreases and the current that can flow in the brushless motor1can be increased by comparison with those in the case in which a single frequency converter is used. Therefore, the torque and output of the motor in the drive method of the above-described Embodiment 1, 2, 3, or 4 can be increased.

As described hereinabove, according to Embodiment 5, as a result of using a plurality of frequency converters it is possible to increase the torque and output of the motor in the drive method of the above-described Embodiment 1, 2, 3, or 4.

In Embodiment 5, the case is explained in which the external AC voltage source is constituted by a single DC voltage source and two frequency converters that convert the DC voltage into respective frequency-variable AC voltage, but such a configuration is not limited, and the resistance voltage can be reduced and the same effect can be obtained also when the number of DC voltage sources is more than one and the number of frequency converters is more than two.

FIG. 28is an enlarged drawing illustrating one pole of the field poles of the brushless motor of Embodiment 6 of the invention. In the brushless motor shown inFIG. 28, a magnet attachment surface is provided on the surface of the field pole core, a permanent magnet is attached to the magnet attachment surface by using an adhesive or the like, and the settings are such that the thickness of the magnet in the center portion and at the ends is h1and h2, respectively.

In the brushless motor shown inFIG. 28, the induced voltage E generated between the terminals of the motor shown inFIG. 4by the rotating field poles is represented by the following Eq. (6) in the case in which the motor has a symmetrical structure for each magnetic pole.

In Eq. (6), the voltage E is represented by a sum of the 1st order fundamental wave component and a (2k+1)-th order harmonic component (k is an integer equal to or greater than 1). Further, E1pand E(2k+1)prepresent a peak of the waveform of each order, and θ(2k+1)erepresents a phase difference between the 1st order fundamental wave component and (2k+1)-th order harmonic component.

Where the induced voltage peak is denoted by EP, as shown inFIG. 4, Epis the maximum value, on the time axis, of the voltage applied from the outside between the phases of the external drive source. Here, the upper limit for the maximum value of the breakdown voltage between the phases of the external drive source is determined by the switching element or circuit element used in the external AC source, and where the breakdown voltage is exceeded, the external drive source can be damaged. Therefore, it is desirable that the peak of the induced voltage be selected small with consideration for the upper-limit voltage. Meanwhile, the torque T of the brushless motor is represented by the following Eq. (7).

Here, ω stands for the angular speed of the brushless motor. As shown in Eq. (7), the torque of the brushless motor is proportional to the fundamental wave component E1pof the induced voltage E. Therefore, in order to increase the torque value, while inhibiting the induced voltage peak, it is necessary to increase the ratio of the fundamental wave peak E1pto the induced voltage peak Ep, that is, increase the E1p/Epratio.

In Embodiment 6, the applied induced voltage E is represented, as shown in Eq. (8) hereinbelow, by a sum of the 1st order fundamental wave component, a 5th order harmonic component, and the other higher harmonic order component.

Here, E1p, E5pare peaks of waveforms of respective orders, Eotherrepresents the other harmonic order, and θ5eis a phase difference between the 5th order harmonic component and the fundamental wave. Changes in E1p/Epoccurring when E5p/E1p, that is, the application ratio of the 5th order harmonic to the fundamental wave of the induced voltage, is changed are considered below.FIG. 29shows the ratio of the fundamental wave of the voltage E1pto the induced voltage peak Epwhen a voltage is induced for which θ5eis set to 180° and E5p/E1pis set to 0% to 15%. It follows from the figure, that E1p/Epis equal to or greater than 1.02 when E5p/E1pis taken as 2% to 12%, and E1p/Epis at a maximum when E5p/E1pis about 6%.

FIG. 30shows the maximum value of Eap/Epwhen θ5is set to 140° to 220° and E5p/E1pis changed within a range of 0% to 15%. It follows from the figure, that E1p/E1pis equal to or greater than 1.02 when θ5is 150° to 210°, and E1p/Epis at a maximum when θ5is 180°. However, in this case, it is assumed that the other harmonic order Eoth, is sufficiently small by comparison with E1pand E5p.

FIG. 31shows the relationship between the peak E5pand phase difference θ5, of the 5th order harmonic component of the induced voltage and the ratio h2/h1of h1and h2in Embodiment 6. Referring toFIG. 29to select the h2/h1ratio such that ensures the following values, it can be found that E5p/E1pis about 6%, θ5eis about 180°, and E1p/Epis at a maximum when h2/h1is about 0.7. It follows from the above that in order to increase E1p/Ep, it is desirable that h2/h1be set to about 0.7 in the brushless motor shown inFIG. 28.

In the above-described example, the case is considered in which the value of the induced voltage E is represented by a sum of the 1st order fundamental wave component, 5th order harmonic component, and the component of other order, as shown in Eq. (8) hereinabove, but the case can be also considered in which the induced voltage is represented by a sum of the 1st order fundamental wave component, 5th order harmonic component, 7th order harmonic component, and the component of other order, as shown in Eq. (9) below.

Here, E1p, E5p, and E7pare peaks of waveforms of respective orders, θ5eis a phase difference between the 5th order harmonic component and the fundamental wave, and θ7eis a phase difference between the 7th order harmonic component and the fundamental wave. Changes in E1p/Epoccurring when (E5p+E7p)/E1p, that is, the sum of the 5th order harmonic application ratio and 7th order harmonic application ratio related to the fundamental wave of the induced voltage, is changed are considered below.FIG. 32shows the ratio of the 1st order fundamental wave E1pof the voltage to the voltage peak Epwhen θ5eand θ7eare set to 180° and (E5p+E7p)/E1pis set to 0% to 40%. It follows fromFIG. 32, that E1p/Epis equal to or greater than 1.02 when (E5p+E7p)/E1pis taken as 2% to 36%, and E1p/Epis at a maximum when (E5p+E7p)/E1pis about 18%.

FIG. 33shows the maximum value of E1p/Epwhen θ5is set to 120° to 240° and E5p/E1pis changed within a range of 0% to 40%. It follows from the figure, that E1p/Epis equal to or greater than 1.02 when θ5is 120° to 240°, and E1p/Epis at a maximum when θ5is 180°. However, in this case, it is assumed that the other harmonic order Eotheris sufficiently small by comparison with E1p, E5pand E7p.

In the case in which h2/h1is set to obtain such values, the effect same as described hereinabove can be obtained. The same effect can be also obtained when a (2m+1)-th order harmonic (m is an integer equal to or greater than 1) is applied to the brushless motor, although such a case is not described hereinabove. Examples of the applied order include the 3rd, 9th, 11th, and 13th order harmonics. However, the problem arising when the induced voltage includes a 3(2k−1)-th order harmonic component (k is an integer equal to or greater than 1) is that a circulating current is generated in the case of the three-phase A connection. Therefore, the Y connection is preferred.

In a brushless motor in which the field pole core has a permanent magnet portion serving as a field pole1and a protruding portion serving as a field pole2with a polarity opposite that of the field pole1, and the field pole1and the field pole2are each produced equidistantly in the circumferential direction, as shown inFIG. 34, the induced voltage E generated between the terminals of the motor shown inFIG. 4by the rotation of the magnetic poles can be represented by the following Eq. (10).

Therefore, by performing full-pitch winding with a winding coil pitch of 180° when a pair of the N pole and S pole of field poles is at an electrical angle of 360°, it is possible to zero the even-order terms represented by Eq. (11) below. Therefore, the same arguments as in the above-described case are valid. Further, torque pulsations and cogging caused by the even-order induced voltage can be reduced.

In Embodiment 6, full-pitch winding with a winding coil pitch of 180° is performed when an angle occupied by a pair of the N pole and S pole of the field poles in the circumferential direction of the field poles is taken as an electrical angle of 360°. Therefore, the harmonic winding factor increases, and the variation amount of the application rate of the 5th and 7th order harmonic realized when h2/h1is changed can be increased. However, the same effect as described hereinabove can be also obtained in the case of concentrated winding in which the windings are wound in a concentrated manner on the teeth and when the coil pitch is set to a value other than 180°.

Further, in Embodiment 6, the 5th order harmonic component V5pis applied to the applied voltage of the brushless motor, but where the induced voltage or applied voltage is sufficiently larger than the drop of voltage on the inductance L or R in the circuit shown inFIG. 4, the induced voltage and applied voltage may be considered to be balanced.

Therefore, where the ratio E5p/E1pof the 5th order harmonic component E5pof the induced voltage to the 1st order fundamental wave component E1pis made substantially equal to the ratio V5p/V1pof the 5th order harmonic component V5pof the harmonic components of the applied voltage to the 1st order fundamental wave component V1p, the voltages E5pand V5pcancel each other, the harmonic current flowing in the current i can be inhibited, and the torque ripples of the motor can be reduced.

In this case, according to Embodiment 1, V5p/V1pis set to 0.02 to 0.12, more desirably to 0.03 to 0.11, and even more desirably to 0.04 to 0.09. Therefore, it is also desirable that E5p/E1pbe set to 2% to 12%, more desirably to 3% to 11%, and even more desirably to 4% to 9%.

Here, considering the results shown inFIG. 29, it can be said that it is desirable that h2/h1in the brushless motor shown inFIG. 28be set to 0.5 to 1.0, more desirably to 0.55 to 0.9, and even more desirably to 0.6 to 0.8. The same effect can be also obtained when an induced voltage of an order other than the 5th order is generated in the brushless motor, a voltage of the same order as the induced voltage is applied, and the ratio of the harmonic component of the induced voltage to the 1st order fundamental wave component is made substantially equal to the ratio of the harmonic component of the applied voltage to the 1st order fundamental wave component, although such a case is not described hereinabove.

FIG. 35shows the amount of magnet necessary for generating a unit torque when h2/h1is likewise changed. The amount of magnet necessary for generating a unit torque which is shown inFIG. 35is normalized by the value obtained when h2/h1is 1.0. It follows fromFIG. 35that the amount of magnet per unit torque is at a minimum when h2/h1is about 1.0.

This is because where the thickness h1of the magnet is equal to h2, the distance between the armature core and the field pole core becomes relatively small, the magnetic resistance of the portions where the permanent magnets are disposed is decreased, the induced voltage is increased, and the motor torque is increased. Therefore, it can be said that in order to reduce the amount of magnet necessary to generate a unit torque, it is desirable to set h2/h1to about 1.0.

FIG. 36shows the amplitude of the cogging torque generated in the brushless motor when h2/h1is likewise changed. The cogging torque inFIG. 36is normalized by the value obtained when h2/h1is 1.0. It follows fromFIG. 36, that where h2/h1is made about 0.65, the cogging torque is minimized. Therefore, it can be said that in order to reduce the cogging torque generated in the brushless motor, it is desirable that h2/h1be set to about 0.65.

Further,FIG. 37shows the amplitude of torque ripples generated in the brushless motor when h2/h1is likewise changed. The torque ripples shown inFIG. 37are normalized by the value obtained when h2/h1is 1.0. It follows fromFIG. 37that where h2/h1is made about 0.2, the torque ripples are minimized. Therefore, it can be said that in order to reduce the torque ripples generated in the brushless motor, it is desirable that h2/h1be set to about 0.2.

Further, the arguments same as in the above-described case are also valid in the case in which the surface of the field pole core has a magnet attachment surface, a permanent magnet is attached by using an adhesive or the like to the magnet attachment surface, the field pole core has a protruding portion between the permanent magnets, the protruding portion projecting from the attachment surface of the permanent magnets in the direction from the center of the field pole core toward the armature core, the thickness in the center of the magnet is set to h1, and the thickness at the ends of the magnet is set to h2, as shown inFIG. 38. This case is, however, different in that the field pole has the saliency of inductance and, therefore, a reluctance torque is generated.

FIG. 39shows the reluctance torque in the case in which h2/h1is changed. The reluctance torque inFIG. 39is normalized by the value obtained when h2/h1is 1.0. It follows fromFIG. 39that when h2/h1is 1.0, the reluctance torque is at a maximum.

Therefore, it can be said that in order to increase the reactance torque in the motor in which the surface of the field pole core has a magnet attachment surface, a permanent magnet is attached by using an adhesive or the like to the magnet attachment surface and the field pole core has a protruding portion between the permanent magnets, the protruding portion projecting from the attachment surface of the permanent magnets in the direction from the center of the field pole core toward the armature core, as shown inFIG. 38, it is desirable to set h2/h1to 1.0.

Since the motor torque is represented by the magnet torque generated by the permanent magnets and the reluctance torque generated irrespectively of the permanent magnets, where the reluctance torque is increased, the amount of magnet necessary for generating a unit torque can be reduced.

It follows from the above that in Embodiment 6, where h2/h1is adequately set, the motor torque is increased while increasing E1p/Epand suppressing the same induced voltage peak, and by setting the harmonic component of the induced voltage to a value substantially equal to the harmonic component of the applied voltage, it is possible to inhibit the harmonic current flowing in the current i, reduce the amount of magnet necessary for generating a unit torque, reduce the cogging torque, and reduce the torque ripples.

Further, in the motor in which the surface of the field pole core has a magnet attachment surface, a permanent magnet is attached by using an adhesive or the like to the magnet attachment surface and the field pole core has a protruding portion between the permanent magnets, the protruding portion projecting from the attachment surface of the permanent magnets in the direction from the center of the field pole core toward the armature core, as shown inFIG. 38, the reluctance torque can be increased and the amount of magnet necessary for generating a unit torque can be reduced.

However, as indicated hereinabove, the value of h2/h1at which the respective effects are remarkably demonstrated differs among the effects. Accordingly, it goes without saying that a more effective and desirable approach to increasing the performance of the brushless motor is in setting the h2/h1value such as to obtain the aforementioned plurality of effects. For example, where h2/h1is set to 0.2 to 1.0, or 0.2 to 0.7, or 0.2 to 0.65, or 0.65 to 0.1, or 0.65 to 0.7, or 0.7 to 1.0, the aforementioned plurality of effects can be demonstrated.

In a brushless motor having the field poles in which a plurality of permanent magnets is fixed to the surface of the field pole core and h2/h1is 0.2 to 1.0, where h1is the thickness of the permanent magnet in the center portion thereof and h2is the thickness of the end portion, in addition to the above-described effects, it is possible to increase the motor torque, while increasing E1p/Epand inhibiting the induced voltage peak, and by setting the harmonic component of the induced voltage to a value substantially equal to the harmonic component of the applied voltage, it is possible to inhibit the harmonic current flowing in the current i, reduce the amount of magnet necessary for generating a unit torque, reduce the cogging torque, and reduce the torque pulsations, which are the effects that could not be obtained in the conventional brushless motors.

Further, when the field pole core has a protruding portion between the permanent magnets, the protruding portion projecting from the attachment surface of the permanent magnets in the direction from the center of the field pole core toward the armature core, in addition to the above-described effects, the reluctance torque is increased and the amount of magnet necessary for generating a unit torque is reduced, which are the effects that could not be obtained in the conventional brushless motors.

Further, when the field pole core has a permanent magnet portion serving as the field pole1and a protruding portion serving as the field pole2with a polarity opposite that of the field pole1, the field pole1and the field pole2are each produced equidistantly in the circumferential direction, the angle in the circumferential direction of the field pole core occupied by the N poles and S poles of the magnets is taken as an electrical angle of 360°, and the winding coil pitch is set to 180°, the even-order induced voltage can be reduced, the motor torque is increased, while E1p/Epis increased and the same induced voltage peak is inhibited, and the cogging and ripples caused by the even-order induced voltage are reduced, which are the effects that could not be obtained in the conventional brushless motors.

Further, when a waveform in which a (2m+1)-th order harmonic (m is an integer equal to or greater than 1) is superimposed under the predetermined phase difference condition and amplitude condition, such that the 1st order fundamental wave peak of the induced voltage is made larger than the induced voltage peak, is obtained for the induced voltage generated in the armature windings between the terminals of the motor by the rotation of the field poles of the motor, the torque motor is increased, while further increasing E1p/Epand inhibiting the same induced voltage peak, which are the effects that could not be obtained in the conventional brushless motors.

Further, when the 5th order harmonic component has a phase difference of 150° to 210° with the 1st order fundamental wave of the voltage in the case in which one period of the harmonic component is taken as 360°, and the ratio of the amplitude of the 5th order component to the amplitude of the 1st order fundamental wave of the voltage is made 2% to 12%, it is possible to increase, by comparison with the above-described case, the torque motor, while further increasing E1p/Epand inhibiting the same induced voltage peak, which are the effects that could not be obtained in the conventional brushless motors.

Further, when the 5th order harmonic component and the 7th order harmonic component have a phase difference of 120° to 240° with the 1st order fundamental wave of the voltage in the case in which one period of the harmonic component is taken as 360°, and the ratio of the sum of the amplitudes of the 5th order component and the 7th order harmonic component to the amplitude of the 1st order fundamental wave of the voltage is made 2% to 36%, it is possible to increase the torque motor by comparison with that in the above-described case, while further increasing E1p/Epand inhibiting the same induced voltage peak, which are the effects that could not be obtained in the conventional brushless motors.

FIG. 40is an enlarged view showing one pole of the rotor of the brushless motor of Embodiment 7 of the invention. InFIG. 40, the brushless motor has a plurality of magnet holes for inserting permanent magnets into the field pole core. Where the permanent magnet is inserted into the magnet hole and the distance from the center of the field pole core to the farthest point of the field pole core is taken as Rc, most of the region of the outer circumferential portion of the field pole core follows a circular arc with a radius of about Rm.

In this case, with respect to the brushless motor shown inFIG. 40, the induced voltage E generated between the terminals of the motor shown inFIG. 4by the rotation of the field poles can be represented by Eq. (6) above when the motor has a symmetrical structure for each magnetic pole.

FIG. 41shows the relationship between the peak E5pand phase difference θ5eof the 5th order harmonic component of the induced voltage and the ratio Rm/Rc in Embodiment 7. In this case, where the Rm/Rc ratio is considered such that the induced voltage E1p/Epis increased, in the same manner as in Embodiment 6, it follows fromFIG. 29that E5p/E1pis 6%, θ5eis about 180 degrees, and E1p/Epis almost at a maximum when Rm/Rc is about 0.7. It follows from the above that E1p/Epcan be increased by setting Rm/Rc to about 0.7 for the brushless motor shown inFIG. 40.

Further, in the above-described example, the case is considered in which the induced voltage E is represented by the sum of the 1st order fundamental wave component, 5th order harmonic component, and other harmonic component, as in Eq. (8), but the where Rm/Rc is adequately set, the same arguments are valid, as explained in Embodiment 6, even when the induced voltage is represented by the sum of the 1st order fundamental wave component, 5th order harmonic component, 7th order harmonic component, and other order component, as in Eq. (9).

Further, although it is not described hereinabove, the same effects can be also obtained when a (2 m+1)-th harmonic (m is an integer equal to or greater than 1) is applied to the brushless motor. Examples of the applied orders are 3rd, 9th, 11th, and 13th order harmonics. However, the problem arising when the induced voltage includes a 3(2k−1)-th order harmonic component (k is an integer equal to or greater than 1) is that a circulating current is generated in the case of the three-phase Δ connection. Therefore, the Y connection is preferred.

In a brushless motor in which the field pole core has a permanent magnet portion serving as the field pole1and a protruding portion serving as the field pole2with a polarity opposite that of the field pole1, and the field pole1and the field pole2are each produced equidistantly in the circumferential direction, as shown inFIG. 42, the induced voltage E generated between the terminals of the motor shown inFIG. 4by the rotation of the magnetic poles can be represented by the Eq. (10) above. Therefore, by performing full-pitch winding with a winding coil pitch of 180° when a pair of the N pole and S pole of field poles is at an electrical angle of 360°, it is possible to zero the even-order terms represented by Eq. (11) above. Thus, the same arguments as in the above-described case are valid.

In Embodiment 7, full-pitch winding with a winding coil pitch of 180° is performed when an angle occupied by a pair of the N pole and S pole of field poles in the circumferential direction of field poles is taken as an electrical angle of 360°. Therefore, the harmonic winding factor increases and the variation amount of the application rate of the 5th and 7th order harmonic of the induced voltage when Rm/Rc is changed can be increased. However, the same effect as described hereinabove can be also obtained in the case of concentrated winding in which the windings are wound in a concentrated manner on the teeth and when the coil pitch is set to a value other than 180°.

Further, in Embodiment 7, the 5th order harmonic component V5pis applied to the applied voltage of the brushless motor, but where the induced voltage or applied voltage is sufficiently larger than the drop of voltage on the inductance L or R in the circuit shown inFIG. 4, the induced voltage and applied voltage may be considered to be balanced.

Therefore, where the ratio E5p/E1pof the 5th order harmonic component E5pof the induced voltage to the 1st order fundamental wave component E1pis made substantially equal to the ratio V5p/V1pof the 5th order harmonic component V5pof the harmonic components of the applied voltage to the 1st order fundamental wave component V1p, the voltages E5pand V5pcancel each other, the harmonic current flowing in the current i can be inhibited, and the torque ripples of the motor can be reduced.

It follows from Embodiment 1 that V5p/V1pis set to 0.02 to 0.12, more desirably to 0.03 to 0.11, and even more desirably to 0.04 to 0.09. Therefore, it is also desirable that E5p/E1pbe set to 2% to 12%, more desirably to 3% to 11%, and even more desirably to 4% to 9%.

Here, considering the results shown inFIG. 29, it can be said that it is desirable that Rm/Rc in the brushless motor shown inFIG. 40be set to 0.5 to 1.0, more desirably to 0.55 to 1.0, and even more desirably to 0.6 to 0.95. The same effect can be also obtained when an induced voltage of an order other than the 5th order is generated in the brushless motor, a voltage of the same order as the induced voltage is applied, and the ratio of the harmonic component of the induced voltage to the 1st order fundamental wave component is made substantially equal to the ratio of the harmonic component of the applied voltage to the 1st order fundamental wave component, although such a case is not described hereinabove.

Further,FIG. 43shows the amount of magnet necessary for generating a unit torque when Rm/Rc is likewise changed. The amount of magnet necessary for generating a unit torque which is shown inFIG. 43is normalized by the value obtained when Rm/Rc is 1.0. It follows fromFIG. 43that the amount of magnet per unit torque is at a minimum when Rm/Rc is 1.0.

This is because where the radius Rm is equal to Rc, the distance between the armature core and the field pole core becomes relatively small, the magnetic resistance of the air gap is decreased, the induced voltage is increased, and the motor torque is increased. Therefore, it can be said that in order to reduce the amount of magnet necessary to generate a unit torque, it is desirable to set Rm/Rc to about 1.0.

FIG. 44shows the amplitude of the cogging torque generated in the brushless motor when Rm/Rc is likewise changed. The cogging torque inFIG. 44is normalized by the value obtained when Rm/Rc is 1.0. It follows fromFIG. 44, that where Rm/Rc is made about 0.5, the cogging torque is substantially at a minimum. Therefore, it can be said that in order to reduce the cogging torque generated in the brushless motor, it is desirable that Rm/Rc be set to 0.5.

Further,FIG. 45shows the amplitude of torque ripples generated in the brushless motor when Rm/Rc is likewise changed. The torque ripples shown inFIG. 45are normalized by the value obtained when Rm/Rc is 1.0. It follows fromFIG. 45that where Rm/Rc is made 0.5, the torque ripples are substantially at a minimum. Therefore, it can be said that in order to reduce the torque ripples generated in the brushless motor, it is desirable that Rm/Rc be set to 0.5.

FIG. 46shows the reluctance torque generated in the brushless motor in the case in which Rm/Rc is changed. The reluctance torque shown inFIG. 46is normalized by the value obtained when Rm/Rc is 1.0. It follows fromFIG. 46that when Rm/Rc is 1,0, the reluctance torque is substantially at a maximum. Therefore, it can be said that in order to increase the reluctance torque, it is desirable that Rm/Rc be set to 1.0.

Since the motor torque is represented by the magnet torque generated by the permanent magnets and the reluctance torque generated irrespectively of the permanent magnets, where the reluctance torque is increased, the amount of magnet necessary for generating a unit torque can be reduced.

FIG. 47shows the inductance Ld in the d-axis direction of the brushless motor in the case in which Rm/Rc is likewise changed. The inductance Ld shown inFIG. 47is normalized by the value obtained when Rm/Rc is 1.0. It follows fromFIG. 47that when Rm/Rc is 1.0, the d-axis component of the inductance is at a maximum.

Typically, where the d-axis component of the inductance increases, it is possible to increase the effect of canceling the induced voltage E generated between the terminals of the motor shown inFIG. 4by the rotation of the field poles when the d-axis current is energized, and the current contributing to the torque which flows in the motor can be increased and the torque in a high-revolution region can be also increased. Therefore, the maximum revolution speed in the idle drive of the motor and the output during high-speed rotation are increased.

Therefore, a large inductance is desirable for increasing the maximum revolution speed in the idle drive of the motor and the output during high-speed rotation. As a consequence, it can be said that it is desirable to set Rm/Rc to 1.0 in order to increase the inductance Ld.

It follows from the above that in Embodiment 7, where Rm/Rc is adequately set, the motor torque is increased while increasing E1p/Epand suppressing the same induced voltage peak, and by setting the harmonic component of the induced voltage to a value substantially equal to the harmonic component of the applied voltage, it is possible to inhibit the harmonic current flowing in the current i, reduce the amount of magnet necessary for generating a unit torque, reduce the cogging torque, reduce the torque ripples, increase the reluctance torque, and increase the d-axis inductance.

However, as indicated hereinabove, the values of Rm/Rc at which the respective effects are remarkably demonstrated differ among the effects. Accordingly, it goes without saying that a more effective and desirable approach to increasing the performance of the brushless motor is in setting the Rm/Rc value such as to obtain the aforementioned plurality of effects. For example, where Rm/Rc is set to 0.5 to 1.0, or 0.5 to 0.7, or 0.7 to 1.0, the aforementioned plurality of effects can be demonstrated.

As described hereinabove, in a brushless motor which has magnet holes for inserting permanent magnets into a field pole core and has field poles obtained by inserting a plurality of permanent magnets into the magnet holes and fixing the permanent magnets, where the distance from the center of the field pole core to the farthest point of the field pole core is taken as Rc, most of the region of the outer circumferential portion of the field pole core follows a circular arc with a radius of about Rm, and where Rm/Rc is set to 0.5 to 1.0, it is possible to increase the motor torque, while increasing E1p/Epand inhibiting the induced voltage peak, reduce the amount of magnet necessary for generating a unit torque, reduce the cogging torque, reduce the torque pulsations, increase the inductance, and increase the maximum revolution speed during idle drive of the motor or the output during high-speed revolution which are the effects that could not be obtained in the conventional brushless motors.

Further, when the field pole core has a permanent magnet portion serving as the field pole1and a protruding portion serving as the field pole2with a polarity opposite that of the field pole1, the field pole1and the field pole2are each produced equidistantly in the circumferential direction, the angle in the circumferential direction of the field pole core occupied by the N poles and S poles of the magnets is taken as an electrical angle of 360°, and the winding coil pitch is set to 180°, the even-order induced voltage can be reduced, the motor torque is increased, while E1p/Epis increased and the same induced voltage peak is inhibited, and the cogging and ripples caused by the even-order induced voltage are reduced, which are the effects that could not be obtained in the conventional brushless motors.

Further, when a waveform in which a (2m+1)-th order harmonic (m is an integer equal to or greater than 1) is superimposed on the 1st order fundamental wave of the induced voltage under the predetermined phase difference condition and amplitude condition, such that the 1st order fundamental wave peak of the induced voltage is made larger than the induced voltage peak, is obtained for the induced voltage generated in the armature windings between the terminals of the motor by the rotation of the field poles of the motor, the torque motor is increased, while further increasing E1p/Epand inhibiting the same induced voltage peak, which are the effects that could not be obtained in the conventional brushless motors.

Further, when the 5th order harmonic component has a phase difference of 150° to 210° with the 1st order fundamental wave of the voltage in the case in which one period of the harmonic component is taken as 360°, and the ratio of the amplitude of the 5th order component to the amplitude of the 1st order fundamental wave of the voltage is made 2% to 12%, it is possible to increase the torque motor by comparison with the above-described case, while further increasing E1p/Epand inhibiting the same induced voltage peak, which are the effects that could not be obtained in the conventional brushless motors.

Further, when the 5th order harmonic component and the 7th order harmonic component have a phase difference of 120° to 240° with the 1st order fundamental wave of the voltage in the case in which one period of the harmonic component is taken as 360°, and the ratio of the sum of the amplitudes of the 5th order component and the 7th order harmonic component to the amplitude of the 1st order fundamental wave of the voltage is made 2% to 36%, it is possible to increase the torque motor by comparison with the above-described case, while further increasing E1p/Epand inhibiting the same induced voltage peak, which are the effects that could not be obtained in the conventional brushless motors.

Where the voltage V shown inFIG. 10is applied between the motor terminals as described in Embodiment 1, the electric current i flowing in each armature winding11of the motor has a waveform including the 5th order harmonic component and the 7th order harmonic component as shown inFIG. 11. Accordingly, when the motor is operated such that the interlinkage magnetic flux represented by Eq. (3) above is interlinked with the armature windings11of the motor in the phase relationship shown inFIG. 12, the torque waveform has the 6th order harmonic component of torque ripples relating to one period of the electrical angle as shown inFIG. 13.

Meanwhile, the magnetic flux φminterlinked with the armature windings11between the motor terminals by the rotation of the field poles of the brushless motor1in Embodiment 8 of the invention is represented by the following Eq. (12).

In Eq. (12), the magnetic flux φmis represented by the sum of the 1st order fundamental wave component, 5th order harmonic component, and 7th order harmonic component.

Further, φm1, φm5, and φm7denote the peaks of the waveforms of respective harmonics, and φm5and φm7denote the phase difference between the 1st order fundamental wave component and the 5th order harmonic component, and between the 1st order fundamental wave component and the 7th order harmonic component, respectively.

When the voltage V including the 5th order harmonic component, which is the applied voltage in the drive method of Embodiment 1 illustrated byFIG. 10, is applied to the brushless motor1having such magnetic flux φminterlinked with the armature windings11, the current i flowing in the armature windings11of the motor is determined by the voltage equation represented by Eq. (1) above and has a waveform including the 5th order harmonic component and 7th order harmonic component such as shown inFIG. 48.

Further, when the motor is operated such that the interlinkage magnetic flux represented by Eq. (12) above interlinks with the armature windings11of the motor with a phase relationship shown inFIG. 49, the torque waveform becomes such as shown inFIG. 50. In this case, the torque waveform shown inFIG. 13and described in Embodiment 1 is also shown at the same scale inFIG. 50. It follows fromFIG. 50that by including φm5and φm7in the magnetic fluxm, it is possible to reduce the 6th order harmonic component of torque ripples relating to one period of the electrical angle.

This can be explained as follows. In the torque waveform shown inFIG. 13, the magnetic flux φmhas only the 1st order fundamental wave component, and the current i includes the 5th and 7th order harmonic components at the amplitudes i5and i7, respectively. Therefore, the 6th order harmonic component T6fof torque ripples is generated in the torque, which is proportional to the magnetic flux φmand current i, whereas in the torque waveform shown inFIG. 50, the 5th and 7th order harmonic components are also present in the magnetic flux φmand, therefore, the 6th order harmonic component T6fis canceled.

More specifically, in the torque waveform shown inFIG. 13, the 6th order harmonic component T6fof torque ripples is represented by the following Eq. (13) for the torque, which is proportional to the magnetic flux φmand current i, whereas in the torque waveform shown inFIG. 50, the 6th order harmonic component T6fof torque ripples is represented by the following Eq. (14) for the torque, which is proportional to the magnetic flux φmand current i.

In Eqs. (13) and (14), θi5and θi7stand for a phase difference between the 5th and 7th order harmonic components and the 1st order fundamental wave component of electric current, and β stands for a phase difference between the 1st order fundamental wave component of the magnetic flux φmand the 1st order fundamental wave component of the electric current i. It follows from Eq. (14) that the 6th order harmonic component T6fof torque ripples of Eq. (13) is canceled in the third and fourth terms.

Thus, it can be said that in order to inhibit the 6th order harmonic component T6fof torque ripples of the motor, it is desirable that the magnetic flux φmbe provided such that reduces the 6th order harmonic component T6fof torque ripples of the motor. As for the method for providing the magnetic flux φm, it is desirable that adequate amounts of φm5/φm1and φm7/φm1and also θm5and θm7be provided.

FIG. 51shows an example of the brushless motor having the interlinkage magnetic flux φmbetween the terminals of the motor provided with the abovementioned adequate amounts of φm5/φm1and φm7/φm1and also θm5and θm7.FIG. 51is an expansion view showing, with linear expansion, one period of the electrical angle for a field pole core14and a permanent magnet13provided in the field pole core14of a motor of a surface magnet type in Embodiment 8 of the invention. A plurality of portions protruding and receding with respect to a reference surface are provided on the surface of the permanent magnet13.

FIG. 52shows the waveform of a spatial magnetic flux density Br in the radial direction in a gap between the field pole and armature in the magnet. As follows fromFIG. 52, the Br waveform can be given by the following Eq. (15).

In Eq. (15), Bb1, Bb5, and Bb7stand for peaks of the waveforms of each other related to the electrical angle, θbstands for an electrical angle phase, θ5band θb7stand for a phase difference between the 5th order harmonic component and the 1st order fundamental wave and between the 7th order harmonic component and the 1st order fundamental wave, respectively, and wt stands for the motor revolution. When such gap Br waveform is provided, where the leak magnetic flux is ignored, the magnetic flux represented by the following Eq. (16) may be found to be interlinked with the armature windings11between the terminals of the brushless motor1.

In Eq. (16), A stands for a proportional coefficient determined by the axial length or pole slot combination. In this case, the Br waveform of Eq. (15) is represented as a sum of triangular functions, and the order with respect to the timing t does not change despite the integration. Therefore, φmin Eq. (16) can be represented as a sum of the 5th order harmonic component and 7th order harmonic component represented by Eq. (12).

Therefore, where Bb5, Bb7and θ105, θb7of the Br waveform are determined, it is possible to determine the corresponding harmonic frequencies of φm. In addition, since the Br waveform between the field pole and the armature is proportional to the magnetomotive force of the permanent magnet13, Bb5, Bb7and θb5, θb7of the Br waveform in Eq. (15) can be determined by the level of a plurality of protrusions and depressions on the surface of the permanent magnet13shown inFIG. 51. It follows from the above that the motor structure shown inFIG. 51makes it possible to apply the adequate amounts of φm5/φm1and θm7/φm1and also θm5and θm7.

As indicated herein above, in Embodiment 8, a plurality of portions protruding and receding with respect to the reference surface is provided on the permanent magnet surface and a harmonic component is included in the gap magnetic flux density in the radial direction in the gap between the field poles and the armature. Therefore, a magnetic flux for which the phase difference and amplitude of the 5th order harmonic component and 7th order harmonic component of the magnetic flux related to the 1st order fundamental wave of the magnetic flux are set under the predetermined conditions such as to cancel the 6th order harmonic component of torque ripples that is generated by the interaction of the harmonic component of the voltage applied to the armature windings and the 5th order harmonic component and 7th order harmonic component of the magnetic flux interlinked with the armature windings is interlinked with the armature windings.

Therefore, since the magnetic flux φmis provided such as to reduce the 6th order harmonic component T6fof torque ripples of the motor, it is possible to inhibit the 6th order harmonic component T6fof torque ripples of the motor, and the motor vibrations and noise can be suppressed, while increasing the torque and output of the motor.

In Embodiment 8, the case is explained in which the 5th order harmonic component is included in the applied voltage, as in the above-described Embodiment 1, but such a feature is not limiting, and the same effect can be also obtained when the 7th order harmonic component is included in the applied voltage, as in the above-described Embodiment 2.

Further, in Embodiment 8, a plurality of portions protruding and receding with respect to the reference surface is provided on the surface of the permanent magnet13and a harmonic component is included in the gap magnetic flux density Br in the radial direction in the gap between the field pole and the armature in order to apply the adequate amounts of φm5/φm1and φm7/φm1and also θm5and θm7. However, such a configuration is not limiting, and the same effect can be also obtained when the magnetization direction or magnetization amount of the magnets is controlled, the motor permeance is changed by changing the shape of the stator and rotor, and a harmonic component is included in the gap magnetic flux density Br in the radial direction in the gap between the field poles and the armature. Further, the motor of a surface magnet type is described by way of example, but the same effect can be also obtained in a motor with embedded magnet, provided that a harmonic component is included in the gap magnetic flux density Br in the radial direction in the gap between the field poles and the armature.

Further, the same effect can be also obtained when the armature windings of all phases are Y connected as shown inFIG. 24, as in the above-described Embodiment 3.

The same effect can be also obtained and the 6th order harmonic component of torque ripples can be inhibited in the operation region in which the applied voltage V is equal to the maximum interphase voltage peak Vmaxthat can be outputted by the external AC voltage source, as in the above-described Embodiment 4.

Further, the same effect can be also obtained in the case in which a combination of one or more DC voltage sources and two or more frequency converters that convert a DC voltage into a frequency-variable AC voltage is used as the external AC voltage source, as in the above-described Embodiment 5.

In Embodiments 1 to 8, the brushless motor and the drive method for the brushless motor are explained, but an electric power steering device may be also configured by generating a torque assisting the steering torque with the brushless motor and the drive method for the brushless motor.

FIG. 53shows an electric power steering device in which an assist torque is generated with the brushless motor and the drive method for the brushless motor of Embodiment 9. The driver steers the front wheels by rotating a handle to the left or to the right.

In the configuration shown inFIG. 53, a torque detection means detects the steering torque of a steering system and outputs the detected torque to a control means. The control means computes a voltage command such that the motor generates a torque assisting the steering torque of the steering system according to the drive method for the brushless motor and outputs the computed voltage command to a voltage application means. The voltage application means applies a voltage to the motor on the basis of the voltage command, and the motor generates a torque assisting the steering torque through a gear.

In the electric power steering device provided with the brushless motor described in Embodiment 9, torque pulsations in the motor are reduced. Therefore, pulsations felt when the handle is steered can be reduced, the steering feeling of the driver can be improved, and noise during the steering can be reduced.

Further, the voltage application device can be designed for a reduced resistance to the induced voltage applied from the motor. Therefore, the voltage application device can be reduced in size and weight and the electric power steering deice can be also reduced in size and weight.

Further, since the output of the motor drive device can be increased and the amount of magnet per unit torque can be reduced, the electric power steering deice can be reduced in size and weight, and the rated torque necessary during the near-end steering can be increased.

Explanation of Reference Numerals