Ghost canceller

A ghost canceller which operates at IF and utilizes a storage mode heterojunction acoustic charge transport device (SM-HACT). The signal delay provided by the SM-HACT is increased by the operation of barrier electrodes which delay the movement of charge packets across the device, thereby eliminating the need for additional digital equalization. The tap weights of the SM-HACT are determined by the operation of a fixed correlator which responds to ghosts in a predetermined training waveform.

BACKGROUND OF THE INVENTION 
This invention relates generally to the field of waveform equalization, and 
more specifically to the field of ghost cancelers for television, and more 
specifically to a storage-mode acoustic charge transport device used as a 
ghost canceller for television. 
Ghosts in television images are caused by multi-path transmission channels. 
In general, they are time-delayed, attenuated, and distorted versions of 
the intended signal. They also can be time-varying when caused by 
reflections from moving objects, for example airplane flutter. In 
conventional NTSC or analog HDTV they are an irritating nuisance which can 
seriously impair picture quality and visibility. In the case of 
terrestrial broadcast of analog HDTV signals, even small amounts of 
multipath can reduce the picture quality to a level which is no better 
than present day NTSC. 
In digital HDTV broadcasts the situation is even more serious, since 
digital schemes do not degrade gracefully under signal impairment as do 
analog transmission systems. This is especially true of the digital HDTV 
broadcast systems which utilize bandwidth compression techniques such as 
Huffman coding and Run-Length Limiting. Such techniques often transmit 
symbols of varying lengths which represent various signal levels. Signal 
levels which repeat often are represented by symbols which are only a few 
bits long, whereas signal levels which occur rarely are represented by 
longer symbols. In order to decipher a stream of variable length symbols, 
the receiver decoder must not make any mistake or it can become completely 
lost. 
The ghost problem is forcing HDTV proponents to develop adaptive 
equalizers, or ghost cancelers, for their systems. They are developing 
digital equalizers because no satisfactory analog equalizer technology has 
existed until now. The basic function of an equalizer, to remove the 
effects of multipath and perhaps other distortions, can be understood in 
either the time or frequency domain. In the time domain, it may be shown 
that the equalizer cancels out the distortion components from the received 
signal by creating a copy of just the distortion components, and then 
subtracting this from the incoming signal. In the frequency domain, the 
undesired signal components cause the frequency response of the 
transmission channel to be distorted, usually having periodic ripples in 
the channel's frequency response. The function of an equalizer may be 
viewed in the frequency domain as an inverse filter, having ripples in its 
frequency response which are exactly opposite to those of the distorted 
transmission channel. When this equalizer is used in the receiver, the 
product of the channel's frequency response and that of the equalizer will 
be flat. The time domain and frequency domain explanations of an equalizer 
are entirely equivalent. 
A key component of an equalizer is a Programmable Transversal Filter or 
PTF. Required characteristics of a PTF include the following: 
its bandwidth must be wide enough to handle the signals of interest; 
it must contain enough independently programmable taps, spaced close enough 
to create an accurate replica of the distortion (ghost) signal; 
its tap weight magnitudes must have sufficient resolution to create an 
accurate replica of the ghost signals; and 
it must be sufficiently long to recreate the expected delays of the 
distortion signals. 
There has been an appreciable effort over the past decade to develop ghost 
cancelers for NTSC television. Most of the approaches have been done at 
baseband, after the video detector. Baseband operation in itself creates 
problems. Envelope detection cannot be used, because it freezes into the 
detected signal artifacts which are caused by interaction between the 
desired signal and the ghost. These artifacts cannot be removed 
subsequently by a linear operation. Synchronous detection must therefore 
be used if baseband ghost cancellation is to be employed. However, carrier 
phase recovery is disturbed by the presence of ghosts. A narrowband 
synchronous detector will detect on the average phase of the received 
signal, which depends on the relative amplitude and carrier phase of any 
ghost signals present. As a result, quadrature components will appear at 
the output of the synchronous detector for all but a few trivial cases. 
Multiple transversal filters must be used to handle in-phase and 
quadrature components. In addition, cross terms in ghosted baseband video 
require four transversal filters in a lattice configuration to completely 
cancel the interfering signals. 
The BTA (Japanese equivalent of the U.S. National Association of 
Broadcasters) has proposed an NTSC ghost canceler based on baseband 
digital signal processing technology. However, digital transversal 
equalizers are expensive and slow. A digitally programmable analog 
transversal equalizer technology is needed, with satisfactory performance 
and affordable cost. 
SUMMARY OF THE INVENTION 
In light of the performance limitations of the existing technology, it is 
an object of this invention to provide a ghost canceller which operates at 
RF or IF frequency. It is a further object of this invention to provide a 
ghost canceller which provides high performance, fast convergence, and 
lower cost when compared to baseband digital approaches. 
A ghost canceller is described herein which has a signal input means, a 
filter means, a summing means, and a signal output means. The output of 
the signal input means and the output of the filter means are inputs to 
the summing means, and the output of the summing means is the input to the 
filter means and the output means. The filter means contains a storage 
mode heterojunction acoustic charge transport (SM-HACT) device which is 
formed with a substrate, a channel means on the substrate, a charge 
injecting means which injects charge packets into the channel means in 
response to the output of the summing means, an acoustic wave means for 
injecting an acoustic wave into the channel means in order to move the 
charge packets along the channel means, a plurality of barrier electrode 
on the channel means which operate when energized to stop the movement of 
the charge packets, a plurality of nondestructive sense electrodes 
interspersed among the barrier electrodes on the channel means and 
operable to sense the charge packets when they are stopped, and a 
differential output circuit having as input the output of the 
nondestructive sense electrodes, where the output of the differential 
output circuit is the output of the filter means. Because the barrier 
electrodes Serve to extend the signal delay available in the SM-HACT 
device, this device can be used as a digitally programmable transversal 
filter in the RF or IF section of a receiver, thereby eliminating the need 
for the more complex digital equalization after A/D conversion in the 
receiver.

DESCRIPTION OF THE PREFERRED EMBODIMENTS 
This invention involves a novel Storage-Mode Heterojunction Acoustic Charge 
Transport (SM-HACT) Digitally Programmable Transversal Filter (DPTF) as an 
equalizer for multi-path signal situations such as radar, communications, 
electronic warfare and television applications, to be used in the RF or IF 
section of the receiver before A/D conversion of the signal. FIG. 1 shows 
a simplified diagram of such a SM-HACT DPTF device. A semiconductor 
substrate 10 has formed on it a channel 11 for conducting mobile charge 
packets 12. The substrate 10, channel 11, and overlaying layer 13 may be 
gallium arsenide, formed with interspersed layers of aluminum gallium 
arsenide 14,15 by techniques well known in the art. Acoustic carrier waves 
16 are generated in the substrate 10 and channel 11 by an RF voltage 
applied to the interdigital Surface Acoustic Wave (SAW) transducer 17. An 
input contact 18 is positioned over the channel 11 as a means for 
injecting a charge into the device. The acoustic waves 16 extract packets 
of electrons 12 from the input contact 18. A signal voltage applied to the 
input contact 18 essentially modulates the amount of charge in each packet 
12 according to the value of the signal voltage at the precise instant 
that the charge packet 12 is formed. Nondestructive sensing (NDS) 
electrodes 19,20 capacitively sense the charge packets 12 as they travel 
through the channel 11. The weight of the signal generated by each NDS 
electrode 19,20 may be determined by the physical length of the electrode 
itself, or the weight may be determined by tap weighting circuitry 21,22, 
which may be for example a programmable digitally controlled multiplying 
D/A converter. The output of the tap weighing circuits 21,22 may be 
connected to the plus and minus inputs of a differential device such as a 
differential output amplifier 23 to form the transversal filter output. 
The device of FIG. 1 operates on RF analog signals over a wide frequency 
range which is determined by the tap spacing and tap weights. Although the 
tap weights may be limited in precision by the tap weighting circuitry, 
the actual multiplication operation at each tap is an analog process, so 
no roundoff error (roundoff noise) occurs as it does in digital filters. 
Also, many more taps can be included on a single HACT DPTF chip than on 
digital filter chips. 
The delay provided by the device of FIG. 1 operated in its standard mode 
could be about 2--3 microseconds. With this amount of delay, additional 
digital equalization after A/D conversion would be required. A preferred 
device would provide about 20 microseconds delay. To accomplish this 
amount of delay, barrier electrodes 24,25 are formed above the channel. 
These barrier electrodes 24,25, preferably interspensed among the NDS 
electrodes 19,20, are pulsed periodically to cause the charge packets 12 
to progress through the device in a series of hops, thereby increasing the 
total delay provided by the device. This makes the device appear 
electrically long, even though it is not physically long. 
In the storage mode of operation, the barrier electrodes 24,25 are driven 
by the waveform of FIG. 2. Most of the time, the voltage on each barrier 
electrode 24,25 is sufficiently negative that each charge packet 12 is 
trapped between two barrier electrodes 24,25, unable to move forward under 
the influence of the acoustic wave 16. Periodically, the voltage is 
reduced briefly, enabling the charge packets 12 to move again. Because the 
charge packets 12 are very close to the top surface in the HACT, the 
amplitude of the barrier waveform can be less than a volt. In one 
embodiment, the barrier voltage is reduced just long enough for a packet 
to move from one storage area to the next, but no farther. The voltage 
applied to the barrier electrodes 24,25 is then increased to stop the 
charge packets 12 from moving again, and the cycle repeats. In this cycle, 
the duration of the part of the barrier waveform corresponding to storage 
time is relatively uncritical, and can be readily increased or decreased. 
The duration of the part of the barrier waveform corresponding to motion 
of the packets is more critical, and the leading and trailing edges of the 
pulse are preferably synchronized with the acoustic wave. It should be 
noted that the device of FIG. 1 can be operated in two modes: if the 
barrier electrodes 24,25 are not pulsed, it can appear approximately 3 
microseconds long; if the barrier electrodes 24,25 are pulsed at the 
proper rate the device can appear much longer, for example 20 microseconds 
long. 
The effect of the barrier electrode action on the signal is sort of an 
"undersample and hold" operation. The repetition rate of the barrier 
pulses sets the sampling period T.sub.samp of the device, which can be an 
integer multiple of the period T.sub.SAW of the acoustic carrier wave. 
(T.sub.SAW would otherwise be the sampling interval, if the barrier 
electrodes were not pulsed.) During each sample interval, the charge 
packets are stopped and held stationary by the barrier electrodes for a 
time T.sub.hold, during which the output is constant. Note that the charge 
packet 12 is being sensed while it is being held. FIGS. 3A and 3B show a 
computer simulation of the effect on a TV IF signal of a storage-mode HACT 
having one tap fully on (tap weight .alpha..sub.1 =1.0) and all other tap 
weights set to zero. In this example, .function.saw=91.5 MHz, which is 
twice the 45.75 MHz standard TV IF picture carrier frequency; T.sub.hold 
=7T.sub.SAW, and T.sub.samp =8T.sub.SAW. The input signal shown in FIG. 3A 
is a 43 MHz RF burst, with a -20 dB ghost separated by 10 .mu.sec. The 
output signal of FIG. 3B shows the effect of the. undersampling operation 
performed by the SM-HACT. 
The corresponding spectra are shown in FIG. 4. The spectrum F(.omega.) of 
the input pulse is bandlimited to the TV IF region which is between 39.75 
MHz and 47.25 MHz. The undersampling of the SM-HACT produces aliased 
copies of the input spectrum, at lower and higher frequencies. For a 
Storage-Mode HACT with N multiple taps of weights .alpha..sub.k, the 
output spectrum is 
##EQU1## 
The sin (.chi.)/.chi. rolloff produces less special energy at the IF band 
than at very low frequencies. If desired, a simple switching circuit may 
be added to the output of the SM-HACT device to narrow the width of each 
output pulse thereby increasing the spectral energy available at higher 
frequencies. 
The SM-HACT can be used in either a feedforward or feedback configuration. 
Because the feedback configuration can in principle cancel the ghosts 
completely, it is the preferred approach. FIG. 5 shows a simplified block 
diagram of a feedback canceller loop employing a Storage-Mode HACT 50. The 
output of the SM-HACT 58 may be bandpass filtered 52 before the summing 
node 54, to remove spectral aliases outside of the TV IF band. This 
assures that the input 51 and output 55 signals of the loop are only IF 
frequencies, so that the ghost canceller loop can be readily inserted into 
standard TV IF sections, with no special design accommodations required. 
Many other variations of the basic loop are possible. For example, it 
should be possible to undersample the input waveform and low-pass filter 
the SM-HACT output, thereby producing a cancellation loop output at low 
frequencies (e.g. 2.89 MHz). This might be used to eliminate the 
synchronous video detector after the loop, allowing direct A/D conversion 
of the output of the ghost canceller loop. 
Any equalizer system requires a training waveform, and some sort of system 
to recognize the echoes of this training signal and set the appropriate 
tap weights in the canceller. Because broadcast HDTV is currently in the 
development stages, such a training signal will likely be standardized at 
the same time as the rest of the signal format. One such signal may be a 
spread spectrum or pulse compression waveform such as a pseudo-noise 
sequence, shown as item 60 in FIG. 6. A matched correlator 62 in the 
receiver, which can be either fixed or programmable, gives a sequence of 
correlation pulses, whose amplitudes and times correspond to the multiple 
paths of the channel, and which are equal to the tap weights to be 
programmed into the equalizer. If no ghosts exist, a single correlation 
spike is produced, as shown by curve 64. Multiple ghosts produce multiple 
correlation peaks 66. This approach has good noise performance; the ghost 
measurement dynamic range is improved by the time-bandwidth product of the 
pulse compression waveform. The matched filter has higher peak power than 
the rest of the HDTV signal, so it can trigger the ghost measurement 
sequence independently of any HDTV decoding or synchronization process. 
This means that even if the ghosting situation is so severe that the TV 
cannot synchronize for operation, the ghost environment can still be 
measured and eliminated. These factors allow a system design which can 
update the equalizer very rapidly. 
FIG. 7 illustrates how this concept can be used in the IF of a proposed 
HDTV approach, using a Storage-Mode HACT equalizer. A tuner 70 provides an 
input signal to an IF section consisting of band pass filter 72, carrier 
recovery circuit 74, and mixer 76. Filter 72 is typically a SAW filter 
which provides the main selectivity of the receiver. The combination of 
carrier recovery circuit 74 and mixer 76 comprise a standard synchronous 
detection scheme. The base band output signal of the mixer 76 provides an 
input signal to A/D converter 78 for processing by digital processor 80. 
The ghost canceller of this invention is inserted between the filter 72 
and the mixer 76. A matched correlator 82 looks at the corrected IF signal 
from downstream of a summation device 84. When the set is first turned on 
or the channel is changed, the tap weights of the SM-HACT 86 device are 
set to zero. In this configuration, the feedback loop is open and the 
corrected IF signal is initially the same as the uncorrected signal. The 
output of the correlator 82 is digitized by an A/D converter 88 to provide 
the coefficients for the SM-HACT 86, with the main correlation peak 
detected by a peak detector 90, triggering the A/D converter 88 which 
feeds a tap weight accumulator 92. The output of the SM-HACT 86 may be 
passed through a band pass filter 94 prior to the summation device 84. The 
first TV frame sets the tap weights very close to the required values, 
with subsequent frames making small corrections as needed. This allows for 
very fast equalizer convergence as well as very accurate cancellation 
after a number of frames. 
FIGS. 8A-8E show time domain waveforms or spectra at various points in the 
system of FIG. 7, as calculated by computer simulation. For this 
simulation, a 60 bit pseudo-noise training waveform was created, along 
with a -20 dB ghost at a time offset of 10 microseconds. This transmit 
waveform was band limited by a cosine-cubed filter, to assure that its 
spectrum fits completely within the allotted 6 MHz TV transmission 
bandwidth. The input IF waveform and its spectrum are shown in FIGS. 8A 
and 8B respectively. The training waveform was designed to be correlated 
with a fixed 13-bit Barker coded biphase correlator (item 73 of FIG. 7), 
to produce a main correlation pulse with very low time sidelobes, as shown 
in FIG. 8C. The A/D sampling starts after the main correlation pulse and 
does not include the main correlation pulse. After downconversion to 2.859 
MHz, which is the center frequency of the lowest alias passband of the 
SM-HACT spectrum that was shown in FIG. 4, the correlation pulse is A/D 
converted to 8 bits of precision at a rate of 45.75 MHz/4=11.4375 
MHz=4.times.2.859 MHz. A peak detector 90 triggers the loading of the 
samples into the register of the tap weight accumulator 92, and from there 
into the tap weight memory of the SM-HACT 86. FIG. 8D shows the frequency 
response of the SM-HACT as programmed with these tap weights. The output 
of the cancellation loop, shown in FIG. 8E still contains a small residual 
ghost that is not completely canceled. This waveform would be applied to 
the correlator 82, creating a large main correlation peak and a smaller 
residual peak corresponding to the residual ghost. The main correlation 
peak would trigger the A/D process again before the second frame. The A/D 
samples from the residual ghost would comprise small corrections that are 
then added to the contents of the tap weight accumulator 92 which 
functions to accumulate tap weight signals generated from successive 
repetition of the training waveform. The contents of this accumulator 92 
would then be loaded into the tap weights of the SM-HACT 86, slightly 
correcting the tap weights which were loaded from the previous training 
pulse. The modified tap weights provide improved ghost cancellation for 
the second frame. Before the third frame, a new ghosted training pulse 
arrives, and the process repeats. 
Because a fixed signal format standard already exists for present-day NTSC 
color TV, the choice of a training signal becomes somewhat restricted, 
depending on the direction that the TV industry takes. Two options exist: 
transmit a special training signal which is approved by the FCC, or use 
the existing TV signal to derive equalizer programming information. The 
simplest situation would be to derive information from the existing signal 
to program the equalizer, because it would not require any FCC mandates. 
Several parts of the video signal, such as the serration pulse edge in the 
center of line 266, the chroma burst, or the horizontal sync pulses can be 
used. Of these, the line 266 edge appears to be the most promising. The 
practical problem with this approach is that all candidates for a training 
pulse are part of the synchronization signal. When a signal is received by 
a local TV station as a network feed, the local station strips off the 
network's sync pulses, and substitutes its own. Accordingly, the ghost 
canceller can only get rid of multipath distortions which are introduced 
in the local broadcast process. Any multipath distortions which are 
introduced in the signal during the network feed cannot be equalized out. 
This consideration may not be very serious under most situations, but it 
is not perfect. 
A better technical solution is for the originator of the signal to 
broadcast a carefully designed video training signal, standardized by the 
FCC, which local broadcasters would leave alone. This would allow an 
adaptive equalizer to clean up multipath distortions introduced anywhere 
in the broadcast chain, including network feeds. Such a training signal 
could be used in a manner similar to the HDTV version discussed in the 
preceding paragraph, provided that a correlator was used which was matched 
to the training signal. 
The above described embodiments are meant as illustration only and should 
not be construed to limit the scope of the invention as claimed below.