Adaptive peak power control

A switching power converter includes a controller configured to transition from a first operating mode to a second operating mode by determining the operating conditions at the transition point between the operation modes. The controller uses the value of the voltage-time product determined at the boundary between the first and second operating modes to predict the voltage to be applied to the primary-side of the transformer. Using the predicted voltage, the controller can adjust the peak-power control threshold on a cycle-by-cycle basis without the transformer reaching saturation.

BACKGROUND

1. Technical Field

The present disclosure relates to controlling a switching power converter to adaptively adjust the output voltage of the power converter during peak power conditions.

2. Description of the Related Art

Power supply demands for electronic devices such as printers, audio devices, and network adapters often vary under different operating conditions. For example, the output power demands of a power converter for a printer may fluctuate in a range of 100% to 400% of the normal output power level. These fluctuations, however, may occur during a relatively short period of time compared to the overall operating period of the power converter. One solution to accommodate these brief power demand fluctuations includes adding design capacity to the power converter to accommodate the additional power supply capacity demands under all operating conditions. Other solutions include increasing the switching frequency of the power converter.

These solutions, like many other solutions for addressing fluctuating power supply demands for electronic devices have drawbacks. Designing for increased capacity often results in a larger and more expensive power converter. Increasing the switching frequency of the power converter increases the amount of undesirable electromagnetic and radio frequency interference (EMI and RFI) generated by the converter. Further, increasing the switching frequency reduces the overall efficiency of the power converter and also results in an undesirable increase in the heat generated by the power converter.

SUMMARY

Embodiments include a power regulation approach for transitioning from a first operating mode to a second operating mode by determining the operating conditions at the transition point between the operation modes. The controller uses the value of the voltage-time product determined at the boundary between the first operating mode and the second operating mode to predict the voltage to be applied to the primary-side of the transformer during a subsequent switching cycle. Using the predicted voltage, the controller can adjust the peak-power control threshold on a cycle-by-cycle basis without the transformer reaching saturation.

The features and advantages described in the specification are not all inclusive and, in particular, many additional features and advantages will be apparent to one of ordinary skill in the art in view of the drawings and specification. Moreover, it should be noted that the language used in the specification has been principally selected for readability and instructional purposes, and may not have been selected to delineate or circumscribe the inventive subject matter.

DETAILED DESCRIPTION OF EMBODIMENTS

Example Switching Power Converter Circuit

FIG. 1is a circuit diagram illustrating a switching power converter100, according to one embodiment. The switching power converter100is a primary-side feedback flyback converter, and includes three principal sections, i.e., a front end104, power stage, and a secondary stage. The front end104is connected to an AC voltage source (not shown) at nodes L, N, and includes a bridge rectifier comprised of inductor L1, resistors R1, F1, diodes D1, D2, D3, D4, and capacitor C2. The rectified input line voltage at node105is input to the supply voltage pin Vcc (pin1) of the controller IC102via resistors R10and R11. The line voltage at node105is also connected to the primary winding106of the power transformer T1-A. The capacitor C5removes high frequency noise from the rectified line voltage. The output of the front end section at node105is an unregulated DC input voltage.

The power stage includes the power transformer T1-A, the switch111, and the controller IC102. The power transformer T1-A includes a primary winding106, a secondary winding107, and an auxiliary winding108. The controller IC102maintains output regulation via control of the ON and OFF states of the switch111. The ON and OFF states of the switch111are controlled via a control signal110output from the OUTPUT pin (pin5) of the controller IC102. The control signal110drives the base (B) of the switch111. The collector (C) of the switch111is connected to the primary winding106, while the emitter (E) of the switch111is connected to ISENSEpin (pin4) of the controller IC102and to ground via the resistor R12. The ISENSEpin senses the current through the primary winding106and the switch111in the form of a voltage across sense the resistor R12. The controller IC102employs the modulation technique (i.e., operation modes) as described below in detail with reference toFIG. 3to control the ON and OFF states of the switch111, the duty cycles of the switch111, and the amplitude of the switch111base current in accordance with varying load conditions at node109. The GND pin (pin2) of the controller IC102is connected to ground. While a BJT switch Q1is used as the switch111in the embodiment ofFIG. 1, a power MOSFET may also be used as the switch111for the switching power converter100according to other embodiments herein.

The secondary stage includes a diode D6functioning as an output rectifier and a capacitor C10functioning as an output filter. The resulting regulated output voltage Vout at node109is delivered to the load (not shown) and a pre-load resistor R14. The pre-load resistor R14stabilizes the output of the power converter at no load conditions. Also, an ESD (Electrostatic Discharge) gap (ESDI) is coupled between the primary winding106and the diode D6.

The output voltage Vout at node109is reflected across the auxiliary winding108, which is input to the VSENSEpin (pin3) of the controller IC102via a resistive voltage divider comprised of resistors R3and R4. Also, although the controller IC102is powered up by the line voltage105at start-up, the controller IC102is powered up by the voltage across the auxiliary winding108after start-up and in normal operation. Thus, diode D5and resistor R2form a rectifier for rectifying the voltage across the auxiliary winding108for use as the supply voltage input to the VCCpin (pin1) of the controller IC102after start-up during normal operation. The capacitor C9is used to hold power from the line voltage at node105at start-up or from the voltage across the auxiliary winding108after start-up between switching cycles.

FIG. 2Ais a diagram illustrating the pin-outs of the controller IC102of the switching power converter100, according to one embodiment. The controller IC102is a 5-pin IC. Pin1(Vcc) is a power input pin for receiving the supply voltage, pin2(Gnd) is a ground pin, pin3(VSENSE) is an analog input pin configured to receive the voltage across the auxiliary winding108of the switching power converter100for primary-side regulation of the output voltage Vout, and pin4(ISENSE) is an analog input pin configured to sense the primary-side current of the flyback switching power converter in the form of an analog voltage, for cycle-by-cycle peak current control and limit. Pin5(Output) is an output pin outputting the control signal110for controlling the on-times and off-times of the switch111as well as the amplitude of the base current or gate current applied to the switch111.

FIG. 2Bis another view of the controller IC102illustrating the internal circuitry of the controller IC102of the switching power converter100. The controller IC102receives analog parameters such as the VSENSEvoltage at pin3and the ISENSEvoltage at pin4, but adaptively processes these parameters using digital circuitry and digital state machines to generate the appropriate control signal at110at pin5(Output). The controller IC102includes several main circuit blocks, including signal conditioning block202, digital logic control204, digital-to-analog converter (DAC)206, Ipeak comparator212, and over-current protection (OCP) comparator210. The controller IC102regulates the output voltage Vout and output current Iout of the switching power converter100by means of adaptive digital, primary-side feedback control. Sensing the primary-side current at the ISENSEpin (pin4) allows cycle-by-cycle peak current control and limit in both CV (Constant Voltage) and CC (Constant Current) modes as well as precise constant current (output current Iout) control that is insensitive to the magnetizing inductance Lm of the transformer T1-A. In particular, the controller IC102uses the sensed primary-side current to predict on a cycle-by-cycle basis a voltage to be applied to the primary-side winding to adjust the peak-power control threshold while the switching power converter is operating under peak power control mode. Sensing the output voltage Vout reflected across the auxiliary winding108at the VSENSEpin (pin3) allows for precise output voltage regulation.

The digital-to-analog converter (DAC)206is configured to receive a digital signal derived from the sensed primary-side switching current and output an analog voltage used as a reference voltage associated with a voltage corresponding to the peak primary-side switching current threshold. The Ipeak comparator212compares the peak primary-side switching current threshold with a voltage signal corresponding to the primary-side switching current sensed at ISENSEpin (pin4) to generate a feedback signal used for peak power control operation in CMM mode. The over-current protection (OCP) comparator210controls the maximum peak power. The OCP comparator210compares the voltage signal corresponding to the primary-side switching current sensed at ISENSEpin (pin4) with a OCP reference voltage threshold. The OCP comparator210generates a logic high signal when the voltage signal corresponding to the primary-side switching current sensed at ISENSEpin (pin4) is higher than the OCP reference voltage threshold.

The controller IC102causes switching power converter100to operate in one of multiple operation modes depending on load conditions, as described below in detail with reference toFIG. 3. According to one embodiment as shown inFIG. 3, the controller IC102employs pulse frequency modulation (PFM), discontinuous conduction mode (DCM), critical discontinuous conduction mode (CDCM), and continuous conduction mode (CCM). In another embodiment, under CCM operation, the controller IC102further employs a peak power mode control. Under the peak power mode control, the controller IC102uses the value of the voltage-time product determined at the boundary of the CDCM and CCM operation to predict the voltage to be applied to the primary-side of the transformer to adjust the peak-power control threshold on a cycle-by-cycle basis. The voltage-time product represents the maximum amount of time that a transformer can have a constant voltage applied to it without the transformer reaching saturation.

Returning toFIG. 2B, the signal conditioning block202receives the VSENSEvoltage and generates voltage and current feedback parameters for use by digital logic control block204. In one example, generated voltage feedback parameters include the feedback voltage value VFBthat represents the value of the VSENSEvoltage sampled at the end of the transformer reset time in each switching cycle. In some cases, the feedback parameter VFBis scaled to a comparable level for comparison with a reference voltage VREF. In one example, VREFis a digital value that represents the target regulated output voltage (e.g., 5V) of the switching power converter100, scaled to a lower value (e.g., 1.538V) according to the turns ratio between the secondary winding107and the auxiliary winding108and the resistive voltage divider (R3/(R3+R4)). Thus, the specific value of the reference voltage VREFis determined according to the target regulated output voltage of the switching power converter100. The generated current feedback signals include, among others, secondary current timing information such as Tp (switching period) and Trst (transformer reset time) output to the digital logic control block204. The voltage feedback values and current feedback values, including VFB, Tp, and Trst may be determined using one of a variety of conventional digital, analog, or combination of digital and analog techniques.

FIG. 3is a graph illustrating the operation modes of a switching power converter100ofFIG. 1, according to one embodiment. For example, as shown inFIG. 3, the controller IC102operates the switching power converter100in PFM operation under low load conditions between corresponding output current levels I1and I2(between operating points A and B). In this example, low load condition occurs when the voltage sensed by the VSENSEpin (pin3) of the controller IC102satisfies a PFM threshold voltage VTH_PFM.

In one embodiment, the threshold voltages used by the controller IC102to determine when to transition to an operation mode are determined based on a percentage of regulation a voltage VREGderived from VSENSE. For example, the PFM threshold voltage VTH_PFMmay be set to a value between 0% and 10% of VREG, and the threshold for PFM to DCM could be set to a value between 10% and 50% of VREG. Other thresholds may also be used without departing from the scope of the disclosure such that the threshold voltage from PFM to DCM is less than the threshold voltage from DCM to CDCM.

The PFM threshold voltage VTH_PFMoccurs when the peak primary-side current is less than the value of VTH_PFMdivided by the value of the sense resistor R12. In PFM operation mode, the duty cycle of the switch111is varied by keeping the conduction pulse width of the control signal110constant, while varying the switching period and thus the switching frequency. For example, in PFM operation, a switch may be turned on for 5 μs of each switching period, but the switching frequency may be varied between 40 kHz (FSW1) and 130 kHz (FSW2). A switching frequency of 40 kHz corresponds to a switching period of 25 μs, and therefore, the duty cycle at this switching frequency is 20% (=5 μs/25 μs). For a switching frequency of 130 kHz, the switching period is 7.7 μs, and therefore, the duty cycle at 130 kHz is 65% (=5 μs/7.7 μs).

In PFM operation, the controller IC102regulates the output voltage by varying the switching period of the control signal110, while keeping the pulse conduction width (TON) of the control signal110constant. As the regulation decreases (i.e., the load decrease), the controller IC102increases the switching period to reduce the output voltage. Increasing the switching period of the switch111causes less energy to be transferred per unit of time (i.e., reduces the duty cycle) to the primary winding106of the power transformer T1-A, which in turn decreases the output voltage of the switching converter100. Conversely, as the regulation increases (i.e., the load increases), the controller IC102decreases the switching period to increase the output voltage of the switching converter100. Decreasing the switching period of the switch111transfers more energy per unit of time to the primary winding106of the power transformer T1-A (i.e., increases the duty cycle), which in turn increases the output voltage of the switching converter power100.

In one embodiment, the controller IC102may employ a regulation scheme for transitioning from a first operating mode to a second operating mode by determining the operating conditions at the transition point using a point where the switch111would have been turned on if operating under the first operating mode as a reference point to determine when to turn on the switch under the second operating mode. In particular, the controller IC102uses the reference point to determine a control period for regulating the switching period of the switch111in a second operating mode as described in U.S. patent application Ser. No. 13/772,202, filed Feb. 20, 2013, which is incorporated by reference in its entirety.

The controller IC102operates the switching power converter100in DCM operation under low load conditions between corresponding output current levels I2and13(between operating points B and C). In this example, low load condition occurs when the voltage sensed by the VSENSEpin (pin3) of the controller IC102is greater than the PFM threshold voltage VTH_PFMbut less than the DCM threshold voltage VTH_DCM. In DCM operation, the controller IC102controls switching of the switch111such that the magnetizing current stored in the power transformer T1-A drops to zero between switching cycles, and the current flowing through the diode D6has completely dropped to zero before the switch111is turned ON.

In DCM operation, the controller IC102may regulate the switching of the switch111using pulse width modulation (PWM) operation mode. In PWM the duty cycle of the switch111is varied by adjusting how long the switch111remains ON (i.e., the conduction pulse width) during each switching period, i.e., using the on-time of the switch111as the control variable, while the switching period remains constant. For example, in PWM operation mode, the switch111may be turned ON at a switching frequency of 100 kHz (and therefore has a switching period of 10 μs). For a duty cycle of 30%, the switch111is controlled to be ON for 3 μs and OFF for 7 μs of each switching period.

In PWM operation mode, controller IC102regulates the output voltage of the switching power converter100by varying the conduction pulse width of control signal110(and thus the on-time of the switch), while keeping the switching period (equal to the inverse of the switching frequency). As the regulation decreases (i.e., the load decrease), the controller IC102reduces the conduction pulse width from to reduce the output voltage. Decreasing the conduction pulse width applied to the switch111causes less energy to be transferred per switching cycle to the primary winding106of power transformer T1-A, which in turn decreases the output voltage of the switching power converter100. Conversely, as the regulation increases (i.e., the load increases), the controller IC102increases the conduction pulse width applied to the switch111to increase the output voltage of the switching power converter100. Increasing the conduction pulse width of the switch111transfers more energy per cycle to the primary winding106of power transformer T1-A, which in turn increases the output voltage of the switching power converter100.

The controller IC102operates the switching power converter100in CDCM operation under load conditions between corresponding output current levels13and14(between operating points C and D). In this example, the controller IC102transitions from DCM operation to CDCM operation when the controller IC102detects a zero voltage switching condition and the voltage sensed by the VSENSEpin (pin3) of the controller IC102is greater than the DCM threshold voltage VTH_DCMbut greater than the reference voltage VTH_PK, which corresponds to the peak primary-side switching current. The zero voltage condition occurs when the voltage across the switch111reaches a local minimum value referred to as a valley. In one embodiment, the controller IC102may include a valley detection circuit to detect features of VSENSEor other feedback signals. In operation, the controller IC102operates in real time to detect and predict the occurrence of valleys on a switching-cycle by switching-cycle basis as described in U.S. patent application Ser. No. 12/642,261, which is incorporated by reference in its entirety. In one embodiment, the valley detection circuit is included in the signal conditional circuit202and provides information about the detected features of VSENSEto the digital logic control block204for further processing.

In CDCM operation, the controller IC102controls the switching of the switch111, where when the switch111is turned ON is determined by zero voltage switching (ZVS), and when the switch111is turned OFF is determined by peak current switching. Peak current switching means that the switch111will be kept ON until the controller IC102detects a peak primary current based on VSENSE. Zero voltage switching means that the switch111will be kept OFF until the reflected voltage detected at VSENSEpin (pin3) falls to zero, at which point the switch111is turned ON.

For example, in CDCM operation, the controller IC102may employ a valley mode switching (VMS) operation mode, where the duty cycle of the switch111is varied by adjusting both the conduction pulse width and switching period of the control signal110. Specifically, in VMS operation mode, the conduction pulse width is varied in accordance with PWM, PFM, or other suitable converter control schemes (i.e., operation modes). The switching period is not predetermined, but instead varies, because the switch is turned on at a valley (local minimum) of VSENSEsignal that occurs immediately subsequent or otherwise subsequent to the desired switch turn on time as calculated by the employed operation mode (PWM or PFM). As previously discussed in conjunction withFIG. 2B, the signal VSENSErepresents the output voltage Vout at node109as reflected across the auxiliary winding108. Accordingly, the VMS operation mode and associated techniques described herein can benefit the switching power converter100that uses any control scheme, regardless of whether PWM or PFM or some other operation mode is used.

At the boundary between the CDCM operation and CCM operation, the controller IC102determines the voltage-time product of the transformer prior to transitioning to the CCM operation, and uses the determined voltage-time product as a reference point to determine a turn-on time of the switch111during the CCM operation. As previously described, the voltage-time product represents the maximum amount of time that a transformer can have a constant voltage applied to it without the transformer reaching saturation. At the boundary between CDCM and CCM operation corresponding output current level14and operation point D, the controller IC102is operating in peak-power control mode and the voltage-time product is a constant that depends on the value of magnetizing inductance Lm of the transformer T1-A and the value of sense resistor R12. That is, the value of the constant associated with the voltage-time product at the boundary between CDCM and CCM operation is user specified resulting from the selection of components. When transitioning from CDCM to CCM operation, at operating point D, the controller IC102registers or stores the value of the voltage-time product at the operating point D, which serves as a reference point to calculate, during a present switching cycle, a voltage level applied to the transformer during subsequent switching cycle under CCM mode to adjust the peak-power control threshold.

During CCM operation, the controller IC102operates in peak-power control mode. The peak load is defined by when the switching power converter100operates at operating point D, which may be 100% load, or 200% load, or greater. As shown in the output voltage VOUTversus output current IOUTgraph, at operating point F, which coincides with operating point D, the switching power converter100delivers maximum current IMAXto the load. The peak load may be user-specified. During CCM operation, the controller IC102regulates the switching of the switch111in accordance with equations (1) and (2) as follows:

tON=(volt-sec)⁢constVin(1)tOFF=Variable(2)
where tONrepresents the conduction pulse duration of the switching signal110applied to the switch111during a switching cycle, (volt-sec) represents the voltage-time product, Vinrepresents the voltage applied to the primary side of the transformer T1-A, and tOFFrepresents the duration that the switch111is not conducting during a switching cycle. The off-time will be regulated by the internal digital control loop within the controller IC102. For example, if the output voltage of the switching power converter100Vout is lower than the target voltage, the off-time will be reduced, if Vout is higher than the target voltage, the off-time will be increased. The on-time or conduction pulse width applied to the switch111during CCM is calculated on a switching-cycle by switching-cycle basis derived from the equation (1). The (volt-sec) during the CCM operation remains constant and maintains the same value registered from the boundary between CDCM and CCM.

Accordingly, in CCM operation, for each switching cycle, the initial primary-side switching current may be high. Given the same Vin*tON(i.e., voltage-time product) applied to the transformer T1-A, the delta current increase is fixed. However, because the initial primary-side switching current may be relatively high compared to the current levels in PFM, DCM, and CDCM operation, the “ending” primary-side switching current during CCM operation may also be relatively high. For example, in CCM operation the controller IC102does not wait for the transformer reset to initiate the next cycle. Thus, in CCM operation the initial current is not zero. Instead, the increased switching current based on Vin*tONis in addition to the initial non-zero current, resulting in a higher ending current. By employing this CCM method, the controller IC102causes the transformer T1-A to output energy that is significantly higher than normal DCM or CDCM operation.

In an embodiment, the controller IC102operates in over current protection mode while operating in CCM operation. For example, the controller IC102will enter OCP operation if the voltage sensed by the VSENSEpin (pin3) of the controller IC102is greater than or equal to the OCP reference threshold value VREF_OCP. In OCP operation, the on-time and off-time will be limited. OCP operation is designed to prevent the primary current from becoming too high, which might damage the switching power converter100or other components. When the controller IC102detects that the primary current reaches a level to trigger OCP threshold, the switch111will be turned off. As shown in the output voltage VOUTversus output current IOUTgraph ofFIG. 3, the curve between the operating points G and H corresponds to the controller IC102operating in OCP mode. Also, during OCP operation, the controller IC102will disable the switching signal110if the OCP condition is satisfied for a specified number (e.g., seven) of consecutive switching cycles. For example, as shown in the output voltage VOUTversus output current IOUTgraph ofFIG. 3, the controller IC102disables the switching signal110at operating point H, which corresponds to when the primary switching current reaches the level represented as ILMT.

FIG. 4illustrates operational waveforms of the switching power converter ofFIG. 1in a peak power operating mode, according to one embodiment. For example,FIG. 4shows an operational waveform of the primary-side switching current and the secondary-side switching current during two consecution switching cycles, n and n+1, respectively. InFIG. 4, the switching cycle n occurs at the boundary between CDCM and CCM, and the switching cycle n+1 occurs in a switching cycle occurring subsequent to switching cycle n. The waveform402represents the primary-side switching current for the switching cycle n, having a conduction pulse width tON. The waveform402increases relatively linearly from an initial value until reaching the voltage threshold VREF_PKthat corresponds to the peak primary-side switching current. At the point when the waveform402reaches the threshold VREF_PK, the conduction pulse width tONends, indicating the switch111transitioned from closed to open (i.e., ON to OFF). At the point when the waveform402reaches the threshold VREF_PK, the waveform402increases until reaching a OCP threshold VREF_OCP, after which the primary-side switching current decreases to its initial value at the beginning of the switching cycle n. The waveform404corresponds to the secondary-side switching current during the portion of the switching cycle n where the switch111is not conducting, represented as tOFF.

The waveform406represents the primary-side switching current for the switching cycle n+1, having a conduction pulse width tON+1. As previously, described with reference toFIG. 3, the conduction pulse duration at the boundary between the CDCM and CCM operation is the same as the same as conduction pulse width during CCM operation. Accordingly, conduction pulse width tON+1is the same as the conduction pulse width tON. The waveform406increases relatively linearly from an initial value until the end of the conduction pulse width tON+1, at which point, the primary switching current is greater than the threshold VREF_PKbut less than OCP threshold VREF_OCP. At the point when the waveform406reaches the end of conduction pulse width tON+1, the waveform406increases beyond the OCP threshold VREF_OCP, after which the primary-side switching current decreases to its initial value at the beginning of the switching cycle n+1. The switching current during the switching cycle n+1 is determined based at least in part on the value of tONfrom the switching cycle n (i.e., previous operation mode) and is limited by whether an OCP condition is detected by the controller IC102. That is, if the controller IC102detects and OCP condition, the value of tON+1will be shorter than the value of tONto prevent the switching current from exceeding a specified OCP level. The waveform408corresponds to the secondary-side switching current during the portion of the switching cycle n+1 where the switch111is not conducting, represented as tOFF.

Upon reading this disclosure, those of skill in the art will appreciate still additional alternative designs for switching power converters. For example, although the controller IC102and its application circuit shown inFIG. 1are based on the primary-side feedback control of flyback converters, the same principles of this disclosure are also applicable to alternative designs based on the secondary-side feedback control. Similar principles can be used with boost type switching power converters or switching power converters with other topologies. Thus, while particular embodiments and applications of the present disclosure have been illustrated and described, it is to be understood that the disclosure is not limited to the precise construction and components disclosed herein and that various modifications, changes and variations which will be apparent to those skilled in the art may be made in the arrangement, operation and details of the method and apparatus of the present disclosure disclosed herein without departing from the spirit and scope of the present disclosure.