Active quasi circulator

An RF quasi circulator circuit is described herein. In accordance with one example of the disclosure the circuit includes a receive port, a transmit port and an antenna port as well as a differential amplifier stage having a first input, a second input and an output that is coupled to the receive port. The circuit further includes a first phase shifting element and a second phase shifting element. The first phase shifting element is coupled between the transmit port and the first input of the differential amplifier and the second phase shifting element is coupled between the transmit port and the second input of the differential amplifier. A tunable impedance is coupled to the differential amplifier, and the antenna port is coupled to the first input of the differential amplifier. The tunable impedance is controlled to tune the damping in a signal path from the transmit port to the receive port.

FIELD

The present disclosure relates to the field of radio frequency (RF) circuits, in particular to the field of active circulator and directional coupler circuits.

BACKGROUND

Radio frequency (RF) receivers and transceivers can be found in numerous applications, particularly in the field of wireless communications and radar sensors. For example in the automotive sector, there is an increasing demand for radar sensors used in so-called “adaptive cruise control” (ACC) or “radar cruise control” systems. Such systems may be used to automatically adjust the speed of an automobile so as to maintain a safe distance from other automobiles ahead.

Modern radar systems make use of highly integrated RF circuits (MMICs, monolithic microwave integrated circuits), which may incorporate all core functions of an RF transmit and receive module (also referred to as “RF font-end”) of a radar transceiver in one single package (single chip transceiver). Such RF front-ends may include, inter alia, a voltage controlled oscillator (VCO), power amplifiers (PA), mixers, and analog-to-digital converters (ADC) and also a circulator or directional coupler.

One characteristic function of transmit and receive modules (RF front-ends) with one or more shared transmit and receive antenna is the separation of the strong transmit signal (TX) from the weak receive signal (RX). Traditionally, this function is implemented by passive devices (circulators, directional couplers), which

have large physical dimensions. Moreover, such passive devices usually exhibit undesired transmission losses. There is a need to replace the mentioned passive devices with active devices, which are a smaller in size and more efficient with regard to transmission losses.

SUMMARY

An RF quasi circulator circuit is described herein. In accordance with one example of the invention the circuit includes a receive port, a transmit port and an antenna port as well as a differential amplifier stage having a first input, a second input and an output that is coupled to the receive port. The circuit further includes a first phase shifting element and a second phase shifting element. The first phase shifting element is coupled between the transmit port and the first input of the differential amplifier and the second phase shifting element is coupled between the transmit port and the second input of the differential amplifier. A tunable impedance is coupled to the differential amplifier, and the antenna port is coupled to the first input of the differential amplifier. The tunable impedance is controlled to tune the damping in a signal path from the transmit port to the receive port

In accordance with another example of the invention the circuit includes a receive port, a transmit port and an antenna port. It further includes a first amplifier stage coupling the transmit port to the antenna port and configured to direct signals received at the transmit port to the antenna port. A second amplifier stage couples the antenna port and the receive port. The second transistor stage is configured to direct signals received at the antenna port to the receive port. A tunable impedance is coupled to the second transistor stage and is controlled to tune the damping in a signal path from the transmit port to the receive port.

DETAILED DESCRIPTION

As mentioned above, a typical function of transmit and receive modules (RF front-ends) with one or more shared transmit and receive antenna is the separation of the strong transmit signal from the weak receive signal. For this purpose, passive devices (e.g., circulators and directional couplers) are commonly used. However, such passive devices often require much space and exhibit undesired transmission losses. One approach to improve the situation is to replace the mentioned passive devices by active devices, which may be designed to be more efficient with regard to losses and smaller in size. Generally, the isolation between an RX port and an TX port of the device is a relevant parameter as it determines the level of the blocker signal (i.e. the portion of the transmit signal which is transmitted from the TX port to the RX port, in an ideal case the blocker signal is zero) of the first devices in the receiver part.

FIG. 1illustrates an example circulator in a schematic diagram. Generally, a circulator for an RF frontend in a radar or communication device has three terminals which are usually referred to as ports. In the example ofFIG. 1, the circulator has three ports PTX, PANT, and PRX, wherein PTXis the transmit port (where the transmit signal TX is applied), PANTis the antenna port (coupled to the antenna), and PRXis the receive port (where the receive signal RX is provided). When designing an active quasi circulator (QC), a general design goal is to obtain a high isolation between the transmit port PTXand the receive port PRXas well as a high gain (no or little losses) in the transmission paths from transmit port PTXto antenna port PANTand from antenna port PANTto receive port PRX. These design goals lead to the following (ideal) matrix of scattering parameters (S-parameters):

One exemplary embodiment of a quasi circulator (QC) circuit with three ports is illustrated inFIG. 2. Accordingly, the QC circuit is composed, inter alia, of two bipolar transistors T1, T2, two λ/4 transmission lines TLλ/4and one resistor R0. The transmit port PTX, at which the transmit signal TX is applied, is formed by the base terminal of bipolar transistor T1and a ground terminal GND coupled to a reference potential (e.g., ground potential). The emitter terminal of bipolar transistor T1is also connected to a ground terminal GND; the collector terminal is connected to a circuit node that is denoted as P1inFIG. 2. Circuit node P1is connected to the base terminal of the second bipolar transistor T2via a first λ/4 transmission line TLλ/4, Circuit node P1is further connected to the emitter terminal of the second transistor T2via a second λ/4 transmission line TLλ/4. The antenna port PANT, to which the antenna is connected, is formed by the emitter terminal of the bipolar transistor T2and a ground terminal GND. The base terminal of bipolar transistor T2is coupled to a supply voltage terminal VDD. The collector terminal of the bipolar transistor T2is connected to the supply voltage terminal VDD via an inductor L1. The receive port PRXis formed by an output terminal, which is connected to the collector of the bipolar transistor via a capacitor C1, and a ground terminal GND. The receive signal RX is provided at the receive port PRX.

In the following description the function of the quasi circulator (QC) circuit is describe in more detail. To keep the explanation simple, the receive path (port PANTto port PRX) and the transmit path (port PTXto PANT) are considered one after the other. The bipolar transistor T1operates as a transconductance amplifier stage, which is configured to amplify the transmit signal TX. At circuit node P1the amplified signal is distributed over two branches and split in two signals TXAand TXB. These two signals TXAand TXBare both subject to a 90 degree phase rotation caused by the two transmission lines TLλ/4. As a result, the voltage drop between the base and the emitter of transistor T2is zero (transistor T2thus remains off). Therefore, no contribution of the transmit signal TX is transmitted to the receive port PRX, and (in an ideal case) the isolation is perfect. The condition for isolation (base-emitter voltage of transistor T2being zero) is satisfied if the transmit signal TX is equally divided into the two signals TXAand TXB(wherein TXA=TXB). In other words, half of the signal power of the transmit signal TX is directed through the first transmission line TLλ/4(to termination resistor R0) and half of the signal power is directed through the second transmission line TLλ/4(to the antenna port). This will be the case when the resistance of resistor R0matches the system impedance Z0(i.e. R0=Z0) and the antenna impedance (present at the antenna port PANT).

An antenna signal ANT received by the antenna is applied to the emitter terminal of the bipolar transistor T2and also directed (with a 180 degree phase rotation) to the base terminal of transistor T2. The signal path from antenna port PANTto base terminal of transistor T2causes a 180 degree phase rotation due to the two λ/4 transmission lines TLλ/4. Thus, the signals at the emitter and at the base of transistor T2have a 180 degree phase shift relative to each other and transistor T2effectively operates as differential amplifier (for signals coming from the antenna), which is generally referred to as differential amplifier AMP. The amplified antenna signal can be tapped at the receive port PRX. The inductor L1and the capacitor C1are used to decouple the receive port PRXfrom the DC supply voltage and to decouple the DC supply terminal VDD from any AC signals. Generally, a differential amplifier has two inputs and is configured to amplify the difference of the signals applied at the two inputs; the amplified difference is provided at the amplifier output. A skilled person is aware of various different possibilities to implement a differential amplifier, which is thus not further discussed herein in more detail. In the example ofFIG. 1(as well as in the examples ofFIGS. 2, and 4-9) a single transistor T2is operated as differential amplifier.

In the following description the circuit ofFIG. 2is analytically analyzed with reference to the VCCS equivalent circuit (VCCS=voltage controlled current source) illustrated inFIG. 3, which represents bipolar transistor T2. InFIG. 3the impedance Zπrepresents the differential base-emitter resistance in the operating point of transistor T2. The current source Qπprovides a current i3proportional to the voltage drop Vπacross impedance Zπ, that is i3=Vπ·gmand gmis the differential transconductance of transistor T2in the operating point. V1denotes the voltage present at the base terminal of transistor T2, V2the voltage present at the emitter of transistor T2, and V3the voltage present at the collector of transistor T2. The equivalent circuit ofFIG. 3is represented by the following matrix YVCCSof admittance parameters (Y-parameters):

The λ/4 transmission lines TLλ/4are represented by the matrix YTLof Y-parameters (Y-matrix)

YTL=[0jZ1jZ10],(3)
wherein Z1denotes the impedance of the transmission line and j is the imaginary unit. Using equations 1 and 2, the Y-matrix YQCof the overall quasi circulator ofFIG. 2can be calculated as follows:

As a result of the matching condition S11=0 (no reflection at port PTX) the impedance Z1of the transmission lines TLλ/4yields Z1=Z0√{square root over (2)}=50 Ω√{square root over (2)} and the S-matrix SQCsimplifies to:

It has been found that the miller capacitance Cμ(seeFIG. 3), which is effective between the base and the collector of the transistor T2, has an even stronger impact on the matrix SQCof S-parameters than capacitance Cπ. The presence of this parasitic capacitance Cπresults in the mentioned isolation condition (S13=S31=0) not being satisfied with Y0= 1/50 Ω, as the circuit is no longer balanced with respect to the circuit node P1(seeFIG. 2). Further parasitic capacitors (e.g. substrate capacitances) may also bring the circuit out-of-balance. Those parasitic capacitances can not be accurately determined and considered in the circuit design. To improve the situation, the termination resistor R0may be replaced by a tuner circuit as illustrated in the examples described below.

FIG. 4illustrates one example implementation of a quasi-circulator in accordance with one embodiment of the present invention. The function of the circuit ofFIG. 4is basically the same as in the previous example ofFIG. 2. However, the present example includes only one differential amplifier AMP (which corresponds to the amplifier stage formed by transistor T2inFIG. 2) for the amplification of the signal ANT incident at the antenna port PANT. The amplifier output is coupled to the receive port PRX, at which the received signal may be tapped.

The differential amplifier AMP has two inputs (an inverting input an a non-inverting input). Each of the two inputs is connected to the transmit port PTXvia a 90° phase shifting elements, which may be, for example, λ/4 transmission lines TLλ/4. In the present example, the non-inverting input of the differential amplifier AMP is coupled to the antenna port PANTand the inverting input of the differential amplifier is coupled to a supply voltage VDDvia an adjustable (tunable) termination impedance ZTUNE. This adjustable termination impedance ZTUNEis basically used for the same purpose as the termination resistor R0shown inFIG. 2. However, the impedance ZTUNEmay be tuned to ensure the overall circuit is (approximately) balanced. A signal TX incident at the transmit port PTXis directed to the antenna port PANTwith a 90 degree phase shift (via one of the 90° phase shift elements). However, the signal TX incident at the transmit port PTXis not directed to the receive port PRXas the difference signal applied to the differential amplifier is zero (both input signals of the differential amplifier AMP are subject to the same phase shift). A signal ANT incident at the antenna port PANTis amplified and output at the receive port PRXdue to the two phase shifting elements, which provide a total phase shift of 180 degree between the two inputs of the differential amplifier AMP. The tunable impedance will be discussed in more detail with reference toFIGS. 6, 7 and 8.

The differential amplifier may be implemented using a bipolar junction transistor T2as shown in the example ofFIG. 5. The base and the emitter of the transistor T2may be regarded as the inputs of the differential amplifier AMP, whereas the collector, which is coupled to the supply voltage VDD by a complex valued impedance L1, may be regarded as output of the differential amplifier AMP. The receive port PRXis coupled to the collector of transistor T2via capacitor C1, which decouples DC signal components from the port PRX. Like in the previous example, a signal TX incident at the transmit port PTXis directed to the antenna port PANTwith a 90 degree phase shift (via one of the 90° phase shift elements), but not directed to the receive port PRXas the difference signal applied between base and emitter of transistor T2is zero (both input signals of the differential amplifier AMP are subject to the same phase shift). A signal ANT incident at the antenna port PANTis amplified and output at the receive port PRXdue to the two phase shifting elements, which provide a total phase shift of 180 degree between the base and emitter of transistor T2.

The circuit ofFIG. 6is similar to the basic circuit ofFIG. 1. Furthermore, the circuit ofFIG. 6is also very similar to the previous example ofFIG. 5. Different from the example ofFIG. 5, the present example has an additional input amplifier stage PRA (like the example shown inFIG. 1), which is implemented as transconductance amplifier stage. As compared to the example ofFIG. 1, the resistor R0(which has a real-valued impedance) is replaced by a circuit representing a tunable impedance ZTUNEas already mentioned in the examples ofFIGS. 4 and 5. When appropriately tuned, the tunable impedance ZTUNEcan (at least partially) compensate for the negative effects of the abovementioned parasitic capacitances (seeFIG. 3) and thus balance the circulator circuit so that half of the power of signal TX is directed to the antenna port and half of the power is dissipated in the termination provided by the tunable impedance ZTUNE. Such a balancing of the quasi-circulator circuit entails an improvement of the isolation between the transmit port PTXand the receive port PRX(i.e. scattering parameter S31is minimized and ideally zero).

The tunable impedance ZTUNEcan also compensate for a mismatch between the antenna impedance and the system impedance Z0. Such an antenna impedance mismatch would also bring the circuit out-of balance and deteriorates the isolation between the transmit port PTXand the receive port PRX. Besides the tunable impedance ZTUNEthe circuit ofFIG. 6operates in the same way as the circuit ofFIG. 1and reference is made to the respective description above. Test measurements have shown that the transmission attenuation between the transmit port PTXand the receive port PRXis improved from 14 dB (without tunable impedance) to 30 dB or even 40-50 dB with appropriate fine tuning.

FIG. 7is one example of a practical implementation of the basic circuit ofFIG. 6. Basically, the circuit ofFIG. 7corresponds to the previous example ofFIG. 6with additional circuitry for biasing the transistors and decoupling DC signal components from the ports PTX, PRX, and PANT. Similar to the example ofFIG. 6, the QC circuit is composed, inter alia, of two bipolar transistors T1, T2, two λ/4 transmission lines TLλ/4for coupling the two transistors T1, T2, the tunable impedance ZTUNEand the mentioned circuitry for biasing the transistors T1, T2and decoupling DC signal components from the ports.

The biasing of transistor T1is provided by a bias voltage source providing a first bias voltage VBIAS1, which is connected to the gate of transistor T1via resistor R1, and a current source CS1connected between the emitter of transistor T1and ground potential (at ground terminal GND). A capacitor C6is connected in parallel to the current source CS1to by-pass RF signals (capacitance C6may be regarded as short-circuit for RF signals). The quiescent point of transistor T1is determined by the current source CS1, resistor R1and the first bias voltage VBIAS1. The transmit port PTX, at which the transmit signal TX is applied, is coupled to the base terminal of bipolar transistor T1via capacitor C1. The delay line TLS1may be used for impedance matching at the transmit port PTX. As in the previous example ofFIG. 6, the collector terminal is connected to a circuit node that is denoted as P1.

The biasing of transistor T2is provided by a bias voltage source providing a second bias voltage VBIAS2, which is connected to the gate of transistor T2via resistor R2. Capacitor C2is connected between the gate of transistor T2and the common circuit node P2of tunable impedance ZTUNEand one of the transmission lines TLλ/4; capacitor C2is thus used to decouple DC signal components (base DC voltage of transistor T2) from circuit node P2. However, the base of transistor T2is (by means of capacitor C2) AC coupled to the supply potential VDDvia the tunable impedance ZTUNE. Capacitor C3is used to decouple DC signal components (emitter DC voltage of transistor T2) from circuit node P3, which is coupled to the antenna port PANT. Capacitor C4provides a DC decoupling of the antenna, and the delay line TLS2(together with capacitor C4and antenna pad P, which acts like a capacitor coupled between ground and the antenna port PANT) may be used for impedance matching.

Circuit node P1(i.e. the output of the first transistor stage formed by transistor T1) is connected to circuit nodes P2and P3via two λ/4-transmission lines TLλ/4. Similar to the basic example ofFIG. 6, the circuit node P1is thus connected (via the two transmission lines TLλ/4) to the base and the emitter of transistor T2. However, this connection is only available for RF signals whereas DC signals are blocked by capacitors C2and C3respectively. A further λ/4-transmission lines TLλ/4is connected between the emitter of transistor T2and ground (ground terminal GND) in order to DC couple the emitter to ground (whereas RF signals are connected to circuit node P3via capacitor C3as mentioned above).

As compared to the basic example ofFIG. 6, the inductor L1is replaced by transmission lines TLL1and TLS3, which are coupled between the collector of transistor T2and a supply voltage terminal providing a second supply voltage VDD2. Accordingly, the common circuit node between the two transmission lines TLL1and TLS3are connected to the receive port via capacitor C5. The purpose of capacitor C5is the same as in the example ofFIG. 6, namely to block DC signals from the receive port PRX. Generally, the present example may be regarded as one practical implementation of the basic circuit ofFIG. 6.

In the present example, the size as well as the bias voltage VBIAS1of transistor T1(transconductance stage) may be designed for an input referred 1 dB compression point of 1 dBm. The base of transistor T1is connected to a circuit node at which the bias voltage VBIAS1is applied. To achieve a linear operation, the emitter of transistor T1is coupled with a bias circuit, which includes the current source CS1as mentioned above.

In the circuit design any parasitic elements between transistor T1and capacitor C6can be considered as they may have an impact on the gain as well as the impedance matching of the transconductance stage formed by transistor T1. The size of transistor T2should be chosen small enough so that its miller capacitance Cμ(seeFIG. 3) does not have a significant impact on the balance of the overall QC circuit. Nevertheless, the biasing of transistor T2may be designed to provide a good signal to noise ratio. The impedance connected at the emitter of transistor T2emitter is low, which may have an adverse effect on the reflection coefficient at the antenna port PANT. Therefore, the capacitor C3should be small.

FIG. 8illustrates one example embodiment of the tunable impedance ZTUNEwhich may be used in the circuit ofFIG. 7. The basic idea is connecting two varactor diodes DV1, DV1′ and DV2, DV2′ in parallel. The delay lines TLS1and TLS2shown inFIG. 8are used to compensate for static capacitances in order to improve the dynamic range of the tunable impedance. The delay lines TL1and TL2are used for impedance transformation of the capacitances of the varactor diodes.

FIG. 9illustrates another example implementation of a quasi-circulator which is an alternative to the previous example ofFIG. 6. The present quasi-circulator is almost identical to the previous example, except that a fixed termination resistor R0is used between the circuit node P2and the supply terminal for the supply voltage VDD. In this case, the tunable impedance ZTUNEis coupled to the antenna port PANT. More precisely, a series circuit of the tunable impedance ZTUNEand a capacitor C1′ is connected parallel to the antenna at the antenna port PANT. In this example, the tunable impedance ZTUNEshould have a wider tuning range than in the previous example ofFIG. 6. Furthermore, the tunable impedance cannot compensate for an arbitrary impedance mismatch of the antenna as it is coupled parallel to the antenna.