Poly-phase AC/DC active power converter

A three-phase AC/DC active power converter provides an H-bridge that is controlled by a DSP (digital signal processing) controller that places the H-bridge in a voltage-boost mode of operation when voltage of a DC-link capacitor maintained by the H-bridge is near than the voltage input from a three-phase power source. The voltage difference between the boosted DC-link voltage and the three-phase power source voltage provides a voltage potential thereby giving the control loops a possible gain value. The gain value provides loop stability to thereby prevent an inrush of electrical current into the power converter upon startup. The converter also allows harmonic distortion to be modified through wave shaping of the normally pure sinus conduction signal.

TECHNICAL FIELD

The present invention is generally directed to AC/DC active power converters. Particularly, the present invention relates to a poly-phase AC/DC active power converter for use with a variable frequency generator. More particularly, the present invention is directed to an AC/DC active power converter that is reconfigured on start-up to operate in a voltage boost mode to generate a gain voltage to reduce electrical current spikes otherwise seen by the power converter. Additionally, the present invention is directed to an AC/DC active power converter that utilizes real-time geometric calculations to identify the conduction phase angle of the input power, so as to output power having a unity power factor.

BACKGROUND ART

Electrical generation systems often provide power consumed by power converters, such as three-phase AC/DC or AC/AC power converters. These power converters modify incoming AC (alternating current) power so that it is output with different electrical specifications. For example, electrical generators may supply power with varying operating frequencies, as in the case of generators used aboard certain aircraft, which are driven directly by the operation of the aircraft's jet engine. Because the jet engine's speed also controls the aircraft thrust, the speed must be varied, which results in AC (alternating current) power having a widely-varying primary operating frequency around the base frequency, such as 400 Hz. Unfortunately, the components that are to be powered in many applications are designed to operate within narrow bands of incoming line frequency and voltages. Additionally, designers of power distribution grids impose demanding specifications that must be met by the loads connected to the grid to enable the safe distribution of power thereto. For example, current and voltage distortion [Total Harmonic Distortion (THD)], as well as voltage and current phasing (Power Factor), are often restricted to maximum levels to protect the power generator and the distribution grid, as well as various electronic components and equipment coupled to the grid.

To overcome the problems associated with the varying frequency of the AC power output by such variable frequency power generators, power converters have utilized various power transformer designs. Specifically, power transformers have been designed to provide multiple phase outputs that are rectified and then fed to a DC-link capacitor for supply to a DC bus. However, the rectification of the incoming AC power by the transformer may result in uncorrected and unavoidable power factor shifts, while injecting an undesirable level of harmonic distortion back onto the power grid. To minimize these drawbacks, poly-phase transformers have been developed to increase the frequency of the generated harmonic distortion to a more acceptable level. Because the performance of such transformer designs tends to be dependent on phase voltage balance, line reactors are required to smooth and balance the capacitor charging currents supplied by the poly-phase transformer, which contributes additional weight, size, and cost to the power supply. Thus, while existing transformer designs provide adequate performance, they are deficient with regard to their large size, significant weight, and excessive cost, and do not compensate for load power factor, thus rendering power converters using such transformers undesirable.

In order to improve upon the deficiencies of the transformer-based converters, all-electronic converters, also referred to as active power converters, have been developed. These converters typically operate by transferring the energy available at the incoming main power source to the DC supply or link capacitor by controlling currents through an inductor. Specifically, the active power converter uses a current loop regulator to control the currents through the inductor in accordance with a control loop system that is adjusted based on the converter's operating conditions, to enable its stable operation. However, upon start up of the converter, the lack of voltage potential between the incoming mains and the DC bus, to which the converter is coupled, compromises control loop stability due to the lack of a forcing function (voltage difference) to provide the required gain for the control loop to correctly operate. This can result in initial startup currents that are largely uncontrolled, which places a significant amount of electrical stress on the components of the power converter and the power grid, resulting in a reduction in their operational reliability. Thus, while active power converters can provide power factor correction in a reduced form factor, that is lightweight, they can suffer from startup current spikes, undesirable even-harmonic line distortion, and errors as the operating frequency of the mains power source changes.

Furthermore, active power converters that provide power factor control require line synchronization, which allows the controller to consume current in phase with the line voltage of the power source. Currently, phase-locked loop (PLL) based line synchronization methods are generally used for controlling the power factor in many power applications. During operation, the PLL operates as a timer where synchronization occurs at a phase zero crossing voltage of the input line power, whereby the PLL generates the required steps between successive zero crossings to presume the phase angle of the incoming line power. While the synchronization established by the PLL mitigates any cumulative error, errors during each cycle of the input line power still occur. Thus, while PLLs generally provide acceptable performance under steady state conditions, they can produce significant errors during transient conditions when the frequency of the input line power is changing, which frequently occurs, when a variable frequency power source is used to supply power to the converter.

Therefore, there is a need for a poly-phase AC/DC active power converter that provides a regulated DC voltage power bus, while maintaining a unity or near-unity power factor and providing low total harmonic distortion on both the current and voltage waveforms carried on the lines from a three-phase power source. In addition, there is a need for a poly-phase AC/DC active power converter that utilizes real-time geometric calculations to monitor frequency changes in the power output by a variable frequency power source in order to correct and provide unity or near-unity power factor. Furthermore, there is a need for a poly-phase AC/DC active power converter that can be initially reconfigured at start-up as a boost regulator to increase voltage on a DC-link capacitor in order to provide gain value so as to reduce or eliminate the uncontrolled inrush of current spikes into the power converter.

SUMMARY OF THE INVENTION

In light of the foregoing, it is a first aspect of the present invention to provide a poly-phase AC/DC active power converter.

It is another aspect of the present invention to provide a poly-phase AC/DC active power converter to convert AC power from a poly-phase variable frequency power source into DC power, comprising an H-bridge converter maintaining a capacitor at its output, the converter adapted to receive each of the phases of AC power source at its input through respective line reactors inline therewith, a controller coupled to the H-bridge converter to control the converter in accordance with a control structure, wherein upon the initial powering of the H-bridge, the controller enters a voltage boost mode, such that the line reactors are charged and discharged into the capacitor to raise its voltage above that of the power source, to generate a gain value used by the control structure to control the current consumed from the power source, so as to maintain a constant output voltage during the start up of a normal AC/DC conversion mode, so as to reduce an inrush of current during the conversion of AC power into DC power.

Yet another aspect of the present invention is a method of controlling an H-bridge to convert AC power from a poly-phase power source into DC power comprising providing an H-bridge power converter maintaining a DC-link capacitor at its output, providing a controller to control the H-bridge in accordance with a control structure, coupling the input of the H-bridge to the power source, operating the H-bridge in a boost mode, so as to charge the DC-link capacitor to a voltage above the voltage of the power source to generate a gain value, executing the control structure with the gain value to control the current consumed by the converter to maintain a constant output voltage, and operating the H-bridge in a normal mode, so as to convert AC power from the power source into DC power.

Still another aspect of the present invention is a method of controlling an AC/DC power converter to convert AC power from a 3-phase power source into DC power comprising providing an H-bridge having a DC output bus, which maintains a DC-link capacitor, providing a controller to control the H-bridge in accordance with a control structure, representing at least 2 of the 3 phase voltages output from the power source as vectors in two-phase rotary coordinates, calculating a conduction angle of the power consumed from the 3-phase power source from the vectors using an arc-tangent function, and controlling the H-bridge to adjust the phase angle of the current consumed from the power source to match the phase angle of the voltage provided by the power source at the DC output bus, to provide unity power factor.

BEST MODE FOR CARRYING OUT THE INVENTION

A poly-phase AC/DC active power converter is generally referred to by the numeral100, as indicated inFIG. 1of the drawings. The power converter100utilizes a plurality of insulated-gate bipolar transistors (IGBT)110A-F and a DC-link capacitor120configured as a three-phase active H-bridge130to convert AC (alternating current) power received from a three-phase variable frequency power source140, which generates power at varying frequencies, into DC (direct current) power. To reduce the inrush of electrical current into the converter100upon a cold start, or initial operation, of the converter100, where the DC-link capacitor voltage is zero, a DSP (digital signal processing) controller200follows a control structure, which controls the IGBTs110A-F in a predetermined sequence, to reconfigure the H-bridge130to operate in a boost regulation mode.

In the boost regulation mode, the converter100operates as a boost rectifier, such that the voltage at the DC-link capacitor120maintained by a DC output bus is raised above that of the voltage of the variable frequency power source140by a predetermined amount. The difference between the two voltages establishes a voltage potential, which forms a gain value that is used to initialize the current controller maintained by the control structure of the DSP controller200. That is, once the voltage difference value (gain) is reached, the control structure reconfigures the H-bridge130to operate in its normal AC/DC conversion mode, and the gain value is used during its initial start-up to provide stability to the control structure, so as to prevent the inrush of uncontrolled electrical current into the H-bridge130, which would otherwise result in the distortion of AC power that is consumed from the power source140, thereby stressing the electrical component converter100. The power converter100also utilizes geometric computations to determine the actual conduction angle of the power consumed by the converter100from the variable frequency power source140in real-time, thus reducing the errors generated by prior art AC/DC converters that utilize phase-locked loops (PLLs). As such, the converter100consumes current in phase with the voltage supplied from the power source140, so as to provide a unity or near-unity power factor with very low total harmonic distortion on both current and voltage throughout a wide range of AC frequencies. Furthermore, while the following discussion relates to the use of the power converter100with three-phase power sources, the converter100may be readily adapted for use with power sources having any number of phases.

The power converter100includes the H-bridge130that is comprised of a network of switching elements250A-F that are comprised of IGBTs110A-F and parallely-coupled diodes300A-F. Specifically, the IGBTs110A-F are coupled to respective diodes300A-F, such that the cathode of each diode300A-F is coupled to the collector (C) terminal of its associated IGBT110A-F to form collector nodes320A-F, while the anode of each diode300A-F is coupled to the emitter (E) terminal of its associated IGBT110A-F to form emitter nodes330A-F. The switching elements250A-C are respectively arranged in series with switching elements250D-F, such that the emitter nodes330A-C are coupled to the collector nodes320D-F via respective node lines360A-C. Furthermore, the collector nodes320A-C of switching elements250A-C are coupled together, while the emitter nodes330D-F of switching elements250D-F are also coupled together. The power converter100also includes the DC-link capacitor120that is coupled at its anode to the collector node320C, and coupled at its cathode to the emitter node330F, which thereby forms the interface of a DC output bus400.

Each phase of the three-phase variable frequency power source140is coupled to the power converter100via power lines500A-C that maintain respective line reactors510A-C in-line therewith. It should be appreciated that the line reactors510A-C comprise an inductor, to provide impedance to the power lines500A-C, so as to reduce input harmonics and to buffer against low-magnitude current spikes generated by the mains power source140. Furthermore, the power source140may comprise any suitable power generator, consisting of any number of phases, such as a turbine driven generator that generates power at varying output frequencies.

To switch each IGBT110A-F between ON and OFF states in the manner to be discussed, the DSP controller200is coupled to the gate terminal (G) of each IGBT110A-F via control lines600A-F. The DSP controller200includes the necessary hardware and/or software needed to carry out the functions to be discussed. It should also be appreciated that the DSP controller200may be replaced with any other suitable controller, microcontroller, or other computing device that provides the necessary functions to carry out the operation of the converter100. In particular, the DSP controller200employs a control structure630, as shown inFIG. 2, that monitors the electrical current consumed by the converter100, and the voltage supplied at the DC bus400by the converter. As such, the DSP controller200applies pulse width modulation (PWM) control signals with suitable duty cycles to control lines600A-F in order to switch the IGBTs110A-F ON and OFF to convert the three-phase variable frequency input power from the power source140into DC power.

Specifically, in the normal AC/DC converter mode, the control structure630implemented by the DSP controller200used to control the switching of the IGBTs110A-F of the H-bridge130is represented by a pair of series coupled proportional integral (PI) controllers632and636that are separated by a summing point640. Specifically, the output of PI controller636is coupled to the converter100via control lines600A-F, while the input of PI controller632is coupled to another summing point646. As such, the summing point640receives the magnitude of the current consumed by the converter100from the power source140, via an inner current loop648, as commanded by the output of the PI controller632, to thereby define a current error value that is utilized by the PI controller636to control the current consumed by the converter100. The control structure630is also configured, whereby the summing point646maintained at the input of the PI controller632receives a predetermined voltage set point value650maintained by the DSP controller200, and the magnitude of the voltage output by the converter100via the DC bus400from an outer voltage loop652to thereby define a voltage error value. Thus, during normal operation, the loop652is used to maintain the voltage output at the DC bus400at a constant value established by the voltage set point value650. In the event the voltage at the DC bus400varies from the voltage set point value650, the current controller632generates a current command value at its output to compensate for the change in voltage at the DC bus400. This commanded current value is used to define the current error that is input into controller636, which modulates the switching of the IGBTs110A-F, so as to adjust the current consumption of the converter100so as to maintain the voltage of the DC bus400at the voltage set point value.

Thus, during normal operation, DSP controller200determines the proper conduction angle of the current consumed by the converter100from the power source140and calculates the voltage error value at the output of summing point646and the current command value at the output of summing point640, which are used to adjust the duty cycle of the PWM signals applied to control lines600A-F to control the amount of current consumed from the variable frequency power source140in order to maintain the DC bus400at the DC voltage established by the voltage set point value650.

However, the control loops652and648of the control structure630require a voltage gain or a voltage potential difference between the peak voltage supplied by the power source140and the voltage at the DC bus400to remain stable and in control, in order to allow the converter100to output DC power as the frequency of the power source140varies. However, this voltage potential between the power source140and the DC bus400or voltage gain is generally not present upon the initial cold start-up of the converter100, thus causing the control structure630to operate in an unstable manner, which results in an unwanted inrush of current into the converter100. That is, on a cold start of the power converter100, when it is initially placed into its normal operating mode, the actual gain, in the form of current error, available at the input the PI controller636is not defined due to the lack of voltage potential difference between the power source140and the DC bus400. As a result, the operation of the PI controller636becomes unstable, causing it to control the H-bridge130in a manner that allows an uncontrolled inrush of current to enter the converter100, thereby generating electrical current spikes and other distortion. Thus, the processes used to enable the converter100to generate a voltage gain needed to generate a current error value at the input of the PI controller636to enable its stable operation on the initial or cold start of the converter100will now be discussed below. It should be appreciated that the voltage gain value is normally some percentage above that of that supplied by the power source140.

Reconfiguration Operation

To provide an amount of voltage gain during the initial start up of the converter100to enable the control loops648and652to operate in a stable manner, the DSP controller200reconfigures the H-bridge130so that it functions in a boost regulator mode, whereby the DC-link capacitor120of the DC bus400is charged to a voltage that exceeds that of the power source140. The operational steps taken by the converter100to carry out the boost regulator mode are generally referred to by the numeral700, as shown inFIG. 3of the drawings. Initially, at step710, the DSP controller200is started. Next, at step720, the DSP controller200determines if the converter100is being cold started. That is, the controller200determines if sufficient voltage exists at the DC bus400to generate the voltage gain needed by the control structure630to enable its normal, stable operation. If the DSP controller200determines that the voltage at the DC bus400is insufficient, whereby it is not greater than that provided at the power source140, the process700continues to step730. At step730, the DSP controller200places the H-bridge130into a boost regulator mode so that the power converter100operates as a boost regulator, as shown inFIG. 4. The DSP controller200switches the appropriate IGBTs110A-110F ON and OFF, such that the line reactors510A-510C are charged by placing them across the lines500A-C of the power source140, and then discharged into the DC-link capacitor120. However, it should be appreciated that there are any scenarios known in the art of transistor switching that would allow current to be built up in the line reactors510.

For example, in boost regulation mode, IGBT110F is turned on at the correct point in time for a predetermined amount of time, during which current from the variable frequency power source140flows through line reactor510B and through line reactor510C, as indicated by the current path732shown inFIG. 4. Continuing, when IGBTs110F is turned off, the energy stored in line reactors510B and510C flows through diodes300B and300F, as shown by the current path734inFIG. 5, thereby transferring this energy into the DC-link capacitor120. In one aspect, the amount of time that the IGBT110F is turned on or off is chosen, so that the current built-up in the line reactor510B and510C remains at safe level to allow the system to charge in a reasonable amount of time.

Thus, by operating the H-bridge130as a boost regulator, the voltage of the DC-link capacitor120is charged to a start-up voltage value that exceeds the peak voltage of the power source140, as indicated at step740. After the DC-link capacitor120has been charged, the process700continues to step750, where the DSP controller200disables the boost mode and places the power converter100into its normal operating mode of AC/DC power conversion, as indicated at step760. Whereby, each phase of the AC power from the variable frequency power source140is supplied to the power converter100to charge the DC link capacitor120of the DC bus400to a peak value of the voltage that has been defined by the voltage set point value650.

However, returning to step720, if the DSP controller200determines that there is a sufficient voltage difference to establish the gain value used by the control structure630, the process proceeds directly to step750, where the converter100is placed in a normal AC/DC conversion mode, as previously discussed.

Synchronization

In order to achieve a unity or near-unity power factor by the power converter100when used with the variable frequency power source140, the DSP controller200utilizes a geometric synchronization technique. This technique tracks the conduction angle of the power supplied by the power source140and commands the current consumed by the converter100in a manner to provide low harmonic distortion and unity or near-unity power factor, as discussed below. Specifically, the DSP controller200utilizes the Park/Clark and Inverse Park/Inverse Clark arithmetic transforms to provide the required current loop control utilized by the controller636of the control structure630. Initially, as shown inFIG. 6, the DSP controller200applies the Clark transform, whereby the voltage of each of the three phases500A-C of the variable frequency power source140, which are initially defined as a three-phase vector state, are converted into a two-phase rotating vector800. The resultant angle of the two-phase rotating vector800is calculated as a conduction angle810of the power supplied from the power source140and is utilized by the DSP controller200and the Inverse Park/Inverse Clark transform to control the IGBTs110A-F of the H-bridge130via control lines600A-F to achieve unity or near-unity power factor, as discussed below.

Specifically, in order to carry out the Park/Clark transform, two out of the three-phase voltages Va, Vb, and Vcof the power source140supplied via respective power lines500A-C are required, such that the two-phase vectors are defined as Vαand Vβ, whereby

Vα=Va,Vβ=(Va+2⁢Vb)3⁢⁢and⁢⁢Va+Vb+Vc=0.
Once each of the phase voltages Va, Vb, and Vcof the power source140are converted into two-phase rotary coordinates, the conduction angle810of the power consumed by the converter100from the variable frequency power source140is calculated using the Arc Tangent function: Line Conduction Angle

Φ=Arc⁢⁢tan⁡(VβVα)+90+offset.
The offset value is used to compensate for delays in signal processing, which may include sampling delay and filter phase lags, associated with the operation of the DSP controller200. Once calculated, the conduction angle (Φ)810is then used by the DSP controller200to compute the inverse Park and the inverse Clark transform so that the current consumed by the converter100from the variable frequency power source140is in phase with the voltage supplied thereby, so as to provide a unity or near-unity power factor.

Specifically, the inverse Park transform is given by:
Vα=Vd*Cos Φ−Vq*Sin Φ
Vβ=Vd*Sin Φ+Vq*Cos Φ, wherebyVdandVqare command voltages for the direct and quadrature axis that are calculated by the PI controller632of the control structure630.

And the inverse Clark transform is given by:

Vr⁢⁢1=VβVr⁢⁢2=(-Vβ+Vα)2Vr⁢⁢3=(-Vβ-Vα)2,
whereby Vr1, Vr2, Vr3represents the rotary voltage commands used by the control structure630to enable the synchronous operation of the controller100.

In other words, the controller200processes the conduction angle (Φ)810via the inverse Park transform to generate Vαand Vβthat are used by the inverse Clark transform to generate the 3 phase rotary voltage commands Vr1, Vr2, and Vr3. The 3 phase rotary voltage commands Vr1, Vr2, and Vr3are used by the controller200to control the switching of the IGBTs110A-F to adjust the phase angle of the current consumed by the H-bridge130so that it matches the phase angle of the voltage output by the power source140, thereby enabling a unity or near-unity power factor to be achieved.

In addition to providing unity power factor operation, the 2-phase vectors Vαand Vβgenerated from the Clark/Park transform can also be used by the DSP controller200to calculate a resultant vector820, which is defined as Vm=√{square root over ((Cβ2+Vα2))}. The resultant vector Vm, allows quick and convenient error checking for both the incoming line voltage range and missing phase detection provided by the variable frequency power source140. Whereby if the calculation of vector820results in a vector length lower than a minimum vector length830, the DSP controller200generates a visual and/or audible prompt via an indicator831maintained to indicate that the incoming power from the power source140has fallen out of predetermined specifications. Similarly, if the calculation of resultant vector820results in a vector, which exceeds a maximum vector length832, the DSP controller200generates a visual and/or audible prompt via the indicator831, to indicate that the incoming power from power source140has fallen out of predetermined specifications. As such, by monitoring the fluctuation of the resultant vector820, various electronic components coupled to the output of the converter100, may be shut down to avoid damage thereto, should the output power of the power source140fall outside predetermined specifications, such as in the case of an overvoltage event.

With reference toFIG. 7, the converter100may be configured whereby the current controller defined by PI controller636generates a modulated current command930to reduce the harmonic distortion associated with power consumed by the converter100from the power source140. As previously discussed, the current controller defined by the PI controller636receives its command signal from PI controller632, and, in concept, PI controller632provides an output for each phase500A-C of the power source140. As such, each phase500A-C of the power source140has an associated current command930generated by the PI controller632that is used to control the switching of the IGBTs110A-F. Specifically, the current command930is modulated by the arithmetic conditions940A-C embodied by the DSP controller200, which are defined respectively as: IcmdA=Icmd*Sin(Φ); IcmdB=Icomd*Sin(Φ+120 degrees); and IcmdC=Sin(Φ+240 degrees). As such, the current command930generated by the current controller632is further modulated by the arithmetic conditions940A-C to generate individual current control commands950A-C that are supplied to the IGBTs110A-F via control lines600A-F, to control current consumed from each of the three phases500A-C of the power source140. Thus, control lines600A/600D provide the modulated current command950A, control lines600B/600E provides the modulated current command950B, and control lines600C/600F provide the modulated current command950C to the H-bridge130to control the consumption of current from the power source140.

However, by varying the Sin function of the arithmetic conditions940A-C via harmonic injection, lookup tables, or conduction angle modulation, the harmonic current content of the power consumed by the converter100from the variable frequency power source140can be altered. In one aspect, instead of modulating the current command930with a pure sinus function Φ, it can be modulated with harmonics that trim the current command ψ to reduce total harmonic distortion of the power consumed from the variable frequency generator140. To achieve this operation, the converter100implements inverse Park and inverse Clark arithmetics, as discussed above, to provide this control, so it is angle fed into the inverse Park transform that is modified to provide the harmonic control.

It is also contemplated that by monitoring the rate of change (dΦ/dt) of the conduction angle810allows for monitoring the frequency of the power received by the H-bridge130from the power source140, while the second differential of (dΦ/dt) allows additional compensation techniques to be employed, such as an acceleration feed forward or time lag compensation, to be performed by the converter100. In particular, feed forward compensation can be used to compensate for delays experienced by the converter100, which may improve line conduction angle810tracking that is achieved by the converter100.

Therefore, one advantage of the present invention is that a poly-phase AC/DC active power converter controls an H-bridge to operate in a boost mode to generate a voltage difference (gain value) for use in a normal mode to prevent the inrush of electrical current from a variable frequency power source coupled thereto. Another advantage of the present invention is that the poly-phase AC/DC active power converter has reduced size and is lightweight.

Yet another advantage of the present invention is that the poly-phase AC/DC active power converter utilizes real-time geometric calculations to track the conduction angle, monitor status, and allow harmonic correction to the incoming frequency of the power output by a variable frequency power source in order to provide unity or near-unity power factor, low or controllable total harmonic distortion, improved frequency change tracking, and simplified error detection.

Thus, it can be seen that the objects of the invention have been satisfied by the structure and methods presented above. While in accordance with the Patent Statutes, only the best mode and preferred embodiment has been presented and described in detail, it is to be understood that the invention is not limited thereto or thereby. Accordingly, for an appreciation of the true scope and breadth of the invention, reference should be made to the following claims.