Controller for power convertor and motor driving system

A controller for a power convertor includes: a torque command value calculation module to calculate a first torque command value to a power convertor; a torque command limit module to receive the first torque command value and generate a second torque command value obtained by correcting the first torque command value so that the first torque command value is limited to a torque limiter range. The torque command limit module sets a width between the upper limit torque value and the lower limit torque value of the torque limiter range to be smaller as a fundamental wave output frequency of the power convertor increases at least in a speed region equal to or higher than a field weakening starting point.

FIELD

The present application relates to a controller for a variable voltage and variable frequency type power convertor and a motor driving system.

BACKGROUND

U.S. Pat. No. 9,281,772 discloses a controller for a power convertor. The controller can put, on a motor, a brake of good response. The following Non Patent Literature 1 discloses a flux observer for safe operation of a motor.

SUMMARY

However, in the systems of the above Patent Literature 1 and Non-Patent Literature 1, no field weakening is implemented. For this reason, an operation of the electric motor can be limited on the high speed range.

As a general technique of field weakening, there is a method of suppressing the influence of voltage saturation when the output frequency of the power converter increases, by lowering the flux command value in inverse proportion to the rotational speed. The fundamental output frequency of the power converter is also referred to as power supply angular frequency.

The flux command method in simply inverse proportion to a frequency does not consider the output limit of the convertor. Therefore, the control stability is potentially degraded when output torque in a field weakening region is increased.

The present application is intended to solve the above-described problem and provide a controller for a power convertor, and a motor driving system that are capable of simultaneously achieving control stability and output torque increase in a field weakening region.

In the control device described in Non-Patent Literature 1, sensorless control is performed by obtaining the rotation speed using the stator flux estimated by the flux observer. However, this stator flux is subject to harmonics caused by PWM control and current measurement noise. For this reason, the stator flux has a low S/N ratio because the flux amplitude decreases particularly in the field weakening region. For these reasons, the accuracy of the speed estimation declines and there is a possibility that stable operation cannot be performed.

The present application is also intended to provide a controller for a power convertor, and a motor driving system that are capable of highly accurately calculating an estimated speed value.

A first controller for a power convertor according to a first aspect of the present application includes: a torque command value calculation module configured to calculate a first torque command value to a power convertor based on a speed command value of a motor driven by the power convertor; a torque command limit module configured to receive the first torque command value and generate a second torque command value obtained by correcting the first torque command value so that the first torque command value is limited to a torque limiter range defined by an upper limit torque value calculated by a predetermined calculation formula for upper limit torque command value calculation and a lower limit torque value obtained by multiplying the upper limit torque value by a predetermined negative coefficient or zero; a flux command generation module configured to generate a stator flux command value in accordance with a fundamental wave output frequency of output from the power convertor; and an output voltage calculation module configured to calculate an output voltage command value of the power convertor based on the second torque command value and the stator flux command value. The torque command limit module calculates the upper limit torque value to be smaller as the fundamental wave output frequency increases at least in a speed region equal to or higher than a field weakening starting point.

A second controller for a power convertor according to a second aspect of the present application includes: a torque command value calculation module configured to calculate a torque command value to a power convertor based on a speed command value of a motor driven by the power convertor; a voltage command value calculation module configured to calculate a voltage command value to the power convertor based on the torque command value calculated by the torque command value calculation module; a flux estimation module configured to calculate estimated values of the stator flux and rotor flux of the motor in a subsequent control period based on the voltage command value to the power convertor and a measured stator current of the motor; and a motor speed estimation module configured to calculate an estimated value of a speed of the motor in a subsequent control period based on the estimated value of the rotor flux calculated by the flux estimation module.

A motor driving system according to a third aspect of the present application includes: a power convertor configured to drive a motor; and one of the first or second controller for a power convertor, which is configured to control the power convertor.

DESCRIPTION OF EMBODIMENTS

Embodiments of the present invention will be described in accordance with the accompanying drawings. It should be noted that in the drawings, the same or corresponding parts are denoted by the same reference signs. Overlapping description of such parts will be simplified or omitted as appropriate.

Symbols are explained below. Complex vectors are as follows.

Scalars are as follows.

P Pole number

P/2 Pole pair number

PstoredTime differentiation of magnetic energy accumulated in the inductance of the induction machine [W]

Keeddy current coefficient

General Description without Specifying Coordinate System (Arbitrary Coordinate System)

θangle [rad] ω=pθ General description without specifying the coordinate system

(Angle of u-Phase Rotor Winding Taken Counterclockwise with Respect to u-Phase Stator Winding)

Rotational Angular Speed of the Output Shaft

Rotational Angular of the Output Shaft

ω=pθ, but if the synchronization angular frequency is constant, it is also expressed as ωt=θ.

Mathematical Elements are as Follows.

j imaginary number

p differential operator

s Laplace operator

Re{ } complex real part

Each symbol of superscript has the following meaning.

fsstationary reference frame

fegeneral synchronous reference frame

frrotor reference frame

frasre-aligned stationary reference frame

f* command value

{dot over (f)} differential value

{circumflex over (f)} estimate value

Each subscript symbol has the following meaning.

First Embodiment

FIG. 1is a configuration diagram of a motor system to which a controller for a power converter according to an embodiment 1 of the present invention is applied.

InFIG. 1, a motor system1includes a motor2and a motor driving system3. For example, the electric motor2is an induction machine.

An output part of the motor2is connected to an input part of a load machine4. For example, the load machine4is an inertia load. A speed sensor119for detecting the rotational speed of the rotor is connected to the electric motor2.

An input part of the motor driving system3is connected to an output part of an AC power supply5. For example, the AC power supply5is a grid.

The motor driving system3includes a diode rectifier6, a capacitor7, an inverter8, a first current detector9a, a second current detector9b, and controller11.

The diode rectifier6converts, into DC power, the three-phase AC power supplied from the AC power supply5. If necessary, the diode rectifier6may be replaced with a PWM converter.

The capacitor7is provided across a DC link on the output side of the diode rectifier6. The capacitor7is used to smooth the DC voltage applied to the DC link.

The inverter8is converts the DC power supplied from the diode rectifier6into three-phase AC power for driving the electric motor2. The inverter8is a voltage source inverter. The inverter8is subjected to variable voltage variable frequency (VVVF) control through pulse width modulation (PWM) control.

The power conversion circuit of the inverter8is formed of three arms. One of the arms includes an upper arm and a lower arm. The upper and lower arms are each formed of at least one switching element.

The first current detector9ais provided at the v-phase of the output side of the inverter8. The first current detector9adetects the v-phase stator current Ivs. The second current detector9bis provided at the w-phase of the output side of the inverter8. The second current detector9bdetects the w-phase stator current Iws.

The controller11includes a speed controller12, a DB-DTFC calculation module14, a first coordinate conversion module15, PWM controller16, a second coordinate conversion module17, a current/flux estimation module20, a speed/phase estimation module21, a torque command limit module13, an appropriate flux command generation module18, a power supply angular frequency calculation module19, and a first slip angular frequency estimation module32.

The speed controller12is a torque command value calculation module. The speed controller12calculates the torque command value Tem1* so that the rotor mechanical angular speed estimated value {circumflex over (ω)}rm-r, which is detected by the speed/phase estimation module21, will follow the rotor mechanical angular speed command value ωrm* obtained from an external device.

Here, the “dqs-axes” is such that the U phase and the q axis coincide of a stator with each other in a stationary axis system, and distributes the three phase components to two phases of the d axis and the q axis orthogonal to each other. The second coordinate conversion module17, which will be described later, is a circuit that performs dq-axes conversion to replace the input signal with the biaxial signal. The first coordinate conversion module15, which will be described later, is an inverse conversion circuit that restores the two phase signals converted into the d axis and the q axis in the stationary axis system to three phase signals in the stationary coordinate system.

For example, a superscript symbol “S” of the stator dqs-axes flux value λqdsSrepresents the stationary coordinate system. A subscript symbol “qds” of the stator dqs-axes flux value λqdsSrepresents “two phase components” of the stator flux. That is, the stator dqs-axes flux value λqdsSrepresents the d-axis component λdsSof the stator flux value and the q-axis component λqsSof the stator flux command value. In the following description, symbols with the subscript suffix “qds” are the symbols representing both the d-axis component and the q-axis component of the stator flux.

On the other hand, a superscript symbol “S” of the rotor dqs-axes flux value λqdrSrepresents the stationary coordinate system. A subscript symbol “qdr” of the rotor dqs-axes flux value ΔqdrSrepresents the d-axis component λdrSof the rotor flux value and the q-axis component λqrSof the rotor flux value. In the following description, symbols with the subscript suffix “qdr” are the symbols representing both the d-axis component and the q-axis component of the rotor flux.

The torque command limit module13calculates the second torque command value Tem* based on the first torque command value Tem1* calculated by the speed control module12, the stator flux command value λs_optgenerated by the appropriate flux command generation module18, and the power supply angular frequency ωecalculated by the power supply angular frequency calculation module19. Details of the torque command limit module13will be described later with reference toFIG. 2.

The DB-DTFC calculation module14calculates the stator dqs-axes voltage command value VqdsS* based on the second torque command value Tem*, the flux command value λs*, the stator dqs-axes flux estimated value {circumflex over (λ)}dsSthe rotor dqs-axes flux estimated value {circumflex over (λ)}qdrS, and the rotor mechanical angular speed estimated value {circumflex over (ω)}rm-restimated by the speed/phase estimation module21which is described later. The flux command value λs* is a stator flux command value λs_optgenerated by the appropriate flux command generation module18. The DB-DTFC calculation module14employs, as a control method, a deadbeat direct torque & flux control (DB-DTFC) method.

The first coordinate conversion module15converts the stator dqs-axes voltage command value VqdsS* into three-phase stator voltage command values Vus*, Vvs*, and Vws*. The conversion performed by the first coordinate conversion module15is an inverse conversion of the dqs-axes transformation.

The PWM controller16converts the three-phase stator voltage command values Vus*, Vvs*, and Vws* into gate pulses for the inverter8based on the pulse width modulation. The PWM controller16outputs the gate pulses to the inverter8.

The second coordinate conversion module17converts the stator currents Ivs, Iws into a stator dqs-axes current measured value iqdss. The conversion performed by the second coordinate conversion module17is “dqs-axes conversion”.

Three-phase to two-phase conversion by dqs-axes conversion is performed based on the following equation. For example, if the currents of three phases are Iu, Iv, Iwand the currents after two-phase conversion are Ids, Iqs, the following equation is obtained.
Iu+Iv+Iw=0
Iqs=Iu
Ids=(Iu+2Iw)/√3

This conversion is somewhat different from Clarke conversion which is a general three-phase to two-phase conversion.

The appropriate flux command generation module18generates the stator flux command value λs_optby performing calculation on an appropriate flux command based on the power supply angular frequency ωecalculated by the power supply angular frequency calculation module19. The stator flux command value λs_opthaving an appropriate value is input to the DB-DTFC calculation module14as the flux command value λs*. The appropriate flux command generation module18will be described later in detail with reference toFIG. 2.

The power supply angular frequency calculation module19calculates the power supply angular frequency ωebased on the rotor mechanical angle estimated value {circumflex over (ω)}rm-restimated by the speed/phase estimation module21and the slip angular frequency estimated value {circumflex over (ω)}slestimated by the first slip angular frequency estimation module32. The relationship between the detected rotational speedωrmfrom the speed sensor119and the estimated value ωrm-rwill be described later with reference toFIG. 3. Details of the power supply angular frequency calculation module19will be described later with reference toFIG. 2.

It should be noted that the output value of the speed sensor119is not the instantaneous value but the average speed between the detection timings, and thus the output value of the speed sensor119is expressed asωrmusing the symbol indicating the average value.

Here, the stator dqs-axes current measured value iqdssand the stator dqs-axes voltage measured value Vqdssare values obtained by dqs converting the three phase measured values. The total sum of the instantaneous values of the three-phase currents, which is the output of the inverter8, is zero. By using this property, in the first embodiment, two-phase current is detected, and for the other phase, a value obtained by inverting the sign of the added value of two-phase current is used.

The current/flux estimation module20calculates the stator dqs-axes flux estimated value {circumflex over (λ)}qdsSand the rotor dqs-axes flux estimated value {circumflex over (λ)}qdrSas a flux estimation module based on the rotor electrical angle estimated value {circumflex over (θ)}restimated by the speed/phase estimation module21, the voltage command value VqdsS*, the stator dqs-axes current measured value iqdss, the rotor mechanical angular speed estimated value {circumflex over (ω)}rm-r.

The speed/phase estimation module21basically performs a calculation such that a torque command value is divided by a moment of inertia of the motor2to calculate an acceleration value, the acceleration value is integrated to calculate a speed, and the speed is integrated to calculate a phase. The delay operator is used in the speed/phase estimation module21.

The rotational speedωrmof the motor2, which is an output of the speed sensor119, is inputted into the subtraction module21a. The subtraction module21acalculates a difference between the rotational speedωrmand an output of the delay calculation module21m. The calculated difference is transmitted to the integral module21band the proportional module21e. The integral module21bintegrates an output of the subtraction module21a. The integrated value by the integral module21bis transmitted to the integral module21cand the proportional module21d.

The integral module21cis an integration circuit having a gain Kio. The integral module21coutputs a calculation result to the adder module21f. The proportional module21dperforms amplification by a gain Kso. The proportional module21doutputs a calculation result to the adder module21f. The proportional module21eperforms amplification by a gain bo. The proportional module21eoutputs a calculation result to the adder module21f.

The adder module21fperforms addition of the output from the integral module21c, the output from the proportional module21d, and the output from the proportional module21e, and outputs the calculation result to the adder module21g. The adder module21gperforms addition of the second torque command value Tem*, which is the output from the torque command limit module13, and the output from the adder module21f, and then outputs a calculation result to the acceleration value calculation module21h.

The acceleration value calculation module21hdivides an inputted value by the moment of inertia estimated value ĵpof the motor2, and then outputs the estimated acceleration value to the integral module21j. The integral module21jintegrates the inputted estimated acceleration value to estimate the rotor mechanical angular speed estimated value {circumflex over (ω)}rm-r.

The rotor mechanical angular speed estimated value {circumflex over (ω)}rm-ris transmitted to the integral module21nand the average value calculation module21k. The rotor mechanical angular speed estimated value {circumflex over (ω)}rm-ris also transmitted to the outside of the speed/phase estimation module21. The outside of the speed/phase estimation module21is for example the DB-DTFC calculation module14. Further, the rotor mechanical angular speed estimated value {circumflex over (ω)}rm-ris also transmitted to the current and flux estimation module20, the power supply angular frequency calculation module19, and the speed control module12.

The integral module21nintegrates the rotor mechanical angular speed estimated value {circumflex over (ω)}rm-rto calculate the estimated phase (mechanical angle) {circumflex over (θ)}rm. The estimated phase (mechanical angle) {circumflex over (θ)}rmis inputted to the proportional module21p. The proportional module21pmultiples the estimated phase (mechanical angle) {circumflex over (θ)}mby the pole number P/2 of the motor2to calculate the estimated phase (electrical angle) {circumflex over (θ)}r. The estimated phase (electrical angle) {circumflex over (θ)}ris outputted to the outside of the speed/phase estimation module21. The outside of the speed/phase estimation module21is for example the current and flux estimation module20.

The average value calculation module21kcalculates the average value {circumflex over (ω)}rmwith respect to the rotor mechanical angular speed estimated value {circumflex over (ω)}rm-rby calculating an average of an estimated value in the present calculation period and another estimated value before one calculation period, and then outputs the calculation result to the delay calculation module21m. The delay calculation module21mperforms a calculation to delay the average value {circumflex over (ω)}rmby one calculation period, and then outputs a calculation result to the subtraction module21aas a subtraction value. It should be noted that, the configuration inFIG. 3may be modified in other system, and the output from the speed sensor119may be a low pass filter, and an output from the low pass filter may be regarded as the rotor mechanical angular speed estimated value {circumflex over (ω)}rm-r.

Voltage, current, and flux always change in the dqs-axes coordinate in the steady state. On the other hand, speed is constant, and the influence of the phase delay need not to be considered. By using the motion observer, phase and speed information with little noise can be obtained without phase delay.

FIG. 2is an enlarged configuration diagram of part of the controller for a power convertor according to the first embodiment. As illustrated inFIG. 2, this controller11includes the speed control module12, the torque command limit module13, the appropriate flux command generation module18, and the power supply angular frequency calculation module19.

The torque command limit module13includes a first block13a, a second block13b, a third block13c, a fourth block13d, a fifth block13e, and a sixth block13f.

The first block13acalculates an upper limit torque command value Te_maxbased on the stator flux command value λs_optin accordance with Expression (30) to be described later. Expression (30) is a predetermined first calculation formula for upper limit torque command value calculation. A second-order polynomial including the second-order term (λs_opt2) of the stator flux command value λs_optand the first-order term (λs_opt) thereof is derived by expanding the expression in the square root in Expression (30).

The second block13bcalculates the upper limit torque command value Te_maxbased on the stator flux command value λs_optin accordance with Expression (36) to be described later. Expression (36) is a predetermined second calculation formula for upper limit torque command value calculation. Expression (36) is a monomial obtained by multiplying the second-order term (λs_opt2) of the stator flux command value by a predetermined coefficient. The predetermined coefficient is

The third block13ccalculates a boundary speed ωe_cin accordance with Expression (34) to be described later. The boundary speed ωe_cis a speed predetermined to be higher than a field weakening starting point, and determined by Expression (34). The field weakening starting point may be, for example, the rated speed of a motor.

The fourth block13ddetermines a speed region based on the power supply angular frequency ωeand the boundary speed ωe_c. The speed region is divided into a normal operation region, Region I, and Region II. Region I is a region between the boundary speed ωe_cand the field weakening starting point. Region II is a region higher than the boundary speed ωe_c.

Specifically, the fourth block13ddetermines the speed region in accordance with the power supply angular frequency ωeas follows. The field weakening starting speed is defined as ωbase. When ωe<ωbase, it is determined the speed is in the normal operation region. When ωbase≤ωe<ωe_c, it is determined that the speed is in Region I (a first field weakening region). When ωe_c≤ωe, it is determined that the speed is in Region II (a second field weakening region). The fourth block13doutputs “0” when the current speed is in the normal operation region or Region I. The fourth block13doutputs “1” when the current speed is in Region II.

When the output from the fourth block13dis “0”, the fifth block13eselects the upper limit torque command value Te_maxoutput from the first block13a. When the output from the fourth block13dis “I”, the fifth block13eselects the upper limit torque command value Te_maxoutput from the second block13b. The fifth block13etransmits the selected upper limit torque command value Te_maxto the sixth block13f.

The sixth block13fdetermines a torque limiter range by using the upper limit torque command value Te_maxtransmitted from the fifth block13e. The torque limiter range is a range defined by an upper limit torque value (+Te_max) and a lower limit torque value (−Te_max). The upper limit torque value is the upper limit torque command value Te_max. The lower limit torque value is a value obtained by multiplying the upper limit torque value by a negative coefficient. The negative coefficient is a predetermined value, and may be “−1” or a value other than minus one. Also, instead of this negative coefficient, the lower limit torque value may be set to zero by multiplying zero by the upper limit torque value.

When the first torque command value Tem1* is in the torque limiter range, the sixth block13fsubstitutes the first torque command value Tem1* directly into the second torque command value Tem*. When the first torque command value Tem1* is larger than the upper limit torque value +Te_max, the sixth block13fsubstitutes the upper limit torque value +Te_maxinto the second torque command value Tem*. When the first torque command value Tem1* is smaller than the lower limit torque value −Te_max, the sixth block13fsubstitutes the lower limit torque value −Te_maxinto the second torque command value Tem*.

In this manner, the sixth block13ffunctions as a filter that allows only a torque command value in the torque limiter range to pass through. Accordingly, the sixth block13fgenerates the second torque command value Tem*. The second torque command value Tem* is a value obtained by correcting the first torque command value Tem1* so that the first torque command value Tem1* is limited to the torque limiter range.

The appropriate flux command generation module18includes a first block18aand a second block18b. The appropriate flux command generation module18generates the stator flux command value λs_optin accordance with the power supply angular frequency ωe.

The first block18acalculates the stator flux command value λs_optin accordance with Expression (29) to be described later. The first block18acalculates a value obtained by dividing a stator voltage maximum value Vmaxby the power supply angular frequency ωe.

The second block18blimits the stator flux command value λs_optcalculated by the first block18ato a certain range. The second block18bemploys a predetermined upper limiter flux value and limits the stator flux command value λs_optto be equal to or smaller than the upper limiter flux value. The upper limiter flux value may be the rated stator flux λrateas one of parameters of the motor in the embodiment.

The power supply angular frequency calculation module19calculates the rotor electrical angular speed estimated value {circumflex over (ω)}rby multiplying the rotor mechanical angular speed estimated value {circumflex over (ω)}rm-rcalculated by the speed and phase estimation module21by P/2. The value P is the pole number of the motor2. The power supply angular frequency calculation module19calculates the power supply angular frequency ωeby adding the rotor mechanical angular speed estimated value {circumflex over (ω)}rm-rand a slip angular frequency estimated value {circumflex over (ω)}sloutputted from a first slip angular estimation module32.

The slip angular frequency estimated value {circumflex over (ω)}slmay be estimated by Expressions (43) or (44), as described later. In the first embodiment, the rotor dqs-axes flux estimated value λqdsSoutputted from the current/flux estimation module20and the rotor dqs-axes current measured value iqdssoutputted from the second the second coordinate conversion module17are inputted to the first slip angular estimation module32. The first slip angular estimation module32performs a calculation based on Expression (43) to estimate the slip angular frequency estimated value {circumflex over (ω)}sl.

Device Operation According to the First Embodiment

The following describes a theory for increasing an output torque in a field weakening region in DB-DTFC control. Expressions (1) to (4) are equations for an induction machine in an optional coordinate system.
Vqds=Rsiqds+jωλqds+pλqds(1)
0=Rriqdr+j(ω−ωr)λqdr+pλqdr(2)
λqds=Lsiqds+Lmiqdr(3)
λqdr=Lriqdr+Lmiqds(4)

A stator-side voltage equation in a synchronous coordinate system (synchronous with the power supply frequency) of the induction machine can be given by Expression (5).
Vqdse=Rsiqdse+jωeλqdse+pλqdse(5)

Any differential term can be omitted in a steady state, and thus Expression (5) becomes Expression (6).
Vqdse=Rsiqdse+jωeλqdse(6)

The term of the power supply angular frequency ωeis dominant in a high speed range, and thus finally, Expression (7) is obtained.
Vqdse=jωeλqdse(7)

It is clear from Expression (7) that a voltage upper limit is reached as the speed increases. In such a case, ωecan be increased by decreasing the flux. This is called field weakening.

One of features of the embodiment is a method of determining a flux command for achieving an increased torque in the high frequency region.

1. Output Torque Increasing Technique

Studies performed by the inventor of the present application have found a novel torque output increasing technology, the contents of which will be described below.

1.1 Stator Flux Limit with the Voltage and Current Limits Taken into Account

1.1.1 Restriction of Stator Voltage (Voltage Limit) by Maximum Output Voltage of Inverter

The stator voltage maximum value Vmaxis limited by a DC voltage Vdcof the inverter and a modulation method. The stator voltage maximum value Vmaxis Vsmax=Vdc/2 for sinusoidal wave pulse width modulation (SPWM), or Vsmax=Vdc/√{square root over (3)} for space vector pulse width modulation (SVPWM). If necessary, in deriving the maximum value Vsmax of the stator voltage, an influence of a forward voltage drop or a dead time of switching elements constituting the inverter8may be taken into consideration.

Another point is, if necessary, a coefficient slightly smaller than 1, for example 0.98, can be used to calculate Vsmax to account for the voltage drop on the stator resistance.

Expression (7) gives a dqs-axes complex vector. Expression (8) is a voltage limit expression obtained from Expression (7).

FIG. 4is a graph illustrating a stator flux locus for each power supply (operation) frequency with a voltage limit taken into account. A voltage limit is illustrated as a circle inFIG. 4. The radius of each circle inFIG. 4represents an allowable stator flux amplitude. The radius (upper limit of the allowable stator flux amplitude) decreases as the power supply (operation) angular frequency ωeincreases. The vertical axis expresses the stator d-axis flux on general synchronous reference frame λdse, and the horizontal axis expresses the stator q-axis flux on general synchronous reference frame λqseinFIG. 10fromFIG. 4respectively.

1.1.2 Limit of Stator Current by Allowable Output Current of Inverter (Current Limit)

In addition to the voltage limit, another limit is placed on the stator flux amplitude. This limit is an allowable output current limit of the inverter8and allowable current through the motor

2. The Maximum Allowable Current Ismaxis Expressed by Expression (9).
(iqse)2+(idse)2≤Ismax2(9)

To use the stator flux in control, a relational expression between the stator current and the stator flux is derived. In the stationary state, Expression (2) as a voltage equation for the rotor becomes Expression (10).
0=Rriqdre+j(ωe−ωr)κqdre(10)

When Expression (10) is rewritten for the q-axis and d-axis components separately, Expressions (11) and (12) below are obtained.
0=Rriqre+(ωe−ωr)λdre(11)
0=Rridre+(ωe−ωr)λqre(12)

Expressions (11) and (12) are simplified by performing coordinate transformation on the rotor flux so that Expressions (13) and (14) are obtained.
λdre=λr(13)
λqre=0  (14)

From Expressions (12) and (14), a rotor-side d-axis current on the synchronous coordinate system is given by Expression (15).
0=idre(15)

Expression (4) is rewritten in a manner divided for the q and d axes.
λqse=Lsiqse+Lmiqre(16)
λdse=Lsidse+Lmidre(17)

From Expressions (15) and (17), Relation Expression (18) between the stator current and the stator flux on the d-axis side can be obtained.
λdse=Lsidse(18)

Then, a relational expression between the stator current and the stator flux on the q-axis side is obtained. Expression (3) is rewritten in a manner divided for the d and q-axes.
λqre=Lmiqse+Lriqre(19)
λdre=Lmidse+Lridre(20)

Expression (21) can be obtained from Expressions (14) and (19).

Lastly, the relational expression between the stator current and the stator flux on the q-axis side can be obtained from Expressions (16) and (21). In the expression, a represents a leakage coefficient listed in a symbol list.

The current-voltage relation expressions (18) and (22) on the d-axis and q-axis sides are substituted into the current limit expression (9) to finally obtain a current limit expression (23).

When illustrated on a stator flux plane, the expression depicts an ellipse as illustrated inFIG. 5.FIG. 5is a graph illustrating a stator flux locus with a current limit taken into account.

1.1.3 Voltage and Current Limits

FIG. 6is a graph illustrating stator flux loci with the voltage and current limits taken into account. As described above in Sections 1.1 and 1.2, the voltage and current limits of the inverter depict a circle and an ellipse, respectively, on the stator flux plane. These results indicate that, in theory, the stator flux exists in a hatched region common to the voltage limit circle and the current limit ellipse illustrated inFIG. 6. Thus, to increase the output torque in the field weakening region, an appropriate combination of d-axis and q-axes flux command vectors need to be selected from the hatched region.

1.2 Stator Flux for Achieving Increased Torque

As described in the previous section, a combination of (d-axis and q-axes) stator flux command vectors are limited by the current and voltage limits of the inverter. Achievable torque differs between stator flux command vectors. The present section derives stator flux for achieving increased torque.

A torque formula is expressed as Expression (24). This formula is expressed only with the stator flux to depict torque on the dqs-axes plane of the stator flux.

Expression (25) can be obtained from Expressions (15), (18), and (20).

Torque Formula (26) expressed only with the stator flux can be obtained from Expressions (14), (24), and (25).

Expression (26) indicates that the stator flux locus depicts a hyperbola on the dqs-axes flux plane when the torque is constant.

FIG. 7is a graph illustrating voltage and current limits and torque curves at a power supply frequency of 1.2 pu.FIG. 7exemplarily illustrates three torque curves having different sizes in addition to voltage and current limits at a power supply frequency of 1.2 pu.

When the power supply frequency is 1.2 pu, the number of combinations of stator fluxes satisfying the voltage and current limits is infinite for TL=1.0 pu. However, the number of such combinations is one for TL=1.7 pu. This is a maximum torque achievable in theory. Both limits cannot be satisfied for TL=2.3 pu, and thus this torque cannot be achieved.

The radius of the voltage limit circle decreases as the power supply frequency increases. Accordingly, no torque curve exists in a region common to the current limit ellipse and the voltage limit circle. A maximum torque is achieved at a contact point between a torque curve and the voltage limit circle. This point is a point of transition from Region I to Region II, which will be described later.

Specifically, in a particular region, the stator flux is limited by two limits, namely, the current and voltage limits. In the other region, however, the stator flux is limited by one limit, namely, the voltage limit. A flux command needs to be determined in each region due to this limit difference.

The above-described two field weakening regions are referred to as Region I and Region II. The following describes Region I, Region II, and a boundary point between these regions. The relation between each region and the speed is listed below with description of terms.

ωe<ωbase: Normal operation region

ωe_c: Boundary speed between Regions I and II

ωbase: Field weakening start speed

The field weakening starting point ωbaseis determined by the stator voltage and the rated flux. In the first embodiment, it is assumed that field weakening automatically starts at voltage saturation.

1.2.2 Stator flux in Region I (ωbase<ωe<ωe_c)

FIG. 8is a graph illustrating a maximum torque curve with the voltage and current limits taken into account at different speeds in Region I. Region I corresponds to a case in which the power supply (operation) angular frequency ωeexceeds a rated speed.

As illustrated inFIG. 8, when the operation frequency is not too high, an intersection point between the current limit ellipse and the voltage limit circle exists above the straight line of λdse=λqse. This operation frequency band is defined to be Region I.

In Region I, a stator flux for achieving the maximum torque is at the intersection point between the current limit ellipse and the voltage limit circle. InFIG. 8, the maximum torque is achieved at operation point A for ωe=1.2 pu and at operation point B for ωe=1.8 pu.

Stator flux command vectors satisfying the intersection point are given by Expressions (27) and (28) due to the voltage and current limits.

The amplitude of the stator flux is limited by Expression (29) as the radius of the voltage limit circle.

Expression (30), which calculates the maximum torque in the field weakening region I, is obtained from Expressions (26) to (29).

Te_max=32⁢P2⁢Lm2σ⁢⁢Ls2⁢Lr⁢λq⁢s_opte⁢λd⁢s_opte=32⁢P2⁢Lm2(1-σ2)⁢Ls2⁢Lr⁢(Ls2⁢Ismax2-λs_opt2)⁢(λs_opt2-σ2⁢Ls2⁢Ismax2)(30)
1.2.3 Stator flux (ωe_c<ωe) in Region II

The intersection point between the ellipse and the circle is below the straight line of λdse=λqseat a higher speed. No intersection point exists at a further higher speed. An operation frequency band in which these phenomena occur is defined to be Region II.

FIG. 9illustrates a condition on the boundary speed between Regions I and II.FIG. 9is a graph indicating the condition on the boundary speed between Regions I and II. The boundary speed ωe_cis calculated by Expressions (31) to (33) below.

Here, among the stator flux λqds_ce, the λqs_cerepresents the q-axis component of the stator flux at the boundary speed, and the λds_cerepresents the d-axis component of the stator flux at the boundary speed.

The boundary speed ωe_cis finally obtained by Expression (34).

FIG. 10is a graph illustrating maximum torque curves with the voltage and current limits taken into account at different power supply angular frequencies in Region II. The maximum torque is achieved at points C and D illustrated inFIG. 10for different operation speeds, respectively. A stator flux command vector for achieving the maximum torque in Region II can be obtained by Expression (35).

Similarly to Region I, the stator flux command amplitude is limited by Expression (29). The maximum torque achievable in the field weakening region H is expressed as Expression (36). Expression (36) is derived from Expressions (26), (29), and (35).

One of differences between Regions I and II is that the current is constantly lower than the current limit (upper limit) due to a condition on the maximum torque achievable in Region II. As a result, a stator flux amplitude for achieving the maximum torque is determined by the radius of the voltage limit circle in both of Regions I and IL

1.2.4 Control Block for Torque Increasing

The entire image of the above-described control is illustrated inFIGS. 1 to 3. The stator flux command is generated by the first block18ainside the appropriate flux command generation module18in accordance with Expression (29) based on the power supply frequency and the stator voltage.

However, a certain limit is placed on the upper limit of the stator flux command by the second block18b. An achievable torque command is calculated by the torque command limit module13by using the stator flux command in accordance with Expressions (30) and (36). The achievable torque differs between the field weakening region I and the field weakening region II as described above.

According to the first embodiment above described, operation described below is achieved.

The first block18aof the appropriate flux command generation module18calculates the stator flux command value λs_optto be smaller as the power supply angular frequency ωeincreases.

The second block18blimits the stator flux command value λs_optto a constant value at the rated flux (λrate) in a low speed range in which the power supply angular frequency ωeis small. However, the second block18bplaces no limit in a high speed range in which the power supply angular frequency ωeis large. The second block18bplaces no limit at least in a speed region equal to or higher than the field weakening starting point. Without the limit by the second block18b, the stator flux command value λs_optis calculated to be smaller as the power supply angular frequency ωeincreases.

As understood from Expressions (30) and (36), the upper limit torque command value Te_maxis calculated to be smaller as the stator flux command value λs_optis smaller. The width between the upper limit torque value (+Te_max) and the lower limit torque value (−Te_max) decreases as the upper limit torque command value Te_maxis smaller.

The torque command limit module13can calculate the upper limit torque command value Te_maxto be smaller as the power supply angular frequency ωeincreases in a somewhat high speed range. When the upper limit torque command value Te_maxis calculated to be smaller, the width between the upper limit torque value (+Te_max) and the lower limit torque value (−Te_max) is set to be smaller.

Through such operation, the torque limiter range can be set to be smaller as the power supply angular frequency ωeincreases at least in the field weakening regions I and II. An allowable torque range in which the control stability can be maintained tends to be smaller as the power supply angular frequency ωeincreases. According to the first embodiment, the torque limiter range is dynamically adjusted in accordance with such a tendency.

Torque increase can be performed without degrading the control stability because a torque command value changes only in the torque limiter range adjusted to an appropriate range. Accordingly, when the motor is operated fast in a field weakening region, the control stability and the output torque increase can be simultaneously achieved.

In the first embodiment, Expression (30) of the first block13ais applied at a speed lower than the boundary speed ωe_c. The upper limit torque command value Te_maxcan be changed at an appropriate tendency in accordance with the second-order polynomial of the stator flux command value λs_opt.

In the first embodiment, Expression (36) of the second block13bis applied at a speed equal to or higher than the boundary speed ωe_c. The upper limit torque command value Te_maxcan be appropriately changed in accordance with the monomial of the second-order term (λs_opt2) of the stator flux command value at a tendency different from that of Expression (30).

Accordingly, the upper limit torque command value Te_maxcan be changed at an appropriate tendency in each of the field weakening regions I and II.

In the embodiment, the torque command limit module13includes the fifth block13e. With this configuration, the upper limit torque command value Te_maxcalculated by the first block13aand the upper limit torque command value Te_maxcalculated by the second block13bcan be selectively switched in accordance with the power supply angular frequency ωe.

In the first embodiment, the fourth block13dis provided. With this configuration, the first block13a, in other words, Expression (30) is applied in both of Region I and the normal operation region. Accordingly, calculation of the upper limit torque command value Te_maxcan be seamlessly performed through the normal operation region and the field weakening region.

In the first embodiment, the appropriate flux command generation module18includes the second block18b. When Expression (29) is applied in a low speed range in which the power supply angular frequency ωeis small, the stator flux command value λs_optis calculated to be excessively large. The second block18bplaces an upper limiter to prevent the stator flux command value λs_optfrom being calculated to be excessively large.

Next, an example of the controller is described with the use ofFIG. 20.FIG. 20is a hardware configuration diagram of the controller for a power converter according to the embodiment 1 of the present invention. The hardware configuration inFIG. 20may also be adapted to the second embodiment and the third embodiment described later.

As illustrated inFIG. 20, each function of the controller9is executed by the processing circuit. The processing circuit includes a processor30aand a memory30b.

For example, the processor30ais a central processing unit (CPU), e.g., a central processing device, a processing device, a microprocessor, a microcomputer, a processor or a digital signal processor (DSP).

For example, the memory30bis a non-volatile or volatile semiconductor memory such as RAM, ROM, flash memory, EPROM, EEPROM, or magnetic disk, flexible disk, optical disk, compact disk, mini-disk or DVD.

In the processing circuit, a program stored in the memory30bis executed by the processor30a.

Second Embodiment

Configuration of Device According to Second Embodiment

FIG. 11is a configuration diagram of a motor system to which a controller for a power convertor according to a second embodiment is applied. The second embodiment differs from the first embodiment in that a motor system in the second embodiment is a sensor-less system including no speed sensor, whereas the motor system according to the first embodiment is a sensor-equipped system including the speed sensor119.

In the second embodiment, the speed sensor119is omitted, and the speed and phase estimation module21is replaced with a speed and phase estimation module121. In addition, the first slip angular frequency estimation module32is replaced by the second slip angular frequency estimation module33. Any other configuration is the same as that of the first embodiment.

The second slip angular frequency estimation module33uses the second torque command value Tem* and the rotor dqs-axes flux estimated value {circumflex over (λ)}qdrSas input values to calculate the slip angular frequency estimation values {circumflex over (ω)}slbased on Equation (44). The slip angular frequency estimated value {circumflex over (ω)}slis inputted to the speed/phase estimation module121and the power supply angular frequency calculation module19.

FIG. 12is a block diagram of the main sections of the controller for the power converter according to the embodiment 1 of the present invention. Next, the current/flux estimation module20and the speed/phase estimation module21will be described with the use ofFIG. 12.

As illustrated inFIG. 12, the current/flux estimation module20includes a current observation module22, a first flux estimation module23and a second flux estimation module24.

The current observation module22calculates a stator dqs-axes current estimated value îqdssin a subsequent control period based on the voltage command value VqdsS*, the stator dqs-axes current measured value iqdss, a rotor mechanical angular speed estimated value {circumflex over (ω)}rm-rand a rotor dqs-axes flux estimated value λqdrS.

The stator dqs-axes current measured value iqdssis the output of the second coordinate conversion module17. The rotor mechanical angular speed estimated value {circumflex over (ω)}rm-ris an output of the speed/phase estimation module121. The rotor dqs-axes flux estimated value {circumflex over (λ)}qdrSis an output of the second flux estimation module24. The voltage command value VqdsS* is an output from the DB-DTFC calculation module14.

On this occasion, the proportional gain K3, the integral gain K4, the estimated value {circumflex over (R)}eqof the equivalent resistance, the estimated value {circumflex over (L)}rof the rotor winding inductance, the estimated value {circumflex over (L)}mof the magnetizing inductance, the rotor resistance {circumflex over (R)}rand the imaginary number j, the rotor mechanical angular speed estimated value {circumflex over (ω)}rm-r, the equivalent time constant τeq, the control period T and the delay operator z−1are used.

Specifically, the current observation module22includes a first block22a, a second block22b, a third block22c, a fourth block22d, a fifth block22e, a sixth block22f, and a seventh block22g.

The first block22acalculates a value obtained by subtracting the stator dqs-axes current estimated value îqdssfrom the stator dqs-axes current measured value iqdss. The stator dqs-axes current estimated value îqdssis the output of the seventh block22g.

The output of the first block22ais inputted into the second block22band the third block22c. The second block22bis a proportional circuit of gain K3. The third block22cis an integrating circuit of gain K4. The fourth block22dcalculates a complemented value from a rotor mechanical angular speed estimated value {circumflex over (ω)}rm-rand a rotor dqs-axes flux estimated value {circumflex over (λ)}qdrS. The rotor mechanical angular speed estimated value {circumflex over (ω)}rm-ris an output of the speed/phase estimation module121. The rotor dqs-axes flux estimated value {circumflex over (λ)}qdrSis an output of the second flux estimation module24.

The fourth block22dcalculates a value obtained by multiplying the rotor dqs-axes flux estimated value {circumflex over (λ)}qdrSby the transfer coefficient G1expressed by the following Expression (37).

The fifth block22ecalculates a value obtained by adding the voltage command value VqdsS*, the value calculated by the second block22b, the value calculated by the third block22c, the value calculated by the fourth block22d.

The sixth block22fcalculates a value obtained by dividing the value calculated by the fifth block22eby the estimated value {circumflex over (R)}eqof the equivalent resistance.

The output of the sixth block22fis input into the seventh block22g. The seventh block22gcalculates a transfer function expressed in the Expression (38). The seventh block22goutputs a stator dqs-axes current estimated value îqds.

The stator dqs-axes current estimated value îqdssis input into the first block22aand the second flux estimation module24.

The first flux estimation module23calculates a rotor dqs-axes flux estimated value λqdrS* based on the rotor electrical angle estimated value {circumflex over (θ)}rand the stator dqs-axes current measured value iqdss. The rotor electrical angle estimated value {circumflex over (θ)}ris an output of the speed/phase estimation module121. The stator dqs-axes current measured value iqdssis an output of the second coordinate conversion module17.

On this occasion, the estimated value {circumflex over (L)}mof the magnetizing inductance of the motor2, the rotor time constant τrof the motor2, the control period T and the delay operator z−1are used.

Specifically, the first flux estimation module23includes a first block23a, a second block23b, and a third block23c.

The first block23aconverts the stator dqs-axes current measured value iqdssto the value of the rotor coordinate system by the rotor electrical angle estimated value {circumflex over (θ)}r.

The output of the first block23ais the input of the second block23b.

The second block23bmultiplies the value calculated by the first block23aby the transfer function G3expressed by the following Expression (39) to calculate the stator dqs-axes flux estimated value of the primary hold.

The output of the second block23bis the input of the third block23c.

The third block23cconverts the output of the second block23bto the value of the rotor coordinate system by the rotor electrical angle estimated value {circumflex over (θ)}r. The output of the second block23bis the rotor dqs-axes flux estimated value {circumflex over (λ)}qdrS*. The rotor dqs-axes flux estimated value λqdrS* is inputted into second flux estimation module24.

The second flux estimation module24calculates a stator dqs-axes flux estimated value {circumflex over (λ)}qdsSand a rotor dqs-axes flux estimated value {circumflex over (λ)}qdrSin a subsequent control period based on the voltage command value VqdsS* outputted from the DB-DTFC calculation module14, the stator dqs-axes current measured value iqdssoutput of the second coordinate conversion module17, the stator dqs-axes current estimated value îqdssestimated by the current observation module22, the rotor dqs-axes flux estimated value λqdrS* estimated by the first flux estimation module23.

On this occasion, the proportional gain K1, the integral gain K2, the estimated value {circumflex over (R)}sof the stator resistance, the leakage factor σ, the estimated value {circumflex over (L)}rof the rotor winding inductance, the estimated value of {circumflex over (L)}sof the stator winding inductance, the estimated value {circumflex over (L)}mof the magnetizing inductance, the control period T and the delay operator z−1are used.

Specifically, the second flux estimation module24includes a first block24a, a second block24b, a third block24c, a fourth block24d, a fifth block24e, a sixth block24f, a seventh block24g, and an eighth block24h, a ninth block24i, and a tenth block24j.

The first block24acalculates a value of the voltage drop by multiplying the stator dqs-axes current measured value iqdssby the estimated value {circumflex over (R)}sof the stator resistance.

The second block24bsubtracts the rotor dqs-axes flux estimated value {circumflex over (λ)}qdrSwhich is the output of the tenth block24jfrom the rotor dqs-axes flux estimated value {circumflex over (λ)}qdrS*.

The third block24ccalculates a value obtained by subtracting the value calculated by the first block24afrom the voltage command value VqdsS*.

The fourth block24d, the fifth block24e, and the sixth block24ffunction as a transition frequency determination unit24k. The transition frequency determining unit24kdetermines the transition frequency between the first flux estimation module23and the second flux estimation module24.

Specifically, the fourth block24dcalculates a value obtained by multiplying the value calculated by the second block24bby the proportional gain K1.

The fifth block24eis an integrating circuit of gain K2. The fifth block24ecalculates an integral of the output of the fourth block24d.

The sixth block24fcalculates a value of the input voltage of the motor2by adding the value calculated by the third block24c, the value calculated by the fourth block24dand the value calculated by the fifth block24e.

The seventh block24gcalculates the stator dqs-axes flux estimated value {circumflex over (λ)}qdsSby integrating the output of the sixth block24f.

The eighth block24hcalculates a value obtained by multiplying the stator dqs-axes current estimated value îqdssby the factor G4expressed by the following Expression (40).
G4=σ{circumflex over (L)}s(40)

The ninth block24icalculates a value obtained by subtracting the value calculated by the eighth block24hfrom the stator dqs-axes flux estimated value {circumflex over (λ)}qdsScalculated by the seventh block24g.

The tenth block24jmultiplies the value calculated by the ninth block24iby the factor G5expressed by the following Expression (41) to calculate the rotor dqs-axes flux estimated value {circumflex over (λ)}qdrS.

For example, in the case where the frequency of the rotor flux of the motor2is lower than the transition frequency, the first flux estimation module23is dominant. For example, in the case where the frequency of the rotor flux of the motor2is higher than the transition frequency, the second flux estimation module24is dominant. As a result, the rotor dqs-axes flux estimated value is accurately calculated.

The speed/phase estimation module21includes a phase estimation module25, a slip angle estimation module26, a flux vector rotation module27, a phase error estimation module28and a speed and position observing module29.

The phase estimation module25calculates an estimated value ejθeof a phase of the flux vector of the motor2from the rotor dqs-axes flux estimated value {circumflex over (λ)}qdrScalculated by the current/flux estimation module20. For example, the phase of flux vector can be the phase of the stator flux vector calculated by the current/flux estimation module20. The rotor dqs-axes flux estimated value {circumflex over (λ)}qdrSwhich is the output of the current/flux estimation module20is input into the phase estimation module25.

In particular, the phase estimation module25calculates an estimated value ejθeof a phase of the stator flux vector of the motor2by the following Expression (42).

Although the power phase can be calculated by using any one of the stator flux {circumflex over (λ)}qdsand the rotor flux {circumflex over (λ)}qdrin the Expression (42), the stator flux is used for the estimation in the first embodiment. In the Expression (42), the suffix “*”, i.e. an asterisk, is “s” when a calculation is done using the stator flux {circumflex over (λ)}qds. On the other hand, in the Expression (42), the suffix “*” is “r” when a calculation is done using the rotor flux {circumflex over (λ)}qdr.

An estimated value {circumflex over (ω)}slof a slip frequency of the motor2is calculated by the second slip angular frequency estimation module33. The estimated value {circumflex over (ω)}slof the slip frequency is inputted to the slip angle estimation module26.

The slip angle estimation module26calculates a slip angle estimated value {circumflex over (θ)}sl. The slip angle estimated value {circumflex over (θ)}slis calculated by integrating the estimated value {circumflex over (ω)}slof the slip frequency. The sine and cosine of the slip angle estimated value {circumflex over (θ)}slis calculated by the sine/cosine calculation unit27aand is inputted to the flux vector rotation module27.

The flux vector rotation module27calculates a first rotor electrical angle estimated value {circumflex over (θ)}r1based on the estimated value ejθeof the phase of the flux vector calculated by the phase estimation module25and the output of the sine/cosine calculation module27a.

The sine and cosine of the rotor electric angle estimated value {circumflex over (θ)}ris calculated by the sine/cosine calculation unit28a. The calculated values of the sine and cosine of the rotor electric angle estimated value {circumflex over (θ)}rare inputted to the phase error estimation module28.

The phase error estimation module28performs additive theorem operation by using the two rotor electrical angles calculated by the flux vector rotation module27and the speed and position observing module29. The estimated value {circumflex over (θ)}errof rotor electrical angle error is calculated by using approximation of sin Δθ≈Δθ when Δθ is minute.

The speed and position observing module29calculates rotor mechanical angular estimated values {circumflex over (ω)}rm-rand {circumflex over (ω)}rm-r, a mechanical angle estimated value {circumflex over (θ)}rm, electrical angle estimated value {circumflex over (θ)}rbased on the estimated value {circumflex over (θ)}errcalculated by phase error estimation module28.

Specifically, the speed and position observing module29includes a first block29a, a second block29b, a third block29c, a fourth block29d, a fifth block29e, a seventh block29f, an eighth block29g, and a ninth block29hand a tenth block29i.

The first block29ais an integrating circuit of gain Kio. The first block29acalculates an integral of the estimated value {circumflex over (θ)}errcalculated by phase error estimation module28. The second block29bcorrects the estimated value {circumflex over (θ)}errestimated by the phase error estimation module28by multiplying the estimated value {circumflex over (θ)}errestimated by the phase error estimation module28and the gain Kso.

The third block29cmultiples the estimated value {circumflex over (θ)}errestimated by the phase error estimation module28and the gain bo.

The fourth block29dcalculates a value obtained by adding the value calculated by the first block29aand the value calculated by the second block29b.

The fifth block29ecalculates a value obtained by adding the value calculated by the fourth block28dand the torque command value Tem*, and outputs the calculated value to the sixth block29f. The sixth block29fdivides the input value by the estimated value Ĵpof moment of inertia and outputs it to the seventh block29g.

The seventh block29gcalculates a rotor mechanical angular speed estimated value {circumflex over (ω)}rm-l. The seventh block29gis an integrating circuit. The seventh block29gcalculates an integral of the output of the sixth block29f.

The eighth block29hcalculates the speed correction value by dividing the value calculated by the third block29cby the estimated value Ĵpof the moment of inertia.

The ninth block29icalculates a rotor mechanical angular speed estimated value {circumflex over (ω)}rm-rby adding the rotor mechanical angular speed estimated value {circumflex over (ω)}rm-lcalculated by the seventh block29gand the speed correction value obtained by the eighth block29h.

The tenth block29jcalculates a rotor mechanical angle estimated value {circumflex over (θ)}rm. The tenth block29jis an integrating circuit. The seventh block29gcalculates an integral of the rotor mechanical angular speed estimated value {circumflex over (ω)}rm-r

The eleventh block29kcalculates the rotor electrical angle estimated value {circumflex over (θ)}rby dividing the value obtained by multiplying the rotor mechanical angle estimated value {circumflex over (θ)}rmcalculated by the tenth block29jby the number of poles P of the motor2by 2.

Operation of Device According to Second Embodiment

1. Field Weakening Region Increase in Self-Sensing

The system according to the second embodiment is a sensor-less system including no speed sensor, and has a self-estimation function of estimating, for example, a speed by using various parameters in the system. The self-estimation function is also referred to as “self-sensing”.

As described above in the first embodiment, the output torque can be increased by selecting a combination of stator flux command vectors in a field weakening region. This allows fast operation when the load torque is large, and accordingly, leads to increase of the field weakening region. The present section describes discussions related to the field weakening region increase by the self-sensing.

In the system according to the second embodiment illustrated inFIGS. 11 and 12, a flux observer is constituted by the first flux estimation module23and the second flux estimation module24. The speed and phase estimation module121performs speed estimation by using a flux observer function of the current and flux estimation module20. This speed estimation using the flux observer function can be performed by using one of a stator flux estimated value and a rotor flux estimated value.

In addition, a slip angular frequency is input to the speed and phase estimation module121. Estimation of the slipping angular frequency can be performed by using one of the stator flux estimated value and the rotor flux estimated value. Thus, either the stator flux estimated value or the rotor flux estimated value can be optionally selected as an input value used in the speed estimation and the slipping angular frequency estimation.

The speed and position observation module29in the speed and phase estimation module121illustrated inFIG. 12is equivalent to the motion observer.

The slipping angular frequency calculated based on the stator flux is expressed by Expression (43). The slipping angular frequency calculated based on the rotor flux is expressed by Expression (44).

When the power supply phase ejθeis estimated by using the stator flux, the slip calculation is performed by the Expression (43). On the other hand, when power supply phase ejθeis estimated by using the rotor flux, the slip calculation is performed by the Expression (44). One of {circumflex over (ω)}sl-sand {circumflex over (ω)}sl-ris eventually used, and the eventually used value is expressed as {circumflex over (ω)}sl.

It should be noted that, in the Expression (43), a superscript symbol “es” of each sign represents a synchronous coordinate system developed with the stator flux as a reference. The stator flux value λqssof the stationary coordinate system is a sine wave signal, and the development is done based on the stator flux as a reference.

In the synchronous coordinate system, the origin of the coordinate system is reset in each control cycle so as to ensure λqses=0. Thereby, other signals are also converted accordingly. The advantageous effect of this coordinate transformation is that specific calculations can be easier.

The rotor flux has less noise than the stator flux. This is because the rotor flux is less affected by a high frequency component included in voltage due to PWM. This will be described below by using Expression (45) as a stator flux equation and Expression (46) as a rotor flux equation.

The stator flux receives an input of voltage as indicated in Expression (45). Thus, the stator flux is directly affected by a high frequency component of voltage due to PWM. However, this influence on the rotor flux is reduced by a filter. This filter effect will be described below.

Expression (46) is provided with Laplace transform and rewritten by replacing coefficients of the stator flux and the rotor flux on the right hand side with A and B, respectively.

As understood from Expression (47), the rotor flux is equivalent to flux obtained by applying a low pass filter to the stator flux. The rotor flux is a value obtained based on the stator flux, and the stator flux is affected by a high frequency component of voltage. However, influence of voltage noise on the rotor flux is reduced by the above-described low pass filter effect.

When the flux amplitude is reduced by field weakening, the S/N ratio degrades. Additionally, in self-sensing in which the stator flux is used in phase estimation, control is likely to be unstable as compared to a case in which the rotor flux is used. This is because the stator flux is directly affected by a high frequency component of voltage.

The above-described low pass filter effect is achieved by the eighth block24h, the ninth block24i, and the tenth block24j, in particular, among the blocks of the second flux estimation module24illustrated inFIG. 12.

4. Test Results

FIGS. 13 to 15are diagrams showing an example of the test results of field weakening in the second embodiment. The response when the speed command is increased from 0.1 pu to 2.0 pu is shown. In both figures, the horizontal axis is the time axis.

The vertical axis inFIG. 12represents the angular speed command value ωrm* of the motor2and the measured speed valueωrm. The vertical axis inFIG. 14represents the estimated torque of the motor2and the torque limit value. The vertical axis inFIG. 15represents the amplitude of the stator flux command value and the estimated stator flux amplitude.

The estimated torque is calculated by multiplying the stator flux estimated value and the rotor flux estimated value. As shown inFIG. 13, the motor2is stably accelerated to a speed of 2.0 PU. According toFIG. 14, the estimated torque of the motor2is equal to the torque limit value during acceleration, but when the acceleration is completed, the estimated torque of the motor2is smaller than the torque limit value.

This performance depends on a bandwidth obtained by internal proportional integral gain of each observer. Suppressing the bandwidth at relatively low level makes it possible that a high frequency band noise component is not included in the estimated speed.

In the field weakening operation, it is preferable to set the bandwidth of the speed and phase estimation module121to be relatively low to reduce noise included in the estimated speed. A speed (phase) for self-sensing can be calculated from various waveforms of current, voltage, and flux, for example.

The second embodiment above described includes at least two characteristic configurations.

The first characteristic configuration is the speed and phase estimation module121. This configuration achieves a sensor-less system equivalent to the system according to the first embodiment. The sensor-less configuration provides a technological advantage such as the torque limit adjustment in a field weakening region achieved according to the first embodiment, and allows any speed sensor to be omitted.

The second characteristic configuration is the usage of the rotor dqs-axes flux estimated value {circumflex over (λ)}qdrSas an input value to the speed and phase estimation module121. The speed and phase estimation module121calculates an estimated value of a speed of the motor in a subsequent control period based on the rotor dqs-axes flux estimated value {circumflex over (λ)}qdrS.

A rotor flux estimated value contains less noise than a stator flux estimated value, which leads to highly accurate calculation of the estimated speed value. As a result, the effect of increasing a field weakening region can be achieved in the self-sensing of the sensor-less system.

Modification of Second Embodiment

FIG. 16is a configuration diagram of a motor system to which a controller for a power convertor according to a modification of the second embodiment is applied. In the modification illustrated inFIG. 16, a switching block122is provided between the speed and phase estimation module121and the current and flux estimation module20. In the modification shown inFIG. 16, the first slip angular frequency estimation module32, the second slip angular frequency estimation module33, and a switching block123connected thereto are provided.

The stator dqs-axes flux estimated value {circumflex over (λ)}qdsSoutputted from the current and flux estimation module20and the stator dqs-axes current measured value iqdssoutputted from the second coordinate conversion module17are inputted to the first slip angular frequency estimation module32. The first slip angular frequency estimation module32performs a calculation based on Expression (43) by using the above inputted values to calculate the slip angular frequency estimated value {circumflex over (ω)}sl_s.

The second torque command value Tem* and the rotor dqs-axes flux estimated value {circumflex over (λ)}qdrSare inputted to the second slip angular frequency estimation module33.

The second slip angular frequency estimation module33performs a calculation based on Expression (44) by using the above inputted values to calculate the slip angular frequency estimated value {circumflex over (ω)}sl_r.

The switching block122selectively transmits one of the estimated value selected from the rotor dqs-axes flux estimated value {circumflex over (λ)}qdrSand the stator dqs-axes flux estimated value {circumflex over (λ)}qdsSto the speed and phase estimation module121.

The switching block123selectively transmits one of the estimated value selected from the slip angular frequency estimated value {circumflex over (ω)}sl_soutputted from the first slip angular frequency estimation module32and the slip angular frequency estimated value {circumflex over (ω)}sl_routputted from the second slip angular frequency estimation module33to the speed and phase estimation module121.

The switching block122works together with the switching block123.

When the switching block122selects the rotor dqs-axes flux estimated value {circumflex over (λ)}qdrS, the switching block123selects the slip angular frequency estimated value {circumflex over (ω)}sl_routputted from the second slip angular frequency estimation module33.

On the other hand, when the switching block122selects the stator dqs-axes flux estimated value {circumflex over (λ)}qdsS, the switching block123selects the slip angular frequency estimated value {circumflex over (ω)}sl_soutputted from the first slip angular frequency estimation module32.

The criterion by which the switching block122and the switching block123switch the above estimated values may be predetermined.

For example, the rotor dqs-axes flux estimated value {circumflex over (λ)}qdrSmay be selected if the power supply angular frequency ωeis equal to or higher than a predetermined value. For example, the stator dqs-axes flux estimated value {circumflex over (λ)}qdsSmay be selected if the power supply angular frequency ωeis lower than the predetermined value.

FIG. 17is a configuration diagram of the controller for a power convertor according to the modification of the second embodiment. The configuration illustrated inFIG. 17does not include the above-described second characteristic configuration included in the second embodiment. Accordingly, in the modification illustrated inFIG. 17, the stator dq-axis flux estimated value {circumflex over (λ)}qdsSis an input value to the speed and phase estimation module121.

The modification illustrated inFIG. 17cannot provide the above-described effects of “improvement of calculation accuracy of the estimated speed value and sensor-less field weakening region increase by the use of the rotor flux”. However, even though those effects are not obtained, the modification illustrated inFIG. 17has advantages of “torque increasing same as that of the first embodiment” and “no need to provide a speed sensor in a sensor-less system”.

The controller11according to the second embodiment may be achieved by using the structure illustrated in the hardware configuration diagram inFIG. 20.

FIG. 21is a graph of relative RMS noise as flux decreases. As illustrated inFIG. 21, relative RMS error included in the estimated speed increases as the flux decreases. This is because that the flux amplitude decreases as the rotor speed increases while the amplitudes of a PWM harmonic wave and a noise of current measurements are constant.

The noise is normalized to 1 when the flux is 1 pu. The noise increases to 13 approximately when the flux becomes near 0.1 pu.

The formula of the noise is rms ({circumflex over (ω)}r−{circumflex over (ω)}avg). In the formula, {circumflex over (ω)}rrepresents an instantaneous speed at each sampling point, and {circumflex over (ω)}avgrepresents an average speed between sampling points.

It is preferable to reduce influence of noise at low amplitude flux (that is, in a high speed range). To achieve this, it is preferable to set controller gains bo, Kso, and Kio to be low when rotor flux base speed estimation is performed by the speed and phase estimation modules21,121illustrated inFIGS. 3, 12 and 17.

InFIGS. 12 and 17, the controller gain bo is the gain of a third block29c. The controller gain Kso is the gain of a second block29b. The controller gain Kio is the gain of a first block29a. The controller gains bo, Kso, and Kio achieve a function as a filter.

Decrease of the controller gains bo, Kso, and Kio is equivalent to decrease of the cutoff frequency of the filter. Accordingly, noise in a high frequency component included in a rotor flux input of an observer can be reduced.

Third Embodiment

FIG. 18is a configuration diagram of a motor system to which a controller for a power convertor according to a third embodiment is applied.

The torque command limit module13, the appropriate flux command generation module18, and the power supply angular frequency calculation module19, which are provided in the first and second embodiments, are omitted in the motor system according to the third embodiment. Any other configuration is the same as that of the second embodiment.

A value input to the DB-DTFC calculation module14is changed due to the configuration omission. In place of the second torque command value Tem*, the first torque command value Tem1* calculated by the speed control module12is input to the DB-DTFC calculation module14. In place of the stator flux command value λs_optgenerated by the appropriate flux command generation module18, the rated stator flux λrateis input to the DB-DTFC calculation module14.

Since the torque command limit module13and the like are omitted, the advantage of “operation at high power in a wider frequency range”, which is achieved in the first embodiment, is restricted in the third embodiment as compared with the first embodiment.

However, similarly to the second embodiment, the speed and phase estimation module121is provided and an input value to the speed and phase estimation module121is the rotor dqs-axes flux estimated value {circumflex over (λ)}qdrSin the third embodiment. Thus, the effects of “improvement of calculation accuracy of the estimated speed value and sensor-less field weakening region increase by the use of the rotor flux”, which are the same as those of the second embodiment, can be obtained in the third embodiment.

Modification of Third Embodiment

FIG. 19is a configuration diagram of a motor system to which a controller for a power convertor according to a modification of the third embodiment is applied.

The configuration shown inFIG. 18is configured with the dead-beat direct torque & flux control (DB-DTFC) method which does not use a PI current controller to generate a voltage command.

However, the configuration in the third embodiment can be implemented not only in the DB-DTFC method but also in a Field Oriented Control method. Field Oriented Control method is also referred to as FOC method hereinafter.

The FOC method is a method in which a current component for generating a torque (rotational force) and a current component for generating a flux are separated from each other, and each current component is independently controlled as a direct current amount.

One of FOC methods is DFOC (Direct Field Oriented Control) method. DFOC method is a method of directly estimating and controlling the flux vector by a flux sensor or a flux observer.FIG. 19illustrates a configuration diagram of a motor system to which a controller, which is configured based on DFOC method, for a power convertor according to a modification of the third embodiment is applied.

Another one of the FOC methods is IFOC (Indirect Field Oriented Control) method. The IFOC method uses indirect type vector control (also called slip frequency type vector control) that controls a slip of an induction machine regardless of flux estimation or flux detection.

In the modification shown inFIG. 19, DB-DTFC calculation module14is replaced with a DFOC calculation module314, and the first coordinate conversion module15is replaced with a fifth coordinate conversion module15a.

The controller11includes a third coordinate conversion module17aand a fourth coordinate conversion module17b. The v-phase stator current Ivsand the w-phase stator current Iwsare inputted to the third coordinate conversion module17a.

The three phase stator voltage command value Vus*, Vvs*, and Vws*, which is outputted from the fifth coordinate conversion module15a, is inputted to the fourth coordinate conversion module17b.

The first torque command value Tem1* calculated by the speed control module12is inputted to the DFOC calculation module314in the system according to DFOC method shown inFIG. 19.

With respect to a flux command, although the rated flux value λrateis inputted to the DFOC calculation module314as the stator flux command value λs*, other system based on DFOC method may perform a control based on the rotor flux command value {circumflex over (λ)}qdrS* instead of the stator flux command value λs*. Since the rotor flux command value {circumflex over (λ)}qdrS* can be calculated from of the stator flux command value λs*, only the stator flux command value λs* is illustrated inFIG. 19.

The speed and phase estimation module121inputs the stator power supply phase estimated value {circumflex over (θ)}eas a reference signal to the third coordinate conversion module17aand the fifth coordinate conversion module15a.

On the basis of the reference signal, the third coordinate conversion module17aconverts the input signal to a signal of the rotating coordinate system of γ and δ components orthogonal to each other.

By appropriately selecting the phase of the reference signal, it is possible to make the γ component in-phase with the reference and to make the8component orthogonal to the reference.

The third coordinate conversion module17aperforms a coordinate conversion to obtain a γ-axis stator current iγand a δ-axis stator current iδ, and outputs the current values to the DFOC calculation module314.

Conversely to the above, the fifth coordinate conversion module15aconverts the voltage command value Vγ* of the γ component and the voltage command value Vδ* of the δ component, which are outputs from the DFOC calculation module314, to the three-phase stator voltage command values Vus*, Vvs*, and Vws*. The output from the fifth coordinate conversion module15ais inputted to the PWM control module16and the fourth coordinate conversion module17b.

The fourth coordinate conversion module17bconverts the inputted three-phase stator voltage command values Vus*, Vvs*, and Vws* which are values in fixed coordinate system to the voltage command value VqdsS* of two-axes components of dqs-axes. The converted values by the fourth coordinate conversion module17bare inputted to the current and flux estimation module20.

The stator flux estimated value {circumflex over (λ)}qdssand the rotor flux estimated value {circumflex over (λ)}qdrsestimated by the current and flux estimation module20are inputted to the DFOC calculation module314.

The DFOC calculation module314internally generates a stator current command value of they component and a stator current command value of the δ component based on the inputted values, and generates the voltage command value Vγ* of the γ component and the voltage command value Vδ* of they component so that the γ-axis stator current iγand the δ-axis stator current iδfollow the command values.

The voltage command value Vγ* of the γ component and the voltage command value Vδ* are inputted to the fifth coordinate conversion module15a.

The voltage command value Vγ* of they component and the voltage command value Vδ* re transmitted to the inverter8as gate pulses via the PWM control module16.

Since the torque command limit module13and the like are omitted, the advantage of “operation at high power in a wider frequency range”, which is achieved in the first embodiment, is restricted in the modification of the third embodiment as compared with the first embodiment.

However, similarly to the second embodiment, the speed and phase estimation module21is provided and an input value to the speed and phase estimation module21is the rotor dqs-axes flux estimated value {circumflex over (λ)}qdrSin the third embodiment. Thus, the effects of “improvement of calculation accuracy of the estimated speed value and sensor-less field weakening region increase by the use of the rotor flux”, which are the same as those of the second embodiment, can be obtained in the third embodiment.

It should be noted that, although the DB-DFTC calculation module14is replaced with the DFOC calculation module314in the modification inFIG. 19, the DB-DFTC calculation module14may be replaced with a FOC calculation module or an IFOC calculation module.

The controller11according to the third embodiment can be achieved by using the structure illustrated in the hardware configuration diagram inFIG. 20.

The features and advantages of the present disclosure (or embodiments) may be summarized as follows.

In the first controller for a power convertor according to the first aspect of the present disclosure, the upper limit torque value is calculated to be smaller as the fundamental wave output frequency of output from the power convertor increases in a field weakening region. In other words, the torque limiter range is set to be smaller as the fundamental wave output frequency increases. The range of the torque command value for maintaining control stability tends to be smaller in a faster operation region in which the fundamental wave output frequency is larger. The torque limiter range is dynamically adjusted in accordance with this tendency. The torque command value changes only in the adjusted torque limiter range, and thus torque increase can be achieved without losing the control stability. As a result, the control stability and the output torque increase can be simultaneously achieved.

According to the second controller for a power convertor according to the second aspect of the present disclosure, the estimated value of the rotor flux is used in calculation. The estimated value of the rotor flux includes less noise than the estimated value of the stator flux, and thus the estimated speed value can be highly accurately calculated with less ripple. The torque command value has less ripple.

According to the third aspect of the present disclosure, the motor driving system including the first controller for a power convertor according to the first aspect can simultaneously achieve the control stability and the output torque increase when the motor is operated fast in the field weakening region. The motor driving system including the second controller for a power convertor according to the second aspect can highly accurately calculate the estimated speed value.