Hand-held, battery operated, doppler ultrasound medical diagnostic device with cordless probe

A hand-held, battery operated, Doppler ultrasound diagnostic device for use in obstetrical applications to, for example, measure fetal heartbeat, and vascular applications to, for example, measure blood flow and pressure, includes a cordless probe, a base unit, and an electric recharging stand. The probe detects the physical response signal being monitored through Doppler ultrasound techniques. The signal is used to frequency modulate a sine wave carrier signal which is transmitted by the probe and received by the base unit. The carrier signal frequency and wave form is chosen so as not to cause interference with other medical equipment that may be nearby. The base unit demodulates the carrier signal and retrieves the human response signal. The base unit displays the human response signal visually with an LED display and audibly with built in speakers. In addition, the base unit includes an output connectors which allow the human response signal to be recorded on a tape recorder or played through earphones. Having a cordless probe enables the base unit to be placed outside the sterile field created around the patient. When not in use, the probe can be stored in a nested position in the base unit and the base unit can be attached to the recharging stand. Both the probe and the base unit are battery operated. The base unit will recharge its batteries when positioned in the recharging stand. The probe will recharge its batteries when the probe is positioned in the base unit and the base unit is positioned in the recharging stand.

BACKGROUND OF THE INVENTION 
1. Field of the Invention 
The present invention is directed generally to a medical diagnostic devices 
that use Doppler ultrasound for obstetrical and vascular monitoring 
applications, and more specifically to a system for wireless transmission 
of signals derived from a Doppler ultrasound probe to a base unit for 
visual display, audible display, and recordation. 
2. Description of the Prior Art 
The diagnostic capabilities of the medical profession have increased 
significantly throughout the years. Two such advancements have been in the 
use of Doppler ultrasound based devices to detect and measure vascular and 
cardial blood flow direction and rate, to detect and measure fetal heart 
rate, and for numerous other diagnostic applications. 
The basic Doppler effect for sound is well-known. When a source of sound 
and a receiver of the sound move in relation to each other, the pitch or 
frequency of the sound perceived or detected at the receiver is different 
from the pitch or frequency of the source. If they are moving toward each 
other, the perceived or received pitch or frequency of the sound is higher 
than the source sound. The classic example is standing near a railroad 
track as a train blowing its whistle passes. As the train approaches, the 
perceived whistle sound is a high pitch, which then changes abruptly to a 
lower pitch as the train passes and goes away from the listener. 
Ultrasound is simply sound that has a higher pitch or frequency than the 
hearing capability of a normal human ear, which is about 20 kilohertz 
(KHz). The Doppler effect for ultrasound is the same as for audible sound, 
but, since ultrasound is at a pitch or frequency beyond the range of human 
ears, electronic equipment is used to detect it. 
The Doppler effect is also produced in echoes, when sound or ultrasound is 
reflected by, or bounced off, a moving object. Thus, sound or ultrasound 
can be produced and projected by a speaker device or ultrasound sender, 
and, if it reflects or bounces off an object or target, the echo or return 
sound can be received and detected. If the ultrasound source, target 
object, and echo receiver are all stationary, the pitch or frequency of 
the echo ultrasound will be the same as the source ultrasound. However, if 
the target object is moving toward the receiver of the ultrasound echo, 
the ultrasound echo received and detected will have a higher pitch than if 
the target object was moving away from the receiver. The speed or velocity 
at which the target object is moving toward or away from the receiver 
determines the pitch or frequency of the echo received. Also, a fluid, 
such as blood, also reflects ultrasound waves, and the velocity or rate of 
blood flow determines the frequency of the echoed ultrasound waves. Thus, 
detecting frequencies of the echoed ultrasound waves can be used to 
measure direction and rate of blood flow. This Doppler effect in echoed 
ultrasound is the principle that is typically utilized in ultrasound 
medical diagnostic devices, where ultrasound signals having frequencies in 
the range between one (1) megahertz (MHz) and twenty (20) MHz are often 
used. 
In medical diagnostic devices using Doppler ultrasound, the source of the 
ultrasound and the receiver of the ultrasound are usually transducers 
mounted in a hand-held probe. The probe is held relatively stationary with 
respect to a target object being detected or measured. Some slow movement 
and positioning of the probe by the physician or technician can be 
accommodated for detecting, if it is substantially slower than the motion 
of the target object. However, where accurate measurements are needed, the 
probe should be held quite stationary. An ultrasound wave stream is 
transmitted by the transducer in the probe in the direction of the target 
object to be detected or measured, and the return echo is received, 
transduced to an electric signal having both a frequency and an amplitude 
that corresponds to the frequency and amplitude of the echoed ultrasound 
waves. For example, in obstetrical applications, such as detecting or 
measuring fetal heart rate, the ultrasound waves from the probe are 
directed so as to intercept the blood flowing in a beating fetal heart. In 
vascular applications, the ultrasound waves from the probe are directed to 
intercept blood moving and circulating in a vein or artery to detect or 
measure blood flow and direction. In both situations, the directed signal 
from the probe is reflected by the flowing blood, which creates Doppler 
shifts from the frequency of the ultrasound by the probe to the 
frequencies of the echoed ultrasound reflected from the flowing blood. The 
reflected ultrasound echoes from the flowing blood is detected by a 
transducer in the probe, which converts ultrasound wave energy to electric 
signals. The Doppler frequency shift between the directed ultrasound and 
the reflected ultrasound echoes returned from the flowing blood varies 
proportionally with the instantaneous velocity of the flowing blood. If 
the blood is flowing away from the directed ultrasound from the probe, the 
reflected ultrasound echoes will have lower frequencies than the directed 
ultrasound. If the blood is flowing toward the directed ultrasound from 
the probe, the reflected ultrasound echoes will have higher frequencies 
than the directed ultrasound. Of course, if the moving target is not 
moving in relation to the directed ultrasound from the probe, the 
reflected ultrasound echo will have the same frequency as the directed 
ultrasound. 
Doppler ultrasound techniques for medical diagnostic purposes are well 
known in the art. For example, see Peter Atkinson & John Woodcock, DOPPLER 
ULTRASOUND AND ITS USE IN CLINICAL MEASUREMENT, published by Academic 
Press of New York City (1982); Matthew Hussey, BASIC PHYSICS AND 
TECHNOLOGY OF MEDICAL DIAGNOSTIC ULTRASOUND, published by Elsevier of New 
York City (1985); and Peter Fish, PHYSICS AND INSTRUMENTATION OF 
DIAGNOSTIC MEDICAL ULTRASOUND, published by John Wiley & Sons of New York 
City (1990). See also, U.S. Pat. Nos. 4,276,491 issued to Daniel; 
4,807,636 issued to Skidmore et al.; 4,850,364 issued to Leavitt; and 
5,394,878 issued to Frazin all of which show medical devices using Doppler 
ultrasound techniques. Furthermore, Doppler ultrasound has become a 
popular method of medical diagnosis because it is non-invasive, painless, 
creates little or no side effects, and is relatively inexpensive. Finally, 
ultrasound frequencies are often used in medical diagnostic applications 
because they reflect well from the boundaries between different organs and 
blood cells without utilizing potentially harmful ionizing radiation. 
In many medical diagnostic applications using Doppler ultrasound, the 
transmitter of the directed signal is placed directly against the human 
skin. For example, when measuring fetal heart rate, the transmitter is 
placed on the midline of the abdomen and aimed downward toward the pubic 
bone. When measuring vascular flow, the transmitter is placed directly 
over the underlying vessel. The direct contact between the transmitter and 
the human skin is necessary to reduce reflections of the directed 
ultrasound and the reflected ultrasound echo caused by the skin, and 
ultrasound does not propagate well in air at the frequencies used in these 
applications. To facilitate ease of use and manual manipulation of 
diagnostic devices using Doppler ultrasound, as described above, it is 
desirable to have a device that is small, portable, and battery operated, 
since the probe must often be placed directly next to the skin of the 
patient being tested. Current Doppler ultrasound probes are connected by a 
cord containing electric wires to a base unit, where the electric signals 
from the receiving transducer are processed for display in useful 
information format. While such current devices are very useful and 
effective, there has been a need for even further improvements-one of 
which is to eliminate the cord between the probe and the base unit. A 
cordless probe would make the probe easier to handle and use, and it would 
reduce the amount of equipment placed in the sterile field around the 
patient. A cordless probe would also enable the person using the 
diagnostic device to place the probe in any desired position, unencumbered 
by a cord and the positioning restrictions that a cord might impose. There 
are some constraints, however, in replacing the cord with some kind of 
wireless signal transmission system between the probe and the base unit. 
For example, the signal transmitted from the probe to the base unit cannot 
interfere with other medical equipment in the room, hospital, or 
ambulance. The signal transmitted from the probe to the base unit may also 
interfere with the operation of the Doppler transceiver, itself, if the 
harmonics coincide with the Doppler signal. Furthermore, since Doppler 
ultrasound in an obstetrical application is often used to reassure the 
mother of the presence of fetal life, it is crucial that the signal from 
the probe be received by the base unit in a very reliable manner, 
regardless of its position in relation to the base receiver and to other 
objects and persons in the room, to avoid alarming the mother. Finally, it 
is desirable to have the batteries used in the probe and the base unit be 
rechargeable during storage of the base unit and the probe so that the 
batteries will be fully charged when the probe and the base unit are used. 
SUMMARY OF THE INVENTION 
Accordingly, it is a general object of this invention to provide a 
hand-held, battery operated, and cordless ultrasound transceiver device 
for use in obstetrical, cardiovascular, and other diagnostic applications. 
It is another general object of this invention to provide ultrasound 
medical diagnostic equipment including a hand-held freely moveable probe 
containing ultrasound producing and receiving components and a base unit 
for processing the Doppler frequency shifted information into visual 
and/or audibly perceivable useful information with wireless transmission 
of signals from the probe to a base unit. 
It is a specific object of this invention to provide a medical diagnostic 
device including a probe and a base unit, wherein the probe transmits a 
signal that is received reliably and consistently by the base unit 
regardless of the orientation, position, and location of the probe in 
relation to the base unit or in relation to other objects and/or persons 
in the room. 
Another specific object of this invention to provide ultrasound medical 
diagnostic equipment including a hand-held freely moveable probe 
containing ultrasound producing and detecting components and a base unit 
wherein signals transmitted by the probe and received by the base unit do 
not interfere with other nearby medical or communications equipment or 
with the ultrasound operations of the probe itself. 
Additional objects, advantages, and novel features of the invention shall 
be set forth in part in the description that follows, and in part will 
become apparent to those skilled in the art upon examination of the 
following or may be learned by the practice of the invention. The objects 
and the advantages may be realized and attained by means of the 
instrumentalities and in combinations particularly pointed out in the 
appended claims. 
To achieve the foregoing and other objects and in accordance with the 
purposes of the present invention, as embodied and broadly described 
therein, the present invention includes a transducer, sensitive to 
physical phenomena in a body, that produces a first electric signal having 
characteristics indicative of the physical phenomena; a signal converter 
for converting the first electric signal into a frequency modulated sine 
wave current signal that drives a light emitter where the light emitted 
has an intensity that is a linear function of the amplitude of the 
frequency modulated sine wave current signal; a light detector spatially 
separate from the light emitter for detecting the emitted light signal and 
producing a corresponding second electric signal; a signal processor for 
extracting the first electric signal from the second electric signal, and 
a audible display, visual display, and a recorder connection for 
displaying and recording the information indicative of the physical 
phenomena. 
To further achieve the foregoing and other objects, the present invention 
further comprises a method of measuring fetal heart rate or vascular or 
cardial blood flow using Doppler ultrasound techniques, by generating a 
first electric signal having a frequency in the ultrasound range; 
generating directed ultrasound waveforms having the same frequency as the 
electric signal; directing the ultrasound waveforms toward flowing blood 
or a beating heart; detecting reflected ultrasound waveforms reflected by 
the flowing blood or the beating heart; producing a second electric signal 
having the same frequency as the reflected ultrasound waveforms; 
heterodyning the first electric signal and the second electric signal to 
create a mixed electric signal having a frequency that is proportional to 
the velocity of the flowing blood or the beating heart; frequency 
modulating a carrier signal with the mixed electric signal to produce a 
frequency modulated sine wave electric signal; driving a light emitter 
with the frequency modulated sine wave electric signal to produce a 
frequency modulated sine wave light signal; detecting the frequency 
modulated sine wave light signal with a detector to produce a third 
electric signal; and frequency demodulating the third electric signal to 
recover the mixed electric signal.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
The ultrasound medical diagnostic device 50 of the present invention shown 
in FIG. 1 includes a base unit 52 for receiving light signals emitted by a 
probe 54, the probe 54 for emitting and receiving ultrasound waves and for 
emitting light signals, and a recharging stand 56 for storing and 
recharging the base unit 52 and the probe 54. The probe 54 can be 
positioned in probe holder 57 in the base unit 52 for storage when the 
medical diagnostic device 50 is not being used. Likewise the base unit 52 
can be placed into the recharging stand 56 for storage when the medical 
diagnostic device 50 is not being used, and for recharging the 
rechargeable battery 107 in the probe 54 and the rechargeable battery 154 
in the base unit 52. 
During use, the detachable nose 60 end of the probe 54 is placed next to 
the skin of a patient, as shown in FIGS. 2 and 3. The probe 54 emits 
ultrasound waves 55 at a constant frequency toward a moving target, for 
example, blood (not shown) flowing in the heart (not shown) of a fetus F 
of a pregnant woman W (see FIG. 2), or the blood (not shown) flowing in an 
artery A, of the patient P being monitored (see FIG. 3). 
Referring to FIG. 2, the blood (not shown) flowing in the heart of the 
fetus F reflects the ultrasound waves 55 transmitted from the probe 54 and 
causes a Doppler frequency shift in the reflected ultrasound waves 59 that 
corresponds to the velocity of flowing blood (not shown). An electric 
signal having a frequency that is in the audible hearing range of a human 
ear and which is still proportional to the velocity of the flowing blood 
being targeted and which can be used to drive a speaker in the base unit 
52 to produce audible sound that can be heard by the operator and by the 
patient, can be created by a conventional heterodyne process. Essentially, 
a constant frequency electric signal is produced to drive or oscillate an 
ultrasound producing or sending transducer in the probe 54, which produces 
and propagates the ultrasound waves 55 at that constant frequency. An 
ultrasound receiving transducer in the probe 54 detects the reflected 
ultrasound echo waves 59 and produces an electric signal that has the same 
frequency as, and an amplitude proportional to, the reflected ultrasound 
waves 59. In the heterodyne process, the electric signal with the sending 
frequency is mixed with the electric signal of the reflected frequency, 
which produces an electric signal that has a beat frequency equal to the 
difference between the respective sending frequency and reflected 
frequency. The beat frequency varies between zero, when there is no 
difference between the sending frequency and the reflected frequency, to 
some frequency in the audible hearing range, when the moving target, for 
example, flowing blood or a beating heart, and the beat frequency is 
proportional to the velocity of the moving target. An audio frequency 
electric signal can be produced from this heterodyne beat frequency for 
driving a speaker that can be heard by the operator and patient, as 
described above, as well as for driving calculations and visual displays 
of heart rate, blood flow velocity, and the like. The audio frequency 
electric signal is transmitted by infrared light from the probe 54 to the 
base unit 52 as will be described in more detail below, where it can be 
processed for visual display, played over speakers, or recorded. 
Propagation of the ultrasound waves 55 and the reflected waves 59 can be 
improved by applying coupling gel 61 (see FIG. 2) to the skin of the 
patient and the detachable nose 60 of the probe 54. The coupling gel 61 
acts as a transfer medium through which the ultrasound waves 55 emitted by 
the sender element 90 and the reflected waves 59 can travel. 
The functional block diagram for the components and electronic circuitry in 
the probe 54 is shown in FIG. 10, while the functional block diagram for 
the components and electronic circuitry in the base unit 52 is shown in 
FIGS. 4 and 5. The functional block diagram for the recharging stand 56 is 
also shown in FIG. 12. 
Referring now to FIG. 10, the components of the probe 54 will now be 
discussed in greater detail. The power of the probe 54 is controlled by 
the on/off control 106. The user activates the probe 54 by placing the 
on/off control 106 in the "on" state by momentarily pressing the on/off 
switch 108, which momentarily places the on/off switch 108 in the "on" 
state. The on/off control 106 is changed to the "off" state the next time 
the on/off switch 108 is momentarily pressed. 
The on/off control 106 provides connection between the battery 107 and the 
power supplies 109 when the on/off control 106 is in the "on" state. The 
power supplies 109 take the voltage supplied by the rechargeable battery 
107 and convert it to create the necessary power, voltages, and voltage 
levels used by the electronic components in the electronic circuitry in 
the probe 54. When the on/off control 106 is in the "off" state, the 
rechargeable battery 107 is disconnected from the electronic circuitry in 
the probe 54. The use of power supplies and power supply circuity to 
create different voltage levels is well known to people having ordinary 
skill in the art and need not be described in any further detail. 
The power supplies 109 also provide an output signal to the low battery 
detector 110, which monitors the charge and energy level of the 
rechargeable battery 107 to determine if the rechargeable battery 107 is 
losing its power or charge. When the on/off control 106 is in the "on" 
state and the rechargeable battery 107 is operating with enough charge to 
drive the electronic circuitry in the probe 54, the low battery detector 
110 provides a signal to the power on LED 111 so as to cause the power on 
LED 111 to emit a continuous visible light. When the on/off control 106 is 
in the "on" state and the rechargeable battery 107 is not operating with 
enough charge, the low battery detector 110 provides a signal to the power 
on LED 111 so as to cause the power on LED 111 to emit a flashing visible 
light as an indicator to the user that the rechargeable battery 107 is low 
on charge. Six V17OR cells manufactured by Varta Batteries, Inc., can be 
used in the rechargeable battery 107 in this invention. The HLMP-M501 
manufactured by Quality Technologies can be used for the "power on" LED 
111 in this invention. This type of power level indication circuitry is 
well known to people having ordinary skill in the art and need not be 
described in any further detail. 
When the on/off control 106 is in the "on" state, the Doppler transceiver 
94 is activated. When the on/off control 106 is in the "off" state, the 
Doppler transceiver 94 is not activated. Doppler transceivers are well 
known in the art. For example, the FP3B manufactured by Medasonics of 
Fremont, Calif., can be used in this invention for the Doppler transceiver 
94. The Doppler transceiver 94 creates the constant frequency electric 
signal which drives the ultrasound sender element or transducer 90 located 
in the detachable nose 60 (see FIGS. 2-5) of the probe 54 to produce and 
propagate the constant frequency ultrasound waves 55. The Doppler 
frequency shifted return ultrasound waves 59 are detected by the receive 
dement or transducer 92, which is also located in the detachable nose 60 
of the probe 54 and controlled by the Doppler transceiver 94. The 
detachable nose 60 provides access to the rechargeable battery 107 for 
replacement. The sender dement 90, receive element 92, and the Doppler 
transceiver 94 can be designed for a specific frequency, depending on how 
the medical diagnostic device 50 is to be used. For example, a two (2) 
megahertz (MHz) or a three (3) MHz signal might be used for obstetric 
applications to detect fetal heartbeats. A two (2) MHz signal is best 
suited for detecting fetal heartbeats after the fetus is twelve weeks old 
and throughout labor and delivery. A three (3) MHz signal has the 
increased sensitivity needed for the early stages of pregnancy. A five (5) 
MHz or an eight (8) MHz signal might be used in vascular applications. The 
five (5) MHz signal is better suited for deep arterial and venous flow 
detection, while the eight (8) MHz signal can be used for superficial 
vessel measurements. 
The Doppler ultrasound waves propagated by the sender element 90 are 
directed outward from, and parallel to, the probe 54 and toward the 
patient being examined. For example, in vascular applications, the 
detachable nose 60 of the probe 54 is placed next to the patient's P skin 
over the underlying vessel A, as shown in FIG. 3. In obstetric 
applications, the detachable nose 60 of the probe 54 is placed next to the 
woman's W skin on the midline of the abdomen, as shown in FIG. 2. In both 
vascular and obstetric applications, the ultrasound waves transmitted by 
the sender element 90 in the detachable nose 60 of the probe 54 are 
directed toward the flowing blood (not shown). 
The flowing blood (not shown) reflects the ultrasound waves and creates a 
Doppler frequency shift between the frequency of the ultrasound signal 55 
from the sender element 90 and the frequency of the reflected ultrasound 
signal 59. The reflected ultrasound signal 59 is detected and received in 
the receive element 92, which is also located in the detachable nose 60 of 
the probe 54. The Doppler frequency shifted electric signal produced by 
the receive element 92 is an input signal to the Doppler transceiver 94. 
The Doppler transceiver 94 amplifies the Doppler frequency shifted 
electric signal and then mixes it with the constant frequency electric 
sender signal in a heterodyne process to produce the audio frequency 
electric signal. For convenience, this audio frequency electric signal 
produced by heterodyning the constant frequency electric sender signal and 
the Doppler frequency shifted electric signal will be referred to as the 
Doppler audio signal, understanding that it is an electric, not an 
acoustic, signal with frequencies in the audio range. The Doppler audio 
signal has a frequency that is proportional to the velocity of the moving 
target or flowing blood. 
The Doppler audio signal is the output signal from the Doppler transceiver 
94 and the input signal to the volume control 96. The volume control 96 is 
used to attenuate the Doppler audio signal for the purpose of reducing the 
amplitude and strength of the Doppler audio signal, and to provide the 
audio volume level preferred by the user. The mount of attenuation done by 
the volume control 96 is controlled by the user through the operation of a 
slide potentiometer 97 (see FIGS. 2-5) located on the probe 54. The user 
adjusts the volume control 96 to produce the desired level of audible 
output from the speakers 134 in the base unit 52. The volume control 96 is 
also shown in FIG. 14 and will be discussed in more detail below. 
Looking again at FIG. 10, the Doppler audio signal that is the output 
signal from the Doppler transceiver 94 is also the input signal to the 
timer control 105. The timer control 105 is used to conserve the energy in 
the rechargeable battery 107 in the probe 54. The timer control 105 
continuously compares the amplitude of the Doppler audio signal to a 
reference value that is predetermined and preset in the timer control 105. 
If the amplitude of the Doppler audio signal is below the preset reference 
value, the timer control 105 initiates a timer clock (not shown) that 
begins running and runs for a predetermined and preset time. If the 
Doppler audio signal remains below the preset reference value for the 
preset time, the timer control 105 provides a signal to the on/off control 
106 that turns off the probe 54 and disconnects the rechargeable battery 
107 from the remainder of the electronic circuitry in the probe 54. If the 
amplitude of the Doppler audio signal falls below the preset reference 
value so as to start the running of the timer clock (not shown), the timer 
clock (not shown) will be reinitialized if the amplitude of the Doppler 
audio signal becomes larger than the reference value before the timer 
clock (not shown) runs for the preset time. Timer clock circuits are 
well-known to persons having ordinary skill in this art, and thus need not 
be shown or described in more detail for purposes of this invention. 
The output signal from the volume control 96 is the input signal to the 
frequency modulator 98, which frequency modulates the Doppler audio signal 
with a 455 kilohertz (KHz) carrier signal. In frequency modulation, a 
modulating signal is used to vary the frequency of a carrier signal to 
create a frequency modulated signal. The frequency modulated signal has a 
constant peak-to-peak amplitude but a varying frequency, where the 
variation in the frequency of the frequency modulated signal is dependent 
on the amplitude of the modulating signal. The frequency of the 
unmodulated carrier signal is called the center frequency. The frequency 
of the frequency modulated signal varies in a range above and below the 
center frequency. 
By way of example and further clarification for the signals involved in the 
medical diagnostic device 50, the modulating signal is the Doppler audio 
signal and is shown in FIG. 19. The unmodulated carrier signal has a 
center frequency of 455 KHz and is shown in FIG. 19. The resulting 
frequency modulated signal is also shown in FIG. 19. All three signals 
shown in FIG. 19 will now be discussed in greater detail. 
The amount of the amplitude of the modulating Doppler audio signal 
determines the amount that the unmodulated carrier signal deviates from 
the center frequency of 455 KHz to produce the frequency modulated signal. 
For example, at time T.sub.0 in FIG. 19, the modulating Doppler audio 
signal has a zero amplitude. Therefore, the frequency modulated signal is 
at the center frequency of 455 KHz. As the amplitude of the modulating 
Doppler audio signal becomes increasingly positive between time T.sub.0 
and time T.sub.1, the frequency of the frequency modulated signal 
increases above 455 KHz. The frequency modulated signal reaches its 
maximum frequency when the modulating Doppler audio signal reaches its 
maximum amplitude at time T.sub.1 in FIG. 19. As the amplitude of the 
modulating Doppler audio signal decreases between time T.sub.1 and time 
T.sub.2, the frequency of the frequency modulated signal decreases but 
remains greater than 455 KHz. At time T.sub.2 in FIG. 19, the modulating 
Doppler audio signal again has a zero amplitude. Therefore, the frequency 
modulated signal is again at the center frequency of 455 KHz. 
As the amplitude of the modulating Doppler audio signal further decreases 
and becomes negative between time T.sub.2 and time T.sub.3, the frequency 
of the frequency modulated signal decreases below 455 KHz. The frequency 
modulated signal reaches its minimum frequency when the modulating Doppler 
audio signal reaches its minimum amplitude at time T.sub.3 in FIG. 19. As 
the amplitude of the modulating Doppler audio signal increases between 
time T.sub.3 and time T.sub.4, the frequency of the frequency modulated 
signal increases but remains less than 455 KHz. At time T.sub.4 in FIG. 
19, the modulating Doppler audio signal again has a zero amplitude. 
Therefore, the frequency modulated signal is again at the center frequency 
of 455 KHz. 
The frequency of the frequency modulated signal continues to change as the 
amplitude of the modulating Doppler audio signal continues to increase and 
decrease. The frequency of the frequency modulated signal returns to the 
455 KHz center frequency each time the amplitude of the modulating Doppler 
audio signal is zero (0). 
The rate of the frequency deviation in the frequency modulated signal is 
determined by the frequency of the modulating Doppler audio signal. For 
example, if the modulating Doppler audio signal has a frequency of 2 KHz, 
the frequency of the frequency modulated signal will swing above and below 
the 455 KHz center frequency of the unmodulated carrier signal two 
thousand (2,000) times a second. Thus, the frequency of the modulating 
Doppler audio signal determines the rate of frequency deviation, but not 
the amount of the frequency deviation. As stated above, the amount of the 
frequency deviation is determined by the amplitude of the modulating 
Doppler audio signal. 
A 455 KHz carrier signal is chosen because the 455 KHz frequency is 
generally above the frequencies of the light noise created by incandescent 
and fluorescent lights, so interference or noise created by such lights 
can be avoided or filtered out. Furthermore, the 455 KHz frequency falls 
within the band of frequencies often used in radio receivers, which allows 
standard and off-the-shelf components to be used in this stage of the 
signal processing of this invention. In the present invention, the 
frequency modulation of the 455 KHz carrier signal by the Doppler audio 
signal will cause the modulated Doppler audio signal to have a frequency 
between 440 KHz and 470 KHz. The 15 KHz limit on the variance from 455 KHz 
is caused by the limits placed on the amplitude of the Doppler audio 
signal by the Doppler transceiver 94. The output signal of the frequency 
modulator 98 is a frequency modulated voltage signal, where the amount of 
frequency modulation is based on the amplitude of the Doppler audio signal 
input from the volume control 96. Frequency modulators and frequency 
modulation of a carrier signal with an audio signal are well known to 
persons having ordinary skill in the art. For example, the XR-2209 
Precision Oscillator manufactured by Exar Corporation of San Jose, Calif., 
can be used in this invention as the frequency modulator 98. The frequency 
modulator 98 and its supporting electronic circuitry are shown in FIG. 14 
and will be discussed in more detail below. 
Looking again at FIG. 10, the output signal from the frequency modulator 98 
is the input signal to the bandpass filter 101. The bandpass filter 101 is 
centered at 455 KHz and passes electric signals having frequencies between 
440 KHz and 470 KHz. The bandpass filter 101 attenuates and filters out 
any electric signals or noise having frequencies below 440 KHz and 
electric signals or noise having frequencies above 470 KHz. The bandpass 
filter 101 is used to filter out unwanted signals or noise that did not 
originate from the reflected Doppler signal and to filter out harmonic 
signals which may be present on the output signal from the modulator 98. 
Bandpass filters are often used in AM and FM commercial radios and are 
well known to persons having ordinary skill in the art. For example, the 
SFG455B Ceramic Filter manufactured by muRata Erie of Smyrna, Ga., can be 
used in this invention for the bandpass filter 101. The bandpass filter 
101 is shown in FIG. 14 and will be discussed in more detail below. 
Looking again at FIG. 10, the output signal from the bandpass filter 101 is 
the input signal to the voltage-to-current converter 102. A significant 
feature of this invention includes the voltage-to-current converter 102, 
which converts the voltage-based and frequency modulated input signal to a 
current-based and frequency modulated output signal that flows through and 
drives the six infrared light emitting diodes (LEDs) 104 LED.sub.1, 
LED.sub.2, LED.sub.3, LED.sub.4, LED.sub.5 and LED.sub.6 (see FIGS. 4-6) 
located on the probe 54. Consequently, according to this invention, the 
frequency of the infrared light signal output of the six infrared LEDs 104 
LED.sub.1, LED.sub.2, LED.sub.3, LED.sub.4, LED.sub.5 and LED.sub.6 can 
have a specific relationship with the Doppler audio signal, as will be 
discussed in more detail below. The six LEDs 104 LED.sub.1, LED.sub.2, 
LED.sub.3, LED.sub.4, LED.sub.5 and LED.sub.6, are sized, shaped, and 
positioned in the probe 54 in a manner that maximizes multidirectional or 
global broadcasting of the infrared carrier medium of the Doppler audio 
signal to enhance the likelihood that it will reach and be received by the 
base unit 52 uninterrupted. 
The LEDs 104 are located at the end 62 of the probe opposite the detachable 
nose 60, as shown in FIGS. 4, 5, and 6. Furthermore, the six infrared LEDs 
104 are positioned so that two (LED.sub.1, LED.sub.4) of the infrared LEDs 
104 emit infrared light in a direction outward and parallel to the 
longitudinal center axis of the probe 54. The four (LED.sub.2, LED.sub.3, 
LED.sub.5, LED.sub.6) remaining infrared LEDs 104 are positioned in 
approximately the same plane so that they emit infrared light outward 
from, and perpendicular to, the probe 54. Furthermore, each of the four 
remaining infrared LEDs 104 LED.sub.2, LED.sub.3, LED.sub.5, and LED.sub.6 
emits infrared light in a direction that is either ninety degrees 
(90.degree.) or one hundred eighty degrees (180.degree.) apart from the 
direction of each of the other three. As previously discussed, with this 
configuration of the six infrared LEDs 104 LED.sub.1, LED.sub.2, 
LED.sub.3, LED.sub.4, LED.sub.5, and LED.sub.6, the infrared light signals 
emitted by the six infrared LEDs 104 LED.sub.1, LED.sub.2, LED.sub.3, 
LED.sub.4, LED.sub.5, and LED.sub.6 are transmitted in a plurality of 
directions. The light signals will reflect off the walls, ceiling, 
equipment, and other objects in the room where the medical diagnostic 
device 50 is being used. Having the six infrared LEDs 104 LED.sub.1, 
LED.sub.2, LED.sub.3, LED.sub.4, LED.sub.5, and LED.sub.6 oriented in 
different directions at the end 62 of the probe 54 ensures that at least 
one of the infrared light signals emitted by the six infrared LEDs 104 
LED.sub.1, LED.sub.2, LED.sub.3, LED.sub.4, LED.sub.5, and LED.sub.6 will 
be detected by the photodetectors 120 DET.sub.1, DET.sub.2, DET.sub.3, and 
DET.sub.4 located on the base unit 52. 
The voltage-to-current converter 102 follows a linear relationship when 
converting the voltage levels of its input signal to the current levels of 
its output signal. A current-based signal is better suited than a 
voltage-based signal to drive the six infrared LEDs 104, LED.sub.1, 
LED.sub.2, LED.sub.3, LED.sub.4, LED.sub.5 and LED.sub.6, because the 
amount of light emitted by each of the six infrared LEDs 104 LED.sub.1, 
LED.sub.2, LED.sub.3, LED.sub.4, LED.sub.5 and LED.sub.6 depends entirely 
on the current flowing through it, and the voltage-to-current ratio in a 
LED is very nonlinear. Therefore, using the voltage-to-current converter 
102 before driving the six infrared LEDs 104 LED.sub.1, LED.sub.2, 
LED.sub.3, LED.sub.4, LED.sub.5 and LED.sub.6, ensures that the amount or 
intensity of infrared light emitted by the LEDs 104 LED.sub.1, LED.sub.2, 
LED.sub.3, LED.sub.4, LED.sub.5 and LED.sub.6 will vary in a linear 
relationship with the voltage level of the input signal to the 
voltage-to-current converter 102. If the voltage-to-current converter 102 
was not used, and the output signal from the bandpass filter 98 was passed 
directly to the six infrared LEDs 104 LED.sub.1, LED.sub.2, LED.sub.3, 
LED.sub.4, LED.sub.5 and LED.sub.6, the amount or intensity of the 
infrared light emitted by the six infrared LEDs 104 LED.sub.1, LED.sub.2, 
LED.sub.3, LED.sub.4, LED.sub.5 and LED.sub.6 would vary in a nonlinear 
relationship with the voltage level of the input signal to the six 
infrared LEDs 104 LED.sub.1, LED.sub.2, LED.sub.3, LED.sub.4, LED.sub.5 
and LED.sub.6. The OEDEL-1L1 infrared light emitting diode manufactured by 
Lumex of Palatine, Illinois, can be used in this invention for LED.sub.1 
and LED.sub.4. The SFH-487P infrared light emitting diode manufactured by 
Siemens Components, Inc., of Cupertino, Calif. can be used in this 
invention for LED.sub.2, LED.sub.3, LED.sub.5, and LED.sub.6. The 
electronic circuitry for the voltage-to-current converter 102 is shown in 
FIG. 15 and will be discussed in more detail below. 
Looking at FIGS. 4 and 5, the components of the base unit 52 will now be 
discussed in greater detail. The user activates the base unit 52 by 
momentarily pressing the on/off switch 152 to place the on/off control 150 
into the "on" state. Placing the on/off control 150 into the "on" state 
provides connection between the rechargeable battery 154 in the base unit 
52 and the remainder of the electronic circuitry located in the base unit 
52. The user deactivates the base unit 52 by momentarily pressing the 
on/off switch 152 again to place the on/off control 150 into the "off" 
state. Placing the on/off control 150 into the "off" state by momentarily 
pressing the on/off switch 152 disconnects the rechargeable battery 154 in 
the base unit 52 from the remainder of the electronic circuitry located in 
the base unit 52. 
As discussed above, the on/off control 150 provides connection between the 
battery 154 and the power supplies 156 when the on/off control 150 is in 
the "on" state. The power supplies 156 take the voltage supplied by the 
rechargeable battery 154 and convert it to create the necessary voltages 
and voltage levels used by the electronic components in the electronic 
circuitry in the base unit 52. The ten V170R cells manufactured by Varta 
Batteries, Inc., can be used in the rechargeable battery 154 in the base 
unit 52. The use of power supplies and power supply circuity to create 
different voltage levels is well known to people having ordinary skill in 
the art and need not be shown or described in more detail. 
The battery 154 in the base unit 52 is rechargeable. The recharger 158 is 
used along with the charge circuit 160 to recharge the rechargeable 
battery 154 in the base unit 52. When the base unit 52 is properly 
positioned (not shown) in the recharging stand 56, and the recharger 158 
is plugged into an electrical outlet (not shown), the recharger 158 drives 
the charge circuit 160 in the base unit 52 to recharge the rechargeable 
battery 154 in the base unit 52. Likewise, the recharger 158 is used along 
with the charge circuit 164 to recharge the rechargeable battery 107 in 
the probe 54. When the probe 54 is correctly positioned in the probe 
holder 57 in the base unit 52 (see FIGS. 1 and 9), the base unit 52 is 
positioned in the recharging stand 56, and the recharger 158 is plugged 
into an electrical outlet (not shown), the recharger 158 drives the charge 
circuit 164 to recharge the rechargeable battery 107 in the probe 54. The 
use of recharging circuitry and rechargeable batteries is well known to 
people having ordinary skill in the art and need not be shown or described 
in further detail for purposes of this invention. 
The user of the medical diagnostic device 50 uses the select controller 170 
and the select switch 172 to signal the microcontroller 146 that the 
medical diagnostic device 50 is operating in the obstetrical mode or in 
the vascular mode. The user of the medical diagnostic device 50 selects 
either the obstetrical function or the vascular function through the 
select controller 170 by pressing the select switch 172 to select either 
the vascular position (not shown) or the obstetrical position (not shown) 
for the purpose or advantages described above. Alternately pressing the 
select switch 172 causes the select controller 170 to provide an signal to 
the microcontroller 146 alternately indicating that the obstetrical 
function or the vascular function has been chosen by the user. Depending 
on whether the vascular mode or the obstetric mode is indicated by the 
select controller 170, the microcontroller 146 processes either the 
vascular waveform signal obtained from the vascular converter 140 or the 
fetal heart rate waveform signal obtained from the fetal heart rate 
generator 142. The microcontroller also provides a visual display signal 
to the display driver 186 which provides a digital signal to the digital 
display 188 on the base unit 52. 
The microcontroller 146 is a standard off-the-shelf microprocessor. For 
example, the 80196 manufactured by the Intel Corporation of Santa Clara, 
Calif., can be used in this invention for the microcontroller 146. The 
microcontroller 146 has an associated memory 182 and an associated voltage 
reference 184. The memory 182 is used to store the program and software 
data used in the microcontroller 146. The voltage reference 184 is used to 
set the amplitude of the signals presented to the ADC 200 from the 
vascular heart rate converter 140 and the fetal heart rate generator 142. 
The voltage reference 184 sets the scale of the digital output of the ADC 
200. Full scale output of the ADC 200 is equal to voltage reference value 
from voltage reference 184. 
The operational block diagram for the electronic circuitry in the base unit 
52, which receives and processes the frequency modulated sine wave 
infrared light signals emitted by the LEDs 104 LED.sub.1, LED.sub.2, 
LED.sub.3, LED.sub.4, LED.sub.5 and LED.sub.6, is shown in FIGS. 4 and 5. 
Looking at FIG. 11, the infrared light signals emitted by the six infrared 
LEDs 104 LED.sub.1, LED.sub.2, LED.sub.3, LED.sub.4, LED.sub.5 and 
LED.sub.6 on the probe 54 are detected by one or more of the 
photodetectors 120 DET.sub.1, DET.sub.2, DET.sub.3, and DET.sub.4 located 
on the base unit 52 (see FIGS. 7-9). The photodetectors 120 DET.sub.1, 
DET.sub.2, DET.sub.3, and DET.sub.4, are each capable of detecting and 
converting the infrared light signals to electric signals, which still 
carry the Doppler audio signals produced in the probe 54, as described 
above. 
The frequency modulated infrared light signals emitted by the LEDs 104 
LED.sub.1, LED.sub.2, LED.sub.3, LED.sub.4, LED.sub.5 and LED.sub.6 on the 
probe 54 have a frequency that varies between 440 KHz and 470 KHz. 
Therefore, the output electric signals from the photodetectors 120 
DET.sub.1, DET.sub.2, DET.sub.3, and DET.sub.4 also have a frequency that 
varies between 440 KHz and 470 KHz. The photodetectors 120 DET.sub.1, 
DET.sub.2, DET.sub.3, and DET.sub.4 are positioned on the base unit 52 so 
as to ensure that the frequency modulated infrared light signals emitted 
by the six infrared LEDs 104 LED.sub.1, LED.sub.2, LED.sub.3, LED.sub.4, 
LED.sub.5 and LED.sub.6 are either directly detected by the photodetectors 
120 DET.sub.1, DET.sub.2, DET.sub.3, and DET.sub.4 or are detected by the 
photodetectors 120 DET.sub.1, DET.sub.2, DET.sub.3, and DET.sub.4 after 
the infrared light signals reflect off of the walls, the ceiling, or other 
objects in the room where the medical diagnostic device 50 is being used. 
This is accomplished by positioning the photodetectors 120 DET.sub.1, 
DET.sub.2, DET.sub.3, and DET.sub.4 on the base unit 52 such that the 
active surfaces of the photodetectors 120 DET.sub.1, DET.sub.2, DET.sub.3, 
and DET.sub.4 are aligned to intercept infrared light from a wide variety 
of angles, including both the horizontal and the vertical planes, as shown 
in FIGS. 7-9. The SFH205 photodetectors manufactured by Siemens 
Components, Inc., of Cupertino, Calif., can be used as the photodetectors 
120 DET.sub.1, DET.sub.2, DET.sub.3, and DET.sub.4 in this invention. 
The infrared light signals detected by the photodetectors 120 DET.sub.1, 
DET.sub.2, DET.sub.3, and DET.sub.4 create electric signals in which the 
current varies linearly with the amplitude or intensity of the infrared 
light signal emitted by the LEDs 104 LED.sub.1, LED.sub.2, LED.sub.3, 
LED.sub.4, LED.sub.5 and LED.sub.6 detected by the photodetectors 120 
DET.sub.1, DET.sub.2, DET.sub.3, and DET.sub.4. The output electric 
signals from the photodetectors 120 DET.sub.1, DET.sub.2, DET.sub.3, and 
DET.sub.4 are the input signals to the resonant circuit 122, which 
converts the current output of the photodetectors 120 DET.sub.1, 
DET.sub.2, DET.sub.3, and DET.sub.4 into a voltage, which is the input 
signal to the preamplifier 124. In addition, the resonant circuit 122 
limits the bandwidth of the electric signal provided to the preamplifier 
124. The resonant circuit 122 and its supporting electronic circuitry are 
shown in FIG. 16 and will be discussed in more detail below. 
The frequency modulated electric output signal from the resonant circuit 
122 is the input signal to the preamplifier 124, which amplifies the 
electric signal to strengthen it by adding more power before proceeding to 
additional signal processing components. The output signal from the 
preamplifier 124 is the input signal to the bandpass filter 126. The 
bandpass filter 126 attenuates and filters electric signals having 
frequencies below 440 KHz and electric signals having frequencies above 
470 KHz. The bandpass filter 126 is used to filter out unwanted signals 
that did not originate from the six infrared LEDs 104 LED.sub.1, 
LED.sub.2, LED.sub.3, LED.sub.4, LED.sub.5 and LED.sub.6 on the probe 54 
that are detected by the photodetectors 120 DET.sub.1, DET.sub.2, 
DET.sub.3, and DET.sub.4, and to filter out electric noise signals that 
may be created by the base unit 52 or by lights and equipment located 
where the medical diagnostic device 50 is being used. In addition, the 
bandpass filter circuit 126 cleans up the signal by removing extraneous 
frequencies or noise that may have produced in the preamplifier circuit 
124. The SFG455B Ceramic Filter manufactured by muRata Erie of Smyrna, 
Ga., can be used in this invention for the bandpass filter 126. The 
bandpass filter 126 and its supporting electronic circuitry are shown in 
FIG. 16 and will be discussed in more detail below. 
The output signal from the bandpass filter 126 is the input signal to the 
frequency demodulator 128. The frequency demodulator 128, which may be an 
integrated circuit (IC), amplifies, demodulates, and filters the Doppler 
audio signal encoded within the frequency modulated infrared light signal 
detected by the photodetectors 120. Frequency demodulators are well known 
by people having ordinary skill in the art. For example, the NE/SA604A 
High-Performance Low-Power FM IF System manufactured by the Phillips 
(formerly the Signetics Company) of Sunnyvale, Calif., can be used in this 
invention for the frequency demodulator 128. The output signal from the 
frequency demodulator 128 is an electric signal that varies in amplitude 
and frequency and the frequency of the output signal is in the audio 
range, that is, the frequency of the output signal is between 100 Hertz 
and one (1) KHz. In general terms, the output signal of the frequency 
demodulator 128 represents the Doppler audio signal discussed above, which 
was the input signal to the frequency modulator 98 in the probe 54. 
Therefore, the output signal from the frequency demodulator 128 is the 
recovered Doppler audio signal, which was provided by the Doppler 
transceiver 94 and volume control 96 in the probe 54. In more specific 
terms, the output signal from the frequency demodulator 128 is an electric 
signal that has a frequency that is a function of the rate of change of 
the frequency of input signal to the frequency demodulator 128, and it has 
an amplitude that is a function of the instantaneous deviation of the 
frequency of the input signal to the frequency demodulator 128 from 455 
KHz. The output signal from the frequency demodulator 128 is the Doppler 
audio signal that was the output signal from the Doppler transceiver 94. 
The frequency demodulator 128 and its supporting electronic circuitry are 
shown in FIG. 17 and will be discussed in more detail below. 
The Doppler audio output signal from the frequency demodulator 128 is the 
input signal to the Doppler audio controller 130. The Doppler audio 
controller 130 includes a low pass filter that further smooths the signal 
created by the frequency demodulator 128 and eliminates high frequencies 
that might be included in the electric signal generated by the frequency 
demodulator 128. The Doppler audio controller 130 also amplifies the 
signal with an amplifier after the signal passes through the low pass 
filter and then filters the signal with a high-pass filter to remove 
signals created by extraneous movement of the probe 54. Audio controllers 
are well known in the art. 
The output signal from the Doppler audio controller 130 is the input signal 
to the audio power amplifier 132, the tape recorder connector 138, the 
vascular converter 140, and the fetal waveform generator 142. The audio 
power amplifier 132 amplifies the Doppler audio signal so that the Doppler 
audio signal can be converted to acoustic sound waves and broadcast by the 
speakers 134 located in the base unit 52 or through headphones (not shown) 
connected to the base unit 52 though the headphone connector 136. The 
audio power amplifier 132 is a standard audio signal power amplifier and 
its use is well known to persons having ordinary skill in the art. For 
example, the TDA7052 manufactured by Phillips (formerly the Signetics 
Company) of Sunnyvale, Calif., can be used in this invention for the audio 
power amplifier 132. The amplified signal output from the audio power 
amplifier 132 is the input signal to both the speakers 134 located in the 
base unit 52 and the headphone connector 136 located in the base unit 52. 
A standard headphone (not shown) can then be plugged into the headphone 
connector 136 and used with the medical diagnostic device 50. The tape 
recorder connector 138 provides a standard connection to a tape recorder 
(not shown) so that the Doppler audio signal can be recorded and stored. 
The vascular converter 140 converts the Doppler audio signal from the 
Doppler audio controller 130 into a signal representing the Doppler audio 
signal and thus, the vascular heart rate of the patient being monitored. 
The vascular converter 140 converts the Doppler audio signal to the 
vascular waveform by the use of a frequency-to-voltage converter whose DC 
output voltage varies linearly with the average frequency of the input 
signal to the vascular converter 140 for the purpose of representing the 
complex audio signal with a relatively simple low frequency waveform. The 
use of frequency-to-voltage converters are well known to people having 
ordinary skill in the art. For example, the LM2907 manufactured by 
National Semiconductor, Inc., can be used for the vascular converter 140 
in this invention. The vascular waveform output signal from the vascular 
converter 140 is an input signal to the microcontroller 146, as shown in 
FIG. 12. 
The fetal heart rate generator 142 converts the Doppler audio signal from 
the Doppler audio controller 130 into a fetal waveform signal that is 
representative of fetal heart rate of the baby of the pregnant woman W 
being monitored. The fetal heart rate generator 142 converts the Doppler 
audio signal to the fetal waveform signal in the following steps. First, 
the amplitude of the Doppler audio signal is adjusted to a predetermined 
level by an automatic gain control. The automatic gain control holds the 
average of the amplitude of the Doppler audio signal at a constant level. 
The output of the automatic gain control is the input to an envelope 
detector which converts the Doppler audio signal into a waveform comprised 
of a DC voltage which varies linearly with the amplitude of the Doppler 
audio signal. The waveform signal from the envelope detector is amplified 
and passed through a low pass filter which further smooths the waveform. 
The function, design, and operation of the fetal heart rate converter 142 
are well known to persons having ordinary skill in the art. The fetal 
waveform output signal from the fetal heart rate generator 142 is an input 
signal to microcontroller 146, as shown in FIG. 12. 
The microcontroller 146 includes an analog-to-digital converter (ADC) 200 
that converts either the vascular waveform signal from the vascular 
converter 140 or the fetal waveform signal from the fetal heart rate 
generator 142 into a twelve bit digital output signal. The twelve bit 
output signal is scaled into a four bit signal which preserves enough 
accuracy to determine the heart rate accurately and enables the heart rate 
calculations to be completed quickly. The analog-to-digital conversion 
done by ADC 200 in the microcontroller 146 is controlled completely by 
software. 
After the microcontroller 146 converts either the analog input signal from 
the vascular converter 140 or the analog signal from the fetal heart rate 
generator 142 into a digital signal, the output digital signal from the 
ADC 200, corresponding to either the vascular waveform from the vascular 
generator 140 or the fetal waveform from the fetal heart rate generator 
142, becomes the input signal to the heart rate calculator 202 in the 
microcontroller 146. The output of the select controller 170 determines 
which digital signal in the ADC 200 becomes the input signal to the heart 
rate calculator 202. 
The function of the heart rate calculator 202 is performed completely in 
software in the microcontroller 142. The heart rate calculator 202 takes 
the digital input signal from the ADC 200 and calculates the corresponding 
heart rate associated with the digital signal. Standard and well known 
mathematical techniques such as, for example, techniques using 
autocorrelation functions or modulus difference functions performed on a 
set of data samples, can be used to determine the period of the digital 
signal. Once the period of the digital signal is determined, a conversion 
to heart beats per minute can be easily made, as is well known in the art. 
The heart rate calculator 202 also has an input from the heart rate average 
switch 204. When the user momentarily presses the heart rate average 
switch 204, the heart rate calculator 202 calculates the average heart 
rate from the digital data gathered during the previous three (3) seconds. 
The average heart rate is displayed for ten (10) seconds and the heart 
rate calculator 202 then returns to sending the current heart rate to the 
display driver 186. The voltage V.sub.S supplied to the heart rate 
selector switch 204 is provided by the power supplies 156 and can be, for 
example, 5.0 volts. 
The output signal from the heart rate calculator 202 is the input signal to 
the display driver 186 which is used to drive the three digit heart rate 
LED display 188 located on the base unit 52. The use of a microcontroller 
to drive a LED display is well known to people having ordinary skill in 
the art. 
More detailed exemplary schematic diagrams for the significant parts of the 
electronic circuitry in the medical diagnostic device 50 are shown in 
FIGS. 7-11. In describing and discussing the electronic components, the 
standard symbols will be used. For example, "R" will be used for 
resistors, "C" will be used for capacitors, "L" will be used for 
inductors, "D" will be used for diodes, "Q" will be used for transistors, 
"OA" will be used for operational amplifiers, "V" will be used for 
voltages, and "I" will be used for currents. 
The electronic components and the electronic circuitry for the volume 
control 96, the frequency modulator 98, and the bandpass filter 101 are 
shown schematically in FIG. 14. The volume control 96, which receives the 
Doppler audio signal on lead 300, includes a fixed value resistor R.sub.1 
for setting the minimum voltage level and a variable resistor R.sub.2 for 
varying the voltage level. The value for the variable resistor R.sub.2 is 
controlled by the user through the operation of the slide potentiometer 97 
located on the probe 54 (see FIGS. 2-5). The user can adjust the slide 
potentiometer 97 to increase or decrease the voltage level of the output 
signal from the volume control 96 on lead 302. 
The output signal of the volume control 96 on lead 302 is the input signal 
of the frequency modulator 98 on lead 304. As discussed above, frequency 
modulator 98 uses the signal from the volume control 96 to frequency 
modulate a carrier signal. The carrier signal can be any convenient 
frequency for purposes of this invention, but a frequency of 455 KHz is 
preferred for the reasons described above. In the resulting frequency 
modulated (FM) output signal from the frequency modulator 98 on lead 306, 
the deviation of the instantaneous frequency of the output signal from the 
frequency of the carrier signal is directly proportional to the 
instantaneous amplitude of the input signal on lead 304 to the frequency 
modulator 98. As discussed above, the XR-2209 Precision Oscillator 98 
manufactured by Exar Corporation of San Jose, Calif., can be used as the 
frequency modulator function. The electrical specifications and the 
operational characteristics for the Exar XR-2209 Precision Oscillator are 
provided in the Exar Databook, copyrighted 1992. In the electronic circuit 
for the frequency modulator 98 shown in FIG. 14, the capacitor C.sub.1 
acts as a DC block to eliminate external DC voltages on the input lead 304 
to the frequency modulator 98. The capacitor C.sub.2 acts as an AC filter 
to ground for the required supply voltage V.sub.DD so that high frequency 
signals are not supplied to the frequency modulator 98. The voltage 
V.sub.DD is supplied by the power supplies 109 in the probe 54 (see FIG. 
10) and can be, for example, 12 volts. The variable resistor R.sub.3 and 
the fixed value resistors R.sub.4 and R.sub.5, along with the capacitor 
C.sub.3, set the frequency of the carrier signal of the frequency 
modulator 98 to the desired frequency, such as 455 Khz. The fixed value 
resistor R.sub.8 and the capacitor C.sub.4 are used to supply the 
necessary bias voltage V.sub.B to the frequency modulator 98. The bias 
voltage V.sub.B is supplied by the power supplies 109 in the probe 54 (see 
FIG. 10), and can be, for example, 5.5 volts. The output from the 
frequency modulator 98 is amplitude adjusted by the fixed value resistor 
R.sub.6 and the variable resistor R.sub.7 configured in a classic voltage 
divider configuration. The variable resistor R.sub.7 is preferably preset 
so that the peak-to-peak voltage of the output signal on lead 306 from the 
frequency modulator 98 is two times V.sub.D1, the voltage drop across the 
diode D.sub.1 shown in FIG. 15. The diode D.sub.1 and the voltage V.sub.D1 
will be discussed in more detail below. 
It is desirable to have the frequency modulated output signal from the 
frequency modulator 98 be a sine wave, because sine wave signals do not 
produce harmonic frequencies and therefore create fewer high frequency 
signals that might cause interference with the Doppler transceiver 94 or 
other equipment located where the medical diagnostic device 50 is being 
used. The Exar XR-2209 Precision Oscillator 98, however, creates only a 
frequency modulated square wave output voltage signal or a frequency 
modulated triangle wave output voltage signal, both of which may contain 
harmonic sine wave signals. The triangle wave frequency modulated output 
signal from the precision oscillator 98 is chosen for this invention 
because a triangle wave is closer to a sine wave than is a square wave, 
and subsequent circuit components described below are used to convert the 
triangle wave output on lead 306 to a sine wave. Therefore, the output 
from the frequency modulator 98 on lead 306 is preferably, but not 
necessarily, a triangle wave voltage signal whose carrier frequency of 455 
KHz is frequency modulated to as much as plus or minus 15 KHz by amplitude 
of the voltage in the Doppler audio signal from the volume control 96. 
Therefore, the output signal from the frequency modulator 98 on lead 306 
is preferably a frequency modulated triangle wave voltage signal that has 
a center frequency of 455 KHz, whose frequency varies between 440 KHz and 
470 KHz, and that has an amplitude that varies between plus V.sub.D1 volts 
and minus V.sub.D1 volts. The frequency of the triangle wave output 
voltage signal from the frequency modulator 98 on lead 306 varies as a 
function of the amplitude of the voltage of the input Doppler audio signal 
to the frequency modulator 98. 
The output signal from the frequency modulator 98 on lead 306 is the input 
signal to the bandpass filter 101 on lead 307 which is used to filter out 
unwanted electric signals that are not part of the Doppler audio signal 
and to filter out electronic noise signals that may be created by the 
probe 54. For example, electronic noise signals might be caused by the 
supply voltage V.sub.DD and the biasing voltage V.sub.B, as well as by the 
electrical equipment in the room where the medical diagnostic device 50 is 
being used. Another function of the bandpass filter 101 is to smooth and 
convert the triangle wave voltage output signal from the frequency 
modulator 98 so that it becomes a sine wave voltage output signal from the 
bandpass filter 101. As discussed above, the SFG455B Ceramic Filter 
manufactured by muRata Erie of Smyrna, Ga., can be used in this invention 
for the bandpass filter 101. The electrical specifications and the 
operational characteristics for the SFG455B Ceramic Filter are given in 
muRata Erie Ceramic Filters, catalog no. P-03-B, page 12. The bandpass 
filter 101 is preferably centered at 455 KHz and preferably passes signals 
having frequencies between 440 KHz and 470 KHz. The bandpass filter 101 
attenuates and filters electric signals having frequencies below 440 KHz 
and electric signals having frequencies above 470 KHz. Therefore, the 
output signal from the bandpass filter 101 on lead 308 is a frequency 
modulated sine wave voltage signal that has a center frequency of 455 KHz, 
whose frequency varies between 440 KHz and 470 KHz, and that has an 
amplitude that is between plus V.sub.D1 volts and minus V.sub.D1 volts. 
Referring now to FIG. 15, the output signal from the bandpass filter 101 on 
lead 308 is the input signal to the voltage-to-current converter circuit 
102, which creates two output currents on leads 308 and 310, respectively, 
that vary in intensity as a function of the voltage amplitude in the 
signal on the input lead 308. The voltage-to-current converter 102 
converts the voltage of the frequency modulated sine wave voltage input 
signal from the bandpass filter 101 into a corresponding frequency 
modulated sine wave current output signal that is used to drive the 
infrared LEDs 104 LED.sub.1, LED.sub.2, LED.sub.3, LED.sub.4, LED.sub.5, 
and LED.sub.6. This conversion allows the LEDs 104 LED.sub.1, LED.sub.2, 
LED.sub.3, LED.sub.4, LED.sub.5, and LED.sub.6 to emit frequency modulated 
sine wave infrared light signals that correspond to the frequency 
modulated triangle wave voltage output signal from the frequency modulator 
98. The capacitor C.sub.10 acts as a block to prevent any DC voltage bias 
from the output of the bandpass filter 101 from existing on the input 
signal to the voltage-to-current converter 102. 
The voltage-to-current converter 102 preferably, but not necessarily, 
includes two operational amplifiers OA.sub.1 and OA.sub.2, because 
separating the LEDs 104 into two groups, one group including LEDs 104 
LED.sub.1, LED.sub.2, and LED.sub.3 and the second group including 
LED.sub.4, LED.sub.5 and LED.sub.6, allows use of lower supply voltages 
V.sub.1 and V.sub.2, and hence lower battery 107 voltage. The LF353 
operational amplifiers manufactured by National Semiconductor, Inc., of 
Santa Clara, Calif., can be used in this invention in the 
voltage-to-current converter circuit 102. The configurations for both 
operational amplifiers OA.sub.1 and OA.sub.2 are identical and serve the 
same function. That is, the voltage V.sub.1 serves the same function and 
has the same value as the voltage V.sub.2, the resistor R.sub.11 serves 
the same function and has the same value as the resistor R.sub.15, the 
resistor R.sub.12 serves the same function and has the same value as the 
resistor R.sub.16, the resistor R.sub.14 serves the same function and has 
the same value as the resistor R.sub.17, the capacitor C.sub.8 has the 
same value and serves the same function as the capacitor C.sub.9, the 
transistor Q.sub.1 is identical to and serves the same function as the 
transistor Q.sub.2. The LED 104 LED.sub.1 is identical to, and serves the 
same function as, the LED 104 LED.sub.4 --both are narrow beam positioned 
as to achieve maximum reflection off of the walls or the ceilings in the 
room where the medical diagnostic device 50 is being used, as will be 
described in more detail below. The LEDs 104 LED.sub.2, LED.sub.3, 
LED.sub.5, and LED.sub.6 are identical, serve the same function, and are 
wide beam positioned so as to maximize coverage in the room where the 
medical diagnostic device 50 is being used, as will also be described in 
more detail below. Since the configurations for the operational amplifiers 
OA.sub.1 and OA.sub.2 are identical and serve the same function, only the 
configuration for operational amplifier OA.sub.1 will be discussed and 
described below. 
The operational amplifier OA.sub.1, the resistor R.sub.12, and the 
transistor Q.sub.1 are arranged in a negative feedback configuration so as 
to function as a voltage-to-current converter with resistor R.sub.11. This 
type of circuit configuration is well known to people having ordinary 
skill in the art. The voltages V.sub.1 and V.sub.2, in addition to the 
voltage V.sub.DD discussed above, are provided by the power supplies 109 
in the probe 54. For example, as shown in FIG. 13, the voltages V.sub.1 
and V.sub.2 can be created by coupling the rechargeable battery 107 to the 
capacitors C.sub.5 and C.sub.6 and the inductors L.sub.1 and L.sub.2 which 
act as an AC noise filter to provide the dean DC voltages V.sub.1 and 
V.sub.2. The voltage V.sub.1 is equal to the voltage V.sub.2, and can be, 
for example, 7.2 volts. The voltage is the required supply voltage to 
power the operational amplifier OA.sub.1, and can be, for example, 12 
volts. The capacitor C.sub.8 provides an AC ground for the voltage 
V.sub.DD so that only DC voltage is supplied to the operational amplifier 
OA.sub.1. 
The voltage signal at the positive terminal 312 of the operational 
amplifier OA.sub.1 includes a DC voltage signal and an AC voltage signal. 
The DC voltage signal at the positive terminal 312 of the operational 
amplifier OA.sub.1 is created by the voltage V.sub.2, the diode D.sub.1, 
the capacitor C.sub.7, and the resistors R.sub.9 and R.sub.10. The AC 
voltage signal at the positive terminal 312 of the operational amplifier 
OA.sub.1 is provided from the frequency modulator 98, the bandpass filter 
101, and the capacitor C.sub.10, as previously discussed. 
The voltage V.sub.D1 is the voltage drop over the diode D.sub.1 created by 
current flowing through the diode D.sub.1. The capacitor C.sub.7 provides 
an AC ground for any AC signals that may be created by the voltage 
V.sub.2. The resistor R.sub.9 is a load resistor for the current flowing 
out of the diode D.sub.1. The resistor R.sub.10 couples the DC bias 
voltage created by the voltage V.sub.2, the capacitor C.sub.7, and the 
resistor R.sub.9 to the positive terminal 312 input of the operational 
amplifier OA.sub.1. The resistor R.sub.10 also provides a high resistance 
for AC signals created by the voltage V.sub.2 from the positive terminal 
312 input of the operational amplifier OA.sub.1. Since the input 
resistance for the operational amplifier OA.sub.1 is very high, there is 
no DC current flowing into the positive terminal 312 of the operational 
amplifier OA.sub.1. There is also no DC current flowing through the 
capacitor C.sub.10. Therefore, no current flows through the resistor 
R.sub.10, and there is no voltage drop over the resistor R.sub.10. 
Therefore, the DC voltage signal at the positive terminal 312 of the 
operational amplifier OA.sub.1 is equal to V.sub.2 minus V.sub.D1 volts. 
As previously discussed, the AC voltage signal at the positive terminal 312 
of the operational amplifier OA.sub.1 is the frequency modulated sine wave 
voltage signal created by the frequency modulator 98, the bandpass filter 
101, and the capacitor C.sub.10. The output signal from the capacitor 
C.sub.10 is a frequency modulated sine wave voltage signal that has a 
center frequency of 455 KHz, whose frequency varies between 440 KHz and 
470 KHz, and that has an amplitude that varies between plus V.sub.D1 volts 
and minus V.sub.D1 volts. Therefore, the total voltage signal at the 
positive terminal 312 of the operational amplifier OA.sub.1, comprised of 
the DC voltage signal plus the AC voltage signal, has a range between 
V.sub.2 volts and V.sub.2 minus 2V.sub.D1 volts. 
Due to the high gain of the operational amplifier OA.sub.1 and the negative 
feedback configuration established with the operational amplifier OA.sub.1 
by the use of resistor R.sub.12 and transistor Q.sub.1, the voltage at the 
negative terminal 313 of the operational amplifier OA.sub.1 is equal to 
the voltage at the positive terminal 312 of the operational amplifier 
OA.sub.1. Therefore, the voltage at the negative terminal 313 of the 
operational amplifier OA.sub.1 also varies between V.sub.2 volts and 
V.sub.2 minus 2V.sub.D1 volts. 
Having a variable voltage level at the negative terminal 313 of the 
operational amplifier OA.sub.1 also causes the current flowing through the 
resistor R.sub.11 to vary. The current I.sub.R11 flowing through the 
resistor R.sub.11 will vary between the current levels provided by the 
following equations: 
##EQU1## 
and Therefore, the AC current I.sub.R11 flowing through the resistor 
R.sub.11 ranges from zero (0) amperes to 2V.sub.D1 /R.sub.11 amperes. 
Consequently, the current I.sub.R11 depends only on the voltage V.sub.D1 
and the 
##EQU2## 
resistor R.sub.11 and does not depend on the voltage V.sub.2. As the 
voltage in the rechargeable battery 107 becomes depleted through use, the 
voltage V.sub.2 will also become reduced. The voltage drop V.sub.D1 across 
the diode D.sub.1, however, will remain approximately constant. With the 
components shown in this embodiment, the voltage V.sub.D1 across the diode 
D.sub.1 will remain approximately 0.55 volts. Therefore, the reduction in 
charge in the rechargeable battery 107 and the reduction of the voltage 
V.sub.2 will not affect either the voltage V.sub.D1 or the current 
I.sub.R11. 
Since the negative terminal 313 of the operational amplifier OA.sub.1 has a 
very high input impedance, none of the current I.sub.R11 will flow into 
the negative terminal 313 of the operational amplifier OA.sub.1. The 
current I.sub.LED flowing through the LEDs 104 LED.sub.1, LED.sub.2, and 
LED.sub.3 is equal to ((.beta.-1)/.beta.).times.I.sub.R11 where .beta. is 
the AC gain of the transistor Q.sub.1 and .beta. is typically equal to 
sixty (60) or greater. For the example components shown in this 
embodiment, the current I.sub.LED varies between zero (0) amperes and 
0.092 amperes. The resistor R.sub.14 is used to provide a means of 
sampling the current signal that flows through the infrared LEDs 104 
LED.sub.1, LED.sub.2, and LED.sub.3. The amount of infrared light emitted 
by the infrared LEDs 104 LED.sub.1, LED.sub.2, and LED.sub.3 is linearly 
related to the amount of the current I.sub.LED that flows through the 
infrared LEDs 104 LED.sub.1, LED.sub.2, and LED.sub.3. The infrared sine 
wave light signal emitted by the infrared LEDs 104 LED.sub.1, LED.sub.2, 
and LED.sub.3 has a varying intensity, a constant peak to peak amplitude, 
a center frequency of 455 KHz, and is frequency modulated between 440 KHz 
and 470 KHz. 
Referring now to FIG. 16, the frequency modulated infrared sine wave light 
signals emitted by the six infrared LEDs 104 LED.sub.1, LED.sub.2, 
LED.sub.3, LED.sub.4, LED.sub.5, and LED.sub.6 are detected in the 
photodetectors 120 DET.sub.1, DET.sub.2, DET.sub.3, and DET.sub.4 located 
on the base unit 52. As discussed above and as shown in FIGS. 7, 8, and 9, 
the photodetectors 120 DET.sub.1, DET.sub.2, DET.sub.3, and DET.sub.4 are 
arranged on the base unit 52 so that at least one of the infrared light 
signals emitted by the LEDs 104 LED.sub.1, LED.sub.2, LED.sub.3, 
LED.sub.4, LED.sub.5, and LED.sub.6 on the probe 54 are detected by the 
photodetectors 120 DET.sub.1, DET.sub.2, DET.sub.3, and DET.sub.4. The 
voltage V.sub.3 is generated by the power supplies 156 located in the base 
unit 52 and can be, for example, 7.5 volts. The resistor R.sub.18 and the 
capacitor C.sub.11 act as a filter to reduce noise in the circuit. 
Likewise, the resistor R.sub.19 and the capacitor C.sub.12 act as a filter 
to reduce noise in the circuit. 
The resistor R.sub.20, the capacitor C.sub.13 and the inductor L.sub.3 act 
together to create the resonant circuit 122 which converts the current 
signal output of the photodetectors 120 DET.sub.1, DET.sub.2, DET.sub.3, 
and DET.sub.4 into a voltage signal which is the input signal to the 
preamplifier 124. Furthermore, the resonant circuit 122 limits the 
bandwidth of the input signals provided to the preamplifier 124. 
The output signal of the resonant circuit 122 on lead 314 is the input 
signal to the preamplifier circuit 124, which includes the resistors 
R.sub.21, R.sub.22, R.sub.23, and R.sub.24, the capacitors C.sub.14 and 
C.sub.15, and the transistors Q.sub.3 and Q.sub.4. The preamplifier 124 is 
configured as a standard cascade amplifier--a type of amplifier that is 
well known to people having ordinary skill in the art. The cascade 
configured preamplifier 124 has high power gain, high input impedance, and 
low noise. The input to the preamplifier 124 is provided at lead 314 shown 
in FIG. 16, while the output from the preamplifier 124 is provided at lead 
316. The resistors R.sub.21 and R.sub.22 provide the necessary voltage 
bias for the transistor Q.sub.3. The resistor R.sub.23 provides the 
necessary voltage bias for the transistor Q.sub.4. The capacitor C.sub.14 
provides an AC ground for the gate of the transistor Q.sub.3. The 
capacitor C.sub.15 provides an AC bypass for the resistor R.sub.23. 
The output signal of the preamplifier 124 at lead 316 is a sine wave 
frequency modulated voltage signal and is the input signal on lead 316 to 
the bandpass filter 126. As discussed above, the SFG455B Ceramic Filter 
manufactured by muRata Erie of Smyrna, Ga., can be used in this invention 
for the bandpass filter 126. The bandpass filter 126 attenuates and 
filters electric signals from the preamplifier 124 that have frequencies 
below 440 KHz and electric signals that have frequencies above 470 KHz. 
The bandpass filter 126 is used to filter out unwanted signals that did 
not originate from the six infrared LEDs 104 LED.sub.1, LED.sub.2, 
LED.sub.3, LED.sub.4, LED.sub.5, and LED.sub.6 on the probe 54 that are 
detected by the photodetectors 120 DET.sub.1, DET.sub.2, DET.sub.3, and 
DET.sub.4 and to filter out electronic noise signals that may be created 
by the base unit 52 or by light and equipment located where the medical 
diagnostic device 50 of this invention is being used. 
The output signal from the bandpass filter 126 on lead 318 is the input 
signal to frequency demodulator 128 shown in FIG. 17. As discussed above, 
the NE/SA604A High-Performance Low-Power FM IF System manufactured by 
Phillips (formerly the Signetics Company) can be used in this invention 
for the frequency demodulator 128. The electrical specifications and 
operational characteristics for the Signetics NE/SA604A High-Performance 
Low-Power FM IF System are provided in the Signetics Linear Data Manual, 
Volume 1: Communications, copyrighted 1988. The voltage V.sub.4 is 
generated by the power supplies 156 located in the base unit 52 and can 
be, for example, 7.5 volts. The resistor R.sub.13 and the capacitor 
C.sub.27 act as a filter to reduce noise in the circuit. 
The purpose of the frequency demodulator 128, as discussed above, is to 
convert the 440 KHz to 470 KHz frequency modulated sine wave signal to an 
audio signal having the same characteristics as the Doppler audio signal 
which is the input to the frequency modulator 98 in probe 54. 
The schematic diagram of the frequency demodulator 128 and its supporting 
electronic circuitry are shown in FIG. 17, and the operational block 
diagram for the frequency demodulator 128 is shown in FIG. 11. Therefore, 
the following description of the frequency demodulator 128 and its 
associated circuits requires references to both FIGS. 4 and 10 
simultaneously. The input signal on pin 16 (see FIG. 17) to the frequency 
demodulator 128 is the output signal on lead 318 from bandpass filter 126 
(see FIG. 16), as just described above. This input signed on lead 318 and 
on pin 16 (see FIG. 17) first passes through the internal high gain 
amplifier 220 (see FIG. 11) before exiting the frequency demodulator 128 
on pin 14 (see FIG. 17). The capacitors C.sub.16 and C.sub.17 shown in 
FIG. 17 comprise the amplifier bypass filter 221 shown in FIG. 11 and 
provide the necessary coupling and filtering for the internal high gain 
amplifier 220. 
The electric signal exiting the frequency demodulator 128 on pin 14 passes 
through the bandpass filter 222 (see FIGS. 4 and 10) before returning as 
an input to the frequency demodulator 128 on pin 12 (see FIG. 17). The 
SFG455B Ceramic Filter manufactured by muRata Erie of Smyrna, Ga., can be 
used in this invention for the bandpass filter 222. The bandpass filter 
222 attenuates and filters electric signals that have frequencies below 
440 KHz and electric signals that have frequencies above 470 KHz. 
The output signal from the bandpass filter 222 is an input signal to the 
frequency demodulator 128 on pin 12 (see FIG. 17). The signal passes 
through the internal limiting amplifier 224 (see FIG. 11) which clips the 
input signal so that the output signal of the internal limiting amplifier 
224 is a square wave signal. The capacitors C.sub.18 and C.sub.19 shown in 
FIG. 17 comprise the limiter bypass filter 225 shown in FIG. 11 and 
provide filtering that the internal limiting amplifier 224 needs for 
stability. 
The square wave output signal from the internal limiting amplifier 224 is 
an output signal on pin 9 of the frequency demodulator 128 (see FIG. 17) 
which then passes through the phase shift circuit 226 (see FIG. 11) before 
re-entering the frequency demodulator 128 on pin 8 (see FIG. 17). The 
phase shift circuit 226, comprising capacitors C.sub.20, C.sub.21, and 
C.sub.22, the resistor R.sub.25, and the inductor L.sub.4 in FIG. 17, 
shifts the phase of the square wave output signal from the internal 
limiting amplifier 224. The amount of the phase shift .PHI. of the square 
wave signal in the phase shift circuit 226 is approximated by: 
where 
##EQU3## 
where f.sub.i is equal to the instantaneous frequency of the input signal 
to the phase shift circuit 226. The phase shifted square wave output 
signal from the phase shift circuit 226 becomes an input signal on pin 8 
(see FIG. 17) to the internal quadrature detector 228 (see FIG. 11). The 
square wave output signal from the internal limiting amplifier 224 is also 
passed directly to the internal quadrature detector 228, as illustrated in 
FIG. 11. For general information on quadrature detectors, see Herbert L. 
Krauss et at., SOLID STATE RADIO ENGINEERING, pages 310-314, published by 
John Wiley & Sons of New York City, (1980). The internal quadrature 
detector 228 multiplies the phase shifted square wave signal and the 
nonphase shifted square wave signal to create an electric signal that 
varies in amplitude at a frequency in the audio range. In other words, the 
output signal of the internal quadrature detector 228 has a frequency that 
is a function of the rate of change of the frequency of the nonphase 
shifted input signal and has an amplitude that is a function of the 
instantaneous deviation from 455 KHz of the frequency of the nonphase 
shifted input signal. 
The output signal from the internal quadrature detector 228 is an input 
signal to the internal mute switch 230 (see FIG. 11). The internal mute 
switch 230 provides the signal from the internal quadrature detector 228 
as an output signal on pin 6 of the frequency demodulator 128 (see FIG. 
17) when the probe 54 is turned "on" and when the amplitude of the 
electric signal from the internal quadrature detector 228 exceeds a 
predetermined and preset threshold value. If the probe 54 is turned "off," 
or if the amplitude of the electric signal that the internal mute switch 
230 receives from the internal quadrature detector 228 does not exceed the 
predetermined and preset threshold value, the internal mute switch 230 
provides no output signal on pin 6 of the frequency demodulator 128. 
Determination of whether the amplitude of the signal received by the 
internal mute switch 230 from the internal quadrature detector 228 exceeds 
the predetermined and preset threshold value is accomplished by the 
internal signal strength indicator 232, the buffer amplifier 234, the mute 
threshold detector 236, and the inverting level shifter 238, all of which 
are shown in FIG. 11. As shown in FIG. 11, the internal high gain 
amplifier 220 and the internal limiting amplifier 224 in the frequency 
demodulator 128 provide an output electric signal to the internal signal 
strength indicator 232. The output signal from the internal signal 
strength indicator 232 on pin 5 of the frequency demodulator 128 (see FIG. 
17) is an input signal on lead 320 to the buffer amplifier 234 in FIG. 18. 
The output signal of the internal signal strength indicator 232 is 
proportional to the signal strength at the input pin 16 of the frequency 
demodulator 128 and is in the form of a DC current. The resistor R.sub.26 
in FIG. 17 converts the DC current output signal on pin 5 of frequency 
demodulator 128 from the signal strength indicator 232 (FIG. 11) to a 
voltage signal. The capacitor C.sub.23 in FIG. 17 eliminates short current 
changes in the output signal from the internal signal strength indicator 
232. 
The buffer amplifier 234 in FIG. 18 lowers the impedance of the signal on 
lead 320 and provides an input signal on lead 322 to the mute threshold 
detector 236 in FIG. 18, which compares the signal strength of the input 
signal on lead 322 to the predetermined and preset threshold value 
discussed above. If the signal strength of the input signal to the mute 
threshold detector 236 is greater than the threshold value, the mute 
threshold detector 236 supplies a negative voltage signal to the inverting 
level shifter 238. The inverting level shifter 238 converts the negative 
voltage signal to a positive logic level signal needed to have the 
internal mute switch 230 provide an output signal on pin 6 of the 
frequency demodulator 128. If the signal strength of the input signal to 
the mute threshold detector 236 is not greater than the threshold value, 
the mute threshold detector 236 provides a positive voltage signal to the 
inverting level shifter 238. The inverting level shifter 238 converts the 
positive voltage signal to a negative logic level signal needed to have 
the internal mute switch 230 not provide an output signal on pin 6 of the 
frequency demodulator 128. 
The output signal from the frequency demodulator 128 on pin 6 (see FIG. 17) 
is low pass filtered by the capacitor C.sub.24. The resistor R.sub.27 
prevents signals from other circuits in the base unit 52 from reaching the 
frequency demodulator 128. The output signal from the low pass filter 240 
on lead 324 is an input signal to the Doppler audio controller 130 shown 
in FIG. 11 and discussed above. 
The schematic diagram for the buffer amplifier 234, mute threshold detector 
236, and the inverting level shifter 238 are shown in FIG. 18. The 
voltages V.sub.5, V.sub.6, and V.sub.7 are supplied by the power supplies 
156 (see FIG. 12) in the base unit 52 and all three can be, for example, 
7.5 volts. The capacitor C.sub.25 acts as an AC ground for the voltage 
V.sub.5. Likewise, the capacitor C.sub.26 acts as an AC ground for the 
voltage V.sub.6. 
The input of lead 320 to the buffer amplifier 234 is on the positive 
terminal of the operational amplifier OA.sub.3. The buffer amplifier 234 
includes the capacitor C.sub.25 and the operational amplifier OA.sub.3 in 
a standard and well-known buffer amplifier configuration to buffer the 
frequency demodulator 128 from the mute threshold detector 236. The LT1013 
operational amplifier manufactured by Linear Technology of Milpitas, 
Calif., can be used in the buffer amplifier 234. 
The output signal from the buffer amplifier 234 on lead 322 is the input 
signal to the mute threshold detector 236, as shown in FIG. 18. The mute 
threshold detector 236 includes the capacitor C.sub.26 and the resistors 
R.sub.28, R.sub.29, R.sub.30, R.sub.31, and R.sub.32. The input signal to 
the mute threshold detector 236 is on the negative terminal of the 
operational amplifier OA.sub.4. The LT1013 operational amplifier 
manufactured by Linear Technology of Milpitas, Calif., can be used in this 
invention in the mute threshold detector 236. The fixed voltage V.sub.8, 
the fixed value resistors R.sub.28 and R.sub.29, and the variable resistor 
R.sub.30 set the signal threshold to be used in the mute threshold 
detector 236. The resistor R.sub.31 is used to create a positive feedback 
signal, which prevents the output of the operational amplifier OA.sub.4 
from changing states if the change in the input signal level is very 
small. The resistor R.sub.32 is used to set the input current level to 
transistor Q.sub.5. The voltage V.sub.8 is supplied by the power supplies 
156 in the base unit 52 and can be, for example 7.5 volts. 
As discussed above, if the signal strength of the output signal from the 
buffer amplifier 234 is greater than the threshold value set by the 
voltage V.sub.8 and the resistors R.sub.28, R.sub.29, and R.sub.30, the 
mute threshold detector 236 supplies a negative voltage output signal on 
lead 324 to the inverting level shifter 238. If the signal strength of the 
output signal from the buffer amplifier 234 is not greater than the 
threshold value set by the voltage V.sub.8 and the resistors R.sub.28, 
R.sub.29, and R.sub.30, the mute threshold detector 236 provides a 
positive voltage output signal on lead 324 to the inverting level shifter 
238. 
The inverting level shifter 238 shown in FIG. 18 includes the resistor 
R.sub.33 and the transistor Q.sub.5. The output lead 326 of the inverting 
level shifter 238 in FIG. 18 is connected to pin 3 of the frequency 
demodulator 128 in FIGS. 4 and 10. The inverting level shifter 238 
converts a negative voltage signal provided by the mute threshold detector 
236 to the logic level signal needed on pin 3 of frequency demodulator 128 
to have the internal mute switch 230 in the frequency demodulator 128 
provide an output signal on pin 6 of the frequency demodulator 128. 
Similarly, the inverting level shifter 238 converts a positive voltage 
signal provided by the mute threshold detector 236 to the logic level 
signal needed to have the internal mute switch 230 in the frequency 
demodulator 128 not provide an output signal on pin 6 of the frequency 
demodulator 128. A positive signal provided by the mute threshold detector 
236 at lead 324 in FIG. 18 activates the transistor Q.sub.5 so that 
current flows through the resistor R.sub.33 which causes the voltage at 
lead 326 to be nearly equal to zero (0). A negative signal provided by the 
mute threshold detector 236 at lead 324 does not activate the transistor 
Q.sub.5. Therefore, no current flows through the resistor R.sub.33, and 
the voltage at lead 326 is equal to V.sub.7. Inverter circuits such as the 
inverting level shifter 238 are well known to people having ordinary skill 
in the art. 
For purposes of examples and not for limitations, the exemplary component 
values for the resistors, capacitors, and inductors listed in the 
following table can be used in the circuits of the diagnostic device 50, 
although other component designs can also be used. 
______________________________________ 
Part 
Component 
Identifier 
Value Manufacturer 
Number 
______________________________________ 
Resistor 
R.sub.1 1,800 
ohms 
Resistor 
R.sub.2 0-10,000 
(variable) ohms 
Resistor 
R.sub.3 0-5,000 
(variable) ohms 
Resistor 
R.sub.4 6,800 
ohms 
Resistor 
R.sub.5 68,000 
ohms 
Resistor 
R.sub.6 3,300 
ohms 
Resistor 
R.sub.7 0-5,000 
(variable) ohms 
Resistor 
R.sub.8 2,200 
ohms 
Resistor 
R.sub.9 10,000 
ohms 
Resistor 
R.sub.10 12,000 
ohms 
Resistor 
R.sub.11 12 
ohms 
Resistor 
R.sub.12 100 
ohms 
Resistor 
R.sub.13 10 
ohms 
Resistor 
R.sub.14 1 
ohm 
Resistor 
R.sub.15 12 
ohms 
Resistor 
R.sub.16 100 
ohms 
Resistor 
R.sub.17 1 
ohm 
Resistor 
R.sub.18 1,000 
ohms 
Resistor 
R.sub.19 100 
ohms 
Resistor 
R.sub.20 10,000 
ohms 
Resistor 
R.sub.21 10,000 
ohms 
Resistor 
R.sub.22 15,000 
ohms 
Resistor 
R.sub.23 1,000 
ohms 
Resistor 
R.sub.24 1,500 
ohms 
Resistor 
R.sub.25 20,000 
ohms 
Resistor 
R.sub.26 100,000 
ohms 
Resistor 
R.sub.27 100 
ohms 
Resistor 
R.sub.28 20,000 
ohms 
Resistor 
R.sub.29 5,600 
ohms 
Resistor 
R.sub.30 0-5,000 
(variable) ohms 
Resistor 
R.sub.31 220,000 
ohms 
Resistor 
R.sub.32 47,000 
ohms 
Resistor 
R.sub.33 47,000 
ohms 
Capacitor 
C.sub.1 0.04 
microfarads 
Capacitor 
C.sub.2 0.047 
microfarads 
Capacitor 
C.sub.3 220 
picofarads 
Capacitor 
C.sub.4 4.7 
microfarads 
Capacitor 
C.sub.5 47 
microfarads 
Capacitor 
C.sub.6 47 
microfarads 
Capacitor 
C.sub.7 0.1 
microfarads 
Capacitor 
C.sub.8 0.047 
microfarads 
Capacitor 
C.sub.9 0.047 
microfarads 
Capacitor 
C.sub.10 0.001 
microfarads 
Capacitor 
C.sub.11 0.1 
microfarads 
Capacitor 
C.sub.12 0.1 
microfarads 
Capacitor 
C.sub.13 120 
picofarads 
Capacitor 
C.sub.14 0.1 
microfarads 
Capacitor 
C.sub.15 0.1 
microfarads 
Capacitor 
C.sub.16 0.1 
microfarads 
Capacitor 
C.sub.17 0.1 
microfarads 
Capacitor 
C.sub.18 0.1 
microfarads 
Capacitor 
C.sub.19 0.1 
microfarads 
Capacitor 
C.sub.20 0.1 
microfarads 
Capacitor 
C.sub.21 180 
picofarads 
Capacitor 
C.sub.22 10 
picofarads 
Capacitor 
C.sub.23 0.1 
microfarads 
Capacitor 
C.sub.24 0.0033 
microfarads 
Capacitor 
C.sub.25 0.1 
microfarads 
Capacitor 
C.sub.26 0.1 
microfarads 
Capacitor 
C.sub.27 10 
microfarads 
Inductor 
L.sub.1 47 
microhenries 
Inductor 
L.sub.2 47 
microhenries 
Inductor 
L.sub.3 680 
microhenries 
Inductor 
L.sub.4 680 
microhenries 
Transistor 
Q.sub.1 Motorola 2N3906 
Transistor 
Q.sub.2 Motorola 2N3906 
Transistor 
Q.sub.3 Siliconix SST309 
Transistor 
Q.sub.4 Siliconix SST309 
Transistor 
Q.sub.5 Motorola 2N3904 
Operational 
OA.sub.1 National LF353 
Amplifier Semiconductor 
Operational 
OA.sub.2 National LF353 
Amplifier Semiconductor 
Operational 
OA.sub.3 Linear LT1013 
Amplifier Technology 
Operational 
OA.sub.4 Linear LT1013 
Amplifier Technology 
______________________________________ 
The remaining circuits for components, including the Doppler audio 
amplifiers and filtering 130, audio power amplifier 132, speakers 134, 
headphone connectors 136, tape recorder output 138, vascular heart rate 
frequency to voltage converter 140, and fetal heart-rate waveform 
generator 142, are all old and well-known in the art and do not form any 
part of this invention. Therefore, the circuits for these components need 
not be described further for purposes of explaining and understanding this 
invention. 
The foregoing description is considered as illustrative only of the 
principles of the invention. Furthermore, since numerous modifications and 
changes will readily occur to those skilled in the art, it is not desired 
to limit the invention to the exact construction and process shown as 
described above. Accordingly, all suitable modifications and equivalents 
may be resorted to falling within the scope of the invention as defined by 
the claims which follow.