AM stereo transmission method and apparatus

This invention discloses an AM stereo transmission method characterized in that when left and right audio signals L and R are transmitted as an AM stereo signal which is represented by S=A cos(.omega.t+.phi.) wherein A denotes an amplitude, .omega. a carrier angular frequency and .phi. a phase angle given by tan.sup.-1 (L-R)/1+L+R, the audio signal of low frequency is transmitted at an amplitude A=(1+L+R), and the audio signal of high frequency is transmitted at an amplitude A=.sqroot.(1+L+R).sup.2 +(L-R).sup.2. The modulator according to the present invention comprises a matrix circuit for providing signals (L+R) and (L-R), an orthogonal modulator to produce an AM stereo signal .sqroot.(1+L+R).sup.2 +(L-R).sup.2 cos(.omega.t+.phi., an amplitude controller, an amplitude control signal generator circuit and a low-pass filter. In the modulator the said AM stereo signal S is changed to the signal (1+L+R)cos(.omega.t+.phi.) for audio signal of low frequency by controlling the amplitude controller. The demodulator according to the present invention comprises an amplitude controller to which the AM signal S=A cos(.omega.t+.phi.) is inputted, an orthogonal demodulator, an matrix circuit for driving signals L and R from signals (L+R) and (L-R) obtained by the orthogonal demodulator, an amplitude control signal generator circuit and a low-pass filter. In the demodulator the said AM stereo signal S=(1+L+R)cos(.omega.t+.phi.) for audio signal of low frequency is changed to the signal .sqroot.(1+L+R).sup.2 +(L-R).sup.2 cos(.omega.t+.phi.) by controlling the amplitude controller.

BACKGROUND OF THE INVENTION 
1. Field of the Invention 
The present invention relates to a method of transmitting AM stereo signals 
and, more particularly, it relates to method and apparatus for AM stereo 
transmission, excellently compatible with the conventional monaural 
receivers and with a narrow side-lobe frequency band. 
2. Prior Art 
Various methods have been proposed and used practically in the United 
States to achieve AM stereo transmission. None of these methods, however, 
are ideal from the viewpoint of their compatibility with conventional 
monaural receivers. One of these methods is to transmit a transmission 
signal S which is represented as follows: 
EQU S=(1+L+R) cos (.omega.t+.phi.) 
wherein L, R and .omega. represent left audio signal, right audio signal an 
carrier angular frequency, respectively, and .phi. is the phase angle 
given by tan.sup.-1 (L-R)/1+L+R. When this signal S is received by the 
conventional monaural receiver, its envelope (1+L+R) is detected. No 
distortion can be found in this detected signal, because only the sum 
signal which represents the monaural signal is present in it. Therefore, 
this transmission method is complete in its compatibility with the 
conventional monaural receivers. However, this signal S has such a 
drawback that a high frequency side-lobe causes the signal to occupy a 
wide band. FIG. 1A shows a frequency spectrum in a case where only the 
left signal of 8 KHz is modulated by 80% according to this method, and it 
is apparent from FIG. 1A that high frequency side-lobe is present. 
Another method is to transmit a signal S which is represented as 
##EQU1## 
Since the difference signal is present in the radical sign of this signal 
envelope, in addition to the sum signal which represents the monaural 
signal, distortion can be found in this envelope-detected signal. 
Therefore, this second method is not entirely compatible with the 
conventional monaural receivers. However, this signal S can be changed as 
follows: 
EQU S=(1+L+R) cos .omega.t-(L-R) sin .omega.t. 
As is apparent from the above equation, it is composed of two AM waves, and 
no side-lobe higher than secondary degree is present accordingly. FIG. 1B 
shows a frequency spectrum in a case where only the left signal of 8 KHz 
is modulated by 80% according to this second method. 
As described above, the first method causes no distortion when the signal 
is envelope-detected, and becomes excellently compatible with the 
conventional monaural receivers, but its side lobe causes it to occupy a 
wide band. On the contrary, the second method maintains the occupied band, 
as narrow as the conventional monaural AM signal, but it causes distortion 
when envelope-detected and it is inferior in its compatibility with the 
conventional monaural receivers. Therefore, neither of these methods is 
ideal for AM stereo transmission. 
SUMMARY OF THE INVENTION 
The object of the present invention is to provide an AM stereo transmission 
method wherein the frequency band occupied by the side-lobe can be kept 
substantially as narrow as in the conventional AM broadcasting and wherein 
excellent audibility can be achieved even when the signal is received by 
the conventional monaural receivers. The object of the present invention 
is also to provide a modulator and a demodulator for achieving the 
transmission method. 
When left and right audio signals L and R are transmitted using an AM 
stereo signal, said AM stereo signal being represented by S=A cos 
(.omega.t+.phi.) wherein A denotes amplitude, .omega. the carrier angular 
frequency, and .phi. the phase angle 
##EQU2## 
it is characterized that transmission is carried out at an amplitude 
A(=1+L+R) in relation to the audio signal of low frequency, and that 
transmission is carried out at an amplitude A=[.sqroot.(1+L+R).sup.2 
+(L-R).sup.2 ] in relation to the audio signal of high frequency. 
FIG. 2A is a fundamental block diagram showing a modulator employed on the 
transmission side of the AM stereo transmission method according to the 
present invention. The modulator comprises a matrix circuit A for 
providing sum and difference signals (L+R) and (L-R) between the left and 
right audio signals L and R, an orthogonal modulator B for orthogonally 
modulating and composing these sum and difference signals to form an AM 
stereo signal S=[.sqroot.(1+L+R).sup.2 +(L-R).sup.2 cos (.omega.t+.phi.)] 
wherein .phi.=tan.sup.-1 (L-R)/1+L+R, an amplitude controller C for 
changing the amplitude of this AM stereo signal S, a circuit (CGAC) E for 
generating an amplitude controlling signal to control this amplitude 
controller C and a low-pass filter D, and it is characterized in that the 
AM stereo signal S is made equal to the output of the orthogonal modulator 
B for high frequency audio signals and to (1+L+R) cos (.omega.t+.phi.) for 
low frequency audio signals by controlling the amplitude controller C with 
the control signal which has passed through the low-pass filter D 
receiving the signal from the CGAC. 
FIG. 2B is a basic block diagram showing a demodulator employed on the 
reception side of the AM stereo transmission method according to the 
present invention. The demodulator comprises an amplitude controller F to 
which the transmitted AM stereo signal S=(1+L+R) cos (.omega.t+.phi.) or 
S=.sqroot.(1+L+R).sup.2 +(L-R).sup.2 cos (.omega.t+.phi.) is inputted, an 
orthogonal demodulator I for orthogonally demodulating the output of this 
amplitude controller F, a matrix circuit J for deriving left and right 
audio signals L and R from sum and difference signals obtained from the 
orthogonal demodulator J, a circuit (CGAC) H for generating a signal to 
control the amplitude controller F, and a low-pass filter G, and it is 
characterized in that the input of the orthogonal demodulator I is the 
same as the input S for high frequency stereo signal and that S=(1+L+R) 
cos (.omega.t+.phi.) of the AM low frequency stereo signals is changed to 
S=.sqroot. (1+L+R).sup.2 +(L-R).sup.2. cos (.omega.t+.phi.) by controlling 
the amplitude controller F with the control signal which has passed 
through the low-pass filter G receiving the signal from the CGAC H.

THE PREFERRED EMBODIMENT OF THE INVENTION 
FIG. 3 shows an example of the modulator used on the transmission side of 
an AM stereo transmission method according to the present invention. 1 and 
1' represent input terminals to which left and right audio signals L and R 
are applied, 2 a matrix circuit for generating sum and difference signals 
(L+R) and (L-R) between the left and right audio signals L and R, 3 an 
amplitude modulator for changing the sum signal (L+R) to a modulated wave, 
4 a balanced modulator for changing the difference signal (L-R) to a 
modulated wave, and 5 an adder for summing outputs of the amplitude and 
balanced modulators 3 and 4. 6 represents a carrier generator for 
generating and unmodulated carrier, and 7 a 90.degree. phase shifter for 
phase-shifting the carrier by 90.degree.. The orthogonal modulator circuit 
shown in FIG. 2A comprises the amplitude modulator 3, balanced modulator 
4, adder 5, carrier generator 6 and 90.degree. phase shifter 7. 8 denotes 
a variable gain amplifier which corresponds to the amplitude controller C 
in FIG. 2A 9 represents an envelope detector for detecting the output 
envelope of the variable gain amplifier 8. 10 represents an adder for 
combining a DC bias voltage (=1) inputted to a terminal 13 and the sum 
signal (L+R) produced through the matrix circuit 2, and 11 a comparator 
for comparing the output of the adder 10 with the output of the envelope 
detector 9. 12 denotes a low-pass filter through which low frequency 
components of the output of the comparator 11 are allowed to pass. The 
amplitude control signal generating circuit (CGAC) shown in FIG. 2A 
comprises the envelope detector 9, adder 10 and comparator 11. 14 
represents an output terminal through which the AM stereo signal is 
outputted. The modulator arranged as described above is operated as 
follows: 
The left and right audio signals L and R provided to the input terminals 1 
and 1' are added and subtracted to form sum and difference signals (L+R) 
and (L-R). The sum signal (L+R) is modulated to a signal (1+L+R) cos 
.omega.t by means of the amplitude modulator 3, while the difference 
signal (L-R) is modulated to a signal-(L-R) sin .omega.t by means of the 
balanced modulator 4. Namely, orthogonal modulation is conducted. In this 
case, the carrier cos .omega.t is generated by the carrier generator 6 and 
inputted to the amplitude modulator 3, while it is phase-shifted by 
90.degree. by the 90.degree. phase shifter 7 and inputted, as-sin 
.omega.t, to the balanced modulator 4. The output of the adder 5 receiving 
the inputs of amplitude modulator 3 and balanced modulator 4 is 
represented by the following AM stereo signal: 
##EQU3## 
wherein 
##EQU4## 
The signal S is introduced to an output terminal 14 through the variable 
gain amplifier 8 and envelope-detected by the envelope detector 9 at the 
same time. Therefore, the output of the envelope detector 9 becomes A in 
the equation (1). The sum signal (L+R) of the matrix circuit 2 is inputted 
into the adder 10 with another input bias "1" through a terminal 13 to 
obtain (1+L+R). 
The output (1+L+R) of the adder 10 and the output A of the envelope 
detector 9 are inputted into the comparator 11 where they are compared. 
The output of the comparator 11 is an error signal between (1+L+R) and A, 
and when the frequency of this error signal, that is, the frequency of 
left and right audio signals is low, the error signal passes through the 
low-pass filter 12 to control the variable gain amplifier 8 until the 
error signal becomes zero by changing the gain. When the error signal is 
zero, the amplitude is given by A=(1+L+R), and an AM stereo signal which 
is represented by 
EQU S=(1+L+R) cos (.omega.t+.phi.). (2) 
can be obtained at the output terminal 14. 
When the frequency of the error signal, or left and right audio signals is 
high, no audio output is passed through the low-pass filter 12 a DC 
component is outputted therefrom, keeping the variable gain amplifier 8 
not controlled, and the output of the adder 5 appears, as it is, at the 
output terminal 14. Namely, 
##EQU5## 
The AM stereo signal of the output terminal 14 is broadcasted as an 
electric wave. 
To summarize the above description, the transmitted AM stereo signal S 
transmitted can be expressed, in the case of audio signals of low 
frequency, by 
EQU S=(1+L+R) cos (.omega.t+.phi.), (4) 
while the transmitted AM stereo signal S can be expressed, in the case of 
audio signals of high frequency, by 
##EQU6## 
Providing that a cut-off frequency of the low-pass filter 12 is 3 KHz, the 
transmitted signal S can be represented by the equation (4) when audio 
signals are less than 3 KHz. This signal has no distortion and is 
compatible with the conventional monaural receivers even when it is 
received by the conventional monaural receivers. 
Although a high-frequency side-lobe is present, the highest frequency of 
the tertiary side-lobe is 9 KHz and most of the signal energy is 
concentrated less than 9 KHz. 
Audio signals higher than 3 KHz are transmitted using the signal S which is 
represented by the equation (5), but since the side-lobe of this signal is 
only primary, no side-lobe higher than 10 KHz is present, providing that 
the highest audio frequency is 10 KHz. When it is assumed that most of the 
energy of the signal represented by the equation (4) can be transmitted 
below the highest frequency 9 KHz of the tertiary side-lobe, therefore, 
the band occupied may be 10 KHz. This is the same as that in the case of 
the conventional monaural AM broadcasting. Although the signal represented 
by the equation (5) causes distortion when is is envelope-detected by the 
conventional monaural receivers, the audio frequency is higher in this 
case than 3 KHz. Therefore, most of higher harmonics are excluded from the 
band, thereby causing no problem in audibility. 
FIG. 4 shows an example of the demodulator used on the reception side of 
the AM stereo transmission method according to the present invention. 
Numeral 15 represents an input terminal to which the transmitted signal S 
is inputted, and 16 a variable gain amplifier which corresponds to the 
amplitude controller in FIG. 2B. Numerals 17 and 18 denote 
synchro-detectors, 19, 20 low-pass filters, 21 a loop filter for detecting 
a DC component from the output of the low-pass filter 20, 22 a voltage 
controlled oscillator for generating carrier for demodulation which is 
used at the time of synchro-detection, and 23 a 90.degree. phase shifter 
for phase-shifting the output of the voltage control oscillator 23. The 
synchro-detector 18, low-pass filter 20, loop filter 21, voltage 
controlled oscillator 22 and 90.degree. phase shifter 23 form a PLL 
circuit as well as a synchro-detector circuit. The PLL circuit, 
synchro-detector 17, and low-pass filter 19 form the orthogonal 
demodulator shown in FIG. 2B. Numeral 24 denotes a matrix circuit for 
separating left and right audio signals L and R from the outputs of the 
low-pass filters 19 and 20. Numerals 28 and 28' denote output terminals 
for the left and right audio signals L and R. 25 represents an envelope 
detector for detecting the envelope of the transmitted signal S. 26 
denotes a comparator for comparing the output of the envelope detector 25 
with the output of the low-pass filter 19, and 27 denotes a low-pass 
filter through which the low frequency component of the output of the 
comparator 26 is allowed to pass to control the variable gain amplifier 
16. 
The envelope detector 25 and comparator 26 form the amplitude control 
signal generator circuit (CGAC) shown in FIG. 2B. 
As will be described later, the difference signal (L-R) is obtained through 
the low-pass filter 20, but when the carrier of the transmitted signal S 
is shifted in phase from the carrier which is the output of the voltage 
controlled oscillator 22 which synchro-detects the signal S, DC a 
component is outputted through the low pass filter 20, in addition to the 
difference signal (L-R). The loop filter 21 allows this DC component to 
pass therethrough, and the voltage controlled oscillator is controlled by 
this DC voltage to generate a carrier B cos .omega.t, having same the 
phase as the carrier of the signal S. 
This carrier B cos .omega.t is inputted to the synchro-detector 17 while it 
is added, as B sin .omega.t, to the synchro-detector 18 through the 
90.degree. phase shifter 23. 
Providing that the AM stereo signal S(=A cos (.omega.t+.phi.)) is added to 
the input terminal 15 and outputted, as it is, through the variable gain 
amplifier 16, this signal S is multiplied by the carrier B cos .omega.t at 
the synchro-detector 17 and by the carrier B sin .omega.t at the 
synchro-detector 18. Let B=2, then respective outputs I and Q are denoted 
by 
##EQU7## 
When these are passed through the low-pass filters 19 and 20, 
respectively, the outputs are given by 
EQU I'=A cos .phi., (8) 
EQU Q'=-A sin .phi.. (9) 
Since 
##EQU8## 
The envelope detector 25 detects the envelope of the signal S and its 
output is denoted by A. This output A and the output I'(=A cos .phi.) of 
the low-pass filter 19 are inputted to the comparator 26 where they are 
compared with each other. The output of the comparator 26 is an error 
signal between A cos .phi. and A, and when the frequency of this error 
signal or of left and right audio signals is low, the error signal passes 
through the low-pass filter 27 to control the variable gain amplifier 16 
and change the amplitude A of the signal S in such a way that the error 
signal becomes zero. Providing that the changed amplitude is A', the 
output of the variable gain amplifier 16 is A' cos (.omega.t+.phi.). 
Therefore, outputs I' and Q' of the low-pass filters 19 and 20 can be 
represented as follows: 
EQU I'=A' cos .phi., (12) 
EQU Q'=-A' sin .phi.. (13) 
When the error signal becomes zero, A=A' cos .phi.. Therefore, 
EQU A'A=A/cos .phi.. (14) 
When the audio signal frequency is low, the AM stereo signal S transmitted 
can be expressed as follows, as already described about the modulator: 
##EQU9## 
and A=(1+L+R). When the equation (14) is replaced by this A and cos .phi. 
of the equation (10), 
##EQU10## 
and the amplitude (1+L+R) of the signal S transmitted is changed to 
##EQU11## 
Therefore, outputs I' and Q' of the low-pass filters 19 and 20 become: 
EQU I'=A' cos .phi.=(1+L+R), (17) 
EQU Q'=-A' cos .phi.=-(L-R). (18) 
and these outputs are added and subtracted through the matrix circuit 24 
after DC the component is removed therefrom, and they are separated to 
left and right audio signals L and R which are supplied to the output 
terminals 28 and 28', respectively. 
In a case where the audio signal frequency is high, the signal S 
transmitted can be expressed as follows: 
##EQU12## 
It is assumed that this signal S is inputted through the input terminal 15 
and outputted, as it is, through the variable gain amplifier 16, the 
outputs I' and Q' of the low pass filters 19 and 20 can be derived from 
the equations (8) and (9) as follows: 
EQU I'=(1+L+R), (20) 
EQU Q'=-(L-R). (21) 
The equation (20) I'=(1+L+R) and the output .sqroot.(1+L+R).sup.2 
+(L-R).sup.2 obtained when the signal S expressed by the equation (19) is 
detected by the envelope detector 25 are compared with each other by the 
comparator 26 to output an error signal. Since the frequency of this error 
signal is high, however, no audio output is generated through the low-pass 
filter 27, and a DC component is outputted therefrom. Therefore, the 
variable gain amplifier 16 is not controlled and the amplitude of the 
signal S is not changed, so that the outputs of the low-pass filters 19 
and 20 are left as expressed as the equations (20) and (21). These outputs 
are added and subtracted through the matrix circuit 24 after the DC 
component is removed therefrom, and they are separated into left and right 
audio signals L and R, which are supplied to the output terminals 28 and 
28'. 
Demodulation of AM stereo signal which is transmitted in the form of 
S=(1+L+R) cos (.omega.t+.phi.) relative to the audio signal of low 
frequency and in the form of S=.sqroot.(1+L+R).sup.2 +(L-R).sup.2 cos 
(.omega.t+.phi.) relative to the audio signal of high frequency can be 
achieved as described above. 
FIG. 5 shows another example of the modulator used on the transmission side 
of the AM stereo transmission method according to the present invention. 
Description will be made leaving the same reference numerals affixed to 
the same parts as those in FIG. 3, and this example in FIG. 5 is different 
from the one shown in FIG. 3 in that the variable gain amplifier 8 is 
replaced by a multiplier 29 and in that the amplitude control signal 
generator circuit (CGAC) comprises the envelope detector 9, adder 10 and 
divider 30. 
Operations of the matrix circuit 2 and orthogonal modulator circuit which 
comprises the amplitude modulator 3, balanced modulator 4, adder 5, 
carrier generator 6 and 90.degree. phase shifter 7 are similar to that in 
FIG. 3. Therefore, the AM stereo signal S outputted from the adder 5 can 
be expressed as follows: 
##EQU13## 
The amplitude .sqroot.(1+L+R).sup.2 +(L-R).sup.2 of this signal S is 
detected by the envelope detector 9 and inputted into the divider 30. 
Applied to another input of the divider 30 is output of the adder 10 which 
is equal to (1+L+R) in FIG. 3. Output C of the divider 30 is as follows: 
##EQU14## 
When the aural signal frequency is low, this output C is passed through 
the low-pass filter 12 and inputted to the multiplier 29 where the 
equations (22) and (23) are multiplied each other, and the following AM 
stereo signal S can be obtained through the output terminal 14: 
EQU S=(1+L+R) cos (.omega.t+.phi.). (24) 
When the audio signal frequency is high, no audio output is generated 
through the low-pass filter 12 and a DC component is outputted therefrom, 
and the output of the adder 5 is supplied, as it is, to the output 
terminal 14. Namely, the AM stereo signal S transmitted is as follows: 
##EQU15## 
The output (1+L+R) of the adder 10 can be obtained by envelope-detecting 
the output of amplitude modulator 3. The output .sqroot.(1+L+R).sup.2 
+(L-R).sup.2 of the envelope detector 9 may be obtained by passing the sum 
of the squares of one output (L-R) of the matrix circuit and the output 
(1+L+R) of the adder 10 through a square root circuit. Or since the output 
of the divider 30 can be expressed by cos .phi. the output of the adder 5 
may be passed through a limiter and multiplied by the output of the 
carrier generator 6, keeping its amplitude certain. 
FIG. 6 shows another example of the demodulator used on the reception side 
of the AM stereo transmission method according to the present invention. 
Description on this example will be made leaving the same reference 
numerals affixed to the same parts as those in FIG. 4. This example is 
different from the one shown in FIG. 4 in that the variable gain amplifier 
14 is replaced by a divider 31 and in that the amplitude control signal 
generator circuit (CGAC) comprises a limiter 32 and a multiplier 33. 
Operations of the matrix circuit 24 and orthogonal demodulator which 
comprises the synchro-detectors 17, 18, low-pass filters 19, 20, loop 
filter 21, voltage controlled oscillator 22 and 90.degree. phase shifter 
23 are similar to those in FIG. 4. The AM stereo signal S transmitted is 
as follows: 
##EQU16## 
This signal S is added to the input terminal 15 and orthogonally 
demodulated through the divider 31 while added to the limiter 32. The 
output of the limiter 32 is inputted into the multiplier 33 as a signal K 
cos (.omega.t+.phi.) where in its amplitude K represents a constant. 
Carrier B cos .omega.t which is the output of the voltage controlled 
oscillator 22 is also inputted to the multiplier 33 and multiplied by the 
output of the limiter 32. As a result, the output D of the multiplier 33 
is as follows: 
EQU D=KB cos (.omega.t+.phi.) cos .omega.t (28) 
If it is assumed that KB=2, equation (28) is given by 
EQU D=cos .phi.+cos .phi. cos 2.omega.t-sin .phi. sin 2.omega.t (29) 
When the audio signal frequency of the AM stereo signal S transmitted is 
low, only cos .phi. is added to the divider 31 by passing this output D 
through the low-pass filter 27 which is same in characteristic as the 
low-pass filter 12 shown in FIGS. 3 or 5. Since the audio signal frequency 
is low this time, the AM stereo signal S transmitted is given by 
EQU S=(1+L+R) cos (.omega.t+.phi.) (30) 
where 
##EQU17## 
Therefore, the output of the divider 31 becomes the following signal S 
which is derived from dividing the equation (30) by the equation (31). 
Namely, 
##EQU18## 
This signal S is demodulated, as expressed by the equations (17) and (18), 
by means of the subsequent orthogonal detector circuit, added and 
subtracted after the DC component is removed therefrom by means of the 
matrix circuit 24, and separated into left and right audio signals L and 
R, which are supplied to the output terminals 28 and 28'. 
As the frequency of a signal becomes higher, its process becomes more 
difficult. Therefore, the following are examples wherein amplitude is 
controlled at low frequency and not controlled at a carrier frequency. 
FIG. 7 shows an example of a modulator which is enabled to control an 
amplitude by changing a high frequency band to a lower band according to 
the present invention. 71 and 72 represent input terminals to which left 
and right audio signals L and R are applied, 73 is the matrix circuit for 
generating sum and difference signals (1+L+R) and (L-R), 74 is the 
amplitude control signal generating circuit (CGAC) which generates an 
amplitude control signal to be multiplied by the sum (1+L+R) and the 
difference (L-R), 77 is the low-pass filter which is enabled to pass the 
low frequency part of the said amplitude control signal, 78 is the first 
multiplier in which the said sum signal (1+L+R) is multiplied by the said 
amplitude control signal, 79 is the second multiplier in which the said 
difference signal (L-R) is multiplied by the said amplitude control 
signal, 80 is the orthogonal modulator which is enabled to modulate the 
outputs of the first and second multipliers orthogonally and to make a 
carrier with modulated and suppressed amplitude, and 81 is the output 
terminal for transmitting the AM stereo signal S. 
The left and right audio signals L and R which are inputted from input 
terminals 71 and 72 are changed to the sum and difference signals (1+L+R) 
and (L-R) in the matrix circuit 73. The inputs 75 and 76 of the CGAC 74 
receive the sum and difference signals (1+L+R) and (L-R) which are 
generated, for examwple, from outputs of the matrix circuit 73, 
respectively. The CGAC 74 which contains a circuit with calculations of 
the square root of a sum of squares and the divider receives the signals 
of the inputs 75 and 76, and outputs the amplitude control signal as 
##EQU19## 
When the frequency of left and right audio signals is low, the said 
amplitude control signal cos .phi., which can pass through the low-pass 
filter 27, is multiplied by the sum signal (1+L+R) inputted into the first 
multiplier and also by the difference signal (L-R) inputted into the 
second multiplier. As a result, the outputs of the first and second 
multipliers become (1+L+R) cos .phi. and (L-R) cos .phi., respectively. In 
the orthogonal modulator 80, the said signals (1+L+R) cos .phi. and (L-R) 
cos .phi. are modulated and summed orthogonally to be a carrier with 
modulated and suppressed amplitude. The AM stereo signal S which is 
outputted from the output terminal 81 is given by 
##EQU20## 
When the frequency of the left and right audio signals L and R is high, the 
amplitude control signal cos .phi. is removed by the low-pass filter 77 
and the filter output is a DC component. Therefore, the outputs of the 
first and second multipliers are the sum and difference signals (1+L+R) 
and (L-R) respectively. In the orthogonal modulator 80, these signals are 
modulated and summed orthogonally to provide a carrier with modulated and 
suppressed amplitude. As a result, the AM stereo signal S which is 
outputted from the output terminal 81 is given by 
##EQU21## 
As described above, since the audio signal S which is given by equation 
(34) for low frequency or by equation (35) for high frequency is 
transmitted as an AM stereo signal, the transmission method is the same as 
the method of the present invention described before. 
FIG. 8 shows an example of demodulator for controlling an amplitude by 
changing a high frequency band to a lower band according to the present 
invention. In the figure, 82 represents the input terminal to receive the 
signal S, 83 is the orthogonal demodulator, 84 and 85 are the first and 
second dividers in which the demodulated signals are divided by the 
amplitude control signal, 86 is the amplitude control signal generating 
circuit (CGAC) which enables generation of an amplitude control signal 
inputting to the first and second dividers 84 and 85, 89 is the low-pass 
filter which passes the lower frequency part of the amplitude control 
signal, 90 is the matrix circuit which adds and subtract the outputs of 
the first and second dividers, respectively, and 91 and 92 are the output 
terminals from which the left and right audio signals L and R are 
outputted. 
When the frequencies of the left and right audio signals L and R are low, 
the signal S represented by equation (34) is inputted to the input 
terminal 82. On the other hand, when the frequencies of the left and right 
audio signals L and R are high, the signal S represented by equation (35) 
is inputted to the terminal 82. To one of the input terminals of the CGAC 
86, the signal given by cos (.omega.t+.phi.) is inputted. This signal is 
obtained, for example, from the output of the limiter receiving the signal 
S=A cos (.omega.t+.phi.) which is inputted from the input terminal 82. The 
another input terminal 88 receives the signal cos .omega.t which is 
generated from, for example, the carrier generator used in demodulation. 
The said two signals which are inputted to the input terminals of the CGAC 
are multiplied by each other and the frequency part being higher than the 
carrier angular frequency .omega. is removed. As the result, the amplitude 
control signal cos .phi. is outputted from the CGAC. 
When the frequency of the left and right audio signals L and R is low, the 
signal S inputted at the input terminal 82 is (1+L+R) cos 
(.omega.t+.phi.). In the orthogonal demodulator 83, the signal S is 
demodulated orthogonally by the carriers cos .omega.t and sin .omega.t, 
and then the demodulated signals represented by (1+L+R) cos .phi. and 
(L-R) cos .phi. are outputted from the orthogonal demodulator 83. The 
demodulated signals are inputted to the first and second dividers, 
respectively. The amplitude control signal cos .phi. which is passed 
through the low-pass filter 89 is inputted to the first and second 
dividers 84 and 85. Therefore, the signals (1+L+R) and (L-R) are outputted 
from the first and second dividers 84 and 85, because the signals (1+L+R) 
cos .phi. and (L-R) cos .phi. are divided by cos .phi., respectively. 
These signals from the dividers are summed and subtracted in the matrix 
circuit 90 and the left and right audio signals L and R are outputted from 
the output terminals 91 and 92. 
When the frequency of left and right audio signals L and R is high, the 
signal S inputted at the input terminal 82 is given by 
.sqroot.(1+L+R).sup.2 +(L-R).sup.2. cos (.omega.t+.phi.) and then the 
signals (1+L+R) and (L-R) are demodulated by means of orthogonal 
demodulation in the orthogonal demodulator 83. The amplitude control 
signal cos .phi. is removed by the low-pass filter 89 and the output 
signal of the filter is a DC component. Therefore, the output signals of 
the orthogonal demodulator are inputted, as is, to the matrix circuit 90 
to be summed and subtracted, and the left and right audio signals L and R 
are outputted from the output terminals 91 and 92, respectively. As 
described above, the AM stereo signal S is demodulated. 
Also, as shown in FIG. 9, the first and second dividers 96 and 97 in the 
demodulator can be placed on the side of the outputs of the matrix circuit 
95. In that case, when the outputs of the orthogonal demodulator 94 are 
(1+L+R) cos .phi. and (L-R) cos .phi. the inputs of the first and second 
dividers 96 and 97 are L cos .phi. and R cos .phi., respectively, and then 
each cos .phi. is removed by the dividers, and the left and right audio 
signals L and R are outputted from the output terminals 100 and 104. 
Consequently, according to the present invention methods shown in the 
examples of FIGS. 7, 8 and 9, because the first and second multipliers in 
the modulator and the first and second dividers in the demodulator are 
placed on the input side of the orthogonal modulator and on the output 
side of the orthogonal demodulator, respectively, low frequency signals 
can be used. 
As described before, FIG. 7 shows a modulator wherein first and second 
multipliers which are amplitude controllers are arranged between the 
matrix circuit and the orthogonal modulator to carry out multiplication 
with audio signals. FIG. 8 shows a demodulator wherein first and second 
dividers are arranged between the orthogonal demodulator and the matrix 
circuit to carry out division with audio signal. FIG. 9 shows a 
demodulator which carries out division after the matrix circuit. 
The following is also an example wherein amplitude is controlled at low 
frequency. FIG. 10 shows a modulator wherein envelope detection is carried 
out after orthogonal modulation to remove the carrier from the audio 
signal, thereby enabling multiplication to be conducted with the audio 
signal, and amplitude modulation is then carried out again. 
The operations of the matrix circuit 107, the orthogonal modulator 108 and 
the amplitude control signal generator circuit (CGAC) 116 are the same as 
those of the modulator shown in FIG. 2A. Therefore, the output of the 
orthogonal modulator 108 is given by 
##EQU22## 
and the outputs of the low-pass filter 114 are cos .phi. and the DC 
component for the audio signals with low and high frequencies, 
respectively. The output of the orthogonal modulator 108 is inputted to 
the envelope detector 110 and the limiter 109. The envelope signal 
.sqroot.(1+L+R).sup.2 +(L-R).sup.2 is outputted from the envelope detector 
110 and cos (.omega.t+.phi.) which has a constant amplitude is outputted 
from the limiter 109. This cos (.omega.t+.phi.) is a phase-modulated 
signal by .phi.=tan (L-R)/1+L+R and is used as a carrier by inputting to 
the amplitude modulator 111. 
When the frequency of left and right audio signals is low, the output 
signal .sqroot.(1+L+R).sup.2 +(L-R).sup.2 of the envelope detector 110 and 
the output cos .phi. of the low-pass filter 114 are multiplied in the 
multiplier 113, and the output of the multiplier 113 becomes a signal 
(1+L+R). This signal (1+L+R) is inputted to the amplitude modulator 111 
which performs the amplitude modulation for the carrier cos 
(.omega.t+.phi.). Therefore, the AM stereo signal outputted from the 
output terminal 112 is given by 
EQU S=(1+L+R) cos (.omega.t+.phi.). (37) 
On the other hand, when the frequency of the left and right audio signals L 
and R is high, the output signal of the low-pass filter 114 is a DC 
component and then the modulating signal for the carrier cos 
(.omega.t+.phi.) is the same as the output signal of the envelope detector 
110. Therefore, the AM stereo signal outputted from the output terminal 
112 is given by 
##EQU23## 
Consequently, by means of this example according to the present invention, 
a low frequency band can be used because the amplitude control is 
performed after the envelope detection. 
FIG. 11 shows a demodulator simplified in construction without using the 
amplitude controlleer. In the figure, 128 is the input terminal inputting 
the AM stereo signal S, 129 is the envelope detector, 130 is the 
synchro-detector, 135 is the carrier generator to generate a carrier for 
demodulation, 131 is the 90.degree. phase shifter to shift the said 
carrier by 90.degree., 132 is the matrix circuit, and 133 and 134 are 
output coterminals for left and right audio signals L.sub.0 and R.sub.0. 
When the frequency of the left and right audio signals L and R is low, the 
AM stereo signal inputted at the input terminal 128 is given by 
EQU S=(1+L+R) cos (.omega.t+.phi.), (39) 
and then the signal (1+L+R) is outputted from the envelope detector 129. 
The carrier generator 135 outputs a carrier cos .omega.t with the same 
phase as that of the input signal S, and the phase shifter 131 outputs a 
signal sin .omega.t with the phase shifted by 90.degree. for the input 
signal S. The synchro-detector 130 which inputs the carrier sin .omega.t 
multiplies the said carrier sin .omega.t and the input signal S, and the 
high frequency part of the product signal is removed by a low-pass filter 
in the synchro-detector 130. Therefore, the output signal of the 
synchro-detector 130 is given by 
EQU (1+L+R) sin .phi.=(L-R) cos .phi.. (40) 
In the matrix circuit 132, the output signal (1+L+R) of the envelope 
detector 129 and the said signal (L-R) cos .phi. are summed and 
subtracted, and then signals without DC part "1" are outputted. Therefore, 
the left and right signal L.sub.0 and R.sub.0 outputted from the output 
terminals 133 and 134 are given respectively by 
EQU L.sub.0 =L(1+cos .phi.)+R(1-cos .phi.), (41) 
EQU R.sub.0 =R(1+cos .phi.)+L(1-cos .phi.). (42) 
If cos .phi. is expanded in series until the second term approximately, 
equation (41) is written by 
##EQU24## 
Note that L.sub.0 is not equal to L and that there exists the second term 
in (43). However, the level of the second term of equation (43) is lower 
than that of the left audio signal L by -12 dB, and also, the average 
level of the difference signal (L-R) is very low generally. As a result, 
the second term of equation (43) has no effect practically. On the other 
hand, the output signal R.sub.0 at the output terminal 134 is given 
similarly by 
##EQU25## 
When the frequency of the left and right signals L and R is high, the AM 
stereo signal S inputted at the terminal 128 is given by 
##EQU26## 
Therefore, the output signals of the envelope detector 129 and the 
synchro-detector 130 are given, respectively, by 
##EQU27## 
Since the equation (46) is rewritten to (1+L+R) sec .phi. in which sec 
.phi. can be expanded in series until the second term, the equation (46) 
is approximately given by 
EQU (1+L+R)(1+.phi..sup.2 /2). (48) 
In the matrix circuit 132, the signals represented by (47) and (48) are 
summed and subtracted with removal of DC part "1". Therefore, the left and 
right demodulated signals L.sub.0 and R.sub.0 which are outputted from 
output terminals 133 and 134 are given respectively by 
##EQU28## 
FIG. 12 shows another demodulator simplified in construction without using 
the amplitude controller. 
When the frequency of the left and right audio signal L and R is low the 
signal S inputting at the input terminal 115 is expressed by equation 
(39). Therefore, the signal (1+L+R) is outputted from the envelope 
detector 116, the signal S is multiplied by the carrier cos .omega.t in 
the synchro-detector 117 to output (1+L+R). cos .phi. and also by the 
carrier sin .omega.t in the synchro-detector 118 to output (L-R) cos 
.phi.. However, since the output signal of the synchro-detector 117 is 
removed by the high-pass filter 121, the matrix circuit performs sum and 
subtract operations between the output signal (1+L+R) passed through the 
low-pass filter 120 from the envelope detector 116 and the output signal 
(L-R) cos .phi. of the synchro-detector 118, and it outputs the signal 
without the DC part "1". Therefore, the left and right signals L.sub.0 and 
R.sub.0 outputted at the output terminals 126 and 127 are the same as 
those given by equations (43) and (44), respectively. 
When the frequency of the left and right audio signals L and R is high, the 
signal S appearing at the input terminal 115 is expressed by (45). 
Therefore, the signals .sqroot.(1+L+R).sup.2 +(L-R).sup.2, (1+L+R) and 
(L-R) are outputted from the envelope detector 129, the synchro-detectors 
117 and 118, respectively. Since the output signal of the envelope 
detector 116 is removed by the low-pass filter 120, the output signal 
(1+L+R) of the synchro-detector 117 passed through the high-pass filter 
and the output signal (L-R) of the synchro-detector 118 are summed and 
subtracted in the matrix circuit 125, and the signals in which the DC part 
"1" is removed are outputted. The left and right signals L.sub.0 and 
R.sub.0 outputted from the output terminals 126 and 127 are given 
respectively by 
EQU L.sub.0 =2L, (51) 
EQU R.sub.0 =2R. (50) 
and then the complete demodulation can be performed. 
As described above, the AM stereo signals are demodulated with practically 
no problem practically though there exists some cross-talk and distortion. 
Consequently, according to examples shown in FIGS. 11 and 12, the 
demodulator can be constructed with a simple structure practically and the 
cost of materials and the production time are reduced. 
According to the present invention, an audio signal of low frequency is 
transmitted using the following AM stereo signal: 
EQU S=(1+L+R) cos (.omega.t+.phi.) 
while an audio signal of high frequency is transmitted using the following 
AM stereo signal: 
##EQU29## 
Therefore, the band occupied by side-lobe can be kept as narrow as that in 
the conventional AM broadcasting. In addition, broad-band broadcasting can 
be practiced. Further, no distortion is caused in the case of an audio 
signal of low frequency even when it is received by the conventional 
monaural receivers. Furthermore, distortion is caused in the case of an 
audio signal of high frequency, but most of its higher harmonics are 
present outside the band, thereby causing no problem in audibility. Still 
further, the band is narrow. Therefore, even when preemphasis is effected 
on the transmission side, little influence is exerted to its adjacent 
channels.