Transmitter and mobile communication terminal using the same

A transmitter employing variable gain amplifiers and operating with both constant and nonconstant envelope modulation systems is contrived to suppress variation in the transmitting power when constant envelope modulation is performed. The transmitter comprises a PM loop, an AM loop, and a variable gain amplifier which is shared by the PM loop and the AM loop and combines phase information that the PM loop outputs and envelope information that the AM loop outputs by gain control. The variable gain amplifier comprises a variable gain amplifier body having a supply voltage terminal and a bias current detection terminal for extracting a bias current corresponding to a gain, wherein the gain changes with a change in the potential of the supply voltage terminal, and a bias control block connected to the supply voltage terminal and the bias current detection terminal. Thereby, a bias control loop is formed to control the bias current so that the gain in the case of constant envelope modulation becomes a predetermined value.

CLAIM OF PRIORITY

The present patent application claims priority from Japanese application JP 2004-368962 filed on Dec. 21, 2004, the content of which is JP 2005-254344 filed on Sep. 2, 2005, the content of which is hereby incorporated by reference into this application.

FIELD OF THE INVENTION

The present invention relates to a transmitter for use in a mobile communication terminal and, in particular, to a dual-mode compatible transmitter that operates with both constant and nonconstant envelope modulation systems.

BACKGROUND OF THE INVENTION

An example of a prior-art transmitter of a dual-mode terminal is configured, including a phase control loop which feeds back and controls the phase of an output signal from a voltage-controlled oscillator and an amplitude control loop which feeds back and controls the gain of a power amplifier which amplifies the above output signal (e.g., see patent document 1: Japanese Patent Laid-open No. 2004-7443).

SUMMARY OF THE INVENTION

In late years, there has been an explosive increase in the number of subscribers of mobile communication, mainly voice services, typified by cellular mobile phones. An example of such a communication system is a Global System for Mobile Communications (GSM) which has been developed mainly in Europe. Meanwhile, lately, there has been an increasing need for high-speed data communication in addition to voice services. In the GSM, the conventional system using a Gaussian Minimum Shift Keying (GMSK) modulation which is a constant envelope modulation is planned to change to an Enhanced Data for Global Evolution (EDGE) system using 8 Phase Shift Keying (8PSK) which is a multilevel, nonconstant envelope modulation. A mobile terminal which will operate with this EDGE system (nonconstant envelope 8PSK modulation) is required to be compatible with the conventional GSM system (constant envelope GMSK modulation). It is therefore essential to be a GSM/EGDE compatible dual mode terminal for future mobile communication.

An example of a prior art transmitter configuration for use in such a dual mode terminal has been disclosed in the above-mentioned patent document 1. In this configuration, the phase control loop is used (the power amplifier gain is fixed) during GSM operation and both the phase control loop and the amplitude control loop are used during EDGE operation. By the amplitude control loop operation, the amplitude of an output signal from the power amplifier is controlled to change with the envelope of the nonconstant envelope 8PSK modulation.

As described above, in the prior art transmitter configuration, the gain of the power amplifier is controlled by the amplitude control loop. As an arrangement to supersede this, prior to the present invention, one of the inventors of the present invention considered a configuration in which a variable gain amplifier for small power is connected in front of the power amplifier with fixed gain.

Circuitry of this variable gain amplifier (hereinafter denoted as “VGA”) is shown inFIG. 22. The VGA is comprised of a VGA body (hereinafter denoted as “VGAB”)200, a switch with two inputs and one output (hereinafter denoted as “DSW”)202a, and an operational amplifier (hereinafter denoted as “OPAMP”)205. The VGAB200is composed of a P-type MOS transistor MP3a, an N-type MOS transistor MN3a, a resistor R3a, and capacitors C3aand C3b.

The transistors MP3aand MN3aconstitute an inverter and are self-biased by the resistor R3a. A signal that is input at an input terminal RFIN is amplified and output from an output terminal RFOUT. A reference voltage DC with a fixed potential is input to one input of the DSW202aand a signal AMSIG whose amplitude changes with the envelope of nonconstant envelope modulation is input to the other input. Output of the DSW202ais connected to a noninverting input terminal of the OPAMP205. Output of the OPAMP205is connected to its inverting input terminal and the OPAMP205constitutes a noninverting amplifier. The output of the OPAMP205is connected to a source terminal of the transistor MP3a, that is, a supply voltage terminal Pt of the VGAB200.

In the case of nonconstant envelope modulation, the DSW202aoutputs the signal AMSIG which changes the potential of the supply voltage terminal Pt, thereby the VGA gain is controlled. As a result, good linearity is maintained over a gain variable range. Besides, the VGA can be realized with a simple structure. On the other hand, in the case of constant envelope modulation, the DSW202aoutputs the reference voltage DC which fixes the potential of the supply voltage terminal Pt. As a result, the VGA operates as a fixed gain amplifier.

The characteristics, e.g., threshold voltages of the above transistors MP3aand/or MN3amay vary due to variance in the manufacturing process or for other reason. This variation has an effect on the gain that should remain constant during constant envelope modulation, when the potential of the supply voltage terminal Pt is kept constant. Such effect on the gain takes place due to a variation in the rate of converting a change in the potential difference between the gate and source of the transistors MP3aand MN3ainvolved during the transmission of an input signal from the input terminal RFIN into a change in the drain current which is output from the output terminal, in short, a variation in transconductance of the transistors MP3aand MN3a. Gain variation is reduced by increasing consumption current and increasing the level of input signals to the VGA to a sufficiently high level. However, the consumption current must be increased more, as the characteristic variance becomes more significant. Needless to explain, the VGA gain variation varies the output power of the transmitter.

An object of the present invention is to provide a transmitter employing variable gain amplifiers, operating with both constant and nonconstant envelope modulation systems, and with little variation in the transmitting power when constant envelope modulation is performed, or a mobile communication terminal using such transmitter.

The above object of the present invention can be achieved effectively by providing a bias control loop in a variable gain amplifier. The variable gain amplifier comprises a variable gain amplifier body having a supply voltage terminal and a bias current detection terminal for extracting a bias current corresponding to a gain, wherein the gain changes with a change in the potential of the supply voltage terminal, and a bias control unit connected to the supply voltage terminal and the bias current detection terminal. Thereby, the bias control loop is provided comprising the variable gain amplifier body and the bias control unit to control the bias current so that the gain in the case of constant envelope modulation becomes a predetermined value. By adopting this means, variation in the gain of the variable gain amplifier is suppressed by the control loop when a transmit signal is modulated by constant envelope modulation. Therefore, a transmitter capable of operating with constant and nonconstant envelope modulation systems and with little variation in the transmitting power can be realized.

These and other objects and many of the attendant advantages of the invention will be readily appreciated as the same becomes better understood by reference to the following detailed description when considered in connection with the accompanying drawings.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

A transmitter of the present invention and a mobile communication terminal using it will be described more fully hereinafter with reference to a number of embodiments shown in the drawings. In all drawings which are used to explain the embodiments, the same reference numerals are used to identify the same or similar elements.

FIG. 1andFIG. 2show a first embodiment of a transmitter of the present invention. The transmitter of the first embodiment shown inFIG. 1includes a VGA (variable gain amplifier) involved in the present invention, shown inFIG. 2.

InFIG. 2, the VGA is comprised of a VGAB (variable gain amplifier body)200, an OPAMP (operational amplifier)205which supplies a gain control signal, which is generated, based on a signal AMSIG whose amplitude changes with the envelope of nonconstant envelope modulation, to the VGAB200, a DSW (a switch with two inputs and one output)202awhich switches the OPAMP205operation between GSM and EDGE, and four circuits which will be mentioned below. Here, the VGAB200has an input terminal RFIN, an output terminal RFOUT, a supply voltage terminal Pt at which the gain control signal is input, and a bias current detection terminal Bt. The four circuits are a current detector (hereinafter denoted as “IDET”)203which detects a bias current flowing in the VGAB200when being connected to the bias current detection terminal Bt of the VGAB200, a switch with one input and one output (hereinafter denoted as “SSW”)201which makes a connection between the bias current detection terminal Bt and the current detector203when in GSM mode and disconnects the connection when in EDGE mode, a current-voltage converter (hereinafter denoted as “IVC”)204which converts the bias current detected by the IDET203into a voltage, and a DSW202bwhich switches connection based on modulation. A bias control block (bias control unit)301is composed of the OPAMP205, the DSW202a, and the above four circuits. A bias control loop300is composed of the VGAB200and the bias control block301.

The DSW202asupplies fixed potential DC to a noninverting input terminal of the OPAMP205when in GSM mode and supplies the signal AMSIG to the same terminal when in EDGE mode. The DSW202bsupplies the output voltage of the IVC204to an inverting input terminal of the OPAMP205when in GSM mode and supplies a gain control signal which is the output signal of the OPAMP205to the same terminal when in EDGE mode.

First, in the case of nonconstant envelope modulation in EDGE mode, the DSW202aoutputs the signal AMSIG, the DSW202bmakes a connection between the output of the OPAMP205and its inverting input terminal, and the SSW201is open. As a result, the gain of the VGAB200is controlled so that the amplitude of the output of the VGA changes with the envelope.

On the other hand, in the case of constant envelope modulation in GSM mode, the DSW202aoutputs the fixed potential DC, the DSW202bmakes a connection between the output of the IVC204and the inverting input terminal of the OPAMP205, the SSW201ais closed, and a feedback loop for the bias current flowing in the VGAB200(bias control loop) is formed. By the feedback loop, the output potential of the IVC204is controlled to be equal to the fixed potential DC. As a result, a constant bias current flows in the VGAB200with a constant transconductance, which reduces variation in the output level of the VGA.

The transmitter of the first embodiment shown inFIG. 1is configured, using the VGA configured as above, involved in the present invention. The transmitter of the first embodiment is comprised of a first carrier frequency generating circuit, a second carrier frequency generating circuit, a phase control loop (hereinafter denoted as “PM loop”), an amplitude control loop for changing amplitude with the envelope (hereinafter denoted as “AM loop”), a quadrature modulator (hereinafter denoted as “MOD”)100, a low-pass filter (hereinafter denoted as “LPF”)110a, a VGA109b, a power amplifier (hereinafter denoted as “PA”)112, an output signal detector113, and a control circuit114.

The first carrier frequency generating circuit is composed of an intermediate-frequency voltage-controlled oscillator (hereinafter denoted as “IFVCO”)105and a divide-by-8 divider (⅛)104d.

The second carrier frequency generating circuit is composed of a radio-frequency (RF) voltage-controlled oscillator (hereinafter denoted as “RFVCO”)106and a divide-by-4 divider (¼)104c.

The PM loop120is composed of a phase comparator (hereinafter denoted as “PD”)107, an LPF101d, a transmitting-frequency voltage-controlled oscillator (hereinafter denoted as “TXVCO”)108, a VGA109a, a mixer102, and an LPF101c.

The AM loop130is composed of an envelope comparator (hereinafter denoted as “AMD”)110, an LPF101e, a voltage-current converter (hereinafter denoted as “VIC”)111, an LPF101f, the VGA109a, the mixer102, and the LPF101c.

A transmitting circuit115comprising the first carrier frequency generating circuit, second carrier frequency generating circuit, PM loop120, AM loop130, LPF101a, MOD100, VGA109aand control circuit114is integrated as a semiconductor integrated circuit (RF-IC: RF-Integrated Circuit). A transmit output circuit116comprising the PA112and output signal detector113is integrated into a module. As above, the transmitter1of the first embodiment consists of the transmitting circuit115and the transmit output circuit116.

In the first embodiment, the VGA shown inFIG. 2is employed as the VGA109a.

Here, the LPF101dconsists of resistors R1a, R1b, capacitors C1a, C1b, and a switch SW1. The switch SW1is used to change the filter's cutoff frequency between GSM and EDGE. The LPF101econsists of a resistor R1cand capacitors C1c, C1d. The LPF101fconsists of a resistor R1dand capacitors C1e, C1f.

Then, how the transmitter configured as above operates will be described. First, in EDGE mode when an in-phase component (hereinafter denoted as “I”) and a quadrature component (hereinafter denoted as “Q”) of baseband signal which are output by a baseband circuit117undergo nonconstant envelope modulation, the I and Q baseband signals (I/Q) are converted into a signal with a first carrier frequency as the center frequency by the quadrature converter100and an unwanted signal is suppressed by the LPF110a. An output signal of the LPF110abecomes a reference signal that is input to the PD107and AMD110.

By the PM loop120, the reference signal and a feedback signal passing via the LPF101cto the PD107are controlled to have the same phase and frequency. That is, the phase control loop synchronizes with the phase of the reference signal generated from the input signal. As a result, at the output of the VGA109a, the phase (or a frequency-modulated component) included in the reference signal is reproduced as phase information and the center frequency is converted into a desired frequency that is determined by the first and second carrier frequencies. When the output frequency of the IFVCO105is, for example, 640 MHz, the first carrier frequency will be 80 MHz and the center frequency of the reference signal also will be 80 MHz. Thus, the center frequency of the feedback signal to the PD107will be 80 MHz. On the other hand, if the frequency of the RFVCO106is 3920 MHz, the second carrier frequency will be 980 MHz. Because the center frequency of the output signal of the VGA109ais converted into 80 MHz by the mixer102, eventually, the center frequency of the output signal of the VGA109awill be 900 MHz.

Similarly, by the AM loop130, the reference signal and the feedback signal passing via the LPF101cto the AMD110are controlled to have the same envelope; that is, the AM loop follows the amplitude of the envelope of the reference signal. As a result, at the output of the VGA109a, the envelope included in the reference signal is reproduced as envelope information. The amplitude of the output signal of the TXVCO108is constant and reproducing the envelope is realized by controlling the gain of the VGA109a. The variable gain range of the VGA109asupports the envelope variation of the modulated signal and this range is about 18 dB in the case of EDGE. The VGA109ais realized with simple circuitry as shown inFIG. 2.

Then, in GSM mode when the I and Q baseband signals which are output by the baseband circuit117undergo constant envelope modulation, circuits in the AM loop130which are necessary for the above AM loop operation and which are not in common to the PM loop120, namely, the AMD110and VIC111are set inoperative. Moreover, the VGA109aoperates as a fixed gain amplifier. The PM loop120operates as described above and the reference signal and the feedback signal passing via the LPF101cto the PD107are controlled to have the same phase and frequency.

As described above, the VGA109ais shared by the PM loop120and AM loop130and combines phase information that the PM loop120outputs and envelope information that the AM loop outputs by gain control.

In GSM and EDGE modes, the antenna output power is required to be variable in a range of at least 40 dB. The output power can be varied in the range of 40 dB by using the VGA109b. The VGA109bis a linear amplifier with a variable gain range of 40 dB or more. The antenna output power is controlled by using the detector113and control circuit114. The output power of the PA112is detected by the detector113and the detector113outputs a detected signal. The detected signal is compared to a reference signal RAMP that is output by the baseband circuit117and the control circuit114generates a gain control signal to the VGA109bso that the detected signal and the reference signal RAMP will match. In this way, the VGA109b, PA112, detector113, and control circuit114form an antenna output power control loop. The reference signal RAMP is not always to be an analog signal and may be a digital signal. In that case, a digital/analog converter is prepared.

In this way, because variable gain range that is required to control the antenna output power is provided by the VGA109b, the PA112is only required to have a sufficient linear characteristic with fixed gain. Therefore, there is no need to add a special function to the PA112for transmitter realization and a general-purpose power amplifier can be employed as the PA112. However, some characteristic of the PA112, for example, gain may change between when the transmitter operates in GSM mode and when it operates in EDGE mode.

The baseband circuit117performs intended control by supplying a control signal (CTRL DATA) to the transmitting circuit115.

Details of the components of the VGA109ashown inFIG. 2will be described below.FIG. 3shows the VGA109aof the first embodiment including examples of the VGAB200, IDET203, and IVC204circuits.

The VGAB200is comprised of a P-type MOS transistor MP3a, an N-type MOS transistor MN3a, capacitors C3a, C3b, and a resistor R3a. The transistors MP3aand MN3aconstitute an inverter circuit and are self-biased by the resistor R3a. A signal that is input at the input terminal RFIN is amplified and output from the output terminal RFOUT. Gain control is performed by changing the source potential of the transistor MP3a(potential of the supply voltage terminal Pt), namely, the supply potential.

The IDET203consists of a P-type MOS transistor MP3b. The transistor MP3band the transistor MP3aconstitute a current mirror via the SSW201and the transistor MP3boutputs a current that is proportional to the current flowing in the VGAB200.

The IVC204consists of a resistor R3band a capacitor C3cconnected in parallel. One terminals of the resistor R3band capacitor C3care grounded and the other terminals are connected to the output of the IDET203and further connected to the DSW202b. The AC component of the output current from the IDET203is eliminated by the capacitor C3cand a voltage that is proportional to the bias current flowing in the VGAB200is generated across the resistor R3b.

Next, a circuit example of the SSW201is shown inFIG. 4. The SSW201is comprised of a P-type MOS transistor MP4, an N-type MOS transistor MN4, and an inverter INV. An Input terminal SWIN is connected to a source terminal of the transistor MP4and a drain terminal of the transistor MN4. An output terminal SWOUT is connected to a drain terminal of the transistor MP4and a source terminal of the transistor MN4. A control terminal SWON is connected to a gate terminal of the transistor MN4and an input terminal of the inverter INV and the output of the inverter INV is connected to a gate terminal of the transistor MP4.

First, when the potential of the control terminal SWON is the supply voltage, the transistor MN4operates in the triode region, whereas the gate of the transistor MP4is brought to the ground potential GND by the inverter INV, and the transistor MP4also operates in the triode region, providing a short circuit between the terminal SWIN and the terminal SWOUT. Conversely, when the potential of the control terminal SWON is GND (ground), both the transistors NN4and MP4do not operate, providing an open condition between the terminal SWIN and the terminal SWOUT. The above-described operation of the SSW201is summarized inFIG. 5.

Then, a circuit example of the DSW202aand DSW202bis shown inFIG. 6. The DSW202aand DSW202bare, respectively, comprised of P-type MOS transistors MP6a, MP6b, N-type MOS transistors MN6a, MN6b, and an inverter INV. One input terminal SWIN1is connected to a source terminal of the transistor MP6aand a drain terminal of the transistor MN6a. The other input terminal SWIN2is connected to a source terminal of the transistor MP6band a drain terminal of the transistor MN6b. An output terminal SWOUT is connected to the drain terminals of the transistors MP6aand MP6band the source terminals of the source terminals of the transistors MN6aand MN6b. A control terminal SWON is connected to the gate terminals of the transistors MN6aand MP6band an input terminal of the inverter INV and the output of the inverter INV is connected to the gate terminals of the transistors MP6aand MN6b.

First, when the potential of the control terminal SWON is the supply voltage, the transistors MN6aand MP6a operate in the triode region, whereas the transistors MN6band MP6bdo not operate, which provides a short circuit between the input terminal SWIN1and the output terminal SWOUT and an open condition between the input terminal SWIN2and the output terminal SWOUT. Conversely, when the potential of the control terminal SWON is GND, the transistors MN6band MP6boperates in the triode region, whereas the transistors MN6aand MP6ado not operate, which provides a short circuit between the input terminal SWIN2and the output terminal SWOUT and a open condition between the input terminal SWIN1and the output terminal SWOUT. The above-described operation of the DSW202aand DSW202bis summarized inFIG. 7.

By the operations of the components described above, in GSM mode when constant envelope modulation is performed, the DSW202aoutputs fixed potential DC, the DSW202bmakes a connection between the output of the IVC203and the inverting input terminal of the OPAMP205, the SSW201ais closed, and a feedback loop for the bias current flowing in the VGAB200is formed. By the feedback loop, the output potential of the IVC204is controlled to be equal to the fixed potential DC. As a result, in the VGAB200, a constant bias current flows through the transistor MP3awhich forms a current mirror with the IDET203and a constant transconductance is obtained, and thereby variation in the output level of the VGA is reduced.

The above switches (SSW201, DSW202a, and DSW202b) are not necessarily embodied in switches and it is needless to say that they are only required to be able to make or break connections between the OPAMP205, VGA200, IDET203, and IVC204, when appropriate. Even if they are embodied in switches, they are not limited to the examples shown inFIGS. 4 and 6.

FIG. 8shows an example of timing when the switches (SSW201, DSW202a, DSW202b) change over. Control signals SWON201, SWON202a, and SWON202bare applied to the SSW201, DSW202a, and DSW202b, respectively.FIG. 8is provided, assuming that: DC and signal AMSIG are applied, respectively, to the input terminals SWIN1and SWIN2of the DSW202a; and the output of the IVC203and the output of the OPAMP205are input, respectively, to the input terminals SWIN1and SWIN2of the DSW202b. The control signals SWON201, SWON202a, and SWON202bsimultaneously change from ground (GND) to supply potential in the middle of changeover from EDGE to GSM.

The above changeover timing is one example. The above switches may be at least required to change over before the transmission state when one modulation system changes to another and do not necessarily change over simultaneously. If one modulation system changes to another during transmission, the switches may be at least required to change over within the transition time.

It is possible to move the circuit element for eliminating the AC component in the IVC204shown inFIG. 3to the side of the gate of the IDET203. The so-modified VGA109ais shown inFIG. 9. InFIG. 9, an LPF consisting of a resistor R10and a capacitor C10is connected to the gate of the transistor MP3bof the IDET203. The IVC204consists entirely of the resistor R3b. For the IVC204shown inFIG. 3, current-voltage conversion gain is determined by the resistance value of the resistor R3band the capacitance value of the capacitor C3cis determined from this resistance value and the cutoff frequency required to eliminate the AC component. InFIG. 9, because the cutoff frequency is determined by the resistance value of the resistor R10and the capacitance value of the capacitor C10, it becomes possible to decrease the capacitance value of the capacitor C10by increasing the resistance value of the resistor R10.

Moreover, it is possible to connect a resistor R13in series to the capacitor C3cin the IVC204shown inFIG. 3. The VGA109ain which the IVC204is so modified is show inFIG. 10. In comparison with the configuration ofFIG. 3, as the result of the provision of the resistor R13, the effect of eliminating the high-frequency component somewhat decreases, but the effect of providing a larger phase margin in the feedback loop for keeping the current flowing in the VGA200constant is obtained.

A second embodiment of a transmitter of the present invention will be described withFIG. 11. In the second embodiment, the VGA shown inFIG. 2, which is employed as the VGA109a, is modified such that an OPAMP205ais configured so that a gain of one or more times can be set. The so-modified VGA109aemployed in the second embodiment is shown inFIG. 11. InFIG. 11, the output of the OPAMP205is connected via a resistor R9ato the input of the DSW202band a terminal on the DSW202bside of the resistor R9ais grounded via a resistor R9b.

In the case of nonconstant envelope modulation in the first embodiment, the gain of the OPAMP205which is a noninverting amplifier is1, whereas, in the second embodiment, the gain becomes 1+RA/RB, where RAand RBdenote the resistance values of the resistors R9aand R9b, respectively. In consequence, the AM loop characteristic that is determined by the product of the gains of the circuits constituting the AM loop can be varied as required.

A third embodiment of a transmitter of the present invention will be described withFIG. 12. In the third embodiment, a VGA in which an N-type MOS transistor is employed in the current detector, which is shown inFIG. 12, is employed as the VGA109a. InFIG. 12, although the VGAB200is the same as that in the first embodiment, the IDET203consists of the N-type MOS transistor MN11. The transistor MN11and the transistor MN3aconstitute a current mirror via the SSW201and the transistor MN11outputs a current that is proportional to the current flowing in the VGAB200. A resistor R11and a capacitor C11constituting the IVC204are connected to the source of the transistor MP3a. The fixed potential DC and the output of the OPAMP205are input to a DSW1100aand the output of the DSW1100ais connected to an inverting input terminal of the OPAMP205. The signal AMSIG and the output of the IVC204are input to a DSW1100band the output of the DSW1100bis connected to noninverting input terminal of the OPAMP205.

In the case of nonconstant envelope modulation, the VGA operates the same as in the first embodiment. On the other hand, in the case of constant envelope modulation, the fixed potential DC is input to the inverting input terminal of the OPAMP205and the output signal of the IVC204is input to the noninverting input terminal of the OPAMP205. Since the output potential of the IVC204decreases as the bias current flowing in the VGA200increases, a feedback loop is formed by inverting the polarity of the OPAMP205, unlike the case of the first embodiment. As a result, in the VGAB200, a constant bias current flows through the transistor MN3awhich forms a current mirror with the IDET203and a constant transconductance is obtained, and thereby variation in the output level of the VGA is reduced.

Although not shown, it is possible to connect a resistor between the output of the OPAMP205and the input of the DSW1100aand connect another resistor between the input of the DSW1100aand the ground, as is the case for the second embodiment. In consequence of this modification, when applied, the AM loop characteristic that is determined by the product of the gains of the circuits constituting the AM loop can be varied as required.

Although not shown, as is the case forFIG. 9for the first embodiment, it is possible to connect a resistor between the SSW201and the gate of the transistor MN11and connect a capacitor between this gate and the ground, while removing the capacitor C11from the IVC204. By this modification, it becomes possible to decrease the capacitance value of the capacitor connected to the gate by increasing the resistance value of the resistor between the SSW201and the gate of the transistor MN11, whereas, for the IVC204shown inFIG. 12, current-voltage conversion gain is determined by the resistance value of the resistor R11and the capacitance value of the capacitor C11is determined from this resistance value and the cutoff frequency required to eliminate the AC component.

Furthermore, although not shown, as is the case forFIG. 10for the first embodiment, it is possible to connect a resistor in series to the capacitor C11. By this modification, the effect of providing a larger phase margin in the feedback loop for keeping the current flowing in the VGA200constant is obtained.

A fourth embodiment of a transmitter of the present invention will be described withFIG. 13. In the fourth embodiment, the VGA109ashown inFIG. 3is modified such that the IDET203is configured with cascaded current mirrors. Other circuits the same as in the first embodiment. The so-modified VGA109aemployed in the fourth embodiment is shown inFIG. 13. InFIG. 13, the IDET203is composed of an N-type MOS transistor MN11and P-type MOS transistors MP3band MP12. The transistor MN11and the transistor MN3aconstitute a current mirror and the transistor MP3band the transistor MP12constitute another current mirror. While the polarity of the OPAMP205is inverted in the third embodiment, a feedback loop is formed in the same way as for the first and second embodiments without inverting the polarity of the OPAMP205in the fourth embodiment. In the VGAB200, a constant bias current flows through the transistor MN3awhich forms a current mirror with the IDET203and a constant transconductance is obtained, and thereby variation in the output level of the VGA is reduced.

Although not shown, it is possible to connect a resistor in series to the capacitor C3cinFIG. 13. By this modification, the effect of providing a larger phase margin in the feedback loop for keeping the current flowing in the VGA200constant is obtained.

A fifth embodiment of a transmitter of the present invention will be described withFIG. 14. In the fifth embodiment, the VGA109ashown inFIG. 3is modified such that the VGAB200is configured as a differential type. Other circuits are the same as in the first embodiment. The so-modified VGA109aemployed in the fifth embodiment is shown inFIG. 14. The VGAB200shown inFIG. 14is configured as a differential amplifier in which a P-type MOS transistor MP14, an N-type MOS transistor MN14, a resistor R14, and capacitors C14aand C14bare added to the VGAB200shown inFIG. 3. Differential input signals are input from terminals RFIN and RFINB and differential output signals are output from terminals RFOUT and RFOUTB. Gain control is performed by the potentials of the sources of the transistor MP3P and the transistor MP14. In comparison with the VGA109aof the first embodiment shown inFIG. 3, common mode noise rejection is enhanced by the differential configuration, though the circuit size increases.

A sixth embodiment of a transmitter of the present invention will be described withFIG. 15. In the sixth embodiment, the VGA109ashown inFIG. 3is modified such that additional transistors are connected in series to the transistors in the VGAB200and a bias circuit is added to the VGAB200. Other circuits are the same as in the first embodiment. The so-modified VGA109aemployed in the sixth embodiment is shown inFIG. 15. In the VGAB200shown inFIG. 15, a resistor R3ais removed and a P-type MOS transistor MP15b, an N-type MOS transistor MN15b, a resistor R15, and a capacitor C15which constitute a bias circuit are added to the VGAB200shown inFIG. 3. Moreover, an N-type MOS transistor MN15ais attached in series to the transistor MN3aand a P-type MOS transistor MP15ais attached in series to the transistor MP3a. In comparison with the first embodiment employing the VGA109ashown inFIG. 3, when the VGA200gain is low in the case of nonconstant envelope modulation, an effect of extending the gain variable range of the VGA109ais expected, which is obtained by reducing signal leakage through gate-drain parasitic capacitances of the transistor MP3aand the transistor MN3aand eliminating signal leakage through a self-bias resistor.

A seventh embodiment of a transmitter of the present invention will be described withFIG. 16. In the seventh embodiment, the VGA109ashown inFIG. 3is modified such that a bias circuit is added to the VGAB200. Other circuits are the same as in the first embodiment. The so-modified VGA109aemployed in the seventh embodiment is shown inFIG. 16. In the VGAB200shown inFIG. 16, a resistor R3ais removed and a P-type MOS transistor MP15b, an N-type MOS transistor MN15b, a resistor R15, and a capacitor C15which constitute a bias circuit are added to the VGAB200shown inFIG. 3. In comparison with the first embodiment shown inFIG. 3, when the VGA200gain is low in the case of nonconstant envelope modulation, an effect of extending the gain variable range of the VGA109ais expected, which is obtained by eliminating signal leakage through a self-bias resistor.

An eighth embodiment of a transmitter of the present invention will be described withFIG. 17. In the eighth embodiment, the VGA109ashown inFIG. 3is modified such that additional transistors are connected in series to the transistors in the VGAB200. Other circuits are the same as in the first embodiment. The so-modified VGA109aemployed in the eighth embodiment is shown inFIG. 17. In the VGAB200shown inFIG. 17, the following transistors are added to the VGAB200shown inFIG. 3; that is, an N-type MOS transistor MN15ais attached in series to the transistor MN3aand a P-type MOS transistor MP15ais attached in series to the transistor MP3a. In comparison with the first embodiment employing shown inFIG. 3, when the VGA200gain is low in the case of nonconstant envelope modulation, an effect of extending the gain variable range of the VGA109ais expected, which is obtained by reducing signal leakage through gate-drain parasitic capacitances of the transistor MP3aand the transistor MN3a.

A ninth embodiment of a transmitter of the present invention will be described withFIG. 18. In the ninth embodiment, the VGA, or the VGA109ashown inFIG. 12is modified such that the transistor MP3ain the VGAB200acts as a load for the transistor MN3aand a bias circuit is added. Other circuits are the same as in the third embodiment. The so-modified VGA employed in the ninth embodiment is shown inFIG. 18. In the VGAB200shown inFIG. 18, a resistor R3ais removed, the gate of the transistor MP3ais grounded, and an N-type MOS transistor MN18, resistors R18a, R18b, R18c, and a capacitor C18which constitute a bias circuit are added to the VGAB200shown inFIG. 12. The transistor MP3aoperates in the deep triode region and acts as a load for the transistor MN3a. Gain control is performed by the potential of the source of the transistor MP3aconnected to the supply voltage of the input bias circuit. In comparison with the third embodiment employing the VGA shown inFIG. 12, when the VGA200gain is low in the case of nonconstant envelope modulation, an effect of extending the gain variable range of the VGA109ais expected, which is obtained by eliminating signal leakage through a self-bias resistor.

As is shown inFIG. 19, it is possible to replace the transistor MP3awith an inductor L19in the VGAB200. Because no voltage drop takes place in the inductor L19, it becomes possible to apply a high voltage to the transistor MN3a.

Moreover, as is shown inFIG. 20, it is possible to replace the inductor L19with a resistor R20in the VGAB200. When circuit elements are integrated on a semiconductor substrate, generally, an area for a resistor is smaller than an area for an inductor. Thus, the size of the VGA109bcan be reduced by adopting the circuitry with the resistor R20.

A mobile communication terminal embodiment of the present invention, which is a tenth embodiment, is shown inFIG. 21. The mobile communication terminal of the tenth embodiment is configured by applying the transmitter of the present invention described for the first through ninth embodiments and dual mode compatible (GSM in which GMSK modulation is performed and EDGE in which 8PSK modulation is performed). Furthermore, this terminal is capable of transmitting and receiving in four frequency bands: that is, a “GSM 850” band (transmitting frequencies of 824-849 MHz, receiving frequencies of 869-894 MHz), a “GSM 900” band (transmitting frequencies of 880-915 MHz, receiving frequencies of 925-960 MHz), a “DCS 1800” band (transmitting frequencies of 1710-1785 MHz, receiving frequencies of 1805-1880 MHz), and a “PCS 1900” band (transmitting frequencies of 1850-1910 MHz, receiving frequencies of 1930-1990 MHz). In GSM and EDGE modes, any of the above four bands is used in accordance with the intended use.

InFIG. 21, reference numeral118denotes a transceiver circuit consisting of a transmitting circuit and a receiving circuit which will be described later and the circuit elements are integrated in a semiconductor integrated circuit (RF-IC). The transmitting circuit is a circuitry to the same extent as the transmitting circuit115shown inFIG. 1. Reference numeral400denotes the entire transceiver circuit including SAW filters (surface acoustic wave filters)401ato401dwhich are band-pass filters and this circuit is integrated as a module. Reference numeral116denotes a transmit output circuit comprising PAs112,112aand an output signal detector113and this circuit is integrate as a module. Reference numeral406denotes a receiver comprising a receiving circuit of the transceiver circuit118and the SAW filters401ato401d.

InFIG. 21, reference numeral117denotes a baseband circuit which is constructed as a Large Scale Integrated (LSI) circuit. The baseband circuit117processes transmit data and received data appropriately and inputs I/Q baseband signals to the quadrature modulator100during transmitting operation and I and Q baseband signals (I/Q) from programmable gain amplifiers (hereinafter denoted as PGAs)404are input to it during receiving operation. The baseband circuit117performs intended control by supplying a control signal (CTRL DATA) to the transceiver circuit118.

In the transmitter410of the mobile communication terminal of the tenth embodiment, a divide-by-2 divider104eand VGAs109aand109dare added to the transmitter according to any of the first through ninth embodiments and a divider104fthat can switch between a divide-by-4 function and a divide-by-2 function is employed instead of the divide-by-4 divider104c.

In GSM 850 and GSM 900 operations, the divide-by-2 divider104e, VGA109a, VGA109b, and PA112operate and the VGA109c, VGA109d, and PA112aare set inoperative. The divider104foperates as a divide-by-4 divider. Other operations are the same as for the first through ninth embodiments.

In DCS 1800 and PCS 1900 operations, the VGA109c, VGA109d, and PA112aoperate and the divide-by-2 divider104e, VGA109a, VGA109b, and PA112are set inoperative. The divider104foperates as a divide-by-2 divider. The VGA109coperates in the same way as the VGA109does in the GSM 850 and GSM 900 operations. The TXVCO108oscillates in a 1.8 GHz band in all of GSM 850, GSM 900, DCS 1800, and PCS 1900 operations.

The receiver406, which is a direct conversion type receiver, is composed of the SAW filters401ato401d, low noise amplifiers (hereinafter denoted as LNAs)402ato402d, direct conversion mixers403ato403h, divide-by-2 dividers104gto104k, and PGAs404aand404bwhich can change their gain discretely. Instead of the PGAs404, VGAs which can change its gain continuously can be used. In the receiver406, the receiving circuit is configured with the circuits except the SAW filters401ato401d.

A received signal is input to a SAW filter that is appropriate for the operating frequency band and the output from the filter is delivered to the baseband circuit117. For example, in GSM 850 operation, a received signal is input to the SAW filter401afrom which the signal is transferred via the LNA402a, the direct conversion mixers403a,403b, and to the PGAs404. A local signal that is input to the direct conversion mixers403a,403bis generated through the divide-by-2 dividers104gand104h.

InFIG. 21, reference numeral405denotes an antenna switch that connects a transmit signal line from the PA112aor PA112to the antenna during transmitting operation and makes a connection between the antenna and an appropriate SAW filter401during receiving operation.

The transceiver circuit118, transmit output circuit116, and entire transceiver circuit400are not limited to those shown in the example ofFIG. 21; for example, the control circuit114may be integrated into the transmit output circuit. While the direct conversion receiver is shown as an example of the receiver406, the receiver is not limited to this type; it is needless to say that, for example, a low-IF receiver and a super-heterodyne receiver can be used.

While the foregoing first through tenth embodiments have illustrated the transmitter and mobile terminal operating in a GSM system using GMSK modulation as a constant envelope modulation system and an EDGE system using 8PSK modulation as a nonconstant envelope modulation system, it is needless to say that the present invention is not so limited and is applicable generally to communication systems in which constant envelope modulation is performed and communication systems in which nonconstant envelope modulation is performed. For example, the invention can also be applied to a wideband Code Division Multiple Access (CDMA) system in which nonconstant envelope modulation is performed.

FIG. 23shows an eleventh embodiment of a transmitter of the present invention. The transmitter of the eleventh embodiment employs a VGA109aas employed in any of the first through ninth embodiments of the present invention. Besides, the transmitter configuration of the eleventh embodiment includes a first carrier frequency generating circuit, a second carrier frequency generating circuits, a PM loop, a quadrature modulator (MOD)100, and an LPF101a. Configured in this manner, the transmitter of the eleventh embodiment can operate only in a system using constant envelope modulation, for example, the GSM system.

The first carrier frequency generating circuit is comprised of an intermediate-frequency IFVCO105and a divide-by-8 divider (⅛)104d. The second carrier frequency generating circuit is comprised of an RFVCO106and a divide-by-4 divider (¼)104c. The PM loop is comprised of a PD107, LPF101d, TXVCO108, VGA109a, mixer102, and LPF101c. The VGA109ais provided as an output buffer of the TXVCO108.

Now, how the transmitter configured as above operates will be described. Output from the baseband circuit, I and Q baseband signals (I/Q) which undergo constant envelope modulation used in GSM are converted into a signal with a first carrier frequency as the center frequency by the quadrature converter100and an unwanted signal is suppressed by the LPF101a. An output signal of the LPF101abecomes a reference signal that is input to the PD107.

By the PM loop, the reference signal and a feedback signal passing via the LPF101cto the PD107are controlled to have the same phase and frequency. That is, the phase control loop synchronizes with the phase of the reference signal generated from the input signal. As a result, at the output of the VGA109a, the phase (or a frequency-modulated component) included in the reference signal is reproduced as phase information and the center frequency is converted into a desired frequency that is determined by the first and second carrier frequencies. If the output frequency of the IFVCO105is, for example, 640 MHz, the first carrier frequency will be 80 MHz and the center frequency of the reference signal also will be 80 MHz. Thus, the center frequency of the feedback signal to the PD107will be 80 MHz. On the other hand, if the frequency of the RFVCO106is 3920 MHz, the second carrier frequency will be 980 MHz. Because the center frequency of the output signal of the VGA109ais frequency converted into 80 MHz by the mixer102, eventually, the center frequency of the output signal of the VGA109awill be 900 MHz.

Provided with the above PM loop, the transmitter of the eleventh embodiment makes an offset PLL transmitter.

Furthermore, in the eleventh embodiment, the internal switches in the VGA109aare placed to the GSM side and the VGA109aoperates as a fixed gain amplifier. That is, the VGA109ais used in the state that a feedback loop is formed to keep the bias current constant and variation in the output level of the offset PLL transmitter is reduced.

Needless to say, it is allowed to remove the internal switches from the VGA109aand use the VGA109as a fixed gain amplifier at all times. This configuration is preferable in terms of cost reduction.

FIG. 24shows a twelfth embodiment of a transmitter of the present invention. In the transmitter of the twelfth embodiment, a divide-by-2 divider104eand a VGA109care added to the transmitter of the eleventh embodiment and a divider104fthat can switch between a divide-by-4 function and a divide-by-2 function is employed instead of the divide-by-4 divider. Configured in this way, the transmitter of the twelfth embodiment is capable of operating in frequency bands that differ by a factor of 2. In consequence, the transmitter can operate in four frequency bands, for example, GSM 850, GSM 900, DCS 1800, and PCS 1900. The VGA109cwhich is also involved in the present invention operates in the same way as the VGA109a, except that it handles different frequency bands from those the VGA109ahandles.

When the transmitter operates in GSM 850 and GSM 900 frequency bands, the divide-by-2 divider104eand VGA109aoperates, the VGA109cis set inoperative, and the divider104foperates as a divide-by-4 divider.

When the transmitter operates in DCS 1800 and PCS 1900 frequency bands, the VGA109coperates, the divide-by-2 divider104eand VGA109aare set inoperative, and the divider104foperates as a divide-by-2 divider.

In the twelfth embodiment, the VGAs109aand109care used in the configuration for constant envelope modulation, that is, in the state that a feedback loop is formed to keep the bias current constant, and variation in the output level of the offset PLL transmitter adapted to transmit in four frequency bands is reduced.

According to the present invention, gain variation is reduced when the variable gain amplifier is used as a fixed gain amplifier. Therefore, an effect is expected in which the transmitter operating with both constant and nonconstant envelope modulation systems and with little variation in the transmitting power can be realized.

It is further understood by those skilled in the art that the foregoing descriptions are preferred embodiments of the disclosed device and that various changes and modifications may be made in the invention without departing from the sprit and scope thereof.