Information compacting method and apparatus, compacted information expanding method and apparatus, compacted information recording/transmitting apparatus, compacted information receiving apparatus and recording medium

Digital audio data of at least two channels are orthogonal-transformed by orthogonal transform for effecting bit compaction. If, when the time length and window shape of the orthogonal transform block is changed, the correlation between channels is found to be high to a certain extent, the time lengths of the orthogonal transform blocks in each channel is selected to be equal to suppress the difference in sound quality between channels to improve a fixed sound image position feeling and hence a high sound quality. This allows to produce a high sound quality for the same bit rate. A lower bit rate may be used for producing the equivalent sound quality. A method for determining the time length of a processing block which is psychoacoustically more desirable processing block may be provided to enable high efficiency compaction and expansion with a psycho-acoustically optimum sound quality.

TECHNICAL FIELD 
This invention relates to a method and apparatus for bit compacting e.g., 
digital audio signals, a method and apparatus for expanding the compacted 
information, a method and apparatus for recording/transmitting the 
compacted information, a recording medium having the compacted information 
recorded thereon, an apparatus for reproducing the compacted information 
from a recording medium, and an apparatus for receiving the compacted 
information. More particularly, it relates to such method and apparatus 
for changing the time duration of a processing block depending on 
amplitude changes of an input signal waveform on the time axis. 
BACKGROUND ART 
The present Assignee has already proposed in e.g., U.S. Pat. No. 5,243,588 
a technology of bit compacting input digital audio signals and recording 
them in a burst fashion on a recording medium with a pre-set data volume 
as a recording unit. 
This technique employs a magneto-optical disc as a recording medium and 
records adaptive differential (AD) PCM audio data as prescribed in the 
audio data formats of the CD-interactive (CD-I) or CD-ROM on the 
magneto-optical disc while reproducing the ADPCM audio data from the 
magneto-optical disc. The ADPCM audio data are recorded in a burst fashion 
on the magneto-optical disc with e.g., 32 sectors of the ADPCM audio data 
and a few linking sectors for an interleaving operation. 
As for the ADPCM audio data in the recording/reproducing apparatus 
employing the magneto-optical disc, several modes can be selected. For 
example, a level A, having a compaction ratio twice that of the 
reproducing time for usual compact disc (CD), with the sampling frequency 
being 37.8 kHz, a level B, having a four-fold compaction ratio, with the 
sampling frequency being 37.8 kHz, and a level C, having an eight-fold 
compaction ratio, with the sampling frequency being 18.9 kHz, are 
prescribed. That is, with the above level B, digital audio data are 
compacted to approximately one-fourth. The reproducing time or play time 
of the disc recorded with the level B is four times that of a standard CD 
format (CD-DA format). This reduces the size of the apparatus because the 
recording/reproducing time approximately equal to that of a smaller sized 
disc having a standard diameter of 12 cm can be achieved. 
Meanwhile, since the speed of disc rotation of the recording/reproducing 
apparatus is the same as that of the standard CD, compacted data 
corresponding to the play time which is four times per unit time can be 
achieved for the level B. Thus the same compacted data is read four times 
in redundancy, with the time units of a sector or a cluster, and only one 
of the four times of reading is routed to audio reproduction. 
Specifically, track jump is carried out for returning to the original 
track position for each revolution during scanning or tracking a 
spirally-shaped recording track, so that the reproduction is carried out 
in such a manner that the same track is repeatedly tracked four times. 
Thus it suffices if regular compacted data is produced for at least one of 
the four times of readout attempts, so that the apparatus is strong 
against errors caused by disturbances or the like and hence may desirably 
be applied to a small-sized portable system. 
It may also be contemplated to use a semiconductor memory as a recording 
medium. Specifically, it may be contemplated to record and/or reproduce 
audio signals using a so-called IC card. Compacted data obtained by bit 
compaction is recorded or reproduced on or from the IC card. Additional 
bit compaction is desirably carried out for further improving the 
compaction efficiency. 
The IC card employing a semiconductor memory, for example, is increased in 
recording capacity or lowered in cost with progress in the semiconductor 
technology. In an initial stage in which the IC card has started to be 
offered to the market, it tends to be insufficient in capacity and 
expensive. Thus it is contemplated to transfer the contents of an other 
recording medium, which is less expensive and larger in capacity, such as 
the above-mentioned magneto-optical disc, to an IC card, a number of 
times, by way of rewriting. Specifically, desired ones of plural airs 
recorded on the magneto-optical disc are dubbed to the IC card and 
occasionally replaced by other airs. By frequently rewriting the contents 
of the IC card, a number of airs may be played outdoors even if only a 
small number of IC cards is owned by the user. 
The present Assignee has proposed in U.S. Pat. No. P 5,197,087 a technique 
of improving the temporal resolution and response characteristics by 
varying processing blocks for compaction responsive to large amplitude 
changes in an input signal. 
This technique resides in changing temporal resolution and frequency 
resolution responsive to characteristics of the input signal. In a 
so-called transform coding employing orthogonal transform, which is among 
the high efficiency compaction methods, the above technique is 
particularly effective against pre-echo which is produced when an input 
signal has rapid amplitude changes. The pre-echo herein means a phenomenon 
in which, if compaction and expansion are carried out while large 
amplitude changes are produced in an orthogonal transform unit, referred 
to hereinafter as an orthogonal transform block, temporally uniform 
quantization noise is produced in the orthogonal transform block and 
raises a problem on the human hearing sense at a small amplitude portion 
of the original signal. 
Meanwhile, with the above-mentioned transform coding, if the time duration 
of the orthogonal transform block is protracted or shortened, the temporal 
resolution or the frequency resolution is lowered, due to characteristics 
of the orthogonal transform, respectively. For example, if the same signal 
is orthogonal transformed using orthogonal transform blocks of different 
time durations, the states of the resulting spectral signals or the 
orthogonal transform coefficients differ appreciably. Such difference is 
naturally reflected in the compacted or expanded signal. It is assumed 
that, for example, the input signal is the two-channel stereophonic music 
signal. While the lengths of the orthogonal transform blocks of respective 
channels are determined in general independently of each other, it may 
occur that the length of the orthogonal transform block of one of the 
channels is selected to be shorter, while that of the other channel is 
selected to be longer, despite the fact that two channels are moderately 
correlated with each other. In such case, a significant variation is 
produced in the states of the spectral signals or the orthogonal transform 
coefficients between the channels, as a result of which the difference in 
the sound quality between the channels becomes outstanding. This causes 
indefinite fixed sound image feeling of the music signals on expansion and 
deteriorated sound quality. 
DISCLOSURE OF THE INVENTION 
In view of the foregoing, it is an object of the present invention to 
provide a technique capable of determining an orthogonal transform block 
size optimally adapted to actual complex input signals and to prevent 
sound quality deterioration under low bit rate conditions as well as to 
improve the sound quality under the constant bit rate condition. 
The present invention, proposed for achieving the above object, provides a 
method for compacting the information in which at least two input signals 
with each one channel are divided into processing blocks having lengths 
adaptively changed to the input signals of the respective channels, and 
information compaction is carried out on the processing block basis. The 
concurrent processing blocks of the respective channels are of equal 
lengths. 
The present invention also provides an apparatus for information compaction 
comprising block dividing means for dividing input signals with at least 
two channels into processing blocks, with the processing block length 
being varied depending on the input signals of the respective channels and 
with the length of the concurrent processing blocks of the respective 
channels being the same, and means for compacting the information for 
processing block based signals. 
The information compacting method and apparatus are designed in the 
following manner. That is, the lengths of the processing blocks for at 
least two of the channels are selected to be equal. With the information 
compacting method and apparatus of the present invention, signal 
correlation between at least two channels is checked and the lengths of 
the processing blocks on the respective channels are selected to be equal 
only when it is judged that the correlation is high. The correlation is 
checked based upon changes in the input signal of a processing block under 
consideration and/or changes in the input signal of an other processing 
block and/or the power or energy or peak information, and/or changes 
and/or the power or energy or peak information in the input signal of a 
concurrent processing block with the processing block under consideration, 
or changes in the input signal of each processing block concurrent with 
the processing block under consideration, and/or changes in the input 
signal of an other processing block and/or power or energy or peak 
information, or changes in the input signal of the processing block under 
consideration or at least one processing block neighboring to the 
processing block under consideration, and/or power or energy or peak 
information. The correlation is checked based upon changes n the input 
signal in the processing block under consideration of each channel and/or 
power or energy or peak information, or changes in the input signal of a 
processing block under consideration in each channel and/or power or 
energy or peak information or on the difference between channels of the 
power or energy or peak information. With the information compaction 
method and apparatus of the present invention, the degree of the pe-set 
masking effect responsive to the input signal is calculated to determine 
the length of the processing block of each channel. The degree of masking 
effect is calculated based upon changes in the input signal of the 
processing block under consideration and/or an other processing block 
and/or the power or energy or peak information, or calculated based upon 
changes in the input signal of the processing block neighboring to the 
processing block under consideration, and/or the power or energy or peak 
information, or calculated based upon changes in the input signal of a 
processing block concurrent to the block under consideration and/or power 
or energy or peak information. With the information compacting method and 
apparatus of the present invention, the ratio of participation in the 
decision of the element for determining the length of the processing block 
under consideration is fixed or adapted to the input signal or rendered 
variable with frequency. With the information compacting method and 
apparatus of the present invention, the time-domain signal is divided by 
orthogonal transform into plural bands on the frequency axis. The 
windowing function employed for orthogonal transform is changed in shape. 
For dividing the time-domain signal into plural bands on the frequency 
axis, the frequency spectrum is divided into plural bands, plural blocks 
each consisting of plural samples are orthogonal transformed to produce 
coefficient data. The bandwidth on the frequency axis are selected to be 
broader with increase in frequency and selected to be equal at at least 
two lowermost bands. With the information compacting method and apparatus 
of the present invention, the allocation of the main information and/or 
subsidiary information of compressed codes is inhibited for signal 
components of the bands approximately higher than the signal passband. 
When determining the time length of the processing blocks using the 
changes in the input signal of the processing block under consideration, 
the boundary value is variable depending on the amplitude and frequency of 
the input signal. 
With a method for expanding the compacted information according to the 
present invention, the information compacted by the information compacting 
method or apparatus is expanded. If orthogonal transform is used for 
information compaction, inverse orthogonal transform is used for 
transforming the signal from plural bands on the frequency axis into 
time-domain signals. For transforming the signal from plural bands on the 
frequency axis into time-domain signals, inverse orthogonal transform is 
applied for each block of each band and the inverse orthogonal transform 
output are synthesized to produce a synthesized time-axis signal. The 
frequency width for synthesis from the plural bands on the frequency axis 
into the time-domain signal is selected to be broader with increase in 
frequency and be equal in at least two lowermost bands. 
The present invention also provides an apparatus for expanding the 
compacted information responsive to the input signal of at least two 
channels and the compacted information of the respective channels is 
expanded on the basis of a concurrent processing block having a pre-set 
length for each channel. The apparatus has expanding means for performing 
expansion which is a counterpart operation of the compression for each 
channel, and synthesizing means for synthesizing the variable length 
processing blocks from the expansion means. The compacted information 
expanding method and apparatus of the present invention expands the 
information compacted by the information compacting method and apparatus. 
In other words, the information compacting method and apparatus and the 
compacted information expanding method and apparatus (high efficiency 
encoding technique and compaction or expansion apparatus) are 
characterized by varying the time length of the orthogonal transform block 
in the course of compaction in conformity to amplitude changes in the 
input signal. The time length of the orthogonal transform block is 
determined based on the energy or power of the frequency band of an other 
channel in addition to the amplitude changes in the time-axis signal in 
the frequency range of the block under consideration, and/or the energy or 
power of the frequency domain of the other channel for calculating the 
masking effect on the orthogonal transform block under consideration if 
the signal correlation between the channels is high. If the masking effect 
can be realized, the total orthogonal transform block sizes of the 
respective channels are temporally long and equal to one another. If the 
masking effect cannot be realized, the total orthogonal transform block 
sizes of the respective channels are temporally short and equal to one 
another. 
The present invention also provides a compacted information 
recording/transmitting apparatus having means for dividing the input 
signal of at least two channels into processing blocks, with the length of 
the processing block being variable responsive to the input signals of the 
respective channels and being constant for each concurrent processing 
block of each channel. The apparatus also has information compacting means 
for compacting the processing block based signal by pre-set information 
compaction, and recording/transmitting means for recording or transmitting 
the compacted information by the information compacting means. The 
compacted information obtained by the information compacting method and 
apparatus is recorded on a recording medium or transmitted on a 
transmission medium. 
The present invention also provides a recording medium having the compacted 
information recorded thereon wherein processing blocks of at least two 
channels are variable in length in conformity to the input signal and the 
compacted information of each channel is recorded based on a processing 
block having the same length for the respective channels. The recording 
medium has recorded thereon the compacted information compacted by the 
information compacting method or apparatus of the present invention. 
The compacted information reproducing apparatus of the present invention 
expands and reproduces the compacted information from the recording medium 
on which the compacted information is pre-recorded by the compacted 
information recording apparatus of the present invention. The compacted 
information receiving apparatus of the present invention receives and 
expands the compacted information transmitted from the compacted 
information transmitting apparatus of the present invention. 
According to the present invention, if, when changing the time size of the 
orthogonal transform block and the window shape against acute changes in 
the input signal, the correlation between channels exist to a certain 
extent, the time length of the orthogonal transform block of each channel 
is selected to be equal to suppress sound quality difference between 
channels to improve a fixed sound image position feeling to provide a 
satisfactory sound quality. 
Thus it becomes possible to achieve better sound quality for the same bit 
rate or to use a lower bit rate to achieve equivalent sound quality.

BEST MODE FOR CARRYING OUT THE INVENTION 
Referring to the drawings, an illustrative embodiment of the present 
invention will be explained in detail. 
In FIG. 1, an arrangement of a compacted data recording/reproducing 
apparatus employing an information compaction method and apparatus and a 
compacted information expanding method and apparatus according to the 
present invention is shown in a block circuit diagram. 
In the compacted data recording/reproducing apparatus, shown in FIG. 1, a 
magneto-optical disc 1, run in rotation by a spindle motor 51, is employed 
as a recording medium. With the present compacted data 
recording/reproducing apparatus, data is recorded along a recording track 
of the magneto-optical disc 1 during data recording on the magneto-optical 
disc 1 by so-called magnetic field modulation in which a magnetic field 
modulated in dependence upon the recording data is applied by a magnetic 
head 54 whilst a laser light is radiated by an optical head 53. With the 
present compacted data recording/reproducing apparatus, the recording 
track on the magneto-optical disc 1 is traced with a laser light of the 
optical head 53 during data reproduction for photomagnetically reproducing 
data. 
Specifically, the optical head 53 is made up of optical components, such as 
a laser light source, e.g., a laser diode, a collimator lens, an objective 
lens, a polarization beam splitter or a cylindrical lens, a photodetector 
having a light receiving section of a pre-set pattern, and so forth. The 
optical head 53 is mounted facing the magnetic head 54 with the 
magneto-optical disc 1 in-between. For recording data on the 
magneto-optical disc 1, the magnetic head 54 is driven by a head driving 
circuit 66 of the recording system as later explained for impressing the 
modulated magnetic field modulated in accordance with the recording data, 
while the laser light is radiated on a target track of the magneto-optical 
disc 1 by the optical head 53 for achieving thermo-magnetic recording in 
accordance with the magnetic field modulation system. On the other hand, 
the optical head 53 detects the laser light reflected from the target 
track after irradiation for detecting the focusing error by e.g. a 
so-called astigmatic method and the tracking error by a so-called 
push-pull method. When reproducing data from the magneto-optical disc 1, 
the optical head 53 detects the focusing error and the tracking error, at 
the same time as it detects the difference in the polarization angle (Kerr 
rotation angle) of the reflected laser light from the target track. 
An output of the optical head 53 is fed to an RF circuit 55. The RF circuit 
55 extracts the focusing error and the tracking error from an output of 
the optical head 53 and transmits the extracted signal to a servo control 
circuit 58. The RF circuit also converts the reproduced signal into a 
hi-level signal which is supplied to a decoder of a reproducing system as 
later explained. 
The servo control circuit 56 is made up e.g., of a focusing servo control 
circuit, a tracking servo control circuit, a spindle motor servo control 
circuit and a thread servo control circuit. The focusing servo control 
circuit focusing controls an optical system of the optical head 53 so that 
the focusing error signal will be reduced to zero. The tracking servo 
control circuit tracking controls the optical system of the optical head 
53 so that the tracking error signal will be reduced to zero. The spindle 
motor servo control circuit controls a spindle motor 51 so that the 
magneto-optical disc 1 will be run in rotation at a pro-set rotational 
velocity, such as at a constant linear velocity. The thread servo control 
circuit causes the optical head 53 and the magnetic head 54 to be moved to 
a target track position of the magneto-optical disc 1 as designated by a 
system controller 57. The servo control circuit 56, effectuating these 
various control operations, transmits the information indicating the 
operating states of various components controlled by the servo control 
circuit 56 to the system controller 57. 
To the system controller 57 are connected a key input unit 58 and a display 
unit 59. The system controller 57 controls the recording system and the 
reproducing system with the operating mode designated by the actuating 
input information entered from the key input unit 58. The system 
controller 57 also controls the recording position and the reproducing 
position on the recording track traced by the optical head 53 and the 
magnetic head 54 based upon the sector-based address information 
reproduced from the recording track of the magneto-optical disc 1 as 
so-called header time or sub-code Q data. The system controller 57 also 
causes the reproducing time to be displayed on a display unit 59 based 
upon the data compaction ratio and the information on the reproducing 
position on the recording track. 
For displaying the reproducing time, the sector-based address information 
(absolute time information) reproduced from the recording track of the 
magneto-optical disc 1 as the header time or the sub-code Q data is 
multiplied by a reciprocal of the data compaction ratio, which is equal to 
4 for the 1/4 compaction, for finding the actual time information, which 
is displayed on the display unit 59. If the absolute time information is 
pre-recorded (pre-formatted) on the recording track of the magneto-optical 
disc 1, the current position may be also displayed in terms of the actual 
recording time during recording by reading out the pre-formatted absolute 
time information. 
The recording system of the present compacted data recording/reproducing 
apparatus is hereinafter explained. 
An analog audio input signal AIN from an input terminal 80 is supplied via 
a low-pass filter 81 to an A/D converter 82. The A/D converter 82 
quantizes the analog audio input signal AIN into a quantized signal, that 
is into e.g., a 16-bit digital audio signal. The digital audio signal from 
the A/D converter 62 is fed to an adaptive transform encoder (ATC) 63. 
To the ATC encoder 63 is fed a digital audio input signal DIN from an input 
terminal 67 via a digital interfacing circuit 68. The ATC encoder 63 
effectuates bit compaction (data compression) on the digital audio signal 
of a pre-set transfer rate which is the analog audio signal AIN quantized 
by the A/D converter 62. Although the following description is made for 
the compaction ratio of 4, the present embodiment is not limited to the 
value of the compaction ratio which may be optionally selected according 
to the usage and application. 
A memory 64, controlled as to data writing and data readout by the system 
controller 57, is employed as a buffer memory for temporally storing a 
compacted digital audio signal from the ATV encoder 63, hereinafter 
referred to as ATC audio data, and for recording the stored data on the 
magneto-optical disc 1 whenever the necessity arises. That is, the ATC 
audio data, supplied from e.g., the ATC encoder 63, has the data transfer 
rate reduced to about one-fourth of the data transfer rate of the standard 
CD-DA format of 75 sectors per second, that is to 18.75 sector per second. 
It is this ATC audio data that is continuously recorded in the memory 64. 
Although it suffices to record the ATC audio data at a rate of one of four 
sectors, as previously explained, sector-continuous recording is made, as 
will be explained later, because recording of every fourth sector is 
virtually infeasible. The recording is made in a burst fashion, with the 
interposition of a non-recording period, at the same data transfer rate of 
75 sectors per second as the standard CD-DA format, with a pre-set 
plurality of sectors, for example, 32 plus a few sectors, or a cluster, as 
a recording unit. That is, in the memory 84, the ATC audio data, 
continuously written at a low transfer rate of 18.75 (=75/4) sectors per 
second corresponding to the above-mentioned bit compaction rate, is read 
out as recording data in a burst fashion at the above-mentioned transfer 
rate of 75 sectors per second. As for the thus read-out and recorded ATC 
audio data, the overall data transfer rate, inclusive of the non-recording 
period, is the above-mentioned low rate of 18.75 sectors per second. 
However, the instantaneous data transfer rate within the time of the 
burst-like recording is the above-mentioned standard rate of 75 sectors 
per second. Thus, if the rotational speed of the disc is equal to that of 
the standard CD-DA format, that is the constant linear velocity, the 
recording may be made at the same recording density or recording pattern 
as that of the CD-DA format. 
The ATC audio data, that is the recorded data, read out from the memory 84 
in a burst fashion at the above-mentioned instantaneous transfer rate of 
75 sectors per second, is supplied to the encoder 65. In the data string 
supplied from the memory 64 to the encoder 65, the unit of continuous 
recording for each recording is a cluster composed of plural (e.g., 32) 
sectors, and a few cluster-linking sectors arrayed ahead and at back of 
the cluster. The cluster-linking sectors are set so as to be longer than 
the interleaving length at the encoder 85, such that data of other 
clusters is not affected by the interleaving operation. 
The encoder 85 effectuates error correction coding, such as parity 
appendage and interleaving, or EHM encoding, on the recording data 
supplied thereto in a burst fashion from the memory 84. The recording data 
thus encoded by the encoder is supplied to a magnetic head driving circuit 
88. The magnetic head driving circuit 88 is connected to the magnetic head 
54 and drives the magnetic head 54 for applying a modulated magnetic field 
corresponding to the recording data to the magneto-optical disc 1. 
The system controller 57, controlling the memory 64 as described above, 
controls the recording position so that the recording data read out in a 
burst fashion from the controlled memory 64 will be continuously recorded 
on the recording track of the magneto-optical disc 1. The recording 
position control is achieved by checking the recording position of the 
recording data read out in a burst fashion from the memory 64 under 
control by the system controller 57 for supplying a control signal 
designating the recording position on the recording track of the 
magneto-optical disc 1 to the servo control circuit 56. 
The reproducing system for the compacted data recording/reproducing 
apparatus is now explained. The reproducing system is designed to 
reproduce the recording data continuously recorded on the recording track 
of the magneto-optical disc 1 by the above-described recording system. 
Thus the reproducing system has the above-mentioned RF circuit 55 for 
converting the playback output, obtained on tracing the recording track of 
the magneto-optical disc 1 by the optical head 53 by a laser light beam, 
into a bi-level signal, and a decoder 71 for decoding the bi-level 
playback signal. Meanwhile, it is possible with the present reproducing 
system to read out not only the magneto-optical disc 1 but also a 
replay-only optical disc, such as a compact disc (CD). 
The decoder 71, which is a counterpart device of the encoder 65 of the 
above-described recording system, effectuates decoding for error 
correction and EFM decoding on the bi-level playback signals from the RF 
circuit 55, and reproduces the ATC audio data at a transfer rate of 75 
sectors per second which is faster than the regular transfer rate. The 
playback data from the decoder 71 is supplied to a memory 72. 
The memory 72 is controlled as to data writing and data readout by the 
system controller 57 such that the playback data supplied from the decoder 
71 at the transfer rate of 75 sectors per second is written therein in a 
burst fashion at the transfer rate of 75 sectors per second. On the other 
hand, the playback data written in a burst fashion in the memory 72 at the 
transfer rate of 75 sectors per second is continuously read out from the 
memory 72 at the regular transfer rate of 75 sectors per second. 
That is, the system controller 57 controls the memory 72 so that the 
playback data will be written therein at the transfer rate of 75 sectors 
per second and so that the playback data will be continuously written 
therefrom at the above-mentioned transfer rate of 18.75 sectors per 
second. The system controller 57, controlling the memory 72 as described 
above, also controls the playback position so that the playback data 
written in a burst fashion from the controlled memory 72 will be 
continuously reproduced from the recording track of the magneto-optical 
disc 1. The playback position control is made by the system controller 57 
controlling the playback position of the playback data read out in a burst 
fashion from the memory 72 for supplying a control signal designating the 
playback position on the recording track of the magneto-optical disc 1 to 
the servo control circuit 
The playback data, that is the ATC audio data, continuously read out from 
the memory 72 at the transfer rate of 18.75 sectors per second, is 
supplied to an ATC decoder 73. The ATC decoder 73 expands the ATC audio 
data four-fold by bit expansion for regenerating e.g., 16-bit a digital 
audio signal (digital audio data), which is supplied to an A/D converter 
74. 
The D/A converter 74 converts the digital audio data supplied from the ATC 
decoder 73 into an analog signal to form an analog audio output signal 
AOUT. The analog audio output signal AOUT from the D/A converter 74 is 
outputted at an output terminal 76 via a low-pass filter 75. 
Meanwhile, the present compacted data recording/reproducing apparatus is 
designed for converting the ATC audio data from the ATC encoder 63 into 
data of a pre-set transmission format by a modulator 77 for transmission 
over an antenna 78. 
The high efficiency compaction encoding, employing the information 
compaction method according to the present invention, is now explained. 
Specifically, the technique of high efficiency encoding an input digital 
signal, such as audio PCM signal, by a sub-band coding (SBC), adaptive 
transform coding (ATC) and adaptive bit allocation, is now explained by 
referring to FIG. 2 ff. 
In the illustrative high efficiency encoding device, shown in FIG. 2, the 
input digital signal is split int plural frequency bands so that two 
lowermost neighboring bands will be of an equal bandwidth and so that the 
bandwidth will become broader with increase in frequency. The input 
digital signal is then orthogonal transformed on the band basis. The 
resulting frequency-domain spectral data is encoded by adaptive bit 
allocation according to so-called critical bands for taking into account 
the psychoacoustic characteristics of the human auditory system for the 
low frequency range and according to sub-bands further divided from the 
critical bands for taking into account the block floating efficiency for 
the mid to high frequency range. These blocks usually become the blocks 
subjected to quantization noise. In the present embodiment, the block size 
for orthogonal transform is adaptively changed before orthogonal transform 
responsive to the input signal, while block-based floating is performed. 
FIG. 2 shows a circuit arrangement for encoding a 1-channel input digital 
signal. 
Referring to FIG. 2, plural-channel audio PCM signals having a frequency 
range of from 0 to 22 kHz, with the sampling frequency of 44.1 kHz, are 
supplied to an input terminal 200. The input signal is divided by a band 
splitting filter 201, such as QMF, into a signal having a range of from 0 
to 11 kHz and a signal having a range of from 11 kHz to 22 kHz. The signal 
in the range of from 0 to 11 kHz is further divided by a band splitting 
filter 202, such as QMF, into a signal having a range of from 0 to 5.5 kHz 
and a signal having a range of from 5.5 kHz to 11 kHz. The 11 to 22 kHz 
range signal from the band-splitting filter 201 is supplied to a modified 
DCT circuit 203, which is a sort of an orthogonal transform circuit, for 
modified DCT processing. Similarly, the 5.5 kHz to 11 kHz range signal 
from the band splitting filter 202 is supplied to a modified DCT circuit 
204, while the 0 to 5.5 kHz range signal from the band-splitting filter 
202 is supplied to a modified DCT circuit 205, for MDCT processing, 
respectively. 
Among the techniques of dividing the input digital signal int plural 
frequency ranges is the above-mentioned QMF filter, as discussed in R. E. 
Crochiere, Digital Coding of Speech in Subbands Bell Tech. J. Vol.55, 
No.8, 1978. The technique of dividing the frequency spectrum into 
frequency bands of an equal bandwidth is discussed in J. H. Rothweiler, 
Polyphase Quadrature Filters- A New Subband Coding Technique, ICASSP 83, 
Boston. Among the techniques for orthogonal transform, there is a 
technique of dividing the input audio signal into frames each of a pre-set 
time duration and processing the resulting blocks with fast Fourier 
transform (FFT) or discrete cosine transform (DCT) for transforming the 
signal from the time axis into the frequency axis. Discussion of MDCT may 
be found in J. P. Princen and A. B. Bradley, University of Surrey Royal 
Melbourne Inst. of Tech., Subband/Transform Coding Using Filter Bank 
Designs Based on Time Domain Aliasing Cancellation, ICASSP 1987. 
FIG. 3 shows illustrative examples of band-based blocks of a standard input 
signal supplied to the respective MDCT circuits 203 to 205, 
In the illustrative example of FIG. 3, each of the signals in the 
respective bands has its unique orthogonal transform block size and may 
have its time resolution switched depending upon time characteristics and 
frequency distribution proper to the signals. If the signal is temporally 
quasi-stationary, the orthogonal transform block size is set to 11.6 mS, 
as shown in FIG. 3A (long mode). If the signal is non-stationary, the 
orthogonal block size is further divided in two or four portions. The 
orthogonal transform block size is equally divided into four portions each 
being 2.6 mS as shown in FIG. 3B for the short mode or the orthogonal 
transform block size is partially divided in two portions each being 5.8 
mS and partially into four portions each being 2.9 mS as shown in FIG. 3C 
for the middle mode A or in FIG. 3D for the middle mode B, respectively, 
in order to cope with the actual complex input signal. Meanwhile, the 
orthogonal transform block size division may be increased in number and/or 
pattern in order to cope with the input signal more flexibly. The 
orthogonal transform block size is determined by block size decision 
circuits 208 to 205 of FIG. 2. The orthogonal transform block size thus 
determined is supplied to the MDCT circuits 203 to 205 and thence 
outputted as the block size information of the respective blocks at output 
terminals 216, 217 and 218, respectively. 
The block size decision circuits 206, 207 and 208 shown in FIG. 2 will now 
be explained. FIG. 4 shows an illustrative circuit arrangement of the 
block size decision circuit 206. Among the outputs of the band splitting 
filter 201 shown in FIG. 2, the signal in the range of from 11 to 22 kHz 
is fed via an input terminal 401 of FIG. 4 to a power calculating circuit 
404. Among the outputs of the band splitting filter 202 shown in FIG. 2, 
the signal in the range of from 5.5 to 11 kHz is fed via an input terminal 
402 of FIG. 4 to a power calculating circuit 405, while the signal in the 
range of from 0 to 5.5 kHz is fed via an input terminal 403 of FIG. 4 to a 
power calculating circuit 406. 
Meanwhile, the operation of the block size decision circuits 207, 208 shown 
in FIG. 2 is the same as that of the block size decision circuit 206, with 
the difference being solely that input signals to input terminals 401, 402 
and 403 differ from those with the block size decision circuit 206. That 
is, in the block decision circuit 207 of FIG. 2, the signal in the range 
of from 5.5 kHz to 11 kHz from the band splitting filter 202 of FIG. 2 is 
supplied to the input terminal 401 of FIG. 4. Similarly, the signal in the 
range of from 11 kHz to 22 kHz from the band splitting filter 201 is 
supplied to the input terminal 402 of FIG. 4, while the signal in the 
range of from 0 to 5.5 kHz from the band splitting filter 202 is supplied 
to the input terminal 403 of FIG. 4. In the block decision circuit 208 of 
FIG. 2, the signal in the range of from 0 to 5.5 kHz from the band 
splitting filter 202 of FIG. 2 is supplied to the input terminal 401 of 
FIG. 4, while the signal in the range of from 11 kHz to 22 kHz from the 
band splitting filter 201 is supplied to the input terminal 403 of FIG. 4. 
The signal in the range of from 5.5 to 11 kHz from the band splitting 
filter 202 of FIG. 2 is supplied to the input terminal 403 of FIG. 4. The 
block size decision circuits 206 to 208 are provided for each channel. 
Meanwhile, the block size decision circuits 206 to 208 may be provided for 
one channel only for determining the orthogonal transform block size for 
plural channels. 
Referring to FIG. 4, the power calculating circuits 404, 405 and 406 
integrate the input time-axis waveform signals for a preset time for 
calculating the power for the respective frequency ranges. The integration 
time needs to be smaller than the smallest of the above-mentioned 
orthogonal transform block sizes. Alternatively, the absolute value of the 
maximum amplitude or the mean amplitude value within the minimum time 
width of the orthogonal transform block size may be employed as a 
representative power. The power information, an output of the power 
calculating circuit 404, is supplied to a memory 410, a inter-channel 
correlative coefficient calculating circuit 411, a variant extraction 
circuit 407 and to a power comparator circuit 409. The power information 
from the power calculating circuits 405, 408 is fed to the power 
comparator circuit 409. Meanwhile, the power calculating circuits 404, 405 
and 406 may be provided on the channel basis so that the power information 
or the respective channels will be calculated in the power calculating 
circuits 404, 405 and 406 for the respective channels. 
The variant extraction circuit 407 differentiates the power information 
supplied from the power calculating circuit 404 in order to find a 
differentiation coefficient which is supplied as the power variant 
information to a primary block size decision circuit 412 and a memory 408. 
The memory 408 stores the power variant information supplied from the 
variant extraction circuit 407 for a time duration not less than the 
maximum time for the above-mentioned orthogonal transform block size. The 
reason is that the temporally neighboring orthogonal transform blocks 
affect one another by the windowing for orthogonal transform and hence the 
primary block size decision circuit 412 is in need of the power variant 
information for a directly temporally previous block. 
Based upon the power variant information of a block under consideration, 
supplied from the variant extraction circuit 407, and the power variant 
information for the block directly temporally previous to the block under 
consideration, supplied from the memory 408, the primary block size 
decision circuit 412 determines the orthogonal transform block size of the 
frequency band under consideration from time change of the power within 
the frequency band under consideration. Specifically, the primary block 
size decision circuit 412 selects a temporally shorter orthogonal 
transform block size if, for example, a time change in excess of a certain 
threshold value is noticed. Although the threshold may be fixed without 
becoming ineffective, it may be set so as to be proportional to frequency 
so that a temporally shorter orthogonal transform block size is selected 
for a larger time change for the high frequency range and a temporally 
shorter orthogonal transform block size is selected for a smaller time 
change for the low frequency range, for utmost effects. The orthogonal 
transform block size thus determined is fed to a secondary block size 
decision circuit 413. 
On the other hand, the power comparator circuit 409 compares the power 
information data of the respective frequency bands supplied from the power 
calculating circuits 404, 405 and 408 for a time length for which the 
concurrent masking effect is produced in order to find the effect on the 
output frequency band of the power calculating circuit 404 of the other 
frequency band, and transmits the resulting masking information to the 
secondary block size decision circuit 413. 
The secondary block size decision circuit 413 corrects the orthogonal 
transform block size from the primary block size decision circuit 412 so 
as to become a temporally longer block size, based upon the masking 
information supplied from the power comparator circuit 409, and transmits 
the corrected orthogonal transform block size to a tertiary block size 
decision circuit 414. That is, the secondary block size decision circuit 
413 corrects the orthogonal transform block size by taking advantage of 
the fact that, if the pre-echo raises the problem in the frequency band 
under consideration, but if there is any signal of a larger amplitude in 
an other frequency band, above all, in a frequency range lower than the 
frequency band under consideration, the pre-echo may not be obstructive to 
the hearing sense, or the ill effect by the pre-echo may be diminished. 
Meanwhile, the masking means a phenomenon in which a signal becomes 
inaudible by another signal due to psychoacoustic characteristics of the 
human hearing system. The masking effect is classified into a time-axis 
masking effect by time-domain signals and a concurrent masking effect by a 
frequency-domain signal. By the masking effect, any noise present in a 
masked portion becomes inaudible. Thus, with actual audio signals, the 
noise in the masked portion is handled as not being obstructive to the 
human hearing system. 
The inter-channel correlative coefficient calculating circuit 411 
calculates the correlative coefficient of the power between neighboring 
channels, using the power information from plural channels from the memory 
410 and the power calculating circuit 404. 
Specifically, the memory 410 is used for supplying the power information of 
plural channels occurring at the same time point as the block under 
consideration to the inter-channel correlative coefficient calculating 
circuit 411. That is, the power information data of plural channels are 
supplied temporally consecutively to the memory 410 from the power 
calculating circuit 404. In the case of two-channel stereo signals, for 
example, the power information data for the left channel of the block 
under consideration, the power information data for the right channel of 
the block under consideration, the power information data for the left 
channel of a block temporally neighboring to the block under consideration 
and the power information data for the right channel of the block 
temporally neighboring to the block under consideration, are consecutively 
supplied in this order from the power calculating circuit 404 to the 
memory 410. The memory 410 holds the power information data of the 
respective channels in order to output the power information data of the 
respective channels of the block concurrent to the block under 
consideration. Thus the memory 410 has a storage capacity proportionate to 
the number of channels. If the memory 410 for two channels has a capacity 
C, a capacity Cn of the memory 410 for n channels is found from the 
equation (1): 
EQU Cn=(n-1)C (1) 
The inter-channel correlative coefficient calculating circuit 411 is fed 
with the power information data of one or plural channels stored in the 
memory 410 and the one-channel power information data from the power 
calculating circuit 404 not stored in the memory 410 in order to calculate 
the correlative coefficient of the power information data of the 
respective channels concurrent with the block under consideration. If the 
number of channels is two, for example, the correlative coefficient r may 
be defined by the equation (2): 
##EQU1## 
where Xi is the power information of the left channel, Yi is the power 
information of the left channel, Ax is a mean value of Xi, Ay is a mean 
value of Yi, Sx is a standard deviation of Xi and Sy is a standard 
deviation of Yi. 
In general, the value of the correlative coefficient E is such that 
-1.ltoreq.r.ltoreq.+1, so that, if Xi and Yi are correlated to each other 
to a larger extent, the value of r is closer to +1, whereas, if Xi and Yi 
are correlated with each other to a lesser extent, the value of r is 
closer to -1. In the above equation (2), b is an integer determining the 
number of blocks taken into account, that is the time range. Although the 
value of b may be fixed without becoming ineffective, it is more effective 
for the value b to be proportionate to the frequency, in such a manner 
that the difference between b and n becomes larger and smaller for the 
lower frequency range and for the higher frequency range, respectively. Ax 
and Ay are mean values of the power information data comprised between b 
and n. If the number of channels is three or more, the correlative 
coefficient is found for the totality of possible pairs and a mean value 
of the correlative coefficients is found as an output of the inter-channel 
correlative coefficient calculating circuit 411. The number of the 
totality of the possible pairs is {N{N-1)}/2, if the number of channels is 
N. 
The tertiary block size decision circuit 414 then re-checks the orthogonal 
transform block size, as determined by the secondary block size decision 
circuit 413, based upon the correlative coefficient r as found by the 
inter-channel correlative coefficient calculating circuit 411, the masking 
information as found by the power comparator circuit 409 and the power 
information of the orthogonal transform block temporally neighboring and 
directly previous to the orthogonal transform block under consideration, 
as held by the memory 408, and ultimately determines the orthogonal 
transform block size of the orthogonal transform block under 
consideration. 
Specifically, the correlative coefficient r sent from the inter-channel 
correlative coefficient calculating circuit 411 is a value ranging from -1 
to +1. The closer the value of the coefficient, the higher is the 
correlation between channels. 
Thus, if the correlation coefficient exceeding a threshold as set by the 
tertiary block size decision circuit 414 enters it, the concurrent masking 
effect can be expected, and the power information has a value exceeding a 
certain threshold value, the tertiary block size decision circuit 414 
unanimously sets the concurrent orthogonal transform block sizes of the 
plural channels to a longer value, for example, to e.g., 11.6 mS which is 
the value for the long mode shown in FIG. 3A. On the other hand, if the 
correlative coefficient has a value larger than a pre-set threshold, the 
concurrent masking effect cannot be expected and the power information 
from the memory 408 has a value smaller than a certain threshold, the 
tertiary block size decision circuit 414 unanimously sets the concurrent 
orthogonal transform block sizes of the plural channels to a shorter 
value, for example, to a value equal to that of the short mode shown in 
FIG. 3B. Although the above threshold values may be fixed without becoming 
ineffective, the threshold values may be variable depending on the 
frequency for utmost effects. 
The values of the power information data of the respective channels may be 
compared to one another in the inter-channel correlative coefficient 
calculating circuit 411, in place of finding the correlative coefficient. 
If the number of channels is two, the absolute value of the difference of 
the power information data is found. If the number of channels is three or 
more, the absolute values of the differences of the totality of possible 
pairs and the mean value thereof is found. The resulting value is 
transmitted to the tertiary block size decision circuit 414. 
The tertiary block size decision circuit 414 determines the size of the 
orthogonal transform block under consideration, based upon the difference 
of the power information data as found by the inter-channel correlative 
coefficient calculating circuit 411, masking information as found by the 
power comparator circuit 409 and the power information of a directly 
temporally previous block as held on the memory 408. For example, if the 
difference of the power information data has a value lower than a certain 
threshold value, concurrent masking effect can be expected and the power 
information of a directly temporally previous orthogonal transform block 
assumes a value larger than a certain threshold value, the tertiary block 
size decision circuit 414 unanimously sets the size of the concurrent 
orthogonal transform blocks of plural channels to a longer value, such as 
a value equal to the value for the long mode shown in FIG. 3A. On the 
other hand, if the difference of the power information data has a value 
lower than a certain threshold value, concurrent masking effect cannot be 
expected and the power information of a directly temporally previous 
orthogonal transform block assumes a value smaller than a certain 
threshold value, the tertiary block size decision circuit 414 unanimously 
sets the size of the concurrent orthogonal transform blocks of plural 
channels to a small value, such as a value equal to the value for the 
short mode shown in FIG. 3B. Although the above threshold values may be 
fixed without becoming ineffective, the threshold values may be variable 
depending on the frequency for utmost effects. 
The size of the orthogonal transform block BS, as determined by the block 
size decision circuit 414, is outputted at an output terminal 416 to the 
MDCT circuit 203 shown in FIG. 2, while being supplied to a window shape 
decision circuit 415. The window shape decision circuit 415 determines the 
window shape based upon the size of the orthogonal transform block BS. 
FIG. 5 shows adjacent blocks and window shapes. The windows employed in the 
orthogonal transform have portions overlapping between neighboring blocks. 
In the present embodiment, the windows have portions overlapping with 
neighboring blocks as far as the mid points thereof. Thus the window shape 
is changed depending upon the size of the neighboring orthogonal transform 
blocks. 
FIG. 6 shows details of the above window shape. In this figure, the window 
functions f(n) and g(n+N) are given as functions satisfying the following 
equations (3) and (4): 
EQU f(n)f(L-1-n)=g(n)g(L-1-n) (3) 
EQU f(n)f(n)+g(n)g(n)=1 (4) 
0.ltoreq.n.ltoreq.L-1 
In the above equation (3), L becomes the orthogonal transform block size if 
the neighboring orthogonal transform block sizes are equal to each other. 
However, if the neighboring orthogonal transform block sizes are 
different, the window functions are given, for the non-overlapping window 
area, by the equations (5) and (6): 
EQU f(n)=g(n)=1 (5) 
K.ltoreq.n.ltoreq.3K/2-L/2 
EQU f(n)=g(n)=0 (6) 
3K/2+L.ltoreq.n.ltoreq.2K 
where L and K denote the orthogonal transform block size of a shorter time 
duration and the orthogonal transform block size of a longer time 
duration, respectively. Thus the frequency resolution in orthogonal 
transform may be improved by setting the length of the overlapping window 
portion to as long a value as possible. It is seen from the above 
description that the shape of the window employed for orthogonal transform 
is determined after determining the temporally consecutive three 
orthogonal block sizes. 
Meanwhile, the power calculating circuits 405,408 and the power comparator 
circuit 409 shown in FIG. 4 may be omitted from the block size decision 
circuits 206, 207 and 208 shown in FIG. 2. In addition, the secondary 
block size decision circuit 413 and/or the tertiary block size decision 
circuit 414 shown in FIG. 4 may be omitted from the block size decision 
circuits 206, 207 and 208. The above-mentioned constitution with a small 
delay may be effectively employed in an example of application in which 
processing delay is not desirable. 
On the other hand, in the tertiary block size decision circuit 414, the 
time durations of all of the concurrent processing blocks may be selected 
to be equal by setting the threshold values to a lower value. This is 
particularly effective for an input signal having high inter-channel 
correlation. 
The illustrative operation of the primary block size decision circuit 412, 
secondary block size decision circuit 413 and the tertiary block size 
decision circuit 414 will now be explained. 
It is assumed that the signals in the respective bands are sine waves, and 
that the signal level (amplitude) in the range of 11 kHz to 22 kHz of an 
input signal shown in FIG. 7A is equal to the signal level (amplitude) in 
the range of 11 kHz to 22 kHz of an input signal shown in FIG. 7B, as 
shown for example in FIGS. 7A and 7B. 
First, if the orthogonal transform block size of a block N under 
consideration is determined solely by amplitude changes in the frequency 
band under consideration, the orthogonal transform block size is selected 
to be equal for the input signal shown in FIG. 7A and that shown in FIG. 
7B. However, if the signals in the bands of 0 to 5.5 kHz or 5.5 kHz to 11 
kHz are taken into consideration, since the power of the signal in a band 
other than the band of 11 kHz to 22 kHz, is lower than that in the band of 
11 kHz to 22 kHz, the pre-echo occurring in the band of 11 kHz to 22 kHz 
is not masked and presents a problem in connection with the human auditory 
sense. Thus, in the present embodiment, the orthogonal transform block 
size of a shorter time duration is selected for the block N of the band of 
from 11 to 22 kHz for the input signal shown in FIG. 7A. 
Conversely, with the input signal shown in FIG. 7B, the power of a signal 
in the band of 0 to 5.5 kHz or the band from 5.5 kHz to 11 kHz is of a 
value sufficient to mask the pre-echo, as compared to the signal power for 
the band of 11 kHz to 22 kHz, the pre-echo occurring in the band of from 
11 kHz to 22 kHz is masked and does not raise a problem in connection with 
the human auditory sense. Thus, in the present embodiment, emphasis is 
placed on frequency resolution for the input signal shown in FIG. 7B, and 
an orthogonal transform block size of a time duration longer than that of 
the input signal shown in FIG. 7A is employed. 
That is, with the present embodiment, different orthogonal block sizes are 
selected for the input signal shown in FIG. 7A and that shown in FIG. 7B 
by the power calculating circuits 404, 405 and 406, power comparator 
circuit 409 and by the secondary block size decision circuit 413 shown in 
FIG. 4. 
It is then assumed that two input signals for a certain band, such as from 
11 kHz to 22 kHz, are sine waves, which are increased in level at a 
different phase, as shown in FIGS. 8A and 8B. It is also assumed that the 
two input signals are two-channel stereo signals, with the input signal 
shown in FIG. 8A being a left channel signal and with the input signal 
shown in FIG. 8B being a right channel signal. Meanwhile, such small phase 
difference between different channels frequently occurs in 
stereophonically recorded music signals. 
First, if the orthogonal transform block size of a block under 
consideration N is determined solely by signal amplitude changes, an 
orthogonal transform block size of a shorter time duration is selected for 
the input signal shown in FIG. 8A, while an orthogonal transform block 
size of a longer time duration is selected for the input signal shown in 
FIG. 8B. These orthogonal block sizes are selected because the relation 
Da&gt;T&gt;Db holds, where Da in FIG. 8A and Db in FIG. 8B are respective 
absolute values of the differences between the maximum amplitude in the 
blocks (N-1) and that in the blocks N and T is a pre-set threshold value, 
although the differences in the above inequality are small. The result is 
that, despite the high correlation between the channels of the input 
signal, the difference in the spectral components or the orthogonal 
transform coefficients in the respective channels resulting from 
orthogonal transform differ significantly thus producing significant 
difference in the sound quality between the channels. For preventing the 
difference in the sound quality from being produced in case such signal is 
entered, the orthogonal transform block size of a longer time duration is 
selected for each channel if the concurrent masking effect or the 
time-domain masking effect acts for the block under consideration N. 
Conversely, if the concurrent masking effect or the time-domain masking 
effect fails to act for the block under consideration N, the orthogonal 
transform block size of a shorter time duration is selected for each 
channel, for preventing the difference in the sound quality from being 
produced between channels. 
With the present embodiment, the tertiary block size decision circuit 414 
equates the orthogonal transform block sizes of the respective channels 
for the input signals of high correlation between the channels, as shown 
in FIG. 8. It is also effective to equate the time durations of the 
processing blocks of at least two of the totality of channels. 
Returning to FIG. 2, the low frequency range spectral data or MDCT 
coefficient data on the frequency axis, resulting from MDCT by the MDCT 
circuits 203 to 205, are grouped according to the critical bands. On the 
other hand, the mid to high range spectral data are grouped according to 
sub-bands divided from the critical bands in view of block floating. The 
spectral data of the respective ranges are supplied to adaptive bit 
allocation encoding circuits 210, 211 and 212. The critical band is the 
band of noise that can be masked by a pure sound that has the same 
intensity as the noise and has a frequency in the vicinity of the 
frequency of the noise. The width of the critical band increases with 
increase in frequency. The entire frequency spectrum of from 0 to 22 kHz s 
split into e.g., 25 critical bands. 
A bit allocation calculating circuit 209 finds the masking quantity for the 
critical bands and respective sub-bands, taking into account the so-called 
masking effect, based upon the spectral data divided into the critical 
bands and the sub-bands. The bit allocation calculating circuit 209 also 
calculates the numbers of allocated bits for the respective bands, based 
upon the masking quantity and an energy peak or signal energy for each 
critical band and each sub-band. The spectral data or MDCT coefficient 
data are re-quantized responsive to the numbers of allocated bits 
allocated to the respective bands by the adaptive bit allocation and 
encoding circuits 210, 211 and 212. The encoded data are outputted at 
output terminals 213, 214 and 215. 
FIG. 9 shows, in a block diagram, an arrangement of an illustrative example 
of the bit allocation calculating circuit 209. 
In this figure, the spectral data or MDCT coefficient data on the frequency 
axis from the MDCT circuits 203, 204 and 205 are supplied via an input 
terminal 900 to a band-based energy calculating circuit 901. The energy 
calculating circuit 901 calculates the sum total of the amplitudes in the 
respective bands in order to find the energy of the respective critical 
bands and the sub-bands and the above-mentioned masking quantity. The 
amplitude peaks or mean values may also be used in place of the band-based 
energy. 
As an output of the energy calculating circuit 901, FIG. 10 shows at SB the 
spectrum of the sums of the respective bands. In FIG. 10, the masking 
quantity and the sub-bands are denoted as sub-bands are denoted as 12 
bands (B1 to B12). 
For scrutinizing the effect of the masking on the spectrum SB, the spectral 
components SB are multiplied by a pre-set weighting function and summed 
together by way of performing convolution. To this end, band-based outputs 
of the energy calculating circuit 901 are transmitted to a convolution 
filter circuit 902. The convolution filter circuit 902 is made up of 
plural delay elements for sequentially delaying input data, plural 
multipliers, such as 25 multipliers in association with the respective 
bands, for multiplying outputs of the delay elements by filter 
coefficients (weighting functions) and an addition unit for summing the 
multiplier outputs. By such convolution, the sum of areas denoted by 
broken lines in FIG. 10 is found. 
Only by way of example, outputs of the delay elements are multiplied by a 
coefficient 1 by a multiplier M for an arbitrary band, a coefficient 0.15 
by a multiplier M-1, a coefficient 0.0019 by a multiplier M-2, a 
coefficient 0.0000086 by a multiplier M+3, a coefficient 0.4 by a 
multiplier M+1, a coefficient 0.06 by a multiplier M+2 and by a 
coefficient 0.007 by a multiplier M+3, by way of performing convolution of 
the spectral components SB, where M is an optional integer of from 1 to 
25. It is noted that the above values are only illustrative examples of 
the multiplication coefficients of the multipliers of the convolution 
filter circuit 902. 
An output of the convolution filter 902 is fed to a subtractive unit 905. 
The subtractive unit 905 is used for finding a level Q corresponding to 
the allowable noise level in the convolved area. Meanwhile, the level Q 
corresponding to the allowable noise level is such a level which becomes 
the allowable noise level for each critical band as a result of the 
deconvolution as later explained. The subtractive unit 905 is supplied 
with a allowance function (a function expressing the masking level) in 
order to find the level .alpha.. The level.gtoreq.is controlled by 
increasing or decreasing the allowance function. The allowance function is 
supplied from a (n-ai) function generating circuit 904 now to be 
explained. 
If the number of the critical bands as counted from the lower range is 
denoted as i, the level .alpha., corresponding to the allowable noise 
level, may be found by the following equation (7): 
EQU .alpha.=S-(n-ai) (7) 
where n and .alpha. are constants, with a &gt;0, and S is intensity of the 
convolved Bark spectrum. In the equation (7), (n-ai) is the allowance 
function. In the present embodiment, n=38 and a=1, for which there is no 
sound quality deterioration and satisfactory encoding may be achieved. 
The level a is found in this manner and routed to the subtractive unit 907 
via a synthesizing circuit 908. The subtractive unit 907 is fed with an 
output of the band-based energy calculating circuit 901, that is the 
above-mentioned spectrum SB, via a delay circuit 908. The subtractive unit 
907 subtracts the masking spectrum from the spectrum SB. Thus the portion 
of the spectrum SB below the level of the masking spectrum MS is masked. 
An output of the subtractive unit 907 is taken out via an allowable nose 
correction circuit 911 and an output terminal so as to be routed to a ROM, 
not shown, in which the information concerning the allocated bit number is 
pre-stored. The ROM is responsive to an output obtained from the 
subtractive unit 907 via the allowable noise correction circuit 911 (the 
level of a difference between the band-based energy and an output of the 
noise level setting means) to output the information concerning the number 
of allocated bits for each band. This information concerning the number of 
allocated bits for each band is routed to the adaptive bit allocation and 
encoding circuits 210, 211 and 212 so that the frequency-domain spectral 
data from the MDCT circuits 203, 204 and 205 will be quantized with the 
numbers of bits allocated for each band. 
In sum, the adaptive bit allocation and encoding circuits 210, 211 and 212 
quantize the band-based spectral data with the number of bits allocated 
responsive to the masking quantity and the level of the difference between 
the energy of the critical bands and the sub-bands and the output of the 
noise level setting means. The delay circuit 908 is employed for delaying 
the spectrum SB from the band-based energy calculating circuit 901 taking 
into account the delay caused in circuits upstream of the synthesizing 
circuit 906. 
Meanwhile, during synthesis by the synthesizing circuit 906, data 
indicating a so-called minimum audibility curve RC from a minimum 
audibility curve generating circuit 909, which stands for psychoacoustic 
characteristics of the human hearing System, as shown in FIG. 12, may be 
combined with the masking spectrum MS. If the absolute noise level is 
below the minimum audibility curve, the noise is not heard. The minimum 
audibility curve becomes different depending on e.g., the playback volume, 
even for the same coding. However, since there is no marked difference in 
the manner of music entering e.g., a 16-bit dynamic range, it may be 
presumed that, if the quantization noise in the frequency range in the 
vicinity of 4 kHz, which is heard most readily by the ear, is not heard, 
no quantization noise in any other frequency range which is below the 
minimum audibility curve is heard. Thus, if it is assumed that the device 
is used so that the noise in the vicinity of 4 kHz of the word length of 
the system is not heard, and the allowable noise level is obtained by 
synthesizing both the minimum audibility curve RC and the masking spectrum 
MS, the allowable noise level can be set so as to be up to the portion 
shown shaded in FIG. 12. In the present embodiment, the 4 kHz level of the 
minimum audibility curve is matched to the lowest level corresponding to 
e.g. 20 bits. Meanwhile, the signal spectrum SS is also shown in FIG. 12. 
The allowable noise correction circuit 911 also corrects the allowable 
noise level in the output of the subtractive unit 907 based upon the 
information of e.g., an equi-loudness curve supplied from a correction 
information outputting circuit 910. The equi-loudness curve is a 
characteristic curve concerning the psychoacoustic characteristics of the 
human auditory system. The equi-loudness curve, also termed an 
equi-loudness sensitivity curve, is obtained on connecting points of the 
sound pressure of tones at various frequencies heard with the same 
loudness as the pure tone at 1 kHz. The equi-loudness curve delineates a 
curve substantially equal to a minimum audibility curve RC shown in FIG. 
12. In the equi-loudness curve, the sound near 4 kHz is heard with the 
same loudness as the sound at 1 kHz despite the sound pressure being lower 
by 8 to 10 dB than that at 1 kHz. Conversely, the sound near 50 Hz cannot 
be heard with the same loudness as the 1 kHz sound unless the sound 
pressure is higher by about 15 dB than that at 1 kHz. Thus it is seen that 
the noise exceeding the level of the minimum audibility curve (allowable 
noise level) preferably has frequency characteristics represented by a 
curve conforming to the equi-loudness curve. From this it follows that 
correction of the allowable noise level in consideration of the 
equi-loudness curve is suited to the psychoacoustic characteristics of the 
human auditory system. 
It is possible for the correction information outputting circuit 910 to 
correct the allowable noise level based upon the information between the 
detection output of the quantization output data quantity at the adaptive 
bit allocation and encoding circuits 210 to 212 and the target bit rate of 
the ultimate encoded data. Specifically, there may be instances wherein 
the total number of bits resulting from previous transient adaptive bit 
allocation to the totality of the unit bit allocation blocks has an error 
with respect to a constant bit rate (target value) determined by the bit 
rate of the ultimate encoded output data. In these instances, bit 
allocation is again carried out for reducing the error to zero. That is, 
if the total number of allocated bits is smaller than the target value, 
the number of bits corresponding to the difference is additively allocated 
to the unit blocks. Conversely, if the total number of allocated bits is 
larger than the target value, the number of bits corresponding to the 
difference is subtractively allocated to the unit blocks. 
To this end, the correction information outputting circuit 910 detects an 
error of the total number of bits from the target value, and outputs 
correction data for correcting the number of bit allocation responsive to 
the error data. If the error data indicates shortage in he number of bits, 
a larger number of bits has been used per unit block so that the data 
volume is in excess of the target value. Conversely, if the error data 
indicates excess in the number of bits, a smaller number of bits per unit 
block suffices, so that the data volume is smaller than the target value. 
Thus the correction information outputting circuit 910 outputs data of the 
above correction value for correcting the allowable noise level in the 
output of the subtractive unit 907 based e.g., on the information data of 
the equi-loudness curve. The allowable noise level from the subtractive 
unit 907 is corrected by the above correction value being supplied to the 
allowable noise correction circuit 911. In the above-described system, the 
data resulting from processing the orthogonal transform output spectrum by 
the subsidiary information, as the main information, and the scaling 
factor indicating the state of the block floating and the word length data 
indicating the word length, as the subsidiary information, are obtained, 
and transmitted from the encoder to the decoder. 
FIG. 13 shows the ATC decoder 73 shown in FIG. 1, that is an illustrative 
constitution of a decoding circuit for re-decoding signals from the 
above-described high efficiency encoding. The band-based quantized MDCT 
coefficients, that is data equivalent to output signals at the output 
terminals 213, 214 and 215 in FIG. 2, are supplied at input terminals 300, 
302 and 304 to decoding circuits 306, 307 and 308, while the information 
on the orthogonal transform block size in use, that is data equivalent to 
output signals at output terminals 216, 217 and 218 in FIG. 2, are 
supplied to decoding circuits 306, 307 and 308. The decoding circuits 306, 
307 and 308 cancel the bit allocation using the adaptive bit allocation 
data. The signal on the frequency axis are then transformed by IMDCT 
circuits 309, 310 and 311 into signals on the time axis. The time-domain 
signals of the partial ranges are decoded into a full range signal by IMQF 
circuits 312 and 313 so as to be outputted via an output terminal 314 to a 
D/A converter 74 shown in FIG. 1. 
The present invention is not limited to the above-described embodiments. 
For example, the recording/reproducing medium need not be integrated to 
the other recording/reproducing medium and may be interconnected via a 
data transfer network. The present invention may also be adapted to a 
signal processing apparatus for processing digital speech signals or 
digital video signals, in addition to the audio PCM signals. The 
above-mentioned synthesis of the minimum audibility curve may also be 
omitted, in which case the minimum audibility curve generating circuit 909 
and the synthesis circuit 908 in FIG. 9 may be omitted and the output of 
the subtractive unit 905 may be directly supplied to the subtractive unit 
907. 
Also a wide variety of bit allocation techniques may be employed. In the 
simplest case, the fixed bit allocation may be employed. In addition, 
simple bit allocation based on the band-based energy or the combination of 
fixed bit allocation and variable bit allocation may be employed. 
From the foregoing it is seen that the present invention provides an 
arrangement in which, if, when the time length of the orthogonal transform 
block and the window shape are changed with respect to acute amplitude 
changes in the input signal, some correlation is found to exist between 
the channels, the same time length may be employed for the orthogonal 
transform blocks of the respective channels for suppressing the difference 
in the sound quality between channels for improving a fixed sound image 
position feeling and satisfactory sound quality. This realizes a more 
satisfactory sound quality for the same bit rate. In addition, the same 
sound quality may be obtained with a lower bit rate. 
That is, the present invention provides means for determining time length 
of the processing blocks which is psychoacoustically desirable for 
compacting temporally fluctuated information signals, thereby enabling 
high efficiency compaction and expansion with a higher sound quality.