Gain control circuit for suppressing Larsen effect in a loudspeaker telephone

A gain control circuit (10) of an amplifier (3) for listening via a loudspeaker (4) in an apparatus, more specifically a telephone set, which includes a microphone (1), in order to suppress the Larsen effect. The gain control circuit comprises first control means (100) which control a first gain reduction of the amplifier (3) when the amplitude of the microphone signal (S) exceeds a first predetermined threshold determined by a first reference voltage VR1, in dependence on a first time constant (RD1+RD2.C). The gain control circuit also includes second control means (200) which start operating only after the said first time constant, and provide a supplementary reduction of the gain depending on a second time constant (RD2.C) which is shorter than the first time constant, as long as the amplitude of the microphone signal (S) exceeds a second threshold, which is lower than the first threshold, determined by a second reference voltage VR2.

BACKGROUND OF THE INVENTION 
The present invention relates to a gain control circuit of an amplifier for 
listening via a loudspeaker in an apparatus further including a microphone 
and its associated amplifier, more specifically with a view to suppressing 
the Larsen effect, the circuit essentially consisting of a negative 
feedback loop reducing the gain of the amplifier for listening via a 
loudspeaker when the signal coming from the microphone exceeds a 
predetermined threshold. 
The invention also relates to the use of this gain control circuit in a 
loudspeaking telephone set. It is a known practice that in all systems 
comprising both a loudspeaker, fed by its amplifier, and a microphone, 
whose signal is applied more or less directly to the amplifier, there is a 
risk of oscillation caused by the Larsen effect due to the acoustic 
coupling between the loudspeaker and the microphone. This is the case, for 
example, in a loudspeaking telephone set, to which case the invention 
refers more particularly, but not exclusively. 
In a loudspeaking telephone set it is necessary to provide means for 
suppressing the state of oscillation due to the Larsen effect and it is 
highly desirable that this suppression occurs automatically by a 
considerable reduction of the gain of the loudspeaker amplifier, that is 
to say without the user of the telephone set having to carry out any 
operation. 
Various technical solutions have already been proposed. Generally speaking, 
it can be stated that these solutions consist in providing on the one hand 
a circuit for detecting the state of oscillation due to the Larsen effect, 
supplying at the output a control signal which can be of the "all or 
nothing" type, or the proportional type, and on the other hand a so-called 
control circuit which influences the variable gain preamplifier stage and 
causes a reduction, or else a total cancelling of the gain when the 
circuit for detecting the Larsen effect has established that a state of 
oscillation exists. 
An embodiment of a loudspeaking telephone set and gain control is 
represented in the French Patent Application FR-A-No. 2 537 810. 
Essentially, however, such an arrangement is still subject to relaxation 
because, after an oscillation, a first gain reduction can make this 
oscillation disappear, whereas a moment later the gain control will 
increase again so that a second oscillation occurs, etc. 
It will be readily understood that there is no simple solution to realize 
the desired function, and the more so as the circuit for detecting the 
Larsen effect must be very selective in distinguishing an oscillation 
signal from a high amplitude but temporary audio signal such as an ambient 
noise or an impact on the microphone. 
SUMMARY OF THE INVENTION 
An object of the invention is to provide a suitable gain control circuit of 
the listening amplifier connected in the loudspeaker channel, which avoids 
the relaxation and which could use circuits that can essentially be 
integrated. 
According to the invention, this object is achieved by means of a control 
circuit in accordance with the opening paragraph, characterized in that 
the negative feedback loop comprises a first control means for controlling 
a gain reduction of the listening amplifier in the loudspeaker channel 
with a first time constant, and the said predetermined threshold being a 
first threshold, also includes second control means for comparing the 
signal coming from the microphone to a second threshold, lower than the 
said first threshold. The second control means control a gain reduction 
with a second time constant lower than the first one, from the moment when 
the gain is reduced to below a reference gain value by the said first 
control means, whereas they remain inactive when the gain remains higher 
than the said reference value. 
When there is no Larsen oscillation, the said first control means are 
generally sufficient to suitably attenuate the amplification of the 
listening signal in the loudspeaker channel, if this is necessary, and to 
ensure the adjustment of the amplification in the usual manner. 
The whole acoustic signal which reaches the microphone at a high level but 
for a short period of time, that is to say a time period which is shorter 
than the first time constant, does not trigger the second control means, 
as opposed to the instability caused by the Larsen effect which produces, 
by acoustic coupling, a high microphone signal and generally for a 
considerable length of time depending on the conditions of the acoustic 
coupling between the loudspeaker and the microphone. The moment when the 
first control means has reduced the gain to below the predetermined value 
taken as a reference, which occurs after the first time constant, the 
second control means come into operation having an adjusted time constant 
which is distinctly lower than the time constant of the first control 
means, and the gain is reduced rapidly to the value required so that the 
adjustment is effected by comparison to the second threshold. As the 
second threshold is chosen to have a lower value than the first threshold, 
relaxation will only occur and the circuit will only return to the normal 
gain conditions when the signal from the microphone becomes lower than the 
second threshold, or in practice, when the cause of the oscillation has 
disappeared. 
In an embodiment which is advantageous in its simplicity, the gain control 
circuit is characterized in that the gain of the listening amplifier 
coupled to the loudspeaker is controlled by the charging voltage of a 
capacitor permanently charged through a current source, and in that the 
first and second control means have outputs which cause this capacitor to 
be discharged via a respective first and second discharge resistor. The 
first discharge resistor has a value which is higher than that of the 
second discharge resistor. 
In addition, in this embodiment the control circuit is characterized in 
that the capacitor is charged from the most positive potential in the 
circuit, in that the said first and the second control means discharge the 
capacitor by their output terminal which is of the NPN transistor type (or 
N-channel MOS), with an open collector (or open drain), and in that the 
first discharge resistor of the capacitor is arranged in series with the 
second discharge resistor. The transistors of this type have the advantage 
of being easy to integrate. 
The use of outputs of the open collector type makes it possible to couple 
the outputs of the first and second control means across their respective 
discharge resistor because the state of high impedance of the output of 
one of the control means has no effect on the outupt of the other control 
means. 
According to the invention, one variant of this embodiment dispenses with 
the use of discharge resistors when the first and second control means 
have their outputs operate at a discharge current which is confined to a 
fixed value. The output current of the first means is then fixed at a 
lower value than the output current of the second control means, and the 
outputs concerned are connected in parallel to the capacitor. 
A particular embodiment of the invention is characterized in that the first 
control means comprise a first comparator whose positive input receives 
the signal coming from the microphone after amplification, whose negative 
input is brought to a first reference voltage which is lower than the 
continuous level of the signal coming from the microphone, and defines 
said first threshold with respect to this level, and whose output 
constitutes the output of the said first control means, in that the second 
control means comprise a second, a third and a fourth comparator, all of 
them having an output of the NPN transistor type having an open collector, 
in that the second comparator has its negative input, which receives the 
signal coming from the microphone after amplification, and its positive 
input, which is brought to a second reference voltage slightly higher than 
the first reference voltage but lower than the continuous level of the 
signal coming from the microphone, and which defines the said second 
threshold with respect to this level, in that the output of the second 
comparator is connected on the one hand to the intermediate point of a 
first impedance bridge which is connected between the end terminals of the 
circuit, and on the other hand to the negative input of the third 
comparator, in that the fourth comparator receives on its positive input a 
voltage substantially equal to the charging voltage of the capacitor and 
has its negative input brought to a third reference voltage level 
corresponding to the voltage offered by the capacitor for the said 
reference gain value, in that the output of the fourth comparator is 
connected to the foot of a second impedance bridge whose head is connected 
to the most positive voltage supply and whose intermediate point is 
connected to the positive input of the third comparator and makes a fourth 
reference voltage apppear, when the output of the fourth comparator is in 
a low state, which voltage is adjusted to a value which is lower than the 
voltage of the intermediate point of the first impedance bridge when the 
output of the second comparator is in the high-impedance state, and in 
that the output of the third comparator constitutes the output of the said 
second control means. 
With the exception of the capacitor whose voltage controls the gain of the 
amplifier for listening via a loudspeaker, the components constituting the 
control circuit can be integrated, which leads to a particularly 
cost-effective circuit.

DESCRIPTION OF THE PREFERRED EMBODIMENTS 
FIG. 1 shows the general circuit diagram of an apparatus compising a 
microphone 1, an amplifier circuit 2 and a signal processing circuit 3, 
and a loudspeaker 4. In the particular case of a telephone set the block 
denoted 2 comprises a preamplifier 5 for the microphone signal, a 
separator module 6, also called a hybrid junction, connected to the 
telephone line L and a listening preamplifier 7 for the receiver 8 of the 
combination. The block denoted 3 ensures the amplification of the 
listening signal in order to feed the loudspeaker 4, and comprises a 
preamplifier stage 3a having a controllable gain and a power amplifier 
stage 3b having a fixed gain. 
Depending on the circumstances, an acoustic coupling can take place between 
the loudspeaker 4 and the microphone 1, which is capable of bringing about 
an instability known by the name of the Larsen effect. In order to 
suppress this undesirable effect or attenuate the consequence thereof 
considerably, there is provided a gain control circuit 10 of the listening 
amplifier 3a contained in the block 3. The gain control circuit 10 roughly 
behaves as a negative feedback loop. Its input signal S can be taken off 
at any point in the amplifier circuit between the microphone 1 and the 
loudspeaker 4, but it should preferably be taken off at the microphone 1, 
more specifically in the case of a telephone set. According to the 
invention, the signal S, after a possible amplification, is applied to the 
input of the first control means 100 which effect a comparison of the 
signal S to a first reference voltage VR1. 
The gain control circuit 10 also includes a capacitor C having one terminal 
connected to grund and the other terminal charged from the positive 
potential V+ through a charging resistor RC. The charging voltage VC of 
this capacitor C can be limited, if desired, by a voltage limiter 11 
constituted, for example, by a specific number of semiconductor junctions 
in the forward direction, a situation which can exist by providing the 
preamplifier stage 3a at the gain control input. The voltage VC serves to 
control the gain of the amplifier 3 and can be converted into the control 
current by means of a resistor R as shown in the Figure. The capacitor C 
can be discharged by the output of the first control means 100 through a 
first discharging resistor RD1, and a second discharging resistor RD2 
connected in series. The quiescent voltage of the signal S determines, 
with respect to the first reference voltage VR1, a first threshold which, 
when not exceeded, maintains the output of the first control means 100 at 
the high-impedance state, while the output of these control means is of 
the open collector type. When the first threshold is exceeded, the output 
of the first control means 100 changes to the low-impedance state and 
causes capacitor C to be discharged through the discharging resistors RD1 
and RD2 in dependence on a first time constant which is equal to 
(RD1+RD2). C. 
The signal S is also applied to the input of the second control means 200, 
and this signal is compared to a second reference voltage VR2. The output 
of second control means 200 is connected to the intermediate point between 
the discharging resistors RD1 and RD2. The second reference voltage VR2 is 
chosen such that, together with the quiescent voltage of the signal S, it 
forms a second threshold which is lower than the said first threshold. As 
will be described in greater detail hereinbelow, the second control means 
200 are also controlled from the charging voltage VC of the capacitor C, 
and the output of the second control means is only activated when the 
voltage VC has dropped below one third of the reference voltage VR3 
corresponding to the voltage carried by the capacitor C for a reduced gain 
value which serves as a reference. Thus, the second control means 200 only 
produces a gain reduction of the power amplifier from the moment when the 
first control means have already reduced the gain to below the said 
reference value. In that case, the output of the second control means 200, 
which is also of the open collector type, causes capacitor C to be 
discharged through the discharging resistor RD2, preferably chosen to have 
a value which is much lower than that of the first discharging resistor 
RD1, and thus in dependence on a second time constant which is much 
shorter than the first time constant of the first control means 100. If 
the signal S of the microphone maintains a relatively high amplitude, that 
is to say higher than the second threshold determined by the second 
reference voltage VR2, the gain of the amplifier for listening 3 will be 
controlled by the second control means 200. 
In contradistinction thereto, when the amplitude of the signal S again 
drops below the second threshold, and thus even further below the first 
threshold, the two outputs of the first and second control means are in 
the high-impedance state and consequently, the capacitor C is recharged 
through the resistor RC and the gain of the amplifier for listening 3 will 
return to a high nominal value. 
It has been indicated above that different thresholds for the operation of 
the control means 100 and 200 are obtained from the reference voltages VR 
1 to VR 3. However, it will be evident that an equivalent solution can be 
obtained from a variant of an embodiment (not shown), operating on a 
current and not on a voltage, in which the thresholds required for the 
operation of the control means 100 and 200 are defined by currents of a 
preset value. 
Analogously, a further variant of the embodiment (not shown) relates to the 
discharging resistors RD 1 and RD 2 which can be suppressed when the 
outputs of the control means 100 and 200 operate with a fixed current 
while being in the conducting state. 
According to this variant, a value has been chosen for the output current 
of the first control means 100 which is preferably much lower than the 
current value of the output of the second control means. 
FIG. 2 a more detailed diagram of an embodiment the control circuit 10 of 
FIG. 1. 
The two terminals of the microphone are connected to two input terminals 13 
and 14 of the control circuit 10. The continuous reference of the 
microphone signal is first eliminated by a pi-filter constituted by the 
series arrangement of the capacitor 15 the resistor 16 and the capacitor 
17. The microphone signal thus filtered is applied to the inputs of a 
differential amplifier 20 via the protective resistors 18 and 19. 
Between the positive supply voltage V+ and the reference voltage (ground) 
is inserted a chain of four resistors 21, 22, 23, 24, which determine a 
first reference voltage VR1 at the junction between the resistors 24 and 
23, a second reference voltage VR2 at the junction between the resistors 
23 and 22 and a main reference voltage VRO at the junction between the 
resistors 22 and 21. The main reference voltage VRO is applied to the 
positive input of the differential amplifier 20 through a protective 
resistor 25 and the negative input of this differential amplifier is 
looped back to its output via a negative feedback resistor 26. 
Finally, the signal which is denoted S' in the Figure, and which is the 
microphone signal amplified by the differential amplifier 20, has a DC 
level which is substantially equal to the main reference voltage VRO. By 
way of a practical example, the gain of the differential amplifier 20 is 
set at around 50, the reference voltages VR0, VR1 and VR2 are set at 
around half the voltage V+, while the reference voltage VR1 is 
approximately 12 millivolts lower than the main reference voltage VR0, 
which difference constitutes the said first threshold, and the second 
reference voltage VR2 is 1.5 millivolts lower than the main reference 
voltage VR0, and constitutes the said second threshold which is eight 
times lower than the first threshold. 
The signal S' coming from the microphone, after amplification, is applied 
to the positive input of a first comparator 101 having an open collector 
output and whose negative input receives the first reference voltage VR1. 
The signal S' is also applied to the negative input of a second comparator 
202 having an open collector, while a positive input thereof receives the 
second reference voltage VR2. The output of the comparator 202 is 
connected to the intermediate point A between a divider constituted by the 
resistor 28, further connected to the positive voltage V+, and the 
resistor 29 connected to ground. The values of the resistors 28 and 29 are 
chosen such that the voltage VA of the point A is slightly higher than 
half the positive voltage V+. A third comparator 203 having an open 
collector output has its negative input connected to the point A. A fourth 
comparator 204 having an open collector output has its negative input 
connected to a third reference voltage VR3 carried by the intermediate 
point B of a resistor divider 30 and 31, which respective resistors are 
connected to the positive voltage V+ and ground. The third reference 
voltage VR3 corresponds to the voltage which the capacitor C has for a 
certain reference gain value below which the second control means have to 
start operating, as will be explained hereinafter. According to the 
explanation given with respect to FIG. 1, the capacitor C is charged 
permanently from the positive voltage supply V+ through the charging 
resistor RC, and is optionally discharged, depending on the circumstances, 
via the outputs of the first comparator 101 and the third comparator 203 
through the discharge resistors RD1 and RD2, respectively, the series 
arrangement of which is connected to the junction point between charging 
resistor RC and the capacitor C. The output of the third comparator 203 
which, when in a high-impedance state, carries a voltage which is 
substantially equal to the charging voltage of the capacitor C, is 
connected via a protective resistor 32 to the positive input of the fourth 
comparator 204. Finally, the output of the fourth comparator 204 is 
connected to the foot of a bridge, constituted by the resistors 34 and 35, 
whose top is connected to the positive voltage supply V+and whose 
intermediate point F makes a fourth reference voltage VR4 appear which, 
when the output of the fourth comparator 204 is in the low-impedance 
state, is adjusted to a value which is lower than the voltage VA of point 
A when the output of the second comparator 202 is in the high-impedance 
state. The intermediate point F of the resistors 34 and 35 is connected to 
the positive input of the third comparator 203. 
With reference to FIG. 3, the operation of the gain control circuit 10 of 
FIG. 2 will now be described. FIG. 3 shows on the one hand the signal S' 
coming from the microphone after amplification, in comparison with the 
first reference voltage VR1 and the second reference voltage VR2, and on 
the other hand the signals: 
VD at the output of the first comparator 101 
VA, the voltage at point A, in comparison with the voltage VR4 
VC, the charging voltage of capacitor C, in comparison with the voltage 
VR3, and 
VE, the voltage at the output of the third comparator 203. 
It should be noted with respect to this Figure that, for convenience sake, 
the signals shown are on a scale totally arbitrary and non-proportioned. A 
first period of time t1 is shown during which the signal S' has a 
momentarily high amplitude which is a characteristic feature of the speech 
signal. It will be remembered that the speech signal can be characterized 
by a peak factor which is substantially in accordance with the following 
law: 
EQU .theta.=(.pi./Ln2) exp(-x.sqroot.3/2) 
in which x represents the ratio between the absolute value of a given 
threshold and the effective voltage of the signal, and .theta. represents 
the sum of the periods of time during which this threshold is exceeded 
relative to the total time of observation. The control circuit 10 utilizes 
this characteristic feature of the speech signal. 
During this period of time t1, the first comparator 101 has an output which 
changes to the low state when the first reference voltage VR1 is exceeded 
and progressively effects through the discharge resistor RD1 having a 
relatively high value, the charge volgage VC of the capacitor C. During 
the same period of time the second reference voltage VR2 is also exceeded 
so that the second comparator 202 provides an output signal VA which, in a 
condition of rest is in the low state, and comprises square waves for the 
corresponding alternations of the signals S'. However, the fourth 
comparator 204, as it has not changed-over the voltage VR4 at the point F, 
is maintained at the highest potential V+. Thus, the third comparator 203 
does not change-over during this period of time t1. 
FIG. 3 also shows another period of time, t2, during which a condition of 
acoustic coupling occurs such that the system oscillates, whereas the 
conditions of oscillation have disappeared for the next period of time t3. 
During a first time interval corresponding to the first discharging time 
constant, due to the operation of the comparator 101 and the first 
discharging resistor RD1, the gain of the amplifier for listening 3 is 
progressively reduced under the influence of the comparator 101, until the 
charging voltage of the capacitor VC drops to below the third reference 
voltage VR3. At this instant, the fourth comparator 204 changes-over, and 
at point F a reference voltage VR4 appears which is lower than the top of 
the positive square waves of the signal VA at the output of comparator 
202. From that instant onwards the third comparator 203 is able to 
change-over in response to the positive square waves of the output of the 
second comparator 202 and in its low-state output leads to an accelerated 
discharging of the capacitor C in response to the discharging resistor RD2 
chosen to have a distinctly lower value than resistor RD1. The gain then 
changes rapidly to a very low value in response to the voltage control of 
the capacitor C, and the circuit then adjusts the value of this gain as a 
function of the conditions and the amplitude of the signal S' as long as 
the latter has an amplitude such that the second reference voltage VR2 
remains exceeded by the negative alternations of this signal. 
During the third period of time t3, during which the acoustic coupling 
conditions have been changed, the signal S' is supposed to be reduced to a 
low value such that neither of the two thresholds is exceeded. The outupts 
of the first comparator 101 and the third comparator 203 are now in the 
high-impedance state so that the capacitor C is recharged through charging 
resistor RC. When the charging voltage VC of capacitor C exceeds the level 
of the third reference voltage VR3 at point B, the output of the fourth 
comparator 204 is again switched to the high-impedance state so that the 
voltage at point F is again increased to the value of the positive voltage 
V+. The unit comprising comparators 202, 203 and 204 returns to its stable 
operating point when there is no high amplitude of the signal S'. 
Finally, it should be observed that the comparator 101, when playing the 
role of first control means, operates according to a first time constant 
which is mainly determined by the discharge resistor RD1 as long as the 
signal S' maintains an amplitude value which can be adjusted by the first 
control means without the gain being reduced to below the reference gain 
corresponding to the third reference voltage VR3. 
When the microphone signal has a high amplitude (the two thresholds being 
exceeded), the capacitor C is discharged according to he said first time 
constant after which the unit comprising the comparators 202, 203 and 204 
starts operating with a time constant caused by the discharge resistor RD2 
which can be chosen, for example, to be 20 times lower than the discharge 
resistor RD1, and this unit of comparators constituting the second control 
means, then adjusts the discharge of the capacitor and thus the gain of 
the amplifier 3 as long as the signal S' maintains an amplitude which 
exceeds the second threshold corresponding to the second reference voltage 
VR2. 
The portion of the circuit 10 of FIG. 2, which is shown inside the broken 
line framework 36, can easily be integrated in the form of a monolithic 
circuit. In order to facilitate this operation, specific resistors of a 
high value can advantageously be exchanged for components having 
equivalent functions. Thus, the first impedance divider, described as 
being constituted by the resistors 28 and 29, can, according to a variant 
which is not shown in the Figure, be constituted by the series arrangement 
of a current source connected to the voltage supply V+ and of a non-linear 
impedance connected to ground, this non-linear impedance being formed by a 
plurality of semiconductor junctions connected in series, which normally 
determine the voltage VA of point A. This also holds for the second 
impedance divider described hereinbefore as formed by the resistors 34 and 
35 and also for the divider formed by the resistors 30 and 31. In a 
similar manner the resistor 21 can be exchanged for a current source and 
the resistor 24 for a series of semiconductor junctions connected in the 
forward direction, while the resistors 22 and 23, having a lower value, 
can be retained. 
Finally, as experts who effect the integration of electronic functions by 
means of monolithic circuits know, variants can be proposed in which the 
functions that have been described in terms of voltage comparisons are 
transformed into equivalent functions using a comparison of currents. It 
will be evident that such variants remain within the scope of the 
invention claimed hereinafter.