Amplifier circuit with bipolar transistors

The invention relates to a circuit comprising an amplifier with bipolar transistors, which amplifier is fed by a signal source with a complex internal resistance and which has a signal negative feedback path connecting the amplifier output to the signal source. For reducing the noise, especially the input noise current which is particularly predominant in circuits of this kind which have a sufficiently large source impedance, a reactance network is arranged between the amplifier and the signal source. The network, in connection with the signal source and the negative feedback path, constitutes a filter which is parallel to the amplifier input and which has at least a series resonance frequency within the signal frequency range.

The invention relates to a circuit comprising an amplifier with bipolar 
transistors. The amplifier is fed by a signal source with a complex 
internal resistance and which has a signal negative feedback path 
connecting the amplifier output to the signal source. Circuits of this 
type are essentially known. 
SUMMARY OF THE INVENTION 
It is an object of the present invention to realize such a circuit in such 
a way that the noise is essentially reduced in the case of where a 
broadband signal source having a complex source impedance is connected to 
the amplifier. According to the invention this object is realized in that 
a reactance network is arranged between the amplifier and the signal 
source, which network, in connection with the signal source and the 
negative feedback path, constitutes a filter which is parallel to the 
amplifier input and which has at least a series resonance frequency within 
the signal frequency range. 
In a transistor amplifier the noise of the input transistor generally has 
the greatest influence on the overall noise of the amplifier. Current 
noise is generally predominant in bipolar transistors. The influence of 
this current noise on the output signal is reduced by the series 
resonance. The circuit arrangement according to the invention can 
therefore be used advantageously as a broadband input stage with low noise 
for sources having an inductive or a capacitive generator impedance. 
The invention will now be described in greater detail by way of example 
with reference to the accompanying drawings.

DESCRIPTION OF THE PREFERRED EMBODIMENT 
FIG. 1 shows a diagram of a noisy amplifier consisting of a noise-free 
amplifier 1 of a noise voltage source u operating in series with the 
(inverting) input of the amplifer and a noise current source i connected 
to this input. The voltage and currents supplied by the noise sources u 
and i, respectively, are generally statistically independent, uncorrelated 
quantities. The amplifier 1, whose structure is not further shown, 
comprises bipolar transistors in which the influence of current noise is 
predominant if the source impedance is sufficiently large. 
The amplifier output from which the output voltage U can be derived is 
connected to the amplifier input through a DC negative feedback path. The 
DC negative feedback path comprises an impedance 2, generally a high-ohmic 
resistor, whilst the noise components thereby produced are assumed to be 
contained in the noise current source i. 
A further negative feedback branch is provided which connects the output 
through an impedance Za to a signal generator having a complex generator 
resistance. The generator is shown in FIG. 1 by means of a signal current 
source supplying a signal current I and a complex resistor Zg connected 
parallel thereto. The ohmic component of this generator resistance should 
be possibly small in the signal frequency range as compared with its 
capacitive or inductive reactive component in order that the influence of 
the noise produced thereby remains small. 
The junction point of the signal source I, Zg and the negative feedback 
impedance Za is connected to the input of the amplifier 1 by a reactance 
network 3. If this reactance network were absent, the amplifier would have 
the same transmission behaviour for the signal current I and the noise 
current i. The reactance network 3 decouples the useful signal source from 
the hoise signal source in such a way that the noise at the output is 
reduced without essentially influencing the signal transmission--and 
without affecting the stability--in the signal transmission range. For 
this purpose the reactance network 3 in connection with the negative 
feedback impedance Za is dimensioned in the case of a given source 
impedance in such a way that a series resonance results in the signal 
frequency range which reduces its influence on the amplifier output 
signal. Since the signal transmission remains essentially uninfluenced by 
the negative feedback through the impedance Za, an improvement of the 
signal-to-noise ratio is obtained. 
This principle will hereinafter be explained in greater detail with 
reference to two concrete embodiments shown in FIGS. 2 and 5 in which 
similar components have the same reference symbols. 
FIG. 2 shows a broadband input stage of a medium wave receiver with an 
inductive generator impedance in the form of a medium wave aerial. The 
generator impedance is shown in FIG. 2 as a substitution circuit 
comprising an inductance L of 0.5 mH and a resistor R of approximately 500 
k0hm arranged parallel thereto. The impedance Za is constituted by a 
capacitor Ca of 5 pF, whilst the reactance network 3 is constituted by a 
capacitor Cp arranged parallel to L and R and a capacitor Cs of 60 pF 
which is arranged between the generator and the amplifier input. The 
amplifier has an input resistance of 22 k0hm and an amplification of 38 dB 
at 1 MHz which decreases by 20 dB/decade in the medium wave range. 
For the noise current source i the arrangement according to FIG. 2 operates 
as a filter which has a transmission minimum (series resonance) at the 
resonance frequency of the inductance L with the sum of Ca, Cs and Cp, 
whilst the frequency for this minimum is lower as the capacitance of 
capacitor Cp is larger. 
A minimum also results for the voltage noise, however at the resonance of 
the inductance L with the sum of Cp and Ca (i.e. this minimum is 
independent of the capacitance of capacitor Cs). Accordingly, the minimum 
for the voltage noise is higher than the minimum for the current noise (at 
the same capacitance Cp). 
Due to the frequency dependence of the transmission of the voltage and 
current noise an overall noise whose minimum is between the minima for the 
current and voltage noise results at the output. This minimum can be 
brought to a desired frequency position in the transmission band by 
suitable choice of the capacitance of capacitor Cp. 
In contrast thereto, the signal transmission for the generator current I is 
substantially not influenced by the capacitance Cp due to the negative 
feedback via the capacitor Ca, and it is even not influenced if this 
capacitance is chosen to be such that together with the inductance L it 
constitutes a resonance in the transmission range. In this case the loop 
gain increases to such an extent that no resonance step-up can occur at 
the output of the amplifier. In the case of a constant generator current I 
the output voltage in the medium wave range, i.e. from 535 to 1605 kHz, 
decreases approximately linearly by approximately 10 dB. This decrease is 
caused by the negative feedback through the capacitor Ca increasing with 
an increasing frequency. 
In FIG. 4a the transmission factor for the noise current i (i.e. the 
voltage produced by the noise current at the output of the amplifier 1, 
divided by the noise current i) is shown on a logarithmic scale as a 
function of the frequency (likewise on a logarithmic scale), and this for 
different capacitances of Cp. The curve a results for Cp=2 pF, b results 
for Cp=30 pF and curve c results for Cp=60 pF. The minimum of the current 
noise is thus shifted to low frequencies with an increasing capacitance. 
Analogously, FIG. 4b shows the transmission factor for the voltage noise 
.mu. as a function of the frequency for the same capacitances. As already 
stated, the minimum value of the voltage noise is higher than the minimum 
value of the current noise, the minimum for Cp=2 pF (curve a) being 
outside the transmission range. 
FIG. 4c shows the overall noise, i.e. the equivalent input noise current 
parallel to the signal current which would produce the same noise at the 
output of an identical, but noise-free arrangement as the noise occurring 
in the present circuit arrangement, as a function of the frequency and 
also with Cp as a parameter. Relatively broad noise minima can be 
recognized which, however, only cover a part of the medium wave range. 
As already stated, the amplification of the useful signal is also dependent 
on the frequency in the circuit arrangement of FIG. 2. To eliminate this 
frequency dependence, the circuit arrangement according to the invention 
is succeeded by a compensation stage 4 having an inverse frequency 
characteristic so that the output signal of this stage is essentially 
frequency-independent--in the case of a constant aerial signal. 
This stage may comprise, for example, a bipolar transistor 41 whose emitter 
resistor 42 is shunted by the series arrangement of a capacitor 43 of 150 
pF and a resistor 45 of 500 Ohm for the medium wave range frequencies. A 
capacitive emitter impedance with an ohmic component results therefrom. 
The base of the transistor 41 is connected to the output of the amplifier 
1 and the emitter of the transistor 41 is connected to the capacitor Ca 
and the DC resistor 2. The resistor 2 may, however, alternatively be 
connected directly to the output of amplifier 1. The collector is 
connected to a supply voltage through a resistor 46 with which a parallel 
resonant circuit 47 tuned to approximately 2 MHz is arranged in parallel. 
The collector impedance thereby becomes inductive with an ohmic component. 
In this circuit arrangement a substantially frequency-independent output 
signal results at the collector which constitutes the output 44 of the 
circuit arrangement. 
FIG. 3 shows a medium wave receiver constructed in accordance with the 
principle shown in FIG. 2. The receiver section 6 comprises the 
compensation circuit 4, the part 5 of the circuit according to the 
invention shown in broken lines in FIG. 2 and further signal processing 
components. The series arrangement of a first capacitor Cp1 and a first 
switch S1 as well as the series arrangement of a second capacitor Cp2 and 
a second switch S2 are arranged parallel to the aerial input 7. The 
capacitor Cp3 arranged parallel to the aerial input represents the winding 
capacitance of the aerial. If Cp1 and Cp2 are each approximately 30 pF, 
either 2 pF, 32 pF or 62 pF are active at the aerial input, dependent on 
the position of the switches S1 and S2. 
The switches S1 and S2 are operated by a logic circuit 9 which is 
controlled by the running unit 8 of the receiver 6, which unit also fixes 
the tuning frequency of the receiver 6. The logic circuit 9 derives the 
switching signals for the switches S1 and S2 from the tuning data of the 
tuning unit 8, for example, the frequency division factors for a PLL 
circuit in the receiver section 6. The circuit 9 is constructed in such a 
way that the two switches are closed at low medium wave frequencies up to 
approximately the frequency f1 (compare FIG. 4c) so that the noise up to 
this frequency varies in accorance with curve c. Between the reception 
frequencies f1 and f2 one of the two switches S1, S2 is open and the other 
disclosed. A capacitance of approximately 32 pF then results at the input 
with a noise variation in accordance with curve b, which in this frequency 
range is lower than the noise according to curve c or a. 
Above the frequency f2 at which the curves a and b intersect each other, 
the two switches S1 and S2 are opened so that only the capacitance Cp3 of 
2 pF is active at the aerial input. The noise then varies in accordance 
with curve a in FIG. 4c. Due to this changeover over it is achieved that 
the capacitance combination producing the lowest noise is always active at 
the input. 
For the purpose of comparison FIG. 5c shows the broken line curve d for the 
overall noise which would result if the capacitor Cs were absent and if 
the switches S1 and S2 were switched in dependence upon the tuning 
frequency such that a noise minimum would result. The improvement realized 
by the invention can clearly be recognized. 
As already stated, the parallel resonance determined by the capacitor Cp in 
connection with the inductive impedance L of the generator cannot lead to 
resonance step-ups due to the negative feedback through Ca. Consequently, 
the currents through the switches S1 and S2 always remain relatively low 
in the circuit arrangement according to FIG. 3 so that their finite 
resistance (particularly if they are formed as semiconductor switches) 
leads to signal losses which are only inconsequential. Moreover, the 
change-over ensures that the parallel resonant circuit consisting of the 
inductance L and the capacitance Cp parallel thereto is always operated 
below its resonance frequency so that resonance effects are therefore also 
avoided. 
The reduction of the noise realized by means of the invention in the medium 
wave receiver is not as large as it would be in a receiver in which the 
aerial input would always be tuned to the reception frequency. Such a 
receiver would, however, either require a relatively expensive variable 
capacitor or a capacitance diode which requires relatively large tuning 
voltages. Moreover, an exact alignment between the oscillator and input 
circuits would be required in such a receiver. In contrast thereto, the 
invention can also be utilized--particularly as an integrated circuit--in 
receivers having a low supply voltage (for example 3 V). An alignment is 
not required. 
Although the circuit according to FIG. 2 has been explained in connection 
with a medium wave receiver, it may also be used in other cases as a 
broadband input stage if only the signal generator has an essentially 
inductive generator resistance as in, for example, the reproducing head of 
a magnetic tape apparatus. In this case the dimensioning of the reactance 
network Cs, Cp (and of the negative feedback capacitor Ca) should be 
adapted to the requirements imposed by such a circuit arrangement. 
FIG. 5 exemplifies the invention with reference to a broadband input stage 
for capacitive signal generators such as, for example, a capacitive 
aerial. In this case the signal source is shown by way of a substitution 
diagram comprising the series arrangement of a signal voltage generator 8 
and a capacitor 9. The reactance network is connected by a first 
inductance LS1 to the input of a first amplifier 1 (whose input noise 
sources are not separately shown). The output of this amplifier is 
connected by a capacitor Ca to the junction point of the generator 8, 9 
and the first inductance LS1. The inductance LS1 is dimensioned in such a 
way that a series resonance results in the signal transmission range which 
reduces the noise caused by the noise current source (not shown) at the 
output 11 of the circuit. 
A parallel resonance does not result in the circuit according to FIG. 5 so 
that the voltage noise is substantially not influenced. Since the current 
noise in an amplifier with bipolar transistors is predominant in the case 
of a sufficiently large generator impedance, this is not a consequential 
drawback. For this reason the circuit is frequency independent due to the 
capacitive negative feedback in connection with the capacitive generator. 
Similarly as in the circuit of FIG. 1 the noise is only influenced in a 
given frequency range around the resonance point. In order to suppress the 
noise in a broader frequency range in a receiver circuit, the reactance 
network in the circuit according to FIG. 5 should be rearranged in a 
similar way as in FIG. 3. For this purpose a second inductance LS2 is 
connected to the input of the first amplifier 1 and its other terminal is 
connected to the inverting input of a further amplifier 1' which has the 
same properties as the amplifier 1. The output of this amplifier is 
connected to the output of the amplifier 1. The power supply terminals of 
the amplifiers 1, 1' are connected through a change-over switch to a power 
supply source so that only one of the two amplifiers is operative. 
When tuning to a lower frequency range, the change-over switch 10 is 
connected to the power supply terminal of the amplifier 1', while the 
reactance network is given by the series circuit of the two inductances 
LS1 and LS2. This reduces the noise in the lower frequency range. When 
tuning to a higher frequency range, the changeover switch 10 is switched 
by a control signal s (which can be generated analogously as the switching 
signals for the switches S1, S2 in the circuit according to FIG. 3) in the 
switching position shown in FIG. 5. In this case only the inductance LS1 
is active. 
Also in this case the signal transmission is substantially not influenced 
by the inductances LS1 and LS2 due to the negative feedback from the 
output 11 through Ca to the signal source. 
The embodiment of FIG. 5 is not limited to receiver circuits having a 
capacitive aerial. It can be used in all those cases in which a generator 
having an essentially capacitive generator resistance is provided, for 
example, a capacitor microphone.