Five-stage neutral point clamped inverter

The multi-level DC/AC converter, comprising: an input (5,7) connectable to a direct voltage source (3), with a first connection (5) and a second connection (7) between which can be applied an input voltage (Vi); a half-bridge with a first controlled switch (21) and a second controlled switch (25) between which is positioned an output (U) of the converter; a first connecting branch (15) between the first controlled switch (21) and the first connection (5) and a second connecting branch (17) between the second controlled switch (25) and the second connection (7); a third controlled switch (59) associated to the first controlled switch (21), connectable in series to the first controlled switch to generate an output voltage exceeding a first limit value (Vi/2); a fourth controlled switch (61) associated to the second controlled switch (25), connectable in series to said second controlled switch to generate an output voltage below a second limit value (−Vi/2).

TECHNICAL FIELD

The present invention relates to improvements to DC/AC converters and more specifically to multi-level PWM converters.

STATE OF THE ART

In many industrial applications, it is required to convert a direct current into alternating current at a predetermined voltage. For this purpose, DC/AC converters or inverters are used. Typical examples of uses of these circuits are found in the field of renewable energies or of alternative energies. Photovoltaic panels, for example, represent sources of direct electrical energy, which is used to supply power to local loads in alternating current, or which is injected into the normal AC electrical energy distribution grid. Other applications in the field of alternative energies contemplate the use of different types of direct current sources, e.g. wind generators, fuel cells, etc.

In general, the direct voltage generated by a photovoltaic panel, or another alternative source, is relatively low and not constant, because of the change in parameters, such as irradiation conditions, which may oscillate slowly and regularly because of the apparent movement of the sun, or rapidly and unpredictably because of transiting clouds and/or of the change in other climate parameters, such as humidity.

On the other hand, the electrical load or the electrical grid to which the source is interfaced through the converter requires alternating voltage power supply with a fixed, well determined frequency and peak value, typically a frequency of 50 or 60 Hz and a peak voltage of 220 V (in the case of single-phase line).

It is therefore necessary to provide DC/AC converters that are able to convert the electrical energy supplied by the direct voltage source into electrical energy with alternating voltage, with a peak value that is typically greater than the value of the direct voltage supplied by the source.

For this purpose, circuits have been designed that comprise a step-up converter interposed between the source and a half-bridge or full-bridge converter.

These circuits have a first drawback due to the complexity and to the cost deriving from the step-up stage. Moreover, the wave form of the voltage output from the converter has a high harmonic content and hence requires a very heavy output filter, with consequent additional costs.

Other known circuits (DE-A-10020537) employ a full-bridge converter connected, through three connections, to a pair of direct current sources connected in series. A chopping module or a DC/DC converter in parallel to the second source enables to generate a second level of voltage connecting the bridge alternatively to only one of the two sources, or to the terminals of the two sources in series. The chopping module is in conduction or interdiction depending on the value of the grid voltage. In this way, a reduced harmonic content of the voltage at the ends of the output inductance is obtained. The electronic switches of the bridge (designed as MOSFET, IGBT or other suitable components) are subject to a high voltage difference when the chopping module or the DC/DC converter is active and they switch at high voltage, with consequent high losses and need to be dimensioned for high voltage ratings.

DE-A-102006010694 describes an additional converter specifically devised for use in combination with photovoltaic panels, which uses a half-bridge structure. In parallel to the electronic switches of the half-bridge are positioned two branches containing a DC/DC converter and an electronic switch controlled as a function of the grid voltage. With this arrangement, the converter can generate five output voltage levels. The arrangement is such that the switches controlled by the two branches of the half-bridge and the two switches of the two branches positioned in parallel to the branches of the half-bridge must be designed for high voltage values, in such a way as to support the voltage of the highest level, with consequent costs and low efficiency because of the high switching losses.

SUMMARY OF THE INVENTION

According to one aspect, the invention provides a DC/AC converter that completely or partially overcomes one or more of the drawbacks of known converters.

Essentially, the invention provides a DC/AC converter with a half-bridge able to be connected at its input to a direct voltage source, wherein to the two controlled switches of the half-bridge are associated respective auxiliary switches that can be driven in such a way as to be inactive, when the output voltage is contained within a range of values around zero, within said range of values the output voltage being generated by driving the two switches of the half-bridge at an appropriate switching frequency, with a PWM driving signal with variable duty cycle, to obtain the gradual increase or the gradual decrease of the output voltage. The two switches are driven with complementary signals, so that when one of the switches is closed the other one is open and vice versa. Vice versa, when an output voltage needs to be generated with a value outside this range of limit values, the two switches of the half-bridge are driven, as a function of the sign of the output voltage, in such a way that one of them is constantly in conduction and in series with one of the auxiliary switches which is vice versa driven with a PWM signal at an appropriate switching frequency and with a variable duty cycle. The sign of the output voltage determines which of the two switches of the half-bridge is kept in continuous conduction and which auxiliary switch is placed in series to it and driven by the PWM signal. Voltage regulators are associated to the auxiliary switches, so that the output voltage can gradually assume positive values above the upper limit value of the aforesaid range, or negative values gradually lower than the lower limit value of said range.

In practice, the portions of the positive and negative half-wave of the output voltage are generated using a series arrangement of switches: depending on the sign of the output voltage, one or the other of the switches of the half-bridge are used in conditions of continuous conduction (always closed) in series with the corresponding auxiliary switch driven with a chopping PWM signal. The voltage rate of the auxiliary switches is thus limited, with a series of advantages that shall be made more readily apparent below.

This concept can be expanded providing more than one auxiliary switch for each of the switches of the half-bridge with the possibility of arranging more than two switches in series depending on the value of the output voltage to be generated, thereby obtaining a converter with a higher number of voltage levels and hence with a reduced harmonic content of the output voltage.

The converter according to the invention enables to obtain a plurality of voltage levels, thereby allowing a reduction of the harmonic content of the output sinusoidal voltage. Moreover, circuit advantages are obtained in terms of reducing losses and reducing component costs.

In some embodiments, the invention provides a DC/AC converter comprising: an input connectable to a source of direct voltage; a half-bridge with a first controlled switch and a second controlled switch; connecting branches between the half-bridge and the connections to the direct voltage source; a third controlled switch associated to the first controlled switch of the half-bridge, connectable in series to said first controlled switch to generate an output voltage exceeding a first limit value; a fourth controlled switch associated to said second controlled switch, connectable in series to the second controlled switch to generate an output voltage below a second limit value.

In advantageous embodiments of the invention, the converter further comprises: a first voltage regulator positioned to regulate the voltage across a first capacitor; a second voltage regulator positioned to regulate the voltage across a second capacitor. The third controlled switch is connected between a first plate of the first capacitor and the first controlled switch of the half-bridge, whilst the fourth controlled switch is connected between a first plate of the second capacitor and the second controlled switch of the half-bridge.

In one embodiment, the invention provides a multi-level DC/AC converter, comprising:an input connectable to a direct voltage source, with a first connection and a second connection across which an input voltage can be applied, a neutral being positioned between the first and the second connection;a half-bridge with a first controlled switch and a second controlled switch between which an output of the converter is positioned;a first connecting branch between the first controlled switch and the first connection and a second connecting branch between the second controlled switch and the second connection;a first voltage regulator positioned to regulate voltage across a first capacitor;a second voltage regulator positioned to regulate voltage across a second capacitor;a third controlled switch connected between a first plate of the first capacitor and the first controlled switch;a fourth controlled switch connected between a first plate of the second capacitor and the second controlled switch.

In some embodiments between the first and the second connection, between which is applied the source of direct voltage, a pair of capacitors are positioned in series, between which the neutral of the circuit is arranged.

Some advantageous embodiments provide that: the first controlled switch and the third controlled switch are connected to a second plate of the first capacitor through the first connecting branch; and the second controlled switch and the fourth controlled switch are connected to a second plate of the second capacitor through the second connecting branch.

In some embodiments the first, second, third and fourth controlled switch are driven in such a way that:the output voltages between a positive limit value and a negative limit value are generated switching said first and said second controlled switch at a switching frequency, maintaining the third and the fourth controlled switch constantly open;output voltages above said positive limit value are generated switching the third controlled switch at a switching frequency and maintaining said first controlled switch constantly in conduction;output voltages below said negative limit value are generated switching the fourth controlled switch at a switching frequency maintaining said second controlled switch constantly in conduction;

Further advantageous features and embodiment of the converter according to the invention are set forth in the appended claims, which are an integral part of the present disclosure.

DETAILED DESCRIPTION OF EMBODIMENTS OF THE INVENTION

FIG. 1shows a first circuit diagram of a converter according to the invention, indicated as a whole with reference number1. Reference number3indicates a direct voltage source, e.g. a group of photovoltaic panels, which supplies an input voltage Vi. The source3is connected to two input connections5and7of the converter1. More in particular, in the diagram ofFIG. 1the positive pole of the source3is connected to the connection5and the negative pole of the source3is connected to the connection7.

Reference numbers9and11indicate two capacitors connected respectively between the connection5and the neutral N and between the connection7and the neutral N of the converter1. The voltage Vi of the source3is split at the ends of the two capacitors9and11, across the plates of each capacitor being present a voltage difference of Vi/2. Therefore, the input voltage of the converter1, referred to the neutral N, is Vi/2.

To the connections5and7a half-bridge is connected, which comprises a first branch15connected between the connection5and the output U of the converter1and a second branch17connected between the connection7and the output U. The first branch comprises a diode19and a controlled electronic switch21, e.g. a MOSFET or an IGBT, arranged in series. The switch21comprises an internal diode illustrated schematically in the drawing. The second branch17comprises a diode23and a second controlled electronic switch25, e.g. a MOSFET with its own internal diode. The centre of the half-bridge is connected with two clamp connections31and33to the neutral N. Each of the two connections31and33comprises a controlled electronic switch, e.g. a MOSFET or an IGBT and a diode, indicated by the references35and37for the branch31and by the references39and41for the branch33.

To the centre or output U of the half-bridge are connected a filtering inductance43and a filtering capacitor45. In the illustrated example, the converter is connected to an electrical energy distribution grid, schematically indicated with reference R, connected between the LC-filter formed by the inductance43and by the capacitor45and the neutral N of the converter1. In this way, the electrical energy supplied by the direct current source3is converted into alternating electrical energy at the grid frequency and injected onto the grid itself, in phase with the grid voltage. Alternatively, the converter1can be connected to a load that is powered (fully or partially, always or only in determined period) by the electrical energy directly supplied by the source3. For the purposes of the description of the invention, it is irrelevant whether the converter1supplies power to a load or supplies energy to the distribution grid. Therefore, hereafter reference shall, be made to this second possibility.

The structure described so far is known in itself.

Grid voltage oscillates sinusoidally and reaches peak values that are typically greater than the value Vi/2. Hence, it is necessary that the converter1be able to boost the voltage to a sufficient value to have an output voltage that reaches the grid voltage values.

For this purpose, the converter1comprises two voltage regulators51and53connected to the neutral N and each to a respective branch55,57in parallel to the branches15and17of the half-bridge. The branch55comprises a controlled switch59, e.g. a MOSFET or an IGBT represented with its internal diode. The branch57comprises a controlled switch61similar to the switch59. A corresponding capacitor63,65is positioned in parallel to each switch59and61. The capacitor63is connected between the branch55and the connection5, whilst the capacitor65is connected between the branch57and the connection7. Moreover, a first plate of the capacitor63is connected, through the branch55, to one end of the switch59, whilst a second plate of the capacitor63is connected through the branch15to a connecting point between the switches59and21. A similar connection is provided for the capacitor65, a first plate whereof is connected through the branch57to one end of the switch61, whilst a second plate is connected through the branch17to a connecting point between the switches25and61.

In the example shown, each voltage regulator51,53comprises two controlled electronic switches52,54in series, connected to the neutral and to the branch55, or to the branch57. The central point between the two switches52,54of each voltage regulator51,53is connected to a respective inductance56, whose second terminal is connected to the connection5for the voltage regulator51and to the connection7for the voltage regulator53.

The voltage regulators51and53are controlled in such a way as to generate a voltage V1across the respective capacitors63and65. In this way, the voltage between the branch55and the neutral is (Vi/2+V1), whilst the voltage between the branch57and the neutral is −(Vi/2+V1). Hence, the circuit is able to generate on the output U, i.e. at the central point of the half-bridge, five levels of voltage equal to 0, Vi/2, (Vi/2+V1), −(Vi/2+V1).

Operation of the circuit described above is as follows.

The switches35,39can be controlled to switch at the grid frequency, e.g. 50 Hz as a function of the sign of the output voltage. More in particular, switches35,39are controlled in such a way that the switch35is closed and the switch39is open in the positive half wave of the output voltage, whilst the switch39is closed and the switch35is open in the negative half wave of the output voltage.

The switches21,25,59,61are controlled to switch at high frequency, e.g. 15 kHz by means of a PWM signal with a duty cycle which is variable according to a control logic that will be described below with reference toFIGS. 2 and 3. Reference shall be made, hereafter, to a converter1applied to the electrical distribution grid to which the converter transfers electrical energy supplied by the source3and in which output current and output voltage are in phase. However, the following considerations also apply if the converter1is connected to a load and if this load introduces a voltage and current phase shift.

FIG. 2Ashows a positive half wave of the grid voltage. t0indicates the zero-crossing point, t1the instant in which the grid voltage reaches the value Vi/2, t2the instant in which the grid voltage after reaching the peak value passes through the value Vi/2 again and t3the subsequent zero-crossing instant.

FIGS. 2B and 2Cshow the driving signal S35and S39respectively of the switches35and39as a function of the output voltage. The signal S35is high (switch39closed) throughout the time interval [t0-t3], whilst the signal S39is low (switch39open). During the negative half wave the situation is reversed: S35is low (switch35open) whilst the signal S39is high (switch39closed).

In the time interval [t0-t1], since the output voltage is lower than Vi/2, the current is transferred to the output through the switch21, whilst the switch59remains open. To generate an output voltage that follows the sinusoidal grid pattern, the switch21is driven with a driving signal S21(FIG. 2D), whose duty cycle changes from 0 to 1 (FIG. 2G). The current entirely flows through the diode19whilst the branch55is not traversed by the current. During the opening intervals (Toff) of the switch21the current flows through the branch31and the diode37.

When the grid voltage reaches the value Vi/2, i.e. the voltage across capacitor9, it is necessary to supply a greater output voltage. For this purpose, the switch21that until the instant t1was choppered with the variable duty cycle signal S21, is kept in fixed conduction, whilst the switch59, which until this point had remained open, is driven with a driving signal S59with variable duty cycle D59, as shown inFIGS. 2E and 2G. The change from 0 to 1 of the duty cycle D59of the driving signal S59of the switch59allows to generate a voltage output that follows the pattern of the grid voltage for the time interval [t1-t2]. The current flows completely in the branch55and through the switches59and21, which are connected in series, during the conduction intervals (Ton) of the switch59, whilst it flows through the branch15and the diode19during the opening intervals (Toff) of the switch59.

When the grid voltage reaches the value Vi/2 at the instant t2and starts to drop below it, the switch59is maintained open and the switch21starts to be choppered by the signal S21with a duty cycle D21again variable from 1 to 0.

FIG. 2Fshows the waveform of the output voltage Vu that is obtained with the driving described above.

A symmetric situation occurs in the negative voltage half wave, in which the output voltage values between 0 and −Vi/2 are generated driving the switch25with a PWM signal with variable duty cycle, whilst the switch61remains open. To generate output voltage values below −Vi/2, the switch25is constantly maintained in conduction, whilst the switch61is driven with a PWM driving signal with variable duty cycle and, hence, it is in series with the switch25.

The curves ofFIG. 3show the corresponding wave forms of the current. Since in the example shown voltage and current are in phase, the pattern of the current matches the pattern of the voltage. In particular,FIG. 3shows: the pattern of the current119through the branch15and the diode19, the current137through the branch31and the diode37, the current159through the switch59, the voltage V1across the capacitor63and the output voltage Vu. As is readily apparent from the aforesaid curves, during the intervals [t0-t1] and [t2-t3] the current circulates through the branch15, the diode19and the switch21in the ON phase of the duty cycle of the switch21, and through the branch31and the diode37in the OFF phase of the duty cycle. In the interval [t1-t2] the current circulates in the branch55, in the switch59and in the switch21in the ON phase of the duty cycle of the switch59and through the branch31and through the branch15and the diode19in the OFF phase of the duty cycle of the switch59.

From the above description, it is clear that during the interval [t1-t2] in which the output voltage exceeds the value Vi/2 the two switches59,21work in series and the switch59must withstand a voltage of V1across its terminals, instead of the entire voltage (V1+Vi/2). Consequently, this switch can be dimensioned with a lower voltage rating than the switch21that, vice versa, must withstand a voltage of (V1+Vi/2) across its terminals.

Since during the negative half wave of the grid voltage there is a mirror-like situation for the switches25and61, the same considerations hold true for these two additional components.

From the above description, it is readily apparent that since the switches59and61must have a lower voltage rating than the switches21and25, their cost is lower than the switches21and25. This enables to obtain a first advantage over known circuits in which the switches of the half-bridge are connected in parallel and hence both must be dimensioned with a voltage rating at the maximum value of the voltage applied at the ends of the half-bridge.

Moreover, since the switching losses are the greater, the greater the voltage across the switch terminals, it is clear that during the time interval [t1-t2] where the switch21is not switched, but instead it remains constantly in conduction, whilst the switch59is choppered, the switching losses will be a function of a voltage V1instead of (V1+Vi/2) and hence far lower. Since within the half wave the interval [t1-t2] longer than half of the half wave, this entails a substantial advantage in terms of switching losses reduction. Not only are the higher losses limited to the intervals [t0-t1] and [t2-t3], but the sum of these intervals is less than [t1-t2].

The same considerations apply in mirror-like manner for the switches25and61.

In the final analysis, with a converter like the one shown inFIG. 1a high number of voltage levels is obtained, in the specific case five levels equal to:
−(V1+Vi/2);−Vi/2;0;Vi/2;(V1+Vi/2)
which allows to have a low harmonic content in the output voltage with a relatively light LC filter. Moreover, contrary to other known circuit solutions that provide the same level of output voltage, a substantial saving is obtained in terms of components cost and an efficient reduction of the switching losses.

FIG. 4shows an additional embodiment of a circuit according to the invention, which enables to obtain, in addition to the advantages already mentioned, a reduction in the number of components as well.

More in particular,FIG. 4shows a converter101connected through connections105and107to a direct voltage source103. The reference letter M indicates earth and the letter U indicates the output of the converter101. Between the connections105and107are positioned a pair of capacitors109A,109B and a half-bridge comprising a first branch115, connected between the connection105and the output U of the converter101, and a second branch117, connected between the connection,107and the output U. The first branch115comprises a controlled electronic switch indicated with IGBT5and an additional controlled electronic switch indicated with the reference IGBT2. Instead of IGBT, the switches can, for example, be constituted by MOSFET. The switches IGBT5and IGBT2are positioned in series between the connection105and the output U. Both switches comprise a respective internal diode schematically shown in the drawing.

A second branch117comprises a controlled electronic switch IGBT9and an additional controlled electronic switch IGBT3, each with its own internal diode as shown in the diagram ofFIG. 4.

To the centre or output U of the half-bridge are connected a filter inductance143and a filter capacitor145. In the example shown, the converter is connected to an electrical energy distribution grid, schematically indicated with the reference R. In this way, the electrical energy supplied by the direct current source103is converted into alternating electrical energy at the grid frequency and injected onto the said grid, in phase with the grid voltage. Alternatively, the converter101can be connected to a load that is powered by the electrical energy supplied by the source103.

The converter101further comprises two voltage regulators151and153connected respectively to the connections105,107and each to a respective branch155,157in parallel to the branches115and117of the half-bridge. The branch155comprises a controlled switch IGBT1, represented with its internal diode. The branch157comprises a controlled switch IGBT4similar to the switch159. In parallel to each switch IGBT1and IGBT4is positioned a respective capacitor163,165. The capacitor163is connected between the branch155and the connection105, whilst the capacitor165is connected between the branch157and the connection107. The voltage regulators can have a construction similar to that described with reference toFIG. 1.

The source103and the voltage regulators151and153generate four voltage levels which, referred to the earth M, have the following values: 0, V1, Vi/2+V1, Vi across the ends of each capacitor109A,109B,163,165.

The operation of the circuit is described hereafter with reference to the diagrams ofFIGS. 5,6and7.FIG. 5shows the waveform of the output voltage between the points U and N of the circuit ofFIG. 4. Voltage has a substantially sinusoidal pattern which is generated by appropriately switching the switches IGBT1, IGBT2, IGBT3, IGBT4, IGBT5, IGBT9.FIG. 6shows the driving signals and the currents in the various components of the circuit andFIG. 7shows the waveform of the voltage and of the current which, in the illustrated example, are in phase.

As in the case of the circuit ofFIG. 1, the switches IGBT2and IGBT3are dimensioned for a higher voltage rating, above Vi/2, where Vi is the voltage across the ends of the connections105and107, whilst switches IGBT1and IGBT3are dimensioned for a lower voltage rating, achieving similar advantages to those described with reference to the example ofFIG. 1.

With reference toFIG. 5, one can observe that the output voltage Vu (between U and N) has a sinusoidal pattern oscillating between a maximum and a minimum that in the example shown have values Vi and −Vi respectively. The chart also shows the values Vi/2 e−Vi/2 of the voltage at the connections105and107. During the positive half wave, the output voltage Vu grows from 0 to Vi/2 in the time interval t0-t1; in the time interval t1-t2grows from Vi/2 to the maximum value (indicated by way of example with the reference Vi) and then drops to Vi/2. In the interval t2-t3, it drops from Vi/2 to 0. In the negative half wave, there is a symmetrical behaviour with reversed signs: in the interval t3-t4the voltage drops from 0 to −Vi/2. In the interval t4-t5it drops to the minimum value Vi and then rises to the value −Vi/2. In the interval t5-t6it rises until it reaches the value 0 again.

In the illustrated hypothesis, with output current and voltage in phase, in the intervals t0-t1and t2-t3the current flows in the switches IGBT5, IGBT2, IGBT3and IGBT9. In the interval t1-t2the current flows in the switches IGBT1, IGBT5, and IGBT2. In the intervals t3-t4and t5-t6the current flows in the switches IGBT5, IGBT2, IGBT3and IGBT9. Lastly, in the interval t4-t5the current flows in the switches IGBT9, ITBT3, and IGBT4.

With reference to the wave forms ofFIG. 6, the current and of the voltage waveforms during a cycle of the output sinusoidal voltage shall now be described in greater detail. In the described example, it is also assumed that the output voltage and current are in phase, although this is not necessary and it does not occur for example if the circuit is connected to a reactive load, instead of to an electrical distribution grid or to a resistive load.

On the x-axis of the diagram inFIG. 6are shown the time instants t1, t2, t3, t4, t5and t6corresponding to the instants t1, . . . t6into which was subdivided the period of the output sinusoidal voltage represented in the diagram ofFIG. 5. The diagram inFIG. 6shows:the output current (in phase with the output voltage,FIG. 7);the currents in each of the switches IGTB1, IGTB2, IGBT3, IGBT4, IGBT5, IGBT9;the driving signals of the switches IGTB1, IGTB2, IGBT3, IGBT4, IGBT5, IGBT9.

The currents and the driving signals of the switches are the following:

In the time interval t0-t1, whilst the output voltage rises from a value 0 to a value Vi/2, and the current (under the assumptions made with respect to the output load) follows the pattern of the voltage, the switch IGBT1is open (driving signal at 0), the switch IGBT2is driven and switches at a frequency, e.g. 15 kHz; the switch IGBT3is driven and switches complementarily to IGBT2, i.e. when the switch IGBT2is closed, the switch IGBT3is open and vice versa. The duty cycle of the switch IGBT2gradually increases from 0 to 1, whilst correspondingly the duty cycle of the switch IGBT3decreases from 1 to 0. The switch IGBT4is open. The switches IGBT5and IGBT9are constantly closed (driving signals fixedly set to the value 1). In the closing time interval (Ton) of the switch IGBT2the current flows through the switches IGBT5and IGBT2(more specifically in the internal diode thereof), whilst during the opening interval (Toff) of the switch IGBT2the current flows through the switches IGBT0and IGBT3(more specifically in the internal diode thereof). Across the terminals of the switch IGBT2there is a voltage difference of Vi.

In the interval t2-t3, whilst the voltage rises above the value Vi/2, the switch IGBT1switches at a frequency, for example, of 15 kHz with a growing duty cycle. The switch IGBT2is constantly closed, the switch IGBT3is constantly open and the switch IGBT5switches in complementary fashion to the switch IGBT1. The switches IGBT9and IGBT4can assume any condition, since the switch IGBT3is open. In the illustrated example, the switch IGBT9is closed and the switch IGBT4is open. As a consequence of the state of the switches, the current flows through the switch IGBT1in the on interval (Ton) of the duty cycle, whilst in the off interval (Toff) of this switch the current flows through the switch IGBT5. As an effect of the gradual duty cycle increase of the switch IGBT1from the instant t1for a period of (t2-t1)/2 there is a gradual increase of the output current. Subsequently, from the intermediate instant of the interval t1-t2until the instant t2there is a gradual reduction of the duty cycle of the switch IGBT1, i.e. of its on time (Ton) and a complementary increase of the duty cycle of the switch IGBT5.

In the subsequent interval t2-t3there is a similar condition to that of the interval t0-t1, but with a gradual reduction of the conduction time (i.e. of the duty cycle) of the switch IGBT2and a consequent complementary gradual increase of the duty cycle of the switch IGBT3.

In the interval t3-t6of the negative half wave of the output current and voltage (assuming, as stated above, that current and voltage are in phase), there is a complementary situation to the one described above.

More in particular, in the interval t3-t4the switch IGBT1is open (driving signal at zero). The switch IGBT2is driven and it switches at a frequency, e.g. 15 kHz, complementarily to the switch IGBT3: i.e. when the switch IGBT2is closed, the switch IGBT3is open and vice versa. The duty cycle of the switch IGBT2gradually decreases, whilst correspondingly the duty cycle of the switch IGBT3increases. The switch IGBT1is open, the switch IGBT5and the switch IGBT9are constantly closed.

The current flows through the switches IGBT9and IGBT3during the closure phase (Ton of the duty cycle) of the switch IGBT3, whilst it flows through the switches IGBT5and IGBT2during the opening interval (Toff of the duty cycle) of the switch IGBT3and the closing interval (Ton of the duty cycle) of the switch IGBT2.

When at instant t4the voltage reaches the value −Vi/2, the state of the switches changes. For the interval t4-t5the switch IGBT2remains open. The switches IGBT1and IGBT5are driven complementarily and in the illustrated example the switch IGBT1is constantly maintained open whilst the switch IGBT5is constantly maintained closed. The switch IGBT3constantly remains in conduction (driving signal at 1), the switch IGBT4switches at high frequency (e.g. 15 kHz) with a variable duty cycle and the switch IGBT9switches complementarily to the switch IGBT4. The duty cycle of the switch IGBT4increases gradually until the minimum voltage is reached and then decreases, whilst the duty cycle of the switch IGBT9follows a complementary pattern.

Current flows constantly through the switch IGBT3and alternatively through the switch IGBT9and IGBT4at the respective conduction intervals, driven complementarily.

In the interval t5-t6the state of the switches returns to being that of the interval t3-t4, but with a gradual increase of the conduction time (i.e. the duty cycle) of the switch IGBT3and a gradual corresponding decrease of the conduction time (i.e. the duty cycle) of the switch IGBT4.

As in the case of the circuit ofFIG. 1, the series arrangement of the switches IGBT1, IGBT2and IGBT3, IGBT4enables to have a particularly favorable condition in terms of reduction of the switching losses by effect of the lower voltage rating on the switches IGBT1and IGBT5, with respect to what takes place in other topologies of multi-level DC/AC converters known in the art.

The advantage is particularly significant considering that the switching time of these switches (t1-t2for IGBT1in the positive half wave and t4-t5for IGBT4in the negative half wave) is longer than the switching time of the switches IGBT2and IGBT3.

It must be understood that the concept whereon the circuits ofFIGS. 1 and 4are designed can also be extended to a configuration with a higher number of voltage levels, since the structure of the circuit is modular.

It is understood that the drawing only shows an example provided by way of a practical demonstration of the invention, which can vary in forms and arrangements without however departing from the scope of the concept underlying the invention. Any reference numerals in the appended claims are provided to facilitate reading of the claims with reference to the description and to the drawing, and do not limit the scope of protection represented by the claims.