Power factor controller for induction motor

A phase-triggered, gate-controlled AC semiconductor switch, in series with an induction motor and its AC supply, optimizes power flow to the motor under changing mechanical load conditions by varying the amount of supply voltage applied to the motor over each half-cycle of the AC supply. The triggering point of the switch relative to the preceding zero crossing point of the supply voltage, that is, the switch firing angle or delay angle, is varied as a function of mechanical loading on the motor by means of a load current-induced feedback voltage augmenting to a varying degree the charging rate of a capacitor which triggers the semiconductor switch into conduction. Under increasing mechanical load conditions, the feedback voltage increases in proportion to the increasing load current, the increasing feedback voltage accelerating the charging rate of the capacitor to trigger the switch into conduction at a reduced firing angle, wherein power flow to the motor is increased. Conversely, under decreasing mechanical load conditions, the feedback voltage decreases in proportion to the decreasing load current, the decreasing feedback voltage decelerating the charge rate of the capacitor to trigger the switch into conduction at an increased firing angle, wherein losses caused by reactive current in the less-than-fully-loaded induction motor are reduced with a resultant optimization of power factor.

BACKGROUND OF INVENTION 
The present invention relates to electronic controllers for motors adapted 
to drive varying or less than full mechanical loads, and more 
particularly, to control circuits for automatically reducing the power 
applied to a less-than-fully-loaded AC induction motor, such reduced power 
application reducing losses caused by reactive current to improve the 
power factor of the induction motor. 
U.S. Pat. No. 4,052,648 to Nola discloses an AC induction motor control 
circuit of the subject type which utilizes a Triac switch (TRIAC is a 
trademark of The General Electric Company of Syracuse, New York) in series 
with an induction motor to lessen the time of supply voltage application 
to the motor, on a half-cycle basis, the time of supply voltage 
application being inversely proportional to the power factor (greater 
current lag; lessening mechanical load) which is sensed by load voltage 
and load current sampling. In effect, Nola continuously senses the phase 
angle between the load voltage and load current, and then uses a phase 
angle-related signal to continuously adjust the firing point of the Triac 
switch relative to the zero crossing point of the line voltage. For a 
sensed increasing phase angle (decreasing power factor) between load 
voltage and load current, Nola shifts the Triac firing point away from the 
line voltage zero crossing point to apply a smaller portion of each 
half-cycle of the line voltage, which inherently decreases the phase angle 
(increasing power factor) and reduces the heat loss (I.sup.2 R) caused by 
the reactive current. 
While Nola recognizes the energy-saving advantages of duty cycle 
controlling an induction motor as a function of load with a 
series-inserted, phase-triggered Triac switch, his phase angle measuring 
requirement and the resultant circuitry are undesirably complex and costly 
as compared to the relative simplicity and low cost of a small, single 
phase induction motor which exhibits the greatest need for reliable power 
factor regulation. 
U.S. application Ser. No. 042,608, filed May 25, 1979, by the inventor in 
the present application discloses an electronic controller which senses 
load current only in providing effective power factor control of an 
induction motor. While this current sensing only controller represents a 
substantial improvement over the earlier-discussed Nola device, it still 
requires a considerable number of components, resulting in costs which 
detract from its advantages in some applications. 
SUMMARY OF THE INVENTION 
In accordance with the present invention, an electronic controller for 
regulating power applied by an AC supply to an AC induction motor is 
provided to improve the power factor of the motor over a range of varying 
mechanical loads. 
A gate-controlled semiconductor AC switching means, connected in electrical 
series relationship with the AC supply and the induction motor, is 
combined with means for detecting current pulse through the motor when the 
AC switching is in a conducting state. The means for detecting includes a 
resistor means in series with the motor, the resistor means providing a 
proportional voltage pulse for each load current pulse. 
The proportional voltage pulse provided by the resistor means is impressed 
across the primary side of a transformer means having a nonlinearly 
responding secondary side. The secondary side provides, in response to the 
proportional voltage, a secondary voltage pulse having a trailing feedback 
portion existent subsequent to the impressed voltage pulse across the 
primary side. 
A control means responsive to the trailing feedback portion amplitude 
triggers the semiconductor switch into a conducting state subsequent to 
the preceding zero crossing point of the supply voltage. The time period 
between the zero crossing point and the triggering is proportional to the 
amplitude of the feedback portion, the switching means switching to a 
nonconducting state generally at the trailing edge of each load current 
pulse.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
Turning to FIG. 1, a conventional, single-phase AC supply 10 of, for 
example, 120 volts AC at 60 hertz is provided. Typically, the supply 10 
takes the form of a pair of commercial power lines 11, 12. The AC supply 
10 provides and applies power to a single-phase AC induction motor 20 of a 
conventional type, the supply 10 and the motor 20 being connected in 
electrical series relationship as illustrated wherein current flow through 
the supply 10 and the motor 20 is substantially equivalent. 
Power applied by the supply 10 to the motor 20 is regulated by a 
series-inserted, gate-controlled semiconductor AC switching means in the 
preferred form of an NPNPN-type thyristor switch 30 commonly known as a 
"Triac" (trademark of The General Electric Company of Syracuse, New York). 
As used herein, the term "thyristor" is intended to cover gate-controlled 
semiconductor switches such as silicon-controlled rectifiers and Triacs, 
which are in effect two silicon-controlled rectifiers connected 
back-to-back with a common gate. Such switches are well known in the art 
and are characterized by their ability to turn themselves off when their 
anode voltage is reduced to a point where a predetermined holding current 
limit through the thyristor is not maintained, resulting in the switch 
returning or recovering to a nonconducting state to block subsequently 
applied anode voltage-induced current flow. 
To complete the series circuit relationship of the supply 10, the motor 20, 
and the gate-controlled switching means 30, a small ohmic value current 
sampling resistor 35 typically of 0.01 ohms is provided. In normal 
operation, with the switch 30 in a fully conducting state, power is 
applied to the motor 20 by the supply 10 wherein generally all alternating 
current flow is through the supply 10, the motor 20, the thyristor switch 
30, and the current sampling resistor 35. In effect, these four elements 
are in electrical series relationship with each other. 
In accordance with known principles, the thyristor switch 30 can be 
phase-triggered on a half-cycle basis to apply more or less power to the 
motor 20, the degree of power application being dependent on the time 
period between the zero crossing point of the supply voltage and the 
firing point of the switch 30, such time period being commonly referred to 
as "the delay angle" or "the firing angle" of the switch 30. When the 
motor 20 is experiencing a light mechanical load, the firing angle of the 
switch 30 is maximized to limit power application to the motor or 
duty-cycle control of the motor to preclude a large current lag (low power 
factor) and resultant I.sup.2 R losses caused by reactive current. 
Conversely, under an increasing mechanical load conditions, the firing 
angle of the switch 30 is gradually and proportionally reduced to apply 
more power to the motor to preclude motor stalling while maintaining an 
acceptable power factor. Such principles are disclosed in the 
heretofore-noted U.S. Pat. No. 4,052,648 to Nola and my earlier-noted 
pending application. 
In accordance with the present invention, the current sampling resistor 35 
serves as a means for detecting alternating direction load current pulses 
through the motor 20 when the switching means 30 is in a conducting state, 
the resistor 35 providing across it a proportional voltage pulse for each 
load current pulsed. 
A step-up transformer 40, preferably in the form of an audio transformer, 
having a low-voltage, primary winding side 42 and a nonlinearly 
responding, high voltage, secondary winding side 44, is provided. The 
proportional voltage pulse generated across the current sampling resistor 
35 is impressed across the primary side 42 of the transformer means 40. In 
accordance with the present invention, the nonlinearly responding 
secondary side 44, in response to the proportional voltage impressed 
across the primary side 42, provides an induced secondary voltage pulse 
having a trailing feedback portion existing subsequent to the impressed 
voltage pulse across the primary side, the amplitude of the trailing 
feedback portion being proportional to the corresponding current pulse 
amplitude. 
A control means responsive to the trailing feedback portion triggers the 
thyristor switch 30 into a conducting state subsequent to the preceding 
zero crossing point of the supply voltage, the time period between the 
zero crossing point and the triggering being proportional to the amplitude 
of the feedback portion. A more detailed discussion of the feedback 
portion control voltage and its application to effect power factor control 
of the motor 20 will be given subsequently in more detail with regard to 
FIG. 2. 
In a preferred form, the control means includes a thyristor firing network 
having, in electrical series relationship, a capacitor 50, a current limit 
resistor 55, and a charging resistor 60. The current limit resistor 55 has 
one end connected to one side of the capacitor 50, while the charging 
resistor 60 also has one end connected to the other side of the capacitor 
50. The other ends of the current limit resistor 55 and the charging 
resistor 60 are connected respectively to the other end 35b of the current 
sampling resistor 35 and a thyristor power lead 32 not connected to an end 
35a of the current sampling resistor 35, the end 35a of the resistor 35 
being connected to another thyristor power lead 33. 
The thyristor firing network further includes a two-lead alternating 
current semiconductor switch, for example, a "Diac" (a trademark of The 
General Electric Company of Syracuse, New York). One lead of the two-lead 
alternating current semiconductor switch 65 is connected to a gate lead 31 
of the thyristor switch 30, while the other lead of the two-lead switch 65 
is connected to the interconnection junction of the current limit resistor 
55 and the capacitor 50. The secondary side 44 of the transformer is 
connected in parallel across the charging resistor 60. 
The operation of the thyristor firing network in conjunction with the 
transformer 40 and the current sampling resistor 35 will now be discussed 
with reference to FIGS. 1 and 2. 
Waveform 2a of FIG. 2 illustrates a sinusoidal AC supply voltage provided 
across the supply 10. FIG. 2b represents the load voltage across the motor 
20 under a steady state less-than-full-load condition, wherein only a 
portion of each half-cycle of the supply voltage is applied to the motor 
20 to effect power factor optimization in accordance with 
earlier-discussed, known techniques to minimize the load current lag 
I.sub.L as illustrated in waveform 2c. The firing angle or phase delay 
angle D.sub.FA, that is, the time period between the zero crossing point 
of the supply voltage and the triggering of the thyristor switch 30 is 
illustrated by waveform 2b. FIG. 2c illustrates the load current pulses 
through the motor 20 and, necessarily, through the switch 30 and the 
current sampling resistor 35. 
In view of waveforms 2a, 2b, and 2c, and with reference to FIG. 1, the 
switch 30 is triggered at the firing angle D.sub.FA, which is determined 
by the rate of charging of the capacitor 50, which upon reaching a 
predetermined voltage breaks down the two-lead alternating current switch 
65 which triggers or fires the triac switch 30 into conduction. The 
circuit configuration and operation of the switches 30, 65, the capacitor 
50, and the current limit resistor 55 are well known in the art. The 
switch 30 returns to a nonconducting condition at the slightly lagging, 
trailing edge (see FIG. 2c) of each load current pulse (generally near 
zero crossing point of supply voltage), as is the characteristic of 
thyristor switches discussed earlier. With the switch 30 in a 
nonconducting state, the capacitor 50 begins charging from the supply 10 
via the current limit resistor 55 and charging resistor 60 (approximately 
560 ohms), the current limit resistor being typically of a high value, 
such as 100 kilohms, to provide the necessary RC time constant for 
establishing the desired thyristor firing angle D.sub.FA. The current 
limiting resistor 55, as its name implies, draws very little load current, 
and hence, as noted earlier, substantially all of the load current passes 
through the switch 30 and the current sampling resistor 35. The supply 
voltage 10 acts as a primary so as to charge the capacitor 50 at a 
predetermined rate. It can be seen that the deletion of the charging 
resistor 60 and the transformer 40 would provide a constant firing angle 
D.sub.FA, regardless of changing load conditions on the motor 20. 
To provide for feedback control of the charging for the capacitor 50 in 
accordance with the present invention, the secondary side 44 of the 
transformer provides control voltage pulses illustrated in FIG. 2d. The 
transformer, as discussed earlier, has a secondary side 44 which is a 
nonlinearly responding step-up winding. The stepped-up secondary voltage 
is desirable to the small voltage induced by the load current cross the 
low ohmic value resistor 35. The step-up function of the transformer 40 
could be eliminated if the ohmic value of the sampling resistor 35 were 
increased. As illustrated, the primary winding side 42 includes a 
polarity-marked lead 43 connected to the interconnection junction of the 
power lead 33 and the end 35a of the resistor 35. The secondary winding 
side 44 also includes a polarity-marked lead 45 connected to the end of 
the charging resistor 60 not connected to the capacitor 50. For given 
polarity voltage pulses through the primary side 42 (and in particular 
nonsinusoidal voltage pulses), the secondary side 44 does not provide 
proportional mirror image secondary voltage pulses but, rather, provides, 
as illustrated in waveform 2d, secondary voltage pulses having trailing 
feedback portions V.sub.F that are existent subsequent to the impressed 
voltage pulse across the primary side corresponding to the respective load 
current pulse (waveform 2c). This feedback portion is applied in proper 
polarity relation, as determined by the earlier-noted interconnection of 
the transformer 40, across the charging resistor 60 to assist as a 
secondary source in the charging of the capacitor 50 during the 
nonconducting period of the switch 30. The charging resistor is desirable 
to provide a primary charging and discharging path for the capacitor 50, 
since the secondary side winding 44 may have a high impedance. 
It can be seen that the supply voltage and the feedback portion of the 
induced secondary voltage are in phase to act in additive fashion as the 
charging source for the capacitor 50. 
In accordance with the present invention, the feedback portion V.sub.F acts 
as a secondary charging source to the capacitor 50 to accelerate or 
decelerate its charging rate to, in effect, shift the firing angle 
D.sub.FA away from or toward the preceding zero crossing point of the 
supply voltage. 
While FIG. 2 illustrates the operation of the circuit of FIG. 1 under a 
steady state, less-than-full-load condition, the response of the circuit 
to a dynamic changing mechanical load on the motor 20 can easily be 
envisioned. 
An increasing mechanical load on the motor 20 increases the amplitude of 
the load current pulses (waveform 2c) through the current sampling 
resistor 35. Under the increasing load condition, the motor is, in effect, 
calling for more power. Since the amplitude of the load current pulses 
through the resistor 35 increases, the proportional voltage impressed 
across the primary side 42 of the transformer proportionally increases to 
provide a secondary induced voltage of increased amplitude having an 
increased amplitude feedback portion V.sub.F as illustrated in waveform 
2d. Since the amplitude of the feedback portion V.sub.F is increased, the 
charging rate of the capacitor 50 is necessarily accelerated wherein it 
reaches the predetermined firing point voltage of the two-lead alternating 
current switch 65 at a point closer in time to the preceding zero crossing 
point of the supply voltage to decrease the firing angle D.sub.FA. In 
turn, decreasing the firing angle D.sub.FA applies more of each half-cycle 
of supply voltage to the motor for increased power application. 
Conversely, the lessening of mechanical load on the motor 20 causes a 
corresponding drop in the amplitude of the load current pulses, which in 
turn causes a corresponding proportional drop in the amplitude of the 
trailing feedback portion V.sub.F wherein the charging rate of the 
capacitor 50 is decelerated so as to increase the firing angle D.sub.FA 
and thus apply less power to the motor. It should also be noted that, 
since the load is a motor 20, as opposed to for example an incandescent 
lamp, a conventional snubber network 70 (dv/dt suppression) parallel to 
the switch 30 is provided to improve commutation and prevent false turn-on 
of the switch 30. 
It can be seen that a relatively simple and fast-acting, positive feedback 
control circuit is provided to effect half-cycle response to power factor 
control of the motor 20. While the illustrated application of the 
invention has been directed to a single-phase circuit, it is clearly 
within the contemplation of the invention that the teaching of the present 
invention also applies to polyphase motor control circuits. 
It should be evident that this disclosure is by way of example and that 
various changes may be made by adding, modifying or eliminating details 
without departing from the fair scope of the teaching contained in this 
disclosure. The invention is therefore not limited to particular details 
of this disclosure except to the extent that the following claims are 
necessarily so limited.