DC offset compensation device

A level detector 2 detects variation of the amplitude of an input signal a to output a level signal b representing HIGH or LOW in order to define the head portion of the input circuit a. A time constant control signal 3 generates a time constant control signal c based on the level signal b to control a time constant of an estimator 4 so as to make the time constant small for a prescribed period from a time when the level signal b varies from HIGH to LOW. The estimator 4 estimates DC offset included in the input signal a with the a time constant variation according to the time constant control signal c to output an estimate d. A compensator 1 subtracts the estimate d from the input signal a to obtain a compensation output. Therefore, in the estimator 4, the speed of estimating the DC offset is different between a period corresponding to the head portion of the input signal a and other periods. Thus, a DC offset compensation device can be configured to be capable of fast DC offset compensation at the head portion of the input signal a and stable DC offset compensation at the other portions.

BACKGROUND OF THE INVENTION 
1. Field of the Invention 
This invention relates to DC offset compensation devices and more 
specifically to a DC offset compensation device automatically eliminating 
a DC offset component which is included in a received signal, a detected 
signal, etc., amplified by an automatic gain control function. 
2. Description of the Background Art 
When a signal with a low voltage level is received, a DC offset component 
caused by an amplifier or an AD converter becomes a big factor in 
interfering with a precise signal receiving. Therefore, it is important to 
eliminate the DC offset component of a received signal. A DC offset 
compensation device for eliminating the DC offset component of the 
received signal is thus installed in a receiver. A conventional DC offset 
compensation device is disclosed in Japanese Patent Laying-Open No. 
62-53023, for example. The conventional DC offset compensation device is 
subsequently described while referring to the drawings. 
FIG. 21 is a block diagram showing a structure of the conventional DC 
offset compensation device. In FIG. 21, the DC offset compensation device 
includes an adder 101, an A/D converter (ADC) 102 and a charge pump 103. 
An analog input signal ain, which is band-limited and whose average of 
amplitude distribution is 0 is inputted to the DC offset compensation 
device. The analog input signal ain is added to a correction signal pout 
at the adder 101, and is converted to a digital output dout at the A/D 
converter 102. The charge pump 103 inputs a most significant bit (MSB) sgn 
of the digital output dout and integrates sgn to obtain the correction 
signal pout. Since the digital output dout is represented by a complement 
on two, the MSB sgn designates a sign of the digital output dout. That is, 
the MSB sgn is 0 when the digital output dout is positive, and the MSB is 
1 when the digital output dout is negative. Therefore, if the digital 
output dout is biased to the negative side on the average, the charge pump 
103 integrates 1 to gradually increase the correction signal pout. In 
response, the voltage level of a signal outputted by the adder 101 moves 
to the positive side. Therefore, the digital output dout also gradually 
moves to the positive side to automatically correct bias of the digital 
output. The DC offset compensation device thus outputs the digital output 
dout without the DC offset component. 
When the DC offset compensation device with the above structure is applied 
to a receiver having an automatic gain control (AGC) amplifier, however, 
the following problems will occur. 
FIGS. 22(a) to 22(d) are diagrams for describing problems when the DC 
offset compensation device in FIG. 21 is applied to the receiver having 
the AGC amplifier. FIG. 22(a) schematically shows a waveform of a signal 
inputted to the AGC amplifier. FIG. 22(b) shows time variation of a gain 
of the AGC amplifier. FIG. 22(c) schematically shows a waveform of a 
signal outputted by the AGC amplifier. FIG. 22(d) shows time variation in 
a voltage level of the DC offset component included in the digital output 
dout when a signal outputted by the AGC amplifier (refer to FIG. 22(c)) is 
inputted to the conventional DC offset compensation device. 
In FIG. 22(a), the input signal inputted to the AGC amplifier is a wide 
band signal which is band-limited and whose average of amplitude 
distribution is 0, as described above. Since the input signal includes a 
DC offset component (refer to a dash-dot line), however, the input signal 
is biased to the positive side. In FIG. 22 (a), only an envelope of the 
input signal (refer to a shaded area) is shown. 
In FIG. 22(b), the gain of the AGC amplifier varies a high gain to a low 
gain at a head portion of a received signal when the received signal rises 
abruptly. The reason for the high gain at the head portion is that the 
gain is controlled so as to maintain the amplitude of the signal received 
by the AGC amplifier in a constant voltage level. The AGC amplifier 
requires some time until it controls the gain to obtain an appropriate 
gain. Therefore, in the signal outputted by the AGC amplifier, its 
amplitude becomes extremely large at the head portion and then converges 
on a prescribed amplitude, as shown in FIG. 22(c). FIG. 22(c) shows only 
an envelope (refer to a shaded area) of the output signal. 
As described above, when the signal inputted to the AGC amplifier includes 
a prescribed DC offset component, the DC offset component appearing in the 
signal outputted from the AGC amplifier profoundly varies according to 
variation of the gain. That is, as shown in FIG. 22(c), the voltage level 
of the DC offset component profoundly varies at the head portion (refer to 
a chain line). Moreover, when signals with various strength are received, 
the magnitude of the DC offset component appearing in the output signal of 
the AGC amplifier varies by signal. 
When the conventional DC offset compensation device is used in this 
situation, since a sufficiently large time constant is given not to 
distort signals, as shown in FIG. 22(d), the DC offset component 
profoundly varied at the head portion of the signal cannot be immediately 
compensated and remains in the digital output dout for a long time. On the 
other hand, if the time constant is set to be small so as to immediately 
compensate the DC offset component at the head portion of the signal, the 
DC offset component follows the variation of the signal itself to distort 
the signal. In addition, at the head portion of the signal, the amplitude 
is extremely large, and may be saturated or distorted. Therefore, the 
small time constant disturbs the operation of the DC offset compensation 
device and causes an erroneous operation. 
Furthermore, DC offset compensation devices for eliminating the DC offset 
component of the input signal other than the DC offset compensation device 
shown in FIG. 21 are disclosed in U.S. Pat. No. 5,241,702 and U.S. Pat. 
No. 5,212,826. Similarly, when these DC offset compensation devices are 
applied to the receiver having the AGC amplifier, the DC offset component 
extremely varied at the head portion of the input signal cannot be 
compensated immediately, and when the time constant is set small so as to 
immediately compensate the DC offset component at the head portion of the 
input signal, as in the case described above, the operation of the DC 
offset compensation device is disturbed and an erroneous operation is 
caused. 
SUMMARY OF THE INVENTION 
An object of the present invention is to provide a DC offset compensation 
device capable of fast DC offset compensation at the head portion of a 
signal and stable DC offset compensation without distortion of the 
following portions of the signal when a receiver having an AGC amplifier 
receives a signal which immediately rises. 
To attain the above object, the present invention is directed to a device 
for compensating DC offset included in an input signal, which comprises: 
a level detecting portion detecting variation in amplitude of the input 
signal to generate and output a level signal which defines a head portion 
of the input signal; 
a time constant control portion generating to output a time constant 
control signal on the basis of the level signal; 
a compensating portion subtracting an estimate of the DC offset from a 
value of the input signal to output a compensation output; and 
an estimating portion inputting the compensation output, estimating the DC 
offset of the input signal on the basis of the compensation output with a 
time constant which varies according to the time constant control signal, 
and outputting the estimate to feedback to the compensating portion. 
As described above, the level detecting portion detects the variation in 
the amplitude of the input signal. The time constant control portion 
generates the time constant control signal for varying the time constant 
during a period corresponding to the head portion of the input signal. In 
response, the estimating portion estimates the DC offset with the time 
constant varying according to the time constant control signal on the 
basis of the inputted compensation output. Therefore, the estimating 
portion can vary rapidly (time constant) for estimating the DC offset 
during the period corresponding to the head portion of the input signal. 
The DC offset compensation device can be thus configured to be capable of 
fast DC offset compensation at the head portion of the input signal and 
stable DC offset compensation at the other portions. 
The level signal is a binary signal having a first value or a second value, 
and a time when the level signal varies from the first value to the second 
value is taken as a first time, while a time lapsed for a prescribed time 
from the first time in a state that the level signal has the second value 
is taken as a second time. In this situation, the time constant control 
portion outputs the time constant control signal for maximizing the time 
constant during a period when the level signal keeps on having the second 
value after the second time. 
Therefore, the estimating portion estimates the DC offset while maximizing 
the time constant when the time constant control signal for maximizing the 
time constant is inputted. The DC offset compensation device can thus 
perform stable DC offset compensation when the DC offset of the input 
signal converges on the vicinity of 0 after the second time. 
The time constant control portion may be adapted to generate and output the 
time constant control signal for minimizing the time constant during a 
period between the first time and the second time, or the time constant 
control signal for monotonically increasing the time constant from the 
minimum to the maximum continuously or stepwise. 
Therefore, the estimating portion estimates the DC offset while minimizing 
the time constant when the time constant control signal for minimizing the 
time constant is inputted. The DC offset compensation device can thus 
perform fast DC offset compensation during a period between the first time 
and the second time. The estimating portion also estimates the DC offset 
while varying the time constant according to variation in the time 
constant control signal for monotonically increasing the time constant 
from the minimum to the maximum continuously or stepwise. The DC offset 
compensation device can thus perform fast and precise DC offset 
compensation during the first time and the second time. 
The estimating portion can be configured, for example, as in the following: 
The estimating portion includes a first integrator integrating the 
compensation signal to output a first integration value, a comparator 
deciding an upper limit reference value and a lower limit reference value 
on the basis of the time constant control signal and comparing the first 
integration value with the decided upper limit reference value and lower 
limit reference value to output a comparison output, and a second 
integrator integrating the comparison output to output the estimate. The 
comparator outputs the comparison output as 0 when the first integration 
value is more than the lower limit reference value and less than the upper 
limit reference value, as +1 when the first integration value is not less 
than the upper limit reference value, and as -1 when the first integration 
value is not more than the lower limit reference value. The first 
integrator resets the first integration value when the comparison output 
is +1 or -1. Therefore, the estimating portion generates different 
comparison outputs according to the time constant control signal, and 
generates an estimate to which these comparison outputs are cumulatively 
added. Thus, the frequency with which the comparator outputs the 
comparison output represented by +1 or -1 is high when the time constant 
control signal has a small value. The estimating portion can hasten the 
speed (time constant) for estimating the DC offset according to the 
frequency of outputting the comparison output represented by +1 or -1. 
The first integrator can be configured, for example, as in the following: 
In a first structure, the first integrator has a quantizer quantizing the 
compensation output according to its amplitude and generating to output a 
quantization signal, a first register holding the first integration value 
and a first adder adding the quantization signal to the first integration 
value held by the first register and taking the addition result as a first 
integration value. 
In a second structure, the first integrator has a quantizer quantizing the 
compensation output according to its amplitude and generating to output a 
quantization signal, and a first up/down counter generating to output the 
first integration value on the basis of the quantization signal. The first 
up/down counter changes a counting direction according to the quantization 
signal. 
The first integrator cumulatively adds the compensation outputs in the 
first or second structure. Furthermore, in the second structure, since the 
first integrator is configured by the first up/down counter, the number of 
its components can be decreased and the circuit configuration of the DC 
offset compensation device can be simplified and downsized, compared with 
the first structure. 
The quantizer is configured to quantize the compensation output, according 
to its amplitude, to two values of {+1, -1} or three values of {+1, 0, -1} 
to output the quantization signal. 
Therefore, the estimating portion estimates the DC offset independently of 
the amplitude of the compensation output. Thus, an erroneous operation 
hardly occurs when DC offset compensation of a signal which has a large 
amplitude at the head portion of the input signal is performed. 
Moreover, the second integrator can be configured, for example, as in the 
following: In a first structure, the second integrator has a second 
register holding the estimate and a second adder adding the comparison 
output to the estimate held by the second register and taking the addition 
result as an estimate. 
In a second structure, the second integrator has a second up/down counter 
generating to output the estimate on the basis of the comparison output. 
The second up/down counter changes a counting direction according to the 
inputted comparison output. 
The second integrator integrates the comparison output and outputs the 
estimate in the first or second structure. Furthermore, in the second 
structure, since the second integrator is configured by the second up/down 
counter, the number of its components can be decreased and the circuit 
configuration of the DC offset compensation device can be simplified and 
downsized, compared with the first structure. 
The estimating portion may be configured a in the following: In another 
first structure, the estimating portion includes a correction constant 
generator generating a correction constant according to the time constant 
control signal, an adder/subtractor selecting addition or subtraction 
according to the compensation output and a register holding the estimate. 
The adder/subtractor adds the estimate held by the register to the 
correction constant to output the addition result as an estimate when the 
compensation output is positive. The adder/subtractor subtracts the 
correction constant from the estimate held by the register to output the 
subtraction result as an estimate when the compensation output is 
negative. 
As described above, since the correction constant generator includes the 
adder/subtractor selecting addition or subtraction on the basis of the 
sign of the compensation output, the correction constants representing 
positive and negative do not have to be generated. Therefore, the number 
of components of the correction constant generator can be decreased and 
the circuit configuration of the DC offset compensation device can be 
simplified and downsized. 
In another second structure, the estimating portion includes a correction 
constant generator generating a positive correction constant and a 
negative correction constant according to the time constant control signal 
to output either one of the correction constants according to the 
compensation output, a register holding the estimate and an adder adding 
the correction constant outputted from the correction constant generator 
to the estimate held by the register to output the addition result as an 
estimate. The correction constant generator outputs the positive 
correction constant when the compensation output is positive. The 
correction constant generator outputs the negative correction constant 
when the compensation output is negative. 
As described above, the correction constant generator generates the 
positive correction constant and negative correction constant according to 
the time constant control signal. Therefore, the positive and negative 
correction constants can be independently set. Furthermore, since the 
estimating portion is configured by the adder, the circuit configuration 
of the DC offset compensation device can be simplified and downsized. 
To attain the above object, the present invention is also directed to a 
device for compensating DC offset included in an input signal, which 
comprises: 
a level detecting portion detecting variation in amplitude of the input 
signal to generate and output a level signal which defines a head portion 
of the input signal; 
a time constant control portion generating to output a time constant 
control signal on the basis of the level signal; 
a compensating portion subtracting an estimate of the DC offset from a 
value of the input signal to output a compensation output; and 
an estimating portion inputting the input signal, estimating the DC offset 
of the input signal with a time constant which varies according to the 
time constant control signal and outputting the estimate to feed forward 
to the compensating portion. 
As described above, the level detecting portion detects the variation in 
the amplitude of the input signal. The time constant control portion 
generates the time constant control signal for varying the time constant 
during a period corresponding to the head portion of the input signal. In 
response, the estimating portion estimates the DC offset with the time 
constant varying according to the time constant control signal from the 
inputted input signal. Therefore, the estimating portion can vary the 
rapidity for estimating the DC offset during the period corresponding to 
the head portion of the input signal. The DC offset compensation device 
can be thus configured to be capable of fast DC offset compensation at the 
head portion of the input signal and stable DC offset compensation at the 
other portions. 
The estimating portion is a low-pass filter varying a tap coefficient 
according to the time constant control signal. 
Therefore, the estimating portion can change the rapidity (time constant) 
for estimating the DC offset component according to the time constant 
control signal. The DC offset compensation device can be thus configured 
to be capable of fast DC offset compensation at the head portion of the 
input signal and stable DC offset compensation at the other portions. 
Further, the level detecting portion described above can be configured, for 
example, as in the following: In a first structure, the level detecting 
portion inputting the input signal includes a high pass filter generating 
to output a high pass signal from which a low frequency component of the 
input signal is eliminated, a rectifier generating to output a 
rectification output obtained by rectifying the high pass signal and a 
smoothing unit generating to output a smooth output with time varying of 
the rectification output reduced. The level detecting portion outputs the 
level signal on the basis of the smooth output. 
In a second structure, the level detecting portion inputting the 
compensation output includes a rectifier generating to output a 
rectification output obtained by rectifying the compensation output and a 
smoothing unit generating to output a smooth output with time variation of 
the rectification output reduced. The level detecting portion outputs the 
level signal on the basis of the smooth output. 
In the first or second structure, as outputting the level signal according 
to the smooth output, the level detecting portion detects the head portion 
of the input signal inputted to the DC offset compensation device from the 
AGC amplifier. Furthermore, in the second structure, the level detecting 
portion generates to output the level signal from the compensation output. 
DC offset is almost eliminated from this compensation output. Therefore, 
in the second structure, the level detecting portion does not require a 
high pass filter. The number of components of the level detecting portion 
can be decreased and the circuit configuration of the DC offset 
compensation device can be simplified and downsized. 
The level signal generated by the level detecting portion having the first 
or second structure is a binary signal having a first value or a second 
value. The level detecting portion compares the smooth output and a 
prescribed reference value, and takes the level signal as the second value 
when the smooth output is continuously larger than the prescribed 
reference value for a prescribed judging time and as the first value when 
otherwise. 
Therefore, if the smooth output exceeds the reference value for a moment by 
a noise or variation of the signal, an erroneous level signal is not 
generated and a rate of an erroneous operation of the level detecting 
portion can be small. 
In a third structure, the level detecting portion inputting the input 
signal includes a high pass filter generating to output a high pass signal 
from which a low frequency component of the input signal is eliminated, a 
rectifier generating to output a rectification output obtained by 
rectifying the high pass signal, a smoothing unit generating to output a 
smooth output with time variation of the rectification output reduced, a 
threshold generating portion time-averaging the smooth output and 
generating to output a threshold formed by multiplying the time-averaged 
smooth output by a prescribed coefficient not less than 1 and a judging 
portion comparing the smooth output inputted by the smoothing unit with 
the threshold and generating to output a judgment output representing 
whether the smooth output is larger or smaller than the threshold. The 
level detecting portion outputs the level signal on the basis of the 
judgment output. 
In the third structure, the threshold generating portion generates to 
output the threshold formed by multiplying the time-averaged smooth output 
by the prescribed coefficient not less than 1. The level judging portion 
outputs the level signal on the basis of the judging result whether the 
smooth output is larger or smaller than the threshold. Therefore, a level 
of the smooth output at the head portion of the signal does not become 
sufficiently large, the head portion of the signal can be precisely 
detected. 
In the level detecting portion having the third structure, the level signal 
is a binary signal having a first value or a second value. The level 
detecting portion takes the level signal as the second value when the 
judgment output continuously becomes a large value for a prescribed 
judging time and as the first value when otherwise. 
In the third structure, the level detecting portion takes the level signal 
as the second value only when the judgment output continuously becomes the 
large value for the prescribed judging time. Therefore, if the smooth 
output is above the reference value for a moment by a noise or variation 
of the signal, an erroneous level signal is not generated and a rate of an 
erroneous operation of the level detecting portion can be small. 
Furthermore, in a fourth structure, the level signal outputted from the 
level detecting portion is a binary signal having a first value or a 
second value. The level detecting portion inputting the input signal 
includes a high pass filter generating to output a high pass signal from 
which a low frequency component of the input signal is eliminated, a 
rectifier generating to output a rectification output obtained by 
rectifying the high pass signal, a smoothing unit generating to output a 
smooth output with time varying of the rectification output reduced, a 
threshold generating portion time-averaging the smooth output and 
generating to output a threshold formed by multiplying the time-averaged 
smooth output by a prescribed coefficient not less than 1, a delay portion 
generating to output a delayed threshold obtained by delaying the 
threshold for not less than a prescribed judging time and a judging 
portion comparing the smooth output with the delayed threshold and 
generating to output a judgment output representing whether the smooth 
output is larger or smaller than the threshold. The level detecting 
portion then takes the level signal as the second value when the judgment 
output continuously becomes a large value for the prescribed judging time 
and as the first value when otherwise. 
In the fourth structure, the threshold generating portion generates to 
output the threshold formed by multiplying the time-averaged smooth output 
by the prescribed coefficient not less than 1. The delay portion generates 
to output the delayed threshold obtained by delaying the threshold for not 
less than the prescribed judging time. The level detecting portion takes 
the level signal as the second value only when the smooth output 
continuously becomes larger than the delayed threshold for the prescribed 
judging time. Therefore, at the head portion of the signal, if the 
threshold varies following the level varying of the signal, since the 
delayed threshold holds a value before varying, without receiving the 
effect of the level varying during the judging time, an erroneous 
operation in detecting the head portion of the signal can be prevented. 
These and other objects, features, aspects and advantages of the present 
invention will become more apparent from the following detailed 
description of the present invention when taken in conjunction with the 
accompanying drawings.

DESCRIPTION OF THE PREFERRED EMBODIMENTS 
(First embodiment) 
FIG. 1 is a block diagram showing a DC offset compensation device according 
to a first embodiment of the present invention. In FIG. 1, the DC offset 
compensation device includes a compensator 1, a level detector 2, a time 
constant control circuit 3 and an estimator 4. The compensator 1 subtracts 
an estimate d inputted from the estimator 4 from an input signal a (refer 
to FIG. 20(a)) inputted from an AGC amplifier (not shown). The compensator 
1 outputs the subtraction result as a compensation output e. The estimator 
4 inputs the compensation output e to generate the estimate d. 
Specifically, the estimator 4 increases the estimate d when a DC offset 
component of the compensation output e is positive, and decreases the 
estimate d when the DC offset component of the compensation output e is 
negative. The compensator 1 and the estimator 4 configure a feedback 
control loop, resulting in that the DC offset component of the 
compensation output e converges on 0. That is, the estimate d converges on 
a value of a DC offset component of the input signal a. The input signal a 
is also provided for the level detector 2. The level detector 2 generates 
a level signal b having a first value representing HIGH or a second value 
representing LOW according to the amplitude of the input signal a, and 
outputs the level signal b to the time constant control circuit 3. HIGH 
and LOW are represented by HIGH and LOW of a logical circuit, 
respectively, for example. The time constant control circuit 3 generates a 
time constant control signal c on the basis of the level signal b and 
provides the time constant control signal c for the estimator 4. The 
estimator 4 varies the time constant of an estimating operation on the 
basis of the time constant control signal c. 
The structure and operation of the DC offset compensation device of the 
first embodiment are subsequently described in more detail referring to 
FIGS. 2 to 8(a)-8(f). FIGS. 2 to 7 are diagrams showing more detailed 
structures of each portion of the DC offset compensation device shown in 
FIG. 1. FIGS. 8(a)-8(f) are diagrams schematically showing waveforms in 
the main portions of the DC offset compensation device shown in FIG. 1. 
FIG. 8(a) shows a waveform of the signal (the input signal a) inputted to 
the DC offset compensation device. In FIG. 8(a), since the input signal a 
is the same as that in FIG. 20(c), the description is omitted. FIGS. 
8(b)-8(f) are subsequently described when required. 
FIG. 2 is a block diagram showing a more detailed structure of the level 
detector 2 shown in FIG. 1. In FIG. 2, the level detector 2 includes a 
high pass filter 21, a rectifier 22, smoothing unit 23, a comparator 24 
(as an example of a to judging means), a shift register 25 and an AND 
circuit 26. The high pass filter 21 eliminates a low frequency component 
in the vicinity of a direct current in the input signal a to generate a 
high pass signal a1. The high pass filter 21 is to prevent an erroneous 
operation of the level detector 2 by the DC offset component of the input 
signal a. The rectifier 22 calculates an absolute value of the high pass 
signal a1 to generate a rectification output a2. The smoothing unit 23 is 
a low pass filter or an integrator which smooths a waveform of the 
rectification output a2 to generate a smooth output a3. The smooth output 
a3 is outputted as a string of sample values by a certain sampling time. 
The comparator 24 compares and judges whether the smooth output a3 is 
larger or smaller than a prescribed reference value by the sampling time. 
As a result of the judgment, the comparator 24 outputs a comparison output 
b0 (judgment output b0) having a value of 1 when the smooth output a3 is 
larger than the reference value or a value of 0 when the smooth output a3 
is smaller than the reference value. The shift register 25 shifts its 
content with the comparison output b0 being inputted by the sampling time, 
and stocks the current three bits of the comparison output b0 to output 
them in parallel. The AND circuit 26 calculates the logical product from 
the three bits of the comparison output b0. The AND circuit 26 generates 
the level signal b (refer to FIG. 8(b)) representing HIGH when the 
obtained logical product is 1 or LOW when the obtained logical product is 
0, and outputs the level signal b to the time constant control circuit 3. 
That is, the level signal b becomes HIGH when the comparison output b0 is 
1 for three consecutive samples, and becomes LOW when otherwise. The 
prescribed reference value is set to a higher value (preferably about 1.5 
to 3 times) than a voltage level of the smooth output a3 obtained as an 
output from the smoothing unit 23 when a signal outputted from the AGC 
amplifier in a state of constant convergence of the gain is taken as the 
input signal a of the level detector 2. 
The shift register 25 and the AND circuit 26 described above are provided 
to prevent an erroneous operation of the level detector 2 due to an effect 
of noise etc. That is, the level detector 2 keeps the level signal b LOW 
when the comparison output b0 becomes 1 for a moment due to the effect of 
noise etc., and makes the level signal b HIGH using a characteristic that 
the comparison output b0 maintains a state of 1 for a certain period of 
time when the level of the signal is truly varied. Thus, the level 
detector 2 does not operate erroneously. While the number of bits of the 
shift register 25 and the AND circuit 26 are three, the number of bits is 
not limited to three and should be defined according to requirements of 
the system design. Further, the shift register 25 and the AND circuit 26 
can be omitted and the smooth output a3 can be taken as the level signal 
b, as it is (in this case, the number of bits is 0). Further, the larger 
the number of bits, the smaller the probability that the level signal b 
erroneously becomes HIGH due to noise etc., but the larger the probability 
that detection of true level varying of the signal fails. 
The level signal b is subsequently described. FIG. 8(b) shows a waveform of 
the level signal b generated by the level detector 2. The input signal a 
is inputted to the level detector 2 (refer to FIG. 8(a)), the amplitude of 
the input signal a becomes large at the head portion, i.e., for a period 
until the gain of the AGC amplifier converges. Therefore, the level signal 
b in FIG. 8(b) becomes HIGH for a short period of time at the head portion 
of the input signal a, and becomes LOW for other periods. 
FIG. 3 is a block diagram showing a detailed structure of the time constant 
control circuit 3 shown in FIG. 1. In FIG. 3, the time constant control 
circuit 3 includes a counter 31. The counter 31 is a counter counting up 
from 0 to the (n-1)-th power of 2 for a certain period of time, and is 
constructed so as to stop when the count value reaches the (n-1)th power 
of 2. In more detail, the counter 31 inputs the level signal b through a 
reset terminal. The counter 31 performs a counting operation when the 
level signal b is LOW, and when the level signal b is HIGH, the counter 31 
resets the counting operation, whereby the count value Q becomes 0. In 
this situation, the counter 31 outputs a significant m bit (m.ltoreq.n-1) 
of the count value Q as a time constant control signal c from a Q 
terminal. At the same time, the most significant bit (MSB) of the count 
value Q is inputted to an enable bar terminal (a bar is on the upper 
portion of ENABLE in FIG. 3 for description). The counter 31 stops the 
counting operation when the MSB of the (n-1)-th power of 2 is inputted to 
the enable bar terminal. As a typical circuit, If m=1, the size of the 
circuit becomes the smallest. In this case, the time constant control 
signal c is a binary signal. The time constant control signal c is a 
signal which controls a time for compensating the DC offset component. 
That is, the smaller a numerical value of a significant m bit outputted as 
the time constant control signal c, the shorter the time for compensating 
the DC offset component. The larger the numerical value of the significant 
m bit, the longer the time for compensating the DC offset component. When 
m.gtoreq.2, the time for compensating the DC offset can be finely 
controlled. 
FIG. 8(c) is a diagram showing the time constant control signal c (m=1). In 
FIG. 8(c), the time constant control signal c is a binary signal which is 
0 or 1. When the level signal b is HIGH, as is clear from the above, the 
counter 31 outputs 0 as the time constant control signal c. The counter 31 
counts up from 0 when the level signal b is LOW. Therefore, the numerical 
value of the significant m bit varies from 0 to 1 after a lapse of a 
certain period of time, and the time constant control signal c becomes 1. 
Therefore, the counter 31 continues to output 0 as the time constant 
control signal c before the lapse of the certain period of time. FIGS. 
8(d) and 8(e) respectively shows the time constant control signal c when 
m=2 and m&gt;&gt;1. Since each of methods of generating the time constant 
control signal c in each case is basically the same as that shown in FIG. 
8(a), the description is omitted. As shown in FIG. 8(d) or FIG. 8(e), when 
the time constant control signal c is monotonically increased from the 
minimum to the maximum stepwise or continuously, the time for compensating 
the DC offset can be finely controlled as described above. 
FIG. 4 is a block diagram showing the detailed structure of the estimator 4 
shown in FIG. 1. In FIG. 4, the estimator 4 includes a first integrator 
41, a comparator 42 and a second integrator 43. The compensation signal e 
is first integrated at the first integrator 41 to become a first 
integration output e1. The comparator 42 compares the first integration 
output e1 with a reference value (described later) defined by the time 
constant control signal c to obtain a comparison output e2. The second 
integrator 43 integrates the comparison output e2 to obtain the estimate 
d. The first integrator 41 is reset when the comparison output e2 is other 
than 0. 
FIG. 5 is a block diagram showing an example of a first structure of the 
first integrator 41 shown in FIG. 4. In FIG. 5, the first integrator 41 
includes an adder 411 and a register 412. The register 412 normally stores 
outputs from the adder 411, and resets its held value e0 to 0 when 
receiving a reset signal. The reset signal is a comparison output e2 
(described later) having a value other than 0 outputted by the comparator 
42. The adder 411 adds the compensation output e to the held value e0 of 
the register 412 to obtain the first integration output e1. Therefore, the 
first integrator 41 cumulatively adds the inputted signals during a period 
of not being reset by the comparison output e2. 
FIG. 6 is a block diagram showing a more detailed structure of the 
comparator 42 shown in FIG. 4. In FIG. 6, the comparator 42 includes a 
first selector 421, a second selector 422, a first comparator 423, a 
second comparator 424 and a synthesizing unit 425. The first selector 421 
selects one of prescribed constants U1 to Uj on the basis of the time 
constant control signal c, and takes the selected constant as an upper 
limit reference value U. The second selector 422 selects one of prescribed 
constants L1 to Lj on the basis of the time constant control signal c, and 
takes the selected constant as a lower limit reference value L. It is 
provided that j is the m-th power of 2. The upper limit reference value U 
is a positive number, and is set to be selected to be a larger value as 
the time constant control signal c is larger. The lower limit reference 
value L is a negative number, and is set to be selected to be a smaller 
value as the time constant control signal c is smaller. The first 
comparator 423 outputs a first comparison output c1 representing 1 when 
the first integration output e1 is not less than the upper limit reference 
value U, and outputs a first comparison output c1 representing 0 when 
otherwise. The second comparator 424 outputs a second comparison output c2 
representing -1 when the first integration output e1 is not more than the 
lower limit reference value L, and outputs a second comparison output c2 
representing 0 when otherwise. The synthesizing unit 425 outputs +1 as a 
comparison output e2 when the first comparator 423 outputs the first 
comparison output c1 representing 1. The synthesizing unit 425 outputs -1 
as a comparison output e2 when the second comparator 424 outputs the 
second comparison output c2 representing -1. Further, the synthesizing 
unit 425 outputs 0 as a comparison output e2 when the first and second 
comparators 423 and 424 output the first and second comparison outputs c1 
and c2 both representing 0. 
FIG. 7 is a block diagram showing an example of the first structure of the 
second integrator 43 shown in FIG. 4. In FIG. 7, the second integrator 43 
includes an adder 431 and a register 432. The second integrator 43 has a 
structure almost the same as that of the first integrator shown in FIG. 5, 
and cumulatively adds the comparison output e2 to obtain the estimate d. 
Through the above structure, in the estimator 4, the smaller the time 
constant control signal c, the smaller the upper limit reference value U, 
and conversely, the larger the lower limit reference value L. Moreover, 
the first integration output e1 has a value obtained by cumulatively 
adding the compensation outputs e from 0. Therefore, when the time 
constant control signal c is small, the first integration output e1 
reaches the upper limit reference value U or the lower limit reference 
value L rapidly. The synthesizing unit 425 thus has the high frequencies 
of outputting +1 or -1 as the comparison output e2. The second integrator 
43 cumulatively adds the comparison outputs e2 to obtain the estimate d. 
Therefore, when the frequencies of outputting +1 or -1 as the comparison 
output e2 becomes high, the second integrator 43 outputs the estimate d 
according to the frequencies. The compensator 1 thus compensates the DC 
offset component rapidly to output the compensation output e when the 
value of the time constant control signal c is small. 
FIG. 8(f) shows a waveform of the DC offset component included in the 
compensation output e. In FIG. 8(f), the DC offset component included in 
the compensation output e approaches 0 rapidly during a period where the 
time constant control signal c is small, as described above. On the other 
hand, during a period where the time constant control signal c is large, 
the DC offset compensation device decreases the DC offset component slowly 
in the vicinity of 0 and keeps it stable. The DC offset compensation 
device can thus perform fast DC offset compensation at the head portion of 
the input signal and stable DC offset compensation at other periods. 
The structure of the first integrator 41 shown in FIG. 4 is not limited to 
that shown in FIG. 5, but may be the one in the following. FIG. 9 is a 
block diagram showing an example of a second structure of the first 
integrator 41 shown in FIG. 4. In FIG. 9, the first integrator 41 is 
different from the structure shown in FIG. 5 in that a quantizer 413 is 
included at an inputting side of the adder 411. Since the other structure 
is the same as the structure for which the same reference numbers are 
provided in FIG. 5, the description is omitted. The quantizer 413 inputs 
the compensation output e, and outputs +1, -1, and 0 when the compensation 
output e is positive, negative, and 0, respectively, as a quantization 
signal q1. Since this structure makes operations of each portion set in 
the rear of the quantizer 413 independent of the amplitude of the 
compensation output e, while there is a disadvantage in the rapidity of 
the estimating operation with respect to a large DC offset compensation, 
compared with the structure of the first integrator 41 shown in FIG. 5, 
there is an advantage in that an erroneous operation due to an input 
signal with a large amplitude at the head portion of the signal or noise 
with other large amplitude hardly occurs. 
FIG. 10 is a block diagram showing an example of a third structure of the 
first integrator 41 shown in FIG. 4. In FIG. 10, the first integrator 41 
is different from the structure shown in FIG. 9 in that an up/down counter 
414 is included instead of the adder 411 and the register 412. Since the 
other structure is the same as the structure for which the same reference 
numbers are provided in FIG. 9, the description is omitted. That is, while 
being formed of the adder 411 and the register 412 in FIG. 9, the 
cumulative adder is formed of the up/down counter 414 in FIG. 10. The 
quantizer 413 outputs the quantization signal q1 as the same described 
above, and controls the counting direction of the up/down counter 414. In 
more detail, the up/down counter 414 performs up-count when the 
quantization signal q1 is +1, down-count when the quantization signal q1 
is -1, and stops counting when the quantization signal q1 is 0. When the 
comparison output e2 outputs other than 0, a held count value of the 
up/down counter is reset to 0. The first integrator 41 shown in FIG. 10 
thus operates the same as the first integrator 41 shown in FIG. 9. 
In the first integrator 41 shown in FIGS. 9 and 10, although the quantizer 
413 outputs 0 as the quantization signal q1 when the compensation output e 
is 0, since the probability that the compensation output e becomes 0 
during actual signal receiving is very small, the quantizer 413 may 
quantize the compensation output e having a value of 0 to +1 or -1. 
Furthermore, in the DC offset compensation device according to the first 
embodiment, the second integrator 43 shown in FIG. 4 is not limited to the 
structure shown in FIG. 7, but may have a structure shown in FIG. 11, for 
example. FIG. 11 is a block diagram showing an example of a second 
structure of the second integrator 43 shown in FIG. 4. In FIG. 11, the 
second integrator 43 includes an up/down counter 433. That is, while being 
formed of the adder 431 and the register 432 in FIG. 7, the cumulative 
adder is formed of the up/down counter 433 in FIG. 11. The up/down counter 
433 performs up-count when the comparison output e2 is +1, down-count when 
the comparison output e2 is -1, and stops counting when the comparison 
output e2 is 0. The second integrator 43 shown in FIG. 11 thus operates 
the same as the second integrator 43 shown in FIG. 7. 
Further, in the DC offset compensation device according to the first 
embodiment, the estimator 4 in FIG. 1 is not limited to the structure 
shown in FIG. 4, but may have the following structure, for example. FIG. 
12 is a block diagram showing an example of a second structure of the 
estimator 4 shown in FIG. 1. In FIG. 12, the estimator 4 includes a 
quantizer 401, a register 402, an adder/subtractor 403 and correction 
constant generator 404. The quantizer 401 quantizes the compensation 
output e inputted to the estimator 4 to ternary values of +1, 0 or 1 to 
output to the adder/subtractor 403 as a quantization signal q2. The 
adder/subtractor 403 changes the operation according to the quantization 
signal q2. Specifically, the adder/subtractor 403 adds a register output 
dd to a correction constant D to obtain the estimate d when the 
quantization signal q2 is +1, subtracts the correction constant D from the 
register output dd to obtain the estimate d when the quantization signal 
q2 is -1, and takes the register output dd as it is as the estimate d when 
the quantization signal q2 is 0. The register 402 holds the estimate d 
outputted from the adder/subtractor 403 at the immediately preceding 
sampling time to output the estimate d as the register output dd. That is, 
the adder/subtractor 403 and the register 402 performs cumulative 
addition/subtraction. The correction constant generator 404 is a selector 
which selects one of a plurality of predetermined positive constants 
according to the time constant control signal c. The correction constant 
generator 404 selects the correction constant D having a smaller value as 
the time constant control signal c is larger, and outputs the selected 
correction constant D to the adder/subtractor 403. 
FIG. 13 is a block diagram showing an example of a third structure of the 
estimator 4 shown in FIG. 1. In FIG. 13, the estimator 4 is different from 
the structure shown in FIG. 12 in that the adder 405 is included instead 
of the adder/subtractor 403 and that the correction constant generator 406 
is included instead of the correction constant generator 404. Since the 
other structure is the same as the structure for which the same reference 
numbers are provided in FIG. 12, the description is omitted. The 
correction constant generator 406 has a first selector 407, a second 
selector 408 and a third selector 409. The first selector 407 selects one 
of a plurality of predetermined positive constants DP1 to DPj (j is the 
m-th power of 2) according to the time constant control signal c, and 
outputs the selected constant to the third selector 409 as a first 
correction constant DP. The first selector 407 selects a constant having a 
larger value as the time constant control signal c is larger. The second 
selector 408 selects one of a plurality of predetermined negative 
constants DN1 to DNj (j is the m-th power of 2) according to the time 
constant control signal c, and outputs the selected constant to the third 
selector 409 as a second correction constant DN. The second selector 408 
selects a constant having a smaller value as the time constant control 
signal c is larger. The third selector 409 selects one among the first 
correction constant DP, the second correction constant DN and 0 according 
to the quantization signal q2, and outputs the selected constant to the 
adder 405 as a correction constant D2. Specifically, the third selector 
409 outputs the first correction constant DP as the correction constant D2 
when the quantization signal q2 is +1. The third selector 409 outputs the 
second correction constant DN as the correction constant D2 when the 
quantization signal q2 is -1. The third selector 409 outputs 0 as the 
correction constant D2 when the quantization signal q2 is 0. In this 
structure, the number of selectors disadvantageously increases compared 
with the structure shown in FIG. 12. However, since the positive 
correction constant DP or negative correction constant DN is added instead 
of the same correction constant D (refer to FIG. 12) being added or 
subtracted, it is advantageous that the adder can be used instead of the 
adder/subtractor and that the positive correction constant DP and negative 
correction constant DN can be separately set. 
(Second embodiment) 
FIG. 14 is a block diagram showing the structure of the DC offset 
compensation device according to a second embodiment of the present 
invention. The DC offset compensation device shown in FIG. 14 is different 
from the DC offset compensation device shown in FIG. 1 in that a level 
detector 20 is provided instead of the level detector 2. Since the other 
structure is the same as the structure for which the same reference 
numbers are provided in FIG. 1, the description is omitted. The level 
detector 20 is different from the level detector 2 shown in FIG. 1 in that 
the level detector 20 inputs the compensation output e to generate the 
level signal b according to the variation of the amplitude of the 
compensation output e. Although the level detector 20 may have the 
structure shown in FIG. 2, it may also have the following structure which 
is subsequently described. 
FIG. 15 is a block diagram showing the structure of the level detector 20 
shown in FIG. 14. The level detector 20 has a structure omitting the high 
pass filter 21, compared with the level detector 2 shown in FIG. 2. Since 
the other structure is the same as the structure for which the same 
reference numbers are provided in FIG. 2, the description is omitted. 
Since the level detector 20 uses the compensation output e as an input, 
the DC components are almost eliminated. It is thus possible to omit the 
high pass filter 21. 
(Third embodiment) 
FIG. 16 is a block diagram showing the structure of the DC offset 
compensation device according to a third embodiment of the present 
invention. Compared with the structure shown in FIG. 1, the DC offset 
compensation device in FIG. 16 is different in that it has a estimator 40 
instead of the estimator 4 in FIG. 1, and that the estimator 40 inputs the 
input signal a while the estimator 4 inputs the compensation output e in 
FIG. 1. Since the other structure is the same as the structure for which 
the same reference numbers are provided in FIG. 1, the description is 
omitted. While the DC offset compensation device shown in FIG. 1 forms a 
feedback control loop with the estimator 4 and the compensator 1, the DC 
offset compensation device shown in FIG. 16 forms a feed-forward control 
loop with the estimator 40 and the compensator 1. In this case, the 
estimator 40 generates an estimate d2 from the input signal a 
independently of the compensation output e from the compensator 1. The 
estimator 40 is formed of a low pass filter 400 whose gain with respect to 
the DC components is 1, and the low pass filter 400 can be realized in the 
following structure subsequently described. 
FIG. 17 is a block diagram showing the structure of the low pass filter 400 
shown in FIG. 16. In FIG. 17, the low pass filter 400 has a shift register 
41, selectors 42 to 46, multipliers 47 and adders 48. The shift register 
41, the multipliers 47 and the adders 48 form a transversal-type low pass 
filter. The characteristic of the filter is decided by a tap coefficient 
which is an output of the selectors 42 to 46. The selectors 42 to 46 
selects one of j pcs (j is the m-th power of 2) of the tap coefficients 
according to the time constant control signal c. These tap coefficients 
are defined so that the larger the time constant control signal c, the 
larger the time constant of the transversal filter (i.e., the lower a 
cut-off frequency). In this structure, the cut-off frequency is raised to 
estimate components in the vicinity of direct current of the input signal 
a when the time constant control signal c is large, while the cut-off 
frequency is lowered to stably estimate components in the vicinity of 
direct current of the input signal a when the time constant control signal 
c is small, and the estimated result is outputted as an estimate d2. 
(Fourth embodiment) 
The DC offset device according to a fourth embodiment of the present 
invention is different from the structure shown in FIG. 1 in that the 
level detector 2 has the structure shown in FIG. 18 instead of the 
structure shown in FIG. 2. Therefore, the illustration of the DC offset 
compensation device according to the fourth embodiment is omitted. 
Further, since the structure other than the above described difference 
from that of FIG. 1 is the same as the structure for which the same 
reference numbers are provided in FIG. 1, the description is omitted. 
The comparator 24 shown in FIG. 2 compares the smooth output a3 with the 
reference value having a fixed value. Therefore, the level detector 2 
shown in FIG. 2 cannot normally generate the level signal b until the 
smooth output a3 at the head portion of the burst-like input signal a 
becomes a sufficiently large value, compared with a state that the gain of 
the AGC amplifier converges. Specifically, at the time of no signal when 
the gain of the AGC amplifier is the maximum, i.e., when the smooth output 
a3 caused only by noise is very small compared with the smooth output a3 
after the gain of the AGC amplifier converges and when the burst-like 
input signal a has a small level such that, at the head portion of the 
input signal a, the smooth output a3 does not reach the fixed reference 
value, the level detector 2 shown in FIG. 2 cannot normally generate the 
level signal b even though an S/N ratio is sufficient, resulting in that 
the compensation of the DC offset component is not rapidly performed and 
the receiving performance is degraded. Therefore, the level detector 2 
shown in FIG. 18 includes a threshold generating circuit 27 and a delay 
unit 28. 
FIG. 18 is a block diagram showing a structure of the level detector 2 of 
the DC offset compensation device according to the embodiment. In FIG. 18, 
the level detector 2 is different from the structure shown in FIG. 2 in 
that the threshold generating circuit 27 and the delay unit 28 are 
included and that the delayed threshold a5 (refer to FIG. 20(d)) is 
inputted as a reference value by the comparator 24. Since the other 
structure is the same as the structure for which the same reference 
numbers are provided in FIG. 2, the description is omitted. 
The threshold generating circuit 27 further decreases time variation of the 
inputted smooth output a3 to generate a threshold a4, and is typically 
realized by a low pass filter or an integrator. FIG. 19 is a block diagram 
showing an example of the structure of the threshold generating circuit 27 
shown in FIG. 18. In FIG. 19, the threshold generating circuit 27 has a 
gain adjusting constant multiplier 271, an adder 272, a register 273 and 
an oblivion constant multiplier 274. The gain adjusting constant 
multiplier 271 multiplies the inputted smooth output a3 by a predetermined 
gain adjusting constant G to output a gain adjusting multiplier output 
a3'. The register 273 holds a threshold a4 outputted from the adder 272 at 
the immediately preceding sampling time to output a register output a4'. 
The oblivion constant multiplier 274 multiplies an oblivion constant 
.omega. (0&lt;.omega.&lt;1) which is a predetermined constant by the register 
output a4' outputted from the register 273. The adder 272 adds the gain 
adjusting multiplier output a3' outputted from the gain adjusting 
multiplier 274 to the register output a4' multiplied by the oblivion 
constant .omega. to obtain a new threshold a4. The gain with respect to 
the DC components of the threshold generating circuit 27 described above 
(=G/(1-.omega.)) is set to not less than 1 (preferably about 1.5 to 3 
times). That is, the threshold a4 is obtained by multiplying the 
time-average of the latest components of the smooth output a3 by the gain 
with respect to the DC components. The effective average time of the 
threshold generating circuit 27, i.e., the time constant, is preferably 
several to tens of times of the symbol time of the input signal a (refer 
to FIG. 20(a)). 
When the threshold a4 generated as described above is inputted as it is to 
the comparator 24 as a reference value, the reference value itself varies 
at the time of level variation at the head portion of the input signal a. 
The reference value thus increases during the time of storing the 
judgement output b0 (hereinafter referred to as a judging time) at the 
shift register 25. Therefore, the comparator 24 fails to detect the level 
variation. The reference value should be generated on the basis of the 
time average value of the smooth output a3 only by noise from inputting 
time of the input signal a at least until the judging time. Therefore, the 
delay unit 28 is set in the rear of the threshold generating circuit 27. 
The delay unit 28 delays the inputted threshold a4 for at least the 
judging time and outputs a delayed threshold a5 to the comparator 24. 
Therefore, the waveforms in the main portions of the level detector 2 shown 
in FIG. 18 are shown in FIGS. 20(a) to 20(d). FIG. 20(a) shows an input 
signal a to be inputted to the DC offset compensation device according to 
the fourth embodiment. In FIG. 20(a), the input signal a has a sufficient 
S/N ratio as described above, but is a burst-like signal, at whose head 
portion the smooth output a3 has a small level. FIG. 20(b) shows the 
smooth output a3 generated on the basis of the input signal a shown in 
FIG. 20(a). In FIG. 20(b), since the smooth output a3 is generated as 
described in the first embodiment, the description is omitted. FIG. 20(c) 
shows a threshold a4 generated on the basis of the smooth output a3 shown 
in FIG. 20(b). In FIG. 20(c), as described above, the threshold a4 is 
obtained by further reducing the time varying of the smooth output a3, and 
is multiplied by the above described gain (=G/(1-.omega.)). FIG. 20(d) 
shows the smooth output a3 inputted to the comparator 24 shown in FIG. 18 
and the delayed threshold a5. In FIG. 20(d), the delayed threshold a5 is 
obtained by delaying the threshold a4 for at least the judging time. 
Therefore, since the comparator 24 compares the smooth output a3 generated 
according to input of the input signal a with the delayed threshold a5 
generated only on the basis of noise at the head portion of the burst-like 
input signal a, the level varying of the head portion of the input signal 
a can be precisely detected to generate the level signal b. 
As described above, the level detector 2 shown in FIG. 18 can generate the 
level signal b by precisely detecting the level varying at the head 
portion of the burst-like input signal a, compared with the level detector 
2 shown in FIG. 2. In the level detector 2 in FIG. 18, however, since the 
reference value itself varies by variation of noise and the input signal a 
compared with the level detector 2 shown in FIG. 2, the detection accuracy 
of the head portion of the burst-like signal increases, while there is a 
fear of generating an erroneous level signal b at the other portions. 
In FIG. 18, when the shift register 25 and the AND circuit 26 are omitted, 
i.e., the judging time is 0, the delay unit 28 can be omitted and the 
threshold a4 is directly taken as a reference value of the comparator 24. 
Further, the structure of level detector 2 can be applied to not only the 
level detector 2 in FIG. 1 but also the level detector in FIG. 16. 
In each of the first to fourth embodiments described above, the input 
signal a may be an analog signal, and may also be a digital signal 
obtained by A/D converting an analog signal. Further, while the input 
signal a has been described assuming that it is an output of the AGC 
amplifier, it may be a detecting signal detecting an output of the AGC 
amplifier. For example, base band signals I, Q generated by quadrature 
demodulation shown in FIG. 10 of Japanese Patent Laying-Open No. 8-032383 
may be used as the input signal a. When the base band signal obtained by 
such detector is an object, if an oscillation signal of a local oscillator 
is leaked to be mixed into the input of the AGC amplifier to cause 
interference, or if an interference signal having the same frequency as 
that of a desired signal is simultaneously received to cause interference, 
the interference appears in the base band signal as the DC offset 
components. Therefore, in these cases, the DC offset compensation device 
also has the effect of reducing interference of an undesired signal. 
While the invention has been described in detail, the foregoing description 
is in all aspects illustrative and not restrictive. It is understood that 
numerous other modifications and variations can be devised without 
departing from the scope of the invention.