AGC circuit particularly for a hearing aid

A high fidelity hearing aid is disclosed for providing high quality sound and is primarily directed to those users whose hearing loss is such that they need some amplification for low level signals, but do not need amplification for high level signals.

BACKGROUND OF THE INVENTION AND STATEMENT OF PRIOR ART 
Certain users of hearing aids have a mild to moderate loss of normal 
sensitivity for low intensity sounds, but retain excellent speech 
discrimination capabilities and good sound quality judgment capabilities 
for sounds which are intense enough to be audible to them. This phenomena, 
commonly referred to as loudness recruitment was, for example, discussed 
by Steinberg and Gardner in "The Dependence of Hearing Impairment on Sound 
Intensity", Journal of the Acoustical Society of America, Vol. 9, Pages 
11-23 (1937). Accordingly, those hearing aid users who have a hearing 
impairment characterized by a mild to moderate loss of sensitivity for low 
intensity sounds require no amplification for high intensity sounds; and, 
amplification of high intensity sounds will generally cause them 
discomfort. This problem is well understood in the art, and a variety of 
solutions have heretofore been employed. However, most of the prior art 
solutions introduce deterioration in the quality of the delivered sound, 
so that traditional hearing aids have often been characterized as "low 
fidelity instruments". 
One prior art approach has been to limit the saturation sound pressure 
output of the hearing aid by means fo a simple peak clipping, which 
produces large amounts of harmonic and intermodulation distortion. More 
recently, relatively low distortion compression limiting has been used to 
limit the saturation sound pressure output of the hearing aid without the 
introduction of large amounts of harmonic or intermodulation distortion. 
With either type of output limiting, the saturation sound pressure level 
out of the hearing aid can be set to a level somewhat below the discomfort 
level of the user. A typical input-output curve for an output limiting 
hearing aid is shown in FIG. 4. One disadvantage to the output-limiting 
approach is that a large percentage of the incoming sounds will be 
amplified to a point which is near the user's discomfort level. 
Another prior art approach has been to use an automatic volume control 
(AVC) circuit which acts to produce a nearly constant output level for a 
wide range of input sound pressure levels. In this case, the nearly 
constant output level can be set to the user's most comfortable listening 
level, so that the majority of useful sounds are presented at the most 
comfortable listening level for the user. A typical input-output curve for 
an automatic volume control hearing aid is shown in FIG. 4. A problem with 
this latter approach is that the normal dynamic range of the desired 
sounds is unduly reduced at the output, so that quiet input sounds and 
background noises are amplified until they appear nearly as loud to the 
user as loud input sounds. This excessive reduction of the dynamic range 
lends an unnatural quality to the sound. 
A more sophisticated solution provided by the prior art has been the use of 
wide dynamic range compression amplification wherein the overall gain of 
the hearing aid is changed smoothly and continuously from a maximum value 
for low level inputs to a minimum value for high level inputs. Compression 
hearing aid amplifiers are commercially available which have a nearly 
constant compression ratio of 2:1 over an input dynamic range of 60 dB or 
more. With a 2:1 compression ratio, for example, such 10 dB increase in 
input level causes only a 5 dB increase in output level. An early version 
of such a compression amplifier suitable for hearing aids was described in 
U.S. Pat. No. 3,229,049, issued to Hyman Goldberg on Jan. 11, 1966. A 
typical input-output curve for prior art logarithmic compression hearing 
aids is shown in FIG. 4. A drawback to such prior art devices, however, is 
that they provide a compression function characterized by an essentially 
constant compression ratio for all sound levels up to those sufficiently 
intense to cause overload of the input circuit, which commonly occurs at 
microphone input levels corresponding to a hearing level of approximately 
80 dB (input sound pressure levels of 90 to 95 dB). For sounds whose sound 
pressure level exceeds 90 to 95 dB, the overload of the input circuitry 
causes a rapid increase in amplifier distortion. Since sounds whose peak 
levels exceed 90 to 95 dB are commonly encountered at concerts, social 
gatherings, etc., the high distortion levels of such prior art amplifiers 
renders them unsuitable for use in a high fidelity hearing aid. 
Also high quality compression circuits or systems for use in broadcast and 
recording studio applications are known in the art, see, for example, U.S. 
Pat. Nos. 3,681,618 and 3,714,462, both to David Blackmer. In these latter 
systems, the compression ratio and the lower and upper threshold of 
compression can be readily adjusted by the user to optimize the overall 
sound quality with different program materials. Such prior art systems 
have generally required power supply voltages and currents which would be 
impractical in a head-worn hearing aid; more specifically, power supply 
requirements of plus and minus 12 volts at a total current drain of 10 mA 
or more is not uncommon for such prior art systems. 
Accordingly, it is a principal object of the present invention to provide a 
high fidelity hearing aid. 
It is another object of the present invention to provide a very low noise, 
low distortion, variable gain preamplifier capable of handling the 
unattenuated output of a subminiature microphone without distortion for 
input sound pressure levels of 110 dB SPL or greater. 
It is another object of the present invention to provide a low distortion, 
low noise, variable compression ratio circuit which requires a supply 
voltage of 1.5 volts, or less, and a supply current of approximately 100 
microamperes, or less. 
It is a further object of the present invention to provide a preamplifier 
whose gain in dB is linearly related to the logarithm of the output signal 
voltage over an input range of at least 40 dB. 
It is yet another object of the present invention to provide a preamplifier 
circuit with the compression ratio and the maximum gain of the 
preamplifier adjustable by the user in a simple manner without affecting 
the ability of the circuit to provide low distortion operation. 
A further object of the present invention is to provide a circuit for 
obtaining wide dynamic range logarithmic compression acting to increase 
circuit gain for low level signals, in combination with fast-acting, low 
distortion output limiting compression acting to attenuate the microphone 
output when output SPL's exceed some predetermined level, and wherein the 
two separate compression actions do not interact in an undesirable manner. 
These and other such objects as may hereinafter appear are attained in the 
embodiments of the invention shown in the accompanying drawings wherein:

THEORETICAL CONSIDERATIONS 
As discussed above, many users with a mild to moderate hearing loss 
experience essentially normal loudness sensation for sounds above a 
certain intensity level. Examination of clinical data and the literature 
indicates that recruitment is typically complete, that is, the loudness 
sensation is essentially normal, for sounds corresponding to a hearing 
level of 80 dB or greater. Consider that for speech sounds, an 80 dB 
hearing level corresponds to a sound pressure level (SPL) in the sound 
field of approximately 95 dB; and for pure tones, 80 dB hearing level 
corresponds to approximately 80 to 90 dB SPL in the sound field in the 
frequency range important for speech perception. 
For a minimum degradation in the perceived sound quality, a high fidelity 
hearing aid should provide sensibly unity acoustical gain over that range 
of sound intensities in which the user has essentially normal loudness 
sensations. An ideal high fidelity hearing aid should thus not overload 
for input sounds above 90 dB SPL, but should produce unity acoustical gain 
up to the 105 to 115 dB peak sound pressure levels commonly encountered at 
concerts, social gatherings, etc. 
Consider now the low level input-output characteristics required of a high 
fidelity hearing aid. A gain numerically equal to the user's hearing loss 
would be required to restore his threshold down to audiometric zero 
levels; however, such a large amount of gain is commonly found to be 
unacceptable. A reason for the foregoing is that under most circumstances, 
the masking produced by the background noise levels commonly encountered 
in residences, offices, etc. render even those persons with unusually 
acute hearing incapable of detecting sounds which are less than 15 to 30 
dB above commonly accepted audiometric zero levels. Similarly, the upper 
limit of what is commonly accepted as the range of normal hearing is set 
at 25 dB above audiometric zero levels. Thus a maximum amplification 
sufficient to reduce the hearing loss to 15 to 20 dB has been generally 
found appropriate. 
It is readily seen, therefore, that a high fidelity hearing aid designed 
for minimum degradation in perceived sound quality should typically have a 
gain-versus-input level which is characterized by a maximum gain which is 
15 to 20 dB less than the hearing loss, and is constant for inputs below a 
15 to 20 dB hearing levels. Further, such a high fidelity hearing aid 
should have a gain which decreases smoothly as the input is increased in 
the range from the 20 dB to 80 dB hearing levels, and has a constant gain 
of unity (0 dB acoustic amplification) for sounds above the 80 dB hearing 
level. In practice, it is also generally desirable for the hearing aid to 
incorporate fast-acting low distortion compression limiting applied to the 
microphone output for sounds above the 100 dB hearing level (110 to 115 dB 
SPL). This latter function not only prevents occasional uncomfortably loud 
sounds from reaching the user, but also functions to avoid any distortion 
or bias shifts in the amplifier during such sound extremes. An example of 
the desired gain-versus-input level characteristic discussed above is 
illustrated in FIG. 4 for an individual with a 45 dB hearing loss. The 
inflections of "knees" occurring at points B and C in the curve of FIG. 4 
will hereafter be respectively referred to as the lower and upper 
thresholds of compression. 
In a hearing aid, a variable compression ratio control is also desirable so 
that one basic hearing aid design can cover a wide variety of hearing 
losses in the mild to moderate range and yet, in each individual case, 
provide the minimum compression ratio required in order for the user to 
have adequate gain for very low level signals. Moreover, if controls for 
adjustment of compression ratio are made available, the user can reduce 
the compression ratio and thus the gain for moderate-level signals when he 
finds himself in a noisy environment so the environmental noises are not 
amplified annoyingly. Being able to chose the minimum compression ratio 
appropriate to each situation is desirable because it generally provides 
the best overall sound quality. 
Experiments using commonly employed speech tests, for example, have shown 
that 2:1 compression ratio with the characteristics shown in FIG. 4 will 
provide adequate gain to compensate for a 45 dB hearing loss, and will 
reduce the 45 dB hearing loss down to approximately 15 dB, which is 
essentially in the normal range. At the same time, such a 2:1 compression 
ratio is low enough so that its operation is frequently unobtrusive enough 
to be nearly inaudible to the untrained listener with normal hearing. For 
example, in the course of conducting various experiments it was found that 
when the inventive circuit is inserted in the amplification chain of a 
home high fidelity system, it makes it possible to enjoy wide dynamic 
range classical recorded music at comfortable listening levels without the 
annoyance created when portions of the music drop below audibility due to 
the masking produced by the noise of household appliances. With the 
appliances turned off, moreover, casual listeners who have been asked to 
judge the overall sound quality of the system have, with rare exception, 
not even noticed the presence of the signal processing provided by the 2:1 
compression circuitry. 
The applicant believes that the combination of the desirable features 
discussed above in a low power, low noise circuit suitable for use in 
head-worn hearing aids is new in the art. More particularly, the inventive 
circuit provides low distortion, low noise, variable compression ratio 
operation while requiring a supply voltage of only 1.5 v or less and 
consuming approximately 100 microamperes of supply current. The circuit 
disclosed herein provides less than a 1.5 micro volt input noise level 
with various setting of the compression ratio including a setting of 1:1 
which corresponds to a fixed gain or unity gain condition, while 
introducting a total harmonic distortion of 2%, or less, for any input 
signal voltage between 1 micro volt and 60 millivolts rms and for any 
compression ratio setting between 1:1 and 3:1. The present circuits 
produce a total harmonic distortion below 5% even with an input signal 
voltage as high as 100 millivolts. Thus, it is seen that the present 
circuit provides an undistorted input dynamic range, that is, the range 
between the input noise level and the input overload level, of nearly 100 
dB. 
DESCRIPTION OF THE INVENTION 
FIG. 1 is a view partly in cross section of a high fidelity hearing aid 10 
in accordance with the invention. As will be appreciated, the structure of 
the high fidelity hearing aid 10 may include the usual microphone, 
receiver and battery mounted in the hearing aid housing. The hearing aid 
10 may include acoustic damping elements in the conduit coupling the 
output of the receiver to the ear mold and thence to the ear of the user. 
High fidelity hearing aid 10 also includes complete amplifier 12 as shown 
in FIG. 2, which includes the variable compression ratio amplifier 11 in 
accordance with the invention which is shown in more detail in FIG. 3. 
FIG. 2 shows the complete amplifier circuitry 12 for adapting or 
interfacing the inventive circuit with the microphone and receiver of the 
high fidelity hearing aid 10, as will be described more fully hereinbelow. 
Refer now to FIG. 3 which shows a schematic diagram of a preferred 
embodiment of the invention. A listing of the values of the various 
components or elements of the circuit 11 of FIG. 3 follows the description 
of the operation of the circuit. 
In FIG. 3, a signal e.sub.in as from a microphone is coupled through a 
series connected capacitor C2 to an amplifier 30 comprising transistors Q1 
and Q2. An important feature of circuit 11 is the feedback network 
provided by resistors R6 and R7 and transistor Q4 connected in an inverted 
configuration to provide an impedance in the feedback circuit configured 
as a voltage controlled resistance (VCR) element. 
A brief explanation of the function of the VCR in the inventive circuit 11 
is now in order. In the desired range of operation, the AC resistance of 
the VCR transistor goes from a few hundred ohms to over 1 megohm and the 
resistance is an approximately inverse exponential function of the control 
voltage. A small signal AC resistance of about 100,000 ohms associated 
with the DC control voltage of 400 mV, with a 10-fold decrease in AC 
resistance resulting from each 60 mV increase in control voltage, is 
typical of some devices. This predictable inverse exponential control 
characteristic allows a simple implementation of a logarithmic compression 
gain control function, as is well known. 
In the circuit of FIG. 3, the VCR transistor is placed in shunt with 
resistor R7 for AC currents by means of the coupling provided by low 
impedance capacitor C3. When the AC resistance of the VCR is high (1 
megohm for example), its shunting effect on R7 will be minimal and so the 
gain of amplifier 30 will be determined by the values of feedback 
resistors R6 and R7. In the preferred embodiment, R6 and R7 are equal 
valued resistances, and the voltage gain of amplifier 30 is approximately 
6 dB under these conditions. When the AC resistance of the VCR is low (a 
few hundred ohms, for example), on the other hand, its shunting effect on 
R7 will be maximal and the gain of amplifier 30 will be maximum. In the 
preferred embodiment, a maximum gain for amplifier 30 of about 46 dB is 
obtained with a control voltage of approximately 600 mV applied to the VCR 
transistor Q4. 
It has been found that the nonlinearity of such a VCR element tends to 
introduce undesirable distortion as the AC voltage is increased, see FIG. 
6. With a 1.5 mV rms AC sine wave, for example, a 1.5% total harmonic 
distortion results, which is about the upper limit for a high fidelity 
instrument. Accordingly, to maintain a low distortion, the AC signal could 
be limited to a maximum 1.5 millivolts across the voltage controlled 
resistance (VCR) when the VCR is a controlling resistance in the circuit 
operation. A disadvantage to this approach in a hearing aid, is that if 
the microphone output level is attenuated by the 32 dB required to bring 
the 60 mV (110 dB SPL equivalent) levels down to 1.5 mV, the intrinsic 
noise level of the circuitry will become objectionable with low level 
inputs. It is, therefore, not advisable to substantially lower the 
microphone level and subsequently amplify it after coupling to the VCR. In 
order to redude the deleterious effect of preamplifier noise levels, it is 
desirable to obtain the highest possible signal voltage levels out of the 
microphone, but in order to reduce the deleterious effect of nonlinear 
distortion at high levels, it is desirable to attenuate the signal voltage 
levels applied to the VCR. 
The inventive circuit 11 provides a solution to the foregoing problem by 
biasing the nonlinear voltage controlled resistance (VCR) element into a 
sensibly nonconducting state under high input signal level conditions, and 
bringing the VCR into effective operation only under low level input 
conditions. Thus under high input signal voltage conditions, where the 
nonlinearity of the VCR transistor Q4 becomes evident, only a negligable 
portion of the signal current flows through VCR transistor Q4 while the 
main port of the signal current flows through (linear) resistor R7. Thus, 
the large signal nonlinearity of the VCR has only a negligable effect on 
the operation of amplifier 30. Furthermore, a maximum of negative feedback 
is applied around amplifier 30 under high signal level conditions so as to 
further minimize the effect of the nonlinearities in the amplifier 
transistors Q1 and Q2. Under low input signal voltage conditions, on the 
other hand, the operation of transistors Q1 and Q2 and the AC resistance 
of VCR transistors Q4 are all sensibly linear, so that undistorted 
operation of amplifier 30 is obtained even under maximum gain conditions 
(i.e., even when the shunting effect of transistor Q4 on feedback resistor 
R7 is maximal). With this approach, an undistorted dynamic range 
improvement of 30 dB or more may be obtained over that obtained when the 
VCR remains the controlling element at all signal levels as shown in FIG. 
6. Note in FIG. 6 that the distortion of circuit 11 is 1% or less for AC 
inputs up to 30 mV, and less than 5% with 100 mV inputs. Careful listening 
tests indicate that the amount of harmonic distortion introduced by the 
operation of circuit 11 (and illustrated in FIG. 6) is undetectable, even 
to a trained listener, under most listening conditions. 
Return now to the description of the circuit of FIG. 3. The signal from 
amplifier 30 is coupled from transistor Q2 to an AC buffer amplifier 32 
comprising transistor Q3 and resistor R8. The emitter of transistor Q3 is 
coupled through capacitor C4 and resistor R9 to the base of transistor Q5 
of logarithmic amplifier 34 to provide a low impedance drive voltage for 
the amplifier 34. The logarithmic amplifier 34 includes transistors Q13, 
Q5, Q6, Q7 and diodes D1, D2 and D3. The emitter of transistor Q3 also 
couples an output to the amplifier of the associated hearing aid or other 
device as indicated. 
The logarithmic AC amplifier 34 has a very high input impedance and roughly 
60 dB of open loop gain. The diode D1 connected in parallel with diodes D2 
and D3 and resistor R13 functions as a nonlinear feedback impedance which 
in cooperation with the input resistor R9 provides a closed loop gain 
which is related to the input signal voltage e.sub.in in such a way that 
the peak voltage at the collector of Q7 is proportional to the logarithm 
of the peak AC voltage e.sub.k developed at the junction 14 of the 
capacitor C4 and resistor R9. The voltage at the collector of Q7 will have 
a DC resting value of roughly 850 mV DC with an AC input of 0.3 mV or less 
and a negative going AC peak value which increases logarithmically at a 
rate of approximately 220 mV for each factor of ten increases in input 
signal voltage e.sub.k. 
With low values of resistor R13, this nearly exact logarithmic relationship 
exists over a 60 dB range of AC input voltages e.sub.k extending from a 
0.3 mV to 300 mV. With higher values of resistor R13, a deviation toward 
linear operation is obtained at higher input levels. The foregoing 
deviation makes possible a more rapid decrease in control voltage to the 
VCR once the upper threshold of compression has been reached in order that 
the nonlinear VCR will have neglible effect on the operation of the 
circuit 11 at high signal levels. 
The sum of the base emitter offset voltages of Q5 and Q7 produces a 
relatively high DC input voltage so that two series diodes (D2 and D3) 
rather than one may be used in the feedback loop without incurring the 
undesirable negative voltage clipping which would otherwise occur due to 
saturation of transistor Q7. The high sensitivity of the logarithmic 
amplifier 34 in turn allows a relatively high compression ratio without 
the need for additional amplification of the control voltage. A wide range 
of operation combined with a high (220 mV/decade) sensitivity of the 
logarithmic amplifier 34 is thus made possible for operation with a 1.3 to 
1.6 volt supply. 
Diode D1 periodically applies an opposite polarity feedback current to that 
which periodically flows through diodes D2 and D3 in order to prevent a 
net DC current flow through capacitor C4 which would otherwise produce an 
undesirable shift in operating bias levels of the logarithmic AC amplifier 
as the input signal changes from a low to a high level. 
Transistor Q8, resistor R17 and capacitor C5 provide a peak detection 
rectifier 35. Transistor Q8 is used in order to provide a high input 
impedance so the rectifier 35 does not load down the output of the 
logarithmic AC amplifier 34. Resistor R17 and capacitor C5 determine the 
attack and release time of the circuit 11 which, in the embodiment shown, 
has an attack time of 3 milliseconds and a release time of about 50 
milliseconds. 
The compression ratio control potentiometer R19 is adjustable to couple a 
selected portion of the output voltage from the emitter of Q8 in rectifier 
circuit 35 to the DC buffer amplifier 36 whose output in turn provides the 
feedback control voltage applied to VCR transistor Q4. The lower terminal 
of potentiometer R19 is connected to the junction of resistor R18 and 
transistor Q11. Resistor R18 has its other terminal connected to supply 
potential, and transistor Q11 is connected through transistor Q12 to 
ground reference. Accordingly, resistor R19 may be adjusted so that a 
reference voltage VR, which is sensibly independent of the rectifier 
output voltage VC at the emitter of transistor Q8 is coupled through the 
buffer amplifier 36 to transistor Q4. The reference voltage VR is adjusted 
to be equal to the DC voltage obtained at the rectifier output when the 
input signal voltage is equal to the upper compression threshold value. 
Transistors Q11 and Q12 are preferably chosen to match the characteristics 
of transistors Q9 and transistor Q4 so that a temperature independent 
operation may be obtained. 
The DC buffer amplifier 36 is formed by transistors Q9 and Q10 and the 
associated resistors provide a high input impedance, low output impedance, 
sensibly unity gain amplifier. Buffer amplifier 36 has a DC offset voltage 
which compensates for the DC offset voltage introduced by rectifier 35. 
The proper level of the DC control voltage VC is obtained by choice of 
resistor R16, which can also easily compensate for small differences in 
these offset voltages. 
With the circuit valves listed hereinafter, the circuit of FIG. 3 will 
provide a maximum compression ratio of nearly 5:1. If a maximum 
compression ratio of only 3:1 is required, transistor Q10 and resistor R21 
may be eliminated from the circuit, and buffer amplifier 36 will have 
noticeably less than unity gain, thus reducing the maximum compression 
ratio available in the circuit. Similarly, D1 D2 and D3 may be 
nongold-doped diodes such as formed by transistors Q11 and Q12 (connected 
as diodes) when a maximum compression ratio of less than 5:1 is 
acceptable; in such case it has been found desirable to insert a series 
resistor at the point labeled RX adjacent resistor R13. 
In general, choosing a lower value for the upper threshold of compression 
will result in reduced distortion at all signal levels. Lowering the upper 
threshold of compression in the circuit of FIG. 3 or FIG. 7 may be 
accomplished most easily by reducing the value of R9 and/or increasing the 
value of R16, followed by appropriate selection of R13 and R18. 
Conversely, increasing the upper threshold of compression may be 
accomplished, at the expense of increased distortion levels, by increasing 
the value of R9 and/or reducing the value of R16. 
As mentioned above, transistor Q4 is connected as a voltage controlled 
resistance VCR. The control voltage from buffer amplifier 36 is impressed 
across the base to collector junction of transistor Q4. The offset voltage 
developed across the emitter collector junction in this inverted 
connection is in the fractional millivolt region and is sensibly 
unchanging over the range of control voltages which is required to give 40 
dB of gain control. An advantage of the low and sensibly unchanging offset 
voltage is that the control voltage is not mixed in with the signal 
voltage, so that as the control voltage is changed, it does not introduce 
any unwanted AC signal resulting in audible "clicks" or "thumps" in the 
output. 
The circuit of FIG. 2 also shows an optional FET Q0 in shunt with the 
microphone output terminal. Note that resistor R2 may not be required in 
those cases where the microphone output impedance is itself high enough. 
The FET Q0 can be used to attenuate the microphone output when the sound 
level input to the microphone is so high that the linear output 
capabilities of the amplifier would otherwise be exceeded or the sound 
pressure level delivered to the user would otherwise be uncomfortable. For 
example, with a typical hearing aid receiver driven from a Class A output 
stange, the upper limit of linear operations of the Class A output stange 
may correspond to 110 dB SPL. Moreover, low current drain hearing aids 
often reproduce an undistorted output only up to 105 dB SPL or less. Even 
though the circuit of FIG. 2 will handle up to the equivalent of 115 dB 
SPL input, therefore, the associated output amplifier of the hearing aid 
would distort the signal under those input SPL conditions unless some 
means such as Q0 were provided to reduce the signal reaching the output 
amplifier. The AC voltage developed at the amplifier output may be sensed 
(by means such as shown in FIG. 2), and used to control the AC resistance 
of Q0 in such a manner as to prevent the above-described difficulties of 
output amplifier distortion and/or user discomfort. Since the input levels 
at which such operation will occur are substantially above the upper 
threshold of compression for the logarithmic compression circuit, 
undesirable interaction between the two circuit functions is avoided. 
The value of capacitor C3 given hereinbelow is chosen so that the frequency 
response of the preamplifier will be sensible independent of gain over the 
audio band. By choosing a smaller value for C3, the frequency response 
will be dependent upon the input signal levels such that the gain and 
frequency response will be related to the input signal voltage in a 
continuous manner over a wide range of output levels due to the wide range 
logarithmic compression operation of the preamplifier. For example, in the 
circuit shown it has been found that a value for C3 which is 0.05 uf will 
often provide improved discrimination of speech signals in a background of 
noise, expecially when the noise has strong low frequency components. 
Capacitor C3 could be selected, or switched in and out of the circuit, to 
provide best sound quality for music, or best discrimination as for 
speech. Such level dependent variation of the frequency response is known 
in the art such as shown in U.S. Pat. No. 3,764,745 issued to Bottcher and 
Heyne; however, the combination with an amplifier providing automatic gain 
control for inputs between a fixed lower threshold and a fixed upper 
threshold, with low distortion constant gain operation for inputs above 
and below the threshold, has been hitherto unknown in the hearing aid art. 
Capacitor C4 may also be chosen to provide a frequency dependent operation 
of the VCR control voltage circuit with again the advantage that the 
operation is continuous over a wide output dynamic range due to the wide 
range logarithmic compression characteristic of the present circuit. 
The input-output characteristics obtained with the circuit of FIG. 3 is 
shown in FIG. 5 for various settings of the compression ratio control 
potentiometer R19. Note in FIG. 5 that the lower threshold of compression, 
which occurs at an input signal voltage of approximately 10 microvolts, 
and the upper threshold of compression, which occurs at an input signal 
voltage of approximately 10 millivolts, are both sensibly unaffected by 
the setting of the compression ratio control potentiometer. Moreover, the 
maximum gain provided by the circuit 11 is automatically determined by the 
setting of the compression ratio control potentiometer so that a high 
compression ratio is accompanied by a high maximum preamplifier gain, and 
a low compression ratio is accompanied by a low maximum preamplifier gain. 
In a hearing aid with commonly available electret or subminiature 
microphones, the lower threshold of compression corresponds to a hearing 
level of approximately 20 dB, while the upper threshold of compression 
corresponds to a hearing level of approximately 80 dB, see FIG. 4. Thus, 
it is seen from an analysis of FIGS. 4 and 5, that when used in a hearing 
aid, the circuit 11 can provide a constant maximum gain for sounds below 
approximately 20 dB hearing level, which sounds are typically composed 
largely of background noises; a smoothly decreasing gain with increasing 
sound intensity up to sounds corresponding to an 80 dB hearing level, and 
then a constant gain which may be chosen as unity acoustical gain by the 
proper selection of output amplifiers in the hearing aid, for sounds above 
80 dB hearing level. Applicant believes that this highly desirable 
operation has hitherto been unavailable in any head-worn hearing aid. 
FIG. 2 shows an application or connection of the inventive circuit in an 
overall hearing aid circuit. Resistor RB is chosen to produce the desired 
DC bias in the output transistor QT. Resistor RE may be chosen to produce 
the desired overall hearing aid gain levels; for example, RE may be chosen 
to produce unity acoustical gain in the hearing aid for sounds above 
approximately 80 dB hearing level, as discussed above. Resistor RF and 
capacitor CF are chosen to provide the desired output limiting as 
discussed above. Capacitor CR is used to control the high frequency 
response of the hearing aid and to insure amplifier stability. User 
operated potentiometer R19 controls the compression ratio in the 20 dB to 
80 dB range of input hearing levels, as described above. An optional 
resistor R23 as shown in FIG. 3 may be used to limit the maximum gain of 
amplifier 11 by placing a resistance in series with VCR transistor Q4. A 
similar result may be obtained by increasing the value of resistor R20, 
thereby limiting the maximum control voltage available to VCR transistor 
Q4. 
The circuit of FIG. 7 is a simplified version of the circuit of FIG. 3. In 
FIG. 7, a compression ratio of approximately 2:1 is all that is desired. 
Accordingly, transistor Q10, Q11 and Q12, and resistors R19 and R21 have 
been removed. Diode D3 has been moved from a position within the feedback 
loop as shown in FIG. 3, to a position where it provides a DC offset 
voltage in the output of the logarithmic AC amplifier. Resistors R15 and 
R16 are connected to form a voltage divider, with the value of resistor 
R16 chosen to produce the desired upper threshold of compression. A switch 
SW is provided and with switch SW open, resistor R23 functions somewhat 
like a volume control in the sense that it controls the maximum gain for 
low level signals. Its operation is illustrated in the curves of FIG. 8. 
With switch SW closed, the VCR is bypassed, so that the preamplifier 11 
functions as a conventional linear amplifier with gain determined by the 
value of R23. It is thus possible to readily change back and forth from a 
conventional linear hearing aid to a wide range compression hearing aid in 
order to determine user preference for one type or the other in different 
listening situations. 
It will be readily appreciated that in addition to application to hearing 
aids, the present circuitry will be useful in other applications where a 
large undistorted input dynamic range and a reduced output dynamic range 
are desirable. The use of low distortion wide range compression circuits 
is now commonplace in the recording industry, for example, as a means for 
reducing the effect of the noise level inherent to the recording medium. 
Use of such circuits has increased the quality of professional tape 
recorded program material. However, prior art low distortion wide dynamic 
range compression circuits suitable for such applications have been 
characterized by a power consumption which was too high to be suitable for 
use in miniature portable high quality tape recorders. Thus, the present 
circuit makes possible the application of previous automatic level control 
and noise reduction techniques to high quality miniature tape recorders 
without noticeably effecting their battery life. For such applications, it 
is possible to choose the circuit values to produce a maximum distortion 
of 1% or less throughout the entire dynamic range of the amplifiers. 
While the invention has been particularly shown and described with 
reference to preferred embodiments thereof, it will be understood by those 
skilled in the art that various changes in form and details may be made 
therein without departing from the spirit and scope of the invention. For 
example, the basic circuit has been specifically designed for ease of 
translation into a monolithic integrated circuit. 
COMPONENT LIST FOR FIG. 3 
Transistors 
Q1, q3, q4, q7, q9, q11, q12=high Beta NPN such as Motorola MMCS 930 
Q5=super Beta NPN such as half of Intersil IT124/D monolithic dual 
Q2, q6, q8, q10, q13--pnp such as Motorola MMCS5089 (Beta not critical in 
PNP transistors) 
Diodes 
D1, d2, d3=gold doped silicon such as Motorola MMCD914 
Resistors 
R3--680k 
R4--1.4 meg 
R5, r22--220k 
R6, r7--56k 
R8, r9--27k 
R11, r12, r13--100k 
R17--1 meg 
R16--selected, approximately 10k 
R18--selected, approximately 5 meg 
R19--2.5 meg potentiometer 
R20--1k 
Resistors 
R21--2 meg 
R23--optional, 100k variable 
Capacitors 
C2--0.01 uf 
C3--1.0 uf 
C4--0.05 uf 
C5--0.33 uf