RF transceiver IC having internal loopback conductor for IP2 self test

An RF transceiver integrated circuit has a novel segmented, low parasitic capacitance, internal loopback conductor usable for conducting IP2 self testing and/or calibration. In a first novel aspect, the transmit mixer of the transceiver is a current mode output mixer. The receive mixer is a passive mixer that has a low input impedance. In the loopback mode, the transmit mixer drives a two tone current signal to the passive mixer via the loopback conductor. In a second novel aspect, only one quadrature branch of the transmit mixer is used to generate both tones required for carrying out an IP2 test. In a third novel aspect, a first calibration test is performed using one quadrature branch of the transmit mixer at the same time that a second calibration test is performed using the other quadrature branch, thereby reducing loopback test time and power consumption.

BACKGROUND INFORMATION

1. Technical Field

The disclosed embodiments relate to internal loopback testing and calibration of RF transceivers.

2. Background Information

Both the transmitter and the receiver of a cellular telephone are ideally linear devices that introduce minimal distortion into the signal being communicated. One type of distortion is referred to as second-order distortion. A linear amplifier generally introduces only a small amount of second order distortion when the amplifier is operating at a low output power level. As the output power increases, however, the output power at the fundamental frequency (the frequency of the input signal) rises at a first rate with respect to overall rising output power, whereas the output power due to second-order distortion rises at a faster rate. When the output power of the amplifier is high enough, the output power of the second-order distortion reaches the output power of the fundamental signal. This point of intersection is referred to as the second order intercept point (IP2). The IP2 point of a system, such as a cellular telephone transmit chain or a cellular telephone receive chain, can be used as a measure of the second-order distortion of the system.

One way to measure the IP2 of a system involves using so-called two-tone analysis. A signal of one pure frequency is referred to as a “tone”. Two tones of equal strength but different frequencies are put through the system. The system will generate an output at each of the two fundamental frequencies, but will also generate an output at other frequencies due to second-order effects. The outputs due to second-order effects will include, for example, an output that has a frequency equal to the sum of the frequencies of the two input tones. The outputs due to second-order effects will also include, for example, an output that has a frequency equal to the difference of the frequencies of the two input tones. The output powers of the output signals that are not at either of the two fundamental frequencies are measured and used to determine the IP2 of the system.

To enhance the operation of the transceiver within a cellular telephone, it is often desired to measure the IP2 of a transceiver and then to calibrate various parts of the transceiver so as to reduce the IP2 exhibited by the transceiver. External signal sources can be used to generate the signals of the two tones for use in the two-tone analysis described above, but such external sources may only be available in the factory during factory calibration. Although such factory calibration may allow the cellular telephone transceiver to be calibrated to account for variations in the semiconductor fabrication process used to make transceiver integrated circuits, such factory calibration cannot account for performance changes that occur due to temperature changes that occur during operation of the cellular telephone. Similarly, such factory calibration cannot account for performance changes that occur due to voltage supply variations that occur during operation of the cellular telephone transceiver. It is therefore desired to be able to monitor IP2 and to calibrate parts of the transceiver during use of the cellular telephone outside the factory such that distortion can remain minimized as operating conditions change.

Several ways have been proposed for using the transmitter of a transceiver to generate the two tones needed for a two-tone IP2 analysis test so that the IP2 measurements and calibrations can be made outside of the factory in a functioning transceiver. One suggestion is set forth in the paper entitled “An IP2 Improvement Technique for Zero-IF Down-Converters” by Darabi et al. This paper describes a long loop approach whereby an external power amplifier (PA) and low-noise amplifier (LNA) are used to generate one tone blocker with AM modulation in every slot. This approach, however, has several drawbacks.

First, an unnecessarily large amount of power that even may exceed a maximum output power rating of the transceiver can be driven back onto the transceiver's antenna during calibration. Usually the power level of the blocker used in calibration testing is higher than the power level of the blocker specified in standards to detect a nonlinear effect. Second, the one tone blocker approach utilizes an operating external power amplifier that generally consumes more power than is necessary. The resulting increased power consumption can reduce talk time due to the power amplifier being turned on in every slot. Third, the one tone blocker approach is not an efficient way to detect the modulated signal in an OFDMA modem.

A second approach is set forth in the WiFi, IEEE 802.11 arts in a paper entitled “A Single-Chip Digitally Calibrated 5.15-5.825-GHz 0.18-um CMOS Transceiver for 802.11a Wireless LAN,” by Bouras et al. The Bouras et al. paper suggests using an on-chip loopback connection to generate one tone for IQ mismatch calibration. If this approach were extended and applied to IP2 calibration of a cellular telephone receiver, several problems would likely occur. First, the loopback circuitry operates in a voltage driven mode. The baseband signal to be detected in the receiver would therefore likely be of an undesirably small amplitude due to the long on-chip conductors that often carry the high frequency RF loopback signals from the transmitter to the receiver. Often the distance between transmitter and receiver within a cellular telephone transceiver integrated circuit is substantial in order to prevent coupling between the receiver and transmitter. This substantial distance means that if the internal loopback connection technique were employed, then the transmitter would have to drive through long conductors to supply the two tones to the receiver circuitry for internal loopback calibration. As a result, the baseband signal as received at the receiver would likely be of such an undesirably small amplitude that calibrating the receiver would be difficult or impossible. Second, the circuits in the loopback path of the WiFi circuit might generate nonlinearities such as intermodulation terms and harmonics. These nonlinearities may interfere with receiver calibration of a cellular telephone transceiver.

SUMMARY

An RF transceiver integrated circuit of a cellular telephone has a loopback conductor circuit usable for conducting IP2 self testing and calibration. The loopback conductor circuit includes a control circuit and a novel segmented, low parasitic capacitance, internal loopback conductor. In one example, a baseband processor integrated circuit can control the control circuit of the loopback conductor circuit in the RF transceiver integrated circuit via a serial bus that extends from the baseband processor integrated circuit to the RF transceiver integrated circuit. The control circuit of the loopback conductor circuit receives control information from the serial bus and in response controls the segmented loopback conductor.

In a first novel aspect, the transmit mixer of the transmit chain of the RF transceiver integrated circuit is a current mode output mixer. The receive mixer of the receive chain of the RF transceiver integrated circuit is a passive mixer that has a relatively low input impedance. Rather than using an active mixer in the receive chain, a passive mixer is used that is followed by a transimpedance amplifier (TIA). The TIA outputs a voltage proportional to its input current received from the passive mixer. In the loopback mode, the transmit mixer drives a two tone current signal to the passive mixer via the segmented loopback conductor. Other portions of the transceiver are controlled to maximize power transfer from the transmit chain to the receive chain during the loopback mode, and to reduce power consumption, and to prevent unwanted RF transmissions from occurring. In a normal operating mode of the RF transceiver integrated circuit, the segments of the loopback conductor are isolated from one another by switch blocks such that parasitic loading and coupling problems that otherwise might occur due to the long segments of the loopback conductor are minimized or avoided.

In a second novel aspect, only one quadrature branch of the transmit mixer is used to generate both tones required for carrying out an IP2 test. Using a single branch reduces power consumption during loopback testing as compared to using two branches in conventional fashion. Using a single branch also facilitates generating two tones that have identical power magnitudes.

In a third novel aspect, a first calibration test is performed using one quadrature branch of the transmit mixer at the same time that a second calibration test is performed using the other quadrature branch. In some situations, performing multiple tests simultaneously reduces loopback test time and reduced loopback test power consumption.

The foregoing is a summary and thus contains, by necessity, simplifications, generalizations and omissions of detail; consequently, those skilled in the art will appreciate that the summary is illustrative only and does not purport to be limiting in any way. Other aspects, inventive features, and advantages of the devices and/or processes described herein, as defined solely by the claims, will become apparent in the non-limiting detailed description set forth herein.

DETAILED DESCRIPTION

FIG. 1is a very simplified high level block diagram of one particular type of mobile communication device1in accordance with one novel aspect. In this particular example, mobile communication device1is a cellular telephone that uses a Code Division Multiple Access (CDMA) or an Orthogonal Frequency Division Multiple Access (OFDMA) cellular telephone communication protocol. The cellular telephone includes (among several other parts not illustrated) an antenna2and two integrated circuits3and4. Integrated circuit4is called a “digital baseband integrated circuit” or a “baseband processor integrated circuit”. Integrated circuit3is an RF transceiver integrated circuit. RF transceiver integrated circuit3is called a “transceiver” because it includes a transmitter as well as a receiver.

FIG. 2is a more detailed block diagram of the RF transceiver integrated circuit3. The receiver includes what is called a “receive chain”5as well as a local oscillator (LO)6. When the cellular telephone is receiving, a high frequency RF signal7is received on antenna2. Information from signal7passes through duplexer8, matching network9, and through the receive chain5. Signal7is amplified by low noise amplifier (LNA)10and is down-converted in frequency by mixer11. The resulting down-converted signal is filtered by baseband filter12and is passed to the digital baseband integrated circuit4. An analog-to-digital converter13in the digital baseband integrated circuit4converts the signal into digital form and the resulting digital information is processed by digital circuitry in the digital baseband integrated circuit4. The digital baseband integrated circuit4tunes the receiver by controlling the frequency of a local oscillator signal (LO) supplied by local oscillator6to mixer11.

If the cellular telephone is transmitting, then information to be transmitted is converted into analog form by a digital-to-analog converter14in the digital baseband integrated circuit4and is supplied to “transmit chain”15. Baseband filter16filters out noise due to the digital-to-analog conversion process. Mixer block17under control of local oscillator18then up-converts the signal into a high frequency signal. Driver amplifier19and an external power amplifier20amplify the high frequency signal to drive antenna2so that a high frequency RF signal21is transmitted from antenna2.

In addition to receive chain5and transmit chain15, RF transceiver integrated circuit3includes a novel loopback conductor circuit23. Loopback conductor circuit23includes a loopback conductor22and a control circuit24. Control circuit24in the example ofFIG. 2includes a bus interface mechanism that interfaces via a SSBI serial bus and conductors25with digital baseband integrated circuit4. Circuitry in the digital baseband integrated circuit4uses the SSBI bus to control the loopback conductor22by sending appropriate loopback control information across SSMI bus25to control circuit24. Control circuit24in turn sends appropriate loopback control signals26to loopback conductor22so that loopback conductor22is controlled to either couple the transmit chain15to the receive chain5or to decouple the transmit chain15from the receive chain5.

FIG. 3is a more detailed diagram of the loopback conductor22of the RF transceiver integrated circuit3ofFIG. 2. The control conductors and control circuit24of the loopback conductor circuit23ofFIG. 2are not illustrated inFIG. 3so that other detail of the circuit can be illustrated. Loopback conductor22includes a programmable mechanism for coupling node27to node28and for coupling node29to node30. Loopback conductor22includes switch blocks31,32,36and37, as well as conductor segments33,34,35,38,39and40. Loopback conductor22also includes four DC blocking capacitors41-44. Each DC blocking capacitor may, for example, have a capacitance of approximately 5 pF. In one advantageous aspect, conductor segments34and39are relatively long so that the transmit chain circuitry can be disposed at a considerable distance away from the receive chain circuitry. Laying out the RF transceiver integrated circuit3such that the transmit chain circuitry is located a considerable distance away from the receive chain helps reduce transmitter leakage and interference between the transmitter and receiver during normal operation. In the illustrated example, conductor segments34and39are each at least two millimeters in length.

There are at least two operating modes of RF transceiver integrated circuit3: 1) a normal operating mode in which the transmit and receive chains are usable to engage in RF wireless communication; and 2) a loopback mode in which the transmit chain is used to drive signals to the receive chain to perform testing and/or calibration.

In the normal operating mode, the N-channel switches of switch block31and switch block32are controlled appropriately by control circuit24such that conductor segment34is isolated and disconnected from both conductor segment33and conductor segment35. Central conductor34is allowed to float. Likewise, conductor segment38is isolated and disconnected from conductor segment39which in turn is isolated and disconnected from conductor segment40. In each of the switch blocks31,32,36and37, the central “T” node is grounded by closing an N-channel switch between the central “T” node and a ground conductor. In the example of switch block31, switch31B is closed to ground central “T” node31A. The other two switches of switch block31are open. The grounding of central “T” node31A prevents signal leakage through the switch block from conductor33to conductor34.

In the normal operating mode, an I quadrature signal path extends through the I quadrature branch of the transmit chain15. The I quadrature signal path extends through portion16A of baseband filter16, through portion17A of mixer17, through a tank circuit45, through driver amplifier19, out of RF transceiver integrated circuit3and to power amplifier20(seeFIG. 2), and through duplexer8to antenna2. The variable capacitor46(C1) on the primary of the tank45and capacitor58(C2) on the secondary of tank45are controlled to tune tank45so that power transmission from mixer17to driver amplifier19is maximized at the desired operating frequency. The I quadrature signal path in receive chain5extends from antenna2(seeFIG. 2), through duplexer8, through matching network9, and into RF transceiver integrated circuit3, through differential low noise amplifier10, through portion11A of mixer11, through transimpedance amplifier (TIA) portion12A of baseband filter12, through the remainder of baseband filter12, and to ADC13in baseband processor integrated circuit4. Variable capacitors47and48(C3) of the differential LNA10are controlled to tune the LNA load of differential LNA10to maximize power transfer from LNA10into receive mixer11. Switch blocks32,37are located as close as possible to receive mixer11and to differential LNA10, thereby minimizing capacitive loading on the conductors that couple the output of LNA10to the receive mixer11. Similarly, switch blocks31and36are located as close as possible to transmit mixer17and tank45, thereby minimizing capacitive loading on the conductors that couple the output of receiver mixer17to tank45. The Q quadrature signal path through and out of the transmitter is similar to the above-described I quadrature transmit signal path, and the Q quadrature signal path into and through the receiver is similar to the above-described Q quadrature receive signal path.

In the loopback mode, the N-channel switches of switch block31and of switch block32are controlled by control circuit24such that conductor segment33is coupled to conductor segment34which in turn is coupled to conductor segment35. Likewise, conductor segment38is coupled to conductor39which in turn is coupled to conductor segment44.

In a first novel aspect, transmit mixer17of transmitter chain15ofFIG. 3is a current mode output mixer. Unlike the receive mixer set forth in the Bouras paper mentioned in the background information section of this patent document, the portions11A and11B of the receive mixer11ofFIG. 3are not active mixers but rather are passive mixers. Rather than each of the portions11A and11B having a high input impedance in the high hundreds of ohms at operating frequency in the Bouras paper, each of the portions11A and11B has a much lower input impedance at the operating frequency. In the example ofFIG. 3, each of portions11A and11B has an input impedance of less than three hundred ohms (for example, in this example, the input impedance is approximately 150 ohms or less). Unlike the transmit mixer set forth in the Bouras paper, each of the portions17A and17B of transmit mixer17is a current mode output mixer. For example, the mixer portion17A of the I quadrature branch of the transmit chain drives a current signal49through the loopback conductor22to passive I quadrature mixer branch11A and to passive Q quadrature mixer branch11B of the receive mixer11. Due to the driving of current signal49and the terminating of the signal path in a low impedance in portions11A and11B, the two tone loopback signal can be driven a long distance of two or more millimeters from the transmit chain to the receive chain while still generating an adequately strong two tone signal in the two portions11A and11B of the receive mixer to perform an IP2 test. For a two tone signal output power of 2 dBm as output from transmit mixer17, the two tone signal is received at receive mixer11at a power of −3 dBm.

In the loopback mode, the capacitance of variable capacitor46(C1) of the primary of tank circuit45is reduced to account for additional parasitic capacitance that is coupled onto the output of transmit mixer17when conductor segments34and39are coupled to the transmit mixer output leads. The switch57is in series with the capacitor58(C2) of the secondary of tank45. Switch57is controlled to be open, and the switch59between the secondary and input lead of driver amplifier19is made to be open. Switch59prevents the input capacitance of driver amplifier19from loading the transmit mixer17during a loopback test. Driver amplifier19is also disabled to reduce current consumption during the loopback test and to prevent undesired strong transmissions from antenna2that might otherwise occur during loopback testing. In one example, the capacitance of variable capacitor46is set to 1.0 pF in the normal operating mode such that tank45resonates at a desired frequency of approximately 2.0 GHz, whereas the capacitance of variable capacitor46is set to approximately 0.5 pF in the loopback mode such that tank45resonates at the same desired frequency of approximately 2.0 GHz. In the illustration ofFIG. 3, the added parasitic capacitances of loopback conductor22are represented by capacitor symbols50,51,52,53,34A and39A. Each of these parasitic capacitances may, for example, be approximately 0.5 pF. During the loopback mode, differential LNA10is disabled so it does not drive the receive chain mixer and interfere with current signal49. Switches in series with LNA load capacitances47and48(C3) are controlled to maximize tank impedance in the loopback mode as well.

In a second novel aspect, only one quadrature branch of the transmit mixer17is used to generate the two tones required for carrying out an IP2 test.FIG. 4is a simplified diagram that illustrates an IP2 test. The two signals F1and F2are of identical power amplitudes. These two tones are driven into receive chain5via the loopback conductor22. The magnitude of any resulting intermodulation distortion coming out of receive chain5is then measured. Various circuit parameters and settings of the transceiver may be changed, and the intermodulation distortion measured again, until the magnitude of the intermodulation distortion power falls below an acceptable level. To perform such an IP2test using the novel loopback conductor22, the I quadrature branch portion14A of the differential current DAC14ofFIG. 1is made to drive a differential current signal to the portion16A of the transmit chain. Although DAC14and portion16A of the I branch are illustrated in simplified form, they are both differential circuits as illustrated more fully in the Q branch. Portion16A includes an RC filter as well as a current mirror. The current mirror outputs a current signal to portion17A of the transmit mixer17such that portion17A outputs current signal49involving two tones. These two tones are communicated via loopback conductor22to the receive chain as explained above. The transistors of the other quadrature mixer branch (the Q quadrature mixer branch17B in this example) are turned off and made nonconductive such that the outputs of the portion16B of baseband filter16are isolated from loopback conductor22. The ground symbols that are illustrated on the gates of the transistors of the Q quadrature mixer portion17B inFIG. 3represent how the transistors are turned off and controlled to be nonconductive. The use of one quadrature branch of the transmit mixer to output the two tones for an IP2 test has several advantages as compared to the use of two mixers to generate the two tones. First, the use of one mixer portion to generate both tones facilitates making both tones of the same amplitude. Second, the DAC14and baseband filter16need only drive one tone trajectory, resulting in more linear output as a function of current consumption. Third, the generation of the two tones does not rely on a mismatch between I and Q quadrature signal paths. Fourth, current consumption is reduced because only one signal path circuit is required as compared to two signal path circuits. This reduction in power consumption is especially advantageous in applications in which IP2 testing and calibration is to be performed in every slot.

In a third novel aspect, a first calibration test is performed using the I quadrature branch16A,17A of the transmit chain at the same time that a second calibration test is performed using the Q quadrature branch16B,17B of the transmit chain. Only one of the two quadrature branches of the transmit chain drives the loopback conductor22during this time. In one example, the I quadrature branch16A,17A of the transmit chain drives the two tone current signal49through loopback conductor22to the two portions11A and11B of receive mixer11in an IP2calibration test. At the same time that this two tone current signal49is being driven across loopback conductor22, a DC offset calibration test is performed in the Q quadrature branch16B,17B of the transmit chain. Control circuit24causes the transistors in portion17B of mixer17to be nonconductive, thereby isolating the outputs of baseband filter portion16B from loopback conductor22. Control circuit24causes switch54to close such that node55is coupled to supply voltage VDD through resistor56. DAC14B of DAC14ofFIG. 1is a current mode output digital-to-analog converter (DAC) in that the magnitude of the current IDAC output by DAC14B should correspond to a digital value received by DAC14B. For a given digital value received, DAC14B should drive the correct amount of current through the baseband filter portion16B such that for a given load on node55, the voltage on node55has a particular voltage. Accordingly, in a DC offset calibration test, the switch54is closed and DAC portion14B is supplied with a digital value, and an analog-to-digital converter portion13C of the ADC13ofFIG. 1reads the voltage on node55. The offset current of DAC14B is then adjusted such that when a particular digital value is supplied to DAC portion14B the voltage on node55is measured to have the desired voltage.

In a cellular telephone, multiple calibration tests including an IP2 test and a DC offset calibration test may be performed periodically during normal operation of the cellular telephone, and/or initially upon power up of the cellular telephone. Allowing multiple ones of these calibration tests to be performed simultaneously as described above with respect to the IP2 test and the DC offset test allows the total amount of time required to carry out testing and calibration to be reduced. Reducing test and calibration time reduces the amount of power consumed to do testing and calibration and also decreases the wait time after a power up condition until the cellular telephone is usable to communicate in the normal operating mode.

FIG. 5is a table that sets forth various characteristics of the novel circuit ofFIGS. 1-3as compared to characteristics of the conventional WiFi loopback prior art set forth in the Bouras paper mentioned in the background section of this patent document.

FIG. 6is a more detailed diagram of one example of branch portion17A or17B of transmit mixer17ofFIG. 3. DCOCEN identifies an active high DC offset calibration enable signal and DCOCENB identifies an active low DC offset calibration enable signal. DCOCEN and DCOCENB are received from control circuit24of the loopback conductor circuit23ofFIG. 2.

FIG. 7is a more detailed diagram of one example of differential LNA10ofFIG. 3.

FIG. 8is a flowchart representation of a method100in accordance with one novel aspect. In one step (step101), in a loopback mode of operation, a current signal is output by a current mode output mixer of a transmit chain. This current signal is supplied through a segmented loopback conductor to a passive mixer of a receive chain. The transmit chain, the loopback conductor, and the receive chain are all parts of the same integrated circuit. In one example, the current mode output mixer of this method is the current mode output mixer17ofFIG. 3. In one example, the passive mixer of this method is the passive mixer11ofFIG. 3. In one example, the segmented loopback conductor of this method is the segmented loopback conductor22ofFIG. 3.

In another step (step102), in a normal operating mode of the integrated circuit, switches in the segmented loopback conductor are maintained in an open state, thereby decoupling the segments of the loopback conductor from one another. The decoupling of segments of the loopback conductor serves to isolate the current mode output mixer of the transmit chain from the passive mixer of the receive chain. In one example, the segments of the loopback conductor of this method include segments33,34,35,38,39and40ofFIG. 3. In one example, the switches that are opened and closed to couple and decouple the segments from one other are the switches of switch blocks31,32,36and37. Steps101and102may be performed in any order, and may be repeated at periodic intervals during operation of the single integrated circuit within a cellular telephone.

Although certain specific embodiments are described above for instructional purposes, the teachings of this patent document have general applicability and are not limited to the specific embodiments described above. Although the switches of the switch blocks31,32,36and37are described above as being N-channel switches, these switches may in other examples be other types of switches such as P-channel switches or transfer gates. Accordingly, various modifications, adaptations, and combinations of the various features of the described specific embodiments can be practiced without departing from the scope of the claims that are set forth below.