Universal ballast control circuit

A universal ballast control circuit allows a universal ballast to accommodate a gas discharge lamp within a relatively wide wattage range using a low-speed microcontroller. The control circuit drives the ballast to start, run and dim a particular lamp type by providing a control voltage signal to a conventional inverter MOSFET driver to effect dynamic and selective changes in the duty cycle and the frequency of the inverter signal. In one aspect of the invention, the control circuit comprises a generator for generating a periodic analog voltage signal, a source for producing a DC voltage signal, a controller which includes a low-speed microcontroller for varying the frequency of the periodic analog voltage signal and the magnitude of the DC voltage signal, and a comparator for comparing the periodic analog voltage signal and the DC voltage signal to produce a control voltage signal. In another aspect of the invention, the control circuit comprises a generator for generating a periodic analog voltage signal, a controller for controlling the shape of the periodic analog voltage signal, and a comparator for comparing waveforms of the periodic analog voltage signal with the waveforms of two DC threshold voltages to generate a control signal. The present invention allows for the control circuit to be powered by a power supply signal either derived from an inverter half-bridge MOSFET driver, extracted from a boost inductor of the boost converter or generated by an appropriately configured dedicated miniature switch mode power supply. Finally, control circuit can be utilized to drive PFC circuity in a feedback configuration to regulate the level of boost converter output voltage signal.

FIELD OF THE INVENTION 
The present invention relates generally to lighting ballasts and in 
particular to a universal ballast control circuit for controlling the 
operation of a wide range of gas discharge lamp types. 
BACKGROUND OF THE INVENTION 
Significant improvements in programmable microcontrollers over the past 
five years as well as the existence of dimming systems which use complex 
algorithms have caused most major electronic ballast companies to develop 
microcontroller driven electronic ballasts. These electronic ballasts 
typically use microcontrollers to adjust the characteristics of the 
inverter voltage signal to accommodate a wide variety of lamps and/or to 
provide dimming functionality. Specifically, by changing the frequency or 
duty cycle of the inverter voltage signal, these electronic ballasts are 
able to start, run and dim a wide variety of gas discharge lamps. 
Some electronic ballasts such as the one disclosed by U.S. Pat. No. 
5,039,921 to Kakitani uses a central processing unit (CPU) to control the 
frequency of the inverter voltage signal to change lamp voltage. The 
Kakitani patent describes a ballast which can be adapted to light and 
drive various types of gas discharge lamps according to each lamp's 
individual rating. The control circuit employs the CPU to detect the 
rating of the discharge lamp based on the lamp's starting voltage and to 
retrieve stored lamp loading data from memory relating to the type of 
discharge lamp detected. The oscillating frequency of the inverter circuit 
voltage signal is then adjusted so that the ballast produces a power 
voltage signal suited to the particular discharge lamp. 
Other electronic ballasts such as the one disclosed by U.S. Pat. No. 
5,569,984 to Holtstag use a microprocessor to control the switching 
frequency and the pulse width of the inverter voltage signal provided to a 
particular lamp to avoid strong acoustic resonances or arc instabilities. 
The microprocessor evaluates deviation of electrical lamp parameters to 
detect arc instabilities and adjusts the frequency and pulse width in 
response. Accordingly, the ballast can operate HID lamps of different 
types, wattages and manufacturers over a broad frequency range despite the 
occurrence of acoustic resonance/arc instabilities among these lamps. 
In order to achieve acceptable levels of accuracy in running and dimming a 
wide variety of gas-discharge lamps, it is necessary to be able to produce 
a wide variety of inverter voltage signals which requires a high 
resolution of control signals. Low-speed microcontrollers cannot provide 
the necessary degree of control to run a lamp within a ballast having 
conventional inverter signal frequencies. In order to achieve the desired 
operation of a typical ballast, expensive high-speed microcontrollers must 
be used which severely limits mass production and consumption of 
microcontroller-based electronic ballasts due to the cost of such 
high-speed microcontrollers. 
Further, since microcontrollers provide discreet output, when digital 
output levels are provided to a lamp, sudden incremental changes in the 
lumen output are produced. These discrete "steps" in light intensity are 
visible to users and are unacceptable in commercial and residential 
environments. Even when the microcontroller is programmed to dim a lamp in 
relatively small increments, dimming a lamp using a digital signal still 
results in visible steps. 
Finally, in order to provide sufficient power supply to the 
microcontroller, either a drop-down resistor or a dedicated off-line power 
supply circuitry is used. The problem with using a simple voltage-drop 
resistor is that the heat and high frequency noise which are generated are 
very difficult to suppress. On the other hand, a separate off-line power 
supply adds substantial expense to the product. 
Thus, there is a need for a universal lighting ballast control circuit 
which can produce a wide range of different control signals to start, run 
and dim a wide variety of gas-discharge lamp types using an inexpensive 
low-speed microcontroller, which can modulate illumination levels on a 
continuously variable basis and which provides power to the 
microcontroller without conventionally known power supply problems and 
associated expense. 
BRIEF SUMMARY OF THE INVENTION 
It is therefore an object of the present invention in one aspect to provide 
a universal ballast control circuit for use with a power circuit coupled 
to an AC source for outputting a high frequency AC signal and a coupling 
circuit coupled to the power circuit for applying the AC signal to any one 
of a plurality of gas discharge lamp types, said control circuit 
comprising: 
(a) a generator for generating a periodic analog voltage signal having a 
first waveform; 
(b) a source for generating a first DC voltage signal having a second 
waveform; 
(c) a controller for controlling the frequency of the periodic analog 
voltage signal; and 
(d) a processor for processing said first DC voltage signal and said 
periodic analog voltage signal to generate a control voltage signal for 
varying the frequency and duty cycle of the AC signal, the frequency and 
duty cycle of said control voltage signal being dependent on said first 
and second waveforms. 
In a second aspect, the present invention provides a universal ballast 
control circuit for use with a power circuit coupled to an AC source for 
outputting a high frequency AC signal and a coupling circuit coupled to 
the power circuit for applying the AC signal to any one of a plurality of 
gas discharge lamp types, said control circuit comprising: 
(a) a generator for generating a periodic analog voltage signal having a 
first waveform; 
(b) a controller for controlling the shape of said first waveform; and 
(c) a comparator for comparing the periodic analog voltage signal with at 
least one DC voltage and for generating a control voltage signal for 
varying the duty cycle and frequency of the AC signal. 
In a third aspect, the present invention provides a method of powering any 
one of a plurality of gas discharge lamp types, each lamp type having a 
predetermined set of lamp characteristics, said method comprising the 
steps of: 
(a) producing a high frequency AC signal; 
(b) applying the AC signal to the lamp; 
(c) generating a periodic analog voltage signal having a first waveform; 
(d) generating a DC voltage signal having a second waveform; 
(e) controlling the frequency of the periodic analog voltage signal; 
(f) controlling the magnitude of the DC voltage signal; 
(g) varying the duty cycle and frequency of the AC signal based on a 
comparison of the first and second waveforms. 
In a fifth aspect, the present invention provides a method of powering any 
one of a plurality of gas discharge lamp types, each lamp type having a 
predetermined set of lamp characteristics, said method comprising the 
steps of: 
(a) producing a high frequency AC signal; 
(b) applying the AC signal to the lamp; 
(c) generating a periodic analog voltage signal having a first waveform; 
(d) controlling the shape of the first waveform; 
(e) varying the duty cycle and frequency of the AC signal based on a 
comparison of first waveform and at least one DC threshold voltage. 
It also an object of the present invention to provide a method of 
controlling the output voltage of a boost converter of a gas-discharge 
lighting ballast, said method comprising the steps of: 
(a) applying a DC signal to a power switch to produce a boost converter 
output voltage; 
(b) generating a periodic AC voltage signal; 
(c) varying the waveform characteristics of the periodic AC voltage signal 
to form a modulated periodic AC voltage signal; 
(d) comparing the modulated periodic AC voltage signal with the boost 
converter output voltage; and 
(e) applying the result to the power switch to change the output voltage of 
the boost converter.

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS 
Reference is first made to FIG. 1, which shows a prior art 
microcontroller-based electronic ballast 10. As is conventionally known, 
ballast 10 includes a bridge rectifier 12, a boost converter 14, an 
inverter 16, resonance network 18, and a microcontroller 20. Ballast 10 is 
used to power a lamp 22 as is conventionally known. 
Bridge rectifier 12 is coupled to a typical AC power line voltage of 
110-120 Volts. A rectifier consisting of diodes provides a full-wave 
rectified DC voltage of about 160 Volts across its output. Bridge 
rectifier 12 may also include an EMI filter for insulating the power lines 
from interference generated by ballast 10. 
Boost converter 14 is coupled to the output of bridge rectifier 12 and is 
used to boost and control the input DC voltage provided by bridge 
rectifier 12 such that appropriate power is provided to lamp 22. Boost 
converter 14 provides regulated voltage to inverter 16. 
Inverter 16 is a voltage-fed half-bridge DC-AC inverter which is used to 
convert the input DC voltage received from boost converter 14 into high 
frequency AC voltage. Half-bridge inverter 16 typically includes a 
half-bridge MOSFET driver 17 and MOSFET transistors Q.sub.A and Q.sub.B at 
its output, although many other implementations are possible (i.e. using 
bipolar transistors). MOSFET driver 17 is typically implemented using an 
integrated circuit such as IR2104 manufactured by International Rectifier. 
Transistors Q.sub.A and Q.sub.B produce an inverter voltage signal which 
is a high frequency generally square wave signal, as is familiar to those 
skilled in the art. The high frequency signal generated by transistors 
Q.sub.A and Q.sub.B is applied to resonance network 18. 
Resonance network 18 is directly coupled to lamp 22 and is commonly used to 
avoid the necessity of an output transformer. Resonance network 18 
typically includes an LCC network of capacitors and inductors which 
provides waveshaping and current limiting to produce a substantial 
sinusoidal lamp current for lamp 22. Ballast designers choose an optimal 
inverter frequency and optimal values of LCC circuit inductance and 
capacitance to create proper currents and voltages across the lamp as well 
as to produce an economical ballast configuration. The LCC network also 
functions as an igniter to ignite the lamp upon initial application of 
power to ballast 10. 
Microcontroller 20 is used to control the operating frequency or duty cycle 
of the inverter voltage signal. In order for ballast 10 to properly 
operate lamp 22, ballast 10 must be able to produce certain voltage and 
current characteristics which are suited to a lamp's particular 
characteristics. When lamp 22 has been struck and is in full operation, 
its running voltage must be within its manufacturer's specified range. 
Typically, ballast 10 would be designed to provide a voltage between 35 
and 130 volts (rms) for running operation of lamp 22. Particular voltages 
must be provided across the filaments of lamp 22 during the course of lamp 
operation. Further, the current flowing through lamp 22 must also be such 
that lamp 22 can be safely run. Finally, a sufficient striking (or 
ignition) voltage must be applied to lamp 22, such that the pressurized 
gas ignites into plasma form and forms a plasma thread. The provision of 
all of these voltage and current characteristics is accomplished by 
controlling the operation of half-bridge MOSFET driver 17 which in turn 
drives transistors Q.sub.I1 and Q.sub.I2 of inverter 16. By controlling 
the duration and frequency that transistors Q.sub.I1 and Q.sub.I2 are 
conductive, microcontroller 20 can ensure that ballast 110 provides the 
proper striking, running and dimming of lamp 22. 
However, in order for ballast 10 to provide the above discussed circuit 
conditions, microcontroller 20 must operate at a high-speed to produce a 
sufficient number of control levels. If microcontroller operates at too 
low a speed, then ballast 10 will not be able to accurately provide the 
various current and voltage characteristics which are necessary for proper 
running and dimming of lamp 22. 
As an illustration, consider a typical low-speed 8-pin microcontroller such 
as the PIC12C508 from Microchip Technology. If this microcontroller 20 is 
configured to directly drive the output half-bridge inverter 16 of ballast 
10, it will result in inaccurate operation of ballast 10. The nominal 
frequency of the PIC12C508 microcontroller 20 is 4 MHz, a typical value 
for this slower class of microcontrollers. 
Since an instruction can only be acted on by microcontroller 20 once every 
four timing cycles, the command cycle time T.sub.C for this device would 
be: 
##EQU1## 
Accordingly, every 1 .mu.sec a digital voltage level can be provided to 
inverter 16 to change the current and voltage characteristics of resonance 
network 18. Typically, fluorescent electronic ballasts operate in a 
frequency range between 20 kHz to 60 kHz. For a ballast having an 
operating frequency of 40 kHz, the duration of the half-cycle pulse 
T.sub.1/2 of the inverter voltage is: 
##EQU2## 
Accordingly, in order to adjust the duty cycle or the frequency of the 
inverter voltage signal, there are only be 12 steps in which to do so. The 
overall accuracy of such control circuitry is approximately 8.3%. This 
accuracy becomes worse when the full range of 50/50 duty cycle oscillation 
cannot be used. When the duty cycle is 20/80 or even 10/90, the driving 
accuracy of the control circuitry will only be about 50-80% which is 
unacceptable for proper operation of ballast 10. 
Consequently, it is necessary to use high-speed microcontrollers which run 
at speeds of between 20 MHz to 40 MHz to properly control the operation of 
typical ballasts running at frequencies between 20 kHz to 60 kHz. Such 
microcontrollers are typically priced at between US$5 to US$10 each, which 
prohibits cost-effective production of ballast 10. Further, when the 
frequency of the digital output pulses produced by high-speed 
microcontrollers 20 is changed, sudden incremental changes in the lumen 
output of lamp 22 result which are visually perceived as light intensity 
"steps". For example, even when microcontroller 20 is programmed to dim 
lamp 22 using 128 light intensity steps, the inventor has found that 
visible steps still occur. 
Reference is now made to FIGS. 2 and 3, which show an improved 
microcontroller-based programmable ballast 110 which includes a control 
circuit 126, according to a preferred embodiment of the invention. Control 
circuit 126 is designed to utilize a relatively inexpensive 
microcontroller 120 to control the inverter voltage signal to start, run 
and dim a wide range of lamp types. Common elements between ballast 110 
and the prior art ballast 10 will be denoted by the same numerals with one 
hundred added thereto. 
Accordingly, ballast 110 includes a bridge rectifier 112, a boost converter 
114, an inverter 116, resonance network 118, a microcontroller 120 as 
previously discussed. Ballast 110 also comprises control circuit 126 which 
uses a low-speed microcontroller 120 for proper operation of ballast 110. 
Control circuit 126 provides an analog control voltage signal V.sub.C to 
half-bridge MOSFET driver 117 which in turn drives MOSFET transistors 
Q.sub.A and Q.sub.B. Control circuit 126 comprises microcontroller 120, 
periodic signal generator 128, a digital-to-analog (D/A) converter 130 and 
a comparator 132. 
Microcontroller 120 of the present invention can be a conventional low-cost 
microprocessor such as PIC12C508 from Microchip Technology. 
Microcontroller 120 generates digital voltage signals V.sub.D1 and 
V.sub.D2 which are input into control circuit 126. 
Periodic voltage signal generator 128 receives digital voltage signal 
V.sub.D2 from microcontroller 120 and generates a periodic voltage signal 
V.sub.P. While any periodic voltage signal can be used within control 
circuit 126, the inventor has determined that a sawtooth waveform is 
preferable as a sawtooth generator can be implemented by a simple and low 
cost circuit. For example, a conventional timer integrated circuit (e.g. a 
555 timer circuit or an IR5155 oscillator circuit) configured with 
appropriate resistive and capacitor elements attached to various pin 
inputs and outputs generates a sawtooth waveform, as is conventionally 
known and as will be discussed. 
D/A converter 130 converts the digital voltage signal V.sub.D1 produced by 
microcontroller 120 into an analog voltage signal V.sub.DC1. D/A converter 
130 is preferably implemented using an integrating capacitor C.sub.I, 
either a series or parallel connected resistor R.sub.I and an 
appropriately oriented diode D.sub.I as shown in FIGS. 4A and 4B, to form 
a conventional integrator circuit. It has been determined that it is 
preferable to use the circuits of FIGS. 4A and 4B instead of 
conventionally available D/A integrated circuits to ensure that a wide 
range of analog signals can be produced cost effectively. 
Referring back to FIGS. 2 and 3, comparator 132 is a general-purpose 
comparator integrated circuit such as the LM393 integrated circuit 
manufactured by Linear Technology. As shown, the DC voltage signal 
V.sub.DC1 being output by the D/A converter 130 is provided to the 
positive input of comparator 132 and the periodic voltage signal V.sub.P 
is provided to the negative input of comparator 132. Comparator 132 
produces a control voltage signal V.sub.C waveform having a duty cycle 
DC.sub.C and a frequency f.sub.C as shown in FIG. 3. 
Control voltage duty cycle DC.sub.C is dependent on the comparative values 
of the DC voltage signal V.sub.DC1 and the periodic voltage signal 
V.sub.P. It will be seen from FIG. 3 that when periodic voltage V.sub.P 
exceeds V.sub.DC1, control voltage V.sub.C goes high and while periodic 
voltage V.sub.P is less than DC voltage signal V.sub.DC1, control voltage 
V.sub.C goes low. Accordingly, control voltage duty cycle DC.sub.C can be 
varied by adjusting the value of the DC voltage signal V.sub.DC1 or by 
changing the digital voltage signal V.sub.D1 generated by microcontroller 
120. 
Further, as can be seen from FIG. 3, the control voltage frequency f.sub.C 
is equivalent to the frequency of the periodic voltage signal V.sub.P. 
Accordingly, the control voltage frequency f.sub.C can be varied by 
controlling the frequency of the periodic voltage signal V.sub.P. One way 
of accomplishing this is by using the DC voltage signal V.sub.DC2 from 
microcontroller 120 to control the current source of the periodic voltage 
signal generator 128 as will be discussed below. 
FIG. 5 shows one possible circuit implementation of control circuit 126 in 
which both the control voltage duty cycle DC.sub.C and the control voltage 
frequency f.sub.C are varied using a periodic voltage signal generator 
128, a controller 129, and a comparator 132. 
Controller 129 includes microcontroller 120 and D/A converters D/A.sub.1 
and D/A.sub.2. Microcontroller 120 outputs two separate digital control 
voltages V.sub.D1 and V.sub.D2 into D/A converters D/A.sub.1 and 
D/A.sub.2, respectively which in turn convert them into DC voltage signals 
V.sub.DC1 and V.sub.DC2. DC voltage signal V.sub.DC1 is input into the 
positive input of comparator 132 and DC voltage signal V.sub.DC2 is used 
to control the current source of the periodic voltage signal generator 
128. By varying DC voltage signal V.sub.DC1, it is possible to control the 
control voltage duty cycle DC.sub.C being output by comparator 132 as 
previously described. By varying the DC voltage signal V.sub.DC2, it is 
possible to control the control voltage frequency f.sub.C, as will be 
discussed. 
Periodic voltage signal generator 128 generates a sawtooth waveform using 
timer IC.sub.T, resistors R.sub.1 and R.sub.2, capacitor C.sub.T, and 
transistor Q.sub.1. Timer IC.sub.T is a conventional timer integrated 
circuit such as a 555 timer circuit or an IR5155 oscillator circuit. RESET 
(pin 4) and VCC (pin 8) are connected to running voltage V.sub.DD. TRIGGER 
(pin 2), THRESHOLD (pin 6) and DISCHARGE (pin 7) of timer IC.sub.T are 
coupled at node A to a grounded timing capacitor C.sub.T. Transistor 
Q.sub.1 has its collector connected to node A and its emitter connected to 
voltage V.sub.cc through resistor R.sub.1. Accordingly, periodic voltage 
signal V.sub.P having a sawtooth waveform is generated at the collector of 
transistor Q.sub.1 as is conventionally known. The current source 
comprising resistor R.sub.1 and transistor Q.sub.1 powered by voltage 
V.sub.cc serves to stabilize the charge current on capacitor C.sub.T. 
When the periodic voltage signal V.sub.P THRESHOLD (pin 6) rises above 2/3 
V.sub.DD, timer IC.sub.T shorts capacitor C.sub.T to ground at DISCHARGE 
(pin 7) through its internal discharge transistor. When the periodic 
voltage signal V.sub.P at THRESHOLD (pin 6) falls below 1/3 V.sub.DD, the 
internal discharge transistor in timer IC.sub.T is disabled and capacitor 
C.sub.T begins to recharge from V.sub.CC through resistor R.sub.1 and 
transistor Q.sub.1. In this way, timer IC.sub.T can be configured to 
operate as an astable multivibrator such that a periodic voltage signal 
V.sub.P is produced across capacitor C.sub.T. Since the current flowing 
through transistor Q.sub.1 is controlled by DC voltage signal V.sub.DC2, 
it is possible to control the frequency of the periodic voltage signal 
V.sub.P by appropriately varying DC signal V.sub.DC2. Transistor Q.sub.1 
operates as a linear modulating amplifier since Q.sub.1 is always biased 
in its active region. Accordingly, as DC voltage signal V.sub.DC2 is 
increased, the current flowing through transistor Q.sub.1 is increased 
(i.e. impedance of transistor Q.sub.1 is decreased) and capacitor C.sub.T 
Will charge at a faster rate. Thus, the set-point of 2/3 V.sub.DD will be 
reached more quickly causing the frequency of periodic voltage signal 
V.sub.P to increase which in turn will increase the control voltage 
frequency f.sub.C. Since the current source comprising resistor R.sub.1, 
transistor Q.sub.1 voltage V.sub.CC can be considered to operate as a 
variable impedance having a linear characteristic when V.sub.DC2 is 
applied to the base of Q.sub.1, the duty cycle of the control voltage 
signal V.sub.C will not be affected by changes in DC signal V.sub.DC2. 
Further, by changing the value of DC voltage signal V.sub.DC1, it is 
possible to change the control voltage duty cycle DC.sub.C in a continuous 
manner. If the integrator capacitor C.sub.I of D/A converter D/A.sub.1 is 
large, a wide range of DC voltage signals V.sub.C, each having a unique DC 
threshold voltage, can be generated for comparison with periodic voltages 
V.sub.P. As DC voltage signal V.sub.DC1 is reduced, the duty cycle of 
control voltage V.sub.C increases and as DC voltage signal V.sub.DC1 
increases, the duty cycle of control voltage V.sub.C decreases. It should 
be noted that the frequency of control voltage V.sub.C will not change as 
DC voltage signal V.sub.DC1 is varied. 
Consequently, it is possible for microcontroller 120 and control circuit 
126 to generate a wide range of control voltages V.sub.C, each with a 
unique frequency f.sub.C and duty cycle DC.sub.C. 
FIG. 6 shows an alternative circuit implementation of control circuit 126 
which uses microcontroller 120 to independently control the control signal 
duty cycle DC.sub.C and the control signal frequency f.sub.C of control 
voltage signal V.sub.C with digital voltage signals V.sub.D1 and V.sub.D2. 
Control circuit 126 comprises controller 129, timer circuit IC.sub.T, 
transistors Q.sub.1 and Q.sub.2, resistor R.sub.1, and timing capacitor 
C.sub.T. 
Controller 129 comprises microcontroller 120 and two D/A converters 
D/A.sub.1 and D/A.sub.2. D/A converters D/A.sub.1 and D/A.sub.2 convert 
digital voltage signals V.sub.D1 and V.sub.D2 from microcontroller 120 
into DC voltage signals V.sub.DC1 and V.sub.DC2, respectively. Each DC 
voltage signal V.sub.DC1 and V.sub.DC2 controls the operation of 
transistors Q.sub.1 and Q.sub.2, respectively, to vary the duty cycle and 
frequency of periodic voltage signal V.sub.P as will be described. 
Timer IC.sub.T is a conventional timer (e.g. a 555 timer) powered by 
voltage V.sub.DD and utilized as a simple oscillator in the present 
circuit. The schematic and written description of the 555 timer circuit 
provided by "Microelectronic Circuits" Third Edition by Adel Sedra and 
Kenneth C. Smith (at pages 875 to 880) is hereby incorporated by 
reference. Timer IC.sub.T compares the periodic voltage signal V.sub.P at 
THRESHOLD (pin 6) with two internally generated threshold voltages namely 
1/3 V.sub.DD and 2/3 V.sub.DD. THRESHOLD (pin 6) of timer IC.sub.T is 
connected to the common collector junction of transistors Q.sub.1 and 
Q.sub.2 and to ground through timer capacitor C.sub.T. OUTPUT (pin 3) of 
timer IC.sub.T produces the control voltage V.sub.C of control circuit 
126. As will be explained, due to the charging and discharging of timer 
capacitor C.sub.T, a periodic voltage signal V.sub.P with a 
triangular-type waveform is generated at the common collector junction. 
When periodic voltage signal V.sub.P is greater than 2/3 V.sub.DD at 
THRESHOLD (pin 6), internal circuitry of timer IC.sub.T will cause OUTPUT 
(pin 3) to go high. When periodic voltage signal V.sub.P is lower than 1/3 
V.sub.DD at THRESHOLD (pin 6), internal circuitry of timer IC.sub.T will 
cause OUTPUT (pin 3) will go low. In this way, control voltage signal 
V.sub.C is controlled by the voltage characteristics (i.e. duty cycle and 
frequency) of periodic voltage signal V.sub.P. 
Transistors Q.sub.1 and Q.sub.2 are coupled to ground through timing 
capacitor C.sub.T and to the output of control circuit 126 through 
resistor R.sub.1. Transistors Q.sub.1 and Q.sub.2 are controlled by DC 
voltage signals V.sub.DC1 and V.sub.DC2, respectively. Resistor R.sub.1 
and control voltage signal V.sub.C act as either a current source for 
transistor Q.sub.1 or a current sink for transistor Q.sub.2, depending on 
the polarity of control voltage signal V.sub.C. Specifically, the 
collectors of transistors Q.sub.1 and Q.sub.2 are coupled to the ground 
through capacitor C.sub.T and the emitters of transistors Q.sub.1 and 
Q.sub.2 are coupled to the output of control circuit 126 through resistor 
R.sub.1. The bases of transistors Q.sub.1 and Q.sub.2 are coupled to the 
DC voltage signal outputs of D/A converters D/A.sub.1 and D/A.sub.2. 
Accordingly, transistors Q.sub.1 and Q.sub.2 operate as amplifiers when 
they are biased in their active region by control voltage signal V.sub.C 
through resistor R.sub.1 and their impedance values can be controlled by 
DC voltage signals V.sub.DC1 and V.sub.DC2, respectively as is 
conventionally known. 
Thus, the control voltage signal V.sub.C produced at OUTPUT (pin 3) of 
timer IC.sub.T, is controlled by the combined operation and relative 
impedance of transistors Q.sub.1 and Q.sub.2. When OUTPUT (pin 3) of timer 
IC.sub.T is high, timer capacitor C.sub.T will charge through resistor 
R.sub.1 and transistor Q.sub.1 until periodic voltage signal V.sub.P 
reaches 2/3 V.sub.DD. When periodic voltage signal V.sub.P at THRESHOLD 
(pin 6) is 2/3 V.sub.DD, timer IC.sub.T will force OUTPUT (pin 3) low and 
capacitor C.sub.T will begin discharging through transistor Q.sub.2 and 
resistor R.sub.1. Once periodic voltage signal V.sub.P at THRESHOLD (pin 
6) decreases to 1/3 V.sub.DD, OUTPUT (pin 3) will be driven high and 
capacitor C.sub.T will start charging through transistor Q.sub.1 again. It 
should be noted that transistors Q.sub.1 and Q.sub.2 will never conduct 
simultaneously, as transistor Q.sub.1 is only on when OUTPUT (pin 3) at 
timer IC.sub.T is high and transistor Q.sub.2 is only on when OUTPUT of 
timer IC.sub.T is low. 
Capacitor C.sub.T will charge or discharge at a rate based on the relative 
impedances of transistors Q.sub.1 and Q.sub.2. That is, if the impedance 
of transistor Q.sub.1 is low, capacitor C.sub.T will charge at a higher 
rate than if the impedance of transistor Q.sub.1 is high. Similarly if the 
impedance of transistor Q.sub.2 is high, capacitor C.sub.T will discharge 
slower than if the impedance of transistor Q.sub.2 is low. That is, the 
duty cycle and the frequency of the periodic voltage signal V.sub.P 
waveform are determined by the direction and rate of current that flows 
through timer capacitor C.sub.T. As the characteristics of periodic 
voltage V.sub.P are changed by DC voltage signals V.sub.DC1 and V.sub.DC2, 
periodic voltage signal V.sub.P at THRESHOLD (pin 6) reaches 2/3 V.sub.DD 
and 1/3V.sub.DD voltage levels at various times which alters the waveform 
characteristics of control voltage signal V.sub.C at OUTPUT (pin 3) of 
timer IC.sub.T. Thus, by modifying the characteristics of periodic voltage 
signal V.sub.P, it is possible to control the pulse duration and pause 
duration of the high and low signals produced by OUTPUT (pin 3) of timer 
IC.sub.T and accordingly the duty cycle and frequency of control voltage 
V.sub.C can be controlled. 
As an illustration of how a control voltage signal V.sub.C is generated by 
the circuit of FIG. 6, a typical periodic voltage signal V.sub.P produced 
at the common collector junction of transistors Q.sub.1 and Q.sub.2 is 
shown in FIG. 7A. When DC voltage signal V.sub.DC1 as shown in FIG. 7B is 
applied to the base of transistor Q.sub.1 and DC voltage signal V.sub.DC2 
as shown in FIG. 7C is applied to the base of transistor Q.sub.2, the 
control voltage signal V.sub.C as shown superimposed on periodic voltage 
signal V.sub.P in FIG. 7A results. Specifically, when DC voltage signal 
V.sub.DC1 is at DC level A (FIG. 7B), periodic voltage signal V.sub.P 
causes control voltage signal V.sub.C at OUTPUT (pin 3) of timer IC.sub.T 
to have a pulse duration of X.sub.1 and a pause duration Y.sub.1 as shown. 
When DC voltage signal V.sub.DC1 is increased to DC level B (FIG. 7B), 
increased current flows through transistor Q.sub.1 when capacitor C.sub.T 
is charging and thus capacitor C.sub.T is charged at an increased rate. 
This causes control voltage signal V.sub.C to have a pulse duration 
X.sub.2 (FIG. 7A) which is less than the initial pulse duration, as shown. 
Similarly, when DC voltage signal V.sub.DC2 (FIG. 7C) is increased from DC 
level A' to B', increased current flows through transistor Q.sub.2 when 
capacitor C.sub.T is discharging and thus capacitor C.sub.T is discharged 
at an increased rate. This results in a shorter pause duration Y.sub.2, as 
shown. 
In this way, a low-speed microcontroller 120 can provide sufficient digital 
voltage signals which can be converted into a wide variety of analog 
signals that can individually control the charge time and discharge time 
of timing capacitor C.sub.T. In this way, it is possible to control the 
duration of the pulses for the control voltage signal V.sub.C and the 
pauses between the pulses to an extremely high degree of resolution. Thus 
both the control signal duty cycle DC.sub.C and the control signal 
frequency f.sub.C can be independently controlled to a wide degree by a 
relatively low-speed microcontroller 120. Further, since the control 
voltage signals V.sub.C are analog, it is possible to modulate 
illumination levels on a continuously variable basis. 
Another aspect of the present invention relates to the ability to power 
microcontroller 120 within a conventional ballast 110 without the 
conventional disadvantages. Typical microcontroller-based ballasts power 
microcontroller and other associated control circuity components either 
through a drop-down resistor which causes problems associated with heat 
and high frequency noise or by using a dedicated off-line power supply 
circuitry which is costly. Accordingly, it is desirable to provide a clean 
high frequency power signal that can be easily filtered and converted to 
DC voltage sufficient to power the microcontroller 120 and associated 
control circuitry 126 without the associated problems 
As illustrated in FIG. 8, the present invention provides power signal 
P.sub.S to microcontroller 120 by extracting energy from half-bridge 
MOSFET driver 117 using a conventionally known bootstrap power supply 142. 
As has been discussed, inverter 116 contains a MOSFET driver 117 which 
drives transistors Q.sub.A and Q.sub.B from a HIGHSIDE MOSFET SIGNAL 
OUTPUT (pin 11) and a LOWSIDE MOSFET SIGNAL OUTPUT (pin 7). Bootstrap 
power supply 142 is connected to bootstrap output (pin 12) and FLOATING 
GROUND POINT (pin 9) of MOSFET driver 117. FLOATING GROUND POINT is 
connected to the common node of transistors Q.sub.A and Q.sub.B. POWER 
SUPPLY (pin 1) of MOSFET driver 117 is fed to the input of bridge 
rectifier 112 through resistor R.sub.BR. Bootstrap power supply 142 
provides power signal P.sub.S to microcontroller 120 through 
reverse-connected diode D.sub.B3 and capacitor C.sub.B3 and through 
forward-connected diode D.sub.B1, as is conventionally known. Bootstrap 
power supply 142 comprises diodes D.sub.B1, D.sub.B2, resistor R.sub.B and 
capacitors C.sub.B1 and C.sub.B2 as is conventionally known. 
FIG. 9 shows an alternative way of providing microcontroller 120 with a 
power supply signal P.sub.S, namely by extracting power from a boost 
inductor L.sub.B of boost converter 114. Boost converter 114 typically 
comprises boost inductor L.sub.B, a PFC MOSFET transistor Q.sub.PFC, a 
bulk capacitor C.sub.B, diode D.sub.P3 and PFC control circuity 143. 
Diode D.sub.P3 acts as a uni-directional switch. When diode D.sub.P3 is 
forward biased (and MOSFET Q.sub.PFC is open), current flowing through 
boost inductor L.sub.B from bridge rectifier 112 will charge bulk 
capacitor C.sub.B to an output voltage level. Diode D.sub.P3 prevents bulk 
capacitor C.sub.B from discharging through MOSFET P.sub.PFC (if closed) or 
through boost inductor L.sub.B. This allows bulk capacitor C.sub.B to be 
charged or "boosted" to exceed the AC input voltage applied to ballast 
110, as is conventionally known. 
Power adaption circuit 144 is shown comprising diodes D.sub.P1, D.sub.P2, 
D.sub.P4, resistors R.sub.P1, and R.sub.P2, and capacitor C.sub.P. Diode 
D.sub.P1, D.sub.P4 and resistor R.sub.P1 are connected in series to 
secondary winding of boost inductor L.sub.B such that current flows to 
microcontroller 120. Schottky diode D.sub.P2 is reverse-connected to 
ensure a stable voltage drop at the node between resistor R.sub.P1 and 
forward-connected diode D.sub.P4. Power supply signal P.sub.S is provided 
to microcontroller 120 from the common node between resistor R.sub.P2 and 
capacitor C.sub.P. Capacitor C.sub.P is used to smooth power signal 
P.sub.S and resistor R.sub.P2 is used as a "start-up" resistor to ensure 
that capacitor C.sub.P undergoes several start-up charging cycles when 
ballast is started. 
Finally, as shown in FIG. 10 microcontroller 120 can be powered by a power 
supply signal P.sub.S which is generated by a dedicated miniature switch 
mode power supply 146 appropriately configured as is conventionally known. 
Switch mode power supply 146 can be restricted to producing between 2 to 3 
watts and is a reliable but somewhat expensive alternative to the previous 
alternatives. Switch mode power supply 146 can be any commercially 
available miniature switch mode power supply 146, such as a TOP210 three 
terminal off-line PWM switch integrated circuit manufactured by Power 
Integrations, Inc. as will be assumed for the following discussion. 
Switch mode power supply 146 can be configured to provide power supply 
signal P.sub.S using transformer T.sub.1, diodes D.sub.P1-P4, capacitors 
C.sub.P1 and C.sub.P2, and resistor R.sub.P as shown. The primary winding 
of transformer T.sub.1 receives the high voltage DC signal from bridge 
rectifier 112 and the other side of the primary is driven by the 
integrated high-voltage MOSFET within power supply 146. Specifically, 
power supply signal P.sub.S is determined by the voltage across CONTROL 
(pin 4) of power supply 146, the voltage drops of diode D.sub.P4 and 
D.sub.P3, and the turns ratio between the bias winding and output windings 
of transformer T.sub.1. Other output voltages can be produced by adjusting 
the turns ratios of transformer T.sub.1. Diodes D.sub.P1 and D.sub.P2 
clamp the voltage spike caused by transformer leakage to a safe value and 
reduce ringing at DRAIN (pin 5) of power supply 146. The power secondary 
winding is rectified and filtered by diode D.sub.P4 and capacitor C.sub.P1 
to create power supply signal P.sub.S. The voltage waveform across bias 
winding is rectified and filtered by diode D.sub.P3, resistor R.sub.P and 
capacitor C.sub.P2 to create a bias voltage to power supply 146. Capacitor 
C.sub.P2 also filters internal MOSFET gate drive charge current spikes on 
the CONTROL pin, determines the auto-restart frequency, and together with 
R.sub.P, compensates the control loop. 
Now referring to FIG. 11, another aspect of the present invention is shown 
whereby a voltage stabilization feedback circuit 150 is used to regulate 
the level of boost converter 114 output voltage signal V.sub.OUT. A 
conventional method of creating power factor correction (PFC) circuity for 
electronic ballasts is by using boost converter circuity. However, it is 
usually impossible to adjust output voltage of the PFC circuity when the 
lamp load changes, during the course of dimming and when input voltage is 
varied. 
Generally, the output voltage signal V.sub.OUT of boost converter 114 
driven in continuous current mode and with constant frequency and supplied 
with an input voltage signal V.sub.IN can be described as follows: 
##EQU3## 
where D is the duty cycle of the operational voltage. However, less well 
recognized is that by changing frequency within a continuous current mode 
of operation, output voltage signal V.sub.OUT can be adjusted within 
certain limits. The expression for output voltage signal V.sub.OUT versus 
switching frequency F can be defined as follows: 
##EQU4## 
where t.sub.ON is the switch on-time and t.sub.OFF is the switch off-time 
which in turn can be defined as follows: 
##EQU5## 
where P.sub.OUT is the output power, L is the inductance of the boost 
inductor, .eta. is the efficiency, V.sub.IN is the input voltage and K is 
the input voltage form coefficient, as is conventionally understood. By 
rearranging these relations, output voltage signal V.sub.OUT can be 
written as follows: 
##EQU6## 
Thus, when output power P.sub.OUT, efficiency of the inverter .eta., boost 
inductance L, and input voltage signal V.sub.IN are fixed, the 
relationship between output voltage signal V.sub.OUT and frequency F has a 
hyperbolic character. Taking these principles into account, the inventor 
has determined that it is possible to adjust boost converter output 
voltage signal V.sub.OUT using microcontroller 120 and a voltage 
stabilizing feedback circuit 150. 
Specifically, feedback circuit 150 comprises capacitors C.sub.F1-F3, diodes 
D.sub.F1-F2, resistors R.sub.1-7, 555 timer circuit IC.sub.T, comparator 
132, and transistors Q.sub.F1-F2. As previously discussed, diode D.sub.F1, 
capacitor C.sub.F1 and resistor R.sub.F1 are configured to form a simple 
D/A converter 130 which serves to convert a digital signal V.sub.D 
produced by microcontroller 120 to a DC signal V.sub.DC. DC signal 
V.sub.DC is used to control the current source of the periodic generator 
128 by triggering transistor Q.sub.F1 which has its emitter connected to 
voltage signal V.sub.DD through resistor R.sub.F3 and to ground through 
capacitor C.sub.F2. Also as previously discussed, the collector of 
transistor Q.sub.P1 is coupled to TRIGGER (pin 2), THRESHOLD (pin 6) and 
DISCHARGE (pin 7) of 555 timer circuit IC.sub.T to produce periodic 
voltage signal V.sub.P as previously described in detail in respect of 
FIG. 5. The periodic voltage signal V.sub.P is input into the negative 
terminal of comparator 132. 
Further, the voltage output of the boost converter 114 V.sub.OUT is applied 
through diode D.sub.F2, capacitor C.sub.F3 and across voltage divider 
comprising resistors R.sub.F6 and R.sub.F7 into the positive terminal of 
comparator 132. The power factor correction signal V.sub.PFC produced by 
comparator 132 is used to control the output voltage signal V.sub.OUT 
produced by boost converter 114 by controlling the operation of PFC MOSFET 
transistor Q.sub.PFC. Specifically, power factor correction signal 
V.sub.PFC is used to control the current source comprising transistor 
Q.sub.F2 and resistor R.sub.F5 driven by voltage V.sub.DD. Thus, by 
appropriately varying the duty cycle of the PFC voltage signal being 
applied to transistor Q.sub.PFC, in such a feedback configuration it is 
possible to stabilize the output of boost converter 114, as is 
conventionally understood. 
As before, microcontroller 120 controls the frequency of power factor 
correction signal V.sub.PFC by controlling the current source connected to 
timer IC.sub.T. Further, the duty cycle of PFC signal V.sub.PFC is 
determined by the difference between boost converter output voltage signal 
V.sub.OUT and the voltage signal V.sub.P and varies itself to maintain 
output voltage when either the input voltage or the output load fluctuate. 
By changing the frequency and duty cycle of the periodic voltage signal 
V.sub.P, the PFC signal V.sub.PFC supplied to transistor Q.sub.PFC can 
regulate the output voltage V.sub.OUT of boost converter 114. Accordingly, 
a relatively low-speed microcontroller 120 can achieve stabilization of 
the boost converter 114. 
In use, control circuit 126 of ballast 110 utilizes a low-speed 
microcontroller 120 to successfully control the operation of MOSFET driver 
117 of a conventional inverter circuit 116 using a control voltage 
V.sub.C. Control circuit 126 can vary the duty cycle DC.sub.C and control 
voltage frequency f.sub.C of a control voltage signal V.sub.C to a high 
degree of resolution. Control circuit 126 generates a periodic voltage 
signal V.sub.P and modulates the periodic voltage V.sub.P so that certain 
DC levels are detected at differing frequencies. These DC levels are used 
to generate the control voltage V.sub.C by either comparing the periodic 
voltage signal V.sub.P with an analog DC signal through a comparator to 
produce control voltage V.sub.C or by passing the periodic voltage signal 
V.sub.P through a timer IC.sub.T to suitably trigger THRESHOLD (pin 6) of 
timer IC.sub.T to generate control voltage V.sub.C at OUTPUT (pin 3) of 
timer IC.sub.T. 
Further, the present invention allows for control circuit 126 to be powered 
using a number of convenient power sources within a conventional ballast 
110. First, microcontroller 120 can be powered by a power supply signal 
P.sub.S derived from half-bridge MOSFET driver 117 of inverter 116, using 
a conventionally known bootstrap power supply 142. Second, microcontroller 
120 can be powered by a power supply signal P.sub.S which is extracted 
from a boost inductor L.sub.B (FIG. 11) of boost converter 114. Finally, 
as microcontroller 120 can be powered by a power supply signal P.sub.S 
generated by a appropriately configured dedicated miniature switch mode 
power supply 146, such as a TOP210 three terminal off-line PWM switch 
integrated circuit manufactured by Power Integrations, Inc. 
Control circuit 126 can also be applied to stabilize the level of boost 
converter 114 output voltage signal V.sub.OUT by providing a feedback 
control signal V.sub.PFC to the PFC MOSFET Q.sub.PFC. Microcontroller 120 
is used to control the frequency and duty cycle of power factor correction 
signal V.sub.PFC by controlling the frequency and duty cycle of the 
periodic voltage signal V.sub.P and comparing the periodic voltage signal 
V.sub.P to the output voltage V.sub.OUT to determine a proper feedback 
control voltage signal V.sub.PFC. 
Accordingly, the present invention provides a universal lighting ballast 
control circuit which generates a wide range of different control signals 
to start, run and dim a wide variety of gas-discharge lamp types using an 
inexpensive low-speed microcontroller. By providing a high umber of 
continuously variable control signals, the present invention an eliminate 
visible steps of light intensity which would otherwise occur when dimming 
a lamp. Further, the microcontroller can be powered within a typical 
ballast without conventionally known power supply problems and associated 
expenses. Finally, the present invention can be used to regulate the boost 
converter output voltage to control and stabilize the operation of the 
power factor correction circuitry. 
As will be apparent to persons skilled in the art, various modifications 
and adaptations of the structure described above are possible without 
departure from the present invention, the scope of which is defined in the 
appended claims.