A contactless electricity-supplying device (20) includes a secondary winding (201) to which electric power is supplied from a primary winding (101), by an AC power supply (6). The impedance characteristic (Z) of Z1 in regard to the frequency has a local maximum (ZMAX) near the frequency (f0) of the fundamental wave component of aforementioned AC power supply (6); and the impedance characteristic (Z) of Z2 in regard to the frequency has the aforementioned frequency (f0) of the fundamental wave component to be between, a frequency (fMAX) that has its local maximum (ZMAX) nearest to aforementioned frequency (f0) of the fundamental wave component, and a frequency (fMIN) that has its local minimum (ZMin) nearest to the frequency (f0) of the fundamental wave component. Z1 indicates that the coupling coefficient (k) between aforementioned primary winding (101) and aforementioned secondary winding (201) is a prescribed value (0.3), and that it is an impedance of just the primary side (Z1) as seen from the output side of aforementioned AC power supply (6); and Z2 indicates that the coupling coefficient (k) between aforementioned primary winding (101) and aforementioned secondary winding (201) is the aforementioned prescribed value (0.3), and that it is an impedance of just the secondary side (Z2) as seen from the side of a load (72) to be connected to aforementioned secondary winding (201).

TECHNICAL FIELD

The present invention relates to a contactless electricity-supplying device.

BACKGROUND ART

A contactless electricity-supplying device is conventionally known which has such a structure that a series capacitor is connected to a primary winding driven by an AC power supply and a parallel capacitor is connected to a secondary winding, where the value of each of the series and parallel capacitors is so set based on a certain expression that a transformer of the known contactless electricity-supplying device is substantially equivalent to an ideal transformer.

CITATION LIST

Patent Literature

SUMMARY OF INVENTION

Technical Problem

However, in the conventional contactless electricity-supplying device, the capacitor and the like are so set as to accomplish high efficiency on the premise that a coupling coefficient between the primary winding and the secondary winding is constant, therefore, when the coupling coefficient changes, the efficiency of the transformer greatly changes, which was a problem.

Therefore, the present invention provides a contactless electricity-supplying device capable of reducing change of efficiency of a transformer even when the coupling state changes.

Solution to Problem

According to the present invention, an impedance characteristic of Z1relative to a frequency has the maximum in a vicinity of a frequency of a fundamental wave component of the alternating current power supply, an impedance characteristic of Z2relative to the frequency has the frequency of the fundamental wave component between, a frequency that has the maximum nearest to the frequency of the fundamental wave component (NA) and a frequency that has the minimum nearest to the frequency of the fundamental wave component, to thereby solve the above problem.

Advantageous Effects of Invention

According to the present invention, the phase characteristic of an impedance (relative to the frequency) viewed from an output side of an alternating current power supply so changes as to rotate around an area in the vicinity of a fundamental wave frequency in accordance with fluctuation of a coupling coefficient. Therefore, when the impedance is set in accordance with the coupling coefficient, the fluctuation band of the phase of the impedance becomes small, as a result, making it possible to suppress decrease of the efficiency.

DESCRIPTION OF EMBODIMENTS

Hereinafter, embodiments of the present invention will be explained based on drawings.

First Embodiment

As an example of a contactless power circuit device according to the first embodiment of the present invention, a contactless electricity-supplying device20used together with a vehicle-oriented cell and a power load of an electric vehicle and the like will be explained.

FIG. 1shows an electric circuit diagram of the contactless electricity-supplying device20. The contactless electricity-supplying device20according to the first embodiment has a high-frequency AC (alternating current) power supply circuit6, a contactless electricity-supplying portion10for contactlessly supplying an electric power outputted from the high-frequency AC power supply circuit6, and a load7to which the electric power is supplied by the contactless electricity-supplying portion10.

The high-frequency AC power supply circuit6has a 3-phase AC power supply64, a rectifier61connected to the 3-phase AC power supply64and adapted to rectify a 3-phase alternating current to a direct current, and a voltage-type inverter63connected to the rectifier61via a smoothing capacitor62and adapted to invert the rectified current to a high-frequency electric power. The rectifier61has such a structure that a pair of a diode61aand a diode61b, a pair of a diode61cand a diode61dand a pair of a diode61eand a diode61fare connected in parallel (three rows) and each of three outputs of the 3-phase AC power supply64is connected to one of respective intermediate connecting points of the above three pairs. The voltage-type inverter63has such a structure that a first series circuit having a switching element63aand a switching element63b(like switching element63a) and a second series circuit having a switching element63c(like switching element63a) and a switching element63d(like switching element63a) are connected in parallel, where each of the switching elements63a,63b,63cand63dhas such a structure that a diode is inversely connected in parallel to a power transistor and the like of MOSFET. The voltage-type inverter63is connected with the rectifier61via the smoothing capacitor62. Then, an intermediate connecting point between the switching element63aand the switching element63band an intermediate connecting point between the switching element63cand the switching element63dare each connected with a power transmission circuit portion100which is a primary side of the contactless electricity-supplying portion10. The voltage-type inverter63supplies an alternating power of about several kHz to 100 kHz to the contactless electricity-supplying portion10.

The contactless electricity-supplying portion10has the power transmission circuit portion100as an input side of a transformer and an incoming circuit portion200as an output side of the transformer. The power transmission circuit portion100has a primary winding101and a capacitor102which is connected in parallel to the primary winding101. The incoming circuit portion200has a secondary winding201, a capacitor202which is connected in parallel to the secondary winding201and a capacitor203which is connected in series to a parallel circuit composed of the secondary winding201and capacitor202.

The load portion7has a rectifying portion71for rectifying into a direct current the alternating power supplied from the contactless electricity-supplying portion10and a load72which is connected to the rectifying portion71. The rectifier71has such a structure that a pair of a diode71aand a diode71bare connected in parallel to a pair of a diode71cand a diode71d. Each of two outputs of the incoming circuit portion200is connected with one of respective intermediate connecting points of the above two pairs. Then, outputs of the rectifying portion71are connected to the load72.

Then, referring toFIG. 2andFIG. 3, an explanation is made about a mutual inductance M of the primary winding101and secondary winding201when the contactless power circuit device (contactless electricity-supplying device20) is provided for a vehicle and a parking area.

According to the first embodiment, the incoming circuit portion200(including the secondary winding201) and the load portion7are provided, for example, for the vehicle while the power transmission circuit portion100(including the primary winding101) and the high-frequency AC power supply circuit6are provided, for example, for the parking area as a ground side. In the case of an electric vehicle, the load72corresponds, for example, to a secondary battery. The secondary winding201is provided for, for example, a chassis of the vehicle. Then, a driver of the vehicle parks the vehicle such that the secondary winding201is positioned on (above) the primary winding101, to thereby supply an electric power from the primary winding101to the secondary winding201, thus charging the secondary battery included in the load72.

FIG. 2aandFIG. 2beach show a plan view and a perspective view of the primary winding101and secondary winding201. InFIG. 2aandFIG. 2b, an X-axis and a Y-axis denote a flat surface direction of the primary winding101and secondary winding201while a z-axis denotes a height direction of the primary winding101and secondary winding201. InFIG. 2aandFIG. 2b, “(a)” denotes the plan view of the primary winding101and secondary winding201, “(b)” denotes the perspective view of the secondary winding201and “(c)” denotes the perspective view of the primary winding101. Now, for convenience sake, the primary winding101and secondary winding201have the same circular configuration. However, it is not necessary to keep such circular configuration and it is not necessary to form the same configuration between the primary winding101and the secondary winding201, according to the first embodiment.

As shown inFIG. 2a, in the X-axis and Y-axis directions which form the flat surface direction, it is preferable that the vehicle be parked such that the secondary winding201coincides with the primary winding101. However, depending on the driver's technique, a position of the primary winding101relative to the secondary winding201is, as the case may be, deviated in the flat surface direction. Moreover, the height of the vehicle differs with type of vehicle, therefore, the height of the primary winding101and the height of the secondary winding201are different from each other with the vehicle height.

FIG. 3ashows changes of the mutual inductance M relative to the deviation of the secondary winding201in the X-axis direction inFIG. 2, andFIG. 3bshows changes of the mutual inductance M relative to the deviation of the secondary winding201in the Z-axis direction inFIG. 2. As shown inFIG. 2a, when the center of the primary winding101coincides with the center of the secondary winding201, a leak magnetic flux between the primary winding101and the secondary winding201is small, thereby the value of the X-axis inFIG. 3acorresponds to zero and the mutual inductance M or a coupling coefficient k is larger. On the other hand, as shown inFIG. 2bcompared withFIG. 2a, when the position of the primary winding101is deviated from the position of the secondary winding201in the X-axis direction, the leak magnetic flux is larger, thereby, as shown inFIG. 3a, the mutual inductance M or the coupling coefficient k is smaller. Moreover, when the deviation of the primary winding101relative to the secondary winding201in the Z-axis (height) direction is larger, as shown inFIG. 3b, the mutual inductance M or the coupling coefficient k is smaller.

Now, a contactless power-supplying device and the like adopted for charging home electric appliances (such as electric toothbrush, shaver and the like) or mobile devices which are made cordless have such a structure that the primary winding101does not move relative to the secondary winding201, therefore, assumption of fluctuation of the mutual inductance M as stated above is not necessary. Thus, on the premise that the mutual inductance M is fixed, the circuit of the capacitors and inductors included in the power transmission circuit portion100and incoming circuit portion200are designed such that the electric power can be efficiently supplied to the incoming circuit portion200under the certain coupling coefficient k.

FIG. 4shows a phase of an input impedance (Zin) viewed from an output side of the AC power supply in the contactless electricity-supplying device in the Patent Document 1, showing a difference in coupling coefficient between the primary winding and the secondary winding. Herein, f0denotes a frequency of a fundamental wave component of the AC power supply (hereinafter, referred to as fundamental wave frequency). When, as a switching power source, an output of, for example, the inverter is connected to the power transmission circuit portion100, the fundamental wave frequency f0depends on a switching frequency of a switching element which drives the inverter. According to the first embodiment, the fundamental wave frequency f0depends on the switching frequencies of switching elements63ato63d.

As showing inFIG. 4, when the coupling coefficient k is 0.1, the phase characteristic of the input impedance is zero in the vicinity of the fundamental wave frequency (f0), therefore, a power factor of supplied power is 1, thus making it possible to efficiently supply the electric power to the load. On the other hand, when the setting of the capacitor-inductor included in the power transmission circuit portion100and incoming circuit portion200are unchanged and the position of the primary winding101is deviated from the secondary winding201to thereby change the coupling coefficient k, the phase in the vicinity of the fundamental wave frequency (f0) is delayed to a large extent when the coupling coefficient k is 0.2. Thus, the power factor of the supplied power is decreased, thereby decreasing efficiency of the power-supplying. Moreover, when the coupling coefficient k is changed to become 0.3, the phase in the vicinity of the fundamental wave frequency (f0) is further delayed, to thereby decrease the power factor of the supplied power, thus decreasing the efficiency of the power-supplying.

That is, when the electric power is inputted in a position (primary winding101and secondary winding201) causing the coupling coefficient k=0.1, the power can be efficiently supplied. However, when the position of the primary winding101is deviated from the position of the secondary winding201to thereby fluctuate the coupling coefficient k, the electric power supplied to the secondary side is remarkably decreased, resulting in decreased efficiency.

According to the first embodiment, when the coupling coefficient k between the primary winding101and the secondary winding201takes a certain value, the phase characteristic of the input impedance (Zin) of the contactless electricity-supplying portion10when viewed from the high-frequency AC power supply circuit6side is in parallel with the frequency axis in the vicinity of the frequency (f0) of the fundamental wave component of the high-frequency AC power supply circuit6. As shown inFIG. 5, with respect to the contactless electricity-supplying device20according to the first embodiment, when the coupling coefficient k is 0.3, the phase characteristic of the input impedance (Zin) is in parallel with the frequency axis in the vicinity of the fundamental wave frequency (f0). In other words, the phase characteristic of the input impedance (Zin) is uneven and is flat in the vicinity of the fundamental wave frequency (f0). In this case, the phase of the input impedance (Zin) is close to zero, thereby the power factor of the supplied power is close to 1, and the contactless electricity-supplying portion10efficiently supplies the power to the secondary side. In addition, it is not necessary that “in parallel with frequency axis” is exactly parallel with the frequency axis and therefore “in parallel with frequency axis” can include a slight inclination.

Then, with respect to the contactless electricity-supplying device20according to the first embodiment, when the coupling coefficient k is fluctuated to become 0.2, the phase characteristic of the input impedance (Zin) changes in such a manner as to rotate around an area in the vicinity of the fundamental wave frequency (f0), and the phase in the vicinity of the fundamental wave frequency (f0) does not change to a large extent compared with when the coupling coefficient k is 0.3, that is, the phase in the vicinity of the fundamental wave frequency (f0) is kept close to zero. Moreover, even when the coupling coefficient k is fluctuated to become 0.1, the phase characteristic of the input impedance (Zin) changes in such a manner as to rotate around the area in the vicinity of the fundamental wave frequency (f0), and the phase in the vicinity of the fundamental wave frequency (f0) does not change to a large extent compared with when the coupling coefficient k is 0.2 or 0.3, that is, the phase in the vicinity of the fundamental wave frequency (f0) is kept close to zero.

In other words, with respect to the phase characteristic of the input impedance (Zin) according to the first embodiment, when the coupling coefficient k between the primary winding101and the secondary winding201takes the certain value (k=0.3 inFIG. 5), the difference between the maximum (ZMAX) and minimum (ZMIN) of the phase characteristic of the input impedance (Zin) of the contactless electricity-supplying portion10is close to zero. Herein, especially, when the phase characteristic has a plurality of maximums (ZMAX), the maximum (ZMAX) denotes a value corresponding to the frequency that is nearest to the fundamental wave frequency (f0). Also, the same is true of the minimum (ZMIN). InFIG. 5, in the case of the frequency (fMAX) denoted by a point P1, the phase has the maximum (ZMAX) and in the case of the frequency (fMIN) denoted by a point P2, the phase has the minimum (ZMIN). Then, as shown inFIG. 5, the difference between the maximum (ZMAX) and the minimum (ZMIN) is close to zero.

In other words, with respect to the phase characteristic of the input impedance (Zin) according to the first embodiment, when the coupling coefficient k between the primary winding101and the secondary winding201takes the certain value (k=0.3 inFIG. 5), the phase characteristic of the input impedance (Zin) of the contactless electricity-supplying portion10has such a feature that an inflection point is in the vicinity of the fundamental wave frequency (f0) and a tangent line of the inflection point is in parallel with the frequency axis. InFIG. 5, Q denotes the inflection point Q which is in the vicinity of the fundamental wave frequency (f0). Moreover, as shown inFIG. 5, the tangent line of the inflection point Q is in parallel with the frequency axis. In addition, it is not necessary that “in parallel with frequency axis” is exactly parallel with the frequency axis and therefore “in parallel with frequency axis” can include a slight inclination.

As stated above, designing the capacitor-inductor included in the contactless electricity-supplying portion10can allow the contactless electricity-supplying device20according to the first embodiment to obtain the above-described phase characteristic of the input impedance (Zin), with the variable coupling coefficient k taking the certain value. Then, explained next referring toFIG. 6is an example of a circuit where the input impedance (Zin) has the above-described phase characteristic.

FIG. 6ashows the impedance characteristic (Z) and phase characteristic (Φ) of the impedance (Z1) of only the primary side relative to the frequency when viewed from the high-frequency AC power supply circuit6side in the contactless electricity-supplying portion10shown inFIG. 1. Moreover,FIG. 6bshows the impedance characteristic (Z) and phase characteristic (Φ) of the impedance (Z2) of only the secondary side relative to the frequency when viewed from the load portion7side in the contactless electricity-supplying portion10shown inFIG. 1. The impedance (Z1) of only the primary side and the impedance (Z2) of only the secondary side may be respectively calculated, with the mutual inductance M as zero.

As shown inFIG. 6a, the impedance characteristic (Z) of the impedance (Z1) of only the primary side has the maximum in the vicinity of the fundamental wave frequency (f0). Moreover, the phase characteristic (Φ) of the impedance (Z1) of only the primary side keeps about +90 degrees up to the area in the vicinity of the fundamental wave frequency (f0), then diverges the phase inclination in the vicinity of the fundamental wave frequency (f0), and then keeps about −90 degrees over the area in the vicinity of the fundamental wave frequency (f0).

As shown inFIG. 6b, the impedance characteristic (Z) of the impedance (Z2) of only the secondary side has the fundamental wave frequency (f0) between the frequency (fMAX) taking the maximum (ZMAX) and the frequency (fMIN) taking the minimum (ZMIN). Herein, especially, in the case of the phase characteristic having a plurality of maximums (ZMAX), the maximum (ZMAX) denotes a value corresponding to the frequency that is nearest to the fundamental wave frequency (f0). Also, the same is true of the minimum (ZMIN). The phase characteristic (Φ) of the impedance (Z2) of only the secondary side has two points (point P1and point P2shown inFIG. 6b) at which the phase inclinations are diverged, and has a portion (between the two points P1and P2) which is parallel to the frequency axis, where the fundamental wave frequency component (f0) is present between the two points P1and P2. In other words, the above phase characteristic (Φ) makes a turn around the area in the vicinity of the fundamental wave frequency (f0) and comes back.

Then, the impedance (Z1) of only the primary side having the characteristic shown inFIG. 6ais set to the power transmission circuit portion100and the impedance (Z2) of only the secondary side having the characteristic shown inFIG. 6bis set to the incoming circuit portion200, to thereby set the contactless electricity-supplying portion10having the above characteristic, as shown inFIG. 5.

As stated above, according to the first embodiment, the position between the primary winding101and the position of the secondary winding201is fluctuated and the coupling coefficient k is fluctuated, however, at the certain value (k=0.3 inFIG. 5), the phase characteristic of the input impedance (Zin) is in parallel with the frequency axis in the vicinity of the fundamental wave frequency (f0). In other words, according to the first embodiment, when the coupling coefficient k takes the certain value, the phase characteristic of the input impedance (Zin) of the contactless electricity-supplying portion10when viewed from the high-frequency AC power supply circuit6side is made parallel to the frequency axis in the vicinity of the fundamental wave frequency (f0) of the high-frequency AC power supply circuit6. Still, in other words, according to the first embodiment, when the coupling coefficient k takes the certain value, the difference between the maximum (ZMAX) and minimum (ZMIN) of the phase characteristic of the input impedance (Zin) of the contactless electricity-supplying portion10is made close to zero. Furthermore, in other words, according to the first embodiment, when the coupling coefficient k takes the certain value, the phase characteristic of the input impedance (Zin) of the contactless electricity-supplying portion10has the inflection point Q in the vicinity of the fundamental wave frequency (f0) and the tangent line of the inflection point Q is in parallel with the frequency axis.

With this, when the coupling coefficient k fluctuates from the certain value, the phase characteristic of the input impedance (Zin) fluctuates in such a manner as to rotate around the point of taking the phase (Φ0) which corresponds to the fundamental wave frequency (f0), thereby making it possible to decrease the fluctuation band of the phase of the input impedance (Zin) relative to the fundamental wave frequency (f0) and to suppress (even when the coupling coefficient k fluctuates) the fluctuation of the phase (Φ0).

According to the first embodiment, the phase characteristic of the input impedance (Zin) having the coupling coefficient k of the certain value has the phase in the vicinity of zero relative to the fundamental wave frequency (f0), thereby making it possible to enhance the efficiency of supplying the electric power to the incoming circuit portion200and also making it possible (when the coupling coefficient k fluctuates from the certain value) to supply the electric power with the high efficiency kept.

According to the first embodiment, with respect to the phase characteristic of the input impedance (Zin) having the coupling coefficient k of the certain value, when the coupling coefficient k is changed within a constant range relative to the certain value, the phase relative to the fundamental wave frequency (f0) fluctuates in the vicinity of zero. With this, even when the coupling coefficient k is fluctuated from the certain value within the constant range, a high power factor can be kept according to the first embodiment, therefore, as a result, making it possible to keep high efficiency against the fluctuation of the coupling coefficient k and to supply the electric power.

In addition, the above constant range can be determined, for example, in the following manner: when the efficiency of supplying the electric power by fluctuating the coupling coefficient k fluctuates, the coupling coefficient k corresponding to an allowable range of the efficiency is set in advance. The allowable range is properly set depending on performance of the used primary winding101, used secondary winding201, standard electric power of the secondary battery as the load72, and the like.

Hereinafter explained referring toFIG. 7toFIG. 10are points and the like that the above impedance characteristic (Z) or phase characteristic (Φ) allows the contactless electricity-supplying device20according to the first embodiment to keep higher electricity-supplying efficiency compared with the conventional contactless electricity-supplying device.

FIG. 7shows electricity-supplying efficiency relative to the coupling coefficient k, with respect to the contactless electricity-supplying device20according to the first embodiment of the present invention compared with the conventional contactless electricity-supplying device. Herein, the efficiency (%) inFIG. 7denotes a ratio of an output power from the contactless electricity-supplying portion10relative to an input power to the contactless electricity-supplying portion10. In addition, the frequency is the fundamental wave frequency of the AC power supply connected to the input side.

With respect to the conventional contactless electricity-supplying device, the circuit of inductor-capacitor included in the contactless electricity-supplying portion is designed focusing on the electricity-supplying efficiency, such that the power factor is improved on the premise of high coupling coefficient k. Thus, the electricity-supplying efficiency is high at the high coupling coefficient k. However, as shown inFIG. 7, when the coupling coefficient k is gradually decreased, the electricity-supplying efficiency is rapidly worsened.

On the other hand, with respect to the contactless electricity-supplying device20according to the first embodiment of the present invention, the coupling coefficient k, when decreased, can keep the electricity-supplying efficiency higher than that of the conventional technology. Moreover, according to the first embodiment, the coupling coefficient k which is small can accomplish a high electricity-supplying efficiency.

FIG. 8shows electricity-supplying efficiency changes when the position of the primary winding101relative to the secondary winding201is deviated in the X-axis direction shown inFIG. 2borFIG. 3a. Herein, the electricity-supplying efficiency is the same as that shown inFIG. 7.

With respect to the conventional contactless electricity-supplying device, when the position of the primary winding101relative to the secondary winding201is deviated in the X-axis direction, the coupling coefficient k is decreased, therefore, when the deviation is enlarged, the efficiency is rapidly decreased at a certain point. On the other hand, with respect to the contactless electricity-supplying device20according to the first embodiment of the present invention, even when the position of the primary winding101relative to the secondary winding201is deviated, the efficiency can be kept high. Then, with the efficiency necessary for the system as 80% (service condition), 80% or over is defined as an allowable efficiency for the system. In this case, the contactless electricity-supplying device20according to the first embodiment can extend the allowability of the decreased efficiency (relative to the deviation in the X-axis direction) by about 1.5 times compared with the conventional technology.

FIG. 9shows the current necessary on the AC power supply side relative to the coupling coefficient k, where such current is observed when obtaining a constant output power is necessary (for example, when a constant power of 10 KW be supplied to the load72). With respect to the conventional contactless electricity-supplying device, when the coupling coefficient k is high, the necessary power can be supplied to the incoming circuit portion200even when the current flowing the power transmission circuit portion100is small, however, when the coupling coefficient k is low, the current flowing the power transmission circuit portion100is large, thereby increasing the loss which occurs to the primary winding101and the like in the circuit. On the other hand, with the contactless electricity-supplying device20according to the first embodiment, when the coupling coefficient k is low, the current flowing the power transmission circuit portion100can be suppressed small, thereby making it possible to efficiently supply the power to the incoming circuit portion200.

In addition, according to the first embodiment, when the coupling coefficient k is 0.3, the primary winding101, capacitor102, secondary winding201, capacitor202and capacitor203included in the contactless electricity-supplying portion10are so set that the phase characteristic or impedance characteristic of the input impedance (Zin) has the above characteristics, however, the coupling coefficient k failing to meet 0.3 is allowed.

That is, under a condition that the position of the secondary winding201relative to the primary winding101is changed, when the phase characteristic or impedance characteristic of the input impedance (Zin) takes the above characteristics in the fluctuation band of the assumed coupling coefficient k, the coupling coefficient k obtained in this case is defined as the certain value. Moreover, designing the circuit such that the phase of the input impedance (Zin) relative to the fundamental wave frequency (f0) in this case is closer to zero can enhance the efficiency.

In addition, the impedance characteristic (Z) of the impedance (Z1) of only the primary side or the impedance characteristic (Z) of the impedance (Z2) of only the secondary side may have an extreme value other than the maximum (ZMAX) or minimum (ZMIN) that is nearest to the fundamental wave frequency (f0). The high-frequency AC power supply circuit6according to the first embodiment corresponds to “alternating current power supply” of the present invention, the capacitor102according to the first embodiment corresponds to “first capacitor” of the present invention, and the capacitor202and capacitor203according to the first embodiment respectively correspond to “third capacitor” and “fourth capacitor” of the present invention.

Second Embodiment

FIG. 10is a circuit portion showing the contactless electricity-supplying device20according to the second embodiment of the present invention. Compared with the first embodiment described above, the second embodiment differs in using a circuit that is different from the circuit of the power transmission circuit portion100inFIG. 1. Other than the above in terms of structure, the second embodiment is substantially the same as the first embodiment, and therefore descriptions of the first embodiment will be properly quoted according to the second embodiment.

As shown inFIG. 10, a power transmission circuit portion311according to the second embodiment has such a structure that an inductor301is connected in series to a parallel circuit having the primary winding101and the capacitor102.

FIG. 11shows the impedance characteristic (Z) and phase characteristic (Φ) relative to the frequency of the impedance (Z1) of only the primary side when viewed from the high-frequency AC power supply circuit6side, with the mutual inductance M as zero.

As shown inFIG. 11, the impedance characteristic (Z) of the impedance (Z1) according to the second embodiment has the maximum (ZMAX) in the vicinity of the fundamental wave frequency (f0) and has the minimum (ZMIN) relative to the frequency higher than the fundamental wave frequency (f0). Moreover, the phase characteristic (Φ) of the impedance (Z1) according to the second embodiment has the phase inclination diverged in the vicinity of the fundamental wave frequency (f0) and has a point at which the phase inclination is further diverged at the frequency higher than the fundamental wave frequency (f0) other than the area in the vicinity of the fundamental wave frequency (f0).

When the contactless electricity-supplying portion10is provided with the power transmission circuit portion311having the impedance characteristic (Z) or phase characteristic (Φ) shown inFIG. 11, the phase characteristic of the input impedance (Zin) has, referring toFIG. 5, such a characteristic as shown according to the first embodiment. With this, in the contactless electricity-supplying device20according to the second embodiment, the fluctuation band of the phase of the input impedance (Zin) relative to the fundamental wave frequency (f0) is small even when the coupling coefficient k fluctuates, thus suppressing the fluctuation of the phase (Φ0), as a result, keeping the power factor high and making it possible to efficiently supply the electric power.

In addition, the inductor301according to the second embodiment corresponds to “first inductor” of the present invention.

Third Embodiment

FIG. 12is a circuit portion showing the contactless electricity-supplying device20according to the third embodiment of the present invention. Compared with the first embodiment described above, the third embodiment differs in using a circuit that is different from the circuit of the power transmission circuit portion100inFIG. 1. Other than the above in terms of structure, the third embodiment is substantially the same as the first embodiment, and therefore descriptions of the first embodiment will be properly quoted according to the third embodiment.

As shown inFIG. 12, a power transmission circuit portion312according to the third embodiment has such a structure that the capacitor302is connected in series to the parallel circuit having the primary winding101and the capacitor102.

FIG. 13shows the impedance characteristic (Z) and phase characteristic (Φ) relative to the frequency of the impedance (Z1) of only the primary side when viewed from the high-frequency AC power supply circuit6side, with the mutual inductance M as zero.

As shown inFIG. 13, the impedance characteristic (Z) of the impedance (Z1) according to the third embodiment has the maximum (mountain) in the vicinity of the fundamental wave frequency (f0) and has the minimum relative to the frequency lower than the fundamental wave frequency (f0) in an area other than the area in the vicinity of the fundamental wave frequency (f0). Moreover, the phase characteristic (Φ) of the impedance (Z1) has the phase inclination diverged in the vicinity of the fundamental wave frequency (f0) and has a point at which the phase inclination is further diverged at a frequency lower than the fundamental wave frequency (f0) other than the area in the vicinity of the fundamental wave frequency (f0).

When the contactless electricity-supplying portion10is provided with the power transmission circuit portion312having the impedance characteristic (Z) and phase characteristic (Φ) shown inFIG. 13, the phase characteristic of the input impedance (Zin) has, referring toFIG. 5, such a characteristic as shown according to the first embodiment. With this, in the contactless electricity-supplying device20according to the third embodiment, the fluctuation band of the phase of the input impedance (Zin) relative to the fundamental wave frequency (f0) is small even when the coupling coefficient k fluctuates, thus suppressing the fluctuation of the phase (Φ0), as a result, keeping the power factor high and making it possible to efficiently supply the electric power.

In addition, the capacitor302according to the third embodiment corresponds to “second capacitor” of the present invention.

Fourth Embodiment

FIG. 14is a circuit portion showing the contactless electricity-supplying device20according to the fourth embodiment of the present invention. Compared with the first embodiment described above, the fourth embodiment differs in using a circuit that is different from the circuit of the power transmission circuit portion100inFIG. 1. Other than the above in terms of structure, the fourth embodiment is substantially the same as the first embodiment, and therefore descriptions of the first embodiment will be properly quoted according to the fourth embodiment.

As shown inFIG. 14, a power transmission circuit portion313according to the fourth embodiment has such a structure that the inductor301is connected to a first end of the parallel circuit having the primary winding101and the capacitor102and the capacitor302is connected to a second end of the above parallel circuit.

FIG. 15shows the impedance characteristic (Z) and phase characteristic (Φ) relative to the frequency of the impedance (Z1) of only the primary side when viewed from the high-frequency AC power supply circuit6side, with the mutual inductance M as zero.

As shown inFIG. 15, the impedance characteristic (Z) of the impedance (Z1) according to the fourth embodiment has the maximum (mountain) in the vicinity of the fundamental wave frequency (f0) and has two minimums in two respective areas other than the area in the vicinity of the fundamental wave frequency (f0). Moreover, the phase characteristic (Φ) of the impedance (Z1) has the phase inclination diverged in the vicinity of the fundamental wave frequency (f0) and has two points at which the phase inclinations are further diverged at respective frequencies other than the area in the vicinity of the fundamental wave frequency (f0).

When the contactless electricity-supplying portion10is provided with the power transmission circuit portion313having the impedance characteristic (Z) or phase characteristic (Φ) shown inFIG. 15, the phase characteristic of the input impedance (Zin) has, referring toFIG. 5, such a characteristic as shown according to the first embodiment. With this, in the contactless electricity-supplying device20according to the fourth embodiment, the fluctuation band of the phase of the input impedance (Zin) relative to the fundamental wave frequency (f0) is small even when the coupling coefficient k fluctuates, thus suppressing the fluctuation of the phase (Φ0), as a result, keeping the power factor high and making it possible to efficiently supply the electric power.

In addition, the inductor301according to the fourth embodiment corresponds to “first inductor” of the present invention and the capacitor302according to the fourth embodiment corresponds to “second capacitor” of the present invention.

Fifth Embodiment

FIG. 16is a circuit portion showing the contactless electricity-supplying device20according to the fifth embodiment of the present invention. Compared with the first embodiment described above, the fifth embodiment differs in using a circuit that is different from the circuit of the incoming circuit portion200inFIG. 1. Other than the above in terms of structure, the fifth embodiment is substantially the same as the first embodiment, and therefore descriptions of the first embodiment will be properly quoted according to the fifth embodiment.

As shown inFIG. 16, an incoming circuit portion411has such a structure that the capacitor401is connected to the secondary winding201in series and the capacitor202is connected in parallel to the serial circuit composed of the secondary winding201and capacitor401.

FIG. 17shows the impedance characteristic (Z) and phase characteristic (Φ) relative to the frequency of the impedance (Z2) of only the secondary side when viewed from the load portion7side, with the mutual inductance M as zero.

As shown inFIG. 17, the impedance characteristic (Z) of the impedance (Z2) has the fundamental wave frequency (f0) between the frequency (fMAX) taking the maximum (ZMAX) and the frequency (fMIN) taking the minimum (ZMIN).

The phase characteristic (Φ) of the impedance (Z2) has two points (point P1and point P2shown inFIG. 17) at which the phase inclinations are diverged and has a portion (between the two points P1, P2) which is parallel to the frequency axis, where the fundamental wave frequency (f0) is present between the two points P1and P2. In other words, the phase characteristic (Φ) makes a turn around the area in the vicinity of the fundamental wave frequency (f0) and comes back.

When the contactless electricity-supplying portion10is provided with the incoming circuit portion411having the impedance characteristic (Z) or phase characteristic (Φ) shown inFIG. 17, the phase characteristic of the input impedance (Zin) has, referring toFIG. 5, such a characteristic as shown according to the first embodiment. With this, in the contactless electricity-supplying device20according to the fifth embodiment, the fluctuation band of the phase of the input impedance (Zin) relative to the fundamental wave frequency (f0) is small even when the coupling coefficient k fluctuates, thus suppressing the fluctuation of the phase (Φ0), as a result, keeping the power factor high and making it possible to efficiently supply the electric power.

In addition, the capacitor401according to the fifth embodiment corresponds to “fifth capacitor” of the present invention and the capacitor202according to the fifth embodiment corresponds to “third capacitor” of the present invention.

Sixth Embodiment

FIG. 18is a circuit portion showing the contactless electricity-supplying device20according to the sixth embodiment of the present invention. Compared with the first embodiment described above, the sixth embodiment differs in using a circuit that is different from the circuit of the incoming circuit portion200inFIG. 1. Other than the above in terms of structure, the sixth embodiment is substantially the same as the first embodiment, and therefore descriptions of the first embodiment will be properly quoted according to the sixth embodiment.

As shown inFIG. 18, an incoming circuit portion412according to the sixth embodiment has such a structure that the capacitor401is connected in series to the secondary winding201and the inductor402is connected in parallel to the serial circuit composed of the secondary winding201and capacitor401.

FIG. 19shows the impedance characteristic (Z) and phase characteristic (Φ) relative to the frequency of the impedance (Z2) of only the secondary side when viewed from the load portion7side, with the mutual inductance M as zero.

As shown inFIG. 19, the impedance characteristic (Z) of the impedance (Z2) has the fundamental wave frequency (f0) between the frequency (fMAX) taking the maximum (ZMAX) and the frequency (fMIN) taking the minimum (ZMIN). Unlike the impedance characteristic (Z) shown inFIG. 6baccording to the first embodiment,FIG. 19of the sixth embodiment shows that the frequency (fMAX) is lower than the fundamental wave frequency (f0) and the frequency (fMIN) is higher than the fundamental wave frequency (f0).

The phase characteristic (Φ) of the impedance (Z2) has two points (point P1and point P2shown inFIG. 19) at which the phase inclinations are diverged and has a portion (between the two points P1, P2) which is parallel to the frequency axis, where the fundamental wave frequency component (f0) is present between the two points P1and P2. In other words, the above phase characteristic (Φ) makes a turn around the area in the vicinity of the fundamental wave frequency (f0) and comes back.

When the contactless electricity-supplying portion10is provided with the incoming circuit portion412having the impedance characteristic (Z) or phase characteristic (Φ) shown inFIG. 19, the phase characteristic of the input impedance (Zin) has, referring toFIG. 5, such a characteristic as shown according to the first embodiment. With this, in the contactless electricity-supplying device20according to the sixth embodiment, the fluctuation band of the phase of the input impedance (Zin) relative to the fundamental wave frequency (f0) is small even when the coupling coefficient k fluctuates, thus suppressing the fluctuation of the phase (Φ0), as a result, keeping the power factor high and making it possible to efficiently supply the electric power.

In addition, the capacitor401according to the sixth embodiment corresponds to “fifth capacitor” of the present invention and the inductor402according to the sixth embodiment corresponds to “third inductor” of the present invention.

Moreover, replacing the capacitor401with the inductor402and thereby connecting the inductor402in series to the secondary winding201while connecting the capacitor401with the series circuit composed of the secondary winding201and inductor402is allowed. In this case, the capacitor401according to the sixth embodiment corresponds to “third capacitor” of the present invention and the inductor402according to the sixth embodiment corresponds to “fourth inductor” of the present invention.

Seventh Embodiment

FIG. 20is a circuit portion showing the contactless electricity-supplying device20according to the seventh embodiment of the present invention. Compared with the first embodiment described above, the seventh embodiment differs in using a circuit that is different from the circuit of the incoming circuit portion200inFIG. 1. Other than the above in terms of structure, the seventh embodiment is substantially the same as the first embodiment, and therefore descriptions of the first embodiment will be properly quoted according to the seventh embodiment.

As shown inFIG. 20, an incoming circuit portion413according to the seventh embodiment has such a structure that the capacitor401is connected in series to the secondary winding201and the capacitor403is connected in an area between a connection (between a first end of the secondary winding201and the capacitor401) and a second end of the secondary winding201. Then, the capacitor202is connected in parallel relative to the serial circuit which has i) the parallel circuit composed of the secondary winding201and capacitor403and ii) the capacitor401.

FIG. 21shows the impedance characteristic (Z) and phase characteristic (Φ) relative to the frequency of the impedance (Z2) of only the secondary side when viewed from the load portion7side, with the mutual inductance M as zero.

As shown inFIG. 21, the impedance characteristic (Z) of the impedance (Z2) of only the secondary side has the fundamental wave frequency (f0) between the frequency (fMAX) taking the maximum (ZMAX) and the frequency (fMIN) taking the minimum (ZMIN).

The phase characteristic (Φ) of the impedance (Z2) has two points (point P1and point P2shown inFIG. 21) at which the phase inclinations are diverged and has a portion (between the two points P1, P2) which is parallel to the frequency axis, where the fundamental wave frequency component (f0) is present between the two points P1and P2. In other words, the above phase characteristic (Φ) makes a turn around the area in the vicinity of the fundamental wave frequency (f0) and comes back.

When the contactless electricity-supplying portion10is provided with the incoming circuit portion413having the impedance characteristic (Z) or phase characteristic (Φ) shown inFIG. 21, the phase characteristic of the input impedance (Zin) has, referring toFIG. 5, such a characteristic as shown according to the first embodiment. With this, in the contactless electricity-supplying device20according to the seventh embodiment, the fluctuation band of the phase of the input impedance (Zin) relative to the fundamental wave frequency (f0) is small even when the coupling coefficient k fluctuates, thus suppressing the fluctuation of the phase (Φ0), as a result, keeping the power factor high and making it possible to efficiently supply the electric power.

In addition, the capacitor403according to the seventh embodiment corresponds to “sixth capacitor” of the present invention.

Eighth Embodiment

FIG. 22is a circuit portion showing the contactless electricity-supplying device20according to the eighth embodiment of the present invention. Compared with the first embodiment described above, the eighth embodiment differs in using a circuit that is different from the circuit of the incoming circuit portion200inFIG. 1. Other than the above in terms of structure, the eighth embodiment is substantially the same as the first embodiment, and therefore descriptions of the first embodiment will be properly quoted according to the eighth embodiment.

As shown inFIG. 22, an incoming circuit portion414according to the eighth embodiment has such a structure that the capacitor202is connected in parallel to the serial circuit composed of the secondary winding201and capacitor401whereas the capacitor203is connected to a connection between the capacitor202and the capacitor401.

FIG. 23shows the impedance characteristic (Z) and phase characteristic (Φ) relative to the frequency of the impedance (Z2) of only the secondary side when viewed from the load portion7side, with the mutual inductance M as zero.

As shown inFIG. 23, the impedance characteristic (Z) of the impedance (Z2) of only the secondary side has the fundamental wave frequency (f0) between the frequency (fMAX) taking the maximum (ZMAX) and the frequency (fMIN) taking the minimum (ZMIN). Herein, especially, in the case of the phase characteristic having a plurality of maximums (ZMAX), the maximum (ZMAX) denotes a value corresponding to the frequency that is nearest to the fundamental wave frequency (f0). Also, the same is true of the minimum (ZMIN).

The phase characteristic (Φ) of the impedance (Z2) of only the secondary side has two points (point P1and point P2shown inFIG. 23) at which the phase inclinations are diverged, and has a portion (between the two points P1, P2) which is parallel to the frequency axis, where the fundamental wave frequency component (f0) is present between the two points P1and P2. In other words, the above phase characteristic (Φ) makes a turn around the area in the vicinity of the fundamental wave frequency (f0) and comes back.

When the contactless electricity-supplying portion10is provided with the incoming circuit portion414having the impedance characteristic (Z) or phase characteristic (Φ) shown inFIG. 23, the phase characteristic of the input impedance (Zin) has, referring toFIG. 5, such a characteristic as shown according to the first embodiment. With this, in the contactless electricity-supplying device20according to the eighth embodiment, the fluctuation band of the phase of the input impedance (Zin) relative to the fundamental wave frequency (f0) is small even when the coupling coefficient k fluctuates, thus suppressing the fluctuation of the phase (Φ0), as a result, keeping the power factor high and making it possible to efficiently supply the electric power.

In addition, the capacitor202according to the eighth embodiment corresponds to “third capacitor” of the present invention, the capacitor401corresponds to “fifth capacitor” of the present invention, and the capacitor203corresponds to “fourth capacitor” of the present invention.

Ninth Embodiment

FIG. 24is a circuit portion showing the contactless electricity-supplying device20according to the ninth embodiment of the present invention. Compared with the first embodiment described above, the ninth embodiment differs in using a circuit that is different from the circuit of the incoming circuit portion200inFIG. 1. Other than the above in terms of structure, the ninth embodiment is substantially the same as the first embodiment, and therefore descriptions of the first embodiment will be properly quoted according to the ninth embodiment.

As shown inFIG. 24, an incoming circuit portion415according to the ninth embodiment has such a structure that the inductor402is connected in parallel to the serial circuit composed of the secondary winding201and capacitor401whereas the capacitor203is connected to a connection between the inductor402and the capacitor401.

FIG. 25shows the impedance characteristic (Z) and phase characteristic (Φ) relative to the frequency of the impedance (Z2) of only the secondary side when viewed from the load portion7side, with the mutual inductance M as zero.

As shown inFIG. 25, the impedance characteristic (Z) of the impedance (Z2) of only the secondary side has the fundamental wave frequency (f0) between the frequency (fMAX) taking the maximum (ZMAX) and the frequency (fMIN1) taking the minimum (ZMIN1). Moreover, the impedance characteristic (Z) of the impedance (Z2) of only the secondary side has the minimum (ZMIN2) other than the minimum (ZMIN1). Herein, the minimum corresponding to the frequency (fMIN1) that is nearest to the fundamental wave frequency (f0) is defined as ZMIN1.

The phase characteristic (Φ) of the impedance (Z2) of only the secondary side has two points (point P1and point P2shown inFIG. 25) at which the phase inclinations are diverged and which sandwich therebetween the fundamental wave frequency component (f0), and has a portion (between the two points P1, P2) which is parallel to the frequency axis. In addition, other than the two points P1, P2at which the phase inclinations are diverged, the phase characteristic (Φ) of the impedance (Z2) has a point (point P3shown inFIG. 25) at which the phase inclination is further diverged. In other words, the above phase characteristic (Φ) makes a turn around the area in the vicinity of the fundamental wave frequency (f0) and comes back.

When the contactless electricity-supplying portion10is provided with the incoming circuit portion415having the impedance characteristic (Z) or phase characteristic (Φ) shown inFIG. 25, the phase characteristic of the input impedance (Zin) has, referring toFIG. 5, such a characteristic as shown according to the first embodiment. With this, in the contactless electricity-supplying device20according to the ninth embodiment, the fluctuation band of the phase of the input impedance (Zin) relative to the fundamental wave frequency (f0) is small even when the coupling coefficient k fluctuates, thus suppressing the fluctuation of the phase (Φ0), as a result, keeping the power factor high and making it possible to efficiently supply the electric power.

In addition, the capacitor401according to the ninth embodiment corresponds to “fifth capacitor” of the present invention and the capacitor203corresponds to “fourth capacitor” of the present invention, and the inductor402corresponds to “third inductor” of the present invention.

In addition, the power transmission circuit portions100,311,312,313and the incoming circuit portions200,411,413,414,415shown according to the first to ninth embodiments may be arbitrarily combined to form the contactless electricity-supplying portion10.

Tenth Embodiment

FIG. 26is a circuit portion showing the contactless electricity-supplying device20according to the tenth embodiment of the present invention. Compared with the first embodiment described above, the tenth embodiment differs in specifying scale of the inductance of each of the primary winding101and the secondary winding201and scale of capacitance of each of the capacitors102,202and203. Other than the above in terms of structure, the tenth embodiment is substantially the same as the first embodiment, and therefore descriptions of the first embodiment will be properly quoted according to the tenth embodiment.

As shown inFIG. 26, on the primary side, there are disposed the primary winding101and the capacitor102which is connected in parallel to the primary winding101. On the secondary side, there are provided the secondary winding201, the capacitor202connected in parallel to the secondary winding201, and the capacitor203connected in series to the parallel circuit composed of the secondary winding201and capacitor202. The above circuit corresponds to the contactless electricity-supplying portion10shown inFIG. 1. Herein, the inductance of the primary winding101is defined as L1, the inductance of the secondary winding201is defined as L2, the capacitance of the capacitor102is defined as C1p, the capacitance of the capacitor202is defined as C2pand the capacitance of the capacitor203is defined as C2s.

The tenth embodiment specifies conditions associated with the scale of the inductance of each of the primary winding101and the secondary winding201and the scale of each of the capacitors102,202and203, sets the fundamental wave frequency (f0) in the vicinity of a resonant frequency (f1) of the impedance (Z1) on the primary side, and sets the fundamental wave frequency (f0) between the first resonant frequency (fa) and second resonant frequency (fb) of the impedance (Z2) on the secondary side.

At first, the capacitance C1pof the capacitor102is explained referring toFIG. 27.FIG. 27shows a circuit on the primary side (transmission side) among the circuits inFIG. 26.

As shown inFIG. 27, the mutual inductance M=0 between the primary winding101and the secondary winding201. Then, the circuit is so designed that the relation between the fundamental wave frequency (f0) supplied from the high-frequency AC power supply circuit6to the circuit on the primary side, the inductance (L1) and the capacitance (C1p) satisfies the following formula 1.
(Expression 1)
C1p=1/(L1(2πf0)2)  (Formula 1)

Then, the impedance characteristic (Z) and phase characteristic (Φ) of the circuit on the primary side shown inFIG. 27is shown inFIG. 28.FIG. 28is a graph showing the impedance characteristic (Z) and phase characteristic (Φ) of the circuit on the primary side relative to the frequency.

The resonant frequency (f1) of the impedance (Z1) corresponds to the frequency showing the maximum of the impedance characteristic (Z) and corresponds to the frequency of the center point of the rotating phase characteristic. Therefore,FIG. 28verifies that the fundamental wave frequency (f0) is positioned in the vicinity of the resonant frequency (f1). That is, designing the circuit such that the fundamental wave frequency (f0) is set in the vicinity of the resonant frequency (f1) satisfies the condition of the formula 1.

With this, the current supplied to the contactless electricity-supplying portion10from the high-frequency AC power supply circuit6can be suppressed low, thus making it possible to enhance the efficiency.

Then, the capacitance C2pof the capacitor202will be explained referring toFIG. 29.FIG. 29shows the parallel circuit composed of the secondary winding201and capacitor202, among the circuits on the secondary side (incoming side) of the circuits inFIG. 26.

As shown inFIG. 29, the mutual inductance M=0 between the primary winding101and the secondary winding201. Then, the circuit is so designed that the relation between the inductance (L1), the capacitance (C1p) and the inductance (L2) and capacitance (C2p) satisfies the following formula 2.
(Expression 2)
C2p<(L1/L2)C1p(Formula 2)

The formula 2 will be explained, while showing inFIG. 30the impedance characteristic (Z) and phase characteristic (Φ) of the circuit on the secondary side of the circuit inFIG. 26.FIG. 30is a graph showing the impedance characteristic (Z) and phase characteristic (Φ) of the circuit on the secondary side relative to the frequency.

As shown inFIG. 30, the second resonant frequency (fb) of the impedance (Z2) corresponds to the frequency (fMAX) showing the maximum (ZMAX) of the impedance characteristic (Z) and corresponds to the frequency of the center point of the rotating phase characteristic (Φ). Moreover, the second resonant frequency (fb) is formed by the resonant circuit (refer toFIG. 29) which is a combination of the inductance (L2) and the capacitance (C2p), where the above resonant circuit and the second resonant frequency (fb) in combination has the following relation (formula 3).

Then, designing the circuit such that the second resonant frequency (fb) is higher than the fundamental wave frequency (f0) establishes the formula 4.
(Expression 4)
f0<fb(Formula 4)

Substituting the formula 1 and formula 3 into the formula 4 leads to the formula 2. That is, designing the circuit such that the fundamental wave frequency (f0) is lower than the second resonant frequency (fb) for satisfying the expression 4 can satisfy the condition of the expression 2.

Then, the capacitance C2sof the capacitor203will be explained referring toFIG. 31.FIG. 31shows a circuit of the secondary side (incoming side) of the circuit inFIG. 26where the circuit inFIG. 31has the parallel circuit (composed of the secondary winding201and capacitor202) and the capacitor203which is connected in series to the parallel circuit.

As shown inFIG. 31, the mutual inductance M=0 between the primary winding101and the secondary winding201. Then, the circuit is so designed that the relation between the inductance (L1), the capacitance (C1p), the inductance (L2), the capacitance (C2p) and the capacitance (C2s) satisfies the following formula 5.
(Expression 5)
(C2s+C2p)>(L1/L2)C1p(Formula 5)

The formula 5 will be explained while showing inFIG. 30the impedance characteristic (Z) and phase characteristic (Φ) of the circuit on the secondary side of the circuit inFIG. 31.

As shown inFIG. 30, the first resonant frequency (fa) of the impedance (Z2) of only the secondary side corresponds to the frequency showing the minimum (ZMIN) of the impedance characteristic (Z) and corresponds to the frequency of the center point of the rotating phase characteristic (101). Moreover, the first resonant frequency (fa) is a resonant frequency of a resonant circuit which is formed by the inductance (L2), the capacitance (C2p) and the capacitance (C2s), where the above resonant circuit and the first resonant frequency (fa) have the following relation (formula 6).

Then, designing the circuit such that the first resonant frequency (fa) is lower than the fundamental wave frequency (f0) establishes the formula 7.
(Expression 7)
f0>fa(Formula 7)

Substituting the formula 1 and formula 6 into the formula 7 leads to the formula 5. That is, designing the circuit such that the fundamental wave frequency (f0) is higher than the first resonant frequency (fa) to satisfy the formula 7 can satisfy the condition of the formula 5.

Then, from the formula 2 and formula 5, the formula 8 can be led as a relation between the inductances L1, L2and the capacities C1p, C2p, C2sof the circuits on the primary and secondary sides.
(Expression 8)
C2p<(L1/L2)C1p<(C2s+C2p)  (Formula 8)

With this, the phase characteristic (Φ) of the impedance (Z2) has two points {corresponding to the first resonant frequency (fa) and second resonant frequency (fb) shown in FIG.30} at which the phase inclinations are diverged, and has a portion {between the two points (fa) and (fb)} which is parallel to the frequency axis, where the fundamental wave frequency component (f0) is present between the first resonant frequency (fa) and the second resonant frequency (fb). As a result, the efficiency of supplying the electric power from the primary side to the secondary side is improved.

Then, an explanation is made about the impedance characteristic (Zin) which is viewed from the output side of the high-frequency AC power supply circuit6in the circuit of the contactless electricity-supplying portion10shown inFIG. 26.

FIG. 32shows an equivalent circuit of the circuit inFIG. 26.

Then, based on the circuit shown inFIG. 32, the impedance characteristic (Zin) viewed from the output side of the high-frequency AC power supply circuit6is subjected to Laplace transformation, as shown in the formula 9.

FIG. 33shows pole tracks of the impedance characteristic (Zin).FIG. 33shows two typical characteristical roots causing a great influence on the circuit characteristic of the poles in formula 9, that is, a pole1that is nearest to an imaginary axis Im (Imaginary) side and a pole2that is second nearest to the imaginary axis Im side. When the coupling coefficient k between the primary winding101and the secondary winding201is increased from an area in the vicinity of 0, the pole1and pole2draw the tracks as shown inFIG. 33. That is, the pole1moves away from the imaginary axis Im in accordance with the increase of the coupling coefficient k, meanwhile the pole2approaches the pole1in accordance with the increase of the coupling coefficient k.

That is, it is assumed that, in accordance with the increase of the coupling coefficient k, the pole1moves away from the imaginary axis Im whereas the pole2approaches the pole1, thus the pole and pole2mutually negate the influence, as a result, suppressing decrease of efficiency. That is, the two poles (pole1and pole2) which are typical characteristical roots draw mutually opposite tracks in accordance with the change of the coupling coefficient k.

On the other hand, in the circuit shown inFIG. 26or the circuit shown inFIG. 32, the inductances L1, L2and capacitances C1p, C2p, C2sof the circuit are so set that the formula 1 and formula 8 are not satisfied, in this case, the impedance characteristic (Zin) viewed from the output side of the high-frequency AC power supply circuit6is denoted as inFIG. 34.FIG. 34shows the pole track of the impedance characteristic (Zin) in a complex plane under a circuit condition where the formula 1 and formula 8 are not satisfied.

As shown inFIG. 34, the pole1that is nearest to the imaginary axis Im side does not move away from the imaginary axis Im in accordance with the increase of the coupling coefficient k, whereas the pole2that is second nearest to the imaginary axis Im side does not approach the imaginary axis Im in accordance with the increase of the coupling coefficient k. Moreover, the pole1and pole2inFIG. 34, compared with those inFIG. 33, draw the tracks in positions away from each other in accordance with the increase of the coupling coefficient k (pole2not being a typical characteristical root), therefore do not cause an influence on each other. Thus, a control root moves away from the imaginary axis Im in accordance with the increase of the coupling coefficient k, thus decreasing the efficiency.

That is, when the pole1which moves away from the imaginary axis Im in accordance with the increase of the coupling coefficient k is present (FIG. 33), the pole1which moves away from the imaginary axis Im and the pole2which approaches the imaginary axis Im {two typical characteristical roots (pole1and pole2) draw mutually opposite tracks in accordance with the change of the coupling coefficient k} are present in the circuit according to the tenth embodiment, and then the control root switches positions from the pole1to the pole2in accordance with the increase of the coupling coefficient k. Thus, according to the tenth embodiment, when the coupling coefficient k is increased, the control root is present in the vicinity of the imaginary axis Im. As a result, the efficiency change in accordance with the change of the coupling coefficient k can be suppressed.

Then, an explain is made about the impedance characteristic (Zin) and phase characteristic (Φin) which are viewed from the output side of the high-frequency AC power supply circuit6.FIG. 35ashows the impedance characteristic (Zin) andFIG. 35bshows the phase characteristic (Φin), in the contactless electricity-supplying portion10according to the tenth embodiment. Moreover,FIG. 35aandFIG. 35bshow respective changes of the impedance characteristic (Zin) and the phase characteristic (Φin) in accordance with the change of the coupling coefficient k.

As shown inFIG. 35b, the tenth embodiment has such a characteristic that the phase rotates around the area in the vicinity of the fundamental wave frequency (f0) in accordance with the increase of the coupling coefficient k. Therefore, even when the coupling coefficient k changes, the phase corresponding to the fundamental wave frequency (f0) takes a value close to 0 degree, thus making it possible to suppress decrease of power factor.

As shown above (especially,FIG. 26), according to the tenth embodiment, the contactless electricity-supplying portion10has such a structure that the primary winding101and the capacitor102are connected in parallel on the primary side whereas the parallel circuit (composed of the secondary winding201and the capacitor202) and the capacitor203connected in series to the parallel circuit are connected on the secondary side. In the contactless electricity-supplying portion10, the fundamental wave frequency (f0) of the alternating power supplied from the high-frequency AC power supply circuit6to the contactless electricity-supplying portion10is set in the vicinity of the resonant frequency (f1) of the impedance (Z1) of only the primary side and is set between the first resonant frequency (fa) and second resonant frequency (fb) of the impedance (Z2) of only the secondary side. With this, in accordance with the change of the coupling coefficient k and in the vicinity of the fundamental wave frequency (f0), the fluctuation of the phase is suppressed, thereby making it possible to suppress decrease of efficiency.

Moreover, according to the tenth embodiment, the circuit is so designed as to satisfy the conditions of the formula 1 and formula 8 in the above circuit. With this, in accordance with the change of the coupling coefficient k and in the vicinity of the fundamental wave frequency (f0), the fluctuation of the phase is suppressed, thereby making it possible to suppress decrease of efficiency.

Moreover, according to the tenth embodiment, when the input impedance characteristic (Zin) is shown by the complex plane in the above circuit, in accordance with the increase of the coupling coefficient k, the pole1that is nearest to the imaginary axis Im moves away from the imaginary axis Im and the pole2that is second nearest to the imaginary axis Im approaches the pole1(FIG. 33). With this, in accordance with the change of the coupling coefficient k and in the vicinity of the fundamental wave frequency (f0), the fluctuation of the phase is suppressed, thereby making it possible to suppress decrease of efficiency.

In addition, according to the tenth embodiment, the pole1corresponds to “first pole” of the present invention and the pole2corresponds to “second pole” of the present invention.

Eleventh Embodiment

FIG. 36shows an electric circuit diagram showing the contactless electricity-supplying device20according to the eleventh embodiment of the present invention. Compared with the first embodiment described above, the eleventh embodiment differs in an output voltage waveform from the high-frequency AC power supply circuit6to the contactless electricity-supplying portion10. Other than the above in terms of structure, the eleventh embodiment is substantially the same as the first embodiment, and therefore descriptions of the first embodiment will be properly quoted according to the eleventh embodiment.

As shown inFIG. 36, the contactless electricity-supplying device20according to the eleventh embodiment is provided with a controlling portion8for controlling switching of the transistors (switching elements)63ato63d. The controlling portion8includes a frequency controlling portion81, a voltage command setting portion82and a voltage command calculating portion83.

Then, an explanation is made about detailed structure of the controlling portion8, referring toFIG. 37.FIG. 37shows a block diagram of the controlling portion8. The frequency controlling portion81has a frequency command setting portion81aand a carrier setting portion81b. The frequency command setting portion81a sets a frequency command value (fref) of the output voltage of the voltage-type inverter63and transmits the frequency command value (fref) to the carrier setting portion81b. The carrier setting portion81bforms an amplitude (Vx) of carrier based on the frequency command value (fref), to thereby form a carrier signal of triangular wave. The carrier setting portion81buses a digital control using for example a microcomputer and forms the amplitude (Vx) from a clock counter which is based on the frequency command value (fref).

The voltage command setting portion82has a voltage amplitude command setting portion82aand a switching pulse setting portion (SW pulse setting portion)82b. The voltage amplitude command setting portion82asets an amplitude command value (Vref) of the output voltage of the voltage-type inverter63, and transmits the amplitude command value (Vref) to the switching pulse setting portion82b. Based on a power command value (Pref) given from an external portion, the voltage amplitude command setting portion82adetermines the amplitude command value (Vref). Comparing the carrier transmitted from the carrier setting portion81bwith the amplitude command value (Vref), the switching pulse setting portion82bsets the switching pulse (SW1) for switching the transistors (switching elements)63ato63d.

Herein, conventionally, the switching pulse (SW1) is inputted to the voltage-type inverter63, and the voltage-type inverter63outputs a supply voltage, for example, sine wave, to the contactless electricity-supplying portion10. According to the eleventh embodiment, setting the voltage command calculating portion83at the controlling portion8outputs, to the contactless electricity-supplying portion10, a supply voltage that is different from the conventional sine wave supply voltage.

Based on the switching pulse (SW1) transmitted from the switching pulse setting portion82b, the voltage command calculating portion83sets a new switching pulse (SW2). When the new switching pulse (SW2) controls the transistors (switching elements)63ato63d, the supply voltage (Vin) supplied from the high-frequency AC power supply circuit6to the contactless electricity-supplying portion10makes a waveform which has, per period, a first rest period (tb1) between a plurality of positive voltage output periods and a second rest period (tb2) between a plurality of negative voltage output periods, as shown inFIG. 38. In each of the first and second rest periods (tb1, tb2), the voltage output has a rest or the voltage is not outputted.FIG. 38shows output characteristic of supply voltage (Vin) relative to time. In addition, according to the eleventh embodiment, the explanation will be made on the premise that the first rest period (tb1) and the second rest period (tb2) have the same length, however, the above two periods (tb1, tb2) failing to have the same length are also allowed.

Hereinafter, referring toFIG. 39, the control operations of the controlling portion8will be set forth.FIG. 39shows the carrier waveform, output waveforms, switching pulse (SW1) waveform, switching pulse (SW2) waveform and supply voltage waveform. InFIG. 39, the abscissa denotes time axis.

At first, the carrier setting portion81btransmits the carrier signal of the amplitude (Vx) to the switching pulse setting portion82b, as shown inFIG. 39(a). The voltage amplitude command setting portion82a, as shown inFIG. 39(a), sets the amplitude command value (Vref) to the amplitude (Vx).

Then, based on the following conditions, the switching pulse setting portion82bforms the switching pulse (SW1) shown inFIG. 39(b).
Switch S1when carrier≦Vx/2
OFF when carrier>Vx/2
Switch S2OFF when carrier≦Vx/2
ON when carrier>Vx/2
Switch S3ON when carrier≧Vrefand carrier≦Vref+Vx/2
OFF when carrier<Vrefor carrier>Vref+Vx/2
Switch S4OFF when carrier≧Vrefand carrier≦Vref+Vx/2
ON when carrier<Vrefor carrier>Vref+Vx/2  (Expression 10)

By the above conditions, the switching pulses (SW1) shown inFIG. 39(b) are formed.

Then, based on the switching pulse (SW1), the voltage command calculating portion83forms the switching pulse (SW2) shown inFIG. 39(c). At first, the voltage command calculating portion83divides the period (T) equivalent to one period of the switching pulse of the switch S3into four sections (Ton1, Ton2, Toff1, Toff2), thus establishing the relation of the formula 10.
(Expression 11)
T=Ton+Toff=Ton1+Ton2+Toff1+Toff2(Formula 10)

Then, with Tonas an ON-period of the transistor63c, Toffas an OFF-period of the transistor63c, and D as a duty ratio, the following formula 11 is established.
(Expression 12)
Ton=Ton1+Ton2, Toff=Toff1+Toff2, D=Ton/T(Formula 11)

The duty ratio D is determined by the amplitude command value (Vref) set by the voltage amplitude command setting portion82aand the period T is determined by the frequency command value (fref) set by the frequency command setting portion81a.

Herein, the rest period tbis for stopping supply voltage to the contactless electricity-supplying portion10by making ON-OFF control of the switching of the transistors (switching elements)63ato63dand is set by the controlling portion8based on the period (T) and duty ratio (D). The period (T), the duty ratio (D) and the rest period (tb) have a predetermined relation and when the period (T) or duty ratio (D) changes, the rest period (tb) also changes. The controlling portion8stores in advance the relation between the period (T), duty ratio (D) and rest period (tb) by, for example, a table and the like.

The section of the rest period tbis different (opposite) from neighboring sections in ON/OFF characteristic. For example, in a section of a certain rest period tb, when the switching pulse (SW2) is ON, the switching pulses (SW2) in the sections neighboring the section of the rest period tb, in other words, the former and latter sections are OFF.

Then, the sections a to h are arranged in order, to thereby form the switching pulse (SW2) as shown inFIG. 39(c). Since the section a and section c are OFF-periods, the section b is an ON-period and since the section e and section g are ON-periods, the section f is an OFF-period.

Moreover, the switching pulse (SW2) of the switch S4is formed in a manner same as that of the switch S3. However, the switching pulse (SW2) of the switch S4is opposite to the switching pulse (SW2) of the switch S3, causing a reversed waveform (symmetrical waveform). The switching pulse (SW2) of each of the switch S1and the switch S2is like the waveform of the switching pulse (SW1).

With this, as shown inFIG. 39(c), the voltage command calculating portion83forms the switching pulse (SW2) based on the switching pulse (SW1). Then, the switching pulse (SW2) operates each of the transistors (switching elements)63ato63dand the power is supplied from the 3-phase AC power supply64. Then, the high-frequency AC power supply circuit6supplies to the contactless electricity-supplying portion10the voltage shown inFIG. 39(d). That is, in the high-frequency AC power supply circuit6according to the eleventh embodiment, the supply voltage which includes, per period (T), a plurality of periods (equivalent to sections a and c) for outputting the positive voltage, a period (equivalent to section b) for stopping the voltage output and disposed between the plurality of periods (sections a and c), a plurality of periods (equivalent to sections e and g) for outputting the negative voltage, and a period (equivalent to section f) for stopping the voltage output and disposed between the plurality of periods (sections e and g) is supplied to the circuit on the primary side of the contactless electricity-supplying portion10.

Then, controlling procedures of the controlling portion8will be explained referring toFIG. 40.FIG. 40is a flowchart showing the controlling procedures of the controlling portion8.

At step1, based on the power command value (Pref), the controlling portion8determines whether or not the duty ratio (D) or the period (T) is changed. When changed (Yes inFIG. 40), the routine proceeds to step2and when not changed (No inFIG. 40), the routine proceeds to step8. In addition, the duty ratio (D) and the period (T) each has an initial value which is set in advance. When implementing the flow inFIG. 40at the first setout, the routine proceeds to step2and in the second flow and thereafter, the routine compares the current duty ratio (D) and period (T) with their initial values or former values, to thereby determine the change.

At step2, the switching pulse setting portion82bsets the switching pulse (SW1).

At step3, based on the formula 10, the voltage command calculating portion83divides the switching pulse (SW1) of the switch S3into four sections (Ton1, Ton2, Toff1, Toff2).

At step4, based on the duty ratio (D) and period (T), the controlling portion8sets the rest period (tb).

At step5, the voltage command calculating portion83divides the switching pulse (SW1) of the switch S3into eight sections (a to h).

At step6, based on the eight sections (a to h) divided at step5, the voltage command calculating portion83sets the switching pulse (SW2) of the switch S3. Moreover, the voltage command calculating portion83reverses the switching pulse (SW2) of the switch S3, to thereby set the switching pulse (SW2) of the switch S4.

Then, at step7, the voltage command calculating portion83sets the switching pulse (SW2) of the switches S1to S4.

Then, at step8, the voltage command calculating portion83outputs the switching pulses (SW2) to the respective transistors (switching elements)63ato63d.

Then, in the circuit of the contactless electricity-supplying portion10shown inFIG. 36, the first situation where the pulse having no rest period (tb) unlike the eleventh embodiment is defined as the supply voltage (hereinafter referred to as example 1) is compared with the second situation that has the supply voltage according to the eleventh embodiment (hereinafter referred to as example 2), while explaining about the EMI (Electro-Magnetic-Interference) level and efficiency.FIG. 41shows the characteristics of the supply voltage and current relative to time according to the example 1, andFIG. 42shows the characteristics of the supply voltage and current relative to time according to the example 2.

Specifically, according to the example 1, as shown inFIG. 41, when an ordinary pulse voltage (Vs) is supplied to the contactless electricity-supplying portion10, the output current (Is) flows from the high-frequency AC power supply circuit6to the contactless electricity-supplying portion10. On the other hand, according to the example 2, when the pulse voltage (Vt) having the rest period (tb) is supplied to the contactless electricity-supplying portion10, the output current (It) flows from the high-frequency AC power supply circuit6to the contactless electricity-supplying portion10. However, according to the example 2, setting the rest period (tb) allows the example 2 to be unchanged from the example 1 in terms of total energy. Sum of integral values (V1+V2) of the supply voltage of the example 2 is made equal to the integral value (V) of the supply voltage of the example 1.

Unlike the example 1, setting the rest period (tb) according to the example 2 allows the raised output current (It) to be lowered once or allows the lowered output current (It) to be raised once. The output current (It) is lowered once or raised once in the rest period (tb), to thereby suppress the peak of the current. Moreover, the inclination (dIt/dt) of the output current (It) is made small.

Then, the EMI (Electro-Magnetic-interference) level will be explained referring toFIG. 43andFIG. 44.FIG. 43andFIG. 44respectively show that the output current (Is) according to the example 1 and the output current (It) according to the example 2 are subjected to FET (Fast Fourier Transformation) analyses, showing characteristics of the EMI level relative to the frequency. The left end of the abscissa corresponds to the fundamental wave frequency component of the fundamental wave frequency (f0).

Herein, the EMI level will be explained. When the output current (Is) of the example 1 or the output current (It) of the example 2 flows in the wiring which connects the high-frequency AC power supply circuit6with the contactless electricity-supplying portion10, the wiring acts like an antenna, thereby causing a possibility that a noise leaks out of the wiring. Then, the noise which corresponds to the EMI occurs at the frequency which is a multiple (in the order of integer) of the fundamental wave frequency component (in other words, having higher frequency component than the fundamental wave frequency component). Then EMI level, that is, the scale of noise depends on the scale of the inclination (dI/dt) of the output current.

When comparing the peak (corresponding to a portion A inFIG. 43) of the EMI level of the example 1 with the peak (corresponding to a portion B inFIG. 44) of the EMI level of the example 2, it is confirmed that the peak of the EMI level according to the example 2 is suppressed. That is, compared with the example 1, the example 2 having the rest period (tb) can make the inclination (dIt/dt) of the output current (It) small, thereby suppressing the EMI level. Then, suppressing the EMI level as shown in the example 2 can prevent noise leak out of the wiring and the example 2 makes the output current (It) small, thereby also suppressing a steady loss.

As stated above, according to the example 2 (eleventh embodiment), the supply voltage including a plurality of positive voltage output periods (sections a and c), the rest period (tb) (section b) between the plurality of positive voltage output periods (sections a and c), a plurality of negative voltage output periods (sections e and g), and the rest period (tb) (section f) disposed between the plurality of negative voltage output periods (sections e and g) is supplied to at least the primary winding101. This makes it possible to suppress the EMI level and enhance the efficiency.

Then, the inverter loss and the efficiency of the example 1 and example 2 will be set forth referring toFIG. 45andFIG. 46.FIG. 45is a graph showing inverter losses of the respective example 1 and example 2, andFIG. 46shows characteristics of the efficiency relative to the coupling coefficient k, according to the example 1 and example 2.

As shown inFIG. 45, the example 2 compared with the example 1 can suppress the steady loss, as stated above. On the other hand, as set forth inFIG. 39(c) andFIG. 39(d), setting the rest period (tb) according to the example 2 increases the number of switching operations of the transistor63cand transistor63dper period (T), thus increasing the switching loss compared with the example 1. Thus, comparing the example 1 with the example 2 concludes that there is not so great a difference in total loss. However, as set forth above, the EMI level contributing to the steady loss causes an influence due to the noise leak out of the circuit. Thus, when the EMI level is high, as the case may be, a noise countermeasure is provided otherwise, as a result, increasing cost and providing an extra circuit space. Therefore, when the inverter loss is equivalent, the example 2 is more preferable in that the steady loss can be more suppressed.

The output current It (refer toFIG. 42) of the example 2 is made smaller than the output current Is (refer toFIG. 41) of the example 1 (Is>It), however, as shown inFIG. 46, the efficiency is not decreased according to the example 2. Thus, compared with the example 1, the example 2 keeps the efficiency while making it possible to suppress the steady loss.

Then, an explanation is made about the number of rest periods (tb) provided per period (T). In the circuit of the contactless electricity-supplying portion10shown inFIG. 36, unlike the example 2 which sets one rest period (tb) per half period (T/2), the third situation (hereinafter referred to as example 3) has the rest period (tb) set twice per half period (T/2). The example 3 is compared with the example 2.

As shown inFIG. 47, according to the example 3, setting the two rest periods (tb) per half period (T/2) decreases the peak of the output current (Iu).FIG. 47shows characteristics of the supply voltage and current relative to time.

Referring toFIG. 48, the inverter loss of each of the example 2 and the example 3 will be set forth.FIG. 48is a graph showing the inverter losses of the respective example 2 and example 3. As shown inFIG. 48, the example 3 (refer toFIG. 47) has the output current (I) smaller than that of the example 2 (refer toFIG. 42) (In<It), thereby, the steady loss according to the example 3 decreases, however, such decrease is small. Moreover, according to the example 3, increase of the number of ON-OFF operations of the transistor63cand transistor63dincreases the switching loss. Then, the increase of the switching loss is larger than the decrease of the steady loss, thereby the example 3 is larger than the example 2 in terms of total inverter loss. That is, in the case of the two or more rest periods (tb) per half period (T/2) (example 3), the effect of suppressing the peak of the output current (I) is smaller compared with one rest period (example 2), meanwhile, the number of switching operations is increased, thus increasing the inverter loss in total.

In addition, in the case of three or more rest periods (tb) per half period (T/2), the effect of suppressing the EMI level is small like the case of two rest periods (tb), thus further enlarging the switching loss, resulting in increased inverter loss.

As set forth above, the example 2 (eleventh embodiment) sets, per period (T), only one rest period (tb) in each of the positive voltage output period and the negative voltage output period. Thus, the EMI level is decreased and thereby the EMI countermeasure is relieved, while making it possible to improve the efficiency.

Then, the rest period (tb) will be set forth referring toFIG. 49andFIG. 50.FIG. 49shows characteristics of the maximum of the EMI (Electro-Magnetic-Interference) level relative to the duty ratio (D), period (T) and rest period (tb), andFIG. 50shows the efficiency relative to the duty ratio (D), period (T) and rest period (tb).

According to the eleventh embodiment, the duty ratio (D) is fixed to a fixed value (D1), the period T is set to a period (T1), a period (T2) and a period (T3), and the rest period (tb) is changed, to thereby provide the maximum of the EMI level (refer toFIG. 49). Moreover, the period (T) is set to a fixed value (T1), the duty ratio (D) is set to a duty ratio (D2) and a duty ratio (D3), and the rest period (tb) is changed, to thereby provide the maximum of the EMI level. Relative to the rest period (tb), the duty ratio (D) and period (T) are fixed, and as shown inFIG. 49, (tb·D)/T is set as an abscissa and the maximum of the EMI level is set as an ordinate. InFIG. 49, the period (T1) and duty ratio (D1) are denoted by graph I, the period (T2) and duty ratio (D1) are denoted by graph II, the period (T3) and duty ratio (D1) are denoted by graph III, the period (T1) and duty ratio (D2) are denoted by graph IV, and the period (T1) and duty ratio (D3) are denoted by graph V.

Moreover, as shown inFIG. 50, likeFIG. 49, the duty ratio (D) is set to a fixed value (D1) and the period (T) is set to a fixed value (T1), to thereby provide the efficiency relative to (tb·D)/T from the graph Ito graph V. In addition, the characteristics shown inFIG. 49andFIG. 50are denoted by approximate curves of the rest period tbby taking the rest period tbwith a plurality of discrete values.

As shown inFIG. 49andFIG. 50, in the case where the duty ratio (D) is fixed to the fixed value (D1) and the period (T) is fluctuated (corresponding to graph I to graph III) and in the case where the period (T) is fixed to the fixed value (T1) and the duty ratio (D) is fluctuated (corresponding to graph IV to graph V), the maximum of EMI level is lowest when the condition of the formula 12 is satisfied, thus causing the highest efficiency.
[Expression 14]
tb=0.015·T/D(Formula 12)

As shown above, according to the eleventh embodiment (example 2), the period (T), the rest period (tb) and the duty ratio (D) satisfy the condition of the formula 12. This makes it possible to decrease the maximum of the EMI level while improving the efficiency.

It is not necessary that the relation of formula 12 is completely equal, that is, the rest period (tb) being close to 0.015·T/D is acceptable.

Moreover, according to the eleventh embodiment, the switching pulse (SW1) of the switch S3is divided to thereby set the switching pulses (SW2) of the switch S1to switch S4. However, the switching pulse (SW1) of any of the switch S1, switch S2and switch S4may be divided.

Moreover, according to the eleventh embodiment, the first rest period (tb1) in the output period of the positive voltage is equal in length to the second rest period (tb2) in the output period of the negative voltage, however, such lengths failing to be the same are allowed.

Moreover, according to the eleventh embodiment, the explanation has been made about the case where only one rest period (tb) is set per half period (T/2) (example 2), however, this does not eliminate the case where two or more rest periods (tb) per half period (T/2) are set (example 3).

Moreover, according to the eleventh embodiment, the explanation has been made that it is preferable that the relation between the period (T), rest period (tb) and duty ratio (D) have the condition of the formula 12. However, failing to meet the formula 12 is allowed.

In addition, according to the eleventh embodiment, the section a and section c correspond to “positive voltage output period” of the present invention, the section b corresponds to “first rest period” of the present invention, the section e and section g correspond to “negative voltage output period” of the present invention and the section f corresponds to “second rest period” of the present invention.

The entire contents of the Japanese Patent Application No. 2009-117527 (filed May 14, 2009) and Japanese Patent Application No. 2010-101755 (filed Apr. 27, 2010) are incorporated herein by reference, in order to protect the above applications from erroneous translations or omitted portions.

As set forth above, the present invention has been described according to the first to eleventh embodiments, however, the present invention is not limited to the above descriptions and various changes or improvements thereof will obviously occur to those skilled in the art.

INDUSTRIAL APPLICABILITY

According to the present invention, the phase characteristic of the impedance (relative to the frequency) viewed from the output side of an AC power supply so changes as to rotate around an area in the vicinity of a fundamental wave frequency in accordance with the fluctuation of a coupling coefficient. Therefore, when the impedance is set in accordance with the coupling coefficient, the fluctuation band of the phase of the impedance becomes small, as a result, making it possible to suppress decrease of efficiency.