Controller for a switch and method of operating the same

A controller for a switch and a method of operating the same. In one embodiment, the controller is configured to measure a voltage of a control terminal of the switch and select a first mode of operation if the voltage of the control terminal is greater than a threshold voltage, and a second mode of operation if the voltage of the control terminal is less than the threshold voltage.

TECHNICAL FIELD

The present invention is directed, in general, to power electronics and, more specifically, to a controller for a switch and method of operating the same.

BACKGROUND

A switched-mode power converter (also referred to as a “power converter”) is a power supply or power processing circuit that converts an input voltage waveform into a specified output voltage waveform. DC-DC power converters convert a direct current (“DC”) input voltage into a DC output voltage. Controllers associated with the power converters manage an operation thereof by controlling conduction periods of power switches employed therein. Generally, the controllers are coupled between an input and output of the power converter in a feedback loop configuration (also referred to as a “control loop” or “closed control loop”).

Typically, the controller measures an output characteristic (e.g., an output voltage, an output current, or a combination of an output voltage and an output current) of the power converter, and based thereon modifies a duty cycle of a power switch of the power converter. The duty cycle “D” is a ratio represented by a conduction period of a power switch to a switching period thereof. In other words, the switching period includes the conduction period of the power switch (represented by the duty cycle “D”) and a non-conduction period of the power switch (represented by the complementary duty cycle (“1-D”). Thus, if a power switch conducts for half of the switching period, the duty cycle for the power switch would be 0.5 (or 50 percent).

The switched-mode power converters can be constructed with different types of power switches such as bipolar transistors, metal-oxide semiconductor field-effect transistors (“MOSFETs”) or insulated gate bipolar transistors (“IGBTs”). At low power levels, for example, an output power less than 100 watts (“W”), the MOSFETs and bipolar transistors are most commonly used for power switches. While MOSFETs can work at higher switching frequency, which enables smaller designs, bipolar transistors are available at lower cost. Additionally, the different switches employ different drivers for their respective control terminals. As a result, separate driver integrated circuits are inventoried to accommodate the use of different switches in a design of a circuit (e.g., a power converter) employing the same.

Accordingly, what is needed in the art is a circuit and related method for a switch that enables a driver to be used for different types of switches such as MOSFETs and bipolar transistors that can be adapted to high-volume manufacturing techniques for a power converter or the like employing the same.

SUMMARY OF THE INVENTION

These and other problems are generally solved or circumvented, and technical advantages are generally achieved, by advantageous embodiments of the present invention, including a controller for a switch and a method of operating the same. In one embodiment, the controller is configured to measure a voltage of a control terminal of the switch and select a first mode of operation if the voltage of the control terminal is greater than a threshold voltage, and a second mode of operation if the voltage of the control terminal is less than the threshold voltage.

Corresponding numerals and symbols in the different figures generally refer to corresponding parts unless otherwise indicated, and may not be redescribed in the interest of brevity after the first instance. The FIGUREs are drawn to illustrate the relevant aspects of exemplary embodiments.

DETAILED DESCRIPTION OF ILLUSTRATIVE EMBODIMENTS

The present invention will be described with respect to exemplary embodiments in a specific context, namely, a controller operable with different types of switches such as a MOSFET or bipolar transistor. While the principles of the present invention will be described in the environment of a power converter, any application that may benefit from the controller as described herein including a power amplifier or a motor controller is well within the broad scope of the present invention.

Turning now toFIG. 1, illustrated is a schematic diagram of an embodiment of a power converter constructed according to the principles of the present invention. The power converter is configured to convert AC mains voltage (designated “Vac in”) to a regulated DC output voltage Vout. A power train (e.g., a flyback power train) of the power converter (also referred to as a “flyback power converter”) includes a power switch Q1coupled to a source of electrical power (e.g., the AC mains) via an input filter (including capacitors C1, C2and an inductor L2) to provide a filtered DC input voltage Vin to a magnetic device (e.g., an isolating transformer or transformer TX1). A resistor R1represents an impedance of the AC mains. Although not illustrated, the power converter may also include an electromagnetic interference filter between the AC mains voltage Vac and a bridge rectifier110. The transformer TX1has a primary winding P1and a secondary winding S1with a turns ratio that is selected to provide the output voltage Vout with consideration of a resulting duty cycle and stress on power train components.

The power switch Q1(e.g., a MOSFET) is controlled by a controller (e.g., an application specific integrated circuit (“ASIC”))120that controls the power switch Q1to be conducting for a duty cycle. The power switch Q1conducts in response to drive signal such as a gate drive voltage dry produced by the controller120with a switching frequency (often designated as “fs”). The duty cycle is controlled (e.g., adjusted) by the controller120to regulate an output characteristic of the power converter such as an output voltage Vout, an output current lout, or a combination thereof. A feedback signal FB traverses a feedback path (a portion of which is identified as130) emanating from a bias winding P2of the transformer TX1to enable the controller120to control the duty cycle to regulate the output characteristic of the power converter proportional to a bias voltage VP from the bias winding P2. A series circuit arrangement of resistors R14, R23provides a voltage divider function to scale the voltage produced for the feedback signal FB by the bias winding P2of the transformer TX1. The bias voltage VP is substantially proportional to a voltage across the secondary winding S1depending on a turns ratio between the primary winding P1and the secondary winding S1.

The voltage produced across the winding P2is rectified by a diode D6and charges a capacitor C4to provide an bias voltage VP for the controller120. A resistor R25provides a current-limit function to limit a charging current into the capacitor C4. A resistor R8provides a start-up charge for the capacitor C4. The AC voltage or alternating voltage appearing on the secondary winding S1of the transformer TX1is rectified by an auxiliary power switch (e.g., diode D7or, alternatively, by a synchronous rectifier, not shown), and the DC component of the resulting waveform is coupled to the output through the low-pass output filter including an output filter capacitor C9to produce the output voltage Vout. A resistor R18is included in the circuit to ensure that there is still power consumption when a load is disconnected from the output terminals out+, out− of the power converter. This ensures that the switching frequency at no load is high enough to react sufficiently to a change in the load. A current sensor R15is coupled to the power switch Q1and provides a voltage that is proportional to a current in the primary switch (Ip≅Ipri, wherein Ipri is a primary current flowing through the primary winding P1of the transformer TX1) for the controller120. This voltage is used to determine the duration of the conduction period of the power switch Q1.

During a first portion of the duty cycle, a primary current Ipri (e.g., an inductor current) flowing through the primary winding P1of the transformer TX1increases as current flows from the input through the power switch Q1. During a complementary portion of the duty cycle (generally co-existent with a complementary duty cycle 1-D of the power switch Q1), the power switch Q1is transitioned to a non-conducting state. Residual magnetic energy stored in the transformer TX1causes conduction of a secondary current Isec through the diode D7when the power switch Q1is off. The diode D7, which is coupled to the output filter capacitor C9, provides a path to maintain continuity of a magnetizing current of the transformer TX1. During the complementary portion of the duty cycle, the magnetizing current flowing through the secondary winding S1of the transformer TX1decreases. In general, the duty cycle of the power switch Q1may be controlled (e.g., adjusted) to maintain a regulation of or regulate the output voltage Vout of the power converter.

In order to regulate the output voltage Vout, a value or a scaled value of the feedback signal FB is compared with a reference voltage within the controller120to control the duty cycle D. A larger duty cycle implies that the power switch Q1is closed for a longer fraction of the switching period of the power converter. Thus, the power converter is operable with a switching cycle wherein an input voltage Vin is coupled to the transformer TX1for a fraction of a switching period by the power switch Q1controlled by controller120.

In a switch-mode power converter constructed with a flyback power train, a voltage produced by the bias winding P2during a flyback portion of a switching cycle can be related to the output voltage Vout by accounting for a turns ratio of the transformer TX1and voltage drops in diodes and other circuit elements. The voltage produced across the bias winding P2is employed to produce an estimate of the output voltage Vout, which in turn is used to regulate the same without crossing the isolation boundary of the transformer TX1.

Turning now toFIG. 2, illustrated is a schematic diagram of another embodiment of a power converter constructed according to the principles of the present invention. The power switch Q2ofFIG. 2is a bipolar transistor in lieu of the MOSFET power switch Q1illustrated inFIG. 1. The controller120ofFIGS. 1 and 2is configured to operate with different types of switches as set forth below. As a result, the controller120can select first and second modes of operation depending on the type of power switch employed in the power converter. For instance, the controller can select the first mode of operation if the power switch is a MOSFET (see, MOSFET power switch Q1inFIG. 1) and a second mode of operation if the power switch is a bipolar transistor (see, bipolar transistor power switch Q2inFIG. 2). It should be understood that the principles of the present invention are not limited to only MOSFETs and bipolar transistors. The power converters ofFIGS. 1 and 2otherwise include like components that operate in similar ways and, as such, will not hereinafter be described again.

Turning now toFIG. 3, illustrated is a schematic diagram of different switches demonstrating the principles of the present invention. The first switch is an npn bipolar transistor Q1with a base terminal Q1-base driven by a drive signal such as a positive drive voltage V1through a resistor R1. The second switch is an n-channel MOSFET Q2with a gate terminal Q2-G driven by the positive drive voltage V1through resistor R2. The resistors R1, R2are each one kilohm (“kΩ”) resistors. Since the bipolar transistor Q1presents a forward-biased junction at its base terminal Q1-base, the voltage of the base terminal does not rise more than about 0.7 volts (“V”). The gate terminal Q2-G of the MOSFET Q2presents a substantially open circuit to a driver, the voltage thereof rises substantially to the voltage of the drive voltage V1, which can be about 10 volts. Accordingly, the voltage at the respective control terminal of each switch can be employed to detect whether the switch is a bipolar transistor or a MOSFET.

Turning now toFIGS. 4 and 5, illustrated are graphical representations illustrating the differences between switches according to the principles of the present invention.FIG. 4illustrates a drive signal such as a drive voltage dry vs. time produced by a pulse-width modulator controller with a drive voltage of 10 volts, and the respective voltages VQ2-G, VQ1-base at the control terminals of a MOSFET and a bipolar transistor, respectively. As demonstrated, the voltage VQ2-G at the control terminal of the MOSFET rises to about 10 volts, and the voltage VQ1-base at the control terminal of the bipolar transistor rises only to about 0.7 volts.

In addition to the drive voltage dry vs. time,FIG. 5illustrates current flowing IQ2-G into the gate terminal of the MOSFET and current IQ1-base flowing into the base terminal of the bipolar transistor. As demonstrated, a brief pulse of current flows into the gate terminal of the MOSFET as its gate-to-source capacitance is charged. Also, a continuous current of about 10 milliamperes (“mA”) flows into the base terminal of the bipolar transistor. Accordingly, the current flowing into the control terminal of a switch can also be employed to detect the type of switch being used in a circuit.

Turning now toFIG. 6, illustrated is a block diagram of an embodiment of a controller (e.g., an application specific integrated circuit (“ASIC”)) constructed according to the principles of the present invention. The controller provides an adaptable drive function dependent on a detected switch embodied in a circuit employing the same (see, e.g., the power converter ofFIGS. 1 and 2). Other types of controllers that provide an adaptable drive function for a switch dependent on the detected switch are well within the broad scope of the present invention.

The controller includes a sample and hold circuit SundH that estimates the output voltage by sampling a voltage of a bias winding of a transformer (e.g., the bias winding P2of the transformer TX1inFIGS. 1 and 2). A comparator circuit Comp includes several comparators to compare a voltage VSuH produced by the sample and hold circuit SundH with a ramp voltage Ref_exp to determine the off time of the drive voltage drv. An output of the comparator circuit is a signal designated Freig. When the signal Freig is high, demagnetization of the transformer has been detected and the drive voltage dry of the controller can be switched on. A timer (designated “Timer”) of the controller produces a pulse-width modulated signal Gin, which determines various conditions under which the drive voltage dry is switched on. Thus, the comparator circuit Comp and timer “Timer” determine when the drive voltage dry can be switched on for a switch. A reference circuit (designated “Reference”) generates various reference voltages used internally by the controller.

A timing circuit SuHclk provides timing when sampling is being performed. The timing circuit SuHclk uses the output of the timer “Timer” to control the timing when a feedback signal FB (e.g., the feedback signal FB produced by the bias winding P2of the transformer Tx1ofFIGS. 1 and 2) is sampled. Various circuit configurations to control timing of a feedback signal FB may be employed to advantage. A current control circuit CC_control calculates when the controller can be switched on to provide a constant output current because the controller can be employed to control a combination of constant voltage/constant current characteristic of a circuit such as a power converter. Thus, the off time of the drive voltage dry for a switch is controlled by a combination of the timing circuit SuHclk and the current control circuit CC_control.

In the controller, the longer of the off times calculated by the timing circuit SuHclk and the current control circuit CC_control is taken as controlling for the off time of the drive voltage dry for a switch. In a voltage-control mode, the calculation of the off time is longer in the timing circuit SuHclk. In a constant-current mode, the timing of the current control circuit CC_control is longer. Thus, the comparator circuit Comp, timing circuit SuHclk and the current control circuit CC_control operate to determine the timing of the drive voltage dry for the switch. An overvoltage protection circuit OVP of the controller provides overvoltage protection for the power converter, and transitions the controller to a safe mode (i.e., the drive voltage dry is switched off), when an abnormal condition of the bias voltage VP is detected. The controller also includes a startup circuit (designated “startup”), a switch detector (designated “switch_detector”) and driver (designated “driver”) that will be described in more detail below.

Turning now toFIGS. 7 to 11, illustrated are diagrams of embodiments of portions of a controller constructed according to the principles of the present invention. Beginning withFIG. 7, illustrated is a startup circuit employable as the startup circuit (designated “startup”) ofFIG. 6. The startup circuit measures the bias voltage VP and when the bias voltage VP is higher than a startup level, a start signal “start” is set high to enable operation of the controller. When the bias voltage VP is lower than an under-voltage lockout level, the start signal “start” is set low to disable operation of the controller. The under-voltage lockout level is dependent on a switch detect signal FET that represents whether a MOSFET or a bipolar transistor was detected in the circuit such as a power converter. Again, the detection of a MOSFET causes the controller to select a first mode of operation, whereas the detection of a bipolar transistor causes the controller to select a second mode of operation. The under-voltage lockout level is set to a higher level when the controller operates in the first mode of operation than when the controller operates in the second mode of operation. In the environment illustrated inFIGS. 1 and 2, the startup level is higher than the under voltage lockout level to ensure that enough energy is stored in the capacitor C4to maintain operation of the controller120after startup until the voltage at the output has risen high enough to power the controller120via the bias winding P2of the transformer TX1.

The circuitry710provides a level shifting function to set the under-voltage lockout level lower when a bipolar transistor is detected. The circuitry710includes comparator U2, inverter U3, 5-volt voltage-reference V1and resistors R2, R3, R4, R5, R6, R7. A MOSFET frequently requires a higher drive voltage at its gate terminal then the base terminal of a bipolar transistor to completely turn the MOSFET on. Accordingly, the under-voltage lockout level at which the controller is enabled to operate is set higher when a MOSFET is detected. The circuit illustrated inFIG. 7is configured to produce a lower switch-off voltage than a switch-on voltage. The circuitry720produces a logical output coupled to the non-inverting input of a comparator E1. The output of the comparator E1is coupled to both inputs of an OR gate U1, the output of which is coupled to the non-inverting input of a comparator E2. The output of the comparator E2produces the start signal “start”. The comparator E2and OR gate U1increase the slope of the start signal during transition between high and low state. The circuitry720represents a simulated current consumed by the controller to improve accuracy of its operation.

Turning now toFIG. 8, illustrated is a switch detector employable as the switch detector (designated “switch_detector”) ofFIG. 6. In the illustrated embodiment, the switch detector detects whether a switch coupled to a drive signal such as the drive voltage dry is, for instance, a MOSFET or a bipolar transistor. When the start signal “start” goes high, which is coupled to a “set” input terminal of latch2through the high-pass network formed with a capacitor C1and a resistor R1, the output Q of latch2is set high to initially signal operation in a MOSFET mode (a first mode of operation). The logic indicated inFIG. 8is operative so that for each pulse, as determined by a pulse-width modulated signal Gin (also referred to as “GIN”), the output Q of latch2can be reset low to indicate a bipolar transistor (for a bipolar mode or second mode of operation) if the drive voltage dry of the driver becomes less than a threshold level (e.g., three volts), when the pulse-width modulated signal GIN is high.

Inversely, the output Q of latch2is left or can be set high to indicate a MOSFET if the drive voltage dry of the driver becomes greater than the threshold level when the pulse-width modulated signal GIN is high. Timing for these operations is controlled by a comparator U1with 3-volt reference Vref coupled to its inverting input. The output of the comparator U1is coupled to the “set” input of latch1, the output of which is coupled to an OR gate U2to signal when the drive voltage dry is greater than three volts. The output of the OR gate U2is coupled to a D flip-flop U5. The output of the D flip-flop U5is coupled to the “reset” input of latch2. Further timing for these operations is controlled by the pulse-width modulated signal GIN that is coupled through the high-pass network formed with the capacitor C2and the resistor R2, the output of which is coupled to the “reset” input of latch1. The pulse-width modulated signal GIN is also coupled to the reset input of the D flip-flop U5.

Turning now toFIG. 9, illustrated is a driver employable as the driver (designated “driver”) ofFIG. 6. The driver produces a series of pulses for the drive signal such as the drive voltage dry to control a switch. The switch detect signal FET indicates whether the switch is a MOSFET (for a first mode of operation) or a bipolar transistor (for a second mode of operation). If the switch detect signal FET is high, the switch has been detected as a MOSFET; otherwise, the switch has been detected as a bipolar transistor. The pulse-width modulated signal Gin is the signal that determines when the drive voltage dry is high or low. When the pulse-width modulated signal Gin is high, the drive voltage dry is high, and vice versa. The complement pulse-width modulated signal GinN is the complement of the pulse-width modulated signal Gin. The start signal “start” is a signal that is set high when the controller is in an active mode. The signal GND represents local circuit ground.

In operation, when the switch detect signal FET is high, a switch S6is off and a switch S5is on. An inverter U2provides signal inversion to control the switches S5, S6. Accordingly, a current limiter “current_limiter” or the voltage limiter “voltage_limiter” is selected by the switch detect signal FET to control a characteristic of the drive voltage drv. When the controller initiates operation at startup, the switch detect signal FET is set high, thereby representing the first mode of operation (i.e., the driven switch is assumed to be a MOSFET). A switch S4is switched on when the start signal “start” is high to enable operation of the driver. The switch S4is configured to connect or disconnect the bias voltage VP from the current limiter current_limiter or the voltage limiter voltage_limiter. A switch S3is to ensure the drive voltage dry is low when the start signal “start” is low, and a switch S1pulls the drive voltage dry low when the complement pulse-width modulated signal GinN is high. Thus, the driver produces the drive voltage dry for the switch based on the pulse-width modulated signal Gin.

Turning now toFIG. 10, illustrated is a schematic drawing of the current limiter “current_limiter” illustrated inFIG. 9that limits a current of the drive voltage dry when a bipolar transistor has been detected by the controller (during the second mode of operation), as indicated by the switch detect signal FET set low. The pulse-width modulated signal Gin is coupled through a resistor R2to the base of a bipolar transistor Q1. The signal Vdd is coupled to the bias voltage VP by switches S4, S6when the switch detect signal FET is set low, as indicated inFIG. 9. An output of the current limiter is the drive voltage drv. The bipolar transistor Q1is an active device to limit a current produced at the output of the current limiter. A pair of diodes D1, D2limit a base voltage of the bipolar transistor Q1with respect to the drive voltage dry to about one diode drop (i.e., to about 0.7 volts). Accordingly, a constant voltage is produced across a resistor R1when the pulse-width modulated signal Gin is high, thereby limiting a current that can flow from the output of the current limiter. Thus, the current limiter is configured to limit a current for the control terminal of the switch (via the drive voltage drv) to a current limit when the controller operates in the second mode of operation.

Turning now toFIG. 11, illustrated is a schematic drawing of the voltage limiter “voltage_limiter” illustrated inFIG. 9that limits a voltage of the drive voltage dry when a MOSFET has been detected by the controller (during the first mode of operation), as indicated by the switch detect signal FET set high. As described previously with reference toFIG. 10, an input to the voltage limiter is the pulse-width modulated signal Gin and an output signal is the drive voltage drv. The signal Vdd is coupled to the bias voltage VP by switches S4, S5when the switch detect signal FET is set high, as indicated inFIG. 9. The level shifter E1shifts the voltage level of the pulse-width modulated signal Gin, which is about five volts, by a factor of three to produce a 15-volt signal on the left terminal of a resistor R1. The resistor R1in conjunction with Zener diode D1(e.g., a 10 volt Zener diode) produces a 10 volt signal at the base of bipolar transistor Q1, the collector of which is coupled through a resistor R2to the signal Vdd. Accordingly the signal Vdd, which is the same as the drive voltage drv, is clamped at the emitter of bipolar transistor Q1to about 10 volts minus a diode drop produced between the base and emitter of transistor Q1. Thus, the circuit illustrated inFIG. 11is operative as a voltage limiter when the switch detect signal FET is set high indicating detection of a MOSFET. Thus, the voltage limiter is configured to limit a voltage for the control terminal of the switch (via the via the drive voltage drv) to a voltage limit when the controller operates in the first mode of operation.

Turning now toFIG. 12, illustrated is another embodiment of a switch detector. While the switch detector ofFIG. 12may be embodied in a controller according to the principles of the present invention, the initial state of a switch detect signal QM-Bis opposite to that of the switch detect signal FET described previously above. In either case, however, the switch detector detects whether a switch coupled to a drive signal such as the drive voltage dry is, for instance, a MOSFET or a bipolar transistor. Upon initial application of bias voltage Vp to the controller, the bias voltage Vp rises, eventually exceeding a threshold voltage of, for instance, two volts. This condition is detected by a comparator C04, which produces an output signal coupled to high pass filter F05. The output of high-pass filter F05is coupled to the reset input of a flip-flop FF03. The flip-flop FF03accordingly resets the switch detect signal QM-Bto a low state, indicating that the switch is initially assumed to be a bipolar transistor. The switch detect signal QM-Bremains in a low state until the drive voltage drv, which is connected to low-pass filter F01, exhibits a voltage greater than two volts, which is detected by comparator C02. The low-pass filter F01is included in the circuit to remove possible extraneous noise from the drive voltage drv. If comparator C02detects the filtered drive voltage dry greater than two volts, its output goes high, which is coupled to the set input of the flip-flop FF03. In this case, the flip-flop FF03sets the switch detect signal QM-Bhigh, indicating the switch is a MOSFET.

Thus, a controller for a switch and a method of operating the same has been introduced herein. In one embodiment, the controller is configured to measure a voltage of a control terminal of the switch and select a first mode of operation (e.g., indicating that the switch is a MOSFET) if the voltage of the control terminal is greater than a threshold voltage, and a second mode of operation (e.g., indicating that the switch is a bipolar transistor) if the voltage of the control terminal is less than the threshold voltage. The controller may include a voltage limiter configured to limit a voltage for the control terminal of the switch to a voltage limit during the first mode of operation. The controller may include a current limiter configured to limit a current for the control terminal of the switch to a current limit during the second mode of operation. An under-voltage lockout level of the controller may be set to a higher level during the first mode of operation than during the second mode of operation. The controller may include a timer configured to produce a pulse-width modulated signal. The controller is configured to control a duty cycle of the switch to regulate an output voltage of a power converter. The controller may initiate operation in the first mode of operation at startup.

Those skilled in the art should understand that the previously described embodiments of a switched-capacitor power converter and related methods of operating the same are submitted for illustrative purposes only. While the principles of the present invention have been described in the environment of a power converter, these principles may also be applied to other systems such as, without limitation, a power amplifier or a motor controller. For a better understanding of power converters, see “Modern DC-to-DC Power Switch-mode Power Converter Circuits,” by Rudolph P. Severns and Gordon Bloom, Van Nostrand Reinhold Company, New York, N.Y. (1985) and “Principles of Power Electronics,” by J. G. Kassakian, M. F. Schlecht and G. C. Verghese, Addison-Wesley (1991).