Methods and apparatus for reducing the average-to-minimum power ratio of communications signals in communications transmitters

Methods and apparatus for reducing the average-to-minimum power ratio (AMPR) of communications signals in communications transmitters. An AMPR-reducing method includes detecting a sample of a baseband signal having a magnitude less than a predetermined low-magnitude threshold. The magnitude and/or angle of the baseband signal is modified in the temporal vicinity of the detected low-magnitude sample, to form a modified baseband signal having a reduced AMPR. The baseband signal is modified by scaling an insertion pulse by a complex pulse insertion vector and combining the resulting scaled complex insertion pulse with the baseband signal in the temporal vicinity of the detected low-magnitude sample. The pulse insertion angle may be set to any angle within a pulse insertion vector range determined by a vector defining the detected low-magnitude sample and a vector that is orthogonal to the trajectory of the baseband signal.

FIELD OF THE INVENTION

The present invention relates to communications systems and methods. More specifically, the present invention relates to methods and apparatus for reducing the average-to-minimum power ratio (AMPR) of communications signals.

BACKGROUND OF THE INVENTION

Radio frequency (RF) communications systems, such as cellular and wireless area networks, are ubiquitous in today's world. A key and essential component of every RF communications system is the RF transmitter. As illustrated inFIG. 1, an RF transmitter100generally comprises a baseband processor102, a frequency upconverter104, a power amplifier (PA)106and an antenna108. The purpose of the baseband processor102is to generate a baseband signal s(t) containing a message to be transmitted and formatted in accordance with a predetermined modulation scheme. The purpose of the frequency upconverter104is to upconvert the baseband signal s(t) to RF, so that the message is capable of being propagated through space (i.e., transmitted over the air) to a remote receiver. The PA106is used to increase the power of the RF signal before it is radiated by the antenna108, to compensate for attenuation of the RF signal as it is transmitted over the air to the remote receiver.

In modern RF transmitters, the message to be transmitted is first digitized in the form of a binary-source data stream. The baseband processor102then groups the data bits in the binary-source data stream into a sequence of N-bit words, where N is some positive integer, and maps the pattern of bits in each N-bit word to one of M=2Npossible symbols. The M symbols are defined by the particular modulation scheme being employed, and affect how the amplitude and/or angle of the RF carrier signal is varied (i.e., modulated) to carry the message in the original binary-source data stream to the remote receiver. By mapping each N-bit word to one of M possible symbols, N=log2M bits can be transmitted in each symbol.

Conceptually, the symbols generated by the baseband processor102can be visualized as a sequence of weighted impulses. These impulses have essentially infinite bandwidth. To limit their bandwidth, the baseband processor102is further configured to shape each symbol by a band-limiting pulse p(t).

Mathematically, the baseband signal s(t) can be expressed as a sequence of pulse-shaped symbols:

s⁡(t)=∑n⁢an⁢p⁡(t-nTs),
where n is a symbol index, anis the nthsymbol in the sequence of symbols, p(t) is the pulse at time t, and Tsis the symbol period. anis either a real or complex number having one of M possible states. For example, in the quadrature phase-shift keying (QPSK) modulation scheme, M=4, and anis given by an=ejπ(2dn+1)/2, where dnis an integer selected from the set {0, 1, 2, 3}.

Because the baseband signal s(t) is in general a complex signal, it is usually expressed in terms of its in-phase (I) and quadrature (Q) components, i.e., as s(t)=I(t)+jQ(t) and the baseband processor102is configured to generate separate pulse-shaped I and Q baseband signals for each of the I and Q channels of the RF transmitter.

FIG. 2is a drawing showing how the baseband signal s(t) is processed in terms of its I and Q components in a practical RF transmitter200. The RF transmitter200comprises a baseband processor202, I-channel and Q-channel digital to analog converters204and206, a transmit local oscillator (Tx-LO)208, a quadrature modulator210; a PA212; and an antenna214. Because of its use of the quadrature modulator210, the RF transmitter200is referred to in the description that follows as the “quadrature-modulator-based” transmitter200.

The quadrature modulator210includes an I-channel mixer216, a Q-channel mixer218, a ninety-degree phase shifter220, and a subtractor222. The I-channel and Q-channel digital to analog converters204and206convert the pulse-shaped I and Q baseband signals from the baseband processor202into analog I and Q baseband signals. The quadrature modulator210then upconverts the analog I and Q baseband signals to RF. Specifically, the I-channel mixer216mixes the analog I baseband signal with an RF carrier signal provided by the Tx-LO208, while the Q-channel mixer218mixes the analog Q baseband signal with a ninety-degree phase-shifted version of the RF carrier signal produced at the output of the ninety-degree phase shifter220. The upconverted I- and Q-channel RF carrier signals are then combined by the subtractor222, to produce the desired modulated RF carrier signal. Finally, the modulated RF carrier signal is amplified by the PA212and radiated over the air to a remote receiver by the antenna214.

One advantage of the quadrature-modulator-based RF transmitter200is that both amplitude and angle (i.e., frequency or phase) modulation can be introduced into the RF carrier signal by simply controlling the amplitudes of the I and Q baseband signals. However, a significant drawback is that it is not very energy efficient, particularly when the modulation scheme being employed is a non-constant envelope modulation scheme.

In an effort to use the RF spectrum as efficiently as possible, modern communications systems commonly employ non-constant envelope modulation schemes in which both the amplitude and angle of the baseband signal s(t) are varied. As illustrated inFIG. 3, use of a non-constant envelope modulation scheme results in a modulated RF carrier signal at the RF input RFin of the PA212having a non-constant (i.e., time varying) envelope. To prevent the PA212from clipping the signal peaks of these signals, the input power of the modulated RF carrier signal must be backed off to ensure that the PA212always operates in its linear region of operation. In other words, the PA212must be operated as a “linear” PA when a quadrature modulator is used.

While employing power back-off does help to ensure PA linearity, it also results in a significant reduction in energy efficiency. The energy efficiency of an RF transmitter is determined in large part by the efficiency of the RF transmitter's PA. The energy efficiency of the PA is defined as the ratio of the PA RF output power to the direct current (DC) power supplied to the PA212from the RF transmitter's constant voltage supply Vs. Energy efficiency is therefore high when the PA is operating at high RF output powers, but low when the PA is operating at low RF output powers. In most applications, the PA operates at high or peak RF output powers only for very short periods of time. For all other times (i.e., most of the time), the RF output power is backed off, resulting in a substantial reduction in energy efficiency.

Low energy efficiency is undesirable in most applications. It is particularly undesirable in battery-powered RF transmitters, such as those used in cellular handsets, since it results in shortened battery life. Fortunately, an alternative type of communications transmitter known as a polar transmitter is available which avoids the linearity versus efficiency tradeoff of the quadrature-modulator-based transmitter200. In a polar transmitter the amplitude information (i.e., the signal envelope) is temporarily removed from the non-constant envelope signal. The remaining signal, which has a constant envelope, is upconverted to RF and applied to the RF input port of the PA while the previously removed signal envelope is used to dynamically control the power supplied to the PA. Because the signal applied to the RF input of the PA has a constant envelope, a more efficient nonlinear PA can be used without the risk of signal peak clipping.

FIG. 4is a drawing showing the basic elements of a polar transmitter400. The polar transmitter400comprises a baseband processor402; a Coordinate Rotation Digital Computer (CORDIC) converter (i.e., rectangular-to-polar converter)404; an amplitude path including an amplitude path DAC406and amplitude modulator408; an angle path including an angle path DAC410and angle modulator412; a PA414; and an antenna416. The purpose of the CORDIC converter404is to convert the digital rectangular-coordinate pulse-shaped I and Q baseband signals from the baseband processor402to digital polar-coordinate amplitude and angle component signals ρ and θ. The amplitude and angle path DACs406and410convert the digital amplitude and angle component signals ρ and θ into analog amplitude and angle modulation signals. In the amplitude path, the amplitude modulator408then modulates a direct current power supply voltage Vsupply (e.g., as provided by a battery) by the amplitude information in the analog amplitude modulation signal. The resulting amplitude-modulated power supply signal Vs(t) is coupled to the power supply port of the PA414. Meanwhile, in the angle path the angle modulator412operates to modulate an RF carrier signal by the angle information in the analog angle modulation signal, to produce an angle-modulated RF carrier signal which is coupled to the RF input port RFin of the PA414.

The angle-modulated RF carrier signal at the RF input port RFin of the PA414has a constant envelope (seeFIG. 5). As alluded to above, this permits the PA414to be configured to operate in its nonlinear region of operation (i.e., as a “nonlinear” PA) without the risk of signal peak clipping. Typically the PA414is implemented as a highly-efficient switch-mode PA (e.g., as a Class D, E or F switch-mode PA) operating between compressed and cut-off states. When configured in this manner, the envelope information in the amplitude-modulated power supply signal Vs(t) is restored at the RF output RFout of the PA414as the PA414amplifies the angle-modulated RF carrier signal. By operating the PA414as a switch and dynamically controlling the power supplied to it, the polar transmitter400is able to achieve significantly higher energy efficiencies than the quadrature-modulator-based RF transmitter200.

Although the polar transmitter400is more energy efficient than the quadrature-modulator-based transmitter200, the amplitude and angle component signals ρ and θ typically have much higher signal bandwidths than the rectangular-coordinate I and Q baseband signals from which they derive. This so-called “bandwidth expansion” phenomenon occurs during the rectangular-to-polar conversion process performed by the CORDIC converter404. The high signal bandwidths are manifested as high-frequency events in the amplitude and angle component signals ρ and θ and are highly undesirable. Not only do the high-frequency events tend to degrade the modulation accuracy of the polar transmitter400, they also cause the transmission spectrum to extend beyond its intended band-limited channel, resulting in adjacent channel interference and an increase in receive band noise. These effects can be very difficult to deal with, especially when modulation accuracy and noise limit standards must be adhered to.

The extent to which high-frequency events appear in the amplitude and angle component signals ρ and θ is very much dependent on the modulation scheme being employed. Modulation schemes that produce signals having a high average-to-minimum power ratio (AMPR) generally have a very large angle component bandwidth. In fact, for modulation schemes that produce signal magnitudes that pass through zero, as illustrated in the signal trajectory diagram inFIG. 6, the signal phase changes very abruptly, by as much as 180 degrees, resulting in an angle component signal θ having essentially infinite bandwidth. Signals of such high bandwidth cannot be accurately processed and transmitted by the polar transmitter400, or by any type of transmitter for that matter.

Various techniques have been proposed to reduce high-frequency events in polar domain signals. One approach, known as “hole blowing,” involves identifying symbols (or samples of symbols) in the baseband signal s(t) during which the magnitude of the signal falls below a predetermined low-magnitude threshold, and then raising the magnitude of the baseband signal s(t) in the temporal vicinity of the identified symbols or samples so that the AMPR of the signal is reduced. The term “hole blowing” is used since the effect of applying the technique is to produce a “hole” in the signal trajectory diagram of the baseband signal s(t). As illustrated inFIG. 7, the “hole” forces the trajectory of the modified baseband signal ŝ(t) to not pass too close to the origin, resulting in a desired reduction in the bandwidth of the signal.

The conventional hole blowing technique is described in detail in U.S. Pat. No. 7,054,385. As explained there, the baseband signal s(t) is modified by adding correction pulses to it, to form the modified baseband signal:

s^⁡(t)=∑n⁢an⁢p⁡(t-nTs)+∑m⁢bm⁢r⁡(t-tm),
where r(t) is the correction pulse, m is the perturbation index, tmrepresents the times when the baseband signal s(t) is perturbed (i.e., the times when the correction pulse r(t) is inserted), and bmis a perturbation sequence representing the amplitude scaling and/or angle shifting applied to the correction pulse r(t).

As shown inFIG. 8, in generating the modified baseband signal ŝ(t) the baseband signal s(t) from the baseband processor102is fed forward to an analyzer802. The analyzer802then determines the perturbation times tmby detecting low-magnitude events in the baseband signal s(t) that are below the predetermined low-magnitude threshold. In response to detected low-magnitude events, the analyzer802generates the perturbation sequence bm. A pulse-shaping filter804generates the correction pulse r(t), scales the pulse by the perturbation sequence bm, and finally adds the scaled correction pulses to the original baseband signal s(t) to produce the desired AMPR-reduced modified baseband signal ŝ(t).

While the conventional hole blowing technique can be helpful in reducing AMPR in some applications, it can be ineffective, deficient or even detrimental in others. The conventional hole blowing technique estimates the signal trajectory of the baseband signal s(t) and identifies low-magnitude events based on only two data points of the baseband signal s(t) at a time. Using only two data points can lead to errors in detecting low-magnitude events and can underestimate the level of AMPR reduction that needs to be performed in order to satisfy a particular design requirement or standards specification. The conventional hole blowing technique also requires that the angle of each correction pulse insertion vector be orthogonal to the signal trajectory of the baseband signal s(t). This rigid requirement on the required angle of the pulse insertion vector can increase the complexity of AMPR-reducing circuitry, and lacks any flexibility that would allow the angle of the pulse insertion vector to be adjusted to satisfy a desired balance or combination of in-band and out-of-band noise performance characteristics. Finally, the conventional hole blowing technique is incapable of taking into account prior modifications to the baseband signal s(t). In some circumstances, this can diminish the overall effectiveness of AMPR reduction. In particular, for those modulation schemes having multiple constellation points near the origin in the complex signal plane, the inability to take into account prior modifications to the baseband signal s(t) can result in the generation of pulse insertion vectors that effectively cancel one another out. Such a result defeats the purpose of hole blowing since the signal trajectory of the baseband s(t) can still pass near or through the origin even after the AMPR reduction process has been applied.

It would be desirable, therefore, to have AMPR-reducing methods and apparatus for communications transmitters that are effective at reducing the AMPR of communications signals but which are not plagued by the drawbacks and limitations associated with conventional hole blowing techniques.

SUMMARY OF THE INVENTION

Methods and apparatus for reducing the average-to-minimum power ratio (AMPR) of communications signals in communications transmitters are disclosed. An exemplary method includes first generating samples of a baseband signal that is formatted in accordance with a non-constant envelope modulation scheme. Next, samples in a first set of samples are analyzed to detect whether any of the samples has a magnitude less than a predetermined low-magnitude threshold. If a low-magnitude event is detected among the samples in the set, the magnitude and/or angle of the baseband signal is modified in the temporal vicinity of the detected low-magnitude sample, thereby forming a modified baseband signal having a reduced AMPR. The baseband signal is modified by scaling an insertion pulse by a complex pulse insertion vector and combining the resulting scaled complex insertion pulse with the baseband signal in the temporal vicinity of the detected low-magnitude sample. Unlike conventional hole blowing techniques, the pulse insertion vector angle is not restricted to being orthogonal to the trajectory of the baseband signal in the temporal vicinity of the detected low-magnitude event. Rather, it may be adjusted and set to any angle within a range of angles determined by the sample vector defining the detected low-magnitude sample and a vector that is orthogonal to the trajectory of the baseband signal in the temporal vicinity of the low-magnitude sample.

The AMPR-reducing methods and apparatus of the present invention also identify low-magnitude samples based on sets of samples that include at least three samples, thereby improving the accuracy and reliability of detecting low-magnitude events in the baseband signal compared to conventional hole blowing techniques, which identify low-magnitude events based on only two data points of the baseband signal at a time.

The AMPR-reducing methods and apparatus of the present invention may be employed in any type of transmitter in which a high AMPR is of concern. Examples of their use in a generalized transmitter, quadrature-modulator-based transmitter, and polar transmitter are illustrated and described.

Further features and advantages of the present invention, including descriptions of the structure and operation of the above-summarized and other exemplary embodiments of the invention, will now be described in detail with respect to accompanying drawings, in which like reference numbers are used to indicate identical or functionally similar elements.

DETAILED DESCRIPTION

Referring toFIG. 9, there is shown a radio frequency (RF) transmitter900including circuitry for reducing the average-to-minimum power ratio (AMPR) of communications signals, according to an embodiment of the present invention. The RF transmitter comprises a baseband processor902, a digital-to-analog converter (DAC)904, a frequency upconverter906, a power amplifier (PA)908, and an antenna910. The baseband processor902includes a baseband modulator912, a pulse-shaping filter914, and an AMPR reduction circuit916. In this and other exemplary embodiments described below, the baseband processor902, including the baseband modulator912, pulse-shaping filter914, and AMPR reduction circuit916, are formed as a digital signal processor (DSP) in one or more integrated circuits. The DSP may implemented as hardware or a combination of hardware and software, such as a microprocessor, microcontroller, field-programmable gate array, or other programmable or nonprogrammable integrated circuit, as will be appreciated by those of ordinary skill in the art.

The baseband modulator912is configured to generate a sequence of symbols at a symbol clock rate from data bits in a digital message to be transmitted. The sequence of symbols is then filtered by the pulse-shaping filter914and sampled by a sample clock (or an oversampling clock if oversampling is used) to provide a sequence of samples representing an unmodified baseband signal s(t)=I(t)+jQ(t), where I(t) and Q(t) are the real (i.e., in-phase) and imaginary (i.e., quadrature phase) components, respectively, of the unmodified baseband signal s(t).

The modulation scheme employed by the baseband modulator912is a non-constant envelope modulation scheme determined by design requirements and/or set by a standard. In one embodiment, the RF transmitter900is configured to operate in a third generation (3G) mobile telecommunications system and employs a non-constant envelope modulation scheme standardized by the 3G Partnership Project (3GPP), such as the Hybrid Phase Shift Keying (HPSK) non-constant envelope modulation scheme used in 3G Universal Mobile Telecommunications System (UMTS) networks or one of the non-constant envelope modulation schemes used by the 3G High-Speed Downlink Packet Access (HSDPA) or High-Speed Uplink Packet Access (HSUPA) communication protocols. In another embodiment, the RF transmitter900is configured for operation in a wireless local area network (LAN) and employs an orthogonal frequency division multiplexing (OFDM) non-constant envelope scheme, such as specified by the Institute of Electrical and Electronics Engineers (IEEE) 802.11 body of standards. While the RF transmitter900is suitable for use in UMTS and wireless LAN applications, those of ordinary skill in the art will appreciate and understand that it is not limited to use in any particular system or application or to any particular modulation scheme or standard. In fact, it may be adapted for use in any communications system in which a non-constant envelope modulation is used and in which a high AMPR is of concern. Further, whereas the exemplary embodiments are described in the context of RF applications, those of ordinary skill in the art will readily appreciate and understand that the methods and apparatus of the present invention are not limited to wireless or RF applications, and may be adapted for use in wired transmitters, such as those configured to transmit over a cable or fiber optic link.

In the exemplary embodiment shown inFIG. 9, the AMPR reduction circuit916is coupled in a feed-forward arrangement between the pulse-shaping filter914and the output of the baseband processor902. In another embodiment, shown inFIG. 10, a similar AMPR reduction circuit1016is coupled between the output of a baseband processor1002and pulse-shaping filter914in a feedback arrangement. Use of the feedback arrangement inFIG. 10is advantageous in that it allows AMPR reduction to be performed on samples of the baseband signal s(t) that may have been previously modified by the AMPR reduction circuit1016, thereby providing more enhanced and accurate control of the signal trajectory, and avoiding the problem of successive modifications of the baseband signal s(t) from possibly canceling one another out.

The AMPR reduction circuit916of the baseband processor902inFIG. 9comprises a local minimum event detector918, an orthogonal vector generator920, a look-up table (LUT)922(labeled as “circle LUT” in the drawing), an AND logic gate924(or other logic equivalent), a pulse generator926, a multiplier928, and a combiner930in the main signal path of the RF transmitter900. The AMPR reduction circuit1016of the RF transmitter1000inFIG. 10is similar, except that it is configured in a feedback arrangement, as explained above.

The local minimum event detector918operates to detect samples in successive three-sample sets of samples of the baseband signal s(t) having magnitudes below a predetermined low-magnitude threshold, and upon detecting such low-magnitude samples, signify the detection of a local minimum event to the AND logic gate924. The samples that are analyzed comprise previously unmodified (in the case of the feed-forward configuration inFIG. 9) or possibly previously modified samples (in the case of the feedback configuration inFIG. 10).

For each three-sample set analyzed by the local minimum event detector918, the orthogonal vector generator920calculates an orthogonal vector that originates from the origin in the complex signal plane (seeFIG. 11) and is orthogonal to a trajectory vector approximating the trajectory of the baseband signal s(t). Using the vector coordinates of the orthogonal vector as a reference, the circle LUT922provides a threshold sample (xc, yc) that intersects with the low-magnitude threshold circle along the direction of the orthogonal vector.

Based on the coordinates of the threshold sample (xc, yc) and the coordinates of the lowest-magnitude sample among the samples in each three-sample set, a pulse insertion vector is calculated. For those three-sample sets in which the local minimum event detector918had detected a local minimum event, the AND logic gate924passes the pulse insertion vector to the multiplier928, which scales an insertion pulse provided by the pulse generator926according to the magnitude and phase of the pulse insertion vector. The AMPR reduction circuit916(or1016) is configured to generate the pulse insertion vector based on the difference between a vector defining the threshold sample (xc, yc) and the sample vector defining the lowest-magnitude sample (xn, yn) (seeFIG. 11). In an alternative embodiment, the AMPR reduction circuit916(or1016) is configured so that it may set the angle of the pulse insertion vector to have any angle α between the direction of the sample vector and the direction of a vector between the lowest-magnitude sample (xn, yn) and the threshold sample (xc, yc), i.e., any angle within a pulse insertion vector range (shaded area inFIG. 11).

Finally, after the insertion pulse has been scaled by the pulse insertion vector, the scaled insertion pulse is combined with the baseband signal s(t) by the combiner930in the main signal path of the RF transmitter900(or1000) to provide the desired AMPR-reduced baseband signals ŝ(t)=Î(t)+j{circumflex over (Q)}(t), where Î(t) is the real (i.e., in-phase) component of the AMPR-reduced signal and {circumflex over (Q)}(t) is the imaginary (i.e., quadrature phase) component of the AMPR-reduced signal.

The flowchart inFIG. 12and the drawings inFIGS. 13-21illustrate in more detail the AMPR reduction method performed by the AMPR reduction circuit916of the baseband processor902inFIG. 9. (The method performed by the AMPR reduction circuit1016of the baseband processor1002inFIG. 10is similar, except that it operates on samples that may have been previously modified, as explained above.) In the first step1202of the AMPR reduction method1200a first three-sample set of samples (xn+1, yn+1), (xn, yn), (xn−1, yn−1) is loaded into the local minimum event detector918. Note that the samples in the first three-sample set (xn+1, yn+1), (xn, yn), xn−1, yn−1), as well as the samples in subsequent three-sample sets, may be temporally adjacent (i.e., consecutive) or sequential but nonconsecutive. Next in steps1204and1206(seeFIGS. 12,13and14) the local minimum event detector918determines which sample among the three samples in the first three-sample set (xn+1, yn+1), (xn, yn), (xn−1, yn−1) has the lowest magnitude1204), and the sample that has the next-lowest magnitude (step1206).

At decision1208, the local minimum event detector918determines whether a local minimum event is present in the first three-sample set (xn+1, yn+1), (xn, yn), xn−1, yn−1). A local minimum event is present if the middle sample (xn, yn) in the three-sample set (xn+1, yn+1), (xn, yn), (xn−1, yn−1) has a magnitude less then the low-magnitude threshold and is the sample among the three samples with the lowest magnitude. If a local minimum event is not detected (“no” at decision1208), at step1210the next three-sample set (xn+2, yn+2), (xn+1, yn+1), (xn, yn) is loaded into the local minimum event detector918and steps1204-1208are repeated. On the other hand, if a local minimum event is detected (“yes” at decision1208), the local minimum event detector918generates a “local minimum event detected” output signal, which is fed to a first input of the AND logic gate924.

FIG. 15Ais a drawing of a local minimum event detection circuit1500that may be used to implement the local minimum event detection portion of the local minimum event detector918. The local minimum event detection circuit1500comprises a group of multipliers1502, a first group of adders1504, a second group of adders1506, and a NOR logic gate1508. The multipliers of the group of multipliers1502and the adders of the first and second groups of adders1504and1506may be formed in a variety of different ways. For example, the multipliers may be formed from logic gates using Wallace trees or a Dadda multipliers and the adders may be formed from logic gates using ripple-carry or carry-lookahead adders, as will be appreciated and understood by those of ordinary skill in the art. The group of multipliers1502and the first group of adders1504operate to form the sums of the squares of the x and y coordinates of each sample of the three-sample set (xn+1, yn+1), (xn, yn), (xn−1, yn−1), i.e., [(xn+1)2+(yn+1)2], [(xn)2+(yn)2], [(xn−1)2+(yn−1)2]. Collectively, these values provide an accurate indication of the relative magnitudes of the three samples. The second group of adders1506operates to subtract the square of the magnitude of the middle sample (xn, yn) from the square of the magnitude of the “next” sample (xn+1, yn+1), and also subtract the square of the magnitude of the middle sample (xn, yn) from the square of the magnitude of the “prior” sample (xn−1, yn−1). The most significant bit (MSB) sign bits at the outputs of the second group of adders1506determine whether the magnitude of the middle sample (xn, yn) is the lowest magnitude among the prior, middle and next samples. If it is, the MSB sign bits at the outputs of both adders of the second group of adders1506are both at a logic “0” and the output of the NOR logic gate1508is a logic “1,” indicating the detection of a local minimum event. Otherwise, the local minimum event detected output of the NOR logic gate1508remains at a logic “0.”

FIG. 15Bis a drawing of an alternative local minimum event detection circuit1510, which may be used to implement the local minimum event detection portion of the local minimum event detector918. The alternative local minimum event detection circuit1510is similar to the local minimum event detection circuit1500inFIG. 15A, except that it incorporates delay (“D”) flip-flops1512and1514. The D flip-flops1512and1514provide a pipelining function for the samples, thereby reducing the number of multipliers and adders that are needed to perform the local minimum event detection.

After the local minimum event detector918has detected the presence of a local minimum event at decision1208, at step1210a trajectory vector (Δx, Δy) approximating the trajectory of the baseband signal s(t) through the three samples is calculated. (For explanation purposes, in the description that follows, it is assumed that a local minimum event was detected in the first three-sample set (xn+1, yn+1), (xn, yn), (xn−1, yn−1).) According to one embodiment, illustrated inFIG. 16, the trajectory vector (Δx, Δy) is defined as the vector (Min1−Min2)=[(xn−xn−1), (yn−yn−1)]=(Δx, Δy) between the lowest magnitude sample (xn, yn) and the next-lowest-magnitude sample (in this example, the prior sample (xn−1, yn−1) is the next-lowest-magnitude sample). In an alternative embodiment, the trajectory vector (Δx, Δy) is defined as the vector between the prior and next samples, i.e., (Δx, Δy)=[(xn+1−xn−1), (yn+1−yn−1)].

FIG. 17Aa drawing of a trajectory vector calculation circuit1700that may be used to compute the trajectory vector in step1212of the AMPR reduction method1200. The trajectory vector calculation circuit1700comprises a group of multipliers1702, a first group of adders1704, an MSB sign bit adder1706, first and second multiplexers1708and1710, and first and second output adders1712and1714. The group of multipliers1702and first group of adders1704operate to determine the squares of the magnitudes of the prior and next samples (xn−1, yn−1) and (xn+1, yn+1). The MSB sign bit adder1706subtracts the square of the magnitude of the next sample (xn+1, yn+1) from the square of the magnitude of the prior sample (xn−1, yn−1). The MSB sign bit at the output of the MSB sign bit adder1706provides an indication of which of the prior and next samples (xn−1, yn−1) and (xn+1, yn+1) has the lowest magnitude. The one that has the lowest magnitude is the sample that has the next-lowest magnitude among the samples of the three-sample set (xn+1, yn+1), (xn, yn), (xn−1, yn−1). The MSB sign bit is inputs of both the first and second multiplexers1708and1710. Accordingly, if the MSB sign bit has a value indicating that the next sample (xn+1, yn+1) is the next-lowest-magnitude sample among the three-sample set (xn+1, yn+1), (xn, yn), (xn−1, yn−1), the xn+1and yn+1coordinates of the next sample (xn−1, yn+1) are passed to the outputs of the first and second multiplexers1708and1710. Otherwise, the xn−1and yn−1coordinates of the prior sample (xn−1, yn−1) are passed to the multiplexer outputs. Finally, the first and second output adders1712and1714subtract the x and y coordinates of the middle sample (xn, yn) from the outputs of the first and second multiplexer1708and1710to produce the trajectory vector (Δx, Δy).

As explained above, in an alternative embodiment, the trajectory vector (Δx, Δy) is defined by the vector difference between the next and prior samples, i.e., (Δx, Δy)=[(xn+1−xn−1), (yn+1−yn−1)].FIG. 17Bis a trajectory vector calculation circuit1720may be used to generate the trajectory vector (Δx, Δy) according to that alternative embodiment.

After the trajectory vector (Δx, Δy) has been determined at step1210, at step1212the orthogonal vector (i.e., the vector that is orthogonal to the trajectory vector) is determined. Given that the dot product of two orthogonal vectors is zero, the orthogonal vector is determined by solving the equation: (trajectory vector)·(orthogonal vector)=0. As shown inFIG. 18, the solution to the equation yields two opposing orthogonal vectors (−Δy, Δx) and (Δy, −Δx). To ensure proper AMPR reduction, it is necessary to select the orthogonal vector that has the appropriate direction. The appropriate direction is the direction that facilitates pushing the signal trajectory of the baseband signal s(t) away from the origin, rather than towards it. This selection process, which is performed in step1214, can be performed in a variety of different ways. An exemplary orthogonal direction selection algorithm is described below.(1) Solve for (x, y) orthogonal direction from intersection of trajectory and orthogonal vectors (seeFIGS. 18 and 19):
by−ax=ci.
basic formula
ynΔx−xnΔy=cii.
equation for trajectory vector through (xn, yn)
y′Δx−x′Δy=ciii.
equation for trajectory vector through (x′, y′)
ynΔx−xnΔy=y′Δx−x′Δyiv.
substitute for constant c
y′=x′Δx/Δyv.
equation for orthogonal vector through (x′, y′)
ynΔx−xnΔy=(x′Δx/Δy)Δx−x′Δyvi.
substitute for y′ in equation vi and solve for x′
−Δy(ynΔx−xnΔy)=(Δx2+Δy2)x′vii.
(Δx2+Δy2) is positive and can be dropped
−Δy(ynΔx−xnΔy)=x′viii.
ynΔx−xnΔy=y′Δx−(y′Δy/Δx)Δyix.
substitute for x′ in equation vi and solve for y′
Δx(ynΔx−xnΔy)=(Δx2+Δy2)y′x.
(Δx2+Δy2) is positive and can be dropped
Δx(ynΔx−xnΔy)=y′xi.(2) Sign of (x, y) used to find appropriate direction of orthogonal vector.

In the example provided here, the orthogonal vector (Δy, −Δx) is determined and selected to be the appropriate orthogonal vector. The orthogonal vector (Δy, −Δx) and its direction are shown inFIG. 19.

After the appropriate orthogonal vector is determined at step1214, at step1216the coordinates of the orthogonal vector are used as a reference into the circle LUT922to retrieve a threshold sample (xc, yc) that intersects with both the low-magnitude threshold circle and the orthogonal vector. The threshold sample (xc, yc) and its relationship to the orthogonal vector are shown inFIG. 20.

After the threshold sample (xc, yc) has been determined at step1216, at step1218the x and y coordinates of the lowest-magnitude middle sample (xn, yn) are subtracted from the x and y coordinates of the threshold sample (xc, yc) to determine the pulse insertion vector (Δx2, Δy2), as shown inFIG. 21. In an alternative embodiment, the AMPR-reduction circuit916(pr1016) is configured to set the pulse insertion vector so that it terminates on the low-magnitude threshold circle but so that it can have any angle a within the range of angles between the sample vector direction and the direction of the vector formed by the difference between the orthogonal vector terminating at the threshold sample (xc, yc) and the vector defining the middle sample (xn, yn), i.e., an angle within the pulse insertion vector range (shaded area inFIG. 21).

Finally, at step1220a complex insertion pulse provided by the pulse generator926is scaled by the pulse insertion vector (Δx2, Δy2) to provide the desired, scaled complex insertion pulse, which is finally added to the baseband signal s(t) in the temporal vicinity of the lowest magnitude middle sample (xn, yn) to locally reduce the AMPR of the baseband signal s(t). (Note that if a local minimum event was not previously detected at decision1208, the output of the AND logic gate924would be zero, in which case a zero-valued insertion pulse would be produced, effectively resulting in a zero-valued insertion pulse being combined with the baseband signal s(t).)

After the scaled complex insertion pulse has been combined with the baseband signal s(t), the next three-sample set of samples (xn+2, yn+2), (xn+1, yn+1), (xn, yn) is loaded into the local minimum event detector918and steps1204and the remaining steps are repeated. The method1200is continuously repeated in the above-described manner until the RF transmitter900(or1000, if it is used) enters a non-transmit or powered down state.

As the AMPR reduction circuit916(or AMPR reduction1016) operates to reduce the AMPR of the baseband signal s(t), a modified baseband signal ŝ(t) having a lower AMPR is produced. The DAC904converts the modified baseband signal ŝ(t) to an analog baseband signal, which is then upconverted to RF by the frequency upconverter906and applied to the RF input RFin of the PA908. Finally, the antenna910radiates the amplified and modulated RF carrier signal over the air to a remote receiver.

The AMPR-reducing methods and apparatus described above can be advantageously employed in any type of transmitter in which a high AMPR is of concern. For example,FIG. 22illustrates how the AMPR-reducing methods and apparatus of the present invention may be used to reduce high-frequency events in the amplitude and angle component signals ρ(t) and θ(t) of a polar transmitter2200. The polar transmitter2200comprises a baseband processor902(or1002) including an AMPR reduction circuit similar to the AMPR reduction circuit916(or1016) inFIGS. 9 and 10; a Coordinate Rotation Digital Computer (CORDIC) converter (i.e., rectangular-to-polar converter)2204; an amplitude path including an amplitude path digital filter2206, amplitude path DAC2208, amplitude path analog filter2210and amplitude modulator2212; an angle path including an angle path digital filter2214, angle path DAC2216, angle path analog filter2218and angle modulator2220; a PA2222; and an antenna2224.

The AMPR reduction circuit916(or1016) operates on the baseband signal s(t)=I(t)+jQ(t) as described above, to provide a modified baseband signal ŝ(t) comprised of modified I and Q signal components Î(t) and {circumflex over (Q)}(t). The modulation scheme employed by the baseband modulator912of the baseband processor902(or1002) is a non-constant envelope modulation scheme. According to one embodiment, the baseband modulator912is configured to generate a baseband signal s(t) that is formatted according to the HPSK non-constant envelope modulation scheme specified by the 3GPP for use in 3G UMTS systems. In another embodiment, the baseband modulator912is configured to employ a non-constant envelope modulation scheme specified for use in the 3G High-Speed Packet Access (HSPA) communication protocols. In yet another embodiment, the polar transmitter2200is configured for operation in a wireless LAN and the baseband modulator912is configured to employ an OFDM non-constant envelope scheme, such as specified by the IEEE 802.11 body of standards.

After the unmodified baseband signal s(t)=I(t)+jQ(t) has been generated, and the AMPR reduction circuit916(or1016) has reduced the AMPR of the baseband signal s(t) to produce the desired AMPR-reduced baseband signal ŝ(t)=Î(t)+j{circumflex over (Q)}(t), the CORDIC converter2204converts the rectangular-coordinate modified Î(t) and {circumflex over (Q)}(t) signal components of the modified baseband signal ŝ(t) to digital polar-coordinate modified amplitude and angle component signals {circumflex over (ρ)}(t) and {circumflex over (θ)}(t).

Due to the prior AMPR-reducing operation performed by the AMPR reduction circuit916(or1016), the digital polar-coordinate amplitude and angle component signals {circumflex over (ρ)}(t) and {circumflex over (θ)}(t) have reduced high-frequency content. The reduced high-frequency content is advantageous for a number of reasons. First, it eliminates the need for, or at least reduces the design specifications of, the amplitude and angle path digital filters2206and2214and the amplitude and angle path analog filters2210and2218. For example, in one embodiment the amplitude and angle path analog filters2210and2218were able to be implemented as 3rdorder Bessel low-pass analog filters having cut-off frequencies of 15 MHz and 30 MHz, respectively, whereas similar performance without the benefit of the AMPR-reducing methods and apparatus of the present invention required 5thorder filters with higher cut-off frequencies and more complex linear and non-linear digital filters. Application of the AMPR-reducing methods and apparatus of the present invention also allowed the design requirements of the PA2222to be relaxed, particularly its required dynamic range.

After the digital polar-coordinate amplitude component signals {circumflex over (ρ)}(t) have been filtered by the amplitude path digital filters2206, converted to an analog amplitude modulation signal by the amplitude path DAC2208and, and filtered by the amplitude path analog filter2210in the amplitude path, the amplitude modulator2212modulates a direct current power supply voltage Vsupply according to the amplitude information in the analog amplitude modulation signal. The resulting amplitude-modulated power supply signal Vs(t) is coupled to the power supply port of the PA2222. Meanwhile, in the angle path the angle modulator2220operates to modulate an RF carrier signal according to the angle information in the analog angle modulation signal provided at the output of the angle path analog filter2218. The resulting angle-modulated RF carrier signal is applied to the RF input RFin of the PA2222.

The PA2222comprises an amplifier having a final-stage switch-mode type of PA (e.g., as a Class D, E or F switch-mode PA) operating between compressed and cut-off states. As the PA2222amplifies the angle-modulated RF carrier signal produced at the output of the angle modulator2220the envelope information in the amplitude-modulated power supply signal Vs(t) from the amplitude modulator2212is restored at the RF output RFout of the PA2222. Finally, the antenna2224radiates the final amplified amplitude- and angle-modulated RF carrier signal over the air to a remote receiver.

FIGS. 23A and 23Bare signal trajectory diagrams obtained from simulations performed on a polar transmitter, similar to the polar transmitter2200inFIG. 22, in which the polar transmitter was configured to process and transmit HSDPA signals. The signal trajectory diagrams illustrate the effectiveness of the AMRP-reducing methods and apparatus of the present invention in reducing the AMPR of the HSDPA signals for pulse insertion vectors at the boundaries of the pulse insertion vector range (shaded area inFIG. 21). In particular, the simulation results inFIG. 23Awere obtained using a pulse insertion vector having the same direction as the sample vector, while the simulation results inFIG. 23Bwere obtained using a pulse insertion vector having an angle defining the other extreme of the pulse insertion vector range, i.e., a pulse insertion vector determined by the vector difference between the vector defining the lowest-magnitude sample (xn, yn) and the orthogonal vector defining the threshold sample (xc, yc), as shown inFIG. 21.

Comparing the signal trajectory diagrams inFIGS. 23A and 23Breveals that the degree to which hole blowing occurs (and AMPR reduced) varies depending on what angle a the pulse insertion vector is set to within the pulse insertion vector range. The signal trajectory diagrams inFIGS. 23A and 23Bboth show hole blowing effects and consequent reductions in AMPR. However, the hole blowing effect is more pronounced inFIG. 23Bthan it is inFIG. 23A, as expected since the magnitude of the pulse insertion vector is larger in the former case.

The dependence of the hole blowing effect on the angle α the pulse insertion vector can be exploited during design to control the amount of AMPR reduction performed on a signal. It can also be used to help satisfy a required or desired balance or combination of in-band and out-of-band noise performance characteristics. For example, for a design imposing strict limits on out-of-band noise, or a design having hardware constraints (for example, a PA with limited dynamic range), a more aggressive hole blowing approach with a pulse insertion vector having a large angle α and large magnitude, such as the pulse insertion vector (Δx2, Δy2) inFIG. 21could be used. On the other hand, for designs focusing more on limiting or controlling in-band noise, a less aggressive hole blowing approach in which a smaller magnitude pulse insertion vector having an angle α closer to zero, i.e., more toward the sample vector direction, could be used.

As explained above, the AMPR-reducing methods and apparatus of the present invention may be exploited in other transmitter topologies.FIG. 24shows, for example, how the AMPR-reducing methods and apparatus of the present invention are used in a quadrature-modulator-based transmitter2400. The quadrature-modulator-based transmitter2400comprises a baseband processor902(or1002) including an AMPR reduction circuit916(or1016) similar to the AMPR reduction circuit916(or1016) inFIG. 9(orFIG. 10); an I-channel DAC2402; a Q-channel DAC2404; a quadrature modulator2406; a PA2408; and an antenna2410. The AMPR reduction circuit916(or1016) operates on the baseband signal s(t)=I(t)+jQ(t) as described above, to provide a modified baseband signal ŝ(t) comprised of modified I and Q signal components Î(t) and {circumflex over (Q)}(t). The modified I and Q signal components Î(t) and {circumflex over (Q)}(t) are converted to analog signals by the I- and Q-channel DACs2402and2404, and then upconverted to RF and combined by the quadrature modulator2406. The PA2408comprises a linear PA (e.g., a Class A, B or AB PA) that operates to amplify the modulated RF carrier signal produced at the output of the quadrature modulator2406. Finally, the antenna2410radiates the amplified and modulated RF carrier signal over the air to a remote receiver. Due to the prior reduction in AMPR of the I and Q signal components, the extent to which power must be backed off to maintain PA linearity is reduced, thereby easing the design requirements of the PA2408, in particular its required dynamic range.

While various embodiments of the present invention have been described above, it should be understood that they have been presented by way of example, and not limitation. It will be apparent to persons skilled in the relevant art that various changes in form and detail may be made therein without departing from the spirit and scope of the invention. The scope of the invention should, therefore, be determined not with reference to the above description, but should instead be determined with reference to the appended claims, along with the full scope of equivalents to which such claims are entitled.