Power switching circuit improved to reduce loss due to reverse recovery current

In a power switching circuit, a second commutation member has a second commutation path electrically connected in parallel to a first commutation path and a second diode provided in the second commutation path and electrically connected antiparallel to a semiconductor switching element. While the semiconductor switching element is off, the second commutation path allows a second current based on the inductive load to flow therethrough in a forward direction of the second diode within a commutation period. The second diode has a second reverse recovery time shorter than a first reverse recovery time of the first diode. A second inductance of the second commutation path is higher than a first inductance of the first commutation path.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is based on Japanese Patent Application 2008-34524 filed on Feb. 15, 2008. The descriptions of the Patent Application are all incorporated herein by reference.

FIELD OF THE INVENTION

The present invention relates to power switching circuits with a switching element for switching on and off the switching element to thereby drive inductive loads. More particularly, the present invention relates to such power switching circuits with a diode connected in antiparallel to the switching element; is diode allows a flywheel current to flow therethrough.

BACKGROUND OF THE INVENTION

Semiconductor switching elements are installed in a power converter, such as an inverter, for driving inductive loads, such as motors. In order to bypass a current flowing in a semiconductor switching element when the semiconductor switching element is turned off, a diode is connected in antiparallel to the semiconductor switching element. Specifically, when the semiconductor switching element is turned off, a current, such as a flywheel current, flowing in the semiconductor switching element continues to flow through the diode.

Normal PN junction diodes have a reverse recovery time between instantaneous switching from a forward bias (conducting state) to a reverse bias (non-conducting state). During the reverse recovery time, a reverse recovery current flows through a PN junction diode due to movement of minority carriers (a few charge carriers) across the p-n junction.

For this reason, when a PN junction diode is used to be connected in antiparallel to such a semiconductor switching element, during the reverse recovery time, the reverse recovery current flows through the diode, resulting in increasing switching loss and noise.

In order to reduce the reverse recovery current, a fast recovery diode (FRD) having the minority carriers each with a reduced lifetime or a Schottky barrier diode having a very small reverse recovery time can be used as a diode connected in antiparallel to a semiconductor switching element installed in such a power converter.

On the other hand, when a MOSFET is used as a semiconductor switching element to be installed in a power converter, because MOSFETs each have an intrinsic diode intrinsically formed in antiparallel thereto, the intrinsic diode is used to cause a flywheel current to continuously flow therethrough.

Specifically, when a MOSFET is used as a semiconductor switching element to be installed in a power converter and the intrinsic diode is used to cause a flywheel current to continuously flow therethrough, during the reverse recovery time, the reverse recovery current flows through the intrinsic diode. This results in increasing switching loss and noise.

Particularly, the higher the breakdown voltage of a MOSFET is, the higher the reverse recovery current flowing trough the intrinsic diode of the MOSFET is. A MOSFET with a super junction structure (SJ) can be used as a semiconductor switching element to be installed in a power converter because it provides both a high breakdown voltage and a low resistance. However, when a MOSFET with the super junction structure (SJ) is used as a semiconductor switching element to be installed in a power converter, the reverse recovery current flowing through the intrinsic diode becomes high and has a steep time characteristic. This results in increasing loss and noise.

In order to reduce loss and/or noise due to the reverse recovery current flowing though the intrinsic diode of a MOSFET, nonpatent document 1 “POWER-MOSFET APPLICATION TECHNOLOGY”, published by Nikkan Kogyo Shimbun Ltd., discloses, on page 139, that a diode with a low breakdown voltage is connected in anti-series to the intrinsic diode of a MOSFET. This prevents a forward current from flowing through the intrinsic diode.

In order to reduce loss and/or noise due to the reverse recovery current flowing though the intrinsic diode of a MOSFET, U.S. Pat. No. 6,058,037 corresponding to Japanese Patent Application Publication No. H10-327585 discloses an external circuit that applies a low reverse voltage to the intrinsic diode of a MOSFET. The low reverse voltage applied to the intrinsic diode causes reverse recovery of the intrinsic diode, thus reducing loss due to the reverse recovery current flowing though the intrinsic diode of the MOSFET.

European Patent Application Publication No. EP1814216 corresponding to Japanese Patent Application Publications No. 2006-141167 and No. 2006-141168 also discloses such an external circuit that applies a low reverse voltage to the intrinsic diode of a MOSFET.

However, the approach disclosed in the nonpatent document 1 may complicate the circuit structure consisting of the MOSFET and the diode. The approach disclosed in the nonpatent document 1 also may create loss due to the forward voltage drop across the anti-series diode upon the MOSFET being on.

The approach disclosed in each of the US Patent and EP Patent Application Publication requires, as the external circuit, a low-voltage source for creating the low voltage, a switch connected between the low-voltage source and the intrinsic diode, and a driver for turning on the switch to thereby apply the low voltage to the intrinsic diode.

Thus, the approach disclosed in each of the US Patent and EP Patent Application Publication may complicate the circuit structure consisting of the MOSFET and the external circuit, and remains loss due to the reverse recovery current based on the low voltage.

SUMMARY OF THE INVENTION

In view of the circumstances set forth above, an object of an aspect of the present invention is to provide power switching circuits having a simple structure and reducing loss and/or noise due to a diode, such as an intrinsic diode, connected in antiparallel to a switching element.

According to one aspect of the present invention, there is provided a power switching circuit. The power switching circuit includes a semiconductor switching element. The semiconductor switching element has a first terminal electrically connected to a power supply source and has a second terminal electrically connected to an inductive load. The power semiconductor switching element is configured to be switched on and off. The power switching circuit includes a first commutation member having a first commutation path and a first diode provided in the first commutation path. The first commutation path is electrically connected across the semiconductor switching element. The first diode is electrically connected antiparallel to the semiconductor switching element. While the semiconductor switching element is off, the first commutation path allows a first current based on the inductive load to flow therethrough in a forward direction of the first diode within a commutation period. The power switching circuit includes a second commutation member having a second commutation path and a second diode provided in the second commutation path. The second commutation path is electrically connected in parallel to the first commutation path. The second diode is electrically connected antiparallel to the semiconductor switching element. While the semiconductor switching element is off, the second commutation path allows a second current based on the inductive load to flow therethrough in a forward direction of the second diode within the commutation period. The first diode has a first reverse recovery time, the second diode has a second reverse recovery time. The second reverse recovery time is shorter than the first reverse recover time. The first commutation path has a first inductance. The second commutation path has a second inductance. The second inductance is higher than the first inductance.

According to the one aspect of the present invention, while the semiconductor switching element is off, the first commutation path allows the first current based on the inductive load to flow therethrough in the forward direction of the first diode within the commutation period. Similarly, while the semiconductor switching element is off, the second commutation path allows the second current based on the inductive load to flow therethrough in the forward direction of the second diode within the commutation period. Because the second inductance is higher than the first inductance, after the first current becomes zero, a current, such as a de-energizing current, continues to flow through the second commutation path based on an electromagnetic energy charged in the second inductance during the commutation period.

Thus, as compared with the structure in which no second inductance is provided in the second commutation path or the second inductance is lower than the first inductance, it is possible to reduce a reverse recovery current through the first diode. This allows the sum of a current flowing through each of the first and second diodes after the commutation period to be immediately reduced. Note that, because the second diode has a reverse recovery characteristic superior than that of the first diode based on the difference between their reverse recovery times, an adverse effect of the reverse recovery characteristic of the second diode is relatively reduced.

In addition, with the configuration of the one aspect of the present invention, a reverse recovery current that conventionally flowed only the first diode is separated to flow both the first diode and the second diode. This makes it possible to reduce the amount of minor carries charged in the first diode by a forward current flowing through the first diode.

Accordingly, the power switching circuit of the one aspect of the present invention has a simple structure in which the second inductance and the second diode are connected across the first diode; this simple structure achieves the effects set forth above. This therefore increases the availability of the power switching circuit.

an actually measured current that rises when the low-side MOSFET is ON in the power switching circuit according to the embodiment;

an actually measured current that rises when the low-side MOSFET is ON in a first comparative power switching circuit; and

an actually measured current that rises when the low-side MOSFET is ON in a second comparative power switching circuit.

DETAILED DESCRIPTION OF EMBODIMENTS OF THE INVENTION

Referring toFIG. 1A, there is provided a power switching circuit1for driving an inductive load47according to the embodiment of the present invention.

The power switching circuit1includes a direct current (DC) battery46and a half-bridge module5.

The half-bridge module5includes a pair (bridge arm) of series-connected high-side and low-side MOSFETs41H and41L serving as switching elements, intrinsic diodes42H and42L of the respective MOSFETs41H and41l, and Schottky barrier diodes (Schottky diodes)43H and43L. Each of the intrinsic diodes42H and42L serves as a first diode (first commutation diode), and each of the Schottky diodes43H and43L serves as a second diode (second commutation diode).

InFIG. 1A, reference character44H represents an inductance of wiring, and reference character44L represents an inductance of wiring.

Specifically, the half-bridge module5consists of an upper arm switch and a lower arm switch.

The upper arm switch consists of a high-side terminal49electrically connected to a positive electrode of the DC battery46, and a load-side terminal48electrically connected to one low-potential side of the inductive load47.

The upper arm switch also consists of the MOSFET41H whose drain is electrically connected to the high-side terminal49, and whose source is electrically connected to the load-side terminal48. Through a main-current path defined from the high-side terminal49to the load-side terminal48through the MOSFET41H, the on and off switchings of the MOSFET41H cause a current based on the DC battery46to intermittently flow.

The first diode42H is contained in a semiconductor chip of the MOSFET41H and connected in antiparallel to the MOSFET41H. The semiconductor chip of the MOSFET41H is illustrated by dashed lines.

A current path defined from the high-side terminal49to the load-side terminal48through the first diode42H will be referred to as “first commutation path P1a” hereinafter.

The second diode (Schottky Barrier Diode)43H is electrically connected to the inductance44H in series. The set of series-connected second diode43H and the inductance44H is electrically connected in antiparallel to the MOSFET41H integrated with the first diode42H.

A current path defined from the high-side terminal49to the load-side terminal48through the second diode43H and the inductance44H will be referred to as “second commutation path P2a” hereinafter.

More specifically, the high-side terminal49of the MOSFET41H consists of a branching point to which the drain D of the MOSFET41H, a positive electrode of the DC battery46, and the cathode of the second diode43H are electrically connected. The other high-potential side of the inductive load47is electrically connected to the positive electrode of the DC battery46.

Similarly, the load-side terminal48of the MOSFET41H consists of a branching point to which the source S of the MOSFET41H, the one low-potential side of the inductive load47, and the anode of the second diode43H.

The lower arm switch consists of a low-side terminal50electrically connected to a negative electrode of the DC battery46, and a load-side terminal51electrically connected to the one low-potential side of the inductive load47.

The lower arm switch also consists of the MOSFET41L whose drain D is electrically connected to the load-side terminal51, and whose source S is electrically connected to the low-side terminal50. Through a main-current path defined from the load-side terminal51to the low-side terminal50through the MOSFET41L, the on and off switchings of the MOSFET41L causes a current to intermittently flow.

The first diode42L is contained in a semiconductor chip of the MOSFET41L and connected in antiparallel to the MOSFET41L. The semiconductor chip of the MOSFET41L is illustrated by dashed lines.

A current path defined from the load-side terminal51to the low-side terminal50through the first diode42L will be referred to as “first commutation path P1b” hereinafter.

The second diode (Schottky diode)43L is electrically connected to the inductance44L in series. The set of series-connected second diode43L and the inductance44L is electrically connected in antiparallel to the MOSFET41L integrated with the first diode42L.

A current path defined from the load-side terminal51to the low-side terminal50through the second diode43L and the inductance44L will be referred to as “second commutation path P2b” hereinafter.

More specifically, the low-side terminal50of the MOSFET41L consists of a branching point to which the source S of the MOSFET41L, the negative electrode of the DC battery46, and the anode of the second diode43L are electrically connected.

Similarly, the load-side terminal51of the MOSFET41L consists of a branching point to which the drain D of the MOSFET41L, the one low-potential side of the inductance47, and the cathode of the second diode43L.

The second diode43H consisting of the Schottky diode has a reverse recovery time shorter than that of the corresponding first diode42H. Similarly, the second diode43L consisting of the Schottky diode has a reverse recovery time shorter than that of the corresponding first diode42L.

Each of the second diodes43H and43L can consist of a PN junction diode. In this modification, doping a breakdown layer close to a p-n junction of a PN junction diode with a heavy metal, such as gold (Au), platinum (Pt), or the like, reduces a lifetime of each of minority carriers of the PN junction diode. This reduces a reverse recovery time of the PN junction diode to thereby create a fast recovery diode to be used as each of the second diodes43H and43K.

The forward threshold voltage value of a Schottky diode is lower than that of a PN junction diode. For this reason, it is possible to reduce a flywheel current that flows through the first diode (intrinsic diode)42H, resulting in reducing a value of the inductance44H. Similarly, it is possible to reduce a flywheel current that flows through the first diode (intrinsic diode)42L, resulting in reducing a value of the inductance44L.

Each of the second diodes43H and43L can consist of the set of a PN junction diode and a Schottky diode that are electrically connected in parallel to each other. Each of the second diodes43H and43L can consist of a junction barrier Schottky diode so as to improve the breakdown characteristic of the junction barrier Schottky diode as compared with that of a normal Schottky diode.

The power switching circuit1further includes a controller10electrically connected to the gate of each of the MOSFETs41H and41L.

The controller10is capable of individually driving on and off each of the high-side switch (MOSFET)41H and the low-side switch (MOSFET)41L to thereby apply a voltage based on a DC voltage across the DC battery46to the inductive load47for driving the inductive load47.

The controller10is operative to variably control a duty (duty cycle) of each of the individual MOSFETs41H and41L, that is, the ratio of on duration of each of the individual MOSFET41H and41L per one cycle (on and off cycle) to thereby control operating conditions of the inductive load47.

Next, operations of the power switching circuit1according to the embodiment will be described hereinafter. Particularly, in the embodiment, operations of the power switching circuit1when the controller10cyclically switches the lower arm switch (low-side switch) on and off will be described hereinafter with reference toFIGS. 1A and 2A.

Referring toFIG. 2A, before time t1, both the MOSFETs41H and41L are off. In other words, there is a dead time during which both the MOSFETs41H and41L are off.

Note that, while the MOSFET41L is ON immediately before the dead time, a current based on the DC battery46flows through a closed loop consisting of the DC battery46, the inductive load47, and the MOSFET41L. This allows electromagnetic energy to be charged in the inductive load47.

Thus, during the dead time before time t1, a load current based on the electromagnetic energy stored in the inductive load47flows as a flywheel current through each of the first and second diodes42H and43H (first and second commutation paths P1aand P2a).

Specifically, the load current is distributed to flow through the first diode42H and flow through the second diode43H.

In a preferred example, the characteristics of the first diode42H and those of the second diode43H can be adjusted such that a distributed current flowing through the first diode42H is equal to that flowing through the second diode43H. In another preferred example, the characteristics of the first diode42H and those of the second diode43H can be adjusted such that a distributed current flowing through the second diode43H is higher than that flowing through the first diode42H.

The flywheel current flowing through the second diode43H allows electromagnetic energy to be charged in the inductance44H.

Thereafter, at time t1, the controller10controls the MOSFET41L to be switched on. This allows a current to flow through the channel of the MOSFET41L. This reduces the amount of the flywheel current flowing through each of the first and second commutation paths P1aand P2a.

Because the inductance44H causes a back EMF (Electro Motive Force) when the flywheel current flowing through the second commutation path P2ais reduced, the rate of reduction in the flywheel current flowing through the second commutation path P2ais lower than that of reduction in the flywheel current flowing through the first commutation path P1a.

For this reason, after time t1, when a forward current through the first diode42H becomes zero at time t2, a flywheel current i2continues to flow through the second diode43H based on the electromagnetic energy stored in the inductance44H.

After the forward current through the first diode42H becomes zero, a reverse recovery current ir flows through the first diode42H from the drain D to the source S from time t2to time t4. The reverse recover current ir is the highest at time t3between time t2and time t4.

As a result, during a period from time t2to time t4, a current component “ir−i2” defined by subtracting the flywheel current i2based on the electromagnetic energy charged in the inductance44H from the reverse recovery current ir flows through the channel of the MOSFET41L.

As illustrated inFIG. 2A, the period between time t2and time t4corresponds to, for example, a turn-on time of the MOSFET41L. For this reason, the channel resistance of the MOSFET41L during the period between time t2and time t4is higher than that during a stable on period of the MOSFET41L after the turn-on time.

Thus, if the inductance44H was negligible so that no flywheel current i2flows through the second commutation path P2a, the reverse recovery current ir flowing trough the MOSFET41L would cause an additional ohmic loss (resistance loss) in the MOSFET41L. The additional ohmic loss is represented by the product of the square of the reverse recovery current ir and the channel resistance.

In other words, a voltage across the channel of the MOSFET41L is reduced from a value identical to the voltage across the DC battery46upon the MOSFET41L being on to nearly zero upon the MOSFET41L being off.

However, if the inductance44H was negligible, during the period from time t2to time t4, the voltage across the channel of the MOSFET41L is sufficient large. This may cause an additional ohmic loss in the MOSFET41L; this additional ohmic loss is represented by the product of the current component and the voltage across the channel of the MOSFET41L.

In contrast, in the embodiment, the current component “ir−i2” defined by subtracting the flywheel current i2based on the electromagnetic energy charged in the inductance44H from the reverse recovery current ir flows trough the MOSFET41L during the period between time t2and time t4.

For this reason, it is possible for the power switching circuit1to reduce, by a level of the second flywheel current i2, a current flowing through the MOSFET41L during the period from time t2to time t4according to the embodiment as compared with a power switching circuit that does not consider the inductance44H in the second commutation path P2a. This results in reducing ohmic loss and noise in the MOSFET41L as compared with the power switching circuit that does not consider the inductance44H in the second commutation path P2a.

As described above, the power switching circuit1is configured to switch on and off the second switch41L while keeping the first switch41H off to thereby drive the inductive load47and reduce ohmic loss and noise in the MOSFET41L.

Modification of the power switching circuit1allows ohmic loss in the MOSFET41H to be reduced.

FIG. 1Bschematically illustrates an example of the structure of a power switching circuit1A according to the modification of the power switching circuit1.

Specifically, as a different point of the structure of the power switching circuit1A from that of the power switching circuit1, the inductive load47is electrically connected at its one low-potential side to the low-side terminal50, and at its other high-potential side to the load-side terminal51.

As well as operations of the power switching circuit1, operations of the power switching circuit1A when the controller10cyclically switches the higher arm switch (high-side switch) on and off will be described hereinafter with reference toFIGS. 1B and 2B.

Referring toFIG. 2B, before time t11, both the MOSFETs41H and41L are off. In other words, there is a dead time during which both the MOSFETs41H and41L are off.

Note that, while the MOSFET41H is ON immediately before the dead time, a current based on the DC battery46flows through a closed loop consisting of the DC battery46, the MOSFET41H, and the inductive load47. This allows electromagnetic energy to be charged in the inductive load47.

Thus, during the dead time before time t11, a load current based on the electromagnetic energy stored in the inductive load47flows as a flywheel current through each of the first and second diodes42L and43L (first and second commutation paths P1band P2b).

Specifically, the load current is distributed to flow through the first diode42L and flow through the second diode43L.

The flywheel current flowing through the second diode43L allows electromagnetic energy to be charged in the inductance44L.

Thereafter, at time t11, the controller10controls the MOSFET41H to be switched on. This allows a current to flow through the channel of the MOSFET41H. This reduces the amount of the flywheel current flowing through each of the first and second commutation paths P1band P2b.

Because the inductance44L causes a back EMF when the flywheel current flowing through the second commutation path P2bis reduced, the rate of reduction in the flywheel current flowing through the second commutation path P2bis lower than that of reduction in the flywheel current flowing through the first commutation path P1b.

For this reason, after time t11, when a forward current through the first diode42L becomes zero at time t12, a flywheel current i2acontinues to flow through the second diode43L based on the electromagnetic energy stored in the inductance44L.

After the forward current through the first diode42L becomes zero, a reverse recovery current ira flows through the first diode42L from the drain D to the source S from time t12to time t14. The reverse recovery current ira is the highest at time t13between time t12and time t14.

As a result, during a period from time t12to time t14, a current component “ira−i2a” defined by subtracting the flywheel current i2abased on the electromagnetic energy charged in the inductance44L from the reverse recovery current ira flows through the channel of the MOSFET41H.

As illustrated inFIG. 2B, the period between time t12and time t14corresponds to, for example, a turn-on time of the MOSFET41H. For this reason, the channel resistance of the MOSFET41H during the period between time t12and time t14is higher than that during a stable on period of the MOSFET41H after the turn-on time.

Thus, if the inductance44L was negligible so that no flywheel current i2aflows through the second commutation path P2b, the reverse recovery current ira flowing through the MOSFET41H would cause an additional ohmic loss in the MOSFET41H; this additional ohmic loss is represented by the product of the square of the reverse recovery current ira and the channel resistance.

In other words, a voltage across the channel of the MOSFET41H is reduced from a value identical to the voltage across the DC battery46upon the MOSFET41H being on to nearly zero upon the MOSFET41H being off.

However, if the inductance44L was negligible, during the period from time t12to time t14, the voltage across the channel of the MOSFET41H is sufficiently large. This may cause an additional ohmic loss in the MOSFET41H; this additional ohmic loss is represented by the product of the current component and the voltage across the channel of the MOSFET41L.

In contrast, in the embodiment, the current component “ira−i2a” defined by subtracting the flywheel current i2abased on the electromagnetic energy charged in the inductance44L from the reverse recovery current ira flows through the MOSFET41H during the period between time t12and time t14.

For this reason, it is possible for the power switching circuit1A to reduce, by a level of the second flywheel current i2a, a current flowing through the MOSFET41H during the period from time t12to time t14according to the embodiment as compared with a power switching circuit that does not consider the inductance44L.

InFIG. 2A, the waveform of a current flowing through the first commutation path P1aaccording to the embodiment is illustrated by solid line with reference character W1, and the waveform of a current flowing through the second commutation path P2aaccording to the embodiment is illustrated by solid line with reference character W2. In addition, the waveform of a current flowing through the MOSFET41L according to the embodiment is illustrated by solid line with reference character W3.

InFIG. 2A, the waveform of a current flowing through the second commutation path P2awhen the inductance44H is negligible is illustrated by dashed line with reference character W2aand the waveform of a current flowing through the MOSFET41L when the inductance44H is negligible is illustrated by dashed line with reference character W3a. Note that, inFIG. 2A, no reverse recovery current flows through the second diode43H because the Schottky diode is used as the second diode43H.

Similarly, InFIG. 2B, the waveform of a current flowing through the first commutation path P1baccording to the embodiment is illustrated by solid line with reference character W11, and the waveform of a current flowing through the second commutation path P2baccording to the embodiment is illustrated by solid line with reference character W12. In addition, the waveform of a current flowing through the MOSFET41H according to the embodiment is illustrated by solid line with reference character W13.

InFIG. 2B, the waveform of a current flowing through the second commutation path P2bwhen the inductance44H is negligible is illustrated by dashed line with reference character W12aand the waveform of a current flowing through the MOSFET41L when the inductance44L is negligible is illustrated by dashed line with reference to the character W13a. Note that, inFIG. 2B, no reverse recovery current flows through the second diode43L because the Schottky diode is used as the second diode43L.

As clearly illustrated inFIG. 2A, if the current through the second diode43H is zero during the reverse recovery current flowing through the first diode42H (see the waveform W2a), a current flowing through the MOSFET41H would be increased (see the waveform W3a). This would increase ohmic loss in the MOSFET41H.

However, in the embodiment, the current through the second diode43H continuously flows during the reverse recovery current flowing through the first diode42H (see the waveform W2). This allows a current flowing through the MOSFET41H to be reduced (see the waveform W3). This reduces ohmic loss and noise in the MOSFET41H.

Similarly, as clearly illustrated inFIG. 2B, if the current through the second diode43L is zero during the reverse recovery current flowing through the first diode42L (see the waveform W12a), a current flowing through the MOSFET41L would be increased (see the waveform W13a). This would increase ohmic loss in the MOSFET41L.

However, in the embodiment, the current through the second diode43L continuously flows during the reverse recovery current flowing through the first diode42L (see the waveform W12). This allows a current flowing through the MOSFET41L to be reduced (see the waveform W13). This reduces ohmic loss and noise in the MOSFET41L.

The principle of the embodiment of the present invention has been described with reference toFIGS. 1A and 1B. In addition to the descriptions, more actual operations of the power switching circuit1using an equivalent circuit of the first and second commutation paths P1aand P2aduring the period from time t2to time t4will be described hereinafter with reference toFIG. 3.

Specifically, inFIG. 3, as described above, reference character ir represents the reverse recovery current that flows through the first commutation path P1aduring the period from time t2to time t4. Reference character i2represents the flywheel current that flows through the second commutation path P2abased on the electromagnetic energy stored in the inductance44H during the period from time t2to time t4.

Reference character r1represents a resistance of the first commutation path P1a, and reference character L1represents an inductance of the first commutation path P1a. Reference character r2represents a resistance of the second commutation path P2a, and reference character L2represents an inductance of the second commutation path P2acorresponding to the inductance44H.

The reverse recovery current ir is designed to flow through the resistance r1and the inductance L1. The flywheel current i2is determined based on: the electromagnetic energy charged in the inductance44H (inductance L2) and a forward voltage drop ΔV across the second diode43H, and the resistance r2. The electromagnetic energy, referred to “E”, charged in the inductance44H (inductance L2) is expressed by the following equation.

Adjustment of the parameters, such as r1, r2, L1, L2, ΔV, allows the average value (integrated value) of the reverse recovery current ir over the period from time t2to time t4to be equal to the average value (integrated value) of the flywheel current i2over the period from time t2to time t4.

Specifically, the area S1of the waveform W1over time t2to time t4corresponding to the integrated value thereover is substantially equal to the area S2of the waveform W2over time t2to time t4corresponding to the integrated value thereover (seeFIG. 2A). Similarly, the area S11of the waveform W11over time t12to time t14corresponding to the integrated value thereover is substantially equal to the area S12of the waveform W12over time t12to time t14corresponding to the integrated value thereover (seeFIG. 2B).

This allows the average value of the difference in current between the reverse recovery current ir and the flywheel current i2to substantially become zero. This effectively reduces ohm loss and noise in the MOSFET41L due to the current difference.

Note that the average value (integrated value) of the reverse recovery current ir over the period from time t2to time t4equal to the average value (integrated value) of the flywheel current i2over the period from time t2to time t4is defined such that:

the average value (integrated value) of the flywheel current i2over the period from time t2to time t4lies within a range from 50 percent of the average value (integrated value) of the reverse recovery current ir over the period from time t2to time t4to 150 percent thereof.

More preferably, the average value (integrated value) of the reverse recovery current ir over the period from time t2to time t4equal to the average value (integrated value) of the flywheel current i2over the period from time t2to time t4is defined such that:

the average value (integrated value) of the flywheel current i2over the period from time t2to time t4is placed within a range from 50 percent of the average value (integrated value) of the reverse recovery current ir over the period from time t2to time t4to 100 percent thereof.

The relationship between the inductance L1and the inductance L2will be described hereinafter with reference toFIG. 4.

FIG. 4illustrates how the ratio, in percent, of the flywheel current i2to the reverse recovery current ir is vaned while the inductances L1and L2are varied. In other words,FIG. 4illustrates the relationships between a variable of the ratio of the flywheel current i2to the reverse recovery current ir, a variable of the inductance L1, and a variable of the inductance L2.

Note that the relationships illustrated inFIG. 4were obtained by, for example, simulations using the power switching circuit1assuming that the forward characteristics of the first diode42H are equal to those of the second diode43H.

Specifically,FIG. 4illustrates the ratio, in percent, of the average value of the flywheel current i2over the period from time t2to time t4to the average value of the reverse recovery current ir over the period from time t2to time t4. As clearly illustrated inFIG. 4, the higher the value of the inductance L2and the lower the value of the inductance L1are, the greater the ratio of the average value of the flywheel current i2to the average value of the reverse recovery current ir is.

Preferably, the value of the inductance L2be set to be two to eight times, more preferably, three to seven times, still more preferably, four to six times, greater than the value of the inductance L1.

For example, because a normal semiconductor switch to be used to the upper arm switch has an inductance of equal to or lower than 10 nanohenries (nH) in its first commutation path, the value of the inductance L2is preferably set to be tens of nanohenries. For this reason, as the inductance44H, an inductance of wiring of the second commutation path P2acan be used by, for example, adjusting the length of the second commutation path P2a.

If the ratio of the value of the inductance L2to the value of the inductance L1was lower than 2, it would be insufficient to reduce the current difference “ir−i2”, reducing the effect of decreasing loss and noise in the MOSFET41L.

In other words, when the ratio of the value of the inductance L2to the value of the inductance L1is equal to or higher than 2, the effect of decreasing loss and noise in the MOSFET41L can be maintained at a certain level.

Specifically, when the ratio of the value of the inductance L2to the value of the inductance L1is equal to or higher than 3, the effect of decreasing loss and noise in the MOSFET41L can be maintained at a higher level.

More specifically, when the ratio of the value of the inductance L2to the value of the inductance L1is equal to or higher than 4, the effect of decreasing loss and noise in the MOSFET41L can be maintained at a more higher level.

Otherwise, if the ratio of the value of the inductance L2to the value of the inductance L1was higher than 8, the inductance44H would prevent the flywheel current i2from flowing through the second diode43H at the start of a commutation period. The commutation period represents a period during which the flywheel current flows through each of the first and second diodes43H and44H upon the MOSFET41L being off; this commutation period corresponds to the period from time t1to time t2.

This would increase the ratio of the flywheel current through the first diode42H to the flywheel current i2through the second diode43H, resulting in increasing the reverse recovery current through the first diode42H during a reverse recovery period from time t2to time t4after the commutation period.

In addition, the flywheel current I2would be reduced, resulting in reducing the electromagnetic energy to be stored in the inductance44H. This would reduce the flywheel current i2during the reverse recovery period from time t2to time t4, resulting in increasing the current difference “ir−i2”. This would increase resistance loss in the MOSFET41L during the reverse recovery period from time t2to time t4.

In other words, when the ratio of the value of the inductance L2to the value of the inductance L1is equal to or lower than 8, it is possible to reduce the adverse effect of the inductance44H on the flywheel current i2through the second diode43H at the start of a commutation period. The effect of decreasing loss and noise in the MOSFET41L can be therefore maintained at a certain level.

When the ratio of the value of the inductance L2to the value of the inductance L1is equal to or lower than 7, it is possible to more reduce the adverse effect of the inductance44H on the flywheel current i2through the second diode43H at the start of a commutation period. The effect of decreasing loss and noise in the MOSFET41L can be therefore maintained at a higher level.

When the ratio of the value of the inductance L2to the value of the inductance L1is equal to or lower than 6, it is possible to still more reduce the adverse effect of the inductance44H on the flywheel current i2through the second diode43H at the start of a commutation period. The effect of decreasing loss and noise in the MOSFET41L can be therefore maintained at a more higher level.

Specifically, it is preferred that the value of the inductance L2be set such that the level of the flywheel current i2reach a steady-state level at the end of the reverse recovery period.

FIGS. 5 and 6schematically illustrate the first packaging example of the MOSFET41H integrated with the first commutation diode42H. In the first packaging example, the MOSFET41H is designed as a rectangular vertical MOSFET chip81.

The source82and gate83are mounted on one surface of the vertical MOSFET chip81, and the drain (not shown) is mounted on the other surface of the vertical MOSFET chip81. Reference character810represents a package with four leads84,85,86, and87located at one side thereof. The package810encapsulates the vertical MOSFET chip81with one ends of the four leads84to87projecting from the one side thereof.

The source82is joined to the other ends of the leads85and87by bonding wires89. Reference character88represents a conductor, such as a conductive substrate, provided in the package810. The drain is joined to the conductor88, and the conductor88is joined to the other end of the lead86by bonding wires89. The gate83is joined to the other end of the lead84by a bonding wire89.

In the first packaging example, the one end of one of the leads85and87to be coupled to the source82, such as the lead85, is electrically connected to the anode of the second diode43H. The one end of the other of the leads85and87to be coupled to the source82, such as the lead87, is joined to one end of a lead located through one side of a package that encapsulates the MOSFET41L and to a terminal to be electrically connected to inductive load47. The other end of the lead of the MOSFET41L to be joined at its one end to the lead87is joined to the drain of the MOSFET41L.

This configuration of the MOSFET41H allows the source82of the MOSFET41H, the anode of the second diode43H, the drain of the MOSFET41L, and the inductive load47to be separated at the source82of the MOSFET41H as the branching point thereamong. This allows the inductance in the first commutation path P1awith the first diode42H (the intrinsic diode) to be set to a minimum value.

In contrast, the second commutation path P2aincluding the second diode43H has an inductance of a conductive path (wire) connecting from the source82of the MOSFET41H to the anode of the second diode43H through the lead87. The inductance of the wire has normally one nH per 1 mm in increase of its longitudinal length although it depends on its shape and arrangement. This is because the inductance of a wire is normally proportional to the length of the wire.

Thus, adjustment of the length of wiring between the lead87and the anode of the second diode43H within a few centimeters allows a required value of the inductance L2to be obtained; this required value of the inductance L2is higher then a value of the inductance L1.

Note that, between a lead coupled to the source of a commercially available discreet vertical MOSFET package as the MOSFET41H and a lead coupled to the drain of the MOSFET41L, a parasitic inductance of the order of 10 nH as the inductance L1of the first commutation path P1ais presented.

In this case, the branching point can be located at the lead to be connected to the source electrode of the commercially available discrete vertical MOSFET package or located close thereto. In addition, to the branching point, the anode of the second diode43H can be joined through an inductance element of 40 to 60 nH.

The inductance element can be formed by a linear wire, a spiral wire, or a specific coil.

Because the branching point is arranged close to the lead of the source of the MOSFET package, it is possible to reduce the wiring inductance in the first commutation path P1a.

FIG. 7schematically illustrates the second packaging example of the MOSFET41H. In the second packaging example, the MOSFET41H is designed as a rectangular vertical MOSFET chip91with which a flywheel diode chip911serving the second diode43H is integrated.

The source92and gate93are mounted on one surface of the vertical MOSFET chip91, and the drain (not shown) is mounted on the other surface of the vertical MOSFET chip91. Reference character913represents a package with five leads94,95,96,97, and98located at one side thereof. The package913encapsulates the vertical MOSFET chip91and the flywheel chip911with one ends of the five leads94to98projecting from the one side thereof.

Reference character99represents a common conductor, such as a common conductive substrate,99provided in the package913. To the conductor99, the drain of the MOSFET chip91and the cathode of the flywheel diode chip911serving as the second diode43H are joined.

The source92is joined to the other ends of the leads95and97by bonding wires910. The other end of the lead94is joined to the gate of the MOSFET chip91by a bonding wire910, and the other end of the lead96is joined to the conductor99by bonding wires910. The lead98is joined to the anode912of the flywheel diode chip911by a bonding wire910.

In the second packaging example, a inductance element as the inductance44H is connected between the lead97and the lead98within the package913.

Because the second diode43H is integrated in the common package913together with the MOSFET chip91, a wiring inductance between the anode912of the second diode43H and the lead98and that between the lead97and the source electrode92are unaffected by external environment. Thus, it is possible to easily determine with high accuracy, variations in the wiring inductance between the anode912of the second diode43H and the lead98and those in the wiring inductance between the lead97and the source electrode92. Variations in the inductance element44H can be therefore reduced.

FIG. 8schematically illustrates the third package example of the MOSFET41H. In the third packaging example, in the configuration ofFIG. 7, a bonding wire110is provided in place of the leads97and98. The bonding wire110is located to directly connect between the source92and the anode912. In the third packaging example, the vertical MOSFET chip91serving as the MOSFET41H and the flywheel diode chip911serving as the second diode43H are integrated in the same package913so that the first and second commutation paths P1aand P2aare formed in the same package913. This allows the inductances L1and L2to be precisely created because they are unaffected by external environment.

FIG. 9schematically illustrates the fourth packaging example of the MOSFET41H. In the fourth packaging example, in the configuration ofFIG. 7, a bonding wire110is provided in place of the lead98. The bonding wire110is located to directly connect between the anode912and the other end of the lead97. To the one end (projecting end) of the lead97, no external elements are joined. The configuration allows the inductance44hbetween the source92and the anode912to increase.

FIG. 10schematically illustrates the fifth packaging example of the MOSFET41H. In the fifth packaging example, in the configuration ofFIG. 9, a conductor1111is located adjacent to the conductor99and integrated in the package913. To the conductor1111, the source92of the vertical MOSFET chip91and the anode912of the flywheel diode chip911serving as the second diode43H are joined.

This allows a value of the inductance44H between the source92and the anode912through the bonding wire1112, the conductor1111, and the bonding wire1110to increase.

This configuration of the MOSFET41H allows the source92of the MOSFET41H, the anode912of the flywheel diode911, the drain of the MOSFET41L, and the inductive load43to be separated at the source92of the MOSFET41H as the branching point thereamong. This allows the inductance in the first commutation path P1awith the first diode42H (the intrinsic diode) to be set to a minimum value.

In addition, the vertical MOSFET chip91serving as the MOSFET41H and the flywheel diode chip911serving as the second diode43H are integrated in the same package913so that the first and second commutation paths P1aand P2aare formed in the same package913. This allows the inductances L1and L2to be precisely created because they are unaffected by external environment.

Change of the shape of the conductor1111allows the value of the inductance L2and the value of the resistance r2to be easily adjusted with high accuracy.

FIG. 11schematically illustrates the sixth packaging example of the MOSFET41H. In the sixth packaging example, in the configuration ofFIG. 10, the conductor99and the conductor1111are mounted on an insulated substrate1112, and the conductor1111is located to extend around three sides and a part of the remaining side of the flywheel diode chip911.

One end and the other end of the conductor1111in its longitudinal (extending) direction are joined to the anode912and the source92by bonding wires1210and1210, respectively. That is, the conductor1111for connecting between the source92and the anode912is designed to sufficiently extend. This achieves an effect of increasing the inductance44H between the source92and the anode912with a compact structure.

In each of the first to sixth packaging examples, the MOSFET41H and the second diode43H are independently formed by separated chips, but can be integrated with each other in a common chip. In this first modification, the source of the MOSFET41H and the anode of the second diode43H are preferred to be separately formed. On the common chip, an inductance element as the inductance44H can be mounted.

In each of the first to sixth packaging examples, a wiring inductance44H or an inductance element44H is arranged between the source of the vertical MOSFET91and the anode of the second diode43H, but can be arranged between the drain of the vertical MOSFET91and the cathode of the second diode43H. Specifically, the inductance L2consists of the inductance of all of the first commutation path P1abetween the branching point at the source side of the vertical MOSFET91and the branching point at the drain side thereof.

In each of the first to sixth package examples, the reverse recovery current through the first diode42H of the vertical MOSFET91is reduced, but the present invention in not limited thereto.

Specifically, the second commutation path can be used to reduce a reverse recovery current through any one of various power semiconductor switching elements each having minor carrier charging effect that is the same as the vertical MOSFET chip91. The various power semiconductor switching elements include DMOSFETs with a high threshold voltage and MOSFETs each with a super junction structure (SJ). Note that, when a MOSFET with a super junction structure is used as the vertical MOSFET chip91(MOSFET41H), because the rate of change in the reverse recovery current is high, it is preferable to reduce the resistance r2of the second commutation circuit to thereby increase the rate of change in the flywheel current i2.

In addition, as illustrated inFIG. 12, when an IGBT is used as the vertical MOSFET chip91(MOSFET41H) and a fast recovery diode is used as the first diode42H and connected in antiparallel to the IGBT, a set of series-connected second diode (Schottky barrier diode)43H and an inductance element44H can be connected in parallel to the first diode42H. This can reduce an adverse effect of the reverse recovery current through the first diode42H like the embodiment.

In the embodiment and each of the first to sixth packaging examples, the present invention is applied to the power switching circuit1consisting of one half-bridge module5, but the present invention is not limited thereto.

Specifically, the present invention can be applied to inverters, DC to DC converters, and the like. For example, an inverter to which the present invention is applied consists of a plurality of half-bridge modules5for, for example, a plurality of multiphase windings. Such an inverter works to individually driving on and off the high-side switch (MOSFET)41H and the low-side switch (MOSFET)41L of each of the half-bridge modules5to thereby convert the DC voltage across the DC battery46to an AC (Alternating Current) voltage to be applied to each of the multiphase windings.

FIG. 13schematically illustrates the first to third graphs:

the first graph G1demonstrates the waveform of an actually measured current that rises when the MOSFET41L is ON in the power switching circuit1according to the embodiment;

the second graph G2demonstrates the waveform of an actually measured current that rises when the MOSFET41L is ON in a first comparative power switching circuit in which no inductance44H is provided in the second commutation path P2a(the second diode43H is only provided therein) in the structure of the power switching circuit1; and

the third graph G3demonstrates the waveform of an actually measured current that rises when the MOSFET41L is ON in a second comparative power switching circuit in which no second commutation path P2ais provided in the structure of the power switching circuit1.

FIG. 13clearly illustrates that the inductance44H provided in the second commutation path P2aallows the reverse recover current through the low-side switch (MOSFET41L) to be effectively reduced.

While there has been described what is at present considered to be the embodiment and its modifications of the present invention, it will be understood that various modifications which are not described yet may be made therein, and it is intended to cover in the appended claims all such modifications as fall within the true spirit and scope of the invention.