Communication receiving unit for the suppression of noise and interference signals

A communication receiving unit for the suppression of noise and jamming signals, which receiving unit comprises: PA0 a plurality of antenna elements (1A-1N); PA0 adaptive weighting filters (7.sub.A,a, . . . , 7.sub.N,k+1) connected thereto; PA0 a combination circuit (4) connected to the weighting filters (7.sub.A,a, . . . , 7.sub.N,k+1); PA0 a detector (5) connected to the combination circuit (4) suited to generate a replica (d) of the information-supplying carrier signal for demodulating the signal (y) of the combination circuit (4) to the information-related frequency band; and PA0 a unit (8) for adjusting the weighting filters (7.sub.A,a, . . . , 7.sub.N,k+1). The detector (5) comprises a circuit (40, 41) for determining a time-averaged power value (W.sub.1) concerning the signal (y) from the combination circuit (4), and a circuit (42, 43) for determining a time-averaged value (W.sub.2) of the cross-correlation concerning the output signal (y) and the replica (d). The adjusting unit (8) is designed for generating modified adjusting signals on the basis of the supplied time-averaged values (W.sub.1 and W.sub.2) for a more accurate adjustment of the weighting filters (7.sub.A,a, . . . , 7.sub.N,k+1).

BACKGROUND OF THE INVENTION 
1. Field of the Invention 
The invention generally relates to adapative communication systems and, in 
particular, to an adaptive communication receiving unit suitable for the 
suppression of noise and jamming signals; which receiving unit comprises: 
a plurality of antenna elements; 
a network of adaptive weighting filters connected to the antenna elements; 
a combination circuit connected to the networks; 
a detector connected to the combination circuit and provided with means for 
generating a replica (d) of the information-supplying carrier signal and 
with means for demodulating the signal (y) of the combination circuit to 
the information-related frequency band on the basis of the replica (d); 
and 
adjusting means for generating adjustment signals for the adaptive 
weighting filters. 
The signal produced by the combination circuit consists of three components 
obtained from the information-carrying communication signal, the jamming 
signal and the noise. 
2. Description of the Prior Art 
A receiving unit of the type as set forth in the opening paragraph is known 
from the article entitled "Adaptive Antenna Systems", by B. Widrow, P. R. 
Mantey, L. J. Griffiths and B. B. Goode, Proceedings of the IEEE, Vol. 55, 
1967, pp. 2143-2159. The communication receiving unit described in this 
article is designed on a transmission characteristic of the combination of 
antenna elements, weighting filters and combination circuit, which 
combination is optimal in respect of the signal distortion; such a 
receiving unit has an antenna pattern with a relatively low sensitivity in 
the direction of the jammer and a relatively high sensitivity in the 
direction of the communication transmission unit. For this reason, an 
iterative process is performed in the receiving unit, comprising the steps 
of: 
a signal measurement per weighting filter; 
the calculation of a weighting factor per weighting filter by the adjusting 
means according to an algorithm adapted to an optimal transmission 
characteristic; 
the generation of a beam-steering command in the form of adjustment signals 
by the adjusting means on the basis of the calculated weighting factors 
for the weighting filters. 
With such a communication receiving unit the signals are however measured 
on a radio-frequency level; this gives rise to problems with respect to 
the phase synchronisation between the various signals, the broadband 
performance of the adjusting means, the dynamics of the receiving part 
subjected to the signal measurement, and the associated hardware 
complexity of this part. 
SUMMARY OF THE INVENTION 
It is an object of the present invention to provide a solution to the above 
problems relating to the signal measurement on a radio frequency level. 
According to the invention, the communication receiving unit, as set forth 
in the opening paragraph, is so designed that the detector comprises first 
means for determining, for the adjusting means, a value (W.sub.1) 
proportional to the time average of the power concerning the signal (y) 
produced by the combination circuit and relating to at least a portion of 
the transmitter frequency band. The detector comprises second means for 
determining, for the adjusting means, a value (W.sub.2) proportional to 
the time average of the cross correlation concerning the output signal (y) 
and the replica (d) and relating to at least a portion of the transmitter 
frequency band. The adjusting means are designed for generating, upon the 
supply of adjusting signals to the weighting filters, modified adjusting 
signals on the basis of the time-averaged values (W.sub.1 and W.sub.2) 
resulting therefrom to adjust the weighting filters corresponding with an 
extreme value of a function f(W.sub.1,W.sub.2) containing the two values 
W.sub.1 and W.sub.2. 
Such a receiving unit stands up well against unknown environmental 
reflections, even with a large relative bandwidth. In view of any jamming 
of the communication between two stations, spread-spectrum modulation and 
demodulation techniques can be applied to advantage, as described in an 
article of R. C. Dixon, entitled "Spread-Spectrum Systems", Wiley 
Interscience, N.Y. etc., 1976. With a communication transmitting unit 
suitable for the application of spread-spectrum modulation on the 
information-conveying signals, the information about the spread-spectrum 
modulation used with the generation of the carrier should be available in 
a communication receiving unit adapted thereto. The means for generating a 
replica (d) of the carrier of the information-supplying signal received by 
the receiving unit --hereinafter called replica signal generator--should 
be provided with memory means for storing the above information about the 
spread-spectrum modulation required with the generation of the carrier of 
the signal received by the receiving unit. With coherent spread-spectrum 
modulation in a communication receiving unit, the carrier and clock 
frequencies for the spread-spectrum code do differ, but are from one and 
the same source. In a communication receiving unit adapted thereto the 
replica-signal generator is able to produce first the complete 
demodulation signal from the carrier frequency and the spread-spectrum 
code, whereafter a single demodulation of the signal from the combination 
circuit takes place. On the other hand, it is possible to carry out a 
stepped demodulation on the basis of individual signals,viz. a signal at 
the carrier frequency and a signal with the spread-spectrum code, such as 
depicted on page 213 of the cited book by R. C. Dixon. In both cases, a 
single feedback loop, based on phase detection, ensures that the carrier 
and code-clock frequencies are brought to the appropriate value. 
With non-coherent spread-spectrum modulation, i.e. with no fixed relation 
between the carrier and the clock frequencies, and under non-severe 
jamming conditions, it is possible to perform the demodulation process as 
follows: The total demodulation signal is generated with the aid of the 
output signal of a frequency generator and that of a spread-spectrum code 
generator. Thereafter, separate feedback loops keep the frequency 
generator and the spread-spectrum code frequency lined up. However, under 
severe jamming conditions it is better to perform the demodulation process 
of the received signal in two steps, that is to line up the 
spread-spectrum code frequency and the carrier frequency in succession. 
With such a communication receiving unit the replica generator is provided 
with a code generator controlled by the memory means and with a frequency 
generator to be tuned to the carrier frequency, while the demodulation 
means is provided with a first demodulator for demodulating the signal (y) 
from the combination circuit by means of the code generator output signal, 
and with a second demodulator connected to the first demodulator for 
demodulating the output signal of the first demodulator by means of the 
frequency generator output signal. 
It is possible to start the adaptation of the antenna diagram during the 
line-up phase of the spread-spectrum code frequency, namely by feeding the 
output signal of the first demodulator to a correlation detector to 
produce an output voltage, depending on the measure of correlation between 
the signals applied to the first demodulator. This output voltage can then 
be supplied to the adjusting means in substitution for the time-average 
value (W.sub.2). 
The invention will now be explained with reference to the accompanying 
drawing figures.

DESCRIPTION OF THE PREFERRED EMBODIMENT 
The communication receiving unit in FIG. 1 consists of a plurality of 
antenna elements 1A-N, a network 2 of adaptive weighting filter circuits 
3A-N, connected to the antenna elements, a combination circuit 4 fed by 
network 2, and a detector 5 connected to combination circuit 4. Antenna 
elements 1A-N may be of any type, and must be positioned at small 
distances from each other, proportionately to the wavelength used, where 
the geometry need not be considered. The weighting filter circuits 3A-N 
must be of the wide-band type. In the embodiment in question each 
weighting filter circuit 3i, where i=A, . . . , N, consists of a circuit 
of series-connected delay lines 6.sub.i,a -6.sub.i,k, while between each 
two successive delay lines and at both sides of this circuit, tappings 
with weighting function elements 7.sub.i,a -7.sub.i,k+1 are incorporated. 
The output signals of all weighting function elements 7.sub.i,j, where 
i=A, . . . , N and j=a, . . . , k+1, are combined in combination circuit 
4. Each weighting function element 7.sub.i,j has a weighting factor 
W(m).sub.i,j, which is set each sample interval m by adjusting means 8. 
With the described combination of antenna elements 1A-N, network 2 and 
adjusting means 8, referred to as an adaptive array in the literature, a 
communication receiving unit requires a transfer characteristic, which is 
optimal in respect of the phase distortion of the detected 
information-carrying signals in view of the dispersiveness of the 
environment, as well as the signal/interference ratio in the receiving 
unit. 
The signal components present at a certain time t on the tappings to the 
weighting function elements 7.sub.A,a, . . . , 7.sub.N,k+1 are designated 
successively by x(t).sub.A,a, . . . , x(t).sub.N,k+1 by which the input 
signal vector X(t).sup.T =[x(t).sub.A,a, . . . , x(t).sub.N,k+1 ].sup.T of 
the weighting function elements is defined by X(t).sup.T, the transpose of 
vector X(t). By a corresponding notation the available weighting factors 
w(m).sub.A,a, . . . , w(m).sub.N,k+1 will define the weighting function 
vector W(m) of network 2 for the weighting function elements 7.sub.A,a, . 
. . , 7.sub.N,k+1, namely in accordance with the definition W(m).sup.T 
=[w(m).sub.A,A, . . . , w(m).sub.N,k+1 ].sup.T, where W(m).sup.T is the 
transpose of the vector W(m). The output signal y.sub.l of combination 
circuit 4 can therefore be represented by the relationship 
y(t)=X(t).sup.T.W(m)=W(m).sup.T.X(t). 
In the embodiment of the receiving unit in question and in view of any 
jamming activities which may be expected with a non-predictable 
dispersiveness of the environment, use is made of the fact that, in 
adapting the antenna pattern and hence in adjusting the weighting factors, 
a characteristic in the signal carrier is provided by the transmitting 
unit, while a priori knowledge about this characteristic is available in 
the receiving unit. In the embodiment in question, the characteristic in 
the carrier relates to the type of modulation used, in this case the 
spread-spectrum type, whereby a greater than normal bandwidth with the 
communication is used to obtain a more favourable transfer characteristic. 
Several types of spread-spectrum modulation are known, such as 
direct-sequence modulation (also called pseudonoise modulation), 
frequency-hopping modulation, chirp modulation, time-hopping modulation, 
as well as various combined types derived therefrom. 
In the case of biphase modulation, a transmitter signal s(t) can be 
represented by s(t)=p.sub.i (t).p.sub.s (t).sin.omega.t, where p.sub.i (t) 
is the biphase coded signal comprising the information, p.sub.s (t) the 
signal produced in accordance with the modulation type, and sin.omega.t 
the carrier signal. It is customary to keep the modulation frequency of 
the p.sub.s (t) signal below the carrier frequency, but much higher than 
the data rate of the p.sub.i (t) signal comprising the information. 
The antenna elements 1A-N receive the s(t) transmitter signal jointly with 
the n(t) noise and j(t) interference signals, resulting in the y(t) signal 
at the output of combination circuit 4. On the ground of the preset 
information about the applied type of modulation and the carrier 
frequency, the y(t) signal is then demodulated. For this purpose the 
detector 5 is provided with means 9 for demodulating the y(t) signal of 
combination circuit 4 to the frequency band (i.e. the information band) 
associated with the information. The demodulation is effected by a signal 
d(t)=p.sub.s (t+.phi..sub.1).sin(.omega.t+.phi..sub.2), where p.sub.s 
(t+.phi..sub.1) and sin(.omega.t+.phi..sub.2) represent replicas of the 
modulation signals generated in the transmitting unit. It is also possible 
to carry out the demodulation in steps, for example first a demodulation 
with the p.sub.s (t+.phi..sub.1) signal and then the resulting 
y(t)*p.sub.s (t+.phi..sub.1) signal with the sin(.omega.t+.phi..sub.2) 
signal. Detector 5 comprises a replica signal generator 10 for generating 
the d(t) demodulation signal or instead thereof the individual p.sub.s 
(t+.phi..sub.1) and sin(.omega.t+.phi..sub.2) signals. The adaptation of 
the antenna pattern relative to the location of the receiving unit, and 
hence the determination of the weighting function vector W(m), is said to 
be ideal only if the y(t) signal to be supplied approaches the d(t) 
demodulation signal (or the composite signal of the individual 
demodulation signals) as well as possible, thus minimising the passage of 
noise and interference signals. The antenna arrangement should therefore 
be so adapted that the error signal .epsilon.(l)=d(l)-y(l) generated is 
minimised in absolute value and regarded over a certain time interval k 
with measuring intervals l=1, . . . , M, so that 
##EQU1## 
assumes a minimum value. From an ergodic ensemble of individual input 
signals X(l) and the desired d(l) modulation signal with the same set of 
weighting factors W(k) it follows that the expected value or the ensemble 
least mean square error: 
##EQU2## 
Since the average values of the above terms are determined on the basis of 
time averaging on sampling values during time interval k, the above 
relationship can be expressed as: 
##EQU3## 
In this expression, G[d.sup.2 (t)] is the time-averaged value of the 
auto-correlation term relative to the replica d(t), equalling the average 
power P.sub.ref relative to d(t), during time interval k. G[y.sup.2 (t)] 
is the time-averaged value of the auto-correlation term relative to the 
output signal y(l) of the combination circuit 4, equalling the output 
power of circuit 4, hence also denoted by S+J+N, representing the average 
value of the S, J and N signal components produced by, respectively, the 
communication, jamming and noise signals at the output of combination 
circuit 4 during time interval k. Further, G[d(t).y(t)] represents the 
time-averaged value of the cross-correlation term relative to replica d(l) 
and output signal y(l) during time interval k and may therefore be 
equalled to YD.sqroot.P.sub.ref. 
In adapting the antenna pattern, it is also possible to use, instead of the 
above algorithm based on the time average of the least mean square error, 
the algorithm, based on the maximization of the signal/noise (S/N) ratio 
and, hence, of the quotient 
##EQU4## 
of the average peak signal power (YD).sup.2.P.sub.ref obtained by 
modulation and the average signal power (S+J+N) from combination circuit 
4. 
In adapting the antenna pattern, also algorithms of a slightly deviating 
form can be used, such as the algorithms: 
##EQU5## 
where the values of .alpha. and .beta. may deviate slightly from the 
values 2 and 1, respectively. This provides an adjustment in accordance 
with an extreme value of a function f(W.sub.1,W.sub.2) obtained through 
partial summation of the two values W.sub.1 and W.sub.2 and of a function 
f(W.sub.1,W.sub.2) containing the quotient of the two values W.sub.1 and 
W.sub.2, respectively. 
A minimum value of f.sub.lmse or a maximum value of f.sub.msnr however 
requires a maximum value of the cross-correlation term 
YD.sqroot.P.sub.ref, but this is possible only if the demodulation signal 
d(l) is optimally aligned with the modulation component in the output 
signal y(l). The cross-correlation term YD.sqroot.P.sub.ref is obtained 
through means 9 for demodulating signal y(l) from the combination circuit 
4 in accordance with replica signal d(l). 
In the embodiment in question, showing a good operability under severe 
jamming conditions, the demodulation is performed in two steps; for this 
purpose, means 9 comprises a first demodulator 11 and a second demodulator 
12. A bandpass filter 13 is inserted between combination circuit 4 and 
first demodulator 11. With a centre frequency equal to the carrier 
frequency, filter 13 has a bandwidth approximately equal to twice the bit 
rate of the spread spectrum code. 
The demodulation signal p.sub.s (t+.phi..sub.1) produced by replica signal 
generator 10 for the first demodulator 11 is generated in a code generator 
14 with the aid of associated memory means containing the information 
about the type of modulation applied with the communication. 
As to the phase angles .phi..sub.1 and .phi..sub.2, it should be noted that 
with a communication over a distance of several kilometers and with a 
synchronous generation of the spread spectrum code in the MHz-field by 
both the transmitting and the receiving units, the spread spectrum signal 
p.sub.s (t+.phi..sub.1) of code generator 14, when supplied to the first 
demodulator 11, leads the corresponding signal component y(t) in the 
output signal of the combination circuit 4 by several bits. Code generator 
14 would therefore have to slow down for a short period, or the generation 
of the signals would have to be delayed over a certain phase .phi..sub.1 
and .phi..sub.2, respectively, to achieve a synchronisation and a phase 
alignment between the spread spectrum code p.sub.s (t+.phi..sub.1) of code 
generator 14 and the corresponding signal component in the output signal 
y(t) of bandpass filter 13. The timing of code generator 14 containing 
memory means still forms an uncertain factor in the demodulation process. 
For this reason, the replica signal generator 10 comprises a first control 
circuit 15, of which a feasible embodiment will be described hereinafter. 
Through a second bandpass filter 16 the output signal y(t)*p.sub.s 
(t+.phi..sub.1) of the first demodulator 11 is supplied to the second 
demodulator 12. With a centre frequency equal to the carrier frequency, 
second bandpass filter 16 has a bandwidth of twice the value of the 
information bit rate. 
The second bandpass filter 16 is followed by the second demodulator 12 for 
further demodulation of the output signal of filter 16 with the aid of a 
frequency generator 17 tuned to the carrier frequency .omega.; frequency 
generator 17 forms part of the replica signal generator 10. 
Also the signal sin(.omega.t+.phi..sub.2) supplied by the frequency 
generator 17 should be phase-synchronised with the corresponding component 
in the signal obtained from the second bandpass filter 16. To this effect 
the replica signal generator 10 contains a second control circuit 18, 
which will be described in more detail herinafter. 
In this embodiment an amplitude detector 20 is connected to the second 
demodulator 12 via a lowpass filter 19 having a limit frequency equal to 
the maximum frequency of the information band; the amplitude detector 20 
produces a pure biphase signal from the supplied signal in so far as the 
latter signal is in any way distorted. Using a decoder 21 connected to 
amplitude detector 20, the information signal is converted into a form 
suitable for observation or recording. 
FIG. 2 shows a feasible embodiment of the first control circuit 15 in 
detail. As to the phase synchronisation of the spread spectrum signal 
p.sub.s (t+.phi..sub.1) with the corresponding component in the output 
signal of the bandpass filter 13, there are two essential steps. 
The first step concerns the time delay of code generator 14, such that a 
synchronous condition is obtained between the output signal p.sub.s 
(t+.phi..sub.1) of generator 14 and the corresponding component of the 
signal supplied by the first bandpass filter 13. The next step concerns 
the phase alignment of the output signal p.sub.s (t+.phi..sub.1) of code 
generator 14 and the above component in the signal from bandpass filter 
13. Through taking these two steps it is achieved that the data signals 
spread over the entire spectrum range are transformed into a narrow 
frequency range about the carrier frequency, while the jamming signals 
usually active in a narrow frequency range about the carrier frequency are 
spread through transformation over the entire spread spectrum range and 
subsequently filtered away to a significant extent by means of the narrow 
bandpass filter 16. 
To carry out the first step, the first control circuit 15 comprises a 
correlation detector 22 connected to the bond pass filter 16 for 
generating a synchronisation signal for code generator 14, containing the 
memory means, at the instant a synchronisation between the signals 
concerned is detected. Other means should then be used to obtain a more 
precise phase alignment between the signals concerned. The correlation 
detector 22 thereto comprises a continuous circuit of an asynchronous 
detector 23, a low-pass filter or integrator 24, and a synchyronisation 
detector 25. Correlation detector 22 also contains a clock generator 26, 
initiated simultaneously with the clock generator of the communication 
transmitting unit and supplying its timing signals to code generator 14 
via switching means 27. It is necessary that the frequency of clock 
generator 26 be set to a slightly lower value, as compared with that for 
the communication transmitting unit, or have the clock generator 26 
periodically perform a small negative frequency jump of short duration, 
while the frequency is otherwise kept the same. The result of the two 
methods is that the time lead of the spread spectrum signal with respect 
to the corresponding component in the output signal of the first bandpass 
filter 13 is eliminated slowly but surely. 
The asynchronous detector 23, which may function as a feedback loop or 
envelope detector (e.g. a rectifying circuit), produces an output signal 
which, after filtering by the lowpass filter or the integration in the 
integrator 24, is a function of the signal level of the supplied input 
signal representing the correlation function. This signal level is 
dependent on the extent to which the correlation between the codes in the 
two supplied signals has been effected. Reaching a predetermined level of 
the output signal of the lowpass filter or integrator 24 therefore implies 
that a synchronisation between the two codes has given a satisfactory 
result. The attainment of this signal level with the correlation between 
the codes is established by the synchronisation detector 25, which 
subsequently sets the switching means 27 to the second position. 
The setting of switching means 27 to the second positipn implements the 
other step, concerning the phase alignment of the output signal of code 
generator 14 with the corresponding component in the output signal of the 
bandpass filter 13. To this effect, the first control circuit 15 comprises 
a feedback loop 28 supplied with the output signal of the lowpass filter 
19 modulated on the carrier. Feedback loop 28 produces a timing signal, 
fed to code generator 14 via switching means 27. In a feasible embodiment, 
feedback loop 28 comprises a bandpass filter 29 and connected thereto, a 
demodulator 30; the demodulator 30 is also fed with the output signal of a 
fixed l.f. square wave generator 31. The signal supplied by demodulator 30 
is fed to an integrator 32 to produce a d.c. signal having the function of 
control signal for a voltage-controlled oscillator 33 (VCO). The output 
signal of VCO 33 and that of the l.f. square-wave generator 31 are jointly 
supplied to an increment phase modulator 34, suitable to shift the phase 
of the other signal to be supplied, viz. the VCO signal, alternately 
forward and backward over a small value in the rhythm of the square wave 
generated. In consequence of this, there will be a change in the extent of 
correlation between the output signal of code generator 14 and the 
corresponding component in the output signal obtained from the first 
bandpass filter 13, causing a shift in the amplitude of the input signal 
of the second demodulator 12. Through the forward and backward shifting of 
the clock phase the signal is amplitude modulated in accordance with the 
phase shifting rhythm particularly with a non-ideal phase alignment, 
causing a change in the d.c. voltage for VCO 33. Only in case of a certain 
setting of the integrator, the VCO 33 will generate a signal with such a 
code frequency that modulation thereon with the square wave does not 
result in an amplitude-modulated signal for filter 29. In such a case, VCO 
33 remains at the set value. Also other embodiments of a second control 
circuit are known, such as from R. C. Dixon, "Spread-Spectrum Systems", 
Wiley Interscience, New York, etc. 1976, pp. 210-212. 
Furthermore, corect working of such a communication receiving unit requires 
a third step for the purpose of the inphase alignment of the frequency of 
generator 17 with the carrier frequency in the output signal of the 
bandpass filter 16. This is realised by the second control circuit 18, of 
which an embodiment of a socalled I.Q. phase-locked loop or "Costas loop" 
is known from the citedbook of R. C. Dixon, pp. 155-158. Frequency 
generator 17, functioning as a voltage controlled oscillator (VCO), 
supplies the demodulation signal to second demodulator 12 and, using a 
phase shifting element 35 with .DELTA..phi.=90.degree., a quadrature 
demodulation signal to a third demodulator 36. The second and the third 
demodulators also receive the filtered output signal of the first 
demodulator 11 to generate, in response, output signals to be supplied to 
a fourth demodulator 38 after filtering in the bandpass filters 19 and 37. 
After filtering in a narrow-bandpass filter 39 with the centre frequency 
equalling the carrier frequency, the output signal of demodulator 38 is 
suitable to function as control signal for VCO 17. 
Embodiments of a spread spectrum communication system are known, where the 
communication transmitting unit uses a single timing unit to control both 
the carrier frequency generator and the code generator, and where there is 
a simple relationship between the carrier frequency and the spread 
spectrum frequency. Hence, there is constantly a coherent relationship 
between the carrier frequency and the spread spectrum frequency. Such an 
embodiment of a communication receiving unit is, for example, described in 
the cited book of R. C. Dixon, pp. 212-214, where a timing unit controls 
both the carrier frequency generator and the code generator. The signal 
supplied by combination unit 4 and then filtered is first demodulated with 
the carrier and then correlated with the spread spectrum signal; the 
resulting signal is subsequently examined for phase deviation in a phase 
detector. After filtering in an l.f. bandpass filter, the signal produced 
by the phase detector is suitable as control signal for the above timing 
unit designed as voltage-controlled oscillator. 
The term S+J+N=G[y.sup.2 (t)] required for the adaptation of the antenna 
pattern refers to the power, as present at the output of combination 
circuit 4. This power can be determined by squaring and subsequent 
averaging, i.e. l.f. filtering of the output signal of combination circuit 
4. As illustrated in FIG. 1, the detector 5 thereto comprises first means 
connected to the first bandpass filter 13, for example a squaring circuit 
40 and, connected thereto, a lowpass filter 41 supplying, conditional upon 
the applied filter characteristic, a value W.sub.1 for adjusting means 8, 
where W.sub.1 is proportional to the time average of the power of the 
signal produced by combination circuit 4 and related to at least a portion 
of the transmitter frequency band. In this connection it is conceivable to 
limit the measurement in its totality to a certain number of frequency 
components. The cross-correlation term YD.sqroot.P.sub.ref represents the 
time average of the cross-correlatin product between the r.f. signal y(t) 
supplied by combination circuit 4 and the singal supplied in a split form 
as r.f. demodulation signal to first and second demodulators 11 and 12. 
The information component is here however considered of being constant in 
amplitude, as provided for in this case, namely by measuring the value of 
YD.sqroot.P.sub.ref only during the time intervals devoid of information, 
the socalled synchronisation blocks. During the synchronisation blocks, 
inserted periodically during the communication, the information component 
P.sub.i (t) is constant, for example 1. It is therefore preferable to 
adapt the antenna pattern continuously during the synchronisation blocks, 
were it not that in general the communication occurs under severe jamming 
conditions and with an unfavourable setting of the communication receiving 
unit, viz. the condition when both the code generator 14 and the carrier 
frequency generator 17 are out of phase alignment. Under such unfavourable 
conditions the cros-correlation term hardly changes in value by varying 
the antenna pattern. This is the reason that in the first instance only 
the code generator 14 is aligned, while for the adaptation of the antenna 
pattern not the cross-correlation term YD.sqroot.P.sub.ref, but 
YD'.sqroot.P.sub.ref is used, where D' refers to the signal produced by 
code generator 14. Here the fact that the cross-correlation term 
YD'.sqroot.P.sub.ref still contains an undesired r.F. component is 
disregarded. This disadvantage is however obviated by selecting the time 
average of the l.f. component of the signal voltage associated with this 
correlation term; this is achieved by using an asynchronous detector (such 
as a phase-locked loop, an envelope detector, etc.) connected to second 
bandpass filter 16 and a subsequent lowpass filter. Using a correlation 
detector 22, as illustrated in FIG. 2, the signal value 
YD'.sqroot.P.sub.ref is obtainable as output voltage of the lowpass filter 
24. Using this output signal, the adjusting signals are determined in the 
adjusting means 8, as part of the initial adaption of the atenna pattern. 
In consequence of this adaption, the cross-correlation term will increase 
progressively during the alignment of code generator 14. Assuming that 
meanwhile the alignment of the frequencyu generator 17 is well under way, 
it may be supposed that when a certain threshold value has been reached by 
the cross correlation term, a usable value of the cross correlation term 
YD.sqroot.P.sub.ref is already available to adapt the antenna pattern by 
using the term YD.sqroot.P.sub.ref instead of YD'.sqroot.P.sub.ref for the 
processing unit. The voltage measurement of YD.sqroot.P.sub.ref should 
however be taken during the periodical synchronisation blocks. Each of 
these blocks are introduced by a very specific code series, On recognising 
this code series, the decoder 21 produces a control signal for a preset 
time, when the then supplied invormation signal P.sub.i (t) assumes a 
constant value, for example "1". This control signal opens a gate swithc 
42, which receives the information signal of the lowpass filter 19. During 
this preset time the gate switch 42 passes the YD.sqroot.P.sub.ref signal 
for an amplitude measuring circuit 43 with a narrow-band l.f. filter 
process, resulting in the measuring value W.sub.2 for adjusting means 8. 
The ensemble average E[.epsilon..sup.2 (l)] to be examined by adjusting 
means 8 is a quadratic semi-definite positive function, which may be 
represented by a concave hyperbolic paraboloidal surface. The adaptation 
of the weighting factores therefore implies the descent along the surface 
to reach the lowest point of the function. To this effect, the gradient 
methods are generally applied; a usable method is known from the article 
of R. Fletcher and C. M. Reeves, "Function minimization by conjugate 
gradients", Electronic Computing Laboratory, The University, Leeds 2; 
Computer Journal, Vol. 7, 1964, pp. 149-153. In this method an iterative 
process is performed a long the conjugate search direction Z.sub.k+1 to 
search for the minimum value of E[.epsilon..sup.2 (l)]. The search 
direction is determined using the formula 
##EQU6## 
and .beta..sub.0 =0 for the first search direction. According to this 
iterative process, the system converges rapidly to the unambiguous minimum 
of E[.epsilon..sup.2 (l)].