Control circuit and method for direct current controlled attenuator

A bridge T attenuator which includes PIN diodes in both the bridge arm and in the shunt arm, and in which the current through the shunt arm is controlled with respect to the current through the bridge arm, so that the product of these two currents equals a constant. This is accomplished by a feedback loop. Thus a relatively simple, inexpensive, and entirely analog solution is provided for a bridge T attenuator to be used with a radio frequency signal, so that a prescribed level of attenuation set by a user may be maintained while the impedance is maintained at a fixed value over the entire attenuation range.

BACKGROUND OF THE INVENTION 
1. Field of the Invention 
This invention relates to a control circuit for a variable bridge T 
attenuator. 
2. Description of the Prior Art 
A bridge T fixed value attenuator of the type shown in FIG. 1A is well 
known. An input signal is applied to the Port 1 terminal; an output signal 
is provided at the Port 2 terminal. The attenuator of FIG. 1A has a main 
arm which includes two resistors R1 and R2. The node between resistors R1 
and R2 is connected to ground via shunt resistor R3. The upper (bridge) 
arm includes resistor R4. To achieve a prescribed level of attenuation 
while maintaining a predetermined input and output impedance R0, resistors 
R1 and R2 are fixed at a value of R0 while the values of resistors R3 and 
R4 are varied so that R3*R4=R.sub.0.sup.2. 
One method for controlling a prior art bridge T attenuator is to use a dual 
potentiometer connected as in FIG. 1B where resistances R3 and R4 are each 
potentiometers. However this use of potentiometers has been found to be 
inferior, because it does not lend itself to high frequencies. Replacing 
resistors R3 and R4 with pin diodes creates an electrically variable 
attenuator which has superior frequency response but now needs an 
electrical drive provided by a simple potentiometer to maintain the 
equality R3*R4=R.sub.0.sup.2. 
A second approach is to use a combination of a microprocessor (or 
microcontroller) and a lookup table stored in a memory to control the 
bridge T attenuator. For various degrees of attenuation, different 
resistance values are stored in the lookup table. When the user selects a 
particular level of attenuation, the microprocessor accesses the lookup 
table, determines the corresponding amount of resistance, and then adjusts 
it accordingly. Such digital control of attenuation is effective but 
requires relatively expensive components, i.e. at least a microcontroller 
and a memory. 
Thus there is not available a bridge T attenuator control circuit or method 
which is satisfactory in terms of functionality by providing a prescribed 
level of attenuation for all values of attenuation while maintaining all 
terminal impedances constant and equal to the characteristic impedance of 
the system and which is also relatively inexpensive and simple. 
SUMMARY 
An electronic circuit controls a variable bridge T attenuator such that the 
terminal impedances remain constant, and equal to the characteristic 
impedance of the system, for all values of attenuation. This control 
circuit causes the resistance of the PIN diodes in the attenuator to 
follow the ideal relationship between the element values with no 
approximation such that (for FIG. 1A) R3.times.R4=R.sub.0.sup.2. Feedback 
in the control circuit forces the product of the currents through the PIN 
diodes to be constant and equal to the square of a reference current. The 
product of the currents is derived by the summation of voltages across the 
PIN diodes of the attenuator. 
The control circuit includes an independent current source, a controlled 
current source and PIN diodes. The independent current source establishes 
the attenuation. The controlled current source is driven by an integrator 
which in turn is driven by the error between the sum of the attenuator PIN 
diode voltages and a fixed reference. Thus a circuit containing a 
multiplier in the feedback loop of an operational amplifier maintains the 
product of currents through the PIN diodes to be constant and equal to the 
square of a reference current. 
Thus a bridge T attenuator in accordance with the invention, instead of 
using a resistor in the bridge arm, uses a diode. Similarly the shunt 
resistor is replaced by a second diode. For control purposes, a master 
(independent) current source provides current to drive the bridge arm and 
the main arm, while a slave (controlled) current source provides current 
to the shunt diode. The product of the currents from the master current 
source and the slave current source are kept constant in order to have 
good return loss. In one embodiment, the product of the two currents is 
kept constant by measuring the master current and adjusting the slave 
current by means of a divider circuit containing a multiplier in a 
feedback loop of an operational amplifier. In another embodiment, instead 
of using a multiplier, one adds a logarithm of the level of the current 
from the master current source to the logarithm level of the current from 
the slave current source, and by feedback maintains the sum of the two 
logarithms at a constant value. 
The diodes are e.g. PIN diodes. This is a type of diode having a resistance 
inversely proportional to the current flowing through the diode. Thus the 
resistance of the bridge arm and the shunt arm are variable in accordance 
with the current supplied thereto. The level of attenuation is user 
controlled by varying the level of the current provided by the master 
current source, for instance by use of a potentiometer. In one embodiment 
the feedback loop is temperature compensated by inclusion of an additional 
PIN diode providing the reference voltage.

DETAILED DESCRIPTION 
The electrical rules governing the operation of bridge T attenuators are 
well known. The following describes these with respect to the prior art 
bridge T attenuator of FIG. 1A and are applicable to the bridge T 
attenuator in accordance with the present invention as described below. If 
K is the ratio between the input and output current and Z is a 
characteristic impedance, then the formulas describing the bridge T 
attenuator are as follows (where the reference number for each component 
in these equations represents the component value): 
EQU R1=R2=Z 
EQU R4=Z(K-1) 
EQU R3=Z/(K-1) 
EQU R3*R4=Z.sup.2 
EQU R4/R3=(K-1).sup.2 
It is known that it is possible to substitute a diode (for instance a PIN 
diode) for certain of the resistors (i.e. resistors R4 and R3) of FIG. 1A. 
As is well known, a PIN diode is a component having a resistance 
proportional to the inverse of the current passing through the diode. PIN 
diodes are commercially available. 
A PIN diode includes a P-N junction with a doping profile tailored so that 
an intrinsic layer, the "I region", is sandwiched between a P (positively 
doped) layer and an N (negatively doped) layer. While certain commercially 
available diodes are designated PIN diodes, the use of this term herein is 
not intended to be limited to such commercially available products, but is 
intended to include any component having the desired current-resistance 
characteristics and also includes a multi-component circuit having more 
than one component which provides the desired characteristics. 
Also, as is well known, a PIN diode at radio frequencies acts as a current 
controlled resistor and a PIN diode will propagate signals over a wide 
range of frequencies. 
Thus as shown in the left hand portion of FIG. 2, one substitutes for the 
bridge arm resistor R4 in FIG. 1A a first PIN diode D4, and similarly 
substitutes for the shunt resistor R3 in FIG. 1A a second PIN diode D3. 
These diodes are then biased by respectively currents I.sub.1 and I.sub.2 
which are depicted in FIG. 2 as being provided by conventional current 
sources also labelled I.sub.1 and I.sub.2. Then applying the above 
equations relating to a bridge T attenuator to the circuit of FIG. 2, 
R.sub.D3 (the resistance through diode D3)=a/I.sub.2 and R.sub.D4 (the 
resistance through diode D4)=a/I.sub.1 , wherein a is the proportionality 
constant of the PIN diode. 
Then by algebra, R.sub.D4 *R.sub.D3 =a.sup.2 /(I.sub.1 *I.sub.2)=Z.sup.2 
and hence I.sub.1 *I.sub.2 =a.sup.2 /Z.sup.2 = constant. 
Therefore in order to have a good return loss, i.e. a properly functioning 
bridge T attenuator, the product of currents I.sub.1 and I.sub.2 is 
maintained constant in accordance with the present invention. 
This is performed in the circuit of FIG. 2 by a control circuit shown in 
the right hand portion of FIG. 2 including a feedback control element 10 
which is connected to the output terminals of current sources I.sub.1 and 
I.sub.2 and which maintains the product of current I.sub.1 (the master) 
with the current I.sub.2 (the slave) as a constant by controlling (in this 
case) the level of current I.sub.2 by means of a feedback control line 12. 
One implementation of feedback control circuit 10 includes measuring the 
level of current I.sub.1, and by means of a divider circuit containing a 
multiplier, maintaining the level of current I.sub.2. The load Ro is not 
shown in FIG. 2, but is connected to Port2. 
An alternative feedback control circuit 10, instead of using a multiplier 
circuit, takes the logarithms of each of currents I.sub.1 and L.sub.2, (or 
proportionate voltages V1, V2) adds them, and provides a feedback to 
current source I.sub.2 to maintain the sum of the two logarithms at a 
constant value. 
The other element shown in FIG. 2 is an inductor Z.sub.1 which while not 
necessary has the advantage of providing a variable impedance, i.e. a high 
impedance at radio frequencies and a low impedance at low frequencies or 
direct current, for the purpose of biasing the diode. 
The circuit of FIG. 2 operates by the user controlling (by means such as by 
a manual potentiometer adjustment) the current level provided by master 
current source I.sub.1. Thus automatically the current level of slave 
current source I.sub.2 is compensated to maintain the product of the two 
current levels as a constant, hence self-adjusting the bridge T 
attenuator. Advantageously the circuit of FIG. 2 can be implemented using 
all analog components (including for instance operational amplifiers) thus 
providing a reliable and inexpensive analog solution to control of a 
bridge T attenuator. Thus this control circuit can be easily integrated 
into another analog circuit without any need for digital-type components 
such as memory or a microprocessor. 
The circuit of FIG. 2 is intended to illustrate the functionality of a 
control circuit in accordance with the present invention. Many different 
circuits in terms of actual components may be implemented which are 
equivalent to and perform the functions of FIG. 2, and all such circuits 
are intended to fall within the scope of the present invention. 
An example of one particular embodiment of a circuit of FIG. 2 is depicted 
in FIG. 3. Similar components in FIG. 3 have the same reference numbers as 
in FIG. 2. In FIG. 3 it can be seen that instead of just the single diode 
D4 in the upper arm, a second diode D2 is connected in series with diode 
D4. Use of two pin diodes D2 and D4 in the upper bridge arm of the bridge 
T attenuator advantageously serves to cancel distortion. 
The master current source in FIG. 3 (corresponding to current source 
I.sub.1 in FIG. 2), includes a voltage source V+ which is series-connected 
to a variable resistor (potentiometer) R6 and a bias fixed resistor R7. 
Adjustment of potentiometer R6 allows DC variance of the attenuation of 
this bridge T attenuator. Other components include capacitors C1, C2, and 
C3 for filtering. The values of components R6, R7, C1, C2, and C3 are not 
critical; the value of component R16 is equal to that of component R7. 
The second current source in FIG. 3 (corresponding to current source 
I.sub.2 in FIG. 2) is the transistor Q1 and associated resistors R8, R9, 
and R10. These in turn are driven by a feedback circuit (corresponding to 
element 10 of FIG. 2) which includes two operational amplifiers 14 and 18. 
In this embodiment, the first operational amplifier 14 is a summer and the 
second operational amplifier 18 is an integrator. These may be operational 
amplifiers of the type commercially available, or equivalent elements 
performing respectively the functions of summing and integrating. 
Resistors R11 and R12 are matched and each connected to the summing 
(inverting) input terminal of operational amplifier 14. The second 
(non-inverting) input terminal of operational amplifier 14 is connected to 
a voltage reference source V.sub.REF which as shown is provided by a 
resistor R15 connected to a voltage supply V+ and the other end of 
resistor R15 connected via a PIN diode D5 to ground. Diode D5 provides the 
desired temperature compensation for the reference voltage. Resistor R14 
connects the output terminal of operational amplifier 14 to its inverting 
terminal, thus providing the summing function. 
The output terminal of operational amplifier 14 is connected by a resistor 
R13 to the inverting input terminal of the second operational amplifier 18 
which here functions as an integrator, due to capacitor C5 being connected 
between its output terminal and its inverting input terminal. The 
noninverting input terminal of operational amplifier 18 is connected to 
the reference voltage V.sub.REF. 
The output terminal of operational amplifier 18 (which is the integrator) 
is connected via feedback line 12 (also shown in FIG. 2) via resistor R10 
to the base (control) terminal of transistor Q1 which is in the second 
current source. Hence, the current I.sub.2 sourced by its current source 
including transistor Q1 and resistor R8 is controlled by a level of the 
voltage on the feedback line 12. 
In this embodiment preferably each of pin diodes D2, D3, D4 and D5 are 
matched, i.e. have similar voltage/current characteristics. The value of 
inductor Z.sub.1 varies with the desired frequency range. The attenuator 
of FIG. 3 has been found to provide satisfactory attenuation over a range 
of at least two dB to 25 dB. 
The bridge T attenuator of FIG. 3 therefore is controlled by the addition 
of two logarithms, each logarithm representing a value of respectively 
currents I.sub.1 and I.sub.2. Thus by adding logarithms, one multiplies 
the two values of which the logarithms have been taken and hence obviates 
the need for an actual multiplier circuit per se. A theoretical 
explanation of this in terms of voltages and currents follows; however, 
understanding of this is not necessary for an appreciation of the 
operation of the present attenuator. 
Let V.sub.1 be the voltage drop across the combination of diodes D2 and D4 
and let V.sub.2 be the voltage drop across the shunt arm diode D3. Let 
I.sub.0 be the reverse bias current of each of diodes D2, D3, and D4. Then 
: 
V.sub.1 =(KT/e)*ln(I.sub.1 /I.sub.0) and 
V.sub.2 =(KT/e)*In(I.sub.2 /I.sub.0). 
K is the Boltzmann constant, T is temperature and e is electron charge. 
Therefore V.sub.1 +V.sub.2 =(KT/e)* ln(I.sub.1 *I.sub.2 /I.sub.0.sup.2)=a 
constant. The circuit of FIG. 3 makes the sum of V.sub.1 and V.sub.2 a 
constant and therefore makes the product of I.sub.1 *I.sub.2 a constant 
using the feedback circuit. Thus in the circuit of FIG. 3 the value of 
I.sub.1 is adjusted as a function of the required attenuation by tuning 
potentiometer R6, while the value of current I.sub.2 is adjusted by the 
feedback loop. 
Therefore 
V.sub.OUT =3*V.sub.REF -V.sub.1 -V.sub.2 and 
V.sub.OUT =V.sub.REF at the integrator input terminal (the noninverting 
terminal of operational amplifier 18) which gives 
EQU V.sub.1 +V.sub.2 =2*V.sub.REF. 
Then replacing the corresponding currents for V.sub.1 and V.sub.2, this 
equation becomes: 
EQU 2*(KT/e) ln (I.sub.REF /I.sub.0)=(KT/e) ln(I.sub.1 /I.sub.0)+(KT/e) 
ln(I.sub.2 /I.sub.0), 
which gives 
EQU I.sub.1 *I.sub.2 =I.sub.REF.sup.2. 
In accordance with a second embodiment of the invention as described above, 
instead of using a sum of the logarithms in the feedback circuit, a 
conceptually simpler (but in circuitry slightly more complex) solution as 
depicted in FIG. 4 is used for the feedback loop. In this case instead of 
using a summer and integrator, an e.g. commercially available multiplier 
circuit 24 receives voltage V.sub.1 and V.sub.2 and multiplies them 
together. This product, by means of comparator 26 which has its inverting 
terminal connected to the output terminal of multiplier 24 and its 
noninverting terminal connected to reference voltage V.sub.REF, provides 
the feedback signal on feedback line 12. Thus the feedback circuit of FIG. 
4 may be substituted in FIG. 3 for the elements 14 and 18 and the 
associated fixed components. 
While in the presently illustrated embodiments the master current source is 
connected to the main arm and bridge arm of the attenuator and the slave 
current source to the shunt arm, these may be reversed. 
This disclosure is illustrative and not limiting; further modifications 
will be apparent to one skilled in the art in light of this disclosure and 
are intended to fall within the scope of the appended claims.