+-------+----------------------------------------------------+--------+
| # Las | # Surge Protection Anthology                       | # Su   |
| t upd |                                                    | rges H |
| ated  | **Part 6 \-- Tutorials, Textbooks, and Reviews**   | appen! |
| April |                                                    |        |
|  2004 | **Text of "Protective Devices" files**             |        |
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FOREWORD

This file contains the text part from seven papers on the following
subjects:

-   Metal oxide varistor: A new way to suppress transients (1979)

-   Transient overvoltage protection: The implications of new techniques
    (1981)

-   Lightning and NEMP transient protection with metal oxide varistors
    (1982)

-   Surge suppressors and clamps (1988)

-   A glimpse at long-term effects of momentary overvoltages on ZnO
    varistors (1989)

-   What are the lights on your surge protector telling you ? (1998)

-   Lingering lead length legacies (2004)

This file is formatted as MS Word, allowing you to do a search for
keywords, but it does not support graphics as it was derived from an OCR
scan of hard-copy archives. However, should you wish to examine the
complete original format, each page in this file has an identifying
header and footer that contain a hyperlink to the pdf file for the
document being displayed on that page, regardless of the font size that
you select for optimum viewing. These headers and footers and hyperlinks
become accessible when you select the "Print Layout" in the "View" mode.

Best wishes and good browsing !

![](media/image1.png){width="0.875in" height="0.4270833333333333in"}

Metal-oxide varistor:

a new way to suppress transients

by J.D. Harnden and F.D. Martzloff, *Corporate Research and Development,
General Electric Co.*

and W.G. Morris and F.B. Golden, *Semiconductor Products Department,
General Electric Co.*

Although it belongs to the established family of reliable varistor
protection devices, the recently introduced metal-oxide type of varistor
is adding a new dimension to the technology of protecting circuits and
components. Trademarked as the GE-MOV and MOV varistor by General
Electric Co., the metal-oxide-type varistor offers the advantages of
nanosecond switching speeds and small size, while being able to handle
current surges on the order of hundreds of amperes.

Like other varistor transient suppressors, the new varistor has a
nonlinear voltage-current characteristic that makes it useful in
voltage-regulation applications. And because its nonlinear V-I curve is
very steep-steeper than that of most other varistors-it can pass widely
varying currents over a narrow voltage range. In some applications,
using the device allows a circuit to be redesigned with fewer
components.

At low applied voltages, the metal-oxide-type varistor looks like an
open circuit because its unique two-phase material assumes the
properties of an insulator. When applied voltage exceeds rated clamping
voltage, the de­vice effectively becomes a short circuit, protecting the
component that it shunts.

The unit, moreover, requires very little standby power, making it useful
for guarding semiconductors. Steady-state power dissipation is typically
a fraction of a milliwatt, as compared to the hundreds of watts
dissi­pated by some other varistor devices.

At present, operating voltage ratings for MOV-brand series VP varistors
range from 140 to 1,400 volts peak, watt-second ratings from 10 to 160
joules, and continuous power ratings range from 0.5 watt to 1.3 watt.
Units are priced from less than \$1 to \$14 in 1,000-unit lots.

# An Inside look

The metal-oxide-type varistor (Fig. 1) has an en­capsulated
polycrystalline ceramic body, with metal contacts and wire leads. Zinc
oxide and bismuth oxide, the essential ingredients, are mixed with
proprietary powdered additives, and then pressed into disks and sintered
at a temperature greater than 1,200^o^C.

Because the bismuth oxide is molten above 825^o^C, it assists in the
formation of a dense polycrystalline ce­ramic through liquid-phase
sintering. During cooling, the liquid phase forms a rigid amorphous
coating around each zinc oxide grain, yielding a microstructure of zinc
oxide grains that are isolated from each other by a thin continuous
intergranular phase. It is this complex two-phase microstructure that is
responsible for the nonlinear characteristics.

The voltage across a metal-oxide-type varistor and the current through
it are related by the power law, I=kVn, where k is a constant. Exponent
n, which is also referred to as alpha (α), typically has a value between
25 and 50 or more, leading to the idealized V-I curve of Fig. 1. Over a
wide current range, the voltage remains within a narrow band that is
commonly called the varistor voltage.

Many properties of the varistor can be directly related to the
microstructure and the properties of the two phases. As shown in Fig. 1,
the unit\'s idealized V-I curve consists of two linear segments for
extremely low and extremely high currents, and a nonlinear segment in
the varistor voltage range.

At low applied voltages, the device\'s linear characteristic can be
attributed primarily to leakage current through the intergranular phase.
Device behavior is roughly that of an insulator, indicating that low
applied voltages cause a high electric field across the intergranular
phase and a low field within the zinc oxide grains.

In the varistor voltage range, the intergranular phase becomes
nonlinear, and current through it increases rapidly as the voltage is
raised slowly. The conduction mechanism that affects the nonlinear
characteristic is now under investigation and is thought to be
space-charge-limiting or tunneling phenomena. For extremely high
currents, the resistance of the intergranular layer becomes less than
that of the zinc oxide grains, causing the V-I curve to tend towards
linearity again.

# Comparing transient suppressors

A comparison of the volt-ampere characteristic of the metal-oxide-type
varistor to that of other voltage suppressors yields the graph shown in
Fig. 2. A number of varistor devices are represented, including silicon,
selenium, silicon-carbide, and metal-oxide types. A point-of-reference
curve for a linear ohmic resistor is also shown. The higher a device\'s
alpha is, the better is its voltage suppression capability.

The parameters of some of the most widely used transient voltage
suppressors are summarized in the table of Fig. 2. (The commercial
voltage ratings reflect peak values; they must be divided by 1.41 to
obtain root-mean-square values.) It should be noted that selenium
suppressors are normally supplied as a single package of several
series-connected plates, so that the selenium voltage rating is not that
of a single plate. The other varistor data, however, reflects
single-unit ratings.

Metal-oxide-type varistors can also be connected in series to increase
clamping-voltage rating. Moreover, they can serve in a structural mode
as well as an electrical one, for instance, by replacing the spacer
between switch contacts to allow approaching arcless switching
commutation, especially in dc applications.

One of the most important considerations when choosing a protection
device is its steady-state power dissipation. In the case of varistors,
the higher the alpha or exponent, the lower will be the standby loss.
For example, if a varistor is shunted across a load, protecting it at a
300-V level in the presence of a 2,000-V transient with a surge
impedance of 10 ohms, a steady-state power dissipation is required of
the varistor, since it is connected continuously to the 117-V rms line.
For the silicon-carbide varistor, which has a typical alpha of 3, the
necessary steady-state dissipation will be 660 watts. On the other hand,
a metal-oxide-type varistor with an exponent of 30 will dissipate only
0.1 milliwatt.

**Suppressor comparison**

V-I curves of various varistor transient suppressors and linear ohmic
resistor show that characteristic for metal-oxide-type varistor is
nearly horizontal because of its high alpha (α) value. (All varistors
obey power law, I = kV^α^) Table compares some key specifications for
several popular surge protectors. Shortly, clamping-voltage range will
be broader for 9 MOV varistors.

# Examining device behavior

Fig. 3a shows the simplified equivalent circuit for the metal-oxide-type
varistor, as well as its schematic symbol; representative capacitance
and resistance values are used. The low-current-level resistance, R-off,
is due primarily to the resistivity of the intergranular phase.
Capacitance C can be attributed to the very thin dielectric of the
intergranular phase; it becomes a fairly significant factor for the
varistor\'s dynamic characteristic. The high-current-level resistance,
R-on, is due to the intrinsic resistivity of the zinc oxide grains; it
represents the unit\'s limiting resistance. The component values in the
figure are for a typical MOV-brand series VP device.

As indicated in Fig. 3b, the device\'s V-I curve ordinarily exhibits a
slightly negative voltage temperature coefficient, in the order of 0.0 1
% /^o^C to 0.05 % /^o^C for the series VP varistors. At very low current
levels, this coefficient is substantially larger but does not affect the
normal operating range. Maximum steady-state power dissipation for a
disk with 3 square centimeters of total surface area is usually 1 watt
in a 70^o^C free-convection ambient environment.

Typical capacitance curves are depicted in Fig. 3c for units with
differing diameters. Device capacitance influences high-frequency
impedance properties, as shown by the plots of Fig. 3d for two
production units\--one with a 130-V rms rating, and the other with a
1,000-V rms rating. A point-of-reference curve for a fixed
0.005-microfarad capacitor is also included.

When a fast-rising surge is applied to a metal-oxide-type varistor, its
capacitance immediately makes it appear as a low impedance. After the
capacitance becomes fully charged, the unit simply operates at the point
predicted by its V-I characteristic.

# Tracing the origin of transients

Transient surges originate from a variety of sources. Regardless of
whether electrical circuits operate from ac or dc sources, they are
often plagued by voltage tran­sients that are either generated within the
circuit itself or transmitted into the circuit from external sources.

One of the most common sources of transients in power distribution
systems is the L(di/dt) voltage caused by transformer magnetizing
currents as the transformers are switched within either a feeder utility
system or an industrial plant\'s own distribution system. Residual
lightning surges are another source of concern. These surges can be
thought of as an overflow on the main lightning arrestors with which
most consumer, commercial, and industrial distribution systems are
protected at the interface between the utility and the user\'s
distribution systems. Still other sources can be found in homes
themselves, resulting from equipment that is connected to the utility
system.

Even a small synchronous line clock in the home can be subjected to a
number of surges within just a 24-hour period. In fact, with the
introduction of more automatic switching functions and complex
electrical equipment into the home, line surges are becoming more
frequent. And oddly, the better and newer the wiring and installation,
the less inherent protection is provided by spill-over occurring in
poorly wired outlets, switches, and fixtures, and the greater the
resulting impressed voltage level on connected equipment.

# Using metal-oxide-type varistors

When a step-down transformer is switched on, it can impress severe
transients on any components connected to its secondary winding due to
its interwinding capacitance. Installing a metal-oxide-type varistor, as
shown in Fig. 4a, can eliminate this startup transient.

Another source of component failure, which is some­times overlooked, can
occur in the conventional transis­tor series-pass voltage regulator (Fig.
4b). When the circuit is turned on, the capacitor appears to be a short
circuit, and the transistor is exposed to the full unregulated bus
voltage. Placing a metal-oxide-type varistor across the transistor
allows a soft current rise to pass through the regulator without the
usual voltage surge.

Semiconductors can be protected with the new varistor, resulting in a
design with fewer components, because the device can improve the
electrical properties of the circuit in which it is installed. As an
example, consider the output stage of small fine-operated radio (Fig.
4c) that requires high-voltage transistors and an associ­ated RC network
to withstand the voltage spikes generated by distortion during overload.

The oscilloscope traces of transistor stress show the voltage transient
that occurs with normal RC suppression but is dramatically reduced with
varistor suppression. Because of this significant transient reduction,
the circuit can be redesigned with fewer parts. (The third scope trace
displays varistor current.)

# Choosing the right varistor

Selecting the correct metal-oxide-type varistor is a simple, logical
procedure. First, find the device with a peak operating voltage rating
that is close to, yet higher than, the normal peak ac-line voltage.
Next, determine or estimate the energy level of the transient to be
sup­pressed. This energy level is usually determined by the energy term,
LI2/2, where I is the peak magnetizing current flowing in inductance L,
which stores the tran­sient energy in its field. In the case of
transformers, I may be considered the peak exciting current.

The expected transient current level must be found next. (Peak feeder
transformer magnetizing current, reflected to the secondary, is often
used to estimate the peak transient current.) Then, the varistor unit
can be selected that has the proper ratings for recurrent voltage,
clamping voltage, and energy level.

In the case of dc applications, varistor steady-state power dissipation
should be checked. Be sure to derate the device\'s energy rating if
power dissipation demands will elevate the varistor case temperature
above 85 ^o^C. For applications where repetitive transients may be
encountered, calculate the expected watt-seconds per pulse and multiply
this figure by the pulse repetition rate to determine the additional
steady-state power dissipation required. And finally, make certain that
the unit can comply with such ambient environmental factors as operating
and storage temperatures.

# Looking ahead

The next major developments for MOV-brand varistors will revolve around
package changes to enhance applications versatility and improve heat
transfer. Future packages will be available in a variety of mounting
schemes-for example, a bolt-down version with or one without
heat-transfer capability that could be directly fastened to chassis,
brackets, bus bars, and heat exchangers. Low-inductance pill-shaped
packages that are compatible with pressure-mounted thyristors are also
being considered. Another possibility is a finned package for direct air
convection and forced cooling.

Since metal-oxide-type varistors can be made rather thick, they should
be available in the near future with kilovolt ratings. MOV-brand series
VP units are currently being developed to cover the clamping voltage
range of 30 V to 10 kV.

# BIBLIOGRAPHY

R. A. Delaney and H. D. Kaiser, Journal of Electrochemical Society,
Solid State Sciences, Vol. 114, pp. 883-842,1967.

M. Matsuoka, T. Masyama, and Y. Iida, Supplementary Journal of Japanese
Society of Applied Physics, Vol. 39, 94-101, 1970.

M. Matsuoka, Japanese Journal of Applied Physics, Vol. 10, 736-746,
1971.

W. G. Morris, "The Electrical Properties of ZnO-Bi~2~O~3~ Ceramics,"
General Electric Co., Corporate Research and Development Report
72CRD069.

F. B. Golden, R. W. Fox, "GE-MOV Varistor Voltage Transient
Suppressors," General Electric Co., Semiconductor Products Dept.,
Application Note 200.60.

F. B. Golden, "A New Voltage Transient Suppressor," IEEE International
Semiconductor Power Converter Conference Record, Publication 72CHO
602-3-1 A, pp. 2-6-1 to 2-6-9.

**Transient Overvoltage Protection:**

**The Implications of New Techniques**

François D. Martzloff

Corporate Research and Development

General Electric Company

# Summary

Reliability problems can occur from the use of modern electronic devices
without applying appropriate protection techniques or using incorrect
installation procedures. Although surge arresters are effective in
limiting overvoltages, a metal oxide varistor can provide a much lower
clamping voltage if installation procedures are taken into
consideration. Sparkover voltage measurements, with a specified time
rise, measured arrester performance. The response of the arresters to a
current impulse was investigated and lead effects were identified. Tests
indicated that the metal oxide varistor, installed with short leads,
provides low clamping voltage.

# Introduction

Incorrect protection for modern electronic devices from lightning
strokes can cause reliability problems which could arise from various
sources:

> · Sensitivity of modern electronic equipment
>
> · Improper procedures of installation
>
> · Complete lack of protective devices.

This paper examines new applications of old concepts which are required
by the constantly increasing use of electronic equipment; the particular
increased sensitivity of these devices; and intense, competitive
pressures. We shall consider first the design and environment of surge
arresters for low-voltage systems and then examine their performance as
a function of installation.

# Surge arrester design for low voltage systems

In the past, typical surge arresters (diverters) for service entrance
duty have been limited to a gap-varistor design. This design involves
gap sparkover voltage with a result of a relatively high clamping
voltage for the arresters. The new, commercial availability of metal
oxide varistors, with current ratings suitable for service entrance
duty, provides a low clamping voltage at the service entrance.

Surge arresters, which have sufficient current discharge capacity,
consist of a gap in series with a non-linear resistor, usually a silicon
carbide block (Figures 1, 2, and 3). These arresters are effective in
limiting overvoltages to levels compatible with solid insulation. In
recognition of this compatibility, the IEC Report 664 \[1\] proposes
voltage levels of 2500 V for a 120 V circuit and 4000 V for a 220 V
circuit (Table 1). However, these voltages are not consistent with the
inherent withstand characteristics of electronic appliances. A much
lower level (indicated by Category I or II of the 664 report) is
required, i.e., 800 or 1500 V for 120 V circuits and 1500 or 2500 V for
220 V circuits. These voltages can be achieved with a 32 mm metal oxide
varistor for which the rated clamping voltage is 550 V and 900 V for
disks suitable for 120 V and 220 V circuits, respectively.

However, the capability for low clamping voltage might not be attained
if installation procedures do not take the connecting lead effects into
consideration. Furthermore the proposed IEC practice of several cascaded
surge protective devices requires careful coordination of the devices
and the intermediate impedance \[2\], a goal which may not be easy to
achieve in routine installation practices.

**Table I**

**Preferred series of values of impulse withstand voltages**

**for rated voltages based on a controlled voltage situation**

## Test procedures and standards

The evaluation of surge arrester performance is accomplished by the
application of standardized tests which are presumably specific to the
operational environment of the arrester.

Performance tests for a low-voltage arrester include sparkover voltage
measurement with a specified rise time and also the use of one or more
current impulses to demonstrate the capability of discharging a surge
either without damage or without the production of excessive discharge
voltage during the surge. Figure 4 shows the relationship between these
parameters of a gap-varistor design. Because damage to semiconductors is
likely to occur during the initial front of the surge before sparkover,
the concern over the following discharge voltage is less significant.

Figure 5, however, shows how the gapless varistor can clamp at lower
voltages. But, there is a risk of an inductive drop which would add a
substantial voltage to the intrinsic clamping voltage due to the long
connecting leads required under some proposed regulations \[3\].

# Sparkover voltage

Figure 6 shows the sparkover voltage of typical arresters in USA
circuits at 120 V line-to-ground and, also, in European circuits at 220
or 440 V between terminals. These sparkover voltages were recorded for a
10 kV/μs rate of rise (Figure 6a). It is apparent that the gap-type
arrester oscillograms exhibit an anomaly at approximately 150 /uus
before the gap sparks over (Figure 6b, c, d).\*

In contrast, the clamping voltage of the varistor (Figures 7a and b) is
not only lower, but it is also free from any interference. In Figure 7c,
the absence of a significant overshoot in varistor clamping is shown:

> · The fast front is the open circuit voltage without the varistor
>
> · The trace to the right illustrates the clamping action of the
> varistor.

## Impulse current

The selection of the current waveform is not obvious. The use of an 8/20
μs waveform to represent surge currents associated with lightning
strokes is well established. Indeed, most standards \[4,5\] call for an
8/20 μs waveform. Levels may be in the range of 3 to 10 kA crest at the
service entrance (Table 2) \[4\].

The selection of an 8/20 μs wave reflects our present day knowledge of
typical lightning currents \[6,71. In addition, the 8/20 μs wave
discharges an appropriate amount of energy in the arrester under test.

The question, then, of the likelihood of a 8 μs front propagating along
a low voltage system can be raised. Figure 8 depicts a possible
distribution of the surge current from a stroke to an overhead system.
Taking 50 kA \[8\] as the median level of lightning stroke, the
resultant 5 kA crest is expected, and, with short distances along the
service drop, a rise time of 8 μs can be maintained.

The explanation of this peculiarity is actually quite simple. In real
time, the gap fires 150 μs before the display records the event, but the
oscilloscope used for these tests has a 150 μs delay line. Therefore,
the anomaly is the interference created in the oscilloscope by the gap.
(Even an EMI option for the oscilloscope is not enough!) This occurrence
exemplifies the objectionable effects that a gap can have upon
electronic devices.

**Table 2**

**Surge voltages and currents deemed to represent the indoor and outdoor
environment**

**and recommended for use in designing protective systems**

Within these parameters, an 8/20 μs waveform for both the conventional
arresters and the candidate metal oxide varistors in service entrance
duty appears reasonable. In addition, it is likely to be demanded in the
performance of test procedures for arresters - either by customers or by
regulatory agencies.

### **Installation of arresters in panels**

Two panels, typical of USA and European hardware (Figures 9 and 10),
were wired in the laboratory and subjected to impinging surges of 5 kA
crest, 8/20 μs (Figure 1la), that were applied between one phase line
and the panel ground. Voltages appearing at the out-going branch
circuits were recorded with oscilloscope probes by using a differential
connection after preliminary checking on signal/noise performance of the
system. Figure 1lb shows the response of the 120 V arrester to this
impinging surge. This response will be discussed in detail with the test
results.

On the USA-type panel, the 120 V arrester was installed externally to
the panel, and the 45 cm long leads were connected to the main entrance
lugs of the panel (as implied by the specifications of the National
Electrical Code and the proposed UL Document \[31). The 220 V arrester
is designed for installation in the panel, and the point-to-point wiring
allows short leads for the connection across line and ground (or
neutral) inside the panel. The 440 V arrester, as indicated by the
manufacturer\'s suggested installation (Figure 12), is intended to be
connected outside at the service entrance rather than at the panel.
Consequently, in the laboratory simulation, it was connected 3 m before
the panel.

The 150 V and 250 V varistors (Figure 13) were installed either outside
or inside the panel. The installation will be discussed with the test
results.

## Test results on discharge voltage

All discharge voltage measurements were made with the surge generator
set for the standard 5 kA crest, 8/20 μs current impulse shown in Figure
11a. The clamping voltage of each device and the impedance of its
connections may reduce the current to some extent (the charging voltage
of the generator was 12 kV), but the same effect would take place under
the assumption of a current division resulting from the ratio of the
impedances offered to the impinging stroke of 50 kA.

Figure 11b shows the discharge voltage of the 120 V arrester which
reflects the applied current wave of Figure 11. In view of the
expectation raised by the low-clamping voltage of the metal oxide
varistors, the discharge voltage of the 150 V varistor recorded in
Figure 14a seems disappointing. This can be explained easily. The
clamping voltage of the varistor is degraded by the addition of the
voltage due to the 45 cm leads (Figure 14b). Setting aside the proposed
installation requirements and seeking optimum performance, the short
connections of Figure 15 produce the remarkably low discharge voltage
shown in Figure 16a. For the 220/380 V panel (Figure 10), the layout of
components and the absence of conflicting specifications, that is, the
promoting of short leads in the standards \[9,10\], makes possible the
equally remarkable low-clamping voltage of the 250 V varistor shown in
Figure 16b.

In contrast, the discharge voltages of conventional arresters are higher
and contain some high frequency oscillations which may be troublesome.
Granted that the voltages are clamped to levels which eliminate the
hazards of flashover in the wiring. That is an accomplishment already.
But these still relatively high discharge voltages may not be low enough
to ensure the survival of electronics connected directly to the mains
protected by these arresters.

Figure 17 shows the response of the integral arrester in the 220/380 V
panel. The short connections made possible by this arrangement eliminate
the problem of added voltage drop. The initial response (17a) of the gap
sparkover is well balanced with the discharge voltage during the full
impulse (17b). There is, however, the problem of unavoidable collapse of
voltage following sparkover, with a possible result of producing
interference in connected electronics as well as direct radiation. (See
footnote under Sparkover voltage.)

Figure 18 shows the response of the arrester installed at the service
entrance. The initial response (Figure 18a) indicates that the
additional leads inductance and capacitance can produce peculiar
resonances. Nevertheless, the complete impulse discharge (18b) is well
balanced with the initial response although the initial collapse reaches
the full amplitude during sparkover.

# Conclusion

Present technology offers two choices for the protection of low-voltage
circuits against atmospheric overvoltages:

> · Conventional arresters
>
> · Metal oxide varistors.

Although conventional arresters provide protection against the hazards
of wiring flashover, they can still allow voltages damaging or
disturbing sensitive electronics. Metal oxide varistors, although not
yet packaged in a manner convenient for panel installation, not only
produce low clamping voltages but they also produce no high frequency
disturbances. These benefits, however, will be obtainable only if proper
installation procedures are followed.

# Acknowledgments

The assistance of Eric Carroll (GETSCO, Paris) and Louis Regez (Landis &
Gyr, Zug) in identifying practices and procuring components, and of Alma
Bongarten, Editor, is gratefully acknowledged.

# References

1\. Insulation coordination within low-voltage systems including
clearances and creepage distances for equip-ent. International
Electrotechnical Commission Report 6640. Geneva, Switzerland 1980

2\. Martzloff, F.D.: \"Coordination of surge protectors in low-voltage
ac power circuits.\" IEEE Transactions on Power Apparatus and Systems.
Vol. PAS-99 No. 1, January/February 1980.

3\. Proposed UL Standard 1449 for low-voltage transient suppressors

4\. Guide for application of valve-type lightning arresters .for
alternative-current systems. ANSI Standard C62.2

5\. Guide on surge voltages in low voltage ac power circuits. IEEE
Standard to be published in 1981

6\. Cianos N. and Pierce E.T.: A ground lightning environment for
engineering usage. Stanford Research Institute Technical Report. August
1972

7\. Golde, R.H., ed: Lightning, 2 vols., Academic Press. London and New
York 1977

8\. Johnson, I.B.: Surge protection in power systems. IEEE Tutorial
Course. IEEE Power Engineering Society. 79EHO144-6-PWR, 1978

9\. Installations électriques à basse tension. French Standard NF
C15-100

10\. Leitsaetze fur den Schutz elektrischer Anlagen gegen
Überspannungen. German Standard VDE 0675

LIGHTNING AND NEMP TRANSIENT PROTECTION

**WITH METAL OXIDE VARISTORS**

François D. Martzloff

General Electric Company

# FUNDAMENTALS ON VARISTORS

The term varistor is derived from its function as a variable resistor.
It is also called a voltage-dependent resistor, but that description
misleadingly implies that the voltage is the independent parameter in
surge protection. Two very different devices have been successfully
developed as varistors: silicon carbide discs have been used for years
in the surge arrester industry, and, more recently, metal oxide varistor
technology has come of age.

## Structure and Packaging

Composition of the varistor consists primarily of zinc oxide with small
additions of selected metal oxides. These materials are sintered at high
temperature to produce a polycrystalline ceramic body. The structure of
the body is a matrix of conductive zinc oxide grains separated by a
highly resistive intergranular boundary (Figure 1).

Figure 1. Photomicrograph of a polished and etched section of a metal
oxide varistor.

Since electrical conduction occurs between zinc oxide grains distributed
throughout the bulk of the device, a varistor is inherently more rugged
than its single junction counterparts, such as zener diodes. In the
varistor, energy is absorbed throughout the body of the device with the
resultant heat spread evenly through its volume. Electrical properties
are controlled mainly by the physical dimensions of the varistor body,
which is generally sintered in the shape of a disc. Energy rating is
determined by volume, voltage rating by thickness, and current
capability by area.

In conventional applications, a varistor device consists of a disc of
the material, with electrodes applied by silk screen, metal spray, etc.,
to opposite, parallel faces. A connection is made to the electrodes for
feeding current in and out of the varistor by attachment of the device
leads to the disc, and thence to the circuit. The attachment can take
many forms (Figure 2),(1) but all involve the insertion of some lead
length, a potential problem that will be discussed later.

## Electrical Parameters

Varistor V-I Characteristics

Varistor electrical characteristics are conve­niently displayed using a
log-log format in order to show the wide range of the V-I curve. The log
format is also clearer than a linear representation which tends to
exaggerate the nonlinearity in proportion to the current scale chosen. A
typical V-I characteristic curve is shown in Figure 3. In order to
illustrate three distinct regions of electrical operation, this plot
shows a wider range of current than is normally provided on varistor
data sheets.

In the leakage region (Figure 3), the predominant parameter is the R_off
resistance (Figure 4), shown as the 109 ohm line in the figure. As the
current density increases, normal varistor operation occurs, with R_x
being the predominant parameter. Finally, in the upturn region, where
R_x has become very small compared to R_on, the latter dominates. Two
other parameters, the capacitance C (reflecting a high effective
dielectric constant of the material) and the lead inductance at high
frequencies.

The leakage region is of little interest to the user; in the normal
operating region, the flatness of the characteristic is a criterion of
clamping effectiveness. Departure from this flat characteristic in the
upturn region could be viewed as undesirable, however, this upturn is
useful in providing current sharing across the section of the device at
high currents.

The capacitance may be objectionable at high frequencies. Adding a low
capacitance avalanche diode in series is a method of reducing the
effective capacitance. The unconventional examples described in the next
section also show means of reducing capacitive effects.

The lead inductance, already mentioned, will be discussed at greater
length in the following subsections.

The V-I characteristic, then, is the basic applica­tion design tool for
selecting a device to perform a protective function. For a successful
application, however, other factors, which are discussed in detail in
the information available from manufacturers, must also be taken into
consideration. Some of these factors are:

> · Selection of the appropriate nominal voltage for the line voltage of
> the application
>
> · Selection of energy-handling capability (including source impedance
> of the transient, waveshape, and number of occurrences)
>
> · Heat dissipation
>
> · Proper installation in the circuit (configuration, lead length)

**\"Overshoot\" - A Lead Effect**

Enough instances of poor installation practices have been observed, and
enough questions have been raised on alleged \"overshoot,\" that a brief
discussion of lead effects is in order.(2) To illustrate the effect of
lead length on the overshoot, two measurement arrangements were used. As
shown in Figures 5a and 5b, respectively, 0.5 cm^2^ and 22 cm^2^ of area
were enclosed by the leads of the varistor and of the voltage probe.

The corresponding voltage measurements are shown in the oscillograms of
Figures 6a and 6b. With a slow current front of 8 μs, there is little
difference in the voltages occurring with a small or large loop area,
even with a peak current of 2.7 kA.

With the steep front of 0.5 μs, the peak voltage recorded with the large
loop is nearly twice the voltage of the small loop. Note (Figure 6b)
that at the current peak, L di/dt = 0, and the two voltage readings are
equal; before the peak, L di/dt is positive, and after, it is negative.

Hence, when one is making measurements as well as when one is designing
a circuit for a protection scheme, it is essential to be alert to the
effects of lead length (or, more accurately, of loop area) for
connecting the varistors. This warning is especially important when the
currents are in excess of a few amperes with rise times of less than one
microsecond.

## Speed of Response

The varistor action depends upon a conduction mechanism similar to that
of other semiconductor devices. For this reason, conduction occurs very
rapidly, with no time lag apparent, even into the nanosecond range.
Figure 7 shows a composite photograph of two voltage traces with and
without a varistor inserted in a very low inductance impulse
generator.(3) The second trace (which is not synchronized with the
first, but merely superimposed on the oscilloscope screen) shows that
the voltage clamping effect of the varistor occurs in less than one
nanosecond.

In the conventional lead-mounted devices, the inductance of the leads
would completely mask the fast action of the varistor. Therefore, the
test circuit for Figure 7 required insertion of a small piece of
varistor material in a coaxial line to demonstrate the intrinsic
varistor response.

Tests made on lead-mounted devices, even with careful attention to
minimizing lead length, show that the voltages induced in the loop
formed by the leads contribute a substantial part of the voltage
appearing across the terminals of a varistor at high current and fast
current rise. In typical power applications where the threat is
lightning or switching surges, the impressed current rise time is seldom
short enough for this effect to become significant. Competitive
technology will make claims that an \"overshoot\" occurs, but that has
been shown to be primarily a lead effect as discussed in the preceding
subsection.

Voltage rate-of-rise is not the best term to use when discussing the
response of a varistor to a fast impulse, unlike spark gaps where a
finite time is involved in switching from a non-conducting to a
conducting state. The response of the varistor to the transient current
delivered by a circuit is the appropriate characteristic to consider.

Applications involving NEMP protection will be fast-rising voltages and
current surges, so that the packaging will become very significant, and
a depar­ture will be required from the conventional packages currently
found in commercial devices. In the next section, these unconventional
applications will be briefly presented.

## Failure Modes

An electrical component is subject to failure either because its
capability was exceeded by the applied stress or because some latent
defect in the component went by unnoticed in the quality control
processes. While this situation is well recognized for ordinary
components, a surge protective device, which is no exception to these
limitations, tends to be expected to perform miracles, or to at least
fail graciously in a \"fail-safe\" mode. The term \"fail to different
users and, therefore, should not be used. To some users, fail-safe means
that the protected hardware must never be exposed to an over-voltage, so
that failure of the protective device must be in the fail-short mode,
even if it puts the system out of operation. To other users, fail-safe
means that the function must be maintained, even if the hardware is left
temporarily unprotected, so that failure of the protective device must
be in the open-circuit mode. It is more accurate and less misleading to
describe failure modes as \"fail-short\" or \"fail-open,\" as the case
may be.(4)

Metal oxide varistors, when subjected to excessive energy (high surge
current), will fail in the \"fail-short\" mode. If the varistor is
connected to a power system with large short-circuit capability, the
fault current may produce melting of a soldered safe,\" however, may
mean different failure modes connection or shattering of the ceramic,
ultimately producing a \"fail-open\" condition.

**UNCONVENTIONAL APPLICATIONS**

For the protection of fast rise transients occurring with NEMP, it is
apparent that lead-mounted devices are likely to be unsatisfactory. To
still benefit from the inherent advantages of metal oxide varistors,
novel packaging is required. In this sec­tion, several examples are
given: bulkhead connectors and coaxial connectors, each with
conventional sintered blocks or with reconstituted material. Both have
been shown to be feasible, although the complete details have not yet
been fully developed.

Two further applications are also described, which are still in the
conceptual stage, but show promise of useful application in the
integrated pro­tection of circuits: as thick film layers between
con­ductor runs, and as substrates.

## Bulkhead Connectors

The inductance of the leads whose effects have been discussed can be
essentially eliminated by construction in a coaxial configuration. This
type of packaging would be especially effective and convenient in
conjunction with connectors for cable entry into shielded boxes. (5)

Figure 8 shows a configuration for a single wire, and Figure 9 for
multiple wires or pairs. In both figures, a varistor bead is inserted
between the conductor(s) and an opening of the bulkhead plate, which is
part of the grounded shell. In this manner, any impinging surge current
flowing in the conductor is diverted to the grounded chassis of the
enclosure, with minimum inductance inserted.

In principle, this construction is quite simple. The details of the
bonding between conductors and varistor will require development, which
should be within the skills of connector manufacturing.

A possible objection to this construction would be-that the whole volume
of the varistor bead acts as the dielectric of a capacitor inserted
between the conductor and its sheath. With an effective dielectric
constant of 1000, and the small radial distance required for low-voltage
clamping, the resulting capacitance might be excessive for
high-frequency applications. We note, however, that some advanced
varistor materials have been fabricated with lower dielectric constants.

An alternate construction is shown in Figure 10, where the varistor disc
has two concentric electrodes that are silk-screened on the side of the
disc, rather than on the inside and outside surfaces, as in Figures 8
and 9. In this manner, the disc can have a substantial distance between
the inside and out side surfaces, reducing the resulting capacitance.
The clamping voltage is then determined by the distance between the
edges of the two electrodes This arrangement offers the advantages of
lower capacitance and greater accuracy in the distance between
electrodes, albeit at the cost of reducing the energy handling
capability since only a thin surface layer of the varistor disc will
conduct the surge current.

Coa**xial Mounting**

A varistor can also be inserted in a shielded cable by a coaxial \"Tee\"
configuration. In fact, that configuration is required for making valid
measurements of the varistor performance in the submicrosecond time
range. (3)

Here again, the capacitance of the varistor might be objectionable at
high frequencies. A possible remedy to this situation is shown in the
configuration of Figure 11, where the capacitance of the varistor is
series-tuned to the inductance of the clamping spring. This achieves
high impedance to the UHF signal frequency, thus low loading, but offers
a lower impedance and good suppression at the lower frequencies
associated with lightning surge currents. (6) This would not, of course,
be suitable for NEMP protection.

# Reconstituted Material

In the two preceding examples, the description of the varistor simply
stated \"varistor disc,\" with the assumption that the material had been
prepared in the conventional sequence of pressing a powder into shape
and sintering the piece in its final form. There is another method of
producing varistor components, especially attractive for parts shaped in
a form different than conventional mass-produced discs.

In this method, the varistor material, obtained by the usual sintering,
is ground into particles of controlled dimension, mixed with a suitable
binder in proper proportions, and molded into the finished part. By this
method, it is possible to embed electrodes and conductors directly into
the varistor material, as well as to fill a cavity such as a conductor
shell. (7)

Figure 12 shows the structure of a portion of a device using
reconstituted material (tracing of a photomicrograph), where the three
constituents interface: electrode, particle of varistor material, and
plastic binder. Here, the particle of varistor material is formed of
several grains of ZnO with the varistor boundary layer between the
individual ZnO grains. If the particles of varistor material were
smaller than the ZnO grains, there would not be any barrier layer left
and the varistor action would be lost. Thus, there is a need to optimize
the grain size, the particle size, and the proportion of binder,
depending on the specific application.

Figure 13 shows a photograph of a connector made by molding the
reconstituted varistor material inside the shell of a connector provided
with two pins. The resulting varistor characteristic is shown in Figure
15 for pin-to-pin and pin-to-case. With a shell that would be especially
made for this application (the sample shown here was a standard shell),
the pin-to-case characteristic would be controlled for lower voltage.

Another shape demonstrated is the plug of Figure 14, where a flat-headed
stud is embedded in a metal cup, illustrating how electrodes can be
molded directly into the reconstituted material. The resulting
characteristic is shown in Figure 16.

## Thick Film

The reconstituted material just described might also be useful for
producing varistor layers with thick-film technology. Currently, the
reconstituted material is obtained by pressing the particles of varistor
material in a cavity. Producing a paint-like film would require some
development in order to maintain the particle/binder proportion leading
to the behavior as a varistor. Should the particles become isolated from
one another by an excess of binder, the varistor action would be lost.

With a suitable paint-like compound, it would be possible to silk screen
varistor layers directly between conductor runs. The clamping voltage
would be controlled by the distance between the runs, while the current
density would be controlled by the combined thickness of the layer and
length along the run.

As an alternate to painting layers of reconstituted varistor material
between runs, a more advanced concept would include the sintering of
varistor layers on a suitable substrate, on which runs and other
components could be added in sub-sequent processing. This concept has
not been reduced to practice, but illustrates the wide range of options
that may be available beyond the commercial packaging of discs.

## Substrates

Another approach to the integration of varistor protection to circuits
would be the use of substrate slabs of sintered varistor material, on
which the circuit runs could be metallized. Two possible concepts are
proposed. In Figure 17, the substrate acts only along a thin layer
between two runs located on the same side of the substrate. Because of
the highly nonlinear property of the material, there is little fringing
of current into the depth of the substrate.

In Figure 18, the substrate acts as an insulating barrier between the
runs on one side and the ground plane on the other side. Under normal
operating voltages, the substrate layer offers a very large resistance
and, thus, is nearly an insulator. Upon development of a high voltage
between any part of the circuit and ground, the surge is immediately
diverted to the ground plane without any additional lead length.

In the latter approach, development of optimum compositions would allow
producing a substrate with sufficient mechanical strength and a voltage
which is still low enough. Here we are dealing with the suppression of
transients of moderate energy at the point where they may be induced
(presumably as the circuit would be illuminated by an EM field).

## Varistor Materials Technology

Many of the unconventional varistor applications listed above required
tailoring the varistor materials properties for optimum implementation
of the system requirements under consideration. Using technology
available or under development, we can modify the varistor material in a
variety of ways. For example, we may:

1\. Increase the varistor breakdown field, thereby decreasing the space
necessary for a given breakdown voltage.

2\. Decrease (or increase) the varistor dielectric constant.

3\. Fabricate screen-printable thin film varistors.

4\. Fabricate large (up to about 103 g weight and 30 cm diameter)
devices for extremely large energy or current carrying capacities.

5\. Directly press varistors in special shapes (e.g., rings or slabs) to
fit specific connector sys­tems.

6\. Fabricate low-voltage varistors with breakdown voltages as low as 15
to 20 volts.

7\. Vary the degree of varistor nonlinearity.

In general, optimization of a particular varistor material parameter
will involve a trade-off with other material properties. The proper
component design will involve the synergistic combination of materials
technology and circuit design expertise.

## CONCLUSIONS

Varistor materials have been shown suitable for applications other than
the conventional lead ­mounted device used in electronic circuit boards
or the large blocks used in power system surge arresters. The
feasibility of some approaches has been shown, with details of
implementation to be developed. For other approaches, the development is
still at the conceptual stage.

# ACKNOWLEDGMENTS

The concepts and technologies discussed in this report represent the
contributions of many others, associated with metal oxide varistor
technologies, some since their early introduction into the United States
by General Electric. The authors listed in the references contributed to
the concepts discussed in the present report and should be acknowledged
as implicit co-authors of this report.

**REFERENCES**

1\. Transient Voltage Suppression Manual, 2nd Edition, edited by J.C.
Hey and W.P. Kram, General Electric Company, Auburn, NY, 1978.

2\. F.A. Fisher, \"\'Overshoot\' - A Lead Effect in Varistor
Characteristics,\" General Electric TIS Report 78CRD201, Schenectady,
NY.

3\. H.R. Philipp and L.M. Levinson, \"ZnO Varistors for Protection
Against Nuclear Electromagnetic Pulses,\" J. Appl. Phys. 52(2), 1981,
pp. 1083-1090.

4\. IEEE Standard C62.33-1981, \"Test Specifications for Varistor Surge
Protective Devices.\"

5\. D.M. Tasca, J.D. Harnden, Jr., and F.D. Martzloff, U.S. Patent
3,711,794 (1973).

6\. F.D. Martzloff, U.S. Patent 3,863,111 (1975).

7\. J.F. Burgess, R.T. Girard, F.D. Martzloff, C.A. Neugebauer, U.S.
Patent 4,103,274 (1978).

SURGE SUPPRESSORS AND CLAMPS

François D. Martzloff

National Bureau of Standards

Gaithersburg MD 20899

# INTRODUCTION

Various devices have been developed for the protection of electrical and
electronic equipment against transient overvoltages. They are often
called \"transient suppressors\" although, for accuracy, they should be
called \"transient limiters,\" \"clamps,\" or \"diverters\" because they
cannot really suppress the transients; rather, they limit the transients
to acceptable levels or make them harmless by diverting them to ground.
The IEEE dictionary has selected the more generic but lengthy term of
"surge protective device."

There are two categories of transient suppressors: those that block
transients, preventing their propagation toward sensitive circuits, and
those that divert transients, limiting residual voltages. Since many of
these transients originate from a current source, blocking them may not
always be possible because the current forced into the high-impedance
blocking path would only result in higher voltages and breakdown.
Therefore, diverting of the transient is more likely to find general
application. A combination of diverting and blocking can be a very
effective approach. This approach generally takes the form of a
multistage circuit, where a first device diverts the transient current
to ground; a second device offers a restricted path for transient
propagation but an acceptable path for the signal or power, and a third
device clamps the residual transient (Figure 1). Thus, we are primarily
interested in diverting devices.

The diverting device can be of two kinds: voltage-clamping or
short-circuiting devices (the latter called \"crowbars\"). Both of them
involve some nonlinearity, either frequency nonlinearity (as in filters)
or, more usually, voltage nonlinearity. This voltage nonlinearity is the
result of two different mechanisms \-- a continuous change in the device
conductivity as the current increases, or an abrupt switching action as
the voltage increases.

Because the technical and trade literature contains many articles on
these devices, we shall limit the discussions of the details but make
some comparisons to point out the significant differences in the
performance . Understanding the relative merits and the limitations of
each technology will allow informed choice and recognition that there is
room for each approach in the wide range of needs and applications.

# CROWBAR DEVICES

The principle of crowbar devices is quite simple: upon occurrence of an
overvoltage, the device changes from a high-impedance state to a
low-impedance state, offering a low-impedance path to divert the surge
to ground. This switching can be inherent to the device, as in the case
of spark gaps involving the breakdown of a gas or be the recently
introduced combination of multijunction solid state devices.

The major advantage of the crowbar device is that its low impedance
allows the flow of substantial surge currents without the development of
high energies within the device itself; the energy has to be spent
elsewhere in the circuit. This \"reflection\" of the impinging surge can
also be a disadvantage in some circuits if the transient disturbance
associated with the gap firing is being considered. Where there is no
power-follow (discussed below), such as in some communication circuits,
the spark gap has the advantage of very simple construction with
potentially low cost.

The crowbar device, however, has three limitations. one is the volt-time
sensitivity of the breakdown process in air gaps or gas tubes. As the
voltage increases across a gap, significant conduction of current --
therefore voltage limitation for the surge -- cannot occur until the
transition to the arc mode of conduction, by avalanche breakdown of the
gas between the two electrodes. The load is unprotected during the
initial rise because of this delay time (typically in microseconds) .
Large variations exist in sparkover voltage attained in successive
operations since the process is statistical in nature. This sparkover
voltage can also be substantially higher after a long period of rest
than after successive discharges. Because of the physics of the process,
it is difficult to produce consistent sparkover voltage for low voltage
ratings. This difficultly is increased by the effect of manufacturing
tolerances on very small gap distances, but can be alleviated by filling
the tube with a gas having lower breakdown voltage than air. The
technology developed by manufacturers of gas tubes has minimized these
effects. One advantage of the solid ­state crowbars over the gas
breakdown tubes is the absence of any practical time delay in firing.

The second limitation is associated with the speed of the sparkover,
which produces fast current rise in the circuits and, thus objectionable
noise. An example is found in hybrid protective systems. Figure 2 shows
the circuit of such a commercially available device. The gap does a very
nice job of diverting impinging high energy surges, but the magnetic
field associated with the high di/dt induces a voltage in the loop
adjacent to the second suppressor, adding a substantial spike to what
was expected to be a low clamping voltage. Some product literature
advocates locating solid-state crowbars on the very circuit board where
components to be protected are mounted -- a perfect recipe for injecting
interference right into the circuits!

A third limitation occurs if power current from the steady-state voltage
source can follow the surge discharge (hence the term \"power-follow\").
In ac circuits, this power follow current may or may not be cleared at a
natural current zero. Additional means, therefore, must be provided to
open the power circuit if the crowbar device is not designed to provide
self-clearing action within specified limits of surge energy, system
voltage, and power-follow current. This combination of a gap with a
current-limiting, nonlinear varistor has been very successful in the
utility industry as high voltage surge arresters.

# VOLTAGE-CLAMPING DEVICES

Voltage-clamping devices have variable impedance, depending on the
current flowing through the device or the voltage across its terminal.
Impedance variation is mono­tonic and does not contain discontinuities,
in contrast to crowbar devices, which show a turn-on action. As far as
their volt-ampere characteristics are concerned, these devices are
time-dependent to a certain degree. However, unlike the sparkover of a
gap or the triggering of a thyristor, time delay is not involved.

When a voltage clamping device is applied, the circuit remains
essentially unaffected by the device before and after the transient for
any steady-state voltage below clamping level. Increased current drawn
through the device as the surge voltage attempts to increase results in
voltage-clamping action. Nonlinear impedance is the result if this
current rise is greater than the voltage increase. The increased voltage
drop in the source impedance (Zs in Figure 3) due to higher current
results in the apparent clamping of the voltage. It must be emphasized
that the device depends\' on the [source impedance]{.underline} to
produce this clamping. The circuit behaves as a voltage divider where
the source impedance (high side of the divider) is constant but the
clamping device impedance (low side of the divider) is changing. If the
impedance of the source is very low, the ratio is low, and eventually
the suppressor could not work at all with a zero source impedance. In
contrast, a crowbar-type device effectively short circuits the transient
toward ground, but, once established, this short-circuit will remain
until the current (the surge current as well as any power-follow current
supplied by the power system) is brought to a low level.

The action of voltage clamping can be performed by any device exhibiting
a nonlinear impedance. Two categories of such devices, having the same
effect but operating quite different physical processes, have found an
acceptance in the industry: polycrystalline varistors and
single-junction avalanche diodes. Another technology, using selenium
rectifiers, has been practically eliminated because of the improved
characteristics of modern varistors.

#### Avalanche diodes

Avalanche diodes were initially applied as voltage clamps in the form of
zener diodes, a natural outgrowth of their application as voltage
regulators. Improved construction, specifically aimed at surge
absorption, has made them very effective suppressors. Large diameter
junction and low thermal impedance connections are used to deal with the
pro­blem of dissipating the heat of the surge, inherent in a thin
single-layer junction.

The advantage of the avalanche diode is the possibility of obtaining
quite low clamping voltages and nearly flat volt-ampere characteristics
over its useful power range. Therefore, these diodes are widely used in
low-voltage electronic circuits to protect 5 V or 15 V logic circuits,
as an example. At higher voltages, the problem of the heat generation
associated with single junctions can be overcome by stacking several
lower voltage junctions.

***Characteristics of Avalanche Diodes*** -- Silicon avalanche diodes
are available with characteristics especially tailored to pro­viding
transient suppression. These special diodes must not be confused with
regulator-type Zener diodes although many engineers tend to use the
generic term \"Zener diode.\" May Zeus help them if they were to
misapply a regulator-type Zener, expecting effective protection!

Figure 4 shows a typical family of the V-I characteristics for one
product. The parameter I is the peak of a specified current waveform,
generally the 10/1000 μs double-exponential test waveform preferred by
communication engineers, rather than a 8/20 μs waveform preferred by
power engineers. The V-I curves are remarkably flat from 1 to 20
amperes, with clamping voltages low enough to protect sensitive
electronics.

Manufacturers of avalanche diodes generally rate their devices in terms
of maximum peak power for a specified pulse duration. Even though the
rated power will decrease with increasing pulse duration, it is not a
case of constant total energy (see Figure 5). A two-decade increase in
pulse duration will reduce the power by only, about one decade,
primarily because of increasing heat transfer for longer pulses.

Since the junction is very thin, the capacitance of an avalanche diode
is appreciable. This capacitance can be a concern in high-frequency
circuits where it would produce an undesirable insertion loss. It is
possible to minimize this effect by using a combination with a
low-capacitance ordinary diode in series with the avalanche diode.
Properly packaged and installed avalanche diodes exhibit a quick
response to steep-front pulses, and have been used for NEMP

However, this quick response protection can be completely obliterated by
improper wiring. The effect of lead length, discussed in detail in the
next section, is applicable to any transient suppressor.

#### Varistors

The term varistor is derived from its function as a variable
[resistor]{.underline}. It is also called a voltage-dependent resistor,
but that description implies that the voltage is the independent
parameter in surge protection, an incorrect perception. Two very
different devices have been successfully developed as varistors: 1)
silicon carbide disks have been used for years in the surge arrester
industry and 2) metal oxide varistors are now widely used.

In silicon carbide varistors, as well as in metal oxide varistors, the
relationship between the current flowing in the device and the voltage
appearing across its terminals can be represented approximately by a
power function I = kV^a^. In this equation, the exponent a, can be
considered as a figure of merit: the higher the value of a, the more
effective the clamping. Hence, there has been a race between
manufacturers and specification writers for higher and higher values of
a. We will see, however, that there are practical limits to this race
and that better performance can be obtained at higher current densities
by departing somewhat from large values of the exponent.

In silicon carbide varistors, the physical process of nonlinear
conduction is not completely understood, and the manufacturing of the
material, successful as it is, has remained an art. It appears that the
process takes place at the tips of the grains of silicon carbide which
are held together by a binder. The story is told that this device\'s
action was found accidentally by having a grinding wheel, on a
disorderly work bench, inadvertently connected to an experimental
circuit; for many years the silicon carbide varistors indeed looked like
grinding wheels, each complete with a hole in the center.

Metal oxide varistors depend on the conduction process occurring at the
boundaries between the grains of oxide (typically zinc oxide) grown
during a carefully controlled sintering process. The physics of the
non-linear conduction have been described in the literature \[1-4\]; in
these application notes, we will be more concerned with the behavior of
the varistors as two-terminal electrical components.

***Electrical Characteristics of Varistors*** -- Because the prime
function of a varistor is to provide the nonlinear effect, the other
parameters are generally the result of a tradeoff in design and inherent
characteristics. Electrical behavior of a varistor can be understood
through the equivalent circuit of Figure 6. The major element is the
varistor proper, Rv, whose V-I characteristic is assumed to be the
perfect power law I = kV^a^. In parallel with this varistor, there is a
capacitor, C, and a leakage resistance R. In series with this
three-component group, there is the bulk resistance of the zinc oxide
grains, RS\$ and the inductance of the leads, L.

Under dc conditions (at low current densi­ties, because obviously no
varistor could stand a high energy produced by dc currents of high
density), only the varistor element and the leakage resistance are
significant. Under pulse conditions at high current den­sities, all but
the leakage resistance are significant: the varistor provides low
impedance to the current flow, but eventually the series resistance
produces an upturn in the V-I characteristic; the lead inductance can
produce spurious overshoot problems if it is not dealt with properly;
the capacitance can offer either a welcome additional path for fast
transients or an objectionable loading for high-frequency signals,
depending on the application.

V-I Characteristics \-\-- When the V-I characteristics are plotted on a
log-log graph, the curve of Figure 7 is obtained. It has three regions
as shown, resulting from the dominance of Rp, Rv, Rs as the current in
the device goes from nanoamperes to kiloamperes.

The V-I characteristic is the basic application design tool for
selecting a device in order to perform a protective function. For
successful application, however, other factors, which are discussed in
detail in information available from manufacturers, must also be taken
into consideration.

Some of these factors are:

\* Selection of the appropriate nominal voltage for the line voltage
(Avoid unnecessarily low clamping.)

\* Selection of energy-handling capability (Consideration of the source
impedance, waveshape, and number of occurrences of the transient \[5\].)

\* Heat dissipation (Ambient temperature, steady state and transient
energy.)

\* Proper installation in the circuit (Lead length).

In fact, enough instances of poor \"installation practices have been
observed that a brief discussion of lead effects is quite in order.

When making voltage measurements across a clamping device for evaluating
its performance, one must recognize possible instrumentation artifacts
which require special precautions to avoid errors:

1\. Use two probes in a differential mode to make a measurement directly
at device terminals, and not a single-ended system.

2\. Avoid contaminating the true device voltage by the additional
voltage caused by magnetic coupling into the probes loop.

Commercial oscilloscope preamplifiers offer a wide choice of
differential mode operation, either through an \[add + invert) mode of
two-channel preamplifiers, or through a differential amplifier built
specifically for high common-mode rejection (sometimes at the expense of
bandwidth). Thus, careful attention must be given to these two aspects
of measurements.

The voltage measured by the two probes (or the voltage that would be
applied to a protected load downstream from the device) is the sum of
the actual clamping voltage at the device terminals and a spurious
voltage caused by magnetic coupling. This spurious voltage is induced
into the loop formed by the clamping device leads and the probes (or the
downstream wiring in the case of an actual protective scheme) by the
changing magnetic field of the surge current flowing in the device.

To illustrate this situation, the measurement circuit shown in Figure 8
was set up in the output circuit of a generator producing a 8/20 μs
impulse. The \"device\" was a hollow conductor, with a hole at the
center through which a twisted pair was fed, one wire of the pair
branching out to each end of the conductor, separated by 10 cm. At the
same 10-cm separation, but outside of the hollow conductor, two thin
wires were also soldered, brought to the midpoint of the hollow
conductor and in close contact with the conductor; from the midpoint
outward, they were twisted in the same manner as the inside pair. A
third pair of wires was soldered at the end points of the hollow
conductor, and arranged to form a rectangle, the hollow conductor being
one side of that rectangle. various widths were set up for the
rectangle, and each time the measured voltage was recorded. Figure 9
shows measured voltage vs. radial distance of the opposite side of the
rectangle. The effect is present even for close proximity, and reaches a
saturation beyond 10 cm.

This example shows that one must not only connect the probes as close as
possible to the clamping devices terminals, but also strive to minimize
the area established by the probes close to the devices.

In this case of a low-voltage suppressor, it would be better to solder
short leads to the device terminals, bring them together while hugging
the device, then twist them into a pair and connect the oscilloscope
probes some distance away from the device. The same conclusion is
applicable to the wiring layout in actual hardware: Creating a loop near
the protective device is an invitation to induce additional voltages in
the output of the protective device, thus losing effectiveness (Figure
3).

Hence, when one is making measurements as well as when one is designing
a circuit for a protection scheme, it is essential to be alert to the
effects of lead length (more accurately of loop area) for connecting the
protective devices. This warning is especially important when the
currents are in excess of a few amperes with rise times of less than 1
μs.

# COMPARISONS OF PROTECTIVE DEVICES

#### Linear Versus Nonlinear Devices

When a protection scheme is designed for an electronic system operating
in an environment which is not completely defined, it is often necessary
to make an assumption about the parameters of the transient expected to
occur. In particular, if an error is made in assuming the source
impedance of this transient, the consequences are dramatically different
between a linear device for which the protective level will be affected
in proportion to the error, and a nonlinear device where the protective
level will be affected approximately in inverse ratio to the exponent of
the V-I characteristic.

#### Spark Gap Versus Varistor

The choice between these two devices will be influenced by the inherent
characteristics of the application. Where power-follow may be a problem,
there is little opportunity to apply a simple gap. Where very
steep-front transients occur, the gap alone may let an excessive voltage
go by the \"protected\" circuit until the voltage is limited by
sparkover. Where the capacitance of a varistor is objectionable, the low
inherent capacitance of a gap will seem attractive. If very high energy
levels can be deposited in a varistor (the power dissipated remains as
heat energy trapped in the bulk material) , compared to the lower levels
inherent to the crowbar action of a gap, then a surge arrester of any
type with high capacity connected at the service entrance may be
combined with a varistor of lower clamping voltage installed further in
the circuit. This combined protection, however, requires proper
coordination between the two suppressors \[6\].

## Avalanche Diode versus varistor

The basic performance characteristics of these two devices are similar,
and therefore the choice may be dictated by clamping voltage
requirements (avalanche diodes are available at lower clamping
voltages), by energy-handling capabilities (avalanche diodes are
generally lower in capability per unit of cost), or by packaging
requirements (varistor material is more flexible and does not require
hermetic packaging).

# FAILURE MODES

Failure of electrical components can occur because their capability was
exceeded by the applied stress or because some latent defect in the
component went by unnoticed in the quality control processes. This
situation is well recognized for ordinary components but a surge
protective device, which is no exception to these limitations, tends to
be expected to perform miracles or at least to fail graciously in a
\"fail-safe\" mode.

The term \"fail-safe,\" however, means different failure modes to
different users, therefore it should not be used. To some users,
fail-safe means that the protected hardware must never be exposed to an
overvoltage, so that failure of the protective device must be in the
fail-short mode, even if it puts the system out of operation. To other
users, fail-safe means that the func­tion must be maintained, even if the
hardware is left temporarily unprotected, so that failure of the
protective device must be in the open-circuit mode. It is more accurate
and less misleading to describe a failure mode \"fail-short\" or
\"fail-open,\" as the case may be.

When the diverting path is a crowbar-type device, little energy is
dissipated in the crowbar, as noted earlier. In a voltage-clamping
device, because more energy is deposited in the device, the energy
handling capability of a candidate protective device is an important
parameter to consider in the design of a protection scheme. With
nonlinear devices, an error made in the assumed value of the current
surge produces little error on the voltage developed across the
protective device and applied to the protected circuit (the error,
however, affects directly the amount of energy which the protective
device has to absorb. At worst, when surge currents in excess of the
protective device capability are imposed by the environment, such as an
error made in the assumptions, a human error in the use of the device,
or because nature tends to support Murphy\'s law, the circuit in need of
protection can generally be protected at the price of failure in the
short-circuit mode of the protective device. However, if substantial
power-frequency currents can be supplied by the power system, a
fail-short protective device generally terminates as fail-open when the
power system fault in the failed device is not quickly cleared by a
series overcurrent protective device (a fuse or a breaker).

When there is a need to eliminate a failed protector at the specific
equipment level, insertion of the fuse in the line provides protection
of the equipment (Figure 10a), while insertion of the fuse in series
with the shunt-connected protector provides pro­tection of the function,
albeit with loss of overvoltage protection (Figure 10b). A fuse must be
selected with suitable i^2^t to withstand the effect of repetitive
surges expected to flow in the application.

## CONCLUSIONS

1\. Surge protective devices are available for protecting low-voltage
electronics. Two basic types offer different advantages:

\* Crowbar devices have a high current capability but they would produce
a power-follow, if applied in a power system; their fast switching
action can be a cause of problems if it is not recognized.

\* Voltage clamping devices, either the avalanche junctions or the
varistors, are free from the problems of power-follow and interference
caused by a fast switching, but the energy which is deposited in these
must be recognized for long-term reliability.

2\. Avalanche diodes offer low clamping voltage, which makes them most
suitable for low-voltage, low-power electronics.

3\. Metal-oxide varistors are now available in a wide range of forms,
clamping voltages and energy-handling capacities.

4\. Each of these devices has its own best field of application for
better reliability of the circuits in the not-quite-defined
electromagnetic environment of power and communication systems.

## REFERENCES

1\. Matsuoka, M., Masa Yama, T. and Lida.,Y. \"Nonlinear Electrical
Properties of Zinc Oxide Ceramics,\" Proc. First Conf. Solid State
Devices, Tokyo, 1969, [J. Japan Soc]{.underline}. [Appl.
Phys.]{.underline}, 39 1970, Suppl. p.94.

2\. Harnden, J.D., Martzloff, F.D., Morris, W.G. and Golden, F.B., \"The
GE-MOV varistor \-- The Super Alpha Varistor,\"
[Electronics]{.underline}, 45, no.21, 1972, p.91.

3\. Sakshaug, E.C., Kresge, J.S. and Miske, S.A., \"A New Concept in
Station Arrester Design,\" [IEEE PAS-96]{.underline} No.2, March-April
1977, p.647.

4\. Mahan, G.D., Levinson, L.M., and Philipp, H.R., \"Theory of
Conduction in ZnO Varistors,\" [Report 78CRD205]{.underline}, General
Electric Company, Schenectady, NY, 1978.

5\. Martzloff, F.D., \"Matching Surge Protective Devices to Their
Environment,\" [IEEE IA-21]{.underline} No.1, Jan/Feb 1985, p.99.

6\. Martzloff, F.D., \"Coordination of Surge Protectors in Low-Voltage
Power Circuits,\" [IEEE PAS-99]{.underline} No.1, Jan/Feb 1980, p.129.

**A GLIMPSE AT LONG-TERM EFFECTS OF MOMENTARY** **OVERVOLTAGES**

**ON ZINC OXIDE VARISTORS**

François D. Martzloff and Thomas F. Leedy

National Institute of Standards and Technology

Gaithersburg, Maryland 20899

**ABSTRACT**

Because the prime function of varistors is the diversion of high energy
surges, most of the attention is directed toward selecting the
appropriate device rating to ensure long life under surge conditions.
Some attention is also given to matching steady-state rating of the
device to the power system voltage. However, during abnormal (and not
well defined) power system conditions, the line voltage can reach values
that will cause substantial current in the varistor. Until the effects
of these momentary overvoltages are better identified and understood,
there will be a risk of near-term failure at worst and accelerated aging
at best.

**SELECTION OF VARISTOR RATINGS**

Selection of a varistor rating is essentially a process of matching the
capabilities of the device to the power system environment, while
obtaining the desired low clamping voltage during surge events. Low
clamping voltage of a varistor may be obtained in one of two ways, or
both: low current density at the surge current, which means a relatively
large area, or low intrinsic voltage, which means a relatively thin
disc. Increasing the area of a disc increases the device cost so that
the natural tendency of designers will be to seek the thinner disc
rather that the larger diameter disc. Therefore, this paper addresses
the concerns raised by excessive reduction of disc thickness, that is,
the use of varistors with rated line voltage lower than what prudent
practice would indicate.

In matching a varistor to its environment, the two principal environment
characteristics to be considered are: (1) the surges expected to occur
on the system \-- the primary reason for using the varistor \-- and (2)
the system rated voltage. In the application of surge arresters to
electric utility systems, considerable attention is given to abnormal
power system conditions, as evident in the rating of these arresters
which takes into consideration the maximum continuous operating voltage
(MCOV) permissible for an arrester. \[1\]

In the case of low-voltage electronic circuits, equipment designers
generally rely on the manufacturer\'s rating of a device for the stated
nominal system voltage. This system voltage is defined with some
expected tolerance, such as 15%, on the maximum voltage that can occur.

**Surge ratings -** Specification sheets published by varistor
manufacturers generally include a \"Pulse Rating\" family of curves
showing the number of permissible surges (pulses) as a function of
current amplitude and duration. This pulse rating makes it possible to
select a device of appropriate dimensions \-- that is, current handling
capability \-- to ensure survival in the known or postulated surge
environment \[2\]. Actually, survival means more than the non-occurrence
of a catastrophic failure. For varistors used in electronic equipment,
the criterion for end of life, or total consumption of the Pulse Rating,
is a change of more than 10% in the device nominal voltage. The actual
process leading to this change in device characteristic is not clearly
described by the manufacturers literature, but is generally believed to
involve microscopic damage to the structure.

**Steady-state vs. momentary ratings -** The steady-state rating is
generally determined by the manufacturers, on the basis of in-house
experience. This process is reflected in the wording of an IEEE Standard
on varistors defining \"Rated RMS Voltage\" as follows: *There is no
single test that can determine the* *voltage rating, but rather an
evaluation process taking into consideration* \.... *7hese*
*considerations are within the realm of the manufacturer\'s and user\'s
application* *engineering functions.* \[3\] In plain English, this means
that there is no clear-cut guidance available; rather, designers are
left on their own to make the bard choices. The manufacturers, however,
do publish an Arrhenius plot of mean life versus temperature, implying
high activation energy levels in the processes leading to the end of
life under steady-state line voltage.

Field experience has shown that the expectation of a + 15% maximum
deviation of line voltage is optimistic. The term \"swell\" has been
proposed \[4\] to describe the occurrence of a momentary increase of the
sinusoidal line voltage amplitude, lasting from a few cycles to a few
seconds. During such a swell, the normal standby current of the varistor
will increase in accordance with the varistor power law. For instance, a
20% increase in line voltage (a factor of 1.2) rather than the 15% often
considered as the \"high-line limit\" by designers, with a varistor
exponent of 32, produces an increase of the standby current of 1.2
elevated to the 32th power \-- a 340-fold increase ! This current level
can no longer be considered a benign \"standby current.\"

Thus, the selection process of a varistor must also include proper
attention to the long-term effects of repeated swells. A search of the
literature reveals that no work has been published on these effects.

Perhaps one of the reasons for this lack of information is that very
little specific information has been collected on the severity of swells
occurring in power systems. The electric utilities and their customers
have recently increased their awareness of the so-called *Power Quality*
issue, as a result of concerns over the performance of increasingly
sophisticated electronic equipment supplied by a disturbance-prone power
system. One of the effects of these concerns has been the development of
disturbance monitors with graphics capability. As more and more of these
monitors are applied, the occurrence and characteristics of swells will
become much better known.

Thus, between the well-defined bounds of short-time surges and the less
defined steady-state voltage rating, neither the occurrence nor the
effects of repeated swells are well defined. Faced with this lack of
information, the authors have initiated a joint project of experimental
and theoretical evaluation of the effect of repeated swells on zinc
oxide varistors. The experimental work consists essentially in applying
repeated swells of various amplitudes and durations, seeking to detect
some change in the electrical characteristics of the varistor that would
herald the onset of objectionable degradation. The theoretical work
consists in modeling the behavior of the varistor, essentially its
temperature, in response to a variety of swells. To fully understand the
science involved, however, this work should be complemented by
fundamental studies on the degradation mechanisms leading to any
observed change in the electrical characteristics. One of the purposes
of this paper is to motivate ceramists toward looking into the effects
of swells from the point of view of ceramic structural changes. In the
meantime, the experimental work, supported by modeling to identify the
impact of various levels of swells, will draw the attention of equipment
designers to the pitfalls of selecting excessively thin discs. \[5\]

**MODELING THE EFFECT OF SWELLS**

A circuit analysis model of a metal-oxide varistor was used to
investigate the behavior of varistors when exposed to swells. The model
consists of two parts: The electrical part of the model predicts the
current-voltage behavior of the varistor and the thermal part of the
model predicts the temperature changes caused by the energy deposited in
the varistor during the swell. The electrical and thermal simulations
were performed using a commercial software system which uses behavioral
equations to mimic nearly any physical device or process.

The model computations were performed for a varistor rated at 130 V rms.
Such a varistor is the lowest rating offered by manufacturers for a 120
V circuit. To represent a possible worst case, the varistor
characteristics used for the model computations were set at the lower
limit of the tolerance band. To simplify the model, no allowance is made
for heat loss by convection, the situation that can be expected in a
tight packaging without any encapsulation. Heat losses by radiation are
negligible at the temperatures involved. These postulates provide worst
case results, a justifiable approach when considering reliability.

Figure 1 shows an example of the output plot of the model computations,
for a 206 V peak line voltage (145 V rms, a 20 % swell). The current
exhibits the highly nonlinear response of a varistor, with 20 mA peaks.
The resulting temperature increase steps are also plotted for each
successive pulse, occurring at the 120 pulse-per-second rate. As the
temperature increases, so will heat losses, until a thermal equilibrium
is reached, unless thermal runaway would occur.

Figure 2 shows the temperature rise predicted for the application of
swells at two voltage levels beyond the 20% swell of Figure 1, each with
a duration of 200 s. This 200 s duration was selected because it
approximates the time required to reach thermal equilibrium in this
model, not because of an inference that power system swells last 200 s.
The lower curve shows the temperature rise of approximately 130 oC
expected from 156 V rms (a 30 % swell). The upper curve shows a
temperature rise of nearly 300 oC for 160 V rms (a 33 % swell). These
curves illustrate that a relatively small change in the swell level,
from 156 V to 160 V rms, produces a drastic increase in the power
dissipation and the temperature rise of a varistor. The 300 oC
temperature rise means assured destruction of the varistor, the 130 oC
temperature could mean some accelerated aging, if repeated enough times.
Thus, the cumulative effect of repeated swells needs to be evaluated.
This evaluation is the subject of the experimental work.

**EXPERIMENTS WITH REPEATED SWELLS**

Several factors are known or suspected in the behavior of actual
varistors exposed to repeated stress, complicating the apparent
simplicity of varistor modeling. Aging \-- a change in the varistor
characteristics in response to overstresses \-- is only one of them.
Even when considering only aging, there is no universally recognized
criterion of degradation. Most varistor specifications refer to a 10%
limit in changes of the nominal voltage of the varistor. However, since
this nominal voltage is defined arbitrarily for a 1 mA current
regardless of the varistor cross section, the criterion varies with
varistor sizes. Another proposed criterion, power dissipation at some ac
voltage, is complicated by the fact that the power dissipation changes
with the duration of application of the steady-state as well as swell ac
voltages.

This effect, which permanently changes the characteristics of the
varistor, sometimes called \"formation,\" was quite apparent when
successive swells were applied. During application of one swell, for
instance, the increasing temperature of the varistor is reflected in the
increased peak amplitude of the current, as shown in Figure 3A. However,
when comparing the pattern of current peaks within the first swell to
the pattern of a later swell, after many additional swells (Figure 3B),
the most obvious change is a reduction of the current peaks between the
original swell and the later swell. Thus, a systematic method to account
for that \"formation\" effect must be developed to define a valid
criterion of aging before conclusions can be drawn on quantifying the
aging effect of swells.

**CONCLUSIONS**

Attempts at improving surge protection by selecting varistors with low
voltage ratings increase the risk of premature aging, even early
failure, of varistors exposed to repeated momentary overvoltages
(\"swells\").

The effects of such repeated swells have not been documented by the
manufacturers, although it is a subject of investigation that would
provide useful results for greater reliability. Additional theoretical
and experimental work is necessary to develop guidance useful to prudent
equipment designers.

While the occurrence of these swells has not been well characterized in
the past, the increasing availability of power line disturbance monitors
will improve this characterization in the future. Combined with a better
understanding of the effects of swells, this knowledge will enhance the
reliable application of metal-oxide varistors.

**REFERENCES**

**\[11** Sakshaug, E.C., Burke, **J.J.,** and Kresge, **J.S.,** Metal
Oxide Arresters on Distribution Systems - Fundamental Considerations,
*IEEE Winter Power* *Meeting Paper,* 1989. (To be published in *IEEE
Transactions on Power Delivery.)*

\[21 Martzloff, F.D., Matching Surge Protective Devices to their
Environment, *IEEE-LAS Transactions,* Vol. IA-21, No.1, pp. 99-106,1986.

\[3\] IEEE Standard Test Specifications for Varistor Surge-Protective
Devices, ANSI/IEEE C62.33-1982.

\[4\] Martzloff, F.D., and Gruzs, T.M., Power Quality Site Surveys:
Facts, Fiction and Fallacies, *IEEE-IAS Transactions,* Vol. 24, No.6,
pp.1005-1017,1988.

\[5\] Martzloff, F.D., and Leedy, T.F., Selecting Varistor Clamping
Voltage: Lower is not Better*! Proceedings, EMC Zurich Symposium,* pp.
137-142, 1989.

**Figure 1.** Computed instantaneous temperature increase of a varistor
during two cycles of a 20% swell

**Figure 2.** Computed integrated temperature increases during a 30%
swell and a 33% swell

**Figure 3.** Samples of varistor current during a 16-second swell with
peak voltage at 117%

**What Are the Lights on Your Surge Protector Telling You?**

** **

François Martzloff

National Institute of Standards and Technology

Gaithersburg, Maryland

 

 

Especially in a free market, sensible standards and rea­sonable
regulations have a role to play toward providing useful, reliable and
safe consumer products, such as the now familiar transient voltage surge
suppressors (TVSS). However, the definition of what is sensible and
reasonable has not been stipulated in any TVSS standard. This article
provides a historical perspective of the evolution of how TVSS standards
address the issue of a disconnector, an emerging concept in TVSS design.

 

A sampling of how this concept has been implemented by the TVSS
indus­try, including overcurrent protective de­vices (fuses) and thermal
runaway pro­tection (thermal cut-out), shows a vari­ety of rather
confusing approaches. To illustrate this point, let's consider three
familiar products: fruit cakes, automo­biles, and TVSSs.

 

We will all agree that our military personnel serving in some desolate
for­eign country are entitled to a high-qual­ity product when the
traditional fruitcake is sent to them at holiday time. To ensure that
this happens, as reported tongue-in-cheek by the media a few years ago,
a lengthy specification had been developed for quality control in the
procurement of fruitcakes.

 

Consider now the plight of the tired traveler, arriving at night in an
unfamil­iar airport and renting a car. The agent behind the counter gives
the traveler the keys and the location of the car in the parking lot,
and smiles with good wishes thrown into the deal. Arriving at the
designated spot in the dark parking lot, everyone expects to find that
the steer­ing is located on the left, controlled by means of a wheel, the
engine is started with an ignition key, generally near the steering
column, and gas and brakes are controlled by the right foot. But what
about finding the headlight control after you have closed the door (it's
cold and windy out)? Worse yet, two miles out of the airport, in heavy
traffic, it suddenly starts to rain and the traveler fumbles to find the
windshield wiper control? Now would it not be lovely if there were
standards mandating the lo­cation of these controls?

 

**No Common Meaning to Lights**

 

Coming now to the point of TVSSs, consider the plight of the consumer
who bought a TVSS, and discarded the bubble package or misplaced the
in­struction sheet. This consumer is now trying to find the meaning of
the pilot light, or pilot lights, indicating the state of the device ---
specifically whether pro­tection is in effect or protection is lost. The
original 1985 version of UL Stan­dard 1449 stipulated that a visible or
other indication of the protection status must be provided when an
overcurrent protection has operated. But that is no help to the computer
user who has in­stalled a plug-in suppressor sandwiched between the wall
and the desk. Even if the TVSS were visible, there might be one. two or
even three pilot lights, with or without clear and reliable indication
of their meaning.

 

To place this undesirable situation in perspective, I visited three
local elec­tronics and do-it-yourself stores and bought 17 different
brands of TVSSs. The question was, "How would the user be informed on
the status of protection after a temporary overvoltage caused failure of
the protective element in the TVSS package?" A temporary overvolt­age was
selected as the method of fail­ure because there is growing evidence that
they cause a large number of the failures that occur --- perhaps the
majority ---rather than "large" surges. Although that state­ment could be
the subject of one entire issue of this magazine, it is only men­tioned
here to describe how the failures were obtained.

 

The point of the inquiry was not to describe how the failure occurred
(in particular whether there was a potential hazard, since the devices
were manu­factured before the 1996 Second Edi­tion of UL 1449 became
effective). The point of the inquiry is to find what in­formation is
clearly conveyed to the user after the TVSS has failed, and its
disconnector has operated. Neither the new UL 1449, nor any IEEE or IEC
standards address that practical ques­tion. In fact, the new 1996 version
of UL 1449 no longer stipulates inclusion of a visible indication. As it
turned out, several of the packages among the 17 specimens had very
little information on the significance of the indicating light or how
the indications should be inter­preted. Furthermore, as mentioned
ear­lier, once the bubble package is dis­carded, the information is lost,
because it is unlikely the typical user will safe­guard the back of the
bubble package where the details --- if any --- might have appeared.

 

**SPD Failure Modes**

 

Surge-protective devices (SPDs) components --- switching away from the
jargon 'TVSS' to the official term of 'SPD' approved by the IEEE and IEC
--- like any electrical component, are subject to failure for a variety
of causes in­cluding severe overstress beyond their rating, natural
ageing, premature failure caused by misapplication, etc. The out­come of
such failures ranges from an open circuit to a very low impedance, but
rarely a zero-impedance short cir­cuit. A final open circuit may be
diffi­cult to achieve in an environment of high available fault current.
If achieved, it implies loss of protection against sub­sequent surges. A
zero-impedance con­dition would mean no energy dissipa­tion (no heating)
in the failed SPD com­ponent and thus a relatively benign fail­ure mode
from the point of view of the SPD environment, but that would re­quire
interruption by an upstream overcurrent protection device.

 

In reality, the nature of the failure mechanism, in particular for
metal-ox­ide varistors, (MOVs) is that the imped­ance of the failed SPD,
which is resis­tive, is low, but not zero. Consequently, the power
circuit will deliver to this medium-range resistance a current that
involves significant energy, depending upon the actual resistance of the
failed device and the upstream im­pedance of the power system.

 

If left unabated, this deposited en­ergy will produce severe overheating
with attendant risk of fire or other ob­jectionable failure modes. It is
to prevent such objectionable scenarios that the concept of an SPD
disconnector emerged, as seen in this historical per­spective.

 

The term "disconnector" is a recent addition to the vocabulary of the
SPD industry, and the spell-checker of most word processors flags it as
unknown (as if your processor had never disconnected on you?). Even the
definition of the term has gone through several itera­tions in recent
years.

 

According to emerging standards, the disconnector function is to
"disconnect an SPD from the power system in case of SPD failure." This
clear and simple concept, however, has not been consis­tently implemented
by industry, in par­ticular by the low-voltage SPD industry. In
particular, two different devices --- a fuse or a thermal cut-off ---
can serve toward that end, each covering one type of SPD failure. In
case of SPD failure resulting in a low impedance mode (a few ohms) an
overcurrent occurs, with amplitudes in excess of the SPD rating, so that
a fuse can clear the circuit. In case of a thermal runaway, the
begin­ning of SPD failure, the current is still too low to operate the
fuse, but the con­sequences of failure (overheating) can be prevented by
the operation of a ther­mal cut-out in close proximity to the SPD
experiencing the beginning of overheating.

 

This ambiguity in the type of disconnector design and the lack of
con­sistency or consensus in the interpreta­tion of the function is
apparent in the wide variety of designs found on the market, while
standard-writing bodies are still working on developing an ac­ceptable
consensus over a broad range of applications. One unresolved issue is
whether the user expects protection after failure --- through a complete
disconnect of the load --- or would rather take a chance by maintaining
continuity of operation, but without protection.

 

Standards development generally lags the development of products in a
competitive, fast-moving technology. Low-voltage SPDs are no exception.
To illustrate this point, Table I presents a chronological tabulation of
the evolu­tion of the disconnector concept, from the introduction of
metal-oxide varistors in the mid-seventies, to the latest amend­ment of
the definition of a disconnector in the late nineties.

 

The tabulation shows the reference document and cites, verbatim or in
summary, the implied need --- and eventu­ally the specification --- for a
suitable disconnector function. Inspection of the chronology shows a
clear trend toward mandating a reliable disconnector func­tion, but also
some concessions to ex­isting product practices. Nevertheless, the
disconnector remains the ultimate protection against objectionable
failure modes. As recognition of this essential role of a disconnector
increases, one can expect that standards are more likely to mandate
provision of a reliable and clearly defined disconnector function for
all low-voltage SPDs. It may he easy for an outsider to make judgments
on the apparent chaos of the marketplace and the present standards, but
there are extenuating circumstances because the concept as well as the
implementation of a satisfactory disconnector are not trivial. Some of
the options, advantages and drawbacks are briefly enumerated below.

 

# Disconnector Provisions

 

## *Integral disconnector*

 

-    Disconnects only the shunt-connected SPD

 

-   Design challenge: requirement of not operating for the expected
    surges but of operating promptly upon failure of the SPD element.

>  

-   User information: necessity to SPD disconnector, the details of the
    \\operation and the interpretation of in­dications provided by all
    the different manufacturers do not offer a systematic. user-friendly
    and standardized situation.

 

To illustrate this point, Table 2 pre­sents a summary of the visible
indica­tions on the 17 blind-purchase of TVSSs, after producing their
failure by applying a temporary overvoltage. In this table, it is
postulated that for the typical consumer, ensuring protection of
expensive equipment is preferable, even if it means that power is cut
off for the equipment; in other words, maintaining continuity is
undesirable.

 

There may be some applications where maintaining continuity would be the
first choice. In either case, a clear indication of which state prevails
should be clearly stated. From this table, it is apparent that the two
possibilities --- maintained continu­ity, or maintained protection ---
are offered by different manufacturers. However, the implications of
this choice were not clearly stated on the package, and in two
instances, the indications were in fact misleading. Hopefully, industry
and standards-writing bodies will rem­edy this situation.

 

Without invoking a bureaucratic sys­tem of regulations, it would seem
that a clear consensus among the manufactur­ers would make the informed
end-user much more appreciative of the industry offerings, and provide
enhanced safety for the public.

 

**Conclusions**

 

-   A survey of the low-voltage SPD (TVSS) products currently on the
    market shows not only a variety of disconnector responses after a
    failure, but in some cases, false indica­tion of the protection
    status.

-   Information provided by several manufacturers, on the product
    pack­(ICC (soon to he discarded after pur­chase by the end-user) or on
    the prod­uct itself fails to give unambiguous information on the
    status of load protection after failure of the SPD component.

-   Anecdotal information indicates that devices currently on the market
    might not be provided with disconnector that meets the undefined but
    expected safety in failure modes. The second edition of UL 1449 is a
    very constructive step toward a remedy, although it does not address
    the question of status indication.

-   This uncoordinated --- not to say chaotic --- situation clearly
    points to the need for accelerating the development of an
    industry-wide consensus on the design, operation. and interpretation
    of the indications of low-voltage SPD disconnectors. Hopefully, this
    article will help the process.

  

**Lingering Lead Length Legacies**

**in Surge-Protective Devices Applications**

François D. Martzloff, Life Fellow, IEEE, and Kermit Phipps, Member IEEE

Abstract---Two experiments are reported to show how some lin­gering
inherited misconceptions about the applications of surge-protective
devices (SPDs) can lead to erroneous or cost-ineffective attempts to
address the issue of lead length. The first experiment demonstrates the
fallacy of the "four-terminal SPD" configuration if taken at face value
without additional precautions on lead dress. The second experiment
provides quantitative information on the actual effect of lead length.
With this information, designers and installers can place the issue in a
realistic perspective and avoid unnecessary effort.

**I. INTRODUCTION**

Many references found (inherited) in the literature imply that lead
length is a significant factor to consider for correct installation and
characterization of surge-protective de­vices (SPDs). This perception has
led to the design of "four-terminal" one-port SPDs where the incoming
power must be routed through the device terminals, rather than using an
SPD with simple shunt connection with a pair of leads of unspecified
length. connected at some point of the system. However, both perception
and design miss the point that the problem is actually one of mutual
inductance as well as "lead length." It is indeed important to recognize
that to achieve an optimum surge protec­tion, connecting leads of SPDs
should be as short as possible. Yet not only long lead lengths, but also
significant loop area formed by the connecting leads, will add an
inductive voltage to the protective voltage of the SPDs, degrading their
performance. The perception was further complicated by concerns about
the "speed of response" issue. After a brief review of some old is­sues,
this paper reports the results of two experiments aimed at placing the
lead length issue in the proper perspective.

**II. HISTORICAL PERSPECTIVE**

Concerns about the undesirable performance degradation of improperly
installed surge-protective devices are not new. In the high-voltage
arena, the concepts of "separation effects" and "connecting lead wires"
have been recognized for more than a half-century (Witzke and Bliss,
1950 \[11); (IEEE Std.C62.2\~19872 \[2\]), and the recent IEC 61 643-12
(2002) \[31 is a comprehensive guide for low-voltage applications.\~ In
the low-voltage arena, the situation has been complicated by misguided
attention to the issue of "speed of response" and concerns about
"overshoot" in the application of SPDs, fueled by debates about relative
merits of emerging technologies such as metal-oxide varistors (MOVs) and
silicon avalanche diodes (SADs). The debates on this issue were quite
active soon after introduction of MOVs and SADs in the seventies. At
that time, so many papers were published that any attempt to recite them
entails a severe risk of offending the authors who might be overlooked
in a list of references, so that we will not take that risk here. The
debates have now abated although lingering perceptions or misperceptions
remain.

For MOVs, a typical application manual published in the seventies \[41
addressed both the issues of "speed of response" and "lead length" as
illustrated by Figs. 1 and 2 (overleaf) excerpted from this manual.
Other documents, such as IEEE Std. C62.33-l982 \[5\] attempted to place
speed of response and overshoot in perspective and de-emphasize the
issues, but were not entirely successful, to wit the claims for
nanosecond response that still appear on the wrapping of some commercial
SPD packages. Some justification for the emphasis on speed of response
can be found in historical context, when interest was arising about
ensuring protection against the nuclear elec­tromagnetic pulse, and might
still be valid for some military applications. In more mundane modern
surge protection for today's industrial, commercial, and residential ac
power cir­cuits, the perspective is different. Reality checks and
economic considerations suggest that these concerns might be an overkill
in surge protection for circuits where the propagation of fast surges is
limited \[61 and typical wiring practices do not allow the
sophistication of perfect connections in those applications.

Therefore, in this paper we will not go beyond a brief review of the old
issues of speed of response and overshoot, but will concentrate on
discussing the lead length that seems to be the lingering and
misunderstood issue.

**A. Speed of Response**

For instance, Fig. 1 shows an oscillogram recorded to doc­ument the
response to fast-rising pulses by an MOV disc in­serted in the coaxial
test fixture necessary to observe this sub-microsecond pulse without
test connection artifacts \[7\]. Indeed, the experiment confirmed the
claim of fast speed of response of the intrinsic MOV material, but
practical devices are generally not fitted in coaxial packages---just
what the tutorial experiment reported later in this paper did require.

**B. Overshoot**

In Fig. 1, one might consider that the "TRACE 2" indeed display an
overshoot that may be significant when concerned with nanosecond
response of an MOV. However, for this context of surge mitigation in ac
power circuits, very short picosecond pulses such as that shown in Fig.
1, or the EFT pulse. defined in IEC 61 000-4-4 \[8\] and IEEE
C62.41-1991 \[91 have to travel only a few meters away from their origin
to have their rise time and duration stretched into tens of nanoseconds
or more \[101.

Fig. 2 shows the test fixture arrangements and resulting per­formance for
two current impulses with rise times, respectively, 8 μs and 0.5 μs
\[11\]. These two rise times were not arbitrarily selected---our readers
will readily recognize what became the IEEE Std. 587-1980 waveforms
\[12\]. However, creating and in­jecting the 2.5-kA, 0.5-μs pulse into
the test circuit required great care, to avoid spurious signals,
illustrating again that such conditions rarely, if ever, do occur in
real-world situations \[13\]. The original figure caption of the GE
Transient Manual included in Fig. 2 mentions both lead length and
overshoot, a regrettable mention---with hindsight---because it focuses
attention on lead length, which is not the direct cause of that
"overshoot."

The perceptive reader at that time might have noticed that the test
fixture sketches in the figure correctly emphasized the issue of loop
area, but the caption provided in the original 1978 figure is perhaps
responsible for launching lead length legacies by fo­cusing on lead
length. Small wonder then, that so many lengthy debates and test
specifications have been concerned with lead length rather than the
direct effect of electromagnetic coupling between the two loops involved
in an installation---the injected surge current loop and the protected
equipment loop. Neverthe­less, there is some redeeming tutorial value in
that figure, which we will now briefly discuss.

**C. Electromagnetic Coupling Between Loops**

In Fig. 2, it should be noted that the traces were superimposed by
multiple exposures and are not synchronous, so no attempt should be made
to correlate exactly the timing of the voltage and current traces.
Nevertheless, the inductive coupling mechanism is apparent in that the
maximum voltage occurs at the begin­ning of the current pulse (maximum
di/dt). Observe the large "overshoot" on the right-side oscillogram for
the 22-cm^2^ area loop and the more modest voltage for the 0.5-cm^2^
area loop. As further evidence, the amount of "overshoot" for the two
traces disappears at the point of current peak (di/dt = 0); at that
point, the voltage trace indicates the "true" level of the voltage
limiting action of the MOV disc, free from spurious inductive effects.

In principle, the coupling aspect is simple, as illustrated in Fig. 3,
which shows how the changing magnetic flux created by the flow of surge
current in the circuit at left (Loop "A") induces a voltage in the
circuit at right (Loop "B"). Thus, the load that was expected to be
protected at the limiting level of the SPD actually sees that limiting
voltage augmented by the induced voltage, as shown by the waveforms
drawn in Fig. 3.

IEC 61 643-12 \[3\] also addresses this effect in Annex K. Figure K-6 of
that annex, redrawn here as Fig. 4, is similar to our Fig. 3, but is
relegated to a distant annex. In contrast, the Fig. 10 of the IEC
document (redrawn here as Fig. 5), is found in the main body under the
heading of "Influence of the connecting lead length" (italics ours) and,
thus, appears to support the four-terminal approach by designating it as
"preferred scheme"---but does not qualify it with a discussion of the
possible electromagnetic coupling.

Note also that this connection arrangement still falls under the
category of "one-port SPD" because there is no impedance between the
"input" and "output" terminals, as would be the case for a "two-port
SPD."^4^ So, the idea of "lead length" as a prime and perhaps sole
factor lingers on into the 21st century. This situation is what
motivated us to conduct the experiments reported in this paper.

To quantify the effect of electromagnetic coupling between the two
loops, we conducted a first experiment using a spe­cially-constructed
coaxial SPD to be inserted in a well-defined configuration of the two
loops---the incoming surge loop "A" and load loop "B" as defined in the
schematic circuit represen­tation of Fig. 3.

**III. NEW EXPERIMENTS**

The first experiment addresses the issue of the inductive cou­pling
between adjacent loops, and the misleading perception that a
four-terminal arrangement, such as that shown in Fig. 5, automatically
circumvents the lead length issue.

The second experiment deals with the real issue of lead length for
shunt-connected ("one port") SPDs, and provides some quan­titative
information on the magnitude of the effect as determined by surge
current parameters and, indeed, lead length.

**A. Coupling Between Adjacent Loops**

For this demonstration, a "coaxial MOV" unit was constructed as shown on
Fig. 6, allowing measurement of the "true" limiting voltage of the MOV
without any significant inductive coupling. A 40-mm-diameter MOV disc
was drilled at its center to allow connecting the core conductor of a
coaxial cable to the top elec­trode, with the shield of the cable
connected to the bottom elec­trode. Fig. 6 is a cross-section view of the
device, which has com­plete rotational symmetry. A cap (cross hatch) is
soldered to the top electrode and a sleeve (cross hatch) is soldered to
the bottom electrode. Connections could be made at any point of the cap
and of the sleeve; for the purpose of showing the device in a
cross-sec­tion, terminal connections are represented on the left and on
the right, but are at the same potential. According to the principle of
the "four-terminal SPD," the incoming power supply (presumed to include
a surge) would be connected at the left of the device, and surge-free
(or at least mitigated) power for the protected load would be available
at the right of the device. The "true" MOV voltage resulting from a
surge can then be observed at the end of the coaxial cable. This
structure was only aimed at the demonstra­tion, not as a practical
low-voltage SPD, but was inspired by old discussions on how to make a
distribution arrester consisting of three parallel-connected stacks of
MOV discs arranged at 120° with the power take-off at the center of the
triangle \[14\].

Using this special coaxial MOV, measurements were per­formed on several
simplified configurations of lead geometry, including the so-called
"four-terminal SPD" suggested by some authors and offered in some
commercial packages. Fig. 7 shows the test circuit and definition of the
terms "span" and "w" (width). The test circuit consists of two
well-defined rectangular loops of conductors with the dimensions shown,
mounted on a common sheet of insulating material. On the left, a loop in
which the applied surge current is driven into the coaxial MOV; on the
right, a loop representing the path of the power supply toward the
protected load---in this case, the voltage input of a digital signal
analyzer (DSA). The surge current imposed on the MOV (maintained at the
same level for all the tests by keeping the same setting on the surge
generator) is monitored by a current-viewing transducer and fed to a
second channel of the DSA. Bonding the conductors of the two loops at
"A" and "B" allows obtaining a configuration equivalent to that of a
conventional two-terminal, one-port, shunt-connected SPD.

Figs. 8 and 9 (overleaf) show the results obtained with two different
standard surge waveforms---the Ring Wave and the Combination Wave of
C62.41-1991 \[91 for fixed dimensions (1 m) of the surge loop (left
square in Fig. 7), and a constant width (w = 1 m) but decreasing span of
the measurement loop (right square in Fig. 7). Maximum decrease of the
span would of course be accomplished by twisting the leads that go to
the DSA, immediately from the point of attachment to the MOV. And the
ultimate minimum span is the coaxial cable output from the MOV.

The oscillograms of Fig. 8 (for the relatively high di/dt of a Ring
Wave) show, from left to right, how the inductive voltage can be reduced
by decreasing the "span" of the loop that feeds power (and the mitigated
surge) to the load. For the first of the oscillograms (at left), the two
loops have a portion of the circuit in common by bonding the two corners
of the squares above ("A") and below ("B") the MOV, clearly the worst
possible case, with a relatively large span representing a very poor
lead ar­rangement.

The next oscillogram toward the right shows the case of the "improved"
four-terminal SPD configuration: the resulting pro­tective voltage is
only reduced from 624 to 618 V (the three-digit values being read from
the numerical display of the DSA, not from the oscillograms). That is
hardly an improvement, and it is perhaps below the variability of the
repeated surges and dig­itizing noise.^5^ This negligible difference
between the two con­figurations demonstrates the fallacy of the
four-terminal con­figuration if not accompanied by attention to the lead
dress and resultant electromagnetic coupling, as can be seen in the
oscil­lograms following these first two. Going on toward the right, the
span is progressively decreased with corresponding decrease
(improvement) of the protective voltage, until the ultimate (but
impractical in the field) idealized configuration of the coaxial MOV
with voltage read from the coaxial cable connection (zero span and zero
loop width, w = 0).

The oscillograms of Fig. 9 show, for a Combination Wave, and from left
to right, the worst case of a large inductive cou­pling (made equivalent
to the common lead configuration by bonding the corners "A" and "B" of
the square loops), the four-terminal coaxial MOV arrangement with a span
of 1 m, and the measurement taken at the output of the coaxial cable.

With that gentler rate of rise (a fifth of that of the Ring Wave for the
current levels injected in this experiment), the effect of inductive
coupling is negligible. (The waveforms are quite sim­ilar, only the
DSA-generated peak values overwritten in the os­cillograms show a
difference). Of course, if the amplitude of the surge current were
higher, the effect of the inductive cou­pling would be higher. Because
this inductive effect is linear, in comparison with the nonlinear
voltage-limiting response of the MOV, it will become increasingly
noticeable for higher surge currents. However, there is a limit to the
di/dt rate of change that can be imposed at the sending end of a branch
circuit, be­cause the voltage necessary at the sending end for driving
such a steep current toward an SPD at the far end would cause a
flashover of the wiring devices in the service panel \[15\]).

**B. Shunt-Connected SPD Installations**

The second experiment demonstrates the effect of installing a separate
shunt-type SPD with long connections to a service panel. For practical
situations, it is often postulated that the con­necting leads can be
represented by an inductance in the order of I RH/in, while the
resistance of these leads can be neglected. A connection involving 30 cm
of leads, a typical length for a careful installation of a
shunt-connected SPD, could add sev­eral hundred volts to the limiting
voltage achieved by the SPD itself in cases of high rates of current
changes in the incoming surge.

SPDs packaged as power strips or plug-in inherently provide a two-port
configuration on which the user has no control, but hopefully include
internal wiring configuration that minimizes the lead effect.
Furthermore, SPDs designed for integrated in­stallation, such as a
meter-base SPD or a panel plug-in SPD, offer a minimum of lead length,
if the grounding lead is kept as short as possible (twisting leads to
reduce the loop area or cancel the coupling is not a possible option in
this case). On the other hand, in the case of separate SPDs permanently
connected in shunt but located some distance from the point of
connection, the length of that connection becomes quite significant, as
our second experiment will show (nothing new about that), but also
quantify.

Again, IEC 61643-12 provides useful information on this subject, but its
Fig. 10-c (redrawn here as Fig. 10) only states that twisting leads is
an acceptable alternative when the pre­ferred four-terminal configuration
(Fig. 5) is not possible. In our second experiment that we are about to
describe, the con­nection of the SPD was made with the closely-spaced but
not twisted pair of conductors encased in their plastic jacket. We did
not attempt to twist the leads, being unaware of a potential sig­nificant
reduction of inductance by twisting the closely-spaced pair (unlike
sensitivity to external electromagnetic fields which, indeed, can be
decreased by twisting leads). Instead, we went to another possibly
greater reduction of inductance by using a coaxial cable connection, as
we will describe later.

The experiment was conducted with a circuit as shown in Fig. 11, using a
20-m-long line of typical residential cable (2 + G conductors, plastic
jacket, 2.5 mm^2^ or #12 AWG). The coaxial varistor was connected at the
far end (right), and a Ring Wave was applied at the near end (left).
With a constant generator setting and an unchanged line length, the
surge current could be maintained at a constant amplitude and waveform,
thus allowing direct comparison of the measured voltages. These voltages
were measured for several distances "d" between the varistor and the
point of connection of the probes. In an actual installation, this point
of connection would be the service panel, and the cable length "d"
between that point and the varistor would be the infamous "lead length"
associated with a real-world installation. Fig. 12 (overleaf) shows the
voltage measured for an ideal (but impractical) coaxial MOV, serving as
baseline reference of the "true" limiting voltage, and the voltages
measured for increasing values of the lead length.

Fig. 12 shows, from left to right, the voltages at the point of
connection of the MOY (Vp) for increasing distances between this point
of connection and the shunt-connected SPD, the ide­alized coaxial MOV in
our experiment (upper trace). The lower traces document the constant
value of the peak of the impinging surge current injected into the
circuit. From left to right, com­pared to the baseline (d = 0) for a
measurement made at the coaxial cable output of the MOV, the Vp caption
shows the in­crease in the effective limiting voltage (decreased
performance) that will be seen by the "protected" load. That increase is
al­ready 240 V for just 1rn of connecting leads and that type of
impinging surge.

Some proposals have been made to decrease this adverse ef­fect by using a
coaxial cable to make the connection of the SPD package to the service
panel. For the last oscillogram to the right of Fig. 12, a readily
available 3-rn length of an RG8 coaxial cable replaced the last 3 m of
the plastic-jacket cable. Interpo­lating the readings on plain cable for
d = 2 m and d = 4 m yields 1150 V (for a 3-rn length of plastic jacketed
cable), com­pared to the 960 V obtained for the coaxial cable connection.
While this might be seen as a significant improvement, it might not be
large enough to justify the costs or the complication of a coaxial cable
lead that would have to be installed by an electri­cian unfamiliar with
methods of connecting such coaxial cables within conventional wiring.

The experiments reported here were conducted using real wiring but
contrived configurations to illustrate the points being made. The
applied surges were delivered by a generator pro­ducing "textbook" C62.41
waveshapes. Being injected in an in­ductive circuit, the actual surge
current had a rise time of about 0.8 μs rather than the C62.41 value of
0.5 μs. Close examina­tion of the original laboratory (larger) current
traces of Fig. 12 revealed an actual maximum rate of rise of about 290
A/μs for a peak value of 220 A (about half the 500 A peak value of the
Category B of C62.41). Such a rate of rise, applied to a connection of 1
m, and rule of thumb of 1 μH/m for the cable inductance, should produce
a voltage drop of about 290 V. The difference in the observed voltages
in Fig. 12 for d = 0 its and d = 1 in is 240 V, not quite the 290 V
computed with the rule of thumb inductance value. Given the wide range
of real-world rates of rise, this difference between the experimental
observations and the rule-of-thumb computation does not affect the
conclusions on the order of magnitude of the effect.

In the previously cited paper \[15\], the "gentle toe" concept was
brought up: the theoretical waveforms, such as those used in numerical
computations, can produce unrealistic values for the maximum rate of
rise at the instant of the surge initiation. In the real world, surge
currents do not have their maximum rate of rise at t = 0, but only some
brief time after. Another difference between theory and reality is that
most practical Combination Wave surge generators have an
"undershoot"---a reversal of po­larity in the surge current---after the
theoretical unidirectional impulse. This polarity reversal is further
enhanced by the in­ductance of practical circuits, as opposed to the dead
short pos­tulated in the definition of the short-circuit current of the
Com­bination Wave. Consequently, for modeling purposes, many re­searchers
use a damped sine wave instead of the standard uni­directional wave
(Hasse et al. \[16\]). However, this damped sine wave has its maximum
di/dt at t = 0 (the derivative of the sine is a cosine); hence, the
"gentle toe" idea to reconcile theory and reality.

For instance, Table I, first compiled in the cited paper \[15\], shows
the maximum values of di/dt for three different nominal rise times of a
damped sine wave with a 5 kA peak. It is note­worthy that the
relationships are not linear, and that the max­imum di/dt is greater than
the value that one would obtain by simply dividing the rise time into
the peak value, a fact that is sometimes overlooked in oversimplified
discussions. These more realistic values of di/dt can then be combined
with the empirical 1 μH/m value of the connecting leads to estimate the
degradation resulting from an excessive lead length.

**IV. CONCLUSIONS**

These experiments have clearly demonstrated that focusing exclusively on
the "lead length" for SPD installations can lead to misconceptions or
unwarranted expectations, such as the "four-terminal SPD" configuration.
On the other hand, a quantitative assessment of the effect of long
connections of a shunt-type de­vice will provide useful guidance on
installation practices.

1\) While there is merit in the concept of a four-terminal shunt SPD,
the benefits can be greatly degraded if proper attention is not given to
the lead configuration. Just using a four-terminal device will not
ensure optimum perfor­mance.

2)  Improper installation of a separate shunt-type SPD via a "long"
    connecting cable to the service panel will degrade the performance
    in case of high rates of current changes in the impinging surge (in
    particular those implied for some commercial packages that propose
    ratings of tens of thousands of amperes). In the case of more
    moderate reasonable rates of change, this effect might have been
    somewhat overemphasized in the literature.

3)  Attention should be given to the lead configuration as well as to
    lead length. For instance:

• For a separately-mounted one-port SPD, twisting the leads or using a
coaxial cable between their point of connection to the protected circuit
and the SPD package will reduce the inductive coupling but not greatly
reduce the inductance of the connection. Lead length remains a
significant factor.

• For one-port SPD packages that are mounted inside or on the side of a
panel, an arrangement that pro­vides a minimum of lead length, twisting
leads (if possible) will help reduce inductive coupling.

**REFERENCES**

\[1\] R. L. Witzke and T. J. Bliss, "Coordination of lightning arrester
location with transformer insulation level," AIEE Trans., vol. 69, pp.
964 -- 975, 1950.

\[2\] IEEE Guide for the Application of Gapped Silicon-Carbide Arresters
for Alternating-Current Systems, IEEE Std, C62.2, 1987.

\[3\] IEC 61643-12(2002) Surge protective devices connected to
low-voltage power distribution systems---Selection and application
principles, 2002.

\[4\] Transient Voltage Suppression Manual, 2nd ed. Auburn, NY: General
Electric Company, 1978.

\[5) IEEE Standard Test Specifications for Varistor Surge-Protective
Devices, IEEE Std. C62.33, 1982.

\[6\] F. D. Martzloff and T. F. Leedy, "Electrical fast transients:
applications and limitations," IEEE Trans. Ind. Applicat,, pp. 151 --
159. Jan/Feb. 1990.

\[7\] L. M. Levinson and H. Philipp, Personal communication and
acknowl­edgment, (General Electric Corporate R&D) performed in the
seventies the experiment that produced the oscillogram of Fig. 1, which
was in­corporated in the GE Transient Suppression Manual \[4\].

\[8\] IEC 61000-4-4 (1995) Electromagnetic Compatibility---Part 4:
Testing and measurement techniques---Section 4: Electrical fast
transient burst immunity tests, 1995.

\[9\] IEEE Recommended Practice on Surge Voltages in AC Power Circuits,
IEEE Std. C62.4l, 1991.

\[10\] F. D. Martzloff and P. F. Wilson, "Fast transient tests---trivial
or terminal pursuit?," in Proc. 7th lot. Zurich Symp. Electromagnetic
Compatibility, 1987.

\[11\] F. A. Fisher, "Overshoot---A Lead Effect in Varistor
Characteristics," General Electric Company, Schenectady, NY, Rep. 78CRD,
1978.

\[12\] IEEE Guide for Surges Voltages in Low-Voltage AC Power Circuits,
IEEE Std. 587, 1980.

\[13\] F. A. Fisher, Personal communication and acknowledgment, (General
Electric Corporate R&D) designed the test generator and performed in the
seventies the experiment that produced the oscillograms of Fig. 2, which
were incorporated in the GE Transient Suppression Manual \[4\].

\[14\] R. E. Koch, "Power Line High Energy Surge Arrester for
Application on a 14.4/24.9kV System,", General Electric Rep. CCR-84-04,
1984.

\[15\] A. Mansoor and F. D. Martzloff, "Driving high surge currents into
long cables: More begets less," IEEE Trans. Power Delivers', vol. 12,
pp. 1176---1183, July 1997.

\[16\] P. Hasse, L. Wiesinger, P. Zahlmann, and W. Zischank, "Principle
for an advanced coordination of surge protective devices in low voltage
sys­tems," in Proc. 22^nd^ Int. Conf Lightning Protection, 1994.

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| ## François Martzloff                                                 |
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| ## April 2004                                                         |
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