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Received: 15 October 2002 **RCG-4**

Source: Document 8B/TEMP/119

Working Party 8B

Draft revision of recommendation ITU-R M.1177-2

Summary of revisions

Revisions to this Recommendation include: New information on reference
bandwidth; additional information on correction factors for normalizing
to reference levels required in Recommendation ITU-R SM 1541 and
Appendix 3 of the Radio Regulations; measurements for multiple pulse or
multimode radars; dynamic range of the measurement system; manually and
automatically controlled measurements; direct measurement method
detailed descriptions of procedures and software; gain and noise figure
calibration using a noise diode; measurement of pulse width and pulse
rise/fall times; measurement of unwanted emissions for low frequency
radars (below 50 MHz);

Draft REVISION of RECOMMENDATION ITU-R M.1177-2[^1]

Techniques for measurement of unwanted\
emissions of radar systems

(Question ITU-R 202/8)

(1995-1997-2000)

Summary

This Recommendation provides two techniques for the measurement of
radiated radar unwanted emissions. It should be used to measure the
spurious emission limits in Appendix 3 (Section II) of the Radio
Regulations (RR), or to measure the level of unwanted emissions falling
within the out‑of-band domain.

The ITU Radiocommunication Assembly,

considering

a\) that both fixed and mobile radar stations in the radiodetermination
service are widely implemented in bands adjacent to and in harmonic
relationship with other services;

b\) that stations in other services are vulnerable to interference from
radar stations with unwanted emissions with high peak power levels;

c\) that many services have adopted or are planning to adopt digital
modulation systems which are more susceptible to interference from radar
unwanted emissions;

d\) that under the conditions stated in a) through c), interference to
stations in other services may be caused by a radar station with
unwanted emissions with high peak power levels;

e\) that ITU-R SM.329 specifies the maximum values of unwanted emissions
in the spurious emission domain from radio transmitters;

f\) that Recommendation ITU-R SM.1541 specifies the generic limits for
unwanted emissions in the out-of-band domain,

recommends

**1** that measurement techniques as described in Annex 1 be used to
provide guidance in quantifying radiated unwanted emission levels from
radar stations operating above 400 MHz;

**2** that measurement techniques as described in either Annex 1 or
Annex 2 are used, as appropriate based upon radar design, to provide
guidance in measuring radiated unwanted emission levels for radar
stations operating between 50 MHz and 400 MHz;

**3** that measurement techniques described in Annex 2 be used to
provide guidance in quantifying radiated unwanted emission levels from
radar stations operating below 50 MHz;

**4** that results of such usage of this Recommendation be reported to
ITU-R, in order to determine any limitations in the techniques, e.g.
tolerances of measurements and repeatability over the required frequency
ranges, so that confidence can be established in the measurement
methods.

Annex 1

Measurement of unwanted emissions of radar systems\
as detailed in Recommends 1 and 2

# 1 Introduction

Techniques have been developed in response to § 1 of Question ITU-R
202/8. Two techniques known as the direct and indirect methods are
described.

The direct measurement method is recommended and measures unwanted
emissions from all radars including those that preclude measurements at
intermediate points within the radar transmitters. Examples include
those which use distributed-transmitter arrays built into (or
comprising) the antenna structure.

The indirect method separately measures the components of the radar and
then combines the results. The recommended split of the radar is to
separate the system after the "rotating joint" (Ro-Jo) and thus to
measure the transmitter output spectrum at the output port of the RoJo
and to combine it with the measured antenna gain characteristics.

.

# 2 Reference Bandwidth

For radar systems, the reference bandwidth (B~ref~) used to define
unwanted emission limits (Rec. ITU-R SM.329, Rec. ITU-R SM.1541 and
Appendix 3) should be calculated for each particular radar system. For
the four general types of radar pulse modulation utilized for
radio­navigation, radiolocation, acquisition, tracking and other
radiodetermination functions, the reference bandwidth values are
determined using the following formulas:

• for fixed-frequency, non-pulse-coded radar, one divided by the radar
pulse length, in seconds (e.g. if the radar pulse length is 1 s, then
the reference bandwidth is 1/1 s  1 MHz);

• for fixed-frequency, phase coded pulsed radar, one divided by the
phase chip length, in seconds (e.g. if the phase coded chip is 2 s long,
then the reference bandwidth is 1/2 s  500 kHz);

• for frequency modulated (FM) or chirped radar, the square root of the
quantity obtained by dividing the chirp bandwidth in MHz by the pulse
length, in seconds (e.g. if the FM is from 1 250 MHz to 1 280 MHz or
30 MHz during the pulse of 10 s, then the reference band­width is
(30 MHz/10 s)1/2 1.73 MHz).

• for radars operating with multiple waveforms the reference bandwidth
is determined empirically from observations of the radar emission. The
empirical observation is performed as follows: The measurement system
receiver is tuned to one of the fundamental frequencies of the radar, or
is tuned to the centre frequency within the chirp range of the radar.
The measurement system bandwidth is set to the widest available value,
and the received power level from the radar in this bandwidth is
recorded. The measurement bandwidth is then progressively narrowed, and
the received power level is recorded as a function of the bandwidth. The
end result is a graph or table showing measured power as a function of
measurement system bandwidth. The required bandwidth is the smallest
bandwidth in which the full power level is still observed and the
reference bandwidth can be calculated from a knowledge of the impulse
response of the measurement receiver using the factor MBR as described
below. If a reduction in power level is observed immediately, then the
widest available bandwidth should be used.

In all cases, where the bandwidths above are greater than 1 MHz, then a
reference bandwidth (B~ref~) of 1 MHz should be used.

# 3 Measurement bandwidth and detector parameters

The measurement bandwidth, B~m~, is defined as the impulse bandwidth of
the receiver and is greater than the IF bandwidth, B~if~, (sometimes
referred to as resolution bandwidth for spectrum analysers). The
measurement bandwidth (B~m~) may be derived from the following equation:

$B_{m} = B_{\text{if}} \times \text{MBR}$ The measurement bandwidth
ratio (MBR) needs to be determined for the measurement receiver being
used. MBR is approximately 3/2 for a --3 dB IF bandwidth Gaussian filter
as typically used in many commercial spectrum analyser receivers. (Note
in some instruments the IF bandwidth is defined at the --6 dB point).

An appropriate receiver IF bandwidth should be selected to give one of
the following recommended measurement bandwidths.

  ------------- -- -------------------------------------------------------
  Measurement      (1/*T *) for fixed-frequency, non-pulse-coded radars,
  bandwidth        where *T* = pulse length.\
  B~m~^1^          (E.g. if radar pulse length is 1 s, then the
                   measurement bandwidth should be 1/(1 s)  1 MHz.)

                   (1/*t *) for fixed-frequency, phase-coded pulsed
                   radars, where *t* = phase-chip length. (E.g. if radar
                   transmits 26-s pulses, each pulse consisting of
                   13 phase coded chips that are 2-s in length, then the
                   measurement bandwidth should be  1/(2 s)  500 kHz.)

                   (*B~c~*/*T *)½, for swept-frequency (FM, or chirp)
                   radars, where *B~c~* = range of frequency sweep during
                   each pulse and *T*: pulse length. (E.g. if radar sweeps
                   (chirps) across frequency range of 1 250-1 280 MHz ( 30
                   MHz of spectrum) during each pulse, and if the pulse
                   length is 10 s, then the measurement bandwidth should
                   be:  ((30 MHz)/(10 s)) ½  MHz 1.73 MHz. In accordance
                   with note 1 a measurement bandwidth close to but less
                   than or equal to 1 MHz should be used in this example.
  ------------- -- -------------------------------------------------------

  ------------- -- -------------------------------------------------------
                   The result of a measurement is as follows: For radars
                   operating with multiple waveforms the measurement
                   bandwidth is determined empirically from observations
                   of the radar emission. The empirical observation is
                   performed as follows: The measurement system receiver
                   is tuned to one of the fundamental frequencies of the
                   radar, or is tuned to the centre frequency within the
                   chirp range of the radar. The measurement system
                   bandwidth is set to the widest available value, and the
                   received power level from the radar in this bandwidth
                   is recorded. The measurement bandwidth is then
                   progressively narrowed, and the received power level is
                   recorded as a function of the bandwidth. The end result
                   is a graph or table showing measured power as a
                   function of measurement system bandwidth. The
                   appropriate measurement

                   bandwidth will be the bandwidth where the first
                   reduction of the full power level is observed. If a
                   reduction in power level is observed immediately, then
                   the widest available measurement bandwidth should be
                   used.

  Video            Measurement system bandwidth
  bandwidth        

  Detector         Positive peak
  ------------- -- -------------------------------------------------------

^1^ Note, in all cases if the above derived measurement bandwidth is
greater than 1 MHz then a measurement bandwidth B~m~ close to 1 MHz
should be used.

**3.1 Measurements within the out-of-band** **(OOB)** **domain**

Within the OOB domain, the limits given in Recommendation ITU-R SM.1541
are defined in dB~pp~. This is a relative power measurement and an IF
bandwidth leading to a measurement bandwidth less than the reference
bandwidth should be used. Even if the measurement bandwidth is less than
the reference bandwidth no correction needs to be done, since both the
peak of the spectrum and the data points within the out-of-band (OOB)
domain are measured using the same measurement bandwidth (B~m~).

Measurements should generally be made using a bandwidth that is close to
but less than the specified reference bandwidth. This approach will
minimize the measurement time but it also causes some broadening of the
measured spectrum. Thus in marginal situations, where measurement of the
true close in spectrum shape may be important, it is recommended that
the close-in region within the OOB domain should be re-measured using a
maximum bandwidth of 0.2/T or 0.2/t as appropriate.

## 3.2 Measurements within the spurious domain

### 3.2.1 Correction of the measurement within the spurious domain

Where the measurement bandwidth (B~m~) is less than the reference
bandwidth (B~ref~), a correction factor needs to be applied to the
measurements conducted within the spurious domain to express the results
in the reference bandwidth. Then the following correction factor should
be applied:

Spurious level (B~ref~)=Spurious level (measured in B~m~)+10xLog(B~ref~
/ B~m~)

Note: This correction factor should be used except where it is known
that the spurious is not noise-like, where a factor between 0 and 20
log(B~ref~/B~m~) may apply and may be derived by measurements in more
than one bandwidth.

### 3.2.2 Correction of the measurement data to the PEP

Within the spurious domain, the limits given in Appendix 3 are defined
in a reference bandwidth (B~ref~) with respect to the peak envelope
power (PEP). Data recorded in dB~pp~ within the spurious domain must be
referenced to the PEP (and not the spectrum peak observed in dB~pp~).

The PEP is approximated using the following correction formula:

For CW and phase coded pulses:

PEP = P~meas~+20xLog(B~pep~/ B~m~), for B~pep~ \> B~m~.

For swept-frequency (FM or Chirp) pulsed radars:

PEP = P~meas~+10xLog(B~c~ /(B~m~^2^xT)), for (B~m~^2^T)/B~c~ \< 1

Where:

###### PEP = peak envelope power

###### P~meas~ = spectrum peak power measured in B~m~ 

B~pep~= the bandwidth calculated according to the following:

• for fixed-frequency, non-pulse-coded radar, one divided by the radar
pulse length, in seconds (e.g. if the radar pulse length is 1 s, then
B~pep~ is 1/1 s  1 MHz);

• for fixed-frequency, phase coded pulsed radar, one divided by the
phase chip length, in seconds (e.g. if the phase coded chip is 2 s long,
then B~pep~ is 1/2 s  500 kHz);

-- for frequency modulated (FM) or chirped radar, the square root of the
quantity obtained by dividing the chirp bandwidth in MHz by the pulse
length, in seconds (e.g. if the FM is from 1 250 MHz to 1 280 MHz or
30 MHz during the pulse of 10 s, then B~pep~ is (30 MHz/10 s)1/2
1.73 MHz).

-- for radars operating with multiple waveforms B~pep~ is determined
empirically from observations of the radar emission. The empirical
observation is performed as follows: The measurement system receiver is
tuned to one of the fundamental frequencies of the radar, or is tuned to
the centre frequency within the chirp range of the radar. The
measurement system bandwidth is set to the widest available value, and
the received power level from the radar in this bandwidth is recorded.
The measurement bandwidth is then progressively narrowed, and the
received power level is recorded as a function of the bandwidth. The end
result is a graph or table showing measured power as a function of
measurement system bandwidth. The required bandwidth is the smallest
bandwidth in which the full power level is still observed and B~pep~ can
be calculated from a knowledge of the impulse response of the
measurement receiver using the criteria described below. If a reduction
in power level is observed immediately, then the widest available
bandwidth should be used. The corrections described in section 3.2 are
illustrated graphically in Figure 1.

FIGURE 1.

Graphical illustration of the correction described in section 3.2

As can be seen in Figure 1, the OOB mask and the measured spectrum have
been referenced to the equivalent PEP level by using the factor
20log(B~pep~/B~m~). The figure shows that the measured spurious is
shifted upwards by an amount equal to the correction factor described in
section 3.2.1 (here taken as10log(B~ref~/B~m~)). In this example, a
measurement bandwidth of 100 kHz was chosen only for illustrative
purposes, even though a bandwidth close to 1 MHz is recommended in this
case. Also for illustrative purposes, the mask is shown offset in
frequency as permitted in Recommendation ITU-R SM.1541.

# 4 Measurements for multiple pulse or multimode radars

For radars with multiple pulse waveforms, the B~-40~ dB bandwidth should
be calculated for each individual pulse type and the maximum B~40~ dB
bandwidth obtained shall be used to establish the shape of the emission
mask (see Rec. ITU-R SM.1541, Annex 8).

For radars with multiple pulse width settings, that can be selected
individually, the setting which results in the widest calculated B~-40~
dB bandwidth (see Rec. ITU-R SM.1541, Annex 8) should be used. Emission
measurements only need to be carried out for this pulse width setting.

For radars using elevation beam scanning, measurements normally need
only be made in the azimuth plane.

# 5 Dynamic range of the measurement system

The measurement system should be able to measure levels of unwanted
emissions as given in Appendix 3. To obtain a complete picture of the
spectrum especially in the spurious emissions domain, it is recommended
to be able to measure levels of emissions 10 dB below the levels given
in Appendix 3.

For a high level of confidence in the results, the measurement dynamic
range of the system should be significantly higher than the required
range of measurement (margin (2) in Figure 2).

The link between the required range of measurement and the recommended
dynamic range of the measurement system is given in Figure 2.

  ------------ ---------- ----- --------------------- ---- --- ---------------------------------- ------------------ --- ----- ----------------- ---- ---------- -------------------- --------------- -------- -- --------- --
                                Levels to measure                                                                                                                                     **Measurement                         
                                                                                                                                                                                      system dynamic                        
                                                                                                                                                                                      range**                               

                                                                                                  **F**. radar                                                                                                    **(2)**   

                                                                                                                                                                                                                            

                                                                                                                                                                                                                            

  \-\-\--                       \<\<\-\--                      -\>                                \<\--B**~n~**-\>       \<-   \-\-\-\-\-\--                     \<\-\--spurious---                                         
  spurious                      O.O.B.\-\-\-\-\-\--                                                                            O.O.B.\-\-\--\>                                                                              
  \-\-\--\>                                                                                                                                                                                                                 

                                                                                                                                                                                                                            

                                                                                                                                                                                                                  **(1)**   

                                                                                                                                                                                                                            

  Spurious                                                                                                                                                       Spurious limit                                             

  limit (App                                                                                                                                                     (App 3)                                                    
  3)                                                                                                                                                                                                                        

  10 dB        Margin                                          \<\-\-\-\-\--B**~-40~**\-\-\--\>                                                                  10 dB                                Margin                

                                                                                                                                                                                                                  **(2)**   

                                                                                                                                                                                                                            

               boundary                                                                                                                               boundary                                                              
  ------------ ---------- ----- --------------------- ---- --- ---------------------------------- ------------------ --- ----- ----------------- ---- ---------- -------------------- --------------- -------- -- --------- --

**(1):** Recommended range of measurement

**(2):** Margin

Figure 2

Relation between the required range of measurement and the\
recommended dynamic range of the measurement system

NOTE - It should be noted that Recommendation ITU-R SM.329, recommends
under category B limits, more stringent limits than those given within
Appendix 3 in some cases. This should be taken into account when
evaluating the required range of measurement and the recommended dynamic
range of the measurement system.

# 6 Direct methods

Two direct methods described below can be used to measure unwanted
emissions (out-of-band and spurious) from radar systems. The first
method is manually controlled and the second method is automatically
controlled. These two methods, have been used to measure the emission
characteristics of radar systems operating at frequencies up to 24 GHz,
transmitter output powers of several megawatts, and equivalent
isotropically radiated power (e.i.r.p.) levels in the gigawatt range.
Taking safety aspects into account, these methods may also be carried
out in an anechoic chamber.

## 6.1 Measurement environment conditions

1 Regarding the measurement distance, either near field or far field
measurements can be made. Variation of the peak received signal should
be made less than 3 dB using the absorber when the receiving antenna is
moved λ×D/2H horizontally or vertically away from the point where
maximum signal is received (H: height of the transmitting point, D:
measurement distance, λ: transmitting wave length).

2 Regarding the measurement site, it is preferable to locate the
transmitting and receiving antennas in a fairly high position such as on
towers. Note that the height should be determined considering the
vertical beam width of the radar and measurement antennas, and no
reflective objects should be between the antennas.

## 6.2 Measurement hardware and software

Block diagram of the type of measurement system required for the two
direct methods are shown in Figure 3(manual method) and Figure 4
(automatic method). The first element to be considered in the system is
the receive antenna. The receive antenna should have a broadband
frequency response, at least as wide as the frequency range to be
measured. A high‑gain response (as achieved with a parabolic reflector)
is usually also desirable. The high gain value permits greater dynamic
range in the measurement; the narrow antenna beamwidth provides
discrimination against other signals in the area; the narrow beamwidth
minimizes problems with multipath propagation from the radar under
measurement; and spectrum data collected with a parabolic antenna
require a minimum of post-measurement correction, as discussed in the
next paragraph. The antenna feed polarization is selected to maximize
response to the radar signal. Circular polarization of the feed is a
good choice for cases in which the radar polarization is not known *a
priori*. The antenna polarization may be tested by rotating the feed (if
linear polarization is used) or by exchanging left and right-hand
polarized feeds, if circular polarization is being used.

Corrections for variable antenna gain as a function of frequency should
be considered. Antenna gain levels are usually specified relative to
that of a theoretically perfect isotropic antenna, in units of dBi. The
effective aperture of an isotropic antenna decreases as 20log(f), where
f is the frequency being measured. This means that, if the measurement
antenna has a constant effective aperture (that is, has an isotropic
gain that increases as 20log(f)), no corrections for variable antenna
gain need be performed. This requirement is met by a theoretically
perfect parabolic reflector antenna, and is one of the reasons that such
an antenna is preferred for a broadband radar spectrum measurement.

Conversely, to the extent to which the gain of the measurement antenna
deviates from a 20log(f) curve (including a less-than-ideal parabolic
antenna), the resulting measurements must be corrected for such
deviation.

The cable connecting the measurement antenna to the measurement system
should also be considered. A length of low-loss radio frequency (RF)
cable (which will vary depending upon the circumstances of measurement
system geometry at each measurement radar site) connects the antenna to
the RF front-end of the measurement system. As losses in this piece of
line attenuate the received radar signal, it is desirable to make this
line length as short, and as low-loss, as possible.

![](media/image2.wmf)

Figure 3

Block diagram for measurement of radiated unwanted emissions\
from radars using the manually controlled direct method

![](media/image3.wmf)

Figure 4

Block diagram for measurement of radiated unwanted emissions from
radars\
using the automatically controlled direct method

The RF front-end is one of the most critical parts of the entire
measurement system. It performs three vital functions. The first is
control and extension of measurement system dynamic range through the
use of variable RF attenuation. The second is bandpass filtering
(preselection) to prevent overload of amplifiers by high-amplitude
signals that are not at the tuned frequency of the

measurement system. The third is low-noise preamplification to provide
the maximum sensitivity to emissions that may be as much as 130 dB below
the peak measured level at the radar fundamental. Each of these sections
in the RF front end is considered below.

The RF attenuator is the first element in the front-end. It provides
variable attenuation (e.g. 0‑70 dB) in fixed increments (e.g. 10
dB/attenuator step). Use of this attenuator during the measurement
extends the dynamic range of the measurement system by the maximum
amount of attenuation available (e.g. 70 dB for a 0‑70 dB attenuator)

### 6.2.1 Manually controlled measurement system

The manually controlled measurement consists to sweep across the
spectrum in fixed increments (equal to the span value). At each
frequency sweep, the attenuator is adjusted to keep the radar peak power
within the dynamic range of the other elements in the measurement system
(often the front‑end amplifier and the spectrum analyser log amplifier
are the limiting elements). With the front-end RF attenuator properly
adjusted at each sweep, a measurement of the radar power at that
frequency is performed.

A bandpass filter manually controlled can be used at this place to avoid
overloaded preamplifier (and thus gain-compressed), if it is necessary
to measure very low spurious (i.e., level fundamental emissions - level
spurious \> instantaneous measurement dynamic)

The final element in the RF front-end is an LNA. An LNA installed as the
next element in the signal path after the preselector. The low-noise
input characteristic of the LNA provides high sensitivity to
low-amplitude spurious radar emissions, and its gain allows for the
noise figure of the rest of the measurement system (e.g. a length of
transmission line and a spectrum analyser).

The sensitivity and dynamic range of the measurement system are
optimized by proper selection of LNA gain and noise figure
characteristics. It is desirable to minimize noise figure while
providing enough gain to allow for all measurement circuitry after the
LNA (essentially the RF line loss after the front end, plus the noise
figure of the spectrum analyser circuitry). Ideally, the sum of the LNA
gain and noise figure (which is the excess noise produced by the LNA
with a 50-ohm termination on its input) should be approximately equal to
the noise figure of the remaining measurement system.

Typical spectrum analyser noise figures are 25‑45 dB (varying as a
function of frequency), and transmission line losses may typically be
5-10 dB, depending upon the quality and the length of the line. As a
result of variation in measurement system noise figure as a function of
frequency, a variety of LNAs used in frequency octaves (e.g. 1-2 GHz,
2-4 GHz, 4-8 GHz, 8-18 GHz, 18‑26 GHz, and 26-40 GHz) may be required.
Each LNA can be optimized for gain and noise figure within each
frequency octave. This also helps match LNAs to octave breaks between
various YIGs (e.g. 0.5-2 GHz, 2-18 GHz, etc.). Use of an LNA after the
preselector (and, if required, a cascaded LNA at the spectrum analyser
input) may reduce the overall measurement system noise figure to about
10‑15 dB. This noise figure range has been found to be adequate for the
measurement of broadband radar emission spectra over a range as large as
130 dB.

The remainder of the RF measurement system is expected to be essentially
a commercially available spectrum analyser or a spectrum analyser with a
preselector or a selective receiver. Any equipment, which can receive
signals over the frequency range of interest, can be used.

### 6.2.2 Automatically controlled measurement system

The key to using the RF front end attenuator, as shown in Figure 3,
effectively in a radar measurement is to tune the measurement system in
fixed-frequency increments (e.g. 1 MHz), called steps, rather than to
sweep across the spectrum, as is more conventionally done with manually
controlled spectrum analysers. At each fixed-frequency step, the
attenuator is adjusted to keep the radar peak power within the dynamic
range of the other elements in the measurement system (often the
front-end amplifier and the spectrum analyser log amplifier are the
limiting elements). With the front-end RF attenuator properly adjusted
at each step, a measurement of the radar power at that frequency is
performed. In this way, a nominal 60 dB dynamic range for the
measurement system is extended by as much as 70 dB, to a total resulting
dynamic range of 130 dB. To minimize measurement time, this attenuator
and the stepped-frequency measurement algorithm that it necessitates can
be controlled by computer.

The next element in the front end, the tunable bandpass filter
preselector is necessary if it is needed to measure low-power spurious
emission levels at frequencies that are adjacent to much higher-level
fundamental emissions (e.g. 130 dB below fundamental). For example, it
may be necessary to measure spurious emissions from an air traffic
control radar at 2 900 MHz that are at a level of\
--120 dBm in the measurement circuitry, while the fundamental emission
level is at + 10 dBm and is only 150 MHz away in frequency (at 2 750
MHz). The measurement system requires an unattenuated low-noise
amplifier (LNA) to measure the spurious emission at 2 900 MHz, but the
amplifier will be overloaded (and thus gain-compressed) if it is exposed
to the unattenuated fundamental emission at 2 750 MHz. For this reason,
attenuation that has frequency‑dependence is required in the front-end
at a position before the LNA input. In practice, this tunable bandpass
filtering is effectively provided by varactor technology (below 500 MHz)
and by yttrium-iron-garnet (YIG) technology (above 500 MHz). The
applicable filters may be procured commercially, and should be designed
to automatically track the tuned frequency of the measurement system.

The final element in the RF front-end is an LNA. An LNA is installed as
the next element in the signal path after the preselector. The low-noise
input characteristic of the LNA provides high sensitivity to
low-amplitude spurious radar emissions, and its gain allows for the
noise figure of the rest of the measurement system (e.g. a length of
transmission line and a spectrum analyser).

Considerations for the sensitivity and dynamic range of the measurement
system, as well as for typical spectrum analyser noise figures, are the
same as stated in clause 4.2.1.

Another option for LNA configuration is one in which LNAs are cascaded.
The first LNA is placed between two stages within the YIG or varactor
bandpass preselector filter. It has a low noise figure, but only enough
gain to allow for the insertion loss of the second YIG stage. A second
(possibly lower‑performance) LNA is placed immediately after the YIG.
This option will provide somewhat lower overall system noise figure
because the second stage of the YIG is allowed for by the first

LNA. However, this option may require more advanced design and
engineering modifications to the preselector filter than an
administration may deem practical.

A third option for the measurement system LNA configuration, and one not
requiring any redesign or retrofitting of the front end preselector
filter, is to place a lower-gain LNA in the front end and a second LNA
at the spectrum analyser signal input. The first LNA is selected to have
very low noise figure and just enough gain to allow for the RF line loss
and the noise figure of the spectrum analyser LNA. The spectrum analyser
LNA, in turn, is selected for a gain characteristic that is just
adequate to allow for the spectrum analyser's noise figure in the
appropriate frequency range of the radar measurement. This set of two
cascaded LNAs may be more easily acquired than a single, extremely
high-performance LNA, and will typically be less susceptible to overload
as the 1 dB compression points can be expected to be higher than those
for individual high-performance LNAs.

The remainder of the RF measurement system is expected to be essentially
a commercially available spectrum analyser. Any spectrum analyser which
can receive signals over the frequency range of interest, and which can
be computer-controlled to perform the stepped-frequency algorithm, can
be used. As noted above, the high noise figure of currently available
spectrum analysers must be allowed for by low‑noise preamplification if
the measurement is to achieve the necessary sensitivity to observe most
spurious emissions.

The measurement system can be controlled via any computer which has a
bus interface (GPIB or equivalent) that is compatible with the computer
controller and interface card(s) being used. In terms of memory and
speed, modern PC‑type computers are quite adequate. The measurement
algorithm (providing for frequency stepping of the spectrum analyser and
the preselector, and control of the front-end variable attenuator) must
be implemented through software. Some commercially available software
may approach fulfilment of this need, but it is likely that the
measurement organization will need to write at least a portion of their
own measurement software. While the development of software requires a
significant resource expenditure, practical experience with such systems
has shown such an investment to be worthwhile if radar emission
measurements are to be performed on a frequent and repeatable basis.

Data may be recorded on the computer's hard drive or on a removable
disk. Ideally, a data record is made for every 100‑200 measurement
steps, so as to keep the size of data files manageable, and to prevent
the loss of an excessive amount of data if the measurement system
computer or other components should fail during the measurement.

## 63 Measurement system calibration

### 6.3.1 Manual direct method

The manually controlled method requires either calibration of all the
measuring components individually or of the whole measuring set with a
calibrated generator (substitution method).

### 6.3.2 Automatic direct method

The measurement system is calibrated by disconnecting the antenna from
the rest of the system, and attaching a noise diode to the RF line at
that point. A 25 dB excess noise ratio (ENR, where ENR = (effective
temperature, ^o^K, of noise diode/ambient temperature, ^o^K)) diode
should be more than adequate to perform a satisfactory calibration,
assuming that the overall system noise figure is less than 20 dB. The
technique is standard Y‑factor measurement, as described in Appendix II
to Annex 1, with comparative power measurements made across the
spectrum, once with the noise diode on and once with the noise diode
off.

The noise diode calibration results in a table of noise figure values
and gain corrections for the entire spectral range to be measured. The
gain corrections may be stored in a look-up table, and are applied to
measured data as those data are collected. Appendix II to Annex 1
describes the calibration procedure in more detail.

The measurement antenna is not normally calibrated in the field.
Correction factors for the antenna (if any) are applied in
post-measurement analysis.

## 6.4 Measurement procedure

### 6.4.1 Manual method

Appendix I to Annex 1 describes the direct method in detail; this
section provides a summary of the method.

Prior to measurement, a spectrum analyser is used to detect the presence
of signals not emitted by the radar : if there are emissions corrupting
the measurement, appropriate filters must be used.

###### Max - Hold function

+---------------+---+----------------------------------------------------+
| > Spectrum    |   | lowest frequency to be measured. (E.g. if radar    |
| > analyser    |   | center frequency is 3 050 MHz, but the spectrum is |
| > center      |   | to be measured across 2‑6 GHz, then initial        |
| > frequency:  |   | spectrum analyser center frequency would be 2      |
|               |   | GHz.)                                              |
+---------------+---+----------------------------------------------------+
| > Spectrum    |   | 10, 20, 50, 100, or 500 MHz.                       |
| > analyser    |   |                                                    |
| > frequency   |   |                                                    |
| > span        |   |                                                    |
+---------------+---+----------------------------------------------------+
| > Spectrum    |   | automatic sweep time                               |
| > analyser    |   |                                                    |
| > sweep time  |   |                                                    |
+---------------+---+----------------------------------------------------+
| > Time        |   | record signal during a minimum of 3 radar beam     |
|               |   | rotation intervals. (E.g. if radar rotates at 40   |
|               |   | r.p.m., or 1.5 s per rotation, then duration       |
|               |   | should be  3\*1.5 s; 4.5 s would be a reasonable   |
|               |   | selection.)                                        |
|               |   |                                                    |
|               |   | Record signal for a sufficient time for spectrum   |
|               |   | to form. Radar antenna may be held stationary and  |
|               |   | aligned for the maximum measurement system         |
|               |   | response.                                          |
+---------------+---+----------------------------------------------------+

*(NOTE* *-* *The setting of the spectrum* *analyser* *sweep time and the
signal record duration should be validated).*

The second measurement point is taken by tuning the measurement system
to the next frequency band to be measured. This frequency is optimally
equal to the first measured frequency band plus the measured span.

In the case where the measurement instrument is a selective receiver,
the measurement is done point by point according to the recommended
bandwidth.

### 6.4.2 Automatic method

Appendix I to Annex 1 describes the direct method in detail; this
section provides a summary of the method. In addition to the parameters
listed in § 2, the spectrum analyser should be set up as follows:

+---------------+---+----------------------------------------------------+
| > Spectrum    |   | lowest frequency to be measured. (E.g. if radar    |
| > analyser    |   | centre frequency is 3 050 MHz, but the spectrum is |
| > centre      |   | to be measured across 2‑6 GHz, then initial        |
| > frequency:  |   | spectrum analyser centre frequency would be 2      |
|               |   | GHz.)                                              |
+---------------+---+----------------------------------------------------+

+---------------+---+----------------------------------------------------+
| > Spectrum    |   | 0 Hz. (Analyser is operated as a time-domain       |
| > analyser    |   | instrument.)                                       |
| > frequency   |   |                                                    |
| > span        |   |                                                    |
+---------------+---+----------------------------------------------------+
| > Spectrum    |   | radar beam rotation interval. (E.g. if radar       |
| > analyser    |   | antenna rotates at 40 r.p.m., or 1.5 s per         |
| > step time   |   | rotation, then step time should be 1.5 s; 2 s      |
|               |   | would be a reasonable selection.) For frequency    |
|               |   | agile radars or radars with vertical scanning      |
|               |   | antenna beams, the step-time may have to be        |
|               |   | several antenna rotation periods. For these more   |
|               |   | complex radar systems, the step time should be     |
|               |   | determined empirically.                            |
+---------------+---+----------------------------------------------------+

With the radar antenna beam scanning normally, and with the measurement
system set up as described above, the first data point is collected. A
data point consists of a pair of numbers: measured power level and the
frequency at which the power level was measured. For example, the first
data point for the above measurement might be 93 dBm at 2 000 MHz. The
data point is collected by monitoring the radar emission at the desired
frequency, in a frequency span of 0 Hz, for an interval (step time)
slightly longer than that of the radar antenna rotation period, or for a
longer step time for complex radar systems. This time-display of the
radar antenna beam rotation will be displayed on the spectrum analyser
screen. The highest point on the trace will normally represent the
received power when the radar beam was aimed in the direction of the
measurement system. That maximum received power value is retrieved
(usually by the control computer, although it could be written down
manually), corrected for measurement system gain at that frequency, and
recorded (usually in a data file on magnetic disk).

The second measurement point is taken by tuning the measurement system
to the next frequency to be measured. This frequency is optimally equal
to the first measured frequency plus the measurement bandwidth (e.g. if
the first measurement was at 2 000 MHz and the measurement bandwidth
were 1 MHz, then the second measured frequency would be 2 001 MHz). At
this second frequency, the procedure is repeated: measure the maximum
power received during the radar beam rotation interval, correct the
value for gain factor(s), and record the resulting data point.

This procedure, which consists of stepping (rather than sweeping) across
the spectrum, continues until all of the desired emission spectrum has
been measured. The stepping process consists of a series of individual
amplitude measurements made at predetermined (fixed-tuned) frequencies
across a spectrum band of interest. The frequency change between steps
is optimally equal to the measurement system IF bandwidth. For example,
measurements across 200 MHz of spectrum might use 200 steps at a 1 MHz
step interval and a 1 MHz IF bandwidth. The step interval may be set
wider in the spurious emission domain to expedite the overall
measurement. However, at frequencies that are integral multiples (e.g.,
2, 3, 4) of the fundamental radar emission, the maximum step interval
should again be about equal to the measurement system IF bandwidth.

The measurement system remains tuned to each frequency for a specified
measurement interval. The interval is called step-time, or dwell. The
dwell time for each step is specified by the measurement system
operator, and is normally slightly longer than the radar beam scanning
interval.

Computer control of the measurement system is desirable if this process
(step, tune, measure, correct for gain, and repeat) is to be performed
with efficiency and accuracy. In order to correctly measure the peak of
the fundamental emission it may be required to use a smaller step
interval of the order of half or less of the measurement bandwidth over
this region.

The stepped time technique is required to enable the insertion of RF
attenuation at the front-end of the measurement system as the
frequencies approach the centre frequency (and any other peaks) of the
radar spectrum. This ability to add attenuation on a frequency-selective
basis makes it possible to extend the dynamic range available for the
measurement to as much as about 130 dB, if a 0‑70 dB RF attenuator is
used with a measurement system having 60 dB of instantaneous dynamic
range. This is of great benefit in identifying relatively low-power
spurious emissions. To achieve the same effect with a swept-frequency
measurement, a notch filter could be inserted at the centre frequency of
the radar, but there would be no practical way to insert a notch filter
for all the other high-amplitude peaks that might occur in the spectrum.

It is *important* to provide adequate bandpass filtering at the
front-end of the measurement system, so that strong off-frequency signal
components do not affect the measurement of low-power spurious
components.

These measurements may be performed without the radar antenna being
rotated, provided that the directions of both maximum fundamental
emission and any unwanted emission are known.

### 6.4.3 Indirect method

Figure 5 illustrates a recommended component separation for the Indirect
method. In this Indirect method, where unwanted emissions are measured
at the Rotating-Joint and then, combined with the antenna
characteristics measured separately at distances of 5 m and 30 m with
appropriate far-field correction, the procedure is:

*Step 1*: Make measurements of a radar transmitter emissions at the
Rotating-Joint (Ro-Jo) with a feeder (as shown in Figure 6).

*Step 2*: Then make separate measurements of a radar antenna maximum
gain at the emission frequencies found in step 1. Here, measurements are
made at the distances of 5 m for frequencies below 5 GHz and 30 m for
frequencies above 5 GHz (as shown in Figure 7).

*Step 3*: Correct the measured gains with an appropriate correction
factor (using a software program or model of the known antenna
performance. In the simplest cases it may be possible to use the
software programme, given in Appendix 3, for the frequencies, at which
the emissions were observed in Step 1).

*Step 4*: Finally, Steps 1 and 3 are combined to obtain the effective
e.i.r.p. radiation at the observed unwanted emission frequencies.

![](media/image4.wmf)

Figure 5

Typical radar system

![](media/image5.wmf)

Figure 6

Measurement at the rotating-joint port

#### 6.4.3.1 Methods of measurement and problems associated with a waveguide

There are two main problems in measuring the transmitter output power
spectrum. The one is accessing the higher frequency components of the
transmitted spectrum without distortion and; the other is measuring very
low level emissions in the presence of the fundamental transmitting
pulse of perhaps 60 kW peak power.

In any waveguide, the propagation mode, TE~10~, can be measured using a
calibrated measuring system. The characteristic of such a system must be
such that it attenuates the powerful fundamental signal sufficiently to
protect the measuring equipment, whilst at other frequencies offers a
negligible attenuation and energy is being measured in the TE~10~ mode.

It should be recognized that, the spurious frequency emissions of the
transmitter output could be in higher order modes and this possibility
should be considered when setting up the measurement system. For simple
radars however this will rarely be of significance as such higher order
modes are generally trapped in a waveguide to coaxial adaptor, or in
antenna feeder and the Ro-Jo connecting to the radar antenna. (i.e.,
waveguide to coaxial adaptors are only designed to couple energy in
TE~10~ mode).

#### 6.4.3.2 The measurement system for the measurement of unwanted emissions in a waveguide

This measuring system allows the measurement of low levels of emissions
accurately in the presence of high power radar pulses.

The main components of the system are a notch filter and a set of
waveguide tapers, from WG 10 to smaller waveguide sizes, to cover the
whole frequency range of interest. The notch filter comprises of a
straight WG 10 waveguide with absorbent elements inside, which
attenuates the fundamental signal while at other frequencies it offers
negligible attenuation. To achieve the required attenuation to protect
the measuring equipment, and to measure emissions at higher frequencies,
linear tapers are used at the output of the notch filter.

The waveguide taper is a high pass filter and thus rejects, by
reflecting back, signals below the cut off frequency. If a taper had
been used directly at an output port of a radar transmitter, the
fundamental would have been reflected back into the transmitter causing
an undesirable mismatch. But with the taper after the notch filter the
reflected signals are absorbed a second time. Thus the return loss at
the fundamental frequency is typically 34 dB, which is low enough to
avoid frequency pulling of the magnetron.

Frequencies above the cut off are transmitted through the transitions
and into the measuring equipment. If possible, a short waveguide
section, should be included to prevent coupling of evanescent modes
between a taper and a waveguide to coaxial transition.

#### 6.4.3.3 Results of measurement at the Ro-Jo port

The measurement technique comprises an exploratory search of a frequency
band of interest to locate and tag significant spurious emissions by
frequency, followed by a revisit to each noted emission for detailed and
accurate measurement of maximum amplitude of that emission.

#### 6.4.3.4 Measurement uncertainty in a waveguide

The system has a measurement accuracy of 1.3 dB across the frequency
band 2 to 18.4 GHz for the waveguide port. Total uncertainties with a
confidence level of not less than 95% can be calculated to be 3.4 dB for
the waveguide port including the spectrum analyser.

#### 6.4.3.5 Measurement of antenna gain characteristic at measured emission frequencies

This indirect method recommends that near-field measurements be made on
the antenna on an open area test site (OATS) at distance of \[5 m\] for
frequency below 5 GHz and 30 m for frequencies above 5 GHz. Correction
factors are then applied to correct the measurement to an equivalent far
field gain, which provide an acceptable correlation with the far field
gain. A typical measurement arrangement is shown in Figure 7.

![](media/image6.wmf)

Figure 7

Near field gain measurement arrangement for 5 m and 30 m distances

#### 6.4.3.6 Near field gain measurement procedure for 5 m and 30 m distances

The measurement of maximum gain of the antenna under test (AUT) shall be
carried out at spurious and out-of-band frequencies measured or
identified, using the method specified in § 4.3. At each measured, or
identified, emission frequency, the gain of AUT shall be maximized by
first rotating through 360% and then further maximized by moving the
test horn up, or down. The gain of the AUT is obtained by measuring
e.i.r.p. at each distance with a known level of power into the AUT at
each frequency of interest. Equations (1) and (2) show details of
calculations to arrive at the equivalent far field gain, *Ga*, of the
AUT from the measured spectrum analyser level, *S*.

*Ga* of the AUT (dBi) measured e.i.r.p. (dBm) -- *Pinput* (dBm) *Gc*
(dB) (1)

Measured e.i.r.p. (dBm) *S* (dBm) 20 log
$\left( \frac{4pd}{\lambda} \right)$(dB) -- *Gr* (dBi) (2)

where:

*Ga*: equivalent far field gain of the AUT (dBi)

*Pinput*: power input into the AUT (dB)

*Gc*: gain correction factors for 5 m and 30 m distances, which can be
calculated for the AUT using a software program specified in Appendix 1

S: measured spectrum analyser level (dBm)

Gr: gain of the receiving test horn antenna (dBi)

d: measuring distance (m)

: wavelength of a frequency of interest (m).

#### 6.4.3.7 Gain correction and reduction factors

The software program given in Appendix 1 gives the far field correction
factors from a near field measurement for a very simple case. The
program derives the correction factor for each distance at the frequency
of interest by considering the phase changes of the received wave across
the linear antenna. (At near distances the wave front is spherical and
not linear.) Therefore, it can be used to infer the maximum antenna gain
at infinity from a near field measurement.

An important point to bear in mind is that the antenna gain pattern is
not addressed. It must be noted that at spurious frequencies the
electrical length of the antenna is different from the mechanical
length; it may well be much shorter. This is due to the different
illumination pattern of the antenna length at frequencies other than the
designed frequency. Thus a more complex software model or data derived
using the direct method may be required to achieve accurate results in
such cases.

#### 6.4.3.8 Near field gain measurement uncertainty with the applied correction factors

The worst-case measurement uncertainty can be calculated to be  6 dB,
which includes, uncertainties due to a spectrum analyser, test horn
gain, cable loss and source and site imperfection. Total uncertainties
with a confidence level of not less than 95% can be calculated to be ±
4.2 dB.

The derivation of the correction factors for these distances assumes the
AUT radiating aperture to be constant at all frequencies.

#### 6.4.3.9 Producing a radar transmitter emission spectrum as an e.i.r.p. by combining measured emissions and antenna gain characteristic

The technique used to obtain a maximum value for omnidirectional
e.i.r.p. is to add, for each emission frequency, the maximum power
generated by radar transmitter (dBm), to the maximum directional gain
(dBi) from the AUT. This means one only has to characterize the AUT at
frequencies at which the radar transmitter emissions were observed.

The effects of the AUT mismatch are considered to be taken into account
automatically in the measurements of gain, because the test equipment is
matched to 50 , the nominal impedance of the coaxial connectors and the
emissions are measured in the 50 measuring receiver.

#### 6.4.3.10 Summary

The indirect method, which is cost effective in time and facilities, is
sensitive enough to allow measurement of low level emission values with
a reasonable accuracy and repeatability. Furthermore, it can be used in
all weather conditions and the measurement frequency range can easily be
extended to 40 GHz or higher. It can also usefully be used in
conjunction with the direct method to assess incremental changes in a
given radar system that has been previously measured

Appendix I

(to Annex 1)

Direct method detailed description of procedures and software

The direct method assumes that the following conditions can be met:

1\) The far field radiation zone of a radar can be accessed by a
measurement system as described in the body of this Annex;

2\) Unwanted feed-through of radar signals directly into the measurement
system hardware (i.e. bypassing the measurement system antenna) can be
minimized to a sufficiently low level to ensure that measurement results
are accurate.

The direct method does not require that the radar operation be
coordinated with the measurement system, although in some cases
cooperative operation may be beneficial in expediting the measurement.

###### The direct method process is as follows:

Step 1. Determine a measurement location

The measurement location should be within or as nearly as possible to
the radar main radiation beam. For surface search radars and some other
radar types, this may be relatively easy, as the radar beam will sweep
across the surface, and the measurement system need only be placed
within this area. For many air search radars, however, the main beam
does not directly illuminate the ground. For these radars, the
measurement system should be located within the maximum coupling zone on
the surface. This zone may be determined by tuning the measurement
system to the radar fundamental frequency and then driving the
measurement system in a vehicle from a position close to the radar to a
position far (on the order of a few kilometres) from the radar. The
measurement system is used to monitor received signal level as a
function of position. This can be done by running a spectrum analyser in
a zero frequency span with a sweep time of 500 s, and watching the peak
level every few seconds when the radar sweeps past the vehicle. The
result is a time display that shows the maximum coupling location(s).

Any place within the maximum coupling zone should be adequate. In
practice, this zone has been found to begin no closer than about 0.75 km
from air search radars, and to extend to no further than about 2 km from
the same radars. There is usually no sharply defined point where maximum
coupling occurs, but rather a broad zone within these limits.

The question of multipath should be considered. Multipath effects have
been observed very rarely. When they have been observed, it has been in
cases in which the radar and the measurement system were separated by
calm, smooth water surfaces. In other cases, irregular intervening
terrain and the use of parabolic reflector antennas by the measurement
system minimize multipath effects to an extent that makes them
negligible. Multipath effects can be checked by repeating the radar
measurement at a second location and comparing the results from the two
measurement locations. Multipath is also believed to be minimized by
raising the measurement antenna on a telescoping mast to a height of
about 10 m above the ground. This also provides a better line-of-sight
between the radar and the measurement system.

Step 2. Set up the measurement system and check for unwanted
feed-through signals

The measurement system is configured with a parabolic reflector antenna
at the top of a 10 m mast (optional), or at a height of at least a few
metres above the ground, to avoid multipath effects and provide
reasonably good line-of-sight propagation. The measurement system should
be tuned to the radar fundamental frequency or maximum emission
frequency, if it is chirped or frequency-hopping.

It is necessary to check for unwanted feed-through (i.e. the unwanted
reception of radar energy within the measurement equipment, bypassing
the measurement antenna). Feed-through is checked by disconnecting the
measurement antenna and terminating the input line with a 50 Ω load. If
feed-through is present, the following options may be exercised:

1\) check to ensure measurement equipment racks (if any) are sealed;

2\) check connectors for firm fittings;

3\) move the radar measurement system to an alternative location, in
which the measurement equipment is shielded from the radar by buildings
or foliage, and in which the antenna is raised above these obstacles on
the telescoping mast;

4\) move the radar measurement system to a larger distance from the
radar.

A well-designed measurement system should minimize the possibility of
unwanted feed-through.

Step 3. Determine radar emissions parameters

The parameters that are most critical to determine before the
measurement begins are beam scanning interval and effective emission
bandwidth. Beam scanning interval and other characteristics are acquired
by tuning the spectrum analyser in a zero span mode and a sweep time
interval of several seconds, and then observing the beam scanning of the
radar.

Determination of the emission bandwidth is accomplished as described in
the main body of this Annex, with the spectrum analyser tuned to the
radar fundamental frequency in a zero span mode, and the IF and video
bandwidths initially set to their widest available values. The IF
bandwidth is then reduced each time the radar beam swings past the
measurement system, and the bandwidth at which the received power level
drops is noted. This is the widest available measurement bandwidth that
is less than the radar emission bandwidth. This will be the measurement
bandwidth used, unless circumstances such as a need to observe the radar
in a particular receiver bandwidth dictate otherwise.

Additional radar emission parameters that should be noted are: pulse
repetition rate, pulse jitter (if any), pulse stagger (if any), and
pulse width. The first three of these parameters may be measured on an
oscilloscope connected to the spectrum analyser's video output. The RF
pulse width (50% voltage points) and rise time (10‑90% voltage points)
should be measured with a peak power meter or suitable wideband RF
detector diode, operated in the square-law region. It should be properly
matched to an oscilloscope having a sufficient bandwidth to enable
display of the pulse waveform without distortion associated with limited
detector bandwidth.

Step 4. Calibrate the measurement system

Manually controlled direct method:

• The manually controlled method requires calibration of all the
measuring components individually or calibration of the whole measuring
set.

Automatically controlled direct method:

See Appendix II of this Annex. Noise diode calibration is recommended,
although alternative methods using signal generators can be used.

Step 5. Configure measurement system software ( automatic method only)

The measurement software must be configured to the desired start
frequency (MHz), stop frequency (MHz), step size (MHz), step interval
(MHz), IF bandwidth (MHz), video bandwidth ( IF bandwidth), detector
(positive peak), spectrum analyser reference level (usually --10 dBm),
initial attenuation at the start frequency (usually zero dB), and
additional data as location, such as radar name, project name for the
measurement, etc.

Step 6. Check for linearity during the measurement

It is critical to maintain the integrity of the measurement by checking
for linearity as the measurement progresses. When measuring, both at the
fundamental frequency and in the spurious emissions, system linearity
should be checked by periodically inserting 10 dB of RF attenuation at
the rf front end, ahead of the LNA. The result should always be a 10 dB
drop in measured signal level. If other than a 10 dB drop is observed
either front-end overload or unwanted feed‑through may be occurring.
Good system design will minimize these potential problems. If they do
occur, it may alternatively be necessary to either take additional steps
to shield the measurement system, or else to move to another location,
as described in Step 2, above.

Step 7. Measure the radar in more than one IF bandwidth (recommended but
not required)

It may be useful to measure radar emissions in several bandwidths. Such
measurements provide an unequivocal indication of the variation in
measured radar power as a function of receiver bandwidth at any given
frequency in the spectrum.

#######  

Appendix II

(to Annex 1)

Gain and noise figure calibration using a noise diode

Measurement system calibration should be performed prior to every radar
emission spectrum measurement. As measurements are performed, gain
corrections may be added automatically to every data point. For
measurement system noise figures of 20 dB or less, noise diode Y-factor
calibration (as described below) may be used. This Appendix describes
the theory and procedure for such calibration.

![](media/image7.png)figure 8

Lumped component diagram of noise diode calibration

The noise diode calibration of a receiver tuned to a particular
frequency may be represented in lumped-component terms as shown in
Figure 8. In this diagram, the symbol represents a power-summing
function that linearly adds any power at the measurement system input to
the inherent noise power of the system. The symbol g represents the
total gain of the measurement system. The measurement system noise
factor is denoted by nf, and the noise diode has an excess noise ratio
denoted as enr. (In this Appendix, all algebraic quantities denoted by
lower-case letters, such as \"g,\" represent linear units. All algebraic
quantities denoted by upper case letters, such as \"G,\" represent
decibel units).

Noise factor is the ratio of noise power from a device, n~device~(W),
and thermal noise,

$\frac{n_{\text{device}}}{\text{kTB}}$

where k is Boltzmann\'s constant (1.38·10^23^J/K), T is system
temperature in Kelvin, and B is bandwidth in hertz. The excess noise
ratio is equal to the noise factor minus one, making it the fraction of
power in excess of kTB. The noise figure of a system is defined as 10
log (noise factor). As many noise sources are specified in terms of
excess noise ratio, that quantity may be used.

In noise diode calibration, the primary concern is the difference in
output signal when the noise diode is switched on and off. For the noise
diode = on condition, the power, P~on~(W), is given by:

$p_{\text{on}} = \left( \text{nf}_{s} + \text{enr}_{d} \right)´\text{gkTB}$

where nf~s~ is system noise factor and enr~d~ is the noise diode enr.

When the noise diode is off, the power, P~off~(W), is given by:

$p_{\text{off}} = \left( \text{nf}_{s} \right)´\text{gkTB}$

The ratio between P~on~ and P~off~ is the Y factor:

$y = \left( \frac{p_{\text{on}}}{p_{\text{off}}} \right) = \frac{\left( \text{nf}_{s} + \text{enr}_{d} \right)}{\text{nf}_{s}}$

$Y = \text{10}\text{log}(y) = \text{10}\text{log}\left( \frac{p_{\text{on}}}{p_{\text{off}}} \right) = P_{\text{on}} - P_{\text{off}}$

Hence the measurement system noise factor can be solved as:

$\text{nf}_{s} = \frac{\text{enr}_{d}}{y - 1}$

The measurement system noise figure is:

$\text{NF}_{s} = \text{10}\text{log}\left( \frac{\text{enr}_{d}}{y - 1} \right) = \text{ENR}_{d} - \text{10}\text{log}(y - 1) = \text{ENR}_{d} - \text{10}\text{log}\left( \text{10}^{\frac{Y}{\text{10}}} - 1 \right)$

Hence:

$g = \frac{p_{\text{on}} - p_{\text{off}}}{\text{enr}_{d} \times \text{kTB}}$

$G = \text{10}\text{log}\left( p_{\text{on}} - p_{\text{off}} \right) - \text{10}\text{log}\left( \text{enr}_{d} \times \text{kTB} \right)$

or

$G = \text{10}\text{log}\left( \text{10}^{\frac{P_{\text{on}}}{\text{10}}} - \text{10}^{\frac{P_{\text{off}}}{\text{10}}} \right) - \text{ENR}_{d} - \text{10}\text{log}\left( \text{kTB} \right)$

In noise diode calibrations, the preceding equation is used to calculate
measurement system gain from measured noise diode values.

Although the equation for NF~s~ may be used to calculate the measurement
system noise figure, software may implement an equivalent equation:

$\text{nf}_{s} = \frac{p_{\text{off}}}{\text{gkTB}}$

$\text{NF}_{s} = \text{10}\text{log}\left( p_{\text{off}} \right) - \text{10}\text{log}\left( \text{gkTB} \right)\  = \ P_{\text{off}} - G - \text{10}\text{log}\left( \text{kTB} \right)$

And substituting the expression for gain into the preceding equation
yields:

$\text{NF}_{s} = P_{\text{off}} + \text{ENR}_{d} - \text{10}\text{log}\left( \text{10}^{\frac{P_{\text{on}}}{\text{10}}} - \text{10}^{\frac{P_{\text{off}}}{\text{10}}} \right)$

The gain and noise figure values determined with these equations may be
stored in look-up tables. The gain values are used to correct the
measured data points on a frequency-by-frequency basis.

Excluding the receive antenna, the entire signal path is calibrated with
a noise diode source prior to a radar spectrum measurement. A noise
diode is connected to the input of the first RF line in place of the
receiving antenna. The connection may be accomplished manually or via an
automated relay, depending upon the measurement scenario. The noise
level in the system is measured at a series of points across the
frequency range of the system with the noise diode turned on. The noise
measurement is accomplished with the IF bandwidth set to 1 MHz and the
video bandwidth set to 1 kHz. The noise diode is then turned off and the
system noise is measured as before, at the same frequencies. The
measurement system computer thus collects a set of P~on~ and P~off~
values at a series of frequencies across the band to be measured. The
values of P~on~ and P~off~ are used to solve for the gain and noise
figure of the measurement system in the equations above.

Appendix III

(to Annex 1)

Measurement of Pulse Width and Pulse Rise/Fall Times

# 1 Introduction

This Annex is intended to provide guidance for measuring radar pulse
parameters needed in applying the emission mask for the out-of-band
(OOB) domain. Recommendation ITU-R SM.1541, Annex 8, address unwanted
emissions in the OOB domain for radar systems. To determine the
necessary bandwidth, B~n~, and the 40 dB bandwidth, B~-40~, the pulse
width, t, and the rise time, t~r~, of pulsed radars must be
measured[^2].

Pulse width, t, is measured at the 6-dB points (50% voltage points) of a
radar pulse. The rise time, t~r~, or fall time, t~f~, is measured
between the 0.9 dB and 20 dB (10%-90% voltage) points on a pulse\'s
leading or trailing edge, respectively. For coded pulses, t~r~ and t~f~
are the rise/fall time of a sub-pulse. If sub-pulses are not
discernable, then it may be assumed that t~r~ is 40% of the time to
switch from one phase or sub-phase to the next.

For some radar designs, pulse width and rise or fall time may be
measured via a hardline connection to a directional coupler. However,
radiated pulse characteristics may differ somewhat from those measured
from directional couplers. Moreover, some radar designs do not provide a
directional coupler. For these radars, pulse width and rise or fall time
can be measured via radiated energy if the measurement system has
sufficient bandwidth (that is, exceeding (1/t~r~)). A potential
impediment to measuring pulse width via radiation is the effect of
multipath energy, which causes a stair-step fall-off on the trailing
edge of each radiated pulse. This effect can be minimized by the use of
a parabolic reflector antenna on the measurement system. If the effect
of multipath can be suppressed enough for the first trailing edge stair
step to occur more than 6 dB below the nominal pulse level, then a
radiated measurement of pulse width is possible if the bandwidth
requirement is met for (1/t~r~).[^3] A broadband diode detector is
required to achieve sufficient bandwidth.

# 2 Measurements for Conventional Radars

## 2.1 Hardline coupled pulse measurements

For hardline-coupled measurements of pulse characteristics, the
measurement setup is shown in Figure 9. A coaxial cable of appropriate
impedance is connected between the directional coupler output and the
input of a wideband (bandwidth exceeding (1/ t~r~)) crystal detector. A
variable attenuator (0-70 dB, for example) is inserted between the
coupler and the detector. Prior to connecting the detector, the
attenuator is initially set to a sufficiently high level as to protect
the crystal from damage.[^4] The maximum permissible detector input
level may be assumed to be +20 dBm, if other data are lacking.

FIGURE 9

Block diagram schematic for measuring radar pulse width and rise time
(or fall time) parameters via a hardline connection to a directional
coupler

The detector output is connected to an oscilloscope having bandwidth
that exceeds (1/ t~r~). Impedances should be matched appropriately; most
modern oscilloscopes having selectable input impedance values. 50 ohms
is typically correct. DC coupling should be used on the oscilloscope
input.

The oscilloscope is adjusted to display and record[^5] radar pulse
envelopes. The setting of the variable attenuator is noted by
measurement personnel.

Next, the line from the radar directional coupler is disconnected from
that device. It is reconnected to the output of a calibrated signal
generator that is capable of producing pulses of approximately the same
width as measured from the radar. The signal generator output is
adjusted to generate an amplitude response on the oscilloscope that is
the same for both envelopes, preferably about +10 dBm.

With this adjustment made, the response of the crystal detector can be
calibrated as follows. The signal generator output is decreased by,
successively, 0.9 dB, 6 dB, and 20 dB.[^6] At each of these levels,
vertical markers are placed on the measured pulse envelope. The
resulting time intervals between the vertical markers provide the pulse
width (delta between the 6-dB points), the rise time (delta between the
0.9 dB and 20 dB points on the leading edge), and the fall time (delta
between the 0.9 dB and 20 dB points on the trailing edge).

## 2.2 Radiated coupled pulse measurements

For radars that do not incorporate directional couplers, pulse
characteristics can only be measured through radiated measurements.
Figure 10 shows the measurement hardware configuration for measuring
radiated pulses.

FIGURE 10

Block diagram schematic for measuring radar pulse width and rise time
(or fall time, if shorter) parameters via radiated pulses

###### The procedure that should be used is as follows:

a\) Position the measurement system at a location with clear
line-of-sight path to the radar transmitter antenna, and as close as
possible without suffering degradation to the measurement system
performance (e.g., feed-through), losing power from the radar by passing
underneath the main beam, or being within the near-field distance of
either the radar antenna or the measurement antenna;

b\) Use a high-gain antenna (e.g., a 1-meter diameter or larger
parabolic at microwave frequencies) on the measurement system to receive
pulses from the radar at the highest possible amplitude and to
discriminate against signals from other transmitters;

c\) At the measurement antenna input, install a bandpass filter that
will pass the radar fundamental-frequency energy and that has bandwidth
that exceeds (1/t~r~ ) of the radar pulses to be measured. Following the
bandpass filter, install a diode detector that has bandwidth and rise
time response speed that exceeds (1/t~r~ ) of the radar pulses to be
measured.[^7]

d\) Connect the detector output to the input of an oscilloscope. The
output impedance of the detector must be matched to the input impedance
of the oscilloscope. The oscilloscope must have single-shot bandwidth
exceeding (1/t~r~) of the radar pulses to be measured. Set the
oscilloscope to a single-sweep mode, with a trigger threshold low enough
to ensure that radar pulses are captured. Wait until a series of pulses
are recorded. Elevate the trigger threshold and wait for another set of
pulses to activate the trigger. Continue this process until the
threshold is sufficiently high that no more pulses are recorded. Reduce
the trigger threshold slightly, and wait for a sequence to be recorded.
This pulse sequence shows the pulse repetition rate.

e\) Measure the pulse width and rise time or fall time on the
oscilloscope using criteria specified above for hard-line coupled
measurements..

## 2.3 Notes on the radiated pulse measurement procedure: By positioning the measurement system in close proximity to the radar, with clear line-of-sight, multipath problems are minimized and the received power in the pulses is maximized. Use of a high-gain measurement antenna further mitigates the multipath problem and increases the received pulse power level.

Care must be taken to ensure that all elements in the measurement system
have bandwidth and time response characteristics that are sufficient to
measure radar pulse rise time. Diode detectors with fast response
characteristics are probably necessary to meet this requirement.

In multi-radar environments, or environments with strong ambient
non-radar signals that are in or near the edges of the spectrum band of
the radar being measured, it may be necessary to take steps to isolate
the pulses of the radar being measured from other signals. The use of a
parabolic antenna for microwave-frequency radars, and also a bandpass
filter at the measurement antenna terminals will help to isolate the
desired pulse waveforms. If these items are not adequate to isolate the
desired pulses, then amplitude-dependent triggering should provide the
necessary isolation, assuming that pulses from the radar being measured
have higher amplitude in the measurement system than any other signals
in the environment.

# 3 Measurements for Advanced Technology Radars

## 3.1 Hardline coupled pulse measurements

In this context, advanced radars are those that utilize pulse
modulation. Either frequency or phase may be modulated. If frequency
modulation (chirping) is being utilized, then the same measurement
techniques may be used as specified above. But the measurement bandwidth
must equal or exceed the total chirped-frequency range. In practice this
may require the use of a broadband diode detector.

Measurements of rise time on chirped pulses are the same as for
non-chirped pulses; the same procedure as specified above may be used.

**Pulse compression ratio (FM pulse systems):** Measurements to
determine pulse compression are described below. This approach is
adequate for determination of pulse compression for all radars,
including advanced systems.

For phase-coded radar pulses, the measurement of pulse width is also
performed as specified above. But measuring the rise time of the
individual phase segments (chips) may be difficult. The first difficulty
arises with ordinary phase coding, in which a phase change of pi may
occur between each chip. Although the phase is shifted, the squared
value of the waveform is observed at the output of the detector, erasing
phase information. This makes the edges of the chips unobservable, in
principle, with a detector of any sort.

In application, transients may occur at the phase transitions between
the chips, and these transients are visible on an oscilloscope. But the
observation of the chip transitions does not result in a measurement of
the chip rise time.

**Total number of subpulses within each pulse (phase-coded systems)**:
Radars employing conventional phase-shifting with instantaneous
switching of ±180° will normally exhibit transients that can be observed
in the envelopes of the detected pulses. In this manner, the number of
chips in each pulse can be determined. However, for radar systems
employing MSK or other phase-shifting technologies that eliminate such
transients, it is impossible to determine the number of chips within
each pulse by measuring the detected pulse envelope. For these radars,
if a pair of hardline connections to monitor the I and Q channels are
not available, the number of chips can only be determined by using
reference materials such as technical manuals, operating manuals, and
specification sheets.

**Chip rise time measurement:** For ordinary phase-coded pulses, the
chip rise time may be measured directly only if the waveform is sampled
prior to detection. This may be done by connecting an IF output[^8] from
a spectrum analyser to a vector signal analyser or similar digital
signal processing device.

Advanced phase-coded pulses do not employ discontinuous phase changes
between chips. Instead they use minimum shift keying (MSK). With MSK
modulation, the requirement for observation of chip rise time is to
separate the I and Q components of the pulse, and observe the rise time
of each component individually. This can be accomplished with an
appropriately programmed vector signal analyser (or dedicated digital
signal processor (DSP) or field programmable gate array (FPGA)) that is
fed the IF output from a spectrum analyser.

If a measurement organization does not have available the
phase-sensitive equipment described above (VSA, DSP, or FPGA with
appropriate software installed), a pulse rise time measurement may be
performed on the rising edge of the pulse instead of a direct chip rise
time measurement. The rise time measurement is performed as described
above. If this is done, the fact should be noted in the resulting data
set.

## 3.2 Radiated coupled pulse measurements

In advanced radars that lack a directional coupler (such as systems
employing multiple transmitter modules), pulse characteristics must be
measured radiatively, as described above. Care must be taken to maintain
adequate bandwidth for measurement of pulse rise time, and the diode
detector input should be at an amplitude that is in the square-law
response of the detector.

## 3.3 Use of reference materials to determine pulse characteristics

Operations manuals, specification sheets, and other radar-specific
references may be presumed to be reasonably accurate for the aggregate
set of all radars within a particular model production line or series,
although it must be recognized that any individual radar may vary
somewhat from the production average. Such variation presumably occurs
as a result of both quality variation in manufacturing and the
maintenance that the radar receives in the field. If one or more of the
required pulse characteristics cannot be measured directly, the
parameter values that are quoted in such references may be used for
emission mask computations.

Appendix IV

(to Annex 1)

Calculation of gain correction factors for a planar antenna array\
using a software program written in BASIC

'\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*

This program is written, in BASIC, to determine the far field from a
near field measurement. Uses only the considerations of the phase
changes of the received wave due to the difference between the spherical
RF wavefront and the planar antenna array. Thus the program should only
be used to determine the boresight or maximum antenna gain at infinity
from a near field measurement. Antenna gain pattern is not addressed
here.

'\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*\*

'Test data for error -.025 pi radians; error \~.3 dB

'freq = 3000

'l = 10

'd = 1

'

CLS

'

INPUT "Enter the antenna frequency in MHz "; freq

INPUT "Now enter the measuring distance in metres from the antenna "; l

INPUT "Enter the maximum dimension of the antenna in metres "; d

'

'

'

CONST c = 300

CONST pi = 3.141592654#

'

'

lamda = c / freq

num = 100

'

'

IF d \< (5 \* lamda) THEN

PRINT "Antenna dimensions should be much greater (\* 5) than";

PRINT " the wavelength for accurate use of this prog"

STOP

END IF

'sum of inphase and quadrature field elements

sumi = 0

sumj = 0

'

' system is symmetrical so integrate from 0 to d/2

FOR i = 0 TO num - 1

dprime = i \* d / (2 \* (num - 1))

phasediff = (l - ((l \^ 2) + (dprime \^ 2)) \^ .5) \* 2 \* pi / lambda

' PRINT " phase diff is ";

' PRINT USING "##.##"; phasediff;

icomp = COS(phasediff)

sumi = sumi + icomp

jcomp = SIN(phasediff)

sumj = sumj + jcomp

NEXT i

PRINT " Max phase error is ";

PRINT USING "##.##"; phasediff / pi;

PRINT «  \* pi rads »

'form final received planar power received from spherical RF wave

res = ((sumj) \^ 2 + (sumi) \^ 2) \^ .5

'PRINT "Result is "; res; "i is "; i; " num is "; num

'Calc gain reduction

gprime = num / res

'

glog = 20 \* (LOG(gprime) / LOG(10#))

PRINT "Gain reduction from infinite far field is ";

PRINT USING "##.### "; glog;

PRINT " dB"

END

Annex 2\
\
Measurement of unwanted emissions of radar systems\
as detailed in Recommends 2 and 3

# 1 Introduction

The techniques recommended are termed direct and indirect. The direct
measurement method accurately measures unwanted emissions from radars
(as detailed in *recommends* 2 and 3) through free space measurement of
the radiated signals. The indirect method measures the signals at
transmitter output then combines it with models of the subsequent system
to estimate the free space field strengths. Comparison of the two
techniques has shown very close agreement; to within 2 dB.

## 2 Reference Bandwidth

In general the rules for determining reference bandwidth for higher
frequency radar (Annex 1) apply to lower frequency radar with suitable
scaling of the waveform parameters.

For radar systems, the reference bandwidth (B~ref~) used to define
unwanted emission limits (SM 329, SM 1541 and Appendix 3) should be
calculated for each particular radar system. For the three general types
of radar pulse modulation utilized for long wavelength radio­navigation,
radiolocation, acquisition, tracking and other radiodetermination
functions, the reference bandwidth values are determined using the
following formulas:

-- for fixed-frequency, non-pulse-coded radar, one divided by the radar
pulse length, in seconds (e.g. if the radar pulse length is 100 s, then
the reference bandwidth is 1/100 s  10 kHz);

-- for fixed-frequency, phase coded pulsed radar, one divided by the
phase chip length, in seconds (e.g. if the phase coded chip is 200 s
long, then the reference bandwidth is 1/200 s  5 kHz);

-- for frequency modulated (FM) or chirped radar, the square root of the
quantity obtained by dividing the chirp bandwidth in MHz by the pulse
length, in seconds (e.g. if the FM is from 1 2.50 MHz to 1 2.51 MHz or
10 kHz during the pulse of 20 ms, then the reference bandwidth is
(10 kHz/20 ms)1/2 7 \*10^2^Hz).

In cases, where the above calculated bandwidths are greater than 1 MHz,
then a reference bandwidth (B~ref~) of 1 MHz should be used.

# 3 Measurement bandwidth and detector parameters

The measurement bandwidth ,B~m~, is defined as the impulse bandwidth of
the receiver and is greater than the IF bandwidth ,B~if~, (sometimes
referred to as resolution bandwidth for spectrum analysers). The
measurement bandwidth (B~m~) may be derived from the following equation:

$B_{m} = B_{\text{if}}´\text{MBR}$

The measurement bandwidth ratio, MBR, needs to be determined for the
measurement receiver being used. MBR is approximately 3/2 for a --3 dB
IF bandwidth Gaussian filter as typically used in many commercial
spectrum analyser receivers. (Note in some instruments the IF bandwidth
is defined at the --6 dB point).

An appropriate receiver IF bandwidth should be selected to give one of
the following recommended measurement bandwidths. ( In general, the
rules for determining measurement bandwidth for higher frequency radar
(Annex 1) apply to lower frequency radar with suitable scaling of the
waveform parameters.)

  ---------------- -- -------------------------------------------------------
  Measurement         (1/*T *) for fixed-frequency, non-pulse-coded radars,
  bandwidth^1,2^      where *T* = pulse length.\
                      (E.g. if radar pulse length is 100 s, then the
                      measurement IF bandwidth should be /(100 s)  10 kHz.)

                      (1/*t *) for fixed-frequency, phase-coded pulsed
                      radars, where *t* = phase-chip length. (E.g. if radar
                      transmits 260 s pulses, each pulse consisting of
                      13 phase coded chips that are 20-ms in length, then the
                      measurement IF bandwidth should be\
                       1/(20 s)  50 kHz.)

                      ( *B*/*T *)½, for swept-frequency (FM or chirp, or
                      FMCW) radars, where *B* : range of frequency sweep
                      during each pulse and *T=* pulse length. (E.g. if radar
                      sweeps (chirps) across frequency range of 1 2.500 -
                      1 2.510 MHz ( 10 kHz of spectrum) during each chirp,
                      and if the chirp length is 20 ms, then the measurement
                      IF bandwidth should be:\
                       ((10 kHz)/(20
                      ms))½  $\sqrt{0\text{.}5}$kHz  7\*10^2^Hz.)
  ---------------- -- -------------------------------------------------------

^1^ Note, in all cases if the above derived measurement bandwidth is
greater than 1 MHz then a measurement bandwidth B~m~ close to 1 MHz
should be used.

The corrections associated with measurement bandwidth transforms to
reference and PEP bandwidths discussed in section 3 of Annex 1, also
apply to long wavelength radars described here in Annex 2.

# 4 Dynamic range of the measurement system

The measurement system should be able to measure levels of unwanted
emissions as given in Appendix 3. To obtain a complete picture of the
spectrum especially in the spurious emissions domain, it is recommended
to be able to measure levels of emissions 10 dB below the levels given
in Appendix 3.

For a high level of confidence in the results, the measurement dynamic
range of the system should be significantly higher than the required
range of measurement (margin (2) in Figure 2).

The link between the required range of measurement and the recommended
dynamic range of the measurement system is given in Figure 2.

5 Direct method

A direct method, described below, can be used to measure unwanted
emissions (out-of-band and spurious) from long wavelength radar systems,
which allow easy access to the main beam of the radar. For instance
where the antenna or array is situated on the ground and is vertically
polarised. This method has been used to measure the emission
characteristics of long wavelength radar systems operating at
frequencies up to 45 MHz, and e.i.r.p. in the Megawatt range.

## 5.1 Measurement hardware and software

Antenna

A block diagram of the type of measurement system required for the two
direct methods are shown in Figure 11. The first element to be
considered in the system is the receive antenna. The receive antenna
should have a broadband frequency response, at least as wide as the
frequency range to be measured. This may require the use of ground
screens. Gain is not usually a problem and so a simple whip antenna with
a ground screen is adequate. Calibration of the antenna gain may be
required for broadband measurement. This can be achieved using a
reference source and a second short (poorly matched) antenna feeding
into a power meter.

The antenna should be located in the far field if practical, for example
at 20 MHz, \>1 km away, although measurement of spectral
characterization has shown no discernible difference in far field and
near field measurements. Many long wavelength radars are arrays which
synthesise a beam that is electronically steerable. In this case the
beam should be steered or measurement antenna positioned such that the
measurement antenna is as close as possible to the peak of the main
beam.

The antenna polarization is selected to maximize response to the radar
signal.

The cable connecting the measurement antenna to the measurement system
may be normal coaxial cable.

Clear Channel Advisor

Because of ionospheric propagation long wavelength transmissions can
travel large distances, and so much of the spectrum that is measured by
the test antenna will, in general, be exposed to external signals. It is
therefore important to have a device advising of occupied channels,
preferably one that can capture this data and give some indication of
the signal strength. The spectrum measuring system can be used for this,
or an independent receiving system. This data can be used to reconcile
any unwanted emissions that may have been caused by external sources. It
should also be used to detect a clear channel for the testing of within
the in-band B~40~ and OOB domains.

![](media/image8.wmf)

Figure 11

Block diagram for measurement of radiated unwanted emissions\
from radars using the manually controlled direct method

# RF Front End

The RF front-end performs two functions. The first is to protect the
front-end of the detection system through the use of variable RF
attenuation. The second is low-noise preamplification to provide the
maximum sensitivity to low power emissions. The RF attenuator is the
first element in the front-end. It provides variable attenuation
(e.g. 0‑70 dB) in fixed increments (e.g. 10 dB/attenuator step).

### 5.1.1 Manually controlled measurement system

The manually controlled measurement consists of sweeping across the
spectrum in fixed increments (equal to the measurement bandwidth). At
each frequency sweep, the attenuator maybe adjusted to keep the radar
peak power within the dynamic range of the measurement system (often the
front‑end amplifier and the spectrum analyser log amplifier are the
limiting elements). With the front-end RF attenuator properly adjusted
at each sweep, a measurement of the radar power at that frequency is
performed.

The final element in the RF front-end is a low noise amplifier (LNA). An
LNA installed as the next element in the signal path after the
preselector. The low-noise input characteristic of the LNA provides high
sensitivity to low-amplitude spurious radar emissions, and its gain
accommodates the noise figure of the rest of the measurement system
(e.g. a length of transmission line and a spectrum analyser/selected
receiver).

The sensitivity and dynamic range of the measurement system are
optimized by proper selection of LNA gain and noise figure
characteristics. It is desirable to minimize noise figure while
providing enough gain to accommodate all measurement circuitry after the
LNA (essentially the RF line loss after the front end, plus the noise
figure of the spectrum analyser /selected receiver circuitry). Ideally,
the sum of the LNA gain and noise figure (which is the excess noise
produced by the LNA with a 50 Ω termination on its input) should be
approximately equal to the noise figure of the remaining measurement
system. For example, assume that the spectrum analyser noise figure is
25 dB and the RF line loss between the RF front end and the analyser is
5 dB. Thus the front-end LNA must accommodate a total noise figure of
30 dB. The sum of the LNA gain and noise figure should therefore be
approximately 30 dB in this example. A combination for such an LNA would
be 3 dB noise figure and 27 dB gain.

The remainder of the RF measurement system is expected to be essentially
a commercially available spectrum analyser or a spectrum analyser with a
pre-selector or a selective receiver. Any equipment, which can receive
signals over the frequency range of interest, can be used. Measurements
have been performed with modern digital receivers which easily
accommodate the frequency and dynamic range requirements, largely
obviating the need for any attenuation or gain in the front end.

# 6 Indirect method

In the indirect method, measurements are made by coupling from the
output of each transmitter. The measurement apparatus is similar to the
direct method from that point. If there are multiple transmitters, then
the complex amplitude must be recorded, then the signals must be
combined together in software taking account of beam steering array
weighting and feeder delays.

Data capture can be readily achieved by connection of the spectrum
analyser or receiver to a laptop PC through a GPIB or equivalent
interface.

\_\_\_\_\_\_\_\_\_\_\_\_

[^1]: ^\*^ This Recommendation should be brought to the attention of the
    International Maritime Organization (IMO), the International Civil
    Aviation Organization (ICAO), the International Maritime Radio
    Association (CIRM), the World Meteorological Organization (WMO) and
    Radiocommunication Study Groups 1, 4 and 9.

[^2]: When the fall time, t~f~, of the radar pulse is less than the rise
    time, t~r~, it should be used in place of the rise time in applying
    equations in Recommendation ITU-R SM.1541.

[^3]: For example, a 1 µs pulse might have a rise time of less than 0.1
    µs. This t~r~ would require bandwidth in excess of 10 MHz for
    accurate measurement. Oscilloscopes are available with bandwidths up
    to 2 GHz. For purposes of measuring radar rise/fall times,
    oscilloscopes with at least 500 MHz should be used. The bandwidth
    must be available in a single-shot (not repetitively sampled) mode,
    as the measurements are made on single radar pulses.

[^4]: The initial attenuator setting may be derived from the radar's
    peak power level and the specified insertion loss of the directional
    coupler.

[^5]: Most oscilloscopes can record data to either an internal disk or
    to an external computer via an IEEE-488 (GPIB) bus. Recording may
    also be achieved by photographing the oscilloscope screen with a
    digital still-frame camera.

[^6]: Crystal detector outputs are not necessarily linear; therefore the
    10 percent, 50 percent, and 90 percent voltage points for the RF
    signal may not appear to be 10 percent, 50 percent, and 90 percent
    voltage points at the DC output of the detector. A calibrated signal
    generator is needed to determine the actual DC output voltages for
    these input voltages.

[^7]: The peak input power to the detector should fall within the
    square-law response region. To obtain the appropriate power input
    level, it may be necessary to install either an attenuator or an
    amplifier between the bandpass filter and the diode detector.

[^8]: The IF output is assumed to be coupled from the spectrum analyser
    prior to the detection and resolution bandwidth stages, so that
    adequate bandwidth is maintained for a pulse rise time measurement.