SNS Project-Wide Beam Current Monitors[^1]

M. Kesselman^\*^, R. Witkover^\*\*^, L. Doolittle^\*\*\*^, J.
Power^\*\*\*\*^

*^\*^BNL, Upton, NY 11973, ^\*\*^ TechSource, Inc., Santa Fe, NM 87594,
**^\*\*\*^** LBNL Berkeley, CA 94720 , ^\*\*\*\*^ LANL Los Alamos, NM
87545*

**Abstract.** A consortium of national laboratories is constructing the
Spallation Neutron Source^1^ (SNS) to be installed at Oak Ridge National
Laboratory. There are signal similarities that exist in the beam
diagnostic instrumentation that could permit common designs. This paper
will focus on the beam current monitoring requirements, and the methods
under consideration to measure beam current in various locations
throughout the SNS facility.

# Requirements

The Front End, Linac and HEBT will have a 57mA peak current pulse of
about 1 ms duration (macro-pulse) chopped at the Ring revolution
frequency of 1.05MHz (mini-pulse). Mini-pulses consist of approximately
a 645nS long bunch and a 300nS gap. The micro-pulse structure of
402.5MHz, which exists in the Front End, Linac, and shortly after
injection into the Ring, will not be observed with the Beam Current
Monitors (BCMs). The Ring will accumulate this current over 1mS, about
1000 mini-pulses, resulting in an average of about 57 amps at the end of
the cycle. The RTBT will carry the current of the last turn in the Ring
to the target. The electronics in the Ring and RTBT must operate over a
signal range of 1000 to 1 (60dB), however, the same basic circuitry used
in the Front End, Linac and HEBT could be used with the addition of gain
switching.

Transformer droop during the macro-pulse, which will cause a baseline
shift, is the result of finite inductance and load resistance limiting
the L/R time constant. If more turns are added to increase inductance
the sensitivity goes down, and resonances with winding capacitance and
delay line effects will limit the high frequency response. Additionally,
the Front End and Linac have little available space, and the
transformers must be mounted in close proximity to DC magnetic fields.
Large, shielded transformers, needed to obtain high inductance and
exclude stray fields, will not fit.

To observe the details of the mini-pulse, a fast response (1nS rise time
with a droop rate of about 0.1%/uS ) is required. For the macro-pulse
current, a rise time of about 50nS and a droop of 0.1%/mS is required.
The distribution of transformers is shown in Table 1.

##### Table 1. Beam Current Monitor Distribution ( ^\*^ not finalized)

+----------------------+------------------------+----------------------+
| ## [L                | ## [Beam Pipe          | ## [Number           |
| ocation]{.underline} |  Diameter]{.underline} | of BCMs]{.underline} |
+----------------------+------------------------+----------------------+
| Front End            | 7cm Flange OD          | 2                    |
+----------------------+------------------------+----------------------+
| Linac                | 2.5cm,3.0cm,8cm^\*^ID  | 13                   |
+----------------------+------------------------+----------------------+
| HEBT                 | 12cm OD                | 4                    |
+----------------------+------------------------+----------------------+
| Ring                 | 21cm OD                | 1                    |
+----------------------+------------------------+----------------------+
| RTBT                 | 21cm OD                | 5                    |
+----------------------+------------------------+----------------------+

.

**Survey of Current Monitoring Techniques**

# Passive:

This is a basic beam current transformer consisting of a
high-permeability tape wound core, with a single winding which acts as
the secondary and the beam as the primary. As described above,
conflicting requirements for rise time and droop make it difficult to
design a transformer suitable to meet both and have adequate sensitivity
for the first turn. A transformer suitable for observing the macro pulse
in the Ring was developed^2^, however the large size and low sensitivity
preclude its use in the Front End and Linac.

Fast Current Transformer (FCT):

A passive transformer but optimized for high frequency response, as a
result there is considerable droop during the macro-pulse. However,
commercial versions are available which are well suited to observing the
mini-pulse. For example, Bergoz^3^ offers a unit with sub-nanosecond
rise time and 0.1%/μS droop. When observing the macro-pulse the baseline
will gradually shift such that the area above zero is equal to the area
below zero (average = zero). For the 645nS pulse with 300nS gap the
steady state transformer response will have a baseline shift nearly 2/3
the pulse. This baseline can be restored on a mini-pulse by mini-pulse
basis using a fast baseline restoration technique, as discussed below,
to make macro-pulse measurements.

Hereward (magnetic feedback):

This approach uses a transformer with active feedback to compensate for
droop by sensing the change in magnetic field in the core and restoring
it through an additional winding. This technique works well, but trades
sensitivity for low frequency response. Estimates indicate that it will
not be appropriate for this application.

Active-Passive^4^:

This is an interesting approach, employing an amplifier to compensate
for droop without affecting the basic transformer high frequency
response, and requiring no additional windings. However, using the
transformer in the feedback loop gives rise to stability problems as a
result of phase shift in the feedback signal. This may show up when the
amplifier is located a distance from the transformer as it is in SNS
(with the exception of the Front End). Bergoz offers an ACCT based on
this approach with a stated bandwidth of 6Hz to 300kHz with cable
lengths hundreds of meters long. This bandwidth will not meet the SNS
requirements.

Direct-Current Current Transformer (DCCT):

This approach involves a magnetic circuit employing multiple windings,
and a modulation technique to force the magnetic core into the
non-linear region, generating harmonics of the modulation frequency. The
second harmonic, which is proportional to the magnetic field passing
through the aperture, is detected and amplified. Response down to DC can
be obtained. By combining this with other high frequency transformers
with suitable matching crossover filters, a wide band system can be
obtained. Several manufacturers offer such a device, however, the high
frequency response is not of sufficient bandwidth and DC response is not
required for SNS. Average current will be obtained numerically from the
pulse response.

Integrating Current Transformer (ICT):

Very fast (sub-nanosecond) current pulses can not be easily detected
with a current transformer due to frequency dependent magnetic losses.
The ICT was designed to avoid this by pre-integrating the signal on an
internal capacitor, then transferring it to the core. This also lessens
the burden on the electronics, which now need only respond to a 10 to
20nS rise time. Bergoz supplies such a unit and associated electronics,
which also provides baseline restoration and an analog output
proportional to the charge. To observe mini-pulses we require nanosecond
rise time response. The sub-nanosecond response of the ICT is not
necessary, and the 20nS response is inadequate.

**DISCUSSION**

The wide band information associated with the mini-pulse and the
information associated with the macro pulse can be extracted from a
single transformer with suitable rise time and droop characteristics. To
analyze the detailed mini-pulse a fast transformer with droop of 0.1%/μS
can be utilized. The macro-pulse data can be observed using the same
transformer if appropriate baseline restoration is provided to restore
the drooping baseline on a turn by turn basis. A general block diagram
is shown in Figure 1.

The input conditioning employs gain adjustment to handle the large
dynamic range of the Ring and RTBT, and need not be employed for the
Front End, Linac and HEBT. The baseline restoration is required in all
cases, and feeds the conditioned signal to various output processors.
High-speed data acquisition with appropriate circular buffers for data
storage process the mini-pulse. Integration of the current for charge
analysis, with an ability to hold the turn-by-turn result and perform a
digital conversion is a common requirement for all lines. Provision for
local viewing and a filtered output to a track and hold amplifier
followed by digital conversion completes the package.

![](media/image1.wmf){width="5.9944444444444445in"
height="3.390277777777778in"}

**Figure 1.** General block diagram

Input Signal Conditioning:

Two problems which must be resolved are: compensating for the signal
droop due to finite L/R, and, noise reduction due to the remote location
of the amplifier from the transformer. A single ended input can easily
match a coaxial transmission line, but may suffer from noise induced on
long cables. If a wide-band high common mode cable is used, a
differential input stage can provide good common mode noise rejection
and properly terminate the cable. A number of input conditioning
amplifiers have been considered.

One approach suggested was to go directly into the negative input
terminal of an op-amp. Since the transformer serves as a current source,
only the feedback resistor determines the op-amp output. In this case
the droop time constant is given by the inductance L divided by R~c~ +
R~a,~ where R~c~ is the coil resistance and R~a~ is the feedback
resistance divided by the open loop gain. This requires the op-amp be
mounted directly at the transformer to avoid the need for cable matching
resistors. Also, voltage noise is amplified by the full open loop
voltage gain of the op-amp. At the high energy end of SNS, the radiation
environment makes this option undesirable.

Gain switching will be necessary to accommodate the large signal range
of the Ring and RTBT. A minimum of two gains is necessary for the RTBT:
single turn capability, and full output capability. The Ring will
require two or three gains, to cover the 1000:1 signal range. This would
allow the signal to swing over a 10:1 range before switching gains and
would maintain a digitizing error between 0.1% and 1%. Additional gain
settings could improve this. Fast electronic switches are available
which would permit gain changes during the gap time, however, the
transient caused by the gain change and any signal filtering could
increase the settling time thereby corrupting data during the next turn.
An alternate solution under consideration would employ the use of dual
variable gain amplifiers, with switching after the amplifiers to allow
observing the output of one while the other is changing gain.

Turn-by-Turn Base Line Restorer (BLR):

The finite L/R of the transformer will cause the baseline to gradually
move away from zero during the mini-pulse (see Figure 2a for HEBT
simulation and Figure 2b for Ring simulation). A circuit is required
which will acquire the signal during the gap time and subtract it from
the signal during the beam time. This can be done either with an analog
or digital circuit such as shown in Figures 3a, 3b, and 3c. The BLR must
respond quickly to permit sufficient settling of the signal electronics
prior to the switched BLR operation. Approximately 7 time constants are
required to settle to 0.1%. For a 50nS rise time response
(Tau=50nS/2.2), about 160nS is required to

achieve 0.1% settling. This provides about 140nS for the BLR to be
switched on and settle which appears practical.

Analog BLR (see Figure 3a):

In the analog implementation the signal baseline is sampled just prior
to the arrival of the first mini-pulse and then prior to successive
mini-pulses, after the falling edge has settled. It is integrated for a
finite time and held. By proper adjustment of the gain, via the
inverting amplifier and the choice of gain in the summing stage, a
voltage exactly opposite in amplitude to the output offset is added to
the input signal. This restores the baseline to zero, and establishes
the appropriate reference for the incoming signal pulse. This process is
repeated for each successive mini-pulse, constantly readjusting the
baseline for the total macro-pulse time. This readjustment helps to
compensate for integrator drift during the macro-pulse. When the
macro-pulse is completed the integrator is reset, and the process
repeats.

Analog/Digital Hybrid BLR (see Figure 3b):

As in the analog BLR, the baseline is sampled just prior to the arrival
of the mini-pulse. An ADC is used together with a DAC to provide a good
representation of the baseline with no droop or drift as with the analog
circuit described above. The digitized DC offset of the baseline is
added to a digital accumulator to store and "remember" the offset level.
By so doing, subsequent adjustments to the baseline offset are
minimized, requiring only small (few bit) corrections. These corrections
are added or subtracted to the previous offset stored in the
accumulator, depending upon the sign of the ADC output. The DAC output
is added to the input, however, due to the phase reversal a subtraction
is actually performed. This process continues for the entire macro-pulse
time, and is repeated at the start of the next macro-pulse. Unlike the
analog version above, the digital accumulator need not be reset at the
end of the macro-pulse, since there is no drift associated with the
digital circuitry.

![](media/image2.wmf){width="2.9in" height="1.7729166666666667in"}

#####  Figure 2a Figure 2b

**Figure 2a.** HEBT Current transformer response showing droop due to
L/R=1mS. Pulse duration is 550nS and gap is 300nS, repeated 1176 times.
The transformer recovers to within 0.1% of the baseline in 7 time
constants. R~L~1 is back termination. **Figure 2b.** Ring Current
transformer response showing droop due to L/R=1mS. Pulse duration is
550nS and gap is 300nS, repeated 1176 times. The transformer recovers to
within 0.1% of the baseline in 7 time constants. R~L~1 is back
termination.

Digital BLR (see Figure 3c):

A fully digital implementation averages the signal during the gap time
in the accumulator/divider. The average base line is stored and held in
the storage register at the end of the gap time, and this is subtracted
from the signal to restore the baseline to zero. The signal must be
delayed in a circular memory to allow time for digital processing and
proper alignment with the averaged base line calculation. The
accumulator is reset prior to the calculation of the next turn's
baseline. This technique suffers no drift, and could include additional
digital processing techniques, such as in-situ calibrated pole-zero
cancellation and other filtering. In addition, it can be implemented in
real-time on an FPGA, or as a post-processing step on a DSP or
general-purpose computer.

# Integrator:

The integrator used to measure the turn-by-turn (mini-pulse) charge must
reset to zero (within a LSB) during the gap time of about 300 nS. This
is insufficient for proper settling of the integrator, and a "ping-pong"
approach is under consideration. This technique employs two integrators,
resetting one while the other is integrating, effectively increasing the
available reset time to 945ns. Circuitry to compensate for charge
injection could be included but may be treated as an additional offset.
The output of the integrator would be acquired during the gap-time by a
fast track and hold and held until digitized by an ADC. The ADC data
will be delivered to the control system and stored in a circular buffer
suitable for holding 10 seconds of data. This buffer can be interrogated
for later analysis.

An alternative design would use a fast 12 - 14 bit ADC operating at 30
MSa/s to 120 MSa/s, sampling the current waveform and accumulating the
result during the mini-pulse. This approach would depend on the
availability of fast, high resolution ADCs at reasonable cost, but might
be preferable to the analog integrator.

![](media/image4.wmf){width="3.05in" height="2.123611111111111in"}

**Figure 3a Figure 3b**

**Figure 3a.** Analog turn by turn baseline restorer using finite time
integration. A general BLR concept that may be implemented in different
ways. **Figure 3b.** Hybrid Analog/Digital turn by turn baseline
restorer. A general BLR concept that may be implemented in different
ways.

![](media/image6.wmf){width="4.65in" height="2.602777777777778in"}

**Figure 3c.** Fully Digital Implementation of BLR

# Mini and Macro-pulse monitoring, and data storage:

In most every case where a current transformer is used, both a
mini-pulse analysis, and a macro-pulse analysis is required. The RTBT
line is an exception, requiring only mini-pulse analysis, and the MEBT
line requiring only macro-pulse analysis. To process the mini-pulse, a
1GSa/S ADC is shown in the block diagram of Figure 1. This converter
will provide the basic data for digital integration to calculate charge,
and will be stored in a circular buffer sufficient to hold 10 seconds of
past historical data for analysis of failures. To accomplish this
requires about 600Mbytes per transformer (ten to fifteen transformers
are under consideration). The transmission of this data will place a
severe burden on the network, even if only exercised off-line, and
further discussion of operational requirements is needed.

Macro-pulse monitoring, depicted in figure 1, is filtered and acquired
at the end of mini-pulses by a fast track and hold for conversion. The
ADC data will be delivered to the control system and stored in a
circular buffer suitable for holding 10 seconds of data. This buffer can
be interrogated for later analysis.

An alternative design under consideration would share the same fast
12-14 bit ADC discussed above. Additional digital processing could
derive macro-pulse properties from that data. Again, values will be
delivered to the control system, and will be stored in a circular buffer
suitable for 10 seconds of data. This buffer can be interrogated for
later analysis.

Local readout can be accomplished by a simple buffer amplifier, which
will make the basic current signal available for viewing locally on an
oscilloscope for diagnostic purposes.

# SUMMARY

A number of approaches have been presented and require additional
investigation. In particular, digital approaches seem to eliminate many
analog difficulties, permitting drift free integration, filtering, and
base line restoration. This seems most encouraging, particularly with
ADCs becoming available which will provide 14 bit resolution at 65MSa/s
(AD6644).

# Acknowledgments

The author wishes to acknowledge the efforts of V. LoDesto and D. Kipp
of BNL for their assistance with the construction and characterization
of a test facility apparatus.

# References

1\. http://www.sns.gov

2\. Kesselman, M., *Study of Ring BCM Requirements, Parameters and
Feasibility*, BNL/SNS Technote 062, May 20,1999.

3\. [http://www.bergoz.com](http://www.bergoz.com/)

4\. Unser, K. B., *Atomkernenergie*, Kerntechnik*,* Vol. 47 (1985) No.1
(pg 48)

[^1]: Work performed under the auspices of the U.S. Department of
    Energy.