FACT SHEET

# US WP8B CONTRIBUTION

DOC. NO: RCG-16

### DATE: 12 August 2002

**TITLE:** WORKING DOCUMENT TOWARDS, PRELIMINARY DRAFT REVISION OF
RECOMMENDATION ITU-R M.1314, Reduction of Spurious Emissions of Radar
Systems Operating in the 3 GHz and 5 GHz Bands

**AUTHOR:** Robert L. Hinkle

> Phone: 202-482-3212
>
> Fax: 202-482-4595
>
> Email: rhinkle@ntia.doc.gov

**CONTRIBUTERS:** Gil Wootton, JSC

**BACKGROUND:** The ITU-R has approved Recommendation ITU-R SM.1541
addressing out-of-band emission masks for the various services. ANNEX 8
of the recommendation contains proposed out-of-band emission masks for
radiodetermination radars. As part of reaching aggreement on out-of-band
emission masks for the radiodetermination service, it was agreed that
further studies on the feasibility of establishing more stringent
emission mask would be completed prioe to the radiocommunication
assembly meeting of 06.

# OBJECTIVE: To provide a working document towards a preliminary draft revision of Recommendation ITU-R M.1314. This document would be further developed as the Working Party 8B Radar Correspondence Group completes it's work program studying the feasibility of establishing more stringent emission masks for radars. 

###  FACT SHEET PREPARER: Bob Hinkle

United States of America

**WORKING DOCUMENT TOWARDS**

**PRELIMINARY DRAFT REVISION OF**

RECOMMENDATION ITU-R M.1314[^1]

REDUCTION OF UNWANTED EMISSIONS OF RADAR SYSTEMS

(Question ITU-R 202/8)

\(1997\)

Rec. ITU-R M.1314

###### Summary

This Recommendation provides information on the design factors affecting
unwanted emission characteristics of radar transmitters to be taken into
account during the design of radars. It also recommends certain types of
transmitter output devices that should be used when practicable to
minimize unwanted emissions.

The ITU Radiocommunication Assembly,

considering

a\) that the radio spectrum available for use by the radiodetermination
service is limited;

b\) that the radionavigation service is a safety service as specified by
No. 953 (S4.10) of the Radio Regulations, and in addition that some
other types of radar systems such as weather radars may perform
safety-of-life functions;

c\) that the necessary bandwidth of emissions from radar stations in the
radiodetermination service is large in order to effectively perform
their function;

d\) that new emerging technology systems may use digital or other
technologies that are more susceptible to interference from radars\'
unwanted emissions due to their high peak envelope power;

e\) that the ITU-R has been studying the question of efficient use of
the radio spectrum by radar;

f\) that unwanted emissions from radar systems may in some cases cause
interference to systems in other radio services operating in the
adjacent and harmonically related bands;

g\) that the ITU-R has approved a radio regulation limiting the spurious
emissions from radiodetermination radars (Appendix 3), and a
recommendation on out-of-band characteristics for radiodetermination
radars (ITU-R SM.1541),

recommends

**1** that the information on radar transmitter design factors affecting
unwanted emission characteristics of radars contained in Annex 1 be used
to reduce unwanted emissions;

**2** that, when practical, linear beam or solid state radar output
devices should be used in radars to reduce non‑harmonic radar spurious
emission levels;

**3** that, when necessary and when possible, radar output filters
should be used to reduce radar unwanted emissions.

ANNEX 1

Reduction of unwanted emissions of radar systems

# 1 Introduction

Whenever two or more systems operate off-tuned from each other,
transmitter unwanted emissions obviously have a major effect on mutual
compatibility. Radar transmitters should therefore be chosen, designed,
and constructed such that the emission spectrum falls off as rapidly as
possible, given the constraints on radar performance, size, cost,
weight, reliability, maintainability, etc. The emission-spectrum skirt
falloff rate (out-of-band emission characteristics), and the emission
floor level (spurious emissions) are determined by the transmitter
hardware and architecture and by the transmitted waveform. Those
influences are discussed below.

# 2 Radar design factors

The function or mission of a radar largely determines the design of the
radar. Radar missions are widely varied (such as: navigation, weather
observation, wind velocity determination, surveillance, imagery and
mapping, terrain following, altimeter, etc.) and generally require
unique performance specifications. These missions determine some
parameters that are not under the control of the radar designer which
directly impacts upon radar design factors such as: required transmitter
power, transmitter waveform selection, transmitter output device
selection, antenna gain, receiver sensitivity, range and azimuth
resolution, and Doppler coverage. The judicious trade-off of radar
design factors to improve emission spectrum control is key in enhancing
compatibility between radar systems and other services.

**2 Waveform selection and shaping**

The choice of pulse waveform type and the way in which the waveform is
shaped can also have important influences on spectrum control and hence
on compatibility. Most radars, especially those using a single power
oscillator or power amplifier, are constrained by considerations of
energy efficiency and heat dissipation to use pulses having essentially
constant-amplitude except during brief transitions between subpulses.
That limits the types of waveforms that can be chosen. Even when that
constraint applies, however, choices remain that can have a major effect
on the emission spectrum.

Radar waveforms can be categorized, at the first level, into
plain-pulse, or unmodulated-pulse, waveforms (having the emission
designator of \"P0\") and intra-pulse-modulated waveforms. Intra-pulse
modulation usually serves as a means for implementing pulse compression,
although an exception occurs in the case of waveforms used to drive
frequency-steered arrays. Intra-pulse modulations can thus be divided
further into the following subcategories:

-   continuous FM, or "chirp" pulses

-   stepped-chirp pulses

-   stepped-frequency pulses used in frequency-steered radars

-   discretely-coded pulses

From the standpoint of emission-spectrum control, a guiding principle in
selecting and shaping a waveform is to remove discontinuities in as many
derivatives of the waveform as possible, since that determines the
ultimate spectrum falloff slope, in dB/decade of frequency offset, that
is achieved. The various pulse waveforms are therefore distinguished by
the differences among their transitions of amplitude, phase, and
frequency within the pulse.

All pulse waveforms, of course, contain rise and fall ramps on the
overall envelope. Other things being equal, it is desirable to have
gradual and smooth rise and fall ramps. However, other things are not
always equal. In particular, pulses generated in crossed-field devices
require quick rise ramps to avoid excitation of spurious oscillatory
modes that would worsen the spectrum. When amplifiers other than
crossed-field devices are used, smooth, gradual rise ramps are helpful
to spectrum control when they can be implemented. Such implementation
might still be difficult because power-amplifier dissipation is usually
high when the amplifiers are not driven close to saturation; that can
motivate use of fast rise and fall ramps even when spurious oscillations
are not a concern.

Continuous frequency modulation, or chirp, waveforms with high
pulse-compression ratio, or bandwidth-pulsewidth product, have very
steep spectrum falloff rates. This applies to both linear FM and
nonlinear FM waveforms. The main contribution to undesired spectral
components of these waveforms arise from the use of short rise ramps on
the pulses.

Stepped-chirp waveforms have piecewise-constant frequencies that
increment or decrement monotonically throughout the pulse. They can be
considered as a subset of continuous-frequency-modulation chirp.
However, stepped-chirp waveforms, as well as non-monotonic
stepped-frequency waveforms that are used with frequency-steered antenna
arrays, have poorer emission spectra than continuous-FM chirp waveforms
have. This is a consequence of discontinuities in the waveform. It might
be feasible to remove those discontinuities by implementing the stepped
chirp in a way that maintains continuity of phase at the junctions
between frequency steps. Even if that is so, however, discontinuities in
the first derivative, which do not occur in true continuous-FM
waveforms, will remain, so the spectrum will not be as good as that of a
continuous-FM pulse with comparable pulse-compression ratio.

There are also certain polyphase-coded waveforms, of which the Frank
polyphase-coded waveform is the prototype, that effectively approximate
chirp waveforms; i.e., they approximate "continuously-coded"
waveforms.[^2] However, these contain abrupt steps in phase, so their
spectra do not fall off nearly as steeply as those of continuous-FM
chirp waveforms.

Herein, discretely-coded radar waveforms refer to those that do not
resemble continuous-FM waveforms in any way. Since that excludes the
polyphase codes, most discretely-coded radar waveforms can be subdivided
into bi-phase-coded and frequency-coded types. Waveforms in either of
these categories can use Barker codes and pseudo-random binary sequence
codes.

In the absence of refinements, discretely phase-coded waveforms have
abrupt transitions between constant-phase \"chips\". (The same is true
of Frank codes and other polyphase codes.) As a consequence, their
spectra fall off at only 20 dB/decade. However, some options are
available that can improve the spectra of phase-coded waveforms.

In principle, the spectrum of the RF drive (excitation) waveform can be
made to fall off arbitrarily fast by filtering the modulating waveforms
or the modulated low-level drive waveforms (either IF or RF) themselves.
However, those gains can be eroded in practice by spectral re-growth
that occurs in both the transmitter power amplifier and in receivers in
the environment. When premodulation filtering is used, the chip-to-chip
transitions are gradual instead of abrupt, but on bi-phase waveforms and
those polyphase waveforms that contain 180-degree phase transitions,
nulls or dimples remain in the waveform envelope because that envelope
passes through zero during transitions from one phase to the other. That
is not a problem in itself, but the advantages gained are reduced by two
factors. One factor is AM-to-PM conversion that occurs in
power-amplifier devices. The extraneous phase modulation that results
widens the spectrum. Another disadvantage is that any limiting that
occurs in either the power-amplifier transmitter stages or in victim
receivers tends to re-introduce abrupt transitions into the dimpled
waveform. Those abrupt steps translate into unwanted spectral sidebands
with spectrum skirts that again fall at only 20 dB/decade.

It is possible to mitigate that spectral regrowth to a considerable
extent. This can be done by constructing exciter (low-level, driver)
waveforms that maintain a nearly constant envelope not only during the
subpulse dwell intervals but also during phase transitions. In such
waveforms, 180-degree phase transitions consist of rotations of carrier
phase through a semicircle in the I-Q, or real-imaginary plane instead
of movements along the I or Q axis that pass through the origin. This
can be implemented by means of quadrature modulators and suitable
waveform-shaping circuitry.

An alternative discretely coded waveform category is continuous-phase
frequency-shift keying (CPFSK). These waveforms are essentially the same
as so-called \"minimum-shift-key\" (MSK) waveforms used in some
communication systems. Although sometimes referred to as
phase-shift-keyed waveforms, these are really frequency-coded because
the phase changes continuously while, in their basic unfiltered form,
the instantaneous frequency changes abruptly and remains constant
throughout each subpulse. There are no discontinuities in the waveform
itself, but there are discontinuities in the first derivative.
Consequently, the spectra approach asymptotes that fall at a rate of
40 dB/decade. Furthermore, these waveforms have constant envelope even
during their subpulse transitions, so they are intrinsically immune to
the spectral-regrowth problems that occur with phase-coded waveforms.
(Since swept-frequency waveforms have no subpulses, they too are immune
from spectral regrowth due to limiting and AM-to-PM conversion.) In
communication systems, pre-modulation-filtering of MSK waveforms is
widely used. It is expected that such filtering could also be applied in
radars, in which event the emission spectrum falloff would theoretically
become steeper than 40 dB/decade.

While steep falloff of the emission spectrum is desirable, it cannot be
pursued without regard for the consequences in range resolution and
Doppler coverage, usually expressed by the shape of the "ambiguity
function". That function represents the magnitude of the output signal
evoked by return from a point target and produced by a filter matched to
the transmitted signal. The ambiguity function is a function of both the
range (time delay) and the Doppler shift of the target return. As one
extreme example, a linear-FM rectangular pulse waveform with infinite
time-bandwidth product (i.e., infinite compression ratio) would have a
perfectly rectangular spectrum, except for the contribution from the
rise and fall ramps. But the response of a matched filter to such a
waveform would have a sin(t)/t response for a constant-Doppler target
return. Such a response has time (i.e., range) sidelobes only about 13
dB below the main response, which is inadequate in some applications
that require a high degree of multiple-target resolution. The
matched-filter response is not simply the Fourier transform of the
emission spectrum. However, there is a tendency for abrupt falloff of
emission spectrum to be accompanied by high range sidelobes in the
response, much as abrupt steps in the time waveform are accompanied by
high sidelobes in the emission spectrum. To some extent, range sidelobe
suppression can be improved by mismatching the receiver signal processor
to the transmitted pulse, but this incurs a loss of sensitivity relative
to that of a matched filter. It is therefore necessary to choose a
waveform that makes a good tradeoff among spectrum control, range
sidelobe suppression, and sensitivity. (Swept-frequency waveforms using
a slightly nonlinear FM profile are good compromises for some
applications.) In general, however, the need for good resolution and
sensitivity narrows the designers' options. In addition, many radar
applications require nearly uniform response over a substantial span of
Doppler frequencies; i.e., they are required to have low "Doppler
sensitivity". This introduces another constraint on the designers'
choice of waveform.

In communication systems, the improvement of spectrum falloff that is
gained by premodulation filtering comes at the expense of worsening
inter-symbol interference. Nevertheless, considerable improvement in
spectrum control can often be achieved before inter-symbol interference
becomes unacceptable. In a radar, the spectrum improvement that can be
gained by use of pre-modulation filtering comes at the cost of degrading
the radar\'s resolution. It is also to be expected that a slight loss of
detection sensitivity will be incurred due to the difficulty of
constructing a perfectly matched filter (or correlation process) for
waveforms containing rounded corners (resulting from premodulation
filtering) instead of sharp discontinuities. As with the analogous
communication-system case, however, it is reasonable to expect that
considerable improvement in spectrum control can often be achieved
before the ambiguity function or sensitivity is degraded significantly.
As indicated above, removing discontinuities in as many derivatives of
the waveform as possible is what determines the ultimate spectrum
falloff slope that is achieved at large frequency offsets. That does not
necessarily require that the filtering have a narrow bandwidth, although
that bandwidth determines the frequency offset at which the ultimate
spectrum slope is attained.

The preceding discussion applies when the waveform has constant
amplitude within its subpulse dwells. The use of multiple
power-amplifier modules (usually solid-state amplifiers) in either
in-guide-combiner or spatial-power-combining architectures, opens up
possibilities for use of smooth amplitude-modulated waveforms. These are
not known to be used in currently-deployed radars, but such waveforms
might be used in the future. This would establish another degree of
design freedom that could be exploited in part to help control radar
emission spectra.

# 3 Selection of radar output devices

The selection of the radar transmitter output device affects the design
of not only the transmitter, but also the radar receiver and antenna
systems. Also, the design of multifunction radar systems can even
further complicate the selection of a radar output device. Other major
design factors in selecting an output device include: energy efficiency
(conversion of DC energy to RF), instantaneous bandwidth (available
tuning bandwidth without adjustments), and pulse-to-pulse coherency
(relative phase of each pulse which is important for Doppler
processing), weight, size, mechanical ruggedness, life of the device and
cost.

Table 1 shows the output device performance for major design factors
considered in the design of radar systems. As seen in Table 1, there is
a wide variation in the output device characteristics for the major
design factors of peak power, instantaneous bandwidth and energy
efficiency. It should be noted that the above design factors must be
given primary consideration in the selection of the radar output device
to ensure that the radar mission(s) can be achieved. Radar output device
spurious emission characteristics are considered only after all mission
objectives are satisfied.

The levels of spurious emissions from radar transmitters are dependent
upon the output device used in the radar transmitter. Knowledge of the
inherent spurious emission characteristics of the various output devices
used in radar transmitters is essential in promoting efficient use of
the spectrum and minimizing interference to services operating in
adjacent bands.

Table 2 lists the spurious emission characteristics (non-harmonic and
harmonic) for output devices used in radar systems operating in the
3 GHz and 5 GHz bands. Radar systems using crossed-field output devices
have inherent non‑harmonic spurious emission levels that would require
filtering if spurious emission limits are greater than about ‑60 dBc.
Both linear-beam tubes and solid-state output devices have inherent
non-harmonic spurious that are below ‑100 dBc. All radar output devices
have harmonic spurious emissions in the range of --15 to --55 dBc, and
thus require filtering to suppress the harmonic spurious emissions. For
radars employing distributed output devices (phased arrays) filtering
may not be practical.

TABLE 1

Radar output device performance for major design factors considered for
radar systems\
operating in the 3 GHz and 5 GHz bands

TABLE 2

Radiodetermination pulsed output device spurious emission\
characteristics for systems in the 3 and 5 GHz bands

# 4 Radar output filters

RF filters at the transmitter output can be very helpful in suppressing
harmonic emissions. RF filters can also be used to suppress
"out-of-band" and non-harmonic spurious emissions that lie closer to the
fundamental emission than the harmonics do. However, their utility for
controlling relatively close-in portions of the emission spectrum is
limited. This is due partly to their additional cost, weight, and size,
and also to the fact that many radars are tunable and/or use multiple
waveforms, some of which have much wider necessary bandwidths than
others have. It is scarcely practical to implement high-power RF filters
that can be re-configured to accommodate changes in carrier frequency or
pulse waveform, especially when it is considered that such changes are
likely to occur within milliseconds.

Transmitter architecture is also an important determinant of the
achievable degree of spectrum control. Where multiple power amplifiers
are used, emission-spectrum falloff rate and level are influenced by
whether the outputs from those power amplifiers are combined within the
transmitter waveguide or only in space after being radiated. In-guide
combining effectively creates a severe impedance mismatch for the
mutually incoherent components of the output waveforms from the power
amplifiers, which can dramatically lower the radiated noise power
relative to the sum of the available noise powers of the amplifiers.
With array radars fed by multiple amplifiers, on the other hand, spatial
power combining is an attractive architecture, but it allows all the
noise power of the amplifiers to be radiated. Opportunities for RF
filtering are also limited in such arrays. This is partly because a
separate filter would be needed for each amplifier, where the number of
amplifiers can be in the hundreds or thousands. It is also partly
because the space between filters would need to be on the order of half
a wavelength, since the radiating elements are normally that close
together to avoid occurrence of unacceptable lobes in the antenna gain
pattern. That space is insufficient for very effective filtering.

As shown in Table 2, the selection of the radar output device has a
major effect on the requirement for filtering non-harmonic spurious
emissions. However, as mentioned earlier, the selection of the radar
output device can not be made entirely on spurious emission
characteristics. Because of the inherently high levels of harmonic
spurious emissions of all output devices, the suppression of harmonic
spurious emissions by the use of harmonic (low pass) filters is
generally performed when practical. To mitigate radar non-harmonic
spurious emissions bands from some moderate and high‑power radars in
bands adjacent to radiodetermination, bandpass filters after the radar
transmitter would also be required for some radar output devices. These
would typically have to be separate from the harmonic filter, since the
wide-stop-band characteristics of harmonic‑suppression filters can not
ordinarily be achieved along with the sharp cutoff characteristic of
adjacent‑band‑suppression filters. The number of filters required can be
much greater than two, however; in active-array radars, one or two
filters would need to be interposed between each power‑output device and
the antenna element or subarray that it feeds. Thousandas of filters
could be required altogether.

The monetary cost of these filters can be significant, since
unconventional filter types, sometimes requiring pressurization or
evacuation, are required to handle the high powers and maintain the
desired suppression over a wide stopband. Use of such filters also
imposes trade-offs in radar system performance. The insertion loss of
transmitter harmonic filters and bandpass filters for radars in these
bands ranges from 0.1 to 0.7 dB. If both harmonic and bandpass filters
are required, the insertion loss would be roughly double. Due to the
many variables in radar operation, the attendant decrement in detection
and tracking performance usually goes unnoticed, but the fact is that
even 0.2 dB represents a major loss of RF power (for example, 47 kW of
peak power in the case of a 1 MW radar). The transmitter would need to
be that much more powerful to recover the performance loss, since it
must be assumed that more cost effective means of improving the
performance would already have been exploited. A loss of 0.4 dB, for
example, corresponds to a 2.3% reduction of detection range, which is
inconsequential for most radars but significant for some. The voltage
standing wave ratio of both types of filters is in the range of 1.1 to
1.3.

Also, power handling, size, and weight of the filter are factors to be
considered in the feasibility of using an output filter on the radar,
particularly in mobile radars. Size and weight can be overriding
considerations in the case of mobile, active-array radars. Filtering
bands close to the radar operating band requires steep selectivity
skirts and hence high energy storage, which raises the risk of breakdown
(or lowers the power-handling capacity) and can also introduce phase
distortion in the passband -- another major consideration for
active-array radars. The higher the radar power, the more attenuation is
needed to suppress spurious outputs to a given level, so the more
sections the filters will need, and hence the higher their insertion
loss, size, and weight will tend to be.

Transmitter filtering is best implemented during the original design of
the radar. The addition of transmitter filters to existing radars has
been achieved with minimal impact on system performance in many cases,
but there have been other cases in which breakdown problems have
occurred when a bandpass filter was added to suppress adjacent‑band
emissions.

# 5 Radar trends

Two major areas of advance driving the direction of the selection of
radar output devices are:

-- digital radar signal processing which is leading to rapid growth in
Doppler radars which require high pulse-to-pulse coherency (linear beam
and solid state output devices),

-- the development of higher power solid-state transmitter devices
(modular/bottled and distributed (phased array) configurations).

These trends will have an influence on reducing the spurious emission
levels of the newer generation of radars.

[^1]: ^\*^ This Recommendation should be brought to the attention of the
    International Maritime Organization (IMO), the International Civil
    Aviation Organization (ICAO), the International Maritime Radio
    Committee (CIRM), and the World Meteorological Organization (WMO)
    and Radiocommunication Study Groups 1 and 9.

[^2]: Chirp waveforms are sometimes referred to as coded waveforms even
    though their "coding" is not discrete.